A8582
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
Description
Features and Benefits
• Wide operating voltage range: 4.7 to 36 V
• UVLO stop threshold is at 3.8 V (typ)
• Supports 40 V input for surge and load dump testing
• Capable of at least 2.0 A steady-state output current
• Adjustable output voltage as low as 0.8 V
• Internal 70 mΩ high-side switching MOSFET
• Adjustable switching frequency, fSW: 0.25 to 2.4 MHz
• Synchronization to external clock: 1.2 × fSW to 1.5 × fSW
• Sleep mode supply current less then 3 μA
• Soft start time externally set via the SS pin
• Very low no-load current, typically 3.5 mA
• Pre-bias startup compatible
• Power OK (POK) output
• Pulse-by-pulse current limiting (OCP)
• Hiccup mode short-circuit protection (HIC)
• Overtemperature protection (TSD)
• Overvoltage protection (OVP)
• Missing asynchronous diode (D1) protection
• Open-circuit and adjacent pin short-circuit tolerant
• Short-to-ground tolerant at every pin
• Externally adjustable compensation
• Stable with ceramic output capacitors
The A8582 is an adjustable frequency, high output current,
PWM regulator that integrates a low resistance, high-side,
N-channel MOSFET. The A8582 incorporates current-mode
control to provide simple compensation, excellent loop
stability, and fast transient response. The A8582 utilizes
external compensation to accommodate a wide range of power
components to optimize transient response without sacrificing
stability.
The A8582 regulates input voltages from 4.7 to 36 V, down to
output voltages as low as 0.8 V, and is able to supply at least
2.0 A of load current. The A8582 features include an externally
adjustable switching frequency, an externally set soft start
time to minimize inrush currents, an EN/SYNC input to either
enable VOUT and/or synchronize the PWM switching
frequency, and a Power OK (POK) output to indicate when
VOUT is within regulation. The A8582 only turns on the
lower FET to charge the boot capacitor when needed, not at
Continued on the next page…
Package: 16-pin TSSOP (suffix LP)
Applications:
•
•
•
•
GPS/infotainment
Automobile audio
Home audio
Network and telecom
Not to scale
Typical Application
V IN
CIN2
50 V
Empty
CIN1
3.3 μF
50 V
1
2
3
5
12
7
4
CSS
22 nF
8
11
RFSET
11.5 kΩ
RZ
15.4 kΩ
CP
10 pF
CZ
820 pF
VIN
VIN
VIN
SW
SW
CBOOT
100 nF
A8582
GND
GND
BOOT
EN/SY NC
FBX
LO
2.2 μH
16
15
14
D1
3 A /40 V
SMA
CO1
10 μF
16 V
RS1
16.5 kΩ
10
CFBX
120 pF
SS
VOUT
RS2
5.23 kΩ
FSET
FB
COMP
9
RFB2
5.23 kΩ
PAD
POK
RFB1
16.5 kΩ
3.3V
RPU
2 kΩ
6
Figure 1. Application schematic, at VIN 5 to 16 V, 3.3 VOUT , at 2 MHz
A8582-DS, Rev. 14
POK
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
Description (continued)
every PWM cycle. This improves light load efficiency and provides
no-load currents as low as 4 mA at 2 MHz. The sleep mode current
of the A8582 control circuitry is less than 3 μA.
diode (D1) protection, open-circuit, adjacent pin short-circuit,
and short-to-ground protection at every pin to satisfy the most
demanding applications.
Protection features include VIN undervoltage lockout (UVLO),
pulse-by-pulse overcurrent protection (OCP), hiccup mode shortcircuit protection (HIC), overvoltage protection (OVP), and thermal
shutdown (TSD). In addition, the A8582 provides unique missing
The A8582 device is available in a 16-pin TSSOP package with
exposed pad for enhanced thermal dissipation. It is lead (Pb) free,
with 100% matte tin leadframe plating.
Selection Guide
Part Number
Packing
A8582KLPTR-T
4000 pieces per 13-in. reel
Absolute Maximum Ratings1
Characteristic
Symbol
VIN Pin to GND
Notes
VIN
SW Pin to GND2
SS Pin
All Other Pins
Unit
–0.3 to 40
V
Continuous
–0.3 to VIN + 0.3
V
Single pulse, tW < 50 ns
–1.0 to VIN + 5.0
V
VBOOT
VSW – 0.3 to
VSW + 7.0
V
VSS
–0.3 to VIN + 0.3
V
VI
–0.3 to 5.5
V
VSW
BOOT Pin Above SW Pin
Rating
Operating Ambient Temperature
TA
–40 to 125
ºC
Maximum Junction Temperature
TJ(max)
150
ºC
Tstg
–55 to 150
ºC
Storage Temperature
K temperature range for automotive
1Stresses
beyond those listed in this table may cause permanent damage to the device. The absolute maximum ratings are
stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the Electrical
Characteristics table is not implied. Exposure to absolute-maximum–rated conditions for extended periods may affect device reliability.
2SW has internal clamp diodes to GND and VIN. Applications that forward bias these diodes should take care not to exceed the IC
package power dissipation limits.
Thermal Characteristics
Characteristic
Symbol
Package Thermal Resistance
RθJA
Test Conditions*
On 4-layer PCB based on JEDEC standard
Value
Unit
34
ºC/W
*Additional thermal information available on the Allegro website
Table of Contents
Specifications
2
Functional Block Diagram
Pin-out Diagram and Terminal List
3
4
Typical Characteristic Performance
8
Functional Description
10
10
14
Overview
Protection Features
Application Information
Design and Component Selection
Package Outline Drawing
16
16
30
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
2
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
Functional Block Diagram
BOOT
VIN
VREG
VREF 0.8 V
POR
Regulator
Slope
Compensation
0.85 V
Typ
SW
∑
Sleep
10 Ω
tOFF(MIN)
–
TSD
–
40 kΩ Typ
VSS –
400 mV
3.5 kΩ
VREF
FAULT A
Fault
Logic
POK
Diode Missing
FBOK
Rising
OVP
85%×VREF
90%×VREF
–
150 nA
FB
+
Clamp 1.7 V Typ
–
+
+
COMP
Error
Amp
OCP
Hiccup reset VSS = 235 mV Typ
10 μA
BOOT
– SW
OFF
POR
Latch reset
EN/SYNC
toggle
EN Digital
UVLO (VIN)
20 μA
1500 Ω
Latched
Hiccup
Protection
HICCUP
B
OVP
–
SS
2.9 V
BS UVLO
+
1.25 V Typ
1.65 V Typ
SW
R Q
EN/SYNC
Comp
EN/SYNC
Diode
Missing
Reset
DOM
S Q
PWM
Comp
PWM
Ramp
Offset
300 mV
–
+
70
mΩ
VREG
PWM Clk
SYNC
Adj
FSET
Current Sense
Amp
OCP
fSW
OSC
Adj
EN/SYNC >1.2 × fSW
FBOK
UVLO
(VIN)
+
VIN
+
VIN
OFF
GND
GND
PAD
OVP
Comp
114%×VREF
FBX
125 ns
A FAULT = 1, if:
EN = 0, or
UVLO = 1, or
OVP = 1, or
Diode Missing = 1
B HICCUP = 1, if Hiccup protection enabled (VFB < 625 mV) and
a net count of > 7 OCP events occur
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
3
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
Pin-out Diagram
VIN 1
16 SW
VIN 2
15 SW
VIN 3
14 BOOT
SS 4
GND 5
POK 6
EN/SYNC 7
FSET 8
PAD
13 NC
12 GND
11 COMP
10 FBX
9 FB
Terminal List Table
1, 2, 3
VIN
Power input for the control circuits and the drain of the internal high-side N-channel
MOSFET. Connect this pin to a power supply of 4.7 to 36 V. A high quality ceramic capacitor
should be placed very close to this pin.
4
SS
Soft-start pin. Connect a capacitor, CSS, from this pin to GND to set the soft-start time. This
capacitor also determines the hiccup period during an overcurrent event.
5, 12
GND
Ground.
6
POK
Power OK output signal. This pin is an open drain output that transitions from low impedance
to high impedance when the output is within the final regulation voltage.
7
EN/SYNC
Enable and synchronization input. This pin is a logic input that turns the converter on or
off. Set this pin to logic high to turn the converter on or set this pin to logic low to turn the
converter off. This pin also functions as a synchronization input to allow the PWM frequency
to be set by an external clock.
8
FSET
9
FB
10
FBX
Remote sense input for the overvoltage protection (OVP) comparator. Connect a resistive
divider from the converter output node, VOUT , to this pin to set the OVP trip threshold. If
OVP protection is not required, this pin should be grounded.
11
COMP
Output of the error amplifier and compensation node for the current-mode control loop.
Connect a series RC network from this pin to GND for loop compensation. See the Design
and Component Selection section of this datasheet for further details.
13
NC
14
BOOT
15, 16
SW
The source of the internal high-side N-channel MOSFET. The external free-wheeling diode
(D1) and output inductor (LO) should be connected to this pin. Both D1 and LO should be
placed close to this pin and connected with relatively wide traces.
–
PAD
Exposed pad of the package providing enhanced thermal dissipation. This pad must be
connected to the ground plane(s) of the PCB with at least 6 vias, directly in the pad.
Frequency setting pin. A resistor, RFSET, from this pin to GND sets the PWM switching
frequency. See figure 10 and/or equation 2 to determine the value of RFSET.
Feedback (negative) input to the Error amplifier. Connect a resistive divider from the
converter output node, VOUT , to this pin to program the output voltage.
No connect.
High-side gate drive boost input. This pin supplies the drive for the high-side N-channel
MOSFET. Connect a 100 nF ceramic capacitor from BOOT to SW.
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
4
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
ELECTRICAL CHARACTERISTICS1 Valid at VIN = 12 V, TA = 25°C ,
indicates specifications guaranteed through
–40°C ≤ TJ ≤ 125°C ; unless otherwise specified
Characteristics
Symbol
Test Conditions
Min.
Typ.2
Max.
Unit
4.7
−
36
V
V
Input Voltage Specifications
Operating Input Voltage Range
VIN
UVLO Start Threshold
VINSTART
VIN rising
−
4.2
4.6
UVLO Stop Threshold
VINSTOP
VIN falling
−
3.8
4.2
V
280
400
520
mV
VEN/SYNC = 5 V, VFB = 1.5 V,
no PWM switching
−
3.0
5.0
mA
VIN = 16 V, VEN/SYNC ≤ 0.4 V,
TA = TJ between –40°C and 85°C
−
−
3.0
μA
VIN = 16 V, VEN/SYNC ≤ 0.4 V, TA = TJ = 125°C
−
5
15
μA
792
800
808
mV
−
–150
–300
nA
UVLO Hysteresis
VUVLOHYS
Input Currents
Input Quiescent Current
Input Sleep Supply Current3
IQ
IQSLEEP
Reference Voltage
Feedback Voltage
VFB
4.7 V < VIN < 36 V, VFB = VCOMP
IFB
VCOMP = 1.5 V, VFB regulated so that
ICOMP = 0 A
Error Amplifier
Feedback Input Bias Current
Open Loop Voltage Gain
Transconductance
Source Current
Sink Current
Maximum Output Voltage
COMP Pull-Down Resistance
AVOL
gm
ICOMP = 0 μA, VSS > 700 mV
–
56
–
dB
550
750
1000
μA/V
0 V < VSS < 700 mV
–
225
–
μA/V
IEA(SRC)
VFB < 0.8 V, VCOMP = 1.5 V
−
–50
−
μA
IEA(SINK)
VFB > 0.8 V, VCOMP = 1.5 V
−
+50
−
μA
1.3
1.7
2.1
V
FAULT = 1
−
1500
−
Ω
VCOMP for 0% duty cycle
−
300
−
mV
VEAVO(max)
RCOMP
Pulse Width Modulation (PWM)
PWM Ramp Offset
VPWMOFFSET
Minimum Controllable On-Time
tON(MIN)
−
65
100
ns
Minimum Switch Off-Time
tOFF(MIN)
−
65
130
ns
COMP to SW Current Gain
gmPOWER
Slope Compensation
SE
−
2.85
−
A/V
fSW = 250 kHz
−
0.19
−
A/μs
fSW = 2.0 MHz
−
1.5
−
A/μs
IDS = 400 mA, VBOOT − VSW = 6 V
−
70
−
mΩ
VIN = 16 V, VEN/SYNC ≤ 0.4 V, VSW = 0 V
TA = TJ between –40°C and 85°C
−
−
10
μA
VIN = 16 V, VEN/SYNC ≤ 0.4 V, VSW = 0 V,
TA = TJ = 125°C
−
50
150
μA
IDS = 10 mA, (VBOOT – VSW) < 4 V
−
10
12
Ω
MOSFET Parameters
Hi-Side MOSFET On Resistance
High-Side MOSFET Leakage Current3
Low-Side MOSFET On Resistance
RDS(on)HS
ILEAK
RDS(on)LS
Continued on the next page…
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
5
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
ELECTRICAL CHARACTERISTICS1 (continued) Valid at VIN = 12 V, TA = 25°C ,
indicates specifications guaranteed through
–40°C ≤ TJ ≤ 125°C ; unless otherwise specified
Characteristics
Symbol
Test Conditions
Min.
Typ.2
Max.
Unit
Oscillator Frequency
Oscillator Frequency
fSW
RFSET = 9.09 kΩ
2.20
2.45
2.70
MHz
RFSET = 24.9 kΩ
0.90
1.00
1.10
MHz
RFSET = 105 kΩ
−
250
−
kHz
Synchronization Timing
Synchronization Frequency Range
fSW_MULT
1.2 × fSW
−
1.5 × fSW
MHz
Synchronized PWM Frequency
fSW_SYNC
−
−
2.9
MHz
Synchronization Input Duty Cycle
DSYNC
−
−
80
%
Synchronization Input Pulse Width
tWSYNC
200
−
−
ns
Synchronization Input Edge Rise Time
trSYNC
−
10
15
ns
Synchronization Input Edge Fall Time
tfSYNC
−
10
15
ns
Enable/Synchronization Input
EN/SYNC High Threshold
VENIH
VEN/SYNC rising
−
1.65
1.80
V
EN/SYNC Low Threshold
VENIL
VEN/SYNC falling
−
1.25
−
V
EN/SYNC Low Threshold (Sleep)
0.40
0.85
−
V
EN/SYNC Hysteresis
VENHYS
VENIH – VENIL
−
400
−
mV
EN/SYNC Digital Delay
tSLEEP
VEN/SYNC transitioning high or low cycles
−
32
−
PWM
cycles
20
40
−
kΩ
Duty cycle = 5%, EN/SYNC = High (no sync)
2.80
3.25
3.70
A
Duty cycle = 90%, EN/SYNC = High (no sync)
EN/SYNC Input Resistance
VENILSLEEP VEN/SYNC falling
REN/SYNC
Overcurrent Protection (OCP) and Hiccup Mode
Pulse-by-Pulse Current Limit
Hiccup Disable Threshold
Hiccup Enable Threshold
OCP / HICCUP Count Limit
ILIM
2.30
2.70
3.15
A
VHICDIS
VFB rising
−
750
−
mV
VHICEN
VFB falling
−
625
−
mV
OCPLIMIT
Hiccup enabled, OCP pulses
−
7
−
counts
VOVPTRIP
VFBX rising, as a percentage of VREF
112
114
116
%
−
125
−
ns
255
330
−
mV
−
235
310
mV
Overvoltage Protection (OVP)
OVP Comparator Threshold
FBX Time Constant
(Filtering)4
τFBX
Soft Start (SS)
SS COMP Release Voltage
VSSRELEASE VSS rising due to ISSSU
SS Fault/Hiccup Reset Voltage
VSSRESET
VSS falling due to ISSHIC
SS Maximum Charge Voltage
VSSCHRG
SS Startup (Source) Current
ISSSU
VSS = 1 V, HICCUP = FAULT = 0
SS Hiccup (Sink) Current
ISSHIC
VSS = 0.5 V, HICCUP = 1
−
3.1
−
V
−10
–20
−30
μA
5
10
20
μA
Continued on the next page…
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
6
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
ELECTRICAL CHARACTERISTICS1 (continued) Valid at VIN = 12 V, TA = 25°C ,
indicates specifications guaranteed through
–40°C ≤ TJ ≤ 125°C ; unless otherwise specified
Characteristics
Symbol
Test Conditions
Min.
Typ.2
Max.
Unit
Soft Start (SS) (continued)
SS Input Resistance
SS to VOUT Delay Time
VOUT Soft Start Ramp Time
SS Switching Frequency
FAULT = 1
−
3.5
−
kΩ
tSSDELAY
RSS
CSS = 22 nF
−
363
−
μs
tSS
CSS = 22 nF
−
880
−
μs
VFB = 0 V
−
fSW / 3
−
MHz
VFB ≥ 600 mV
−
fSW
−
MHz
fSS
Power OK (POK) Output
VPOK
IPOK = 4 mA
−
−
0.4
V
POK Leakage
IPOKLEAK
VPOK = 5 V
−
−
1
μA
POK Comparator Threshold
VPOKTHRESH
VFB rising, as a percentage of VREF
87
90
93
%
POK Hysteresis
VPOKHYS
VFB falling, as a percentage of VREF
2
5
6
%
−
PWM
cycles
POK Output Voltage
tdPOK
VFB rising only
Thermal Shutdown Threshold4
TTSD
Temperature rising
Thermal Shutdown Hysteresis4
TTSDHYS
POK Digital Delay
−
7
150
165
−
°C
−
20
−
°C
Thermal Protection (TSD)
1For input and output current specifications, negative current is defined as coming out of the node or pin (sourcing), positive current is defined as
going into the node or pin (sinking).
2Typical specifications are at T = 25ºC.
A
3For T = T between –40°C and 85°C, ensured by design and characterization, not production tested.
A
J
4Ensured by design and characterization, not production tested.
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
7
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
Typical Characteristic Performance
Switching Frequency versus Temperature
Switching
Frequency,
Switching
Frequency,
fSW f(MHz)
SW (MHz)
Reference Voltage versus Temperature
803
802
801
800
799
798
797
796
-50
-25
0
25
50
75
100
125
150
175
262
1.01
261
1.00
260
259
0.99
258
0.98
257
256
0.97
255
0.96
254
-50
-50
-25
-25
0
0
25
25
50
50
75
75
100
100
125
125
150
150
175
175
Ambient Temperature, TA (°C)
Soft Start (Source) Current versus Temperature
Soft Start Hiccup (Sink) Current versus Temperature
–16
–17
–18
–19
–20
-50
-25
0
25
50
75
100
125
150
175
10.0
9.9
9.8
9.7
9.6
9.5
-50
-25
Ambient Temperature, TA (°C)
750
700
650
-25
0
25
50
75
100
Ambient Temperature, TA (°C)
25
50
75
100
125
150
175
Error Amplifier Voltage Gain versus Temperature
125
150
175
Open Loop Voltage Gain, AVOL (dB)
800
600
-50
0
Ambient Temperature, TA (°C)
Error Amplifier Transconductance versus Temperature
Transconductance, gm (μA/V)
1.02
263
Ambient Temperature, TA (°C)
SS Hiccup (Sink) Current, ISSHIC (μA)
SS Startup (Source) Current, ISSSU (μA)
Reference Voltage, VREF (mV)
804
VIN = 4.7 V
58
57
56
55
54
53
52
-50
-25
0
25
50
75
100
125
150
175
Ambient Temperature, TA (°C)
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
8
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
UVLO Threshold Voltage versus Temperature
Enable Threshold Voltage versus Temperature
4.2
VINSTART (VIN rising)
4.1
4.0
3.9
VINSTOP (VIN falling)
3.8
3.7
-50
Input Quiescent Current, IQ (μA)
EN/SYNC Threshold, VENx (V)
1.4
-25
0
25
50
75
100
125
150
175
1.3
VENIH (Run: VEN/SYNC rising)
1.2
1.1
1.0
0.9
VENILSLEEP (Sleep: VEN/SYNC falling)
0.8
0.7
0.6
-50
-25
0
25
50
75
100
125
150
175
Ambient Temperature, TA (°C)
Ambient Temperature, TA (°C)
Sleep Input Current versus Temperature
SW Leakage Output Current versus Temperature
VIN = 16 V, EN/SYNC = Low
VIN = 16 V, EN/SYNC = Low
5
60
High-Side MOSFET Leakage,
ILEAK (μA)
UVLO Threshold, VINx (V)
4.3
4
3
2
1
0
–1
-50
-25
0
25
50
75
100
Ambient Temperature, TA (°C)
125
150
175
50
40
30
20
10
0
–10
-50
-25
0
25
50
75
100
125
150
175
Ambient Temperature, TA (°C)
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
9
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
Functional Description
Overview
The A8582 is an asynchronous PWM regulator that incorporates
all the control and protection circuitry necessary to satisfy a wide
range of applications. The A8582 employs current mode control
to provide fast transient response, simple compensation, and
excellent stability. The features of the A8582 include a precision
reference, an adjustable switching frequency, a transconductance
error amplifier, an enable/synchronization input, an integrated
high-side N-channel MOSFET, adjustable soft-start time, pre-bias
startup, low current sleep mode, and a Power OK (POK) output. The protection features of the A8582 include undervoltage
lockout (UVLO), pulse-by-pulse over current protection (OCP),
hiccup mode short-circuit protection (HIC), overvoltage protection (OVP), and thermal shutdown (TSD). In addition, the A8582
provides open-circuit, adjacent pin short-circuit, and pin-toground short circuit protection.
Reference Voltage
The A8582 incorporates an internal reference that allows output
voltages as low as 0.8 V. The accuracy of the internal reference
is ±1% through the operating temperature range. The output voltage of the regulator is adjusted by connecting a resistor divider
(RFB1 and RFB2 in figure 1) from VOUT to the FB pin of
the A8582.
with two positive and one negative inputs. The negative input is
simply connected to the FB pin and is used to sense the feedback
voltage for regulation. The two positive inputs are used for soft
start and regulation. The error amplifier performs an “analog OR”
selection between the two positive inputs. The error amplifier
regulates to either the soft start pin voltage (minus 400 mV) or
the A8582 internal reference, whichever is lower.
To stabilize the regulator, a series RC compensation network
(RZ and CZ) must be connected from the error amplifier output
(COMP pin) to GND as shown in figure 1. In some applications,
an additional, a low value capacitor (CP) may be connected in
parallel with the RC compensation network to reduce the loop
gain at higher frequencies. However, if the CP capacitor is too
large, the phase margin of the converter may be reduced. If the
regulator is disabled or a fault occurs, the COMP pin is immediately pulled to GND via approximately 1500 Ω, and PWM
switching is inhibited.
Slope Compensation
The A8582 incorporates internal slope compensation to allow
PWM duty cycles above 50% for a wide range of input/output
voltages, switching frequencies, and inductor values. As shown in
the Functional Block Diagram, the slope compensation signal is
added to the sum of the current sense and PWM ramp offset. The
amount of slope compensation is scaled directly with the switching frequency.
Oscillator/Switching Frequency
The PWM switching frequency of the A8582 is adjustable from
250 kHz to 2.4 MHz and has an accuracy of ±12% through
the operating temperature range. Connecting a resistor from
the FSET pin to GND, as shown in figure 1, sets the switching
frequency. An FSET resistor with 1% tolerance is recommended.
A graph of switching frequency versus FSET resistor value is
shown in the Design and Component Selection section of this
datasheet.
Transconductance Error Amplifier
The primary function of the transconductance error amplifier
is to regulate the converter output voltage. The error amplifier
is shown in figure 2. It is shown as a 3-terminal input device
400 mV
A8582
SS
Error Amplifier
+
+
VREF
800 mV
COMP
-
FB
Figure 2. The A8582 transconductance error amplifier
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Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
Sleep Mode
If the voltage at the EN/SYNC pin is pulled below 400 mV
(VENILSLEEP ) the A8582 will enter a sleep mode where the internal control circuits will be shut off and draw less than 3 μA from
VIN . However, the total current drawn by the VIN pin will be the
sum of the current drawn by the control circuitry (<3 μA) plus
any leakage due to the high-side MOSFET (<10 μA at 25°C).
Note that, when used as a synchronization input, soft start is at
the base frequency set by the FSET resistor. Synchronization
to the external clock occurs after soft start is completed (when
VFB > VPOKTHRESH). When being used as a synchronization
input, the applied clock pulses must satisfy the pulse width, dutycycle, and rise/fall time requirements shown in the Electrical
Characteristics table in this datasheet.
To automatically enable the A8582, the EN/SYNC input pin may
be connected to a voltage rail, such as VIN , via a resistor and a
Zener diode as shown in figure 4.
Enable/Synchronization (EN/SYNC) Input
The enable/synchronization (EN/SYNC) input provides three
functions:
• A control input that commands the sleep mode of the A8582.
When EN/SYNC is very low (VEN/SYNC< VENILSLEEP ), most
of the internal circuits are de-biased to provide the sleep mode
current of less than 3 μA.
There is a short delay between when EN/SYNC transitions low
and when PWM switching stops. This is necessary because the
enable circuitry must distinguish between a constant logic level
and synchronization pulses at the lowest switching frequency.
The nominal delay from when EN/SYNC transitions low and
PWM switching stopping is 32 PWM clock cycles. The shut-
• A simple logic input. If EN/SYNC is a logic low (VEN/SYNC <
VENIL ), then the A8582 and VOUT will be off. If EN/SYNC is
a logic high (VEN/SYNC > VENIH ), the A8582 will turn on and,
provided there are no fault conditions, soft start will be initiated
and VOUT will ramp to its final voltage in a time set by the soft
start capacitor (CSS). (The operating modes of the A8582 based
on EN/SYNC voltage are summarized in figure 3.)
• A synchronization input that accepts an external clock to turn on
the A8582 and (after soft starting) will scale the PWM switching frequency from 1.2X to 1.5X above the base frequency set
by the FSET resistor.
V IN
R
EN/SYNC
A8582
2.2 V < V Z < 4.7 V
Figure 4. Automatically enabling the A8582 from VIN or some other
power rail
VEN > 1.15 V
VEN > 1.65 V
SLEEP
WAKE
RUN
(iIN < 3 μA
PWM = Off )
(iIN ≈ 2 mA
PWM = Off )
(iIN ≈ 3 mA
PWM = On )
VEN < 0.85 V
for 32 cycles
VEN > 1.15 V
VSS < 0.2 V
Discharge
Soft-start capacitor
Wait up to
32 cycles
(PWM = Off)
(PWM = On)
VEN < 0.85 V
Timer expired
Figure 3. EN/SYNC voltage and A8582 operating modes
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11
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
down transition delay from switching to sleep mode is shown in
figure 5.
above the error amplifier voltage, the comparator will reset the
PWM flip-flop and the upper MOSFET will be turned off. If the
output voltage of the error amplifier drops below the PWM Ramp
Offset (VPWMOFFSET) then zero PWM duty-cycle (pulse skipping)
operation is achieved.
Power MOSFETs
The A8582 includes a low RDS(on) , high-side N-channel
MOSFET capable of delivering up to 2.0 A of current at high
duty cycles. The A8582 also includes a 10 Ω, low-side MOSFET
to insure the boot capacitor (CBOOT) is always charged.
Current Sense Amplifier
Unlike other typical asynchronous regulators, the A8582 only
turns on the lower MOSFET when the boot capacitor must be
charged. This minimizes negative currents in the output inductor
and improves the light load efficiency. When the EN/SYNC input
is low or a fault occurs, the A8582 is disabled and the regulator output stage is tristated by turning off both the upper and
lower MOSFETs.
Pulse Width Modulation (PWM)
A high-speed PWM comparator, capable of pulse widths less
than 100 ns, is included in the A8582. The inverting input of
the comparator is connected to the output of the error amplifier.
The noninverting input is connected to the sum of the current
sense signal, the slope compensation, and a PWM Ramp Offset
(VPWMOFFSET, nominally 300 mV). At the beginning of each
PWM cycle, the CLK signal sets the PWM flip-flop and the
upper MOSFET is turned on. When the summation of the DC
offset, the slope compensation, and the current sense signal rises
A high-bandwidth current sense amplifier monitors the current in
the upper MOSFET. The PWM comparator, the pulse-by-pulse
current limiter, and the hiccup mode up/down counter require the
current signal.
Soft Start (Startup) and Inrush Current Control
Inrush currents to the converter are controlled by the soft start
function of the A8582. When the A8582 is enabled and all faults
are cleared, the soft start (SS) pin will source approximately
20 μA (ISSSU) and the voltage on the soft start capacitor (CSS)
will ramp upward from 0 V. When the voltage on the soft start
pin exceeds the Soft Start COMP Release Voltage threshold
(VSSRELEASE , 330 mV typical, measured at the soft start pin) the
output of the error amplifier is released, and shortly thereafter the
upper and lower MOSFETs will begin switching. As shown in
figure 6, there is a short delay (tSSDELAY) to initiate PWM switching, between when the EN/SYNC pin transitions high and when
the soft start voltage reaches 330 mV.
VOUT
C1
VCOMP
32 cycles delay
C2
VEN/SYNC
C3
t
Figure 5. PWM switching stops and sleep mode begins approximately
32 cycles after EN/SYNC transitions low; shows VOUT (ch1, 1 V/div.),
VCOMP (ch2, 1 V/div.), VEN/SYNC (ch3, 2 V/div.), t = 50 μs/div.
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Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
When the A8582 begins PWM switching, the error amplifier
regulates the voltage at the FB pin to the soft start pin voltage
minus the Soft Start PWM Threshold voltage (VSSPWM). When
PWM switching starts, the voltage at the soft start pin rises from
330 mV to 1.13 V (a difference of 800 mV), the voltage at the FB
pin rises from 0 V to 800 mV, and the regulator output voltage
rises from 0 V to the required set-point determined by the feedback resistor divider (RFB1 and RFB2).
When the voltage at the soft start pin reaches approximately
1.13 V, the error amplifier will “switch over” and begin regulating to the A8582 internal reference, 800 mV. The voltage at the
soft start pin will continue to rise to about 3.3 V. The soft start
functionality is shown in figure 6.
If the A8582 is disabled or a fault occurs, the internal fault latch
is set and the soft start pin is pulled to GND via approximately
3.5 kΩ. The A8582 will clear the internal fault latch when the
voltage at the soft start pin decays to approximately 235 mV
(VSSRESET).
If the A8582 enters hiccup mode, the capacitor on the soft start
pin is discharged by a 10 μA current sink (ISSHIC ). Therefore, the
soft start pin capacitor value (CSS) controls the time between soft
start attempts. Hiccup mode operation is discussed in more detail
in the Output Short Circuit (Hiccup Mode) Protection section of
this datasheet. During startup, the PWM switching frequency is
scaled linearly from fSW / 3 to fSW as the voltage at the FB pin
ramps from 0 V to 600 mV. This is done to minimize the peak
current in the output inductor when the input voltage is high and
VEN/SYNC
C1
C2
C3
tSS
If the output capacitors are pre-biased to some voltage, the A8582
will modify the normal startup routine to prevent discharging
the output capacitors. Normally, the COMP pin is released and
PWM switching starts when the voltage at the soft start pin
reaches 330 mV. In the case with pre-bias at the output, the prebias voltage will be sensed at the FB pin. The A8582 will not
start switching until the voltage at the soft-start pin increases to
approximately VFB + 330 mV. At this soft start pin voltage, the
error amplifier output is released, the voltage at the COMP pin
rises, PWM switching starts, and VOUT will ramp upward starting
from the pre-bias level. Figure 7 shows startup when the output
voltage is pre-biased to 2.0 V.
Power OK (POK) Output
The Power OK (POK) output is an open drain output, so an
external pull-up resistor must be connected. An internal comparator monitors the voltage at the FB pin and controls the open drain
device at the POK pin. POK remains low until the voltage at the
FB pin is within 10% of the final regulation voltage. The POK
output is pulled low if: (1) the EN/SYNC pin transitions low for
more than 32 PWM cycles, (2) UVLO occurs, (3) TSD occurs, or
(4) OVP occurs.
VEN/SYNC
5V
VOUT increases
monotonically
C1
2V
VOUT
VOUT
C2
1.13 V
VCOMP
C3
0.330 V
C4
C5
Pre-Biased Startup
5V
tSSDELAY
VSS
the output of the regulator is either shorted, or soft starting a
relatively high output capacitance.
COMP pin
released
VCOMP
VSS
Switching delayed until
VSS = VFB + 0.330 V
0.330 V
C4
IL
C5
t
Figure 6. Startup to VOUT = 5 V, 2.0 A, with CSS = 22 nF; shows VEN/SYNC
(ch1, 2 V/div.), VOUT (ch2, 2 V/div.), VCOMP (ch3, 500 mV/div.), VSS (ch4,
500 mV/div.), IL (ch5, 2 A /div.), t = 200 μs/div.
IL
t
Figure 7. Startup to VOUT = 5 V, with VOUT pre-biased to 2 V; shows
VEN/SYNC (ch1, 2 V/div.), VOUT (ch2, 2 V/div.), VCOMP (ch3, 500 mV/div.),
VSS (ch4, 500 mV/div.), IL (ch5, 2 A /div.), t = 200 μs/div.
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Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
If the A8582 is running and EN/SYNC transitions low, then after
32 PWM cycles, POK will transition low and remain low only as
long as the internal rail is able to enhance the open drain output
device. After the internal rail collapses, POK will return to the
high impedance state. The POK comparator incorporates hysteresis to prevent chattering due to voltage ripple at the FB pin.
Protection Features
Undervoltage Lockout (UVLO)
An Undervoltage Lockout (UVLO) comparator monitors the voltage at the VIN pin and keeps the regulator disabled if the voltage
is below the lockout threshold (VINSTART). The UVLO comparator incorporates enough hysteresis (VUVLOHYS) to prevent on/off
cycling of the regulator due to IR drops in the VIN path during
heavy loading or during startup.
Thermal Shutdown (TSD)
The A8582 protects itself from over-heating, with an internal
thermal monitoring circuit. If the junction temperature exceeds
the upper thermal shutdown threshold (TTSD , nominally 165°C)
the voltages at the soft start and COMP pins will be pulled to
GND and both the upper and lower MOSFETs will be shut off.
The A8582 will stop PWM switching and stay in WAKE state
(see figure 3). It will automatically restart when the junction
temperature decreases more than the thermal shutdown hysteresis
(TTSDHYS , nominally 20°C).
3.6
Maximum
Pulse-by-Pulse Current LImit
ILIM, D (%)
3.4
Overvoltage Protection (OVP)
The A8582 provides a remote sense input pin (FBX) to protect
the system from an overvoltage condition. An overvoltage condition will occur if the FB pin is inadvertently grounded, the series
feedback resistor (RFB1 in figure 1) is missing, the FB pin is not
soldered, the FB trace is broken, or the COMP pin is shorted to
a voltage higher than approximately 1.6 V. When an overvoltage
condition is detected: (1) the fault is latched, (2) PWM switching
stops, and (3) POK, SS, and COMP are pulled low. An OVP fault
may be cleared by either toggling the EN/SYNC input or cycling
power to the VIN pin.
The FBX pin should be connected to VOUT using a feedback
resistor divider as shown in figure 1. To prevent nuisance trips it
is recommended that a capacitor (CFBX) be included from FBX
to ground to place a pole at approximately 2X to 5X the system
crossover frequency, fC (see the Compensation Components
section of this datasheet). For optimal protection the trace that
connects the FBX resistor divider should be separate from the
trace that connects the FB resistor divider. If the OVP function is
not required, the FBX pin can be grounded to essentially disable
the OVP comparator.
Usually, the FBX resistor divider will be identical to the FB
resistor divider and the OVP threshold will be equal to VOVPTRIP
shown in the Electrical Characteristics table (nominally 114% of
VREF). However, if nuisance trips occur during transient situations, the OVP trip threshold can be scaled slightly higher by
using a resistor divider that provides less voltage at the FBX pin.
Reducing the signal at the FBX pin essentially desensitizes the
OVP circuit, so care should be taken not to increase the OVP trip
threshold beyond a reasonable amount.
Table 1. Pulse-by-Pulse Current Limit
versus Duty Cycle
ILIM
(A)
Typical
D
(%)
5
2.80
3.20
3.70
Minimum
20
2.69
3.09
3.49
40
2.58
2.98
3.38
60
2.47
2.87
3.27
2.2
80
2.36
2.76
3.16
2.0
90
2.30
2.70
3.10
3.2
3.0
2.8
2.6
2.4
5
Min.
Typ.
Max.
10 15 20 25 30 35 40 45 50 55 60 65 70 75 80 85 90
Duty Cycle, D (%)
Figure 8. Pulse-by-pulse current limit versus duty cycle
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Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
Disable Threshold (VHICDIS , nominally 750 mV), Hiccup mode
protection is disabled.
Pulse-by-Pulse Overcurrent Protection (OCP)
The A8582 monitors the current in the upper MOSFET and if the
current exceeds the pulse-by-pulse overcurrent threshold (ILIM)
then the upper MOSFET is turned off. Normal PWM operation
resumes on the next clock pulse from the oscillator. The A8582
includes leading edge blanking to prevent falsely triggering the
pulse-by-pulse current limit when the upper MOSFET is turned
on. Pulse-by-pulse current limiting is always active.
Hiccup Mode overcurrent protection monitors the number of
overcurrent events using an up/down counter: an overcurrent
pulse increases the count by one, and a PWM cycle without an
overcurrent pulse decreases the count by one. If the total count
reaches more than 7 (while Hiccup mode is enabled) then the
Hiccup latch is set and PWM switching is stopped. The Hiccup
signal causes the COMP pin to be pulled low with a relatively
low resistance (1500 Ω). Hiccup mode also enables a current sink
connected to the soft start pin (nominally 10 μA) so, when Hiccup first occurs, the voltage at the soft start pin ramps downward.
Hiccup mode operation is shown in figure 9.
The A8582 is conservatively rated to deliver 2.0 A for most
applications. However, the exact current it can support is heavily
dependent on duty cycle, ambient temperature, thermal resistance of the PCB, airflow, component selection, and nearby heat
sources. The A8582 is designed to deliver more current at lower
duty cycles and slightly less current at higher duty cycles. For
example, the pulse-by-pulse limit at 20% duty cycle is ≥ 2.69 A
but at 80% duty cycle the pulse limit is ≥ 2.36 A. Use table 1 and
figure 8 to determine the real current limit, given the duty cycle
required for each application. Take care to do a careful thermal
solution or thermal shutdown will occur.
When the voltage at the soft start pin decays to a low level (VSSRESET , 235 mV typical), the Hiccup latch is cleared and the 10 μA
soft start pin current sink is turned off. The soft start pin will
resume charging the soft start capacitor with 20 μA and the voltage at the soft start pin will ramp upward. When the voltage at the
soft start pin exceeds the COMP release threshold (VSSRELEASE ,
330 mV typical), the low resistance pull-down at the COMP pin
will be turned off and the Error amplifier will force the voltage
at the COMP pin to ramp up quickly, and PWM switching will
begin. If the short circuit at the converter output remains, another
Hiccup cycle will occur. Hiccups will repeat until the short circuit
is removed or the converter is disabled. If the short circuit is
removed, the A8582 will soft start normally and the output voltage will be ramped to the required level as shown in figure 9.
Output Short Circuit (Hiccup Mode) Protection
Hiccup mode protects the A8582 when the load is either too high
or when the output of the converter is shorted to ground. When
the voltage at the FB pin is below the Hiccup Enable Threshold (VHICEN , nominally 625 mV), Hiccup mode protection is
enabled. When the voltage at the FB pin is above the Hiccup
Short removed
C2
VSS
330 mV
VOUT
235 mV
C1
VCOMP
C3
≈ 6.5 A
IL
C4
t
Figure 9. Hiccup mode operation and recovery ; shows VSS (ch1, 200 mV/div.),
VOUT (ch2, 2 V/div.), VCOMP (ch3, 1 V/div.), IL (ch4, 5 A/div.), t = 500 μs/div.
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15
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
Application Information
Design and Component Selection
RFSET = 26730 – 1.8
fSW
Setting the Output Voltage (VOUT, RFB1, RFB2)
The output voltage of the A8582 is determined by connecting
a resistor divider from the output node (VOUT) to the FB pin,
as shown in figure 10. There are trade-offs when choosing the
value of the feedback resisters. If the series combination (RFB1
+ RFB2) is relatively low, the light load efficiency of the regulator will be reduced. So to maximize the efficiency, it is best to
choose high values for the resistors. On the other hand, if the parallel combination (RFB1 // RFB2) is too high, then the regulator
may be susceptible to noise coupling into the FB pin. In general,
the feedback resisters must satisfy the ratio shown in equation 1
to produce a required output voltage.
RFB1
VOUT
RFB2 = 0.8 V – 1
When the PWM switching frequency is chosen, the designer
should be aware of the minimum controllable PWM on-time,
tON(MIN) of the A8582. If the system required on-time is less than
the A8582 minimum controllable on-time, then switch node jitter
will occur, and the output voltage will have increased ripple or
oscillations. The PWM switching frequency should be calculated
using equation 3, where VOUT is the output voltage, tON(MIN) is
the minimum controllable on-time of the A8582 (worst case of
Table 2. Recommended Feedback Resistor Values
VOUT
(V)
RFB1
VOUT to FB pin
(kΩ)
RFB2
FB pin to GND
(kΩ)
1.2
6.04
12.1
1.5
7.50
8.45
1.8
9.09
7.15
2.5
12.4
5.76
3.3
16.5
5.23
5.0
24.9
4.75
7.0
34.8
4.53
8.0
40.2
4.42
9.6
47.5
4.32
(1)
Table 2 shows the most common output voltages and recommended feedback resistor values, assuming less than 0.2% efficiency loss at light load of 100 mA and a parallel combination of
4 kΩ presented to the FB pin. For optimal system accuracy, it is
recommended that the feedback resistors have ≤1% tolerances.
PWM Switching Frequency (RFSET)
VOUT
A8582
RFB1
FB
RFB2
2400
PWM Switching Frequency, fSW (kHz)
The PWM switching frequency is set by connecting a resistor
from the FSET pin to ground. Figure 11 is a graph showing the
relationship between the typical switching frequency (y axis)
and the FSET resistor, 1/RFSET (x axis). For a given switching
frequency (fSW), the FSET resistor can be calculated using equation 2, where fSW is in kHz and RFSET is in kΩ.
(2)
2200
2000
1800
1600
1400
1200
1000
800
600
400
200
0.01
0.02
0.03
0.04
0.05
0.06
0.07
0.08
0.09
0.10
0.11
1/ RFSET Resistance, 1/RFSET (kΩ)
Figure 10. Connecting the feedback divider
Figure 11. PWM switching frequency versus 1/RFSET
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Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
100 ns), and VIN(MAX) is the maximum required operational input
voltage to the A8582 (not the peak surge voltage).
fSW <
VOUT
tON(MIN) × VIN(MAX)
(3)
If the A8582 synchronization function is employed, the base
switching frequency should be chosen such that jitter will not
result at the maximum synchronized switching frequency according to equation 3, that is, 1.5 × fSW < fSW calculated by equation 2.
Output Inductor (LO)
The value of the output inductor (LO) is usually calculated to set
a particular peak-to-peak ripple current in the inductor. However,
the inductor physical size and cost will be directly proportional
to the peak current or saturation specification. There are tradeoffs
among: peak-to-peak ripple current, system efficiency, transient
response, and cost. If the peak-to-peak inductor ripple is chosen
to be relatively high, then the inductor value will be low, the system efficiency will be reduced, the transient response will be fast,
the inductor physical size will be small, and the cost reduced. If
the peak-to-peak inductor ripple is chosen to be relatively low,
then the inductor value will be high, the system efficiency will be
higher, the transient response will be slow, the inductor physical
size will be larger, and the cost will be increased.
Equation 4 can be used to estimate the inductor value, given a
particular peak-to-peak ripple current (ΔIL ), input voltage (VIN ),
output voltage (VOUT), and switching frequency (fSW). The reference designs in this datasheet use a peak-to-peak ripple current of
25% of the 2.0 A, DC rating of the A8582, or 0.5 APP .
V
LO ≥ f OUT
SW × ∆IL
V
1 – VOUT
IN
(4)
If the preceding equation yields an inductor value that is not a
standard value, the next higher available value should be used.
After choosing a standard inductor value, equation 5 should be
used to make sure the A8582 slope compensation is adequate.
In this equation VIN(MIN) is the minimum required input voltage,
VOUT is the output voltage, fSW is the switching frequency, and
Vf is the forward voltage of the asynchronous Schottky diode.
LO ≥ 1.3 ×
VOUT + Vf
fSW
1–
0.18 × (VIN(MIN)+ Vf )
VOUT + Vf
(5)
Ideally, the rated saturation current of the inductor should be
higher than the maximum current capability of the A8582 at
the expected duty cycle. Unfortunately this usually results in
a physically larger, more costly inductor. At a minimum, the
saturation current of the inductor should support the DC rating
of the A8582 (2.0 A), plus ½ of the inductor peak-to-peak ripple
current (usually 0.5 APP ), the capacitive startup current (ICO ),
and some margin for component, frequency, and voltage tolerances. For example, an inductor with a 2.7 A rating allows 2.0 A
of load current, 0.25 APEAK of ripple current, 0.25 A of capacitive startup current (ICO ), along with a 20% frequency decrease,
a 20% inductance decrease, and a 10% input voltage increase (at
5.0 VOUT , 12 VIN , 2 MHz ).
After an inductor is chosen, it should be tested during output
short circuit conditions. The inductor current should be monitored
using a current probe. A good design should ensure the inductor
or the regulator are not damaged when the output is shorted to
GND at maximum input voltage and the highest expected ambient temperature
Output Capacitors (COUT)
The output capacitors filter the output voltage to provide an
acceptable level of ripple voltage and they store energy to help
maintain voltage regulation during a load transient. The voltage
rating of the output capacitors must support the output voltage
with sufficient design margin.
The output voltage ripple (ΔVOUT ) is a function of the output
capacitor parameters: ESRCO , ESLCO , and CO , as follows:
ΔVOUT = ΔVESR + ΔVESL + ΔVCO
(6)
It is commonly known that, for a constant load on the regulator, the current in the output inductor is equal to the DC output
current plus ΔIL . Therefore, using Kirchoff’s Current law, it can
be shown that the current in the output capacitors is equal to the
ripple current in the output inductor, or IC = ΔIL . Knowing this,
we can determine the first term in equation 6:
ΔVESR = ΔIL × ESRCO
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(7)
17
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
To calculate the second term in equation 6, ΔVESL , we must
determine the slope of the output inductor current, di/dt, which is
(VIN – VOUT) / LO:
∆VESL = LO
di
V –V
= ESLCO × IN OUT
dt
LO
(8)
To calculate the third term in equation 6, we must understand
that, over a single PWM cycle, the amount of charge into the
output capacitors must equal the amount of charge out of the
capacitors, or the capacitor output voltages would drift. What this
means is the output inductor current (ΔIL) flows in and out of the
output capacitor and is centered at 0 A, as shown in figure 12.
For any capacitor, the voltage is:
∆VCO ≥
1
COUT
i × dt
In this case, the integral term can be graphically calculated by
examining the 2 areas, A1 and A2, shown in figure 12:
∆I
DTS ∆IL DTS
A1 = 1 × L ×
=
2
2
2
8
∆I
(1 –D)TS
∆IL TS
∆IL DTS
A2 = 1 × L ×
=
–
2
2
2
8
8
∆I T
i × dt = A1 + A2 = L S
8
Substituting this into the equation for ΔVCO results in:
TS
ICO (A)
DTS
(1 – D)TS
∆IL / 2
0
A1
DTS /2
A2
[(1 – D)TS ]/2
–∆IL / 2
Time
Figure 12. Output capacitor current waveform
∆VCO =
∆IL TS
∆IL
=
8 COUT
8 fSW COUT
(9)
Combining equations 7, 8, and 9 results in an expression for the
total output voltage ripple:
∆VOUT = ∆IL× ESRCO +
∆IL
VIN – VOUT
× ESLCO +
8 fSW COUT
LO
(10)
The type of output capacitors will determine which terms of
equation 10 are dominant.
For ceramic output capacitors the ESR and ESL are extremely
low, so the output voltage ripple will be dominated by the third
term of equation 10:
∆IL
∆VOUT =
(10a)
8 fSW COUT
To reduce the voltage ripple of a design using ceramic output
capacitors, simply: increase the total capacitance, reduce the
inductor current ripple (that is, increase the inductor value), or
increase the switching frequency.
For electrolytic output capacitors the value of capacitance will be
relatively high, so the third term in equation 10 will be minimized
and the output voltage ripple will be determined primarily by the
first two terms of equation 10:
V – VOUT
∆VOUT = ∆IL× ESRCO + IN
× ESLCO
(10b)
LO
To reduce the voltage ripple of a design using electrolytic output
capacitors, simply: decrease the equivalent ESR and ESL by
using a high(er) quality capacitor, and/or add more capacitors in
parallel, or reduce the inductor current ripple (that is, increase
the inductor value). The ESR of some electrolytic capacitors
can be quite high, so Allegro recommends choosing a quality
capacitor that clearly documents the ESR or the total impedance
in the datasheet. Also, the ESR of electrolytic capacitors usually increases significantly at cold ambient, which increases the
output voltage ripple and, in many cases, reduces the stability of
the system.
To reduce the output voltage ripple and save PCB area, a design
could combine both ceramic and electrolytic capacitors in parallel. If this is done, the ceramic capacitors should be placed and
grounded as close as possible to the load to be most effective. AC
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Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
ripple voltage measurements should be made differentially across
the ceramic capacitors with a very short ground lead.
The transient response of the A8582 depends on the number and
type of output capacitors. In general, minimizing the ESR of the
output capacitance will result in a better transient response. The
ESR can be minimized by simply: adding more capacitors in
parallel, or by using higher quality capacitors. At the instant of a
fast load transient (di/dt), the output voltage will change by the
amount:
∆VOUT = ∆ILOAD × ESRCO + di ESLCO
(11)
dt
After the load transient occurs, the output voltage will deviate
for a short time depending on the system bandwidth, the output
inductor value, and output capacitance. After a short delay, the
Error amplifier will bring the output voltage back to its nominal
value. The speed at which the Error amplifier brings the output
voltage back to its set point will depend mainly on the closedloop bandwidth of the system. A higher bandwidth usually results
in a shorter time to return to the nominal voltage. However, a
higher bandwidth system may be more difficult to obtain acceptable gain and phase margins. Selection of the compensation
components (RZ, CZ, CP) are discussed in more detail in the
Compensation Components section of this datasheet.
Input Capacitors (CIN)
Three factors should be considered when choosing the input
capacitors. First, they must be chosen to support the maximum
expected input voltage with adequate design margin. Second,
their rms current rating must be higher than the expected rms
input current to the regulator. Third, they must have enough
capacitance and a low enough ESR to limit the input voltage
dV/dt to something much less than the hysteresis of the UVLO
circuitry (nominally 400 mV for the A8582) at maximum loading
and minimum input voltage.
The input capacitors must deliver the rms current according to
equation 12, where the duty cycle, D ≈ (VOUT + Vf ) / (VIN + Vf )
and Vf is the forward voltage of the asynchronous diode (D1 in
figure 1):
Irms = IO D×(1– D)
(12)
duty cycle (D) on the x axis and determine the input/output
current multiplier on the y axis. For example, at a 20% duty
cycle, the input/output current multiplier is 0.400. Therefore,
if the regulator is delivering 2.0 A of steady-state load current,
the input capacitor(s) must support 0.400 × 2.0 A or 0.8 Arms . A
single capacitor may support the rms input current requirement or
several capacitors may have to be paralleled. Ceramic capacitors
can deliver quite a bit of current but their total capacitance will be
relatively low. For example, a 4.7 μF, 16 V, 1206, X7R ceramic
capacitor can easily deliver 3 to 4 Arms .
Electrolytic capacitors can typically deliver 100 to 500 mArms of
current so 2 or 3 of these may be required to support the ripple
current. Electrolytic capacitors will typically offer much more
capacitance than the same quantity of ceramic capacitors. So,
electrolytic capacitors are typically able to provide more current
over extended periods of time where VIN would otherwise droop.
However, ceramic capacitors have very low ESR and inductance,
so they are best for filtering the high frequency switching noise.
A good design will employ both types of capacitors with the
ceramic capacitors placed closest to the input pin of the A8582.
The input capacitors must limit the voltage deviations at the VIN
pin to something significantly less than the A8582 UVLO hysteresis during maximum load and minimum input voltage. Equation
13 allows us to calculate the minimum input capacitance:
IOUT × D × (1 – D )
CIN ≥
(13)
fSW(MIN) × (∆VIN(MIN) – IOUT × ESRCIN)
Where ΔVIN(MIN) is chosen to be much less than the hysteresis of the VIN UVLO comparator (ΔVIN(MIN) ≤ 100 mV is
Irms / IOUT (A)
A8582
0.55
0.50
0.45
0.40
0.35
0.30
0.25
0.20
0.15
0.10
0.05
0
0
10
20
30
40
50
60
70
80
90
100
Duty Cycle, D (%)
Figure 13 shows the normalized input capacitor rms current
versus duty cycle. To use this graph, simply find the operational
Figure 13. Normalized input capacitor ripple current versus duty cycle
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Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
recommended), fSW(MIN) is the lowest expected PWM frequency, and ESRCIN is the equivalent series resistance of the
input capacitor(s).
If we choose ceramic input capacitors (ESR < 5 mΩ), the IOUT ×
ESRCIN term can be neglected in equation 13. Also, the
D × (1 – D) term has an absolute maximum value of 0.25 at
50% duty cycle. So, for a conservative design, based on
IOUT = 2.0 A, fSW(MIN) = 1.6 MHz (2 MHz – 20%), D × (1 – D) =
0.25, and ΔVIN =100 mV:
CIN ≥
2.0 (A) × 0.25
= 3.1 μF
1.6 (MHz) × 100 (mV)
A good design should consider the DC-bias effect on a ceramic
capacitor: as the applied voltage approaches the rated value, the
capacitance value decreases. This effect is very pronounced with
the Y5V and Z5U temperature characteristic devices (as much as
90% reduction) so these types should be avoided. The X5R and
X7R type capacitors should be the primary choices due to their
stability versus both DC bias and temperature.
For all ceramic capacitors, the DC-bias effect is even more
pronounced on smaller case sizes, so a good design will use the
largest affordable case size (such as 1206 or 1210). Also, it is
advisable to select input capacitors with plenty of design margin
in the voltage rating, to accommodate the worst case transient
input voltage (for example, load dump as high as 40 V for automotive applications).
Asynchronous Diode (D1)
There are three requirements for the asynchronous diode. First,
the asynchronous diode must be able to withstand the regulator
input voltage when the high-side MOSFET is on. Therefore, the
design should have a diode with a reverse voltage rating ( Vr )
higher than the maximum expected input voltage (that is, the
surge voltage). Second, the forward voltage of the diode (Vf )
should be minimized or the regulator efficiency will suffer. Also,
if Vf is too high, the missing diode protection in the A8582 could
be falsely activated. A Schottky-type diode, which can maintain
a very low Vf when the converter output is shorted to ground at
the coldest ambient temperature, is highly recommended. Third,
the asynchronous diode must conduct the output current when
the high-side MOSFET is off. Therefore, the average forward
current rating of this diode (If(av) ) must be high enough to deliver
the load current according to equation 14, where D is the duty
cycle (VOUT + Vf ) / (VIN + Vf ) and IOUT(max) is the maximum
continuous ouput current of the regulator:
If(av) ≥ IOUT(max) (1 – D(min))
(14)
To save cost and PCB area, the designer might be tempted to use
a diode with a relatively low current rating and the smallest PCB
footprint. However, doing this usually results in a hotter diode
and lower system efficiency. For the asynchronous converter, the
majority of losses can occur in this diode. To optimize efficiency,
one should use a higher rated, physically larger diode. Also,
diodes with very high reverse voltage ratings usually have higher
forward voltages, which reduces system efficiency. Therefore, a
diode with the lowest possible reverse voltage rating should be
used. However, care should be taken to be sure this diode is not
destroyed during input voltage transients or surge events.
Bootstrap Capacitor (CBOOT)
A bootstrap capacitor must be connected between the BOOT and
SW pins to provide floating gate drive to the high-side MOSFET.
For most applications 100 nF is sufficient. This should be a
high-quality ceramic capacitor, such as an X5R or X7R, with a
voltage rating of at least 16 V. The A8582 incorporates a low-side
MOSFET to insure that the bootstrap capacitor is always charged,
even when the converter is lightly loaded.
Soft Start and Hiccup Mode Timing (CSS)
The soft start time of the A8582 is determined by the value of
the capacitance on the SS pin. When the A8582 is enabled, the
voltage at the SS pin will start from 0 V and will be charged by
the soft start current, ISSSU (nominally 20 μA). However, PWM
switching will not begin instantly because the voltage at the
SS pin must rise above the COMP release voltage, VSSRELEASE
(nominally 0.33 V). The soft start delay (tSSDELAY) can be calculated using equation 15:
0.33 (V)
tSSDELAY = CSS ×
ISSSU
(15)
If the A8582 is starting into a full load (nominally 2.0 A) and
the soft start time (tSS) is too fast, the pulse-by-pulse overcurrent threshold may be exceeded and Hiccup mode protection
triggered. This occurs because the total of the full load current,
the inductor ripple current, and the additional current required to
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Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
charge the output capacitors (ICO = CO × dVOUT /dtSS) is higher
than the pulse-by-pulse current threshold, as shown in figure 14.
This phenomena is more pronounced when using high value
electrolytic type output capacitors.
To avoid prematurely triggering hiccup mode the soft start
capacitor, CSS, should be calculated using the following formula:
20 (μA) × VOUT × COUT
CSS ≥
(16)
0.8 (V) × ICO
Where VOUT is the output voltage, COUT is the output capacitance, ICO is the amount of current allowed to charge the output
capacitance during soft start (Allegro recommends 0.125 A < ICO
< 0.375 A). Higher values of ICO result in faster soft start times.
However, lower values of ICO insure that Hiccup mode is not
falsely triggered as components vary.
Components can easily change due to initial tolerances, aging, or
temperature (output capacitance, soft start capacitance, soft start
charging currents, and so forth). Allegro recommends starting the
design with an ICO of 0.125 A and increasing it only if the soft
start time is too slow. If a non-standard capacitor value for CSS is
calculated, the next larger value should be used.
The output voltage ramp time, tSS , can be calculated by using
either of the following formulas:
tSS = VOUT ×
or
COUT
ICO
tSS = 0.8 (V) ×
(17a)
CSS
20 (μA)
(17b)
}
ILOAD
tSS
Time
Figure 14. Output capacitor current (ICO) during startup
Compensation Components (RZ, CZ, CP)
To compensate the system it is important to understand where the
buck power stage, load resistance, and output capacitance form
their poles and zeros in frequency. Also, it is important to understand that the compensated Error amplifier introduces a zero and
two more poles, and also where these should be placed to maximize system stability, provide a high bandwidth, and optimize the
transient response.
First, we will take a look at the power stage of the A8582, the
output capacitors, and the load resistance. This circuitry is commonly referred as the “control to output” transfer function. The
low frequency gain of this section depends on the COMP to
SW current gain (gmPOWER), and the value of the load resistor
(RLOAD). The DC gain of the control-to-output is:
GCO = gmPOWER × RLOAD
(18)
The control-to-output transfer function has a pole (fP1) formed by
the output capacitance (COUT) and load resistance (RLOAD) at:
1
fP1 =
(19)
2 × RLOAD × COUT
IOUT (A)
ILIM
When the A8582 is in Hiccup mode, the CSS capacitor is used as
a timing capacitor and sets the hiccup period. The SS pin charges
the CSS capacitor with ISSSU (nominally 20 μA) during a startup
attempt and discharges the CSS capacitor with ISSHIC (nominally
10 μA) between startup attempts. Because the ratio of the SS pin
currents is 2:1, the time between hiccups will be at least twice
as long as the startup time. Therefore, the effective duty-cycle of
the A8582 will be very low when the output is shorted to ground.
With such a low duty cycle, the junction temperature of the
A8582 will be maintained at an extremely low value, compared
to other short circuit protection techniques.
ICO
The control-to-output transfer function also has a zero (fZ1)
formed by the output capacitance (COUT) and its associated ESR:
1
fZ1 =
(20)
2 × ESR × COUT
For a design with very low-ESR type output capacitors (for example, ceramic or OSCON output capacitors), the ESR zero (fZ1 )
is usually at a high frequency, so it can be ignored. On the other
hand, if the ESR zero falls below or near the 0 dB crossover frequency of the system (such as with electrolytic output capacitors),
then it should be cancelled by the pole formed by the CP capacitor
and the RZ resistor (discussed and identified later as fP3).
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Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
A Bode plot of the control-to-output transfer function for figure 1
(VOUT = 3.3 V, RLOAD = 2.0 Ω) is shown in figure 15. The pole
at fP1 can be seen at 8.8 kHz, while the ESR zero, fZ1 , occurs at
a very high frequency, 5.9 MHz (this is typical for a design using
ceramic output capacitors).
Next, we will take a look at the feedback resistor divider, (RFB1
and RFB2), the Error amplifier (gm), and its compensation network RZ/CZ/CP. It greatly simplifies the transfer function derivation if RO >> RZ, and CZ >> CP. In most cases, RO > 2 MΩ,
1 kΩ < RZ < 50 kΩ, 220 pF < CZ < 47 nF, and CP <100 pF, so
the following analysis should be very accurate. The low frequency gain of the control section (GC) is formed by the feedback
resistor divider and the Error amplifier. It can be calculated using
equation 21, where VOUT is the output voltage, VFB is the reference voltage (0.8 V), gm is the Error amplifier transconductance
(750 μA / V), and RO is the Error amplifier output impedance
(AVOL /gm):
GC =
VFB
= V
× gm × RO
OUT
The transfer function of the compensated Error amplifier has a
(very) low frequency pole (fP2) dominated by the output Error
amplifier output impedance (RO) and the CZ compensation
capacitor:
fP2 =
fP1 = 8.8 kHz
Gain (dB)
GCO = 24.8 dB
0
fZ1 = 5.9 MHz
Phase (°)
90
0
102
103
104
105
Frequency (Hz)
1
2 × R Z × CZ
(23)
106
Figure 15. Control-to-output Bode plot for circuit in figure 1
1
2 × RZ × C P
(24)
A Bode plot of the Error amplifier and its compensation network
is shown in figure 16. fP2, fP3, and fZ2 are indicated on the gain
(magnitude) plot. Notice that the zero (fZ2 at 13.2 kHz) has been
placed so that it is in the vicinity of the pole at fP1 (8.8 kHz) previously shown in the control-to-output Bode plot, figure 15.
-25
-90
101
(22)
The transfer function of the compensated Error amplifier also has
a low frequency zero (fZ2) dominated by the RZ resistor and the
CZ capacitor:
fP3 =
-50
180
1
2 × RO × CZ
Lastly, the transfer function of the compensated Error amplifier
has a higher frequency pole (fP3) dominated by the RZ resistor
and the CP capacitor:
75
25
(21)
VFB
= V
× AVOL
OUT
fZ2 =
50
RFB2
g
R
RFB1 + RFB2 × m × O
107
Finally, we take a look at the combined Bode plot of both the
control-to-output and the compensated Error amplifier in figure 17. Careful examination of this plot shows that the magnitude
and phase of the entire system (red curve) are simply the sum
of the Error amplifier response (blue curve, figure 16) and the
control-to-output response (green curve, figure 15). As shown in
figure 17, the bandwidth of this system is 135 kHz and the phase
margin is approximately 82 degrees.
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Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
A Generalized Tuning Procedure
1) Choose the system bandwidth, fC , the frequency at which the
magnitude of the gain will cross 0 dB. Recommended values for
fC based on the PWM switching frequency are: fSW /20 < fC <
fSW /10. A higher value of fC will generally provide a better transient response, while a lower value of fC will be easier to obtain
higher gain and phase margins.
2) Calculate the RZ resistor value to set the required system
bandwidth (fC):
V
2 × × COUT
RZ = fC × OUT ×
(25)
VFB
gmPOWER × gm
3) Determine the frequency of the pole (fP1) formed by COUT and
RLOAD by using equation 19 (repeated here):
1
fP1 =
2 × RLOAD × COUT
4) Calculate the CZ capacitor value by setting fZ2 at 1.5 × fP1:
1
CZ =
(26)
2 × × RZ × 1.5 × fP1
5) Calculate the frequency of the ESR zero (fZ1) formed by the
output capacitor(s) by using equation 20 (repeated here):
1
fZ1 =
(20)
2 × ESR × COUT
5a) If fZ1 is at least 1 decade higher than the target crossover frequency (fC) then fZ1 can be ignored. This is usually the case
for a design using ceramic output capacitors. Use equation 24
to calculate the value of CP by setting fP3 to either 10 × fC or
fSW / 2, whichever is higher.
5b) On the other hand, if fZ1 is near or below the target crossover
frequency (fC) then use equation 24 to calculate the value of
CP by setting fP3 equal to fZ1. This is usually the case for a
design using high ESR electrolytic output capacitors.
75
75
25
fP2 = 183 Hz
0
fZ2 = 13.2 kHz
-25
fP3 = 1.0 MHz
0
Combined
-50
180
Phase (°)
Phase (°)
25
-25
-50
180
90
0
-90
101
fC = 135 kHz
50
Gain (dB)
Gain (dB)
50
GC = 45.6 dB
102
103
104
105
Frequency (Hz)
Figure 16. Compensated Error amplifier Bode plot
106
107
90
Phase Margin = 82°
Combined
0
-90
101
102
103
104
105
Frequency (Hz)
106
107
Figure 17. Bode plot for the complete system (combined = red curve)
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Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
A Simple PSpice® Model for the A8582
Show in figure 18 is a very simple, first-order model for a current
mode buck converter. This model allows a designer to easily
modify the Error amplifier compensation, produce the Bode
plot, and estimate the gain and phase margins. It should shorten
the design time by allowing the designer to quickly examine the
effects and trade-offs of modifying the system variables.
In the PSpice model, the transconductance Error amplifier is
modelled by the GEA block with a gain of gm . Its output impedance, RO , is calculated as AVOL/gm (nominally 1.06 MΩ for the
A8582). The compensation components of interest are Rz, Cz,
and Cp shown at the COMP node. The PWM modulator and current control loop are simply modelled as the COMP to SW gain,
gmPOWER, documented in the electrical characteristics of this
datasheet. RLOAD is the load resistance and COUT is the output
capacitance with its equivalent ESR.
The component labelled Lac (10 GH) is used to maintain a closed
loop so PSpice can perform a DC bias point calculation, yet
effectively “break” the loop for AC analysis. Also, the compo-
nents labelled Cac (10 GF) and source V2 are used to inject a
1 V, AC signal for frequency response analysis. This model will
predict the magnitude of the gain and 0 dB crossover frequency
(fC) fairly accurately, provided that fSW / 20 < fC < fSW / 10. It
will be optimistic when predicting the phase margin because the
the PWM current control is approximated as a simple gain. The
designer should try to obtain at least 60 degrees of phase margin
with the model and then verify the bandwidth and gain/phase
margins with a network analyzer on the actual circuit.
To produce the control-to-output Bode plot use:
dB(V(Vout)/V(VC)) and P(V(Vout)/V(VC))
To produce the Bode plot of the error amplifier, its compensation,
and the feedback resistor divider use:
dB(V(COMP)/V(Vout)) and P(V(COMP)/V(Vout))
To produce the overall system Bode plot use:
dB(V(COMP)/V(VC)) and P(V(COMP)/V(VC))
GEA
GAIN = {gm}
FB
VREF
0.8V
RFB1
16.5K
0
+
-
RFB2
5.23K
COMP
0
0
PARAMETERS:
AVOL = 794
Ro
{AVOL/gm}
0
1
Rz
15.4K
Cz
820pF
IC = 0
Lac
10GH
VC
2
Cp
10pF
Cac
10GF
GCL
GAIN = {gm_power}
+
-
0
0
V2
1Vac
0Vdc
0
gm = 750u
Vout
0
Rload
2.0
0
0
Cout
9uF
IC = 0
ESR
3m
0
gm_power = 2.85
Figure 18. A simple PSpice model for the A8582 current mode buck converter
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Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
Power Dissipation and Thermal Calculations
PCOND = I 2rms(FET) × RDS(on)HS
The power dissipated in the A8582 is the sum of the power dissipated from the VIN supply current (PIN), the power dissipated due
to the switching of the internal power MOSFET (PSW), the power
dissipated by the internal gate driver (PDRIVER), and the power
dissipated due to the rms current being conducted by the internal
MOSFET (PCOND).
The power dissipated from the VIN supply current can be calculated using equation 27, where VIN is the input voltage and IQ
is the input quiescent current drawn by the A8582 (nominally
3 mA):
PIN = VIN × IQ + QG × fSW × (VIN – VGS)
(27)
The power dissipated by the internal high-side MOSFET while
it is switching can be calculated using equation 28, where VIN is
the input voltage, IOUT is the regulator output current, fSW is the
PWM switching frequency, and tr and tf are the rise and fall times
measured at the SW node. The exact rise and fall times at the SW
node will depend on the external components and PCB layout, so
each design should be measured at full load. Approximate values
for both tr and tf range from 5 ns to 10 ns.
PSW =
VIN × IOUT× (tr + tf) × fSW
2
(28)
The power dissipated by the internal gate driver can be calculated
using equation 29, where VGS is the internal gate drive voltage
(nominally 5 V), QG is the total gate charge to get to VGS (typically about 4 nC), and fSW is the switching frequency.
PDRIVER = QG × VGS × fSW
(29)
The power dissipated by the internal high-side MOSFET while
it is conducting can be calculated using equation 30, where IOUT
is the regulator output current, ΔIL is the peak-to-peak inductor ripple current, RDS(on)HS is the drain-to-source on-resistance
of the high-side MOSFET, and Vf is the forward voltage of the
asynchronous diode, D1.
=
VOUT + Vf
VIN + Vf
× I 2OUT +
∆I 2L
× RDS(on)HS
12
(30)
The RDS(on) of the high-side MOSFET will have some part-topart tolerance plus an increase from self-heating and elevated
ambient temperatures. A conservative design should accomodate
an RDS(on) with at least a 25% initial tolerance plus 0.4% / °C
increase due to temperature.
Finally, the total power dissipated (PTOT) is the sum of the previous four equations:
PTOT = PIN + PSW + PDRIVER + PCOND
(31)
The average junction temperature can be calculated with equation
32, where PTOT is the total power dissipated, RθJA is the junctionto-ambient thermal resistance (34 °C/W on a 4-layer PCB), and
TA is the ambient temperature:
TJ = PTOT × RθJA + TA
(32)
The maximum junction temperature will be dependent on how
efficiently heat can be transferred from the PCB to ambient air.
The thermal pad on the bottom of the IC should be connected
to a at least one ground plane using multiple vias for optimum
performance. A small amount of airflow can improve the thermal
performance considerably.
As with any regulator, there are limits to the amount of power
that can be delivered and heat that can be dissipated before risking thermal shutdown. There are tradeoffs between ambient operating temperature, input voltage, output voltage, output current,
switching frequency, PCB thermal resistance, and airflow.
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25
A8582
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
PCB Component Placement and Routing
A good PCB layout is critical if the regulator is to provide clean,
stable output voltages. Follow these guidelines to insure good
PCB layout. Figure 19(a) shows an example component placment
and routing. Figure 19(b) shows the three critical current loops
that should be minimized and connected by relatively wide traces.
6) To have the highest output voltage accuracy, the regulation
sense trace (from VOUT to RFB1) should be connected as close
as possible to the load.
7) For optimal system reliability, its best to have two independent
traces for regulation (FB, RFB1, RFB2) and overvoltage protection (FBX, RS1, RS2).
1) By far, the highest di/dt occurs at the instant the upper FET
turns on and the asynchronous diode (D1) undergoes reverse
recovery. The ceramic input capacitors (CIN) must deliver this
high frequency current. Therefore, the loop from the ceramic
input capacitors through the upper FET and asynchronous diode
to ground should be minimized. Ideally this connection is made
on both the top (component) layer and via the ground plane.
8) Place the frequency setting resistor (RFSET) as close as possible to the FSET pin (pin 8). Place a via to the GND plane as
close as possible to the resistor solder pad.
2) When the upper FET is on, current flows from the input supply/capacitors, through the upper FET, into the load via the output
inductor, and back to ground. This loop should be minimized and
have relatively wide traces. Ideally this connection is made on
both the top (component) layer and via the ground plane.
10) Place the soft start capacitor (CSS) as close as possible to the
SS pin (pin 4). Place a via to the GND plane as close as possible
to this component.
3) When the upper FET is off, “free-wheeling” current flows
from ground through the asynchronous diode, into the load via
the output inductor, and back to ground. This loop should be
minimized and have relatively wide traces. Ideally this connection is made on both the top (component) layer and via the
ground plane.
4) The voltage on the SW node (pins 15 and 16) transitions from
0 V to VIN very quickly and is the root cause of many noise
issues. Its best to place the asynchronous diode and output inductor close to the A8582 to minimize the size of the SW polygon.
Also, keep low level analog signals (like FB, FBX, COMP, and
FSET) away from the SW polygon.
5) Place the feedback resistor divider (RFB1 and RFB2) very
close to the FB pin (pin 9). Place the overvoltage sense resistor divider (RS1 and RS2) very close to the FBX pin (pin 10).
Ground both of these resistor dividers as close as possible to the
A8582 and to each other.
9) Place the compensation components (RZ, CZ, and CP) as close
as possible to the COMP pin (pin 11). Place vias to the GND
plane as close as possible to these components.
11) Place the boot strap capacitor (CBOOT) near the BOOT pin
(pin 14) and keep the routing to this capacitor as short as possible.
12) When routing the input and output ceramic capacitors (CIN,
COUT), use multiple vias to GND and place the vias as close as
possible to the component solder pads.
13) To minimize PCB losses and improve system efficiency, the
input (VIN) and output (VOUT) traces should be as wide as possible and be duplicated on multiple layers, if possible.
14) To improve thermal performance, place multiple vias to the
GND plane around the anode of the asynchronous diode.
15) The thermal pad under the A8582 must connect to the GND
plane using multiple vias; more vias will insure the lowest operating temperature and highest efficiency. For even better thermal
performance, the thermal via pattern can be extended beyond
(above and below) the footprint of the A8582 as shown in figure 22(a).
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
26
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
VOUT
A
CO1
LO
CO2
C
B
GND
CIN1 CIN2
U1
D1
C1
CSS
RFSET
RPU
A
+
SW
RFB1
VIN
CBOOT
CP
RFB2
E
RZ
D
CZ
RS1
RS2
CFBX
D
C
EN/SYNC
PCB outline
Ground circuit
Ground vias
Ground plane (opposite side)
Other circuits
Thermal vias
A. VOUT, VIN on multiple layers
B. SW polygon minimized
C. VOUT sense trace
D. Feedback and compensation components
E. Exposed pad under device soldered to GND
CO1, CO2 output capacitors
C1 input bulk capacitor
CIN1, CIN2 input ceramic capacitors
CBOOT boot capacitor
D1 Asynchronous diode
Figure 19(a). Example PCB component placement and routing
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
27
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
VOUT
LO
CO1
CO2
GND
CIN1 CIN2
D1
U1
C1
CSS
RFSET
RPU
+
SW
CBOOT
CP
RFB2
RFB1
VIN
RZ
CZ
RS1
RS2
CFBX
EN/SYNC
Upper FET on
Free-Wheeling
Reverse Recovery
Figure 19(b). Current loops that should be minimized and connected by wide traces
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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28
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
Application Circuit and Performance
V IN
1
2
3
CIN2
50 V
Empty
CIN1
3.3 μF
50 V
5
12
7
4
CSS
22 nF
8
11
RFSET
11.5 kΩ
SW
SW
VIN
VIN
VIN
CBOOT
100 nF
A8582
GND
GND
FBX
CO1
10 μF
16 V
RS1
16.5 kΩ
10
RS2
5.23 kΩ
CFBX
120 pF
SS
FSET
9
FB
COMP
CZ
820 pF
RFB1
16.5 kΩ
RFB2
5.23 kΩ
PAD
3.3V
RPU
2 kΩ
POK
6
POK
95
0.10
90
80
Deviation form VOUT at 50 mA (%)
VIN = 8 V
VIN = 12 V
VIN = 16 V
85
Efficiency (%)
Recommended Components
L1: 2.2 μH, 50 mΩ, 12.54 ASAT , 5.2 × 5.5 × 2.0 mm
Vishay: IHLP2020BZER2R2M01
D1: Schottky, 3 A, 40 V, SMA
Diodes, Inc.: B340A-13-F
CO1: 10 μF, 10%, 16 V, X7R, 1206
Murata: GRM31CR71C106KAC7L, or
TDK: C3216X7R1C106K
CIN1: 3.3 μF, 10% or 20%, 50 V, X5R or X7R, 1210
Murata: GRM55DR71H335MA01L, or
TDK: C3225X7R1H335M
VOUT
D1
3 A /40 V
SMA
14
BOOT
EN/SY NC
RZ
15.4 kΩ
CP
10 pF
LO
2.2 μH
16
15
75
70
65
60
0
–0.10
VIN = 8 V
VIN = 12 V
–0.20
VIN = 16 V
–0.30
55
500
1000
1500
2000
2500
–0.40
0
250
500
750
Efficiency versus Output Current, fSW = 2 MHz, and VOUT = 3.3 V
1250
1500
1750
2000
2250
2500
Load Regulation versus Output Current, fSW = 2 MHz, and VOUT = 3.3 V
0.50
60
0.40
48
0.30
Phase Margin = 60°
36
200
160
120
0.20
24
80
0.10
12
40
Gain (dB)
Line Regulation (%)
1000
Output Current, IOUT (mA)
Output Current, IOUT (mA)
0
–0.10
–0.20
0
0
Gain = 0 dB
–12
–24
–40
–80
Gain Margin = 14 dB
–0.30
–36
–120
–0.40
–48
–160
–0.50
–60
10–1
4
5
6
7
8
9
10
11
12
13
14
15
16
17
18
Input Voltage, VIN (V)
Line Regulation versus Output Current, fSW = 2 MHz, ILOAD = 2.0 A,
and VOUT = 3.3 V
100
101
102
Frequency (kHz)
Phase Margin (°)
0
fc = 140 kHz
50
–200
103
Bode Plot
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
29
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
VEN/SYNC
VEN/SYNC
C1
C1
VOUT
VOUT
C2
C2
VCOMP
VSS
C3
C3
VCOMP
VSS
C4
C4
t
Startup at 2.2 A; shows VEN/SYNC (ch1, 5 V/div.), VOUT (ch2, 1 V/div.), VCOMP
(ch3, 500 mV/div.), VSS (ch4, 1 V/div.), , t = 500 μs/div.
t
Shutdown at 2.2 A; shows VEN/SYNC (ch1, 5 V/div.), VOUT (ch2, 1 V/div.),
VCOMP (ch3, 500 mV/div.), VSS (ch4, 1 V/div.), , t = 20 μs/div.
C1
C1
VOUT
VOUT
VCOMP
VLX
C2
C2
C3
VLX
VCOMP
C3
IL
IL
C4
C4
t
t
PWM at 165 mA Load; shows VOUT (ch1, 1 V/div.), VLX (ch2, 5 V/div.),
VCOMP (ch3, 500 mV/div.), IL (ch4, 1 A/div.), t = 200 ns/div.
PWM at 2.2 A Load; shows VOUT (ch1, 1 V/div.), VLX (ch2, 5 V/div.), VCOMP
(ch3, 500 mV/div.), IL (ch4, 1 A/div.), t = 200 ns/div.
VOUT
C1
VOUT
C1
VSS
C2
VCOMP
VCOMP
C3
IOUT
C3
C4
IL
C2
t
0.4 to 1.5 A (1.1 A) Transient Response; shows VOUT (ch1, 100 mV/div.),
VCOMP (ch2, 200 mV/div.), IOUT (ch3, 1 A /div.), t = 20 μs/div.
t
Hiccup Mode Operation; shows VOUT (ch1, 500 mV/div.), VSS (ch2,
500 mV/div.), VCOMP (ch3, 1 V/div.), IL (ch4, 2 A/div.), t = 500 μs/div.
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
30
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
Package LP, 16-Pin TSSOP with Exposed Thermal Pad
0.45
5.00±0.10
16
0.65
16
8º
0º
0.20
0.09
1.70
B
3 NOM
4.40±0.10
3.00
6.40±0.20
6.10
0.60 ±0.15
A
1
1.00 REF
2
3 NOM
0.25 BSC
Branded Face
16X
SEATING
PLANE
0.10 C
0.30
0.19
SEATING PLANE
GAUGE PLANE
C
3.00
C
PCB Layout Reference View
For Reference Only; not for tooling use (reference MO-153 ABT)
Dimensions in millimeters
Dimensions exclusive of mold flash, gate burrs, and dambar protrusions
Exact case and lead configuration at supplier discretion within limits shown
1.20 MAX
0.65 BSC
1 2
0.15
0.00
A Terminal #1 mark area
B
Exposed thermal pad (bottom surface); dimensions may vary with device
C Reference land pattern layout (reference IPC7351
SOP65P640X110-17M);
All pads a minimum of 0.20 mm from all adjacent pads; adjust as
necessary to meet application process requirements and PCB layout
tolerances; when mounting on a multilayer PCB, thermal vias at the
exposed thermal pad land can improve thermal dissipation (reference
EIA/JEDEC Standard JESD51-5)
Allegro MicroSystems, Inc.
115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
1.508.853.5000; www.allegromicro.com
31
Wide Input Voltage, 2.4 MHz , 2.0 A
Asynchronous Buck Regulator
A8582
Revision History
Revision
Revision Date
Rev. 14
December 13, 2012
Description of Revision
Update Functional Block Diagram
PSpice® is a registered trademark of Cadence® Design Systems, Inc.
Copyright ©2011-2012, Allegro MicroSystems, Inc.
Allegro MicroSystems, Inc. reserves the right to make, from time to time, such departures from the detail specifications as may be required to permit improvements in the performance, reliability, or manufacturability of its products. Before placing an order, the user is cautioned to verify that the
information being relied upon is current.
Allegro’s products are not to be used in life support devices or systems, if a failure of an Allegro product can reasonably be expected to cause the
failure of that life support device or system, or to affect the safety or effectiveness of that device or system.
The information included herein is believed to be accurate and reliable. However, Allegro MicroSystems, Inc. assumes no responsibility for its use;
nor for any infringement of patents or other rights of third parties which may result from its use.
For the latest version of this document, visit our website:
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115 Northeast Cutoff
Worcester, Massachusetts 01615-0036 U.S.A.
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32