ETC AB-179

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®
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VIDEO OPERATIONAL AMPLIFIER
By Klaus Lehmann, Burr-Brown International GmbH
CURRENT OR VOLTAGE FEEDBACK?
THAT’S THE QUESTION HERE.
In analog technology, important advances are being made in
video operational amplifiers, both in processing and in
circuitry. The complementary bipolar processes used by
“Bell Laboratories” should be emphasized, which have vertically structured and approximately equal electric NPN and
PNP transistors on one substrate. These processes help to
create the basis for an effective complementary-symmetric
circuit technique, which is currently obtaining its best results
with the so-called “Diamond” structure. This structure plays
a key role in transimpedance or current-feedback amplifiers.
The following article is concerned exclusively with explaining these key topics and the new circuit designs.
R
C
1
+VIN
TA ∞
2
B
7
R2
3
VOUT
R23
FIGURE 2. OPA as Series Connection between TA and B.
The following discussion will deal first with practical circuits and then with theoretical models. It is based on publications [5] and [6], and essays [15] and [16], written at the
Institute for Semiconductor Technology of the Darmstadt
Technical College, with the assistance of Diplom Engineer
R.B. Steck.
VCC
I
+VIN
VOUT
CONVENTIONAL CIRCUIT TECHNIQUES
Figure 1 illustrates the conventional structure of a feedback
amplifier, including the wide-band operational amplifier
OPA discussed here. As can be seen in Figure 2, this kind of
OPA consists of a differential amplifier TA with highimpedance output (7) and an impedance converter B inserted
afterwards. The elements R and C function between them,
determining, among other things, the open-loop gain GOL,
slew rate, and bandwidth f–3dB. The circuit techniques of B
are not discussed here, but it could have a structure such as
the push-pull one shown in Figure 3.
I
VEE
FIGURE 3. Buffer with Push-pull Structure.
Figure 4 illustrates what is probably the simplest and most
well-known structure of the differential amplifier TA. It can
be seen in Figure 4 that the capacitor C can be charged with
2
R2
OPA
–SRMAX = I/C
I
C
1
VIN
+SRMAX = I/C
VCC
+VIN
3
1
1
7
2
VOUT
B
R23
2
3
VOUT
R2
R23
2I
VEE
FIGURE 1. Voltage-Feedback Operational Amplifier.
©
1993 Burr-Brown Corporation
FIGURE 4. Conventional Structure of a Differential
Amplifier.
AB-179
Printed in U.S.A. May, 1993
bias current I for rising and falling signals. The following
equations result from the positive slew rate SR + max. or
negative rate SR – max:
SR + max =
∆V
I
=
∆t
C
V
s
SR – max =
∆V
I
=
∆t
C
V
s
current-feedback amplifier. Differing input impedances can
result in adverse performance under certain conditions, but
these are not discussed here. The structure’s slew rate
performance shown in Figure 7 is worth noting. In theory,
the transfer current of C is not limited for a rising signal. In
practice, however, a current limitation appears at 10-20-fold
bias current, dependent upon the dimensioning in the current
mirror D1/Tr.4.
[]
[]
For sinusiodal signals, the largest signal change occurs at the
zero point. The following equation results:
VCC
(√ )( )
2
2π
f–3dB =
SRmax
VpO
+SRMAX = I/C
D1
–SRMAX = I/C
4
ˆ
[Hz]; VpO =ˆ V
C
B
7
Because SR + max. ≠ SR – max, a asymmetric limited
frequency response results during large signal modulation.
The cascaded circuit varitations also have current-limited
modulation behavior as shown in Figure 5.
+VIN
1
1
2
R23
2
VOUT
R2
2I
VCC
–SRMAX = I/C
3
FIGURE 6. Differential Amplifier with Signal Current
Mirror.
VCONST.
7
1
1
2
+SRMAX ≈ 10I/C
VCC
C
+VIN
I
VEE
+SRMAX = I/C
2I
3
B
4
I
R23
2
–SRMAX = I/C
D1
C
3
B
7
VOUT
R2
1
2I
I
+VIN
VEE
1
5
3
VOUT
I
FIGURE 5. Cascaded Differential Amplifier.
R23
2
I
R2
VEE
CURRENT FEEDBACK CONCEPT
FIGURE 7. The Current-Feedback Concept.
The introduction of the signal current mirror diode D1/
Transistor Tr.4, as shown in Figure 6, does not yet result in
improvement of the SR behavior, but will prove to be useful
later on. VBE is compensated by Tr.1 according to the
differential amplifier principle with transistor Tr.2. The VBE
compensation of Tr.1 is illustrated in Figure 7, along with
the previously inserted complementary emitter follower Tr.5.
The feedback loop can now be directly connected to the
emitter of Tr.1. The adaptors 1 and 2 are still the inputs of
the differential amplifier. Its impedance has now been
changed.
DIAMOND STRUCTURE
Asymmetric SR performance is caused by the asymmetric
circuit structure. A complementary-symmetric structure, as
shown in Figure 8, attains symmetric SR values at least 10
times higher than those of conventional structures with the
same quiescent current. The analogies to CMOS technology
are worth noting.
Parameters such as frequency response, modulation capability, and distortion can be attained with drastically reduced
quiescent current. This advantage plays an important role in
portable devices, simplifies power supply equipment, and
reduces inhouse heating, especially with respect to the development of increasingly small devices. The circuit shown
in Figure 8 without B, R2, and R23, is called a Diamond
1 = high impedance
2 = low impedance
The conventional voltage feedback has now become current
feedback. This structure is designated a transimpedance or
2
circuit, and is typical for a current-feedback amplifier. Processes such as correct transmission of coded color television
signals require low phase delays dependent upon the signal
modulation. Such phase delays can arise in circuits such as
the one shown in Figure 7 through the voltage-dependent
collector substrate capacity of the Tr.4.
The circuit in Figure 10, for example, delivers the following
as voltage feedback:
In the circuit shown in Figure 8, the varicap of the Tr.4 is
largely compensated through the varicap of the complementary Tr.8. The advantages of the current-feedback concepts,
however, are counterbalanced by several disadvantages.
Another problem is the size of the input voltage offset.
±1mV is typical in circuits with voltage-feedback differential amplifiers (Figure 6). Current-feedback circuits like the
ones shown in Figures 7 and 8 are typical at ±50mV.
Currently, there is no better serial “matching” between NPN
and PNP transistors. Various suggestions have attained small
offset voltages with circuit variations, but these resulted in
poorer transmission characteristics. Differing early voltages
in NPN and PNP transistors cause different mirroring in D1/
Tr.4 and D2/Tr.8 and thus bring about an output bias current
at point 7. This effect is a general weakness of the Diamond
structure.
GC100kHz = –32dB; GOL100kHz = +52dB
As current feedback, it delivers:
GC100kHz = +5dB; GOL100kHz = +58dB
Generally stated, applications do exist in which two highimpedance differential inputs are necessary. Poorer common-mode gain is a direct result of the unequal inputs.
VCC
+SRMAX ≈ 10I/C
D1
–SRMAX ≈ 10I/C
4
I
C
1
1
5
VOLTAGE FEEDBACK
WITH THE DIAMOND STRUCTURE
B
7
R23
2
In order to make the best of the advantages of both ideas, it
now makes sense to combine the Diamond structure and
voltage feedback together in one structure. The current
feedback shown in Figure 8 can be transferred to voltage
feedback illustrated in Figure 9 by inserting the buffer B2.
The desired combination would be attainable with a buffer
which was “ideal” in respect to its amplitude and phase
behavior. The amplitude behavior of the buffer shown in
Figure 3 (with real current source I) reaches this condition
with f–3dB ≈ 3.0 GHz (I = 1.9 mA). The phase delay continues
to be damaged inside the control loop through the lengthening of the signal delay time by approximately TDB2 ≈ 25ps.
The principle illustrated in Figure 9 is shown in more detail
in Figure 10. The contrasting results of PSPICE simulations
3
6
VOUT
+VIN
R2
7
8
I
D2
VEE
FIGURE 8. Current-Feedback Amplifier with Diamond
Structure.
VCC
+SRMAX ≈ 10I/C
D1
–SRMAX ≈ 10I/C
4
I
C
1
+VIN
1
5
R89
9
8
6
B1
7
B2
R23
2
VOUT
R2
7
8
I
D2
VEE
FIGURE 9. Voltage-Feedback Amplifier with Diamond Structure.
3
3
conducted with current and voltage feedback are summarized in Figure 11. All simulations were carried out with
fully equipped current sources. To give the user an overview, the f–3dB frequencies are illustrated in Figure 12 dependent upon the closed-loop gain for both feedback types. The
relatively low frequency response differences shown here do
not concur with the assertions in many publications about
current-feedback amplifiers. This discrepancy can be explained by the fact that these articles place an unrealistic
emphasis on the ability of the current-feedback concept—
they maintain that this concept alone brought about the
improved bandwidth. Improved technology and the Diamond structure play a more decisive role than the currentfeedback concept.
The first section of this article deals with the basics of
current and voltage-feedback video operational amplifiers
and their corresponding circuits. These practically oriented
considerations are then followed by extensive theoretical
analysis. The point of departure for the following models is
the performance of Diamond structure with open loop gain
GOL+ as shown in Figures 8 and 10. The measurement circuit
shown in Figure 13 is used for analysis. The results are
presented in Figure 14. The curves follow a simple model up
to f ≈ 300 MHz, formed by the parameters gm, R2, R, and
C. Simple modelling of the amplitude response with R and
C is not enough to describe the phase delay. Through a row
of additional, partially very small RC parts inside the control
loop, more phase delays result than would occur alone with
Buffer B2
VCC
I
I
I
R89
1
+VIN
8
2
9
7
3
VOUT
I
I
I
VEE
R2
R23
FIGURE 10. Developed Voltage-Feedback Amplifier.
Amplification (+GLC)
40
f–3dB
1.0
vf
20
(GHz)
0.8
cf
cf
vf
vf
vf
0
–20
–40
10
30
100
300M
1.0G
3.0G
cf
0.6
0.4
–CGL
16
(dB)
24 18 12
8
4
0.2
2
2
4
6
6
12 18 24
8
16 +CGL
(dB)
FIGURE 12. Bandwidth with Current and Voltage Feedback.
10G
Frequency (Hz)
FIGURE 11. Closed Gain Progress with Current and Voltage Feedback.
4
R and C. An additional phase delay with variation of the
amplitude response is attained through insertion of the delay
time TD. The following equation applies to the delay time of
a RC part:
to form the new current source gm’ (Figure 16). The voltage-feedback shown in Figure 16 varies from the currentfeedback model (Figure 17) only in the use (or non-use) of
the buffer in the feedback loop.
T = [arc tg (ωRC)]/ω.
For a better overview, the output conditions, various equations, and optimal operating conditions are summarized in
Table I. The derivations are described in detail [5].
As already mentioned, if ωRC remains sufficiently small, it
will be T ≈ RC. All of these small time constants are
summarized in the following model to the total delay time
TD. To be able to imagine this concretely, it is important to
remember that small time constants always turn the phase,
but do not necessarily damage the amplitude response considered here.
R
C
TD
3
–1
7
VOUT
R23
VIN
R
C
+VIN
1
2
C2
TA
B
7
1
gm
3
2
gm
R2
R89
VOUT
8
9
R23
R2
1F
FIGURE 15. Model of a Differential Amplifier Type.
10kΩ
gm = 2I/VT
gm’ = 1/(R2 – 1/gm)
G+OL = VOUT/V+IN = gm’ + R
FIGURE 13. Circuitry for Adjustment of G+OL.
R
C
TD
–1
7
60
Amplification (+GLC)
40
0.01
10
30
100
+VIN
1
3
VOUT
R23
g'm
+1
2
300
20
R2
R89
0
FIGURE 16. Model of the Voltage Feedback.
–20
–40
100
1M
10M
100M
1.0G
10G
Frequency (Hz)
R
FIGURE 14. Open-Loop Frequency Response.
+VIN
1
TD
–1
7
DELAY TIME MODEL
The circuits with voltage feedback as shown in Figures 4, 5,
6, and 9 are described with the model shown in Figure 15.
The signal inversion of the current mirror in Figures 6 and
9 is viewed in the output buffer with G = –1. The delay time
TD is inserted here. The dependence of the transconductance
gm on the quiescent current I varies according to circuit
structure. In the case shown in Figure 4, for example, gm =
I/2VΤ , while in the more interesting Figures 8 and 9, gm =
2I/VΤ. In the next formal step, the two controlled current
sources gm and the resistor R89 (see Figure 15) are combined
C
3
VOUT
R23
gm'
2
R2
FIGURE 17. Model of the Current Feedback.
5
OPTIMAL FREQUENCY
RESPONSE ADJUSTMENT
DIFFERENCES BETWEEN
CURRENT AND VOLTAGE FEEDBACK
The equation for calculation of |AN| and the graphic values
in Figure 18 describe the various operating conditions of the
four amplifier variations. Normally, the optimal adjustment
is installed. This has the following conditions:
GCLMAX is reached for R89 → 0 or R2 || R23 → 0, which is
practically unattainable with current feedback. Current feedback usually obtains lower values for GCLMAX with otherwise
comparable parameters. GOL0 is dependent upon GCLO. Amplifiers with adjustable GCLO, such as those for gain control,
simultaneously require correction of GOLO. Figure 19 shows
the frequency response differences between current and
voltage feedback according to the delay time model. In the
case of voltage feedback, a change in R2 means a change in
R89 as well. With current feedback, the gains GCLO and GOLO
are simultaneously adjusted to one another during changes
in R2, which can be an important advantage. Feedback
buffers (GCL+O = +1) have no inverting input. While the
disadvantage of the low impedance inverting input of the
current-feedback concept is not significant, here the low
circuit expenditure and quiescent current are important advantages.
C/gm = 2 kO TD
Internally compensated amplifiers use an integrated compensation capacitor C corresponding to the already mentioned condition for the poorest case i.e. the smallest closed
loop gain, for example kO = 1. This method is not well-suited
to the wide-band amplifiers discussed here. The slew rate
and large-signal frequency response are significantly reduced through the compensation capacitor. From this adjustment, deviating, i.e. larger, closed-loop gain leads to reduced frequency response. Amplifiers adjustable externally
with C allow optimal adjustment with kO, but do not significantly improve large-signal performance. Optimal adjustment with gm avoids these disadvantages. One possible
compromise is to insert amplifiers compensated internally
over gm for various closed-loop gain ranges (e.g. GCL =
1-3, 3-10, 10-30). Making the open-loop gain GOL externally
adjustable over gm for optimal frequency response is the
most effective solution. This can be adjusted with R89 using
voltage feedback and with R2/R236 using current feedback.
30
Amplification (+GLC)
20
30
Amplitude Response \AN\
20
0
R89
11.2
0.01
5.9
–4
25.6
–2
47.13
–1
74
21.5
74
–10
–30
10M
0
R89
|AN|opt.
12dB/Okt.
–10
–20
30M
100M
300M
3.0G
30M
100M
300M
1.0G
3.0G
10G
FIGURE 19. Model Frequency Response with Current and
Voltage Feedback.
470
1.0G
780MHz
640MHz
Frequency (Hz)
48
53
70
163
with internal
C-compensation
5dB/Okt.
f–3dB
f–3dB
45.1
–20
10
–30
10M
10
R2||R23
–12
10G
Frequency (Hz)
FIGURE 18. Adjustment of the Frequency Response.
DIMENSIONING
Inside the optimal adjustment range, the four amplifier
variations have a bandwidth f–3dB, which is dependent only
upon the additional delay time TD. The bandwidth corresponds to the model and is independent of the closed-loop
gain GCL (Figure 19). The deviations of the model frequency
response (Figure 19) from the simulated frequency response
(Figure 11) result from simplified modelling (see Figure 14).
This performance is not limited to the current feedback and
has become practically visible through technological advances. The first priority goes to achieving the shortest
possible delay time TD inside the control loop. The second
priority goes to small capacity C due to high slew rate and
wide large-signal frequency response, as well as higher, if
possible closed-loop gain GCLMAX.
6
FEEDBACK MODE
VOLTAGE FEEDBACK
OPERATION MODE
NON-INVERTING
CURRENT FEEDBACK
INVERTING
NON-INVERTING
INVERTING
MODEL CIRCUITRY
R
C
R
TD
3
–1
7
C
R
TD
VOUT
3
–1
7
C
R
TD
VOUT
7
–1
3
R23
gm'
1
R23
gm'
1
2
+1
R2
R23
gm'
1
–1
R23
2
R2
R2
R2
–VIN
–VIN
CLOSED-LOOP GAIN DC
+
GCLO
= 1 / k 0+ = 1 + R 23 / R 2
+
GCL
CLOSED-LOOP GAIN AC
OPEN-LOOP GAIN DC & AC
TRANSCONDUCTANCE
= VOUT /
+
GCLO
= 1 / k 0+ = 1 + R 23 / R 2
–
GCLO
= 1 / k 0– = 1 + R 23 / R 2
+
V IN
= VOUT /
–
GCL
+
GCL
–
V IN
g m = 2 I / V T ; T C = C / g′m
OPERATING
TRANSCONDUCTANCE
g′m = 1 / (R89 + 2 / g′m )
AGREEMENTS
+
+
+
–
V IN = V IN
; GCL = GCL
; GCLO = GCLO
= GCLO
+ 1; k 0 = k 0+ = k 0– / (k 0– + 1)
OUTPUT EQUATIONS
(V IN – V 2 ) / V 7 = jωC / g′m ; V 2 = k 0 ⋅ VOUT ; VOUT = V 7 ⋅ exp(– jωT D )
CLOSED-LOOP
GAIN COMPLEX
GCL = [k 0 – ωT C ⋅ sin(ωT D ) + jωT C ⋅ cos(ωT D )]–1/2
CLOSED-LOOP
GAIN (Amount)
GCL = [k 20 + (ωT C )2 – 2k 0ωT C ⋅ sin(ωT D )]–1/2
APPROXIMATION
sin(ωT D ) ≈ ωT D – (ωT D )3 / 6
AMOUNT OF NORMALIZED
CLOSED-LOOP GAIN
GCL = [k 20 + T C (T C – 2k 0 T D )ω 2 + k 0 T CT3D ⋅ ω 4 / 3]–1/2
AN =
GCL
GCLO
 T

T T3
= 1 + C2 (T C – 2k 0 T D )ω 2 + C D ω 4 
k0
3k 0


AMOUNT OF OPTIMIZED
CLOSED-LOOP GAIN
GCL OPT = [k 20 + k 0 T CT3D ⋅ ω 4 / 3]–1/2
AMOUNT OF NORMALIZED
OPTIMIZED CLOSED-LOOP
GAIN
A N OPT =
GCL OPT
GCLO
 C / g′m 3 4 
= 1 +
T Dω 
3k 0


–1/2
 2 4

= 1 + T D
⋅ ω4 
3


–1/2
I; R; C; TD
R89
BLOCK DIAGRAM
R89
8
+VIN
8
1
VFA
2
R2
R23
+
G OL0 = G CL0
RC
2TD
+VIN
1
VOUT
9
VOUT
9
3
VFA
2
–VIN R2
3
1
RC
2T D
2
R89
R89
2T
2T
≈ + D
GCLO ⋅ C
R89 ≈
CLO
R89
MAX ATTAINABLE OPTIMIZED
CLOSED-LOOP GAIN
+
GCLO
max =
CLO
2I ⋅ T D
VT ⋅ C
CLO
2T
2T D
+ 1)C
–
(GCLO
–
GCLO
max =
2I ⋅ T D
–1
VT ⋅ C
f–3dB = 0.176/TD
TABLE I. Summary of the Basis and Result of the Delay Time Model.
7
3
R23
+
G OL0 = G CL0
D
2T D
V
=
– T
–
(GCLO
+ 1)C
I
1
VOUT
CFA
R2
R23
–
G OL0 = (G CL0
+ 1)
2T
V
= + D – T
GCLO ⋅ C
I
METHOD OF APPROXIMATION
FOR GCLO ➞ 1
BANDWIDTH OF THE
OPTIMIZED RESPONSE
–1/2
GCLO
GCLO  C 
=
2  g′m  4πf ⋅ GOL
T C – 2k 0 T D = 0; T D =
ADJUSTMENT FOR OPTIMAL
FREQUENCY RESPONSE
–
–
GCL
= VOUT / V IN
g′m = 1 / (R 2 / / R 23 + 1 / g m )
CONDITIONS FOR OPTIMIZED
FREQUENCY RESPONSE
OPEN-LOOP GAIN DC FOR
OPTIMIZED FREQUENCY
RESPONSE
= VOUT /
–
GCLO
= 1 / k 0– = 1 + R 23 / R 2
+
V IN
–
GOL0
= g′m ⋅ R; GOL = g′m / ωC | f > 1MHz
ABBREVIATIONS
AMPLIFIER
CHARACTERISTIC VALUES
VOUT
3
gm'
1
2
2
+1
TD
VOUT
7
+VIN
+VIN
C
RC
2T D
2
–VIN R2
R 23 ≈
2T D
C
C
+
GCLO
max =
4I ⋅ T D
VT ⋅ C
3
R23
–
G OL0 = (G CL0
+ 1)
RC
2T D
D
D
VT
2T D
+
R 23 =
– GCLO
C
2I
VOUT
CFA
2T D
V
–
– (GCLO
+ 1) T
C
2I
C
2I
2T
2T D
R 23 ≈
C
C 4
4I ⋅ T D
–
–1
GCLO max =
VT ⋅ C
R 23 =
LITERATURE
[1] Friedrich, H.;
Elektronik 87,vol. 26
Das Zauberwort heißt Transimpedanz-Verstärker
(The Magic Word is Transimpedance Amplifiers)
[9] Nelson, D.;
US-Patent, No. 4 502 020; Feb. 26, ’85
[10] Nelson, D.;
US-Patent, No. 4 628 279, Dec. 9, ’86
[11] Nelson, D.;
US-Patent, No. 4 713 628; Dec. 15, ’87
[2] Goodenough, F.;
Electronic Design 87, April 16, pp. 59…
[12] Palmer, W.;
Electronics 88, Jan. 7,pp. 151…
Transimpedance amps: fast yet accurate
[3] Goodenough, F.;
Electronic Design 87, Oct. 29, pp. 67…
Exotic ICs put 200MHz signals, 15ns settling in
everyday jobs
[13] Saller, K.;
US-Patent, No. 4 639 685, Jan. 27, ’87
[4] Goodenough, F.;
Electronic Design 88, vol. 2, pp. 29…
A slew of new high performance op amps shatters
speed limits
[14] Shattock, R.;
Electronic Engineering 86, Mar., pp. 59…
A novel design approach to high frequency op amps
[5] Lehmann, K.;
Elektronik 80, vol. 24, pp. 101…
Berechnung des Frequenzganges von VideoOperationsverstärkern (Calculation of the Frequency
Response of Video Operational Amplifiers)
[15] Schwehr, S.; Sibrai, A.;
Studienarbeit p. 48, TH Darmstadt 88, Inst. f.
Halbleitertechnik Entwurf und Layouterstellung eines
Video-Operationsverstärkers in komplementärer
Bipolartechnologie. (Design and Layout of a Video
Operational Amplifier in Complementary Bipolar
Technology)
[6] Lehmann, K.;
Elektronik 80, vol. 26, pp. 81…
Schaltungstechnik von Video Operationsverstärkern
(Circuit Techniques with Video Operational
Amplifiers)
[16] Sibrai, A.;
Diplomarbeit D56, TH-Darmstadt 1/89,
Inst. f. Halbleitertechnik
Makromodellierung von Video-Operationsverstärkern
in komplementärer Bipolartechnologie (Macro Model
of Video Operational Amplifiers in Complementary
Bipolar Techonolgy)
[7] Moser, K.D.;
eee-Bauelemente 87,vol. 23, pp. 22…
Neuartige Operationsverstärker (New Kinds of
Operational Amplifiers)
[17] Yee, S.;
US-Patent, No. 3 418; Dec. 24, ’68
[8] Nelson, D.;
US-Patent, No. 4 358 739, Nov. 9,’82
8