ETC TPS61041

TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
LOW POWER DC/DC BOOST CONVERTER IN SOT-23 PACKAGE
FEATURES
D 1.8 V to 6 V Input Voltage Range
D Adjustable Output Voltage Range up to 28 V
D 400 mA (TPS61040) and 250 mA (TPS61041)
D
D
D
D
D
DESCRIPTION
The TPS61040/41 is a high-frequency boost converter
dedicated for small to medium LCD bias supply and
white LED backlight supplies. The device is ideal to
generate output voltages up to 28 V from a dual cell
NiMH/NiCd or a single cell Li-Ion battery. The part can
also be used to generate standard 3.3 V/5 V to 12 V
power conversions.
Internal Switch Current
Up to 1 MHz Switching Frequency
28 µA Typical No Load Quiescent Current
1 µA Typical Shutdown Current
Internal Softstart
Available in a Tiny 5-Pin SOT23 Package
The TPS61040/41 operates with a switching frequency
up to 1 MHz. This allows the use of small external
components using ceramic as well as tantalum output
capacitors. Together with the tiny SOT23 package, the
TPS61040/41 gives a very small overall solution size.
The TPS61040 has an internal 400 mA switch current
limit, while the TPS61041 has a 250 mA switch current
limit, offering lower output voltage ripple and allows the
use of a smaller form factor inductor for lower power
applications. The low quiescent current (typically
28 µA) together with an optimized control scheme,
allows device operation at very high efficiencies over
the entire load current range.
APPLICATIONS
D LCD Bias Supply
D White-LED Supply for LCD Backlights
D Digital Still Camera
D PDAs, Organizers and Handheld PCs
D Cellular Phones
D Internet Audio Player
D Standard 3.3 V/5 V to 12 V Conversion
TYPICAL APPLICATION
90
L1
10 µH
5 V
IN
SW
FB
D1
1
R1
VI = 5 V
86
84
CFF
CO
1 µF
3
VO = 18 V
88
VOUT
VIN to 28 V
Efficiency – %
VIN
1.8 V to 6.0 V
CIN
4.7 µF
EFFICIENCY
vs
OUTPUT CURRENT
VI = 3.6 V
82
80
VI = 2.4 V
78
76
4
EN
GND
2
74
R2
72
70
0.10
1
10
IO – Output Current – mA
100
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
Copyright  2002, Texas Instruments Incorporated
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
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1
TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
DBV PACKAGE
(TOP VIEW)
SW
1
GND
2
FB
3
5
VIN
4
EN
Ordering Information†
TA
SWITCH CURRENT LIMIT
SOT23 PACKAGE (DBV)
PACKAGE MARKING
400 mA
TPS61040DBV
PHOI
250 mA
TPS61041DBV
–40
40 to 85°C
PHPI
† The DBV package is available in tape & reel. Add “R” suffix (DBVR) to order quantities of 3000 parts.
functional block diagram
SW
Under Voltage
Lockout
Bias Supply
VIN
400 ns Min
Off Time
Error Comparator
–
FB
S
+
RS Latch
Logic
Gate
Driver
Power MOSFET
N-Channel
VREF = 1.233 V
R
Current Limit
6 µs Max
On Time
EN
+
_
RSENSE
Soft
Start
GND
Terminal Functions
TERMINAL
NAME
NO.
I/O
DESCRIPTION
I
Connect the inductor and the Schottky diode to this pin. This is the switch pin and is connected to the drain of the internal
power MOSFET.
SW
1
GND
2
FB
3
I
This is the feedback pin of the device. Connect this pin to the external voltage divider to program the desired output
voltage.
EN
4
I
This is the enable pin of the device. Pulling this pin to ground forces the device into shutdown mode reducing the supply
current to less than 1 µA. This pin should not be left floating and needs to be terminated.
VIN
5
I
Supply voltage pin
2
Ground
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TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
detailed description
operation
The TPS61040/41 operates with an input voltage range of 1.8 V to 6 V and can generate output voltages up
to 28 V. The device operates in a pulse frequency modulation (PFM) scheme with constant peak current control.
This control scheme maintains high efficiency over the entire load current range, and with a switching frequency
up to 1 MHz, the device enables the use of very small external components.
The converter monitors the output voltage, and as soon as the feedback voltage falls below the reference
voltage of typically 1.233 V, the internal switch turns on and the current ramps up. The switch turns off as soon
as the inductor current reaches the internally set peak current of typically 400 mA (TPS61040) or 250 mA
(TPS61041). Refer to the section peak current control for more information. The second criteria that turns off
the switch is the maximum on-time of 6 µs (typical). This is just to limit the maximum on-time of the converter
to cover for extreme conditions. As the switch is turned off the external Schottky diode is forward biased
delivering the current to the output. The switch remains off for a minimum of 400 ns (typical), or until the feedback
voltage drops below the reference voltage again. Using this PFM peak current control scheme the converter
operates in discontinuous conduction mode (DCM) where the switching frequency depends on the output
current, which results in very high efficiency over the entire load current range. This regulation scheme is
inherently stable, allowing a wider selection range for the inductor and output capacitor.
peak current control
The internal switch turns on until the inductor current reaches the typical dc current limit (ILIM) of 400 mA
(TPS61040) or 250 mA (TPS61042). Due to the internal propagation delay of typical 100 ns, the actual current
exceeds the dc current limit threshold by a small amount. The typical peak current limit can be calculated:
) Vin
100 ns
L
I
+ 400 mA ) Vin
100 ns for the TPS61040
peak(typ)
L
I
+ 250 mA ) Vin
100 ns for the TPS61041
peak(typ)
L
I
peak(typ)
+I
LIM
The higher the input voltage and the lower the inductor value, the greater the peak.
By selecting the TPS61040 or TPS61041, it is possible to tailor the design to the specific application current
limit requirements. A lower current limit supports applications requiring lower output power and allows the use
of an inductor with a lower current rating and a smaller form factor. A lower current limit usually has a lower output
voltage ripple as well.
softstart
All inductive step-up converters exhibit high inrush current during start-up if no special precaution is made. This
can cause voltage drops at the input rail during start up and may result in an unwanted or early system shut
down.
I
The TPS61040/41 limits this inrush current by increasing the current limit in two steps starting from LIM for
4
256 cycles to
I LIM
2
for the next 256 cycles, and then full current limit (refer to Figure 14).
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3
TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
detailed description (continued)
enable
Pulling the enable (EN) to ground shuts down the device reducing the shutdown current to 1 µA (typical). Since
there is a conductive path from the input to the output through the inductor and Schottky diode, the output
voltage is equal to the input voltage during shutdown. The enable pin needs to be terminated and should not
be left floating. Using a small external transistor disconnects the input from the output during shutdown as shown
in Figure 18.
undervoltage lockout
An undervoltage lockout prevents misoperation of the device at input voltages below typical 1.5 V. When the
input voltage is below the undervoltage threshold the main switch is turned off.
absolute maximum ratings over operating free-air temperature (unless otherwise noted)†
Supply voltages on pin VIN (see Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to 7 V
Voltages on pins EN, FB (see Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V to VIN + 0.3 V
Switch voltage on pin SW (see Note 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30 V
Continuous power dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . See Dissipation Rating Table
Operating junction temperature, TJ . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to 150°C
Storage temperature, TStg . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . –65°C to 150°C
Lead temperature (soldering 10 seconds) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 260°C
† Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and
functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not
implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
NOTE 1: All voltage values are with respect to network ground terminal.
DISSIPATION RATING TABLE
PACKAGE
TA ≤ 25°C
POWER RATING
DERATING FACTOR
ABOVE TA = 25°C
TA = 70°C
POWER RATING
TA = 85°C
POWER RATING
DBV
357 mW
3.5 mW/°C
192 mW
140 mW
NOTE: The thermal resistance junction to ambient of the 5-pin SOT23 is 250°C/W.
recommended operating conditions
MIN
Input voltage range, Vin
TYP
1.8
Output voltage range, VOUT
Inductor (see Note 2), L
2.2
V
V
1
MHz
µH
µF
µF
1
Operating ambient temperature, TA
Operating junction temperature, TJ
NOTE 2: Refer to application section for further information
4
6
4.7
Output capacitor (see Note 2), COUT
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UNIT
28
10
Switching frequency (see Note 2), f
Input capacitor (see Note 2), Cin
MAX
–40
85
°C
–40
125
°C
TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
electrical characteristics, Vin = 2.4 V, EN = VIN, TA = –40°C to 85°C, typical values are at TA = 25°C
(unless otherwise noted)
supply current
PARAMETER
Vin
Input voltage range
IQ
ISD
Operating quiescent current
VUVLO
Under-voltage lockout threshold
Shutdown current
TEST CONDITIONS
MIN
TYP
MAX
6
V
28
50
µA
0.1
1
µA
1.5
1.7
V
TYP
MAX
1.8
IOUT = 0 mA, not switching, VFB = 1.3 V
EN=GND
UNIT
enable
PARAMETER
VIH
VIL
EN high level input voltage
II
EN input leakage current
TEST CONDITIONS
MIN
1.3
UNIT
V
EN low level input voltage
0.4
V
0.1
1
µA
MIN
TYP
MAX
30
V
250
400
550
ns
4
6
7.5
µs
EN = GND or VIN
power switch and current limit
PARAMETER
TEST CONDITIONS
UNIT
Vsw
Maximum switch voltage
toff
ton
Minimum off time
RDS(ON)
MOSFET on-resistance
Vin = 2.4 V; Isw = 200 mA; TPS61040
600
1000
mΩ
RDS(ON)
MOSFET on-resistance
Vin = 2.4 V; Isw = 200 mA; TPS61041
750
1250
mΩ
MOSFET leakage current
Vsw = 28 V
TPS61040
1
10
µA
MOSFET current limit
350
400
450
mA
MOSFET current limit
TPS61041
215
250
285
mA
ILIM
ILIM
Maximum on time
output
PARAMETER
VOUT
Vref
Adjustable output voltage range
IFB
VFB
Feedback input bias current
TEST CONDITIONS
MIN
Internal voltage reference
Feedback trip point voltage
TYP
Vin
MAX
V
1
µA
1.258
V
1.233
VFB = 1.3 V
1.8 V ≤ Vin ≤ 6.0 V
1.208
1.233
UNIT
28
V
Line regulation (see Note 3)
1.8 V ≤ Vin ≤ 6.0 V; Vout = 18 V; Iload = 10 mA
Cff = not connected
0.05
%/V
Load regulation (see Note 3)
Vin = 2.4 V; Vout = 18 V; 0 mA ≤ Iout ≤ 30 mA
0.15
%/mA
NOTE 3: The line and load regulation depend on the external component selection. Refer to the application section for further information.
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5
TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
TYPICAL CHARACTERISTICS
Table of Graphs
FIGURE
vs Load current
1, 2, 3
vs Input voltage
4
Quiescent current
vs Input voltage and temperature
5
Feedback voltage
vs Temperature
6
Switch current limit
vs Temperature
7
vs Supply voltage, TPS61041
8
η
Efficiency
IQ
VFB
ISW
ICL
RDSon
Switch current limit
RDSon
vs Supply voltage, TPS61040
9
vs Temperature
10
vs Supply voltage
12
Load transient response
13
Start-up behavior
14
EFFICIENCY
vs
OUTPUT CURRENT
EFFICIENCY
vs
LOAD CURRENT
90
90
VO = 18 V
88
88
VI = 5 V
86
86
82
80
VI = 2.4 V
78
74
72
72
Figure 1
6
78
74
100
TPS61041
80
76
1
10
IO – Output Current – mA
TPS61040
82
76
70
0.10
L = 10 µH
VO = 18 V
84
VI = 3.6 V
Efficiency – %
Efficiency – %
84
11
Line transient response
70
0.10
1
10
IL – Load Current – mA
Figure 2
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100
TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
TYPICAL CHARACTERISTICS
EFFICIENCY
vs
LOAD CURRENT
EFFICIENCY
vs
INPUT VOLTAGE
90
90
88
VO = 18 V
86
IO = 10 mA
86
L = 10 µH
84
IO = 5 mA
84
L = 3.3 µH
82
Efficiency – %
Efficiency – %
L = 10 µH
VO = 18 V
88
80
78
82
80
78
76
76
74
74
72
72
70
70
0.10
1
10
IL – Load Current – mA
1
100
2
3
4
5
6
VI – Input Voltage – V
Figure 3
Figure 4
TPS61040
FEEDBACK VOLTAGE
vs
FREE-AIR TEMPERATURE
QUIESCENT CURRENT
vs
INPUT VOLTAGE
1.24
40
TA = 85°C
35
30
VFB – Feedback Voltage – V
Quiescent Current – µ A
1.238
TA = 27°C
25
TA = –40°C
20
15
10
1.236
VCC = 2.4 V
1.234
1.232
5
0
1.8
2.4
3
3.6
4.2
4.8
5.4
6
VI – Input Voltage – V
Figure 5
1.23
–40
–20
0
20
40
60
80
TA – Temperature – °C
100
120
Figure 6
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7
TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
TYPICAL CHARACTERISTICS
TPS61040/41
TPS61041
SWITCH CURRENT LIMIT
vs
FREE-AIR TEMPERATURE
CURRENT LIMIT
vs
SUPPLY VOLTAGE
260
430
256
390
I (CL) – Current Limit – mA
I (SW) – Switch Current Limit – mA
258
TPS61040
410
370
350
330
310
290
254
250
248
246
244
270
TPS61041
250
242
230
–40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90
TA – Temperature – °C
240
1.8
TPS61040
rDS(on) – Static Drain-Source On-State Resistance – mΩ
420
I (CL) – Current Limit – mA
415
410
TA = 27°C
400
395
390
385
380
1.8
2.4
3
3.6
4.2
3
3.6
4.2
4.8
5.4
6
Figure 8
CURRENT LIMIT
vs
SUPPLY VOLTAGE
405
2.4
VCC – Supply Voltage – V
Figure 7
4.8
5.4
6
VCC – Supply Voltage – V
Figure 9
8
TA = 27°C
252
TPS61040/41
STATIC DRAIN-SOURCE ON-STATE RESISTANCE
vs
FREE-AIR TEMPERATURE
1200
1000
TPS61041
800
TPS61040
600
400
200
0
–40 –30 –20 –10 0 10 20 30 40 50 60 70 80 90
TA – Temperature – °C
Figure 10
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TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
rDS(on) – Static Drain-Source On-State Resistance – mΩ
TYPICAL CHARACTERISTICS
TPS61040/41
STATIC DRAIN-SOURCE ON-STATE RESISTANCE
vs
SUPPLY VOLTAGE
1000
900
800
TPS61041
700
600
TPS61040
500
400
300
200
100
0
1.8
2.4
3
3.6
4.2
4.8
5.4
6
VCC – Supply Voltage – V
Figure 11
VO = 18 V
VI
2.4 V to 3.4 V
VO
100 mV/div
200 µS/div
Figure 12. Line Transient Response
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9
TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
TYPICAL CHARACTERISTICS
VO = 18 V
VO
100 mA/div
VO
1 mA to 10 mA
200 µS/div
Figure 13. Load Transient Response
VO = 18 V
VO
5 V/div
EN
1 V/div
II
50 mA/div
Figure 14. Start-Up Behavior
10
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TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
APPLICATION INFORMATION
inductor selection, maximum load current
Since the PFM peak current control scheme is inherently stable, the inductor value does not affect the stability
of the regulator. The selection of the inductor together with the nominal load current, input and output voltage
of the application determines the switching frequency of the converter. Depending on the application, inductor
values between 2.2 µH up to 47 µH are recommended. The maximum inductor value is determined by the
maximum on time of the switch, typically 6 µs. The peak current limit of 400 mA/250mA (typically) should be
reached within this 6-µs period for proper operation.
The inductor value determines the maximum switching frequency of the converter. Therefore, select the
inductor value that ensures the maximum switching frequency at the converter maximum load current is not
exceeded. The maximum switching frequency is calculated by the following formula:
fS max +
Vin
min
I
L
P
(Vout–Vin)
Vout
Where:
IP = Peak current as described in the previous peak current control section
L = Selected inductor value
Vinmin = The highest switching frequency occurs at the minimum input voltage
If the selected inductor value does not exceed the maximum switching frequency of the converter, the next step
is to calculate the switching frequency at the nominal load current using the following formula:
ǒ loadǓ +
fS I
2
I
(Vout–Vin ) Vd)
load
I
2
P
L
Where:
IP = Peak current as described in the previous peak current control section
L = Selected inductor value
Iload = Nominal load current
Vd = Rectifier diode forward voltage (typically 0.3V)
A smaller inductor value gives a higher converter switching frequency, but lowers the efficiency.
The inductor value has less effect on the maximum available load current and is only of secondary order. The
best way to calculate the maximum available load current under certain operating conditions is to estimate the
expected converter efficiency at the maximum load current. This number can be taken out of the efficiency
graphs shown in Figures 1, 2, 3, and 4. The maximum load current can then be estimated as follows:
I
I 2 L fS max
+h P
load max
2 (Vout * Vin)
Where:
IP = Peak current as described in the previous peak current control section
L = Selected inductor value
fSmax = Maximum switching frequency as calculated previously
η = Expected converter efficiency. Typically 70% to 85%
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11
TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
APPLICATION INFORMATION
The maximum load current of the conveter is the current at the operation point where the coverter starts to enter
the continuous conduction mode. Usually the converter should always operate in discontinuous conduction
mode.
Last, the selected inductor should have a saturation current that meets the maximum peak current of the
converter (as calculated in the peak current control section). Use the maximum value for ILIM for this calculation.
Another important inductor parameter is the dc resistance. The lower the dc resistance, the higher the efficiency
of the converter. Refer to the Table 1 and the typical applications for the inductor selection.
Table 1. Recommended Inductor for Typical LCD Bias Supply (see Figure 15)
DEVICE
INDUCTOR VALUE
TPS61040
TPS61041
COMPONENT SUPPLIER
COMMENTS
10 µH
Sumida CR32-100
High efficiency
10 µH
Sumida CDRH3D16-100
High efficiency
10 µH
Murata LQH4C100K04
High efficiency
4.7 µH
Sumida CDRH3D16-4R7
Small solution size
4.7 µH
Murata LQH3C4R7M24
Small solution size
10 µH
Murata LQH3C100K24
High efficiency
Small solution size
setting the output voltage
The output voltage is calculated as:
V out + 1.233 V
ǒ1 ) R1
Ǔ
R2
For battery powered applications a high impedance voltage divider should be used with a typical value for R2
of ≤200 kΩ and a maximum value for R1 of 2.2 MΩ. Smaller values might be used to reduce the noise sensitivity
of the feedback pin.
A feedforward capacitor across the upper feedback resistor R1 is required to provide sufficient overdrive for the
error comparator. Without a feedforward capacitor, or one whose value is too small, the TPS61040/41 shows
double pulses or a pulse burst instead of single pulses at the switch node (SW), causing higher output voltage
ripple. If this higher output voltage ripple is acceptable, the feedforward capacitor can be left out.
The lower the switching frequency of the converter, the larger the feedforward capacitor value required. A good
starting point is to use a 10 pF feedforward capacitor. As a first estimation, the required value for the feedforward
capacitor at the operation point can also be calculated using the following formula:
C
FF
+
2
p
1
fS
20
R1
Where:
R1 = Upper resistor of voltage divider
fS = Switching frequency of the converter at the nominal load current (See previous section for calculating the
switching frequency)
CFF= Choose a value that comes closest to the result of the calculation
12
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TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
APPLICATION INFORMATION
The larger the feedforward capacitor the worse the line regulation of the device. Therefore, when concern for
line regulation is paramount, the selected feedforward capacitor should be as small as possible. See the next
section for more information about line and load regulation.
line and load regulation
The line regulation of the TPS61040/41 depends on the voltage ripple on the feedback pin. Usually a 50 mV
peak-to-peak voltage ripple on the feedback pin FB gives good results.
Some applications require a very tight line regulation and can only allow a small change in output voltage over
a certain input voltage range. If no feedforward capacitor CFF is used across the upper resistor of the voltage
feedback divider, the device has the best line regulation. Without the feedforward capacitor the output voltage
ripple is higher because the TPS61040/41 shows output voltage bursts instead of single pulses on the switch
pin (SW), increasing the output voltage ripple. Increasing the output capacitor value reduces the output voltage
ripple.
If a larger output capacitor value is not an option, a feedforward capacitor CFF can be used as described in the
previous section. The use of a feedforward capacitor increases the amount of voltage ripple present on the
feedback pin (FB). The greater the voltage ripple on the feedback pin (≥50 mV), the worse the line regulation.
There are two ways to improve the line regulation further:
1. Use a smaller inductor value to increase the switching frequency which will lower the output voltage ripple,
as well as the voltage ripple on the feedback pin.
2. Add a small capacitor from the feedback pin (FB) to ground to reduce the voltage ripple on the feedback
pin down to 50 mV again. As a starting point, the same capacitor value as selected for the feedforward
capacitor CFF can be used.
output capacitor selection
For best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low
ESR value but tantalum capacitors can be used as well, depending on the application.
Assuming the converter does not show double pulses or pulse bursts on the switch node (SW), the output
voltage ripple can be calculated as:
I
DV out + out
C out
ǒ
Ǔ
I
L
1
P
–
)I
P
fS(Iout) Vout ) Vd–Vin
ESR
Where:
IP = Peak current as described in the previous peak current control section
L = Selected inductor value
Iout = Nominal load current
fS (Iout) = Switching frequency at the nominal load current as calculated previously
Vd = Rectifier diode forward voltage (typically 0.3V)
Cout = Selected output capacitor
ESR = Output capacitor ESR value
Refer to Table 2 and typical applications section for choosing the output capacitor.
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TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
APPLICATION INFORMATION
Table 2. Recommended Input and Output Capacitors
DEVICE
TPS61040/41
CAPACITOR
VOLTAGE RATING
COMPONENT SUPPLIER
4.7 µF/X5R/0805
6.3 V
Tayo Yuden JMK212BY475MG
CIN/COUT
COMMENTS
10 µF/X5R/0805
6.3 V
Tayo Yuden JMK212BJ106MG
CIN/COUT
1.0 µF/X7R/1206
25 V
Tayo Yuden TMK316BJ105KL
COUT
1.0 µF/X5R/1206
35 V
Tayo Yuden GMK316BJ105KL
COUT
4.7 µF/X5R/1210
25 V
Tayo Yuden TMK325BJ475MG
COUT
input capacitor selection
For good input voltage filtering, low ESR ceramic capacitors are recommended. A 4.7 µF ceramic input
capacitor is sufficient for most of the applications. For better input voltage filtering this value can be increased.
Refer to Table 2 and typical applications for input capacitor recommendations.
diode selection
To achieve high efficiency a Schottky diode should be used. The current rating of the diode should meet the peak
current rating of the converter as it is calculated in the section peak current control. Use the maximum value
for ILIM for this calculation. Refer to Table 3 and the typical applications for the selection of the Schottky diode.
Table 3. Recommended Schottky Diode for Typical LCD Bias Supply (see Figure 15)
DEVICE
TPS61040/41
REVERSE VOLTAGE
COMPONENT SUPPLIER
COMMENTS
30 V
ON Semiconductor MBR0530
20 V
ON Semiconductor MBR0520
20 V
ON Semiconductor MBRM120L High efficiency
30 V
Toshiba CRS02
layout considerations
Typical for all switching power supplies, the layout is an important step in the design; especially at high peak
currents and switching frequencies. If the layout is not carefully done, the regulator might show noise problems
and duty cycle jitter.
The input capacitor should be placed as close as possible to the input pin for good input voltage viltering. The
inductor and diode should be placed as close as possible to the switch pin to minimize the noise coupling into
other circuits. Since the feedback pin and network is a high impedance circit the feedback network should be
routed away from the inductor. The feedback pin and feedback network should be shielded with a ground plane
or trace to minimize noise coupling into this circuit.
Wide traces should be used for connections in bold as shown in Figure 15. A star ground connection or ground
plane minimizes ground shifts and noise.
14
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TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
APPLICATION INFORMATION
D1
L1
VO
VIN
VIN
CFF
R1
SW
CO
FB
CIN
EN
R2
GND
Figure 15. Layout Diagram
VIN
1.8 V to 6 V
L1
10 µH
VOUT
18 V
TPS61040
VIN
C1
4.7 µF
D1
CFF
22 pF
R1
2.2 MΩ
SW
C2
1 µF
FB
EN
GND
L1:
D1:
C1:
C2:
R2
160 kΩ
Sumida CR32–100
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden TMK316BJ105KL
Figure 16. LCD Bias Supply
L1
10 µH
D1
VO
18 V
TPS61040
VIN
1.8 V to 6 V
C1
4.7 µF
VIN
SW
R1
2.2 MΩ
FB
EN
GND
R2
160 kΩ
CFF
22 pF
C2
1 µF
DAC or Analog Voltage
0 V = 25 V
1.233 V = 18 V
L1:
D1:
C1:
C2:
Sumida CR32-100
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden GMK316BJ105KL
Figure 17. LCD Bias Supply With Adjustable Output Voltage
www.ti.com
15
TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
APPLICATION INFORMATION
R3
200 kΩ
BC857C
L1
10 µH
VIN
1.8 V to 6 V
D1
VOUT
18 V / 10 mA
TPS61040
VIN
R1
2.2 MΩ
SW
C2
1 µF
FB
C1
4.7 µF
EN
CFF
22 pF
C3
0.1 µF
(Optional)
R2
160 kΩ
GND
L1:
D1:
C1:
C2:
Sumida CR32-100
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden TMK316BJ105KL
Figure 18. LCD Bias Supply With Load Disconnect
D3
V2 = –10 V/15 mA
D2
L1
6.8 µH
C4
4.7 µF
C3
1 µF
D1
V1 = 10 V/15 mA
TPS61040
VIN = 2.7 V to 5 V
C1
4.7 µF
VIN
SW
R1
1.5 MΩ
CFF
22 pF
FB
EN
GND
C2
1 µF
L1:
D1, D2, D3:
C1:
C2, C3, C4:
R2
210 kΩ
Murata LQH4C6R8M04
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden EMK316BJ105KF
Figure 19. Positive and Negative Output LCD Bias Supply
16
www.ti.com
TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
APPLICATION INFORMATION
L1
6.8 µH
D1
VO = 12 V/35 mA
TPS61040
VIN
VIN 3.3 V
CFF
4.7 pF
R1
1.8 MΩ
SW
C2
4.7 µF
FB
C1
10 µF
EN
GND
L1:
D1:
C1:
C2:
R2
205 kΩ
Murata LQH4C6R8M04
Motorola MBR0530
Tayo Yuden JMK212BJ106MG
Tayo Yuden EMK316BJ475ML
Figure 20. Standard 3.3-V to 12-V Supply
D1
3.3 µH
5 V/45 mA
TPS61040
VIN
1.8 V to 4 V
SW
CFF
3.3 pF
R1
620 kΩ
C2
4.7 µF
FB
C1
4.7 µF
EN
GND
R2
200 kΩ
L1:
D1:
C1, C2:
Murata LQH4C3R3M04
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Figure 21. Dual Battery Cell to 5V/50mA Conversion Efficiency ≈ 84% at Vin = 2.4 V to Vo = 5 V/45 mA
L1
10 µH
VCC = 2.7 V to 6 V
VIN
C1
4.7 µF
PWM
100 Hz to 500 Hz
SW
D1
D2
24 V
(Optional)
FB
EN
GND
C2
1 µF
RS
82 Ω
L1:
D1:
C1:
C2:
Murata LQH4C100K04
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Tayo Yuden TMK316BJ105KL
Figure 22. White LED Supply With Adjustable Brightness Control Using a PWM Signal on the Enable Pin
Efficiency ≈ 86% at Vin = 3 V, ILED = 15 mA
www.ti.com
17
TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
L1
10 µH
VCC = 2.7 V to 6 V
C1
4.7 µF
VIN
SW
D1
MBRM120L
C2†
100 nF
D2
24 V
(Optional)
FB
EN
R1
120 kΩ
GND
Analog Brightness Control
3.3 V≅ Led Off
0 V≅ Iled = 20 mA
R2 160 kΩ
RS
110 Ω
L1:
D1:
C1:
C2:
Murata LQH4C3R3M04
Motorola MBR0530
Tayo Yuden JMK212BY475MG
Standard Ceramic Capacitor
† A smaller output capacitor value for C2 will cause a larger LED ripple
Figure 23. White LED Supply With Adjustable Brightness Control
Using an Analog Signal on the Feedback Pin
18
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TPS61040
TPS61041
SLVS413A – FEBRUARY 2002 – REVISED OCTOBER 2002
MECHANICAL DATA
DBV (R-PDSO-G5)
PLASTIC SMALL-OUTLINE
0,50
0,30
0,95
5
0,20 M
4
1,70
1,50
1
0,15 NOM
3,00
2,60
3
Gage Plane
3,00
2,80
0,25
0°–8°
0,55
0,35
Seating Plane
1,45
0,95
0,05 MIN
0,10
4073253-4/F 10/00
NOTES: A.
B.
C.
D.
All linear dimensions are in millimeters.
This drawing is subject to change without notice.
Body dimensions do not include mold flash or protrusion.
Falls within JEDEC MO-178.
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19
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