AD ADP2105ACPZ-1.8-R7

1 Amp/1.5 Amp/2 Amp Synchronous,
Step-Down DC-to-DC Converters
ADP2105/ADP2106/ADP2107
FEATURES
GENERAL DESCRIPTION
Extremely high 97% efficiency
Ultralow quiescent current: 20 μA
1.2 MHz switching frequency
0.1 μA shutdown supply current
Maximum load current:
ADP2105: 1 A
ADP2106: 1.5 A
ADP2107: 2 A
Input voltage: 2.7 V to 5.5 V
Output voltage: 0.8 V to VIN
Maximum duty cycle: 100%
Smoothly transitions into low dropout (LDO) mode
Internal synchronous rectifier
Small 16-lead 4 mm × 4 mm LFCSP_VQ package
Optimized for small ceramic output capacitors
Enable/Shutdown logic input
Undervoltage lockout
Soft start
The ADP2105/ADP2106/ADP2107 are low quiescent current,
synchronous, step-down dc-to-dc converters in a compact 4 mm ×
4 mm LFCSP_VQ package. At medium-to-high load currents,
these devices use a current-mode, constant-frequency pulse
width modulation (PWM) control scheme for excellent stability
and transient response. To ensure the longest battery life in
portable applications, the ADP2105/ADP2106/ADP2107 use a
pulse frequency modulation (PFM) control scheme under light
load conditions that reduces switching frequency to save power.
The ADP2105/ADP2106/ADP2107 run from input voltages of
2.7 V to 5.5 V, allowing single Li+/Li− polymer cell, multiple
alkaline/NiMH cells, PCMCIA, and other standard power sources.
The output voltage of ADP2105/ADP2106/ADP2107-ADJ is
adjustable from 0.8 V to the input voltage, while the ADP2105/
ADP2106/ADP2107-xx are available in preset output voltage
options of 3.3 V, 1.8 V, 1.5 V, and 1.2 V. Each of these variations is
available in three maximum current levels, 1 A (ADP2105), 1.5 A
(ADP2106), and 2 A (ADP2107). The power switch and synchronous rectifier are integrated for minimal external part count
and high efficiency. During logic-controlled shutdown, the
input is disconnected from the output, and it draws less than
0.1 μA from the input source. Other key features include
undervoltage lockout to prevent deep-battery discharge and
programmable soft start to limit inrush current at startup.
APPLICATIONS
Mobile handsets
PDAs and palmtop computers
Telecommunication/Networking equipment
Set top boxes
Audio/Video consumer electronics
TYPICAL PERFORMANCE CHARACTERISTICS
TYPICAL OPERATING CIRCUIT
0.1μF
VIN
10Ω
INPUT VOLTAGE = 2.7V TO 5.5V
100
VIN = 3.3V
10μF
VOUT = 2.5V
VIN = 3.6V
FB
95
OFF
15
14
13
GND
IN
PWIN1
OUTPUT VOLTAGE = 2.5V
LX2 12
1
EN
2
GND
3
GND
LX1 10
4
GND
PWIN2 9
2μH
90
85
06079-001
80
0
200
400
PGND 11
ADP2107-ADJ
VIN = 5V
75
16
FB
600
800
COMP
SS
5
6
70kΩ
85kΩ
10μF
FB
VIN
40kΩ
AGND NC
7
8
4.7μF
10μF
LOAD
0A TO 2A
1nF
120pF
1000 1200 1400 1600 1800 2000
LOAD CURRENT (mA)
Figure 1. Efficiency vs. Load Current for the ADP2107 with VOUT = 2.5 V
NC = NO CONNECT
06079-002
EFFICIENCY (%)
ON
Figure 2. Circuit Configuration of ADP2107 with VOUT = 2.5 V
Rev. 0
Information furnished by Analog Devices is believed to be accurate and reliable. However, no
responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other
rights of third parties that may result from its use. Specifications subject to change without notice. No
license is granted by implication or otherwise under any patent or patent rights of Analog Devices.
Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.461.3113
©2006 Analog Devices, Inc. All rights reserved.
ADP2105/ADP2106/ADP2107
TABLE OF CONTENTS
Features .............................................................................................. 1
Setting the Output Voltage........................................................ 15
Applications....................................................................................... 1
Inductor Selection ...................................................................... 16
General Description ......................................................................... 1
Output Capacitor Selection....................................................... 17
Typical Performance Characteristics ............................................. 1
Input Capacitor Selection.......................................................... 17
Typical Operating Circuit................................................................ 1
Input Filter................................................................................... 18
Revision History ............................................................................... 2
Soft Start ...................................................................................... 18
Specifications..................................................................................... 3
Loop Compensation .................................................................. 18
Absolute Maximum Ratings............................................................ 5
Bode Plots.................................................................................... 19
Thermal Resistance ...................................................................... 5
Load Transient Response .......................................................... 20
Boundary Condition.................................................................... 5
Efficiency Considerations ......................................................... 21
ESD Caution.................................................................................. 5
Thermal Considerations............................................................ 21
Pin Configuration and Function Descriptions............................. 6
Design Example.......................................................................... 22
Typical Performance Characteristics ............................................. 7
External Component Recommendations.................................... 24
Theory of Operation ...................................................................... 12
Circuit Board Layout Recommendations ................................... 26
Control Scheme .......................................................................... 12
Evaluation Board ............................................................................ 27
PWM Mode Operation.............................................................. 12
Evaluation Board Schematic (ADP2107-1.8V)...................... 27
PFM Mode Operation................................................................ 12
Recommended PCB Board Layout
(Evaluation Board Layout)........................................................ 27
Pulse-Skipping Threshold ......................................................... 12
100% Duty Cycle Operation (LDO Mode) ............................. 12
Slope Compensation .................................................................. 13
Features ........................................................................................ 13
Application Circuits ....................................................................... 29
Outline Dimensions ....................................................................... 31
Ordering Guide .......................................................................... 31
Applications Information .............................................................. 15
External Component Selection................................................. 15
REVISION HISTORY
7/06—Revision 0: Initial Version
Rev. 0 | Page 2 of 32
ADP2105/ADP2106/ADP2107
SPECIFICATIONS
VIN = 3.6 V @ TA = 25°C, unless otherwise noted. 1 Bold values indicate −40°C ≤ TJ ≤ +125°C.
Table 1.
Parameter
INPUT CHARACTERISTICS
Input Voltage Range
Undervoltage Lockout Threshold
Undervoltage Lockout Hysteresis 2
OUTPUT CHARACTERISTICS
Output Regulation Voltage
Load Regulation
Line Regulation 3
Output Voltage Range
FEEDBACK CHARACTERISTICS
OUT_SENSE Bias Current
FB Regulation Voltage
FB Bias Current
INPUT CURRENT CHARACTERISTICS
IN Operating Current
IN Shutdown Current
LX (SWITCH NODE) CHARACTERISTICS
LX On Resistance 4
LX Leakage Current4
LX Peak Current Limit4
Conditions
Min
VIN rising
VIN falling
2.7
2.2
2.0
ADP210x-3.3, load = 10 mA
ADP210x-3.3, VIN = 3.5 V to 5.5 V, no load to full load
ADP210x-1.8, load = 10 mA
ADP210x-1.8, VIN = 2.7 V to 5.5 V, no load to full load
ADP210x-1.5, load = 10 mA
ADP210x-1.5, VIN = 2.7 V to 5.5 V, no load to full load
ADP210x-1.2, load = 10 mA
ADP210x-1.2, VIN = 2.7 V to 5.5 V, no load to full load
ADP2105
ADP2106
ADP2107
Measured in servo loop
ADP210x-ADJ
ADP210x-1.2
ADP210x-1.5
ADP210x-1.8
ADP210x-3.3
ADP210x-ADJ
ADP210x-ADJ
3.267
3.201
1.782
1.746
1.485
1.455
1.188
1.164
Typ
2.4
2.2
200
3.3
3.3
1.8
1.8
1.5
1.5
1.2
1.2
0.4
0.5
0.6
0.1
0.8
0.784
−0.1
3
4
5
10
0.8
Max
Unit
5.5
2.6
2.5
V
V
V
mV
3.333
3.399
1.818
1.854
1.515
1.545
1.212
1.236
V
V
V
V
V
V
V
V
%/A
%/A
%/A
%/V
V
0.3
VIN
6
8
10
20
0.816
+0.1
μA
μA
μA
μA
V
μA
ADP210x-ADJ, VFB = 0.9 V
ADP210x-xx, output voltage 10% above regulation voltage
VEN = 0 V
20
20
0.1
30
30
15
μA
μA
μA
P-channel switch
N-channel synchronous rectifier
VIN = 5.5 V, VLX = 0 V, 5.5 V
P-channel switch, ADP2107
P-channel switch, ADP2106
P-channel switch, ADP2105
In PWM mode of operation, VIN = 5.5 V
100
90
0.1
2.9
2.25
1.5
165
140
15
mΩ
mΩ
μA
A
A
A
ns
2.6
2.0
1.3
3.3
2.6
1.8
100
LX Minimum On-Time4
ENABLE CHARACTERISTICS
EN Input High Voltage
EN Input Low Voltage
EN Input Leakage Current
VIN = 2.7 V to 5.5 V
VIN = 2.7 V to 5.5 V
VIN = 5.5 V, VEN = 0 V, 5.5 V
2
−1
−0.1
0.4
+1
OSCILLATOR FREQUENCY
SOFT START PERIOD
VIN = 2.7 V to 5.5 V
CSS = 1 nF
1
750
1.2
1000
1.4
1200
Rev. 0 | Page 3 of 32
V
V
μA
MHz
μs
ADP2105/ADP2106/ADP2107
Parameter
THERMAL CHARACTERISTICS
Thermal Shutdown Threshold
Thermal Shutdown Hysteresis
COMPENSATOR TRANSCONDUCTANCE (Gm)
CURRENT SENSE AMPLIFIER GAIN (GCS)2
Conditions
Min
Typ
Max
140
40
50
1.875
2.8125
3.625
ADP2105
ADP2106
ADP2107
1
All limits at temperature extremes are guaranteed via correlation using standard statistical quality control (SQC). Typical values are at TA = 25°C.
Guaranteed by design.
The ADP2015/ADP2106/ADP2107 line regulation was measured in a servo loop on the ATE that adjusts the feedback voltage to achieve a specific comp voltage.
4
All LX (switch node) characteristics are guaranteed only when the LX1 and LX2 pins are tied together.
5
These specifications are guaranteed from −40°C to +85°C.
2
3
Rev. 0 | Page 4 of 32
Unit
°C
°C
μA/V
A/V
A/V
A/V
ADP2105/ADP2106/ADP2107
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
IN, EN, SS, COMP, OUT_SENSE/FB to
AGND
LX1, LX2 to PGND
PWIN1, PWIN2 to PGND
PGND to AGND
GND to AGND
PWIN1, PWIN2 to IN
Operating Junction Temperature Range
Storage Temperature Range
Soldering Conditions
THERMAL RESISTANCE
Rating
−0.3 V to +6 V
θJA is specified for the worst-case conditions, that is, a device
soldered in a circuit board for surface-mount packages.
−0.3 V to (VIN + 0.3 V)
−0.3 V to +6 V
−0.3 V to +0.3 V
−0.3 V to +0.3 V
−0.3 V to +0.3 V
−40°C to +125°C
−65°C to +150°C
JEDEC J-STD-020
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Table 3. Thermal Resistance
Package Type
16-Lead LFCSP_VQ/QFN
Maximum Power Dissipation
1
θJA 1
40
1
Unit
°C/W
W
θJA is specified for the worst-case conditions; that is, θJA is specified for device
soldered in circuit board for surface mount packages.
BOUNDARY CONDITION
Natural convection, 4-layer board, exposed pad soldered to
the PCB.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the
human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. 0 | Page 5 of 32
ADP2105/ADP2106/ADP2107
13 PWIN1
14 IN
15 GND
16 OUT_SENSE/FB
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
PIN 1
INDICATOR
12 LX2
EN 1
TOP VIEW
(Not to Scale)
11 PGND
10 LX1
9
PWIN2
NC = NO CONNECT
06079-003
NC 8
COMP 5
AGND 7
GND 4
SS 6
GND 3
ADP2105/
ADP2106/
ADP2107
GND 2
Figure 3. Pin Configuration
Table 4. Pin Function Descriptions
Pin No.
1
Mnemonic
ADP210x-xx ADP210x-ADJ
EN
EN
2, 3, 4,
15
GND
GND
5
COMP
COMP
6
SS
SS
7
AGND
AGND
8
9, 13
NC
PWIN2,
PWIN1
NC
PWIN2, PWIN1
10, 12
LX1, LX2
LX1, LX2
11
PGND
PGND
14
IN
IN
16
OUT_SENSE
FB
Description
Enable Input. Drive EN high to turn on the ADP2105/ADP2106/ADP2107. Drive EN low to turn
it off and reduce the input current to 0.1 μA.
Test Pins. These pins are used by Analog Devices, Inc. for internal testing and are not ground
return pins. Tie these pins to the AGND plane as close to the ADP2105/ADP2106/ADP2107 as
possible.
Feedback Loop Compensation Node. COMP is the output of the internal transconductance
error amplifier. Place a series RC network from COMP to AGND to compensate the converter.
See the Loop Compensation section.
Soft Start Input. Place a capacitor from SS to AGND to set the soft start period. A 1 nF capacitor
sets a 1 ms soft start period.
Analog Ground. Connect the ground of the compensation components, soft start capacitor,
and the voltage divider on the FB pin to the AGND pin as close as possible to the ADP2105/
ADP2106/ADP2107. Also connect AGND to the exposed pad of ADP2105/ADP2106/ADP2107.
No Connect. Not internally connected. Can be connected to other pins or left unconnected.
Power Source Inputs. The source of the PFET high-side switch. Bypass each PWIN pin to the nearest
PGND plane with a 4.7 μF or greater capacitor as close as possible to the ADP2105/ADP2106/
ADP2107. See the Input Capacitor Selection section.
Switch Outputs. The drain of the P-channel power switch and N-channel synchronous rectifier.
Tie the two LX pins together and connect the output LC filter between LX and the output
voltage.
Power Ground. Connect the ground return of all input and output capacitors to PGND pin,
using a power ground plane as close as possible to the ADP2105/ADP2106/ADP2107. Also
connect PGND to the exposed pad of the ADP2105/ADP2106/ADP2107.
ADP2105/ADP2106/ADP2107 Power Input. The power source for the ADP2105/ADP2106/
ADP2107 internal circuitry. Connect IN and PWIN1 with a 10 Ω resistor as close as possible to
the ADP2105/ADP2106/ADP2107. Bypass IN to AGND with a 0.1 μF or greater capacitor. See
the Input Filter section.
Output Voltage Sense or Feedback Input. For fixed output versions, connect OUT_SENSE to the
output voltage. For adjustable versions, FB is the input to the error amplifier. Drive FB through
a resistive voltage divider to set the output voltage. The FB regulation voltage is 0.8 V.
Rev. 0 | Page 6 of 32
ADP2105/ADP2106/ADP2107
TYPICAL PERFORMANCE CHARACTERISTICS
100
100
95
95
90
VIN = 2.7V
80
VIN = 5.5V
75
VIN = 4.2V
70
65
80
75
VIN = 4.2V
70
55
1
10
06079-004
60
INDUCTOR: SD14, 2.5µH
DCR: 60mΩ
TA = 25°C
50
1000
100
VIN = 5.5V
55
1
10
LOAD CURRENT (mA)
Figure 7. Efficiency—ADP2105 (1.8 V Output)
100
100
VIN = 3.6V
95
90
90
85
85
80
EFFICIENCY (%)
VIN = 5.5V
75
VIN = 4.2V
70
VIN = 2.7V
VIN = 3.6V
80
VIN = 4.2V
75
70
VIN = 5.5V
65
65
55
1
10
06079-052
INDUCTOR: CDRH5D18, 4.1μH
DCR: 43mΩ
TA = 25°C
50
1000
100
INDUCTOR: D62LCB, 2µH
DCR: 28mΩ
TA = 25°C
55
1
10
1000
10000
LOAD CURRENT (mA)
LOAD CURRENT (mA)
Figure 5. Efficiency—ADP2105 (3.3 V Output)
Figure 8. Efficiency—ADP2106 (1.2 V Output)
100
100
VIN = 3.6V
95
90
95
90
VIN = 2.7V
85
EFFICIENCY (%)
85
VIN = 4.2V
80
75
VIN = 5.5V
70
65
VIN = 5.5V
80
VIN = 4.2V
75
70
65
60
60
INDUCTOR: D62LCB, 2µH
DCR: 28mΩ
TA = 25°C
55
1
10
100
1000
06079-062
EFFICIENCY (%)
100
06079-008
60
60
VIN = 3.6V
INDUCTOR: D62LCB, 3.3µH
DCR: 47mΩ
TA = 25°C
55
50
10000
LOAD CURRENT (mA)
1
10
100
1000
LOAD CURRENT (mA)
Figure 6. Efficiency—ADP2106 (1.8 V Output)
Figure 9. Efficiency—ADP2106 (3.3 V Output)
Rev. 0 | Page 7 of 32
06079-053
EFFICIENCY (%)
95
50
1000
100
LOAD CURRENT (mA)
Figure 4. Efficiency—ADP2105 (1.2 V Output)
50
INDUCTOR: SD3814, 3.3µH
DCR: 93mΩ
TA = 25°C
06079-061
65
60
50
VIN = 3.6V
85
VIN = 3.6V
EFFICIENCY (%)
EFFICIENCY (%)
85
VIN = 2.7V
90
10000
ADP2105/ADP2106/ADP2107
100
100
95
95
90
VIN = 2.7V
EFFICIENCY (%)
VIN = 4.2V
75
70
VIN = 5.5V
65
VIN = 4.2V
80
VIN = 5.5V
75
70
65
60
55
1
10
100
1000
06079-010
60
INDUCTOR: SD12, 1.2µH
DCR: 37mΩ
TA = 25°C
INDUCTOR: D62LCB, 1.5µH
DCR: 21mΩ
TA = 25°C
55
50
10000
1
10
LOAD CURRENT (mA)
Figure 13. Efficiency—ADP2107 (1.8 V)
1.23
100
95
1.22
VIN = 5.5V
VIN = 4.2V
75
70
65
2.7V, +25°C
3.6V, +25°C
5.5V, +25°C
2.7V, +125°C
3.6V, +125°C
5.5V, +125°C
1.21
1.20
1.19
INDUCTOR: CDRH5D28, 2.5µH
DCR: 13mΩ
TA = 25°C
55
1
10
100
1000
1.18
1.17
0.01
10000
06079-082
60
50
2.7V, –40°C
3.6V, –40°C
5.5V, –40°C
VIN = 3.6V
06079-054
EFFICIENCY (%)
OUTPUT VOLTAGE (V)
90
80
10000
1000
LOAD CURRENT (mA)
Figure 10. Efficiency—ADP2107 (1.2 V)
85
100
06079-063
EFFICIENCY (%)
85
80
50
VIN = 2.7V
90
VIN = 3.6V
85
VIN = 3.6V
0.1
1
10
100
1000
10000
LOAD CURRENT (mA)
LOAD CURRENT (mA)
Figure 11. Efficiency—ADP2107 (3.3 V)
Figure 14. Output Voltage Accuracy—ADP2107 (1.2 V)
1.85
3.38
3.36
3.6V, –40°C
5.5V, –40°C
3.6V, +25°C
5.5V, +25°C
3.6V, +125°C
5.5V, +125°C
OUTPUT VOLTAGE (V)
1.81
1.79
3.34
3.32
3.30
3.28
3.26
1.77
1.75
0.1
1
2.7V, +25°C
3.6V, +25°C
5.5V, +25°C
10
2.7V, +125°C
3.6V, +125°C
5.5V, +125°C
100
1000
3.24
3.22
0.01
10000
LOAD CURRENT (mA)
06079-081
2.7V, –40°C
3.6V, –40°C
5.5V, –40°C
06079-064
OUTPUT VOLTAGE (V)
1.83
0.1
1
10
100
1000
LOAD CURRENT (mA)
Figure 12. Output Voltage Accuracy—ADP2107 (1.8 V)
Figure 15. Output Voltage Accuracy—ADP2107 (3.3 V)
Rev. 0 | Page 8 of 32
10000
ADP2105/ADP2106/ADP2107
120
10000
PMOS POWER SWITCH
100
SW ON RESISTANCE (mΩ)
+25°C
–40°C
10
1.2
40
20
06079-016
+125°C
1
0.8
NMOS SYNCHRONOUS RECTIFIER
60
1.6
2.0
2.4
2.8
3.2
3.6
4.0
4.4
4.8
0
2.7
5.2
TA = 25°C
3.0
3.3
3.6
0.802
1260
0.801
1250
0.800
0.799
0.798
0.797
20
40
60
80
100
+125°C
1220
+25°C
–40°C
1210
06079-021
1190
2.7
120 125
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
INPUT VOLTAGE (V)
Figure 20. Switching Frequency vs. Input Voltage
2.35
1.70
2.30
1.65
2.25
PEAK CURRENT LIMIT (A)
1.75
1.60
1.55
ADP2105 (1A)
1.45
1.40
ADP2106 (1.5A)
2.20
2.15
2.10
2.05
2.00
1.95
1.35
1.30
TA = 25°C
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
06079-073
PEAK CURRENT LIMIT (A)
5.4
1230
Figure 17. Feedback Voltage vs. Temperature
1.25
2.7
5.1
1240
TEMPERATURE (°C)
1.50
4.8
1200
06079-017
0.796
0
4.5
Figure 19. Switch On Resistance vs. Input Voltage
SWITCHING FREQUENCY (kHz)
FEEDBACK VOLTAGE (V)
Figure 16. Quiescent Current vs. Input Voltage
–20
4.2
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
0.795
–40
3.9
06079-018
100
80
1.90
1.85
2.7
5.7
TA = 25°C
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
Figure 21. Peak Current Limit of ADP2106
Figure 18. Peak Current Limit of ADP2105
Rev. 0 | Page 9 of 32
5.4
5.7
06079-072
INPUT CURRENT (µA)
1000
ADP2105/ADP2106/ADP2107
135
2.95
2.85
ADP2107 (2A)
2.80
2.75
2.70
2.65
06079-071
2.60
2.55
2.50
2.7
TA = 25°C
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
120
105
90
VOUT = 1.2V
75
60
45
VOUT = 1.8V
15
TA = 25°C
0
2.7
5.7
3.0
3.3
3.6
Figure 22. Peak Current Limit of ADP2107
4.5
4.8
5.1
5.4
5.7
195
PULSE SKIPPING THRESHOLD CURRENT (mA)
135
120
105
VOUT = 1.2V
90
75
60
VOUT = 2.5V
VOUT = 1.8V
30
06079-067
PULSE SKIPPING THRESHOLD CURRENT (mA)
4.2
Figure 25. Pulse Skipping Threshold vs. Input Voltage for ADP2105
150
15
0
2.7
3.9
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
45
VOUT = 2.5V
30
TA = 25°C
3.0
3.3
3.6
3.9
4.2
4.5
4.8
5.1
5.4
180
VOUT = 1.2V
165
150
135
VOUT = 1.8V
120
105
90
VOUT = 2.5V
75
60
45
30
15
TA = 25°C
0
2.7
5.7
3.0
3.3
3.6
INPUT VOLTAGE (V)
3.9
4.2
4.5
4.8
5.1
5.4
06079-068
PEAK CURRENT LIMIT (A)
2.90
06079-066
PULSE SKIPPING THRESHOLD CURRENT (mA)
3.00
5.7
INPUT VOLTAGE (V)
Figure 23. Pulse Skipping Threshold vs. Input Voltage for ADP2106
Figure 26. Pulse Skipping Threshold vs. Input Voltage for ADP2107
140
LX NODE (SWITCH NODE)
SWITCH ON RESISTANCE (mΩ)
120
INDUCTOR CURRENT
Δ: 260mV
@: 3.26V
1
OUTPUT VOLTAGE
CH1 1V
CH3 5V
CH4 1AΩ
M 10µs
T 45.8%
A CH1
80
NMOS SYNCHRONOUS RECTIFIER
60
40
20
06079-074
4
PMOS POWER SWITCH
100
0
–40
1.78V
06079-083
3
–20
0
20
40
60
80
100
JUNCTION TEMPERATURE (°C)
Figure 24. Short Circuit Response at Output
Figure 27. Switch On Resistance vs. Temperature
Rev. 0 | Page 10 of 32
120
ADP2105/ADP2106/ADP2107
LX NODE (SWITCH NODE)
LX NODE
(SWITCH NODE)
3
3
1
1
OUTPUT VOLTAGE (AC-COUPLED)
OUTPUT VOLTAGE (AC-COUPLED)
CH4 200mAΩ
M 2µs
T 6%
A CH3
06079-031
INDUCTOR CURRENT
CH1 50mV
CH3 2V
INDUCTOR CURRENT
06079-030
4
4
3.88V
CH1 20mV
CH3 2V
Figure 28. PFM Mode of Operation at Very Light Load (10 mA)
M 1µs
T 17.4%
CH4 1AΩ
A CH3
3.88V
Figure 31. PWM Mode of Operation at Medium/Heavy Load (1.5 A)
LX NODE (SWITCH NODE)
3
CHANNEL 3
FREQUENCY
= 336.6kHz
3
Δ: 2.86A
@: 2.86A
LX NODE (SWITCH NODE)
1
INDUCTOR CURRENT
OUTPUT VOLTAGE (AC-COUPLED)
OUTPUT VOLTAGE
1
CH4 200mAΩ
M 400ns
T 17.4%
A CH3
3.88V
06079-032
INDUCTOR CURRENT
CH1 50mV
CH3 2V
4
06079-033
4
CH1 1V
CH3 5V
Figure 29. DCM Mode of Operation at Light Load (100 mA)
M 4µs
T 45%
CH4 1AΩ
A CH3
1.8V
Figure 32. Current Limit Behavior of ADP2107 (Frequency Foldback)
LX NODE (SWITCH NODE)
ENABLE VOLTAGE
3
OUTPUT VOLTAGE
3
1
1
INDUCTOR CURRENT
OUTPUT VOLTAGE (AC-COUPLED)
CH4 1AΩ
M 2µs
T 13.4%
A CH3
1.84V
Figure 30. Minimum Off Time Control at Dropout
06079-035
4
CH1 20mV
CH3 2V
4
06079-034
INDUCTOR CURRENT
CH1 1V
CH3 5V
CH4 500mAΩ
M 400µs
T 20.2%
A CH1
1.84V
Figure 33. Startup and Shutdown Waveform (CSS = 1 nF → SS Time = 1 ms)
Rev. 0 | Page 11 of 32
ADP2105/ADP2106/ADP2107
THEORY OF OPERATION
The ADP2105/ADP2106/ADP2107 are step-down, dc-to-dc
converters that use a fixed frequency, peak current-mode
architecture with an integrated high-side switch and low-side
synchronous rectifier. The high 1.2 MHz switching frequency
and tiny 16-lead, 4 mm × 4 mm LFCSP_VQ package allow for
a small step-down dc-to-dc converter solution. The integrated
high-side switch (P-channel MOSFET) and synchronous rectifier
(N-channel MOSFET) yield high efficiency at medium-toheavy loads. Light load efficiency is improved by smoothly
transitioning to variable frequency PFM mode.
The ADP2105/ADP2106/ADP2107-ADJ operate with an input
voltage from 2.7 V to 5.5 V and regulate an output voltage down
to 0.8 V. The ADP2105/ADP2106/ADP2107 are also available with
preset output voltage options of 3.3 V, 1.8 V, 1.5 V, and 1.2 V.
CONTROL SCHEME
The ADP2105/ADP2106/ADP2107 operate with a fixed
frequency, peak current-mode PWM control architecture at
medium-to-high loads for high efficiency, but shift to a variable
frequency PFM control scheme at light loads for lower quiescent current. When operating in fixed frequency PWM mode,
the duty cycle of the integrated switches is adjusted to regulate
the output voltage, but when operating in PFM mode at light
loads, the switching frequency is adjusted to regulate the output
voltage.
The ADP2105/ADP2106/ADP2107 operate in the PWM mode
only when the load current is greater than the pulse-skipping
threshold current. At load currents below this value, the converter
smoothly transitions to the PFM mode of operation.
PFM MODE OPERATION
The ADP2105/ADP2106/ADP2107 smoothly transition to the
variable frequency PFM mode of operation when the load current
decreases below the pulse-skipping threshold current, switching
only as necessary to maintain the output voltage within regulation.
When the output voltage dips below regulation, the ADP2105/
ADP2106/ADP2107 enter PWM mode for a few oscillator cycles
to increase the output voltage back to regulation. During the wait
time between bursts, both power switches are off, and the output
capacitor supplies all the load current. Because the output voltage
dips and recovers occasionally, the output voltage ripple in this
mode is larger than the ripple in the PWM mode of operation.
PULSE-SKIPPING THRESHOLD
The output current at which the ADP2105/ADP2106/ADP2107
transition from variable frequency PFM control to fixed frequency
PWM control is called the pulse-skipping threshold. The pulseskipping threshold has been optimized for excellent efficiency
over all load currents. The variation of pulse-skipping threshold
with input voltage and output voltage is shown in Figure 23,
Figure 25, and Figure 26.
100% DUTY CYCLE OPERATION (LDO MODE)
As the input voltage drops, approaching the output voltage,
the ADP2105/ADP2106/ADP2107 smoothly transition to 100%
duty cycle, maintaining the P-channel MOSFET switch on continuously. This allows the ADP2105/ADP2106/ADP2107 to regulate
the output voltage until the drop in input voltage forces the
P-channel MOSFET switch to enter dropout, as shown in the
following equation:
VIN(MIN) = IOUT × (RDS(ON) − P + DCRIND) + VOUT(NOM)
PWM MODE OPERATION
In PWM mode, the ADP2105/ADP2106/ADP2107 operate at
a fixed frequency of 1.2 MHz set by an internal oscillator. At the
start of each oscillator cycle, the P-channel MOSFET switch is
turned on, putting a positive voltage across the inductor. Current
in the inductor increases until the current sense signal crosses
the peak inductor current level that turns off the P-channel
MOSFET switch and turns on the N-channel MOSFET synchronous rectifier. This puts a negative voltage across the inductor,
causing the inductor current to decrease. The synchronous
rectifier stays on for the rest of the cycle, unless the inductor
current reaches zero, which causes the zero-crossing comparator
to turn off the N-channel MOSFET, as well. The peak inductor
current is set by the voltage on the COMP pin. The COMP pin
is the output of a transconductance error amplifier that compares
the feedback voltage with an internal 0.8 V reference.
The ADP2105/ADP2106/ADP2107 achieve 100% duty cycle
operation by stretching the P-channel MOSFET switch on-time
if the inductor current does not reach the peak inductor current
level by the end of the clock cycle. Once this happens, the oscillator remains off until the inductor current reaches the peak
inductor current level, at which time the switch is turned off and
the synchronous rectifier is turned on for a fixed off-time. At
the end of the fixed off-time, another cycle is initiated. As the
ADP2105/ADP2106/ADP2107 approach dropout, the switching
frequency decreases gradually to smoothly transition to 100%
duty cycle operation.
Rev. 0 | Page 12 of 32
ADP2105/ADP2106/ADP2107
SLOPE COMPENSATION
Short Circuit Protection
Slope compensation stabilizes the internal current control loop
of the ADP2105/ADP2106/ADP2107 when operating beyond
50% duty cycle to prevent sub-harmonic oscillations. It is implemented by summing a fixed scaled voltage ramp to the current
sense signal during the on-time of the P-channel MOSFET switch.
The ADP2105/ADP2106/ADP2107 include frequency foldback
to prevent output current run-away on a hard short. When the
voltage at the feedback pin falls below 0.3 V, indicating the possibility of a hard short at the output, the switching frequency is
reduced to 1/4 of the internal oscillator frequency. The reduction
in the switching frequency gives more time for the inductor to
discharge, preventing a runaway of output current.
The slope compensation ramp value determines the minimum
inductor that can be used to prevent sub-harmonic oscillations
at a given output voltage. The slope compensation ramp values
for ADP2105/ADP2106/ADP2107 follow. For more information,
see the Inductor Selection section.
For the ADP2105:
Slope Compensation Ramp Value = 0.72 A/μs
For the ADP2106:
Slope Compensation Ramp Value = 1.07 A/μs
Undervoltage Lockout (UVLO)
To protect against deep battery discharge, undervoltage lockout
circuitry is integrated on the ADP2105/ADP2106/ADP2107.
If the input voltage drops below the 2.2 V UVLO threshold, the
ADP2105/ADP2106/ADP2107 shut down, and both the power
switch and synchronous rectifier turn off. Once the voltage rises
again above the UVLO threshold, the soft start period is initiated,
and the part is enabled.
Thermal Protection
For the ADP2107:
Slope Compensation Ramp Value = 1.38 A/μs
FEATURES
Enable/Shutdown
Drive EN high to turn on the ADP2105/ADP2106/ADP2107.
Drive EN low to turn off the ADP2105/ADP2106/ADP2107,
reducing input current below 0.1 μA. To force the ADP2105/
ADP2106/ADP2107 to automatically start when input power
is applied, connect EN to IN. When shut down, the ADP2105/
ADP2106/ADP2107 discharge the soft start capacitor, causing
a new soft start cycle every time they are re-enabled.
Synchronous Rectification
In addition to the P-channel MOSFET switch, the ADP2105/
ADP2106/ADP2107 include an integrated N-channel MOSFET
synchronous rectifier. The synchronous rectifier improves
efficiency, especially at low output voltage, and reduces cost and
board space by eliminating the need for an external rectifier.
In the event that the ADP2105/ADP2106/ADP2107 junction
temperatures rise above 140°C, the thermal shutdown circuit turns
off the converter. Extreme junction temperatures can be the
result of high current operation, poor circuit board design, and/or
high ambient temperature. A 40°C hysteresis is included so that
when thermal shutdown occurs, the ADP2105/ADP2106/
ADP2107 do not return to operation until the on-chip temperature drops below 100°C. When coming out of thermal
shutdown, soft start is initiated.
Soft Start
The ADP2105/ADP2106/ADP2107 include soft start circuitry
to limit the output voltage rise time to reduce inrush current at
startup. To set the soft start period, connect the soft start
capacitor (CSS) from SS to AGND. When the ADP2105/ADP2106/
ADP2107 are disabled, or if the input voltage is below the undervoltage lockout threshold, CSS is internally discharged. When the
ADP2105/ADP2106/ADP2107 are enabled, CSS is charged through
an internal 0.8 μA current source, causing the voltage at SS to rise
linearly. The output voltage rises linearly with the voltage at SS.
Current Limit
The ADP2105/ADP2106/ADP2107 have protection circuitry to
limit the direction and amount of current flowing through the
power switch and synchronous rectifier. The positive current
limit on the power switch limits the amount of current that can
flow from the input to the output, while the negative current
limit on the synchronous rectifier prevents the inductor current
from reversing direction and flowing out of the load.
Rev. 0 | Page 13 of 32
ADP2105/ADP2106/ADP2107
COMP 5
SS 6
14 IN
SOFT
START
9 PWIN2
REFERENCE
0.8V
CURRENT SENSE
AMPLIFIER
13 PWIN1
FB1 16
OUT_SENSE1 16
GM ERROR
AMP
PWM/
PFM
CONTROL
AGND 7
GND 2
FOR PRESET
VOLTAGES
OPTIONS ONLY
DRIVER
AND
ANTISHOOT
THROUGH
GND 3
GND 4
CURRENT
LIMIT
10 LX1
12 LX2
SLOPE
COMPENSATION
NC 8
GND 15
OSCILLATOR
ZERO CROSS
COMPARATOR
11 PGND
1FB FOR ADP210x-ADJ (ADJUSTABLE VERSION) AND OUT_SENSE FOR ADP210x-xx (FIXED VERSION).
Figure 34. Block Diagram of the ADP2105/ADP2106/ADP2107
Rev. 0 | Page 14 of 32
06079-037
THERMAL
SHUTDOWN
EN 1
ADP2105/ADP2106/ADP2107
APPLICATIONS INFORMATION
into account when calculating resistor values. The FB bias
current can be ignored for a higher divider string current, but
this degrades efficiency at very light loads.
EXTERNAL COMPONENT SELECTION
The external component selection for the ADP2105/ADP2106/
ADP2107 application circuits shown in Figure 35 and Figure 36
depend on input voltage, output voltage, and load current
requirements. Additionally, tradeoffs between performance
parameters like efficiency and transient response can be made
by varying the choice of external components.
To limit output voltage accuracy degradation due to FB bias
current to less than 0.05% (0.5% maximum), ensure that the
divider string current is greater than 20 μA. To calculate the
desired resistor values, first determine the value of the bottom
divider string resistor, RBOT, by
SETTING THE OUTPUT VOLTAGE
RBOT =
The output voltage of ADP2105/ADP2106/ADP2107-ADJ is
externally set by a resistive voltage divider from the output
voltage to FB. The ratio of the resistive voltage divider sets the
output voltage, while the absolute value of those resistors sets
the divider string current. For lower divider string currents, the
small 10 nA (0.1 μA maximum) FB bias current should be taken
0.1μF
VIN
10Ω
VFB
I STRING
where:
VFB = 0.8 V, the internal reference.
ISTRING is the resistor divider string current.
INPUT VOLTAGE = 2.7V TO 5.5V
CIN1
VOUT
16
15
OUT_SENSE GND
ON
14
13
IN
PWIN1
LX2 12
1 EN
OFF
OUTPUT VOLTAGE = 1.2V, 1.5V, 1.8V, 3.3V
L
2 GND
ADP2105/
ADP2106/
ADP2107
3 GND
COUT
COMP
SS
5
6
VIN
CIN2
AGND NC
7
LOAD
LX1 10
PWIN2 9
4 GND
VOUT
PGND 11
8
CSS
RCOMP
06079-065
CCOMP
NC = NO CONNECT
Figure 35. Typical Applications Circuit for Fixed Output Voltage Options (ADP2105/ADP2106/ADP2107-xx)
0.1μF
VIN
10Ω
INPUT VOLTAGE = 2.7V TO 5.5V
CIN1
FB
OFF
16
15
14
13
FB
GND
IN
PWIN1
LX2 12
1
EN
2
GND
3
GND
4
GND
OUTPUT VOLTAGE
= 0.8V TO VIN
L
ADP2105/
ADP2106/
ADP2107
SS
5
6
RTOP
LX1 10
PWIN2 9
COMP
RCOMP
PGND 11
VIN
AGND NC
7
COUT
LOAD
FB
CIN2
RBOT
8
CSS
CCOMP
NC = NO CONNECT
06079-038
ON
Figure 36. Typical Applications Circuit for Adjustable Output Voltage Option (ADP2105/ADP2106/ADP2107-ADJ)
Rev. 0 | Page 15 of 32
ADP2105/ADP2106/ADP2107
Ensure that the maximum rms current of the inductor is greater
than the maximum load current, and the saturation current of
the inductor is greater than the peak current limit of the converter
used in the application.
Once RBOT is determined, calculate the value of the top resistor,
RTOP, by
⎡V − VFB ⎤
RTOP = RBOT ⎢ OUT
⎥
⎦
⎣ VFB
The ADP2105/ADP2106/ADP2107-xx (where xx represents
the fixed output voltage) include the resistive voltage divider
internally, reducing the external circuitry required. Connect the
OUT_SENSE to the output voltage as close as possible to the
load for improved load regulation.
INDUCTOR SELECTION
The high switching frequency of ADP2105/ADP2106/ADP2107
allows for minimal output voltage ripple even with small inductors.
The sizing of the inductor is a trade-off between efficiency and
transient response. A small inductor leads to larger inductor
current ripple that provides excellent transient response but
degrades efficiency. Due to the high switching frequency of
ADP2105/ADP2106/ADP2107, shielded ferrite core inductors
are recommended for their low core losses and low EMI.
As a guideline, the inductor peak-to-peak current ripple, ΔIL,
is typically set to 1/3 of the maximum load current for optimal
transient response and efficiency.
ΔI L =
VOUT × (V IN − VOUT ) I LOAD (MAX )
≈
3
V IN × f SW × L
⇒ LIDEAL =
Table 5. Minimum Inductor Value for Common Output
Voltage Options for the ADP2105 (1 A)
VOUT
1.2 V
1.5 V
1.8 V
2.5 V
3.3 V
2.7 V
1.67 μH
1.68 μH
2.02 μH
2.80 μH
3.70 μH
3.6 V
2.00 μH
2.19 μH
2.25 μH
2.80 μH
3.70 μH
VIN
4.2 V
2.14 μH
2.41 μH
2.57 μH
2.80 μH
3.70 μH
5.5 V
2.35 μH
2.73 μH
3.03 μH
3.41 μH
3.70 μH
Table 6. Minimum Inductor Value for Common Output
Voltage Options for the ADP2106 (1.5 A)
VOUT
1.2 V
1.5 V
1.8 V
2.5 V
3.3 V
2.7 V
1.11 μH
1.25 μH
1.49 μH
2.08 μH
2.74 μH
3.6 V
2.33 μH
1.46 μH
1.50 μH
2.08 μH
2.74 μH
VIN
4.2 V
2.43 μH
1.61 μH
1.71 μH
2.08 μH
2.74 μH
5.5 V
1.56 μH
1.82 μH
2.02 μH
2.27 μH
2.74 μH
Table 7. Minimum Inductor Value for Common Output
Voltage Options for the ADP2107 (2 A)
2.5 × VOUT × (VIN − VOUT )
μH
VIN × I LOAD (MAX )
where fSW is the switching frequency (1.2 MHz).
The ADP2105/ADP2106/ADP2107 use slope compensation in
the current control loop to prevent subharmonic oscillations
when operating beyond 50% duty cycle. The fixed slope compensation limits the minimum inductor value as a function of
output voltage.
VOUT
1.2 V
1.5 V
1.8 V
2.5 V
3.3 V
2.7 V
0.83 μH
0.99 μH
1.19 μH
1.65 μH
2.18 μH
3.6 V
1.00 μH
1.09 μH
1.19 μH
1.65 μH
2.18 μH
VIN
4.2 V
1.07 μH
1.21 μH
1.29 μH
1.65 μH
2.18 μH
5.5 V
1.17 μH
1.36 μH
1.51 μH
1.70 μH
2.18 μH
Table 8. Inductor Recommendations for the ADP2105/
ADP2106/ADP2107
For the ADP2105:
L > (1.12 μH/V) × VOUT
Vendor
Sumida
For the ADP2106:
L > (0.83 μH/V) × VOUT
For the ADP2107:
Toko
L > (0.66 μH/V) × VOUT
Also, 4.7 μH or larger inductors are not recommended because
they may cause instability in discontinuous conduction mode
under light load conditions.
Finally, it is important that the inductor be capable of handling
the maximum peak inductor current, IPK, determined by the
following equation:
Coilcraft
Cooper
Bussmann
⎛ ΔI ⎞
I PK = I LOAD ( MAX ) + ⎜ L ⎟
⎝ 2 ⎠
Rev. 0 | Page 16 of 32
Small-Sized Inductors
( < 5 mm × 5 mm)
CDRH2D14, 3D16,
3D28
1069AS-DB3018,
1098AS-DE2812,
1070AS-DB3020
LPS3015, LPS4012,
DO3314
SD3110, SD3112,
SD3114, SD3118,
SD3812, SD3814
Large-Sized Inductors
( > 5 mm × 5 mm)
CDRH4D18, 4D22,
4D28, 5D18, 6D12
D52LC, D518LC,
D62LCB
DO1605T
SD10, SD12, SD14, SD52
ADP2105/ADP2106/ADP2107
18
17
16
15
14
13
12
11
10
9
8
7
6
5
4
3
2
1
0
15
20
0
–20
1
–40
2
3
–60
–80
0
2
4
06079-060
–100
14.7µF 0805 X5R MURATA GRM21BR61A475K
210µF 0805 X5R MURATA GRM21BR61A106K
322µF 0805 X5R MURATA GRM21BR60J226M
6
VOLTAGE (VDC)
Figure 38. % Drop-In Capacitance vs. DC Bias for Ceramic Capacitors
(Information Provided by Murata Corporation)
For example, to get 20 μF output capacitance at an output voltage
of 2.5 V, based on Figure 38, as well as giving some margin for
temperature variance, it is suggested that a 22 μF and a 10 μF
capacitor be used in parallel to ensure that the output capacitance
is sufficient under all conditions for stable behavior.
Table 9. Recommended Input and Output Capacitor Selection
for the ADP2105/ADP2106/ADP2107
06079-070
% OVERSHOOT OF OUTPUT VOLTAGE
The output capacitor selection affects both the output voltage
ripple and the loop dynamics of the converter. For a given loop
crossover frequency (the frequency at which the loop gain
drops to 0 dB), the maximum voltage transient excursion
(overshoot) is inversely proportional to the value of the output
capacitor. Therefore, larger output capacitors result in improved
load transient response. To minimize the effects of the dc-to-dc
converter switching, the crossover frequency of the compensation
loop should be less than 1/10 of the switching frequency. Higher
crossover frequency leads to faster settling time for a load transient
response, but it can also cause ringing due to poor phase
margin. Lower crossover frequency helps to provide stable
operation but needs large output capacitors to achieve competitive
overshoot specifications. Therefore, the optimal crossover
frequency for the control loop of ADP2105/ADP2106/ADP2107
is 80 kHz, 1/15 of the switching frequency. For a crossover
frequency of 80 kHz, Figure 37 shows the maximum output
voltage excursion during a 1A load transient, as the product of
the output voltage and the output capacitor is varied. Choose
the output capacitor based on the desired load transient
response and target output voltage.
It is also important, while choosing output capacitors, to
account for the loss of capacitance due to output voltage dc bias.
Figure 38 shows the loss of capacitance due to output voltage dc
bias for a few X5R MLCC capacitors from Murata.
CAPACITANCE CHANGE (%)
OUTPUT CAPACITOR SELECTION
20
25
30
35
40
45
50
55
60
65
70
OUTPUT CAPACITOR × OUTPUT VOLTAGE (μC)
For example, if the desired 1A load transient response (overshoot)
is 5% for an output voltage of 2.5 V, then from Figure 37
Output Capacitor × Output Voltage = 50 μC
50 μ C
2 .5
Vendor
Murata
Taiyo Yuden
GRM21BR61A475K
LMK212BJ475KG
GRM21BR61A106K
LMK212BJ106KG
GRM21BR60J226M
JMK212BJ226MG
INPUT CAPACITOR SELECTION
Figure 37. % Overshoot for a 1 A Load Transient Response vs.
Output Capacitor × Output Voltage
⇒ Output Capacitor =
Capacitor
4.7 μF 10 V
X5R 0805
10 μF 10 V
X5R 0805
22 μF 6.3 V
X5R 0805
≈ 20 μ F
The ADP2105/ADP2106/ADP2107 have been designed for
operation with small ceramic output capacitors that have low
ESR and ESL, thus comfortably able to meet tight output voltage
ripple specifications. X5R or X7R dialectrics are recommended
with a voltage rating of 6.3 V or 10 V. Y5V and Z5U dialectrics
are not recommended, due to their poor temperature and dc
bias characteristics. Table 9 shows a list of recommended MLCC
capacitors from Murata and Taiyo Yuden.
The input capacitor reduces input voltage ripple caused by the
switch currents on the PWIN pins. Place the input capacitors as
close as possible to the PWIN pins. Select an input capacitor
capable of withstanding the rms input current for the maximum
load current in your application.
For the ADP2105, it is recommended that each PWIN pin be
bypassed with a 4.7 μF or larger input capacitor. For the ADP2106,
bypass the PWIN pins with a 10 μF and a 4.7 μF capacitor, and
for the ADP2107, bypass each PWIN pin with a 10 μF capacitor.
As with the output capacitor, a low ESR ceramic capacitor is
recommended to minimize input voltage ripple. X5R or X7R
dialectrics are recommended, with a voltage rating of 6.3 V or
10 V. Y5V and Z5U dialectrics are not recommended, due to
their poor temperature and dc bias characteristics. Refer to
Table 9 for input capacitor recommendations.
Rev. 0 | Page 17 of 32
ADP2105/ADP2106/ADP2107
INPUT FILTER
The IN pin is the power source for the ADP2105/ADP2106/
ADP2107 internal circuitry, including the voltage reference and
current sense amplifier that are sensitive to power supply noise.
To prevent high frequency switching noise on the PWIN pins from
corrupting the internal circuitry of the ADP2105/ADP2106/
ADP2107, a low-pass RC filter should be placed between the IN
pin and the PWIN1 pin. The suggested input filter consists of
a small 0.1 μF ceramic capacitor placed between IN and AGND
and a 10 Ω resistor placed between IN and PWIN1. This forms
a 150 kHz low-pass filter between PWIN1 and IN that prevents
any high frequency noise on PWIN1 from coupling into the
IN pin.
The transconductance error amplifier drives the compensation
network that consists of a resistor (RCOMP) and capacitor (CCOMP)
connected in series to form a pole and a zero, as shown in the
following equation:
⎛
1
ZCOMP (s) = ⎜⎜ RCOMP +
sC
COMP
⎝
⎞ ⎛ 1 + sRCOMP CCOMP
⎟=⎜
⎟ ⎜
sCCOMP
⎠ ⎝
⎞
⎟
⎟
⎠
At the crossover frequency, the gain of the open loop transfer
function is unity. This yields the following equation for the
compensation network impedance at the crossover frequency:
⎛ (2π )FCROSS ⎞⎛ COUTVOUT
⎟⎜
ZCOMP (FCROSS ) = ⎜
⎜ G G
⎟⎜ V
m CS
REF
⎝
⎠⎝
⎞
⎟
⎟
⎠
SOFT START
where:
The ADP2105/ADP2106/ADP2107 include soft start circuitry
to limit the output voltage rise time to reduce inrush current at
startup. To set the soft start period, connect a soft start capacitor
(CSS) from SS to AGND. The soft start period varies linearly
with the size of the soft start capacitor, as shown in the
following equation:
FCROSS = 80 kHz, the crossover frequency of the loop.
COUTVOUT is determined from the Output Capacitor Selection
section.
TSS = CSS × 109 ms
F
(2 π)⎛⎜ CROSS
⎝ 4
To get a soft start period of 1 ms, a 1 nF capacitor must be
connected between SS and AGND.
LOOP COMPENSATION
The ADP2105/ADP2106/ADP2107 utilize a transconductance
error amplifier to compensate the external voltage loop. The
open loop transfer function at angular frequency, s, is given by
⎛Z
(s) ⎞⎛ V
H (s) = GmGCS ⎜⎜ COMP ⎟⎟⎜⎜ REF
⎝ sCOUT ⎠⎝ VOUT
To ensure that there is sufficient phase margin at the crossover
frequency, place the Compensator Zero at 1/4 of the crossover
frequency, as shown in the following equation:
⎞R
⎟ COMP CCOMP = 1
⎠
Solving the above two simultaneous equations yields the value
for the compensation resistor and compensation capacitor, as
shown in the following equation:
⎞
⎟
⎟
⎠
⎛ (2 π)FCROSS
RCOMP = 0.8 ⎜⎜
⎝ GmGCS
CCOMP =
where:
VREF is the internal reference voltage (0.8 V).
VOUT is the nominal output voltage.
ZCOMP(s) is the impedance of the compensation network at the
angular frequency, s.
COUT is the output capacitor.
Gm is the transconductance of the error amplifier (50 μA/V
nominal).
GCS is the effective transconductance of the current loop.
GCS = 1.875 A/V for the ADP2105.
GCS = 2.8125 A/V for the ADP2106.
GCS = 3.625 A/V for the ADP2107.
Rev. 0 | Page 18 of 32
2
πFCROSS RCOMP
⎞⎛ COUT VOUT
⎟⎜
⎟⎜ V
REF
⎠⎝
⎞
⎟
⎟
⎠
ADP2105/ADP2106/ADP2107
BODE PLOTS
60
60
ADP2106
0
CROSSOVER
OUTPUT VOLTAGE = 1.8V
FREQUENCY = 87kHz
–10 INPUT VOLTAGE = 5.5V
LOAD CURRENT = 1A
–20 INDUCTOR = 2.2µH (LPS4012)
OUTPUT CAPACITOR = 22µF + 22µF
–30 COMPENSATION RESISTOR = 180kΩ
COMPENSATION CAPACITOR = 56pF
–40
1
10
100
(kHz)
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.
180
10
300
Figure 42. ADP2105 Bode Plot at VIN = 5.5 V, VOUT = 1.2 V and Load = 1 A
60
ADP2106
ADP2107
50
180
CROSSOVER
OUTPUT VOLTAGE = 1.8V
–10 INPUT VOLTAGE = 3.6V
FREQUENCY = 83kHz
LOAD CURRENT = 1A
–20 INDUCTOR = 2.2µH (LPS4012)
OUTPUT CAPACITOR = 22µF + 22µF
–30 COMPENSATION RESISTOR = 180kΩ
COMPENSATION CAPACITOR = 56pF
–40
1
10
100
(kHz)
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.
90
LOOP PHASE
135
0
180
CROSSOVER
OUTPUT VOLTAGE = 2.5V
–10 INPUT VOLTAGE = 5V
FREQUENCY = 76kHz
LOAD CURRENT = 1A
–20 INDUCTOR = 2µH (D62LCB)
OUTPUT CAPACITOR = 10µF + 4.7µF
–30 COMPENSATION RESISTOR = 70kΩ
COMPENSATION CAPACITOR = 120pF
–40
1
10
100
(kHz)
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
10% OVERSHOOT FOR A 1A LOAD TRANSIENT.
300
Figure 40. ADP2106 Bode Plot at VIN = 3.6 V, VOUT = 1.8 V, and Load = 1 A
60
10
45
PHASE
MARGIN = 65°
20
LOOP PHASE (Degrees)
135
LOOP PHASE
0
0
LOOP GAIN
30
LOOP GAIN (dB)
90
LOOP PHASE (Degrees)
45
PHASE
MARGIN = 52°
20
40
0
LOOP GAIN
06079-056
LOOP GAIN (dB)
180
CROSSOVER
OUTPUT VOLTAGE = 1.2V
FREQUENCY = 79kHz
INPUT VOLTAGE = 5.5V
LOAD CURRENT = 1A
–20 INDUCTOR = 3.3µH (SD3814)
OUTPUT CAPACITOR = 22µF + 22µF + 4.7µF
–30 COMPENSATION RESISTOR = 267kΩ
COMPENSATION CAPACITOR = 39pF
–40
1
10
100
(kHz)
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.
300
30
10
135
0
50
40
90
LOOP PHASE
–10
Figure 39. ADP2106 Bode Plot at VIN = 5.5 V, VOUT = 1.8 V and Load = 1 A
60
20
LOOP PHASE (Degrees)
135
06079-058
90
LOOP PHASE
45
PHASE
MARGIN = 49°
300
06079-059
10
0
LOOP GAIN
30
LOOP GAIN (dB)
45
PHASE
MARGIN = 48°
06079-055
LOOP GAIN (dB)
30
20
40
0
LOOP GAIN
LOOP PHASE (Degrees)
40
ADP2105
50
50
Figure 43. ADP2107 Bode Plot at VIN = 5 V, VOUT = 2.5 V and Load = 1 A
60
ADP2105
50
ADP2107
50
LOOP PHASE
0
CROSSOVER
OUTPUT VOLTAGE = 1.2V
FREQUENCY = 71kHz
–10
INPUT VOLTAGE = 3.6V
LOAD CURRENT = 1A
–20 INDUCTOR = 3.3µH (SD3814)
OUTPUT CAPACITOR = 22µF + 22µF + 4.7µF
–30 COMPENSATION RESISTOR = 267kΩ
COMPENSATION CAPACITOR = 39pF
–40
1
10
100
(kHz)
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
5% OVERSHOOT FOR A 1A LOAD TRANSIENT.
90
135
180
20
10
LOOP PHASE
0
CROSSOVER
OUTPUT VOLTAGE = 3.3V
–10 INPUT VOLTAGE = 5V
FREQUENCY = 67kHz
LOAD CURRENT = 1A
–20 INDUCTOR = 2.5µH (CDRH5D28)
OUTPUT CAPACITOR = 10µF + 4.7µF
–30 COMPENSATION RESISTOR = 70kΩ
COMPENSATION CAPACITOR = 120pF
–40
1
10
100
(kHz)
NOTES
1. EXTERNAL COMPONENTS WERE CHOSEN FOR A
10% OVERSHOOT FOR A 1A LOAD TRANSIENT.
300
Figure 41. ADP2105 Bode Plot at VIN = 3.6 V, VOUT = 1.2 V, and Load = 1 A
45
PHASE
MARGIN = 70°
90
135
180
LOOP PHASE (Degrees)
10
0
LOOP GAIN
30
300
Figure 44. ADP2107 Bode Plot at VIN = 5 V, VOUT = 3.3 V, and Load = 1 A
Rev. 0 | Page 19 of 32
06079-069
PHASE
MARGIN = 51°
20
40
LOOP GAIN (dB)
45
LOOP PHASE (Degrees)
0
30
06079-057
LOOP GAIN (dB)
LOOP GAIN
40
ADP2105/ADP2106/ADP2107
LOAD TRANSIENT RESPONSE
OUTPUT CURRENT
OUTPUT CURRENT
3
3
CH2 LOW
–51mV
CH2 LOW
–93mV
OUTPUT VOLTAGE (AC-COUPLED)
OUTPUT VOLTAGE (AC-COUPLED)
2
2
CH2 50mV~
M 10µs
CH3 1A
A CH3
06079-076
LX NODE (SWITCH NODE)
CH1 2V
1
06079-075
1
LX NODE (SWITCH NODE)
0.5A
CH1 2V
CH2 50mV~
M 10µs
CH3 1A
A CH3
0.5A
OUTPUT CAPACITOR: 22µF + 22µF + 4.7µF
INDUCTOR: SD14, 2.5µH
COMPENSATION RESISTOR: 270kΩ
COMPENSATION CAPACITOR: 39pF
OUTPUT CAPACITOR: 22µF + 4.7µF
INDUCTOR: SD14, 2.5µH
COMPENSATION RESISTOR: 135kΩ
COMPENSATION CAPACITOR: 82pF
Figure 45. 1 A Load Transient Response for ADP2105-1.2
with External Components Chosen for 5% Overshoot
Figure 48. 1 A Load Transient Response for ADP2105-1.2
with External Components Chosen for 10% Overshoot
OUTPUT CURRENT
OUTPUT CURRENT
3
3
CH2 LOW
–164mV
CH2 LOW
–112mV
2
2
OUTPUT VOLTAGE (AC-COUPLED)
OUTPUT VOLTAGE (AC-COUPLED)
CH2 100mV~ M 10µs
CH3 1A
A CH3
06079-078
LX NODE (SWITCH NODE)
CH1 2V
1
06079-077
1
LX NODE (SWITCH NODE)
CH1 2V
0.5A
CH2 100mV~ M 10µs
CH3 1A
A CH3
0.5A
OUTPUT CAPACITOR: 22µF + 22µF
INDUCTOR: SD3814, 3.3µH
COMPENSATION RESISTOR: 270kΩ
COMPENSATION CAPACITOR: 39pF
OUTPUT CAPACITOR: 10µF + 10µF
INDUCTOR: SD3814, 3.3µH
COMPENSATION RESISTOR: 135kΩ
COMPENSATION CAPACITOR: 82pF
Figure 46. 1 A Load Transient Response for ADP2105-1.8
with External Components Chosen for 5% Overshoot
Figure 49. 1 A Load Transient Response for ADP2105-1.8
with External Components Chosen for 10% Overshoot
OUTPUT CURRENT
OUTPUT CURRENT
3
3
CH2 LOW
–178mV
2
CH2 LOW
OUTPUT VOLTAGE (AC-COUPLED) –308mV
2
LX NODE (SWITCH NODE)
CH1 2V
CH2 100mV~ M 10µs
CH3 1A
1
06079-079
1
A CH3
06079-080
OUTPUT VOLTAGE (AC-COUPLED)
LX NODE (SWITCH NODE)
0.5A
CH1 2V
CH2 200mV~ M 10µs
CH3 1A
A CH3
0.5A
OUTPUT CAPACITOR: 22µF + 4.7µF
INDUCTOR: CDRH5D18, 4.1µH
COMPENSATION RESISTOR: 270kΩ
COMPENSATION CAPACITOR: 39pF
OUTPUT CAPACITOR: 10µF + 4.7µF
INDUCTOR: CDRH5D18, 4.1µH
COMPENSATION RESISTOR: 135kΩ
COMPENSATION CAPACITOR: 82pF
Figure 47. 1 A Load Transient Response for ADP2105-3.3
with External Components Chosen for 5% Overshoot
Figure 50. 1 A Load Transient Response for ADP2105-3.3
with External Components Chosen for 10% Overshoot
Rev. 0 | Page 20 of 32
ADP2105/ADP2106/ADP2107
EFFICIENCY CONSIDERATIONS
The amount of power loss can by calculated by
Efficiency is defined as the ratio of output power to input power.
The high efficiency of the ADP2105/ADP2106/ADP2107 has
two distinct advantages. First, only a small amount of power is
lost in the dc-to-dc converter package that reduces thermal
constraints. In addition, high efficiency delivers the maximum
output power for the given input power, extending battery life
in portable applications.
There are four major sources of power loss in dc-to-dc
converters like the ADP2105/ADP2106/ADP2107.
•
•
•
•
PSW = (CGATE − P + CGATE − N) × VIN2 × fSW
where:
(CGATE − P + CGATE − N) ~ 600 pF.
fSW = 1.2 MHz, the switching frequency.
Transition Losses
Transition losses occur because the P-channel MOSFET power
switch cannot turn on or turn off instantaneously. At the middle of
a LX node transition, the power switch is providing all the inductor
current, while the source to drain voltage of the power switch is
half the input voltage, resulting in power loss. Transition losses
increase with load current and input voltage and occur twice for
each switching cycle.
Power switch conduction losses
Inductor losses
Switching losses
Transition losses
Power Switch Conduction Losses
The amount of power loss can be calculated by
Power switch conduction losses are caused by the flow of output
current through the P-channel power switch and the N-channel
synchronous rectifier, which have internal resistances (RDS(ON))
associated with them. The amount of power loss can be approximated by
PTRAN =
VIN
× I OUT × (tON + tOFF ) × f SW
2
where tON and tOFF are the rise time and fall time of the LX node,
which are approximately 3 ns.
THERMAL CONSIDERATIONS
PSW − COND = [RDS(ON) − P × D + RDS(ON) − N × (1 − D)] × IOUT2
where D = VOUT/VIN.
The internal resistance of the power switches increases with
temperature but decreases with higher input voltage. Figure 19
in the Typical Performance Characteristics section shows the
change in RDS(ON) vs. input voltage, while Figure 27 in the
Typical Performance Characteristics section shows the change
in RDS(ON) vs. temperature for both power devices.
Inductor Losses
Inductor conduction losses are caused by the flow of current
through the inductor, which has an internal resistance (DCR)
associated with it. Larger sized inductors have smaller DCR,
which can improve inductor conduction losses.
Inductor core losses are related to the magnetic permeability of
the core material. Because the ADP2105/ADP2106/ADP2107
are high switching frequency dc-to-dc converters, shielded ferrite
core material is recommended for its low core losses and low EMI.
The total amount of inductor power loss can be calculated by
PL = DCR × IOUT2 + Core Losses
Switching Losses
Switching losses are associated with the current drawn by the
driver to turn on and turn off the power devices at the
switching frequency. Each time a power device gate is turned on
and turned off, the driver transfers a charge ΔQ from the input
supply to the gate and then from the gate to ground.
In most applications, the ADP2105/ADP2106/ADP2107 do not
dissipate a lot of heat due to their high efficiency. However, in
applications with high ambient temperature, low supply voltage,
and high duty cycle, the heat dissipated in the package is large
enough that it can cause the junction temperature of the die to
exceed the maximum junction temperature of 125°C. Once the
junction temperature exceeds 140°C, the converter goes into
thermal shutdown. It recovers only after the junction temperature
has decreased below 100°C to prevent any permanent damage.
Therefore, thermal analysis for the chosen application solution
is very important to guarantee reliable performance over all
conditions.
The junction temperature of the die is the sum of the ambient
temperature of the environment and the temperature rise of the
package due to the power dissipation, as shown in the following
equation:
TJ = TA + TR
where:
TJ is the junction temperature.
TA is the ambient temperature.
TR is the rise in temperature of the package due to power
dissipation in it.
Rev. 0 | Page 21 of 32
ADP2105/ADP2106/ADP2107
The rise in temperature of the package is directly proportional
to the power dissipation in the package. The proportionality
constant for this relationship is defined as the thermal
resistance from the junction of the die to the ambient
temperature, as shown in the following equation:
2.
See whether the output voltage desired is available as a
fixed output voltage option. Because 2 V is not one of the
fixed output voltage options available, choose the adjustable
version of ADP2106.
3.
The first step in external component selection for an
adjustable version converter is to calculate the resistance of
the resistive voltage divider that sets the output voltage.
TR = θJA × PD
where:
RBOT =
TR is the rise in temperature of the package.
PD is the power dissipation in the package.
θJA is the thermal resistance from the junction of the die to the
ambient temperature of the package.
For example, consider an application where the ADP2107-1.8
is used with an input voltage of 3.6 V and a load current of 2 A.
Also, assume that the maximum ambient temperature is 85°C.
At a load current of 2 A, the most significant contributor of
power dissipation in the dc-to-dc converter package is the
conduction loss of the power switches. Using the graph of
switch resistance vs. temperature (see Figure 27), as well as the
equation of power loss given in the Power Switch Conduction
Losses section, the power dissipation in the package can be
calculated by
⎡ 2 V − 0.8 V ⎤
⎡V
− VFB ⎤
RTOP = RBOT ⎢ OUT
⎥ = 60 kΩ
⎥ = 40 kΩ × ⎢
⎢⎣ 0.8 V ⎦⎥
⎣ VFB
⎦
4.
Calculate the minimum inductor value as follows:
For the ADP2106:
L > (0.83 μH/V) × VOUT
Ö L > 0.83 μH/V × 2 V
Ö L > 1.66 μH
Next, calculate the ideal inductor value that sets the
inductor peak-to-peak current ripple, ΔIL, to1/3 of the
maximum load current at the maximum input voltage.
PSW − COND = [RDS(ON) − P × D + RDS(ON) − N × (1 − D)] × IOUT2 =
[109 mΩ × 0.5 + 90 mΩ × 0.5] × (2 A)2 ~ 400 mW
The θJA for the LFCSP_VQ package is 40°C/W, as shown in
Table 3. Thus, the rise in temperature of the package due to
power dissipation is
LIDEAL =
2.5 × VOUT × (VIN − VOUT )
μH =
VIN × I LOAD (MAX )
2.5 × 2 × (4.2 − 2)
μH = 2.18 μH
4. 2 × 1 .2
TR = θJA × PD = 40°C/W × 0.40 W = 16°C
The junction temperature of the converter is
The closest standard inductor value is 2.2 μH. The
maximum rms current of the inductor should be greater
than 1.2 A, and the saturation current of the inductor
should be greater than 2 A. One inductor that meets these
criteria is the LPS4012-2.2 μH from Coilcraft.
TJ = TA + TR = 85°C + 16°C = 101°C
which is below the maximum junction temperature of 125°C.
Thus, this application operates reliably from a thermal point
of view.
5.
DESIGN EXAMPLE
Consider an application with the following specifications:
Input Voltage = 3.6 V to 4.2 V.
Output Voltage = 2 V.
Typical Output Current = 600 mA.
Maximum Output Current = 1.2 A.
Soft Start Time = 2 ms.
Overshoot ≤ 100 mV under all load transient conditions.
1.
0.8 V
VFB
=
= 40 kΩ
I STRING 20 μA
Choose the output capacitor based on the transient
response requirements. The worst-case load transient is
1.2 A, for which the overshoot must be less than 100 mV,
which is 5% of the output voltage. Therefore, for a 1 A load
transient, the overshoot must be less than 4% of the output
voltage. For these conditions, Figure 37 gives
Output Capacitor × Output Voltage = 60 μC
⇒ Output Capacitor =
Choose the dc-to-dc converter that satisfies the maximum
output current requirement. Because the maximum output
current for this application is 1.2 A, the ADP2106 with a
maximum output current of 1.5 A is ideal for this
application.
Rev. 0 | Page 22 of 32
60 μC
2 .0 V
≈ 30 μF
Next, taking into account the loss of capacitance due to dc
bias, as shown in Figure 38, two 22 μF X5R MLCC capacitors
from Murata (GRM21BR60J226M) are sufficient for this
application.
ADP2105/ADP2106/ADP2107
6.
Because the ADP2106 is being used in this application, the
input capacitors are 10 μF and 4.7 μF X5R Murata capacitors
(GRM21BR61A106K and GRM21BR61A475K).
7.
The input filter consists of a small 0.1 μF ceramic capacitor
placed between IN and AGND and a 10 Ω resistor placed
between IN and PWIN1.
8.
Choose a soft start capacitor of 2 nF to achieve a soft start
time of 2 ms.
9.
Finally, the compensation resistor and capacitor can be
calculated as
⎛ (2 π)FCROSS
RCOMP = 0.8 ⎜⎜
⎝ GmGCS
⎞
⎟
⎟
⎠
⎛
⎞⎛ 30 μF × 2 V ⎞
(2 π) × 80 kHz
⎟⎜
⎟ = 215 kΩ
= 0 .8 ⎜
⎜ 50 μA / V × 2.8125 A / V ⎟⎜ 0.8 V ⎟
⎝
⎠⎝
⎠
CCOMP =
Rev. 0 | Page 23 of 32
⎞⎛ COUT VOUT
⎟⎜
⎟⎜ V
REF
⎠⎝
2
2
=
= 39 pF
πFCROSS RCOMP π × 80 kHz × 215 kΩ
ADP2105/ADP2106/ADP2107
EXTERNAL COMPONENT RECOMMENDATIONS
Table 10. Recommended External Components for Popular Output Voltage Options at 80 kHz Crossover Frequency with
10% Overshoot for a 1 A Load Transient (Refer to Figure 35 and Figure 36)
Part
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2105-1.2
ADP2105-1.5
ADP2105-1.8
ADP2105-3.3
ADP2106-1.2
ADP2106-1.5
ADP2106-1.8
ADP2106-3.3
ADP2107-1.2
ADP2107-1.5
ADP2107-1.8
ADP2107-3.3
VOUT (V)
0.9
1.2
1.5
1.8
2.5
3.3
0.9
1.2
1.5
1.8
2.5
3.3
0.9
1.2
1.5
1.8
2.5
3.3
1.2
1.5
1.8
3.3
1.2
1.5
1.8
3.3
1.2
1.5
1.8
3.3
CIN1 1 (μF)
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
10
10
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
CIN2 2 (μF)
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
10
10
10
10
10
10
10
10
4.7
4.7
4.7
4.7
10
10
10
10
10
10
10
10
COUT 3 (μF)
22 + 10
22 + 4.7
10 + 10
10 + 10
10 + 4.7
10 + 4.7
22 + 10
22 + 4.7
10 + 10
10 + 10
10 + 4.7
10 + 4.7
22 + 10
22 + 4.7
10 + 10
10 + 10
10 + 4.7
10 + 4.7
22 + 4.7
10 + 10
10 + 10
10 + 4.7
22 + 4.7
10 + 10
10 + 10
10 + 4.7
22 + 4.7
10 + 10
10 + 10
10 + 4.7
L (μH)
2.0
2.5
3.0
3.3
3.6
4.1
1.5
1.8
2.0
2.2
2.5
3.0
1.2
1.5
1.5
1.8
1.8
2.5
2.5
3.0
3.3
4.1
1.8
2.0
2.2
3.0
1.5
1.5
1.8
2.5
1
4.7 μF 0805 X5R 10 V Murata–GRM21BR61A475KA73L.
10 μF 0805 X5R 10 V Murata–GRM21BR61A106KE19L.
2
4.7 μF 0805 X5R 10 V Murata–GRM21BR61A475KA73L.
10 μF 0805 X5R 10 V Murata–GRM21BR61A106KE19L.
3
4.7 μF 0805 X5R 10 V Murata–GRM21BR61A475KA73L.
10 μF 0805 X5R 10 V Murata–GRM21BR61A106KE19L.
22 μF 0805 X5R 6.3 V Murata–GRM21BR60J226ME39L.
4
0.5% accuracy resistor.
5
0.5% accuracy resistor.
Rev. 0 | Page 24 of 32
RCOMP (kΩ)
135
135
135
135
135
135
90
90
90
90
90
90
70
70
70
70
70
70
135
135
135
135
90
90
90
90
70
70
70
70
CCOMP (pF)
82
82
82
82
82
82
100
100
100
100
100
100
120
120
120
120
120
120
82
82
82
82
100
100
100
100
120
120
120
120
RTOP 4 (kΩ)
5
20
35
50
85
125
5
20
35
50
85
125
5
20
35
50
85
125
-
RBOT 5 (kΩ)
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
-
ADP2105/ADP2106/ADP2107
Table 11. Recommended External Components for Popular Output Voltage Options at 80 kHz Crossover Frequency with
5% Overshoot for a 1 A Load Transient (Refer to Figure 35 and Figure 36)
Part
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2105-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2106-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2107-ADJ
ADP2105-1.2
ADP2105-1.5
ADP2105-1.8
ADP2105-3.3
ADP2106-1.2
ADP2106-1.5
ADP2106-1.8
ADP2106-3.3
ADP2107-1.2
ADP2107-1.5
ADP2107-1.8
ADP2107-3.3
VOUT (V)
0.9
1.2
1.5
1.8
2.5
3.3
0.9
1.2
1.5
1.8
2.5
3.3
0.9
1.2
1.5
1.8
2.5
3.3
1.2
1.5
1.8
3.3
1.2
1.5
1.8
3.3
1.2
1.5
1.8
3.3
CIN1 1 (μF)
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
10
10
4.7
4.7
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
CIN2 2
(μF)
4.7
4.7
4.7
4.7
4.7
4.7
10
10
10
10
10
10
10
10
10
10
10
10
4.7
4.7
4.7
4.7
10
10
10
10
10
10
10
10
COUT 3 (μF)
22 + 22 + 22
22 + 22 + 4.7
22 + 22
22 + 22
22 + 10
22 + 4.7
22 + 22 + 22
22 + 22 + 4.7
22 + 22
22 + 22
22 + 10
22 + 4.7
22 + 22 + 22
22 + 22 + 4.7
22 + 22
22 + 22
22 + 10
22 + 4.7
22 + 22 + 4.7
22 + 22
22 + 22
22 + 4.7
22 + 22 + 4.7
22 + 22
22 + 22
22 + 4.7
22 + 22 + 4.7
22 + 22
22 + 22
22 + 4.7
L (μH)
2.0
2.5
3.0
3.3
3.6
4.1
1.5
1.8
2.0
2.2
2.5
3.0
1.2
1.5
1.5
1.8
1.8
2.5
2.5
3.0
3.3
4.1
1.8
2.0
2.2
3.0
1.5
1.5
1.8
2.5
1
4.7μF 0805 X5R 10V Murata – GRM21BR61A475KA73L
10μF 0805 X5R 10V Murata – GRM21BR61A106KE19L
2
4.7μF 0805 X5R 10V Murata – GRM21BR61A475KA73L
10μF 0805 X5R 10V Murata – GRM21BR61A106KE19L
3
4.7μF 0805 X5R 10V Murata – GRM21BR61A475KA73L
10μF 0805 X5R 10V Murata – GRM21BR61A106KE19L
22μF 0805 X5R 6.3V Murata – GRM21BR60J226ME39L
4
0.5% Accuracy Resistor
5
0.5% Accuracy Resistor
Rev. 0 | Page 25 of 32
RCOMP (kΩ)
270
270
270
270
270
270
180
180
180
180
180
180
140
140
140
140
140
140
270
270
270
270
180
180
180
180
140
140
140
140
CCOMP (pF)
39
39
39
39
39
39
56
56
56
56
56
56
68
68
68
68
68
68
39
39
39
39
56
56
56
56
68
68
68
68
RTOP 4 (kΩ)
5
20
35
50
85
125
5
20
35
50
85
125
5
20
35
50
85
125
-
RBOT 5 (kΩ)
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
40
-
ADP2105/ADP2106/ADP2107
CIRCUIT BOARD LAYOUT RECOMMENDATIONS
Good circuit board layout is essential in obtaining the best
performance from the ADP2105/ADP2106/ADP2107. Poor
circuit layout degrades the output ripple, as well as the
electromagnetic interference (EMI) and electromagnetic
compatibility (EMC) performance.
Figure 52 and Figure 53 show the ideal circuit board layout for
the ADP2105/ADP2106/ADP2107. Use this layout to achieve
the highest performance. Refer to the following guidelines if
adjustments to the suggested layout are needed.
•
Use separate analog and power ground planes. Connect
the ground reference of sensitive analog circuitry (such as
compensation and output voltage divider components) to
analog ground; connect the ground reference of power
components (such as input and output capacitors) to power
ground. In addition, connect both the ground planes to the
exposed pad of the ADP2105/ADP2106/ADP2107.
•
For each PWIN pin, place an input capacitor as close to the
PWIN pin as possible and connect the other end to the closest
power ground plane.
•
Place the 0.1 μF, 10 Ω low-pass input filter between the IN
pin and the PWIN1 pin, as close to the IN pin as possible.
•
Ensure that the high current loops are as short and as wide
as possible. Make the high current path from CIN through
L, COUT, and the PGND plane back to CIN as short as possible.
To accomplish this, ensure that the input and output
capacitors share a common PGND plane.
Also, make the high current path from PGND pin of the
ADP2105/ADP2106/ADP2107 through L and COUT back
to the PGND plane as short as possible. To do this, ensure
that the PGND pin of the ADP2105/ADP2106/ADP2107
is tied to the PGND plane as close as possible to the input
and output capacitors.
•
Place the feedback resistor divider network as close as
possible to the FB pin to prevent noise pickup. Try to
minimize the length of trace connecting the top of the
feedback resistor divider to the output while keeping away
from the high current traces and the switch node (LX) that
can lead to noise pickup. To reduce noise pickup, place an
analog ground plane on either side of the FB trace. For the
low fixed voltage options (1.2 V and 1.5 V), poor routing
of the OUT_SENSE trace can lead to noise pickup, adversely
affecting load regulation. This can be fixed by placing a 1 nF
bypass capacitor close to the OUT_SENSE pin.
•
The placement and routing of the compensation components
are critical for proper behavior of the ADP2105/ADP2106/
ADP2107. The compensation components should be placed
as close to the COMP pin as possible. It is advisable to use
0402-sized compensation components for closer placement,
leading to smaller parasitics. Surround the compensation
components with analog ground plane to prevent noise
pickup. Also, ensure that the metal layer under the
compensation components is the analog ground plane.
Rev. 0 | Page 26 of 32
ADP2105/ADP2106/ADP2107
EVALUATION BOARD
EVALUATION BOARD SCHEMATIC (ADP2107-1.8)
C7
0.1µF
VCC
R3
10Ω
VIN
VCC
C1
10µF1
OUT
U1
J1
GND
16
15
OUT_SENSE
1
EN
2
GND
INPUT VOLTAGE = 2.7V TO 5.5V
14
13
GND IN
PWIN1
LX2 12
EN
PGND 11
ADP2107-1.8
3
GND
LX1 10
4
GND
PWIN2 9
1
L12
2µH
VCC
6
7
17
R4
0Ω
OUT
8
GND
R5
NS
R1
140kΩ
C6
68pF
C4
22µF1
C3
22µF1
C2
10µF1
COMP SS AGND PADDLE NC
5
OUTPUT VOLTAGE = 1.8V, 2A
VOUT
2
1 MURATA
X5R 0805
10μF: GRM21BR61A106KE19L
22μF: GRM21BR60J226ME39L
2 2μH INDUCTOR D62LCB TOKO
C5
1nF
NC = NO CONNECT
06079-044
R2
100kΩ
Figure 51. Evaluation Board Schematic of the ADP2107-1.8 (Bold Traces Are High Current Paths)
RECOMMENDED PCB BOARD LAYOUT (EVALUATION BOARD LAYOUT)
JUMPER TO ENABLE
ENABLE
GROUND
VIN
100kΩ PULL-DOWN
GROUND
INPUT
INPUT CAPACITOR
POWER GROUND
PLANE
PLACE THE FEEDBACK RESISTORS AS
CLOSE TO THE FB PIN AS POSSIBLE.
RTOP RBOT
CONNECT THE GROUND RETURN OF
ALL POWER COMPONENTS SUCH AS
INPUT AND OUTPUT CAPACITORS TO
THE POWER GROUND PLANE.
OUTPUT CAPACITOR
CIN
COUT
LX
OUTPUT
PGND
ADP2105/ADP2106/ADP2107
VOUT
LX
RCOMP
CIN
CCOMP
PLACE THE COMPENSATION
COMPONENTS AS CLOSE TO
THE COMP PIN AS POSSIBLE.
INDUCTOR (L)
COUT
OUTPUT CAPACITOR
CSS
ANALOG GROUND PLANE
POWER GROUND
INPUT CAPACITOR
06079-045
CONNECT THE GROUND RETURN OF ALL
SENSITIVE ANALOG CIRCUITRY SUCH AS
COMPENSATION AND OUTPUT VOLTAGE
DIVIDER TO THE ANALOG GROUND PLANE.
Figure 52. Recommended Layout of Top Layer of ADP2105/ADP2106/ADP2107
Rev. 0 | Page 27 of 32
ADP2105/ADP2106/ADP2107
ENABLE
VIN
GND
GND
ANALOG GROUND PLANE
POWER GROUND PLANE
INPUT VOLTAGE PLANE
CONNECTING THE TWO
PWIN PINS AS CLOSE
AS POSSIBLE.
VIN
VOUT
CONNECT THE PGND PIN
TO THE POWER GROUND
PLANE AS CLOSE TO THE
ADP2105/ADP2106/ADP2107
AS POSSIBLE.
FEEDBACK TRACE: THIS TRACE CONNECTS THE TOP OF THE
RESISTIVE VOLTAGE DIVIDER ON THE FB PIN TO THE OUTPUT.
PLACE THIS TRACE AS FAR AWAY FROM THE LX NODE AND HIGH
CURRENT TRACES AS POSSIBLE TO PREVENT NOISE PICKUP.
Figure 53. Recommended Layout of Bottom Layer of ADP2105/ADP2106/ADP2107
Rev. 0 | Page 28 of 32
06079-046
CONNECT THE EXPOSED PAD OF
THE ADP2105/ADP2106/ADP2107
TO A LARGE GROUND PLANE TO
AID POWER DISSIPATION.
ADP2105/ADP2106/ADP2107
APPLICATION CIRCUITS
0.1μF
VIN
10Ω
INPUT VOLTAGE = 5V
10μF1
VOUT
16
14
GND IN
13
PWIN1
LX2 12
1
EN
2
GND
2.5μH2
PGND 11
ADP2107-3.3
3
GND
LX1 10
4
GND
PWIN2 9
COMP
SS
5
6
10μF1
8
1nF
70kΩ
OUTPUT VOLTAGE = 3.3V
4.7μF1
LOAD
0A TO 2A
VIN
AGND NC
7
VOUT
10μF1
1 MURATA
X5R 0805
10μF: GRM21BR61A106KE19L
4.7μF: GRM21BR61A475KA73L
2 SUMIDA CDRH5D28: 2.5μH
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 10% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
120pF
06079-047
OFF
15
OUT_SENSE
ON
Figure 54. Application Circuit—VIN = 5 V, VOUT = 3.3 V, LOAD = 0 A to 2 A
0.1μF
VIN
10Ω
INPUT VOLTAGE = 3.6V
10μF1
VOUT
16
1
EN
2
GND
14
GND IN
13
PWIN1
LX2 12
1.5μH2
PGND 11
ADP2107-1.5
3
GND
LX1 10
4
GND
PWIN2 9
COMP
SS
5
6
10μF1
8
1nF
140kΩ
OUTPUT VOLTAGE = 1.5V
22μF1
LOAD
0A TO 2A
VIN
AGND NC
7
VOUT
22μF1
1 MURATA
X5R 0805
10μF: GRM21BR61A106KE19L
22μF: GRM21BR60J226ME39L
2 TOKO D62LCB OR COILCRAFT LPS4012
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 5% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
68pF
06079-048
OFF
15
OUT_SENSE
ON
Figure 55. Application Circuit—VIN = 3.6 V, VOUT = 1.5 V, LOAD = 0 A to 2 A
0.1μF
VIN
10Ω
INPUT VOLTAGE = 2.7V TO 4.2V
4.7μF1
VOUT
16
14
GND IN
13
PWIN1
LX2 12
1
EN
2
GND
2.7μH2
PGND 11
ADP2105-1.8
3
GND
LX1 10
4
GND
PWIN2 9
COMP
SS
5
6
270kΩ
39pF
AGND NC
1nF
7
VOUT
22μF1
8
OUTPUT VOLTAGE = 1.8V
22μF1
LOAD
0A TO 1A
VIN
4.7μF1
1 MURATA
X5R 0805
4.7μF: GRM21BR61A475KA73L
22μF: GRM21BR60J226ME39L
2 TOKO 1098AS-DE2812: 2.7μH
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 5% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
Figure 56. Application Circuit—VIN = Li-Ion Battery, VOUT = 1.8 V, LOAD = 0 A to 1 A
Rev. 0 | Page 29 of 32
06079-049
OFF
15
OUT_SENSE
ON
ADP2105/ADP2106/ADP2107
0.1μF
INPUT VOLTAGE = 2.7V TO 4.2V
VIN
10Ω
4.7μF1
VOUT
16
15
13
PWIN1
LX2 12
1
EN
2
GND
2.4μH2
VOUT
PGND 11
22μF1
ADP2105-1.2
3
GND
LX1 10
4
GND
PWIN2 9
COMP
SS
5
6
4.7μF1
1 MURATA
X5R 0805
4.7μF: GRM21BR61A475KA73L
22μF: GRM21BR60J226ME39L
2 TOKO 1069AS-DB3018HCT OR
TOKO 1070AS-DB3020HCT
8
1nF
135kΩ
LOAD
0A TO 1A
VIN
AGND NC
7
OUTPUT VOLTAGE = 1.2V
4.7μF1
82pF
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 10% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
06079-050
OFF
14
GND IN
OUT_SENSE
ON
Figure 57. Application Circuit—VIN = Li-Ion Battery, VOUT = 1.2 V, LOAD = 0 A to 1 A
0.1μF
VIN
10Ω
INPUT VOLTAGE = 5V
10μF1
FB
OFF
16
15
14
13
FB
GND
IN
PWIN1
LX2 12
1
EN
2
GND
2.5μH2
ADP2106-ADJ
85kΩ
3
GND
LX1 10
4
GND
PWIN2 9
COMP SS
5
180kΩ
56pF
OUTPUT VOLTAGE = 2.5V
PGND 11
AGND NC
6
1nF
7
10μF1
22μF1
LOAD
0A TO 1.5A
FB
VIN
40kΩ
4.7μF1
8
1 MURATA
X5R 0805
4.7μF: GRM21BR61A475KA73L
10μF: GRM21BR61A106KE19L
22μF: GRM21BR60J226ME39L
2 COILTRONICS SD14: 2.5μH
NOTES
1. NC = NO CONNECT.
2. EXTERNAL COMPONENTS WERE
CHOSEN FOR A 5% OVERSHOOT
FOR A 1A LOAD TRANSIENT.
Figure 58. Application Circuit—VIN = 5 V, VOUT = 2.5 V, LOAD = 0 A to 1.5 A
Rev. 0 | Page 30 of 32
06079-051
ON
ADP2105/ADP2106/ADP2107
OUTLINE DIMENSIONS
4.00
BSC SQ
PIN 1
INDICATOR
0.65 BSC
TOP
VIEW
12° MAX
3.75
BSC SQ
0.75
0.60
0.50
(BOTTOM VIEW)
13
12
PIN 1
INDICATOR
16
1
2.25
2.10 SQ
1.95
EXPOSED
PAD
9
8
4
5
0.25 MIN
1.95 BSC
0.80 MAX
0.65 TYP
0.05 MAX
0.02 NOM
SEATING
PLANE
0.35
0.30
0.25
0.20 REF
COPLANARITY
0.08
COMPLIANT TO JEDEC STANDARDS MO-220-VGGC
010606-0
1.00
0.85
0.80
0.60 MAX
0.60 MAX
Figure 59. 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
4 mm × 4 mm Body, Very Thin Quad
(CP-16-4)
Dimensions shown in millimeters
ORDERING GUIDE
Model
ADP2105ACPZ-1.2-R7 1
ADP2105ACPZ-1.5-R71
ADP2105ACPZ-1.8-R71
ADP2105ACPZ-3.3-R71
ADP2105ACPZ-R71
ADP2106ACPZ-1.2-R71
ADP2106ACPZ-1.5-R71
ADP2106ACPZ-1.8-R71
ADP2106ACPZ-3.3-R71
ADP2106ACPZ-R71
ADP2107ACPZ-1.2-R71
ADP2107ACPZ-1.5-R71
ADP2107ACPZ-1.8-R71
ADP2107ACPZ-3.3-R71
ADP2107ACPZ-R71
ADP2105-1.8-EVAL
ADP2105-EVAL
ADP2106-1.8-EVAL
ADP2106-EVAL
ADP2107-1.8-EVAL
ADP2107-EVAL
1
Output
Current
1A
1A
1A
1A
1A
1.5 A
1.5 A
1.5 A
1.5 A
1.5 A
2A
2A
2A
2A
2A
Junction
Temperature
Range
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
−40°C to +125°C
Output Voltage
1.2 V
1.5 V
1.8 V
3.3 V
ADJ
1.2 V
1.5 V
1.8 V
3.3 V
ADJ
1.2 V
1.5 V
1.8 V
3.3 V
ADJ
1.8 V
Adjustable, but set to 2.5 V
1.8 V
Adjustable, but set to 2.5 V
1.8 V
Adjustable, but set to 2.5 V
Z = Pb-free part.
Rev. 0 | Page 31 of 32
Package Description
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
16-Lead LFCSP_VQ
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
Evaluation Board
Package Option
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
CP-16-4
ADP2105/ADP2106/ADP2107
NOTES
©2006 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D06079-0-7/06(0)
Rev. 0 | Page 32 of 32