AD ADP1610

1.2 MHz DC-DC Step-Up Switching Converter
ADP1610
FEATURES
GENERAL DESCRIPTION
Fully integrated 1.2 A , 0.2 Ω, power switch
Pin-selectable 700 kHz or 1.2 MHz PWM frequency
92% efficiency
Adjustable output voltage up to 12 V
3% output regulation accuracy
Adjustable soft start
Input undervoltage lockout
MSOP 8-lead package
The ADP1610 is a dc-to-dc step-up switching converter with an
integrated 1.2 A, 0.2 Ω power switch capable of providing an
output voltage as high as 12 V. With a package height of less that
1.1 mm, the ADP1610 is optimal for space-constrained
applications such as portable devices or thin film transistor
(TFT) liquid crystal displays (LCDs).
The ADP1610 operates in pulse-width modulation (PWM)
current mode with up to 92% efficiency. Adjustable soft start
prevents inrush currents at startup. The pin-selectable switching
frequency and PWM current-mode architecture allow excellent
transient response, easy noise filtering, and the use of small,
cost-saving external inductors and capacitors.
APPLICATIONS
TFT LC bias supplies
Portable applications
Industrial/instrumentation equipment
The ADP1610 is offered in the Pb-free 8-lead MSOP and
operates over the temperature range of −40°C to +85°C.
FUNCTIONAL BLOCK DIAGRAM
COMP
IN
1
6
ERROR
AMP
REF
ADP1610
gm
BIAS
FB 2
5 SW
F/F
R Q
S
RAMP
GEN
DRIVER
COMPARATOR
SS 8
SD
OSC
SOFT START
CURRENT
SENSE
AMPLIFIER
3
4
GND
04472-001
RT 7
Figure 1.
Rev. 0
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However, no responsibility is assumed by Analog Devices for its use, nor for any
infringements of patents or other rights of third parties that may result from its use.
Specifications subject to change without notice. No license is granted by implication
or otherwise under any patent or patent rights of Analog Devices. Trademarks and
registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781.329.4700
www.analog.com
Fax: 781.326.8703
© 2004 Analog Devices, Inc. All rights reserved.
ADP1610
TABLE OF CONTENTS
Specifications..................................................................................... 3
Choosing the Input and Output Capacitors ........................... 11
Absolute Maximum Ratings............................................................ 4
Diode Selection........................................................................... 12
ESD Caution.................................................................................. 4
Loop Compensation .................................................................. 12
Pin Configuration and Function Descriptions............................. 5
Soft Start Capacitor .................................................................... 13
Typical Performance Characteristics ............................................. 6
Application Circuits ................................................................... 13
Theory of Operation ...................................................................... 10
DC-DC Step-Up Switching Converter with True Shutdown14
Current-Mode PWM Operation .............................................. 10
TFT LCD Bias Supply ................................................................ 14
Frequency Selection ................................................................... 10
Sepic Power Supply .................................................................... 14
Soft Start ...................................................................................... 10
Layout Procedure ........................................................................... 15
On/Off Control........................................................................... 10
Outline Dimensions ....................................................................... 16
Setting the Output Voltage ........................................................ 10
Ordering Guide .......................................................................... 16
REVISION HISTORY
10/04—Revision 0: Initial Version
Rev. 0 | Page 2 of 16
ADP1610
SPECIFICATIONS
VIN = 3.3 V, TA = −40°C to +85°C, unless otherwise noted.
All limits at temperature extremes are guaranteed by correlation and characterization using standard statistical quality control (SQC),
unless otherwise noted.
Table 1.
Parameter
SUPPLY
Input Voltage
Quiescent Current
Nonswitching State
Shutdown
Switching State1
OUTPUT
Output Voltage
Load Regulation
Overall Regulation
REFERENCE
Feedback Voltage
Line Regulation
ERROR AMPLIFIER
Transconductance
Voltage Gain
FB Input Bias Current
SWITCH
SW On Resistance
SW Leakage Current
Peak Current Limit2
OSCILLATOR
Oscillator Frequency
Maximum Duty Cycle
SHUTDOWN
Shutdown Input Voltage Low
Shutdown Input Voltage High
Shutdown Input Bias Current
SOFT START
SS Charging Current
UNDERVOLTAGE LOCKOUT3
UVLO Threshold
UVLO Hysteresis
1
2
3
Symbol
Conditions
VIN
Min
Typ
2.5
Max
Unit
5.5
V
IQ
IQSD
VFB = 1.3 V, RT = VIN
VSD = 0 V
390
0.01
600
10
µA
µA
IQSW
fSW = 1.23 MHz, no load
1
2
mA
12
V
mV/mA
%
1.248
+0.15
V
%/V
VOUT
VIN
ILOAD = 10 mA to 150 mA, VOUT = 10 V
Line, load, temperature
VFB
VIN = 2.5 V to 5.5 V
gm
AV
0.05
±3
1.212
−0.15
∆I = 1 µA
100
60
10
VFB = 1.23 V
RON
ISW = 1.0 A
VSW = 12 V
ICLSET
fOSC
DMAX
RT = GND
RT = IN
COMP = open, VFB = 1 V, RT = GND
0.49
0.89
78
VIL
VIH
ISD
Nonswitching state
Switching state
VSD = 3.3 V
2.2
This parameter specifies the average current while switching internally and with SW (Pin 5) floating.
Guaranteed by design and not fully production tested.
Guaranteed by characterization.
Rev. 0 | Page 3 of 16
µA/V
dB
nA
200
0.01
2.0
400
20
mΩ
µA
A
0.7
1.23
83
0.885
1.6
90
MHz
MHz
%
0.6
V
V
µA
0.01
VSS = 0 V
VIN rising
1.230
1
3
2.2
2.4
220
µA
2.5
V
mV
ADP1610
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
IN, COMP, SD, SS, RT, FB to GND
SW to GND
RMS SW Pin Current
Operating Ambient Temperature Range
Operating Junction Temperature Range
Storage Temperature Range
θJA, Two Layers
θJA, Four Layers
Lead Temperature Range (Soldering, 60 s)
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only and functional operation of the device at these or
any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability. Absolute maximum ratings apply individually
only, not in combination. Unless otherwise specified, all other
voltages are referenced to GND.
Rating
−0.3 V to +6 V
14 V
1.2 A
−40°C to +85°C
−40°C to +125°C
−65°C to +150°C
206°C/W
142°C/W
300°C
IN
RC
CC
VOUT
COMP
1
6
ERROR
AMP
R1
REF
L1
ADP1610
BIAS
gm
FB
CIN
IN
2
R2
SW
VIN
1.2MHz
VOUT
COUT
R Q
RAMP
GEN
D1
5
F/F
S
DRIVER
COMPARATOR
RT
7
OSC
700kHz
SD 3
CSS
8
SOFT START
CURRENT
SENSE
AMPLIFIER
4
GND
Figure 2. Block Diagram and Typical Application Circuit
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on
the human body and test equipment and can discharge without detection. Although this product features
proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy
electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance
degradation or loss of functionality.
Rev. 0 | Page 4 of 16
04472-002
SS
ADP1610
COMP 1
FB 2
ADP1610
SD 3
TOP VIEW
(Not to Scale)
GND 4
8
SS
7
RT
6
IN
5
SW
04472-003
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
Figure 3. Pin Configuration
Table 3. Pin Function Descriptions
Pin No.
1
Mnemonic
COMP
2
FB
3
4
5
SD
GND
SW
6
IN
7
RT
8
SS
Description
Compensation Input. Connect a series resistor-capacitor network from COMP to GND to compensate the
regulator.
Output Voltage Feedback Input. Connect a resistive voltage divider from the output voltage to FB to set the
regulator output voltage.
Shutdown Input. Drive SD low to shut down the regulator; drive SD high to turn it on.
Ground.
Switching Output. Connect the power inductor from the input voltage to SW and connect the external rectifier
from SW to the output voltage to complete the step-up converter.
Main Power Supply Input. IN powers the ADP1610 internal circuitry. Connect IN to the input source voltage.
Bypass IN to GND with a 10 µF or greater capacitor as close to the ADP1610 as possible.
Frequency Setting Input. RT controls the switching frequency. Connect RT to GND to program the oscillator to
700 kHz, or connect RT to IN to program it to 1.2 MHz.
Soft Start Timing Capacitor Input. A capacitor from SS to GND brings up the output slowly at power-up.
Rev. 0 | Page 5 of 16
ADP1610
TYPICAL PERFORMANCE CHARACTERISTICS
100
VIN = 2.5V
VIN = 5.5V
VIN = 3.3V
VIN = 2.5V
80
EFFICIENCY (%)
70
VOUT = 7.5V
FSW = 1.2MHz
L = 4.7µH
90
VIN = 3.3V
80
EFFICIENCY (%)
100
VIN = 5.5V
VOUT = 10V
FSW = 700kHz
L = 10µH
90
60
50
40
30
70
60
50
20
0
1
10
100
LOAD CURRENT (mA)
04472-008
40
04472-005
10
30
1000
1
Figure 4. Output Efficiency vs. Load Current
100
90
CURRENT LIMIT (A)
VIN = 2.5V
70
60
50
40
30
20
VIN = 5.5V
2.0
VIN = 3.3V
1.8
VIN = 2.5V
1.6
0
1
10
100
LOAD CURRENT (mA)
1.2
–40
1000
Figure 5. Output Efficiency vs. Load Current
10
35
AMBIENT TEMPERATURE (°C)
60
85
1.4
VOUT = 7.5V
FSW = 700kHz
L = 10µH
OSCILLATORY FREQUENCY (MHz)
VIN = 5.5V
VIN = 3.3V
VIN = 2.5V
80
70
60
50
30
10
100
LOAD CURRENT (mA)
1.2
0.8
0.6
Figure 6. Output Efficiency vs. Load Current
RT = GND
0.4
0
–40
1000
RT = VIN
1.0
0.2
04472-007
40
1
–15
Figure 8. Current Limit vs. Ambient Temperature, VOUT = 10 V
100
90
04472-009
04472-006
1.4
10
EFFICIENCY (%)
2.2
VOUT = 10V
VIN = 3.3V
–15
04472-010
EFFICIENCY (%)
2.4
VIN = 3.3V
80
1000
Figure 7. Output Efficiency vs. Load Current
VIN = 5.5V
VOUT = 10V
F = 1.2MHz
L = 4.7µH
10
100
LOAD CURRENT (mA)
10
35
AMBIENT TEMPERATURE (°C)
60
Figure 9. Oscillatory Frequency vs. Ambient Temperature
Rev. 0 | Page 6 of 16
85
ADP1610
0.50
FSW = 700kHz
VFB = 1.3V
1.2
RT = VIN
QUIESCENT CURRENT (mA)
0.45
1.0
0.8
0.6
RT = GND
0.4
3.0
3.5
4.0
4.5
SUPPLY VOLTAGE (V)
5.0
0.35
VIN = 3.3V
0.30
VIN = 2.5V
0.20
–40
5.5
60
85
FSW = 1.23kHz
VFB = 1.3V
VIN = 2.5V
0.55
QUIESCENT CURRENT (mA)
300
250
VIN = 3.3V
200
VIN = 5.5V
150
100
0.50
VIN = 5.5V
0.45
VIN = 3.3V
0.40
VIN = 2.5V
04472-012
10
35
AMBIENT TEMPERATURE (°C)
60
0.30
–40
85
2.0
1.24
1.8
SUPPLY CURRENT (mA)
1.245
1.235
1.23
1.225
1.22
5
20
35
50
65
80
AMBIENT TEMPERATURE (°C)
95
110
60
85
FSW = 1.23kHz
VFB = 1V
1.6
VIN = 5.5V
1.4
1.2
VIN = 3.3V
1.0
VIN = 2.5V
0.8
04472-013
–25 –10
10
35
AMBIENT TEMPERATURE (°C)
Figure 14. Quiescent Current vs. Ambient Temperature
Figure 11. Switch Resistance vs. Ambient Temperature
1.215
–15
0.6
–40
125
–15
10
35
AMBIENT TEMPERATURE (°C)
60
Figure 15. Supply Current vs. Ambient Temperature
Figure 12. FB Regulation Voltage vs. Ambient Temperature
Rev. 0 | Page 7 of 16
04472-016
–15
04472-015
0.35
50
1.21
–40
10
35
AMBIENT TEMPERATURE (°C)
0.60
350
0
–40
–15
Figure 13. Quiescent Current vs. Ambient Temperature
Figure 10. Oscillatory Frequency vs. Supply Voltage
SWITCH RESISTANCE (mΩ)
VIN = 5.5V
04472-014
VOUT = 10V
0
2.5
FB REGULATION VOLTAGE (V)
0.40
0.25
0.2
04472-011
OSCILLATORY FREQUENCY (MHz)
4.4
85
ADP1610
1.4
CH1 = IL 200mA/DIV
CH2 = VSW 5V/DIV
FSW = 700kHz
VFB = 1V
1.3
VIN = 3.3V
VOUT = 10V
ILOAD = 20mA
FSW = 700kHz
L = 10µH
SUPPLY CURRENT (mA)
1.2
VIN = 5.5V
1.1
1.0
2
0.9
0.8
VIN = 3.3V
0.7
VIN = 2.5V
–15
10
35
AMBIENT TEMPERATURE (°C)
60
CH1 10.0mVΩ CH2 5.00V
85
Figure 16. Supply Current vs. Ambient Temperature
M400ns
A CH2
T
136.000ns
10.0V
Figure 19. Switching Waveform in Discontinuous Conduction
3.5
VIN = 3.3V, VOUT = 10V
COUT = 10µF, L = 10µH, RC = 130Ω
CC = 270pF, FSW = 700kHz
CH1 = VOUT, 200mV/DIV
CH2 = IOUT, 200mA/DIV
VIN = 3.3V
SD = 0.4V
3.0
SUPPLY CURRENT (µA)
04472-020
0.5
0.4
–40
1
04472-017
0.6
2.5
1
2.0
1.5
1.0
15
70
TEMPERATURE (°C)
CH1 200mV
125
Figure 17. Supply Current in Shutdown vs. Ambient Temperature
CH1 = IL 500mA/DIV
CH2 = VSW 5V/DIV
04472-021
0
–40
2
04472-018
0.5
CH2 10.0mVΩ M200µs
A CH2
7.60mV
Figure 20. Load Transient Response, 700 kHz , VOUT = 10 V
VIN = 3.3V, VOUT = 10V
COUT = 10µF, L = 4.7µH, RC = 220kΩ
CC = 150pF, FSW = 1.2MHz
CH1 = VOUT, 200mV/DIV
CH2 = IOUT, 200mA/DIV
VIN = 3.3V
VOUT = 10V
ILOAD = 200mA
FSW = 700kHz
L = 10µH
1
2
CH1 10.0mVΩ CH2 5.00V
M400ns
A CH2
T
136.000ns
10.0V
04472-022
2
04472-019
1
CH1 200mV
Figure 18. Switching Waveform in Continuous Conduction
CH2 10.0mVΩ M200µs
A CH2
7.60mV
Figure 21. Load Transient Response, 1.2 MHz, VOUT = 10 V
Rev. 0 | Page 8 of 16
ADP1610
2
2
4
4
VIN = 3.3V
VOUT = 10V
IOUT = 0.2A
CSS = 0nF
CH1 10.0mVΩ CH2 2.00V
CH3 10.0V
CH4 10.00V
M100µs
T
A CH2
680mV
CH1 10.0mVΩ CH2 2.00V
CH3 10.0V
CH4 10.00V
414.800µs
2
4
4
VIN = 3.3V
VOUT = 10V
IOUT = 0.2A
CSS = 10nF
3
CH1 10.0mVΩ CH2 2.00V
CH3 10.0V
CH4 10.00V
M100µs
T
A CH2
A CH2
1.72V
405.600µs
CH1 = IL 1A/DIV
CH2 = SD
CH3 = VOUT
CH4 = SW, FSW = 700kHz
IOUT = 0.2A
VIN = 3.3V
VOUT = 10V
CSS = 10nF
1
04472-024
1
VIN = 3.3V
VOUT = 10V
IOUT = 0.2A
CSS = 0nF
Figure 24. Start-Up Response from Shutdown, SS = 0 nF
2
CH1 = IL 1A/DIV
CH2 = VIN
CH3 = VOUT
CH4 = SW, FSW = 700kHz
M100µs
T
Figure 22. Start-Up Response from VIN, SS = 0 nF
3
CH1 = IL 1A/DIV
CH2 = VIN
CH3 = VOUT
CH4 = SW, FSW = 700kHz
1
04472-023
1
3
04472-025
CH1 = IL 1A/DIV
CH2 = VIN
CH3 = VOUT
CH4 = SW, FSW = 700kHz
04472-026
3
CH1 10.0mVΩ CH2 2.00V
CH3 10.0V
CH4 10.00V
680mV
M100µs
T
414.800µs
A CH2
1.72V
405.600µs
Figure 25. Start-Up Response from Shutdown, SS = 10 nF
Figure 23. Start-Up Response from VIN, SS = 10 nF
Rev. 0 | Page 9 of 16
ADP1610
THEORY OF OPERATION
The ADP1610 current-mode step-up switching converter
converts a 2.5 V to 5.5 V input voltage up to an output voltage as
high as 12 V. The 1.2 A internal switch allows a high output
current, and the high 1.2 MHz switching frequency allows tiny
external components. The switch current is monitored on a
pulse-by-pulse basis to limit it to 2 A.
CURRENT-MODE PWM OPERATION
The ADP1610 uses current-mode architecture to regulate the
output voltage. The output voltage is monitored at FB through a
resistive voltage divider. The voltage at FB is compared to the
internal 1.23 V reference by the internal transconductance error
amplifier to create an error current at COMP. A series resistorcapacitor at COMP converts the error current to a voltage. The
switch current is internally measured and added to the stabilizing ramp, and the resulting sum is compared to the error
voltage at COMP to control the PWM modulator. This currentmode regulation system allows fast transient response, while
maintaining a stable output voltage. By selecting the proper
resistor-capacitor network from COMP to GND, the regulator
response is optimized for a wide range of input voltages, output
voltages, and load conditions.
FREQUENCY SELECTION
The ADP1610’s frequency is user-selectable to operate at either
700 kHz to optimize the regulator for high efficiency or to
1.2 MHz for small external components. Connect RT to IN for
1.2 MHz operation, or connect RT to GND for 700 kHz
operation. To achieve the maximum duty cycle, which might be
required for converting a low input voltage to a high output
voltage, use the lower 700 kHz switching frequency.
SOFT START
To prevent input inrush current at startup, connect a capacitor
from SS to GND to set the soft start period. When the ADP1610
is in shutdown (SD is at GND) or the input voltage is below the
2.4 V undervoltage lockout voltage, SS is internally shorted to
GND to discharge the soft start capacitor. Once the ADP1610 is
turned on, SS sources 3 µA to the soft start capacitor at startup.
As the soft start capacitor charges, it limits the voltage at COMP.
Because of the current-mode regulator, the voltage at COMP is
proportional to the switch peak current, and, therefore, the
input current. By slowly charging the soft start capacitor, the
input current ramps slowly to prevent it from overshooting
excessively at startup.
ON/OFF CONTROL
The SD input turns the ADP1610 regulator on or off. Drive SD
low to turn off the regulator and reduce the input current to
10 nA. Drive SD high to turn on the regulator.
When the dc-dc step-up switching converter is turned off, there
is a dc path from the input to the output through the inductor
and output rectifier. This causes the output voltage to remain
slightly below the input voltage by the forward voltage of the
rectifier, preventing the output voltage from dropping to zero
when the regulator is shut down. Figure 28 shows the application circuit to disconnect the output voltage from the input
voltage at shutdown.
SETTING THE OUTPUT VOLTAGE
The ADP1610 features an adjustable output voltage range of VIN
to 12 V. The output voltage is set by the resistive voltage divider
(R1 and R2 in Figure 2) from the output voltage (VOUT) to the
1.230 V feedback input at FB. Use the following formula to
determine the output voltage:
VOUT = 1.23 × (1 + R1/R2)
(1)
Use an R2 resistance of 10 kΩ or less to prevent output voltage
errors due to the 10 nA FB input bias current. Choose R1 based
on the following formula:
⎛ VOUT − 1.23 ⎞
R1 = R2 × ⎜⎜
⎟⎟
1.23
⎝
⎠
(2)
INDUCTOR SELECTION
The inductor is an essential part of the step-up switching
converter. It stores energy during the on-time, and transfers that
energy to the output through the output rectifier during the offtime. Use inductance in the range of 1 µH to 22 µH. In general,
lower inductance values have higher saturation current and
lower series resistance for a given physical size. However, lower
inductance results in higher peak current that can lead to
reduced efficiency and greater input and/or output ripple and
noise. Peak-to-peak inductor ripple current at close to 30% of
the maximum dc input current typically yields an optimal
compromise.
For determining the inductor ripple current, the input (VIN) and
output (VOUT) voltages determine the switch duty cycle (D) by
the following equation:
D=
Rev. 0 | Page 10 of 16
VOUT − V IN
VOUT
(3)
ADP1610
Table 4. Inductor Manufacturers
Vendor
Sumida
847-956-0666
www.sumida.com
Coilcraft 847-639-6400
www.coilcraft.com
Toko 847-297-0070
www.tokoam.com
Part
CMD4D11-2R2MC
CMD4D11-4R7MC
CDRH4D28-100
CDRH5D18-220
CR43-4R7
CR43-100
DS1608-472
DS1608-103
D52LC-4R7M
D52LC-100M
L (µH)
2.2
4.7
10
22
4.7
10
4.7
10
4.7
10
Using the duty cycle and switching frequency, fSW, determine the
on-time by the following equation:
tON =
D
f SW
(4)
The inductor ripple current (∆IL) in steady state is
∆IL =
V IN × t ON
L
(5)
Solving for the inductance value, L,
L=
V IN × t ON
∆I L
(6)
Make sure that the peak inductor current (the maximum input
current plus half the inductor ripple current) is below the rated
saturation current of the inductor. Likewise, make sure that the
maximum rated rms current of the inductor is greater than the
maximum dc input current to the regulator.
For duty cycles greater than 50%, which occur with input
voltages greater than one-half the output voltage, slope
compensation is required to maintain stability of the currentmode regulator. For stable current-mode operation, ensure that
the selected inductance is equal to or greater than LMIN:
L > L MIN =
VOUT − V IN
1.8 A × f SW
Max DC Current
0.95
0.75
1.00
0.80
1.15
1.04
1.40
1.00
1.14
0.76
The ADP1610 requires input and output bypass capacitors to
supply transient currents while maintaining constant input and
output voltage. Use a low ESR (equivalent series resistance),
10 µF or greater input capacitor to prevent noise at the
ADP1610 input. Place the capacitor between IN and GND as
close to the ADP1610 as possible. Ceramic capacitors are
preferred because of their low ESR characteristics. Alternatively,
use a high value, medium ESR capacitor in parallel with a 0.1 µF
low ESR capacitor as close to the ADP1610 as possible.
Height (mm)
1.2
1.2
3.0
2.0
3.5
3.5
2.9
2.9
2.0
2.0
The output capacitor maintains the output voltage and supplies
current to the load while the ADP1610 switch is on. The value
and characteristics of the output capacitor greatly affect the
output voltage ripple and stability of the regulator. Use a low
ESR output capacitor; ceramic dielectric capacitors are
preferred.
For very low ESR capacitors such as ceramic capacitors, the
ripple current due to the capacitance is calculated as follows.
Because the capacitor discharges during the on-time, tON, the
charge removed from the capacitor, QC, is the load current
multiplied by the on-time. Therefore, the output voltage ripple
(∆VOUT) is
∆VOUT =
QC
I ×t
= L ON
C OUT
C OUT
(8)
where:
COUT is the output capacitance,
IL is the average inductor current,
t ON =
D
f SW
(9)
and
D=
(7)
CHOOSING THE INPUT AND OUTPUT CAPACITORS
Max DCR (mΩ)
116
216
128
290
109
182
60
75
87
150
VOUT − V IN
(10)
VOUT
Choose the output capacitor based on the following equation:
C OUT ≥
I L × (VOUT − V IN )
(11)
f SW × VOUT × ∆VOUT
Table 5. Capacitor Manufacturers
Vendor
AVX
Murata
Sanyo
Taiyo–Yuden
Rev. 0 | Page 11 of 16
Phone No.
408-573-4150
714-852-2001
408-749-9714
408-573-4150
Web Address
www.avxcorp.com
www.murata.com
www.sanyovideo.com
www.t-yuden.com
ADP1610
The regulator loop gain is
DIODE SELECTION
The output rectifier conducts the inductor current to the output
capacitor and load while the switch is off. For high efficiency,
minimize the forward voltage drop of the diode. For this reason,
Schottky rectifiers are recommended. However, for high voltage,
high temperature applications, where the Schottky rectifier
reverse leakage current becomes significant and can degrade
efficiency, use an ultrafast junction diode.
Make sure that the diode is rated to handle the average output
load current. Many diode manufacturers derate the current
capability of the diode as a function of the duty cycle. Verify
that the output diode is rated to handle the average output load
current with the minimum duty cycle. The minimum duty cycle
of the ADP1610 is
D MIN =
VOUT − V IN − MAX
(12)
VOUT
Table 6. Schottky Diode Manufacturers
Phone No.
602-244-3576
805-446-4800
310-322-3331
where:
AVL is the loop gain.
VFB is the feedback regulation voltage, 1.230 V.
VOUT is the regulated output voltage.
VIN is the input voltage.
GMEA is the error amplifier transconductance gain.
ZCOMP is the impedance of the series RC network from COMP to
GND.
GCS is the current sense transconductance gain (the inductor
current divided by the voltage at COMP), which is internally set
by the ADP1610.
Web Address
www.mot.com
www.diodes.com
www.irf.com
To determine the crossover frequency, it is important to note
that, at that frequency, the compensation impedance (ZCOMP) is
dominated by the resistor, and the output impedance (ZOUT) is
dominated by the impedance of the output capacitor. So, when
solving for the crossover frequency, the equation (by definition
of the crossover frequency) is simplified to
| AVL | =
LOOP COMPENSATION
The ADP1610 uses external components to compensate the
regulator loop, allowing optimization of the loop dynamics for a
given application.
The step-up converter produces an undesirable right-half plane
zero in the regulation feedback loop. This requires compensating the regulator such that the crossover frequency occurs well
below the frequency of the right-half plane zero. The right-half
plane zero is determined by the following equation:
⎛ V
FZ (RHP) = ⎜⎜ IN
⎝ VOUT
V
V FB
× IN × G MEA × Z COMP × G CS × Z OUT (14)
VOUT VOUT
ZOUT is the impedance of the load and output capacitor.
where VIN-MAX is the maximum input voltage.
Vendor
Motorola
Diodes, Inc.
Sanyo
AVL =
VFB VIN
1
×
× GMEA× RCOMP× GCS ×
=1
VOUT VOUT
2π × fC × COUT
where:
fC is the crossover frequency.
RCOMP is the compensation resistor.
Solving for RCOMP,
R COMP =
2
⎞ R LOAD
⎟ ×
⎟
2π × L
⎠
(13)
(15)
2π × f C ×C OUT × VOUT × VOUT
V FB × V IN × G MEA × G CS
(16)
For VFB = 1.23, GMEA = 100 µS, and GCS = 2 S,
RCOMP =
where:
2.55 × 10 4 × f C × COUT × VOUT × VOUT
VIN
(17)
FZ(RHP) is the right-half plane zero.
Once the compensation resistor is known, set the zero formed
by the compensation capacitor and resistor to one-fourth of the
crossover frequency, or
RLOAD is the equivalent load resistance or the output voltage
divided by the load current.
To stabilize the regulator, make sure that the regulator crossover
frequency is less than or equal to one-fifth of the right-half
plane zero and less than or equal to one-fifteenth of the
switching frequency.
C COMP =
2
π × f C × RCOMP
where CCOMP is the compensation capacitor.
Rev. 0 | Page 12 of 16
(18)
ADP1610
Table 7. Recommended External Components for Popular Input/Output Voltage Conditions
VIN (V)
3.3
5
VOUT (V)
5
5
9
9
12
12
9
9
12
12
fSW (MHz)
0.700
1.23
0.700
1.23
0.700
1.23
0.700
1.23
0.700
1.23
L (µH)
4.7
2.7
10
4.7
10
4.7
10
4.7
10
4.7
COUT (µF)
10
10
10
10
10
10
10
10
10
10
CIN (µF)
10
10
10
10
10
10
10
10
10
10
R1 (kΩ)
30.9
30.9
63.4
63.4
88.7
88.7
63.4
63.4
88.7
88.7
R2 (kΩ)
10
10
10
10
10
10
10
10
10
10
RComp (kΩ)
50
90.9
71.5
150
130
280
84.5
178
140
300
Ccomp (pF)
520
150
820
180
420
100
390
100
220
100
IOUT_MAX (mA)
600
600
350
350
250
250
450
450
350
350
ERROR AMP
REF
gm
COMP
1
Table 8. Typical Soft Start Period
FB 2
RC
VIN (V)
3.3
C2
04472-004
CC
Figure 26. Compensation Components
The capacitor, C2, is chosen to cancel the zero introduced by
output capacitance ESR.
5
Solving for C2,
C2 =
ESR × C OUT
RCOMP
(19)
For low ESR output capacitance such as with a ceramic capacitor, C2 is optional. For optimal transient performance, the RCOMP
and CCOMP might need to be adjusted by observing the load
transient response of the ADP1610. For most applications, the
compensation resistor should be in the range of 30 kΩ to
400 kΩ, and the compensation capacitor should be in the range
of 100 pF to 1.2 nF. Table 7 shows external component values
for several applications.
VOUT (V)
5
5
9
9
12
12
9
9
12
12
COUT (µF)
10
10
10
10
10
10
10
10
10
10
CSS (nF)
20
100
20
100
20
100
20
100
20
100
tSS (ms)
0.3
2
2.5
8.2
3.5
15
0.4
1.5
0.62
2
Conversely, if fast startup is a requirement, the soft start
capacitor can be reduced or even removed, allowing the
ADP1610 to start quickly, but allowing greater peak switch
current (see Figure 22 to Figure 25).
APPLICATION CIRCUITS
The circuit in Figure 27 shows the ADP1610 in a step-up
configuration. The ADP1610 is used here to generate a 10 V
regulator with the following specifications: VIN = 2.5 V to 5.5 V,
VOUT = 10 V, and IOUT ≤ 400 mA.
4.7µH
SOFT START CAPACITOR
The voltage at SS ramps up slowly by charging the soft start
capacitor (CSS) with an internal 3 µA current source. Table 8
listed the values for the soft start period, based on maximum
output current and maximum switching frequency.
L
D1
ADP1610
3.3V
6
IN
3
SD
7
RT
8
SS
10V
SW 5
R1
71.3kΩ
ON
FB 2
CIN
10µF
CSS
22nF
A 20 nF soft start capacitor results in negligible input current
overshoot at startup, and so is suitable for most applications.
However, if an unusually large output capacitor is used, a longer
soft start period is required to prevent input inrush current.
R2
10kΩ
COMP 1
GND
4
RCOMP
220kΩ
CCOMP
150pF
COUT
10µF
04472-030
The soft start capacitor limits the rate of voltage rise on the
COMP pin, which in turn limits the peak switch current at
startup. Table 8 shows a typical soft start period, tSS, at
maximum output current, IOUT_MAX, for several conditions.
Figure 27. 3.3 V to 10 V Step-Up Regulator
The output can be set to the desired voltage using Equation 2.
Use Equation 16 and 17 to change the compensation network.
Rev. 0 | Page 13 of 16
ADP1610
DC-DC STEP-UP SWITCHING CONVERTER WITH
TRUE SHUTDOWN
Some battery-powered applications require very low standby
current. The ADP1610 typically consumes 10 nA from the
input, which makes it suitable for these applications. However,
the output is connected to the input through the inductor and
the rectifying diode, allowing load current draw from the input
while shut down. The circuit in Figure 28 enables the ADP1610
to achieve output load disconnect at shutdown. To shut down
the ADP1610 and disconnect the output from the input, drive
the SD pin below 0.4 V.
R4
200Ω
BAV99 C5
10nF
C6
D8
10µF
VGL
–5V
D9
BZT52C5VIS
L
SD
D1
ADP1610
3.3V
10V
SW 5
6
IN
3
SD
7
RT
8
SS
R1
71.3kΩ
ON
D1
10V
R2
10kΩ
COMP 1
SW 5
CSS
22nF
R1
71.3kΩ
GND
4
RCOMP
220kΩ
CCOMP
150pF
COUT
10µF
04472-033
3
C2
1µF
FB 2
CIN
10µF
7
RT
8
SS
R2
10kΩ
COMP 1
OFF
CSS
22nF
GND
4
RCOMP
220kΩ
CCOMP
150pF
Figure 29. TFT LCD Bias Supply
COUT
10µF
SEPIC POWER SUPPLY
04472-031
Q1 B
Figure 28. Step-Up Regulator with True Shutdown
TFT LCD BIAS SUPPLY
Figure 29 shows a power supply circuit for TFT LCD module
applications. This circuit has +10 V, −5 V, and +22 V outputs.
The +10 V is generated in the step-up configuration. The −5 V
and +22 V are generated by the charge-pump circuit. During the
step-up operation, the SW node switches between 10 V and
ground (neglecting forward drop of the diode and on resistance
of the switch). When the SW node is high, C5 charges up to
10 V. C5 holds its charge and forward-biases D8 to charge C6
to −10 V. The Zener diode, D9, clamps and regulates the output
to −5 V.
The circuit in Figure 30 shows the ADP1610 in a single-ended
primary inductance converter (SEPIC) topology. This topology
is useful for an unregulated input voltage, such as a batterypowered application in which the input voltage can vary
between 2.7 V to 5 V, and the regulated output voltage falls
within the input voltage range.
The input and the output are dc-isolated by a coupling capacitor, C1. In steady state, the average voltage of C1 is the input
voltage. When the ADP1610 switch turns on and the diode
turns off, the input voltage provides energy to L1, and C1
provides energy to L2. When the ADP1610 switch turns off and
the diode turns on, the energy in L1 and L2 is released to charge
the output capacitor, COUT, and the coupling capacitor, C1, and
to supply current to the load.
4.7µH
The VGH output is generated in a similar manner by the
charge-pump capacitors, C1, C2, and C4. The output
voltage is tripled and regulated down to 22 V by the
Zener diode, D5.
L1
C1
10µF
ADP1610
2.5V–5.5V
6
IN
3
SD
ON
CIN
10µF
CSS
22nF
7
RT
8
SS
4.7µH
L2
R1
16.8kΩ
FB 2
COMP 1
GND
4
RCOMP
60kΩ
CCOMP
1nF
Figure 30. 3.3 V DC-DC Converter
Rev. 0 | Page 14 of 16
3.3V
SW 5
COUT
10µF
R2
10kΩ
04472-032
R3
10kΩ
VGH
22V
D5
BZT52C22
FB 2
ADP1610
IN
D4
C3
10µF
C1
10nF D2
BAV99
4.7µH
CIN
10µF
L
6
D5
BAV99
D3
D7
4.7µH
3.3V Q1 A FDC6331
R3
200Ω
C4
10nF
ADP1610
LAYOUT PROCEDURE
To get high efficiency, good regulation, and stability, a welldesigned printed circuit board layout is required. Where
possible, use the sample application board layout as a model.
•
Keep the low ESR input capacitor, CIN, close to IN and
GND.
•
Keep the high current path from CIN through the inductor
L1 to SW and PGND as short as possible.
•
Keep the high current path from CIN through L1, the
rectifier D1, and the output capacitor COUT as short as
possible.
•
Keep high current traces as short and as wide as possible.
•
Place the feedback resistors as close to the FB pin as
possible to prevent noise pickup.
•
Place the compensation components as close as possible to
COMP.
•
Avoid routing high impedance traces near any node
connected to SW or near the inductor to prevent radiated
noise injection.
04472-028
Follow these guidelines when designing printed circuit boards
(see Figure 1):
04472-029
Figure 32. Sample Application Board (Top Layer)
04472-027
Figure 33. Sample Application Board (Silkscreen Layer)
Figure 31. Sample Application Board (Bottom Layer)
Rev. 0 | Page 15 of 16
ADP1610
Preliminary Technical Data
OUTLINE DIMENSIONS
3.00
BSC
8
3.00
BSC
1
5
4.90
BSC
4
PIN 1
0.65 BSC
1.10 MAX
0.15
0.00
0.38
0.22
COPLANARITY
0.10
0.23
0.08
8°
0°
0.80
0.60
0.40
SEATING
PLANE
COMPLIANT TO JEDEC STANDARDS MO-187AA
Figure 34. 8-Lead Mini Small Outline Package [MSOP]
(RM-8)
Dimensions shown in millimeters
ORDERING GUIDE
Model
ADP1610ARMZ-R71
1
Temperature Range
−40°C to +85°C
Package Description
8-Lead Mini Small Outline Package [MSOP]
Z = Pb-free part.
© 2004 Analog Devices, Inc. All rights reserved. Trademarks and
registered trademarks are the property of their respective owners.
D04472–0–10/04(0)
Rev. 0 | Page 16 of 16
Package Option
RM-8
Branding
P03