a 16-Bit High Speed Current-Output DAC DAC16 FEATURES ⴞ1 LSB Differential Linearity (max) Guaranteed Monotonic Over Temperature Range ⴞ2 LSB Integral Linearity (max) 500 ns Settling Time 5 mA Full-Scale Output TTL/CMOS Compatible Low Power: 190 mW (typ) Available in Die Form FUNCTIONAL BLOCK DIAGRAM IREF DAC16 BUFFER CCOMP REF GND VCC AGND VEE DAC IOUT DGND DB0 (LSB) DB15 (MSB) APPLICATIONS Communications ATE Data Acquisition Systems High Resolution Displays 0.1 The DAC16 is a 16-bit high speed current-output digital-toanalog converter with a settling time of 500 ns. A unique combination of low distortion, high signal-to-noise ratio, and high speed make the DAC16 ideally suited to performing waveform synthesis and modulation in communications, instrumentation, and ATE systems. Input reference current is buffered, with fullscale output current of 5 mA. The 16-bit parallel digital input bus is TTL/CMOS compatible. Operating from +5 V and –15 V supplies, the DAC16 consumes 190 mW (typ) and is available in a 24-lead epoxy DIP, epoxy surface-mount small outline (SOL), and in die form. PERCENT OF FULL-SCALE – % GENERAL DESCRIPTION VLOGIC = +5V TURNING OFF VLOGIC = 0V TURNING ON 0.01 IFS = 4mA TA = 258C 0.001 0 100 200 300 400 500 SETTLING TIME – ns 600 700 800 Figure 1. DAC16 Settling Time Accuracy vs. Percent of Full Scale REV. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781/329-4700 World Wide Web Site: http://www.analog.com Fax: 781/326-8703 © Analog Devices, Inc., 1999 DAC16–SPECIFICATIONS(@ V CC = +5.0 V, VEE = –15.0 V, IREF = 0.5 mA, CCOMP = 47 F, TA = Full Operating Tem- ELECTRICAL CHARACTERISTICS perature Range unless otherwise noted. See Note 1 for supply variations.) Parameter Conditions Min Typ Max Units –2 –4 –1 –1 –4 –6 –1 –1.5 ± 1.2 ± 1.6 ± 0.5 ± 0.7 ± 1.4 ±2 ± 0.5 ± 0.6 +2 +4 +1 +1.5 +4 +6 +1.5 +2 1 LSB LSB LSB LSB LSB LSB LSB LSB LSB ppm/°C % FS ppm/°C Integral Linearity “G” Integral Linearity “G” Differential Linearity “G” Differential Linearity “G” Integral Linearity “F” Integral Linearity “F” Differential Linearity “F” Differential Linearity “F” Zero Scale Error Zero Scale Drift Gain Error Gain Drift INL INL DNL DNL INL INL DNL DNL ZSE TCZSE GE TCGE TA = +25°C REFERENCE2 Reference Input Current IREF Note 2 350 OUTPUT CHARACTERISTICS Output Current Output Capacitance Settling Time IOUT COUT tS Note 2 2.8 LOGIC CHARACTERISTICS Logic Input High Voltage Logic Input Low Voltage Logic Input Current Logic Input Current Logic Input Current Input Capacitance VINH VINL IINH IINH IINL CIN TA = +25°C TA = +25°C VIN = 5.0 V, DB0–DB10 VIN = 5.0 V, DB11–DB15 VIN = 0 V, DB0–DB15 SUPPLY CHARACTERISTICS Power Supply Sensitivity Positive Supply Current Positive Supply Current Negative Supply Current Power Dissipation PSS ICC ICC IEE PDISS VCC = 4.5 V to 5.5 V, VEE = –13 V to –17 V All Bits HIGH All Bits LOW TA = +25°C TA = +25°C TA = +25°C 0.025 ± 0.225 5 625 µA 5.0 mA pF ns 10 500 0.003% of Full Scale 2.4 0.8 7.5 100 1 V V µA µA µA pF 20 22 7.5 10 260 ppm/V mA mA mA mW 8 15 6 7.5 188 NOTES 1 All supplies can be varied ± 5% and operation is guaranteed. Device is tested with nominal supplies. 2 Operation is guaranteed over this reference range, but linearity is neither tested not guaranteed (see Figures 7 and 8). Specifications subject to change without notice. WAFER TEST LIMITS (@ V CC = +5.0 V, VEE = –15.0 V, IREF = 0.5 mA, CCOMP = 47 F, TA = +25ⴗC unless otherwise noted.) Parameter Symbol Integral Nonlinearity Differential Nonlinearity Zero Scale Error Gain Error Logic Input High Voltage Logic Input Low Voltage Logic Input Current Positive Supply Current Negative Supply Current Power Dissipation INL DNL ZSE GE VINH VINL IIN ICC IEE PDISS Conditions DAC16G Limit Units ±3 ±1 ±1 ± 12 2.4 0.8 75 20 10 250 LSB max LSB max LSB max % FS max V min V max µA max mA max mA max mW max NOTE Electrical tests are performed at wafer probe to the limits shown. Due to variations in assembly methods and normal yield loss, yield after packaging is not guaranteed for standard product dice. Consult factory to negotiate specifications based on dice lot qualification through sample lot assembly and testing. –2– REV. B DAC16 ABSOLUTE MAXIMUM RATINGS CAUTION (TA = +25°C unless otherwise noted) 1. Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress rating only and functional operation at or above this specification is not implied. Exposure to the above maximum rating conditions for extended periods may affect device reliability. 2. Digital inputs and outputs are protected; however, permanent damage may occur on unprotected units from high energy electrostatic fields. Keep units in conductive foam or packaging at all times until ready to use. Use proper antistatic handling procedures. 3. Remove power before inserting or removing units from their sockets. VCC to VEE . . . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V, +25.0 V VCC to DGND . . . . . . . . . . . . . . . . . . . . . . . . . –0.3 V, +7.0 V VEE to AGND . . . . . . . . . . . . . . . . . . . . . . . . +0.3 V, –18.0 V DGND to AGND . . . . . . . . . . . . . . . . . . . . . . –0.3 V, +0.3 V REF GND to AGND . . . . . . . . . . . . . . . . . . . –0.3 V, +1.0 V IREF . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1 mA Analog Output Current . . . . . . . . . . . . . . . . . . . . . . . . . . 8 mA Digital Input Voltage to DGND . . . . . . . . . . . . . . . . . . . ≤VCC Operating Temperature Range FP, FS . . . . . . . . . . . . . . . . . . . . . . . . . . . . –40°C to +85°C GS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 0°C to +70°C Dice Junction Temperature . . . . . . . . . . . . . . . . . . . . . +150°C Storage Temperature . . . . . . . . . . . . . . . . . . –65°C to +150°C Power Dissipation . . . . . . . . . . . . . . . . . . . . . . . . . . 1000 mW Lead Temperature (Soldering, 60 sec) . . . . . . . . . . . . +300°C Package Type θJA1 θJC Units 24-Lead Plastic DIP (P) 24-Lead Plastic SOL (S) 62 70 32 22 °C/W °C/W PIN CONFIGURATION 24-Lead DIP (P, S) NOTE 1 θJA is specified for worst case mounting conditions, i.e., θJA is specified for device in socket. IREF 1 24 DGND 2 23 IOUT VCC 3 22 AGND DB15 (MSB) 4 21 REF GND DB14 5 20 VEE DB13 6 DB12 7 DICE CHARACTERISTICS VCC DGND IREF CCOMP IOUT AGND REF GND DB15 (MSB) DAC16 CCOMP TOP VIEW 19 DB0 (LSB) (Not to Scale) 18 DB1 DB11 8 17 DB2 DB10 9 16 DB3 DB9 10 15 DB4 DB8 11 14 DB5 DB7 12 13 DB6 VEE DB14 DB0 (LSB) PIN DESCRIPTION DB13 DB12 DB1 DB11 DB2 DB10 DB3 DB9 DB8 DB7 DB6 DB5 DB4 Die Size 0.129 x 0.153 inch, 19,737 sq. mils (3.277 x 3.886 mm, 12.73 sq. mm) The DAC16 Contains 330 Transistors. Substrate is VEE Polarity. Pin (P, S) Name Description 1 2 3 4–19 20 21 22 23 24 Reference Current Input Digital Ground +5 V Digital Supply 16-Bit Digital Input Bus. DB15 is the MSB. –15 V Analog Supply Reference Current Return Analog Ground/Output Reference Current Output Current Ladder Compensation IREF DGND VCC DB15–DB0 VEE REF GND AGND IOUT CCOMP ORDERING GUIDE Model Grade DNL (max) Temperature Ranges Package Descriptions Package Options DAC16GS DAC16FP DAC16FS DAC16GBC ±1 ±2 ±2 ±1 0°C to +70°C –40°C to +85°C –40°C to +85°C +25°C 24-Lead SOL 24-Lead PDIP 24-Lead SOL Die R-24 N-24 R-24 REV. B –3– DAC16 minimizing the deleterious effects of digital feedthrough while allowing the user to tailor the digital interface to the speed requirements and bus configuration of the application. +5V 10kV 1 24 2 23 3 22 4 21 5 20 6 19 7 18 8 17 9 16 10 15 11 14 12 13 NC Equivalent Circuit Analysis An equivalent circuit for static operation of the DAC16 is illustrated in Figure 4. IREF is the current applied to the DAC16 and is set externally to the device by VREF and RREF. The output capacitance of the DAC16 is approximately 10 pF and is code independent. Its output resistance RO is code dependent and is given by: –15V 1 = 1 + DB9 + DB10 + X RO 8 kΩ 288 kΩ 144 kΩ 72 kΩ where Figure 2. Burn-In Diagram DB9 = State of Data Bit 9 = 0 or 1; OPERATION Novel DAC Architecture DB10 = State of Data Bit 10 = 0 or 1; and The DAC16 was designed with a compound DAC architecture to achieve high accuracy, excellent linearity, and low transition errors. As shown in Figure 3, the DAC’s five most-significant bits utilize 31 identical segmented current sources to obtain optimal high speed settling at major code transitions. The lower nine bits utilize an inverted R-2R ladder network which is lasertrimmed to ensure excellent differential nonlinearity. The middle two bits (DB9 and DB10) arc binary-weighted and scaled from the MSB segments. Note that the flow of output current is into the DAC16—there is no signal inversion. As shown, the switches for each current source are essentially diodes. It is for this reason that the output voltage compliance of the DAC16 is limited to a few millivolts. The DAC16 was designed to operate with an operational amplifier configured as an I–V converter; therefore, the DAC16’s output must be connected to the sum node of an operational amplifier for proper operation. Exceeding the output voltage compliance of the DAC16 will introduce linearity errors. The reference current buffer assures full accuracy and fast settling by controlling the MSB reference node. The 16-bit parallel digital input is TTL/CMOS compatible and unbuffered, X = Decimal representation of the 5 MSBs (DB11–DB15) = 0 to 31. IOUT IDAC CO RO IOUT = 8 • IREF 65,535 Digital Code 65,536 RO = SEE TEXT CO = 10pF Figure 4. Equivalent Circuit for the DAC16 Table I provides the relationship between the input digital code and the output resistance of the DAC16. Table I. DAC16 Output Resistance vs. Digital Code Hex Digital Code Scale Output Resistance FFFF BFFF 7FFF 3FFF 0 Zero 1/4 1/2 3/4 Full – 1 LSB 8 kΩ 4.2 kΩ 2.9 kΩ 2.2 kΩ 1.8 kΩ IOUT DB0 – DB8 AGND DB11 – DB15 DB10 8kV DB9 4kV 18kV IREF SW SW SW SW SW10 SW9 SW8 8kV 4kV SW7 4kV SW6 4kV 4kV SW0 DB0 – DB15 SWITCH DETAIL +5V FROM SWITCH DECODER 31 CURRENT SOURCES 125mA EACH 62.5mA 31.25mA 9 CURRENT SOURCES 15.63mA EACH CCOMP Figure 3. DAC16 Architecture –4– REV. B Typical Performance Characteristics–DAC16 Digital Input Considerations The threshold of the DAC16’s digital input circuitry is set at 1.4 V, independent of supply voltage. Hence, the digital inputs can interface with any type of 5 V logic. Illustrated in Figure 5 is the equivalent circuit of the digital inputs. Note that the individual input capacitance is approximately 7 pF. +5V This input capacitance can be used in conjunction with an external R-C circuit for digital signal deskewing, if required. In applications where some of the DAC16’s digital inputs are not used, the recommended procedure to turn off one or more inputs is to connect each input line to +5 V as shown in Figure 6. +0.7V +5V DBX R2 75kV Q1 Q2 DAC16 DB0 Q3 R1 20kV R3 28kV –15V TO DAC SWITCH DB1 –0.7V Figure 6. Handling Unused DAC16 Digital Inputs Figure 5. Equivalent Circuit of a DAC16 Digital Input 2.0 3 2 +INL 0 –1 –INL –2 –3 –4 0.2 0.3 0.4 0.5 0.6 REFERENCE CURRENT – mA 0.7 Figure 7. Integral Nonlinearity vs. IREF 0.5 –0.5 –DNL 0 –5 –10 –15 –40 –20 0 20 40 60 TEMPERATURE – 8C Figure 10. Gain Error vs. Temperature REV. B 80 0.4 0.2 –1.5 –2.0 0.2 0.3 0.4 0.5 0.6 REFERENCE CURRENT – mA 0 0.7 Figure 8. Differential Nonlinearity vs. IREF 80 1.5 +INL 2 –INL 0 VCC = +5V VEE = –15V IREF = 0.5mA –2 –4 0 20 40 60 –40 –20 TEMPERATURE – 8C Figure 9. Zero Scale Output vs. Temperature 4 5 0.6 –1.0 INTEGRAL NONLINEARITY – LSB GAIN ERROR – LSB 10 +DNL 0 15 VCC = +5V VEE = –15V IREF = 0.5mA VCC = +5V VEE = –15V IREF = 0.5mA 0.8 1.0 –40 –20 0 20 40 60 TEMPERATURE – 8C 80 Figure 11. Integral Nonlinearity vs. Temperature –5– DIFFERENTIAL NONLINEARITY – LSB 1 1.0 VCC = +5V VEE = –15V TA = +258C 1.5 ZERO SCALE – LSB VCC = +5V VEE = –15V TA = +258C DIFFERENTIAL NONLINEARITY – LSB INTEGRAL NONLINEARITY – LSB 4 VCC = +5V VEE = –15V IREF = 0.5mA 1.0 0.5 +DNL 0 –0.5 –DNL –1.0 –1.5 –40 –20 0 20 40 60 TEMPERATURE – 8C 80 Figure 12. Differential Nonlinearity vs. Temperature DAC16–Typical Performance Characteristics 20 20 50 15 15 10 I CC – mA SUPPLY CURRENT – mA ICC, LOGIC BITS = HIGH IEE , LOGIC BITS = LOW LOGIC BIT CURRENT – mA VEE = –15V VCC = +5V TA = +258C 10 IEE , LOGIC BITS = HIGH 5 5 40 DB0 – DB4 30 20 10 ICC, LOGIC BITS = LOW 0 –40 –20 0 20 40 60 TEMPERATURE – 8C DB5 – DB15 0 80 Figure 13. Supply Current vs. Temperature 5 TYPICAL + DNL 0 –0.5 TYPICAL – DNL –1.0 WORST CASE – DNL 4 TYPICAL + INL 2 1 TYPICAL – INL 0 –1 WORST CASE – INL 200 400 600 800 1000 BURN-IN TIME – Hours 1200 Figure 16. Differential Nonlinearity vs. Time Accelerated by Burn-In WORST CASE + GAIN ERROR 110 0 TYPICAL GAIN ERROR –10 –2 0 200 400 600 800 1000 BURN-IN TIME – Hours 1200 Figure 17. Integral Nonlinearity vs Time Accelerated by Burn-In APPLICATIONS Power Supplies, Bypassing, and Grounding All precision converter products require careful application of good grounding practices to maintain full-rated performance. As is always the case with analog circuits operating in digital environments, digital noise is prevalent; therefore, special care must be taken to ensure that the DAC16’s inherent precision is maintained. This means that particularly good engineering judgment should be exercised when addressing the power supply, grounding, and bypassing issues using the DAC16. The DAC16 was designed to operate from +5 V and –15 V supplies. The +5 V supply primarily powers the digital portion of the DAC16 and can consume 20 mA, maximum. Although very little +5 V supply current is used by the reference amplifier, large amounts of digital noise present on the +5 V supply can introduce analog errors. It is, therefore, very important that the +5 V supply be well filtered and regulated. The –15 V supply provides most of the current for the reference amplifier and all of the current for the internal DAC. Although the maximum current in this supply is 10 mA, it must provide a low impedance path for the DAC switch currents. Therefore, it too must be well filtered and regulated. WORST CASE – GAIN ERROR –20 VCC = +5V, VEE = –15V TA = +258C, IREF = 0.5mA –3 –4 0 VCC = +5V, VEE = –15V TA = +258C, IREF = 0.5mA 120 3 80 130 5 GAIN ERROR – LSB 0.5 –40 –20 0 20 40 60 TEMPERATURE – 8C Figure 15. Digital Input Current vs. Temperature WORST CASE + INL INTEGRAL NONLINEARITY – LSB DIFFERENTIAL NONLINEARITY – LSB 1 2 3 4 LOGIC INPUT VOLTAGE – V ALL DATA BITS 0 6 VCC = +5V, VEE = –15V TA = +258C, IREF = 0.5mA WORST CASE + DNL –1.5 0 Figure 14. VCC Supply Current vs. Logic Input Voltage, All Data Bits 1.5 1.0 VCC = +5V VEE = –15V VIN = +5V –30 0 200 400 600 800 1000 BURN-IN TIME – Hours 1200 Figure 18. Gain Error vs. Time Accelerated by Burn-In The DAC16 includes two ground connections in order to minimize system accuracy degradation arising from grounding errors. The two ground pins are designated DGND (Pin 2) and AGND (Pin 22). The DGND pin is the return for the digital circuit sections of the DAC and serves as their input threshold reference point. Thus, DGND should be connected to the same ground as the circuitry that drives the digital inputs. Pin 22, AGND, serves as the reference point for the 9-bit lower-order DAC as well as the common for the reference amplifier, REFGND (Pin 21). This pin should also serve as the reference point for all analog circuitry associated with the DAC16. Therefore, to minimize any errors, it is recommended that AGND connection on the DAC16 be connected to a high quality analog ground. If the system contains any analog signal path carrying a significant amount of current, then that path should have its own return connection to Pin 22. It is often advisable to maintain separate analog and digital grounds throughout a complete system, tying them common to one place only. If the common tie point is remote and an accidental disconnection of that one common tie point were to occur due to card removal with power on, a large differential voltage between the two commons could develop. To protect devices that interface to both digital and analog parts of the –6– REV. B DAC16 system, such as the DAC16, it is recommended that common ground tie points be provided at each such device. If only one system ground can be connected directly to the DAC16, it is recommended that the analog common be used. If the system’s AGND has suitable low impedance, then the digital signal currents flowing in it should not seriously affect the ground noise. The amount of digital noise introduced by connecting the two grounds together at the device will not adversely affect system performance due to loss of digital noise immunity. +5V 10mF FB 0.1mF VCC DAC16 VEE AGND 0.1mF Generous bypassing of the DAC’s supplies goes a long way in reducing supply-line induced errors. Even with well-filtered, well-regulated supplies, local bypassing consisting of 10 µF tantalum electrolytic shunted by a 0.1 µF ceramic is recommended. The decoupling capacitors should be connected between the DAC’s supply pins (Pin 3 for +5 V, Pin 20 for –15 V) and the analog ground (Pin 22). Figure 19 shows how the DGND, AGND, and bypass connections should be made to the DAC16. +5V VCC 10mF 0.1mF AGND FB –15V Figure 20. Using a Ferrite Bead as a High Frequency Filter Reference Amplifier Considerations The reference input current buffer is a high performance amplifier optimized for high accuracy and linearity. The design of the reference amplifier ensures fast settling times by tightly controlling the node common to all the current sources internal to the DAC with an external compensation capacitor (CCOMP). Since the primary design goal of the DAC16 is to achieve 16-bit performance, proper operation of the reference amplifier requires a DB0 – DB15 DAC16 VEE DGND IOUT 0.1mF 47 µF tantalum electrolytic capacitor shunted by a 0.1 µF ceramic capacitor, as shown in Figure 21. Increasing the capacitance at this node above the recommended values does not further reduce the analog transition current noise spikes at the output of the reference amplifier. Reducing the value of compensation, however, is not recommended as DAC linearity will degrade as a result. In most systems, the VEE supply offers sufficiently low impedance to maintain a quiet return point for the reference amplifier. If this is not the case, the AGND point can also be used for the compensation capacitor return, as shown in Figure 21. 10mF TO OTHER TO ANALOG POWER CIRCUITS GROUND –15V Figure 19. Recommended Grounding and Bypassing Scheme for the DAC16 Using the Right Capacitors Probably the most important external components associated with high speed design are the capacitors used to bypass the power supplies and to provide compensation. Both selection and placement of these capacitors can be critical and, to a large extent, dependent upon the specifics of the system configuration. The dominant consideration in selection of bypass and compensation capacitors for the DAC16 is minimization of series resistance and inductance. Many capacitors begin to look inductive at 20 MHz and above—the very frequencies where rejection of interference is needed. Ceramic and film-type capacitors generally feature lower series inductance than tantalum or electrolytic types. A few general rules are of universal use when approaching the issue of compensation or bypassing. CCOMP DAC16 VEE 0.1mF IOUT AGND 47mF –15V Figure 21a. Recommended Compensation Scheme to VEE CCOMP 0.1mF Bypass capacitors should be installed on the printed circuit board with the shortest possible leads consistent with reliable construction. This helps to minimize series inductance in the leads. Chip capacitors are optimal in this respect. Where illustrated in the applications section, large tantalum electrolytic capacitors are shunted by low self-inductance ceramic capacitors. This technique reduces the self-resonance of the electrolytic while shifting the resonant frequency of the ceramics out-of-band. DAC16 IOUT AGND 47mF Figure 21b. Recommended Compensation Scheme to AGND In applications where 16-bit multiplying performance is required, the DAC16 might appear to be a viable solution. However, the compensation capacitor network would have to be removed in these applications. The DAC16’s reference amplifier was specifically designed for low frequency operation, with a compensation capacitor network. In fact, this network serves not only as a charge reservoir for the DAC’s internal current sources but also as a wideband noise filter for the Some series inductance between the DAC supply pins and the power supply plane often helps to filter out high frequency power supply noise. This inductance can be generated using a small ferrite bead as shown in Figure 20. REV. B 10mF –7– DAC16 reference amplifier. Completely removing the compensation network would introduce large linearity errors, reference amplifier instability, wideband reference amplifier noise, and poor settling time. Because the DAC exhibits an internal current scaling factor of eight times (8×), the reference amplifier requires only 500 µA input current from the user-supplied precision reference for a 4 mA full-scale output current. In applications that do not require such high output currents, good accuracy can be achieved with input reference currents in the range of 350 µA ≤ IREF ≤ 625 µA. The best signal-to-noise ratios, of course, will be achieved with a 625 µA reference current which yields a maximum 5 mA output current. Figure 22 illustrates how to form the reference input current with a REF02 and a 10 kΩ precision resistor. +15V thermally well-matched. Thin-film resistor networks work well here. In this circuit, the parallel combination of R1 and R2 forms a 3 Hz low-pass filter with C1. The only noise source that remains is the thermal noise of R2 which can be a significantly lower noise generator than the voltage reference. Input Coding The unipolar digital input coding of the DAC16 employs negative logic to control the output current; that is, an all zero input code (0000H) yields an output current 1 LSB below full scale. Conversely, an all 1s input code (FFFFH) yields a zero analog current output. An expression for the DAC16’s transfer equation can be expressed by: 65,535 – Digital Code IOUT = 8 × IREF × 65,536 Table II provides the relationship between the digital input codes and the output current of the DAC16. RREF 10kV 0.1mF IREF REF02 Table II. Unipolar Code Table DAC16 IOUT REF GND IREF = VREF RREF Figure 22. Generating the DAC16’s Reference Input Current Reducing Voltage Reference Noise In data converters of 16-bit and greater resolution, noise is of critical importance. Surprisingly, the integrated voltage reference circuit used may contribute the dominant share of a system’s noise floor, thereby degrading system dynamic range and signal-to-noise ratio. To maximize system dynamic range and SNR, all external noise contributions should be effectively much less than 1/2 LSB. For example, in a 5 V DAC16 application, one LSB is equivalent to 76 µV. This means that the total wideband noise contribution due to a voltage reference and all other sources should be less than 38 µV rms. These noise levels are not easy targets to hit with standard off-the-shelf reference devices. For example, commercially available references might exhibit 5 µV rms noise from 0.1 Hz to 10 Hz: but, over a 100 kHz bandwidth, its 300 µV rms of noise can easily swamp out a 16-bit system. Such noisy behavior can degrade a DAC’s effective resolution by increasing its differential nonlinearity which, in turn, can lead to nonmonotonic behavior or analog errors. The easiest way to reduce noise in the reference circuit is to band-limit its noise before feeding it to the converter. In the case of the DAC16, the reference is not a voltage, but a current. Illustrated in Figure 23 is a simple way of hand-limiting Digital Input Word (Hex) DAC16 Output Current IOUT Comment 0000 7FFE 7FFF 8000 FFFF 8 × (2 – 1)/2 × IREF 8 × (215 + 1)/216 × IREF 8 × (215/216) × IREF 8 × (215– 1)/216 × IREF 0 Full Scale Midscale + 1 LSB Midscale Midscale – 1 LSB Zero Scale 16 16 Since the DAC16 exhibits a small output voltage compliance on the order of a few millivolts, a high accuracy operational amplifier must be used to convert the DAC’s output current to a voltage. Refer to the section on selecting operation amplifiers for the DAC16. The circuit shown in Figure 24 illustrates a unipolar output configuration. In symbolic form, the transfer equation for this circuit can be expressed by: 65,535 – Digital Code VO = R3 × 8 × IREF 65,536 In this example, the reference input current was set to 500 µA which produces a full-scale output current of 4 mA – 1 LSB. The DAC’s output current was scaled by R3, a 1.25 kΩ resistor, to produce a 5 V full-scale output voltage. Bear in mind that to ensure the highest possible accuracy, matched thin-film resistor networks are almost a necessity, not an option. The resistors used in the circuit must have close tolerance and tight thermal tracking. Table III illustrates the relationship between the input digital code and the circuit’s output voltage for the component values shown. +15V Table III. Unipolar Output Voltage vs. Digital Input Code R1 5kV 0.1mF REF02 R2 5kV IREF C1 22mF DAC16 REF GND AGND Figure 23. Filtering a Reference’s Wideband Noise voltage reference noise by splitting RREF into two equal resistors and bypassing the common node with a capacitor. To minimize thermally induced errors, R1 and R2 must be electrically and Digital Input Word (Hex) Decimal Number in in DAC Decoder Analog Output Voltage (V) 0000 7FFE 7FFF 8000 FFFF 65,535 32,769 32,768 32,767 0 4.999924 2.500076 2.500000 2.499924 0 –8– REV. B DAC16 DIGITAL INPUT WORD DB0 – DB7 DB8 – DB15 CLK 74AC11377 EN R3 2.5kV (5kV42) 74AC11377 +15V 0.1mF +15V 8 0.1mF R2 5kV R1 5kV REF02 8 10mF IOUT IREF 22mF OP97A DAC16 REF GND 0.1mF CCOMP +5V 10mF AGND 100nF CERAMIC 47mF DIGITAL +5V PIN 3, DAC16 10mF 0.1mF 10mF –15V 0.1mF VOUT 0V TO +10V FS –15V RESISTORS: CADDOCK T912–5K–010–02 (OR EQUIVALENT) 5kV, 0.01%, TC TRACK = 2 ppm/8C –15V Figure 24. Unipolar Circuit Configuration DIGITAL INPUT WORD DB8 – DB15 DB0 – DB7 CLK 74AC11377 EN R3 2.5kV (5kV42) R4 2.5kV (5kV42) 74AC11377 +15V 0.1mF +15 V 0.1mF +5VREF 8 +5VREF REF02 R2 5kV R1 5kV 22mF 8 10mF IOUT IREF REF GND OP97A DAC16 0.1mF CCOMP +5V 10mF DIGITAL +5V PIN 3, DAC16 0.1mF AGND 100nF CERAMIC 47mF VOUT 65V FS 10mF 0.1mF 10mF –15V –15V –15V RESISTORS: CADDOCK T912–5K–010–02 (OR EQUIVALENT) 5kV , 0.01%, TC TRACK = 2 ppm/8C Figure 25. Bipolar Circuit Configuration Bipolar Configuration Table IV. Bipolar Output Operation vs. Digital Input Code For applications that require a bipolar output voltage, the circuit in Figure 24 can be modified slightly by adding a resistor from the reference to the inverting sum node of the output amplifier to level shift the output signal. The transfer equation for the circuit now becomes: 65,535 – Digital Code R4 VO = R4 × 8 × IREF – VREF × R3 65,536 The circuit has the form shown in Figure 25, and Table IV provides the relationship between the digital input code and the circuit’s output voltage for the component values shown. REV. B –9– Digital Input Word (Hex) Decimal Number in DAC Decoder Analog Output Voltage (V) 0000 7FFE 7FFF 8000 FFFF 65,535 32,769 32,768 32,767 0 4.999848 152E-6 0 –152E-6 –5.00000 DAC16 DIGITAL INPUT WORD DB8 – DB15 DB0 – DB7 CLK 74AC11377 EN R3 1.25kV (2.49kV42) 74AC11377 +15V 0.1mF +15V 8 0.1mF REF02 8 10mF R2 5kV R1 5kV IOUT IREF 22mF REF GND OP27A DAC16 0.1mF C1 47mF +5V 10mF DIGITAL +5V PIN 3, DAC16 CCOMP AGND C2 100nF CERAMIC 0.1mF 10mF 10mF –15V VOUT 0V TO +5V FS –15V RESISTORS: CADDOCK T912–5K–010–02 (OR EQUIVALENT) 5kV, 0.01%, TC TRACK = 2 ppm/8C 0.1mF –15V Figure 26. DAC16 Noise Measurement Test Circuit at different speeds, then the DAC output current will momentarily take on some incorrect value. This effect is particularly troublesome at the “carry points,” where the DAC output is to change by only one LSB, but several of the larger current sources must be switched to realize this change. Data skew can allow the DAC output to move a substantial amount towards full scale or zero (depending upon the direction of the skew) when only a small transition is desired. The glitch-sensitive user should be equally diligent about minimizing the data skew at the DAC16’s inputs, particularly the five most significant bits. This can be achieved by using the proper logic family and gate to drive the DAC inputs, and keeping the interconnect lines between the latches and the DAC inputs as short and as well matched as possible. Logic families that were empirically determined to operate well with the DAC16 are devices from the 74AC11xxx and 74ACT11xxx advanced CMOS logic families. These devices have been purposely designed with improved layout and tailored rise times for minimizing ground bounce and digital feedthrough. DAC16 Noise Performance The novel architecture employed in the DAC16 yields very low wideband noise. Figure 26 illustrates the circuit configuration for evaluating the DAC16’s noise performance. An OP27 is used as the DAC16’s output I–V converter which is configured to produce a 5 V full-scale output voltage. The output of the OP27 was then capacitively coupled to an OP37 stage configured in a gain of 101. Note that the techniques for reducing wideband noise of the voltage reference and the DAC’s internal reference amplifier were used. As a result of these techniques, the DAC16 exhibited a full-scale output noise spectral density of 31 pA/√Hz at 1 kHz. Digital Feedthrough and Data Skew The DAC16 features a compound DAC architecture where the 5 most significant bits utilize 31 identical, segmented current sources to obtain optimal high speed settling at major code transitions. Although every effort has been made to equalize the speeds at which the DAC switches operate, there exists finite skew in the MSB DAC switches. As with any converter product, a high speed digital-to-analog converter is forced to exist on the frontier between the noisy environment of high speed digital logic and the sensitive analog domain. The problems of this interlace are particularly acute when demands of high speed (greater than 10 MHz switching times) and high precision are combined. No amount of design effort can perfectly isolate the analog portions of a DAC from the spectral components of a digital input signal with a 2 ns rise time. Inevitably, once this digital signal is brought onto the chip, some of its higher frequency components will find their way to the sensitive analog nodes, producing a digital feedthrough glitch. To minimize the exposure to this effect, the DAC16 was designed to omit intentionally the on-board latches that are usually included in many slower DACs. This not only reduces the overall level of digital activity on chip, it also avoids bringing a latch clock pulse onto the IC, whose opposite edge inevitably produces a substantial glitch, even when the DAC is not supposed to be changing codes. The DAC16 uses each digital input line to switch each current segment in the DAC between the output diode-connected transistor and the logic control transistor. If the input bits are not changed simultaneously, or if the different DAC bits switch Deglitching The output glitch of the DAC16 at the major carry (7FFEH to 7FFFH) is a not-insignificant 360 pA-sec, manifested as a momentary output transition to the negative rail for approximately 200 ns. Due to the inherent low-pass or time-sampled nature of many systems, this behavior in the DAC16 is not noticeable and does not detract from overall performance. Some applications however may prove so sensitive to glitch impulse that reduction by an order of magnitude or more is required. In order to realize low glitch impulses, some sort of sample-andhold amplifier-based deglitching scheme must be used. There are high speed SHAs available with specifications sufficient to deglitch the DAC16; however, most are hybrid in topology at costs which can be prohibitive. A high performance, low cost alternative shown in Figure 27 is a discrete SHA utilizing a high speed monolithic op amp and high speed DMOS FET switches. This SHA circuit uses the inverting integrator structure. A 300 MHz gain-bandwidth product op amp, the AD841, is the heart of this fast SHA. The time constant formed by the 200 Ω resistor and the 100 pF capacitor determines the acquisition time and also hand limits the output signal to eliminate slewinduced distortion. –10– REV. B DAC16 200V 200V INPUT +15V 14 IN4735 13 M1 16 11 12 M2 9 100pF 360V 360V +5V 6 MC10124 Q1 T/H 249V –15V 169V 169V 510V –5V –5V Q2 20kV AD841 OUTPUT 249V –15V 3 4 M4 1 Q1, Q2 = MPS571 M1 – M4 = SD5000 0.39mF 75V 500pF –5V TO PIN 2 SD5000 1.6kV 8 M3 5 Figure 27. A High Performance Deglitching Circuit A discrete drive circuit is used to achieve the best performance from the SD5000 quad DMOS switch. This switch-driving cell is composed of MPS571 RF NPN transistors and an MC10124 TTL-to-ECL translator. Using this technique provides both high speed and highly symmetrical drive signals for the SD5000 switches. The switches arc arranged in a single-pole, doublethrow (SPDT) configuration. The 500 pF “flyback” capacitor is switched to the op amp summing junction during the hold mode to keep switching transients from feeding to the output. This capacitor is grounded during sample mode to minimize its effect on acquisition time. Careful circuit layout of the high speed SHA section is almost as important as the design itself. Double-sided printed circuit board, a compact layout, and short critical signal paths all ensure best performance. Op Amp Selection When selecting the amplifier to be used for the DAC16’s I–V converter, there are two main application areas; those requiring high accuracy, and those seeking high speed. In high accuracy applications, three parameters are of prime importance: (1) input offset voltage. VOS; (2) input bias current, –IB; and (3) offset voltage drift, TCVOS. In these applications where 16-bit performance must be maintained with an external reference at +5 V, an op amp’s input offset voltage must be less than 15 µV (≈0.1 LSB) with a bias current less than 6 nA. The op amp must also exhibit high open-loop gain to keep the offset voltage below this limit over the specified full-scale output range. Thus, for a maximum output of 5 V, the op amp’s open loop gain must be greater than 1300 V/mV. For low frequency, high accuracy applications, Table IV lists selected compatible operational amplifiers available from Analog Devices. These operational amplifiers satisfy all the above requirements and in most all cases will not require offset voltage nulling. Table V. Precision Operational Amplifier the DAC16 Model VOS TCVOS IB AVOL OP177 OP77 OP27 OP97 10 µV 25 µV 25 µV 25 µV 0.3 µV/°C 0.6 µV/°C 0.3 µV/°C 2 µV/°C 2 nA 2.8 nA 80 nA 0.15 nA 12000 V/mV 2000 V/mV 1500 V/mV 2000 V/mV REV. B In high speed applications where resolution is more important than absolute accuracy, operational amplifiers such as the AD843 offer the requisite settling time. Although these amplifiers are not specified for 16-bit performance, their settling times are two to three times faster than the DAC16 and will introduce negligible error to the overall circuit’s settling time. It is possible to estimate the 16-bit settling time of an operational amplifier if its 12-bit settling time is known. Assuming that the op amp can be modeled by a single-pole response, then the ratio of the op amp’s 16-bit settling time to its 12-bit settling can be expressed as: ts (16 − bit ) = 1.33 ts (12 − bit ) Since many operational amplifier data sheets provide charts illustrating 0.01% settling time versus output voltage step size, all that is required to estimate an op amp’s 16-bit settling time is to multiply the 12-bit settling time for the required full-scale voltage by 1.33. The circuit’s overall settling time can then be approximated by the root-sum-square method: 2 2 t S = (t DAC ) + (tOA ) where tDAC = DAC16’s specified full-scale settling time tOA = Op amp full-scale settling time As a design aid, Table VI illustrates a high speed operational amplifier selector guide for devices compatible with the DAC16 for high speed applications. All these devices exhibit the requisite settling time, input offset voltage, and input bias current consistent with maximum performance. Table VI. High Speed Operational Amplifiers for the DAC16 Model tS to % VOS TCVOS IB AVOL OP467 AD817 AD829 AD841 AD843 AD845 AD847 0.5 mV 2 mV 0.5 mV 1 mV 1 mV 0.25 mV 1 mV 3.5 µV/°C 10 µV/°C 0.3 µV/°C 35 µV/°C 12 µV/°C 5 µV/°C 15 µV/°C 0.5 µA 6.6 µA 7 µA 5 µA 0.001 µA 0.001 µA 5 µA 20 V/mV 6 V/mV 100 V/mV 45 V/mV 25 V/mV 500 V/mV 5.5 V/mV –11– 200 ns –0.01 70 ns –0.01 90 ns –0.1 110 ns –0.01 135 ns –0.01 350 ns –0.01 120 ns –0.01 DAC16 The choice of amplifier depends entirely on the required system accuracy, the required temperature range, and the operating frequency. CFB DAC16 IDAC RFB RO CO CIN VOUT C1883b–1–4/99 In using high speed op amps, the output capacitance of the DAC16 appears across the inputs of the op amp where it and the op amp’s input capacitance will set an additional pole in the op amp’s loop gain response. The pole is formed with the feedback resistance and the output resistance of the DAC. This additional pole may adversely affect the transient response of the circuit due to the added phase shift. Placing a small capacitor across the feedback resistance, as shown in Figure 28, compensates for the additional pole. The value of the capacitor can be determined by setting RFBCFB = RO (CO + CIN) and should be adjusted for optimum transient response. Figure 28. Compensating for the Feedback Pole OUTLINE DIMENSIONS Dimensions shown in inches and (mm). 24-Lead Epoxy DIP (P) (N-24) 24-Lead Epoxy SOL (S) (R-24) 1.275 (32.30) 1.125 (28.60) 13 12 1 0.280 (7.11) 0.240 (6.10) PIN 1 0.060 (1.52) 0.015 (0.38) 0.210 (5.33) MAX 0.200 (5.05) 0.125 (3.18) 0.150 (3.81) MIN 0.022 (0.558) 0.014 (0.356) 0.100 (2.54) BSC 0.070 (1.77) SEATING 0.045 (1.15) PLANE 24 13 0.325 (8.25) 0.300 (7.62) 0.2992 (7.60) 0.2914 (7.40) 0.195 (4.95) 0.115 (2.93) 1 0.015 (0.381) 0.008 (0.204) PIN 1 0.0118 (0.30) 0.0500 0.0040 (0.10) (1.27) BSC 12 0.4193 (10.65) 0.3937 (10.00) 0.1043 (2.65) 0.0926 (2.35) 88 08 0.0192 (0.49) SEATING 0.0125 (0.32) 0.0138 (0.35) PLANE 0.0091 (0.23) 0.0291 (0.74) 3 458 0.0098 (0.25) 0.0500 (1.27) 0.0157 (0.40) PRINTED IN U.S.A. 24 0.6141 (15.60) 0.5985 (15.20) –12– REV. B