ETC EUP3411

芯美电子
EUP3411
2A,25V,380KHz Step-Down Converter
with Soft Start
DESCRIPTION
FEATURES
The EUP3411 is a current mode step-down switching
regulator with a built in internal power MOSFET. It
achieves 2A continuous output current over a wide
input supply range with excellent load and line
regulation.
Current mode operation allows for fast dynamic
response and instantaneous duty cycle adjustment as
the input varies.
The EUP3411 has an internal soft-start circuit to
minimize the inrush current and the output overshoot at
start-up. Cycle-by-cycle current limiting and thermal
shutdown are provided. It draws 0.45mA of supply
current in standby mode, and only 16µA in shutdown
mode.
The EUP3411 is stable with low ESR output ceramic
capacitors.
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2A Output Current
0.17ΩInternal Power MOSFET Switch
4.75V to 25V Wide Operating Input Range
Output Adjustable from 1.2V to 16V
Up to 95% Efficiency
Low 16µA Shutdown Current
Fixed 380KHz Frequency
Thermal Shutdown and Overcurrent Protection
Programmable Under Voltage Lockout
Available in MSOP-10 with Exposed Pad
Package
RoHS Compliant and 100% Lead(Pb)-Free
APPLICATIONS
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DSL Modems
Broadband Networking Products
Commercial Low Power Systems
Typical Application Circuit
Figure 1.
DS3411
Ver1.0
Jan. 2008
1
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EUP3411
Pin Configurations
Package Type
Pin Configurations
MSOP-10
Pin Description
PIN
NAME
1
NC
2
BS
3
NC
4
VIN
5
SW
6
GND
7
FB
8
COMP
9
EN
10
SS
DS3411
Ver1.0
Jan. 2008
DESCRIPTION
No Connect.
Bootstrap. A small capacitor (C5) is needed to drive the power switch’s gate above
the supply voltage. It is connected between the SW and BS pins to form a floating
supply across the power switch driver. The voltage across C5 is about 5V and is
supplied by the internal 5V supply when the SW pin voltage is low.
No Connect.
Supply Voltage. The EUP3411 operates from a 4.75V to 25V unregulated input. C1
is needed to prevent large voltage spikes from appearing at the input.
Switch. This is the source of the internal switching MOSFET.
Ground.
Feedback. An external resistor divider from the output to GND, tapped to the FB pin,
sets the output voltage. To prevent current limit runaway during a short circuit fault
condition the frequency foldback comparator lowers the oscillator frequency when
the FB voltage is below 400mV.
Compensation. This node is the output of the transconductance error amplifier and is
used for frequency compensation of the feedback loop. Frequency compensation is
done at this node by connecting a series R-C to ground.
Enable/UVLO. A voltage greater than 3V enables operation. Leave EN unconnected
for automatic startup. This pin actually has two thresholds. If it is taken below the
first threshold voltage, the switch turns off and the output falls to zero. The internal
circuitry remains active. If the voltage on this pin is lowered below the second
threshold, the IC enters shutdown mode drawing only 16µA from the input.
An Under Voltage Lockout (UVLO) function can be implemented by the addition of
a resistor divider from VIN to GND.
Soft-Start. Connect SS to an external capacitor to program the soft-start. If unused,
leave it open.
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EUP3411
Ordering Information
Order Number
EUP3411MIR1
Package Type
Marking
Operating Temperature range
MSOP-10
xxxxx
3411A
-40 °C to 85°C
EUP3411 □ □ □ □
Lead Free Code
1: Lead Free 0: Lead
Packing
R: Tape & Reel
Operating temperature range
I: Industry Standard
Package Type
M: MSOP
Block Diagram
Figure 2.
DS3411
Ver1.0
Jan. 2008
3
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EUP3411
Absolute Maximum Ratings
Supply Voltage (VIN) --------------------------------------------------------------- -0.3V to 26V
Switch Node Voltages (VSW) --------------------------------------------------- -1V to VIN+0.3V
Bootstrap Voltage (V BS ) -------------------------------------- V SW -0.3V to V SW +6V
Feedback Voltage (VFB) ---------------------------------------------------------- -0.3V to 6V
Enable/UVLO Voltage (VEN) ---------------------------------------------------- -0.3V to 6V
Comp Voltage (VCOMP) ----------------------------------------------------- -0.3V to 6V
SS Voltage (VSS) ---------------------------------------------------------------- -0.3V to 6V
Junction Temperature -------------------------------------------------------------------- 150°C
Lead Temperature ------------------------------------------------------------------------ 260°C
Storage Temperature -------------------------------------------------------- -65°C to 150°C
„
„
„
„
„
„
„
„
„
„
Operating Ratings
Supply Voltage (VIN) ------------------------------------------------------------- 4.75V to 25V
Operating Temperature ---------------------------------------------------------- -40°C to 85°C
Thermal Resistance θJA (MSOP-10) ----------------------------------------------- 61.11°C/W
„
„
„
Electrical Characteristics
Unless otherwise specified, VIN=12V ,TA=25°C.
Parameter
Feedback Voltage
Symbol
Conditions
VFB
4.75V ≤ VIN ≤ 25V
EUP3411
Min
Typ Max.
1.162
1.200
1.236
Unit
V
Upper Switch On Resistance
RDS(ON)1
0.17
Ω
Lower Switch On Resistance
RDS(ON)2
6.8
Ω
Upper Switch Leakage
VEN=0V, VSW=0V
Current Limit
Current Sense Transconductance Output
Current to Comp Pin Voltage
Error Amplifier Voltage Gain
Error Amplifier Transconductance
Oscillation Frequency
GEA
Fosc1
∆IC=±10uA
Short Circuit Frequency
Maximum Duty Cycle
Fosc2
VFB=0V
VFB=1.0V
A
GCS
2
A/V
AVEA
400
V/V
700
380
uA/V
KHz
320
0.7
VEN=0V
VEN Rising
Operating Supply Current
Soft-Start Current
Thermal Shutdown
Jan. 2008
440
45
90
VFB=1.5V
EN Pull Up Current
EN Under Voltage Lockout Threshold
Voltage
EN Under Voltage Lockout Hysteresis
Shutdown Supply Current
Ver1.0
uA
3
Minimum Duty Cycle
EN Shutdown Threshold Voltage
DS3411
5
0.95
KHz
%
0
1.4
4
2
2.5
VEN=0V
110
16
VFB=1.4V
VSS=0V
0.45
5.2
%
V
uA
3
V
30
mV
uA
0.7
160
4
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mA
uA
℃
芯美电子
EUP3411
Typical Characteristics
C1=10uF, C2=22uF, L=15uH, TA=25℃.
Efficiency vs Output Current
95
5.0V
90
EFFICIENCY(%)
3.3V
85
2.5V
80
75
70
VIN=12V
L=15uH
65
60
0
500
1000
1500
2000
OUTPUT CURRENT(mA)
DS3411
Ver1.0
Jan. 2008
5
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EUP3411
Typical Characteristics (continued)
C1=10uF, C2=22uF, L=15uH, TA=25℃.
DS3411
Ver1.0
Jan. 2008
6
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EUP3411
Inductor Selection
The inductor is required to supply constant current to
the output load while being driven by the switched
input voltage. A larger value inductor will result in
less ripple current that will result in lower output
ripple voltage. However, the larger value inductor
will have a larger physical size, higher series
resistance, and/or lower saturation current. A good
rule for determining the input voltage and inductance
to use is to allow the peak-to- peak ripple current in
the inductor to be approximately 30% of the
maximum switch current limit. Also, make sure that
the peak inductor current is below the maximum
switch current limit. The inductance value can be
calculated by:

 V
V
L = OUT ∗  1 − OUT 

VIN 
fS ∗ ∆I L 
Functional Description
The EUP3411 is a current-mode step-down
switch-mode regulator. It regulates input voltages
from 4.75V to 25V down to an output voltage as low
as 1.2V, and is able to supply up to 2A of load current.
The EUP3411 uses current-mode control to regulate
the output voltage. The output voltage is measured at
FB through a resistive voltage divider and amplified
through the internal error amplifier. The output
current of the transconductance error amplifier is
presented at COMP where a network compensates the
regulation control system. The voltage at COMP is
compared to the switch current measured internally to
control the output voltage. Slope compensation
provides stability in constant frequency architectures
by preventing subharmonic oscillations at high duty
cycles. It is accomplished internally by adding a
compensating ramp to the inductor current signal.
Normally, this results in a reduction of maximum
inductor peak current for high duty cycles.
The converter uses an internal n-channel MOSFET
switch to step down the input voltage to the regulated
output voltage. Since the MOSFET requires a gate
voltage greater than the input voltage, a boost
capacitor connected between SW and BS drives the
gate. The capacitor is internally charged while the
switch is off. An internal 6.8Ω switch from SW to
GND is used to insure that SW is pulled to GND
when the switch is off to fully charge the BS
capacitor.
Where fS is the switching frequency, ∆IL is the
peak-to-peak inductor ripple current and VIN is the
input voltage.
Choose an inductor that will not saturate under the
maximum inductor peak current. The peak inductor
current can be calculated by:
 V
VOUT
∗  1 − OUT
I LP = I LOAD +
VIN
2 ∗ f S ∗ L 
Where ILOAD is the load current.
Output Rectifier Diode
Application Information
The output rectifier diode supplies the current to the
inductor when the high-side switch is off. To reduce
losses due to the diode forward voltage and recovery
times, use a Schottky diode.
Choose a diode whose maximum reverse voltage
rating is greater than the maximum input voltage, and
whose current rating is greater than the maximum
load current.
Setting the Output Voltage
The output voltage is set using a resistive voltage
divider from the output voltage to FB pin. The
voltage divider divides the output voltage down to the
feedback voltage by the ratio:
R2
V
=V
∗
FB
OUT R1 + R 2
Input Capacitor
Where VFB is the feedback voltage and VOUT is the
output voltage.
Thus the output voltage is :
VOUT = 1.20 ∗
The input current to the step-down converter is
discontinuous, therefore a capacitor is required to
supply the AC current to the step-down converter
while maintaining the DC input voltage. Use low
ESR capacitors for the best performance. Ceramic
capacitors are preferred, but tantalum or low-ESR
electrolytic capacitors may also suffice.
Since the input capacitor absorbs the input switching
current it requires an adequate ripple current rating.
The RMS current in the input capacitor can be
estimated by:
R1 + R 2
R2
A typical value for R2 can be as high as 100kΩ, but a
typical value is 10kΩ. Using that value, R1 is
determined by :
(
R1 = 8.33 ∗ VOUT − 1.20
)
For example, for a 3.3V output voltage, R2 is 10kΩ,
and R1 is 17.5kΩ.
I CIN = I LOAD ∗
DS3411
Ver1.0
Jan. 2008




V
VOUT 
∗  1 − OUT

VIN
VIN





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EUP3411
The worst-case condition occurs at VIN = 2VOUT,
where:
Compensation Components
The EUP3411 employs current mode control for easy
compensation and fast transient response. The system
stability and transient response are controlled through
the COMP pin. COMP pin is the output of the internal
transconductance
error
amplifier.
A series
capacitor-resistor combination sets a pole-zero
combination to control the characteristics of the
control system.
I
I CIN = LOAD
2
For simplification, choose the input capacitor whose
RMS current rating greater than half of the maximum
load current.
The input capacitor can be electrolytic, tantalum or
ceramic. When using electrolytic or tantalum
capacitors, a small, high quality ceramic capacitor, i.e.
0.1µF, should be placed as close to the IC as possible.
When using ceramic capacitors, make sure that they
have enough capacitance to provide sufficient charge
to prevent excessive voltage ripple at input. The input
voltage ripple caused by capacitance can be estimated
by:
V
I
 V

∆V
= LOAD ∗ OUT ∗  1 − OUT 
IN f ∗ C


V
V
IN 
IN
S
IN

The DC gain of the voltage feedback loop is given by:
A
Output Capacitor
The output capacitor is required to maintain the DC
output voltage. Ceramic, tantalum, or low ESR
electrolytic capacitors are recommended. Low ESR
capacitors are preferred to keep the output voltage
ripple low. The output voltage ripple can be estimated
by:
 V
 

1
V

∆ V OUT = OUT ∗  1 − OUT  ∗  R ESR +


V
8 ∗ f S ∗ C O 
fS∗L 
IN  
f
CS
∗A
VEA
∗
V
FB
V
OUT
P1
P2
=
G
EA
2 π ∗ C3 ∗ A
=
2π ∗ C
O
VEA
1
∗R
LOAD
Where GEA is the error amplifier transconductance.
The system has one zero of importance, due to the
compensation capacitor (C3) and the compensation
resistor (R3). This zero is located at:
f
Z1
1
=
2 π ∗ C3 ∗ R 3
The system may have another zero of importance, if
the output capacitor has a large capacitance and/or a
high ESR value. The zero, due to the ESR and
capacitance of the output capacitor, is located at:
f
ESR
=
1
2π ∗ C o ∗ R
ESR
In this case, a third pole set by the compensation
capacitor (C6) and the compensation resistor (R3) is
used to compensate the effect of the ESR zero on the
loop gain. This pole is located at:
f
The characteristics of the output capacitor also affect
the stability of the regulation system. The EUP3411 can
be optimized for a wide range of capacitance and ESR
values.
Jan. 2008
LOAD
f
Where L is the inductor value, RESR is the equivalent
series resistance (ESR) value of the output capacitor
and CO is the output capacitance value.
In the case of ceramic capacitors, the impedance at the
switching frequency is dominated by the capacitance.
The output voltage ripple is mainly caused by the
capacitance. For simplification, the output voltage
ripple can be estimated by:
 V

V
OUT
∆V
=
∗  1 − OUT 
OUT 8 ∗ 2 ∗ L ∗


V
CO 
fS
IN 
In the case of tantalum or electrolytic capacitors, the
ESR dominates the impedance at the switching
frequency. For simplification, the output ripple can be
approximated to:

 V
V
∆V
= OUT ∗  1 − OUT  ∗ R
OUT
ESR

f ∗L 
V
S
IN 

Ver1.0
∗G
Where RLOAD is the load resistor value, GCS is the
current sense transconductance and AVEA is the error
amplifier voltage gain.
The system has two poles of importance. One is due
to the compensation capacitor (C3) and the output
resistor of error amplifier, and the other is due to the
output capacitor and the load resistor. These poles are
located at:
Where CIN is the input capacitance value.
DS3411
VDC
=R
P3
=
1
2π ∗ C6 ∗ R 3
The goal of compensation design is to shape the
converter transfer function to get a desired loop gain.
The system crossover frequency where the feedback
loop has the unity gain is important.
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EUP3411
Lower crossover frequencies result in slower line and
load transient responses, while higher crossover
frequencies could cause the system to become unstable.
A good rule of thumb is to set the crossover frequency
to below one-tenth of the switching frequency. To
optimize the compensation components, the following
procedure can be used:
1. Choose the compensation resistor (R3) to set the
desired crossover frequency. Determine the R3 value
by the following equation:
R3 =
2π ∗ C
∗f
V
O C ∗ OUT
V
G
∗G
FB
EA
CS
Figure 3.
Power Dissipation and Temperature Rise
Where fC is the desired crossover frequency, which is
typically less than one tenth of the switching frequency.
2. Choose the compensation capacitor (C3) to achieve
the desired phase margin. For applications with typical
inductor values, setting the compensation zero, fZ1, to
below one forth of the crossover frequency provides
sufficient phase margin. Determine the C3 value by the
following equation:
2
C3 >
π ∗ R3 ∗ f
The power dissipation of the EUP3411 is mostly from
the conduction loss of the internal main switch. This
power loss is estimated to be:
V
2 ∗R
P
≅ OUT ∗ I
∗ 1 .3
LOSS
OUT
DS(ON)
V
IN
Where 1.3 is a temperature coefficient factor that
reflects the increase in the RDS(ON) resistance at
elevated temperatures.
For example: for VIN = 12V, VOUT = 3.3V and IOUT =
2A:
C
Where R3 is the compensation resistor value.
3. Determine if the second compensation capacitor (C6)
is required. It is required if the ESR zero of the output
capacitor is located at less than half of the switching
frequency, or the following relationship is valid:
1
2π ∗ C
O
∗R
ESR
3 .3 V
P
≅
∗ (2 A )2 ∗ 0.17 Ω ∗ 1.3 = 0.24 W
LOSS
12 V
f
< S
2
The junction temperature of the EUP3411 can be
further determined by:
If this is the case, then add the second compensation
capacitor (C6) to set the pole fP3 at the location of the
ESR zero. Determine the C6 value by the equation:
C6 =
C
O
∗R
TJ = TA + θ JA ∗ PLOSS
θJA is the thermal resistance from junction to ambient.
Its value is a function of the IC package, the
application layout and the air cooling system.
Because the thermal resistance θ JA is 61.11°C/W,
the resulting rise in temperature between junction and
ambient is approximately 15°C.Therefore, caution
must be exercised when using the EUP3411 in
applications with high duty cycles.
ESR
R3
External Bootstrap Diode
For applications with large duty cycles, it is
recommended that an external boost diode be
connected to a fixed 5V. This helps improve the
efficiency of the EUP3411 regulator and also avoids
the problems caused by the decrease of BS voltage
with large duty cycles. The fixed 5V can be pulled
from the input of the system the output generated by
the power supply.
DS3411
Ver1.0
Jan. 2008
9
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EUP3411
Layout Guidelines:
In order to achieve optimal electrical and thermal
performance, special attention must be paid to the
PCB layouts. The following guidelines should be
used to ensure proper operation of the converters.
1. A ground plane is suggested to minimize
switching noises and trace losses and maximize
heat transferring.
2. Start the PCB layout by placing the power
components first. Arrange the power circuit to
achieve a clean power flow route. Put al1 power
connections on one side of the PCB with wide
copper filled areas if possible.
3. The VIN bypass capacitor should be placed next
to the VIN and GND pins.
4. The trace connecting the feedback resistors to the
output should be short, direct and far away from
any noise sources such as switching node and
switching components.
5. Minimize the loop including input capacitor, the
EUP3411 and Schottky diode. Make sure the
trace width is wide enough to reduce copper
losses in this loop.
6. Maximize the trace width of the loop connecting
the inductor, Schottky diode and the output
capacitor.
7. Connect the ground of the feedback divider and
the compensation components directly to the
GND pin of the EUP3411 by using a separate
ground trace.
8. Connect GND to a large copper area to remove
the IC heat and increase the power capability of
the EUP3411. A few feedthrough holes are
required to connect this large copper area to a
ground plane to further improve the thermal
environment of the EUP3411. The traces attached
to other pins should be as wide as possible for the
same purpose.
DS3411
Ver1.0
Jan. 2008
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EUP3411
Packaging Information
MSOP-10
SYMBOLS
A
A1
D
E1
E
L
b
e
D1
E2
DS3411
Ver1.0
Jan. 2008
MILLIMETERS
MIN.
MAX.
1.10
0.00
0.15
3.00
3.00
4.70
5.10
0.40
0.80
0.17
0.33
0.50
1.80
1.66
INCHES
MIN.
0.000
MAX.
0.043
0.006
0.118
0.118
0.185
0.016
0.006
0.201
0.031
0.013
0.020
0.071
0.065
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