LT1306 Synchronous, Fixed Frequency Step-Up DC/DC Converter U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO Output Disconnected from Input During Shutdown Output Voltage Remains Regulated When VIN > VOUT Controlled Input Current During Start-Up 300kHz Current Mode PWM Operation Can Be Externally Synchronized Internal 2A Switches Operates with VIN as Low as 1.8V Automatic Burst Mode Operation at Light Loads Quiescent Current: 160µA Shutdown Current: 9µA Typ U APPLICATIO S ■ ■ ■ ■ Satellite Phones Portable Instruments Personal Digital Assistants Palmtop Computers , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. The LT®1306 is a fully integrated, fixed frequency synchronous boost converter capable of generating 5V at 1A from a Li-Ion cell. The device contains both the main power switch and synchronous rectifier on chip and automatically disconnects the output from the input in shutdown, eliminating the need for external load disconnect circuitry. Additionally, the output remains regulated when VIN exceeds VOUT, allowing difficult step-up/stepdown converter functions to be easily realized using a single inductor. The internal 300kHz oscillator of the LT1306 can be easily synchronized to an external clock from 425kHz to 500kHz. This allows switching harmonics to be tightly controlled and eliminates any beat frequencies that may result from a multifrequency system. The LT1306 automatically shifts into power saving Burst ModeTM operation at light loads. At heavy loads the LT1306 operates in fixed frequency current mode. No-load quiescent current is 160µA and reduces to 9µA in shutdown mode. The LT1306 is available in an SO-8 package. U TYPICAL APPLICATION D1 Efficiency SW S/S + CAP R1 768k CIN2 0.1µF FB VC R3 118k CZ 68nF 5V 1A OUT LT1306 CIN1 22µF 85 + GND CP 68pF CO1 220µF R2 249k 1306 F01 CO2 1µF CIN1: AVX TAJC226M010 CO1: AVX TPSE227M010R0100 CIN1, CO2: CERAMIC C1: AVX TAJA105K020 D1: MMBD914LT1 L1: CTX10-2 EFFICIENCY (%) VIN + L1 10µH 1-CELL Li-Ion 90 C1 1µF VIN = 4.2V 80 VIN = 3.6V VIN = 2.6V 75 70 VO = 5V L1 = 10µH (FIGURE 1) 65 60 1 10 100 LOAD CURRENT (mA) 1000 1306 TA01 Figure 1. Single Li-Ion Cell to 5V Converter 1 LT1306 W U U W W W (Note 1) U ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION VIN Voltage ............................................................. 10V S/S Voltage ............................................................... 7V FB Voltage .............................................................. 10V VOUT Voltage .......................................................... 5.5V Junction Temperature .......................................... 125°C Operating Temperature Range (Note 2) .. – 40°C to 85°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER TOP VIEW VC 1 8 S/S FB 2 7 VIN LT1306ES8 VOUT 3 6 CAP GND 4 5 SW S8 PART MARKING S8 PACKAGE 8-LEAD PLASTIC SO 1306 TJMAX = 125°C, θJA = 90°C/ W Consult factory for Industrial and Military grade parts. ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 2.5V, VS/S = VIN, VC open unless otherwise noted. PARAMETER CONDITIONS Reference Voltage Measured at the FB Pin Reference Line Regulation 1.8V ≤ VIN ≤ 7V FB Input Bias Current VFB = VREF Error Amplifier Transconductance ∆I = ±0.2µA Error Amplifier Output Source Current MIN MAX UNITS 1.24 1.26 V 0.002 0.1 %/V 10 25 nA 80 150 220 µΩ –1 VFB = 1V, VC = 0.8V 5 7.5 11 µA Error Amplifier Output Sink Current VFB = 1.5V, VC = 0.8V 5 7.5 11 µA Error Amplifier Output Clamp Voltage VFB = 1V 1.18 1.28 1.38 V 1.8 V 6 15 µA –3 µA 260 225 310 305 415 390 kHz kHz 90 80 ● ● VIN Undervoltage Lockout Threshold Idle Mode Output Leakage Current 1.22 TYP 1.55 VFB = 1.5V, VOUT = 5.5V, VSW = 1.7V ● Output Source Current in Shutdown VOUT = 0V, VIN = VSW = 7V, VCAP = 7.2V, VS/S = 0V ● Switching Frequency 1.8V ≤ VIN ≤ 7V, 0°C ≤ TA ≤ 85°C 1.8V ≤ VIN ≤ 7V, TA = – 40°C ● Maximum Duty Cycle VFB = 1V, 0°C ≤ TA ≤ 85°C VFB = 1V, TA = – 40°C 80 65 Switch Current Limit Duty Cycle = 0.1 (Note 3) Duty Cycle = 0.8 (Note 3) 2.3 2.0 Burst Mode Operation Switch Current Limit A A 250 Switch VCESAT ISW = 2A Rectifier VCESAT ISW = 2A Stepdown Mode Rectifier Voltage VOUT = 0V, ISW = 1A VOUT = 2.2V, ISW = 1A Switch and Rectifier Leakage Current VOUT = 0V, VIN = VSW = 7V, VCAP = 7.2V, VS/S = 0V 2 % % 0.45 0.49 0.3 + VIN 1.3 ● 0.1 mA 0.575 V 0.675 V 0.7 + VIN 1.8 V V 20 µA LT1306 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 2.5V, VS/S = VIN, VC open unless otherwise noted. PARAMETER CONDITIONS S/S Pin Current VS/S = VIN VS/S = 0V MIN Shutdown Pin Input High Voltage Shutdown Pin Input Low Voltage 1.2 Shutdown Delay 12 Synchronization Frequency Range 425 Operating Supply Current Quiescent Supply Current VS/S = VIN, VFB = 1.5V ● Shutdown Supply Current VS/S = 0V CAP Pin Leakage Current VIN = VCAP = 7V, VS/S = 2.5V, ISW = 0 TYP MAX 20 UNITS 6 –3 µA µA 0.45 V V 50 µs 500 kHz 4.5 8 mA 160 250 µA 16 µA 10 µA 9 ● Output Boost-to-Stepdown Threshold VIN V Output Stepdown-to-Boost Threshold VIN – 0.1 V Note 1: Absolute Maximum Ratings are those values beyond which the life to the device may be impaired. Note 2: The LT1306E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the – 40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 3: Switch current limit guaranteed by design/correlation to static tests. U W TYPICAL PERFORMANCE CHARACTERISTICS Reference Voltage vs Temperature Maximum Load Current vs Input Voltage REFERENCE VOLTAGE (V) ILOADMAX (A) VO = 3.3V VO = 5V 1.0 0.5 L = 10µH TJ = 125°C TA = 25°C TA = 50°C 0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 VIN (V) 1306 • G01 S/S Pin Current vs S/S Pin Voltage 1.239 5 1.238 4 TA = –40°C TA = 25°C 3 1.237 2 1.236 IS/S (µA) 1.5 1.235 1.234 TA = 85°C 1 0 –1 –2 1.233 –3 1.232 1.231 –40 –4 –5 –20 0 20 40 60 80 100 TEMPERATURE (°C) 1306 • G02 0 1 2 3 VS/S (V) 4 5 1306 • G03 3 LT1306 U W TYPICAL PERFORMANCE CHARACTERISTICS Shutdown Supply Current vs Input Voltage 40 155 VS/S = 2.5V TA = 25°C S/S CURRENT (µA) 30 25 20 TA = 85°C 15 TA = –40°C IDLE-MODE SUPPLY CURRENT (µA) 5.0 35 SUPPLY CURRENT (µA) Idle-Mode Supply Current vs Temperature S/S Pin Current vs Temperature 2.5 0 10 VS/S = 0V 5 0 4 6 8 INPUT VOLTAGE (V) 2 10 –2.5 – 40 12 0 20 40 60 TEMPERATURE (°C) –20 80 150 145 140 135 –40 100 –20 0 20 40 60 80 1306 • G04 1306 • G06 1306 • G05 Oscillator Frequency Line Regulation Frequency vs Temperature Maximum Duty Ratio 95 315 320 100 TEMPERATURE (°C) 310 VIN = 2.5V 90 305 310 305 85 300 DUTY RATIO (%) FREQUENCY (kHz) FREQUENCY (kHz) 315 295 290 285 280 80 75 70 275 65 270 300 0 1 2 3 4 5 6 7 8 9 265 –40 10 40 0 60 20 TEMPERATURE (°C) –20 VIN (V) 80 1306 • G07 100 Switch Saturation Voltage vs Current 0.7 3.0 TA = 25°C 0.6 500 400 300 200 SWITCH VOLTAGE (V) 2.8 CURRENT LIMIT (A) MAXIMUM RISE TIME (ns) 80 1306 • G09 Current Limit vs Duty Cycle 600 2.6 2.4 2.2 100 1 3.5 1.5 2.0 2.5 3.0 SYNCHRONIZING PULSE AMPLITUDE (V) 1306 • G10 4 40 20 60 0 TEMPERATURE (°C) 1306 • G08 Maximum Allowable Rise Time of Synchronizing Pulse 0 60 –40 –20 100 0.5 TA = 25°C 0.4 TA = 85°C 0.3 TA = –40°C 0.2 0.1 0 2.0 0 10 20 30 40 50 60 DUTY CYCLE (%) 70 80 90 1306 • G11 0 0.5 1.0 2.0 1.5 SWITCH CURRENT (A) 2.5 1306 • G12 LT1306 U W TYPICAL PERFORMANCE CHARACTERISTICS Rectifier Saturation Voltage vs Current 0.7 1.90 TA = 85°C 0.5 0.4 TA = –40°C VIN = 6V VOUT = 5V TA = 25°C 1.85 RECTIFIER VOLTAGE (V) 0.6 RECTIFIER VOLTAGE (V) Continuous-Conduction Mode Switching Waveforms in Boost Operation Stepdown-Mode Rectifier Voltage vs Current TA = 25°C 0.3 0.2 0.1 1.80 VSW 5V/DIV 1.75 IL 0.5A/DIV 1.70 1.65 VO 0.1V/DIV AC 1.60 0 1.55 0 0.5 1.0 2.0 1.5 RECTIFIER CURRENT (A) 2.5 0 0.5 1.0 1.5 RECTIFIER CURRENT (A) 1306 • G13 VO 5V/DIV VIN = 2.5V 1ms/DIV 2µs/DIV Transient Response of the Converter in Figure 1 with a 50mA to 800mA Load Step Continuous-Conduction Mode Switching Waveforms in Stepdown Mode VS/S 5V/DIV IL 2A/DIV VIN = 4.2V VO = 5V 1306 • G14 Start-Up to Shutdown Transient Response* VSW 5V/DIV 2.0 VSW 5V/DIV LOAD CURRENT 0.5A/DIV DC IL 0.5V/DIV INDUCTOR CURRENT 1A/DIV VO 50mV/DIV AC OUTPUT 0.1V/DIV AC VIN = 6V VO = 5V 2µs/DIV VIN = 3.6V VO = 5V 1ms/DIV *Notice that the Input Start-Up Current is well Controlled and the Output Voltage Falls to Zero in Shutdown. 5 LT1306 U U U PIN FUNCTIONS VC (Pin 1): Compensation Pin for Error Amplifier. VC is the output of the transconductance error amplifier. Loop frequency compensation is done by connecting an RC network from the VC pin to ground. FB (Pin 2): Inverting Input of the Error Amplifier. Connect the resistor divider tap here. Set output voltage according to VOUT = 1.24V (1 + R1/R2). VOUT (Pin 3): Output of the Switching Regulator and Emitter of the Synchronous Rectifier. Connect appropriate output capacitor from here to ground. VOUT must be kept below 5.5V. GND (Pin 4): Ground. Connect to local ground plane. CAP (Pin 6): Power Supply to the Synchronous Rectifier Driver. The bootstrap capacitor and the blocking diode are tied to this pin. The CAP voltage switches between a low level of VIN – VD to a high level determined by the VSW high level. VIN (Pin 7): Supply or Battery Input Pin. Must be closely bypassed to ground plane. S/S (Pin 8): Shutdown and Synchronization Pin. Shutdown is active low with a typical threshold of 0.9V. For normal operation, the S/S pin is tied to VIN. To externally synchronize the switching regulator, drive the S/S pin with a pulse train. SW (Pin 5): Switch Pin. The collectors of the grounded power switch and the synchronous rectifier. Keep the SW trace as short as possible to minimize EMI. W BLOCK DIAGRA VIN VC – 7 1 + A1 gm FB 2 – – X3 X4 A3 VB DCM CONTROL CAP 6 + 1.65V 1.24V IRECT > 0 UVLO A5 X5 IDLE + 5 SW A4 + S/S 8 SYNC Q X4 + A2 SENSE AMP CLK REF/BIAS SHUTDOWN DELAY Figure 2. LT1306 Block Diagram 6 Q1 X2 R PWM CONTROL SHDN RECTIFIER IRECT + VCE2 – RS – 300kHz OSC Σ Q2 S + + RAMP COMPENSATION OUT X1 – 4 GND 1306 F02 3 LT1306 U OPERATIO The LT1306 is a fixed frequency current mode PWM regulator with integrated power transistor Q1 and synchronous rectifier Q2. In the Block Diagram, Figure 2, the PWM control circuit is enclosed within the dashed line. It consists of the current sense amplifier (A2), the oscillator, the compensating ramp generator, the PWM comparator (A4), the logic (X1 and X2), the power transistor driver (X4) and the main power switch (Q1). Notice that the clock (CLK) “blanks” Q1 conduction. The internal oscillator frequency is 300kHz. The pulse width of the clock determines the maximum on duty ratio of Q1. In the LT1306 this is set to 88%. Q1 turns on at the trailing edge of the clock pulse. To prevent subharmonic oscillation above 50% duty ratio, a compensating ramp (generated from the oscillator sawtooth) is added to the sensed Q1 current. Q1 is turned off when this sum exceeds the error amplifier A1 output, VC. Q1’s absolute current limit is reached when VC’s upward excursion is clamped internally at 1.28V. The error amplifier output, VC, determines the peak switch current required to regulate the output voltage. VC is a measure of the output power. At heavy loads, the average and the peak inductor currents are both high. VC moves to the upper end of its operating range and the LT1306 operates in continuous conduction mode (CCM). As load decreases, the average inductor current decreases. In CCM, the peak-to-peak inductor current ripple to the first order depends only on the inductance, the input and the output voltages. When the average inductor current falls below 1/2 of the peak-to-peak inductor current ripple, the converter enters discontinuous conduction mode (DCM). The switching frequency remains constant except that the inductor current always returns to zero within each switching cycle. In both CCM and DCM, the output voltage is regulated with negative feedback. A1 amplifies the error voltage between the internally generated 1.24V reference and the attenuated output voltage. The RC network from the VC pin to ground provides the loop compensation. Further reduction in the load moves VC towards the lower end of its operating range. Both the peak inductor current and switch Q1’s on-time decrease. Hysteretic comparator A3 determines if VC is too low for the LT1306 to operate efficiently. As VC falls below the trip voltage VB, the output of A3 goes high. All circuits except the error amplifier, comparators A3 and A5, and the rectifier driver control X5, are turned off. After the remaining energy stored in the inductor is delivered to the output through the synchronous rectifier Q2, the LT1306 stops switching. In this idle state, the LT1306 draws only 160µA from the input. With switching stopped and the load being powered by the output filter capacitor, the output voltage decreases. VC then starts to increase. Q1 does not start to switch until VC rises above the upper trip point of A3. The LT1306 again delivers power to the output as a current mode PWM converter except that the switch current limit is only about 250mA due to the low value of VC. If the load is still light, the output voltage will rise and VC will fall, causing the converter to idle again. Power delivery therefore occurs in bursts. The on-off cycle frequency, or burst frequency, depends on the operating conditions, the inductance and the output filter capacitance. The output voltage ripple in Burst Mode operation is usually higher than either CCM or DCM operation. Burst Mode operation increases light load efficiency because it delivers more energy to the output during each clock cycle than is possible with DCM operation’s extremely low peak switch current. This allows fewer switching cycles per unit time to maintain a given output. Chip supply current therefore becomes a small fraction of the total input current. The synchronous rectifier is represented as NPN transistor, Q2, in the Block Diagram (Figure 2). A rectifier drive circuit, X5, supplies variable base drive to Q2 and controls the voltage across the rectifier. The supply voltage, VCAP, for the driver is generated locally with the bootstrap circuit, D1 and C1 (Figure 1). When Q1 is on, the bootstrap capacitor C1 is charged from the input to the voltage VIN – VD1(ON) – VCESAT1. The charging current flows from the input through D1, C1 and Q1 to ground. After Q1 is switched off, the node SW goes above VO by the rectifier drop VCESAT2. D1 becomes back-biased and the CAP voltage is pushed up to VO + VCESAT2 + VIN – VD1(ON) – VCESAT1. C1 supplies the base drive to Q2. The consumed charge is replenished during the Q1 on interval. 7 LT1306 U OPERATIO In boost operation, X5 drives the rectifier Q2 into saturation. The voltage across the rectifier is VCESAT. As the inductor current decreases, Q2’s base drive also decreases. X5 ceases supplying base current to Q2 when the inductor current falls to zero. If VIN > VO, Q2 will no longer be driven into saturation. Instead the voltage across Q2 is allowed to increase so that the inductor voltage reverses polarity as Q1 switches. Since the inductor voltage is bipolar, volt-second balance can be maintained regardless of the input voltage. The LT1306 is therefore capable of operating as a step-down regulator with the basic boost topology. Input start-up current is also well controlled since the inductor current cannot increase during Q1’s off-time with negative inductor voltage. The rectifier voltage drop depends on both the input and the output voltages. Efficiency in the step-down mode is less than that of a linear regulator. For sustained stepdown operation, the maximum output current will be limited by the package thermal characteristics. A hysteretic comparator in driver X5 controls the mode of operation. DC transfer characteristics of the comparator are shown in Figure 3 and Figure 4. A logic low at the S/S pin (Pin 8) initiates shutdown. First, all circuit blocks in the LT1306 are switched off. The synchronous rectifier Q2 and its driver are kept on to allow stored inductive energy to flow to the output. As VO drops below VIN, the voltage across the rectifier Q2 increases so that the inductor voltage reverses. Inductor current continues to fall to zero. Driver X5 then turns off and the rectifier, Q2, becomes an open circuit. The LT1306 dissipates only 9µA in shutdown. The LT1306 is guaranteed to start with a minimum VIN of 1.8V. Comparator A5 senses the input voltage and generates an undervoltage lockout (UVLO) signal if VIN falls below this minimum. In UVLO, VC is pulled low and Q1 stops switching. The LT1306 draws 160µA from the input. MODE MODE BOOST BOOST STEPDOWN STEPDOWN 1306 F03 0 VIN – 0.1V VIN VO Figure 3. DC Transfer Characteristics of the Mode Control Comparator Plotted with VO as an Independent Variable. VIN is Considered Fixed. 8 1306 F04 VO VO + 0.1V VIN Figure 4. DC Transfer Characteristics of the Mode Control Comparator Plotted with VIN as an Independent Variable. VO is Considered Fixed. LT1306 U W U U APPLICATIONS INFORMATION Output Voltage Setting The output voltage of the LT1306 is set with a resistive divider, R1 and R2 (Figure 1 and Figure 5), from the output to ground. The divider tap is tied to the FB pin. Current through R2 should be significantly higher than the FB pin input bias current (≤ 25nA). With R2 = 249k, the input bias current of the error amplifier is 0.5% of the current in R1. VO R1 FB PIN R2 ( ) ( ) VO = 1.24V 1 + R1 R2 VO –1 R1 = R2 1.24 1306 F05 Figure 5. Feedback Resistive Divider The inductor should be able to handle the full load peak inductor current without saturation. The peak inductor current can be as high as 2A. This places a lower limit on the core size of the inductor. Powder iron cores have unacceptable core losses and are not suitable for high efficiency applications. Most ferrite core materials have manageable core losses and are recommended. Inductor DC winding resistance (DCR) also needs to be considered for efficiency. Usually there are trade-offs between core loss, DCR, saturation current, cost and size. For EMI sensitive applications, one may want to use magnetically shielded or toroidal inductors to contain field radiation. Table 1 lists a number of inductors suitable for LT1306 applications. Table 1. Inductors Suitable for Use with the LT1306 VENDOR PART NO. VALUE MAX DCR (µH) (Ω) CORE TYPE HEIGHT (mm) Toroid 4.8 Synchronization and Shutdown BH Electronics 511-0033 The S/S pin (Pin 8) can be used to synchronize the oscillator or disconnect the load from the input. The S/S pin is tied to the input (VIN > 1.8V) for normal operation. The oscillator in the LT1306 can be externally synchronized by driving the S/S pin with a pulse train (See the graph “Maximum Allowable Rise Time of Synchronizing Pulse” in the Typical Performance Characteristics). The synchronization is positive edge triggered. The recommended frequency of the external clock ranges from 425kHz to 500kHz. If synchronization results in switching jitter, reducing the rising edge dv/dt of the external clock pulse usually cures the problem. Coilcraft Shutdown will be activated if the S/S pin voltage stays below the shutdown threshold (0.45V) for more than 50µs. This shutdown delay is reset whenever the S/S pin goes above the shutdown threshold. The output filter capacitor is usually chosen based on its equivalent series resistance (ESR) and the acceptable change in output voltage as a result of load transients. The output voltage ripple at the switching frequency can be estimated by considering the peak inductor current and the capacitor ESR. Inductor The value of the energy storage inductor L1 (Figure 1) is usually selected so that the peak-to-peak ripple current is less than 40% of the average inductor current. For 1- or 2-cell alkaline or single Li-Ion to 5V applications, 10µH to 20µH is recommended for the LT1306 running at 300kHz. A 5µH to 10µH inductor can be used if the LT1306 is externally synchronized at 500kHz. 5.0 0.023 DO3308-103 10 0.09 Open 3.0 DO3316-472 4.7 0.018 Open 5.2 DO3316-103 10 0.029 Open 5.2 DO3316-153 15 0.046 Open 5.2 CTX5-2 5 0.021 Toroid 6.0 CTX10-2 10 0.032 Toroid 6.0 Murata LQN6C4R7 4.7 0.034 Open 5.0 Sumida CDRH73-100 10 0.072 Magnetic Shielding 3.4 CD43-4R7 4.7 0.109 Open 3.2 Coiltronics Capacitors IPEAK ≈ IIN ≈ (IO)(VO ) VIN output ripple ≅ (ESR)(IPEAK) = (ESR)(IO )(VO ) VIN 9 LT1306 U W U U APPLICATIONS INFORMATION Since a boost converter produces high output current ripple, one also needs to consider the maximum ripple current rating of the output capacitor. Capacitor reliability will be affected if the ripple current exceeds the maximum allowable ratings. This maximum rating is usually specified as the RMS ripple current. In the LT1306 the RMS output capacitor ripple current is: IO VO – VIN VIN For 2-cell to 5V applications, 220µF low ESR solid tantalum capacitors (AVX TPS series or Sprague 593D series) work well. To reduce output voltage ripple due to heavy load transients or Burst Mode operation, higher capacitance may be used. For through-hole applications, Sanyo OS-CON capacitors are also good choices. In a boost regulator, the input capacitor ripple current is much lower. Maximum ripple current rating and input voltage ripples are not usually of concern. A 22µF tantalum capacitor soldered near the input pin is generally an adequate bypass. Bootstrap Supply Diode D1 and capacitor C1 generate a pulsating supply voltage, VCAP, which is higher than the output. The rectifier drive circuit runs off this supply. During rectifier on-time, the rectifier base current drains C1. Q2 base current and the maximum allowable VCAP ripple voltage determine the size of C1. A 1µF capacitor is sufficient to keep VCAP ripple below 0.3V. For a 2-cell input (VIN > 1.8V) over an extended temperature range, a BAT54 Schottky diode may be used for D1. The use of a Schottky diode increases the bootstrap voltage and the operating headroom for the rectifier driver, X5. Diodes like a 1N4148 or 1N914 work well for 2-cell inputs over the 0°C to 70°C commercial temperature range. The charge drawn from C1 during the rectifier on-time has to be replenished during the switch on-interval. As duty cycle decreases, the amplitude of the C1 charging current can increase dramatically especially when delivering high power to the load. This charging current flows through the 10 switch and can cause the current limit comparator to trip erratically. For boost applications where VIN is a few tenths of a volt below VO, a 1µF or 2.2µF tantalum capacitor (such as AVX TAJ series) can be used for C1. The ESR of the tantalum capacitor limits the charging current. A low value resistor (2Ω to 5Ω) can also be added in series with C1 for further limiting the charging current although this tends to lower the converter efficiency slightly. Frequency Compensation Current mode switching regulators have two feedback loops. The inner current feedback loop controls the inductor current in response to the outer loop. The outer or overall feedback loop tightly regulates the output voltage. The high frequency gain asymptote of the inner current loop rolls off at – 20dB/decade and crosses the unity gain axis at a frequency ωc between 1/6 to 2/3 of the switching frequency. The current loop is stable and is wideband compared to the overall voltage feedback loop. The low frequency current loop gain is not high (usually between unity and 10) but it increases the low frequency impedance of the inductor as seen by the output filter capacitor. (In a boost regulator, the inductor is connected to the output during the switch off-time.) Current mode control introduces an effective series resistance (>> DCR) to the inductor that damps the LC tank response. The complex high-Q poles of the LC filter are now separated, resulting in a dominant pole determined by the filter capacitance and the load resistance and a second high frequency pole. For a boost regulator the control to output transfer function can be shown to have a dominant pole at the load corner frequency ωP = 1 RL (CO ) 2 and a moving right-half plane (RHP) zero with a minimum value of R (1 – DMAX ) ωZ = L L 2 LT1306 U U W U APPLICATIONS INFORMATION where Output Voltage Maximum DC Load Current = Maximum Converter Duty Cycle The low frequency zero 1/R3CZ of the compensation network is placed at ωP/2. RL = Maximum Load = DMAX = VO + 0.1 There is also a second pole at the current loop crossover frequency ωC (Figure 6). ωZ is much lower in frequency than ωC. The loop is compensated by adjusting the midband gain with resistor R3 (Figure 7) so that the overall loop gain crosses 0dB before the minimum frequency RHP zero (i.e., corresponding to the highest duty ratio). The value of R3 can be estimated with the fromula: 390 VO(1 – DMAX )CORL L The amplitude response of the error amplifier with the compensation network shown is: ( ) 1 + S • R3 • CZ R2 VˆC = gm VˆO R1 + R2 S • CZ 1 + S • R3 • CP [ ( CP = 1 3ω ZR3 Higher output filter capacitance rolls off the gain response from a lower corner frequency so higher midband gain is required in the compensation network to make the overall loop gain cross 0dB just below ωZ. Layout Consideration Due to the low transconductance of the error amplifier, the gain setting resistor R3 is AC-coupled with capacitor CZ. This prevents R3 from inducing an offset to the input of the error amplifier. It also creates a pole at DC and a low frequency zero. CZ >> CP 2 R3ωP The capacitor CP ensures adequate gain margin beyond the RHP zero. The high frequency pole 1/R3CP of the amplifier frequency response is placed beyond ωZ. VO – VIN(MIN) + 0.5 R3 = CZ = To minimize EMI and high frequency resonances, it is essential to keep the SW and the CAP trace leads as short as possible. The input and the output bypass capacitors CIN and COUT should be placed close to the IC package and soldered to the ground plane. A ground plane under the switching regulator is highly recommended. Figure 8 shows a suggested component placement and PC board layout. )] 11 LT1306 U U W U APPLICATIONS INFORMATION GAIN (dB) MIDBAND GAIN = (gm)(R3)(R2) R1 + R2 VˆC Vˆ O R (1 – DMAX)2 RHP ZERO = L L AMPLITUDE RESPONSE OF THE ERROR AMPLIFIER AMOUNT OF MIDBAND GAIN NEEDED 0 CURRENT LOOP CROSSOVER FREQUENCY ωC ωZ ωP 1 1 (R3)(CZ) RL (CO) 2 () ω OVERALL LOOP GAIN AFTER COMPENSTION LOOP GAIN CROSSOVER FREQUENCY ≈ 1 ωZ 3 AMPLITUDE RESPONSE OF CONTROL-TO-OUTPUT TRANSFER FUNCTION BEFORE COMPENSATION VˆO Vˆ C 1306 F06 Vˆ O Vˆ C Figure 6. Gain Asymptotes of the Control-to-Output ˆ and Error Amplifier ˆ Transfer Function VC VO SW L VIN Q2 PWM CONTROL LOGIC VO Q1 R1 RECTIFIER IO FB – gm + LT1306 1.24V R2 CO GND VC R3 CP CZ 1306 F07 Figure 7. Current Mode Boost Converter Overall-Loop Compensation 12 RL LT1306 U U W U APPLICATIONS INFORMATION GROUND PLANE CZ R3 S/S VIN CP VC R2 R1 LT1306 8 2 7 3 6 4 5 CIN2 + CO1 + VOUT CO2 D1 + 1 C1 CIN1 VIAS GND L1 1306 F08 Figure 8. Recommended Component Placement for LT1306. Notice That the Input and the Output Capacitors Are Grounded at the Same Point. A Ground Plane Under the DC/DC Converter Is Highly Recommended. Use Multiple Vias to Tie Pin 4 Copper to the Ground Plane 13 LT1306 U TYPICAL APPLICATIO S 2-Cell NiMH to 3.3V Output D1 C1 1µF L1 4.7µH + 2V TO 3V VIN 2V/500kHz SW S/S CIN1 0.1µF CERAMIC + CAP 3.3V 1A OUT R1 412k LT1306 CIN2 22µF FB VC R3 95k CZ 5.6nF + GND R2 249k CP 39pF 1306 F09 Efficiency 90 VO = 3.3V L1 = 4.7µH EFFICIENCY (%) 85 VIN = 3V 80 VIN = 2.5V VIN = 1.8V 75 70 65 60 1 10 100 LOAD CURRENT (mA) 1000 2000 1306 F09a 14 CO 220µF CIN1: AVX TAJC226M010 CO1: AVX TPSE227M010R0100 C1: AVX TAJA105K020 D1: CMDSH-3 L1: LQN6C4R7 LT1306 U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 8 7 6 5 0.150 – 0.157** (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) 1 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0.053 – 0.069 (1.346 – 1.752) 0°– 8° TYP 0.016 – 0.050 (0.406 – 1.270) 0.014 – 0.019 (0.355 – 0.483) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 2 3 4 0.004 – 0.010 (0.101 – 0.254) 0.050 (1.270) BSC Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. SO8 1298 15 LT1306 U TYPICAL APPLICATIO S 4-Cell NiMH to 5V Output Efficiency D1 90 C1 1µF L1 10µH SW S/S + CAP CIN2 0.1µF CERAMIC 5V 1A OUT R1 768k LT1306 CIN1 22µF EFFICIENCY (%) VIN FB VC R3 75k CZ 15nF VIN = 4.8V 85 + 3.6V TO 6.5V VO = 5V L1 = 10µH + GND CP 22pF CO1 220µF R2 249k 1306 F09 CO2 1µF CERAMIC VIN = 3.6V 80 75 VIN = 6V 70 65 CIN1: AVX TAJC226M010 CO1: AVX TPSE227M010R0100 C1: AVX TAJA105K020 D1: MMBD914LT1 L1: CTX10-3 60 1 10 100 LOAD CURRENT (mA) 1000 2000 1306 F10a Transient Response with Step Input (4V to 6V) VIN 5V/DIV VSW 5V/DIV IL 500mA/DIV VO 0.1V/DIV AC 0.5ms/DIV RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LT1302 High Output Current Micropower DC/DC Converter 5V/600mA from 2V, 2A Internal Switch, 200µA IQ LT1304 2-Cell Micropower DC/DC Converter 5V/200mA, Low-Battery Detector Active in Shutdown LT1307/LT1307B Single Cell, Micropower, 600kHz PWM DC/DC Converters 3.3V at 75mA from One Cell, MSOP Package LT1308A/LT1308B High Output Current Micropower DC/DC Converter 5V at 1A from Single Li-Ion Cell LT1316 Burst Mode Operation DC/DC with Programmable Current Limit 1.5V Minimum, Precise Control of Peak Current Limit LT1317/LT1317B Micropower, 600kHz PWM DC/DC Converters 100µA IQ, Operate with VIN as Low as 1.5V LT1610 Single-Cell Micropower DC/DC Converter 3V at 30mA from 1V, 1.7MHz Fixed Frequency LT1613 1.4MHz Switching Regulator in 5-Lead SOT-23 5V at 200mA from 4.4V Input, Tiny SOT-23 package LT1615 Micropower Step-Up DC/DC in 5-Lead SOT-23 20µA IQ, 36V/350mA Internal Switch, VIN as Low as 1.2V LTC1624 High Efficiency N-channel Switching Regulator Controller VOUT = 1.19V to 30V in Stepdown; VIN = 3.5V to 36V SO-8 Package LT1949 600kHz, 1A Switch PWM DC/DC Converter 1.1A, 0.5Ω/30V Internal Switch, VIN as Low as 1.5V 16 Linear Technology Corporation 1306f LT/TP 0400 4K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 1999