Supertex inc.
HV9910B
Universal High Brightness
LED Driver
Features
General Description
►
►
►
►
►
►
►
The HV9910B is an open loop, current mode, control LED
driver IC. The HV9910B can be programmed to operate in
either a constant frequency or constant off-time mode. It
includes an 8.0 - 450V linear regulator which allows it to work
from a wide range of input voltages without the need for an
external low voltage supply. The HV9910B includes a PWM
dimming input that can accept an external control signal with a
duty ratio of 0 - 100% and a frequency of up to a few kilohertz.
It also includes a 0 - 250mV linear dimming input which can
be used for linear dimming of the LED current.
Switch mode controller for single switch LED drivers
Enhanced drop-in replacement to the HV9910
Open loop peak current controller
Internal 8.0 to 450V linear regulator
Constant frequency or constant off-time operation
Linear and PWM dimming capability
Requires few external components for operation
Applications
►
►
►
►
►
►
The HV9910B is ideally suited for buck LED drivers. Since
the HV9910B operates in open loop current mode control, the
controller achieves good output current regulation without the
need for any loop compensation. PWM dimming response is
limited only by the rate of rise and fall of the inductor current,
enabling very fast rise and fall times. The HV9910B requires
only three external components (apart from the power stage)
to produce a controlled LED current making it an ideal solution
for low cost LED drivers.
DC/DC or AC/DC LED driver applications
RGB backlighting LED driver
Back lighting of flat panel displays
General purpose constant current source
Signage and decorative LED lighting
Chargers
Typical Application Circuit
CIN
CO
D1
CDD
VDD
L1
VIN
HV9910B
LD
PWMD
RT
RT
Supertex inc.
GATE
Q1
CS
GND
RCS
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
HV9910B
Ordering Information
Pin Description
Package Options
Device
HV9910B
8-Lead SOIC
16-Lead SOIC
HV9910BLG-G
HV9910BNG-G
-G indicates package is RoHS compliant (‘Green’)
VIN 1
16
NC
NC 2
15
NC
NC 3
14
RT
CS 4
13
LD
VDD
VIN 1
8
RT
GND 5
12
CS 2
7
LD
NC 6
11
NC
GND 3
6
VDD
NC 7
10
NC
GATE 4
5
PWMD
GATE 8
9
PWMD
8-Lead SOIC (LG)
Absolute Maximum Ratings
Product Marking
Parameter
Value
VIN to GND
-0.5V to +470V
VDD to GND
12V
CS, LD, PWMD, GATE, RT to GND
-40°C to +150°C
Storage temperature range
-65°C to +150°C
Continuous power dissipation (TA = +25°C)
8-Lead SOIC
16-Lead SOIC
Y = Last Digit of Year Sealed
WW = Week Sealed
L = Lot Number
= “Green” Packaging
YWW
9910B
LLLL
-0.3V to (VDD +0.3V)
Junction temperature range
Package may or may not include the following marks: Si or
8-Lead SOIC (LG)
Top Marking
HV9910BNG
630mW
1300mW
YWW
Stresses beyond those listed under “Absolute Maximum Ratings” may cause
permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in
the operational sections of the specifications is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect device reliability.
LLLLLLLL
Bottom Marking
CCCCCCCCC AAA
Y = Last Digit of Year Sealed
WW = Week Sealed
L = Lot Number
C = Country of Origin*
A = Assembler ID*
= “Green” Packaging
*May be part of top marking
Package may or may not include the following marks: Si or
Thermal Resistance
16-Lead SOIC (NG)
Package
θja
8-Lead SOIC
128OC/W
16-Lead SOIC
82OC/W
Electrical Characteristics (The specifications are at T = 25°C and V
A
Sym
16-Lead SOIC (NG)
Description
IN
= 12V, unless otherwise noted.)
Min
Typ
Max
Units
Conditions
Input
VINDC
Input DC supply voltage range1
*
8.0
-
450
V
IINSD
Shut-down mode supply current
*
-
0.5
1.0
mA
DC input voltage
Pin PWMD to GND
Notes:
1. Also limited by package power dissipation limit, whichever is lower.
† VDD load current external to the HV9910B.
*
Denotes the specifications which apply over the full operating ambient temperature range of -40°C < TA < +125°C.
# Guaranteed by design.
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
2
HV9910B
Electrical Characteristics (cont.) (The specifications are at T = 25°C and V
A
Sym
Description
IN
= 12V, unless otherwise noted.)
Min
Typ
Max
Units
Conditions
Internal Regulator
Internally regulated voltage
-
7.25
7.5
7.75
V
VIN = 8.0V, IDD(ext)(†) = 0, 500pF at
GATE; RT = 226kΩ, PWMD = VDD
Line regulation of VDD
-
0
-
1.0
V
VIN = 8.0 - 450V, IDD(ext) = 0, 500pF at
GATE; RT = 226kΩ, PWMD = VDD
Load regulation of VDD
-
0
-
100
mV
IDD(ext) = 0 - 1.0mA, 500pF at GATE;
RT = 226kΩ, PWMD = VDD
UVLO
VDD undervoltage lockout threshold
*
6.45
6.7
6.95
V
VDD rising
∆UVLO
VDD undervoltage lockout hysteresis
-
-
500
-
mV
VDD falling
Current that the regulator can
supply before IC goes into UVLO
#
5.0
-
-
mA
VIN = 8.0V
VDD
ΔVDD, line
Internal Regulator
ΔVDD, load
IIN,MAX
PWM Dimming
VEN(lo)
Pin PWMD input low voltage
*
-
-
0.8
V
VIN = 8.0 - 450V
VEN(hi)
Pin PWMD input high voltage
*
2.0
-
-
V
VIN = 8.0 - 450V
Pin PWMD pull-down resistance
at PWMD
-
50
100
150
kΩ
225
250
275
213
250
287
REN
VPWMD = 5.0V
Current Sense Comparator
VCS,TH
Current sense pull-in threshold
voltage
-
VOFFSET
Offset voltage for LD comparator
*
-12
-
12
-
150
215
280
TBLANK
Current sense blanking interval
mV
mV
ns
-40°C < TA < +85°C
TA < +125°C
--0 < TA < +85OC, VLD = VDD,
VCS = VCS,TH + 50mV after TBLANK
-40 < TA < +125OC, VLD = VDD,
VCS = VCS,TH + 50mV after TBLANK
-
145
215
315
-
-
80
150
-
20
25
30
-
80
100
120
GATE sourcing current
-
165
-
-
mA
VGATE = 0V, VDD = 7.5V
ISINK
GATE sinking current
-
165
-
-
mA
VGATE = VDD, VDD = 7.5V
tRISE
GATE output rise time
-
-
30
50
ns
CGATE = 500pF, VDD = 7.5V
tFALL
GATE output fall time
-
-
30
50
ns
CGATE = 500pF, VDD = 7.5V
tDELAY
Delay to output
ns
VLD = VDD,
VCS = VCS,TH + 50mV after TBLANK
Oscillator
fOSC
Oscillator frequency
kHz
RT = 1.00MΩ
RT = 226kΩ
GATE Driver
ISOURCE
Notes:
1. Also limited by package power dissipation limit, whichever is lower.
† VDD load current external to the HV9910B.
*
Denotes the specifications which apply over the full operating ambient temperature range of -40°C < TA < +125°C.
# Guaranteed by design.
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
3
HV9910B
Application Information
The HV9910B is optimized to drive buck LED drivers using
open-loop peak current mode control. This method of control
enables fairly accurate LED current control without the need
for high side current sensing or the design of any closed loop
controllers. The IC uses very few external components and
enables both Linear and PWM dimming of the LED current.
A resistor connected to the RT pin programs the frequency
of operation (or the off-time). The oscillator produces pulses
at regular intervals. These pulses set the SR flip-flop in the
HV9910B which causes the GATE driver to turn on. The same
pulses also start the blanking timer which inhibits the reset
input of the SR flip flop and prevent false turn-offs due to the
turn-on spike. When the FET turns on, the current through
the inductor starts ramping up. This current flows through
the external sense resistor RCS and produces a ramp voltage
at the CS pin. The comparators are constantly comparing
the CS pin voltage to both the voltage at the LD pin and
the internal 250mV. Once the blanking timer is complete, the
output of these comparators is allowed to reset the flip flop.
When the output of either one of the two comparators goes
high, the flip flop is reset and the GATE output goes low. The
GATE goes low until the SR flip flop is set by the oscillator.
Assuming a 30% ripple in the inductor, the current sense
resistor RCS can be set using:
RCS =
and any external resistor dividers needed to control the IC.
The VDD pin must be bypassed by a low ESR capacitor to
provide a low impedance path for the high frequency current
of the output GATE driver.
The HV9910B can also be operated by supplying a voltage
at the VDD pin greater than the internally regulated voltage.
This will turn off the internal linear regulator of the IC and the
HV9910B will operate directly off the voltage supplied at the
VDD pin. Please note that this external voltage at the VDD
pin should not exceed 12V.
Although the VIN pin of the HV9910B is rated up to 450V,
the actual maximum voltage that can be applied is limited
by the power dissipation in the IC. For example, if an 8-pin
SOIC (junction to ambient thermal resistance Rθ,j-a = 128°C/
W) HV9910B draws about IIN = 2.0mA from the VIN pin, and
has a maximum allowable temperature rise of the junction
temperature limited to about ΔT = 100°C, the maximum voltage at the VIN pin would be:
0.25V (or VLD)
1.15 • ILED (A)
Constant frequency peak current mode control has an inherent disadvantage – at duty cycles greater than 0.5, the
control scheme goes into subharmonic oscillations. To prevent this, an artificial slope is typically added to the current
sense waveform. This slope compensation scheme will affect the accuracy of the LED current in the present form.
However, a constant off-time peak current control scheme
does not have this problem and can easily operate at duty
cycles greater then 0.5 and also gives inherent input voltage rejection making the LED current almost insensitive to
input voltage variations. But, it leads to variable frequency
operation and the frequency range depends greatly on the
input and output voltage variation. HV9910B makes it easy
to switch between the two modes of operation by changing
one connection (see oscillator section).
In these cases, to operate the HV9910B from higher input
voltages, a Zener diode can be added in series with the VIN
pin to divert some of the power loss from the HV9910B to
the Zener diode. In the above example, using a 100V zener
diode will allow the circuit to easily work up to 450V.
The input current drawn from the VIN pin is a sum of the
1.0mA current drawn by the internal circuit and the current
drawn by the GATE driver (which in turn depends on the
switching frequency and the GATE charge of the external
FET).
IIN ≈ 1.0mA + QG • fS
In the above equation, fS is the switching frequency and QG
is the GATE charge of the external FET (which can be obtained from the datasheet of the FET).
Input Voltage Regulator
The HV9910B can be powered directly from its VIN pin and
can work from 8.0 - 450VDC at its VIN pin. When a voltage
is applied at the VIN pin, the HV9910B maintains a constant
7.5V at the VDD pin. This voltage is used to power the IC
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
4
HV9910B
Current Sense
The current sense input of the HV9910B goes to the noninverting inputs of two comparators. The inverting terminal
of one comparator is tied to an internal 250mV reference
whereas the inverting terminal of the other comparator is
connected to the LD pin. The outputs of both these comparators are fed into an OR GATE and the output of the OR
GATE is fed into the reset pin of the flip-flop. Thus, the comparator which has the lowest voltage at the inverting terminal
determines when the GATE output is turned off.
The outputs of the comparators also include a 150-280ns
blanking time which prevents spurious turn-offs of the external FET due to the turn-on spike normally present in peak
current mode control. In rare cases, this internal blanking
might not be enough to filter out the turn-on spike. In these
cases, an external RC filter needs to be added between the
external sense resistor (RCS) and the CS pin.
Please note that the comparators are fast (with a typical
80ns response time). Hence these comparators are more
susceptible to be triggered by noise than the comparators
of the HV9910. A proper layout minimizing external inductances will prevent false triggering of these comparators.
Oscillator
The oscillator in the HV9910B is controlled by a single resistor connected at the RT pin. The equation governing the
oscillator time period tOSC is given by:
tOSC(μs) =
RT(kΩ) + 22
25
The GATE output of the HV9910B is used to drive an external FET. It is recommended that the GATE charge of the
external FET be less than 25nC for switching frequencies
≤100kHz and less than 15nC for switching frequencies >
100kHz.
Supertex inc.
The Linear Dimming pin is used to control the LED current.
There are two cases when it may be necessary to use the
Linear Dimming pin.
► In some cases, it may not be possible to find the exact
RCS value required to obtain the LED current when the
internal 250mV is used. In these cases, an external voltage divider from the VDD pin can be connected to the LD
pin to obtain a voltage (less than 250mV) corresponding to
the desired voltage across RCS.
► Linear dimming may be desired to adjust the current
level to reduce the intensity of the LEDs. In these cases,
an external 0-250mV voltage can be connected to the LD
pin to adjust the LED current during operation.
To use the internal 250mV, the LD pin can be connected to
VDD.
Note:
Although the LD pin can be pulled to GND, the output current will not go to zero. This is due to the presence of a minimum on-time (which is equal to the sum of the blanking time
and the delay to output time) which is about 450ns. This will
cause the FET to be on for a minimum of 450ns and thus the
LED current when LD = GND will not be zero. This current is
also dependent on the input voltage, inductance value, forward voltage of the LEDs and circuit parasitics. To get zero
LED current, the PWMD pin has to be used.
PWM Dimming
If the resistor is connected between RT and GND, HV9910B
operates in a constant frequency mode and the above equation determines the time-period. If the resistor is connected
between RT and GATE, the HV9910B operates in a constant
off-time mode and the above equation determines the offtime.
GATE Output
Linear Dimming
PWM Dimming can be achieved by driving the PWMD pin
with a low frequency square wave signal. When the PWM
signal is zero, the GATE driver is turned off and when the
PWMD signal if high, the GATE driver is enabled. Since the
PWMD signal does not turn off the other parts of the IC,
the response of the HV9910B to the PWMD signal is almost
instantaneous. The rate of rise and fall of the LED current is
thus determined solely by the rise and fall times of the inductor current.
To disable PWM dimming and enable the HV9910B permanently, connect the PWMD pin to VDD.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
5
HV9910B
Block Diagram
VIN
Regulator
VDD
POR
LD
+
Blanking
CS
250mV
+
-
R Q
GATE
S
Oscillator
RT
GND
PWMD
Pin Description
Pin #
Function
Description
8-Lead SOIC
16-Lead SOIC
1
1
VIN
This pin is the input of an 8.0 - 450V linear regulator.
2
4
CS
This pin is the current sense pin used to sense the FET current by means
of an external sense resistor. When this pin exceeds the lower of either the
internal 250mV or the voltage at the LD pin, the GATE output goes low.
3
5
GND
Ground return for all internal circuitry. This pin must be electrically connected to the ground of the power train.
4
8
GATE
This pin is the output GATE driver for an external N-channel power
MOSFET.
5
9
PWMD
This is the PWM dimming input of the IC. When this pin is pulled to GND,
the GATE driver is turned off. When the pin is pulled high, the GATE driver
operates normally.
6
12
VDD
7
13
LD
This pin is the linear dimming input and sets the current sense threshold as
long as the voltage at the pin is less than 250mV (typ).
This is the power supply pin for all internal circuits.
It must be bypassed with a low ESR capacitor to GND (≥0.1μF).
8
14
RT
This pin sets the oscillator frequency. When a resistor is connected between RT and GND, the HV9910B operates in constant frequency mode.
When the resistor is connected between RT and GATE, the IC operates in
constant off-time mode.
-
2, 3, 6, 7, 10,
11, 15, 16
NC
No connection
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
6
HV9910B
8-Lead SOIC (Narrow Body) Package Outline (LG)
4.90x3.90mm body, 1.75mm height (max), 1.27mm pitch
D
θ1
8
E
E1
L2
Note 1
(Index Area
D/2 x E1/2)
L
1
θ
L1
Top View
Gauge
Plane
Seating
Plane
View B
A
View B
Note 1
h
h
A A2
Seating
Plane
b
e
A1
A
Side View
View A-A
Note:
1. This chamfer feature is optional. A Pin 1 identifier must be located in the index area indicated. The Pin 1 identifier can be: a molded mark/identifier;
an embedded metal marker; or a printed indicator.
Symbol
Dimension
(mm)
A
A1
A2
b
MIN
1.35*
0.10
1.25
0.31
NOM
-
-
-
-
MAX
1.75
0.25
1.65*
0.51
D
E
E1
4.80* 5.80* 3.80*
4.90
6.00
3.90
5.00* 6.20* 4.00*
e
1.27
BSC
h
L
0.25
0.40
-
-
0.50
1.27
L1
1.04
REF
L2
0.25
BSC
JEDEC Registration MS-012, Variation AA, Issue E, Sept. 2005.
* This dimension is not specified in the JEDEC drawing.
Drawings are not to scale.
Supertex Doc. #: DSPD-8SOLGTG, Version I041309.
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
7
θ
θ1
0O
5O
-
-
8O
15O
HV9910B
16-Lead SOIC (Narrow Body) Package Outline (NG)
9.90x3.90mm body, 1.75mm height (max), 1.27mm pitch
D
16
θ1
E1 E
Note 1
(Index Area
D/2 x E1/2)
L2
1
L
Top View
View B
A
A A2
e
A1
View
B
h
h
Seating
Plane
Seating
Plane
θ
L1
Gauge
Plane
Note 1
b
Side View
View A-A
A
Note:
1. This chamfer feature is optional. If it is not present, then a Pin 1 identifier must be located in the index area indicated. The Pin 1 identifier can be:
a molded mark/identifier; an embedded metal marker; or a printed indicator.
Symbol
Dimension
(mm)
A
A1
A2
b
D
E
E1
MIN
1.35*
0.10
1.25
0.31
9.80*
5.80* 3.80*
NOM
-
-
-
-
9.90
6.00
MAX
1.75
0.25
1.65*
0.51
3.90
10.00* 6.20* 4.00*
e
1.27
BSC
h
L
0.25
0.40
-
-
0.50
1.27
L1
L2
1.04 0.25
REF BSC
θ
θ1
0O
5O
-
-
8O
15O
JEDEC Registration MS-012, Variation AC, Issue E, Sept. 2005.
* This dimension is not specified in the JEDEC drawing.
Drawings are not to scale.
Supertex Doc. #: DSPD-16SONG, Version G041309.
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline
information go to http://www.supertex.com/packaging.html.)
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
Supertex inc.
©2010 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
Doc.# DSFP-HV9910B
B061209
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
8
Supertex inc.
HV9910BDB1
Off-Line, High Brightness, 1.4A
LED Driver Demo Board
General Description
Specifications
The HV9910BDB1 demoboard is an offline, high current
LED driver designed to drive a 40V LED string at 1.4A from
a 110V input. The demoboard uses Supertex’s HV9910B
LED driver IC to drive a buck converter.
Input voltage
90 - 135Vrms, 50/60Hz
Output voltage
10 - 40V
Output current
1.4A max
Output current ripple (typ)
The HV9910BDB1 has a typical full load efficiency of 88%,
with the buck converter efficiency (excluding the diode
bridge rectifier and EMI filter) at 93%. The demoboard also
meets CISPR-15 conducted EMI standards.
The output current can be adjusted in two ways - either
with linear dimming using the onboard potentiometer or
with PWM dimming by applying a TTL-compatible square
wave signal at the PWMD terminal. Using linear dimming,
the output current of the HV9910DB1 can be lowered to
about 0.1A (note: zero output current can be obtained only
by PWM dimming).
40% (peak-peak)
Full load efficiency (@110V)
88%
Power factor (@110V)
0.64
Input current (@110V)
0.83A rms
Input current THD (@110V)
117%
Switching frequency (typ)
50kHz
0.1A
Minimum output current (@110V)
Conducted EMI
Meets CISPR-15
Temperature rise of heatsink
(@110V input and full load)
50°C
Open LED protection
yes
Output short circuit protection
Dimensions
Connection Diagram
WARNING!!!
no
86.4mm X 58.4mm
10 - 40V, 1.4A max.
90 - 135VAC
50/60Hz
Do not connect earth-grounded test instruments. Doing so will short the AC line, resulting in damage to the
instrument and/or the HV9910BDB1. Use floating high voltage differential probes or isolate the demoboard
by using an isolating transformer.
There is no galvanic isolation. Dangerous voltages are present when connected to the AC line.
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
1
HV9910BDB1
Connections
1. Connect the input AC voltage between the AC IN terminals as shown in the connection diagram.
2. Connect the LED string between LED+ (anode of LED string) and LED- (cathode of LED string).
3. Connect the PWMD terminal to the VDD terminal using the jumper provided to enable the LED driver.
4. The current level can be adjusted using the on-board potentiometer.
PWM Dimming
The HV9910BDB1 is capable of being PWM dimmed by applying a square wave TTL compatible signal between PWMD
and GND terminals. However, since there is no galvanic isolation on the board, care must be taken to prevent damage to
the PWM dimming source and/or the HV9910BDB1. One simple way is to isolate the LED driver from the AC line using an
isolation transformer. Another approach is to use an opto-isolator to drive the PWMD pin as shown in the figure below.
3.8kΩ
VDD
5.0V
square wave signal
(<1.0kHz)
Opto-isolator
(eg: LTV-814 from Lite-On)
PWMD
Typical Results
Full Load Efficiency vs. Input Voltage
89
Efficiency vs. Load Voltage (@VIN=110V rms)
90
Efficiency (%)
Efficiency (%)
86
88
82
87
78
86
90
100
110
120
130
140
74
5
15
Change in output current (%)
Change in output current (%)
15
2.5
0
-2.5
-5
90
100
110
120
Input Voltage (Vrms)
Supertex inc.
35
45
Load Regulation of Load Current
(@ VIN = 110V rms)
Line Regulation of Load Current
5
25
Load Voltage (V)
Input Voltage (Vrms)
130
140
12.5
10
7.5
5
2.5
0
5
15
25
35
45
Load Voltage (V)
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
2
HV9910BDB1
Waveforms
Drain Voltage
Input Voltage
Input
Current
LED Current
Steady State waveforms at 150Vdc input and full
load output
PWMD Voltage
LED Current
PWM Dimming at 100Hz
Steady State waveforms at 150Vdc input and full
load output
PWMD Voltage
LED Current
Rising Edge of LED Current during PWM Dimming
PWMD Voltage
LED Current
Falling Edge of LED Current during PWM Dimming
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
3
HV9910BDB1
Conducted EMI Measurements at Full Load and 110V AC input
CISPR-15 Limit
Silk Screen
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
4
1
J1
2
1
MOV1
R
ERZ-V05D221
F1
2A
125V
4
C4
0.1µF
250V
3
2
L2
4
1
C5
0.1µF
250V
1
4
+
Supertex inc.
2-
t
3
D2
DF1504S-T
RT1
CL-130
C3
68µF
200V
C7
68µF
200V
R3
5k
C6
2.2µF
16V
0
C1
1.0µF
250V
R6
1k
L3
1
J3A
22µH
R2
178k
1
VIN
1
RT
J3C
J3B
0
5
CS
GATE
PWMD
GND
LD
HV9910B
VDD
U1
3
R5
1k
9
13
12
0
C10
1.0µF
250V
2
C8
0.1µF
16V
2
4
8
14
R7
0.27
0.25W
C1
1.0µF
250V
R4
0.0
R1
464k
2A
1000µH
L1
R7
0.27
0.25W
Q1
FQD8N25
1
D1
MURS240
2
C2
0.47µF
250V
1
2
J2
HV9910BDB1
Schematic Diagram
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
5
HV9910BDB1
Bill of Materials
Package
Manufacturer
Manufacturer’s Part
Number
1µF, 250V metallized polyester capacitor
Radial
EPCOS Inc
B32522C3105J
0.47µF, 250V metallized polyester capacitor
Radial
EPCOS Inc
B32521C3474J
C3, C7
68µF, 250V electrolytic capacitor
Radial
Panasonic
EEU-EE2D680
2
C4, C5
0.1µF, 250V metallized polyester X2
capacitor
Radial
Panasonic
ECQ-U2A104MV
5
1
C6
2.2µF, 16V, X7R ceramic chip capacitor
SMD0805
---
---
6
1
C8
0.1µF, 50V, X7R ceramic chip capacitor
SMD0805
---
---
7
1
C9
open
---
---
---
8
1
D1
400V, 2A ultra fast recovery diode
SMB
ON Semi
MURS240T3
9
1
D2
400V, 1.5A single phase diode bridge
DF-S
Diodes Inc
DF1504S-T
10
1
F1
2A, 125V slow blow fuse
SMT
Littelfuse Inc
0452002.MRL
11
2
J1, J2
2 position 0.156” header
Thru-Hole
Molex
26-48-1021
12
1
J3
3 position, 0.1” pitch vertical header
Thru-Hole
Molex
22-28-4030
13
1
L1
1000µH, 2A rms, 2A sat inductor
Radial
Coilcraft
PCV-2-105-02L
-
-
---
Cross Reference
Radial
Coiltronics
CTX01-17784G-R
14
1
L2
0.6mH, 1A rms common mode choke
Thru-Hole
Coilcraft
BU9-6011R0BL
15
1
L3
22µH, 2.1A sat, 1.9A rms inductor
Radial
Coilcraft
RFB0807-220L
16
1
MOV1
220V, 600A surge absorber
Radial
Panasonic
ERZ-V05D221
17
1
Q1
250V, 0.55Ω, N-channel FET
DPAK
Fairchild Semi
FQD8N25
18
1
RT1
2A rms, 50Ω inrush current limiter
Radial
GE Sensing
CL-130
19
1
R1
464KΩ, 1/8W, 1% chip resistor
SMD0805
---
---
20
1
R2
178KΩ, 1/8W, 1% chip resistor
SMD0805
---
---
21
1
R3
5KΩ 6mmsq single turn potentiometer
SMT
Bourns Inc
3361P-1-502GLF
22
1
R4
0.0Ω, 1/8W chip resistor
SMD0805
---
---
23
2
R5, R6
1KΩ, 1/8W, 1% chip resistor
SMD0805
---
---
24
2
R7, R8
0.27Ω, 1/4W, 1% chip resistor
SMD1206
---
---
25
1
U1
SO-16
Supertex
HV9910BNG-G
Item
#
Qty
RefDes
Description
1
2
C1, C10
2
1
C2
3
2
4
Universal LED Driver
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
Supertex inc.
©2010 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
100410
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
6
HV9910BDB2
Universal, Off-Line, High Brightness,
350mA LED Driver Demo Board
General Description
Specifications
The Supertex HV9910BDB2 demo board is a HighBrightness LED power driver to supply a string of LEDs
using the HV9910B IC from a universal AC input voltage.
The HV9910BDB2 can supply a maximum output current
of 350mA to drive 10 - 40V LED strings from a wide input
voltage - 90 to 265VAC, 50/60Hz.
The power conversion stage of the HV9910BDB2 consists
of a diode bridge rectifier followed by a current-controlled
buck converter operating at a switching frequency of
50kHz. The nominal output current of the demo board can
be adjusted to any value between 30 and 350mA using
the on-board trimming potentiometer. PWM dimming can
be achieved by applying a pulse-width-modulated square
wave signal between the PWMD and GND pins. Zero
output current can be obtained only by PWM dimming.
The HV9910BDB2 is not CISPR-15 compliant. Additional
filtering is required to make the board meet CISPR-15
limits.
Parameter
Value
Input voltage
90 - 265Vrms, 50/60Hz
Output voltage
10 - 40V
Output current
350mA max
Output current ripple (typ)
@110V input
25% (peak-peak)
@40V output, 350mA
88% @110VAC
Full load efficiency
86% @230VAC
0.70 @110VAC
Power factor
0.48 @230VAC
0.20A @110VAC
Input current (rms)
0.14A @230VAC
94% @110VAC
Input current THD
95% @230VAC
Switching frequency (typ)
50kHz
20mA @110VAC
Minimum output current
30mA @230VAC
Open LED protection
yes
Output short circuit protection
Dimensions
no
68.6mm X 49.6mm
Connection Diagram
10 - 40V,
350mA max.
WARNING!!!
90 - 265VAC
50/60Hz
Do not connect earth-grounded test instruments. Doing so will short the AC line, resulting in damage to the instrument and/or
the HV9910BDB2. Use floating high voltage differential probes or isolate the demoboard by using an isolating transformer.
There is no galvanic isolation. Dangerous voltages are present when connected to the AC line.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
1
HV9910BDB2
Connections
PWM Dimming
1. Connect the input AC voltage between the AC IN
terminals as shown in the connection diagram.
2. Connect the LED string between LED+ (anode of LED
string) and LED- (cathode of LED string).
3. Connect the PWMD terminal to the VDD terminal using
the jumper provided to enable the LED driver.
4. The current level can be adjusted using the on-board
potentiometer.
The HV9910BDB2 is capable of being PWM dimmed by
applying a square wave TTL compatible signal between
PWMD and GND terminals. However, since there is no
galvanic isolation on the board, care must be taken to
prevent damage to the PWM dimming source and/or the
HV9910BDB2. One simple way is to isolate the LED driver
from the AC line using an isolation transformer. Another
approach is to use an opto-isolator to drive the PWMD pin
as shown in the figure below.
3.8kΩ
VDD
5.0V
square wave signal
(<1.0kHz)
Opto-isolator
(eg: LTV-814 from Lite-On)
PWMD
Typical Results
Efficiency vs.Input Voltage
90
40V output
80
75
10V output
70
65
60
110VAC Input
85
Efficiency (%)
Efficiency (%)
85
Efficiency vs. LED String Voltage
90
80
230VAC Input
75
70
65
80
130
180
230
Input Voltage (VAC)
280
60
0
10
Current Regulation vs. Input Voltage
5
10V Output
0
40V Ouput
-5
-10
80
130
180
230
Input Voltage (VAC)
280
40
50
Current Regulation vs. LED String
Voltage
20
% change in LED Current
% change in LED Current
10
20
30
LED String Voltage (V)
15
230VAC Input
10
110VAC Input
5
0
0
10
20
30
LED String Voltage (V)
40
50
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
2
HV9910BDB2
Waveforms
LED Current
LED Current
Drain Voltage
Steady State waveforms at 110VAC input and full
load output
Drain Voltage
Steady State waveforms at 230VAC input and full
load output
LED Current
LED Current
Drain Voltage
Steady State waveforms at 90VAC input and
40V, 350mA output
Drain Voltage
Steady State waveforms at 230VAC input and
10V, 350mA output
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
3
HV9910BDB2
Silk Screen
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
4
2
1
J1
C4
2A, 250VAC
F1
AC2
AC1
U1
NEG
POS
C1
C8
C3
0.47uF, 400V
2.2uF, 16V
R7
1K
5K
R3
J3A
C6
R2
178K
1
0.1uF, 305VAC
t
CL-140
0.1uF
J3B
R5
1k
9
13
12
EN
HD
Rosc
J3C
CS
Gate
HV9910
Vdd
U2
3
1
Vin
Gnd
5
RT1
4
8
14
R4
1k
100pF
C7
464K
R1
R6
0.56
STD7NM50N
Q1
L1
2
4.7mH
RL-1292-4700
1
D1
STTH2R06U
C5
1
2
J2
HV9910BDB2
Schematic Diagram
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
5
0.47uF, 400V
2
22uF, 400V
HV9910BDB2
Bill of Materials
Package
Manufacturer
Manufacturer’s
Part Number
22µF, 400V electrolytic capacitor
Radial
Nichion
UVR2G220MHD
0.47µF, 400V metal film capacitor
Radial
EPCOS Inc
B32522C6474K
C4
0.1µF, 305VAC EMI suppresion
capacitor
Radial
EPCOS Inc
B32922C3104M
1
C6
0.1µF, 16V X7R ceramic chip capacitor
SMD0805
Panasonic
ECJ-2VB1C104K
5
1
C7
100pF, 50V C0G ceramic chip
capacitor
SMD0805
TDK Corp
C2012C0G1H101J
6
1
C8
2.2µF, 16V X7R ceramic chip capacitor
SMD0805
TDK Corp
C2012X7R1C225K
7
1
D1
600V, 2A ultrafast diode
SMB
ST Micro
STTH2R06U
8
1
F1
2A, 250VAC time lag fuse
Radial
Cooper
Bussman
SR-5-2A-BK
9
1
H1
15C/W DPAK heatsink
SMT
Aavid
7106PD
10
2
J1,J2
2 position, 0.156” pitch, vertical header
Thru-Hole
Molex
26-48-1021
11
1
J3
3 position, 0.100” pitch, vertical header
Thru-Hole
Molex
22-03-2031
12
1
L1
4.7mH, 400mA rms,
470mA sat inductor
Axial
Renco USA
13
1
Q1
550V, 0.7Ω N-channel FET
DPAK
ST Micro
14
1
RT1
50Ω NTC inrush limiter
Thru-Hole
GE Sensing
15
1
R1
464KΩ, 1/8W, 1% chip resistor
SMD0805
---
---
16
1
R2
178KΩ, 1/8W, 1% chip resistor
SMD0805
---
---
17
1
R3
5KΩ top adjust trimpot
SMT
Bourns Inc
18
3
SMD0805
---
---
19
1
R6
0.56Ω, 1/4W, 1% chip resistor
SMD1206
---
---
20
1
U1
400V, 1A single phase diode bridge
DF-S
Diodes Inc
21
1
U2
Universal LED Driver
SO-16
Supertex
Item
#
Qty
RefDes
1
1
C1
2
2
C3,C5
3
1
4
Description
R4,R5,R7 1KΩ, 1/8W, 1% chip resistor
RL-1292-4700
STD5NM50
CL-140
3361P-1-502G
DF04S
HV9910BNG-G
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives an
adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability to the
replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and specifications
are subject to change without notice. For the latest product specifications refer to the Supertex inc. website: http//www.supertex.com.
©2009
012309
All rights reserved. Unauthorized use or reproduction is prohibited.
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
6
HV9910BDB3
Low Voltage, High Current, LED Driver Demoboard
General Description
Specifications
The HV9910BDB3 demoboard is a high current LED driver
designed to drive one LED or two LEDs in series at currents
up to 1.0A from a 10 – 30V DC input. The demoboard uses
Supertex’s HV9910B Universal LED driver IC to drive a
buck converter.
Parameter
Value
Input voltage
The HV9910BDB3 can be configured to operate in either a
constant frequency mode (for driving a single LED) or in a
constant off-time mode (for driving two LEDs).
10 - 30VDC
Output voltage constant frequency mode
2.0 - 4.5V
Output voltage constant off-time mode
4.0 - 8.0V
Maximum output current
Output current ripple (typ)
The output current can be adjusted in two ways – either
with linear dimming using the onboard potentiometer or
with PWM dimming by applying a TTL – compatible square
wave signal at the PWMD terminal. Using linear dimming,
the output current of the HV9910DB1 can be lowered to
about 0.01A (note: zero output current can be obtained
only by PWM dimming).
Efficiency (@ 12V input)
1.0A ± 10%
20% (peak-peak)
86% (for one LED)
93% (for two LEDs)
Open LED protection
yes
Output short circuit protection
Dimensions
no
48.2mm X 29.0mm
Connection Diagram
+
-
+
Short for
constant frequency mode
Short for
constant off-time mode
Connections
1. Input Connection: Connect the input DC voltage between VIN and GND terminals of connector J1 as shown in the
connection diagram.
2. Output Connection: Connect the LEDs between LED+ (anode of LED string) and LED- (cathode of LED string) of
connector J2.
a. If the load is one LED, short the RT and FREQ terminals of connector J4 using a jumper.
b. If the load is two LEDs, short the RT and OFFT terminals of connector J4 using a jumper.
3. PWM Dimming Connection:
a. If no PWM dimming is required, short PWMD and VDD terminals of connector J3.
b. If PWM dimming is required, connect the TTL-compatible PWM source between PWMD and GND terminals of
connector J3. The recommended PWM dimming frequency is ≤ 1.0kHz.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
1
HV9910BDB3
Frequently Asked Questions
1. Why does the demoboard have two operating modes?
Constant frequency mode limits the maximum output voltage to less then 50% of the minimum input voltage. So, in
this case, if we use only the constant frequency mode, the maximum output voltage will have to be less than 5V. Constant off-time mode removes this limitation and allows the output voltage become higher. However, in order to achieve
reasonable noise immunity and to limit the switching frequency variation over the input voltage range, it is not recommended to operate the HV9910DB3 with the output voltage exceeding 80% of the input voltage, even in the constant
off-time mode. Please refer to application note AN-H50 on the Supertex website for more details.
2. If the minimum input voltage in my application is higher (say 20V), does that mean I can drive a 9V LED string
in the constant frequency mode or an 16V LED string in the constant off-time mode using the
demoboard?
Although a larger LED string can be driven using the demoboard in these conditions, the demoboard will not be able to
drive the LED at 1A.
The HV9910B is a constant peak current controller. The average LED current is equal to the peak current set (using the
sense resistor) minus one-half of the ripple current in the inductor.
Higher output voltages lead to larger ripple current values, which will reduce the maximum LED current the board can
deliver.
3. How can I compute the maximum LED current the demoboard can deliver if I use a higher input voltage and a
higher LED string voltage?
Parameters
Minimum input voltage
Maximum LED string voltage
Switching frequency (constant frequency mode)
Off-Time (constant off-time mode)
HV9910B CS threshold voltage
Sense Resistor
Inductor
Constant Off-Time Mode
Δl =
= VIN,MIN
= VO,MAX
= fS
(100kHz)
= TOFF
(5.1μs)
= VCS
(0.25V)
= RCS
(0.22Ω)
=L
(220μH)
Constant Frequency Mode
VO,MAX • TOFF
ILED =
L
VCS
RCS
–
VO,MAX •
Δl
2
Maximum Switching Frequency =
Δl =
{
1–
VO, MAX
VIN,MIN
{
{
1–
VO, MAX
VIN,MIN
{
L • fS
ILED =
VCS
RCS
–
Δl
2
TOFF
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
2
HV9910BDB3
Frequently Asked Questions (cont.)
4. If the constant off-time mode allows a wider LED voltage range, why not use that mode exclusively? Why do
we need the constant frequency mode?
Although the constant off-time mode allows the demoboard to operate at a higher output voltage, the LED ripple current
is directly proportional to the output voltage in this mode. This makes it difficult to get a good load regulation of the LED
current in the constant off-time mode with a wide variation in the LED string voltage (in this case it will be a 1:4 variation).
At lower LED voltage values, the ripple will be lower and the LED current would be higher.
By switching between the two modes depending on the load, we can get a better current accuracy without having to
adjust the LD voltage or the sense resistor.
Load Regulation (@ VIN = 12V)
10
Change in current (%)
Constant off-time mode
8
With mode change
6
4
Constant Off-time Mode
Constant Frequency Mode
2
0
2
4
6
8
Load Voltage (V)
Constant Off-Time Mode
5. Why is the efficiency of the demoboard higher with a load of two LEDs compared to a single LED load?
Losses in the HV9910BDB3 occur due mainly due to two factors:
a. Conduction losses in the FET and diode
b. Switching losses in the FET
Switching losses are dependent on the switching frequency, input voltage and total parasitic capacitance at the node.
At higher switching frequencies, the switching losses are higher.
Conduction losses are dependent on the duty cycle. Since the voltage drop on the FET is smaller than the voltage drop
on the diode (the on-resistance of the FET is very small), the higher the duty cycle, the smaller is the conduction loss.
Please note that we are ignoring the losses in the inductor, which will be identical in both cases.
Also, efficiency = POUT / PIN = POUT / (POUT + losses) = 1/ (1 + losses/POUT), where POUT is the output power and PIN is the
input power. So, if the output power is higher, the fixed switching losses are a smaller fraction of the output power and
thereby the efficiency is higher.
Comparing the operation of the converter in both modes at 12V input for this particular demoboard, the following
are the differences:
a. Output power is higher with 2 LEDs as the load
b. Switching frequency in the constant off-time mode is 55kHz, whereas it is 100kHz in the constant frequency
mode
c. Duty cycle of operation is about higher in the constant off-time mode by a factor of 2 than in the constant
frequency mode
All the above factors favor the higher load voltage and thus the demoboard has a higher efficiency when the load is larger.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
3
HV9910BDB3
Frequently Asked Questions (cont.)
6. Why are the LED current rise and fall times during PWM dimming different when the load changes from
one LED to two LEDs?
The LED current rise time is directly proportional to VIN - VOUT and the fall time is proportional to VOUT (where VIN is the
input voltage and VOUT is the output voltage). Since VOUT is higher with two LEDs, the rise time will be larger and the fall
time will be smaller.
Typical Results
Constant Frequency Mode:
The HV9910BDB3 is designed to be operated in the constant frequency mode when the load is a single LED. In this
mode, the line regulation of the LED current is less than 2% and full-load efficiency greater than 80%.
Efficiency vs. Input Voltage (@ VO = 4V)
Line Regulation (@ VO = 4V)
2
Change in current (%)
Efficiency (%)
88
86
84
1
0
-1
82
8
12
16
20
24
28
32
-2
8
Input Voltage (V)
16
20
24
28
32
Input Voltage (V)
Fig. 1. Efficiency vs. Input Voltage Plot
Fig. 2. Line Regulation of LED Current Plot
Efficiency vs. Load Voltage
(@ VIN = 12V)
Load Regulation (@ VIN = 12V)
3
Change in current (%)
90
Efficiency (%)
12
85
80
2
1
0
75
2
3
4
Load Voltage (V)
Fig. 3. Efficiency vs. Load Voltage Plot
5
2
3
4
Load Voltage (V)
Fig. 4. Load Regulation of LED Current Plot
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
4
5
HV9910BDB3
Constant Off-Time Mode:
The HV9910BDB3 is designed to be operated in the constant off-time mode when the load is two LEDs in series. In this
mode, the line regulation of the LED current is less than 2% and the efficiency greater than 80%.
Efficiency vs. Input Voltage
(@VO = 7.8V)
95
Line Regulation (@ VO = 7.8V)
2
Change in current (%)
Efficiency (%)
94
93
92
1
0
-1
91
-2
90
8
12
16
20
24
28
8
32
16
20
24
28
32
Fig. 6. Line Regulation of LED Current Plot
Fig. 5. Efficiency vs. Input Voltage Plot
Load Regulation (@ VIN = 12V)
Efficiency vs. Load Voltage
(@VIN = 12V)
10
Change in current (%)
95
Efficiency (%)
12
Input Voltage (V)
Input Voltage (V)
90
85
80
8
6
4
2
0
75
2
4
6
Load Voltage (V)
Fig. 7. Efficiency vs. Load Voltage Plot
8
2
4
6
Load Voltage (V)
Fig. 8. Load Regulation of LED Current Plot
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
5
8
HV9910BDB3
The variation in the switching frequency, when the HV9910BDB3 is operated in the constant off-time mode, is shown in
Figs. 9 and 10.
Switching Frequency vs. Load
Voltage (@VIN =12V)
Switching Frequency vs. Input Voltage
(@VO =7.8V)
140
Switching Frequency (kHz)
Switching Frequency (kHz)
140
120
120
100
100
80
60
40
20
8
12
16
20
24
28
32
Input Voltage (V)
Fig. 9. Switching Frequency vs. Input Voltage Plot
80
60
40
2
4
6
8
Load Voltage (V)
Fig. 10. Switching Frequency vs. Load Voltage Plot
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6
HV9910BDB3
Waveforms
Constant Frequency mode (LED Voltage = 3.3V):
LED Current
LED Current
Drain Voltage
Drain Voltage
(a) 10V input
(b) 12V input
LED Current
LED Current
Drain Voltage
Drain Voltage
(c) 24V input
(d) 30V input
Fig.11. Steady State Waveforms in Constant Frequency Mode
C1 (Yellow)
C4 (Green)
Time Base
:
:
:
Drain Voltage (10V/div)
LED Current (200mA/div)
5μs/div
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7
HV9910BDB3
Waveforms (cont.)
PWM
Dimming
Input
LED Current
(a)
PWM Dimming Performance
Time Scale
:
500μs/div
LED Current
PWM Dimming Input
PWM Dimming Input
LED Current
(a)
PWM Dimming Rise Time
Time Scale
:
(b) PWM Dimming Fall Time
10μs/div
Time Scale
:
10μs/div
Fig.12. PWM Dimming Performance in Constant Frequency Mode
C1 (Yellow)
C4 (Green)
:
:
PWMD Input Voltage (2V/div)
LED Current (200mA/div)
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8
HV9910BDB3
Waveforms (cont.)
Constant Off-time mode (LED Voltage = 6.4V):
LED Current
LED Current
Drain Voltage
Drain Voltage
(b) 12V input
(a) 10V input
LED Current
LED Current
Drain Voltage
Drain Voltage
(d) 30V input
(c) 24V input
Fig.13. Steady State Waveforms in Constant Frequency Mode
C1 (Yellow)
C4 (Green)
Time Base
:
:
:
Drain Voltage (10V/div)
LED Current (200mA/div)
10μs/div
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9
HV9910BDB3
Waveforms (cont.)
PWM
Dimming
Input
LED Current
(a)
PWM Dimming Performance
Time Scale
:
500μs/div
PWM Dimming Input
PWM Dimming Input
LED Current
LED Current
(b) PWM Dimming Rise Time
Time Scale
:
(a)
10μs/div
PWM Dimming Fall Time
Time Scale
:
10μs/div
Fig.14. PWM Dimming Performance in Constant Frequency Mode
C1 (Yellow)
C4 (Green)
:
:
PWMD Input Voltage (2V/div)
LED Current (200mA/div)
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10
C6
2
R6
R2
C1
100uF, 35V
5K
147K
C2
100uF, 35V
C5
R3
1k
5
7
6
J3
U1
ROSC
CS
GATE
HV9910B
PWMD
LD
VDD
3
3
1
2
2
0.1uF, 16V
1
1
1
VIN
GND
3
2
4
8
J4
R5
1 1
226k
3 3
R7
105k
J1
J1
C3
220uH
L1
R4
0.22
Si2318DS
Q1
1
D1
B140-13
2
C4
1
2
J2
J
2
HV9910BDB3
Schematic Diagram
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11
2.2uF, 50V
2.2uF, 50V
2 2
2.2uF, 16V
HV9910BDB3
Bill of Materials
Item
#
Quantity
Ref
Des
1
2
C1,C2
100µF, 35V, electrolytic capacitor
2
2
C3,C4
3
1
4
Description
Manufacturer’s Part
Number
Package
Manufacturer
SMT
Panasonic
2.2µF, 50V, X7R ceramic chip
capacitor
SMD1206
Murata
C5
0.1µF, 16V X7R ceramic chip
capacitor
SMD0805
Panasonic
ECJ-2VB1C104K
1
C6
2.2µF, 16V X7R ceramic chip
capacitor
SMD0805
TDK Corp
C2012X7R1C225K
5
1
D1
40V, 1A schottky diode
SMA
Diodes Inc
B140-13
6
2
J1,J2
2 position, 5mm pitch, vertical
header
Thru-Hole
On Shore
Tech
EDSTL130/02
7
2
J3,J4
3 position, 0.100” pitch, vertical
header
Thru-Hole
Molex
8
1
L1
220uH, 1.3A rms, 2.4A sat inductor
SMT
Coiltronics
9
1
Q1
40V, 45mΩ, 10nC N-channel FET
SOT-23
Vishay
10
1
R2
147KΩ, 1/8W, 1% chip resistor
SMD0805
-
-
11
1
R3
1kΩ, 1/8W, 1% chip resistor
SMD0805
-
-
12
1
R4
0.22Ω, 1/4W, 1% chip resistor
SMD1206
-
-
13
1
R5
226kΩ, 1/8W, 1% chip resistor
SMD0805
-
-
14
1
R6
5KΩ top adjust trimpot
SMT
Bourns Inc
15
1
R7
105kΩ, 1/8W, 1% chip resistor
SMD0805
-
16
1
U1
Universal LED Driver
SO-8
Supertex
EEV-FK1V101P
GRM31CR71H225KA88L
22-03-2031
DR127-221-R
Si2318DS
3361P-1-502G
HV9910BLG-G
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives an
adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability to the
replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and specifications
are subject to change without notice. For the latest product specifications refer to the Supertex inc. website: http//www.supertex.com.
©2008
082608
All rights reserved. Unauthorized use or reproduction is prohibited.
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
12
HV9910BDB7
HV9910B PFC 40W
LED Driver Demoboard
General Description
The Supertex HV9910BDB7 demonstrates the use of an
HV9910B control IC in an off-line, High Brightness LED driver
application. The board incorporates power factor correction
(PFC) and satisfies the limits for harmonic currents according
to the EN61000-3-2 Class C standard having total harmonic
distortion (THD) less than 20%. The board features a low
component count and long life operation due to the absence
of electrolytic capacitors. The board is designed to supply a
string of LEDs with a current of 350mA and a voltage in the 65
to 105V range from a 220/230VAC line.
The conversion stage draws line current throughout the AC
line cycle, partly using a charge pumping and partly using a
boost conversion technique to charge the bulk energy storage
capacitors. The LED current is provided with a continuous
mode buck stage giving a DC current with about 30% peak-topeak ripple. A patent for this conversion technique is pending.
Please inquire with the Supertex applications department for
design guidance, should change of input line voltage, output
voltage, or output current be desired.
An effort was made to satisfy the requirements of CISPR
15 (EN55015), limits and methods of measurement of radio
disturbance characteristics of electrical lighting and similar
equipment.
Specifications
Parameter
Value
Input voltage
190~265VAC, 50Hz
Power factor
0.95
<20%;
EN 61000-3-2 Class C
Total harmonic distortion
EMI limits
CISPR 15
Output voltage
65~105V
Output current
350mA±10%
Output power
40W
Efficiency
90%
Load regulation
<3%
AC Line regulation
<1.5%
Output ripple
<30% peak-peak
Life time
Non-Electrolytic
Output short circuit
protection
No
Output open circuit
protection
Yes
Dimensions
3.0” x 2.3” x 1.1”
(76mm x 58mm x 28mm)
The connection diagram details the hookup of the board to
the AC line. Note that the load is NOT galvanically isolated,
and that measurements to the board require measurement
techniques in common use with non-isolated off-line power
supplies (isolation transformers, differential probes, etc).
Board Layout
Top View
Bottom View
Actual Size: 3.0” x 2.3” x 1.1”
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HV9910BDB7
Connection Diagram
105VDC
350mADC
Connections
Input Voltage - Connect the AC line input voltage to AC
VIN as shown.
220VAC
LED String - Connect the LED strings between LED+ and
LED- as shown (anode of the string to LED+ and cathode
to LED-).
Schematic Diagram
J1
HDR
AC
L2
560μH
CL
CL-140
190 … 265VRMS
F1
1A F
L3
4.7mH
C10
220nF
BR1 1
DF04S
3
D3
STTH1L06A
D9
S1J
J2
HDR
D2
STTH1L06A
4
C12
100nF
R11A
10kΩ
D1
STTH1R06A
POS
C5
470nF
NEG
VIN
8
1
IC1
HV9910BNG-G
1
R12A
499kΩ
R1
464kΩ
C2
2.2μF
C3
10nF
AC
R12B
499kΩ
C1
2.2μF
C20 R20
100pF 2.2kΩ
2
C11
100nF
105VDC
350mADC
D4
STTH1L06A
GATE
HV9910B
CS
ROSC
GND
PWM
VDD
3
5
6
LD
7
CT1
1:1 2
L1
10mH
3
4
2
R4
1.0kΩ
C7
100pF
4
R16
620mΩ
M1
SPP04N50C3
R6
620mΩ
C8
10μF
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2
HV9910BDB7
Typical Characteristics
Figure 1.
Efficiency at full load.
Figure 2.
Efficiency (VLED = 105V, ILED = 350mA)
0.91
Efficiency at nominal AC line voltage.
Efficiency (VIN = 230VAC, ILED = 350mA)
0.91
0.9
0.905
EFF
EFF
0.89
0.9
0.88
0.895
0.87
0.89
0.86
0.885
190
Figure 3.
210
230
VAC
250
270
Load regulation.
Figure 4.
ILED Load Regulation (VIN = 230VAC)
0.366
0.355
0.358
0.354
0.356
0.353
0.354
0.352
0.352
0.351
0.35
0.35
0.348
104.3
84.4
VO
67.5
Harmonic Distortion vs. AC Line Voltage.
Harmonic Distortion (VO = 105V, ILED = 350mA)
0.349
190
Figure 6.
0.96
35
0.95
30
0.94
25
210
230
VAC
250
270
Power Factor vs. AC Line Voltage.
Power Factor (VO = 105V, ILED = 350mA)
PF
0.93
THD,%
0.92
20
0.91
15
0.9
10
0.89
5
0
190
ILED
0.356
0.36
40
67.5
ILED Line Regulation (VO = 105V)
0.357
0.362
Figure 5.
84.4
VO
AC line regulation.
0.358
ILED
0.364
0.85
104.3
0.88
210
230
VAC
250
270
0.87
190
210
230
VAC
250
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3
270
HV9910BDB7
Figure 7.
EMI Characterization - Conducted Emissions vs CISPR 15 Limits.
Figure 8.
CT1 Construction Diagram.
4T : 4T
AWG24
1
4
2
3
Mfr: Ferroxcube
PN: TN/10/6/4
1
4
3
2
TOP VIEW
Vertical Toroid Mount
Mfr: Lodestone Pacific
PN: VTM455-4 CT1
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4
HV9910BDB7
Bill of Materials
Qty
Ref Des
Description
Manufacturer
Part Number
1
BR1
Rect Bridge, DFS, 400V, 1.5A
Fairchild Semi
DF04S
2
C1, C2
Panasonic ECG
ECQ-E2W225KH
1
C3
Cap, MZPEF, 630VDC, 5%, 10NF
EPCOS Inc
B32529C8103J
1
C5
Cap, MZPEF, 400V, 10%, .47UF
EPCOS Inc
B32522C6474K
1
C7
Cap, CER, NP0, 50V, 10%, 0805, 100PF
Kemet
C0805C101K5GACTU
1
C8
Cap, CER, X7R, 10V, 10%, 1206, 10UF
Murata
GRM31CR71A106KA01L
1
C20
Cap, CER, NP0, 1000V, 5%, 0805, 100PF
Vishay/Vitramon
VJ0805A101JXGAT5Z
1
C10
Cap, MKP, 220NF, 305VAC, X2, 125C, 20%
EPCOS Inc
B32922T2224M
2
C11, C12
Cap, MKP, 100NF, 305VAC, X2, 125C, 20%
EPCOS Inc
B32922A2104M
1
CL1
Inrush current limiter, 50/0.89Ω, 1.1A
GE Sensing
CL-140
Yageo / Ferroxcube
TN10/6/4-3E25
Toroidal core mount, 0.455Dia, 4PIN
Lodestone Pacific
VTM455-4T
Magnet wire, MW28C, SPN AWG24
MWS Wire Industries
SPN AWG24
Diode, ultrafast, 600V, 1A, SMA
STMicroelectronics
STTH1R06A
Diode fast, 600V, 1A ,SMA
STMicroelectronics
STTH1L06A
Diodes Inc
S1J-13-F
Wickmann USA
37011000410
Aavid Thermalloy
574502B03700G
Supertex
HV9910BLG-G
Header, 2POS, .156, VERT TIN
Molex
26-48-1021
1
1
Cap, MZPEF, 450VDC, 10%, 2.2UF
Core, toroidal, TN10/6/4-3E25
CT1
AR
1
D1
3
D2, D3, D4
1
D9
Rectifier, GPP, 600V, 1A, SMA
1
F1
Fuse fast, 1.00A, IEC, Short, TR5
1
HS1
Heatsink, TO-220, Ver MNT W/Tab, H75 21K
1
IC1
IC, LED Driver, 8-Lead SOIC
2
J1, J2
1
L1
Choke AXL,14mm, 10mH, 10%, 350mA
Renco
RL-1292-10000
1
L2
Choke SH RAD,16mm, 10%, 560µH, 1.1A
Sumida
RCR1616NP-561K
1
L3
Choke SH RAD,13mm, 15%, 4.7mH, 370mA
Sumida
RCP1317NP-472L
1
M1
MOSFET, N-CH, 560V, 4.5A, TO-220AB
Infineon Technologies
SPP04N50C3
1
R1
Resistor 1/8W, 1%, 0805, 464KΩ
Panasonic ECG
ERJ-6ENF4643V
1
R4
Resistor 1/8W, 1%, 0805, 1.00KΩ
Panasonic ECG
ERJ-6ENF1001V
1
R11
Resistor 1/8W, 1%, 0805, 10.0KΩ
Panasonic ECG
ERJ-6ENF1002V
R12A, R12B Resistor 1/4W, 1%, 1206, 499KΩ
Panasonic ECG
ERJ-8ENF4993V
Resistor 1/2W, 5%, 2010, 2.2KΩ
Panasonic ECG
ERJ-12ZYJ222U
Resistor 1/2W, 1%, 1206, .62Ω
Susumu Co Ltd
RL1632R-R620-F
2
1
R20
2
R6, R16
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
Supertex inc.
©2009 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
010510
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
5
Supertex inc.
AN-H48
Application Note
Buck-based LED Drivers
Using the HV9910B
Fundamental Buck Converter topology is an excellent
choice for LED drivers in off-line (as well as low-voltage)
applications as it can produce a constant LED current at
very high efficiencies and low cost. A peak-current-controlled
buck converter can give reasonable LED current variation
over a wide range of input and LED voltages and needs
little effort in feedback control design. Coupled with the fact
that these converters can be easily designed to operate at
above 90% efficiency, the buck-based driver becomes an
unbeatable solution to drive High Brightness LEDs.
The Supertex HV9910B provides a low-cost, low component
count solution to implement the continuous mode buck
converter. HV9910B has two current sense threshold voltages
– an internally set 250mV and an external voltage at the
LD pin. The actual threshold voltage will be the lower of the
internal 250mV and the voltage at the LD pin. The low sense
voltage allows the use of low current sense resistor values.
HV9910B operates down to 8V input, which is required for
automobile applications, and can take a maximum of 450V
input, which makes it ideal for off-line applications. It also
has an internal regulator that supplies power to the IC from
the input voltage, eliminating the need for an external low
voltage power supply. It is capable of driving the external
FET directly, without the need for additional driver circuitry.
Linear or PWM dimming can also be easily implemented
using the HV9910B.
This Application Note discusses the design of a buck-based
LED driver using the HV9910B with the help of an off-line
application example. The same procedure can be used to
design LED drivers with any other lower voltage AC or DC
input; 12V for example.
The information in this Application Note also applies to the
Supertex HV9910.
Circuit Diagram
D1
D2
LED(s)
L1
VIN
C1
VDD
C2
RT
HV9910B
LD
C3
PWMD
GND
CS
R1
NTC1
AC Input Voltage Range
Expected LED string voltage
VNOM,AC = 120V rms
VO,MIN = 20V
VMIN,AC = 90V rms
VMAX,AC = 135V rms
Q1
GATE
VO,MAX = 40V
R2
Stabilized LED current
Expected Efficiency
IO,MAX 350mA
η = 0.9
freq = 60Hz
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
AN-H48
Step 1: Switching Frequency and resistor (R1)
The switching frequency determines the size of the inductor L1 and size or type of input filter capacitor C1. A larger
switching frequency will result in a smaller inductor, but will
increase the switching losses in the circuit. For off-Line applications, typical switching frequencies should be in range
20KHz-150KHz. The higher the input voltage range (for example in Europe 230VAC), the lower the frequency should
be to avoid extensive capacitive losses in the converter. For
North America AC line a frequency of fS = 100kHz is a good
compromise. From the datasheet, the oscillator resistor
needed to achieve this is 228kΩ.
The hold-up and input filter capacitor required at the the diode bridge output have to be calculated at the minimum AC
input voltage. The minimum capacitor value can be calculated as:
VO,MAX x IO,MAX
C1 ≥
(5)
2 x V2MIN,AC - V2MIN,DC x η x freq
In this example, C1 ≥ 26.45µF.
Note: Equation (5) yields a conservative estimate to
for the least amount of capacitance required. It means
that the capacitor filter will normally care large ripple
content. Some electrolytic capacitors may not be able
Step 2: Choose the Input Diode Bridge (D1) to withstand such ripple current and minimum value of
C1 capacitor may not be met, forcing the design to use
and the thermistor (NTC1)
The voltage rating of the diode bridge will depend on the larger value capacitor. In the case where the allowable
maximum value of the input voltage. The current rating will ripple at the input of the buck converter is large, the
depend on the maximum average current drawn by the con- capacitor C1 can be reduced significantly. See the Appendix for a more accurate calculation of the required
verter.
capacitor value.
VBRIDGE = 1.5 • (√2 • VMAX,AC) (1)
The voltage rating of the capacitor should be more than the
VO,MAX • IO,MAX
peak input voltage with 10-12% safety margin.
IBRIDGE =
(2)
V
•
η
MIN, DC
VMAX,CAP ≥ √2 • VMAX,AC → VMAX,CAP ≥ 191V
(6)
The 1.5 factor in equation (1) a 50% safety margin is more
Choose a 250V, 33µF electrolytic capacitor.
than enough. For this design, choose a 400V, 1.0A diode
bridge.
Such electrolytic capacitors have a sizable ESR component.
The large ESR of these capacitors makes it inappropriate
Placing a thermistor (or resistor) in series with input bridge
to absorb the high frequency ripple current generated by
rectifier will effectively limit the inrush charging current to inthe buck converter. Thus, adding a small MLCC capacitor
put bulk capacitor C1 during the initial start-up of the convertin parallel with the electrolytic capacitor is a good option to
er. Except this useful action during very short time interval,
absorb the high frequency ripple current. The required high
such a series element creates a unnecessary power loss
frequency capacitance can be computed as:
dissipation during normal operation of the converter, and
IO,MAX • 25
must be minimized. A good rule of thumb is that the thermisC2 =
(7)
tor should limit the inrush current to not more than five times
(fS • 0.05 • VMIN,DC )
the steady state current as given by equation (2), assuming
maximum voltage is applied. The required cold resistance is:
In this design example, the high frequency capacitance reVBRIDGE
quired is about 250V, 22µF.
RCOLD =
(3)
5
•
I
BRIDGE
Step 4: Choose the Inductor (L1)
Step 3: Choose the Input Capacitors (C1/C2)
The first design criterion to meet is that the maximum LED
string voltage is should be less than half the minimum input
voltage to avoid having to implement a special loop compensation technique. For this example, the minimum rectified
voltage should be:
VMIN,DC = 2 • VO,MAX = 80V
Supertex inc.
(4)
The inductor value depends on the ripple current in the
LEDs. Assume a +/- 15% ripple (a total of 30%) in the LED
current, an aggressive assumption would go up to +/-30% to
reduce the size of the inductor more than twice at the price
of reduced efficiency and, possibly, reduced LED lifetime.
Then, the inductor L1 can be computed at the rectified value
of the nominal input voltage as:
L1 =
]
This gives us a 200Ω resistance at 25°C. Choose a thermistor with a resistance around 200Ω and rms current greater
than 0.2A for that application.
VO,MAX x 1-
VO,MAX
√2 x VAC,NOM
]
0.3 x IO,MAX x fS
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2
(8)
AN-H48
Step 6: Choose the Sense Resistor (R2)
In this example, L1 = 2.9mH.
The sense resistor value is given by:
The peak current rating of the inductor will be:
IP = 0.35 • 1.15 = 0.4A 0.25
R2 =
(14)
1.15 • IO,MAX (9)
The rms current through the inductor will be the same as
the average current for the chosen 30% ripple.Right inductor
for this application is an off-the-shelf 2.7mH, 0.54A (peak),
0.33A (rms) inductor.
if the internal voltage threshold is being used. Otherwise,
substitute the voltage at the LD pin instead of the 0.25V in
equation (14).
For this design, R2 = 0.55Ω. Also calculate the resistor power
dissipation:
Step 5: Choose the FET (Q1) and Diode (D2)
The peak voltage seen by the FET is equal to the maximum
input voltage. Using a 50% safety rating,
VFET = 1.5 • (√2 • 135) = 286V PR2 = (IO,MAX)2 • R2 = 0.067W
(15)
A 0.1W resistor is good for this application.
(10)
The maximum rms current through the FET depends on the
maximum duty cycle, which is 50% by design. Hence, the
current rating of the FET is:
Note:
Capacitor C3 is a bypass capacitor. A typical value of 1.0 to
2.2µF, 16V is recommended.
Design for DC/DC Applications
(11)
The same procedure can be used for DC/DC applications
(like the HV9910DB3). The only modifications are that the
input diode bridge and input hold-up capacitor are not required. A small input capacitance to absorb high frequency
ripple current is all that is required. This capacitance can be
computed using equation (7).
Typically a FET with about 3 times the current is chosen to
minimize the resistive losses in the switch.
For this application chose a 300V, <1A MOSFET, such as a
BSP130 from Phillips. Actual MOSFET type should be determined by the transistor permitted power dissipation on
printed board. For example, a BSP130 SOT-223 package
limits the dissipation to less than a Watt at 50+ Celsius, even
if the MOSFET peak current capability is 1.5A. A good rule
of thumb is to limit overall MOSFET power dissipation to
not more than 3-5% of total output power, by making a right
transistor choice. In choosing MOSFET transistors for such
LED drivers, going bigger does not mean getting better, just
the opposite. Using TO-220 transistor 500/4A/2W instead
of SOT-223 transistor 300V/0.5A/6W does more harm than
good, reducing overall efficiency by several percent.
Appendix
The more accurate equations for computing the required capacitance values are:
]
2 x VO,MAX x IO,MAX x t1 +
The peak voltage rating of the diode is the same as the FET.
Hence,
VDIODE = VFET = 286V
(12)
C1 ≥
IDIODE = 0.5 • IO,MAX = 0.175A 1
4 x freq
]
2 x V2MIN,AC - V2MIN,DC x η
(16)
]
(17)
For the example in this application note, the actual minimum
capacitance required from the above equations is 19µF (as
compared to 26µF from equation (5)).
The average current through the diode is:
]
VMIN,DC
1
sin-1
t1 =
2 x π x freq
√2 x VMIN,AC
]
]
IFET ≈ IO,MAX • √0.5 = 0.247A (13)
Choose a 300V, 1A ultra-fast diode.
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
Supertex inc.
©2011 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
022509
3
Supertex inc.
AN-H50
Application Note
Constant, Off-time, Buck-based, LED Drivers
Using the HV9910B
Constant frequency, peak current controlled buck converters
(Fig. 1) are an excellent choice for driving LEDs for a number
of reasons:
►► Reasonable regulation of LED current over wide
variations in input and output voltages.
►► Simple to design as no feedback compensation
is required.
►► PWM dimming response of the converters is
almost instantaneous.
However, peak current controlled buck converters go into
sub-harmonic oscillations at duty cycles over 50%. These
oscillations cause the average output current to drop, while
the output ripple current increases. The only way to avoid
these problems is by adding slope compensation circuitry
externally. The slope compensation adds an upward slope
on to the current sense signal and the converter can be
stabilized by varying the slope of the added ramp (Fig.
2). This added ramp causes an error between the sensed
current (as seen at the CS pin of the HV9910B) and the
actual LED current.
Although this error can be compensated for by changing the
sensed resistor appropriately, the converter’s rejection of
the input and output voltage variations will be significantly
degraded. Thus, changing the input or output voltage will
significantly change the LED current, without additional
feedback circuitry for regulating the LED current. This
makes the peak current controlled buck converter practically
useless for cases where the input voltage is less than twice
the output voltage.
This problem can be overcome by changing the control
method to a constant off-time operation. In this case, the offtime is fixed by design, the on-time is based on the current
sense signal and the switching time-period adjusts to be
equal to the on-time plus the off-time. This change will allow
the converter to work with greater than 50% duty cycles and
still have the advantages of the peak current controlled buck
converter given above.
The information in this Application Note also applies to the
Supertex HV9910.
Fig. 1. Constant Frequency, Peak Current Controlled LED Driver
LED(s)
D1
R1
VIN
VDD
+
-
RT
L1
HV9910B
C1
LD
GATE
PWMD
C2
Supertex inc.
Q1
CS
GND
R2
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AN-H50
Fig. 2. Slope Compensation to Eliminate Subharmonic Oscillation
IR2
Current Through
Sense Resistor
VCS
Error due to
Slope Compensation
Added
External Slope
Voltage Across
Sense Resistor
t
t
The unique design of the oscillator in the HV9910B allows
the IC to be configured for either constant frequency or constant off-time based on how one resistor, connected to the
RT pin, is wired. For normal operation as constant frequency
converter, the resistor at the RT pin is connected to GND
(Fig. 3a). For operation as a constant off-time converter, the
resistor is connected between the RT and GATE pins (Fig.
3b). In both cases, the equation to determine the resistor is
given by:
RT(kΩ) + 22
TOSC(µs) =
25
For constant frequency operation TOSC is set to the switching
time period and for constant off-time operation, TOSC is set to
the required off-time.
Fig. 3b. Constant Off-Time Operation
Fig. 3a. Constant Frequency Operation
Connected
to GATE
Connected
to GND
VIN
VDD
VIN
R1
VDD
RT
HV9910B
HV9910B
LD
CS
GND
C2
R2
Q1
GATE
PWMD
CS
GND
C2
LD
Q1
GATE
PWMD
RT
R2
Fig. 4 Circuit Diagram
LED(s)
D1
VIN
VDD
+
-
R1
RT
L1
HV9910B
C1
LD
GATE
PWMD
C2
Supertex inc.
Q1
CS
GND
R2
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2
AN-H50
An example detailing the design of a constant off-time buck
converter is shown in Fig. 4.
Input Voltage:
VIN,MIN = 9V
ripple desired in the output current. Assuming a 30% peak to
peak ripple in the output current,
VO,NOM • tOFF
L1 =
(4)
0.3 • IO Output Voltage (corresponds to two 1W LEDs):
The peak current rating of the inductor should be greater
than 1.3xIO and the rms current rating of the inductor should
be at least IO. For this example, the closest inductor available is a 330μH inductor with a 0.6A rms current rating and
a 0.6A saturation current rating.
VO,MIN = 4.6V
Step 4: Choose the Sense Resistor (R2)
VIN,NOM = 12V
VIN,MAX = 16V
VO,NOM = 6.8V
VO,MAX = 8V
LED current:
The peak current sensed by the HV9910 corresponds to the
average output current plus one half of the actual current
ripple. The peak current is given by:
IPK = IO +
IO = 350mA
Expected Efficiency:
Step 1: Choose the Nominal Switching Frequency
Although the switching frequency is variable, a nominal
switching frequency can be chosen. The actual frequency
will vary around this nominal value based on the actual input
and output conditions. A larger switching frequency will typically result in a smaller inductor, but will increase the switching losses in the circuit.
A typical switching frequency: fS,NOM = 100kHz is a good compromise, which corresponds to a time period of:
1
TS,NOM =
= 10μs
fS,NOM (1)
x TS,NOM
(2)
This off-time will then be set by the resistor R1 based on the
following equation:
R1 = (tOFF (μs) • 25) - 22 (kΩ)
R2 =
0.25
IPK
(6) if the internal voltage threshold is being used. Otherwise,
substitute the voltage at the LD pin instead of the 0.25V in
equation (6). The power rating required for the sense resistor can be computed using:
V
P
= (IO)2 X O,MAX × R2
SENSE
V
IN,MIN
(7)
For this design,
IPK = 0.394A, R2 = 0.633Ω, and PSENSE = 0.069W
Step 5: Choose the FET (Q1) and Diode (D1)
The peak voltage seen by the FET is equal to the maximum
input voltage. Using a 50% safety rating:
VFET = 1.5 • VIN,MAX = 24V (3)
In this case, tOFF = 4.33μs and R1 = 86.25kΩ.
Note that in this case, the converter is operating at 56.7%
duty cycle.
Step 3: Choose the required Inductor L1
The value of the inductor L1 will depend on the peak-to-peak
Supertex inc.
Note: Capacitor C2 is a bypass capacitor. A typical value of
1μF, 16V ceramic capacitor is recommended.
Step 2: Compute the Off-Time and Resistor R1
The off-time can be calculated as:
(5)
2 • L1 The sense resistor can be then be computed as:
η = 0.85
V
tOFF = 1 − O,NOM
V
IN,NOM
VO,NOM • tOFF
(8)
The maximum rms current through the FET is:
= IO x
IFET
√
VO,MAX
VIN,MIN
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3
(9)
AN-H50
Typically a FET with about 3 times the current is chosen to
minimize the resistive losses in the switch. For this application, choose a 40V, 1Ω FET (TN2504 from Supertex in a
SOT-89 package).
Step 6: Analysis of the Switching Frequency Variation
The peak voltage rating of the diode is the same as the FET.
Hence:
VDIODE = VFET = 24V The average current through the diode is:
V
IDIODE = Io × 1 − O,MIN
VIN,MAX
= 0 .25 A
The two extremes of the switching frequency can be approximately computed as:
fS,MIN =
(10)
1
TS,MAX
V
1 − O,MAX
V
IN,MIN
=
tOFF
V
1 − O,MIN
V
1
IN,MAX
fS,MAX =
=
TS,MIN
tOFF
(11)
(12)
(13)
In this case, the switching frequency varies from:
Choose a 30V, 1A schottky diode.
25kHz (VIN = 9V, VO = 8V) to
164kHz (VIN = 16V, VO = 4.6V)
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
Supertex inc.
©2011 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
111011
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
4
Supertex inc.
AN-H64
Application Note
Compatibility and Functional Differences between
the HV9961 and HV9910B LED Drivers
Figure 1. Typical application circuit of HV9910B and HV9961.
VIN
VO
1
5 PWMD
VIN
L
GATE 4
HV9961
or CS
HV9910B
6 VDD
VLD
7 LD
GND
2
RT 8
3
Peak-Current Control
Current Control
vs.
RCS
RT
Average-Mode
Peak-current control of a buck converter used in the
HV9910B, while being the most economical and simplest way
to regulate the LED current, suffers accuracy and regulation
problems. These problems arise from the so-called peak-toaverage current error, contributed by the current ripple in the
output inductor and by the propagation delay in the current
sense comparator.
The peak-to-average current error ∆IL(ERR) is inherent to the
HV9910B, since the IC is controlling the peak inductor cur-
rent IL(PK), whereas the intent is to regulate the average current IL(AVG). The difference between the two currents equals
one-half of the inductor current ripple ∆IL, which can be expressed by the following equation:
1
VOtOFF
∆IL =
(1)
2
2L
In this equation, VO is the LED voltage, tOFF is the off-time
of the GATE output of the HV9910B (the lower waveform in
Fig.2), and L is the inductance value. Note that all parameters in right side of Equation 1 can vary from one part to
another and depend on the operating temperature.
Figure 2. Peak-to-average current error produced by the peak-current control method of
HV9910B.
ΔtCS
IL(PK)
IL(CS)
IL(AVG)
ΔIL(ERR)
ON
Supertex inc.
ΔIL
OFF
ON
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AN-H64
Another source of error produced by a peak-current controller such as the HV9910B is associated with the currentsense comparator delay ∆tCS. The actual peak current IL(PK)
is higher than the comparator threshold reference IL(CS) because of this propagation delay. Therefore, the total peak-toaverage error can be expressed as:
∆IL(ERR) =
VOtOFF - 2VIN∆tCS
2L
(2)
where VIN is the input DC power supply voltage.
As one could see from Equation 2, the average inductor current IL(AVG) also suffers poor load and line regulation, since it
is dependent on the input voltage VIN and the output voltage
VO.
Lastly, there is a significant part-to-part variation in the LED
current that occurs due to the CS input offset voltage VOS.
Although this offset voltage is only ±25mV at -40°C < TA
< +85°C, it contributes as much as ±10% variation of the
LED current even at the maximum CS threshold voltage of
250mV.
The HV9961 overcomes the above drawbacks by means of
Supertex’s average-mode constant current control method.
The IC regulates the average inductor current IL(AVG) directly and accurately within ±3% over a wide GATE duty cycle
range of at least 0.1 < D < 0.75. It also includes an auto-zero
circuit at the CS input that cancels the propagation and offset errors.
Linear Dimming
When the LD voltage is VLD ≥1.5V, the output LED current is
simply programmed with the HV9961 as:
IL(AVG) =
272mV ±3%
RCS
(3)
where 272mV is the internally fixed reference voltage. Otherwise:
IL(AVG) =
VLD ±3%
5.5 • RCS
(4)
Unlike the HV9910B, which has the LD range from 0 to 0.25V,
the active LD input voltage range of the HV9961 is from 0
to 1.5V. Moreover, for the HV9910B, VLD = GND does not
produce ILED = 0A due to the DMIN limitation. There is always
some residual LED current remaining despite connecting LD
to GND. The HV9961 overcomes this issue by disabling the
GATE output when VLD < 150mV. The GATE switching resumes when VLD > 200mV.
Note that the latter feature of the HV9961 allows a mixedmode PWM/linear dimming mode. A single square-wave input signal can be applied at LD, where both the signal duty
cycle and its amplitude are modulated in order to expand the
dimming range.
Figure 3. Typical output voltage regulation characteristic of LED current.
0.50
LED Current (A)
0.45
0.40
HV9961
0.35
0.30
HV9910B
0.25
0.20
0
10
20
30
40
50
60
Output Voltage (V)
Supertex inc.
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2
AN-H64
Figure 4. Effect of the output short circuit on the inductor current.
HV9910B
ILIM
HV9961
400µs
SHORT
Short-Circuit Protection
Both the HV9910B and HV9961 are characterized by a minimum on-time of the GATE output. This minimum on-time
includes the leading-edge blanking delay and the currentsense comparator propagation delay. The minimum ontime is 0.47µs(max) for the HV9910B and 1.0µs(max) for
the HV9961. When a short circuit is applied at the output of
the buck converter, the only voltage available to reset the
magnetic flux in the inductor during tOFF is the rectifier diode voltage drop. When the converter keeps switching at
the same frequency rate this may not be enough. Therefore,
the inductor current will keep rising every switching cycle.
(See Figure 4.)
The HV9961 is protecting the LED driver from such “staircase” saturation of the inductor by introducing a second
threshold ILIM = 0.44V/RCS. When this threshold is reached,
the GATE output becomes disabled for 400µs, thus letting
the inductor current ramp down to a safe level.
Constant-Frequency and Constant Off-Time
Operating Modes
The HV9910B can be configured for operating in either
switching mode. When RT is connected to GND, it maintains
a constant switching frequency. Wiring RT to GATE yields a
fixed tOFF mode. The corresponding timing equations are:
tOSC = 40pF • RT + 0.88µs tOFF = 40pF • RT + 0.88µs (5a)
(5b)
where tOSC is the switching period with RT wired to GND, and
tOFF is the off-time with RT connected to GATE.
The HV9961 does not support the fixed frequency mode.
Moreover, the RT resistor must be wired to GND in all cases.
Supertex inc.
Therefore, the HV9961 cannot be used as a direct drop-in
replacement in the applications of the HV9910B wired for
the fixed tOFF operation, and a layout change is required. The
HV9961 tOFF is given by:
tOFF = 40pF • RT + 0.3µs (5c)
If the HV9910B is wired for the fixed frequency operation,
the conversion to the HV9961 will merely require the RT resistor value change. Since tOFF = (1-VO / VIN) • tOSC, Equations
5a and 5c can be solved for the new RT value:
( (
RT(HV9961) = 1 -
VO
• (RT(HV9910B) + 22kΩ) - 7.5kΩ (6)
VIN
Duty Cycle Range
The duty cycle is determined by the equation D = tON/tOSC
= tON/(tOFF + tON). Both the HV9910B and the HV9961 have
their minimum duty cycle Dmin limited by the minimum ontime. However, with the HV9961 the guaranteed ±3% accuracy of the LED current can only be achieved with the duty
cycle Dmin>0.08~0.1.
The maximum duty cycle of the HV9910B operating with
the fixed frequency is limited to Dmax = 0.5. Exceeding D =
0.5 with this operating mode causes sub-harmonic oscillation at ½ of the switching frequency. When the HV9910B is
operated with fixed tOFF, there is no theoretical limit of Dmax.
However, due to parasitic resistances in the circuit and large
switching frequency variation, it is not recommended that a
Dmax = 0.8 be exceeded with this operating mode.
With the HV9961, regulation of the average inductor current
is limited to Dmax ≤ 0.75. When D = 0.75 (125oC) or D = 0.8
(105oC) is exceeded, the functionality of the HV9961 will begin approaching that of the HV9910B, and the LED current
will drop accordingly.
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3
AN-H64
Table 1. Functional comparison of HV9910B and HV9961.
Feature
HV9910B
HV9961
Fixed-Frequency Mode
Resistor from RT to GND
N/A
Fixed Off-Time Mode
Resistor from RT to GATE
Resistor from RT to GND
(value adjustment needed for
conversion from HV9910B)
250mV or VLD (peak)
272mV or VLD /5.5 (average)
10%
Auto-zero
Depends on inductance and
switching frequency variation
Independent of inductance and
switching frequency variation
Poor. LED current depends on input
and output voltage
Good
0 to 250mV
0.2V(0.15V) to 1.5V
5% (typ.) of ILED @ VLD = 250mV
0A
none
440mV
N/A
400µs
465ns
1000ns
0.5 (fixed freq.), 0.8 (fixed TOFF)
0.75
Current Threshold
Current Threshold Accuracy
LED Current Accuracy
LED Current Regulation
LD Input Range
Residual LED Current at VLD = GND
Current Limit Threshold
Hiccup Time
Minimum On-Time
Maximum Duty Cycle
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
Supertex inc.
©2011 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
081109
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
4
DN-H01
Design Note
Isolated LED Driver
Using the HV9910B
Design Parameters
Introduction
In a few general lighting applications, there is a need to isolate
the LEDs from the AC input line. These are cases when
the driver terminals of the LED strings are exposed to the
external environmental conditions, or the LED strings are user
accessible for maintenance during operation. In these cases,
an isolated LED driver is needed for safety considerations.
Parameter
Value
Input voltage
90 – 256VAC, 50/60Hz
LED string voltage
4.0 – 16V
LED current
This design note provides the circuit schematic, bill of
materials, and transformer design for an isolated LED driver
using Supertex’s HV9910B. The power stage is a flyback
converter with an isolated secondary side feedback, using an
opto coupler, to ensure a very good line and load regulation
(typically <1% over line and load). Below are the design
parameters which are the target specification for this LED
driver circuit. This LED driver will also meet CISPR-15 EMI
limits for general lighting.
350mA
Initial regulation
<5%
Line and load regulation
<1%
Over voltage protection
20V
Switching Frequency
100kHz
The information in this datasheet also applies to the Supertex
HV9910.
Circuit Schematic
1
NEG
2
C1
RT
HV9910B
LD
GATE
PWMD
CS
14
8
228k
MMBT2222A
5.49k
4
5
R13
100
R7
20k
U3
97.6k
C11
1k
D9
0.1uF
LMV431
1
R12
4.99k
J1
LED-1
C8
R9
R11
20V
1.78,1/4W 1.78,1/4W
H11A817A
VDD
0.01uF, 250VAC
F1
2A, 250VAC
R2
R1
9
R6
4
R8
0.56
D8
1n, 250VAC
9.1V
STD1NK60T4
O
BAV20W-7
3
C6
C12
D7
Q3
O
C5
3
3
9
B1100-13
2
1
13
VDD
GND
L1
C9
12
5
4
BU9HS-153R15B
0.01uF, 250VAC
1uF, 16V
C4
Q1
1
R5
1uF, 50V
VDD
VIN
AC2
POS
AC1
R3
1k
U2
O
D3
MUR140RL
D4
U1
10
4.7uF, 25V
C3
10uF, 400V
C2
2.2uF, 400V
2.2uF, 400V
C13
D1
1N4764ADO41
LED+1
D2
T1
CL-140
4.7uF, 25V
t
RT1
J2
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DN-H01
Bill of Materials
Item
Qty
#
Ref
Description
Package Manufacturer
Manufacturer’s Part
Number
0.01uF, 250VAC metal polypropylene capacitors
Thru-Hole
EPCOS Inc
B81122A1103M
Thru-Hole
Panasonic
ECQ-E4225KF
Nichion
UVR2G100MHD
1
2
C1, C4
2
2
C2, C13 2.2uF, 400V metal film capacitors
3
1
C3
4
2
C5, C6
5
1
6
10uF, 400V electrolytic capacitors
Radial
4.7uF, 25V X7R ceramic chip capacitor
SMD1210
TDK Corporation
C3225X7R1E475M
C8
1uF, 50V X7R ceramic chip capacitor
SMD1206
TDK Corporation
C3216X7R1H105K
1
C9
1uF, 16V X7R ceramic chip capacitor
SMD0805
TDK Corporation
C2012X7R1C105K
7
1
C11
0.1uF, 16V X7R ceramic chip capacitor
SMD0805
Kemet
C0805C104K4RACTU
8
1
C12
1n, 250VAC ceramic capacitor Y2/X1
Thru-Hole
Panasonic
ECK-NVS102ME
9
1
D1
100V, 1W zener diode
DO-41
Micro Semi
1N4764ADO41
10
1
D2
100V, 1A schottky diode
SMA
Diodes Inc
B1100-13
11
1
D3
400V, 1A ultrafast switching diode
On Semi
MUR140RL
12
2
D4
150V, 400mA switching diode
SOD123
Diodes Inc
BAV20W-7
13
1
D7
9.1V, 500mW zener diode
SOD123
Diodes Inc
BZT52C9V1-7
14
1
D8
20V, 500mW zener diode
SOD123
Diodes Inc
BZT52C20-7-F
15
1
D9
1.24V, precision shunt regulator
SOT-23
National Semi
LMV431
16
1
F1
2A, 250VAC fuse
Thru-Hole
Cooper/Bussmann BK/PCB-2
17
1
L1
15mH (300uH differential),
0.15A rms common mode choke
Thru-Hole
Coilcraft
BU9HS-153R15B
18
1
Q1
40V, 600mA NPN transistor
SOT-23
ST Micro
MMBT2222A
19
1
Q3
600V, 1A N-Channel MOSFET
DPAK
ST Micro
STD1NK60T4
20
1
RT1
50ohm Inrush current limiter
Thru-Hole
GE Infrastructure
CL-140
21
2
R1, R2
1.78,1/4W, 1% chip resistor
SMD0805
Yageo
9C12063A1R78FGHFT
22
2
R3, R9
1k, 1/8W, 1% chip resistor
SMD0805
Yageo
9C08052A1001FKHFT
23
1
R5
226k, 1/8W, 1% chip resistor
SMD0805
Yageo
9C08052A2263FKHFT
24
1
R6
5.49k, 1/8W, 1% chip resistor
SMD0805
Yageo
9C08052A5491FKHFT
25
1
R7
20k, 1/8W, 1% chip resistor
SMD0805
Yageo
9C08052A2002FKHFT
26
1
R8
0.56, 1/8W, 1% chip resistor
SMD0805
Panasonic
ERJ-6RQFR56V
27
1
R11
97.6k, 1/8W, 1% chip resistor
SMD0805
Yageo
9C08052A9762FKHFT
28
1
R12
4.99k, 1/8W, 1% chip resistor
SMD0805
Yageo
9C08052A4991FKHFT
29
1
R13
100 ohm, 1/8W, 1% chip resistor
SMD0805
Yageo
9C08052A1000FKHFT
30
1
T1
Flyback Transformer
-
---
31
1
U1
400V, 1A Single Phase diode bridge rectifier
DF-S
Diodes Inc
DF04S
32
1
U2
Universal LED Driver
SO-16
Supertex
HV9910BNG-G
33
1
U3
Single Channel Optoisolator
4-DIP
Fairchild
H11A817A
DO-41
-
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
2
DN-H01
Flyback Transformer Details
Component
Description
Core :
EFD20/10/7 – 3C90 – A250 from Ferroxcube
(EFD 20 core with 160µm gap in the center leg)
Bobbin:
CPHS – EFD20 – 1S – 10P from Ferroxcube
Primary:
66 turns of AWG#32 magnet wire
Secondary:
13 turns of AWG#24 equivalent triple-insulated litz wire
Auxiliary:
32 turns of AWG#32 magnet wire
Insulation:
3M 1928 Polyester Film, 2.0 mil thick tape
Schematic Diagram of the Transformer
T1
1
10
Primary
O
O
O
Secondary
3
4
9
Auxiliary
5
LPRIMARY = 1.1mH ± 8%
Leakage inductance = 8% of LPRIMARY
Winding Diagram
Secondary Side
Primary Side
AWG #24
Pins 9, 10
Pins 6, 7
Pin 4
Pin 5
2 Layers Tape
2 Layers Tape
AWG #32
2 Layers Tape
Pin 3
Pin 1
AWG #32
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives an
adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability to the
replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and specifications
are subject to change without notice. For the latest product specifications refer to the Supertex inc. website: http//www.supertex.com.
©2009
022009
All rights reserved. Unauthorized use or reproduction is prohibited.
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
3
Selector Guide
LED Driver ICs
Switching Converters
Device
VIN
Topology
min
(V)
max
(V)
Output
Current
(mA)
Dimming
Package
Options
Demoboards
Application/
Design
Notes
PWM /
Linear
24-Lead TSSOP (TS)
---
---
Automotive (AEC-Q100 Certified) LED Drivers
AT9917
Boost, Sepic
5.3
40
External FET
AT9919
Buck
4.5
40
External FET
PWM
8-Lead DFN (K7)
AT9919DB1
---
24-Lead TSSOP (TS)
---
---
AT9932
Boost-Buck (Ćuk)
5.3
40
External FET
PWM /
Linear
AT9933
Boost-Buck (Ćuk)
9.0
75
External FET
PWM
8-Lead SOIC (LG)
AT9933DB1
AN-H51
AN-H58
General Purpose LED Drivers
HV9801A
Buck
15
450
External FET
4-Level
Switch
8-Lead SOIC (LG)
16-Lead SOIC (NG)
---
---
HV9861A
Buck
12
450
External FET
PWM /
Linear
8-Lead SOIC (LG)
16-Lead SOIC (NG)
HV9861ADB1
---
HV9910B
Buck
8.0
450
External FET
PWM /
Linear
8-Lead SOIC (LG)
16-Lead SOIC (NG)
HV9910BDB2
HV9910BDB3
HV9910BDB7
HV9910DB6
AN-H48
AN-H50
AN-H64
DN-H01
HV9918
Buck
4.5
40
Integrated FET
PWM
8-Lead DFN (K7)
HV9918DB1
---
HV9919B
Buck
4.5
40
External FET
PWM
8-Lead DFN (K7)
HV9919BDB1
---
HV9921
Buck
20
400
20
No
3-Lead TO-92 (N3)
3-Lead SOT-89 (N8)
HV9921DB1
---
HV9922
Buck
20
400
50
No
3-Lead TO-92 (N3)
3-Lead SOT-89 (N8)
HV9922DB1
HV9922DB2
DN-H02
DN-H03
HV9923
Buck
20
400
30
No
3-Lead TO-92 (N3)
3-Lead SOT-89 (N8)
HV9923DB1
---
HV9925
Buck
20
400
20 - 50
PWM
8-Lead SOIC (SG)
w/ Heat Slug
HV9925DB1
---
HV9930
Hysteric
8.0
200
External FET
PWM
8-Lead SOIC (LG)
HV9930DB1
HV9930DB2
AN-H51
AN-H58
AN-H52
DN-H04
DN-H05
DN-H06
HV9931
Single-Switch PFC
8.0
450
External FET
PWM
8-Lead SOIC (LG)
HV9931DB1v2
HV9931DB2v1
HV9931DB5
HV9961
Buck
8.0
450
External FET
PWM /
Linear
8-Lead SOIC (LG)
16-Lead SOIC (NG)
HV9961DB1
AN-H64
HV9967B
Buck
8.0
60
External FET
PWM /
Linear
8-Lead DFN (K7)
8-Lead MSOP (MG)
---
---
HV9971
Flyback
-
-
External FET
PWM
8-Lead SOIC (LG)
HV9971DB1
---
Backlight LED Drivers
HV9860
Boost
10
40
External FET
PWM
16-Lead SOIC (NG)
HV9860DB1
---
HV9861A
Buck
12
450
External FET
PWM /
Linear
8-Lead SOIC (LG)
16-Lead SOIC (NG)
HV9861ADB1
---
HV9911
Boost, Sepic,
Buck-Boost
9.0
250
External FET
PWM
16-Lead SOIC (NG)
HV9911DB1v2
HV9911DB2
HV9911DB3
HV9911DB4
AN-H55
HV9912
Boost, Sepic,
Buck-Boost
9.0
100
External FET
PWM
16-Lead SOIC (NG)
HV9912DB1
---
HV9957
Boost
2.7
28
30 x 6-Channel
PWM
24-Lead QFN (K7)
HV9957DB1
---
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: (408) 222-8888 ● www.supertex.com
Selector Guide
LED Driver ICs
Switching Converters
Device
VIN
Topology
min
(V)
max
(V)
Output
Current
Dimming
Package
Options
Demoboards
Application/
Design
Notes
(mA)
Backlight LED Drivers (cont.)
HV9961
Buck
8.0
450
External FET
PWM /
Linear
8-Lead SOIC (LG)
16-Lead SOIC (NG)
HV9961DB1
AN-H64
HV9963
Boost, Sepic,
Buck-Boost
8.0
40
External FET
PWM /
Linear
16-Lead SOIC (NG)
---
---
HV9967B
Buck
8.0
60
External FET
PWM /
Linear
8-Lead DFN (K7)
8-Lead MSOP (MG)
---
---
HV9980
Buck
100
160
70
PWM /
Linear
24-Lead SOW (WG)
HV9980DB1
---
HV9982
Boost, SEPIC
10
40
External FET
PWM /
Linear
40-Lead QFN (K6)
HV9982DB1
---
HV9985
Boost, SEPIC
10
40
External FET
PWM /
Linear
40-Lead QFN (K6)
44-Lead QSOP (QP)
HV9985DB1
---
HV9986
Boost, SEPIC
10
40
External FET
PWM /
Linear
40-Lead QFN (K6)
---
---
HV9989
Boost, SEPIC
10
40
External FET
PWM /
Linear
40-Lead QFN (K6)
---
---
8-Lead SOIC (LG)
HV9931DB1v2
HV9931DB2v1
HV9931DB5
AN-H52
DN-H04
DN-H05
DN-H06
Offline PFC
Buck, BIBRED
HV9931
8.0
450
External FET
PWM
Linear Regulators
VOUT
VIN
min
max
min
max
Output
Current
Dimming
Parallelable
Package
Options
Features
CL2
5.0
90
5.0
90
20
External
FET
Yes
3-Lead TO-252 (K4)
3-Lead TO-92 (N3)
3-Lead SOT-89 (N8)
---
CL25
5.0
90
5.0
90
25
External
FET
Yes
3-Lead TO-92 (N3)
3-Lead SOT-89 (N8)
---
CL220
5.0
220
5.0
220
20
External
FET
Yes
3-Lead TO-252 (K4)
3-Lead TO-220 (N5)
---
CL320
6.5
90
4.0
90
20
PWM
Yes
8-Lead SOIC (SG)
w/ Heat Slug
OTP, separate ENABLE pin
CL325
6.5
90
4.0
90
25
PWM
Yes
8-Lead SOIC (SG)
w/ Heat Slug
OTP, separate ENABLE pin
CL330
6.5
90
4.0
90
30
PWM
Yes
8-Lead SOIC (SG)
w/ Heat Slug
OTP, separate ENABLE pin
CL520
4.75
90
1.0
90
20
-
Yes
3-Lead TO-252 (K4)
3-Lead TO-92 (N3)
---
CL525
4.75
90
1.0
90
25
-
Yes
3-Lead TO-252 (K4)
3-Lead TO-92 (N3)
---
CL6
6.5
90
4.0
90
100
No
Yes
3-Lead TO-252 (K4)
3-Lead TO-220 (N5)
Reverse polarity protection, OTP
CL7
6.5
90
4.0
90
100
PWM
Yes
8-Lead SOIC (SG)
w/ Heat Slug
Reverse polarity protection, OTP
Device
(V)
(V)
(V)
(V)
(mA)
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: (408) 222-8888 ● www.supertex.com