ETC HV9931

HV9931
HV9931 Unity Power Factor LED Lamp Driver
Features









General Description
The HV9931 is a ﬁxed frequency PWM controller IC designed
to control an LED lamp driver using a single-stage PFC
buckboost-buck topology. It can achieve a unity power factor
and a very high step-down ratio that enables driving a single
high-brightness LED from the 85-264VAC input without a
need for a power transformer. This topology allows reducing
the ﬁlter capacitors and using non-electrolytic capacitors to
improve reliability. The HV9931 uses open-loop peak current
control to regulate both the input and the output current. This
control technique eliminates a need for loop compensation,
limits the input inrush current, and is inherently protected from
input under-voltage condition.
Constant output current
Large step-down ratio
Unity power factor
Low input current harmonic distortion
Fixed frequency or ﬁxed off-time operation
Internal 450V linear regulator
Input and output current sensing
Input current limit
Enable, PWM and phase dimming
Applications




Capacitive isolation protects the LED Lamp from failure of the
switching MOSFET. HV9931 provides a low-frequency PWM
dimming input that can accept an external control signal with a
duty ratio of 0-100% and a frequency of up to a few kilohertz.
The PWM dimming capability enables HV9931 phase control
solutions that can work with standard wall dimmers.
Ofﬂine LED lamps and ﬁxtures
Street lamps
Trafﬁc signals
Decorative lighting
Typical Application Circuit
D4
VIN
D1
L1
C1
L2
D2
~AC
~AC
Rref1
D3
Q1
CIN
VO
RS2
RS1
RCS2
RCS1
VIN
GATE
RT
+
Rref2
RT
PWMD
CS1
CS2
GND
VDD
C2
HV9931
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HV9931
Ordering Information
8-Lead SOIC (Narrow Body)
Device
4.90x3.90mm body
1.75mm height (max)
1.27mm pitch
HV9931
HV9931LG-G
-G indicates package is RoHS compliant (‘Green’)
Pin Conﬁguration
Absolute Maximum Ratings
Parameter
Value
VIN to GND
-0.5V to +470V
VDD to GND
-0.3V to +13.5V
CS1, CS2, PWMD, GATE, RT to GND
Operating temperature range
-0.3V to (VDD +0.3V)
-40°C to +85°C
Storage temperature range
VIN
1
8
RT
CS1
2
7
CS2
GND
3
6
VDD
GATE
4
5
PWMD
8-Lead SOIC (LG)
-65°C to +150°C
Continuous power dissipation (TA = +25°C)
(top view)
630mW
Stresses beyond those listed under “Absolute Maximum Ratings” may cause
permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated
in the operational sections of the speciﬁcations is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device
reliability.
Product Marking
Y = Year Sealed
WW = Week Sealed
L = Lot Number
= “Green” Packaging
YWW
H9931
LLLL
Thermal Resistance
8-Lead SOIC (LG)
Package
θja
8-Lead SOIC
128OC/W
Electrical Characteristics
(The * denotes the speciﬁcations which apply over the full operating junction temperature range of
-40°C < TA < +85°C, otherwise the speciﬁcations are at TA = 25°C, VIN = 12V, unless otherwise noted)
Sym
Parameter
Min
Typ
Max
Units
Conditions
VINDC
Input DC supply voltage range*
8.0
-
450
V
IINSD
Shut-down mode supply current*
-
0.5
1.0
mA
PWMD connected to GND
7.12
7.50
7.88
V
VIN = 8.0, IDD(EXT) = 0,
GATE = 500pF, RT = 226KΩ
0
-
1.0
V
VIN = 8.0 - 450V, IDD(ext) = 0,
GATE = 500pF, RT = 226kΩ,
VDD rising
Input
DC input voltage
Internal Regulator
VDD
ΔVDD, line
Internally regulated voltage
Line regulation of VDD
UVLO
VDD undervoltage lockout threshold
6.45
6.70
6.95
V
∆UVLO
VDD undervoltage lockout hysteresis
-
500
-
mV
---
PWM Dimming
VPWMD(lo)
PWMD input low voltage
-
-
1.0
V
VIN = 8.0 - 450V
VPWMD(hi)
PWMD input high voltage
2.4
-
-
V
VIN = 8.0 - 450V
PWMD pull-down resistance
50
100
150
kΩ
RPWMD
VPWMD = 5.0V
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2
HV9931
Electrical Characteristics (cont.) (The * denotes the speciﬁcations which apply over the full operating junction temperature
range of -40°C < TA < +85°C, otherwise the speciﬁcations are at TA = 25°C, VIN = 12V, unless otherwise noted)
Sym
Parameter
Min
Typ
Max
Units
Conditions
GATE
VGATE(hi)
GATE high output voltage*
VDD -0.3
-
VDD
V
IGATE = 10mA, VDD = 7.5V,
VIN open
VGATE(lo)
GATE low output voltage*
0
-
0.3
V
IGATE = -10mA, VDD = 7.5V,
VIN open
TRISE
GATE output rise time
-
30
50
ns
CGATE = 500pF, VDD = 7.5V,
VIN open
TFALL
GATE output fall time
-
30
50
ns
CGATE = 500pF, VDD = 7.5V,
VIN open
TDELAY
Delay from CS trip to GATE
-
150
300
ns
VCS1, VCS2 = -100mV
TBLANK
Blanking delay
150
215
280
ns
VCS1, VCS2 = -100mV
Oscillator frequency
80
100
120
kHz
RT = 226KΩ
-15
-
15
mV
---
Oscillator
FOSC
Comparators
VOFFSET1
VOFFSET2
Comparator input offset voltage*
Functional Block Diagram
VIN
Regulator
VDD
7.5V
Osc
CS1
Leading
Edge
Blanking
RT
S
R Q
GATE
CS2
AGND
PWMD
HV9931
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3
HV9931
Functional Description
Power Topology
The HV9931 is optimized to drive Supertex’s proprietary
single-stage, single-switch, non-isolated topology, cascading
an input power factor correction (PFC) buck-boost stage
and an output buck converter power stage. This power
converter topology offers numerous advantages useful
for driving high-brightness light emitting diodes (HB LED).
These advantages include unity power factor, low harmonic
distortion of the input AC line current, and low output current
ripple. The output load is decoupled from the input voltage
with a capacitor making the driver inherently failure-safe for
the output load. The power converter topology also permits
reducing the size of a ﬁlter capacitor needed, enabling use
of non-electrolytic capacitors. The latter advantage greatly
improves reliability of the overall solution.
The HV9931 is a peak current-mode controller that is
speciﬁcally designed to drive a constant current buckboost-buck power converter. This patent pending control
scheme features two identical current sense comparators
for detecting negative current signal levels. One of the
comparators regulates the output LED current, while the
other is used for sensing the input inductor current. The
second comparator is mainly responsible for the converter
start-up. The control scheme inherently features low inrush
current and input under-voltage protection. The HV9931 can
operate with programmable constant frequency or constant
off-time. In many cases, the constant off-time operating mode
is preferred, since it improves line regulation of the output
current, reduces voltage stress of the power components
and simpliﬁes regulatory EMI compliance. (See Application
Note AN-H52.)
Input Voltage Regulator
The HV9931 can be powered directly from its VIN pin, and
takes a voltage from 8V to 450V. When a voltage is applied
at the VIN pin, the HV9931 seeks to maintain a constant
7.5V at the VDD pin. The VDD voltage can be also used as a
reference for the current sense comparators. The regulator
is equipped with an under-voltage protection circuit which
shuts off the HV9931 when the voltage at the VDD pin falls
below 6.2V.
The VDD pin must be bypassed by a low ESR capacitor
(≥ 0.1µF) to provide a low impedance path for the high
frequency current of the output GATE driver.
The HV9931 can also be operated by supplying a voltage
at the VDD pin greater than the internally regulated voltage.
This will turn off the internal linear regulator and the HV9931
will function by drawing power from the external voltage
source connected to the VDD pin.
PWM Dimming and Wall Dimmer Compatibility
PWM Dimming can be achieved by applying a TTLcompatible square wave signal at the PWMD pin. When the
PWMD pin is pulled high, the GATE driver is enabled and the
circuit operates normally. When the PWMD pin is left open
or connected to GND, the GATE driver is disabled and the
external MOSFET turns off. The HV9931 is designed so that
the signal at the PWMD pin inhibits the driver only, and the
IC need not go through the entire start-up cycle each time
ensuring a quick response time for the output current.
The power topology requires little ﬁlter capacitance at
the output, since the output current of the buck stage is
continuous, and since AC line ﬁltering is accomplished
through the middle capacitor rather than the output one.
Therefore, disabling the HV9931 via its PWMD or VIN pins
can interrupt the output LED current in accordance with
the phase-controlled voltage waveform of a standard wall
dimmer.
Oscillator
Connecting an external resistor from RT pin to GND programs
switching frequency:
FS [kHz ] =
25000
RT [K Ω ]+ 22
Connecting the resistor from the RT pin to the GATE
programs constant off-time:
TOFF [µ s ] =
RT [K Ω ] + 22
25
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4
HV9931
Input and Output Current Feedback
Two current sense comparators are included in the HV9931.
Both comparators have their non-inverting inputs internally
connected to ground (GND). The CS1 and CS2 inputs are
inverting inputs of the comparators. Connecting a resistor
divider into either of these inputs from a positive reference
voltage and a negative current sense signal programs the
current sense threshold of the comparator. The VDD voltage
of the HV9931 can be used as the reference voltage. If more
accuracy is needed, an external reference voltage can be
applied. When either the CS1 or the CS2 pin voltage falls
below GND, the GATE pulse is terminated. A leading edge
blanking delay of 215ns (typ) is added. The GATE voltage
becomes high again upon receiving the next clock pulse of
the oscillator circuit.
Referring to the Functional Circuit Diagram, the CS2
comparator is responsible for regulating output current. The
output LED current can be programmed using the following
equation:
RCS 2 =
1
∆ I L2
2
⋅ RREF 2 ⋅ RS 2
7.5V
Io +
where ∆IL2 is the peak-to-peak current ripple in L2. The CS1
comparator limits the current in the input inductor L1. There
is no charge in the capacitor C1 upon the start-up of the
converter. Therefore, L2 cannot develop the output current,
and the HV9931 starts-up in the input current limiting mode.
The CS1 current threshold must be programmed such that no
input current limiting occurs in normal steady-state operation.
The CS1 threshold can be programmed in accordance with
a similar equation:
RCS 1 =
I L1( PK )
7.5V
⋅ RREF 1 ⋅ RS 1
where IL1(PK) is the maximum peak current in L1.
MOSFET Gate Driver
Typically, the GATE driving capability of the HV9931 is limited
by the amount of power dissipation in its linear regulator.
Thus, care must be taken selecting a switching MOSFET
to be used in the circuit. An optimal trade-off must be found
between the GATE charge and the on-resistance of the
MOSFET to minimize the input regulator current.
Switching Waveform
GATE
VDD
0
t
0
t
iL2
iL1
0
t
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5
HV9931
Functional Circuit Diagram
D1
L1
D4
VIN
iL1
CIN
~AC
+
VC1
L2
D2
_
-
iL2
D3
Q1
~AC
RCS1
C1
RS1
VO
RS2
_ VS1 +
+
RT
VS2
+
_
RCS2
PWMD
GATE
RT
OSC
S Q
R
CS2
CS1
Rref1
Rref2
RE G
VIN
7.5V
VDD
GND
HV9931
CDD
Pin Description
Pin #
Pin Name
Description
1
VIN
This pin is the input of a high voltage regulator.
2
CS1
This pin is used to sense the input and output currents of the converter. It is the inverting input
of the internal comparator.
3
GND
Ground return for all the internal circuitry. This pin must be electrically connected to the ground
of the power train.
4
GATE
This pin is the output GATE driver for an external N-channel power MOSFET.
5
PWMD
When this pin is pulled to GND, switching of the HV9931 is disabled. When the PWMD pin
is released, or external TTL high level is applied to it, switching will resume. This feature is
provided for applications that require PWM dimming of the LED lamp.
6
VDD
This is a power supply pin for all internal circuits. It must be bypassed with a low ESR capacitor
to GND.
7
CS2
This pin is used to sense the input and output currents of the converter. It is the inverting input
of the internal comparator.
8
RT
Oscillator control. A resistor connected between this pin and GND sets the PWM frequency. A
resistor connected between this pin and GATE sets the PWM off-time.
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6
HV9931
8-Lead SOIC (Narrow Body) Package Outline (LG)
4.90x3.90mm body, 1.75mm height (max), 1.27mm pitch
D
θ1
8
E
E1
L2
Note 1
(Index Area
D/2 x E1/2)
L
1
Top View
View B
A
Note 1
A
θ
L1
Seating
Plane
View B
h
h
A2
Gauge
Plane
Seating
Plane
b
e
A1
A
Side View
View A-A
Note:
1. This chamfer feature is optional. A Pin 1 identiﬁer must be located in the index area indicated. The Pin 1 identiﬁer can be: a molded mark/identiﬁer;
an embedded metal marker; or a printed indicator.
Symbol
Dimension
(mm)
A
A1
A2
b
MIN
1.35*
0.10
1.25
0.31
NOM
-
-
-
-
MAX
1.75
0.25
1.65*
0.51
D
E
E1
4.80* 5.80* 3.80*
4.90
6.00
3.90
5.00* 6.20* 4.00*
e
1.27
BSC
h
L
0.25
0.40
-
-
0.50
1.27
L1
1.04
REF
L2
0.25
BSC
θ
θ1
0O
5O
-
-
8O
15O
JEDEC Registration MS-012, Variation AA, Issue E, Sept. 2005.
* This dimension is not speciﬁed in the original JEDEC drawing. The value listed is for reference only.
Drawings are not to scale.
Supertex Doc. #: DSPD-8SOLGTG, Version H101708.
(The package drawing(s) in this data sheet may not reﬂect the most current speciﬁcations. For the latest package outline
information go to http://www.supertex.com/packaging.html.)
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives an
adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability to the
replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and specifications
are subject to change without notice. For the latest product specifications refer to the Supertex inc. website: http//www.supertex.com.
©2008
Doc.# DSFP-HV9931
A102108
All rights reserved. Unauthorized use or reproduction is prohibited.
7
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Tel: 408-222-8888
www.supertex.com
Supertex inc.
HV9931DB2v1
LED Driver Demo Board
Input 230VAC // Output 350mA, 40V (14W)
General Description
The HV9931 LED driver is primarily targeted at low to
medium power LED lighting applications where galvanic
isolation of the LED string is not an essential requirement.
The driver provides near unity power factor and constant
current regulation using a two stage topology driven by
a single MOSFET and control IC. Triac dimming of this
design is possible with the addition of some components for
preloading and inrush current shaping.
The DB1 and DB2 demo boards were designed for a ﬁxed
string current of 350mA and a string voltage of 40V for a load
power of about 14W. The boards will regulate current for an
output voltage down to 0V.
Nominal input voltage for the DB1 is 120VAC, for the DB2
230VAC. Design for universal input (85 to 265VAC) is by
all means possible but does increase cost and size while
lowering efﬁciency.
The input EMI ﬁlter was designed to suppress the differential
mode switching noise to meet CISPR15 requirements.
No speciﬁc components were added to suppress currents
of common mode nature. Common mode current can be
controlled in many ways to satisfy CISPR 15 requirements.
featured are output current soft start and protections from
line overvoltage, load overvoltage and open circuit. The
driver is inherently short circuit proof by virtue of the peak
current regulation method.
Speciﬁcations
Input voltage:
200VRMS to 265VRMS, 50Hz
Output voltage:
0 to 40V
Output current:
350mA +/-5%
Output power:
14W
Power factor
98%
Total harmonic distortion
EN61000-3-2 Class C
EMI limits
CISPR 15 (see text)
Efﬁciency
83%
Output current ripple
30%PP
Input overvoltage protection
265VRMS, Non-Latching
Output overvoltage protection 46V, Latching
Switching frequency
80kHzNOM
Dimensions:
3.5” x 3.0” x 1.25”
The board is ﬁtted with a number of optional circuits; a
schematic of a simpliﬁed driver is given as well. The circuits
Board Layout and Connections
A
V
V
A
Supertex inc.
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1
HV9931DB2v1
 Warning!
Working with this board can cause serious bodily harm or
death. Connecting the board to a source of line voltage will
result in the presence of hazardous voltage throughout the
system including the LED load.
The board should only be handled by persons well aware of
the dangers involved with working on live electrical equipment. Extreme care should be taken to protect against electric shock. Disconnect the board before attempting to make
any changes to the system conﬁguration. Always work with
another person nearby who can offer assistance in case of
an emergency. Wear safety glasses for eye protection.
Special Note:
The electrolytic capacitor carries a hazardous voltage for an
extended time after the board is disconnected. The board
includes a 1MΩ resistor placed across the electrolytic capacitor which will slowly discharge the capacitor after disconnection from line voltage. The voltage will fall more or
less exponentially to zero with a time constant of about 100
seconds. Check the capacitor voltage before handling the
board.
Connection Instructions
Step 1.
Carefully inspect the board for shipping damage, loose
components, etc, before making connections.
Step 2.
Attach the board to the line and load as shown in the diagram.
Be sure to check for correct polarity when connecting the
LED string to avoid damage to the string. The board is short
circuit and open circuit proof. The LED string voltage can
be anything between zero and 40V, though performance will
suffer when the string voltage is substantially lower than the
target of 40V. See the typical performance graphs.
voltage and LED string voltage are more or less constant
as well. Duty cycle and bus voltage do adjust in response to
changes in line or load voltage but are otherwise constant
over the course of a line cycle. With the HV9931, OFF time is
ﬁxed by design, being programmed by an external resistor,
whereas ON time adjusts to a more or less constant value,
being under control of the HV9931 peak current regulator.
Principles of Operation
The input or buck-boost stage is designed for operation
in discontinuous conduction mode (DCM) throughout the
range of line and load voltage anticipated. This can be
accomplished by making the input inductor sufﬁciently small.
A well known property of the DCM buck-boost stage, when
operated with constant ON time and constant OFF time, is
that input current is proportional to input voltage, whether
in peak value or average value. This results in sinusoidal
input current when the input voltage is sinusoidal, thereby
giving unity power factor operation when operating from the
rectiﬁed AC line voltage.
The output or buck stage is designed for operation in
continuous conduction mode (CCM), operating with about 20
to 30% inductor current ripple. This amount of ripple serves
the needs of the HV9931 peak current controller which relies
on a sloping inductor current for setting ON time, and is of an
acceptable level to high brightness LEDs. Duty cycle is more
or less constant throughout the line cycle as the DC bus
When operated in the anticipated range of line and load
voltage, the MOSFET ON time will be under control of the
output stage current controller, which turns the MOSFET
off when sensing that the output inductor current has
reached the desired peak current level as programmed by
a resistive divider at the CS2 pin. Under certain abnormal
circumstances such as initial run-up and line undervoltage,
which both could lead to the draw of abnormally high line
current, ON time is further curtailed by the action of the CS1
comparator, which monitors the input stage inductor current
against a threshold. This threshold can be a simple DC level
or be shaped in time as is performed on the demo board. In
particular, when shaping the CS1 threshold with the shape of
the rectiﬁed AC line input voltage waveform, the line current
will be bounded by a more or less sinusoidal line current
envelope which results in sinusoidal input current for low line
and other abnormal conditions.
Step 3.
Energize the mains supply. The board can be connected to
mains directly. Alternatively voltage can be raised gradually
from zero to full line voltage with the aid of an adjustable AC
supply such as a Variac or a programmable AC source.
The HV9931 topology can be viewed as a series connection
of two basic power supply topologies, (1) a buck-boost
stage as ﬁrst or input stage, for purpose of converting AC
line power into a source of DC power, commonly known as
the DC bus, having sufﬁcient capacitive energy storage to
maintain the bus voltage more or less constant throughout
the AC line cycle, and (2) a buck stage as second or output
stage for powering the LED string, stepping down the DC
bus voltage to the LED string voltage in order to produce a
steady LED string current.
Supertex inc.
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2
HV9931DB2v1
The design exercise of an HV9931 LED driver revolves
around establishing component values for (1) the input and
output stage inductors, (2) a value for the bus capacitor, and
(3) a value for switching cycle OFF time, which together
result in (1) acceptable current ripple at the output stage
(say 30%), (2) an acceptable bus voltage ripple (say 5%),
and (3) an input stage which maintains DCM operation over
the desired line and load voltage range.
For a given HV9931 design, the bus voltage rises and falls
with like changes in line and load voltage. This is unlike a
two stage design having two transistors and control ICs,
where the bus voltage can be set independent of line and
load voltage variation. If the desired ranges of line and load
voltage are particularly large then the latter topology may be
preferable so as to avoid large variation in bus voltage.
The design of an HV9931 based LED driver is not further
discussed here, except for noting that a semi-automatic
design tool is available in Mathcad form, based on behavioral
Simpliﬁed Schematic Diagram
F11
250mA
AC2
L21
2.2mH
L11
2.2mH
C11
47nF
1
A Simpliﬁed Version of the Design
The demo board can be simpliﬁed signiﬁcantly. Below is a
schematic showing the essential elements of the driver.
Contact Supertex Applications Engineering for guidance in
simplifying the design or for adding functions such as triac
dimmability.
D32
STTH108A
L31
1.2mH
E31
22μF
D31
STTH108A
+
R37
6.8kΩ
C21
47nF
D42
STTH1R06A
M31
SPA02N80C3
R51
205kΩ
1
R62
2.43kΩ
THROV
BT168GW
ZOV
BZX84C43
ANO
A
R61
270mΩ
C
ROV
10kΩ
CAT
C37
100pF
4
Optional Output
Overvoltage Protection
L41
3.9mH
D41
STTH1R06A
C
BR11
RH06-T
2
Mathcad design data can be found at the end of this
document. The data tends to be in good agreement with the
actual demo board despite the omission of switching losses
in the model. For this design we can see that the calculated
efﬁciency is off by say 5 percent likely due underestimation
of switching losses and inductor core and winding losses.
3
C12
47nF
AC1
simulation, which, allows components to be adjusted in an
iterative manner, starting from an initial guess. The tool allows
quick evaluation of nine standard test cases, exercising the
design over line voltage variation and tolerance variation of
three component parameters.
VIN
2
R68
75kΩ
4
8
RT
GATE
IC51
CS1
R71
680mΩ
R72
2.67kΩ
CS2
HV9931LG
GND
VDD
PWM
3
6
5
7
R73
75kΩ
A
C51
10µF
Note on Inductors:
This board was ﬁtted with standard (COTS) inductors. These
are not necessarily an optimal choice but present an expedient way to go when evaluating a design. Custom engineered
parts generally give better performance, particularly with respect to efﬁciency.
Drum core style inductors, whether in radial or axial leaded
versions, are popular for their ready availability and low cost.
Drum core styles have particularly simple construction and
Supertex inc.
can be wound for lowest cost without coil former (bobbin).
They may serve well during the development stage, but may
not be the best choice for ﬁnal design. Keep these type of
inductors away form any metallic surface such as heatsinks,
PCB copper planes, metallic enclosures, and capacitors, as
these unshielded parts can create high eddy current losses
in these parts. For tightly packaged designs or where inductor losses are an issue, drum core style inductors are not
recommended.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
3
Supertex inc.
AC1
AC2
F11
250mA
R82
13.0kΩ
R83
1MΩ
R84
1MΩ
C12
47nF
Q82
MMBT2907A
MOV11 C11
430V
47nF
L11
2.2mH
C81
10nF
2
TVS11
SMAJ
440CA
1
R80
200kΩ
Q81
MMBT2222A
R81
10kΩ
3
1
REC
BR11
RH06-T
DN65
BAV99
C65
10µF
2
4
3
L21
2.2mH
L1D
R68
1MΩ
R88
10MΩ
R87
200kΩ
2
R37
6.8kΩ
IC51
Q84
MMBT2907A
VDD
6
3
C51
10µF
HV9931
VDD
ENA
R51
205kΩ
GATE
4
GATE
CS2
8
5
PWM
RT
D42
MMDB914
7
L41
3.9mH
R90
200kΩ
C72
100pF
R79
100Ω
D42
STTH1R06A
SN2
D79
MMBD914
D41
STTH1R06A
M31
SPA02N80C3
R31
1MΩ
+
E31
22μF
GND
CS1
VIN
1
IDD
R39
100Ω
C37
100pF
D31
STTH108A
R99
1kΩ
C62
100pF
R62
2.43kΩ
R86
100kΩ
Q83
MMBT2222A
R85
100kΩ
Z61
BZX84C7V5
R64
1.3MΩ
R63
75kΩ
R65
1.3MΩ
R61
270mΩ
RS1
C21
47nF
D37
STTH108A
L31
1.2mH
D32
STTH108A
R73
75kΩ
R72
2.67kΩ
R71
680mΩ
RS2
C41
10nF
Z90
BZX84C7V5
Z91
BZX84C47
GND2
GND1
ANO
CAT
HV9931DB2v1
Schematic Diagram
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
4
HV9931DB2v1
Typical Characteristics
String Current [mA] vs. String Voltage [V]
1000
100
900
90
800
80
70
700
135VRMS
600
60
120VRMS
500
50
400
40
300
30
200
0
10
(100VRMS, 120VRMS, 135VRMS)
virtually the same
20
100VRMS
10
100
0
Efficiency [%] vs. String Voltage [V]
20
30
40
50
0
0
10
20
30
40
50
THD [%] vs. String Voltage [V]
PF [%] vs. String Voltage [V]
30
100
90
25
80
135VRMS
70
20
120VRMS
60
100VRMS
15
50
40
100VRMS
10
30
20
120VRMS
135VRMS
5
10
0
0
10
20
30
Supertex inc.
40
50
0
0
10
20
30
40
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
5
50
HV9931DB2v1
Typical Waveforms (1)
Line Voltage and Current at nominal load (350mA, 40V)
200VRMS
230VRMS
265VRMS
IAC
VAC
Line Voltage and Current at half load (350mA, 20V)
200VRMS
230VRMS
265VRMS
Output Current and Drain Voltage at nominal load (350mA, 40V)
VDRAIN
ILED (Peak)
ILED (Valley)
Output Current and Drain Voltage at half load (350mA, 20V)
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
6
HV9931DB2v1
Typical Waveforms (2) (120VRMS, 40V, 350mA)
Drain Voltage and LED Current
40µs per div
400µs per div
4µs per div
350mAAVE
ILED
VDRAIN
Drain Voltage and Gate Voltage
40ns per div
4µs per div
40ns per div
VG @ IC51
VGATE
VG @ M31
Turn-ON
VDRAIN
Turn-OFF
Recovery of D41
Recovery of D42
Drain Voltage and Current Sense Voltages of Stages 1 and 2
VRS1
Recovery of D42
VRS2
VDRAIN
Recovery of D41
Drain Voltage and Voltages at Test Points REC, SN3, SN2
VREC
Supertex inc.
VSN3
VSN2
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
7
HV9931DB2v1
Typical Waveforms (3) (120VRMS, 40V, 350mA)
Drain Voltage and Voltage at the Test Point L1D (3 points along the AC line cycle)
AT ~ 90°
AT ~ 30°
AT ~ 10°
Clamping action of D37
VDRAIN
VL1D
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
8
HV9931DB2v1
EMI Signature
Board suspended about 3” above reference plane.
Limit Line:
Detector:
IF Bandwidth:
Shielding:
CISPR 15 Quasi Peak (9kHz to 30MHz)
Peak Hold
9kHz
2 copper shields, surrounding the power section on top and bottom of the board, terminated at the source
of the MOSFET.
Without shielding :
110dBµV
100dBµV
90
80
66
60
56
50dBµV
10kHz
100kHz
10MHz
1MHz
With shielding :
110dBµV
100dBµV
90
80
66
60
56
50dBµV
10kHz
100kHz
The performance graphs above were obtained from the
board not having speciﬁc measures to suppress common
mode emissions, such as inclusion of a common mode inductor in the AC line input circuitry. The above graphs show
how shielding can signiﬁcantly reduce emissions, particu-
Supertex inc.
1MHz
10MHz
larly in the upper frequency range. The shielding also was
instrumental in reducing the lower frequency emissions by
reducing magnetic ﬁeld coupling from the main inductors
to the EMI ﬁlter inductors (EMI ﬁlter section kept outside of
shielded area).
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
9
HV9931DB2v1
Mathcad Design Data
Corner
x
0
0
1
2
3
4
5
6
7
8
Corner
L1
uH
0
616
560
504
616
560
504
616
560
504
L1
-
-
-
-
RL1
mR
0
2000
2000
2000
2000
2000
2000
2000
2000
2000
RL1
-
-
-
-
L2
mH
0
3300
3300
3300
3300
3300
3300
3300
3300
3300
L2
-
-
-
-
RL2
mR
0
3000
3000
3000
3000
3000
3000
3000
3000
3000
RL2
-
-
-
-
ILRF2
%
0
32
32
32
32
32
32
32
32
32
ILRF2
-
-
-
-
C2
uF
0.0
37.6
47.0
56.4
37.6
47.0
56.4
37.6
47.0
56.4
C2
-
-
-
-
NF
x
0
2
2
2
2
2
2
2
2
2
NF
-
-
-
-
LF
uH
0
1000
1000
1000
1000
1000
1000
1000
1000
1000
LF
-
-
-
-
RLF
mR
0
2000
2000
2000
2000
2000
2000
2000
2000
2000
RLF
-
-
-
-
CF
nF
0
100
100
100
100
100
100
100
100
100
CF
-
-
-
-
C1
nF
0
100
100
100
100
100
100
100
100
100
C1
-
C2V
135
-
RS
mR
0
1000
1000
1000
1000
1000
1000
1000
1000
1000
RS
-
C2R
1345
-
VD
mV
0
1000
1000
1000
1000
1000
1000
1000
1000
1000
VD
-
-
-
-
TF
us
0.0
8.7
8.7
8.7
8.7
8.7
8.7
8.7
8.7
8.7
TF
-
-
-
-
RT
kR
0
196
196
196
196
196
196
196
196
196
RT
-
-
-
-
FM
Hz
0
50
50
50
50
60
50
50
50
50
FM
-
-
-
-
VMRMS
V
0
100
100
100
120
120
120
135
135
135
VMRMS
-
-
-
-
IMRMS
mA
0
167
160
153
137
133
130
122
118
115
IMRMS
130
137
115
167
IMMAX
mA
0
246
232
221
200
191
185
176
169
163
IMMAX
185
200
163
246
V3AVG
V
0
40
40
40
40
40
40
40
40
40
V3AVG
40
40
40
40
I3AVG
mA
0
361
350
335
361
350
339
361
350
339
I3AVG
339
361
335
361
PM
W
0.0
16.5
15.9
15.3
16.3
15.8
15.5
16.2
15.8
15.3
PM
15.5
16.3
15.3
16.5
P3
W
0.0
14.4
14.0
13.4
14.4
14.0
13.6
14.4
14.0
13.6
P3
13.6
14.4
13.4
14.4
EFF
%
0.0
87.5
88.0
87.8
88.7
88.5
87.7
88.9
88.7
88.3
EFF
87.7
88.7
87.5
88.9
PF
%
0.0
98.7
99.3
99.6
98.9
99.3
99.5
98.8
99.1
99.3
PF
98.9
99.5
98.7
99.6
THD
%
0.0
9.0
5.3
3.3
6.4
3.8
2.5
5.1
3.1
2.1
THD
2.5
6.4
2.1
9.0
H3
%
0.0
8.7
5.1
3.1
6.2
3.6
2.3
5.0
2.9
1.9
H3
2.3
6.2
1.9
8.7
H5
%
0.0
1.7
1.0
0.7
1.1
0.7
0.6
0.8
0.6
0.5
H5
0.6
1.1
0.5
1.7
TAMIN
us
0.0
4.6
4.8
4.8
3.7
3.9
3.9
3.2
3.4
3.4
TAMIN
3.7
3.9
3.2
4.8
TAMAX
us
0.0
5.8
5.4
5.2
4.3
4.2
4.2
3.7
3.6
3.6
TAMAX
4.2
4.3
3.6
5.8
TFMIN
us
0.0
7.0
8.7
10.5
7.0
8.7
10.5
7.0
8.7
10.5
TFMIN
7.0
10.5
7.0
10.5
TFMAX
us
0.0
7.0
8.7
10.5
7.0
8.7
10.5
7.0
8.7
10.5
TFMAX
7.0
10.5
7.0
10.5
DAMIN
%
0.0
39.6
35.5
31.6
34.7
30.8
27.4
31.8
28.0
24.7
DAMIN
27.4
34.7
24.7
39.6
DAMAX
%
0.0
45.3
38.4
33.1
38.3
32.6
28.4
34.5
29.4
25.5
DAMAX
28.4
38.3
25.5
45.3
DC1MAX
%
0.0
98.6
79.7
65.2
87.1
70.0
57.7
80.4
64.3
52.4
DC1MAX
57.7
87.1
52.4
98.6
FSMIN
kHz
0.0
78.4
70.6
63.9
88.4
77.3
68.4
93.9
81.0
71.2
FSMIN
68.4
88.4
63.9
93.9
FSMAX
kHz
0.0
86.5
74.0
65.4
93.6
79.4
69.4
97.8
82.6
71.9
FSMAX
69.4
93.6
65.4
97.8
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
10
HV9931DB2v1
Mathcad Design Data (cont.)
Corner
x
0
0
1
2
3
4
5
6
7
8
Corner
IL1RMS
mA
0
428
426
423
383
384
388
359
361
363
IL1RMS
383
388
359
428
IL1MAX
mA
0
1121
1233
1345
1063
1184
1318
1036
1161
1291
IL1MAX
1063
1318
1036
1345
IL2RMS
mA
0
362
351
338
362
351
341
362
351
341
IL2RMS
341
362
338
362
IL2MAX
mA
0
406
406
406
406
406
406
406
406
406
IL2MAX
406
406
406
406
I2RMS
mA
0
389
367
345
356
337
322
337
319
304
I2RMS
322
356
304
389
V2MIN
V
0
94
110
127
111
130
149
123
144
166
V2MIN
111
149
94
166
V2MAX
V
0
107
119
134
122
137
154
133
151
171
V2MAX
122
154
107
171
V2RELPPR
%
0.0
13.1
7.9
4.8
9.7
5.8
3.7
8.1
4.8
3.0
V2RELPPR
4
10
3
13
ISRMS
mA
0
504
492
480
455
446
442
428
420
414
ISRMS
442
455
414
504
ISMAX
mA
0
1526
1639
1750
1469
1590
1723
1442
1567
1696
ISMAX
1469
1723
1442
1750
VSMAX
V
0
241
254
270
285
301
319
317
336
357
VSMAX
285
319
241
357
IDL1AVG
mA
0
300
271
245
253
229
211
228
206
188
IDL1AVG
211
253
188
300
IDF1AVG
mA
0
152
128
108
131
111
96
120
101
86
IDF1AVG
96
131
86
152
IDR2AVG
mA
0
152
129
108
131
111
94
119
100
85
IDR2AVG
94
131
85
152
IDF2AVG
mA
0
209
221
227
230
239
244
242
250
254
IDF2AVG
230
244
209
254
IRS1RMS
mA
0
295
303
310
260
270
282
242
252
262
IRS1RMS
260
282
242
310
IRS2RMS
mA
0
235
213
192
218
198
180
208
188
171
IRS2RMS
180
218
171
235
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
11
HV9931DB2v1
Simulated Waveforms (Mathcad)
Corner 0 (100VAC) (High Duty)
Corner 1 (100VAC) (Nom Duty)
Corner 2 (100VAC) (Low Duty)
Corner 3 (120VAC) (High Duty)
Corner 4 (120VAC) (Nom Duty)
Corner 5 (120VAC) (Low Duty)
Corner 6 (135VAC) (High Duty)
Corner 7 (135VAC) (Nom Duty)
Corner 8 (135VAC) (Low Duty)
Drain Voltage Envelope
Rectified Line Voltage
Bus Voltage
Input Inductor
Peak Current
Envelope
Line Voltage
Supertex inc.
Line Current
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
12
HV9931DB2v1
Bill of Materials
Qty
REF
Description
Manufacturer
Product Number
1
BR11
RECT BRIDGE GP MINIDIP 600V 0.5A
Diodes Inc
RH06-T
2
C62, C72
CAP CER NP0 50V 10% 0805 100PF
Kemet
C0805C101K5GACTU
2
C41, C81
CAP CER X7R 100V 10% 0805 10NF
Kemet
C0805C103K1RACTU
1
C37
CAP CER NP0 1000V 5% 0805 100PF
Vishay/Vitramon
VJ0805A101JXGAT5Z
2
C51, C65
CAP CER X7R 16V 10% 1206 10µF
Murata
GRM31CR71C106KAC7L
3
C11, C12, C21
CAP MKP 305VAC X2 125C 20% 47NF
EPCOS Inc
B32921A2473M
3
D31, D32, D37
DIODE ULTRAFAST 800V 1A SMA
STMicroelectronics
STTH108A
2
D41, D42
DIODE ULTRAFAST 600V 1A SMA
STMicroelectronics
STTH1R06A
2
D39, D79
DIODE ULTRAFAST HI COND SOT-23
Fairchild Semiconductor
MMBD914
1
DN65
DIODE SW DUAL 75V 350MW SOT23
Diodes Inc
BAV99-7-F
1
E31
CAP ALEL ED RAD10X20 250V 20% 22µF
Panasonic ECG
EEU-ED2E220
1
F11
FUSE SLOW IEC TR5 250MA
Littelfuse Wickmann
37202500411
1
HS
HEATSINK TO220 W/TAB W86 D40 H75 21K
Aavid Thermalloy
574502B03700G
1
IC51
IC LED DRIVER 8L SOIC
Supertex
HV9931LG-G
2
L11, L21
CHOKE SH RAD13MM 15% 2.2MH 520MA
Sumida
RCP1317NP-222L
1
L31
CHOKE RAD 450D 710L 10% 1200µH
Renco
RL-5480-4-1200
1
L41
CHOKE RAD 625D 700L 10% 3.9MH
Renco
RL-5480-5-3900
1
M31
MOSFET N-CH 800V 2A 2.7R TO-220FP
Inﬁneon Technologies
SPA02N80C3
1
MOV11
SUR ABSORBER 10MM 430VDC 2500A ZNR
Panasonic ECG
ERZ-V10D431
2
Q81, Q83
TRANSISTOR GP NPN AMP SOT-23
Fairchild Semiconductor
MMBT2222A
2
Q82, Q84
TRANSISTOR GP PNP AMP SOT-23
Fairchild Semiconductor
MMBT2907A
1
R99
RES 1/8W 0805 1% 1.00KΩ
Panasonic ECG
ERJ-6ENF1001V
2
R39, R79
RES 1/8W 0805 1% 100Ω
Panasonic ECG
ERJ-6ENF1000V
1
R62
RES 1/8W 0805 1% 2.43KΩ
Panasonic ECG
ERJ-6ENF2431V
1
R72
RES 1/8W 0805 1% 2.67KΩ
Panasonic ECG
ERJ-6ENF2671V
1
R81
RES 1/8W 0805 1% 10.0KΩ
Panasonic ECG
ERJ-6ENF1002V
1
R82
RES 1/8W 0805 1% 13.0KΩ
Panasonic ECG
ERJ-6ENF1302V
1
R63, R73
RES 1/8W 0805 1% 75.0KΩ
Panasonic ECG
ERJ-6ENF7502V
2
R85, R86
RES 1/8W 0805 1% 100KΩ
Panasonic ECG
ERJ-6ENF1003V
1
R51
RES 1/8W 0805 1% 205KΩ
Panasonic ECG
ERJ-6ENF2053V
3
R80, R87, R90
RES 1/8W 0805 1% 200KΩ
Panasonic ECG
ERJ-6ENF2003V
2
R64, R65
RES 1/8W 0805 1% 1.30MΩ
Panasonic ECG
ERJ-6ENF1304V
3
R68, R83, R84
RES 1/8W 0805 1% 1.00MΩ
Panasonic ECG
ERJ-6ENF1004V
1
R88
RES 1/8W 0805 1% 10.0MΩ
Vishay/Dale
CRCW080510M0FKEA
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
13
HV9931DB2v1
Bill of Materials (cont.)
Qty
REF
Description
Manufacturer
Product Number
1
R37
RES 1/4W 1206 5% 6.8KΩ
Panasonic ECG
ERJ-8GEYJ682V
1
R31
RES 1/4W 1206 1% 10.0MΩ
Vishay/Dale
CRCW120610M0FKEA
1
R61
RES 1/4W 0805 1% .27Ω
Susumu Co Ltd
RL1220S-R27-F
1
R71
RES 1/4W 0805 1% .68Ω
Susumu Co Ltd
RL1220S-R68-F
1
TVS11
DIODE TVS BIDIR SMA 400W 5% 440V
Littelfuse Inc
SMAJ440CA
2
Z61, Z90
DIODE ZENER 350MW SOT-23 7.5V
Diodes Inc
BZX84C7V5-7-F
1
Z91
DIODE ZENER 350MW SOT-23 47V
Diodes Inc
BZX84C47-7-F
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
Supertex inc.
©2010 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
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110510
14
Supertex inc.
HV9931DB1v2
LED Driver Demo Board
Input 120VAC // Output 350mA, 40V (14W)
General Description
The HV9931 LED driver is primarily targeted at low to
medium power LED lighting applications where galvanic
isolation of the LED string is not an essential requirement.
The driver provides near unity power factor and constant
current regulation using a two stage topology driven by
a single MOSFET and control IC. Triac dimming of this
design is possible with the addition of some components for
preloading and inrush current shaping.
The DB1 and DB2 demo boards were designed for a ﬁxed
string current of 350mA and a string voltage of 40V for a load
power of about 14W. The boards will regulate current for an
output voltage down to 0V.
Nominal input voltage for the DB1 is 120VAC, for the DB2
230VAC. Design for universal input (85 to 265VAC) is by
all means possible but does increase cost and size while
lowering efﬁciency.
The input EMI ﬁlter was designed to suppress the differential
mode switching noise to meet CISPR15 requirements.
No speciﬁc components were added to suppress currents
of common mode nature. Common mode current can be
controlled in many ways to satisfy CISPR 15 requirements.
featured are output current soft start and protections from
line overvoltage, load overvoltage and open circuit. The
driver is inherently short circuit proof by virtue of the peak
current regulation method.
Speciﬁcations
Input voltage:
100VRMS to 135VRMS, 60Hz
Output voltage:
0 to 40V
Output current:
350mA +/-5%
Output power:
14W, Max
Power factor
98%
Total harmonic distortion
EN61000-3-2 Class C
EMI limits
CISPR 15 (see text)
Efﬁciency
83%
Output current ripple
30%PP
Input overvoltage protection
140VRMS, Latching
Output overvoltage protection 43V, Latching
Switching frequency
73kHz
Dimensions:
3.5” x 3.0” x 1.25”
The board is ﬁtted with a number of optional circuits; a
schematic of a simpliﬁed driver is given as well. The circuits
Board Layout and Connections
A
V
V
A
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1
HV9931DB1v2
 Warning!
Working with this board can cause serious bodily harm or
death. Connecting the board to a source of line voltage will
result in the presence of hazardous voltage throughout the
system including the LED load.
The board should only be handled by persons well aware of
the dangers involved with working on live electrical equipment. Extreme care should be taken to protect against electric shock. Disconnect the board before attempting to make
any changes to the system conﬁguration. Always work with
another person nearby who can offer assistance in case of
an emergency. Wear safety glasses for eye protection.
Special Note:
The electrolytic capacitor carries a hazardous voltage for an
extended time after the board is disconnected. The board
includes a 1MΩ resistor placed across the electrolytic capacitor which will slowly discharge the capacitor after disconnection from line voltage. The voltage will fall more or
less exponentially to zero with a time constant of about 100
seconds. Check the capacitor voltage before handling the
board.
Connection Instructions
Step 1.
Carefully inspect the board for shipping damage, loose
components, etc, before making connections.
Step 2.
Attach the board to the line and load as shown in the diagram.
Be sure to check for correct polarity when connecting the
LED string to avoid damage to the string. The board is short
circuit and open circuit proof. The LED string voltage can
be anything between zero and 40V, though performance will
suffer when the string voltage is substantially lower than the
target of 40V. See the typical performance graphs.
voltage and LED string voltage are more or less constant
as well. Duty cycle and bus voltage do adjust in response to
changes in line or load voltage but are otherwise constant
over the course of a line cycle. With the HV9931, OFF time is
ﬁxed by design, being programmed by an external resistor,
whereas ON time adjusts to a more or less constant value,
being under control of the HV9931 peak current regulator.
Principles of Operation
The input or buck-boost stage is designed for operation
in discontinuous conduction mode (DCM) throughout the
range of line and load voltage anticipated. This can be
accomplished by making the input inductor sufﬁciently small.
A well known property of the DCM buck-boost stage, when
operated with constant ON time and constant OFF time, is
that input current is proportional to input voltage, whether
in peak value or average value. This results in sinusoidal
input current when the input voltage is sinusoidal, thereby
giving unity power factor operation when operating from the
rectiﬁed AC line voltage.
The output or buck stage is designed for operation in
continuous conduction mode (CCM), operating with about 20
to 30% inductor current ripple. This amount of ripple serves
the needs of the HV9931 peak current controller which relies
on a sloping inductor current for setting ON time, and is of an
acceptable level to high brightness LEDs. Duty cycle is more
or less constant throughout the line cycle as the DC bus
When operated in the anticipated range of line and load
voltage, the MOSFET ON time will be under control of the
output stage current controller, which turns the MOSFET
off when sensing that the output inductor current has
reached the desired peak current level as programmed by
a resistive divider at the CS2 pin. Under certain abnormal
circumstances such as initial run-up and line undervoltage,
which both could lead to the draw of abnormally high line
current, ON time is further curtailed by the action of the CS1
comparator, which monitors the input stage inductor current
against a threshold. This threshold can be a simple DC level
or be shaped in time as is performed on the demo board. In
particular, when shaping the CS1 threshold with the shape of
the rectiﬁed AC line input voltage waveform, the line current
will be bounded by a more or less sinusoidal line current
envelope which results in sinusoidal input current for low line
and other abnormal conditions.
Step 3.
Energize the mains supply. The board can be connected to
mains directly. Alternatively voltage can be raised gradually
from zero to full line voltage with the aid of an adjustable AC
supply such as a Variac or a programmable AC source.
The HV9931 topology can be viewed as a series connection
of two basic power supply topologies, (1) a buck-boost
stage as ﬁrst or input stage, for purpose of converting AC
line power into a source of DC power, commonly known as
the DC bus, having sufﬁcient capacitive energy storage to
maintain the bus voltage more or less constant throughout
the AC line cycle, and (2) a buck stage as second or output
stage for powering the LED string, stepping down the DC
bus voltage to the LED string voltage in order to produce a
steady LED string current.
Supertex inc.
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2
HV9931DB1v2
The design exercise of an HV9931 LED driver revolves
around establishing component values for (1) the input and
output stage inductors, (2) a value for the bus capacitor, and
(3) a value for switching cycle OFF time, which together
result in (1) acceptable current ripple at the output stage
(say 30%), (2) an acceptable bus voltage ripple (say 5%),
and (3) an input stage which maintains DCM operation over
the desired line and load voltage range.
For a given HV9931 design, the bus voltage rises and falls
with like changes in line and load voltage. This is unlike a
two stage design having two transistors and control ICs,
where the bus voltage can be set independent of line and
load voltage variation. If the desired ranges of line and load
voltage are particularly large then the latter topology may be
preferable so as to avoid large variation in bus voltage.
The design of an HV9931 based LED driver is not further
Simpliﬁed Schematic Diagram
F11
250mA
AC2
L21
1mH
L11
1mH
C11
100nF
1
2
Mathcad design data can be found at the end of this
document. The data tends to be in good agreement with the
actual demo board despite the omission of switching losses
in the model. For this design we can see that the calculated
efﬁciency is off by say 5 percent likely due underestimation
of switching losses and inductor core and winding losses.
A Simpliﬁed Version of the Design
The demo board can be simpliﬁed signiﬁcantly. Below is a
schematic showing the essential elements of the driver.
D32
STTH1L06A
L31
560μH
E31
47μF
D31
STTH1L06A
C
BR11
RH06-T
R37
6.8kΩ
C21
100nF
D42
STTH102A
M31
SPA04N50C3
R51
196kΩ
C
ZOV
BZX84C43
1
R62
2.43kΩ
THROV
BT168GW
ANO
A
R61
180mΩ
ROV
10kΩ
CAT
C37
100pF
4
Optional Output
Overvoltage Protection
L41
3.3mH
D41
STTH1R06A
+
3
C12
100nF
AC1
discussed here, except for noting that a semi-automatic
design tool is available in Mathcad form, based on behavioral
simulation, which, allows components to be adjusted in an
iterative manner, starting from an initial guess. The tool allows
quick evaluation of nine standard test cases, exercising the
design over line voltage variation and tolerance variation of
three component parameters.
VIN
2
R68
75kΩ
4
8
RT
GATE
IC51
CS1
R71
680mΩ
R72
2.67kΩ
CS2
HV9931LG
GND
VDD
PWM
3
6
5
7
R73
75kΩ
A
C51
10µF
Contact Supertex Applications Engineering for guidance in
simplifying the design or for adding functions such as triac
dimmability.
Note on Inductors:
This board was ﬁtted with standard (COTS) inductors. These
are not necessarily an optimal choice but present an expedient way to go when evaluating a design. Custom engineered
parts generally give better performance, particularly with respect to efﬁciency.
Drum core style inductors, whether in radial or axial leaded
Supertex inc.
versions, are popular for their ready availability and low cost.
Drum core styles have particularly simple construction and
can be wound for lowest cost without coil former (bobbin).
They may serve well during the development stage, but may
not be the best choice for ﬁnal design. Keep these type of
inductors away form any metallic surface such as heatsinks,
PCB copper planes, metallic enclosures, and capacitors, as
these unshielded parts can create high eddy current losses
in these parts. For tightly packaged designs or where inductor losses are an issue, drum core style inductors are not
recommended.
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3
Supertex inc.
AC1
AC2
F11
250mA
R82
24.3kΩ
R83
1MΩ
R84
1MΩ
C12
100nF
Q82
MMBT2907A
MOV11 C11
275V
100nF
L11
1mH
C81
10nF
2
TVS11
SMAJ
440CA
1
R80
100kΩ
Q81
MMBT2222A
R81
10kΩ
3
1
REC
BR11
RH06-T
DN65
BAV99
C65
10µF
2
4
3
L21
1mH
L1D
R68
1MΩ
R88
10MΩ
R87
200kΩ
2
R37
6.8kΩ
IC51
Q84
MMBT2907A
VDD
6
3
C51
10µF
HV9931
VDD
ENA
R51
196kΩ
GATE
4
GATE
CS2
8
5
PWM
RT
D42
MMDB914
7
L41
3.3mH
R90
150kΩ
C72
100pF
R79
100Ω
D42
STTH102A
SN2
D79
MMBD914
D31
STTH1R06A
M31
SPA04N50C3
R31
1MΩ
+
E31
47μF
GND
CS1
VIN
1
IDD
R39
100Ω
C37
100pF
D31
STTH1L06A
R99
1kΩ
C62
100pF
R62
2.43kΩ
R86
100kΩ
Q83
MMBT2222A
R85
100kΩ
Z61
BZX84C7V5
R63
75kΩ
R64
634kΩ
R65
634kΩ
R61
180mΩ
RS1
C21
100nF
D37
STTH1L06A
L31
560μH
D32
STTH1L06A
R73
75kΩ
R72
2.67kΩ
R71
680mΩ
RS2
C41
10nF
Z90
BZX84C7V5
Z91
BZX84C47
GND2
GND1
ANO
CAT
HV9931DB1v2
Schematic Diagram
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4
HV9931DB1v2
Typical Characteristics
String Current [mA] vs. String Voltage [V]
1000
100
900
90
800
80
700
70
135VRMS
600
60
120VRMS
500
50
400
40
300
30
200
(100VRMS, 120VRMS, 135VRMS)
virtually the same
20
100VRMS
100
0
Efficiency [%] vs. String Voltage [V]
10
0
10
20
30
40
50
0
0
10
20
30
40
50
THD [%] vs. String Voltage [V]
PF [%] vs. String Voltage [V]
100
30
90
25
80
135VRMS
70
20
120VRMS
60
100VRMS
50
15
40
100VRMS
10
30
20
120VRMS
135VRMS
5
10
0
0
10
20
Supertex inc.
30
40
50
0
0
10
20
30
40
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5
50
HV9931DB1v2
Typical Waveforms (1)
Line Voltage and Current at nominal load (350mA, 40V)
100VRMS
120VRMS
135VRMS
IAC
VAC
Line Voltage and Current at half load (350mA, 20V)
100VRMS
120VRMS
135VRMS
Output Current and Drain Voltage at nominal load (350mA, 40V)
ILED (Peak)
VDRAIN
ILED (Valley)
Output Current and Drain Voltage at half load (350mA, 20V)
Supertex inc.
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6
HV9931DB1v2
Typical Waveforms (2) (120VRMS, 40V, 350mA)
Drain Voltage and LED Current
400µs per div
40µs per div
4µs per div
ILED
350mAAVE
VDRAIN
Drain Voltage and Gate Voltage
4µs per div
40µs per div
40µs per div
VGATE
Turn-ON
Turn-OFF
Recovery of D42
Recovery of D41
VDRAIN
Drain Voltage and Current Sense Voltages of Stages 1 and 2
VRS1
VRS2
Recovery of D41
Recovery of D42
VDRAIN
Drain Voltage and Voltages at Test Points REC, SN3, SN2
VSN3
VREC
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VSN2
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7
HV9931DB1v2
Typical Waveforms (3) (120VRMS, 40V, 350mA)
Drain Voltage and Voltage at the Test Point L1D (3 points along the AC line cycle)
AT ~ 90°
AT ~ 30°
AT ~ 10°
Clamping action of D37
VDRAIN
VL1D
Supertex inc.
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8
HV9931DB1v2
EMI Signature
Board suspended about 3” above reference plane.
Limit Line:
Detector:
IF Bandwidth:
Shielding:
CISPR 15 Quasi Peak (9kHz to 30MHz)
Peak Hold
9kHz
2 copper shields, surrounding the power section on top and bottom of the board, terminated at the source
of the MOSFET.
Without shielding :
110dBµV
100dBµV
90
80
66
56
60
50dBµV
10kHz
100kHz
10MHz
1MHz
With shielding :
110dBµV
100dBµV
90
80
66
56
60
50dBµV
10kHz
100kHz
The performance graphs above were obtained from the
board not having speciﬁc measures to suppress common
mode emissions, such as inclusion of a common mode inductor in the AC line input circuitry. The above graphs show
how shielding can signiﬁcantly reduce emissions, particu-
Supertex inc.
1MHz
10MHz
larly in the upper frequency range. The shielding also was
instrumental in reducing the lower frequency emissions by
reducing magnetic ﬁeld coupling from the main inductors
to the EMI ﬁlter inductors (EMI ﬁlter section kept outside of
shielded area).
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9
HV9931DB1v2
Mathcad Design Data
Corner
x
0
0
1
2
3
4
5
6
7
8
Corner
L1
µH
0
616
560
504
616
560
504
616
560
504
L1
-
-
-
-
RL1
mR
0
2000
2000
2000
2000
2000
2000
2000
2000
2000
RL1
-
-
-
-
L2
mH
0
3300
3300
3300
3300
3300
3300
3300
3300
3300
L2
-
-
-
-
RL2
mR
0
3000
3000
3000
3000
3000
3000
3000
3000
3000
RL2
-
-
-
-
ILRF2
%
0
32
32
32
32
32
32
32
32
32
ILRF2
-
-
-
-
C2
uF
0.0
37.6
47.0
56.4
37.6
47.0
56.4
37.6
47.0
56.4
C2
-
-
-
-
NF
x
0
2
2
2
2
2
2
2
2
2
NF
-
-
-
-
LF
µH
0
1000
1000
1000
1000
1000
1000
1000
1000
1000
LF
-
-
-
-
RLF
mR
0
2000
2000
2000
2000
2000
2000
2000
2000
2000
RLF
-
-
-
-
CF
nF
0
100
100
100
100
100
100
100
100
100
CF
-
-
-
-
C1
nF
0
100
100
100
100
100
100
100
100
100
C1
-
C2V
135
-
RS
mR
0
1000
1000
1000
1000
1000
1000
1000
1000
1000
RS
-
C2R
1345
-
VD
mV
0
1000
1000
1000
1000
1000
1000
1000
1000
1000
VD
-
-
-
-
TF
us
0.0
8.7
8.7
8.7
8.7
8.7
8.7
8.7
8.7
8.7
TF
-
-
-
-
RT
kR
0
196
196
196
196
196
196
196
196
196
RT
-
-
-
-
FM
Hz
0
50
50
50
50
60
50
50
50
50
FM
-
-
-
-
VMRMS
V
0
100
100
100
120
120
120
135
135
135
VMRMS
-
-
-
-
IMRMS
mA
0
167
160
153
137
133
130
122
118
115
IMRMS
130
137
115
167
IMMAX
mA
0
246
232
221
200
191
185
176
169
163
IMMAX
185
200
163
246
V3AVG
V
0
40
40
40
40
40
40
40
40
40
V3AVG
40
40
40
40
I3AVG
mA
0
361
350
335
361
350
339
361
350
339
I3AVG
339
361
335
361
PM
W
0.0
16.5
15.9
15.3
16.3
15.8
15.5
16.2
15.8
15.3
PM
15.5
16.3
15.3
16.5
P3
W
0.0
14.4
14.0
13.4
14.4
14.0
13.6
14.4
14.0
13.6
P3
13.6
14.4
13.4
14.4
EFF
%
0.0
87.5
88.0
87.8
88.7
88.5
87.7
88.9
88.7
88.3
EFF
87.7
88.7
87.5
88.9
PF
%
0.0
98.7
99.3
99.6
98.9
99.3
99.5
98.8
99.1
99.3
PF
98.9
99.5
98.7
99.6
THD
%
0.0
9.0
5.3
3.3
6.4
3.8
2.5
5.1
3.1
2.1
THD
2.5
6.4
2.1
9.0
H3
%
0.0
8.7
5.1
3.1
6.2
3.6
2.3
5.0
2.9
1.9
H3
2.3
6.2
1.9
8.7
H5
%
0.0
1.7
1.0
0.7
1.1
0.7
0.6
0.8
0.6
0.5
H5
0.6
1.1
0.5
1.7
TAMIN
µs
0.0
4.6
4.8
4.8
3.7
3.9
3.9
3.2
3.4
3.4
TAMIN
3.7
3.9
3.2
4.8
TAMAX
µs
0.0
5.8
5.4
5.2
4.3
4.2
4.2
3.7
3.6
3.6
TAMAX
4.2
4.3
3.6
5.8
TFMIN
µs
0.0
7.0
8.7
10.5
7.0
8.7
10.5
7.0
8.7
10.5
TFMIN
7.0
10.5
7.0
10.5
TFMAX
µs
0.0
7.0
8.7
10.5
7.0
8.7
10.5
7.0
8.7
10.5
TFMAX
7.0
10.5
7.0
10.5
DAMIN
%
0.0
39.6
35.5
31.6
34.7
30.8
27.4
31.8
28.0
24.7
DAMIN
27.4
34.7
24.7
39.6
DAMAX
%
0.0
45.3
38.4
33.1
38.3
32.6
28.4
34.5
29.4
25.5
DAMAX
28.4
38.3
25.5
45.3
DC1MAX
%
0.0
98.6
79.7
65.2
87.1
70.0
57.7
80.4
64.3
52.4
DC1MAX
57.7
87.1
52.4
98.6
FSMIN
kHz
0.0
78.4
70.6
63.9
88.4
77.3
68.4
93.9
81.0
71.2
FSMIN
68.4
88.4
63.9
93.9
FSMAX
kHz
0.0
86.5
74.0
65.4
93.6
79.4
69.4
97.8
82.6
71.9
FSMAX
69.4
93.6
65.4
97.8
Supertex inc.
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10
HV9931DB1v2
Mathcad Design Data (cont.)
Corner
x
0
0
1
2
3
4
5
6
7
8
Corner
IL1RMS
mA
0
428
426
423
383
384
388
359
361
363
IL1RMS
383
388
359
428
IL1MAX
mA
0
1121
1233
1345
1063
1184
1318
1036
1161
1291
IL1MAX
1063
1318
1036
1345
IL2RMS
mA
0
362
351
338
362
351
341
362
351
341
IL2RMS
341
362
338
362
IL2MAX
mA
0
406
406
406
406
406
406
406
406
406
IL2MAX
406
406
406
406
I2RMS
mA
0
389
367
345
356
337
322
337
319
304
I2RMS
322
356
304
389
V2MIN
V
0
94
110
127
111
130
149
123
144
166
V2MIN
111
149
94
166
V2MAX
V
0
107
119
134
122
137
154
133
151
171
V2MAX
122
154
107
171
V2RELPPR
%
0.0
13.1
7.9
4.8
9.7
5.8
3.7
8.1
4.8
3.0
V2RELPPR
4
10
3
13
ISRMS
mA
0
504
492
480
455
446
442
428
420
414
ISRMS
442
455
414
504
ISMAX
mA
0
1526
1639
1750
1469
1590
1723
1442
1567
1696
ISMAX
1469
1723
1442
1750
VSMAX
V
0
241
254
270
285
301
319
317
336
357
VSMAX
285
319
241
357
IDL1AVG
mA
0
300
271
245
253
229
211
228
206
188
IDL1AVG
211
253
188
300
IDF1AVG
mA
0
152
128
108
131
111
96
120
101
86
IDF1AVG
96
131
86
152
IDR2AVG
mA
0
152
129
108
131
111
94
119
100
85
IDR2AVG
94
131
85
152
IDF2AVG
mA
0
209
221
227
230
239
244
242
250
254
IDF2AVG
230
244
209
254
IRS1RMS
mA
0
295
303
310
260
270
282
242
252
262
IRS1RMS
260
282
242
310
IRS2RMS
mA
0
235
213
192
218
198
180
208
188
171
IRS2RMS
180
218
171
235
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
11
HV9931DB1v2
Simulated Waveforms (Mathcad)
Corner 0 (100VAC) (High Duty)
Corner 1 (100VAC) (Nom Duty)
Corner 2 (100VAC) (Low Duty)
Corner 3 (120VAC) (High Duty)
Corner 4 (120VAC) (Nom Duty)
Corner 5 (120VAC) (Low Duty)
Corner 6 (135VAC) (High Duty)
Corner 7 (135VAC) (Nom Duty)
Corner 8 (135VAC) (Low Duty)
Drain Voltage Envelope
Rectified Line Voltage
Bus Voltage
Input Inductor
Peak Current
Envelope
Line Voltage
Supertex inc.
Line Current
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
12
HV9931DB1v2
Bill of Materials
Qty
REF
Description
Manufacturer
Product Number
1
BR11
RECT BRIDGE GP MINIDIP 600V 0.5A
Diodes Inc
RH06-T
2
C62, C72
CAP CER NP0 50V 10% 0805 100PF
Kemet
C0805C101K5GACTU
2
C41, C81
CAP CER X7R 100V 10% 0805 10NF
Kemet
C0805C103K1RACTU
1
C37
CAP CER NP0 1000V 5% 0805 100PF
Vishay/Vitramon
VJ0805A101JXGAT5Z
2
C51, C65
CAP CER X7R 16V 10% 1206 10µF
Murata
GRM31CR71C106KAC7L
3
C11, C12, C21
CAP MKP 305VAC X2 125C 20% 100NF
EPCOS Inc
B32921C3104M
1
D42
DIODE ULTRAFAST 200V 1A SMA
STMicroelectronics
STTH102A
3
D31, D32, D37
DIODE FAST 600V 1A SMA
STMicroelectronics
STTH1L06A
1
D41
DIODE ULTRAFAST 600V 1A SMA
STMicroelectronics
STTH1R06A
2
D39, D79
DIODE ULTRAFAST HI COND SOT-23
Fairchild Semiconductor
MMBD914
1
DN65
DIODE SW DUAL 75V 350MW SOT23
Diodes Inc
BAV99-7-F
1
E31
CAP ALEL ED RAD12X20 200V 20% 47µF
Panasonic ECG
EEU-ED2D470
1
F11
FUSE SLOW IEC TR5 250MA
Littelfuse Wickmann
37202500411
0
HS
HEATSINK TO220 W/TAB W86 D40 H75 21K
Aavid Thermalloy
574502B03700G
1
IC51
IC HV9931 LED DRIVER 8L SOIC
Supertex
HV9931LG-G
2
L11, L21
CHOKE SH RAD13MM 15% 1.0MH 820MA
Sumida
RCP1317NP-102L
1
L31
CHOKE RAD 450D 710L 10% 560µH
Renco
RL-5480-4-560
1
L41
CHOKE RAD 625D 700L 10% 3.3MH
Renco
RL-5480-5-3300
1
M31
MOSFET N-CH 560V 4.5A 0.95R TO-220FP
Inﬁneon Technologies
SPA04N50C3
1
MOV11
MOV 10MM 430VDC 2500A ZNR
Panasonic ECG
ERZ-V10D431
2
Q81, Q83
TRANSISTOR GP NPN SOT-23
Fairchild Semiconductor
MMBT2222A
2
Q82, Q84
TRANSISTOR GP PNP SOT-23
Fairchild Semiconductor
MMBT2907A
2
R90, R99
RES 1/8W 0805 1% 1.00KΩ
Panasonic ECG
ERJ-6ENF1001V
2
R39, R79
RES 1/8W 0805 1% 100Ω
Panasonic ECG
ERJ-6ENF1000V
1
R62
RES 1/8W 0805 1% 2.43KΩ
Panasonic ECG
ERJ-6ENF2431V
1
R72
RES 1/8W 0805 1% 2.67KΩ
Panasonic ECG
ERJ-6ENF2671V
1
R81
RES 1/8W 0805 1% 10.0KΩ
Panasonic ECG
ERJ-6ENF1002V
1
R82
RES 1/8W 0805 1% 24.3KΩ
Panasonic ECG
ERJ-6ENF2432V
1
R63, R73
RES 1/8W 0805 1% 75.0KΩ
Panasonic ECG
ERJ-6ENF7502V
2
R80, R85, R86
RES 1/8W 0805 1% 100KΩ
Panasonic ECG
ERJ-6ENF1003V
1
R90
RES 1/8W 0805 1% 150KΩ
Panasonic ECG
ERJ-6ENF1503V
1
R51
RES 1/8W 0805 1% 196KΩ
Panasonic ECG
ERJ-6ENF1963V
1
R87
RES 1/8W 0805 1% 200KΩ
Panasonic ECG
ERJ-6ENF2003V
2
R64, R65
RES 1/8W 0805 1% 634KΩ
Panasonic ECG
ERJ-6ENF6343V
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
13
HV9931DB1v2
Bill of Materials (cont.)
Qty
REF
Description
Manufacturer
Product Number
3
R68, R83, R84
RES 1/8W 0805 1% 1.0MΩ
Panasonic ECG
ERJ-6ENF1004V
1
R88
RES 1/8W 0805 1% 10.0MΩ
Vishay/Dale
CRCW080510M0FKEA
1
R37
RES 1/4W 1206 5% 6.8KΩ
Panasonic ECG
ERJ-8GEYJ682V
1
R31
RES 1/4W 1206 5% 1.0MΩ
Panasonic ECG
ERJ-8GEYJ105V
1
R61
RES 1/4W 0805 1% .18Ω
Susumu Co Ltd
RL1220S-R18-F
1
R71
RES 1/4W 0805 1% .68Ω
Susumu Co Ltd
RL1220S-R68-F
1
TVS11
DIODE TVS BIDIR SMA 400W 5% 440V
Littelfuse Inc
SMAJ440CA
2
Z61, Z90
DIODE ZENER 350MW SOT-23 7.5V
Diodes Inc
BZX84C7V5-7-F
1
Z91
DIODE ZENER 350MW SOT-23 47V
Diodes Inc
BZX84C47-7-F
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
Supertex inc.
©2010 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
041410
14
Supertex inc.
HV9931DB5
Universal Input, Single High Brightness,
LED Driver Demoboard
General Description
The Supertex HV9931DB5 demoboard is a high brightness
(HB) LED power driver to supply one HB LED, using the
HV9931 IC from either a 110 or 220VAC supply. The
HV9931DB5 is ideal for incandescent retroﬁt applications,
as it features a very small size and a low component count.
Speciﬁcations
Parameter
Value
Input
90 – 265V AC, 50/60Hz
LED current set point
350mA ± 10%
Maximum output voltage
The HV9931DB5 avoids the use of electrolytic capacitors,
which reduce the lifetime of the circuit in high ambient
temperatures (which would be found in the base of a bulb).
The demo board can be used to test the performance of the
HV9931 as a constant current driver to power LEDs.
4.0V
variable
(constant off-time,
TOFF = 8.0μs)
Switching frequency
Board dimensions
OD = 29mm, HT = 15mm
The HV9931DB5 uses a unique cascaded converter circuit,
with a single active switch, to achieve the high step down
conversion ratio required for operating low voltage LEDs
from a high input voltage. This circuit allows the converter
to operate at a high switching frequency, about 120kHz,
while still regulating the output current at all times. The
HV9931DB5 supplies 350mA to a 4.0V(max) LED with input
voltages ranging from 90 – 265VAC 50/60Hz.
Board Layout and Connections
C1
D2
D3
Q1
R12
U1
C4
C5
C3
BR1
R1
L3
C2
R9
R2
L2
R3
R8
R10
L1
D1
D6
R11
AC1
MOV1 AC2
R4
LEDLED+
POS
AC Line
90 -260 VAC
Connections:
1. Connect the universal input to the AC IN terminals.
2. Connect the output to the LED terminals:
- Red wire to anode of LED
- Black wire to cathode of LED.
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
HV9931DB5
Testing the HV9931DB5:
Place an ammeter in series with the LED to measure the
LED current. The LED should glow when the AC power is
turned on.
Note on Current Measurement:
The HV9931DB5 is designed to regulate the output current
at 350mA (the recommended current level for most 1.0W
HB LEDs). This can easily be veriﬁed by applying a DC
voltage greater than 50V at the input of the demo board.
However, when the output current is measured with an
AC waveform, the measured current is typically around
Fig. 1: Output Current at 120V Input Voltage
300mA. This drop in the current is due to the demo board
turning off when the instantaneous input voltage is less
than 40V. This dropout at low voltages causes the average
current to drop by about 50mA. The output current can be
increased or decreased by increasing the value of resistor
R10 proportionally.
Open LED Protection:
The HV9931DB5 is not protected against open LED
conditions. Leaving the LED terminals open while applying
an input voltage will damage the circuit.
Fig. 2: Output Current at 240V Input Voltage
Fig. 3: Input Voltage and Current Waveforms at 120VAC Input
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
2
HV9931DB5
Fig. 4a: Conducted EMI test (CISPR 15) at 120VAC
Fig. 4b: Conducted EMI test (CISPR 15) at 240VAC
Schematic Diagram
D1
AC1
R1
200Ω
BR1
AC2
MOV1
430V
R2
200Ω
-
L3
L1
2200µH
470µH
+
C2
0.033µF
400V
R3
10Ω
1
2
VIN
CS1
4
3
RT
GATE
GND
CS2
6
VDD
PWMD
R10
1.0kΩ
7
5
R11
19.1kΩ
HV9931DB5 PCB Layers
Top Layer
Supertex inc.
C4
4.7µF
10V
LED+
HV9931
R8
50kΩ
C5
0.1µF
50V
D3
BYD77D
R4
1.0Ω
8
LED-
470µH
R12
178kΩ
D6
33V
R8
35.7kΩ
BYD57J
C1
0.15µF
250V
Q1
IRFRC20
C3
0.1µF
400V
L2
D2
BYD57J
Bottom Layer
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
3
HV9931DB5
Bill of Materials
Quan
Ref
Des
Description
Package
Manufacturer
Manufacturer’s Part
Number
1
BR1
Rectiﬁer Bridge GP 600V 0.8A
MiniDIP
Diodes Inc
HD06-T
1
C1
Capacitor 150nF 250VDC polyester ﬁlm
TH
Panasonic
ECQ-E2154KB
1
C2
Capacitor 33nF 400VDC polyester ﬁlm
TH
Panasonic
ECQ-E4333KF
1
C3
Capacitor 100nF 400VDC polyester ﬁlm
TH
Panasonic
ECQ-E4104KF
1
C4
Capacitor 4.7μF 10VDC ceramic X7R
1206
Murata Electronics
GRM31CR71A475MA01L
1
C5
Capacitor 0.1μF 50VDC ceramic X7R
1206
Kemet
C0805C103K5RACTU
2
D1, D2
Diode ultra fast SW 600V 1A
SOD87
Philips
BYD57J
1
D3
Diode ultra fast SW 200V 2A
SOD87
Philips
BYD77D
1
D6
Diode Zener 33V 500mW
SOT-123
Diodes Inc
BZT52C33-7
1
Q1
MOSFET 600V 2A I-PAK
TH
IR
IRFUC20
1
U1
LED Driver IC
SO-8
Supertex Inc
HV9931LG
1
MOV1
Varistor 275V RMS
TH
Littelfuse Inc
V430MA7B
2
L1, L2
Inductor radial 470μH
TH
C&D Technologies
17474
1
L3
2.2mH, 64mA, axial
TH
Central Technologies
CTH6-222K
2
R1, R2
Resistor 200Ω 1/4W 5% Surge
1206
Panasonic
9C12063A2000FKHFT
1
R3
Resistor 10Ω 1/8W 1%
0805
Yageo America
RC0805FR-0710L
1
R4
Resistor 1.0Ω 1/4W 1% 1206 SMD
1206
Yageo America
9C12063A1R0FKHFT
1
R9
Resistor 50.0kΩ 1/8W 1%
0805
Yageo America
RC0805FR-0750KL
1
R11
Resistor 19.1kΩ 1/8W 1%
0805
Yageo America
RC0805FR-0719K1L
1
R8
Resistor 35.7kΩ 1/8W 1%
0805
Yageo America
RC0805FR-0735K7L
1
R10
Resistor 1.0kΩ 1/8W 1%
0805
Yageo America
RC0805FR-071K0L
1
R12
Resistor 178kΩ 1/8W 1%
0805
Yageo America
RC0805FR-07178KL
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
Supertex inc.
©2010 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
110910
4
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
Supertex inc.
DN-H05
Design Note
56W Off-line, 120VAC with PFC, 160V, 350mA Load,
Dimmer Switch Compatible LED Driver
Specifications
AC line voltage
100 - 135 VAC
LED (string) voltage
20 – 160V
LED current
350mA
Switching frequency
63kHz - @ VOUT = 160VDC
92kHz - @ VOUT = 20VDC
Efficiency*
> 88 % - @ VOUT = 160VDC
Open circuit protection
Latches off @ VOUT = 180VDC
Other protections
See text
AC line undervoltage
AC line and output power fall off
gradually below 100 VAC
Dimmer switch
compatibility
Yes
THD*
~ 12% - @ VOUT = 160VDC
Power Factor*
> 98% - @ VOUT = 160VDC
NOTE:
* Measurements taken with the damper switch bypassed. Expect a
slight degradation in efficiency, THD, etc, when the damper switch is
enabled.
General Description
This Design Note describes the results of a 56W LED Driver
Design. The driver allows smooth dimming of the LED light
when the driver is connected to a regular (TRIAC based)
dimmer switch.
Equipment). The driver is able to maintain very good line
regulation for an AC input voltage ranging between 90 and
140VAC. Below 90VAC, input power and output power fall
gradually as AC line voltage falls.
Topology
The design is an example of the Bibred topology, specifically geared to LED driving. The HV9931 is suited for driving
the Buck-Boost-Buck (BBB) topology, described in detail in
AN-H52, and the Bibred Topology, as shown in this design
note. The BBB serves applications needing large voltage
step-down ratio, whereas the Bibred serves applications
with modest step-down ratio.
Common to both topologies is operation of the input stage
in discontinuous conduction mode (DCM) and operation of
the output stage in continuous conduction mode (CCM). In
both cases, The output stage is configured as a buck stage,
which is supplied from a bulk energy storage capacitor, sufficiently large to provide a more or less constant supply voltage when considered over a AC line cycle. Constant supply
voltage entails a constant switch duty cycle when supplying
the LED load. Without entering in more detail, both the DCM
input stages of the BBB and the Bibred respond with a more
or less sinusoidal AC line input current when driven from a
switch operating at constant duty cycle.
Dimmer Switch Compatibility
This design drives a string of series connected LEDs with
a fixed current of 350mA and a string voltage of 160V max.
This same design can be operated at a lower string voltages
as well, with slight loss of efficiency or degradation of AC line
current THD, see the performance graphs.
Efficiency can be increased by using components having
less equivalent resistance, particularly L1, L2 and M1, and
by lowering of the switching frequency. All the common tradeoffs in power supply design, that is, cost versus size versus
efficiency, apply to this driver design as well.
The input line current features low harmonic distortion, satisfying the requirements of EN 61000-3-2 Class C (Lighting
Supertex inc.
The following links provide helpful information regarding the
regular domestic dimmer switch:
http://home.howstuffworks.com/dimmer-switch.htm
http://www.epanorama.net/documents/lights/lightdimmer.html.
The driver design contains two extra circuits to provide dimmer switch compatibility: a damper circuit and a bleeder circuit.
The damper circuit provides damped charging of the driver’s
input filter circuit. Resistive damping is required to prevent
AC line input current oscillations, due to the sudden rise of
the AC line voltage when the dimmer switch TRIAC comes
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
DN-H05
into conduction. The damper circuit contains two major components, (1) a damper resistor (R81), and (2) a MOSFET
(M81) for purpose of bypassing R81 shortly after charging of
the EMI filter capacitors is accomplished, thus carrying the
AC line current for the remainder of the AC line half-cycle,
without major power loss.
lation of energy on the bulk capacitor E31. The build-up of
energy may raise the capacitor voltage to a destructive level.
The high valued bleeder resistors R31 and R32 only serve
the purpose of discharging E31 following a complete turnoff, in order to provide touch-safety given some delay (RC
time constant = 44s).
The bleeder circuit provides a nominal 1.0kΩ load to the
rectified AC line to suppress a voltage rise at the input capacitors C21 thru C23 when the TRIAC in the light dimmer
is off. A typical dimmer switch contains an EMI suppression
capacitor, in the 10 to 100nF range, which is located in parallel to the TRIAC, thereby allowing significant current to flow
to the input capacitors. When the voltage rises above the
undervoltage threshold of the HV9931, several switching cycles may occur, causing the flow of output current, which will
be perceived as flicker. The bleeder circuit removes the 1kΩ
loading when the rectified line voltage exceeds about 12V
in order to suppress power dissipation in the 1kΩ bleeder
resistor when the TRIAC is on.
Output Short Circuit Protection
Protection Circuits
Short circuit protection can be added by monitoring the output current at R71, and providing a latched shut-off similar to
the one provided for output overvoltage protection.
A number of circuits can be added to the basic LED driver
circuit to provide protection against:
•
•
•
•
The output current is well regulated, except for very low output voltages; below a VOUT of about 10V control is gradually
lost, and current may rise to about 600mA at about 2V (see
performance graph). Further lowering of the output voltage
will cause the voltage on E31 to rise to a dangerous level as
output loading is barely present.
Note that the HV9931 can not reduce duty-cycle to an arbitrarily low level; leading edge blanking sets a lower limit to
the duty-cycle. Operation at minimum duty-cycle causes a
certain amount of power to flow which such be drained by
the load or other circuitry, or should lead to a shut-off of the
driver.
AC Line Overvoltage
Output Overvoltage
Output Short Circuit
AC line Overvoltage
Bulk Capacitor Overvoltage
The driver design provides latching shut-off protection
against overvoltage, which may occur in the open load condition. The need for other protection circuits depends on the
intended use of the driver.
Overvoltage Protection
The overvoltage protection circuit provides latch-off protection. Overvoltage at the output causes conduction of the
zener diodes Z71 and Z72, thereby triggering the two-transistor thyristor structure, which disables the HV9931 by pulling the PWM pin low. An alternative implementation of the
discrete two-transistor structure is the use of a true thyristor
device or a dual transistor device (MMDT2227).
AC line overvoltage protection can be attained in a manner very similar to output voltage protection. In this case
non-latching protection may be preferred, so as to avoid
nuisance shut-down due to short-lived transients. A zener
diode, transistor combination, which can pull down the PWM
pin, is all that is required.
Bulk Capacitor Overvoltage Protection
As mentioned under overvoltage protection, a non latch-off
protection scheme may allow sustained energy accumulation on the bulk capacitor.
Non latch-off protection requires active monitoring/limiting
of the bulk capacitor voltage, which represents a significant
amount of circuitry, and may not be worth the added expense. An alternate method is to provide output loading in
the form of a zener diode clamp placed across the bulk capacitor or the output circuit.
Protection circuits that do not provide latch-off should be
avoided since the existence of any switching cycles, when
no output loading is present, will cause sustained accumu-
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
2
DN-H05
Miscellaneous Notes
EMI, Common Mode Filtering
The magnitude and frequency dependency of the common
mode conducted interference current depends heavily on
physical layout, actual component choice, component orientation, location of the LED driver circuit with respect to the
LED load and enclosure, and many other factors. As such
the design may or may not require the addition of the common mode choke ahead of the bridge rectifier.
VDD and the VDD Capacitor
The capacitor on the VDD pin (C51) is purposely chosen to
be small, 220nF, so that the HV9931 shuts off near the zerocrossing of the AC line voltage.
This behavior is desired in a dimmer switch compatible design. Without this provision, the HV9931 will keep switching when the TRIAC is off, sustained by the energy stored
on a large VDD capacitor, thereby losing the dimming effect
and depleting the energy stored in the electrolytic capacitor
needed for operation as a dimmable driver.
LED Current at Zero Crossing
With a small VDD capacitor, the LED current drops out near
the zero-crossing due to the HV9931 VDD voltage dropping
out.
The LED current drop-out causes a small drop in the average LED current, which shows up as line regulation error.
Drop-out increases as AC line voltage drops.
Note that if dimmer switch compatibility is not desired, than
the VDD capacitor can made large, say 10µF, which prevents this drop-out from occurring.
Efficiency, THD, PF Measurements
Measurements of efficiency, power factor and harmonic
distortion were taken with the damper circuit removed and
a large VDD capacitor (10µF), in order to provide the best
numbers possible for this design.
The addition of the damper circuit (Dimmer switch compatible design) does not have any major effect on the measurement results, since the damper circuit primarily affects operation during the zero-crossings only, where little if any AC
Supertex inc.
current flows. The effect of the on-resistance of the bypass
switch can be accounted for in a straightforward manner in
efficiency calculations.
CS1 Programming
Control of M1 should, under regular circumstances, be governed by the action of comparator CS2, which provides regulation of the LED current. CS1 should regulate only if limitation of input stage current is necessary, which may be the
case during start-up, during AC line undervoltage and during
certain transient conditions. The programming of the CS1
comparator should present an envelope for the input stage
current, which prevents CS1 from interfering with the regulation of the output current under normal operating conditions.
A simple DC threshold, set at, say, 120% of the maximum
current at normal operating conditions, will suffice. This design employs a somewhat more sophisticated envelope for
the purpose of limiting the AC line current when undervoltage
occurs. The threshold is a scaled version of the input voltage,
thus reducing the input current envelope as input voltage
reduces. By proper choice of values, CS1 will thus become
active for input voltages lower than 80VAC, thus programming an approximately sinusoidal current waveform. For line
voltages larger than 80V, this scaled threshold is limited to a
DC threshold of fixed value.
Inductors L1 and L2
An effort was made to select low-cost off-the-shelf inductors for this design. A more compact design having higher
efficiency can be accommodated by the use of custom inductors.
A major disadvantage of the drum core inductors in this design is their large ambient field. Particularly the AC field of L1
may cause large eddy current losses in nearby conductive
elements, such as copper planes, heatsinks, capacitor foils,
etc., and may also cause modification of control signals on
the board.
Mounting L1 about 2 inches away from the board decreased
losses by about 1.75W, corresponding to a rise in efficiency
from 85.8% to 88.1%. Furthermore, the setpoint value of
output current shifted by about 10mA.
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3
DN-H05
EMI Filter
The EMI filter should be considered a best effort approach,
given the uncertainty regarding the final environment, layout
and choice of components. The EMI characteristics of individual components, pcb layout techniques and many other
factors affect to what extent low and high frequency energy
couple to the AC terminals of the driver.
Particularly the unshielded inductors L1 and L2, should be
kept well away from the inductors and the traces of the EMI
filter in order to avoid magnetic (transformer) coupling.
Capacitive coupling between traces, heatsinks, etc may
have a significant effect on circuit operation and EMI performance as well.
Dimmable vs Non-dimmable Setup
Note that certain measurements are taken with a non-dimmable version of the driver design. The design is turned into
the non-dimmable version by bypassing the damper circuit
(add of a wire jumper between test points P15 and P61), and
by increasing the VDD capacitor C51 from 220nF to 10µF.
It goes without saying that the non-dimmable version is not
to be used on a AC line circuit with attached dimmer switch.
Regular oscilloscope probes, i.e. with grounding clips, which
are non-isolated from safety ground, may affect circuit behavior adversely, particularly when dimming, even if the rest
of the experimental setup is isolated from safety ground by
isolation transformers and the like. Regular probes should
be used with caution.
Current waveforms were generally taken with active current
probes. The schematic shows in a number of places a pair
of adjoining testpoints for purpose of breaking the trace and
inserting a wire loop.
IOUT Regulation versus Output Voltage
Note that output current increases with decreasing output
voltage, see performance graph. Output rises from 350mA
to 450mA, when the output voltage drops from 160V to 10V,
a difference of about 100mA.
This result is inherent to the control scheme in use: peak
current control. Although it is desired to regulate the average
LED current to a fixed value, peak current control is preferred
due to its lower cost. The resulting peak to average error is
a function of the output voltage, which can be compensated
for with additional circuitry.
No damage but substantial flicker will result.
Measurement Techniques
A number of voltages of interest, such as the AC line voltage waveform, the voltage on the bulk energy storage capacitor VE31, were taken with the aid of a differential voltage
probe.
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
4
Supertex inc.
M81
SPP02
N50C3
P84
P83
P82
P81
CM11
7mH
0.6A
C83
100pF
630V
P85
C82
10nF
R84
100kΩ
Q84
2907A
P15
Damper Circuit
P13
P14
R83
100kΩ
MOV11
10mm
390V
F11
1A
TR5
R81
120kΩ
5W
AC
P11
AC
P12
INPUT
120V
500mA FL
P86
P16
Z85
12V
BR11
600V
1A
R86
100kΩ
R85
100kΩ
R87
100kΩ
Z87
12V
L23
390µH
C23
470nF
250V
R93
39kΩ
Q91
2222A
R92
1.0kΩ
R94
1.0MΩ
R62
1.0MΩ
R63
5.49kΩ
M91
ST1
NK60Z
P91
R91
1.0kΩ
C62
100pF
P62
2
Z61
7.5V
CS1
R65
604kΩ
R66
604kΩ
GATE
VIN
VDD
6
RT
8
C51
220nF
25V
3
GND
HV9931
4
5
PWM
+
E31
22µF
450V
CS2
7
IC51
C72
100pF
P72
R73
2.43kΩ
C71
R71
470mΩ 100nF
P71
D41
STTH
1R06A
R72
100kΩ
2xS
5.6mH
P33 P41 L41+L42
11.2mH
R31
1MΩ
[R51 = 191kΩ][TOFF = 8.52µs]
P51
M31
SPP04
N50C3
P32
R51
191kΩ
1
P01 P02
P52
D31
L31
P21 P31 STTH
1R06A 560µH
R64
100kΩ
P61
C21
470nF
250V
C61
R61
100nF 22mΩ
C22
470nF
250V
L21
390µH
Bleeder Circuit
L22
390µH
R31
1MΩ
P43
R75
10Ω
Q51
2222A
R54
100kΩ
Z72
91V
Z71
91V
P42
C41
10nF
250V
R52
100kΩ
R53
100kΩ
Q52
2907A
POS
P45
NEG
P44
OUTPUT
350mA
160V max
(56W FL)
DN-H05
Schematic Diagram
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5
DN-H05
Performance Graphs, AC Line Voltage
IOUT (Line Regulation)
450
mA
Efficiency
120
%
VOUT = 160VDC
IOUT = 350mADC
VOUT = 160VDC
IOUT = 350mADC
100
350
80
VRMS
250
60
80
100
VAC
120
140
160
THD
30
VRMS
60
60
80
100
120
%
VOUT = 160VDC
IOUT = 350mADC
140
160
PF
120
%
VAC
VOUT = 160VDC
IOUT = 350mADC
110
20
100
10
90
0
60
VRMS
80
100
VAC
Supertex inc.
120
140
160
80
60
VRMS
80
100
VAC
120
140
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
6
160
DN-H05
Performance Graphs, Output Voltage
IOUT
800
Efficiency
120
mA
%
VAC = 120VRMS
700
VAC = 120VRMS
100
88%
600
80
500
400
60
300
200
VDC
0
20
40
60
80
100
120
140
160
180
40
200
VDC
0
20
40
60
80
VOUT
THD
100
VAC = 120VRMS
120
140
160
180
200
PF
120
%
90
100
VOUT
%
VAC = 120VRMS
110
80
100
70
60
90
50
80
40
30
70
20
60
10
0
VDC
0
20
40
60
80
100
120
VOUT
Supertex inc.
140
160
180
200
50
VDC
0
20
40
60
80
100
120
140
160
VOUT
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7
180
200
DN-H05
Performance Graphs, Dimmer Switch Controlled
IOUT
500
mADC
600W Leviton Dimmer Switch
VAC = 120VRMS
400
300
200
100
°
0
0
20
40
60
80
100
120
140
160
180
Dimmer Conduction Angle
Supertex inc.
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8
DN-H05
VAC, IAC, IOUT (1/8) non-dimmable, 120VAC
VAC
IAC
IOUT
VAC, IAC, IOUT (2/8) dimmable, 120VAC
IAC
VAC
IOUT
VAC: 120VRMS IAC: 556mARMS THD: 12.2% PF: 98.2% VOUT: 160VDC
IOUT: 363mADC
VAC: 120VRMS IAC: 560mARMS THD: 13.3% PF: 98.2% VOUT: 160VDC
IOUT: 363mADC
VAC, IAC, IOUT (3/8) non-dimmable, 140VAC
VAC, IAC, IOUT (4/8) dimmable, 140VAC
VAC: 141VRMS IAC: 486mARMS THD: 13.9% PF: 97.7% VOUT: 161VDC
IOUT: 365mADC
VAC: 141VRMS IAC: 489mARMS THD: 14.6% PF: 97.7% VOUT: 161VDC
IOUT: 364mADC
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
9
DN-H05
VAC, IAC, IOUT (5/8) non-dimmable, 100VAC
VAC, IAC, IOUT (6/8) dimmable, 100VAC
VAC: 100VRMS IAC: 663mARMS THD: 12.5% PF: 98.4% VOUT: 159VDC
IOUT: 360mADC
VAC: 100VRMS IAC: 668mARMS THD: 13.8% PF: 98.4% VOUT: 159VDC
IOUT: 366mADC
VAC, IAC, IOUT (7/8) non-dimmable, 60VAC
VAC, IAC, IOUT (8/8) non-dimmable, 60VAC
VAC: 60VRMS IAC: 800mARMS THD: 4.8% PF: 99.8% VOUT: 132VDC
IOUT: 300mADC (output regulation is lost at 60VRMS)
VAC: 60VRMS IAC: 668mARMS THD: 8.5% PF: 99.6% VOUT: 122VDC
IOUT: 278mADC (output regulation is lost at 60VRMS)
Supertex inc.
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10
DN-H05
VDRAIN, VVIN, IL1, IL2
(1/4)
VDRAIN, VVIN, IL1, IL2
(2/4)
IL2
~100mAPP
~15VPP
VVIN
IL1
VDRAIN
VAC: 120VRMS (non-dimmable setup)
VAC: 120VRMS (non-dimmable setup)
VDRAIN, VVIN, IL1, IL2
VDRAIN, VVIN, IL1, IL2
(3/4)
VAC: 120VRMS (non-dimmable setup)
Supertex inc.
(4/4)
VAC: 120VRMS (non-dimmable setup)
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11
DN-H05
VGATE
VRS1, VRS2 Current Sense
VGATE
IL1
IL1
VRS1
VDRAIN
VRS2
VGATE
VAC: 120VRMS (non-dimmable setup)
VAC: 120VRMS (non-dimmable setup)
MOSFET Turn-on
MOSFET Turn-off
IL1
VDRAIN
VGATE
VAC: 120VRMS (non-dimmable setup) ) (RS1 = R61) (RS2 = R71)
Supertex inc.
(non-dimmable setup)
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12
DN-H05
CS1 Programming (1/7),
VAC = 140VRMS
CS1 Programming (2/7),
VAC = 120VRMS
VDRAIN
VIN
IAC
VREF,CS1
7.5VDC
(non-dimmable setup)
CS1 Programming (3/7),
(non-dimmable setup)
VAC = 100VRMS
(non-dimmable setup)
Supertex inc.
CS1 Programming (4/7),
VAC = 80VRMS
(non-dimmable setup)
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
13
DN-H05
CS1 Programming (5/7),
VAC = 60VRMS
VAC = 40VRMS
VAC: 120VRMS (non-dimmable setup) (RS1 = R61) (RS2 = R71)
(non-dimmable setup)
CS1 Programming (7/7),
CS1 Programming (6/7),
VAC = 20VRMS
VAC: 100VRMS IAC: 6686mARMS THD: 8.5% PF: 99.6% v VOUT: 122VDC
IOUT: 278mADC
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
14
DN-H05
VAC, IAC, VVIN, VDRAIN (1/4),
Angle = 165º
VAC, IAC, VVIN, VDRAIN (2/4),
Angle = 110º
VAC
IAC
VDRAIN
VVIN
IOUT: 335mADC
IOUT: 230mADC
VAC, IAC, VVIN, VDRAIN (3/4),
Angle = 65º
IOUT: 130mADC
VAC, IAC, VVIN, VDRAIN (4/4),
Angle = 20º
IOUT: 25mADC
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
15
DN-H05
IOUT Regulation (1/4),
Angle = 165º
IOUT Regulation (2/4),
Angle = 105º
VDRAIN
IAC
350mA
IOUT
IOUT: 335mADC
IOUT: 215mADC
IOUT Regulation (3/4),
Angle = 45º
IOUT: 70mADC
IOUT Regulation (4/4),
Angle = 20º
IOUT: 10mADC
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
16
DN-H05
VAC, VDAMPER, VVIN, Angle = 30º
VAC, VDAMPER, VVIN, VGATE,M81 (1/3),
Angle = 165º
VDAMPER
VAC
VAC
VVIN
VVIN
12VDC
VGATE,M81
VDAMPER
IOUT: 335mADC
IOUT: 335mADC
VAC, VDAMPER, VVIN, VGATE,M81 (2/3),
Angle = 120º
VAC, VDAMPER, VVIN, VGATE,M81 (3/3),
Angle = 25º
IOUT: 30mADC
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
17
DN-H05
Bill Of Materials
Qty
Ref
Description
Manufacturer
Mfr. Part Number
1
C41
Cap .01µF 250V Metal Polypro
EPCOS Inc
B32621A3103J
3
C21, C22, C23
Cap .47µF 250V Metal Polypro
EPCOS Inc.
B32652A3474J
2
C61, C71
Cap Ceramic .19 µF 16V 10% X7R 0805
Murata
GRM219R71C104KA01D
1
C82
Cap Ceramic 10000PF 50V 5% C0G 0805
Murata
GRM2195C1H103JA01D
1
C51
Cap .22µF 25V Ceramic X7R 0805
Panasonic ECG
ECJ-2YB1E224K
1
C83
Cap Ceramic 100PF 630V C0G 5% 1206
TDK Corporation
C3216C0G2J101J
2
C62, C72
Kemet
C0805C101K5GACTU
1
E31
Panasonic ECG
EEU-EB2W220
2
D31, D41
Diode Fast 600V 1A SMA
STMicroelectronics
STTH1R06A
2
Z85, Z87
Diode Zener 225MW 12V SOT23
ON Semiconductor
BZX84C12LT1
1
Z61
Diode Zener 225MW 7.5V SOT23
ON Semiconductor
BZX84C7V5LT1
2
Z71, Z72
Diode Zener 225MW 91V SOT23
ON Semiconductor
MMBZ5270BLT1
1
CM11
Filter Line 7MH 0.6A TYPE 16M
Panasonic ECG
ELF-16M060A
1
F11
Fuse T-LAG 1.00A 250V UL TR5
Wickmann USA
37411000410
2
HS81, HS31
Aavid Thermalloy
574602B03700
1
IC51
Supertex
HV9931LG
2
L41, L42
Inductor 5.6MH 0.45ARMS Axial
Renco
RL-1292-5600
1
L31
Inductor 560UH 0.8ARMS Radial
Renco
RL-1256-1-560
3
L21, L22, L23
Inductor HI Current Radial 390µH
JW Miller
6000-391K-RC
2
M31, M81
MOSFET N-CH 560V 4.5A TO-220AB
Infineon Technologies
SPP04N50C3
1
M91
MOSFET N-CH 600V 250MA SOT223
STMicroelectronics
STN1NK60Z
1
BR11
Rectifier Bridge 1AMP 600V DFS
Gen. Semiconductor/
Vishay
DF06S-E3\45
1
R61
Resistor .22Ω 1/4W 1% 0805 SMD
Susumu Co Ltd
RL1220S-R22-F
1
R71
Resistor .47Ω 1/4W 1% 0805 SMD
Susumu Co Ltd
RL1220S-R47-F
1
R75
Resistor 10.0Ω 1/8W 1% 0805 SMD
Panasonic ECG
ERJ-6ENF10R0V
1
R73
Resistor 2.43KΩ 1/8W 1% 0805 SMD
Panasonic ECG
ERJ-6ENF2431V
Cap Ceramic 100PF 50V NP0 0805
Cap 22µF 450V Elect EB Radial
Heatsink TO220 VER MNT W/TAB.69”
IC LED Driver SOIC-8
Supertex inc.
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18
DN-H05
Bill Of Materials (cont.)
Qty
Ref
Description
1
R63
7
Manufacturer
Mfr. Part Number
Resistor 5.49KΩ 1/8W 1% 0805 SMD
Panasonic ECG
ERJ-6ENF5491V
R52, R53, R54,
R64, R72, R84, Resistor 100KΩ 1/8W 1% 0805 SMD
R85
Panasonic ECG
ERJ-6ENF1003V
1
R51
Resistor 191KΩ 1/8W 1% 0805 SMD
Panasonic ECG
ERJ-6ENF1913V
2
R65, R66
Resistor 604KΩ 1/8W 1% 0805 SMD
Panasonic ECG
ERJ-6ENF6043V
1
R62
Resistor 1.00MΩ 1/8W 1% 0805 SMD
Panasonic ECG
ERJ-6ENF1004V
1
R93
Resistor 39KΩ 1/8W 5% 0805 SMD
Panasonic ECG
ERJ-6GEYJ393V
1
R91
Resistor 1.00KΩ 1/4W 1% 1206 SMD
Panasonic ECG
ERJ-8ENF1001V
3
R83, R86, R87
Resistor 100KΩ 1/4W 1% 1206 SMD
Panasonic ECG
ERJ-8ENF1003V
4
R31, R32, R92,
Resistor 1.00MΩ 1/4W 1% 1206 SMD
R94
Panasonic ECG
ERJ-8ENF1004V
Yageo Corporation
SQP500JB-120R
SUR Absorber 10MM 390VDC 2500A ZNR
Panasonic ECG
ERZ-V10D391
1
R81
Resistor 120Ω 5W 5% Wirewound
1
MOV11
2
Q51, Q91
Transistor GP NPN AMP SOT-23
Fairchild
Semiconductor
MMBT2222A
1
Q52, Q84
Transistor GP PNP AMP SOT-23
Fairchild
Semiconductor
MMBT2907A
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
©2012 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
011112
Supertex inc.
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
Supertex inc.
DN-H04
Design Note
Charting a HV9931 Driver Design
This application note allows you to generate or check a
HV9931 based driver design by using a set of graphs and
scaling rules. The graphs describe a base design for a range
of possible output voltages. Simple scaling rules allow you to
adapt the graphical data to the output current and switching
frequency of your target design. The driver design features
AC input with power factor correction, satisfying IEC harmonic limits for lighting equipment (EN61000-3-2 Class C).
The worksheet contains a sample design for your guidance.
Subsequently, scale the base design data to the desired output current and off-time of your target design. The worksheet
contains the scaling instructions for all parameters; either
multiply (M), divide (D) or leave the parameter unchanged
(ü) using the ratio of target current and target T-off time. A
sample calculation for a target of 500mA and 15µs is provided in the worksheet.
Please refer to application note AN-H52 for detailed information on HV9931 based LED Driver design. The data presented here is a graphical representation of the information
given in AN-H52.
Supplementary specifications of the base design are as follows:
►►
►►
►►
►►
Estimated Efficiency: 75%.
Output Current Switching Ripple: 30%.
Input Current 3rd Harmonic: 10%.
AC Line Frequency: 50Hz (240V); 60Hz (120V);
50Hz/60Hz (Universal).
►► RS1, RS2 Trip Voltage: 500mV.
►► RREF: 100kΩ.
Graphs are attached for the following three common design
cases:
1. 120VAC input
2. 240VAC input
3. Universal input
(85V…135V)
(200V…265V)
(85V…265V)
The graphs provide design information on the:
The graphs represent design data, such as component values, stress ratings, duty cycle, etc for a driver design at an
output voltage of your choice (up to 100V). As higher output
voltage represents higher output power, choose the lowest
output voltage compatible with your needs. Read the design
data from the curves and enter the values into the associated worksheet on the next page. These values correspond
to a base design with 1A of output current and an off-time of
10µs.
►► Components: (L1, L2, C1), (M1, D1, D2, D3, D4),
(RS1, RCS1, RS2, RCS2)
►► Timing: Duty Cycle, Switching Frequency
►► AC line: Line Current, Line Power
L2
D2
50/60Hz
D1
D4
L1
+
C1
100/120Hz
D3
M1
RCS1
CS1
Supertex inc.
RS1
RS2
RREF
RREF
VREF
HV9931
LEDS
RCS2
CS2
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DN-H04
Hints and Comments
Input Voltage Range:
cally, commercially available inductors are specified with DC
current ratings only, and the designer is left to guess the
performance under AC conditions. A good starting point for
design is to assume that the AC Rating of the inductor is four
times less than its DC rating.
Input Voltage Range, Lower Input Voltage:
L1 deserves particular attention because of its high current
swing. When designing inductors, the related magnetic flux
density swing can be adjusted to an acceptable level by
proper choice of core geometry and winding detail.
The base design will operate at voltages higher than 135V
(265V) without major change. The voltage ratings of the
MOSFET, the diodes and C1 should be increased accordingly.
Operation at input voltages lower than 85V (200V) is possible as well. This requires lowering of the inductance of L1
in order to avoid continuous current mode operation of the
input stage, which leads to severe input current waveform
distortion (see AN-H52 for general theory of operation). Do
not lower the inductance of L1 unnecessarily, as the reduction in L1 will bring about a need for higher voltage and current ratings of the power stage components. AN-H52 allows
you to study the impact of this change.
L1, Nominal Range:
Stay close to the calculated value when finalizing the design.
The computed value is a maximum value; using a larger value
results in CCM operation at low AC line voltage, thus causing the input current to become severely distorted. Using a
small value for L1 causes the current and voltage stress on
a number of components to increase. Keep in mind that the
standard tolerance of inductors is in the 10 to 20% range,
and that therefore the nominal value of the inductor should
be adjusted accordingly.
Should the initial target inductance differ considerably from
commercially available inductance values, then a commercially available value can nevertheless be accommodated by
adjusting the switching frequency, which is accomplished by
adjusting TOFF. The worksheet shows in which way L1 (and
couple of other parameters) can be changed by a change
in TOFF.
L1, Construction:
Particular attention should be paid to the design or rating of
inductor L1. Inductor L1 operates in discontinuous current
mode (DCM), that is, the current swings between zero and
the peak value within a single switching cycle. This large
current swing at high frequency (50 … 100kHz) may cause
significant losses, if not addressed properly.
The current of L1 swings 100% within a single switching
cycle; in contrast, the current of L2 swings about 30%. Typi-
Supertex inc.
Powdered iron cores and ferrite cores have been used with
success. Low cost drum core types (surface mount or leaded type) can be used as well, and are especially convenient
during the prototyping stage due to their widespread availability. The ambient magnetic field of these unshielded types
may induce voltage in nearby circuits and other elements
casuing shift in operating point and eddy current losses. This
effect can be quite noticeable, and forces placement of such
inductors well away form control circuitry, copper planes,
heatsinks, capacitors, etc. The ambient field can cause excessive EMI as well, which may cause non-compliance with
EMI standards.
If space is at a premium, cores with a closed magnetic core
or having shielding, such as toroids or EE cores should be
used, which tend to be more expensive, but cause lower
losses and allow tighter packaging.
Switching Frequency, Efficiency, Size:
Switching frequency can be scaled up or down based on
the typical trade-offs between cost, size and efficiency. An
efficiency of 75% was assumed. Higher efficiencies are attainable by lowering switching losses and conduction losses
which generally means the use of larger / more expensive
components.
RS2:
Note the assumed circuit location of RS2; in older schematics RS2 is located in the path between D3 and CO and
thus carries the load current at all times; in newer schematics RS2 is located in the path between D3 and circuit ground
and carries current only during the on-time of the MOSFET.
The new location leads to significantly less power dissipation and is therefore preferred. The graph of RS2 dissipated
power reflects the new circuit location.
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2
DN-H04
Line Frequency:
When the line frequency differs from 60Hz (50Hz), then the
capacitance of C1 should be adjusted. C1 is inversely related to the line frequency; e.g. at a line frequency of 400Hz,
C1 can be 6.7 (8) times smaller. Note that the RMS Current
Rating of commercially available capacitors may not allow
you to fully exploit this potential reduction.
RS1, RS2:
RS1 (or RS2) can be scaled up or down to match a commercially available value; RCS1 (or RCS2) should be scaled up
or down with the same factor.
A Current Threshold voltage of 500mV provides a good
starting point for the majority of applications. The threshold
voltage can be increased in order to lower noise sensitivity
or reduce the impact of the CS1 and CS2 offset voltage, or
can be decreased in order to lower sense resistor power
dissipation.
Non-Electrolytic Capacitor Designs:
This procedure documents design which does not incorporate the T-off modulation technique as described in AN-H52.
T-off modulation allows further reduction of AC line current
harmonics, at the expense of a few (low cost) components.
An alternate use of the modulation technique is reduction of
C1 capacitance, while maintaining a similar level of harmonics. Reduction of capacitance on the order of five times or
more is viable. The capacitance reduction may warrant replacement of an electrolytic capacitor with a film or a ceramic
capacitor.
Supertex inc.
Non-electrolytic capacitors are preferable in situations where
high temperature operation or long life is desired. Note that
a switch to non-electrolytic capacitors does not necessarily
mean that physical size reduces as well, which is an area
where electrolytic capacitors excel.
Another price to pay is that less capacitance brings about
an increase in the 100/120Hz ripple on the C1 capacitor,
which requires corresponding increases of the voltage rating
of all surrounding components. E.g., a reduction by a factor
of five results in five times more 100/120Hz ripple. Note that
the calculation of L1 assumes that C1 is fairly large, corresponding to a C1 voltage that is quasi DC. This assumption
may not be valid anymore, and with large enough ripple, the
capacitor voltage in the ripple valley may be low enough so
as to cause continuous conduction mode operation during
part of the AC line cycle, which is undesirable. This can be
remedied by a reduction in value of L1, which increases the
margin to CCM operation for a given capacitor voltage, and
at the same time raises the DC operating point for the capacitor giving additional margin.
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3
DN-H04
Fig. 1 - AC Line Power (W) vs Output Voltage (V)
120
110
120, 240, UNI
Output Voltage (V)
100
90
80
70
60
50
40
30
20
10
0
25
50
75
AC Line Power (W)
100
125
150
Fig. 2 - Max RMS Line Current (A) vs Output Voltage (V)
120
110
240
Output Voltage (V)
100
120, UNI
90
80
70
60
50
40
30
20
10
0
0
0.25
0.50
0.75
1.00
1.25
1.50
1.75
Max RMS Line Current (A)
Fig. 3 - L1 Inductance (µH) vs Output Voltage (V)
120
110
Output Voltage (V)
100
120, UNI
240
90
80
70
60
50
40
30
20
10
0
Supertex inc.
(300)
300
(707)
350
400
450
500
550
L1 Inductance (µH)
600
650
700
750
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4
DN-H04
Fig. 4 - L1 Peak Current - LL (A) vs Output Voltage (V)
120
110
Output Voltage (V)
120, UNI
240
100
90
80
70
60
50
40
30
20
10
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
L1 Peak Current LL (A)
5.5
6.0
6.5
7.0
7.5
Fig. 5 - L1 RMS Current - LL (A) vs Output Voltage (V)
120
110
240
Output Voltage (V)
100
120, UNI
90
80
70
60
50
40
30
20
10
0
0.5
1.0
1.5
2.0
2.5
L1 RMS Current LL (A)
3.0
Fig. 6 - L2 Inductance (µH) vs Output Voltage (V)
120
110
120, 240, UNI
Output Voltage (V)
100
90
80
70
60
50
40
30
20
10
0
500
1000
1500
2000
2500
3000
3500
4000
L2 Inductance (µH)
Supertex inc.
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5
DN-H04
Fig. 7 - L2 Peak Current (A) vs Output Voltage (V)
120
110
120, 240, UNI
100
Output Voltage (V)
90
80
70
60
50
40
30
20
(1.15)
10
0
0.25
0.5
0.75
1
1.25
1.5
1.75
2
2.25
1.50
1.75
2.00
2.25
L2 Peak Current (A)
Fig. 8 - L2 RMS Current (A) vs Output Voltage (V)
120
110
120, 240, UNI
Output Voltage (V)
100
90
80
70
60
50
40
30
20
(1.12)
10
0
0.25
0.50
0.75
1.00
1.25
L2 RMS Current (A)
Fig. 9 - C1 Capacitance (µF) vs Output Voltage (V)
120
110
Output Voltage (V)
100
240
120, UNI
90
80
70
60
50
40
30
20
10
0
Supertex inc.
35
40
45
50
C1 Capacitance (µF)
55
60
65
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6
DN-H04
Fig. 10 - C1 RMS Current - LL (A) vs Output Voltage (V)
120
110
Output Voltage (V)
240
120, UNI
100
90
80
70
60
50
40
30
20
10
0
0.25
0.50
0.75
1.00
1.25
1.50
1.75
2.00
C1 RMS Current LL (A)
Fig. 11 - C1 Max Voltage - HL (V) vs Output Voltage (V)
120
110
120
Output Voltage (V)
100
240
UNI
90
80
70
60
50
40
30
20
10
0
50
100
150
200
250
300
C1 Max Voltage HL (V)
350
400
450
Fig. 12 - C1 Min Voltage - LL (V) vs Output Voltage (V)
120
110
120, UNI
Output Voltage (V)
100
240
90
80
70
60
50
40
30
20
10
0
25
50
75
100
125
150
175
200
225
C1 Min Voltage LL (V)
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
7
DN-H04
Fig. 13 - M1 Peak120
Current - LL (A) vs Output Voltage (V)
110
240
100
120, UNI
Output Voltage (V)
90
80
70
60
50
40
30
20
10
0
0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 8.0 8.5 9.0
M1 Peak Current LL (A)
Fig. 14 - M1 Max RMS Current - LL (A) vs Output Voltage (V)
120
110
Output Voltage (V)
120, UNI
240
100
90
80
70
60
50
40
30
20
10
0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
M1 Max RMS Current LL (A)
Fig. 15 - M1 Peak Drain Voltage - HL (V) vs Output Voltage (V)
120
110
Output Voltage (V)
240
120
100
UNI
90
80
70
60
50
40
30
20
10
0
150
200
250
300
350
400
450
500
550
600
650
700
750
800
850
M1 Peak Drain Voltage HL (V)
Supertex inc.
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8
DN-H04
Fig. 16 - D1 Ave Forward Current - LL (A) vs Output Voltage (V)
120
110
240
Output Voltage (V)
100
120, UNI
90
80
70
60
50
40
30
20
10
0
0.05 0.10 0.15 0.20 0.25 0.30 0.35 0.40 0.45 0.50 0.55 0.60 0.65 0.70 0.75
D1 Avg Forward Current LL (A)
Fig. 17 - D1 Max Reverse Voltage - HL (V) vs Output Voltage (V)
120
110
120
Output Voltage (V)
100
240
UNI
90
80
70
60
50
40
30
20
10
0
200
250
300
350
400
450
500
550
600
650
D1 Max Reverse Voltage HL (V)
700
750
800
850
Fig. 18 - D2 Ave Forward Current - LL (A) vs Output Voltage (V)
120
110
240
Output Voltage (V)
100
120, UNI
90
80
70
60
50
40
30
20
10
0
0.10 0.15
0.20 0.25
0.30 0.35
0.40 0.45
0.50 0.55
0.60 0.65 0.70
D2 Avg Forward Current LL (A)
Supertex inc.
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9
DN-H04
Fig. 19 - D2 Max Reverse Voltage - HL (V) vs Output Voltage (V)
120
110
120
Output Voltage (V)
100
240, UNI
90
80
70
60
50
40
30
20
(191)
10
0
50
100
150
(375)
200
250
300
350
400
D2 Max Reverse Voltage HL (V)
Fig. 20 - D3 Ave Forward Current - HL (A) vs Output Voltage (V)
120
110
Output Voltage (V)
100
240
120
UNI
90
80
70
60
50
40
30
20
10
0
0.55
0.60
0.65
0.70
0.75
0.80
0.85
0.90
D3 Ave Forward Current HL (A)
0.95
1.00
Fig. 21 - D3 Max Reverse Voltage - HL (V) vs Output Voltage (V)
120
110
120
Output Voltage (V)
100
240
UNI
90
80
70
60
50
40
30
20
10
0
Supertex inc.
50
100
150
200
250
300
350
D3 Max Reverse Voltage HL (V)
400
450
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10
DN-H04
Fig. 22 - D4 Ave Forward Current - LL (A) vs Output Voltage (V)
120
110
240
Output Voltage (V)
100
120, UNI
90
80
70
60
50
40
30
20
10
0
0.25
0.50
0.75
1.00
1.25
1.50
1.75
2.00
2.25
D4 Ave Forward Current LL (A)
Fig. 23 - RS1, RS2: Resistance (mΩ) vs Output Voltage (V)
Output Voltage (V)
200
RS1
100
70
50
RS2
30
20
10
7
5
3
2
(435)
1
0
100
200
300
400
500
600
700
800
RS1, RS2 Resistance (mΩ)
900
1000 1100 1200
Fig. 24 - RS1, RS2: Power Dissipation - LL (mW) vs Output Voltage (V)
120
110
Output Voltage (V)
RS1
RS2
100
90
80
70
60
50
120, 240, UNI
40
30
20
10
0
0
25
Supertex inc.
50
75
100 125 150 175 200 225 250 275 300 325 350 375 400
RS1, RS2 Power Dissipation LL (mW)
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11
DN-H04
Fig. 25 - RCS1, RCS2: Resistance (kΩ) vs Output Voltage (V)
120
110
RCS2
Output Voltage (V)
100
RCS1
90
80
70
60
50
40
30
20
(6.65)
10
0
6.00
6.25
6.50
6.75
(8.00)
7.00
7.25
7.50
7.75
8.00
8.25
8.50
RCS1, RCS2 Resistance (kΩ)
Fig. 26 - RT Resistance (kΩ) vs Output Voltage (V)
120
110
120, 240, UNI
Output Voltage (V)
100
90
80
70
60
50
40
30
20
(228)
10
0
220
221
222
223
224
225
226
RT Resistance (kΩ)
227
228
229
230
Fig. 27 - Min Duty Cycle - HL (%) vs Output Voltage (V)
120
110
UNI
Output Voltage (V)
100
240
120
90
80
70
60
50
40
30
20
10
0
0
Supertex inc.
5
10
15
20
25
30
35
Min Duty Cycle HL (%)
40
45
50
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12
DN-H04
Fig. 28 - Max Duty Cycle - LL (%) vs Output Voltage (V)
120
110
240
Output Voltage (V)
100
120, UNI
90
80
70
60
50
40
30
20
10
0
0
5
10
15
20
25
30
35
40
45
Max Duty Cycle LL (%)
50
55
60
65
70
Fig. 29 - Min Switching Frequency - LL (kHz) vs Output Voltage (V)
120
110
120, UNI
Output Voltage (V)
100
240
90
80
70
60
50
40
30
20
10
0
30
35
40
45
50
55
60
65
70
75
80
85
Min Switching Frequency LL (kHz)
90
95
100
Fig. 30 - Max Switching Frequency - HL (kHz) vs Output Voltage (V)
120
110
Output Voltage (V)
100
120
UNI
240
90
80
70
60
50
40
30
20
10
0
50
Supertex inc.
55
60
65
70
75
80
85
90
Max Switching Frequency HL (kHz)
95
100
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13
DN-H04
Design Worksheet
Base Design
IO
TOFF
Target Design
120V 24V
0.50
1.5
0.5A 15µs
31.8
M
ü
15.9
W
LL
0.35
M
ü
0.18
A
3
-
300
D
M
954
µH
Peak I
4
LL
2.58
M
ü
1.29
A
RMS I
5
LL
1.05
M
ü
0.52
A
L
6
-
915
D
M
2745
µH
Peak I
7
-
1.15
M
ü
0.575
A
RMS I
8
-
1.12
M
ü
0.56
A
C
9
-
57.1
M
ü
28.55
µF
RMS I
10
LL
1.02
M
ü
0.51
A
Max V
11
HL
105
ü
ü
105
V
Min V
12
LL
62
ü
ü
62
V
Peak ID
13
LL
3.69
M
ü
1.85
A
RMS ID
14
LL
1.5
M
ü
0.75
A
Peak VD
15
HL
295
ü
ü
295
V
Ave I
16
LL
0.41
M
ü
0.205
A
Peak VR
17
HL
295
ü
ü
295
V
Ave I
18
LL
0.38
M
ü
0.19
A
Peak VR
19
HL
191
ü
ü
191
V
Ave I
20
HL
0.72
M
ü
0.36
A
Peak VR
21
HL
104
ü
ü
104
V
Ave IF
22
LL
0.72
M
ü
0.36
A
R
23
-
188
D
ü
3.76
mΩ
P
24
LL
89
M
ü
44.5
mW
R
23
-
435
D
ü
348
mΩ
P
24
-
174
M
ü
87
mW
RCS1
R
25
-
8.00
ü
ü
8.00
kΩ
RCS2
R
25
-
6.65
ü
ü
6.65
kΩ
RT
26
-
228
ü
M
342
kΩ
Min D
27
HL
31.5
ü
ü
31.5
%
Max D
28
LL
39.9
ü
ü
39.9
%
Min FS
29
LL
61.8
ü
D
41.2
kHz
Max FS
30
HL
69.0
ü
D
46.0
kHz
Item
Parameter
Fig
AC
Line
AC
Line
P
1
-
Max RMS I
2
L
L1
L2
C1
M1
D1
D2
D3
D4
RS1
RS2
Timing
Units
LL (Low Line), HL (High Line), Base IO (1A), Base TOFF (10µs), M (Multiply), D (Divide), ü (No Change)
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
©2012 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
011212
Supertex inc.
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com
Supertex inc.
AN-H52
Application Note
HV9931 Unity Power Factor
LED Lamp Driver
Introduction
Development of high-brightness light emitting diodes
(LED) revolutionized the lighting industry in the recent
years. Semiconductor light sources replace incandescent
bulbs in an increasing number of applications due to their
unsurpassed reliability and efficiency. Such applications
include traffic signals, emergency lighting, hard-to-reach
lighting fixtures, automotive lighting, accent and decorative
lighting. Many of these applications demand off-line power
drivers capable of regulated DC output current, low DC
output voltage and input unity power factor.
A flyback converter can become a simple solution for these
types of applications. When operating in discontinuous
conduction mode, a flyback converter inherently provides
a good power factor since the peak current in its inductor
is proportional to the instantaneous input voltage. However,
a very large electrolytic smoothing capacitor is needed at
the load in order to attenuate the rectified AC line ripple
component of the output current. Low dynamic resistance
of LEDs aggravates the problem even further. There are
power topologies that can resolve this problem by cascading
converter stages using a single active switch. Most of
these topologies include an input boost converter stage for
shaping the input current. Hence they require a transformer
with a high step-down turn ratio in order to drive low voltage
LEDs. A power transformer would be needed even when
galvanic isolation of the output in not required. Overall power
efficiency, cost and reliability can be improved by using a
step-down buck-boost input stage.
Fig 1: Power conversion topology*
*This topology includes intellectual property of Supertex, Inc. A paid up license is offered for application of the HV9931 product.
C1
+
-
D4
+
D3
L1
CO
+
LED
M1
A simple transformerless power converter is shown in Fig.1.
Its input buck-boost stage consisting of L1, C1, D1 and D4 is
cascaded with an output buck stage including L2, D2, D3 and
CO. Both converter stages share a single power MOSFET
M1. The input buck-boost stage operates in discontinuous
conduction mode (DCM), while the output buck stage runs
in continuous conduction mode (CCM). Both converter
stages can operate as step-down voltage converters. The
overall step-down ratio is a product of the step-down ratios
of the two converter stages. Thus a high step-down ratio is
achieved without using a transformer. Steady-state voltage
and current waveforms of this converter are shown in Fig.2.
Switching the MOSFET M1 on applies the rectified AC line
voltage across L1. Current in L1 rises linearly. At the same
time, the bulk capacitor C1 powers the output buck stage.
Supertex inc.
L2
D2
D1
(Note the negative polarity of the voltage across C1 with
respect to ground when M1 is on.) The current in L2 ramps
up. The current paths for this switching state are shown in
Fig.3a.
When M1 turns off, D1 becomes forward-biased. The input
inductor current diverts into C1. At the same time, the
current in the output inductor L2 finds its way through D3.
(See Fig. 3b.). The current in L1 ramps down. As soon as the
current reaches zero, the diode D1 becomes reverse-biased
and prevents the current in L1 from reversing. (The reverse
current flow back into the input source would otherwise
cause harmonic distortion of the input current and reduction
in the overall efficiency.) Fig.3c depicts this switching state.
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Fig 2: Voltage and current switching waveforms
ON
M1
OFF
VIN
VL1
0
-VC3
ILI(PK)
IL1
0
V0
0
VL2
-VC3
ILI(PK)
ID1
0
IL2(PK)
IL2
ID2
0
IL2(PK)
I0
0
VIN + VC3
VIN
VDS(M1)
0
ILI(PK) + IL2(PK)
IM1
IL2(PK)
ID3
0
I0
0
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Reff = 2.L1.FS/D2 is the effective input resistance of the
converter. This feature of the switching converter of Fig. 1
ensures low harmonic distortion of the input AC current and
near-unity power factor. Other techniques using the HV9931
that can reduce harmonic distortion even further will be
discussed below.
The value of the bulk capacitor C1 needs to be large enough
to attenuate rectified AC line ripple. Then the duty cycle D
and the switching frequency FS can be assumed constant
over the AC line cycle. In this case, both the peak current
IL1(PK) in L1 and the average input current IIN are directly
proportional to the input voltage VIN. (See Fig. 4.) The factor
Fig 3: Switching states of the converter:
(a) energizing L1 and L2, (b) de-energizing L1 and L2, (c) dead time of L1.
D1
D2
L2
+
L1
D4
+
C1
CO
+
D3
-
LED
M1
(a)
D2
D1
L2
+
+
-
D4
C1
CO
L1
LED
+
D3
M1
(b)
L2
+
+
D4
-
C1
CO
L1
D3
LED
+
M1
(c)
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Fig 4: Waveforms explaining the unity power factor feature of the HV9931.
VIN
IL1(PK) =
D • VIN
L1 • FS
IIN =
=
=
1
• D • IL1(PK) =
2
D2
• VIN =
2L1 • FS
VIN
REFF
D=
TON
TS =
TON
TS
= TON • FS
1
FS
LED Current Control Loop
The HV9931 is a peak current control IC that is specifically
designed for optimally controlling the non-isolated singlestage PFC converter described above. A typical application
circuit of the HV9931 is shown in Fig. 5.
where α = 40pF, τO = 880ns. Connecting the resistor from RT
to GATE programs constant off-time:
Upon application of 12 - 450V at VIN, the built-in high
voltage regulator circuit seeks to regulate 7.5V ± 5% at VDD.
The circuit is equipped with an under-voltage protection
comparator (UVLO) that inhibits switching until a threshold
voltage is reached at VDD. A 0.5V hysteresis is included to
prevent oscillation.
It can be shown that the fixed off-time operating mode: a)
reduces the voltage stress at C1; b) improves input AC ripple
rejection; c) inherently introduces frequency jitter that can
help reduce the size of the input EMI filter required. Hence,
we will assume the fixed off-time mode for the purpose of
this discussion.
TOFF = α • RT + τO
(2)
As soon the start-up threshold is reached at VDD, an The control circuit further includes two comparators for
internal oscillator circuit is enabled. The output signal of the programming peak currents in L1 and L2. Both comparators
oscillator triggers a PWM latch. The GATE output becomes use the ground potential (GND) as a reference and can
high, the power MOSFET Q1 switches ON. The oscillator be used to monitor voltage signals of negative polarity
circuit can be programmed with a single resistor connected with respect to GND. A blanking delay of 215ns is added
to RT for either constant switching frequency or fixed off- to prevent false tripping the comparators due to the circuit
time operation. In the fixed off-time mode, the oscillator will parasitics. The currents iL1 and iL2 that trip the comparators
set the PWM latch after a programmed time period following can be computed as:
the turn-off of the GATE output. In order to program the
VREF • RCS
HV9931 for constant frequency operation, the timing resistor iL(PK) =
(3)
RREF • RS
needs to be connected between RT and GND. The switching frequency in this case can be calculated using the following where VREF is an external reference voltage. We will use
equation:
VREF = VDD as an example. When either of the comparators
1
FS =
(1) detects negative input voltage at its CS input, the PWM latch
α • RT + τO
resets, the GATE output becomes low, and the MOSFET Q1
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turns off. Note, that since L2 is assumed to operate in CCM:
1/2ΔiL2 that needs to be accounted for when programming
the resistor divider RREF2/RCS2. Fortunately, this error is
nearly constant for any input voltage at fixed TOFF and it is
relatively small compared to iL2 (15% typ.) Hence the ripple
will have a minimal effect on the overall regulation of the
output current. The error is however a function of the output
voltage variation and the inductance value tolerances of L2.
1
iL2(PK) = iL2 +
• ∆iL2
(4)
2
where iL2 is the average current, and ΔiL2 is the peak-to-peak
current ripple in L2. Thus the constant peak current control
used in the HV9931 introduces a peak-to-average error
Fig 5: Typical HV9931 off-line PFC LED Driver application circuit
D1
D4
VIN
L1
C1
iL1
+ VC1 -
CIN
~AC
~AC
L2
D2
-
iL2
D3
M1
VO
RS1
RS2
- VS1 +
- VS2 +
RT
+
RCS2
GATE
PWMD
RCS1
RT
OSC
S Q
R
CS1
RREF1
VIN
-
-
+
+
REG
RREF2
VDD
7.5V
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GND
HV9931
CDD
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Power Converter Design
Designing L1
We need to design the input buck-boost stage to operate Since our discussion is limited to the constant off-time case,
in DCM for any given line and load condition to ensure low let us express equation 8 in terms of TOFF = TON • (1-D) / D:
distortion of the input current and stability of the control loop.
VAC • √2 • TOFF
D
Therefore, let us assume that the current in L1 becomes IL1(PK) =
•
(9)
L1
1 - D
critically continuous at full load and some minimum operating AC line voltage VAC(MIN). Naturally, this boundary conduction Finally, combining the equations (5), (6), (7) and (9) and
mode (BCM) condition occurs at the peak of each half-wave solving for the inductance value gives:
of the input AC current. If we assume a unity power factor (PF
VAC(MIN) • √2 • TOFF
= 1), this boundary condition will then coincide with the peak L1 =
(10)
4 • IO input voltage VAC(min) • √2. Since both converter stages are in CCM, the ratio between the output and the input voltage (Note, that the critical inductance L1 corresponding to the
can be expressed as:
boundary conduction at VAC(min) and Io is independent of the
output voltage or the efficiency of the converter.)
VO
DMAX • η1
=
• DMAX • η2
(5)
VAC(MIN) • √2
1 - DMAX
The designer must be careful when considering standard
inductors for L1 or designing a custom one. Since L1
conducts discontinuous current, magnetic flux excursion in
the core material can be quite significant. Hence the design
of L1 is limited by the power dissipation in the magnetic core
material rather than by the saturation current of the inductor
selected.
D2MAX • η
=
1 - DMAX
where η1 and η2 are the corresponding efficiencies of the
input buck-boost stage and the output buck stage. The
overall converter efficiency equals η = η1 • η2. The duty ratio
D of the switch M1 is the greatest at this condition. (Duty
ratio is defined as D = TON / TS, where TON is the on-time of
M1, and TS is the switching period.)
Designing C1
Selecting the capacitance value for C1 is based on the input
harmonics limits required for a specific application. Lighting
products are sold in large quantities, and thus these high
The input AC line current can be obtained from the output volume products can potentially have a high impact on the
LED current IO and the output voltage VO as:
low voltage public supply system. The European EN 610003-2 Class C limits are comparable to the limits imposed
VO • IO
IAC =
(6) by ANSI C82.77 standards in the U.S. market, and restrict
VAC • η
overall current harmonics to approximately 33%. Both the
Class C and ANSI standards limit the 3rd harmonic current
On the other hand,
of lighting products to ~ 30%. The regulations for LED-based
traffic signal heads are generally stricter and require total
D
IAC • √2 =
• IL1(PK) (7) harmonic distortion (THD) to be less than 20% (ITE VTCSH
2
Part 2).
where the peak current IL1(PK) in L1 can be given as:
The prevalent component of the AC ripple voltage across C1
is the 2nd AC line harmonic. This ripple causes modulation of
the duty cycle according to:
VAC • √2 • TON
IL1(PK) =
(8) VO
L1
D(t) =
(11)
η
•
V
2
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where VC is voltage across C1. On the other hand, the input
AC current can be expressed as:
Recalling that D = VO / (η2 • VC) and using (16), we can
determine the voltage at C1 for a given VAC:
VO
VC =
• (1 + √1 + δ ) (18)
VAC • √2 • D(t)2 • TOFF
2 • η2
IAC(t) =
• sin(2π • FAC • t) (12) 2L1 • (1 -D(t))
We have assumed that KC = íC/VC << 1. This condition is met
where FAC is the AC line frequency. Let us assume a small by selecting C1 large enough so that the AC ripple voltage at
2nd harmonics ripple voltage νC across C1, so that the C1 is low. Therefore, C1 decouples the bulk of the AC ripple
current at the output of the input converter stage. Averaged
voltage at C1 can be written as:
over a switching cycle, this current can be written as:
VC(t) = VC - νC • sin(4π • FAC • t)
(13)
VAC(t) • IAC(t) • η1
I2(t) =
(19)
VC
where í << V . Substituting (11) and (13) in (12) will produce C
C
a displaced fundamental term and a 3rd harmonic term in the
AC line current. It can be shown from the resulting equation
that the 3rd harmonic distortion of the input AC line current
for a given relative 2nd harmonic ripple KC = íC/VC <<1 is:
where Vc is determined from (18). The AC line current IAC(t)
is given by the equation (12). Then, under the assumptions
made above, the AC component of I2(t) contains 2nd harmonic
current only. This AC current in C1 can be expressed as:
2
ΔI3rd
1 2-D
D2 η1 • V AC • TOFF
• cos(4π • FAC • t) K3 =
≈
•
• KC
(14) IC(t) = - 1 - D •
2 • L1 • VC IAC
2 1 - D (20)
Thus every 1% of 2nd harmonic ripple at C1 will generate at Substituting D and V from (16) and (18) gives:
C
least 1% of 3rd harmonic component in the AC line current
2 • IO
even when the duty cycle is small.
IC(t) = • cos(4π • FAC • t)
(21)
1 + √1 + δ
Let us determine the capacitance value of C1 needed to limit
the 3rd harmonic distortion to some given K3. Equations (6), Relative ripple voltage at C1 can be calculated as KC = IC(PK)
(7) and (9) together can be solved for the duty cycle D at any • ZC/VC, where IC(PK) is the amplitude of IC(t) and ZC = (4π
• FAC • C1)-1 is the impedance of C1 at 2 • FAC. Substituting
VAC within the operating range.
VC from (18), we obtain:
VO • IO • L1
V2AC • TOFF • η
D=
1+
-1
1
η2 • IO
•
VO • IO • L1
V2AC • TOFF • η
IC(t) =
•
(22)
2
(1 + √1 + δ)
π • FAC • C1 • VO
Let us introduce a parameter δ as follows:
Solving the equation (22) for C1 and substituting KC from
2 • V2AC • TOFF • η
δ=
(15) (17) we get:
L1 • VO • IO η2 • IO
1
C1
=
(23)
π • FAC • K3 • VO
1
Then the duty cycle can be expressed as:
δ• 1+
1+δ
2 • (√1 + δ - 1)
D=
(16)
δ
We can rewrite the equation (14) now as:
[
]
1
K3 = KC •
(17)
1 - 1
√1 + δ Supertex inc.
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of íC(t). In order to determine the modulation needed, we
can expand the equation (24) in Taylor series in íC(t). Then
we can negate the 1st order term of the resulting expansion
in íC(t) by modulating TOFF inverse proportionally. This
technique can achieve very good results since the linear
term is responsible for the displaced fundamental and the
bulk of 3rd harmonic in IAC(t).
The RMS value of the switching current in C1 can be
calculated using the following equation:
64
VO
IC(SW) = IO •
+D
•
9 • π • η • η1 VAC • 2
(23a)
The RMS value of the second AC line harmonic is derived
from (21):
One circuit implementation of this ripple cancellation
IO • √2
IC(LINE) =
(23b) feedback technique is shown in Fig.6. A charge pump circuit
1 + √1 + δ
consisting of the capacitor CA and the diodes D5 and D6
performs level translation of VC to the ground potential. The
voltage at C1 is reconstructed across the capacitor CB. The
Using Non-Electrolytic Capacitors for C1
The lifetime and the reliability of high brightness LEDs is values of CB and the bleeder resistor RB are selected such
remarkable. However, unlike incandescent light sources, that (2πRBCB)-1 >> 2 • FAC to preserve the ripple voltage
LEDs generate conducted heat that needs to be dissipated íC(t). Capacitor CFF decouples the DC component of VC. The
within the lighting fixture. A power supply will be expected to back-to-back connected Zener diodes D8 and D9 clamp the
function at elevated temperatures and match the lifetime of feedback voltage during initial charging of CFF. A proportional
the LEDs when such power supply is integrated within the LED AC current íC(t)/RFF then modulates the off-time programmed
fixture. In many cases, this requirement rules out electrolytic by the RT pin of the HV9931.
capacitors commonly used in power supplies. As a “rule of
α • (VRT - VD)
+ τO (25)
thumb”, electrolytic capacitors suffer two times reduction TOFF(t) =
V
V
v
(t)
RT
D
of their life with every 10°C operating temperature rise. - C R
RFF
T
Therefore, it is desirable to be able to use a non-electrolytic
capacitor for C1. Metallized polyester or PEN film capacitors can be considered for C1 as the most size and cost efficient where VD = 0.7V, VRT ≈ 6.5V. The capacitance value of CFF
replacement of aluminum electrolytic capacitors. However, is selected such that (2πRFFCFF)-1 << 2 • FAC. Substituting
they contribute a substantially higher cost per microfarad TOFF(t) given by (25) in the 1st order Taylor series term of
compared to electrolytic capacitors having similar voltage the equation (24) and solving it for RFF gives the feedback
ratings. Thus, our design goal is to minimize the value of C1 resistor needed to cancel harmonic distortion of the input AC
current.
while retaining low harmonic distortion.
α • RT2 • VO
δ
•
As C1 becomes smaller, the condition of KC << 1 is no longer RFF =
4 • √1 + δ η2 • (VRT - VD ) • (α • RT + τO ) met. Thus, we cannot use the equation (14) for calculating
(26)
the 3rd harmonic distortion coefficient K3. However, the equations (11) and (12) are still valid. We will combine these
two equations and use VC(t) = VC + íC(t), where íC(t) is the (The derivation of the equation (26) has been omitted for the
sake of simplicity.)
AC ripple voltage at C1.
Note, the circuit of Fig.6 contributes a positive feedback
VAC • √2 • VO2 • TOFF
whose gain must not exceed the negative feedback gain
IAC(t) =
2L1
•
η
•
(V
+
v
(t))[(V
+
v
(t))
•
η
V
]
2
C
C
C
C
2
O
imposed by the equation (11) to avoid loop oscillation!
• sin(2π • FAC • t)
(24)
We can see from the equation (24) that harmonic distortion
of IAC(t) can be reduced by modulating TOFF as a function
Supertex inc.
Therefore, perfect cancellation of harmonic distortion can
only be achieved at a single point corresponding to the
highest VO and VAC. Thus, the equation (26) must use
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Fig 6: Feedback circuit improving the power factor and THD
VIN
D4
D1
D2
L1
HV9931
RT
VO
~AC
+
RS2
RS1
GATE
RT
D5
CA
CFF
RFF
D8
D9
D6
RB
CB
VO(max) and VAC(max). Nevertheless, a dramatic reduction of
C1 can still be achieved (up to several times depending on
the input AC voltage range required).The designer must not
forget another constraint limiting the minimum value of C1.
The voltage at C1 must not fall below the output voltage (VC
> VO) in order to avoid interruptions of the output current.
D3
M1
CIN
~AC
L2
C1
D5
must satisfy:
1
ISAT > IO +
∆IL2
2
(29) Power Semiconductor Components
Let us calculate the voltage and the current ratings of the
MOSFET M1 and the rectifiers D1-D4. The current in M1
VC(MIN) - VO
KC(MAX) <
(27) is composed from the currents in the inductors L1 and L2.
VC(MIN)
Hence, the RMS current in M1 can be computed as:
Calculating L2
DMAX • IL1(PK)2
ID(M1) =
+ DMAXIO2
Calculating the value of the output filter inductor L2 is simple. (30)
6
The designer must decide on the amount of switching ripple
current in L2. Then:
where IL1(PK) and Dmax are calculated from (9) and (16)
at
VAC(min). We disregarded the ripple current in L2 in
VO • TOFF
L2 =
(28) the equation (30). The drain voltage rating of M1 can be
∆IL2 • η2 determined as:
VDS(M1) = VAC(MAX) √2 + VC(MAX) (1 +KC )
(31)
where ∆IL2 is the peak-to-peak current ripple in L2. Larger values of L2 will produce smaller ripple ∆IL2, and therefore where VC(max) and KC are calculated at VAC(max) using (18)
smaller peak-to-average error in the output current control and (22). It is very important to find a good balance between
loop. However, it would also make the output current sense the total gate charge Qg and the on resistance RDS(ON) of
comparator more susceptible to noise. It is a good practice the power MOSFET M1. Using the MOSFET with lower
to design L2 for ∆IL2 = 0.2~0.3. An output capacitor can be RDS(ON) will not necessarily achieve greater efficiency. The
added to reduce the output ripple current further if needed.
HV9931 has a gate driving capability mainly limited by the
Unlike the input inductor L1, design of L2 is typically limited power dissipation in the high voltage regulator. In addition to
by the saturation flux of its magnetic material. However, generating higher switching power loss, MOSFETs with high
power dissipation due to the core loss may also need to Qg will require more current from the regulator. Non-optimal
be considered. The saturation current rating of the inductor selection of M1 may cause the HV9931 to overheat.
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correspondingly. Due to a finite reverse recovery time of D1,
the input inductor L1 develops certain reverse current in the
beginning of the dead time. Since L1 runs in DCM, reverse
IO
4 • √2
ID1 =
•
(32) recovery of D1 is negligible from the overall power efficiency
π
η1 • (1 + √1+ δMIN )
standpoint. However, even a small reverse current in L1
can cause a very high voltage spike across D4 when both
diodes stop conducting. Thus, ultra fast recovery rectifier is
2 • (√1 + δMIN - 1)
ID2 = DMAX • IO =
• IO
(33) recommended for D1.
δMIN
The highest currents in D1-D4 averaged over the AC line
cycle can be calculated as:
Since Cj4<<COSS typically, the post-conduction oscillation
occurs mainly across D4. The drain voltage of M1 will remain
(√1 + δMAX - 1)2
ID3 = (1 - DMIN) • IO =
• IO
(34) almost unchanged throughout the dead time. Besides
δMIN
causing the high voltage stress across D4, this oscillation
may affect the EMI performance of the circuit. Thus, adding
an RC snubber circuit across D4 is recommended. If the
2 • √2
1
4 • √2
+
• IO (35) snubber capacitance value is greater than (COSS+Cj1), the
ID4 =
•
δMIN
η1 • (1 + √1+ δMIN )
π
reverse voltage rating of D4 can be reduced significantly.
A fast 400V rectifier can be used for D4 in a universal 90where δMAX and δMIN are calculated from (15) at VAC(MAX) and 260VAC LED driver with adequate selection of the RC
VAC(MIN) correspondingly. Peak currents in D1 and D4 equal snubber components.
to IL1(PK) determined from (9) at VAC(MIN). Peak currents in
D2 and D3 are computed as IO + ½∆IL2. The voltage ratings Using ultra-fast recovery rectifiers for D2 and D3 is essential
for D1-D3 are given as:
for good efficiency of the LED driver. Both diodes operate at
high current and are subjected to fast transitions and high
VR(D1) = VAC(MAX) √2 + VC(1 + KC )
(36) reverse voltage.
VR(D2) = VAC(MAX) √2
PWM And Linear Dimming
Many LED applications require dimming. Two types of
dimming are available: analog and PWM. With analog (or
linear) dimming, 50% brightness is achieved by applying
VR(D3) = VC (1 + KC )
(38) 50% of the maximum current to the LED. Drawbacks to this
method include LED color shift and the need for an analog
control signal, which is not sometimes readily available.
where VC and KC are calculated at VAC(MAX) from the PWM dimming is achieved by applying full current to the
equations (18) and (22).
LED at a reduced duty cycle. For 50% brightness, full
current is applied at a 50% duty cycle. The frequency of the
The required reverse voltage rating of D4 depends on PWM signal must be above 100 Hz to ensure that the PWM
several factors. The dead time switching state of Fig.3(c) is pulsing is not visible to the human eye. The maximum PWM
characterized by a post-conduction resonance. The LC tank frequency depends upon the power-supply startup and
is formed by L1 and the parasitic capacitance of D1, D4 and response times. The HV9931 features a PWMD enable input
M1. The resonant period can be estimated as:
that accepts a PWM dimming control logic signal. The GATE
output is disabled when this signal is low. At the same time,
L1 • Cj4(COSS + Cj1 )
(39) since M1 is off and the rectifier D4 is reverse biased there is
TR = 2π
no discharge path for C1. Hence the current in L2 will recover
COSS + Cj1 + Cj4
within a single switching cycle back to its original level with
where COSS is the output capacitance of M1, Cj1 and Cj4 no overshoots as soon as the PWMD signal becomes high
are reverse-biased junction capacitances of D1 and D4 again.
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In some cases, however, linear dimming is preferred for
simplicity and component count reduction when the PWM
control signal is not available. On the first glance at the
HV9931, merely programming the divider ratio of RREF2
and RCS2 can control the output LED current proportionally.
However, this method would affect the required voltage
ratings of C1, M1 and D1-D4. The problem can be explained
by the power imbalance between the input DCM and the
output CCM converter stages. The DC voltage conversion
ratio of the output buck stage is given by (11). Hence the
duty cycle of the CCM buck stage is independent of the
output current. On the other hand, the input buck-boost stage
delivers an amount of energy every switching cycle that is a
function of the duty ratio and the switching frequency. The
balance is achieved by increased voltage at C1 for smaller
output currents. The voltage stress can be very significant
for universal 90 - 260VAC input designs.
the Design Example section of this application note. (Note
that VC exhibits no further increase as the output buck
stage enters DCM.) Thus, the linear dimming method will
require significantly higher voltage ratings of the switching
components.
The voltage stress problem at light load can be resolved
by making TOFF proportional to IO. The equation (18) will
no longer be load dependent since δ = const when TOFF/
IO = const. One possible implementation of this dimming
technique is depicted in Fig. 8. The timing resistor is altered
in proportion with the output divider ratio. In order to maintain
constant VC, the resistor values must satisfy:
Ra
Rb
=
(40)
R
R
CS2
T
However, the designer must be careful when using this
technique, since, for example, linear dimming to 33% of
the nominal load will cause the switching frequency of the
converter to triple.
Fig.7 shows VC as a function of the output current based
on the equation (18) for the universal LED driver given in
Fig 7. Voltage at C1 as a function of the output current in the case of linear dimming.
500
DCM
C1 Voltage (V)
400
300
CCM
200
100
0
0.2
0.4
0.6
0.8
LED Current (A)
Fig 8. Linear dimming circuit maintaining constant voltage at C1
to M1 (GATE)
RB
RT
SW1
Supertex inc.
GATE
RCS2
CS2
HV9931
RT
RREF2
VDD
SW2
to +VO
RA
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AN-H52
Phase-Control Dimming
However, some additional circuitry may be needed depending
on the topology and the power ratings of the phase-control
dimmer or in order to make the HV9931 work with any
dimmer. Most dimmers include an EMI filter for attenuating
RF interference caused by the thyristor switching. A typical
2-wire 600W dimmer is shunted by a 0.047~0.1µF capacitor
that causes substantial AC leakage current. This current can
develop significant voltage at the input of the LED driver
while it is off. Hence multiple premature startup attempts
may occur causing the LEDs to flicker. In order to resolve
this problem, a bleeder resistor can be connected across
the LED driver input while the HV9931 is off. The resistor
can be disconnected from the input as soon as the HV9931
resumes switching. The circuit diagram of Fig. 9 shows one
simple implementation of this technique.Inrush charging of
the capacitance at the AC input of the LED driver needs to
be considered also. The thyristor may turn off due to a zerocurrent condition created by a resonance in the LC circuit
formed by the filter inductor of the dimmer (a few tens of
µH typically) and the input capacitance of the LED driver.
Although RBL will help to damp this resonance, an additional
resistor may be needed in series with the AC input of the
LED driver.
One of the main advantages of the HV9931 LED driver
solution is its inherent compatibility with phase-control
dimmers. Solid-state light dimmers have been around since
the 1960’s. They work by varying the duty cycle of the full
AC voltage that is applied to the lights being controlled.
Typical light dimmers are built using thyristors, and the exact
time when the thyristor is triggered is relative to the zero
crossings of the AC power. When the thyristor is triggered
it keeps conducting until the current passing though it goes
to zero.
Typical switch-mode LED drivers do not work well with phasecontrol dimmers because of the large output capacitance
that they must use to filter the 2nd AC line harmonic ripple.
Interruptions of the voltage at the output of a phase dimmer
would have no effect on the output current of these LED
drivers.
The HV9931 LED driver cuts the output current naturally as
soon as its GATE output switching halts. The energy stored
in C1 is preserved until the switching resumes. Merely
selecting a VDD bypass capacitor CDD small enough (about
0.1µF typically) will disable the HV9931 switching when
the input AC line voltage drops out. Adding a Zener diode
in series with VIN (VZ ≈ 50V for 120VAC) is recommended
for reliable phase-control operation and under-voltage
protection. Alternatively, the PWMD pin can be used to
disable switching when the input voltage is low.
Input power consumption of the LED driver needs to be
taken into account too. Lower power LED drivers (10W or
less) may draw input current that is smaller than the holding
current of the thyristors. Use phase-control dimmers having
the adequate power ratings, or connect more than one LED
driver to the dimmer output to avoid this problem.
Fig 9. Bleeder circuit for use with 120VAC phase-control dimmers
VIN
D1
D4
L1
C1
M1
CIN
~AC
~AC
L2
D2
-
D3
VO
RS2
RS1
+
D10
47V
VIN
CDD
0.01µF
VDD
CF
0.01µF
RBL
1k
D11
BAV99
GATE
CG
0.01µF
HV9931
Supertex inc.
RG
2.7k
M2
DN3545N8
(Depl. NFET)
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AN-H52
Output Open Circuit And Input Under Voltage In addition to D10, an improved input under voltage
protection circuit is shown in Fig.10a that can achieve better
Protection
performance compared to the simple fixed input current
limiting. The reference for the CS1 comparator is derived
from the input rectified AC waveform. The voltage divider
ratio of R1 :RREF1 is programmed such that the Zener diode
ZREF1 clamps the divider voltage at any input greater than
VAC(min), i.e.:
HV9931 is a constant output current source. Hence it can
generate destructive voltage at its output in the case of an
output open circuit condition. A simple circuit shown in Fig.
10 protects the HV9931 LED driver from the output overvoltage. Zener voltage of D12 greater than the maximum
output voltage must be selected. Resistor ROV is typically
100~200Ω. However, it still may affects the output current
divider ratio and needs to be included in the calculations by
replacing RCS2 by (RCS2 + ROV) in the equation (3). Note,
that the open circuit can create an over-voltage condition
across C1. This voltage stress can be limited by connecting
a Zener diode or TVS across C1 limiting the voltage to some
acceptable level greater than VC(max). The power dissipation
in this voltage clamp device is usually small, since the
HV9931 operates at minimum duty cycle during the open
circuit condition.
R1 = RREF1 •
VAC(MIN) • √2 - VZREF1
(41)
VZREF1
VREF = VZREF1 should be used with the equation (3) to set
the peak current limit for L1 within the normal operating
input AC voltage range. When the input voltage falls below
VAC(min), the reference voltage will reduce too preventing
the inductor L1 from entering continuous conduction mode
(CCM). (Operating L1 in CCM can cause undesirable LED
flickering, audible noise and excessive heat dissipation due
to the loop oscillation.) RBIAS creates a positive offset voltage
to maintain the reference above 0V in the zero crossings of
the AC line voltage, and thereby prevents interruptions of the
output current.
The HV9931 inherently protects the LED driver from an input
under voltage condition by limiting the input current. However,
increased input current may generate excessive power
dissipation in L1, D4, M1 and RCS1. Additional protection is
recommended by connecting a Zener diode in series with
the VIN pin of the HV9931. (See D10 in Fig. 9)
Fig 10. Output open circuit protection
CS2
HV9931
RCS2
RREF2
to +VO
ROV
D12
VDD
VZ > VO
to -VO
Fig 10a. Input under voltage protection
AC Bridge “+”
D10
VIN
RREF1
CS1
HV9931
VDD
Supertex inc.
R1
RBIAS
RCS1
VZREF1
ZREF1
to RS1 “-”
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13
AN-H52
Fig 11. Transient voltage protection.
HV surge
to CIN “+”
VIN
2.7k
VDD
HV9931
MOV1
430V
10k
PWMD
MMBT2222A
ZD1
5.6V
100k
Surge Immunity and EMI Considerations
minimized. The first loop including of M1, C1, D2 and D3 can
significantly degrade the overall EMI performance due to the
reverse recovery current in D3. Using soft recovery diode is
recommended for D3. Adding an RC snubber circuit across
D3 can be useful (not shown in Fig.11). The second loop
consists of CIN, D1, C1, M1 and RS1. Since the input buckboost stage runs in DCM, the reverse recovery current in D1
is insignificant. However, charging its junction capacitance
can generate fast current transients. The large physical
dimensions of C1 can complicate optimal routing of these
loops. Auxiliary low ESR/ESL capacitors Caux1 and Caux2 can
be used for optimizing the printed circuit board routing. Caux1
and Caux2 are responsible for the fast switching transition
currents only and hence are typically very small. When these
bypass capacitors are used, the areas formed by Caux1, D1,
Caux2, M1, RS1 and M1, Caux2, D2, D3 need to be considered
mainly.
High voltage surges occur on the AC power mains as a
result of switching operations in the power grid and from
nearby lightning strikes. LED lighting and signal equipment
may be subjected to surge immunity compliance testing in
accordance with various standards (EN61000-4-5, NEMA
TS-2 2.1.8 etc.) to insure its continued reliable operation if
subjected to realistic levels of surge voltages. The HV9931
LED driver circuit relies mainly on the transient suppressors
(MOV, TVS) to protect it from the input AC line surge. There
is little capacitance available at the AC input of the LED driver
to absorb high surge energy. Thus a transient suppressor
needs to be connected across the AC input terminals.
Additional protection circuitry may be also required to
protect M1, D1, D2 and the HV9931 itself. A simple circuit
shown in Fig.11 clamps the voltage at VIN in the case of
an input over-voltage transient. At the same time, it disables
the GATE output of the HV9931 to protect the switching
components. When HV9931 is disabled, M1 and D1 will only
have to withstand the input surge voltage VSURGE rather than
(VSURGE + VC).
Optimal routing of the HV9931 gate output loop can be
important for EMI performance as well as for preventing
destructive oscillations of the M1 gate voltage. The gate
driver loop area must be minimized. The trace connecting
the source terminal of M1 with the GND pin of the HV9931
must be as short as possible. The VDD bypass capacitor CDD
must have low ESR and needs to be placed in the immediate
proximity of the HV9931.
As with all switching converters, selection of the input filter is
critical to obtaining good EMI. The HV9931 solution provides
an inherent advantage of the frequency dither due to the AC
voltage ripple across C1 when the fixed off-time operating
mode is used. The C1 voltage feedback introduces additional
frequency dither when utilized. Hence the required noise
attenuation can be lowered yielding a smaller EMI filter.
Post-conduction oscillation across D4 during the dead time
of L1 can be another substantial source of RF emission.
Adding a snubber circuit (Rd and Cd in Fig.11) can help
significantly. In addition, this snubber is needed to reduce
the voltage stress at D4 as it has been discussed in the
previous sections.
Some important guidelines must be followed for optimal
EMI performance of the HV9931 power converter. The
area of the fast switching loops shown in Fig.11 must be
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2.7k
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14
AN-H52
Fig 12. Fast switching current loops.
RD
CD
VIN
CAUX22
D1
L2
L1
D4
C1
CAUX1
CIN
D2
D3
M1
M
~AC
~AC
VO
+
RS2
RS1
GATE
GND
HV9931
CDD
VDD
LED DRIVER DESIGN EXAMPLE
Let us design a power converter for driving LEDs with the following characteristics:
Input AC Line Voltage
80 - 260VAC, 50-60Hz
Output Current
750mA
Output Current Ripple
±15%
Output Voltage
25V (max.)
THD
<20% at 120VAC
OFF Time
10µs
Predicted Efficiency
76%
The value of L2 can be calculated from the equation (28).
We will assume that the efficiencies of the input buck-boost
stage and the output buck stage are η1 = 0.85 and η2 = 0.9
correspondingly. The efficiency of a DCM buck-boost stage
is typically lower compared to the CCM buck stage. The
overall efficiency η = η1η 2 ≈ 0.76.
The DC current rating of L2 equals to IO = 0.75A. The
saturation current must satisfy the condition (29) resulting
in ISAT > 0.86A.
Step 1. Using the equation (2), we will calculate the timing
resistor RT value for TOFF = 10µs. The resulting timing
resistor:
RT = 228K.
Step 3. Assuming 0.25W power dissipation in the output
current sense resistor RS2, we can calculate its value.
Step 2. We will allow 30% peak-to-peak switching current
ripple in L2, or ∆iL2 = 0.3iL2 = 0.225A. Then according to the
equation (4), the peak current in L2 is:
RS2 =
0.25W
IO2
≈ 0.44Ω
We will select a 0.47Ω 1/2W resistor for RS2. Let us use the
VDD pin as a reference voltage (VDD = 7.5V).
iL2(PK) = 0.86A.
Supertex inc.
L2 ≈ 1.2mH
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AN-H52
(Note, that although VDD is relatively precise, it may exhibit
certain dropouts near the AC line voltage cusps when there
is no input voltage available at VIN. An external voltage
reference is needed for better accuracy.) Selecting RREF2 =
100K, we can calculate the value of RCS2 using the equation
(3).
RCS2 = 5.4KΩ
Solving this equation for RS1, we obtain:
Let us select RS1 0.47Ω 1/4W. To calculate RCS1, we will use
the equation (3) assuming VREF = VDD and RREF1 = 100K as
before. We will program the peak input current limit as 120%
of iL1(PK). Then:
Step 4. The input inductor L1 is assumed to reach boundary
conduction mode (BCM) at VAC(MIN) at the peak of the input
voltage hump. Using the equation (10), we can calculate the
critical inductance value that meets this condition.
RS1 ≈ 0.47Ω
RCS1 = 15.8KΩ
Step 8. Let us assume the third harmonic distortion coefficient
K3 = 0.15 at VAC = 120VAC. Then, the equation (23) gives
the value of C1.
L1 = 377µH
Step 5. Let us calculate the parameter δ and the duty cycle
D at VAC(MIN), VAC(MAX) and VAC using equations (15) and
(16):
C1 ≈ 31µF
Using the same equations (18) at VAC = 260VAC, we can
calculate the required voltage rating of C1.
1) δmin = 14, Dmax = 0.41 at 80VAC;
2) δmax = 146, Dmin = 0.15 at 260VAC;
3) δ = 31, D = 0.3 at 120VAC.
VC(MAX) = 182V
Step 6. The maximum peak current in L1 will occur at
VAC(min). It can be calculated from the equation (9).
The voltage ripple at C1 is small at high input voltage. The
equation (22) gives KC(MIN) = 0.032. Thus, the peak voltage
at C1 is:
iL1(PK) = 2.1A
Note that most “off-the-shelf” 330µH DC chokes may be not
suitable for L1. Since the current in L1 cycles from 0 to as
high as iL1(PK) every switching cycle, there may be excessive
power dissipated in the magnetic core of L1 due to large
magnetic flux excursion. On the other hand, the wire gauge
used in such inductors is selected based on its DC current
rating, whereas the RMS current in L1 is substantially lower
than its peak current. Thus, custom designing of L1 is likely
to produce a more size efficient solution.
The switching ripple current rating is calculated using the
equations (23a):
IC(sw)(MAX) = 0.82A(rms) at 80VAC,
IC(sw) = 0.68A(rms) at 120VAC.
The 120Hz ripple current rating is calculated using the
equations (23b):
Step 7. The next step is calculating the input current sense
and divider resistors RS1 and RCS1. Let us allow 0.1W of
power dissipation in RS1 at VAC(MIN). Power dissipation in
RS1 can be calculated as:
WRS1 =
VC(PK) = (1 + KC(MIN)) VC(MAX) = 188V
IC(LINE)(MAX) = 0.22A(rms) at 80VAC,
IC(LINE) = 0.16A(rms) at 120VAC.
An electrolytic capacitor 33uF,200V can be selected for C1.
DMAX • IL1(PK)2 • RS1
6
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16
AN-H52
Step 9. If a smaller film capacitor is desired for C1, the circuit
of Figure 6 can be used. The value of RFF is calculated at
VAC = VAC(MAX) = 260VAC and VO = VO(MAX) = 25V using the
equation (26).
Step 10. Optimal selecting of the switching MOSFET M1
is based on finding a good balance between the total gate
charge Qg and the on-resistance RDS(ON). The drain voltage
rating is given by the equation (31).
RFF = 3MΩ
Select RFF 3.3MΩ to avoid loop oscillation at high AC line
voltage. CFF is selected such that:
Acceptable Qg is limited by the allowed power dissipation in
the HV9931. The power dissipation can be estimated as:
1
CFF >>
2π • RFF • 100Hz
WREG(max) =
Select CFF 4.7nF, 200V. The minimum value of C1 is limited by
(27). Calculation of the C1 value needed to meet the desired
harmonic distortion of IAC is very complex. The designer
may want to experiment with different capacitance values of
C1 to find the optimal one. Experimental verification shows,
however, that THD<20% at 120VAC is possible with less
than 1/3 of the C1 value calculated above. Two 4.7uF 250V
metalized polyester film capacitors connected in parallel
were used.
•
2√2
π
• VAC(max) - VZ
Qg • (1 - Dmin )
TOFF
+
VREF
RREF1
+
VREF
RREF2
+ 1mA
where VZ is Zener voltage of D10.
Let us select SPP03N60C3 for M1. This is a 650V, 3.2A
MOSFET by Infineon Technologies with RDS(ON) = 1.26Ω
and Qg(max) ≈ 13nC at VDS = 420V, VGS = 7.5V.
Then:
WREG(MAX) ≈ 400mW(MAX)
The RB value is selected based on the desired power
dissipation such that RB << RFF. A 330K resistor will dissipate
0.1W at VC(max) = 182V. The value of CB is calculated from:
The maximum RMS current in M1 is calculated from the
equation (30) as ID(M1) = 0.73A. The peak current in M1 is
IL1(PK)+IL2(PK) ≈ 3A. (Note that the maximum power dissipation
in HV9931LG (SO-8) must be derated 6.3mW/°C above
25°C. Thus, the maximum operating ambient temperature
needs to be less than 60°C. Using HV9931P (DIP-8) will be
limited to TA < 80°C.) A larger VZ can be selected to reduce
power dissipation in the HV9931.
1
CB >>
≈ 4.0nF
2π
•
R
•
120Hz
B
The flying capacitor CA must be selected such that:
TOFF
CA >>
≈ 25pF
RB
Step 11. In accordance with the equations (32)-(35), the
average currents in D1-D4 are:
We can select CB = CA = 4700pF for simplicity. Both capacitors
must be rated to withstand VC(pk). Zener diodes D8 and D9
must not distort the AC ripple waveform at the output of CFF.
In other words, their breakdown voltage must be set higher
than the C1 voltage ripple amplitude at 120VAC. Leaving
D8 and D9 out or selecting the diodes with excessively high
breakdown voltage may increase the start-up time of the
LED driver.
Supertex inc.
VDS(max) = 556V
ID1 = 0.33A, ID2 = 0.31A, ID3 = 0.64A, ID4 = 0.6A.
Peak currents in D1 and D4 equal to the peak current in L1
or:
ID1(PK) = ID4(PK) = IL1(PK) = 2.1A.
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AN-H52
The equations (36)-(38) give the reverse voltage across D1D3, resulting in:
BYD57K data by Philips shows Cj < 2pF at VR > 100V. By
choosing Cd = 200pF and Rd = 2.7KΩ, we can use a 400V
rectifier for D4, for example, BYD57G (400V, 1A, trr = 30ns)
by Philips.
VR(D1) = 562V, VR(D2) = 368V, VR(D3) = 188V.
Adding an RC snubber is recommended across D4. Reverse
voltage across D4 depends on the capacitance value of
CDselected for this RC snubber. The snubber capacitor CDd
needs to be greater than COSS+ Cj1, where COSS is drainto-source capacitance of M1, and Cj1 is the reverse biased
junction capacitance of D1. Usually, Cj1 can be disregarded
compared to the COSS. The typical data by Infineon shows
COSS < 20pF at VDS > 100V for SPP03N60C3. BYD57K by
Philips (800V,1A, trr = 75ns) can be selected for D1. The
Fast switching rectifiers are needed for D2 and D3. We
can select D2 STTA106A (600V, 1.0A, trr = 20ns) and D3
STTH102A (200V, 1.0A, trr = 30ns) by STMicroelectronics.
Step 12. Output filter capacitor CO of a few hundred
nanofarads will be needed for improved EMI performance.
Alternatively, a larger value of this capacitor can be used to
reduce the switching ripple current in the LEDs further.
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
Supertex inc.
©2011 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
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Tel: 408-222-8888
www.supertex.com
110711
18
Supertex inc.
DN-H06
14W Off-line LED Driver,
120VAC, PFC, 14V, 1.0A Load
Specifications
Parameter
Value
AC line voltage
100 - 135VAC
LED (string) voltage
0 - 14V
LED current
1.0A
Switching frequency
70 - 120kHz
Design Note
The input line current features low harmonic distortion, satisfying the requirements of EN 61000-3-2 Class C (Lighting
Equipment). Open circuit and in short circuit at the output
can be sustained indefinitely. The AC line current is limited
to an input voltage range from zero to 135VAC. Both the
output current and line current drop gradually as AC line
voltage falls below 100VAC.
Open circuit protection
Yes (output voltage 33V)
Please refer to application note AN-H52 for a detailed description of and design guidance for the HV9931LED driver
control IC.
Short circuit protection
Yes (output current 1.0A)
Miscellaneous Notes
Efficiency
74% (@ 14V)
AC line undervoltage
LED and AC line current fall
off gradually below 100 VAC
Light dimmer compatible
No
THD
~16% (LED voltage 14V)
Power factor
>95% (LED voltage 14V)
General Description
This Design Note describes the results of a 14W LED Driver
Design. The design specifically forgoes the use of electrolytic capacitors, which form a point of weakness in high reliability and high ambient temperature applications.
The design drives one or more high brightness LEDs, in parallel or series combinations, at a current of 1.0A and up to a
voltage of 14V. This same design can be operated at lower
voltage/power levels as well, with slight loss of efficiency
and THD.
The results, in particular the waveforms, documented in this
note apply more broadly, i.e. at other output currents and
voltage levels when appropriate adjustments are made to
the size and value of certain components.
Efficiency can be increased by using components having
less ohmic resistance, particularly L1 and M1, and by lowering the switching frequency.
Supertex inc.
EMI, Common Mode Filtering:
The magnitude and frequency dependency of the common
mode current on the line input depends heavily on physical layout and location of the LED driver circuit and the attached load. As such, the design may or may not require
the addition of a common mode choke ahead of the bridge
rectifier.
Open Circuit Operation
During open circuit operation the HV9931 is made to run
at minimum duty cycle through the action of CS2 and ZOV.
Some energy transfer, as small as it may be, still occurs,
which causes the voltage on C1, and thereby the peak drain
voltage on M1, to rise to a higher level. Circuit losses keep
this raise in check. From experimental data: peak VC1 rises
by about 35V from 120 to 155V, and maximum VDS rises by
60V from 270 to 330V. If this rise is undesirable, a zener
diode or a bleeder resistor can be placed across C1 to limit
the voltage rise across C1 and M1, or more sophisticated
circuitry can be added to further limit switching activity.
M1 Turn Off
An external pull down transistor was added to the gate drive
circuit to speed up the turn off transition. Note that M1’s
drain current, which is more or less triangular in shape, is
largest at turn off. Figures 19 and 20 illustrate the gain in
turnoff speed that can be attained by this simple addition.
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DN-H06
Measurements showed an increase in efficiency by 0.5%
from 73.2 to 73.7%, corresponding to a reduction in switching loss of 100mW. The small gain in efficiency may not warrant the addition of the pull down transistor, but may nevertheless be interesting when power levels are higher or a
larger MOSFET having more gate and reverse transfer capacitance is in use.
VDD at Zero Crossing
VDD may drop out at the AC line zero crossings, and cause
a short lived drop in LED current if the capacitor at the VDD
pin is made small. If this effect is undesirable, then the CDD
should be chosen sufficiently large. Figures 5 and 6 demonstrate this effect.
CS1 Programming
Control of M1 should, under regular circumstances, be governed by the action of comparator CS2, which provides
regulation of the LED current. CS1 should regulate only if
limitation of input stage current is necessary, as during AC
line undervoltage or during transient conditions. CS1 is to
remain inactive by programming an envelope for the input
stage current with an adequate margin, such that CS1 does
not interfere with the regulation of the output current under
normal circumstances. A simple DC threshold of adequate
value will suffice.
active for input voltages lower than 100VAC, and take over
regulation by limiting input stage current to an approximate
sinusoidal waveform. For line voltages larger than 100V, this
scaled threshold will become unnecessarily accommodative, and zener diode ZREF1 will limit its rise.
Diodes D1 and D2
D1 and D2 are part of the RT oscillator circuit which determines the switching frequency, or more precisely, the off-time
(TOFF) of the switching period. The off time is determined by
the oscillator discharge current which should appear when
M1 is turned off, i.e. when the GATE pin is low. The main
contribution to the discharge current is due to current in RT
when the voltage at the GATE pin is low. Current originating
at the RFF resistor is meant to modulate this discharge current in order to affect an increase or decrease of TOFF. Note
that RFF is driven by the ripple voltage across C1. As such,
RFF carries an alternating current, which is present regardless of the timing needs of the RT pin. D1 and D2 resolve two
issues depending on the polarity of the RFF current. When
RFF sources current, it will overdrive the pin when GATE is
high, which is undesirable. Diode D2 blocks this current, and
the current will follow an the path through RT and the GATE
pin. When RFF sinks current, diode D1 sources this current
during the time that the RT discharge current should be zero
(GATE pin high).
This design employs a somewhat more sophisticated envelope for the purpose of limiting the AC line current when
undervoltage occurs. The threshold is a scaled version of the
input voltage, thus reducing input current as input voltage
reduces. By proper choice of values, CS1 will thus become
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2
DN-H06
Schematic 1
D1
ES1J
CFA
1mH
B1
600V
0.8A
CF
100nF
250V
CFB
1mH
C1
10μF
250V
D4
ES1J
L1
120μH
RS2
220mΩ
QG
2N2907
MOV1
220VDC
RCS1
6.49kΩ
H1
DN1
MMBD3004S
RREFA
604kΩ
RREFZ
1MΩ
CA
1nF
500V
ROV
200Ω
CFF
10nF
RREFB
604kΩ
RREF2
100kΩ
RT
158kΩ
CB RB
1nF
500V 1MΩ
ZREF1
7.5V
D2
1N914
D1
1N914
2
1
CS1
VIN
4
8
GATE
ZOV
33V
RCS2
3.16kΩ
RFF
1.5MΩ
RREF1
100kΩ
CO
1μF
50V
D3
ES1D
M1
SPD08N60C3
RG
10Ω
H2
L2
390μH
CD
220pF RD
500V 2.7kΩ
CIN
100nF
250V
RS1
100mΩ
F1
0.5A
D2
ES1J
RT
7
IC1
CS2
HV9931LG
PWM
VDD
5
6
GND
3
CDD
100μF
10V
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
3
DN-H06
Schematic 2
D1
C1
D2
D4, L1
L2
C1
D1
D2
H20
D4
L71
B10
600V
0.8A
L72
L1
CD
C71
L2
RD
D3
CIN
M1
DG
RS1
MOV10
220VDC
CO
D3
RS2
QG
CIN
M1
RS2
CA
H10
ROV
DN1
F1
0.5A
ZOV
RREFA
CFF
RCS1
RREFB
RREFZ
RCS2
RFF
RREF1
RREF2
CB
RT
RB
ZREF1
D1
2
1
CS1
VIN
4
D2
8
GATE
RT
7
IC1
CS2
HV9931LG
PWM
VDD
5
6
GND
3
CDD
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
4
DN-H06
Schematic 3
C1
D1
D2
H20
D4
L71
B10
600V
0.8A
L72
L1
CD
C71
L2
RD
CIN
CO
D3
M1
DG
RS1
RS2
QG
MOV10
220VDC
CA
H10
ROV
DN1
F1
0.5A
ZOV
RREFA
CFF
RCS1
RREFB
RREFZ
RCS2
RFF
RREF1
RREF2
CB
RT
RB
ZREF1
D1
2
1
CS1
VIN
4
D2
8
GATE
RT
7
IC1
CS2
HV9931LG
PWM
VDD
5
6
GND
3
CDD
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
5
DN-H06
Fig 1. (VAC, IAC), Nominal (120V)
Fig 2. (VAC, IAC), Low Line (100V)
IAC
VAC
THD: 15.3%
THD: 22.5%
Fig 3. (VAC, IAC), High Line (135V)
Fig 4. (VAC, IAC, Undervoltage (90V)
THD: 9.2%
THD: 35.6%
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
6
DN-H06
Fig 5. (VDD, ILED) with Large CDD (100μF)
ACI
Fig 6. (VDD, ILED) with Small CDD (1μF)
IAC
VAC
Small CDD
VAC
ILED
ILED
VDD
VDD
Much smaller CDD is feasible, if slight loss of regulation at the zero crossings
is acceptable.
Fig 7. (VAC, IAC), RFF Removed, Low Line
Fig 8. (VAC, IAC), RFF Removed, Nominal
THD: Increases to 54.1% (from 22.5%)
THD: Increases to 31.5% (from 15.3%)
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
7
DN-H06
Fig 9. (ILED, IAC, VAC) when RREFZ Removed
IAC
Fig 10. (VIN, VC1, IL1) Detail, (2ms/div)
RREFZ Removed
ILED
VAC
RREF3 is needed to prevent loss of ILED regulation at the zero crossings,
where CS1 not receive adequate bias from VIN.
Fig 11. (VIN, VC1, IL1) Detail, (1ms/div)
VC1
VIN
Fig 12. (VIN, IAC, IL1) Detail, (100μs/div)
VIN
IAC
IAC
IL1
Supertex inc.
IL1
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
8
DN-H06
Fig 13. (VIN, IAC, IL1) Detail, (10μs/div)
Fig 14. (VIN, IAC, IL1) Detail, Time Base (2μs/div)
VIN
VIN
IAC
IAC
IL1
IL1
Fig 15. M1 Drain Voltage, (2μs/div)
Fig 16. M1 Drain Voltage, (20μs/div)
VIN
VDS
IL1
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
9
DN-H06
Fig 17. M1 Drain Voltage, (2ms/div)
Fig 18. M1 Turn on
VDS
1.
2.
VGS
1. VGS rises steadily; Diode D3 recovers.
2. VGS plateaus; Miller effect due to falling VDS.
Fig 19. M1 Turn Off
Fig 20. M1 Turn Off, Q1 Removed
VGS plateau about 25ns.
Gate turn-off assisted by external PNP pull down transistor
Slower turn-off. VGS plateau about 70ns.
Pull down transistor Q1 removed, and ZeroΩ RG.
Supertex inc.
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
10
DN-H06
Fig 21. Input Filter Detail, (VIN, VBR, IBR)
Fig 22. CS1 Programming Detail, VZREF1
VIN
VBR
VIN
VZREF1
IBR
No ripple visible on VBR; Filter rejects voltage ripple of VIN.
No Ripple visible on IBR; Filter rejects current ripple of IL1.
Supertex inc.
Limit defined in part by DC level, in part by VIN.
Limit never exceeds DC level,
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
11
DN-H06
Fig 23. Line Regulation
Fig 24. Efficiency vs Line Voltage
100
1050
Efficiency [%]
LED Current [mA]
90
1000
80
70
60
950
90
100
110
120
130
140
150
50
90
100
110
120
130
140
150
Line Voltage [V]
Line Voltage [V]
Fig 25. THD vs Line Voltage
Fig 26. Power Factor vs Line Voltage
50
1.00
40
Power Factor
THD [%]
0.95
30
20
0.85
10
0
0.90
90
100
110
120
130
Line Voltage [V]
Supertex inc.
140
150
0.80
90
100
110
120
130
140
Line Voltage [V]
● 1235 Bordeaux Drive, Sunnyvale, CA 94089 ● Tel: 408-222-8888 ● www.supertex.com
12
150
DN-H06
Fig 27. Load Regulation
Fig 28. Efficiency vs Load Voltage
1300
100
90
1250
80
70
Efficiency [%]
LED Current [mA]
1200
1150
1100
1050
1000
50
40
30
20
950
900
60
10
0
5
10
LED Voltage [V]
0
15
Fig 29. THD vs Load Voltage
0
5
10
15
LED Voltage [V]
Fig 30. Power Factor vs Load Voltage
50
1.00
40
Power Factor [%]
THD [%]
0.95
30
20
0.85
10
0
0
0.90
5
10
LED Voltage [V]
0.80
15
0
5
10
LED Voltage [V]
15
Supertex inc. does not recommend the use of its products in life support applications, and will not knowingly sell them for use in such applications unless it receives
an adequate “product liability indemnification insurance agreement.” Supertex inc. does not assume responsibility for use of devices described, and limits its liability
to the replacement of the devices determined defective due to workmanship. No responsibility is assumed for possible omissions and inaccuracies. Circuitry and
specifications are subject to change without notice. For the latest product specifications refer to the Supertex inc. (website: http//www.supertex.com)
Supertex inc.
©2012 Supertex inc. All rights reserved. Unauthorized use or reproduction is prohibited.
010512
13
1235 Bordeaux Drive, Sunnyvale, CA 94089
Tel: 408-222-8888
www.supertex.com

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