NSC LM363

LM363 Precision Instrumentation Amplifier
General Description
The LM363 is a monolithic true instrumentation amplifier. It
requires no external parts for fixed gains of 10, 100 and
1000. High precision is attained by on-chip trimming of offset voltage and gain. A super-beta bipolar input stage gives
very low input bias current and voltage noise, extremely low
offset voltage drift, and high common-mode rejection ratio.
A two-stage amplifier design yields an open loop gain of
10,000,000 and a gain bandwidth product of 30 MHz, yet
remains stable for all closed loop gains. The LM363 operates with supply voltages from g 5V to g 18V with only
1.5 mA current drain.
The LM363’s low voltage noise, low offset voltage and offset voltage drift make it ideal for amplifying low-level, lowimpedance transducers. At the same time, its low bias current and high input impedance (both common-mode and
differential) provide excellent performance at high impedance levels. These features, along with its ultra-high common-mode rejection, allow the LM363 to be used in the
most demanding instrumentation amplifier applications, replacing expensive hybrid, module or multi-chip designs. Because the LM363 is internally trimmed, precision external
resistors and their associated errors are eliminated.
The 16-pin dual-in-line package provides pin-strappable
gains of 10, 100 or 1000. Its twin differential shield drivers
eliminate bandwidth loss due to cable capacitance. Compensation pins allow overcompensation to reduce bandwidth and output noise, or to provide greater stability with
capacitive loads. Separate output force, sense and reference pins permit gains between 10 and 10,000 to be programmed using external resistors.
On the 8-pin metal can package, gain is internally set at 10,
100 or 500 but may be increased with external resistors.
The shield driver and offset adjust pins are omitted on the
8-pin versions.
The LM363 is rated for 0§ C to 70§ C.
Features
Y
Y
Y
Y
Y
Y
Y
Y
Offset and gain pretrimmed
12 nV/0Hz input noise (G e 500/1000)
130 dB CMRR typical (G e 500/1000)
2 nA bias current typical
No external parts required
Dual shield drivers
Can be used as a high performance op amp
Low supply current (1.5 mA typ)
Typical Connections
16-Pin Package
8-Pin Package
G e 10 2, 3, 4, open
G e 100 3–4 shorted
G e 1000 2–4 shorted
TL/H/5609 – 33
TL/H/5609 – 1
Connection Diagrams
Metal Can Package
Order Number LM363H-10,
LM363H-100 or LM363H-500
See NS Package Number H08C
C1995 National Semiconductor Corporation
TL/H/5609
16-Pin Dual-In-Line Package
TL/H/5609 – 2
Order Number 363D
See NS Package Number D16C
RRD-B30M115/Printed in U. S. A.
LM363 Precision Instrumentation Amplifier
April 1991
Absolute Maximum Ratings (Note 5)
Input Voltage
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales
Office/Distributors for availability and specifications.
Supply Voltage
g 18V
Differential Input Voltage
Input Current
g 10V
Equal to Supply Voltage
Reference and Sense Voltage
g 25V
Lead Temp. (Soldering, 10 sec.)
ESD rating to be determined.
300§ C
g 20 mA
LM363 Electrical Characteristics (Notes 1 and 2)
LM363
Parameter
Conditions
Typ
Tested
Limit
(Note 3)
Design
Limit
(Note 4)
150
250
2.5
400
700
6
mV
mV
mV
4
8
75
mV/§ C
mV/§ C
mV/§ C
Units
FIXED GAIN (8-PIN)
Input Offset Voltage
G e 500
G e 100
G e 10
30
50
0.5
Input Offset Voltage Drift
G e 500
G e 100
G e 10
1
2
20
Gain Error
( g 10V Swing, 2 kX Load)
G e 500
G e 100
G e 10
0.1
0.07
0.05
0.8
0.7
0.6
0.9
0.8
0.7
%
%
%
Input Offset Voltage
G e 1000
G e 100
G e 10
50
100
1
250
450
3.5
500
900
8
mV
mV
mV
Input Offset Voltage Drift
G e 1000
G e 100
G e 10
1
2
10
5
10
100
mV/§ C
mV/§ C
mV/§ C
Gain Error
( g 10V Swing, 2 kX Load)
G e 1000
G e 100
G e 10
2.0
0.1
0.6
3.5
0.8
2.3
%
%
%
Gain Temperature Coefficient
G e 1000
G e 500
G e 100, 10
40
20
10
Gain Non-Linearity
( g 10V Swing, 2 kX Load)
G e 10, 100
G e 500, 1000
PROGRAMMABLE GAIN (16-PIN)
3.0
0.7
2.0
FIXED GAIN AND PROGRAMMABLE
0.01
0.01
2
ppm/§ C
ppm/§ C
ppm/§ C
0.03
0.05
0.04
0.06
%
%
LM363 Electrical Characteristics (Continued) (Notes 1 and 2)
LM363
Parameter
Conditions
Typ
Tested
Limit
(Note 3)
Design
Limit
(Note 4)
Units
Common-Mode Rejection
Ratio (b10VsVCMs10V)
G e 1000, 500
G e 100
G e 10
130
120
105
114
94
90
104
84
80
dB
dB
dB
Positive Supply Rejection
Ratio (5V to 15V)
G e 1000, 500
G e 100
G e 10
130
120
100
110
100
85
100
95
78
dB
dB
dB
Negative Supply Rejection
Ratio (b5V to b15V)
G e 1000, 500
G e 100
G e 10
120
106
86
100
85
70
90
75
60
dB
dB
dB
Input Bias Current
2
10
20
nA
Input Offset Current
1
3
5
100
8
Common-Mode Input
Resistance
Differential Mode Input
Resistance
G e 1000, 500
G e 100
G e 10
0.2
2
20
Input Offset Current Change
b 11V s VCM s 13V
20
Reference and Sense
Resistance
GX
GX
GX
100
300
pa/V
30
80
27
83
kX
kX
kX
50
Min
Max
nA
GX
Open Loop Gain
GCL e 1000, 500
10
1
Supply Current
Positive
Negative
1.2
1.6
2.4
2.8
V/mV
3.0
3.4
mA
mA
Note 1: These conditions apply unless otherwise noted; V a e 15V, Vb eb 15V, VCM e 0V, RL e 2 kX, reference pin grounded, sense pin connected to output and
Tj e 25§ C.
Note 2: Boldface limits are guaranteed over full temperature range. Operating ambient temperature range is 0§ C to 70§ C for the LM363.
Note 3: Guaranteed and 100% production tested.
Note 4: Guaranteed but not 100% tested. These limits are not used in determining outgoing quality levels.
Note 5: Maximum rated junction temperature is 100§ C for the LM363. Thermal resistance, junction to ambient, is 150§ C/W for the TO-99(H) package and 100§ C/W
for the ceramic DIP (D).
3
Typical Performance Characteristics TA e 25§ C
Fixed Gain and Programmable
Parameter
1000/500
Units
100
10
Input Voltage Noise, rms, 1 kHz
12
18
90
Input Voltage Noise (Note 6)
0.4
1.5
10
nV/ SHz
mVp-p
Input Current Noise, rms, 1 kHz
0.2
0.2
0.2
pA/ SHz
pAp-p
Input Current Noise (Note 6)
40
40
40
Bandwidth
30
100
200
kHz
Slew Rate
1
0.36
0.24
V/ms
Settling Time, 0.1% of 10V
70
25
20
ms
Offset Voltage Warm-Up Drift (Note 7)
5
15
50
mV
Offset Voltage Stability (Note 8)
5
10
100
mV
0.01
0.005
0.05
%
Gain Stability (Note 8)
Note 6: Measured for 100 seconds in a 0.01 Hz to 10 Hz bandwidth.
Note 7: Measured for 5 minutes in still air, V a e 15V, Vb eb 15V. Warm-up drift is proportionally reduced at lower supply voltages.
Common-Mode Input
Voltage Limit
Supply Current vs Supply
Voltage
Input Bias Current vs
Temperature
Output Swing Referred to
Supplies
Supply Current vs
Temperature
Input Offset Current vs
Temperature
TL/H/5609 – 3
4
Typical Performance Characteristics
(Continued)
Output Current Limit
Input Noise Voltage
Input Current Noise
Input Current vs Voltage
Overdrive
Gain Non-Linearity
Gain Error vs Frequency*
*Trimmed to zero at 100 Hz
Gain Error vs Frequency*
Positive Power Supply
Rejection
Negative Power Supply
Rejection
Negative Power Supply
Rejection
Negative Power Supply
Rejection
*Trimmed to zero at 100 Hz
Negative Power Supply
Rejection
TL/H/5609 – 4
5
Typical Performance Characteristics
(Continued)
CMRR with Balanced
Source Resistance
CMRR with Balanced
Source Resistance
CMRR with Balanced
Source Resistance
CMRR with Unbalanced
Source Resistance
CMRR with Unbalanced
Source Resistance
CMRR with Unbalanced
Source Resistance
CMRR with Balanced
Source Resistance
CMRR with Balanced
Source Resistance
CMRR with Balanced
Source Resistance
CMRR with Unbalanced
Source Resistance
CMRR with Unbalanced
Source Resistance
CMRR with Unbalanced
Source Resistance
TL/H/5609 – 5
6
Typical Performance Characteristics (Continued)
Shield Driver Bias Voltage
Shield Driver Loading Error
Shield Driver Loading Error
Shield Driver Loading Error
Small Signal Transient
Response
Small Signal Transient
Response
Small Signal Transient
Response
Small Signal Transient
Response
Large Signal Transient
Response
Large Signal Transient
Response
Large Signal Transient
Response
Large Signal Transient
Response
TL/H/5609 – 6
7
Simplified Schematic (pin numbers in parentheses are for 8-pin package)
TL/H/5609 – 7
Theory of Operation
This voltage divided by the attenuation factor
R2
R4
e
R3 a R4 R1 a R2
is equal to the output-to-reference voltage. Hence, the overall gain is given by
VOUT R3 a R4 RE3-4
e
c
.
Ge
VIN
R4
RE1-2
Referring to the Simplified Schematic, it can be seen that
the input voltage is applied across the bases of Q1 and Q2
and appears between their emitters. If RE1-2 is the resistance across these emitters, a differential current equal to
VIN/RE1-2 flows from Q1’s emitter to Q2’s. The second
stage amplifier shown maintains Q1 and Q2 at equal collector currents by negative feedback to Q4. The emitter currents of Q3 and Q4 must therefore be unbalanced by an
amount equal to the current flow across RE1-2. Defining
RE3-4 e R5 a R6, the differential voltage across the emitters
of Q4 to Q3 is equal to
VIN
c RE3-4.
RE 1-2
8
Application Hints
overdrives these diodes conduct, greatly increasing input
currents. This behavior is illustrated in the IIN vs VIN plot in
the Typical Performance Characteristics. (The graph is not
symmetrical because at large input currents a portion of the
current into the device flows out the V b terminal.)
The input protection resistors allow a full 10V differential
input voltage without degradation even at G e 1000. At input
voltages more than one diode drop below Vb or two diode
drops above V a input, current increases rapidly. Diode
clamps to the supplies, or external resistors to limit current
to 20 mA, will prevent damage to the device.
The LM363 was designed to be as simple to use as possible, but several general precautions must be taken. The differential inputs are directly coupled and need a return path
to power supply common. Worst-case bias currents are only
10 nA for the LM363, so the return impedance can be as
high as 100 MX. Ground drops between signal return and IC
supply common should not be ignored. While the LM363
has excellent common-mode rejection, signals must remain
within the proper common-mode range for this specification
to apply. Operating common-mode range is guaranteed
from b10V to a 10V with g 15V supplies.
The high-gain (500 or 1000) versions have large gain-bandwidth products (15 MHz or 30 MHz) so board layout is fairly
critical. The differential input leads should be kept away
from output force and sense leads, especially at high impedances. Only 1 pF from output to positive input at 100 kX
source impedance can cause oscillations. The gain adjust
leads on the 16-pin package should be treated as inputs
and kept away from the output wiring.
REFERENCE AND SENSE INPUTS
The equivalent circuit is shown in the schematic diagram.
Limitations for correct operation are as follows. Maximum
differential swing between reference and sense pins is typically g 15V ( g 10V guaranteed). If this limit is exceeded, the
sense pin no longer controls the output, which then pegs
high or low. The negative common-mode limit is 1.5V below
Vb. (This is permissible because R2 and R4 are returned to
a node biased higher than Vb.) If large positive voltages are
applied to the reference and sense pins, the common-mode
range of the signal inputs begins to suffer as the drop
across R13 and R16 increases. For example, at g 15V supplies, VREF e VSENSE e 0V, signal input range is typically
b 12V to a 13.5V. At VREF e VSENSE e 15V, signal input
range drops to b11V to a 13.5V. The reference and sense
pins can be as much as 10V above V a as long as a restricted signal common-mode range (b10V min) can be tolerated.
For maximum bipolar output swing at g 15V supplies, the
reference pin should be returned to a voltage close to
ground. At lower supply voltages, the reference pin need
not be halfway between the supplies for maximum output
swing. For example, at V a e a 12V and Vb e b5V,
grounding the reference pin still allows a a 11V to b4V
swing. For single-supply systems, the reference pin can be
tied to either supply if a single output polarity is all that is
required. For a bipolar input and output, create a low impedance reference with an op amp and voltage divider or a
regulator (e.g., LM336, LM385, LM317L). This forms the reference for all succeeding signal-processing stages. (Don’t
connect the reference terminal directly to a voltage divider;
this degrades gain error.) See Figure 1 .
POWER SUPPLY
The LM363 may be powered from split supplies from g 5V
to g 18V (or single-ended supplies from 10V to 36V). Positive supply current is typically 1.2 mA independent of supply
voltage. The negative supply current is higher than the positive by the current drawn through the voltage dividers for the
reference and sense inputs (typ 600 mA total). The LM363’s
excellent PSRR often makes regulated supplies unnecessary. Actually, supply voltage can be as low as 7V total but
PSRR is severely degraded, so that well-regulated supplies
are recommended below 10V total. Split supplies need not
be balanced; output swing and input common-mode range
will simply not be symmetrical with unbalanced supplies. For
example, at a 12V and b5V supplies, input common-mode
range is typically a 10.5V to b2V and output swing is a 11V
to b4V.
When using ultra-low offset versions, best results are obtained at g 15V supplies. For example, the LM363-500’s offset voltage is guaranteed within 150 mV at g 15V at 25§ C.
Running at g 5V results in a worst-case negative PSRR error of 10V (b15V to b5V) multiplied by 3.2X10b6 (110 dB)
or 32 mV, increasing the worst-case offset. Positive PSRR
results in another 10 mV worst-case change.
INPUTS
The LM363 input circuitry is depicted in the Simplified Schematic. The input stage is run relatively rich (50 mA) for low
voltage noise and wide bandwidth; super-beta transistors
and bias-current cancellation (not shown) keep bias currents low. Due to the bias-current cancellation circuitry, bias
current may be either polarity at either input. While input
current noise is high relative to bias current, it is not significant until source resistance approaches 100 kX.
Input common-mode range is typically from 3V above V b to
1.5V below V a , so that a large potential drop between the
input signal and output reference can be accommodated.
However, a return path for the input bias current must be
provided; the differential input stage is not isolated from the
supplies. Differential input swing in the linear region is equal
to output swing divided by gain, and typically ranges from
1.3V at G e 10 to 13 mV at G e 1000.
Clamp diodes are provided to prevent zener breakdown and
resulting degradation of the input transistors. At large input
a. Usual configuration maximizes bipolar output swing.
TL/H/5609 – 8
b. Unequal supplies, output ground referred. Full output swing preserved referred to supplies.
FIGURE 1. Reference Connections
9
Application Hints (Continued)
TL/H/5609 – 9
c. Single Supply, Unipolar Output
d. Single Supply, Bipolar Output
FIGURE 1. Reference Connections (Continued)
duce an offset shift. A simple low-pass RC filter will usually
cure this problem (Figure 2 ). Use film type resistors for their
low thermal EMF. In highly noisy environments, LC filters
can be substituted for increased RF attenuation.
OUTPUTS
The LM363’s output can typically swing within 1V of the
supplies at light loads. While specified to drive a 2 kX load
to g 10V, current limit is typically 15 mA at room temperature. The output can stably drive capacitive loads up to
400 pF. For higher load capacitance, the amplifier may be
overcompensated (see COMPENSATION section, following). The output may be continuously shorted to ground
without damaging the device.
OFFSET VOLTAGE
The LM363’s offset voltage is internally trimmed to a very
low value. Note that data sheet values are given at
Tj e 25§ C, VCM e 0V and V a e Vb e 15V. For other conditions, warm-up drift, temperature drift, common-mode rejection and power supply rejection must be taken into account.
Warm-up drift, due to chip and package thermal gradients, is
an effect separate from temperature drift. Typical warm-up
drift is tabulated in the Electrical Characteristics; settling
time is approximately 5 minutes in still air. At load currents
up to 5 mA, thermal feedback effects are negligible
(DVOSs2mV at G e 1000).
Care must be taken in measuring the extremely low offset
voltages of the high gain amplifiers. Input leads must be
held isothermal to eliminate thermocouple effects. Oscillations, due to either heavy capacitive loading or stray capacitance from input to output, can cause erroneous readings.
In either case, overcompensation will help. High frequency
noise fed into the inputs may be rectified internally, and pro-
TL/H/5609 – 10
FIGURE 2. Low Pass Filter Prevents RF Rectification
Instrumentation amplifiers have both an input offset voltage
(VIOS) and an output offset voltage (VOOS). The total inputreferred offset voltage (VOSRTI) is related to the instrumentation amplifier gain (G) as follows: VOSRTI e VIOS a VOOS/
G. The offset voltage given in the LM363 specifications is
the total input-referred offset. As long as only one gain is
used, offset voltage can be nulled at either input or output
as shown in Figures 3a and 3b . When the 16-pin device is
used at multiple gain settings, both VIOS and VOOS should
be nulled to get minimum offset at all gains, as shown in
Figure 3c . The correct procedure is to trim VOOS for zero
output at G e 10, then trim VIOS at G e 1000.
TL/H/5609 – 11
FIGURE 3. Offset Voltage Trimming
10
Application Hints (Continued)
Because the LM363’s offset voltage is so low to begin with,
offset nulling has a negligible effect on offset temperature
drift. For example, zeroing a 100 mV offset, assuming external
resistor TC of 200 ppm/§ C and worst-case internal resistor
TC, results in an additional drift component of 0.08 mV/§ C.
For this reason, drift specifications are guaranteed, with or
without external offset nulling.
worst-case output offset of 50 mV, creating an input-referred error of 5 mV at G e 10 or 50 mV at G e 1000.
Increasing gain this way increases output offset error. An
LM363H-100 may have an output offset of 5 mV, resulting in
input referred offset component of 50 mV. Raising the gain
to 200 yields a 10 mV error at the output and changes input
referred error by an additional 50 mV.
External resistors connected to the reference and sense
pins can only increase the gain. If ultra-low output impedance is not critical, the technique in Figure 5 can be used to
trim the gain to nominal value. Alternatively, the VOS adjustment terminals on the 16-pin package may be used to trim
the gain (Figure 10b ).
GAIN ADJUSTMENT
Gain may be increased by adding an external voltage divider between output force and sense and reference; the preferred connection is shown in Figure 4 . Since both the
sense and reference pins look like 50 kX ( g 20 kX) to Vb,
impedances presented to both pins must be equal to avoid
offset error. For example, a 100X imbalance can create a
R1 and R2 should be as low as possible to avoid errors due to 50 kX
input impedance of reference and sense pins. Total resistance
(R2 a 2R1) should be above 4 kX, however, to prevent excessive load
on the LM363 output. The exact formula for calculating gain (G) is:
#
G e GO 1 a
2R1 R1
a
R2 50k
GO e preset gain
J
The last term may be ignored in applications where gain accuracy is not
critical. The table below gives suggested values for R1 and R2 along
with the calculated error due to ‘‘closest value’’ standard 1% resistors.
Total gain error tolerance includes contributions from LM363 GO error
and resistor tolerance ( g 1%) and works out to approximately 2.5% in
every case.
Pinout shown is for 16-pin package. This same technique can also be
used with 8-pin versions.
TL/H/5609–12
Gain Increase
1.5
2
2.5
3
4
5
6
7
8
9
10
R1
1.21k
1.21k
R2
5k
2.49k
2k
2k
1.78k
2k
2.49k
2.94k
3.48k
3.92k
4.42k
2.74k
2.05k
1.21k
1k
1k
1k
1k
1k
Error (typ)
a 0.6%
b 0.2%
0
1k
b 0.3%
b 0.6%
a 0.8%
a 0.5%
b 0.9%
a 0.4%
b 0.9%
b 0.7%
FIGURE 4. Increasing Gain
Pinout shown is for 8-pin versions.
This same technique can also be used
with 16-pin version.
TL/H/5609 – 13
FIGURE 5. Adjusting Gain, Alternate Technique
11
Application Hints (Continued)
Heavy Miller overcompensation on the 16-pin package can
degrade AC PSRR. A large capacitor between pins 15 and
16 couples transients on the positive supply to the output
buffer. Since the amplifier bandwidth is severely rolled off it
cannot keep the output at the correct state at moderate
frequencies. Hence, for good PSRR, either keep the Miller
capacitance under 1000 pF or use the pin 15-to-ground
compensation shown in Table I.
COMPENSATION AND OUTPUT CLAMPING
The LM363 is internally compensated for unity feedback
from output to sense. Increasing gain with external dividers
will decrease the bandwidth and increase stability margin.
Without external compensation, the amplifier can stably
drive capacitive loads up to 400 pF. When used as an op
amp (sense and reference pins grounded, feedback to inverting input), the LM363 is stable for gains of 100 or more.
For greater stability, the device may be over-compensated
as in Figure 6 . Tables I and II depict suggested compensation components along with the resulting changes in large
and small signal bandwidth for the 8-pin and 16-pin packages, respectively.
Note that the RC network from pin 8 of the 8-pin device to
ground has a large effect on power bandwidth, especially at
low gains. The Miller capacitance utilized for overcompensating the 16-pin device permits higher slew rate and larger
load capacitance for the same bandwidth, and is preferred
when bandwidth must be greatly reduced (e.g., to reduce
output noise).
TL/H/5609 – 14
FIGURE 6. Overcompensation
TABLE I. Overcompensation on 8-Pin Package
Compensation Network
(Pin 8 to Ground) ²
Gain
Ð
100 pF, 15k
1000 pF, 5k
0.01 mF,500X
0.1 mF
Ð
100 pF, 15k
1000 pF, 5k
0.01 mF, 500X
0.1 mF
Ð
100 pF, 15k
1000 pF, 5k
0.01 mF, 500X
0.1 mF
500
100
10
Small Signal
3 dB
Bandwidth
(kHz)
125
95
45
10
1
240
170
80
20
2
240
170
90
20
2
Power
Bandwidth
( g 10V Swing)
(Hz)
100k
15k
1.8k
200
20
100k
15k
1.8k
200
20
100k
15k
1.8k
200
20
Maximum
Capacitive
Load
(pF)
400
600
800
1000*
1000*
400
900
1200
1600*
2000*
400
900
1200
1600*
2000*
*Also stable for CL t 0.05 mF
² Pin 15 to ground on 16-pin package
TABLE II. Overcompensation on 16-Pin Package
Gain
1000
100
10
Compensation
Capacitor
(Pin 15 to 16)
Ð
10 pF
100 pF
1000 pF
0.01 mF
Ð
10 pF
100 pF
1000 pF
0.01 mF
Ð
10 pF
100 pF
1000 pF
0.01 mF
Small Signal
3 dB
Bandwidth
(Hz)
45k
16k
2.5k
250
25
140k
50k
7.5k
750
75
180k
60k
9k
900
90
*Also stable for CL t 0.05 mF
12
Power
Bandwidth
( g 10V Swing)
(Hz)
45k
16k
2.5k
250
25
100k
50k
7.5k
750
75
90k
50k
9k
900
90
Maximum
Capacitive
Load
(pF)
1000*
2000*
2500*
3000*
3000*
900
1600
2000*
2000*
2000*
600
1100
1600
2000*
2000*
Application Hints (Continued)
50 pF to ground at both shield driver outputs. Do not use
only one shield driver for a single-ended signal as oscillations can result; shield driver to input capacitance must be
roughly balanced ( g 30%). To further reduce noise pickup,
the shielded signal lines may be enclosed together in a
grounded shield. If a large amount of RF noise is the problem, the only sure cure is a filter capacitor at both inputs;
otherwise the RFI may be internally rectified, producing an
offset.
DC loading on the shield drivers should be minimized. The
drivers can only source approximately 40 mA; above this
value the input stage bias voltages change, degrading VOS
and CMRR. While the shield drivers can sink several mA,
VOS may degrade severely at loads above 100 mA (see
Shield Driver Loading Error curve in Typical Performance
Characteristics). Because the shield drivers are one diode
drop above the input levels, unbalanced leakage paths from
shield to input can produce an input offset at high source
impedances. Buffering with emitter-followers (Figure 8b ) reduces this leakage current by reducing the voltage differential and eliminates any loading on the amplifier.
Because the LM363’s output voltage is approximately one
diode drop below the voltage at pin 15 (pin 8 for the 8-pin
device), this point may be used to limit output swing as seen
in Figure 7a . Current available from this pin is only 50 mA, so
that zeners must have a sharp breakdown to clamp accurately. Alternatively, a diode tied to a voltage source could
be used as in Figure 7b .
TL/H/5609 – 15
FIGURE 7. Output Clamp
SHIELD DRIVERS
When differential signals are sent through long cables, three
problems occur. First, noise, both common-mode and differential, is picked up. Second, signal bandwidth is reduced by
the RC low-pass filter formed by the source impedance and
the cable capacitance. Finally, when these RC time constants are not identical (unbalanced source impedance
and/or unbalanced capacitance), AC common-mode rejection is degraded, amplifying both induced noise and
‘‘ground’’ noise. Either filtering at the amplifier inputs or
slowing down the amplifier by overcompensating will indeed
reduce the noise, but the price is slower response. The
LM363D’s dual shield drivers can actually increase bandwidth while reducing noise.
The way this is done is by bootstrapping out shield capacitance. The shield drivers follow the input signal. Since both
sides of the shield capacitance swing the same amount, it is
effectively out of the circuit at frequencies of interest.
Hence, the input signal is not rolled off and AC CMRR is not
degraded (Figure 8 ). The LM363D’s shield drivers can handle capacitances (shield to center conductor) as high as
1000 pF with source resistances up to 100 kX.
For best results, identical shielded cables should be used
for both signal inputs, although small mismatches in shield
driver to ground capacitance (s500 pF) do not cause problems. At certain low values of cable capacitance (50 pF –
200 pF), high frequency oscillations can occur at high
source resistance (t 10 kX). This is alleviated by adding
TL/H/5609 – 16
FIGURE 8. Driving Shielded Cables
MISCELLANEOUS TRIMMING
The VOS adjust and shield driver pins available on the 16pin package may be used to trim the other parameters besides offset voltage, as illustrated in Figure 10 . The bias-current trim relies on the fact that the voltage on the shield
driver and gain setting pins is one diode drop respectively
above and below the input voltage. Input bias current can
be held to within 100 pA over the entire common-mode
range, and input offset current always stays under 30 pA.
The CMRR trims use the shield driver pins to drive the VOS
adjust pins, thus maintaining the LM363’s ultra-high input
impedance.
13
Application Hints (Continued)
VOS. Both the gain and DC CMRR trims can degrade positive PSRR; the positive PSRR can then be nulled out if desired. The correct order of trimming from first to last is bias
current, gain, CMRR, negative PSRR, positive PSRR and
VOS.
If power supply rejection is critical, frequently only the negative PSRR need be adjusted, since the positive PSRR is
more tightly specified. Any or all of the trim schemes of
Figure 10 can be combined as desired. As long as the center tap of the 100k trimpot is returned to a voltage 200 mV
below V a , the trim schemes shown will not greatly affect
Top Trace: Cable Shield Grounded
Bottom Trace: Cable Shield Bootstrapped
TL/H/5609–17
TL/H/5609 – 18
FIGURE 9. Improved Response using Shield Drivers
TL/H/5609 – 19
FIGURE 10. Other Trims for 16-Pin Package
14
Typical Applications
4 mA-20 mA Two Wire Current Transmitter
TL/H/5609 – 20
The LM329 reference provides excellent line regulation and gain stability. When bridge is balanced
(IOUT e 4 mA), there’s no drop across R3 and R4, so that gain and offset adjustments are non-interactive. The LM334 configured as a zero-TC current source supplies quiescent current to circuit.
R11 provides current limiting.
Design Equations
IOS e (IR6 a IR7)
Gain e
#1
a
J
R2
e 4 mA
R1
DIOUT AV R2 a R3 a R4 10 mA
j X
j
DVIN R1
R3 a R4
mV
when AV e LM363 voltage gain
Pick I334 e
0.68V 68 mV
j 3.8 mA
a
R9
R10
VZ b 2.4V
e 26 mA
IMAX e I334 a
R11
IBRIDGE(MAX) j I334-I363-IZ j 1.5mA
Precision Current Source (Low Output Current)
R1 e R2
IOUT e
VIN
, l VIN l s 10V
GR1
TL/H/5609 – 21
Precision Voltage to Current Converter (Low Input Voltage)
R1 e R2
Req e R1 ll 50 kX
IOUT e
G VIN
G VIN
e
Req
1 kX
TL/H/5609 – 22
15
Typical Applications (Continued)
Curvature Corrected Platinum RTD Thermometer
*70k and 2k should track to 5 ppm/§ C
**Less than 5 ppm/§ C drift
² Less than 100 ppm/§ C drift
² ² These resistors should track to 20 ppm/§ C
³ Equivalent circuit, showing lead resistance
This thermometer is capable of 0.01§ C accuracy over b 50§ C to
a 150§ C. A unique trim arrangement eliminates cumbersome trim interactions so that zero, gain, and nonlinearity correction can be trimmed in
one oven trip. Extra op amps provide full Kelvin sensing on the sensor
without adding drift and offset terms found in other designs. A2 is configured as a Howland current pump, biasing the sensor with a fixed
current.
TL/H/5609 – 23
Resistors R2, R3, R4 and R5 from a bridge driven into balance by A1. In
balance, both inputs of A1 are at the same voltage. Since R6 e R7, A1
draws equal currents from both legs of the bridge. Any loading of the
R4/R5 leg by the sensor would unbalance the bridge; therefore, both
bridge taps are given to the sensor open circuit voltage and no current
is drawn.
Precision Temperature Controller
TL/H/5609 – 24
*Ultronix 105A wirewound
Thermistor e Yellow Springs Ý44032
Setpoint stability e 2.5X10b4§ C/Hr
16
Typical Applications (Continued)
Low Frequency Rolloff (AC Coupling)
f1 e
1
e 1 Hz
2qC1(50 kX)
f2 e 100 f1 e 100Hz
Reduced DC voltage gain
attenuates offset error and
1/f noise by a factor of 100.
TL/H/5609 – 25
Precision Comparator with Balanced Inputs and Variable Offset
Boosted Current Source with Limiting
R1 e R2
IO e
G VIN
R2
IMAX e
VBE
R2
j 60 mA
tpd j 15 mS at 1 mV overdrive
DVOUT e V2 a 0.6V
Hysteresis e
DVOUT
e 2 mV
G(R1 a R2)
Offset e VSENSE/G
g 1.3V range
TL/H/5609 – 26
Thermocouple Amplifier with Cold Junction Compensation
Input protection circuitry allows
thermocouple to short to 120 VAC without
damaging amplifier.
Calibration:
1) Apply 50 mV signal in place of thermocouple.
Trim R3 for VOUT e 12.25V.
2) Reconnect thermocouple. Trim R9 for correct
output.
TL/H/5609 – 27
17
Typical Applications (Continued)
Synchronous Demodulator
TL/H/5609 – 28
*Use square wave drive produced by optical chopper to run LF13333 switch inputs.
Pulsed Bridge Driver/Amplifier
TL/H/5609 – 29
18
Typical Applications (Continued)
Precision Barometer
**Parallel trim for 28.00× Hg e 0V
² Parallel trim for 32.00× Hg e 4V out
*B.L.H. Electronics ÝDHF-444114
Pressure Transducer,
350X input impedance.
Output e 1 mV/volt excitation/psi
TL/H/5609 – 30
Removing Large DC Offsets
*Optional bandlimiting to reduce noise.
Pick R1C1 e R2C2 e R3C3/10
e
1
2 q fl
TL/H/5609 – 31
fl e 0.1 Hz for values shown. Integrator nulls out offset error to
LM363 bias currents flowing into R1 and R2.
Removing Small DC Offsets
*Optional bandlimiting to reduce noise.
Low frequency break
frequency fl e
1
e 0.01 Hz
2qR1C1
Accommodates out referred offset of several volts. Limit is set by max
differential between reference and sense terminals.
TL/H/5609 – 32
19
20
Physical Dimensions inches (millimeters)
Metal Can Package (H)
Order Number LM363H-10, LM363H-100 or LM363H-500
NS Package Number H08C
21
LM363 Precision Instrumentation Amplifier
Physical Dimensions inches (millimeters) (Continued)
Hermetic Dual-In-Line Package (D)
Order Number LM363D
NS Package Number D16C
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