LT1769 Constant-Current/ Constant-Voltage 2A Battery Charger with Input Current Limiting DESCRIPTIO U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ The LT®1769 current mode PWM battery charger is a simple, efficient solution to fast charge modern rechargeable batteries including lithium-ion (Li-Ion), nickel-metalhydride (NiMH) and nickel-cadmium (NiCd) that require constant-current and/or constant-voltage charging. The internal switch is capable of delivering 2A** DC current (3A peak current). Charge current can be programmed by resistors or a DAC to within 5%. With 0.5% reference voltage accuracy, the LT1769 meets the critical constant-voltage charging requirement for Li-Ion cells. Simple Solution to Charge NiCd, NiMH and Lithium Rechargeable Batteries—Charging Current Programmed by Resistors or DAC Adapter Current Limit Allows Maximum Possible Charging Current During System Use* Precision 0.5% Accuracy for Voltage Mode Charging High Efficiency Current Mode PWM with 3A Internal Switch 5% Charge Current Accuracy Adjustable Undervoltage Lockout Automatic Shutdown When AC Adapter is Removed Low Reverse Battery Drain Current: 3µA Current Sensing Can Be at Either Terminal of the Battery Charging Current Soft Start Shutdown Control Available in 28-Lead Narrow SSOP Package A third control loop is provided to regulate the current drawn from the input AC adapter. This allows simultaneous operation of the equipment and battery charging without overloading the adapter. Charge current is reduced to keep the adapter current below specified levels. The LT1769 can charge batteries ranging from 1V to 20V. Ground sensing of current is not required and the battery’s negative terminal can be tied directly to ground. A saturating switch running at 200kHz gives high charging efficiency and small inductor size. A blocking diode is not required between the chip and the battery because the chip goes into sleep mode and drains only 3µA when the wall adapter is unplugged. U APPLICATIO S ■ ■ Chargers for NiCd, NiMH, Lead-Acid, Lithium Rechargeable Batteries Switching Regulators with Precision Current Limit , LTC and LT are registered trademarks of Linear Technology Corporation. *US patent number 5,723,970 **See LT1510 for 1.5A charger; see LT1511 for 3A charger U TYPICAL APPLICATIO C2 0.47µF †† D1 SS24 GND CLP SW CLN VCC C1 1µF UV COMP1 2nF 10k VC SPIN RS3 200Ω 1% BAT VIN (ADAPTER INPUT) 11V TO 28V TO MAIN SYSTEM LOAD R5† UNDERVOLTAGE LOCKOUT R6 5k PROG OVP SENSE NOTE: COMPLETE LITHIUM-ION CHARGER, NO TERMINATION REQUIRED. RS4, R7 AND C1 ARE OPTIONAL FOR IIN LIMITING *TOKIN OR UNITED CHEMI-CON/MARCON CERAMIC SURFACE MOUNT **22µH SUMIDA CDRH125 † SEE APPLICATIONS INFORMATION FOR INPUT CURRENT LIMIT AND UNDERVOLTAGE LOCKOUT †† GENERAL SEMICONDUCTOR. FOR TJ LESS THEN 100°C MBRS130LT3 CAN BE USED RS4† ADAPTER CURRENT SENSE CIN* 15µF BOOST LT1769 L1** 22µH D2 1N4148 D3†† SS24 R7† 500Ω 300Ω 0.33µF CPROG 1µF RPROG 4.93k 1% RS2 200Ω 1% RS1 0.05Ω BATTERY CURRENT SENSE R3 + 390k 0.25% BATTERY VOLTAGE SENSE VBAT COUT 22µF TANT 8.4V Li-Ion 1511 • F01 R4 162k 0.25% Figure 1. 2A Lithium-Ion Battery Charger 1 LT1769 U W U U W W W ABSOLUTE MAXIMUM RATINGS PACKAGE/ORDER INFORMATION (Note 1) Supply Voltage (VCC, CLP and CLN Pin Voltage) ......................... 30V BOOST Pin Voltage with Respect to VCC ................. 25V IBAT (Average)........................................................... 2A Operating Junction Temperature Range Commercial ........................................... 0°C to 125°C Industrial ......................................... – 40°C to 125°C Operating Ambient Temperature Commercial ............................................ 0°C to 70°C Industrial ........................................... – 40°C to 85°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C ORDER PART NUMBER TOP VIEW GND** 1 28 GND** GND** 2 27 GND** GND** 3 26 GND** SW 4 25 VCC1* BOOST 5 24 VCC2* UV 6 23 VCC3* GND** 7 22 GND** GND** 8 21 PROG OVP 9 20 VC CLP 10 19 UVOUT CLN 11 18 COMP2 COMP1 12 17 BAT SENSE 13 16 SPIN GND** 14 15 GND** LT1769CGN LT1769IGN *ALL VCC PINS SHOULD BE CONNECTED TOGETHER CLOSE TO THE PINS ** ALL GND PINS ARE FUSED TO INTERNAL DIE ATTACH PADDLE FOR HEAT SINKING. CONNECT THESE PINS TO EXPANDED PC LANDS FOR PROPER HEAT SINKING. 35°C/W THERMAL RESISTANCE ASSUMES AN INTERNAL GROUND PLANE DOUBLING AS A HEAT SPREADER GN PACKAGE 28-LEAD PLASTIC SSOP TJMAX = 125°C, θJA = 35°C/ W** Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, VBAT = 8V, RS2 = RS3 = 200Ω (see Block Diagram), VCLN = VCC. No load on any outputs unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS 4.5 4.6 6.8 7.0 mA mA 100 10 107 12 12 mV mV mV 110 14 13 mV mV mV Overall Supply Current VPROG = 2.7V, VCC ≤ 20V VPROG = 2.7V, 20V < VCC ≤ 25V Sense Amplifier CA1 Gain and Input Offset Voltage (With RS2 = 200Ω, RS3 = 200Ω) (Measured across RS1)(Note 2) 8V ≤ VCC ≤ 25V , 0V ≤ VBAT ≤ 20V RPROG = 4.93k RPROG = 49.3k TA < 0°C VCC = 28V, VBAT = 20V RPROG = 4.93k RPROG = 49.3k TA < 0°C ● ● ● ● 93 8 7 ● ● 90 6 7 6 VCC Undervoltage Lockout (Switch OFF) Threshold Measured at UV Pin ● 7 8 V UV Pin Input Current 0.2V ≤ VUV ≤ 8V ● 0.1 5 µA UV Output Voltage at UVOUT Pin In Undervoltage State, IUVOUT = 70µA ● 0.1 0.5 V UV Output Leakage Current at UVOUT Pin 8V ≤ VUV, VUVOUT = 5V ● 0.1 3 µA Reverse Current from Battery (When VCC Is Not Connected, VSW Is Floating) VBAT ≤ 20V, VUV ≤ 0.4V 3 15 µA 2 LT1769 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, VBAT = 8V, RS2 = RS3 = 200Ω (see Block Diagram), VCLN = VCC. No load on any outputs unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS 0.1 0.25 6 8 10 20 9 12 µA µA mA mA 0.15 0.25 Ω 25 35 mA/A 2 4 100 200 µA µA 20 µA Overall Boost Pin Current VCC = 20V, VBOOST = 0V VCC = 28V, VBOOST = 0V 2V ≤ VBOOST – VCC < 8V (Switch ON) 8V ≤ VBOOST – VCC ≤ 25V (Switch ON) Switch Switch ON Resistance 8V ≤ VCC ≤ VMAX, ISW = 2A, VBOOST – VSW ≥ 2V ∆IBOOST/∆ISW During Switch ON VBOOST = 24V, ISW ≤ 2A Switch OFF Leakage Current VSW = 0V, VCC ≤ 20V 20V < VCC ≤ 28V ● ● ● Minimum IPROG for Switch ON ● 2 4 Minimum IPROG for Switch OFF ● 1 2.4 Maximum VBAT for Switch ON ● mA VCC – 2 V – 125 µA Current Sense Amplifier CA1 Inputs (Sense, BAT) Input Bias Current ● Input Common Mode Low ● Input Common Mode High ● – 50 – 0.25 SPIN Input Current V VCC – 2 V – 100 – 200 µA 2.465 2.477 V 2.489 2.489 V V Reference Reference Voltage (Note 3) Reference Voltage RPROG = 4.93k, Measured at OVP with VA Supplying IPROG and Switch OFF All Conditions of VCC, TA ≥ 0°C TA < 0°C (Note 4) 2.448 ● ● 2.441 2.43 Oscillator Switching Frequency Switching Frequency All Conditions of VCC, TA ≥ 0°C TA < 0°C 180 200 220 kHz ● ● 170 160 200 230 230 kHz kHz 90 85 93 ● 150 250 Maximum Duty Cycle % % Current Amplifier CA2 Transconductance VC = 1V, IVC = ±1µA Maximum VC for Switch OFF IVC Current (Out of Pin) ● VC ≥ 0.6V VC < 0.45V 550 µmho 0.6 V 100 3 µA mA 3 LT1769 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 16V, VBAT = 8V. No load on any outputs unless otherwise noted. PARAMETER CONDITIONS MIN TYP MAX UNITS Transconductance (Note 3) Output Current from 50µA to 500µA 0.25 0.6 1.3 mho Output Source Current VOVP = VREF + 10mV, VPROG = VREF + 10mV 1.1 OVP Input Bias Current VA Output Current at 0.5mA VA Output Current at 0.5mA, TA > 90°C Voltage Amplifier VA ● ● mA ±3 ±10 25 nA nA –15 Current Limit Amplifier CL1, 8V ≤ Input Common Mode Turn-On Threshold 0.5mA Output Current 93 100 107 mV Transconductance Output Current from 50µA to 500µA 0.5 1 2 mho CLP Input Current 0.5mA Output Current, VUV ≥ 0.4V 0.3 1 µA CLN Input Current 0.5mA Output Current VUV ≥ 0.4V 0.8 2 mA Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: Tested with Test Circuit 1. Note 3: Tested with Test Circuit 2. Note 4: A linear interpolation can be used for reference voltage specification between 0°C and – 40°C. U W TYPICAL PERFORMANCE CHARACTERISTICS 100 98 ICC vs VCC ICC vs Duty Cycle Efficiency of Figure 1 Circuit 7.0 8 VIN = 16.5 VBAT = 8.4V 7 6 94 TJ = 25°C CHARGER EFFICIENCY 90 88 5 TJ = 0°C TJ = 125°C 4 TJ = 25°C 3 86 INCLUDES LOSS IN DIODE D3 84 ICC (mA) 92 ICC (mA) EFFICIENCY (%) TJ = 0°C 6.5 96 2 6.0 TJ = 125°C 5.5 5.0 1 82 4.5 0 80 0.2 0.6 1.0 1.4 IBAT (A) 1.8 2.2 1769 G01 4 MAXIMUM DUTY CYCLE VCC = 16V 0 10 20 30 40 50 60 DUTY CYCLE (%) 70 80 1769 G02 0 5 10 15 VCC (V) 20 25 30 1769 G03 LT1769 U W TYPICAL PERFORMANCE CHARACTERISTICS IVA vs ∆VOVP (Voltage Amplifier) VREF Line Regulation 0.003 4 0.002 3 ALL TEMPERATURES ∆VOVP (mV) ∆VREF (V) 0.001 0 2 TJ = 125°C –0.001 1 –0.002 –0.003 TJ = 25°C 0 10 5 15 VCC (V) 20 25 0 30 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 IVA (mA) 1769 G04 1769 G05 Maximum Duty Cycle VC Pin Characteristics 98 –1.20 –1.08 97 –0.96 –0.84 –0.72 95 IVC (mA) DUTY CYCLE (%) 96 94 –0.60 –0.48 93 –0.36 92 –0.24 –0.12 91 0 90 0 20 60 40 80 100 120 JUNCTION TEMPERATURE (°C) 0.12 140 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0 VC (V) 1769 G06 1769 G07 Reference Voltage vs Temperature PROG Pin Characteristics 2.470 6 REFERENCE VOLTAGE (V) 2.468 IPROG (mA) TJ = 125°C TJ = 25°C 0 2.466 2.464 2.462 2.460 2.458 –6 0 1 2 3 VPROG (V) 4 5 1769 G08 0 25 125 50 75 100 JUNCTION TEMPERATURE 150 1769 G09 5 LT1769 U U U PIN FUNCTIONS GND (Pins 1 to 3, 7, 8, 14, 15, 22, 26 to 28): Ground Pins. Must be connected to expanded PC lands for proper heat sinking. See Applications Information section for details. SW (Pin 4): Switch Output. The Schottky catch diode must be placed with very short lead length in close proximity to SW pin and GND. BOOST (Pin 5): This pin is used to bootstrap and drive the switch power NPN transistor to a low on-voltage for low power dissipation. In Figure 1, VBOOST = VCC + VBAT when switch is on. For lowest IC power dissipation, connect boost diode D1 to a 3V to 6V at 30mA voltage source (see Figure 10). UV (Pin 6): Undervoltage Lockout Input. The rising threshold is at 6.7V with a hysteresis of 0.5V. Switching stops in undervoltage lockout. When the input supply (normally the wall adapter output) to the IC is removed, the UV pin must be pulled down to below 0.7V (a 5k resistor from adapter output to GND is required) otherwise the reverse battery current drained by the IC will be approximately 200µA instead of 3µA. Do not leave the UV pin floating. When connected to VIN with no resistor divider, the builtin 6.7V undervoltage lockout will be effective. OVP (Pin 9): This is the input to amplifier VA with a threshold of 2.465V. Typical bias current is about 3nA out of this pin. For charging lithium-ion batteries, VA monitors the battery voltage and reduces charging when battery voltage reaches the preset value. If it is not used, the OVP pin should be grounded. CLP (Pin 10): This is the positive input to the input current limit amplifier CL1. The threshold is set at 100mV. When used to limit supply current, a filter is needed to filter out the 200kHz switching noise. CLN (Pin 11): This is the negative input to the input current limit amplifier CL1. COMP1 (Pin 12): This is the compensation node for the input current limit amplifier CL1. At input adapter current limit, this node rises to 1V. By forcing COMP1 low with an external transistor, amplifier CL1 will be defeated (no adapter current limit). COMP1 can source 200µA. If this function is not used, the resistor and capacitor on COMP1 pin, shown on the Figure 1 circuit, are not needed. 6 SENSE (Pin 13): Current Amplifier CA1 Input. Sensing can be at either terminal of the battery. SPIN (Pin 16): This pin is for the current amplifier CA1 bias. It must be connected to RS1 as shown in the 2A Lithium Battery Charger (Figure 1). BAT (Pin 17): Current Amplifier CA1 Input. COMP2 (Pin 18): This is also a compensation node for amplifier CL1. Voltage on this pin rises to 2.8V at input adapter current limit and/or at constant-voltage charging. UVOUT (Pin 19): This is an open-collector output for undervoltage lockout status. It stays low in undervoltage state. With an external pull-up resistor, it goes high at valid VCC. Note that the base drive of the open-collector NPN comes from CLN pin. UVOUT stays low only when CLN is higher than 2V. Pull-up current should be kept under 100µA. VC (Pin 20): This is the inner loop control signal for the current mode PWM. Switching starts at 0.7V. In normal operation, a higher VC corresponds to higher charge current. A capacitor of at least 0.33µF to GND filters out noise and controls the rate of soft start. To stop switching, pull this pin low. Typical output current is 30µA. PROG (Pin 21): This pin is for programming the charge current and for system loop compensation. During normal operation, VPROG stays close to 2.465V. If it is shorted to GND switching will stop. When a microprocessor controlled DAC is used to program charge current, it must be capable of sinking current at a compliance up to 2.465V. VCC1, VCC2, VCC3 (Pins 23 to 25): Input Supply. For good bypass, a low ESR capacitor of 15µF or higher is required, with the lead length kept to a minimum. VCC should be between 8V and 28V and at least 3V higher than VBAT. Undervoltage lockout starts and switching stops when VCC goes below 7V typical. Note that there is an internal parasitic diode from SW pin to VCC pin. Do not force VCC below SW by more than 0.7V with battery present. All three VCC pins should be shorted together close to the pins. LT1769 W BLOCK DIAGRAM – UV UVOUT + + 7V 200kHz OSCILLATOR + SHUTDOWN 0.7V + VCC – VSW S BOOST – VCC R R + + SLOPE COMPENSATION SW 1.5V VCC SPIN VBAT – PWM C1 – + B1 + GND QSW R R2 + SENSE RS3 – BAT RS2 CA1 IPROG R1 1k R3 0VP – VA CA2 + 75k RS1 BAT + VC IBAT gm = 0.64 VREF Ω – VREF 2.465V 100mV + CLP + CL1 – CLN COMP1 COMP2 PROG CPROG IPROG RPROG )(R ) (I IBAT = PROG S2 RS1 RS2 = 2.465V RPROG RS1 1769 BD ( )( ) (RS3 = RS2) 7 LT1769 TEST CIRCUITS Test Circuit 1 SPIN LT1769 CA1 CA2 1k + 0.047µF + – VC 60k – SENSE RS3 200Ω BAT RS2 200Ω RS1 100Ω + VBAT VREF PROG 1µF RPROG 300Ω + LT1006 1k + 1769 TC01 – ≈ 0.65V 20k Test Circuit 2 LT1769 OVP + VA – VREF PROG 10k IPROG 10k 0.47µF RPROG – + + LT1013 2.465V 1769 TC02 U OPERATION The LT1769 is a current mode PWM step-down (buck) switcher. The battery DC charge current is programmed by a resistor RPROG (or a DAC output current) at the PROG pin (see Block Diagram). Amplifier CA1 converts the charge current through RS1 to a much lower current IPROG fed into the PROG pin. Amplifier CA2 compares the output of CA1 with the programmed current and drives the PWM control loop to force them to be equal. High DC accuracy is achieved with averaging capacitor CPROG. Note that IPROG has both AC and DC components. IPROG goes through R1 and generates a ramp signal that is fed to the PWM control comparator C1 through buffer B1 and level 8 shift resistors R2 and R3, forming the current mode inner loop. The BOOST pin drives the switch NPN QSW into saturation and reduces power loss. For batteries like lithium-ion that require both constant-current and constant-voltage charging, the 0.5%, 2.465V reference and the amplifier VA reduce the charge current when battery voltage reaches the preset level. For NiMH and NiCd, VA can be used for overvoltage protection. When the input voltage is removed, the VCC pin drops to 0.7V below the battery voltage, forcing the charger into a low battery drain (3µA typical) sleep mode. To shut down the charger, simply pull the VC pin low with a transistor. LT1769 U U W U APPLICATIONS INFORMATION Input and Output Capacitors In the 2A Lithium-Ion Battery Charger (Figure 1), the input capacitor (CIN) is assumed to absorb all input switching ripple current in the converter, so it must have adequate ripple current rating. Worst-case RMS ripple current will be equal to one half of the output charge current. Actual capacitance value is not critical. Solid tantalum capacitors such as the AVX TPS and Sprague 593D series have high ripple current rating in a relatively small surface mount package, but caution must be used when tantalum capacitors are used for input bypass. High input surge currents are possible when the adapter is hot-plugged to the charger and solid tantalum capacitors have a known failure mechanism when subjected to very high turn-on surge currents. Selecting a high voltage rating on the capacitor will minimize problems. Consult with the manufacturer before use. Alternatives include new high capacity ceramic (5µF to 20µF) from Tokin or United Chemi-Con/ Marcon, et al. Sanyo OS-CON can also be used. The output capacitor (COUT) is also assumed to absorb output switching ripple current. The general formula for capacitor ripple current is: ( ) V 0.29 (VBAT) 1 – BAT VCC IRMS = (L1)(f) For example, VCC = 16V, VBAT = 8.4V, L1 = 20µH, and f = 200kHz, IRMS = 0.3A. EMI considerations usually make it desirable to minimize ripple current in the battery leads. Beads or inductors can be added to increase battery impedance at the 200kHz switching frequency. Switching ripple current splits between the battery and the output capacitor depending on the ESR of the output capacitor and the battery impedance. If the ESR of COUT is 0.2Ω and the battery impedance is raised to 4Ω with a bead or inductor, only 5% of the ripple current will flow into the battery. Soft-Start and Undervoltage Lockout The LT1769 is soft-started by the 0.33µF capacitor on the VC pin. On start-up, the VC pin voltage will quickly rise to 0.5V, then ramp at a rate set by the internal 45µA pull-up current and the external capacitor. Charge current starts ramping up when VC pin voltage reaches 0.7V and full current is achieved with VC at 1.1V. With a 0.33µF capacitor, the time to reach full charge current is about 10ms and it is assumed that input voltage to the charger will reach full value in less than 10ms. The capacitor can be increased up to 1µF if longer input start-up times are needed. In any switching regulator, conventional time-based softstarting can be defeated if the input voltage rises much slower than the time out period. This happens because the switching regulators in the battery charger and the computer power supply are typically supplying a fixed amount of power to the load. If the input voltage comes up slowly compared to the soft-start time, the regulators will try to deliver full power to the load when the input voltage is still well below its final value. If the adapter is current limited, it cannot deliver full power at reduced output voltages and the possibility exists for a quasi “latch” state where the adapter output stays in a current limited state at reduced output voltage. For instance, if maximum charger plus computer load power is 25W, a 15V adapter might be current limited at 2A. If adapter voltage is less than (25W/2A = 12.5V) when full power is drawn, the adapter voltage will be pulled down by the constant 25W load until it reaches a lower stable state where the switching regulators can no longer supply full load. This situation can be prevented by utilizing undervoltage lockout, set higher than the minimum adapter voltage where full power can be achieved. A fixed undervoltage lockout of 7V is built into the LT1769. This 7V threshold can be increased by adding a resistive divider to the UV pin as shown in Figure 2. Internal lockout is performed by clamping the VC pin low. The VC pin is released from its clamped state when the UV pin rises above 7V and is pulled low when the UV pin drops below 6.5V (0.5V hysteresis). At the same time UVOUT goes high with an external pull-up resistor. This signal can be used to alert the system that charging is about to start. The charger will start delivering current about 4ms after VC is released, as set by the 0.33µF capacitor. A resistor divider is used to set the desired VCC lockout voltage as shown in Figure 2. A typical value for R6 is 5k and R5 is found from: 9 LT1769 U U W U APPLICATIONS INFORMATION R5 = ally, batteries will automatically be charged at the maximum possible rate of which the adapter is capable. R6(VIN – VUV ) VUV VUV = Rising lockout threshold on the UV pin VIN = Charger input voltage that will sustain full load power Example: With R6 = 5k, VUV = 6.7V and setting VIN at 12V; R5 = 5k (12V – 6.7V)/6.7V = 4k The resistor divider should be connected directly to the adapter output as shown, not to the VCC pin, to prevent battery drain with no adapter voltage. If the UV pin is not used, connect it to the adapter output (not VCC) and connect a resistor no greater than 5k to ground. Floating this pin will cause reverse battery current to increase from 3µA to 200µA. If connecting the unused UV pin to the adapter output is not possible, it can be grounded. Although it would seem that grounding the pin creates a permanent lockout state, the UV circuitry is arranged for phase reversal with low voltages on the UV pin to allow the grounding technique to work. 100mV + + – Charge Current Programming The basic formula for charge current is (see Block Diagram): IBAT = IPROG ( )( RS2 2.465V = RS1 RPROG )( ) RS2 RS1 where RPROG is the total resistance from PROG pin to ground. CLP 1µF CL1 This is accomplished by sensing total adapter output current and adjusting the charge current downward if a preset adapter current limit is exceeded. True analog control is used, with closed-loop feedback ensuring that adapter load current remains below the limit. Amplifier CL1 in Figure 2 senses the voltage across RS4, connected between the CLP and CLN pins. When this voltage exceeds 100mV, the amplifier will override the programmed charge current to limit adapter current to 100mV/RS4. A lowpass filter formed by 500Ω and 1µF is required to eliminate switching noise. If the input current limit is not used, both CLP and CLN pins should be connected to VCC. CLN 500Ω RS4* VCC AC ADAPTER OUTPUT VIN + R5 LT1769 UV *RS4 = 100mV ADAPTER CURRENT LIMIT R6 For the sense amplifier CA1 biasing purpose, RS3 should have the same value as RS2 and SPIN should be connected directly to the sense resistor (RS1) as shown in the Block Diagram. For example, 2A charge current is needed. For low power dissipation on RS1 and enough signal to drive the amplifier CA1, let RS1 = 100mV/2A = 0.05Ω. This limits RS1 power to 0.2W. Let RPROG = 5k, then: 1769 F02 Figure 2. Adapter Input Current Limiting Adapter Current Limiting An important feature of the LT1769 is the ability to automatically adjust charge current to a level which avoids overloading the wall adapter. This allows the product to operate at the same time the batteries are being charged without complex load management algorithms. Addition- 10 )(R ) (I )(R RS2 = RS3 = BAT PROG S1 2.465V (2A)(5k)(0.05) = = 200Ω 2.465V Charge current can also be programmed by pulse width modulating IPROG with a switch Q1 to RPROG at a frequency higher than a few kHz (Figure 3). Charge current will be proportional to the duty cycle of the switch with full current at 100% duty cycle. LT1769 U U W U APPLICATIONS INFORMATION LT1769 PROG 300Ω RPROG 4.7k 5V 0V CPROG 1µF Q1 VN2222 PWM IBAT = (DC)(2A) 1769 F03 Figure 3. PWM Current Programming When power is on, there is about 200µA of current flowing out of the BAT and SENSE pins. If the battery is removed during charging, and total load including R3 and R4 is less than 200µA, VBAT could float up to VCC even though the loop has turned switching off. To keep VBAT regulated to the battery voltage in this condition, R3 and R4 can be chosen to draw 0.5mA and Q3 can be added to disconnect them when power is off (Figure 4). R5 isolates the OVP pin from any high frequency noise on VIN. An alternative method is to use a Zener diode with a breakdown voltage two or three volts higher than battery voltage to clamp the VBAT voltage. Lithium-Ion Charging The 2A Lithium-Ion Battery Charger (Figure 1) charges at a constant 2A until battery voltage reaches a limit set by R3 and R4. The charger will then automatically go into a constant-voltage mode with current decreasing to near zero over time as the battery reaches full charge. This is the normal regimen for lithium-ion charging, with the charger holding the battery at “float” voltage indefinitely. In this case no external sensing of full charge is needed. Battery Voltage Sense Resistors Selection To minimize battery drain when the charger is off, current through the R3/R4 divider is set at 15µA. The input current to the OVP pin is 3nA and the error can be neglected. With divider current set at 15µA, VBAT = 8.4V, R4 = 2.465/15µA = 162k and, R3 = (R4)(VBAT − 2.465) = 162k (8.4 − 2.465) 2.465 2.465 = 390k Li-Ion batteries typically require float voltage accuracy of 1% to 2%. Accuracy of the LT1769 OVP voltage is ±0.5% at 25°C and ±1% over full temperature. This leads to the possibility that very accurate (0.1%) resistors might be needed for R3 and R4. Actually, the temperature of the LT1769 will rarely exceed 50°C in float mode because charging currents have tapered off to a low level, so 0.25% resistors will normally provide the required level of overall accuracy. LT1769 Q3 VN2222 R5 220k VBAT + R3 12k 0.25% 8.4V VIN OVP R4 4.99k 0.25% 1769 F04 Figure 4. Disconnecting Voltage Divider Some battery manufacturers recommend terminating the constant-voltage float mode after charge current has dropped below a specified level (typically around 10% of the full current) and a further time out period of 30 to 90 minutes has elapsed. This may extend battery life, so check with the manufacturer for details. The circuit in Figure 5 will detect when charge current has dropped below 270mA. This logic signal is used to initiate a timeout period, after which the LT1769 can be shut down by pulling the VC pin low with an open collector or drain. Some external means must be used to detect the need for additional charging or the charger may be turned on periodically to complete a short float-voltage cycle. Current trip level is determined by the battery voltage, R1 through R3 and the sense resistor (RS1). D2 generates hysteresis in the trip level to avoid multiple comparator transitions. 11 LT1769 U U W U APPLICATIONS INFORMATION IBAT R1 = RS3 200Ω SENSE RS1 0.05Ω LT1769 RS2 200Ω BAT VBAT ADAPTER OUTPUT BAT R1* C1 1.6k 0.1µF 3.3V OR 5V D1 1N4148 3 8 – 7 LT1011 2 R2 560k + R4 470k NEGATIVE EDGE TO TIMER 4 ILOW R2 = (2.465)(2000 ) IHI − ILOW All battery chargers with fast charge rates require some means to detect full charge in the battery and terminate the high charge current. NiCd batteries are typically charged at high current until a temperature rise or battery voltage decrease is detected as an indication of near full charge. The charging current is then reduced to a much lower value and maintained as a constant trickle charge. An intermediate “top off” current may also be used for a fixed time period to reduce total charge time. 1 D2 1N4148 R3 430k * TRIP CURRENT = = R1(VBAT) (R2 + R3)(RS1) (1.6k)(8.4V) ≈ 270mA (560k + 430k)(0.05Ω) 1769 F04 Figure 5. Current Comparator for Initiating Float Time Out Nickel-Cadmium and Nickel-Metal-Hydride Charging The 2A Lithium-Ion Battery Charger shown in Figure 1 can be modified to charge NiCd or NiMH batteries. For example, if a 2-level charge is needed; 1A when Q1 is on and 100mA when Q1 is off. PROG 300Ω 1µF NiMH batteries are similar in chemistry to NiCd but have two differences related to charging. First, the inflection characteristic in battery voltage as full charge is approached is not nearly as pronounced. This makes it more difficult to use – dV/dt as an indicator of full charge, and an increase in battery temperature is more often used with a temperature sensor in the battery pack. Secondly, constant trickle charge may not be recommended. Instead, a moderate level of current is used on a pulse basis (≈ 1% to 5% duty cycle) with the time-averaged value substituting for a constant low trickle. Please contact the Linear Technology Applications department about charge termination circuits. If overvoltage protection is needed, R3 and R4 can be calculated according to the procedure described in LithiumIon Charging section. The OVP pin should be grounded if not used. LT1769 R1 49.3k R2 5.49k Q1 1769 F05 Figure 6. 2-Level Charging For 1A full current, the current sense resistor (RS1) should be increased to 0.1Ω so that enough signal (10mV) will be across RS1 at 0.1A trickle charge to keep charging current accurate. For a 2-level charger, R1 and R2 are found from: 12 (2.465)(2000 ) When a microprocessor DAC output is used to control charge current, it must be capable of sinking current at a compliance up to 2.5V if connected directly to the PROG pin. Thermal Calculations If the LT1769 is used for charging currents above 1A, a thermal calculation should be done to ensure that junction temperature will not exceed 125°C. Power dissipation in the IC is caused by bias and driver current, switch resistance and switch transition losses. The GN package, with a thermal resistance of 35°C/W, can provide a full 2A charging current in many situations. A graph is shown in the Typical Performance Characteristics section. LT1769 U U W U APPLICATIONS INFORMATION PBIAS = (3.5mA )(VIN) + 1.5mA(VBAT ) (VBAT ) 2 + VIN SW [7.5mA + (0.012)(IBAT )] LT1769 C2 L1 (IBAT )(VBAT )2 1+ VBAT 30 PDRIVER = 55(VIN) BOOST D2 SPIN VX IVX (I ) (RSW )(VBAT ) + t V I f PSW = BAT ( OL)( IN)( BAT )( ) VIN 1769 F07 + 10µF 2 RSW = Switch ON resistance ≈ 0.16Ω tOL = Effective switch overlap time ≈ 10ns f = 200kHz Example: VIN = 19V, VBAT = 12.6V, IBAT = 2A: ( (12.6) + )( ) ( ) PBIAS = 3.5mA 19 + 1.5mA 12.6 2 19 [7.5mA + (0.012)(2000mA)] = 0.35W 2 PDRIVER The average IVX required is: PDRIVER 0.09 W = = 28mA 3.3V VX The previous example shows the dramatic drop in driver power dissipation when the boost diode (D2) is connected to an external 3.3V source instead of the 12.6V battery. PDRIVER drops from 0.43W to 0.09W resulting in an approximately 12°C drop in junction temperature. −9 Fused-lead packages conduct most of their heat out the leads. This makes it very important to provide as much PC board copper around the leads as is practical. Total thermal resistance of the package-board combination is dominated by the characteristics of the board in the immediate area of the package. This means both lateral thermal resistance across the board and vertical thermal resistance through the board to other copper layers. Each layer acts as a thermal heat spreader that increases the heat sinking effectiveness of extended areas of the board. 2 19 = 0.42 + 0.08 = 0.5W Total Power in the IC is: 0.35 + 0.43 + 0.5 = 1.3W Temperature rise will be (1.3W)(35°C/W) = 46°C. This assumes that the LT1769 is properly heat sunk by connecting the eleven fused ground pins to expanded traces and that the PC board has a backside or internal plane for heat spreading. The PDRIVER term can be reduced by connecting the boost diode D2 (see Figure 7) to a lower system voltage (lower than VBAT) instead of VBAT. (IBAT )(VBAT )(VX ) 1+ V30X Then PDRIVER = 55(VIN ) For example, VX = 3.3V then: (2A)(12.6V)(3.3V) 1+ 330.3V = = 0.09 W 55(19V ) (2)(12.6) 1+ 1230.6 = 0.43W = 55(19) (2) (0.16)(12.6) + 10 (19)(2)(200kHz) = PDRIVER PSW Figure 7. Lower VBOOST Total board area becomes an important factor when the area of the board drops below about 20 square inches. The graph in Figure 8 shows thermal resistance vs board area for 2-layer and 4-layer boards with continuous copper planes. Note that 4-layer boards have significantly lower thermal resistance, but both types show a rapid increase for reduced board areas. Figure 9 shows actual measured lead temperatures for chargers operating at full current. 13 LT1769 U U W U APPLICATIONS INFORMATION STANDARD CONNECTION THERMAL RESISTANCE (°C/W) 45 40 SW C3 0.47µF 35 C3 0.47µF BOOST LT1769 D2 30 HIGH DUTY CYCLE CONNECTION D2 SPIN SENSE VX 3V TO 6V BAT SENSE BAT CX 10µF 4-LAYER BOARD 20 VBAT MEASURED FROM AIR AMBIENT TO DIE USING COPPER LANDS AS SHOWN ON DATA SHEET 15 BOOST LT1769 SPIN 2-LAYER BOARD 25 SW VBAT + + 10 0 5 20 15 25 10 BOARD AREA (IN2) 30 35 1769 F10 1769 F08 Figure 10. High Duty Cycle Figure 8. LT1769 Thermal Resistance HIGH DUTY CYCLE CONNECTION VIN LEAD TEMPERATURE ON PINS 1, 2, 3 (°C) 70 60 VIN = 19V VBAT = 12.3V VBOOST = 5V 2-LAYER BOARD ROOM TEMP = 24°C 50 Q2 5 IN2 BOARD RX 50k D1 C2 0.47µF D2 SW VCC BOOST LT1769 SPIN 40 25 IN2 BOARD Q1 = Si4435DY Q2 = TP0610L 30 20 + Q1 NOTE: PEAK DIE TEMPERATURE WILL BE ABOUT 15°C HIGHER AT 2A CHARGE CURRENT VX 3V TO 6V SENSE BAT CX 10µF + 0 1 0.5 1.5 CHARGE CURRENT (A) VBAT 2 1769 F09 1769 F11 Figure 9. LT1769 Lead Temperature Battery voltage and input voltage will affect device power dissipation, so the data sheet power calculations must be used to extrapolate these readings to other situations. Vias should be used to connect board layers together. Planes under the charger area can be cut away from the rest of the board and connected with vias to form both a low thermal resistance system and to act as a ground plane for reduced EMI. Glue-on, chip-mounted heat sinks are effective only in moderate power applications where the PC board copper cannot be used, or where the board size is small. They offer very little improvement in a properly laid out multilayer board of reasonable size. 14 Figure 11. Replacing the Input Diode Higher Duty Cycle for the LT1769 Battery Charger Maximum duty cycle for the LT1769 is typically 90%, but this may be too low for some applications. For example, if an 18V ±3% adapter is used to charge ten NiMH cells, the charger must put out approximaly 15V. A total of 1.6V is lost in the input diode, switch resistance, inductor resistance and parasitics, so the required duty cycle is 15/16.4 = 91.4%. The duty cycle can be extended to 93% by restricting boost voltage to 5V instead of using VBAT as is normally done. This lower boost voltage also reduces power dissipation in the LT1769, so it is a win-win decision. Connect an external source of 3V to 6V at VX node in Figure 10 with a 10µF CX bypass capacitor. LT1769 U U W U APPLICATIONS INFORMATION Lower Dropout Voltage For even lower dropout and/or reducing heat on the board, the input diode D3 can be replaced with a FET (see Figure 11). Connect a P-channel FET in place of the input diode with its gate connected to the battery causing the FET to turn off when the input voltage goes low. The problem is that the gate must be pumped low so that the FET is fully turned on even when the input is only a volt or two above the battery voltage. Also there is a turn-off speed issue. The FET should turn off instantly when the input is dead shorted to avoid large current surges from the battery back through the charger into the FET. Gate capacitance slows turn-off, so a small P-channel (Q2) is added to discharge the gate capacitance quickly in the event of an input short. The Q2 body diode creates the necessary pumping action to keep the gate of Q1 low during normal operation. Note that Q1 and Q2 have a VGS spec limit of 20V. This restricts VIN to a maximum of 20V. For low dropout operation with VIN > 20V consult factory. Optional Diode Connections The typical application in Figure 1 shows a single diode (D3) to isolate the VCC pin from the adaptor input and to block reverse input voltage (both steady state and transient). This simple connection may be unacceptable in situations where the system load must be powered from the battery when the adapter input power is removed. As shown in Figure 12, a parasitic diode exists from the SW R7 500Ω + CLN INTERNAL PARASITIC DIODE RS1 SWITCH NODE L1 VBAT RS4 TO SYSTEM LOAD + CIN Switch rise and fall times are under 10ns for maximum efficiency. To minimize radiation, the catch diode, SW pin and input bypass capacitor leads should be kept as short as possible. A ground plane should be used under the switching circuitry to prevent interplane coupling and to act as a thermal spreading path. All ground pins should be connected to expanded traces for low thermal resistance. The fast-switching high current ground path, including the switch, catch diode and input capacitor, should be kept very short. Catch diode and input capacitor should be close to the chip and terminated to the same point. This path contains nanosecond rise and fall times with several amps of current. The other paths contain only DC and/or 200kHz tri-wave and are less critical. Figure 13 indicates the high speed, high current switching path. Figure 14 shows critical path layout. Contact Linear Technology for the LT1769 circuit PCB layout or Gerber file. ADAPTER IN VCC SW L1 C1 1µF Layout Considerations D3 CLP LT1769 pin to the VCC pin in the LT1769. When the input power is removed, this diode will become forward biased and will provide a current path from the battery to the system load. Because of diode power limitations, it is not recommended to power the system load through the internal parasitic diode. To safely power the system load from the battery, an additional Schottky diode (D4) is needed. For minimum losses, D4 could be replaced by a low RDS(ON) MOSFET which is turned on when the adapter power is removed. D4 VIN CIN HIGH FREQUENCY CIRCULATING PATH D1 COUT BAT + 1769 F13 1769 F12a Figure 12. Modified Diode Connection Figure 13. High Speed Switching Path Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 15 LT1769 U U W U APPLICATIONS INFORMATION GND CIN D1 L1 TO GND GND GND GND SW BOOST UV GND GND OVP CLP CLN COMP1 SENSE GND GND GND GND VCC1 VCC2 VCC3 GND PROG VC UVOUT COMP2 BAT SPIN GND TO GND RS1 COUT GND NOTE: CONNECT ALL GND PINS TO EXPANDED PC LANDS FOR PROPER HEAT SINKING 1769 F14 Figure 14. Critical Electrical and Thermal Path Layout U PACKAGE DESCRIPTION Dimensions in inches (millimeters) unless otherwise noted. GN Package 28-Lead Plastic SSOP (Narrow 0.150) (LTC DWG # 05-08-1641) 0.015 ± 0.004 × 45° (0.38 ± 0.10) 0.0075 – 0.0098 (0.191 – 0.249) 0.053 – 0.069 (1.351 – 1.748) 0.386 – 0.393* (9.804 – 9.982) 0.004 – 0.009 (0.102 – 0.249) 28 27 26 25 24 23 22 21 20 19 18 17 1615 0.033 (0.838) REF 0° – 8° TYP 0.016 – 0.050 (0.406 – 1.270) 0.008 – 0.012 (0.203 – 0.305) 0.229 – 0.244 (5.817 – 6.198) 0.0250 (0.635) BSC * DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 0.150 – 0.157** (3.810 – 3.988) GN28 (SSOP) 1098 1 2 3 4 5 6 7 8 9 10 11 12 13 14 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC®1325 Microprocessor-Controlled Battery Management System Can Charge, Discharge and Gas Gauge NiCd and Lead-Acid Batteries with Software Charging Profiles LT1372/LT1377 500kHz/1MHz Step-Up Switching Regulators High Frequency, Small Inductor, High Efficiency Switchers, 1.5A Switch LT1376 500kHz Step-Down Switching Regulator High Frequency, Small Inductor, High Efficiency Switcher, 1.5A Switch LT1505 High Current, High Efficiency Battery Charger 94% Efficiency, Synchronous Current Mode PWM LT1510 Constant-Voltage/Constant-Current Battery Charger Up to 1.5A Charge Current for Lithium-Ion, NiCd and NiMH Batteries LT1511 Constant-Voltage/Constant-Current Battery Charger Up to 3A Charge Current for Lithium-Ion, NiCd and NiMH Batteries LT1512/LT1513 SEPIC Battery Chargers VIN Can Be Higher or Lower Than Battery Voltage LTC1729 Li-Ion Battery Charger Termination Controller Preconditioning If Cell < 2.7V, 3hr Time-Out, C/10 Detection, Temp Sensor Pin, Charger and Battery Detection LTC1759 SMBus Smart Battery Charger 94% Efficiency with Input Current Limiting, Up to 8A ICHG 16 Linear Technology Corporation 1769f LT/TP 0999 4K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 1999