LINER LTC4090-5

LTC4090/LTC4090-5
USB Power Manager with
2A High Voltage Bat-Track
Buck Regulator
DESCRIPTION
FEATURES
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The LTC®4090/LTC4090-5 are USB power managers plus
high voltage Li-Ion/Polymer battery chargers. The devices
control the total current used by the USB peripheral for
operation and battery charging. Battery charge current is
automatically reduced such that the sum of the load current
and the charge current does not exceed the programmed
input current limit. The LTC4090/LTC4090-5 also accommodate high voltage power supplies, such as 12V AC/DC
wall adapters, FireWire, or automotive power.
Seamless Transition Between Power Sources: LiIon Battery, USB, and 6V to 36V Supply (60V Max)
2A Output High Voltage Buck Regulator with BatTrackTM Adaptive Output Control (LTC4090)
Internal 215mΩ Ideal Diode Plus Optional External
Ideal Diode Controller Provides Low Loss Power
Path When External Supply / USB Not Present
Load Dependent Charging from USB Input Guarantees Current Compliance
Full Featured Li-Ion Battery Charger
1.5A Maximum Charge Current with Thermal Limiting
NTC Thermistor Input for Temperature Qualified
Charging
Tiny (3mm × 6mm × 0.75mm) 22-Pin DFN Package
The LTC4090 provides a Bat-Track adaptive output that
tracks the battery voltage for high efficiency charging from
the high voltage input. The LTC4090-5 provides a fixed 5V
output from the high voltage input to charge single cell
Li-Ion bateries. The charge current is programmable and an
end-of-charge status output (⎯C⎯H⎯R⎯G) indicates full charge.
Also featured are programmable total charge time, an NTC
thermistor input used to monitor battery temperature while
charging and automatic recharging of the battery.
APPLICATIONS
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HDD-Based Media Players
Personal Navigation Devices
Other USB-Based Handheld Products
Automotive Accessories
, LT, LTC and LTM are registered trademarks of Linear Technology Corporation.
Bat-Track is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
TYPICAL APPLICATION
0.47μF
HIGH (6V-36V)
VOLTAGE INPUT
HVIN
LTC4090/LTC4090-5 High Voltage
Battery Charger Efficiency
6.8μH
SW
BOOST
22μF
1μF
90
HVOUT
5V WALL
ADAPTER
IN
LTC4090
OUT
VC
LOAD
4.7μF
TIMER
270pF
0.1μF
BAT
CLPROG
RT
59k
40.2k
2k
GND
PROG
100k
+
Li-Ion BATTERY
70
LTC4090-5
60
50
40
HVIN = 8V
HVIN = 12V
HVIN = 24V
HVIN = 36V
30
VOUT (TYP)
VBAT + 0.3V
5V
5V
VBAT
LTC4090
1k
EFFICIENCY (%)
4.7μF
USB
FIGURE 12 SCHEMATIC
WITH RPROG = 52k
80 NO OUTPUT LOAD
HVPR
AVAILABLE INPUT
HV INPUT (LTC4090)
HV INPUT (LTC4090-5)
USB ONLY
BAT ONLY
20
2.0
4090 TAO1
2.5
3.5
3.0
VBAT (V)
4.0
4.5
4090 TA01b
4090fa
1
LTC4090/LTC4090-5
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Notes 1, 2, 3, 4)
HVIN, HVEN (Note 9) ................................................60V
BOOST ......................................................................56V
BOOST above SW .....................................................30V
PG, SYNC ..................................................................30V
IN, OUT, HVOUT
t < 1ms and Duty Cycle < 1% .................. –0.3V to 7V
Steady State............................................. –0.3V to 6V
BAT, HPWR, SUSP, VC, ⎯C⎯H⎯R⎯G, ⎯H⎯V⎯P⎯R ........... –0.3V to 6V
NTC, TIMER, PROG, CLPROG ..........–0.3V to VCC + 0.3V
IIN, IOUT, IBAT (Note 5) ..............................................2.5A
Operating Temperature Range ..................... –40 to 85°C
Junction Temperature ........................................... 110°C
Storage Temperature Range....................... –65 to 125°C
TOP VIEW
SYNC
1
22 HVEN
PG
2
21 HVIN
RT
3
20 SW
19 BOOST
VC
4
NTC
5
VNTC
6
18 HVOUT
23
17 TIMER
HVPR
7
16 SUSP
CHRG
8
15 HPWR
PROG
9
14 CLPROG
GATE 10
13 OUT
BAT 11
12 IN
DJC PACKAGE
22-LEAD (6mm × 3mm) PLASTIC DFN
TJMAX = 110°C, θJA = 47°C/W
EXPOSED PAD (PIN 23) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC4090EDJC#PBF
LTC4090EDJC#TRPBF
4090
22-Lead (6mm × 3mm) Plastic DFN
–40°C to 85°C
LTC4090EDJC-5#PBF
LTC4090EDJC-5#TRPBF
40905
22-Lead (6mm × 3mm) Plastic DFN
–40°C to 85°C
Consult LTC Marketing for parts specified with wider operating temperature ranges.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. HVIN = HVEN = 12V, BOOST = 17V, VIN = HPWR = 5V, VBAT = 3.7V,
RPROG = 100k, RCLPROG = 2k and SUSP = 0V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
USB Input Current Limit
USB Input Supply Voltage
●
IIN
Input Bias Current
IBAT = 0 (Note 6)
Suspend Mode; SUSP = 5V
●
●
ILIM
Current Limit
HPWR = 5V
HPWR = 0V
●
●
IIN(MAX)
Maximum Input Current Limit
(Note 7)
VIN
RON
On-Resistance VIN to VOUT
IOUT = 80mA
VCLPROG
CLPROG Servo Voltage in Current Limit
RCLPROG = 2k
RCLPROG = 1k
ISS
Soft-Start Inrush Current
4.35
475
90
5.5
V
0.5
50
1
100
mA
μA
500
100
525
110
mA
mA
2.4
A
Ω
0.215
●
●
0.98
0.98
1.00
1.00
10
1.02
1.02
V
V
mA/μs
4090fa
2
LTC4090/LTC4090-5
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. HVIN = HVEN = 12V, BOOST = 17V, VIN = HPWR = 5V, VBAT = 3.7V,
RPROG = 100k, RCLPROG = 2k and SUSP = 0V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
VCLEN
Input Current Limit Enable Threshold
Voltage (VIN - VOUT)
(VIN - VOUT) Rising
(VIN - VOUT) Falling
20
–80
50
–50
80
–20
mV
mV
VUVLO
Input Undervoltage Lockout
VIN Rising
3.6
3.8
4
ΔVUVLO
Input Undervoltage Lockout Hysteresis
VIN Rising – VIN Falling
●
130
V
mV
High Voltage Regulator
HVIN Supply Voltage
●
6
VOVLO
HVIN Overvoltage Lockout Threshold
●
36
IHVIN
HVIN Bias Current
Shutdown; HVEN = 0.2V
Not Switching, HVOUT = 3.6V
VOUT
Output Voltage with HVIN Present
Assumes HVOUT to OUT Connection,
0 ≤ VBAT ≤ 4.2V (LTC4090)
VOUT
Output Voltage with HVIN Present
fSW
Switching Frequency
tOFF
Minimum Switch Off-Time
ISW(MAX)
Switch Current Limit
Duty Cycle = 5%
VSAT
Switch VCESAT
ISW = 2A
SW = 10V, HVOUT = 0V
VHVIN
IR
Boost Schottky Reverse Leakage
VB(MIN)
Minimum Boost Voltage (Note 8)
IBST
BOOST Pin Current
60
V
41.5
45
V
0.01
130
0.5
200
μA
μA
3.45
VBAT + 0.3
4.6
V
Assumes HVOUT to OUT Connection
(LTC4090-5)
4.85
5
5.15
V
RT = 8.66k
RT = 29.4k
RT = 187k
2.1
0.9
160
2.4
1.0
200
2.7
1.15
240
MHz
MHz
kHz
60
150
ns
3.5
4.0
A
●
●
3.0
500
mV
0.02
2
μA
1.5
2.1
V
22
35
mA
15
22
60
27
35
100
μA
μA
μA
4.165
4.158
4.200
4.200
4.235
4.242
V
V
465
900
500
1000
535
1080
mA
mA
●
ISW = 1A
Battery Management
IBAT
Battery Drain Current
VBAT = 4.3V, Charging Stopped
Suspend Mode, SUSP = 5V
VIN = 0V, BAT Powers OUT, No Load
VFLOAT
VBAT Regulated Output Voltage
IBAT = 2mA
IBAT = 2mA; 0 ≤ TA ≤ 85°C
ICHG
Constant-Current Mode Charge Current, RPROG = 100k
No Load
RPROG = 50k, 0 ≤ TA ≤ 85°C
ICHG(MAX)
Maximum Charge Current
●
●
●
●
1.5
VPROG
PROG Pin Servo Voltage
RPROG = 100k
RPROG = 50k
●
●
kEOC
Ratio of End-of-Charge Indication
Current to Charge Current
VBAT = VFLOAT (4.2V)
●
ITRKL
Trickle Charge Current
BAT = 2V
VTRKL
Trickle Charge Threshold Voltage
BAT Rising
VCEN
Charge Enable Threshold Voltage
(VOUT – VBAT) Falling; VBAT = 4V
(VOUT – VBAT) Rising; VBAT = 4V
ΔVRECHRG
Recharge Battery Threshold Voltage
Threshold Voltage Relative to VFLOAT
●
A
0.98
0.98
1.00
1.00
1.02
1.02
V
V
0.085
0.1
0.11
mA/mA
35
50
60
2.75
2.9
3.0
55
80
●
–65
–100
mA
V
mV
mV
–135
mV
4090fa
3
LTC4090/LTC4090-5
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. HVIN = HVEN = 12V, BOOST = 17V, VIN = HPWR = 5V, VBAT = 3.7V,
RPROG = 100k, RCLPROG = 2k and SUSP = 0V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
tTIMER
TIMER Accuracy
VBAT = 4.3V
–10
Recharge Time
Percent of Total Charge Time
50
%
Low Battery Trickle Charge Time
Percent of Total Charge Time,
VBAT <2.9V
25
%
105
°C
TLIM
Junction Temperature in Constant
Temperature Mode
TYP
MAX
10
UNITS
%
Internal Ideal Diode
RFWD
Incremental Resistance, VON Regulation IOUT = 100mA
125
mΩ
RDIO, ON
On-Resistance VBAT to VOUT
IOUT = 600mA
215
mΩ
VFWD
Voltage Forward Drop (VBAT – VOUT)
IOUT = 5mA
IOUT = 100mA
IOUT = 600mA
●
10
30
55
160
50
mV
mV
mV
VOFF
Diode Disable Battery Voltage
2.7
V
IFWD
Load Current Limit for VON Regulation
550
mA
ID(MAX)
Diode Current Limit
2.2
A
20
mV
External Ideal Diode
VFWD, EXT
External Diode Forward Voltage
Logic (⎯C⎯H⎯R⎯G, ⎯H⎯V⎯P⎯R, TIMER, SUSP, HPWR, HVEN, PG, SYNC)
●
0.14
VTIMER = 0V
●
5
Output Low Voltage
(⎯C⎯H⎯R⎯G, ⎯H⎯V⎯P⎯R); ISINK = 5mA
●
VIH
Input High Voltage
SUSP, HPWR
VIL
Input Low Voltage
SUSP, HPWR
VHVEN, H
HVEN High Threshold
VHVEN, L
HVEN Low Threshold
IPULLDN
Logic Input Pull-Down Current
SUSP, HPWR
2
IHVEN
HVEN Pin Bias Current
HVEN = 2.5V
5
VPG
PG Threshold
HVOUT Rising
2.8
V
ΔVPG
PG Hysteresis
35
mV
IPGLK
PG Leakage
VCHG, SD
Charger Shutdown Threshold Voltage
on TIMER
ICHG, SD
Charger Shutdown Pull-Up Current on
TIMER
VOL
IPG
PG Sink Current
VSYNC, L
SYNC Low Threshold
VSYNC, H
SYNC High Threshold
ISYNC
SYNC Pin Bias Current
0.4
14
0.1
μA
0.4
1.2
2.3
PG = 0.4V
100
0.3
V
10
μA
μA
1
900
μA
μA
0.5
V
0.8
VSYNC = 0V
V
V
0.1
●
V
V
0.4
PG = 5V
V
0.1
V
μA
4090fa
4
LTC4090/LTC4090-5
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. HVIN = HVEN = 12V, BOOST = 17V, VIN = HPWR = 5V, VBAT = 3.7V,
RPROG = 100k, RCLPROG = 2k and SUSP = 0V, unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
1.4
2.5
3.5
4.4
4.85
UNITS
IVNTC
VNTC Pin Current
VNTC = 2.5V
●
VVNTC
VNTC Bias Voltage
IVNTC = 500μA
●
INTC
NTC Input Leakage Current
NTC = 1V
VCOLD
Cold Temperature Fault Threshold
Voltage
Rising NTC Voltage
Hysteresis
0.738 • VNTC
0.02 • VNTC
V
V
VHOT
Hot Temperature Fault Threshold
Voltage
Falling NTC Voltage
Hysteresis
0.29 • VNTC
0.01 • VNTC
V
V
VDIS
NTC Disable Threshold Voltage
Falling NTC Voltage
Hysteresis
NTC
V
0
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC4090/LTC4090-5 are guaranteed to meet performance
specifications from 0°C to 85°C. Specifications over the –40°C to 85°C
operating temperature range are assured by design, characterization and
correlation with statistical process controls.
Note 3: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperatures will exceed 110°C when overtemperature protection is
active. Continuous operation above the specified maximum operating
junction temperature may result in device degradation or failure.
●
75
mA
±1
100
35
μA
125
mV
mV
Note 4: VCC is the greater of VIN, VOUT, and VBAT
Note 5: Guaranteed by long term current density limitations.
Note 6: Total input current is equal to this specification plus 1.002 • IBAT
where IBAT is the charge current.
Note 7: Accuracy of programmed current may degrade for currents
greater than 1.5A.
Note 8: This is the minimum voltage across the boost capacitor needed to
guarantee full saturation of the switch.
Note 9: Absolute Maximum Voltage at HVIN and HVEN pins is for nonrepetative 1 second transients; 40V for continuous operation.
TYPICAL PERFORMANCE CHARACTERISTICS
Battery Regulation (Float)
Voltage vs Temperature
VFLOAT Load Regulation
4.30
4.220
RPROG = 34k
4.215
4.25
Battery Current and Voltage vs
Time (LTC4090)
5
VIN = 5V
IBAT = 2mA
4
VFLOAT (V)
VFLOAT (V)
4.10
4.205
4.200
4.195
4.190
3
2
1250mAh
CELL
HVIN = 12V
RPROG = 50k
4.185
4.00
0
200
400
600
IBAT (mA)
800
1000
4090 G01
4.180
–50
–25
0
50
25
TEMPERATURE (°C)
75
100
0
900
600
C/10
1
4.05
1200
VBAT
VOUT
VCHRGB
IBAT
0
50
IBAT (mA)
4.15
VBAT, VOUT, VCHRGB (V)
4.210
4.20
1500
TERMINATION 300
100
150
0
200
TIME (MIN)
4090 G02
4090 G03
4090fa
5
LTC4090/LTC4090-5
TYPICAL PERFORMANCE CHARACTERISTICS
Charging from USB, IBAT vs VBAT
600
VIN = 5V
VOUT = NO LOAD
500 RPROG = 100k
RCLPROG = 2k
1000
500
800
VBAT = 3.7V
900 VIN = 0V
700
300
200
HPWR = 0V
100
0
0
0.5
1
1.5
2 2.5
VBAT (V)
3
3.5
4
4.5
IOUT (mA)
400
IBAT (mA)
IBAT (mA)
600
HPWR = 5V
400
300
300
200
0
100
2000
–50°C
0°C
50°C
100°C
60
40
VFWD (mV)
80
90
90
85
85
80
75
70
55
50
100
0
0.2
IOUT (A)
2.4
2.2
MINIMUM
2.0
1.8
5
10
15
20
25
HVIN (V)
30
35
4090 G09
0.6
0.4
IOUT (A)
0.8
HVIN = 8V
HVIN = 12V
HVIN = 24V
HVIN = 36V
60
55
50
1.0
0
0.2
0.6
0.4
IOUT (A)
0.8
1.0
4090 G29
High Voltage Regulator Switch
Voltage Drop
140
700
120
600
VOLTAGE DROP (mV)
MINIMUM SWITCH ON TIME (ns)
TYPICAL
70
High Voltage Regulator Minimum
Switch On-Time vs Temperature
High Voltage Regulator Maximum
Load Current
2.6
75
4090 G08
4090 G07
2.8
80
65
HVIN = 8V
HVIN = 12V
HVIN = 24V
HVIN = 36V
60
FIGURE 12 SCHEMATIC
VBAT = 4.21V (IBAT = 0)
200
150
FIGURE 12 SCHEMATIC
95 VBAT = 4.21V (IBAT = 0)
65
1500
20
100
VFWD (mV)
LTC4090-5 High Voltage Regulator
Efficiency vs Output Load
EFFICIENCY (%)
EFFICIENCY (%)
IOUT (mA)
2500
50
4090 G06
100
3500
3.0
0
125
100
FIGURE 12 SCHEMATIC
95 VBAT = 4.21V (IBAT = 0)
3000
–50°C
0°C
50°C
100°C
100
LTC4090 High Voltage Regulator
Efficiency vs Output Load
VBAT = 3.7V
4500 VIN = 0V
Si2333 PFET
4000
0
400
4090 G05
5000
500
500
RPROG = 2.1k
100 VIN = 5V
VBAT = 3.5V
θJA = 40°C/W
0
50
25
75
–50 –25
0
TEMPERATURE (°C)
4090 G04
1000
600
200
Ideal Diode Current vs Forward
Voltage and Temperature with
External Device
0
Ideal Diode Current vs Forward
Voltage and Temperature (No
External Device)
Charge Current vs Temperature
(Thermal Regulation)
100
80
60
40
20
0
–50 –25
500
400
300
200
100
0
25 50 75 100 125 150
TEMPERATURE (˚C)
4090 G10
0
0
500
1500
1000
2000
SWITCH CURRENT (mA)
2500
4090 G11
4090fa
6
LTC4090/LTC4090-5
TYPICAL PERFORMANCE CHARACTERISTICS
High Voltage Regulator
Frequency Foldback
High Voltage Regulator Switch
Frequency
High Voltage Regulator Soft-Start
1000
1100
4.0
900
900
800
700
600
3.5
800
SWITCH CURRENT LIMIT (A)
SWITCHING FREQUENCY (kHz)
FREQUENCY (kHz)
1000
700
600
500
400
300
200
0
0
0
25 50 75 100 125 150
TEMPERATURE (°C)
2
3
HVOUT (V)
1
2.5
2.0
DUTY CYCLE = 10 %
6.5
3.5
100
3.0
2.0
5.0
3.5
3.0
0
25 50 75 100 125 150
TEMPERATURE (°C)
1
2.90
HVOUT THRESHOLD VOLTAGE (V)
1.2
2.00
CURRENT LIMIT CLAMP
VC VOLTAGE (V)
1.0
1.50
1.00
SWITCHING THRESHOLD
0.4
0.50
0.2
0
–50 –25
1000
High Voltage Regulator Power
Good Threshold
2.50
4090 G18
10
100
LOAD CURRENT (mA)
4090 G17
High Voltage Regulator VC
Voltages
2.0
TO RUN
4.0
1.0
4090 G16
1.4
0.5
1.0
1.5
BOOST DIODE CURRENT (A)
TO START
4.5
1.5
High Voltage Regulator Boost
Diode VF vs IF
0
3.5
5.5
DUTY CYCLE = 90 %
2.5
4090 G15
0
3
6.0
0
–50 –25
1.0
0.6
1
2
2.5
1.5
RUN/SS PIN VOLTAGE (V)
High Voltage Regulator Minimum
Input Voltage
0.5
0.8
0.5
4090 G14
HVIN (V)
SWITCH CURRENT LIMIT (A)
SWITCH CURRENT LIMIT(A)
3.0
1.5
BOOST DIODE Vf (V)
0
7.0
4.0
3.5
80
1.0
0
4
4.5
4.0
60
40
DUTY CYCLE (%)
1.5
High Voltage Regulator Switch
Current Limit
High Voltage Regulator Switch
Current Limit
20
2.0
4090 G13
4090 G12
0
2.5
0.5
100
500
–50 –25
3.0
0
25 50 75 100 125 150
TEMPERATURE (°C)
4090 G19
2.85
2.80
2.75
2.70
2.65
–50 –25
0
25 50
75 100 125 150
TEMPERATURE (°C)
4090 G20
4090fa
7
LTC4090/LTC4090-5
TYPICAL PERFORMANCE CHARACTERISTICS
LTC4090 Input Connect
Waveforms
LTC4090 Input Disconnect
Waveforms
LTC4090 Response to Suspend
VIN
5V/DIV
VIN
5V/DIV
SUSP
5V/DIV
VOUT
5V/DIV
IIN
0.5A/DIV
VOUT
5V/DIV
IIN
0.5A/DIV
VOUT
5V/DIV
IIN
0.5A/DIV
IBAT
0.5A/DIV
IBAT
0.5A/DIV
IBAT
0.5A/DIV
1ms/DIV
VBAT = 3.85V
IOUT = 100mA
1ms/DIV
4090 G21
LTC4090 High Voltage Input
Connect Waveforms
VHVIN
5V/DIV
VOUT
5V/DIV
IHVIN
1A/DIV
VOUT
5V/DIV
IHVIN
1A/DIV
IBAT
1A/DIV
IBAT
1A/DIV
2ms/DIV
4090 G24
LTC4090 Response to HPWR
HPWR
IIN
0.5A/DIV
IBAT
0.5A/DIV
2ms/DIV
4090 G25
VBAT = 3.85V
IOUT = 100mA
100μs/DIV
VBAT = 3.85V
IOUT = 50mA
4090 G26
LTC4090 High Voltage Regulator
Load Transient
HVOUT
50mV/DIV
HVOUT
50mV/DIV
IOUT
1A/DIV
IL
1A/DIV
25μs/DIV
4090 G23
5V/DIV
LTC4090 High Voltage Regulator
Load Transient
ILOAD = 500mA
VBAT = 3.85V
IOUT = 50mA
LTC4090 High Voltage Input
Disconnect Waveforms
VHVIN
10V/DIV
VBAT = 3.85V
IOUT = 100mA
1ms/DIV
4090 G22
VBAT = 3.85V
IOUT = 100mA
4090 G27
ILOAD = 500mA
25μs/DIV
4090 G28
4090fa
8
LTC4090/LTC4090-5
PIN FUNCTIONS
SYNC (Pin 1): External Clock Synchronization Input. See
synchronizing section in the Applications Information
section. Ground pin when not used.
PG (Pin 2): The PG pin is the open collector output of an
internal comparator. PG remains low until the HVOUT pin
is above 2.8V. PG output is valid when HVIN is above 3.6V
and HVEN is high.
RT (Pin 3): Oscillator Resistor Input. Connecting a resistor
to ground from this pin sets the switching frequency.
VC (Pin 4): High Voltage Buck Regulator Control Pin. The
voltage on this pin controls the peak switch current in the
high voltage regulator. Tie an RC network from this pin to
ground to compensate the control loop.
NTC (Pin 5): Input to the NTC Thermistor Monitoring
Circuits. The NTC pin connects to a negative temperature
coefficient thermistor which is typically co-packaged with
the battery pack to determine if the battery is too hot or too
cold to charge. If the battery temperature is out of range,
charging is paused until the battery temperature re-enters
the valid range. A low drift bias resistor is required from
VNTC to NTC and a thermistor is required from NTC to
ground. If the NTC function is not desired, the NTC pin
should be grounded.
VNTC (Pin 6): Output Bias Voltage for NTC. A resistor from
this pin to the NTC pin will bias the NTC thermistor.
⎯ ⎯V⎯P⎯R (Pin 7): High Voltage Present Output (Active Low).
H
A low on this pin indicates that the high voltage regulator
has sufficient voltage to charge the battery. This feature
is enabled if power is present on HVIN, IN, or BAT (i.e.,
above UVLO thresholds).
⎯ H
⎯ R
⎯ G
⎯ (Pin 8): Open-Drain Charge Status Output. When the
C
battery is being charged, the ⎯C⎯H⎯R⎯G pin is pulled low by an
internal N-channel MOSFET. When the timer runs out or
the charge current drops below 10% of the programmed
charge current or the input supply is removed, the ⎯C⎯H⎯R⎯G
pin is forced to a high impedance state.
PROG (Pin 9): Charge Current Program Pin. Connecting
a resistor from PROG to ground programs the charge
current:
ICHG( A) =
50, 000 V
RPROG
GATE (Pin 10): External Ideal Diode Gate Connection. This
pin controls the gate of an optional external P-channel
MOSFET transistor used to supplement the internal ideal
diode. The source of the P-channel MOSFET should be
connected to OUT and the drain should be connected to
BAT. When not in use, this pin should be left floating. It
is important to maintain high impedance on this pin and
minimize all leakage paths.
BAT (Pin 11): Single-Cell Li-Ion Battery. This pin is used
as an output when charging the battery and as an input
when supplying power to OUT. When the OUT pin potential
drops below the BAT pin potential, an ideal diode function
connects BAT to OUT and prevents OUT from dropping
more than 100mV below BAT. A precision internal resistor
divider sets the final float (charging) potential on this pin.
The internal resistor divider is disconnected when IN and
HVIN are in undervoltage lockout.
IN (Pin 12): Input Supply. Connect to USB supply, VBUS.
Input current to this pin is limited to either 20% or 100%
of the current programmed by the CLPROG pin as determined by the state of the HPWR pin. Charge current (to the
BAT pin) supplied through the input is set to the current
programmed by the PROG pin but will be limited by the
input current limit if charge current is set greater than the
input current limit or if the sum of charge current plus load
current is greater than the input current limit.
OUT (Pin 13): Voltage Output. This pin is used to provide
controlled power to a USB device from either USB VBUS
(IN), an external high voltage supply (HVIN), or the battery
(BAT) when no other supply is present. The high voltage
supply is prioritized over the USB VBUS input. OUT should
be bypassed with at least 4.7μF to GND.
4090fa
9
LTC4090/LTC4090-5
PIN FUNCTIONS
CLPROG (Pin 14): Current Limit Program and Input Current Monitor. Connecting a resistor, RCLPROG, to ground
programs the input to output current limit. The current
limit is programmed as follows:
1000 V
ICL ( A) =
RCLPROG
In USB applications, the resistor RCLPROG should be set
to no less than 2.1k. The voltage on the CLPROG pin is
always proportional to the current flowing through the
IN to OUT power path. This current can be calculated as
follows:
IIN( A) =
VCLPROG
• 1000
RCLPROG
HPWR (Pin 15): High Power Select. This logic input is used
to control the input current limit. A voltage greater than
1.2V on the pin will set the input current limit to 100% of
the current programmed by the CLPROG pin. A voltage
less than 0.4V on the pin will set the input current limit to
20% of the current programmed by the CLPROG pin. A
2μA pull-down current is internally connected to this pin
to ensure it is low at power up when the pin is not being
driven externally.
SUSP (Pin 16): Suspend Mode Input. Pulling this pin
above 1.2V will disable the power path from IN to OUT.
The supply current from IN will be reduced to comply
with the USB specification for suspend mode. Both the
ability to charge the battery from HVIN and the ideal diode
function (from BAT to OUT) will remain active. Suspend
mode will reset the charge timer if OUT is less than BAT
while in suspend mode. If OUT is kept greater than BAT,
such as when the high voltage input is present, the charge
timer will not be reset when the part is put in suspend. A
2μA pull-down current is internally connected to this pin
to ensure it is low at power up when the pin is not being
driven externally.
TIMER (Pin 17): Timer Capacitor. Placing a capacitor,
CTIMER, to GND sets the timer period. The timer period
is:
tTIMER (hours) =
CTIMER • RPROG • 3hours
0.1µF • 100k
Charge time is increased if charge current is reduced
due to load current, thermal regulation and current limit
selection (HPWR low).
Shorting the TIMER pin to GND disables the battery
charging functions.
HVOUT (Pin 18): Voltage Output of the High Voltage
Regulator. When sufficient voltage is present at HVOUT,
the low voltage power path from IN to OUT will be disconnected and the ⎯H⎯V⎯P⎯R pin will be pulled low to indicate
that a high voltage wall adapter has been detected. The
LTC4090 high voltage regulator will maintain just enough
differential voltage between HVOUT and BAT to keep the
battery charger MOSFET out of dropout (typically 300mV
from OUT to BAT). The LTC4090-5 high voltage regulator
will provide a 5V output to the battery charger MOSFET.
HVOUT should be bypassed with at least 22μF to GND.
BOOST (Pin 19): This pin is used to provide drive voltage,
higher than the input voltage, to the internal bipolar NPN
power switch.
SW (Pin 20): The SW pin is the output of the internal high
voltage power switch. Connect this pin to the inductor,
catch diode and boost capacitor.
HVIN (Pin 21): High Voltage Regulator Input. The HVIN pin
supplies current to the internal high voltage regulation and
to the internal high voltage power switch. The presence
of a high voltage input takes priority over the USB VBUS
input (i.e., when a high voltage input supply is detected,
the USB IN to OUT path is disconnected). This pin must
be locally bypassed.
HVEN (Pin 22): High Voltage Regulator Enable Input. The
HVEN pin is used to disable the high voltage input path.
Tie to ground to disable the high voltage input or tie to at
least 2.3V to enable the high voltage path. If this feature
is not used, tie HVEN to the HVIN pin. This pin can also
be used to soft-start the high voltage regulator; see the
Applications Information section for more information.
Exposed Pad (Pin 23): Ground. The exposed package pad
is ground and must be soldered to the PC board for proper
functionality and for maximum heat transfer (use several
vias directly under the LTC4090/LTC4090-5).
4090fa
10
LTC4090/LTC4090-5
BLOCK DIAGRAM
C2
BOOST
10
HVIN
INTERNAL
REFERENCE
10
D1
+
HVEN
+
IL
–
SOFT-START
R
S
–
10
RT1
RT
Q
Q
DRIVER
OSCILLATOR
200kHz - 2.4MHz
SYNC
10
HVOUT
–
VC
GM
CF
C1
VSET
3.6V (LTC4090)
5V (LTC4090-5)
+
+
–
VC CLAMP
CC
+
–
PG
22
+
2.8V
IN
IIN
1000
CLPROG
HVPR
4.25V (RISING)
3.15V (FALLING)
–
19
CURRENT LIMIT
1V
+
SOFT-START
ILIM
CURRENT
CONTROL
CL
–
DIE
TEMP
RCLPROG
13
+
+
–
10
+
–
75mV (RISING)
25mV (FALLING)
IN
ILIM CNTL
OUT
ENABLE
CC/CV REGULATOR
CHARGER
ENABLE
105°C
HPWR 500mA/100mA
+
–
2μA
30mV
20mV
IN OUT BAT
–
+
BAT
21
21
+
SOFT-START2
1V
EDA
BAT
ICHG
CHARGE CONTROL
21
GATE
IDEAL
DIODE
TA
+
OUT
+
–
10
350mV
(LTC4090)
+
–
10
RC
L1
SW
Q1
–
0.25V
+
2.9V
BATTERY
UVLO
CHG
–
23
PROG
–
RPROG
VOLTAGE DETECT
15
VNTC
–
10k
14
+
UVLO
RECHRG
NTCERR
+
NTC
–
BAT UV
TOO
COLD
T
–
TIMER
OSCILLATOR
21
CONTROL LOGIC
HOLD
10k
4.1V
RECHARGE
RESET
TOO
HOT
CHRG
CLK
COUNTER
CTIMER
18
STOP
+
C/10
+
NTC ENABLE
0.1V
EOC
2μA
–
16
GND
11
SUSP
4090 BD
4090fa
11
LTC4090/LTC4090-5
OPERATION
Introduction
The LTC4090/LTC4090-5 are complete PowerPathTM
controllers for battery powered USB applications. The
LTC4090/LTC4090-5 are designed to receive power from
a low voltage source (e.g., USB or 5V wall adapter), a
high voltage source (e.g., FireWire/IEEE1394, automotive
battery, 12V wall adapter, etc.), and a single-cell Li-Ion
battery. They can then deliver power to an application
connected to the OUT pin and a battery connected to the
BAT pin (assuming that an external supply other than
the battery is present). Power supplies that have limited
current resources (such as USB VBUS supplies) should
be connected to the IN pin which has a programmable
current limit. Battery charge current will be adjusted to
ensure that the sum of the charge current and load current does not exceed the programmed input current limit
(see Figure 1).
An ideal diode function provides power from the battery
when output / load current exceeds the input current limit or
when input power is removed. Powering the load through
the ideal diode instead of connecting the load directly to
the battery allows a fully charged battery to remain fully
charged until external power is removed. Once external
power is removed the output drops until the ideal diode is
forward biased. The forward biased ideal diode will then
provide the output power to the load from the battery.
The LTC4090/LTC4090-5 also include a high voltage
switching regulator which has the ability to receive power
from a high voltage input. This input takes priority over the
USB VBUS input (i.e., if both HVIN and IN are present, load
current and charge current will be delivered via the high
voltage path). When enabled, the high voltage regulator
regulates the HVOUT voltage using a constant frequency,
current mode regulator. An external PFET between HVOUT
(drain) and OUT (source) is turned on via the ⎯H⎯V⎯P⎯R pin
allowing OUT to charge the battery and/or supply power
to the application. The LTC4090’s Bat-Track maintains
approximately 300mV between the OUT pin and the BAT
pin, while the LTC4090-5 provides a fixed 5V output.
PowerPath is a trademark of Linear Technology Corporation
HVIN
SW
L1
Q1
D1
HIGH VOLTAGE
BUCK REGULATOR
HVOUT
C1
+
4.25V (RISING)
3.15V (FALLING)
–
HVPR
19
+
–
IN
+
–
ENABLE
LOAD
75mV (RISING)
25mV (FALLING)
OUT
21
USB CURRENT LIMIT
+
–
+
–
30mV
CC/CV REGULATOR
CHARGER
30mV
+
EDA
IDEAL
DIODE
OUT
21
GATE
–
BAT
21
4090 F01
BAT
+
Li-Ion
Figure 1. Simplified PowerPath Block Diagram
4090fa
12
LTC4090/LTC4090-5
OPERATION
USB Input Current Limit
The input current limit and charge control circuits of the
LTC4090/LTC4090-5 are designed to limit input current as
well as control battery charge current as a function of IOUT.
OUT drives the external load and the battery charger.
If the combined load at OUT does not exceed the programmed input current limit, OUT will be connected to IN
through an internal 215mΩ P-channel MOSFET.
If the combined load at OUT exceeds the programmed input
current limit, the battery charger will reduce its charge current by the amount necessary to enable the external load
to be satisfied while maintaining the programmed input
current. Even if the battery charge current is set to exceed
the allowable USB current, a correctly programmed input
current limit will ensure that the USB specification is never
violated. Furthermore, load current at OUT will always be
prioritized and only excess available current will be used
to charge the battery.
The input current limit, ICL, can be programmed using the
following formula:
⎛ 1000
⎞ 1000 V
ICL = ⎜
• VCLPROG ⎟ =
⎝ RCLPROG
⎠ RCLPROG
where VCLPROG is the CLPROG pin voltage (typically 1V)
and RCLPROG is the total resistance from the CLPROG pin
to ground. For best stability over temperature and time,
1% metal film resistors are recommended.
The programmed battery charge current, ICHG, is defined
as:
⎛ 50, 000
⎞ 50, 000 V
ICHG = ⎜
• VPROG ⎟ =
⎝ RPROG
⎠ RPROG
Input current, IIN, is equal to the sum of the BAT pin output
current and the OUT pin output current. VCLPROG will track
the input current according to the following equation:
IIN = IOUT + IBAT =
VCLPROG
• 1000
RCLPROG
In USB applications, the maximum value for RCLPROG
should be 2.1k. This will prevent the input current from
exceeding 500mA due to LTC4090/LTC4090-5 tolerances
and quiescent currents. A 2.1k CLPROG resistor will give a
typical current limit of 476mA in high power mode (when
HPWR is high) or 95mA in low power mode (when HPWR
is low).
When SUSP is driven to a logic high, the input power
path is disabled and the ideal diode from BAT to OUT will
supply power to the application.
High Voltage Step Down Regulator
The power delivered from HVIN to HVOUT is controlled by
a constant frequency, current mode step down regulator.
An external P-channel MOSFET directs this power to OUT
and prevents reverse conduction from OUT to HVOUT (and
ultimately HVIN).
An oscillator, with frequency set by RT, enables an RS flipflop, turning on the internal power switch. An amplifier and
comparator monitor the current flowing between HVIN and
SW pins, turning the switch off when this current reaches
a level determined by the voltage at VC. An error amplifier
servos the VC node to maintain approximately 300mV
between OUT and BAT (LTC4090). By keeping the voltage
across the battery charger low, efficiency is optimized because power lost to the battery charger is minimized and
power available to the external load is maximized. If the
BAT pin voltage is less than approximately 3.3V, then the
error amplifier will servo the VC node to provide a constant
HVOUT output voltage of about 3.6V (LTC4090). An active
clamp on the VC node provides current limit. The VC node
is also clamped to the voltage on the HVEN pin; soft-start
is implemented by generating a voltage ramp at the HVEN
pin using an external resistor and capacitor.
The switch driver operates from either the high voltage
input or from the BOOST pin. An external capacitor and
internal diode are used to generate a voltage at the BOOST
pin that is higher than the input supply. This allows the
driver to fully saturate the internal bipolar NPN power
switch for efficient operation.
To further optimize efficiency, the high voltage buck regulator automatically switches to Burst Mode® operation in
light load situations. Between bursts, all circuitry associated
with controlling the output switch is shut down reducing
the input supply current.
4090fa
13
LTC4090/LTC4090-5
OPERATION
IIN
500
IIN
100
500
IIN
ILOAD
300
200
100
400
ILOAD
60
40
20
IBAT
(CHARGING)
(CHARGING)
0
0
4090 F02a
100
200
300
400
500
IBAT
ILOAD(mA)
(IDEAL DIODE)
(a) High Power Mode/Full Charge
RPROG = 100k and RCLPROG = 2k
ILOAD
300
IBAT = ICHG
200
IBAT = ICL = IOUT
100
IBAT
0
CURRENT (mA)
80
CURRENT (mA)
CURRENT (mA)
400
IBAT
(CHARGING)
0
0
20
40
4090 F02b
60
100
IBAT
ILOAD(mA)
(IDEAL DIODE)
0
80
(a) Low Power Mode/Full Charge
RPROG = 100k and RCLPROG = 2k
4090 F02c
100
200
300
400
500
IBAT
ILOAD (mA)
(IDEAL DIODE)
(a) High Power Mode with
ICL = 500mA and ICHG = 250mA
RPROG = 100k and RCLPROG = 2k
Figure 2. Input and Battery Currents as a Function of Load Current
The oscillator reduces the switch regulator’s operating
frequency when the voltage at the HVOUT pin is low (below 2.95V). This frequency foldback helps to control the
output current during start-up and overload.
The high voltage regulator contains a power good comparator which trips when the HVOUT pin is at 2.8V. The PG
output is an open-collector transistor that is off when the
output is in regulation, allowing an external resistor to pull
the PG pin high. Power good is valid when the switching
regulator is enabled and HVIN is above 3.6V.
Ideal Diode From BAT to OUT
The LTC4090/LTC4090-5 have an internal ideal diode as
well as a controller for an optional external ideal diode. If
a battery is the only power supply available, or if the load
current exceeds the programmed input current limit, then
the battery will automatically deliver power to the load via
an ideal diode circuit between the BAT and OUT pins. The
ideal diode circuit (along with the recommended 4.7μF
capacitor on the OUT pin) allows the LTC4090/LTC4090-5
to handle large transient loads and wall adapter or USB
VBUS connect/disconnect scenarios without the need for
large bulk capacitors. The ideal diode responds within
a few microseconds and prevents the OUT pin voltage
from dropping significantly below the BAT pin voltage.
A comparison of the I-V curve of the ideal diode and a
Schottky diode can be seen in Figure 3.
If the desired input current increases beyond the programmed input current limit additional current will be drawn
from the battery via the internal ideal diode. Furthermore,
if power to IN (USB VBUS) or HVIN (high voltage input) is
removed, then all of the application power will be provided
by the battery via the ideal diode. A 4.7μF capacitor at
OUT is sufficient to keep a transition from input power
to battery power from causing significant output voltage
droop. The ideal diode consists of a precision amplifier that
enables a large P-channel MOSFET transistor whenever the
voltage at OUT is approximately 20mV (VFWD) below the
voltage at BAT. The resistance of the internal ideal diode
is approximately 215mΩ.
If this is sufficient for the application then no external
components are necessary. However if more conductance
is needed, an external P-channel MOSFET can be added
from BAT to OUT. The GATE pin of the LTC4090/LTC4090-5
drives the gate of the external PFET for automatic ideal
diode control. The source of the external MOSFET should
be connected to OUT and the drain should be connected
to BAT. In order to help protect the external MOSFET in
overcurrent situations, it should be placed in close thermal
contact to the LTC4090/LTC4090-5.
Burst Mode is a registered trademark of Linear Technology Corporation
4090fa
14
LTC4090/LTC4090-5
OPERATION
Suspend Mode
Battery Charger
trickle charge mode to bring the cell voltage up to a safe
level for charging. The charger goes into the fast charge
constant current mode once the voltage on the BAT pin
rises above 2.9V. In constant current mode, the charge
current is set by RPROG. When the battery approaches the
final float voltage, the charge current begins to decrease
as the LTC4090/LTC4090-5 switch to constant voltage
mode. When the charge current drops below 10% of the
programmed value while in constant voltage mode the
⎯C⎯H⎯R⎯G pin assumes a high impedance state.
The battery charger circuits of the LTC4090/LTC4090-5
are designed for charging single cell lithium-ion batteries.
Featuring an internal P-channel power MOSFET, the charger
uses a constant current / constant voltage charge algorithm
with programmable charge current and a programmable
timer for charge termination. Charge current can be
programmed up to 1.5A. The final float voltage accuracy
is ±0.8% typical. No blocking diode or sense resistor is
required when powering either the IN or the HVIN pins.
The ⎯C⎯H⎯R⎯G open-drain status output provides information
regarding the charging status of the LTC4090/LTC4090-5
at all times. An NTC input provides the option of charge
qualification using battery temperature.
An external capacitor on the TIMER pin sets the total
minimum charge time. When this time elapses, the
charge cycle terminates and the ⎯C⎯H⎯R⎯G pin assumes a
high impedance state, if it has not already done so. While
charging in constant current mode, if the charge current
is decreased by thermal regulation or in order to maintain
the programmed input current limit, the charge time is
automatically increased. In other words, the charge time is
extended inversely proportional to the actual charge current
delivered to the battery. For Li-Ion and similar batteries that
require accurate final float potential, the internal bandgap
reference, voltage amplifier and the resistor divider provide
regulation with ±0.8% accuracy.
The charge cycle begins when the voltage at the OUT
pin rises above the battery voltage and the battery voltage is below the recharge threshold. No charge current
actually flows until the OUT voltage is 100mV above
the BAT voltage. At the beginning of the charge cycle, if
the battery voltage is below 2.9V, the charger goes into
Trickle Charge and Defective Battery Detection
When SUSP is pulled above VIH the LTC4090/LTC4090-5
enter suspend mode to comply with the USB specification.
In this mode, the power path between IN and OUT is put
in a high impedance state to reduce the IN input current to
50μA. If no other power source is available to drive HVIN,
the system load connected to OUT is supplied through the
ideal diodes connected to BAT.
CONSTANT
I0N
LTC4090/LTC4090-5
CURRENT (A)
IMAX
SLOPE: 1/RDIO(ON)
CONSTANT
R0N
IFWD
SLOPE: 1/RFWD
SCHOTTKY
DIODE
CONSTANT
V0N
At the beginning of a charge cycle, if the battery voltage
is below 2.9V, the charger goes into trickle charge reducing the charge current to 10% of the full-scale current.
If the low battery voltage persists for one quarter of the
programmed total charge time, the battery is assumed
to be defective, the charge cycle is terminated and the
⎯C⎯H⎯R⎯G pin output assumes a high impedance state. If
for any reason the battery voltage rises above ~2.9V the
charge cycle will be restarted. To restart the charge cycle
(i.e., when the dead battery is replaced with a discharged
battery), simply remove the input voltage and reapply it
or cycle the TIMER pin to 0V.
Programming Charge Current
The formula for the battery charge current is:
0
FORWARD VOLTAGE (V)
VFWD
4090 F03
Figure 3. LTC4090/LTC4090-5 Versus Schottky Diode Forward
Voltage Drop
ICHG = IPROG • 50, 000 =
VPROG
• 50, 000
RPROG
4090fa
15
LTC4090/LTC4090-5
OPERATION
where VPROG is the PROG pin voltage and RPROG is the
total resistance from the PROG pin to ground. Keep in
mind that when the LTC4090/LTC4090-5 are powered
from the IN pin, the programmed input current limit takes
precedence over the charge current. In such a scenario,
the charge current cannot exceed the programmed input
current limit.
For example, if typical 500mA charge current is required,
calculate:
RPROG =
1V
• 50, 000 = 100k
500mA
For best stability over temperature and time, 1% metal
film resistors are recommended. Under trickle charge
conditions, this current is reduced to 10% of the fullscale value.
The Charge Timer
The programmable charge timer is used to terminate the
charge cycle. The timer duration is programmed by an
external capacitor at the TIMER pin. The charge time is
typically:
tTIMER (hours) =
CTIMER • RPROG • 3hours
0.1µF • 100k
The timer starts when an input voltage greater than the
undervoltage lockout threshold level is applied or when
leaving shutdown and the voltage on the battery is less than
the recharge threshold. At power-up or exiting shutdown
with the battery voltage less than the recharge threshold,
the charge time is a full cycle. If the battery is greater than
the recharge threshold the timer will not start and charging
is prevented. If after power-up the battery voltage drops
below the recharge threshold, or if after a charge cycle
the battery voltage is still below the recharge threshold,
the charge time is set to one-half of a full cycle.
The LTC4090/LTC4090-5 have a feature that extends charge
time automatically. Charge time is extended if the charge
current in constant current mode is reduced due to load
current or thermal regulation. This change in charge time
is inversely proportional to the change in charge current.
As the LTC4090/LTC4090-5 approach constant voltage
mode the charge current begins to drop. This change in
charge current is due to normal charging operation and
does not affect the timer duration.
Consider, for example, a USB charge condition where
RCLPROG = 2k, RPROG = 100k and CTIMER = 0.1μF. This
corresponds to a three hour charge cycle. However, if the
HPWR input is set to a logic low, then the input current
limit will be reduced from 500mA to 100mA. With no additional system load, this means the charge current will
be reduced to 100mA. Therefore, the termination timer
will automatically slow down by a factor of five until the
charger reaches constant voltage mode (i.e. VBAT approaches 4.2V) or HPWR is returned to a logic high. The
charge cycle is automatically lengthened to account for
the reduced charge current. The exact time of the charge
cycle will depend on how long the charger remains in
constant current mode and/or how long the HPWR pin
remains logic low.
Once a time-out occurs and the voltage on the battery is
greater than the recharge threshold, the charge current
stops, and the ⎯C⎯H⎯R⎯G output assumes a high impedance
state if it has not already done so.
Connecting the TIMER pin to ground disables the battery
charger.
⎯C⎯H⎯R⎯G Status Output Pin
When the charge cycle starts, the ⎯C⎯H⎯R⎯G pin is pulled to
ground by an internal N-channel MOSFET capable of driving an LED. When the charge current drops below 10%
of the programmed full charge current while in constant
voltage mode, the pin assumes a high impedance state,
but charge current continues to flow until the charge
time elapses. If this state is not reached before the end
of the programmable charge time, the pin will assume a
high impedance state when a time-out occurs. The ⎯C⎯H⎯R⎯G
current detection threshold can be calculated by the following equation:
IDETECT =
0.1V
5000 V
• 50, 000 =
RPROG
RPROG
4090fa
16
LTC4090/LTC4090-5
OPERATION
For example, if the full charge current is programmed
to 500mA with a 100k PROG resistor the ⎯C⎯H⎯R⎯G pin will
change state at a battery charge current of 50mA.
Note: The end-of-charge (EOC) comparator that monitors the charge current latches its decision. Therefore,
the first time the charge current drops below 10% of the
programmed full charge current while in constant voltage mode, it will toggle ⎯C⎯H⎯R⎯G to a high impedance state.
If, for some reason the charge current rises back above
the threshold, the ⎯C⎯H⎯R⎯G pin will not resume the strong
pull-down state. The EOC latch can be reset by a recharge
cycle (i.e., VBAT drops below the recharge threshold) or
toggling the input power to the part.
Automatic Recharge
After the battery charger terminates, it will remain off
drawing only microamperes of current from the battery. If
the product remains in this state long enough, the battery
will eventually self discharge. To ensure that the battery is
always topped off, a charge cycle will automatically begin
when the battery voltage falls below VRECHRG (typically
4.1V). To prevent brief excursions below VRECHRG from
resetting the safety timer, the battery voltage must be
below VRECHRG for more than a few milliseconds. The
charge cycle and safety timer will also restart if the IN
UVLO cycles low and then high (e.g. IN, is removed and
then replaced).
Thermal Regulation
To prevent thermal damage to the IC or surrounding
components, an internal thermal feedback loop will
automatically decrease the programmed charge current
if the die temperature rises to approximately 105°C.
Thermal regulation protects the LTC4090/LTC4090-5
from excessive temperature due to high power operation
or high ambient thermal conditions and allows the user
to push the limits of the power handling capability with a
given circuit board design without risk of damaging the
LTC4090/LTC4090-5 or external components. The benefit
of the LTC4090/LTC4090-5 thermal regulation loop is that
charge current can be set according to actual conditions
rather than worst-case conditions with the assurance that
the battery charger will automatically reduce the current
in worst-case conditions.
Undervoltage Lockout
An internal undervoltage lockout circuit monitors the input
voltage (IN) and the output voltage (OUT) and disables
either the input current limit or the battery charger circuits
or both. The input current limit circuitry is disabled until
VIN rises above the undervoltage lockout threshold and VIN
exceeds VOUT by 50mV. The battery charger circuits are disabled until VOUT exceeds VBAT by 50mV. Both undervoltage
lockout comparators have built-in hysteresis.
NTC Thermistor
The battery temperature is measured by placing a negative temperature coefficient (NTC) thermistor close to
the battery pack. To use this feature connect the NTC
thermistor, RNTC, between the NTC pin and ground and a
bias resistor, RNOM, from VNTC to NTC. RNOM should be
a 1% resistor with a value equal to the value of the chosen
NTC thermistor at 25°C (denoted R25C).
The LTC4090/LTC4090-5 will pause charging when the
resistance of the NTC thermistor drops to 0.41 times the
value of R25C or approximately 4.1k (for a Vishay “Curve
2” thermistor, this corresponds to approximately 50°C).
The safety timer also pauses until the thermistor indicates
a return to a valid temperature. As the temperature drops,
the resistance of the NTC thermistor rises. The LTC4090/
LTC4090-5 are also designed to pause charging (and timer)
when the value of the NTC thermistor increases to 2.82
times the value of R25C. For a Vishay “Curve 2” thermistor
this resistance, 28.2k, corresponds to approximately 0°C.
The hot and cold comparators each have approximately
3°C of hysteresis to prevent oscillation about the trip point.
Grounding the NTC pin disables all NTC functionality.
4090fa
17
LTC4090/LTC4090-5
APPLICATIONS INFORMATION
USB and 5V Wall Adapter Power
Although the LTC4090/LTC4090-5 are designed to draw
power from a USB port, a higher power 5V wall adapter
can also be used to power the application and charge the
battery (higher voltage wall adapters can be connected
directly to HVIN). Figure 4 shows an example of combining
a 5V wall adapter and a USB power input. With its gate
grounded by 1k, P-channel MOSFET MP1 provides USB
power to the LTC4090/LTC4090-5 when 5V wall power is
not available. When 5V wall power is available, diode D1
supplies power to the LTC4090/LTC4090-5, pulls the gate
of MN1 high to increase the charge current (by increasing
the input current limit), and pulls the gate of MP1 high to
disable it and prevent conduction back to the USB port.
Setting the Switching Frequency
The high voltage switching regulator uses a constant
frequency PWM architecture that can be programmed to
switch from 200kHz to 2.4MHz by using a resistor tied
from the RT pin to ground. A table showing the necessary
RT value for a desired switching frequency is in Table 1.
5V WALL
ADAPTER
850mA ICHG
ICHG
BAT
D1
LTC4090
USB POWER
500mA ICHG
IN
SWITCHING FREQUENCY (MHz)
RT VALUE (kΩ)
0.2
187
0.3
121
0.4
88.7
0.5
68.1
0.6
56.2
0.7
46.4
0.8
40.2
0.9
34.0
1.0
29.4
1.2
23.7
1.4
19.1
1.6
16.2
1.8
13.3
2.0
11.5
2.2
9.76
2.4
8.66
CLPROG
1k
MN1
2.87k
2k
Li-Ion
BATTERY
59k
4090 F04
Figure 4. USB or 5V Wall Adapter Power
Operating Frequency Tradeoffs
Selection of the operating frequency for the high voltage
buck regulator is a tradeoff between efficiency, component
size, minimum dropout voltage, and maximum input voltage. The advantage of high frequency operation is that
smaller inductor and capacitor values may be used. The
disadvantages are lower efficiency, lower maximum input
voltage, and higher dropout voltage. The highest acceptable
switching frequency (fSW(MAX)) for a given application can
be calculated as follows:
fSW(MAX ) =
Table 1. Switching Frequency vs RT Value
+
PROG
MP1
VD + VHVOUT
tON(MIN) • ( VD + VHVIN – VSW )
where VHVIN is the typical high voltage input voltage,
VHVOUT is the output voltage of the switching regulator, VD
is the catch diode drop (~0.5V), and VSW is the internal
switch drop (~0.5V at max load). This equation shows
that slower switching frequency is necessary to safely
accommodate high VHVIN/VHVOUT ratio. Also, as shown in
the next section, lower frequency allows a lower dropout
voltage. The reason input voltage range depends on the
switching frequency is because the high voltage switch
has finite minimum on and off times. The switch can turn
on for a minimum of ~150ns and turn off for a minimum
of ~150ns. This means that the minimum and maximum
duty cycles are:
DCMIN = fSW • tON(MIN)
DCMAX = 1 – fSW • tOFF(MIN)
where fSW is the switching frequency, tON(MIN) is the
minimum switch-on time (~150ns), and tOFF(MIN) is the
4090fa
18
LTC4090/LTC4090-5
APPLICATIONS INFORMATION
minimum switch-off time (~150ns). These equations show
that duty cycle range increases when switching frequency
is decreased.
A good choice of switching frequency should allow adequate input voltage range (see next section) and keep
the inductor and capacitor values small.
HVIN Input Voltage Range
The maximum input voltage range for the LTC4090/
LTC4090-5 applications depends on the switching frequency, the Absolute Maximum Ratings of the VHVIN and
BOOST pins, and the operating mode.
may enter pulse skipping operation where some switching pulses are skipped to maintain output regulation. In
this mode the output voltage ripple and inductor current
ripple will be higher than in normal operation. Above 41.5V,
switching will stop.
The minimum input voltage is determined by either the high
voltage regulator’s minimum operating voltage of ~6V or by
its maximum duty cycle (see equation in previous section).
The minimum input voltage due to duty cycle is:
VHVIN(MIN) =
VHVOUT + VD
−V +V
1− fSW tOFF(MIN) D SW
The high voltage switching regulator can operate from
input voltages up to 36V, and safely withstand input voltages up to 60V. Note that while VHVIN > 41.5V (typical),
the LTC4090/LTC4090-5 will stop switching, allowing the
output to fall out of regulation.
where VHVIN(MIN) is the minimum input voltage, and
tOFF(MIN) is the minimum switch off time (150ns). Note
that higher switching frequency will increase the minimum
input voltage. If a lower dropout voltage is desired, a lower
switching frequency should be used.
While the high voltage regulator output is in start-up,
short-circuit, or other overload conditions, the switching
frequency should be chosen according to the following
discussion.
Inductor Selection and Maximum Output Current
For safe operation at inputs up to 60V the switching frequency must be low enough to satisfy VHVIN(MAX) ≥ 45V
according to the following equation. If lower VHVIN(MAX)
is desired, this equation can be used directly.
VHVIN(MAX ) =
VHVOUT + VD
–V +V
fSW • tON(MIN) D SW
where VHVIN(MAX) is the maximum operating input voltage,
VHVOUT is the high voltage regulator output voltage, VD is
the catch diode drop (~0.5V), VSW is the internal switch
drop (~0.5V at max load), fSW is the switching frequency
(set by RT), and tON(MIN) is the minimum switch-on time
(~150ns). Note that a higher switching frequency will depress the maximum operating input voltage. Conversely,
a lower switching frequency will be necessary to achieve
safe operation at high input voltages.
If the output is in regulation and no short-circuit, startup, or overload events are expected, then input voltage
transients of up to 60V are acceptable regardless of the
switching frequency. In this mode, the LTC4090/LTC4090-5
A good choice for the inductor value is L = 6.8μH (assuming a 800kHz operating frequency). With this value the
maximum load current will be ~2.4A. The RMS current
rating of the inductor must be greater than the maximum
load current and its saturation current should be about
30% higher. Note that the maximum load current will be
programmed charge current plus the largest expected
application load current. For robust operation in fault
conditions, the saturation current should be ~3.5A. To
keep efficiency high, the series resistance (DCR) should
be less than 0.1Ω. Table 2 lists several vendors and types
that are suitable.
Table 2. Inductor Vendors
VENDOR URL
PART SERIES
TYPE
Murata
www.murata.com
LQH55D
Open
TDK
www.componenttdk.com
SLF7045
SLF10145
Shielded
Shielded
Toko
www.toko.com
D62CB
D63CB
D75C
D75F
Shielded
Shielded
Shielded
Open
Sumida
www.sumida.com
CR54
CDRH74
CDRH6D38
CR75
Open
Shielded
Shielded
Open
4090fa
19
LTC4090/LTC4090-5
APPLICATIONS INFORMATION
Catch Diode
The catch diode conducts current only during switch-off
time. Average forward current in normal operation can
be calculated from:
ID( AVG) = IHVOUT •
( VHVIN – VHVOUT )
VHVIN
where IHVOUT is the output load current. The only reason to
consider a diode with a larger current rating than necessary
for nominal operation is for the worst-case condition of
shorted output. The diode current will then increase to the
typical peak switch current. Peak reverse voltage is equal
to the regulator input voltage. Use a Schottky diode with a
reverse voltage rating greater than the input voltage. The
overvoltage protection feature in the high voltage regulator
will keep the switch off when VHVIN > 45V which allows
the use of 45V rated Schottky even when VHVIN ranges
up to 60V. Table 3 lists several Schottky diodes and their
manufacturers.
Table 3. Diode Vendors
PART NUMBER
VR
(V)
IAVE
(A)
VF AT 1A
(MV)
VF AT 2A
(MV)
On Semiconductor
MBRM120E
MBRM140
20
40
1
1
530
550
595
Diodes Inc.
B130
B220
B230
B360
DFLS240L
30
20
30
60
40
1
2
2
3
2
500
International Rectifier
10BQ030
20BQ030
30
30
1
2
420
500
COUT =
100
VOUT fSW
where fSW is in MHz, and COUT is the recommended output
capacitance in μF. Use X5R or X7R types. This choice will
provide low output ripple and good transient response.
Transient performance can be improved with a higher value
capacitor if the compensation network is also adjusted
to maintain the loop bandwidth. A lower value of output
capacitor can be used to save space and cost but transient
performance will suffer. See the High Voltage Regulator
Frequency Compensation section to choose an appropriate
compensation network.
When choosing a capacitor, look carefully through the
data sheet to find out what the actual capacitance is under
operating conditions (applied voltage and temperature).
A physically larger capacitor, or one with a higher voltage
rating, may be required. High performance tantalum or
electrolytic capacitors can be used for the output capacitor.
Low ESR is important, so choose one that is intended for
use in switching regulators. The ESR should be specified
by the supplier, and should be 0.05Ω or less. Such a
capacitor will be larger than a ceramic capacitor and will
have a larger capacitance, because the capacitor must be
large to achieve low ESR.
Ceramic Capacitors
500
500
550
500
470
470
High Voltage Regulator Output Capacitor Selection
The high voltage regulator output capacitor has two essential functions. Along with the inductor, it filters the
square wave generated at the switch pin to produce the
DC output. In this role it determines the output ripple, and
low impedance at the switching frequency is important.
The second function is to store energy in order to satisfy
transient loads and stabilize the LTC4090/LTC4090-5’s
control loop. Ceramic capacitors have very low equivalent series resistance (ESR) and provide the best ripple
performance. A good starting value is:
Ceramic capacitors are small, robust and have very low
ESR. However, ceramic capacitors can cause problems
when used with the high voltage switching regulator
due to their piezoelectric nature. When in Burst Mode
operation, the LTC4090/LTC4090-5’s switching frequency
depends on the load current, and at very light loads the
LTC4090/LTC4090-5 can excite the ceramic capacitor at
audio frequencies, generating audible noise. Since the
LTC4090/LTC4090-5 operate at a lower current limit during
Burst Mode operation, the noise is typically very quiet to a
casual ear. If this is unacceptable, use a high performance
tantalum or electrolytic capacitor at the output.
High Voltage Regulator Frequency Compensation
The LTC4090/LTC4090-5 high voltage regulator uses
current mode control to regulate the output. This simplifies loop compensation. In particular, the high voltage
4090fa
20
LTC4090/LTC4090-5
APPLICATIONS INFORMATION
regulator does not require the ESR of the output capacitor
for stability, so you are free to use ceramic capacitors to
achieve low output ripple and small circuit size. Frequency
compensation is provided by the components tied to the
VC pin, as shown in Figure 1. Generally a capacitor (CC)
and a resistor (RC) in series to ground are used. In addition, there may be a lower value capacitor in parallel.
This capacitor (CF) is not part of the loop compensation
but is used to filter noise at the switching frequency, and
is required only if a phase-lead capacitor is used or if the
output capacitor has high ESR.
Loop compensation determines the stability and transient
performance. Designing the compensation network is a bit
complicated and the best values depend on the application
and in particular the type of output capacitor. A practical
approach is to start with the front page schematic and tune
the compensation network to optimize performance. Stability should then be checked across all operating conditions,
including load current, input voltage and temperature. The
LTC1375 datasheet contains a more thorough discussion
of loop compensation and describes how to test the stability using a transient load. Figure 5 shows the transient
response when the load current is stepped from 500mA
to 1500mA and back to 500mA.
Low Ripple Burst Mode Operation and Pulse-Skip
Mode
The LTC4090/LTC4090-5 are capable of operating in either
low ripple Burst Mode operation or pulse-skip mode which
are selected using the SYNC pin. Tie the SYNC pin below
VSYNC,L (typically 0.5V) for low ripple Burst Mode operation
or above VSYNC,H (typically 0.8V) for pulse-skip mode.
To enhance efficiency at light loads, the LTC4090/LTC4090-5
can be operated in low ripple Burst Mode operation which
keeps the output capacitor charged to the proper voltage
while minimizing the input quiescent current. During Burst
Mode operation, the LTC4090/LTC4090-5 deliver single
cycle bursts of current to the output capacitor followed
by sleep periods where the output power is delivered to
the load by the output capacitor. Because the LTC4090/
LTC4090-5 deliver power to output with single, low current
pulses, the output ripple is kept below 15mV for a typical
application. As the load current decreases towards a no
load condition, the percentage of time that the high voltage regulator operates in sleep mode increases and the
average input current is greatly reduced resulting in high
efficiency even at very low loads. See Figure 6.
At higher output loads (above 70mA for the front page
application) the LTC4090/LTC4090-5 will be running at
the frequency programmed by the RT resistor, and will be
operating in standard PWM mode. The transition between
PWM and low ripple Burst Mode operation is seamless,
and will not disturb the output voltage.
If low quiescent current is not required, the LTC4090/
LTC4090-5 can operate in pulse-skip mode. The benefit
of this mode is that the LTC4090/LTC4090-5 will enter full
frequency standard PWM operation at a lower output load
current than when in Burst Mode operation. The front page
application circuit will switch at full frequency at output
loads higher than about 60mA.
VIN = 12V; FIGURE 12 SCHEMATIC
ILOAD = 10mA
IL
0.5A/DIV
FIGURE 12 SCHEMATIC
VSW
5V/DIV
HVOUT
50mV/DIV
VOUT
10mV/DIV
IL
1A/DIV
5μs/DIV
25μs/DIV
4090 F05
4090 F06
Figure 6. High Voltage Regulator Burst Mode Operation
Figure 5. Transient Load Response of the LTC4090 High Voltage
Regulator Front Page Application as the Load Current is Stepped
from 500mA to 1500mA.
4090fa
21
LTC4090/LTC4090-5
APPLICATIONS INFORMATION
Boost Pin Considerations
Capacitor C2 (see Block Diagram) and an internal diode
are used to generate a boost voltage that is higher than
the input voltage. In most cases, a 0.47μF capacitor will
work well. The BOOST pin must be at least 2.3V above
the SW pin for proper operation.
High Voltage Regulator Soft-Start
The HVEN pin can be used to soft-start the high voltage
regulator of the LTC4090/LTC4090-5, reducing maximum
input current during start-up. The HVEN pin is driven
through an external RC filter to create a voltage ramp at
this pin. Figure 7 shows the start-up and shutdown waveforms with the soft-start circuit. By choosing a large RC
time constant, the peak start-up current can be reduced
to the current that is required to regulate the output, with
no overshoot. Choose the value of the resistor so that it
can supply 20μA when the HVEN pin reaches 2.3V.
IL
1A/DIV
RUN
15k
HVEN
0.22μF
VRUN/SS
2V/DIV
GND
VOUT
2V/DIV
2ms/DIV
4090 F07
Figure 7. To Soft-Start the High Voltage Regulator, Add a Resistor
and Capacitor to the HVEN Pin
Synchronization and Mode
The SYNC pin allows the high voltage regulator to be
synchronized to an external clock.
Synchronizing the LTC4090/LTC4090-5 internal oscillator to an external frequency can be done by connecting a
square wave (with 20% to 80% duty cycle) to the SYNC
pin. The square wave amplitude should be such that the
valleys are below 0.3V and the peaks are above 0.8V (up
to 6V). The high voltage regulator may be synchronized
over a 300kHz to 2MHz range. The RT resistor should be
chosen such that the LTC4090/LTC4090-5 oscillate 25%
lower than the external synchronization frequency to ensure
adequate slope compensation. While synchronized, the
high voltage regulator will turn on the power switch on
positive going edges of the clock. When the power good
(PG) output is low, such as during start-up, short-circuit,
and overload conditions, the LTC4090/LTC4090-5 will disable the synchronization feature. The SYNC pin should be
grounded when synchronization is not required.
Alternate NTC Thermistors and Biasing
The LTC4090/LTC4090-5 provide temperature qualified
charging if a grounded thermistor and a bias resistor are
connected to NTC (see Figure 8). By using a bias resistor
whose value is equal to the room temperature resistance
of the thermistor (R25C) the upper and lower temperatures
are pre-programmed to approximately 50°C and 0°C,
respectively (assuming a Vishay “Curve 2” thermistor).
The upper and lower temperature thresholds can be adjusted by either a modification of the bias resistor value
or by adding a second adjustment resistor to the circuit.
If only the bias resistor is adjusted, then either the upper
or the lower threshold can be modified but not both. The
other trip point will be determined by the characteristics
of the thermistor. Using the bias resistor in addition to an
adjustment resistor, both the upper and the lower temperature trip points can be independently programmed with
the constraint that the difference between the upper and
lower temperature thresholds cannot decrease. Examples
of each technique are given below.
NTC thermistors have temperature characteristics which
are indicated on resistance-temperature conversion tables.
The Vishay-Dale thermistor NTHS0603N02N1002J, used
in the following examples, has a nominal value of 10k
and follows the Vishay “Curve 2” resistance-temperature
characteristic. The LTC4090/LTC4090-5’s trip points are
designed to work with thermistors whose resistance-temperature characteristics follow Vishay Dale’s “R-T Curve 2.”
The Vishay NTHS0603N02N1002J is an example of such
a thermistor. However, Vishay Dale has many thermistor
products that follow the “R-T Curve 2” characteristic in a
variety of sizes. Furthermore, any thermistor whose ratio
of RCOLD to RHOT is about 7.0 will also work (Vishay Dale
R-T Curve 2 shows a ratio of 2.815/0.409 = 6.89).
4090fa
22
LTC4090/LTC4090-5
APPLICATIONS INFORMATION
In the explanation below, the following notation is used.
R25C = Value of the Thermistor at 25°C
RNTC|COLD = Value of Thermistor at the Cold Trip Point
RNTC|HOT = Value of the Thermistor at the Hot Trip Point
rHOT= Ratio of RNTC|HOT to R25C
RNOM = Primary Thermistor Bias Resistor (see Figure 8)
R1 = Optional Temperature Range Adjustment resistor
(see Figure 9)
The trip points for the LTC4090/LTC4090-5’s temperature qualification are internally programmed at 0.29 •
VNTC for the hot threshold and 0.74 • VNTC for the cold
threshold.
Therefore, the hot trip point is set when:
RNOM + RNTCHOT
|
• VNTC = 0.29 • VNTC
RNOM + RNTC|COLD
By setting RNOM equal to R25C, the above equations result
in rHOT = 0.409 and rCOLD = 2.815. Referencing these ratios
to the Vishay Resistance-Temperature Curve 2 chart gives
a hot trip point of about 50°C and a cold trip point of about
0°C. The difference between the hot and cold trip points
is approximately 50°C.
By using a bias resistor, RNOM, different in value from
R25C, the hot and cold trip points can be moved in either
direction. The temperature span will change somewhat due
to the non-linear behavior of the thermistor. The following
equations can be used to easily calculate a new value for
the bias resistor:
rHOT
•R
0.409 25C
r
RNOM = COLD • R25C
2.815
• VNTC = 0.74 • VNTC
VNTC
VNTC
NTC BLCOK
6
RNOM
10k
NTC
and
RNOM =
and the cold trip point is set when:
RNTC|COLD
RNTC|HOT = 0.409 • RNOM
RNTC|COLD = 2.815 • RNOM
rcold = Ratio of RNTC|COLD to R25C
RNTCHOT
|
Solving these equations for RNTC|COLD and RNTC|HOT results
in the following:
NTC BLCOK
6
0.738 • VNTC
–
TOO_COLD
RNOM
13.2k
NTC
0.738 • VNTC
–
TOO_COLD
5
+
5
+
RNTC
10k
–
R1
1.97k
–
TOO_HOT
0.29 • VNTC
TOO_HOT
0.29 • VNTC
+
RNTC
10k
+
+
+
NTC_ENABLE
0.1V
–
4090 F08
Figure 8. Typical NTC Thermistor Circuit
NTC_ENABLE
0.1V
–
4090 F09
Figure 9. NTC Thermistor Circuit with Additional Bias Resistor
4090fa
23
LTC4090/LTC4090-5
APPLICATIONS INFORMATION
where rHOT and rCOLD are the resistance ratios at the desired hot and cold trip points. Note that these equations
are linked. Therefore, only one of the two trip points can
be chosen, the other is determined by the default ratios
designed in the IC. Consider an example where a 40°C
hot trip point is desired.
In general, if the LTC4090/LTC4090-5 is being powered from
IN the power dissipation can be calculated as follows:
PD = (VIN – VBAT) • IBAT + (VIN – VOUT) • IOUT
where PD is the power dissipated, IBAT is the battery
charge current, and IOUT is the application load current.
For a typical application, an example of this calculation
would be:
From the Vishay Curve 2 R-T characteristics, rHOT is 0.5758
at 40°C. Using the above equation, RNOM should be set
to 14.0k. With this value of RNOM, the cold trip point is
about -7°C. Notice that the span is now 47°C rather than
the previous 50°C. This is due to the increase in “temperature gain” of the thermistor as absolute temperature
decreases.
This examples assumes VIN = 5V, VOUT = 4.75V, VBAT =
3.7V, IBAT = 400mA, and IOUT = 100mA resulting in slightly
more than 0.5W total dissipation.
The upper and lower temperature trip points can be independently programmed by using an additional bias resistor
as shown in Figure 9. The following formulas can be used
to compute the values of RNOM and R1:
If the LTC4090 is being powered from HVIN, the power
dissipation can be estimated by calculating the regulator
power loss from an efficiency measurement, and subtracting the catch diode loss.
rCOLD – rHOT
• R25C
2.815
R1= 0.409 • RNOM – rHOT • R25C
RNOM =
For example, to set the trip points to -5°C and 55°C with
a Vishay Curve 2 thermistor choose
RNOM =
3.532 – 0.3467
• 10k = 13.2k
2.815 – 0.409
the nearest 1% value is 13.3k.
R1 = 0.409 • 13.3k – 0.3467 • 10k = 1.97k
the nearest 1% value is 1.96k. The final solution is shown
in Figure 9 and results in an upper trip point of 55°C and
a lower trip point of -5°C.
Power Dissipation and High Temperature
Considerations
The die temperature of the LTC4090/LTC4090-5 must be
lower than the maximum rating of 110°C. This is generally
not a concern unless the ambient temperature is above
85°C. The total power dissipated inside the LTC4090/
LTC4090-5 depend on many factors, including input voltage
(IN or HVIN), battery voltage, programmed charge current,
programmed input current limit, and load current.
PD = (5V – 3.7V) • 0.4A + (5V – 4.75V) • 0.1A
= 545mW
PD = (1− η) • ⎡⎣ VHVOUT •(IBAT + IOUT )⎤⎦
⎛ V
⎞
− VD • ⎜ 1− HVOUT ⎟ • (IBAT + IOUT ) + 0.3V • IBAT )
VHVIN ⎠
⎝
where η is the efficiency of the high voltage regulator and
VD is the forward voltage of the catch diode at I = IBAT
+ IOUT. The first term corresponds to the power lost in
converting VHVIN to VHVOUT, the second term subtracts
the catch diode loss, and the third term is the power dissipated in the battery charger. For a typical application,
an example of this calculation would be:
PD = (1− 0.87) • [ 4V •(1A + 0.6 A)]
4V ⎞
⎛
−0.4V • ⎜ 1−
• (1A + 0.6 A ) + 0.3V • 1A = 0.7 W
⎝ 12V ⎟⎠
This example assumes 87% efficiency, VHVIN = 12V, VBAT
= 3.7V (VHVOUT is about 4V), IBAT = 1A, IOUT = 600mA
resulting in about 0.7W total dissipation. If the LTC4090-5
is being powered from HVIN, the power dissipation can
be estimated by calculating the regulator power loss from
an efficiency measurement, and subtracting the catch
diode loss.
4090fa
24
LTC4090/LTC4090-5
APPLICATIONS INFORMATION
)( (
))
(IBAT + IOUT ) + (5V – VBAT ) • IBAT
(
PD = 1 – η • 5V • IBAT + IOUT
⎛
5V ⎞
– VD • ⎜ 1 –
⎟•
⎝ VHVIN ⎠
thermal resistance from die (i.e., junction) to ambient can
be reduced to θJA = 40°C/W.
Board Layout Considerations
The difference between this equation and that for the
LTC4090 is the last term, which represents the power
dissipation in the battery charger. For a typical application,
an example of this calculation would be:
⎛
5V ⎞
PD = 1 – 0.87 • 5V • 1A + 0.6 A – 0.4V • ⎜ 1 –
•
⎝ 12V ⎟⎠
)( (
))
(1A + 0.6A ) + (5V – 3.7V ) • 1A = 1.97W
(
Like the LTC4090 example, this examples assumes 87%
efficiency, VHVIN = 12V, VBAT = 3.7V, IBAT = 1A and IOUT =
600mA resulting in about 2W total power dissipation.
It is important to solder the exposed backside of the package to a ground plane. This ground should be tied to other
copper layers below with thermal vias; these layers will
spread the heat dissipated by the LTC4090/LTC4090-5.
Additional vias should be placed near the catch diode.
Adding more copper to the top and bottom layers and
tying this copper to the internal planes with vias can
reduce thermal resistance further. With these steps, the
C1 AND D1
GND PADS
SIDE-BY-SIDE
AND SEPERATED
WITH C3 GND PAD
As discussed in the previous section, it is critical that
the exposed metal pad on the backside of the LTC4090/
LTC4090-5 package be soldered to the PC board ground.
Furthermore, proper operation and minimum EMI requires
a careful printed circuit board (PCB) layout. Note that large,
switched currents flow in the power switch (between the
HVIN and SW pins), the catch diode and the HVIN input
capacitor. These components, along with the inductor and
output capacitor, should be placed on the same side of
the circuit board, and their connections should be made
on that layer. Place a local, unbroken ground plane below
these components. The loop formed by these components
should be as small as possible.
Additionally, the SW and BOOST nodes should be kept
as small as possible. Figure 10 shows the recommended
component placement with trace and via locations.
High frequency currents, such as the high voltage input
current of the LTC4090/LTC4090-5, tend to find their way
along the ground plane on a mirror path directly beneath
the incident path on the top of the board. If there are slits or
cuts in the ground plane due to other traces on that layer,
the current will be forced to go around the slits. If high
frequency currents are not allowed to flow back through
their natural least-area path, excessive voltage will build
up and radiated emissions will occur. See Figure 11.
MINIMIZE D1, L1,
C3, U1, SW PIN LOOP
U1 THERMAL PAD
SOLDERED TO PCB.
VIAS CONNECTED TO ALL
GND PLANES WITHOUT
THERMAL RELIEF
4090 F11
MINIMIZE TRACE LENGTH
4090 F10
Figure 10. Suggested Board Layout
Figure 11. Ground Currents Follow Their Incident Path at High
Speed. Slices in the Ground Plane Cause High Voltage and Increased
Emissions.
4090fa
25
LTC4090/LTC4090-5
APPLICATIONS INFORMATION
IN and HVIN Bypass Capacitor
Battery Charger Stability Considerations
Many types of capacitors can be used for input bypassing;
however, caution must be exercised when using multilayer
ceramic capacitors. Because of the self-resonant and high
Q characteristics of some types of ceramic capacitors,
high voltage transients can be generated under some
start-up conditions, such as from connecting the charger
input to a hot power source. For more information, refer
to Application Note 88.
The constant-voltage mode feedback loop is stable without
any compensation when a battery is connected with low
impedance leads. Excessive lead length, however, may add
enough series inductance to require a bypass capacitor
of at least 1μF from BAT to GND. Furthermore, a 4.7μF
capacitor with a 0.2Ω to 1Ω series resistor to GND is
recommended at the BAT pin to keep ripple voltage low
when the battery is disconnected.
TYPICAL APPLICATIONS
HIGH
(6V TO 36V)
VOLTAGE
INPUT
HVIN
C1
1μF
50V
1206
BOOST
L1
6.8μH
0.47μF
16V
SW
C3
22μF
6.3V
1206
D1
HVEN
IN
USB
680Ω
4.7μF
6.3V
59k
1%
LTC4090
HPWR
HVOUT
VC
270pF
SUSP
0.1μF
2.1k
1%
HVPR
Q1
1k
TIMER
LOAD
OUT
4.7μF
6.3V
CLPROG
71.5k
1%
40.2k
1%
GATE
Q2
PROG
BAT
RT
+
VNTC
PG
10k
1%
Li-Ion
BATTERY
NTC
SYNC
T 10k
680Ω
D: DIODES INC. B360A
L: SUMIDA CDR6D28MN-GR5
Q1, Q2: SILICONIX Si2333DS
CHRG
4090 F12
Figure 12. 800kHz Switching Frequency
L
10μH
0.47μF
HIGH (6V TO 36V)
TRANSIENT TO 60V*
L
2.2μH
0.47μF
SW
BOOST
HVIN
4.7μF
1μF
HIGH (6V TO 16V)
VOLTAGE INPUT
HVOUT
SW
BOOST
HVIN
22μF
1μF
HVOUT
IN
USB
35k
88.7k
330pF
Q1
HVPR
LTC4090
4.7μF
1k
VC
OUT
RT
TIMER
BAT
CLPROG
0.1μF
2.1k
GND
IN
USB
1k
LOAD
VC
4.7μF
RT
TIMER
PROG
71.5k
Q1
HVPR
LTC4090
4.7μF
+
Li-Ion BATTERY
30k
GND
11.5k
330pF
L: SUMIDA CDRH8D28/HP-100
* USE SCHOTTKY DIODE RATED AT VR > 45V
CLPROG
OUT
BAT
PROG
0.1μF
2.1k
71.5k
LOAD
4.7μF
+
Li-Ion BATTERY
L: SUMIDA CDRH4D22/HP-2R2
4090 TAO4
4090 TAO3
Figure 13. 400kHz Switching Frequency
Figure 14. 2MHz Switching Frequency
4090fa
26
LTC4090/LTC4090-5
PACKAGE DESCRIPTION
DJC Package
22-Lead Plastic DFN (6mm × 3mm)
(Reference LTC DWG # 05-08-1714)
0.889
0.70 ±0.05
R = 0.10
0.889
3.60 ±0.05
1.65 ±0.05
2.20 ±0.05 (2 SIDES)
PACKAGE
OUTLINE
0.25 ± 0.05
0.50 BSC
5.35 ± 0.05
(2 SIDES)
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
NOTE:
1. DIMENSIONS ARE IN MILLIMETERS
2. APPLY SOLDER MASK TO AREAS THAT
ARE NOT SOLDERED
3. DRAWING IS NOT TO SCALE
6.00 ±0.10
(2 SIDES)
0.889
R = 0.10
TYP
3.00 ±0.10
(2 SIDES)
R = 0.115
TYP
0.40 ± 0.05
12
22
0.889
1.65 ± 0.10
(2 SIDES)
PIN 1
TOP MARK
(NOTE 6)
11
0.200 REF
1
0.25 ± 0.05
0.50 BSC
0.75 ±0.05
0.00 – 0.05
5.35 ± 0.10
(2 SIDES)
PIN #1 NOTCH
R0.30 TYP OR
0.25mm × 45°
CHAMFER
(DJC) DFN 0605
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING PROPOSED TO BE MADE VARIATION OF VERSION (WXXX)
IN JEDEC PACKAGE OUTLINE M0-229
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON TOP AND BOTTOM OF PACKAGE
4090fa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
27
LTC4090/LTC4090-5
RELATED PARTS
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DESCRIPTION
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1.2A, 750kHz Step-Down Switching
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Controller and Li-Ion Charger
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USB Power Manager with Ideal Diode
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ThinSOT is a trademark of Linear Technology Corporation.
4090fa
28 Linear Technology Corporation
LT 0208 REV A • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2007