LINER LTC1871EMS-1

LTC1871-1
Wide Input Range, No RSENSE™
Current Mode Boost,
Flyback and SEPIC Controller
DESCRIPTION
FEATURES
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High Efﬁciency (No Sense Resistor Required)
Wide Input Voltage Range: 2.5V to 36V
Current Mode Control Provides Excellent
Transient Response
High Maximum Duty Cycle (92% Typ)
±2% RUN Pin Threshold with 100mV Hysteresis
±1% Internal Voltage Reference
Ultra Low Pulse Skip Threshold for Wide Input
Range Applications
Micropower Shutdown: IQ = 10μA
Programmable Operating Frequency
(50kHz to 1MHz) with One External Resistor
Synchronizable to an External Clock Up to 1.3 × fOSC
User-Controlled Pulse Skip or Burst Mode® Operation
Internal 5.2V Low Dropout Voltage Regulator
Output Overvoltage Protection
Capable of Operating with a Sense Resistor for High
Output Voltage Applications
Small 10-Lead MSOP Package
The LTC®1871-1 is a wide input range, current mode, boost,
ﬂyback or SEPIC controller that drives an N-channel power
MOSFET and requires very few external components. It
eliminates the need for a current sense resistor by utilizing
the power MOSFET’s on-resistance, thereby maximizing
efﬁciency. Higher output voltage applications are possible
with the LTC1871-1 by connecting the SENSE pin to a
resistor in the source of the power MOSFET.
The IC’s operating frequency can be set with an external
resistor over a 50kHz to 1MHz range, and can be synchronized to an external clock using the MODE/SYNC pin.
The LTC1871-1 differs from the LTC1871 by having a
lower pulse skip threshold, making it ideal for applications requiring constant frequency operation at light
loads. The lower pulse skip threshold also helps maintain
constant frequency operation in applications with a wide
input voltage range. For applications requiring primaryto-secondary side isolation, please refer to the LTC1871
datasheet.
APPLICATIONS
The LTC1871-1 is available in the 10-lead MSOP package.
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L, LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology Corporation.
No RSENSE is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners.
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Telecom Power Supplies
Portable Electronic Equipment
TYPICAL APPLICATION
VIN
3.3V
Efﬁciency of Figure 1
L1
1μH
100
D1
ITH
RC
22k
CC1
6.8nF
CC2
47pF
R2
37.4k
1%
VIN
+
LTC1871-1
R1
12.1k
1%
FB
FREQ
RT
80.6k
1%
90
SENSE
MODE/SYNC
INTVCC
GATE
GND
CVCC
4.7μF
X5R
+
CIN
22μF
6.3V
×2
M1
COUT1
150μF
6.3V
×4
VOUT
5V
7A
(10A PEAK)
COUT2
22μF
6.3V
X5R
×2
GND
Burst Mode
OPERATION
80
EFFICIENCY (%)
RUN
70
60
PULSE-SKIP
MODE
50
40
18711 F01a
CIN:
TAIYO YUDEN JMK325BJ226MM
COUT1: PANASONIC EEFUEOJ151R
COUT2: TAIYO YUDEN JMK325BJ226MM
D1: MBRB2515L
L1: SUMIDA CEP125-H 1R0MH
M1: FAIRCHILD FDS7760A
30
0.001
0.01
0.1
1
OUTPUT CURRENT (A)
10
18711 F01b
Figure 1. High Efﬁciency 3.3V Input, 5V Output Boost Converter (Bootstrapped)
18711fb
1
LTC1871-1
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
VIN Voltage ............................................... – 0.3V to 36V
INTVCC Voltage............................................ –0.3V to 7V
INTVCC Output Current .......................................... 50mA
GATE Voltage ............................ –0.3V to VINTVCC + 0.3V
ITH, FB Voltages ....................................... –0.3V to 2.7V
RUN, MODE/SYNC Voltages ....................... –0.3V to 7V
FREQ Voltage ............................................ –0.3V to 1.5V
SENSE Pin Voltage .................................... –0.3V to 36V
Operating Junction Temperature Range (Note 2)
LTC1871E-1 ......................................... –40°C to 85°C
LTC1871I-1 ........................................ –40°C to 125°C
Junction Temperature (Note 3) ............................ 125°C
Storage Temperature Range................... –65°C to 150°C
Lead Temperature (Soldering, 10 sec) .................. 300°C
TOP VIEW
RUN
ITH
FB
FREQ
MODE/
SYNC
10
9
8
7
6
1
2
3
4
5
SENSE
VIN
INTVCC
GATE
GND
MS PACKAGE
10-LEAD PLASTIC MSOP
TJMAX = 125°C, θJA = 120°C/W
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC1871EMS-1#PBF
LTC1871EMS-1#TRPBF
LTCTV
10-Lead Plastic MSOP
–40°C to 85°C
LTC1871IMS-1#PBF
LTC1871IMS-1#TRPBF
LTCTV
10-Lead Plastic MSOP
–40°C to 125°C
LEAD BASED FINISH
TAPE AND REEL
PART MARKING
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC1871EMS-1
LTC1871EMS-1#TR
LTCTV
10-Lead Plastic MSOP
–40°C to 85°C
LTC1871IMS-1
LTC1871IMS-1#TR
LTCTV
10-Lead Plastic MSOP
–40°C to 125°C
Consult LTC Marketing for parts speciﬁed with wider operating temperature ranges.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel speciﬁcations, go to: http://www.linear.com/tapeandreel/
ELECTRICAL CHARACTERISTICS
The l denotes the speciﬁcations which apply over the full operating
temperature range, otherwise speciﬁcations are at TA = 25°C. VIN = 5V, VRUN = 1.5V, RFREQ = 80k, VMODE/SYNC = 0V, unless otherwise speciﬁed.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
Main Control Loop
VIN(MIN)
Minimum Input Voltage
I-Grade (Note 2)
IQ
Input Voltage Supply Current
Continuous Mode
V
V
550
1000
μA
550
1000
μA
250
500
μA
●
250
500
μA
10
20
μA
●
10
20
μA
●
VMODE/SYNC = 0V, VITH = 0V (Note 5)
VMODE/SYNC = 0V, VITH = 0V (Note 5),
I-Grade (Note 2)
Shutdown Mode
2.5
2.5
(Note 4)
VMODE/SYNC = 5V, VFB = 1.4V, VITH = 0.75V
VMODE/SYNC = 5V, VFB = 1.4V, VITH = 0.75V,
I-Grade (Note 2)
Burst Mode Operation, No Load
●
VRUN = 0V
VRUN = 0V, I-Grade (Note 2)
18711fb
2
LTC1871-1
ELECTRICAL CHARACTERISTICS
The l denotes the speciﬁcations which apply over the full operating
temperature range, otherwise speciﬁcations are at TA = 25°C. VIN = 5V, VRUN = 1.5V, RFREQ = 80k, VMODE/SYNC = 0V, unless otherwise speciﬁed.
SYMBOL
PARAMETER
VRUN+
Rising RUN Input Threshold Voltage
VRUN–
Falling RUN Input Threshold Voltage
VRUN(HYST)
RUN Pin Input Threshold Hysteresis
IRUN
RUN Input Current
VFB
Feedback Voltage
CONDITIONS
●
VITH = 0.4V (Note 5)
VITH = 0.4V (Note 5), I-Grade (Note 2)
IFB
FB Pin Input Current
VITH = 0.4V (Note 5)
ΔVFB
Line Regulation
2.5V ≤ VIN ≤ 30V
ΔVIN
Load Regulation
ΔVITH
TYP
MAX
1.348
I-Grade (Note 2)
ΔVFB
MIN
●
UNITS
V
1.223
1.198
1.248
1.273
1.298
50
100
150
mV
35
100
175
mV
1
60
nA
1.230
1.242
1.248
V
V
1.255
V
●
1.218
1.212
●
1.205
V
V
18
60
nA
0.002
0.02
%/V
0.002
0.03
%/V
2.5V ≤ VIN ≤ 30V, I-Grade (Note 2)
●
VMODE/SYNC = 0V, VITH = 0.5V to 0.9V (Note 5)
●
–1
–0.1
%
VMODE/SYNC = 0V, VITH = 0.5V to 0.9V (Note 5)
I-Grade (Note 2)
●
–1
–0.1
%
2.5
6
ΔVFB(OV)
ΔFB Pin, Overvoltage Lockout
VFB(OV) – VFB(NOM) in Percent
gm
Error Ampliﬁer Transconductance
ITH Pin Load = ±5μA (Note 5)
VITH(BURST)
Burst Mode Operation ITH Pin Voltage
Falling ITH Voltage (Note 5)
VSENSE(MAX)
Maximum Current Sense Input Threshold
Duty Cycle < 20%
ISENSE(ON)
SENSE Pin Current (GATE High)
VSENSE = 0V
35
ISENSE(OFF)
SENSE Pin Current (GATE Low)
VSENSE = 30V
Oscillator Frequency
RFREQ = 80k
Duty Cycle < 20%, I-Grade (Note 2)
10
650
μmho
195
120
●
150
%
mV
180
mV
200
mV
50
μA
0.1
5
μA
250
300
350
kHz
300
350
kHz
100
Oscillator
fOSC
RFREQ = 80k, I-Grade (Note 2)
●
250
50
1000
kHz
I-Grade (Note 2)
●
50
1000
kHz
87
92
97
%
I-Grade (Note 2)
●
87
92
97
%
1.25
1.30
1.25
1.30
Oscillator Frequency Range
DMAX
Maximum Duty Cycle
fSYNC/fOSC
Recommended Maximum Synchronized
Frequency Ratio
tSYNC(MIN)
MODE/SYNC Minimum Input Pulse Width
VSYNC = 0V to 5V
25
ns
tSYNC(MAX)
MODE/SYNC Maximum Input Pulse Width
VSYNC = 0V to 5V
0.8/fOSC
ns
VIL(MODE)
Low Level MODE/SYNC Input Voltage
VIH(MODE)
fOSC = 300kHz (Note 6)
fOSC = 300kHz (Note 6), I-Grade (Note 2)
●
I-Grade (Note 2)
●
I-Grade (Note 2)
●
High Level MODE/SYNC Input Voltage
RMODE/SYNC
MODE/SYNC Input Pull-Down Resistance
VFREQ
Nominal FREQ Pin Voltage
0.3
V
0.3
V
1.2
V
1.2
V
50
0.62
kΩ
V
18711fb
3
LTC1871-1
ELECTRICAL CHARACTERISTICS
The l denotes the speciﬁcations which apply over the full operating
temperature range, otherwise speciﬁcations are at TA = 25°C. VIN = 5V, VRUN = 1.5V, RFREQ = 80k, VMODE/SYNC = 0V, unless otherwise speciﬁed.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
VIN = 7.5V
5.0
5.2
5.4
5.0
UNITS
Low Dropout Regulator
VINTVCC
INTVCC Regulator Output Voltage
5.2
5.4
V
ΔVINTVCC
INTVCC Regulator Line Regulation
7.5V ≤ VIN ≤ 15V
8
25
mV
ΔVIN1
ΔVINTVCC
INTVCC Regulator Line Regulation
15V ≤ VIN ≤ 30V
70
200
mV
VLDO(LOAD)
INTVCC Load Regulation
0 ≤ IINTVCC ≤ 20mA, VIN = 7.5V
VDROPOUT
INTVCC Regulator Dropout Voltage
INTVCC Load = 20mA
280
IINTVCC
Bootstrap Mode INTVCC Supply
RUN = 0V, SENSE = 5V
10
●
VIN = 7.5V, I-Grade (Note 2)
V
ΔVIN2
–2
–0.2
●
I-Grade (Note 2)
%
mV
20
μA
30
μA
GATE Driver
tr
GATE Driver Output Rise Time
CL = 3300pF (Note 7)
17
100
ns
tf
GATE Driver Output Fall Time
CL = 3300pF (Note 7)
8
100
ns
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The LTC1871E-1 is guaranteed to meet performance speciﬁcations
from 0°C to 85°C junction temperature. Speciﬁcations over the – 40°C
to 85°C operating junction temperature range are assured by design,
characterization and correlation with statistical process controls. The
LTC1871I-1 is guaranteed over the full –40°C to 125°C operating junction
temperature range.
Note 3: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formula:
TJ = TA + (PD • 110°C/W)
Note 4: The dynamic input supply current is higher due to power MOSFET
gate charging (QG • fOSC). See Applications Information.
Note 5: The LTC1871-1 is tested in a feedback loop which servos VFB to
the reference voltage with the ITH pin forced to the midpoint of its voltage
range (0.3V ≤ VITH ≤ 1.2V, midpoint = 0.75V).
Note 6: In a synchronized application, the internal slope compensation
gain is increased by 25%. Synchronizing to a signiﬁcantly higher ratio will
reduce the effective amount of slope compensation, which could result in
subharmonic oscillation for duty cycles greater than 50%.
Note 7: Rise and fall times are measured at 10% and 90% levels.
TYPICAL PERFORMANCE CHARACTERISTICS
FB Voltage vs Temp
FB Voltage Line Regulation
FB Pin Current vs Temperature
60
1.231
1.25
50
1.23
FB PIN CURRENT (nA)
FB VOLTAGE (V)
FB VOLTAGE (V)
1.24
1.230
40
30
20
1.22
10
1.21
–50 –25
1.229
0
25 50 75 100 125 150
TEMPERATURE (°C)
18711 G01
0
5
10
15
20
VIN (V)
25
30
35
18711 G02
0
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
18711 G03
18711fb
4
LTC1871-1
TYPICAL PERFORMANCE CHARACTERISTICS
Shutdown Mode IQ vs VIN
Shutdown Mode IQ vs Temperature
20
30
Burst Mode IQ vs VIN
600
VIN = 5V
20
10
Burst Mode IQ (μA)
SHUTDOWN MODE IQ (μA)
SHUTDOWN MODE IQ (μA)
500
15
10
400
300
200
5
100
0
0
10
20
VIN (V)
30
0
–50 –25
40
0
18711 G04
10
0
20
VIN (V)
30
18711 G05
Burst Mode IQ vs Temperature
18
Gate Drive Rise and
Fall Time vs CL
60
CL = 3300pF
IQ(TOT) = 550μA + Qg • f
16
40
18711 G06
Dynamic IQ vs Frequency
500
50
400
14
12
200
40
TIME (ns)
300
IQ (mA)
Burst Mode IQ (μA)
0
25 50 75 100 125 150
TEMPERATURE (°C)
10
8
6
RISE TIME
30
20
FALL TIME
4
100
10
2
0
–50 –25
0
0
0
25 50 75 100 125 150
TEMPERATURE (°C)
0
200
400
800
600
FREQUENCY (kHz)
18711 G07
1200
RUN Thresholds vs VIN
1.2
0
10
20
VIN (V)
30
40
18711 G10
4000
6000 8000
CL (pF)
10000 12000
RT vs Frequency
1000
1.35
RT (kΩ)
RUN THRESHOLDS (V)
1.3
2000
18711 G09
RUN Thresholds vs Temperature
1.40
1.4
0
18711 G08
1.5
RUN THRESHOLDS (V)
1000
1.30
100
1.25
1.20
–50 –25
0
25 50 75 100 125 150
TEMPERATURE (°C)
18711 G11
10
0 100 200 300 400 500 600 700 800 900 1000
FREQUENCY (kHz)
18711 G12
18711fb
5
LTC1871-1
TYPICAL PERFORMANCE CHARACTERISTICS
SENSE Pin Current
vs Temperature
Maximum Sense Threshold
vs Temperature
Frequency vs Temperature
325
35
160
GATE HIGH
VSENSE = 0V
GATE FREQUENCY (kHz)
315
310
305
300
295
290
285
SENSE PIN CURRENT (μA)
MAX SENSE THRESHOLD (mV)
320
155
150
145
30
280
275
–50 –25
0
140
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
18711 G13
0
25 50 75 100 125 150
TEMPERATURE (°C)
18711 G14
INTVCC Load Regulation
18711 G15
INTVCC Dropout Voltage
vs Current, Temperature
INTVCC Line Regulation
500
5.4
VIN = 7.5V
450
5.1
DROPOUT VOLTAGE (mV)
INTVCC VOLTAGE (V)
5.2
INTVCC VOLTAGE (V)
25
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
5.3
5.2
150°C
400
125°C
350
75°C
300
25°C
250
200
0°C
150
–50°C
100
50
5.0
0
10
20
30 40
50 60
INTVCC LOAD (mA)
70
80
5.1
0
5
10
15
18711 G16
20 25
VIN (V)
30
0
35
40
18711 G17
0
5
10
15
INTVCC LOAD (mA)
20
18711 G18
PIN FUNCTIONS
RUN (Pin 1): The RUN pin provides the user with an
accurate means for sensing the input voltage and programming the start-up threshold for the converter. The
falling RUN pin threshold is nominally 1.248V and the
comparator has 100mV of hysteresis for noise immunity.
When the RUN pin is below this input threshold, the IC
is shut down and the VIN supply current is kept to a low
value (typ 10μA). The Absolute Maximum Rating for the
voltage on this pin is 7V.
ITH (Pin 2): Error Ampliﬁer Compensation Pin. The
current comparator input threshold increases with this
control voltage. Nominal voltage range for this pin is 0V
to 1.40V.
FB (Pin 3): Receives the feedback voltage from the external
resistor divider across the output. Nominal voltage for
this pin in regulation is 1.230V.
FREQ (Pin 4): A resistor from the FREQ pin to ground
programs the operating frequency of the chip. The nominal
voltage at the FREQ pin is 0.6V.
18711fb
6
LTC1871-1
PIN FUNCTIONS
MODE/SYNC (Pin 5): This input controls the operating
mode of the converter and allows for synchronizing the
operating frequency to an external clock. If the MODE/
SYNC pin is connected to ground, Burst Mode operation
is enabled. If the MODE/SYNC pin is connected to INTVCC,
or if an external logic-level synchronization signal is applied to this input, Burst Mode operation is disabled and
the IC operates in a continuous mode.
INTVCC (Pin 8): The Internal 5.20V Regulator Output.
The gate driver and control circuits are powered from
this voltage. Decouple this pin locally to the IC ground
with a minimum of 4.7μF low ESR tantalum or ceramic
capacitor.
VIN (Pin 9): Main Supply Pin. Must be closely decoupled
to ground.
SENSE (Pin 10): The Current Sense Input for the Control
Loop. Connect this pin to the drain of the power MOSFET
for VDS sensing and highest efﬁciency. Alternatively, the
SENSE pin may be connected to a resistor in the source
of the power MOSFET. Internal leading edge blanking is
provided for both sensing methods.
GND (Pin 6): Ground Pin.
GATE (Pin 7): Gate Driver Output.
BLOCK DIAGRAM
RUN
+
BIAS AND
START-UP
CONTROL
SLOPE
COMPENSATION
1
C2
–
1.248V
VIN
FREQ
V-TO-I
4
OSC
9
0.6V
IOSC
MODE/SYNC
INTVCC
5
85mV
+
1.230V
S
Q
GND
R
+
0.175V
FB
–
3
1.230V
+
7
LOGIC
OV
–
GATE
PWM LATCH
50k
EA
+
BURST
COMPARATOR
CURRENT
COMPARATOR
SENSE
+
10
C1
–
–
gm
ITH
V-TO-I
2
INTVCC
8
5.2V
ILOOP
LDO
RLOOP
1.230V
SLOPE
1.230V
–
2.00V
+
UV
TO
START-UP
CONTROL
GND
BIAS
VREF
6
18711 BD
VIN
18711fb
7
LTC1871-1
OPERATION
Main Control Loop
The LTC1871-1 is a constant frequency, current mode
controller for DC/DC boost, SEPIC and ﬂyback converter
applications. The LTC1871-1 is distinguished from conventional current mode controllers because the current
control loop can be closed by sensing the voltage drop
across the power MOSFET switch instead of across a
discrete sense resistor, as shown in Figure 2. This sensing
technique improves efﬁciency, increases power density,
and reduces the cost of the overall solution.
L
D
VIN
VOUT
VIN
+
SENSE
COUT
VSW
GATE
GND
GND
2a. SENSE Pin Connection for
Maximum Efﬁciency (VSW < 36V)
L
D
VIN
VOUT
VIN
VSW
GATE
+
SENSE
GND
GND
COUT
RS
18711 F02
2b. SENSE Pin Connection for Precise
Control of Peak Current or for VSW > 36V
Figure 2. Using the SENSE Pin On the LTC1871-1
For circuit operation, please refer to the Block Diagram of
the IC and Figure 1. In normal operation, the power MOSFET
is turned on when the oscillator sets the PWM latch and
is turned off when the current comparator C1 resets the
latch. The divided-down output voltage is compared to an
internal 1.230V reference by the error ampliﬁer EA, which
outputs an error signal at the ITH pin. The voltage on the
ITH pin sets the current comparator C1 input threshold.
When the load current increases, a fall in the FB voltage
relative to the reference voltage causes the ITH pin to rise,
which causes the current comparator C1 to trip at a higher
peak inductor current value. The average inductor current
will therefore rise until it equals the load current, thereby
maintaining output regulation.
The nominal operating frequency of the LTC1871-1 is
programmed using a resistor from the FREQ pin to ground
and can be controlled over a 50kHz to 1000kHz range. In
addition, the internal oscillator can be synchronized to
an external clock applied to the MODE/SYNC pin and can
be locked to a frequency between 100% and 130% of its
nominal value. When the MODE/SYNC pin is left open, it
is pulled low by an internal 50k resistor and Burst Mode
operation is enabled. If this pin is taken above 2V or an
external clock is applied, Burst Mode operation is disabled
and the IC operates in continuous mode. With no load (or
an extremely light load), the controller will skip pulses in
order to maintain regulation and prevent excessive output
ripple.
The RUN pin controls whether the IC is enabled or is in a low
current shutdown state. A micropower 1.248V reference
and comparator C2 allow the user to program the supply
voltage at which the IC turns on and off (comparator C2
has 100mV of hysteresis for noise immunity). With the
RUN pin below 1.248V, the chip is off and the input supply
current is typically only 10μA.
An overvoltage comparator OV senses when the FB pin
exceeds the reference voltage by 6.5% and provides a
reset pulse to the main RS latch. Because this RS latch is
reset-dominant, the power MOSFET is actively held off for
the duration of an output overvoltage condition.
The LTC1871-1 can be used either by sensing the voltage
drop across the power MOSFET or by connecting the
SENSE pin to a conventional shunt resistor in the source
of the power MOSFET, as shown in Figure 2. Sensing the
voltage across the power MOSFET maximizes converter
efﬁciency and minimizes the component count, but limits
the output voltage to the maximum rating for this pin (36V).
By connecting the SENSE pin to a resistor in the source
of the power MOSFET, the user is able to program output
voltages signiﬁcantly greater than 36V.
18711fb
8
LTC1871-1
OPERATION
Programming the Operating Mode
For applications where maximizing the efﬁciency at very
light loads (e.g., <100μA) is a high priority, the current
in the output divider could be decreased to a few microamps and Burst Mode operation should be applied (i.e.,
the MODE/SYNC pin should be connected to ground).
In applications where ﬁxed frequency operation is more
critical than low current efﬁciency, or where the lowest
output ripple is desired, pulse-skip mode operation should
be used and the MODE/SYNC pin should be connected
to the INTVCC pin. This allows discontinuous conduction
mode (DCM) operation down to near the limit deﬁned
by the chip’s minimum on-time (about 175ns). Below
this output current level, the converter will begin to skip
cycles in order to maintain output regulation. Figures 3
and 4 show the light load switching waveforms for Burst
Mode and pulse-skip mode operation for the converter
in Figure 1.
Burst Mode Operation
Burst Mode operation is selected by leaving the MODE/
SYNC pin unconnected or by connecting it to ground. In
normal operation, the range on the ITH pin corresponding to
no load to full load is 0.30V to 1.2V. In Burst Mode operation, if the error ampliﬁer EA drives the ITH voltage below
0.525V, the buffered ITH input to the current comparator
C1 will be clamped at 0.525V (which corresponds to 25%
of maximum load current). The inductor current peak is
then held at approximately 30mV divided by the power
VIN = 3.3V
VOUT = 5V
IOUT = 500mA
MOSFET RDS(ON). If the ITH pin drops below 0.175V, the
Burst Mode comparator B1 will turn off the power MOSFET
and scale back the quiescent current of the IC to 250μA
(sleep mode). In this condition, the load current will be
supplied by the output capacitor until the ITH voltage rises
above the 50mV hysteresis of the burst comparator. At
light loads, short bursts of switching (where the average
inductor current is 20% of its maximum value) followed
by long periods of sleep will be observed, thereby greatly
improving converter efﬁciency. Oscilloscope waveforms
illustrating Burst Mode operation are shown in Figure 3.
Pulse-Skip Mode Operation
With the MODE/SYNC pin tied to a DC voltage above 2V,
Burst Mode operation is disabled. The internal, 0.525V
buffered ITH burst clamp is removed, allowing the ITH
pin to directly control the current comparator from no
load to full load. With no load, the ITH pin is driven below
0.175V, the power MOSFET is turned off and sleep mode
is invoked. Oscilloscope waveforms illustrating this mode
of operation are shown in Figure 4.
When an external clock signal drives the MODE/SYNC
pin at a rate faster than the chip’s internal oscillator, the
oscillator will synchronize to it. In this synchronized mode,
Burst Mode operation is disabled. The constant frequency
associated with synchronized operation provides a more
controlled noise spectrum from the converter, at the expense of overall system efﬁciency of light loads.
MODE/SYNC = 0V
(Burst Mode OPERATION)
VIN = 3.3V
VOUT = 5V
IOUT = 500mA
VOUT
50mV/DIV
VOUT
50mV/DIV
IL
5A/DIV
IL
5A/DIV
10μs/DIV
18711 F03
Figure 3. LTC1871-1 Burst Mode Operation
(MODE/SYNC = 0V) at Low Output Current
MODE/SYNC = INTVCC
(PULSE-SKIP MODE)
2μs/DIV
18711 F04
Figure 4. LTC1871-1 Low Output Current Operation with
Burst Mode Operation Disabled (MODE/SYNC = INTVCC)
18711fb
9
LTC1871-1
APPLICATIONS INFORMATION
The external clock signal must exceed 2V for at least 25ns,
and should have a maximum duty cycle of 80%, as shown
in Figure 5. The MOSFET turn on will synchronize to the
rising edge of the external clock signal.
0.6V, and the current that ﬂows into the FREQ pin is used
to charge and discharge an internal oscillator capacitor. A
graph for selecting the value of RT for a given operating
frequency is shown in Figure 6.
1000
RT (kΩ)
When the oscillator’s internal logic circuitry detects a
synchronizing signal on the MODE/SYNC pin, the internal oscillator ramp is terminated early and the slope
compensation is increased by approximately 30%. As
a result, in applications requiring synchronization, it is
recommended that the nominal operating frequency of
the IC be programmed to be about 75% of the external
clock frequency. Attempting to synchronize to too high an
external frequency (above 1.3fO) can result in inadequate
slope compensation and possible subharmonic oscillation
(or jitter).
100
10
0 100 200 300 400 500 600 700 800 900 1000
FREQUENCY (kHz)
18711 F06
2V TO 7V
MODE/
SYNC
Figure 6. Timing Resistor (RT) Value
tMIN = 25ns
0.8T
GATE
T
T = 1/fO
D = 40%
IL
18711 F05
Figure 5. MODE/SYNC Clock Input and Switching
Waveforms for Synchronized Operation
Programming the Operating Frequency
The choice of operating frequency and inductor value is
a tradeoff between efﬁciency and component size. Low
frequency operation improves efﬁciency by reducing
MOSFET and diode switching losses. However, lower
frequency operation requires more inductance for a given
amount of load current.
The LTC1871-1 uses a constant frequency architecture that
can be programmed over a 50kHz to 1000kHz range with
a single external resistor from the FREQ pin to ground, as
shown in Figure 1. The nominal voltage on the FREQ pin is
INTVCC Regulator Bypassing and Operation
An internal, P-channel low dropout voltage regulator produces the 5.2V supply which powers the gate driver and
logic circuitry within the LTC1871-1, as shown in Figure 7.
The INTVCC regulator can supply up to 50mA and must be
bypassed to ground immediately adjacent to the IC pins
with a minimum of 4.7μF tantalum or ceramic capacitor.
Good bypassing is necessary to supply the high transient
currents required by the MOSFET gate driver.
For input voltages that don’t exceed 7V (the absolute
maximum rating for this pin), the internal low dropout
regulator in the LTC1871-1 is redundant and the INTVCC
pin can be shorted directly to the VIN pin. With the INTVCC
pin shorted to VIN, however, the divider that programs the
regulated INTVCC voltage will draw 10μA of current from
the input supply, even in shutdown mode. For applications
that require the lowest shutdown mode input supply current, do not connect the INTVCC pin to VIN. Regardless of
whether the INTVCC pin is shorted to VIN or not, it is always
necessary to have the driver circuitry bypassed with a
4.7μF tantalum or low ESR ceramic capacitor to ground
immediately adjacent to the INTVCC and GND pins.
In an actual application, most of the IC supply current is
used to drive the gate capacitance of the power MOSFET.
18711fb
10
LTC1871-1
APPLICATIONS INFORMATION
INPUT
SUPPLY
2.5V TO 30V
VIN
1.230V
–
P-CH
+
CIN
R2
R1
5.2V INTVCC
+
LOGIC
DRIVER
GATE
CVCC
4.7μF
M1
GND
18711 F07
GND
PLACE AS CLOSE AS
POSSIBLE TO DEVICE PINS
Figure 7. Bypassing the LDO Regulator and Gate Driver Supply
As a result, high input voltage applications in which a
large power MOSFET is being driven at high frequencies
can cause the LTC1871-1 to exceed its maximum junction temperature rating. The junction temperature can be
estimated using the following equations:
IQ(TOT) ≈ IQ + f • QG
PIC = VIN • (IQ + f • QG)
TJ = TA + PIC • RTH(JA)
The total quiescent current IQ(TOT) consists of the static
supply current (IQ) and the current required to charge and
discharge the gate of the power MOSFET. The 10-pin MSOP
package has a thermal resistance of RTH(JA) = 120°C/W.
As an example, consider a power supply with VIN = 5V and
VO = 12V at IO = 1A. The switching frequency is 500kHz,
and the maximum ambient temperature is 70°C. The power
MOSFET chosen is the IRF7805, which has a maximum
RDS(ON) of 11mΩ (at room temperature) and a maximum
total gate charge of 37nC (the temperature coefﬁcient of
the gate charge is low).
IQ(TOT) = 600μA + 37nC • 500kHz = 19.1mA
PIC = 5V • 19.1mA = 95mW
TJ = 70°C + 120°C/W • 95mW = 81.4°C
This demonstrates how signiﬁcant the gate charge current
can be when compared to the static quiescent current in
the IC.
To prevent the maximum junction temperature from being
exceeded, the input supply current must be checked when
operating in a continuous mode at high VIN. A tradeoff
between the operating frequency and the size of the power
MOSFET may need to be made in order to maintain a reliable
IC junction temperature. Prior to lowering the operating
frequency, however, be sure to check with power MOSFET
manufacturers for their latest-and-greatest low QG, low
RDS(ON) devices. Power MOSFET manufacturing technologies are continually improving, with newer and better
performance devices being introduced almost yearly.
Output Voltage Programming
The output voltage is set by a resistor divider according
to the following formula:
R2 VO = 1.230V • 1+ R1
The external resistor divider is connected to the output
as shown in Figure 1, allowing remote voltage sensing.
The resistors R1 and R2 are typically chosen so that the
18711fb
11
LTC1871-1
APPLICATIONS INFORMATION
error caused by the current ﬂowing into the FB pin during normal operation is less than 1% (this translates to a
maximum value of R1 of about 250k).
Programming Turn-On and Turn-Off Thresholds
with the RUN Pin
The LTC1871-1 contains an independent, micropower voltage reference and comparator detection circuit that remains
active even when the device is shut down, as shown in
Figure 8. This allows users to accurately program an input
voltage at which the converter will turn on and off. The
falling threshold voltage on the RUN pin is equal to the
internal reference voltage of 1.248V. The comparator has
100mV of hysteresis to increase noise immunity.
The turn-on and turn-off input voltage thresholds are
programmed using a resistor divider according to the
following formulas:
R2 VIN(OFF) = 1.248V • 1+ R1
R2 VIN(ON) = 1.348V • 1+ R1
The resistor R1 is typically chosen to be less than 1M.
For applications where the RUN pin is only to be used as
a logic input, the user should be aware of the 7V Absolute
Maximum Rating for this pin! The RUN pin can be connected to the input voltage through an external 1M resistor,
as shown in Figure 8c, for “always on” operation.
VIN
+
R2
RUN
+
RUN
COMPARATOR
BIAS AND
START-UP
CONTROL
6V
INPUT
SUPPLY
–
OPTIONAL
FILTER
CAPACITOR
R1
1.248V
μPOWER
REFERENCE
GND
–
18711 F8a
Figure 8a. Programming the Turn-On and Turn-Off Thresholds Using the RUN Pin
VIN
+
R2
1M
RUN
COMPARATOR
RUN
+
RUN
+
6V
INPUT
SUPPLY
–
6V
EXTERNAL
LOGIC CONTROL
1.248V
–
18711 F08b
RUN
COMPARATOR
–
GND
1.248V
18711 F08c
Figure 8b. On/Off Control Using External Logic
Figure 8c. External Pull-Up Resistor On
RUN Pin for “Always On” Operation
18711fb
12
LTC1871-1
APPLICATIONS INFORMATION
Application Circuits
A basic LTC1871-1 application circuit is shown in
Figure 1. External component selection is driven by the
characteristics of the load and the input supply. The
ﬁrst topology to be analyzed will be the boost converter,
followed by SEPIC (single ended primary inductance
converter).
Boost Converter: Duty Cycle Considerations
For a boost converter operating in a continuous conduction
mode (CCM), the duty cycle of the main switch is:
V +V –V D = O D IN VO + VD where VD is the forward voltage of the boost diode. For
converters where the input voltage is close to the output
voltage, the duty cycle is low and for converters that develop
a high output voltage from a low voltage input supply,
the duty cycle is high. The maximum output voltage for a
boost converter operating in CCM is:
VO(MAX) =
VIN(MIN)
(1– DMAX )
– VD
The maximum duty cycle capability of the LTC1871-1 is
typically 92%. This allows the user to obtain high output
voltages from low input supply voltages.
Boost Converter: The Peak and Average Input Currents
The control circuit in the LTC1871-1 is measuring the input
current (either by using the RDS(ON) of the power MOSFET
or by using a sense resistor in the MOSFET source), so
the output current needs to be reﬂected back to the input
in order to dimension the power MOSFET properly. Based
on the fact that, ideally, the output power is equal to the
input power, the maximum average input current is:
IIN(MAX) =
IO(MAX)
1– DMAX
The peak input current is:
IO(MAX)
IIN(PEAK) = 1+ •
2 1– DMAX
The maximum duty cycle, DMAX, should be calculated at
minimum VIN.
Boost Converter: Ripple Current ΔIL and the ‘χ’ Factor
The constant ‘χ’ in the equation above represents the
percentage peak-to-peak ripple current in the inductor,
relative to its maximum value. For example, if 30% ripple
current is chosen, then χ = 0.30, and the peak current is
15% greater than the average.
For a current mode boost regulator operating in CCM,
slope compensation must be added for duty cycles above
50% in order to avoid subharmonic oscillation. For the
LTC1871-1, this ramp compensation is internal. Having an
internally ﬁxed ramp compensation waveform, however,
does place some constraints on the value of the inductor
and the operating frequency. If too large an inductor is
used, the resulting current ramp (ΔIL) will be small relative
to the internal ramp compensation (at duty cycles above
50%), and the converter operation will approach voltage
mode (ramp compensation reduces the gain of the current
loop). If too small an inductor is used, but the converter
is still operating in CCM (near critical conduction mode),
the internal ramp compensation may be inadequate to
prevent subharmonic oscillation. To ensure good current
mode gain and avoid subharmonic oscillation, it is recommended that the ripple current in the inductor fall in the
range of 20% to 40% of the maximum average current.
For example, if the maximum average input current is
1A, choose a ΔIL between 0.2A and 0.4A, and a value ‘χ’
between 0.2 and 0.4.
Boost Converter: Inductor Selection
Given an operating input voltage range, and having chosen
the operating frequency and ripple current in the inductor,
the inductor value can be determined using the following
equation:
L=
VIN(MIN)
IL • f
• DMAX
where:
IL = •
IO(MAX)
1– DMAX
18711fb
13
LTC1871-1
APPLICATIONS INFORMATION
Remember that boost converters are not short-circuit
protected. Under a shorted output condition, the inductor
current is limited only by the input supply capability. For
applications requiring a step-up converter that is shortcircuit protected, please refer to the applications section
covering SEPIC converters.
The minimum required saturation current of the inductor
can be expressed as a function of the duty cycle and the
load current, as follows:
IO(MAX)
IL(SAT) 1+ •
2 1– DMAX
The saturation current rating for the inductor should be
checked at the minimum input voltage (which results
in the highest inductor current) and maximum output
current.
Boost Converter: Operating in Discontinuous Mode
Discontinuous mode operation occurs when the load current is low enough to allow the inductor current to run out
during the off-time of the switch, as shown in Figure 9.
Once the inductor current is near zero, the switch and diode
capacitances resonate with the inductance to form damped
ringing at 1MHz to 10MHz. If the off-time is long enough,
the drain voltage will settle to the input voltage.
Depending on the input voltage and the residual energy
in the inductor, this ringing can cause the drain of the
power MOSFET to go below ground where it is clamped
by the body diode. This ringing is not harmful to the IC
and it has not been shown to contribute signiﬁcantly to
EMI. Any attempt to damp it with a snubber will degrade
the efﬁciency.
Boost Converter: Inductor Core Selection
Once the value for L is known, the type of inductor must
be selected. High efﬁciency converters generally cannot
afford the core loss found in low cost powdered iron cores,
forcing the use of more expensive ferrite, molypermalloy
or Kool Mμ® cores. Actual core loss is independent of core
size for a ﬁxed inductor value, but is very dependent on
the inductance selected. As inductance increases, core
losses go down. Unfortunately, increased inductance
requires more turns of wire and therefore, copper losses
will increase. Generally, there is a tradeoff between core
losses and copper losses that needs to be balanced.
Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can
concentrate on copper losses and preventing saturation.
Ferrite core material saturates “hard,” meaning that the
inductance collapses rapidly when the peak design current
is exceeded. This results in an abrupt increase in inductor
ripple current and consequently, output voltage ripple. Do
not allow the core to saturate!
Molypermalloy (from Magnetics, Inc.) is a very good,
low cost core material for toroids, but is more expensive
than ferrite. A reasonable compromise from the same
manufacturer is Kool Mμ.
Boost Converter: Power MOSFET Selection
MOSFET DRAIN
VOLTAGE
2V/DIV
The power MOSFET serves two purposes in the LTC1871-1:
it represents the main switching element in the power path,
and its RDS(ON) represents the current sensing element
for the control loop. Important parameters for the power
MOSFET include the drain-to-source breakdown voltage
(BVDSS), the threshold voltage (VGS(TH)), the on-resistance
(RDS(ON)) versus gate-to-source voltage, the gate-to-source
and gate-to-drain charges (QGS and QGD, respectively),
the maximum drain current (ID(MAX)) and the MOSFET’s
thermal resistances (RTH(JC) and RTH(JA)).
INDUCTOR
CURRENT
2A/DIV
The gate drive voltage is set by the 5.2V INTVCC low drop
regulator. Consequently, logic-level threshold MOSFETs
should be used in most LTC1871-1 applications. If low
input voltage operation is expected (e.g., supplying power
VIN = 3.3V IOUT = 200mA
VOUT = 5V
2μs/DIV
18711 F09
Figure 9. Discontinuous Mode Waveforms
18711fb
14
LTC1871-1
APPLICATIONS INFORMATION
from a lithium-ion battery or a 3.3V logic supply), then
sublogic-level threshold MOSFETs should be used.
Pay close attention to the BVDSS speciﬁcations for the
MOSFETs relative to the maximum actual switch voltage in
the application. Many logic-level devices are limited to 30V
or less, and the switch node can ring during the turn-off of
the MOSFET due to layout parasitics. Check the switching
waveforms of the MOSFET directly across the drain and
source terminals using the actual PC board layout (not
just on a lab breadboard!) for excessive ringing.
During the switch on-time, the control circuit limits the
maximum voltage drop across the power MOSFET to about
150mV (at low duty cycle). The peak inductor current
is therefore limited to 150mV/RDS(ON). The relationship
between the maximum load current, duty cycle and the
RDS(ON) of the power MOSFET is:
RDS(ON) VSENSE(MAX) •
1– DMAX
1+ 2 •IO(MAX) • T
It is worth noting that the 1 – DMAX relationship between
IO(MAX) and RDS(ON) can cause boost converters with a
wide input range to experience a dramatic range of maximum input and output current. This should be taken into
consideration in applications where it is important to limit
the maximum current drawn from the input supply.
Calculating Power MOSFET Switching and Conduction
Losses and Junction Temperatures
In order to calculate the junction temperature of the
power MOSFET, the power dissipated by the device must
be known. This power dissipation is a function of the
duty cycle, the load current and the junction temperature
itself (due to the positive temperature coefﬁcient of its
RDS(ON)). As a result, some iterative calculation is normally
required to determine a reasonably accurate value. Since
the controller is using the MOSFET as both a switching
and a sensing element, care should be taken to ensure
that the converter is capable of delivering the required
load current over all operating conditions (line voltage
and temperature), and for the worst-case speciﬁcations
200
2.0
ρT NORMALIZED ON RESISTANCE
MAXIMUM CURRENT SENSE VOLTAGE (mV)
The VSENSE(MAX) term is typically 150mV at low duty
cycle, and is reduced to about 100mV at a duty cycle of
92% due to slope compensation, as shown in Figure 10.
The ρT term accounts for the temperature coefﬁcient of
the RDS(ON) of the MOSFET, which is typically 0.4%/°C.
Figure 11 illustrates the variation of normalized RDS(ON)
over temperature for a typical power MOSFET.
Another method of choosing which power MOSFET to
use is to check what the maximum output current is for a
given RDS(ON), since MOSFET on-resistances are available
in discrete values.
1– DMAX
IO(MAX) = VSENSE(MAX) •
1+ 2 • RDS(ON) • T
150
100
50
0
0
0.2
0.5
0.4
DUTY CYCLE
0.8
1.0
18711 F10
Figure 10. Maximum SENSE Threshold Voltage vs Duty Cycle
1.5
1.0
0.5
0
–50
50
100
0
JUNCTION TEMPERATURE (°C)
150
18711 F11
Figure 11. Normalized RDS(ON) vs Temperature
18711fb
15
LTC1871-1
APPLICATIONS INFORMATION
for VSENSE(MAX) and the RDS(ON) of the MOSFET listed in
the manufacturer’s data sheet.
The power dissipated by the MOSFET in a boost converter
is:
2
IO(MAX) PFET = • RDS(ON) • DMAX • T
1– DMAX IO(MAX)
+k • VO1.85 •
•C
•f
(1– DMAX ) RSS
The ﬁrst term in the equation above represents the
losses in the device, and the second term, the switching
losses. The constant, k = 1.7, is an empirical factor inversely
related to the gate drive current and has the dimension
of 1/current.
From a known power dissipated in the power MOSFET, its
junction temperature can be obtained using the following
formula:
TJ = TA + PFET • RTH(JA)
The RTH(JA) to be used in this equation normally includes
the RTH(JC) for the device plus the thermal resistance from
the case to the ambient temperature (RTH(CA)). This value
of TJ can then be compared to the original, assumed value
used in the iterative calculation process.
Boost Converter: Output Diode Selection
To maximize efﬁciency, a fast switching diode with low
forward drop and low reverse leakage is desired. The output
diode in a boost converter conducts current during the
switch off-time. The peak reverse voltage that the diode
must withstand is equal to the regulator output voltage.
The average forward current in normal operation is equal
to the output current, and the peak current is equal to the
peak inductor current.
IO(MAX)
ID(PEAK) =IL(PEAK) = 1+ •
2 1– D
The power dissipated by the diode is:
PD = IO(MAX) • VD
and the diode junction temperature is:
Remember to keep the diode lead lengths short and to
observe proper switch-node layout (see Board Layout
Checklist) to avoid excessive ringing and increased
dissipation.
Boost Converter: Output Capacitor Selection
I2R
MAX
The RTH(JA) to be used in this equation normally includes
the RTH(JC) for the device plus the thermal resistance from
the board to the ambient temperature in the enclosure.
Contributions of ESR (equivalent series resistance), ESL
(equivalent series inductance) and the bulk capacitance
must be considered when choosing the correct component
for a given output ripple voltage. The effects of these three
parameters (ESR, ESL and bulk C) on the output voltage
ripple waveform are illustrated in Figure 12e for a typical
boost converter.
The choice of component(s) begins with the maximum
acceptable ripple voltage (expressed as a percentage of
the output voltage), and how this ripple should be divided
between the ESR step and the charging/discharging ΔV.
For the purpose of simplicity we will choose 2% for the
maximum output ripple, to be divided equally between the
ESR step and the charging/discharging ΔV. This percentage ripple will change, depending on the requirements
of the application, and the equations provided below can
easily be modiﬁed.
For a 1% contribution to the total ripple voltage, the ESR
of the output capacitor can be determined using the following equation:
ESRCOUT 0.01• VO
IIN(PEAK)
where:
IO(MAX)
IIN(PEAK)= 1+ •
2 1– DMAX
For the bulk C component, which also contributes 1% to
the total ripple:
COUT IO(MAX)
0.01• VO • f
TJ = TA + PD • RTH(JA)
18711fb
16
LTC1871-1
APPLICATIONS INFORMATION
For many designs it is possible to choose a single capacitor
type that satisﬁes both the ESR and bulk C requirements
for the design. In certain demanding applications, however,
the ripple voltage can be improved signiﬁcantly by connecting two or more types of capacitors in parallel. For
example, using a low ESR ceramic capacitor can minimize
the ESR step, while an electrolytic capacitor can be used
to supply the required bulk C.
Once the output capacitor ESR and bulk capacitance have
been determined, the overall ripple voltage waveform
should be veriﬁed on a dedicated PC board (see Board
Layout section for more information on component placement). Lab breadboards generally suffer from excessive
series inductance (due to inter-component wiring), and
these parasitics can make the switching waveforms look
signiﬁcantly worse than they would be on a properly
designed PC board.
The output capacitor in a boost regulator experiences high
RMS ripple currents, as shown in Figure 12. The RMS
output capacitor ripple current is:
IRMS(COUT) IO(MAX) •
tested for use in switching power supplies. An excellent
choice is AVX TPS series of surface mount tantalum. Also,
ceramic capacitors are now available with extremely low
ESR, ESL and high ripple current ratings.
Boost Converter: Input Capacitor Selection
The input capacitor of a boost converter is less critical
than the output capacitor, due to the fact that the inductor
is in series with the input and the input current waveform
is continuous (see Figure 12b). The input voltage source
impedance determines the size of the input capacitor,
L
VIN
D
SW
VOUT
COUT
RL
12a. Circuit Diagram
IIN
IL
VO – VIN(MIN)
VIN(MIN)
Note that the ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life. This
makes it advisable to further derate the capacitor or to
choose a capacitor rated at a higher temperature than
required. Several capacitors may also be placed in parallel
to meet size or height requirements in the design.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest product of
ESR and size of any aluminum electrolytic, at a somewhat
higher price.
In surface mount applications, multiple capacitors may
have to be placed in parallel in order to meet the ESR or
RMS current handling requirements of the application.
Aluminum electrolytic and dry tantalum capacitors are
both available in surface mount packages. In the case of
tantalum, it is critical that the capacitors have been surge
12b. Inductor and Input Currents
ISW
tON
12c. Switch Current
ID
tOFF
IO
12d. Diode and Output Currents
ΔVCOUT
VOUT
(AC)
ΔVESR
RINGING DUE TO
TOTAL INDUCTANCE
(BOARD + CAP)
12e. Output Voltage Ripple Waveform
Figure 12. Switching Waveforms for a Boost Converter
18711fb
17
LTC1871-1
APPLICATIONS INFORMATION
Table 1. Recommended Component Manufacturers
VENDOR
AVX
BH Electronics
COMPONENTS
TELEPHONE
WEB ADDRESS
Capacitors
(207) 282-5111
avxcorp.com
Inductors, Transformers
(952) 894-9590
bhelectronics.com
Coilcraft
Inductors
(847) 639-6400
coilcraft.com
Coiltronics
Inductors
(407) 241-7876
coiltronics.com
Diodes, Inc
Diodes
(805) 446-4800
diodes.com
MOSFETs
(408) 822-2126
fairchildsemi.com
Fairchild
General Semiconductor
Diodes
(516) 847-3000
generalsemiconductor.com
International Rectiﬁer
MOSFETs, Diodes
(310) 322-3331
irf.com
IRC
Sense Resistors
(361) 992-7900
irctt.com
Kemet
Tantalum Capacitors
(408) 986-0424
kemet.com
Toroid Cores
(800) 245-3984
mag-inc.com
Microsemi
Diodes
(617) 926-0404
microsemi.com
Murata-Erie
Inductors, Capacitors
(770) 436-1300
murata.co.jp
Capacitors
(847) 843-7500
nichicon.com
Magnetics Inc
Nichicon
On Semiconductor
Diodes
(602) 244-6600
onsemi.com
Panasonic
Capacitors
(714) 373-7334
panasonic.com
Sanyo
Capacitors
(619) 661-6835
sanyo.co.jp
Sumida
Inductors
(847) 956-0667
sumida.com
Taiyo Yuden
Capacitors
(408) 573-4150
t-yuden.com
TDK
Capacitors, Inductors
(562) 596-1212
component.tdk.com
Thermalloy
Heat Sinks
(972) 243-4321
aavidthermalloy.com
Tokin
Capacitors
(408) 432-8020
nec-tokinamerica.com
Toko
Inductors
(847) 699-3430
tokoam.com
United Chemicon
Capacitors
(847) 696-2000
chemi-com.com
Vishay/Dale
Resistors
(605) 665-9301
vishay.com
Vishay/Siliconix
MOSFETs
(800) 554-5565
vishay.com
Vishay/Sprague
Capacitors
(207) 324-4140
vishay.com
Small-Signal Discretes
(631) 543-7100
zetex.com
Zetex
which is typically in the range of 10μF to 100μF. A low ESR
capacitor is recommended, although it is not as critical as
for the output capacitor.
The RMS input capacitor ripple current for a boost converter is:
VIN(MIN)
IRMS(CIN) = 0.3 •
• DMAX
L•f
Please note that the input capacitor can see a very high
surge current when a battery is suddenly connected to
the input of the converter and solid tantalum capacitors
can fail catastrophically under these conditions. Be sure
to specify surge-tested capacitors!
Burst Mode Operation and Considerations
The choice of MOSFET RDS(ON) and inductor value also
determines the load current at which the LTC1871-1 enters Burst Mode operation. When bursting, the controller
clamps the peak inductor current to approximately:
IBURST(PEAK) =
30mV
RDS(ON)
which represents about 20% of the maximum 150mV
SENSE pin voltage. The corresponding average current
depends upon the amount of ripple current. Lower inductor
values (higher ΔIL) will reduce the load current at which
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18
LTC1871-1
APPLICATIONS INFORMATION
Burst Mode operations begins, since it is the peak current
that is being clamped.
The output voltage ripple can increase during Burst Mode
operation if ΔIL is substantially less than IBURST. This can
occur if the input voltage is very low or if a very large
inductor is chosen. At high duty cycles, a skipped cycle
causes the inductor current to quickly decay to zero.
However, because ΔIL is small, it takes multiple cycles
for the current to ramp back up to IBURST(PEAK). During this inductor charging interval, the output capacitor
must supply the load current and a signiﬁcant droop in
the output voltage can occur. Generally, it is a good idea
to choose a value of inductor ΔIL between 25% and 40%
of IIN(MAX). The alternative is to either increase the value
of the output capacitor or disable Burst Mode operation
using the MODE/SYNC pin.
Burst Mode operation can be defeated by connecting the
MODE/SYNC pin to a high logic-level voltage (either with
a control input or by connecting this pin to INTVCC). In
this mode, the burst clamp is removed, and the chip can
operate at constant frequency from continuous conduction
mode (CCM) at full load, down into deep discontinuous
conduction mode (DCM) at light load. Prior to skipping
pulses at very light load (i.e., < 5% of full load), the
controller will operate with a minimum switch on-time
in DCM. Pulse skipping prevents a loss of control of
the output at very light loads and reduces output voltage ripple.
Efﬁciency Considerations: How Much Does VDS
Sensing Help?
The efﬁciency of a switching regulator is equal to the
output power divided by the input power (×100%).
Percent efﬁciency can be expressed as:
% Efﬁciency = 100% – (L1 + L2 + L3 + …),
where L1, L2, etc. are the individual loss components
as a percentage of the input power. It is often useful to
analyze individual losses to determine what is limiting
the efﬁciency and which change would produce the most
improvement. Although all dissipative elements in the
circuit produce losses, four main sources usually account
for the majority of the losses in LTC1871-1 application circuits:
1. The supply current into VIN. The VIN current is the
sum of the DC supply current IQ (given in the Electrical
Characteristics) and the MOSFET driver and control
currents. The DC supply current into the VIN pin is typically about 550μA and represents a small power loss
(much less than 1%) that increases with VIN. The driver
current results from switching the gate capacitance
of the power MOSFET; this current is typically much
larger than the DC current. Each time the MOSFET is
switched on and then off, a packet of gate charge QG
is transferred from INTVCC to ground. The resulting
dQ/dt is a current that must be supplied to the INTVCC
capacitor through the VIN pin by an external supply. If
the IC is operating in CCM:
IQ(TOT) ≈ IQ = f • QG
PIC = VIN • (IQ + f • QG)
2. Power MOSFET switching and conduction losses. The
technique of using the voltage drop across the power
MOSFET to close the current feedback loop was chosen
because of the increased efﬁciency that results from
not having a sense resistor. The losses in the power
MOSFET are equal to:
2
IO(MAX) PFET = • RDS(ON) • DMAX • T
1– DMAX IO(MAX)
+k • VO1.85 •
•C
•f
(1– DMAX ) RSS
The I2R power savings that result from not having a
discrete sense resistor can be calculated almost by
inspection.
2
IO(MAX) PR(SENSE) = • RSENSE • DMAX
1– DMAX To understand the magnitude of the improvement with
this VDS sensing technique, consider the 3.3V input,
5V output power supply shown in Figure 1. The maximum load current is 7A (10A peak) and the duty cycle
is 39%. Assuming a ripple current of 40%, the peak
inductor current is 13.8A and the average is 11.5A.
With a maximum sense voltage of about 140mV, the
sense resistor value would be 10mΩ, and the power
18711fb
19
LTC1871-1
APPLICATIONS INFORMATION
dissipated in this resistor would be 514mW at maximum output current. Assuming an efﬁciency of 90%,
this sense resistor power dissipation represents 1.3%
of the overall input power. In other words, for this application, the use of VDS sensing would increase the
efﬁciency by approximately 1.3%.
For more details regarding the various terms in these
equations, please refer to the section Boost Converter:
Power MOSFET Selection.
IOUT
2V/DIV
VIN = 3.3V
VOUT = 5V
MODE/SYNC = INTVCC
(PULSE-SKIP MODE)
VOUT (AC)
100mV/DIV
100μs/DIV
3. The losses in the inductor are simply the DC input current squared times the winding resistance. Expressing
this loss as a function of the output current yields:
18711 F13
Figure 13. Load Transient Response for a 3.3V Input,
5V Output Boost Converter Application, 0.7A to 7A Step
2
IO(MAX) PR(WINDING) = • RW
1– DMAX 4. Losses in the boost diode. The power dissipation in the
boost diode is:
PDIODE = IO(MAX) • VD
The boost diode can be a major source of power loss
in a boost converter. For the 3.3V input, 5V output at
7A example given above, a Schottky diode with a 0.4V
forward voltage would dissipate 2.8W, which represents
7% of the input power. Diode losses can become signiﬁcant at low output voltages where the forward voltage
is a signiﬁcant percentage of the output voltage.
5. Other losses, including CIN and CO ESR dissipation and
inductor core losses, generally account for less than
2% of the total additional loss.
Checking Transient Response
The regulator loop response can be veriﬁed by looking at
the load transient response. Switching regulators generally
take several cycles to respond to an instantaneous step
in resistive load current. When the load step occurs, VO
immediately shifts by an amount equal to (ΔILOAD)(ESR),
and then CO begins to charge or discharge (depending on
the direction of the load step) as shown in Figure 13. The
regulator feedback loop acts on the resulting error amp
output signal to return VO to its steady-state value. During
this recovery time, VO can be monitored for overshoot or
ringing that would indicate a stability problem.
A second, more severe transient can occur when connecting loads with large (> 1μF) supply bypass capacitors.
The discharged bypass capacitors are effectively put in
parallel with CO, causing a nearly instantaneous drop in
VO. No regulator can deliver enough current to prevent
this problem if the load switch resistance is low and it is
driven quickly. The only solution is to limit the rise time
of the switch drive in order to limit the inrush current
di/dt to the load.
Boost Converter Design Example
The design example given here will be for the circuit shown
in Figure 1. The input voltage is 3.3V, and the output is 5V
at a maximum load current of 7A (10A peak).
1. The duty cycle is:
V + V – V 5 + 0.4 – 3.3
D = O D IN =
= 38.9%
5 + 0.4
VO + VD 2. Pulse-skip operation is chosen so the MODE/SYNC pin
is shorted to INTVCC.
3. The operating frequency is chosen to be 300kHz to
reduce the size of the inductor. From Figure 5, the
resistor from the FREQ pin to ground is 80k.
4. An inductor ripple current of 40% of the maximum load
current is chosen, so the peak input current (which is
also the minimum saturation current) is:
7
IO(MAX)
IIN(PEAK) = 1+ •
= 1.2 •
= 13.8A
2 1– DMAX
1– 0.39
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20
LTC1871-1
APPLICATIONS INFORMATION
The inductor ripple current is:
IO(MAX)
7
IL = •
= 0.4 •
= 4.6A
1– DMAX
1– 0.39
And so the inductor value is:
VIN(MIN)
3.3V
L=
• DMAX =
• 0.39 = 0.93μH
IL • f
4.6A • 300kHz
The component chosen is a 1μH inductor made by
Sumida (part number CEP125-H 1ROMH) which has
a saturation current of greater than 20A.
5. With the input voltage to the IC bootstrapped to the
output of the power supply (5V), a logic-level MOSFET
can be used. Because the duty cycle is 39%, the maximum SENSE pin threshold voltage is reduced from its
low duty cycle typical value of 150mV to approximately
140mV. Assuming a MOSFET junction temperature of
125°C, the room temperature MOSFET RDS(ON) should
be less than:
1– DMAX
RDS(ON) VSENSE(MAX) •
1+ 2 •IO(MAX) • T
1– 0.39
= 0.140V •
= 6.8m
0.4 1+ 2 • 7A • 1.5
The MOSFET used was the Fairchild FDS7760A, which
has a maximum RDS(ON) of 8mΩ at 4.5V VGS, a BVDSS
of greater than 30V, and a gate charge of 37nC at 5V
VGS.
6. The diode for this design must handle a maximum
DC output current of 10A and be rated for a minimum
reverse voltage of VOUT, or 5V. A 25A, 15V diode from
On Semiconductor (MBRB2515L) was chosen for its
high power dissipation capability.
ESR ceramic. Based on a maximum output ripple voltage
of 1%, or 50mV, the bulk C needs to be greater than:
COUT IOUT(MAX)
0.01• VOUT • f
=
7A
= 466μF
0.01• 5V • 300kHz
The RMS ripple current rating for this capacitor needs
to exceed:
VO – VIN(MIN)
IRMS(COUT) IO(MAX) •
=
VIN(MIN)
7A •
5V – 3.3V
= 5A
3.3V
To satisfy this high RMS current demand, four 150μF
Panasonic capacitors (EEFUEOJ151R) are required.
In parallel with these bulk capacitors, two 22μF, low
ESR (X5R) Taiyo Yuden ceramic capacitors (JMK325BJ226MM) are added for HF noise reduction.
Check the output ripple with a single oscilloscope
probe connected directly across the output capacitor
terminals, where the HF switching currents ﬂow.
8. The choice of an input capacitor for a boost converter
depends on the impedance of the source supply and the
amount of input ripple the converter will safely tolerate. For this particular design and lab setup a 100μF
Sanyo Poscap (6TPC 100M), in parallel with two 22μF
Taiyo Yuden ceramic capacitors (JMK325BJ226MM)
is required (the input and return lead lengths are kept
to a few inches, but the peak input current is close to
20A!). As with the output node, check the input ripple
with a single oscilloscope probe connected across the
input capacitor terminals.
7. The output capacitor usually consists of a high valued
bulk C connected in parallel with a lower valued, low
18711fb
21
LTC1871-1
APPLICATIONS INFORMATION
PC Board Layout Checklist
the power MOSFET or the bottom terminal of the sense
resistor, 4) the negative terminal of the input capacitor
and 5) at least one via to the ground plane immediately
adjacent to Pin 6. The ground trace on the top layer of
the PC board should be as wide and short as possible
to minimize series resistance and inductance.
1. In order to minimize switching noise and improve output
load regulation, the GND pin of the LTC1871-1 should
be connected directly to 1) the negative terminal of the
INTVCC decoupling capacitor, 2) the negative terminal
of the output decoupling capacitors, 3) the source of
VIN
L1
JUMPER
R3
RC
R4
CC
J1
CIN
PIN 1
R2
LTC1871-1
R1
RT
CVCC
PSEUDO-KELVIN
SIGNAL GROUND
CONNECTION
SWITCH NODE IS ALSO
THE HEAT SPREADER
FOR L1, M1, D1
M1
COUT
COUT
D1
VIAS TO GROUND
PLANE
VOUT
TRUE REMOTE
OUTPUT SENSING
BULK C
18711 F14
LOW ESR CERAMIC
Figure 14. LTC1871-1 Boost Converter Suggested Layout
VIN
R3
CC
R1
R2
R4
1
RC
2
SENSE
VIN
ITH
J1
SWITCH
NODE
9
LTC1871-1
3
4
RT
RUN
L1
10
5
FB
FREQ
INTVCC
GATE
MODE/
SYNC
GND
D1
8
7
M1
6
+
CVCC
CIN
GND
BOLD LINES INDICATE HIGH CURRENT PATHS
+
PSEUDO-KELVIN
GROUND CONNECTION
COUT
VOUT
18711 F15
Figure 15. LTC1871-1 Boost Converter Layout Diagram
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22
LTC1871-1
APPLICATIONS INFORMATION
2. Beware of ground loops in multiple layer PC boards.
Try to maintain one central ground node on the board
and use the input capacitor to avoid excess input ripple
for high output current power supplies. If the ground
plane is to be used for high DC currents, choose a path
away from the small-signal components.
7. If a sense resistor is used in the source of the power
MOSFET, minimize the capacitance between the SENSE
pin trace and any high frequency switching nodes. The
LTC1871-1 contains an internal leading edge blanking
time of approximately 180ns, which should be adequate
for most applications.
3. Place the CVCC capacitor immediately adjacent to the
INTVCC and GND pins on the IC package. This capacitor carries high di/dt MOSFET gate drive currents. A
low ESR and ESL 4.7μF ceramic capacitor works well
here.
8. For optimum load regulation and true remote sensing,
the top of the output resistor divider should connect
independently to the top of the output capacitor (Kelvin
connection), staying away from any high dV/dt traces.
Place the divider resistors near the LTC1871-1 in order
to keep the high impedance FB node short.
+
VIN
D1
SW
VOUT
+
L2
RL
COUT
•
16a. SEPIC Topology
VIN
+
•
VOUT
+
VIN
RL
•
16b. Current Flow During Switch On-Time
VIN
+
•
D1
+
6. Place the small-signal components away from high
frequency switching nodes. In the layout shown in
Figure 14, all of the small-signal components have
been placed on one side of the IC and all of the power
components have been placed on the other. This also
allows the use of a pseudo-Kelvin connection for the
signal ground, where high di/dt gate driver currents
ﬂow out of the IC ground pin in one direction (to the
bottom plate of the INTVCC decoupling capacitor) and
small-signal currents ﬂow in the other direction.
C1
L1
•
+
5. Check the stress on the power MOSFET by measuring
its drain-to-source voltage directly across the device
terminals (reference the ground of a single scope probe
directly to the source pad on the PC board). Beware
of inductive ringing which can exceed the maximum
speciﬁed voltage rating of the MOSFET. If this ringing
cannot be avoided and exceeds the maximum rating
of the device, either choose a higher voltage device
or specify an avalanche-rated power MOSFET. Not all
MOSFETs are created equal (some are more equal than
others).
9. For applications with multiple switching power converters connected to the same input supply, make sure
that the input ﬁlter capacitor for the LTC1871-1 is not
shared with other converters. AC input current from
another converter could cause substantial input voltage ripple, and this could interfere with the operation
of the LTC1871-1. A few inches of PC trace or wire
(L ≈ 100nH) between the CIN of the LTC1871-1 and the
actual source VIN should be sufﬁcient to prevent current
sharing problems.
+
4. The high di/dt loop from the bottom terminal of the
output capacitor, through the power MOSFET, through
the boost diode and back through the output capacitors
should be kept as tight as possible to reduce inductive
ringing. Excess inductance can cause increased stress
on the power MOSFET and increase HF noise on the
output. If low ESR ceramic capacitors are used on the
output to reduce output noise, place these capacitors
close to the boost diode in order to keep the series
inductance to a minimum.
VOUT
+
VIN
RL
•
16c. Current Flow During Switch Off-Time
Figures 16. SEPIC Topology and Current Flow
18711fb
23
LTC1871-1
APPLICATIONS INFORMATION
SEPIC Converter Applications
The LTC1871-1 is also well suited to SEPIC (single-ended
primary inductance converter) converter applications. The
SEPIC converter shown in Figure 16 uses two inductors.
The advantage of the SEPIC converter is the input voltage
may be higher or lower than the output voltage, and the
output is short-circuit protected.
The ﬁrst inductor, L1, together with the main switch,
resembles a boost converter. The second inductor, L2,
together with the output diode D1, resembles a ﬂyback or
buck-boost converter. The two inductors L1 and L2 can be
IL1
IIN
SW
ON
SW
OFF
17a. Input Inductor Current
IO
IL2
independent but can also be wound on the same core since
identical voltages are applied to L1 and L2 throughout the
switching cycle. By making L1 = L2 and winding them on
the same core the input ripple is reduced along with cost
and size. All of the SEPIC applications information that
follows assumes L1 = L2 = L.
SEPIC Converter: Duty Cycle Considerations
For a SEPIC converter operating in a continuous conduction
mode (CCM), the duty cycle of the main switch is:
VO + VD D= VIN + VO + VD where VD is the forward voltage of the diode. For converters where the input voltage is close to the output voltage
the duty cycle is near 50%.
The maximum output voltage for a SEPIC converter is:
VO(MAX) = ( VIN + VD )
17b. Output Inductor Current
DMAX
1
– VD
1– DMAX
1– DMAX
IIN
The maximum duty cycle of the LTC1871-1 is typically
92%.
IO
SEPIC Converter: The Peak and Average
Input Currents
IC1
17c. DC Coupling Capacitor Current
ID1
IO
17d. Diode Current
VOUT
(AC)
ΔVCOUT
The control circuit in the LTC1871-1 is measuring the input
current (either using the RDS(ON) of the power MOSFET
or by means of a sense resistor in the MOSFET source),
so the output current needs to be reﬂected back to the
input in order to dimension the power MOSFET properly.
Based on the fact that, ideally, the output power is equal
to the input power, the maximum input current for a SEPIC
converter is:
D
IIN(MAX) =IO(MAX) • MAX
1– DMAX
The peak input current is:
ΔVESR
RINGING DUE TO
TOTAL INDUCTANCE
(BOARD + CAP)
17e. Output Ripple Voltage
Figure 17. SEPIC Converter Switching Waveforms
D
IIN(PEAK) = 1+ •IO(MAX) • MAX
2
1– DMAX
The maximum duty cycle, DMAX, should be calculated at
minimum VIN.
18711fb
24
LTC1871-1
APPLICATIONS INFORMATION
The constant ‘χ’ represents the fraction of ripple current in
the inductor relative to its maximum value. For example, if
30% ripple current is chosen, then χ = 0.30 and the peak
current is 15% greater than the average.
It is worth noting here that SEPIC converters that operate
at high duty cycles (i.e., that develop a high output voltage from a low input voltage) can have very high input
currents, relative to the output current. Be sure to check
that the maximum load current will not overload the input
supply.
SEPIC Converter: Inductor Selection
For most SEPIC applications the equal inductor values
will fall in the range of 10μH to 100μH. Higher values will
reduce the input ripple voltage and reduce the core loss.
Lower inductor values are chosen to reduce physical size
and improve transient response.
Like the boost converter, the input current of the SEPIC
converter is calculated at full load current and minimum
input voltage. The peak inductor current can be signiﬁcantly
higher than the output current, especially with smaller inductors and lighter loads. The following formulas assume
CCM operation and calculate the maximum peak inductor
currents at minimum VIN:
V +V
IL1(PEAK) = 1+ •IO(MAX) • O D
2
VIN(MIN)
VIN(MIN) + VD
IL2(PEAK) = 1+ •IO(MAX) •
2
VIN(MIN)
The ripple current in the inductor is typically 20% to 40%
(i.e., a range of ‘χ’ from 0.20 to 0.40) of the maximum
average input current occurring at VIN(MIN) and IO(MAX) and
ΔIL1 = ΔIL2. Expressing this ripple current as a function of
the output current results in the following equations for
calculating the inductor value:
VIN(MIN)
• DMAX
L=
IL • f
where:
IL = •IO(MAX) •
DMAX
1– DMAX
By making L1 = L2 and winding them on the same core,
the value of inductance in the equation above is replace
by 2L due to mutual inductance. Doing this maintains the
same ripple current and energy storage in the inductors. For
example, a Coiltronix CTX10-4 is a 10μH inductor with two
windings. With the windings in parallel, 10μH inductance is
obtained with a current rating of 4A (the number of turns
hasn’t changed, but the wire diameter has doubled). Splitting the two windings creates two 10μH inductors with a
current rating of 2A each. Therefore, substituting 2L yields
the following equation for coupled inductors:
VIN(MIN)
L1= L2 =
•D
2 • IL • f MAX
Specify the maximum inductor current to safely handle
IL(PK) speciﬁed in the equation above. The saturation
current rating for the inductor should be checked at the
minimum input voltage (which results in the highest
inductor current) and maximum output current.
SEPIC Converter: Power MOSFET Selection
The power MOSFET serves two purposes in the LTC1871-1:
it represents the main switching element in the power path,
and its RDS(ON) represents the current sensing element
for the control loop. Important parameters for the power
MOSFET include the drain-to-source breakdown voltage
(BVDSS), the threshold voltage (VGS(TH)), the on-resistance
(RDS(ON)) versus gate-to-source voltage, the gate-to-source
and gate-to-drain charges (QGS and QGD, respectively),
the maximum drain current (ID(MAX)) and the MOSFET’s
thermal resistances (RTH(JC) and RTH(JA)).
The gate drive voltage is set by the 5.2V INTVCC low dropout
regulator. Consequently, logic-level threshold MOSFETs
should be used in most LTC1871-1 applications. If low
input voltage operation is expected (e.g., supplying power
from a lithium-ion battery), then sublogic-level threshold
MOSFETs should be used.
The maximum voltage that the MOSFET switch must
sustain during the off-time in a SEPIC converter is equal
to the sum of the input and output voltages (VO + VIN).
As a result, careful attention must be paid to the BVDSS
speciﬁcations for the MOSFETs relative to the maximum
actual switch voltage in the application. Many logic-level
18711fb
25
LTC1871-1
APPLICATIONS INFORMATION
devices are limited to 30V or less. Check the switching
waveforms directly across the drain and source terminals
of the power MOSFET to ensure the VDS remains below
the maximum rating for the device.
During the MOSFET’s on-time, the control circuit limits
the maximum voltage drop across the power MOSFET to
about 150mV (at low duty cycle). The peak inductor current
is therefore limited to 150mV/RDS(ON). The relationship
between the maximum load current, duty cycle and the
RDS(ON) of the power MOSFET is:
RDS(ON) VSENSE(MAX)
IO(MAX)
•
1
1
•
V +V 1+ 2 • T O D + 1
VIN(MIN) The VSENSE(MAX) term is typically 150mV at low duty cycle
and is reduced to about 100mV at a duty cycle of 92% due
to slope compensation, as shown in Figure 8. The constant
‘χ’ in the denominator represents the ripple current in the
inductors relative to their maximum current. For example,
if 30% ripple current is chosen, then χ = 0.30. The ρT term
accounts for the temperature coefﬁcient of the RDS(ON) of
the MOSFET, which is typically 0.4%/°C. Figure 9 illustrates
the variation of normalized RDS(ON) over temperature for
a typical power MOSFET.
Another method of choosing which power MOSFET to
use is to check what the maximum output current is for a
given RDS(ON) since MOSFET on-resistances are available
in discrete values.
IO(MAX) VSENSE(MAX)
RDS(ON)
•
1
1
•
V +V 1+ 2 • T O D + 1
VIN(MIN) Calculating Power MOSFET Switching and Conduction
Losses and Junction Temperatures
In order to calculate the junction temperature of the
power MOSFET, the power dissipated by the device must
be known. This power dissipation is a function of the
duty cycle, the load current and the junction temperature
itself. As a result, some iterative calculation is normally
required to determine a reasonably accurate value. Since
the controller is using the MOSFET as both a switching
and a sensing element, care should be taken to ensure
that the converter is capable of delivering the required
load current over all operating conditions (load, line and
temperature) and for the worst-case speciﬁcations for
VSENSE(MAX) and the RDS(ON) of the MOSFET listed in the
manufacturer’s data sheet.
The power dissipated by the MOSFET in a SEPIC converter
is:
2
D
PFET = IO(MAX) • MAX • RDS(ON) • DMAX • T
1– DMAX (
+ k • VIN(MIN) + VO
•C
•f
)1.85 •IO(MAX) • 1–DMAX
DMAX RSS
The ﬁrst term in the equation above represents the I2R
losses in the device and the second term, the switching
losses. The constant k = 1.7 is an empirical factor inversely
related to the gate drive current and has the dimension
of 1/current.
From a known power dissipated in the power MOSFET, its
junction temperature can be obtained using the following
formula:
TJ = TA + PFET •RTH(JA)
The RTH(JA) to be used in this equation normally includes
the RTH(JC) for the device plus the thermal resistance from
the board to the ambient temperature in the enclosure.
This value of TJ can then be used to check the original
assumption for the junction temperature in the iterative
calculation process.
SEPIC Converter: Output Diode Selection
To maximize efﬁciency, a fast-switching diode with low
forward drop and low reverse leakage is desired. The output
diode in a SEPIC converter conducts current during the
switch off-time. The peak reverse voltage that the diode
must withstand is equal to VIN(MAX) + VO. The average
forward current in normal operation is equal to the output
current, and the peak current is equal to:
V +V
ID(PEAK) = 1+ •IO(MAX) • O D + 1
2
VIN(MIN) The power dissipated by the diode is:
PD = IO(MAX) • VD
18711fb
26
LTC1871-1
APPLICATIONS INFORMATION
and the diode junction temperature is:
SEPIC Converter: Output Capacitor Selection
For many designs it is possible to choose a single capacitor
type that satisﬁes both the ESR and bulk C requirements
for the design. In certain demanding applications, however,
the ripple voltage can be improved signiﬁcantly by connecting two or more types of capacitors in parallel. For
example, using a low ESR ceramic capacitor can minimize
the ESR step, while an electrolytic or tantalum capacitor
can be used to supply the required bulk C.
Because of the improved performance of today’s electrolytic, tantalum and ceramic capacitors, engineers need
to consider the contributions of ESR (equivalent series
resistance), ESL (equivalent series inductance) and the
bulk capacitance when choosing the correct component
for a given output ripple voltage. The effects of these three
parameters (ESR, ESL, and bulk C) on the output voltage
ripple waveform are illustrated in Figure 17 for a typical
coupled-inductor SEPIC converter.
Once the output capacitor ESR and bulk capacitance have
been determined, the overall ripple voltage waveform
should be veriﬁed on a dedicated PC board (see Board
Layout section for more information on component placement). Lab breadboards generally suffer from excessive
series inductance (due to inter-component wiring), and
these parasitics can make the switching waveforms look
signiﬁcantly worse than they would be on a properly
designed PC board.
The choice of component(s) begins with the maximum
acceptable ripple voltage (expressed as a percentage of
the output voltage), and how this ripple should be divided
between the ESR step and the charging/discharging ΔV.
For the purpose of simplicity we will choose 2% for the
maximum output ripple, to be divided equally between the
ESR step and the charging/discharging ΔV. This percentage
ripple will change, depending on the requirements of the
application, and the equations provided below can easily
be modiﬁed.
The output capacitor in a SEPIC regulator experiences
high RMS ripple currents, as shown in Figure 17. The
RMS output capacitor ripple current is:
TJ = TA + PD • RTH(JA)
The RTH(JA) to be used in this equation normally includes
the RTH(JC) for the device plus the thermal resistance from
the board to the ambient temperature in the enclosure.
For a 1% contribution to the total ripple voltage, the ESR
of the output capacitor can be determined using the following equation:
ESRCOUT 0.01• VO
IIN(PEAK)
where:
V +V
ID(PEAK) = 1+ •IO(MAX) • O D + 1
2
VIN(MIN) For the bulk C component, which also contributes 1% to
the total ripple:
COUT IO(MAX)
0.01• VO • f
IRMS(C1) =IO(MAX) •
VO + VD
VIN(MIN)
Note that the ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life. This
makes it advisable to further derate the capacitor or to
choose a capacitor rated at a higher temperature than
required. Several capacitors may also be placed in parallel
to meet size or height requirements in the design.
Manufacturers such as Nichicon, United Chemicon and
Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric
capacitor available from Sanyo has the lowest product of
ESR and size of any aluminum electrolytic, at a somewhat
higher price.
In surface mount applications, multiple capacitors may
have to be placed in parallel in order to meet the ESR or
RMS current handling requirements of the application.
Aluminum electrolytic and dry tantalum capacitors are
both available in surface mount packages. In the case of
tantalum, it is critical that the capacitors have been surge
tested for use in switching power supplies. An excellent
18711fb
27
LTC1871-1
APPLICATIONS INFORMATION
choice is AVX TPS series of surface mount tantalum. Also,
ceramic capacitors are now available with extremely low
ESR, ESL and high ripple current ratings.
SEPIC Converter: Input Capacitor Selection
The input capacitor of a SEPIC converter is less critical
than the output capacitor due to the fact that an inductor
is in series with the input and the input current waveform
is triangular in shape. The input voltage source impedance
determines the size of the input capacitor which is typically in the range of 10μF to 100μF. A low ESR capacitor
is recommended, although it is not as critical as for the
output capacitor.
The RMS input capacitor ripple current for a SEPIC converter is:
1
• IL
IRMS(CIN) =
12
Please note that the input capacitor can see a very high
surge current when a battery is suddenly connected to
the input of the converter and solid tantalum capacitors
can fail catastrophically under these conditions. Be sure
to specify surge-tested capacitors!
SEPIC Converter: Selecting the DC Coupling Capacitor
The coupling capacitor C1 in Figure 16 sees nearly a rectangular current waveform as shown in Figure 17. During the
switch off-time the current through C1 is IO(VO/VIN) while
approximately – IO ﬂows during the on-time. This current
waveform creates a triangular ripple voltage on C1:
VC1(PP) =
IO(MAX)
C1• f
•
VO
VIN + VO + VD
The maximum voltage on C1 is then:
VC1(MAX) = VIN +
VC1(PP)
2
which is typically close to VIN(MAX). The ripple current
through C1 is:
IRMS(C1) =IO(MAX) •
VO + VD
VIN(MIN)
The value chosen for the DC coupling capacitor normally
starts with the minimum value that will satisfy 1) the RMS
current requirement and 2) the peak voltage requirement
(typically close to VIN). Low ESR ceramic and tantalum
capacitors work well here.
SEPIC Converter Design Example
The design example given here will be for the circuit shown
in Figure 18. The input voltage is 5V to 15V and the output
is 12V at a maximum load current of 1.5A (2A peak).
1. The duty cycle range is:
VO + VD = 45.5% to 71.4%
D= VIN + VO + VD 2. The operating mode chosen is pulse skipping, so the
MODE/SYNC pin is shorted to INTVCC.
3. The operating frequency is chosen to be 300kHz to
reduce the size of the inductors; the resistor from the
FREQ pin to ground is 80k.
4. An inductor ripple current of 40% is chosen, so the peak
input current (which is also the minimum saturation
current) is:
V +V
IL1(PEAK) = 1+ •IO(MAX) • O D
2
VIN(MIN)
12 + 0.5
0.4 = 1+
• 1.5 •
= 4.5A
2 5
The inductor ripple current is:
DMAX
1– DMAX
0.714
= 0.4 • 1.5 •
= 1.5A
1– 0.714
IL = •IO(MAX) •
18711fb
28
LTC1871-1
APPLICATIONS INFORMATION
And so the inductor value is:
VIN(MIN)
5
L=
• DMAX =
• 0.714 = 4μH
2 • IL • f
2 • 1.5 • 300k
1
1
•
IO(MAX)
V +V 1+ 2 • T O D + 1
VIN(MIN) 1
1
0.12
= 12.7m
•
•
=
1.5 1.2 • 1.5 12.5 5 + 1
The component chosen is a BH Electronics BH5101007, which has a saturation current of 8A.
5. With an minimum input voltage of 5V, only logic-level
power MOSFETs should be considered. Because the
maximum duty cycle is 71.4%, the maximum SENSE
pin threshold voltage is reduced from its low duty
cycle typical value of 150mV to approximately 120mV.
Assuming a MOSFET junction temperature of 125°C,
the room temperature MOSFET RDS(ON) should be less
than:
2
RC
33k
CC1
R1
6.8nF 12.1k
1%
R2
105k
1%
L1*
SENSE
RUN
VIN
ITH
4
RT
80.6k
1%
5
FB
10
CDC
10μF
25V
X5R
D1
FREQ
GATE
MODE/SYNC
GND
VOUT
12V
1.5A
(2A PEAK)
9
+
INTVCC
8
7
M1
6
CVCC
4.7μF
X5R
+
CIN
47μF
L2*
COUT1
47μF
20V
×2
COUT2
10μF
25V
X5R
×2
•
GND
18711 F018a
CIN, COUT1: KEMET T495X476K020AS
CDC, COUT2: TAIYO YUDEN TMK432BJ106MM
D1:
INTERNATIONAL RECTIFIER 30BQ040
L1, L2: BH ELECTRONICS BH510-1007 (*COUPLED INDUCTORS)
M1:
INTERNATIONAL RECTIFIER IRF7811W
Figure 18a. 4.5V to 15V Input, 12V/2A Output SEPIC Converter
100
95
90
85
EFFICIENCY (%)
CC2
47pF
VIN
4.5V to 15V
•
LTC1871-1
3
•
For a SEPIC converter, the switch BVDSS rating must be
greater than VIN(MAX) + VO, or 27V. This comes close to
an IRF7811W, which is rated to 30V, and has a maximum
room temperature RDS(ON) of 12mΩ at VGS = 4.5V.
R3
1M
1
VSENSE(MAX)
RDS(ON) 80
75
VIN = 12V
VIN = 4.5V
VIN = 15V
70
65
60
55
50
45
0.001
VO = 12V
MODE = INTVCC
0.01
0.1
1
OUTPUT CURRENT (A)
10
18711 F18b
Figure 18b. SEPIC Efﬁciency vs Output Current
18711fb
29
LTC1871-1
APPLICATIONS INFORMATION
VIN = 4.5V
VOUT = 12V
VIN = 15V
VOUT = 12V
VOUT (AC)
200mV/DIV
VOUT (AC)
200mV/DIV
IOUT
0.5A/DIV
IOUT
0.5A/DIV
50μs/DIV
50μs/DIV
18711 F19
Figure 19. LTC1871-1 SEPIC Converter Load Step Response
6. The diode for this design must handle a maximum
DC output current of 2A and be rated for a minimum
reverse voltage of VIN + VOUT, or 27V. A 3A, 40V diode
from International Rectiﬁer (30BQ040) is chosen for its
small size, relatively low forward drop and acceptable
reverse leakage at high temp.
7. The output capacitor usually consists of a high valued
bulk C connected in parallel with a lower valued, low ESR
ceramic. Based on a maximum output ripple voltage of
1%, or 120mV, the bulk C needs to be greater than:
IOUT(MAX)
COUT =
0.01• VOUT • f
1.5A
= 41μF
0.01• 12V • 300kHz
The RMS ripple current rating for this capacitor needs
to exceed:
VO
IRMS(COUT) IO(MAX) •
=
VIN(MIN)
1.5A •
12V
= 2.3A
5V
To satisfy this high RMS current demand, two 47μF
Kemet capacitors (T495X476K020AS) are required. As a
result, the output ripple voltage is a low 50mV to 60mV.
In parallel with these tantalums, two 10μF, low ESR (X5R)
Taiyo Yuden ceramic capacitors (TMK432BJ106MM) are
added for HF noise reduction. Check the output ripple
with a single oscilloscope probe connected directly
across the output capacitor terminals, where the HF
switching currents ﬂow.
8. The choice of an input capacitor for a SEPIC converter
depends on the impedance of the source supply and the
amount of input ripple the converter will safely tolerate. For this particular design and lab setup, a single
47μF Kemet tantalum capacitor (T495X476K020AS) is
adequate. As with the output node, check the input ripple
with a single oscilloscope probe connected across the
input capacitor terminals. If any HF switching noise is
observed it is a good idea to decouple the input with
a low ESR, X5R ceramic capacitor as close to the VIN
and GND pins as possible.
9. The DC coupling capacitor in a SEPIC converter is chosen based on its RMS current requirement and must be
rated for a minimum voltage of VIN plus the AC ripple
voltage. Start with the minimum value which satisﬁes
the RMS current requirement and then check the ripple
voltage to ensure that it doesn’t exceed the DC rating.
IRMS(CI) IO(MAX) •
= 1.5A •
VO + VD
VIN(MIN)
12V + 0.5V
= 2.4A
5V
For this design a single 10μF, low ESR (X5R) Taiyo
Yuden ceramic capacitor (TMK432BJ106MM) is
adequate.
18711fb
30
LTC1871-1
TYPICAL APPLICATIONS
2.5V to 3.3V Input, 5V/2A Output Boost Converter
VIN
2.5V to 3.3V
L1
1.8μH
D1
1
2
RC
22k
VIN
ITH
10
CC1
R1
6.8nF 12.1k
1%
R2
37.4k
1%
4
RT
80.6k
1%
5
FB
+
INTVCC
FREQ
MODE/SYNC
GATE
GND
VOUT
5V
2A
9
LTC1871-1
3
8
7
6
M1
CVCC
4.7μF
X5R
+
CIN
47μF
6.3V
COUT1
150μF
6.3V
×2
COUT2
10μF
6.3V
X5R
×2
GND
18711 TA01a
CIN:
COUT1:
COUT2:
CVCC:
SANYO POSCAP 6TPA47M
SANYO POSCAP 6TPB150M
TAIYO YUDEN JMK316BJ106ML
TAIYO YUDEN LMK316BJ475ML
D1: INTERNATIONAL RECTIFIER 30BQ015
L1: TOKO DS104C2 B952AS-1R8N
M1: SILICONIX/VISHAY Si9426
Output Efﬁciency at 2.5V and 3.3V Input
100
95
90
EFFICIENCY (%)
CC2
47pF
SENSE
RUN
85
80
75
70
65
60
55
50
0.001
0.01
0.1
1
OUTPUT CURRENT (A)
10
18711 TA01b
18711fb
31
LTC1871-1
TYPICAL APPLICATIONS
18V to 27V Input, 28V Output, 400W 2-Phase, Low Ripple, Synchronized RF Base Station Power Supply (Boost)
VIN
18V to 27V
R1
93.1k
1%
L1
5.6μH
1
2
CC1
47pF
SENSE
RUN
VIN
ITH
10
RT1
150k
5%
CFB1
47pF
5
FB
INTVCC
FREQ
GATE
MODE/SYNC
GND
1
2
D1
VIN
ITH
7
M1
6
CVCC1
4.7μF
X5R
CIN2
2.2μF
35V
X5R
CFB2
47pF
R3
12.1k
1%
R4
CC3
261k
6.8nF 1%
4
RT2
150k
5%
5
FB
INTVCC
FREQ
GATE
MODE/SYNC
GND
+
COUT2
330μF
50V
RS1
0.007Ω
1W
10
GND
COUT5*
330μF
50V
×4
L4
5.6μH
COUT6*
2.2μF
35V
X5R
D2
VOUT
28V
14A
9
LTC1871-1
3
COUT1
2.2μF
35V
X5R
×3
8
L3
5.6μH
SENSE
RUN
CIN1
330μF
50V
9
EXT CLOCK
INPUT (200kHz)
CC2
47pF
+
LTC1871-1
3
4
RC
22k
L2
5.6μH
+
R2
8.45k
1%
8
7
M2
6
CVCC2
4.7μF
X5R
CIN3
2.2μF
35V
X5R
COUT3
2.2μF +
35V
X5R
×3
L5*
0.3μH
COUT4
330μF
50V
*L5, COUT5 AND
COUT6 ARE AN
OPTIONAL SECONDARY
FILTER TO REDUCE
OUTPUT RIPPLE FROM
<500mVP-P TO <100mVP-P
RS2
0.007Ω
1W
18711 TA04
CIN1:
CIN2, 3:
COUT2, 4, 5:
COUT1, 3, 6:
CVCC1, 2:
SANYO 50MV330AX
TAIYO YUDEN GMK325BJ225MN
SANYO 50MV330AX
TAIYO YUDEN GMK325BJ225MN
TAIYO YUDEN LMK316BJ475ML
L1 TO L4:
L5:
D1, D2:
M1, M2:
SUMIDA CEP125-5R6MC-HD
SUMIDA CEP125-0R3NC-ND
ON SEMICONDUCTOR MBR2045CT
INTERNATIONAL RECTIFIER IRLZ44NS
5V to 12V Input, ±12V/0.2A Output SEPIC Converter with Undervoltage Lockout
R2
54.9k
1%
R1
127k
1%
L1*
1
2
RC
22k
CC2
100pF
VIN
5V to 12V
•
SENSE
RUN
VIN
ITH
10
CDC1
4.7μF
16V
X5R
D1
VOUT1
12V
0.4A
9
LTC1871-1
3
CC1
R4
6.8nF 127Ω
1%
R3
1.10k
1%
4
RT
60.4k
1%
5
FB
INTVCC
FREQ
GATE
MODE/SYNC
NOTE:
1. VIN UVLO+ = 4.47V
VIN UVLO– = 4.14V
GND
COUT1
4.7μF
16V
X5R
×3
8
7
6
L2*
M1
CVCC
4.7μF
10V
X5R
CIN1
1μF
16V
X5R
+
CIN2
47μF
16V
AVX
D1, D2: MBS120T3
L1 TO L3: COILTRONICS VP1-0076 (*COUPLED INDUCTORS)
M1:
SILICONIX/VISHAY Si4840
•
RS
0.02Ω
CDC2
4.7μF
16V
X5R
D2
L3*
•
GND
COUT2
4.7μF
16V
X5R
VOUT2
×3
–12V
18711 TA03
0.4A
18711fb
32
LTC1871-1
TYPICAL APPLICATIONS
4.5V to 28V Input, 5V/2A Output SEPIC Converter with Undervoltage Lockout and Soft-Start
R2
54.9k
1%
R1
115k
1%
C1
4.7nF
•
L1*
1
RC
12k
SENSE
RUN
2
VIN
ITH
10
VIN
4.5V to 28V
CDC
2.2μF
25V
X5R
×3
D1
+
LTC1871-1
3
CC1
8.2nF
R3
154k
1%
CC2
47pF
R4
49.9k
1%
FB
4
RT
162k
1%
INTVCC
FREQ
5
GATE
MODE/SYNC
GND
VOUT
5V
2A
(3A TO 4A PEAK)
9
8
7
6
CVCC
4.7μF
10V
X5R
CIN1
2.2μF
35V
X5R
M1
+
CIN2
22μF
35V
L2*
•
COUT1
330μF
6.3V
COUT2
22μF
6.3V
X5R
GND
18711 TA02a
R5
100Ω
Q1
C2
1μF
X5R
NOTES:
1. VIN UVLO+ = 4.17V
VIN UVLO– = 3.86V
2. SOFT-START dVOUT/dt = 5V/6ms
R6
750Ω
CIN1, CDC:
CIN2:
COUT1:
COUT2:
CVCC:
TAIYO YUDEN GMK325BJ225MN
AVX TPSE226M035R0300
SANYO 6TPB330M
TAIYO YUDEN JMK325BJ226MN
LMK316BJ475ML
D1:
L1, L2:
M1:
Q1:
INTERNATIONAL RECTIFIER 30BQ040
BH ELECTRONICS BH510-1007 (*COUPLED INDUCTORS)
SILICONIX/VISHAY Si4840
PHILIPS BC847BF
Soft-Start
Load Step Response at VIN = 4.5V
VOUT
100mV/DIV
(AC)
VOUT
1V/DIV
IOUT
1A/DIV
(DC)
2.2A
0.5A
18711 TA02b
1ms/DIV
250μs/DIV
18711 TA02c
Load Step Response at VIN = 28V
VOUT
100mV/DIV
(AC)
IOUT
1A/DIV
(DC)
2.2A
0.5A
250μs/DIV
18711 TA02d
18711fb
33
LTC1871-1
TYPICAL APPLICATIONS
5V to 15V Input, – 5V/5A Output Positive-to-Negative Converter with Undervoltage Lockout and Level-Shifted Feedback
•
R1
154k
1%
R2
68.1k C1
1% 1nF
2
SENSE
RUN
VIN
ITH
4
5
CC2
330pF
FB
FREQ
MODE/SYNC
INTVCC
GATE
GND
9
COUT
100μF
6.3V
X5R
×2
CDC
22μF
25V
X7R
M1
8
7
6
CIN
47μF
16V
X5R
CVCC
4.7μF
10V
X5R
RT
80.6k
1%
R4
10k
1%
6
1
–
R3
10k
1%
L2*
10
LTC1871-1
3
RC
10k
CC1
10nF
•
L1*
1
VIN
5V to 15V
VOUT
–5V
5A
4
D1
GND
C2
10nF
R5
40.2k
1%
LT1783
3
18711 TA05
+
2
CIN:
CDC:
COUT:
CVCC:
TDK C5750X5R1C476M
TDK C5750X7R1E226M
TDK C5750X5R0J107M
TAIYO YUDEN LMK316BJ475ML
D1:
ON SEMICONDUCTOR MBRB2035CT
L1, L2: COILTRONICS VP5-0053 (*3 WINDINGS IN PARALLEL
FOR THE PRIMARY, 3 IN PARALLEL FOR SECONDARY)
M1:
INTERNATIONAL RECTIFIER IRF7822
18711fb
34
LTC1871-1
PACKAGE DESCRIPTION
MS Package
10-Lead Plastic MSOP
(Reference LTC DWG # 05-08-1661 Rev E)
0.889 ± 0.127
(.035 ± .005)
5.23
(.206)
MIN
3.20 – 3.45
(.126 – .136)
3.00 ± 0.102
(.118 ± .004)
(NOTE 3)
0.50
0.305 ± 0.038
(.0197)
(.0120 ± .0015)
BSC
TYP
RECOMMENDED SOLDER PAD LAYOUT
0.254
(.010)
10 9 8 7 6
3.00 ± 0.102
(.118 ± .004)
(NOTE 4)
4.90 ± 0.152
(.193 ± .006)
DETAIL “A”
0.497 ± 0.076
(.0196 ± .003)
REF
0° – 6° TYP
GAUGE PLANE
1 2 3 4 5
0.53 ± 0.152
(.021 ± .006)
DETAIL “A”
0.86
(.034)
REF
1.10
(.043)
MAX
0.18
(.007)
SEATING
PLANE
0.17 – 0.27
(.007 – .011)
TYP
0.50
(.0197)
BSC
0.1016 ± 0.0508
(.004 ± .002)
MSOP (MS) 0307 REV E
NOTE:
1. DIMENSIONS IN MILLIMETER/(INCH)
2. DRAWING NOT TO SCALE
3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS.
MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS.
INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE
5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX
18711fb
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
35
LTC1871-1
TYPICAL APPLICATION
High Power SLIC Supply with Undervoltage Lockout
(Also See the LTC3704 Data Sheet)
GND
•
D2
10BQ060
4
R1
49.9k
1%
VIN
7V TO 12V
R2
150k
1%
CR
1nF
+
•
T1*
1, 2, 3
D4
10BQ060
C5
10μF
25V
X5R
•
•
FB
INTVCC
FREQ
GATE
MODE/SYNC
GND
IRL2910
+
f = 200kHz
C1
4.7μF
X5R
*COILTRONICS VP5-0155
(PRIMARY = 3 WINDINGS IN PARALLEL)
C2
4.7μF
50V
X5R
6
RF1
10k
1%
RS
0.012Ω
COUT
3.3μF
100V
6
10k
1
VOUT2
–72V
200mA
RF2
196k
1%
4
+
LTC1871-1
RT
120k
C4
10μF
25V
X5R
VIN
ITH
CC1
1nF
D3
10BQ060
VOUT1
–24V
200mA
SENSE
RUN
RC
82k
5
–
CC2
100pF
CIN
220μF
16V
TPS
C3
10μF
25V
X5R
3
LT1783
C8
0.1μF
2
18711 TA06
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
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Current Mode PWM Controller
300kHz Fixed Frequency, Boost, SEPIC, Flyback Topology
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Current Mode DC/DC Controller
SO-8; 300kHz Operating Frequency; Buck, Boost, SEPIC Design;
VIN Up to 36V
LTC1700
No RSENSE Synchronous Step-Up Controller
Up to 95% Efﬁciency, Operation as Low as 0.9V Input
LTC1871
Wide Input Range Controller
No RSENSE, 5V Gate Drive, Current Mode Control
®
LTC1871-7
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LTC1872
SOT-23 Boost Controller
Delivers Up to 5A, 550kHz Fixed Frequency, Current Mode
LT1930
1.2MHz, SOT-23 Boost Converter
Up to 34V Output, 2.6V ≤ VIN ≤ 16V, Miniature Design
LT1931
Inverting 1.2MHz, SOT-23 Converter
Positive-to-Negative DC/DC Conversion, Miniature Design
LTC3401/LTC3402
1A/2A 3MHz Synchronous Boost Converters
Up to 97% Efﬁciency, Very Small Solution, 0.5V ≤ VIN ≤ 5V
LTC3704
Positive-to-Negative DC/DC Controller
No RSENSE, Current Mode Control, 50kHz to 1MHz
LT3782
2-Phase Step-Up DC/DC Controller
6V ≤ VIN ≤ 40V; 4A Gate Drive, 150kHz to 500kHz
18711fb
36 Linear Technology Corporation
LT 0108 REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507
●
www.linear.com
© LINEAR TECHNOLOGY CORPORATION 2007

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