LTC3631 High Efficiency, High Voltage 100mA Synchronous Step-Down Converter DESCRIPTION FEATURES n n n n n n n n n n n n n n Wide Input Voltage Range: Operation from 4.5V to 45V Overvoltage Lockout Provides Protection Up to 60V Internal High Side and Low Side Power Switches No Compensation Required 100mA Output Current Low Dropout Operation: 100% Duty Cycle Low Quiescent Current: 12μA 0.8V ±1% Feedback Voltage Reference Adjustable Peak Current Limit Internal and External Soft-Start Precise RUN Pin Threshold with Adjustable Hysteresis 3.3V, 5V and Adjustable Output Versions Only Three External Components Required for Fixed Output Versions Low Profile (0.75mm) 3mm × 3mm DFN and Thermally-Enhanced MS8E Packages APPLICATIONS n n n n n n 4mA to 20mA Current Loops Industrial Control Supplies Distributed Power Systems Portable Instruments Battery-Operated Devices Automotive Power Systems The LTC®3631 is a high voltage, high efficiency step-down DC/DC converter with internal high side and synchronous power switches that draws only 12μA typical DC supply current at no load while maintaining output voltage regulation. The LTC3631 can supply up to 100mA load current and features a programmable peak current limit that provides a simple method for optimizing efficiency in lower current applications. The LTC3631’s combination of Burst Mode® operation, integrated power switches, low quiescent current, and programmable peak current limit provides high efficiency over a broad range of load currents. With its wide 4.5V to 45V input range and internal overvoltage monitor capable of protecting the part from 60V surges, the LTC3631 is a robust converter suited for regulating a wide variety of power sources. Additionally, the LTC3631 includes a precise run threshold and soft-start feature to guarantee that the power system start-up is well-controlled in any environment. The LTC3631 is available in the thermally enhanced 3mm × 3mm DFN and MS8E packages. L, LT, LTC, LTM, Burst Mode, Linear Technology and the Linear logo are registered trademarks and ThinSOT is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. TYPICAL APPLICATION Efficiency and Power Loss vs Load Current 100 90 5V, 100mA Step-Down Converter EFFICIENCY 1000 VIN SW LTC3631-5 RUN VOUT SS HYST ISET GND 3631 TA01a 70 100 60 POWER LOSS 50 10 POWER LOSS (mW) VIN 5V TO 45V 2.2μF VOUT 5V 10μF 100mA EFFICIENCY (%) 80 100μH 40 30 20 0.1 VIN = 12V VIN = 36V 1 10 LOAD CURRENT (mA) 1 100 3631 TA01b 3631fb 1 LTC3631 ABSOLUTE MAXIMUM RATINGS (Note 1) VIN Supply Voltage ..................................... –0.3V to 60V SW Voltage (DC) ........................... –0.3V to (VIN + 0.3V) RUN Voltage .............................................. –0.3V to 60V HYST, ISET, SS Voltages ............................... –0.3V to 6V VFB ............................................................... –0.3V to 6V VOUT (Fixed Output Versions)....................... –0.3V to 6V Operating Junction Temperature Range (Note 2).................................................. –40°C to 125°C Storage Temperature Range................... –65°C to 150°C Lead Temperature (Soldering, 10 sec) MS8E ................................................................ 300°C PIN CONFIGURATION TOP VIEW TOP VIEW SW VIN ISET SS 1 2 3 4 9 GND 8 7 6 5 SW 1 GND HYST VOUT/VFB RUN VIN 2 ISET 3 9 GND SS 4 8 GND 7 HYST 6 VOUT/VFB 5 RUN MS8E PACKAGE 8-LEAD PLASTIC MSOP DD PACKAGE 8-LEAD (3mm s 3mm) PLASTIC DFN TJMAX = 125°C, θJA = 40°C/W, θJC = 5°-10°C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB TJMAX = 125°C, θJA = 43°C/W, θJC = 3°C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3631EMS8E#PBF LTC3631EMS8E#TRPBF LTFDT 8-Lead Plastic MSOP –40°C to 125°C LTC3631EMS8E-3.3#PBF LTC3631EMS8E-3.3#TRPBF LTFFP 8-Lead Plastic MSOP –40°C to 125°C LTC3631EMS8E-5#PBF LTC3631EMS8E-5#TRPBF LTFFR 8-Lead Plastic MSOP –40°C to 125°C LTC3631IMS8E#PBF LTC3631IMS8E#TRPBF LTFDT 8-Lead Plastic MSOP –40°C to 125°C LTC3631IMS8E-3.3#PBF LTC3631IMS8E-3.3#TRPBF LTFFP 8-Lead Plastic MSOP –40°C to 125°C LTC3631IMS8E-5#PBF LTC3631IMS8E-5#TRPBF LTFFR 8-Lead Plastic MSOP –40°C to 125°C LTC3631EDD#PBF LTC3631EDD#TRPBF LFDV 8-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LTC3631EDD-3.3#PBF LTC3631EDD-3.3#TRPBF LFFN 8-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LTC3631EDD-5#PBF LTC3631EDD-5#TRPBF LFFQ 8-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LTC3631IDD#PBF LTC3631IDD#TRPBF LFDV 8-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LTC3631IDD-3.3#PBF LTC3631IDD-3.3#TRPBF LFFN 8-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C LTC3631IDD-5#PBF LTC3631IDD-5#TRPBF LFFQ 8-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3631fb 2 LTC3631 ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are for TA = 25°C (Note 2). VIN = 10V, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Input Supply (VIN) VIN Input Voltage Operating Range 4.5 UVLO VIN Undervoltage Lockout VIN Rising VIN Falling Hysteresis OVLO VIN Overvoltage Lockout VIN Rising VIN Falling Hysteresis IQ DC Supply Current (Note 3) Active Mode Sleep Mode Shutdown Mode l l 45 V 3.80 3.75 4.15 4.00 150 4.50 4.35 47 45 50 48 2 52 50 V V V 125 12 3 220 22 6 μA μA μA VRUN = 0V V V mV Output Supply (VOUT/VFB) VOUT Output Voltage Trip Thresholds LTC3631-3.3V, VOUT Rising LTC3631-3.3V, VOUT Falling l l 3.260 3.240 3.310 3.290 3.360 3.340 V V LTC3631-5V, VOUT Rising LTC3631-5V, VOUT Falling l l 4.940 4.910 5.015 4.985 5.090 5.060 V V VFB Rising l 0.792 0.800 0.808 l 3 5 7 mV –10 0 10 nA VFB Feedback Comparator Trip Voltage VHYST Feedback Comparator Hysteresis IFB Feedback Pin Current Adjustable Output Version, VFB = 1V ΔVLINEREG Feedback Voltage Line Regulation VIN = 4.5V to 45V LTC3631-5, VIN = 6V to 45V VRUN Run Pin Threshold Voltage RUN Rising RUN Falling Hysteresis 1.17 1.06 1.21 1.10 110 1.25 1.14 V V mV IRUN Run Pin Leakage Current RUN = 1.3V –10 0 10 nA VHYSTL Hysteresis Pin Voltage Low RUN < 1V, IHYST = 1mA 0.07 0.1 V IHYST Hysteresis Pin Leakage Current VHYST = 1.3V –10 0 10 nA ISS Soft-Start Pin Pull-Up Current VSS < 1.5V 4.5 5.5 6.5 tINTSS Internal Soft-Start Time SS Pin Floating IPEAK Peak Current Trip Threshold ISET Floating 500k Resistor from ISET to GND ISET Shorted to GND 0.001 V %/V Operation RON Power Switch On-Resistance Top Switch Bottom Switch ISW = –25mA ISW = 25mA Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: The LTC3631 is tested under pulsed load conditions such that TJ ≈ TA. The LTC3631E is guaranteed to meet specifications from 0°C to 85°C junction temperature. Specifications over the –40°C to 125°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3631I is guaranteed over the full –40°C to 125°C operating junction 0.75 l 200 40 225 120 50 3.0 1.5 μA ms 280 65 mA mA mA Ω Ω temperature range. Note that the maximum ambient temperature consistent with these specifications is determined by specific operating conditions in conjunction with board layout, the rated package thermal impedance and other environmental factors. The junction temperature (TJ, in °C) is calculated from the ambient temperature (TA, in °C) and power dissipation (PD, in Watts) according to the formula: TJ = TA + (PD • θJA) where θJA (in °C/W) is the package thermal impedance. Note 3: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. 3631fb 3 LTC3631 TYPICAL PERFORMANCE CHARACTERISTICS Efficiency vs Load Current, VOUT = 5V 95 95 VOUT = 5V FIGURE 10 CIRCUIT 90 Efficiency vs Load Current, VOUT = 3.3V VIN = 12V 90 VOUT = 3.3V FIGURE 10 CIRCUIT 90 VOUT = 2.5V 85 FIGURE 10 CIRCUIT VIN = 12V VIN = 24V 75 80 VIN = 24V 70 65 65 1 10 LOAD CURRENT (mA) VIN = 36V 75 70 60 0.1 VIN = 24V 70 65 55 60 0.1 100 VIN = 36V 75 60 1 10 LOAD CURRENT (mA) 50 0.1 100 Efficiency vs Input Voltage 0.20 VOUT = 5V FIGURE 10 CIRCUIT Load Regulation 5.05 ILOAD = 100mA FIGURE 10 CIRCUIT VIN = 10 VOUT = 5V FIGURE 10 CIRCUIT 5.04 90 ILOAD = 1mA 75 OUTPUT VOLTAGE (V) ILOAD = 10mA 80 5.03 0.10 ΔVOUT/VOUT (%) EFFICIENCY (%) ILOAD = 100mA 100 3631 G03 Line Regulation 85 1 10 LOAD CURRENT (mA) 3631 G02 3631 G01 95 EFFICIENCY (%) EFFICIENCY (%) VIN = 36V 80 VIN = 12V 80 85 85 EFFICIENCY (%) Efficiency vs Load Current, VOUT = 2.5V 0 –0.10 5.02 5.01 5.00 4.99 4.98 70 4.97 65 10 15 30 25 35 20 INPUT VOLTAGE (V) 40 45 4.96 –0.20 5 10 15 20 25 30 35 INPUT VOLTAGE (V) 3631 G04 0.799 –10 20 50 80 TEMPERATURE (°C) 3631 G06 Peak Current Trip Threshold vs Temperature and ISET 110 LTC1144 • TPC06 5.6 250 VIN = 10V PEAK CURRENT TRIP THRESHOLD (mA) FEEDBACK COMPARATOR HYSTERESIS (mV) FEEDBACK COMPARATOR TRIP VOLTAGE (V) 0.800 10 20 30 40 50 60 70 80 90 100 LOAD CURRENT (mA) 0 Feedback Comparator Hysteresis vs Temperature VIN = 10V 0.798 –40 45 3631 G05 Feedback Comparator Trip Voltage vs Temperature 0.801 40 5.4 5.2 5.0 4.8 4.6 4.4 –40 –10 20 50 80 TEMPERATURE (°C) 110 3631 G08 VIN = 10V 225 ISET = OPEN 200 175 150 125 RSET = 500k 100 75 ISET = GND 50 25 0 –40 –10 50 80 20 TEMPERATURE (°C) 110 3631 G09 3631fb 4 LTC3631 TYPICAL PERFORMANCE CHARACTERISTICS Peak Current Trip Threshold vs RISET 180 160 140 120 100 80 60 40 20 200 400 600 800 RISET (kΩ) 1000 12 200 175 150 RSET = 500k 125 100 75 ISET = GND 50 5 SHUTDOWN 4 10 15 20 25 30 35 40 45 50 INPUT VOLTAGE (V) 15 5 25 35 45 VIN VOLTAGE (V) 3631 G12 Switch On-Resistance vs Temperature 5 4.5 VIN = 10V VIN = 10V SLEEP 10 8 6 4 SHUTDOWN 2 SWITCH 0N-RESISTANCE (Ω) 4.0 SWITCH ON-RESISTANCE (Ω) VIN SUPPLY CURRENT (μA) 6 Switch On-Resistance vs Input Voltage 12 3.5 TOP 3.0 2.5 2.0 BOTTOM 1.5 1.0 4 TOP 3 BOTTOM 2 1 0.5 0 –40 –10 20 50 80 TEMPERATURE (°C) 0 110 0 10 30 20 VIN VOLTAGE (V) 50 80 20 TEMPERATURE (°C) 110 Operating Waveforms 1.300 RUN COMPARATOR THRESHOLD (V) 0.5 0.4 0.3 0.2 SW = 0V SW = 45V –10 –10 3631 G15 RUN Comparator Threshold Voltages vs Temperature VIN = 45V 0 –40 0 –40 3631 G14 Switch Leakage Current vs Temperature 0.1 50 40 3631 G13 SWITCH LEAKAGE CURRENT (μA) 8 3631 G11 Quiescent Supply Current vs Temperature 0.6 10 0 0 3631 G10 14 SLEEP 2 25 0 1200 14 ISET OPEN 225 VIN SUPPLY CURRENT (μA) 200 0 Quiescent Supply Current vs Input Voltage 250 VIN = 10V 220 PEAK CURRENT TRIP THRESHOLD (mA) PEAK CURRENT TRIP THRESHOLD (mA) 240 Peak Current Trip Threshold vs Input Voltage 50 80 20 TEMPERATURE (°C) 110 3631 G16 1.250 SWITCH VOLTAGE 20V/DIV RISING 1.200 OUTPUT VOLTAGE 50mV/DIV 1.150 FALLING 1.100 INDUCTOR CURRENT 100mA/DIV 1.050 1.000 –40 –10 20 50 80 TEMPERATURE (°C) 110 3631 G17 VIN = 36V VOUT = 5V ILOAD = 35mA 20μs/DIV 3631 G18 3631fb 5 LTC3631 TYPICAL PERFORMANCE CHARACTERISTICS Soft-Start Waveform Load Step Transient Response Short-Circuit Response OUTPUT VOLTAGE 50mV/DIV OUTPUT VOLTAGE 2V/DIV LOAD CURRENT 50mA/DIV INDUCTOR CURRENT 100mA/DIV OUTPUT VOLTAGE 1V/DIV CSS = 0.047μF 5ms/DIV 3631 G19 VIN = 24V VOUT = 5V 1ms/DIV 3631 G20 VIN = 10V VOUT = 5V 200μs/DIV 3631 G21 PIN FUNCTIONS SW (Pin 1): Switch Node Connection to Inductor. This pin connects to the drains of the internal power MOSFET switches. VIN (Pin 2): Main Supply Pin. A ceramic bypass capacitor should be tied between this pin and GND (Pin 8). ISET (Pin 3): Peak Current Set Input. A resistor from this pin to ground sets the peak current trip threshold. Leave floating for the maximum peak current (225mA). Short this pin to ground for the minimum peak current (50mA). A 1μA current is sourced out of this pin. SS (Pin 4): Soft-Start Control Input. A capacitor to ground at this pin sets the ramp time to full current output during start-up. A 5μA current is sourced out of this pin. If left floating, the ramp time defaults to an internal 0.75ms soft-start. VOUT/VFB (Pin 6): Output Voltage Feedback. For the fixed output versions, connect this pin to the output supply. For the adjustable version, an external resistive divider should be used to divide the output voltage down for comparison to the 0.8V reference. HYST (Pin 7): Run Hysteresis Open-Drain Logic Output. This pin is pulled to ground when RUN (Pin 5) is below 1.2V. This pin can be used to adjust the RUN pin hysteresis. See Applications Information. GND (Pin 8, Exposed Pad Pin 9): Ground. The exposed pad must be soldered to the printed circuit board ground plane for optimal electrical and thermal performance. RUN (Pin 5): Run Control Input. A voltage on this pin above 1.2V enables normal operation. Forcing this pin below 0.7V shuts down the LTC3631, reducing quiescent current to approximately 3μA. 3631fb 6 LTC3631 BLOCK DIAGRAM VIN 1μA ISET 2 C2 3 – PEAK CURRENT COMPARATOR + RUN LOGIC AND SHOOTTHROUGH PREVENTION + 5 1.2V – SW L1 VOUT 1 C1 HYST 7 + REVERSE CURRENT COMPARATOR FEEDBACK COMPARATOR GND GND 8 9 – VOLTAGE REFERENCE + + – 0.800V 5μA SS 4 R1 VOUT/VFB 6 R2 PART NUMBER R1 R2 LTC3631 LTC3631-3.3 LTC3631-5 0 2.5M 4.2M d 800k 800k 3631 BD IMPLEMENT DIVIDER EXTERNALLY FOR ADJUSTABLE VERSION 3631fb 7 LTC3631 OPERATION (Refer to Block Diagram) The LTC3631 is a step-down DC/DC converter with internal power switches that uses Burst Mode control, combining low quiescent current with high switching frequency, which results in high efficiency across a wide range of load currents. Burst Mode operation functions by using short “burst” cycles to ramp the inductor current through the internal power switches, followed by a sleep cycle where the power switches are off and the load current is supplied by the output capacitor. During the sleep cycle, the LTC3631 draws only 12μA of supply current. At light loads, the burst cycles are a small percentage of the total cycle time which minimizes the average supply current, greatly improving efficiency. Main Control Loop The feedback comparator monitors the voltage on the VFB pin and compares it to an internal 800mV reference. If this voltage is greater than the reference, the comparator activates a sleep mode in which the power switches and current comparators are disabled, reducing the VIN pin supply current to only 12μA. As the load current discharges the output capacitor, the voltage on the VFB pin decreases. When this voltage falls 5mV below the 800mV reference, the feedback comparator trips and enables burst cycles. At the beginning of the burst cycle, the internal high side power switch (P-channel MOSFET) is turned on and the inductor current begins to ramp up. The inductor current increases until either the current exceeds the peak current comparator threshold or the voltage on the VFB pin exceeds 800mV, at which time the high side power switch is turned off, and the low side power switch (N-channel MOSFET) turns on. The inductor current ramps down until the reverse current comparator trips, signaling that the current is close to zero. If the voltage on the VFB pin is still less than the 800mV reference, the high side power switch is turned on again and another cycle commences. The average current during a burst cycle will normally be greater than the average load current. For this architecture, the maximum average output current is equal to half of the peak current. The hysteretic nature of this control architecture results in a switching frequency that is a function of the input voltage, output voltage and inductor value. This behavior provides inherent short-circuit protection. If the output is shorted to ground, the inductor current will decay very slowly during a single switching cycle. Since the high side switch turns on only when the inductor current is near zero, the LTC3631 inherently switches at a lower frequency during start-up or short-circuit conditions. Start-Up and Shutdown If the voltage on the RUN pin is less than 0.7V, the LTC3631 enters a shutdown mode in which all internal circuitry is disabled, reducing the DC supply current to 3μA. When the voltage on the RUN pin exceeds 1.21V, normal operation of the main control loop is enabled. The RUN pin comparator has 110mV of internal hysteresis, and therefore must fall below 1.1V to disable the main control loop. The HYST pin provides an added degree of flexibility for the RUN pin operation. This open-drain output is pulled to ground whenever the RUN comparator is not tripped, signaling that the LTC3631 is not in normal operation. In applications where the RUN pin is used to monitor the VIN voltage through an external resistive divider, the HYST pin can be used to increase the effective RUN comparator hysteresis. An internal 1ms soft-start function limits the ramp rate of the output voltage on start-up to prevent excessive input supply droop. If a longer ramp time and consequently less supply droop is desired, a capacitor can be placed from the SS pin to ground. The 5μA current that is sourced out of this pin will create a smooth voltage ramp on the capacitor. If this ramp rate is slower than the internal 1ms soft-start, 3631fb 8 LTC3631 OPERATION (Refer to Block Diagram) then the output voltage will be limited by the ramp rate on the SS pin instead. The internal and external soft-start functions are reset on start-up and after an undervoltage or overvoltage event on the input supply. from the ISET pin to ground. The 1μA current sourced out of this pin through the resistor generates a voltage that is translated into an offset in the peak current comparator, which limits the peak inductor current. In order to ensure a smooth start-up transition in any application, the internal soft-start also ramps the peak inductor current from 50mA during its 1ms ramp time to the set peak current threshold. The external ramp on the SS pin does not limit the peak inductor current during start-up; however, placing a capacitor from the ISET pin to ground does provide this capability. Input Undervoltage and Overvoltage Lockout Peak Inductor Current Programming The offset of the peak current comparator nominally provides a peak inductor current of 225mA. This peak inductor current can be adjusted by placing a resistor The LTC3631 implements a protection feature which disables switching when the input voltage is not within the 4.5V to 45V operating range. If VIN falls below 4V typical (4.35V maximum), an undervoltage detector disables switching. Similarly, if VIN rises above 50V typical (47V minimum), an overvoltage detector disables switching. When switching is disabled, the LTC3631 can safely sustain input voltages up to the absolute maximum rating of 60V. Switching is enabled when the input voltage returns to the 4.5V to 45V operating range. 3631fb 9 LTC3631 APPLICATIONS INFORMATION The basic LTC3631 application circuit is shown on the front page of this data sheet. External component selection is determined by the maximum load current requirement and begins with the selection of the peak current programming resistor, RISET. The inductor value L can then be determined, followed by capacitors CIN and COUT. Peak Current Resistor Selection The peak current comparator has a maximum current limit of 225mA nominally, which results in a maximum average current of 112mA. For applications that demand less current, the peak current threshold can be reduced to as little as 50mA. This lower peak current allows the use of lower value, smaller components (input capacitor, output capacitor and inductor), resulting in lower input supply ripple and a smaller overall DC/DC converter. The threshold can be easily programmed with an appropriately chosen resistor (RISET) between the ISET pin and ground. The value of resistor for a particular peak current can be computed by using Figure 1 or the following equation: RISET = IPEAK • 4.5 • 106 where 50mA < IPEAK < 225mA. The peak current is internally limited to be within the range of 50mA to 225mA. Shorting the ISET pin to ground programs the current limit to 50mA, and leaving it floating sets the current limit to the maximum value of 225mA. When selecting this resistor value, be aware that the 1100 1000 900 RISET (kΩ) 800 700 600 500 400 300 200 100 0 20 30 40 50 60 70 80 90 MAXIMUM LOAD CURRENT (mA) 100 3631 F01 Figure 1. RISET Selection maximum average output current for this architecture is limited to half of the peak current. Therefore, be sure to select a value that sets the peak current with enough margin to provide adequate load current under all foreseeable operating conditions. Inductor Selection The inductor, input voltage, output voltage and peak current determine the switching frequency of the LTC3631. For a given input voltage, output voltage and peak current, the inductor value sets the switching frequency when the output is in regulation. A good first choice for the inductor value can be determined by the following equation: V V L = OUT • 1– OUT VIN f • IPEAK The variation in switching frequency with input voltage and inductance is shown in the following two figures for typical values of VOUT. For lower values of IPEAK, multiply the frequency in Figure 2 and Figure 3 by 225mA/IPEAK. An additional constraint on the inductor value is the LTC3631’s 100ns minimum on-time of the high side switch. Therefore, in order to keep the current in the inductor well controlled, the inductor value must be chosen so that it is larger than LMIN, which can be computed as follows: L MIN = VIN(MAX ) • tON(MIN) IPEAK(MAX ) where VIN(MAX) is the maximum input supply voltage for the application, tON(MIN) is 100ns, and IPEAK(MAX) is the maximum allowed peak inductor current. Although the above equation provides the minimum inductor value, higher efficiency is generally achieved with a larger inductor value, which produces a lower switching frequency. For a given inductor type, however, as inductance is increased DC resistance (DCR) also increases. Higher DCR translates into higher copper losses and lower current rating, both of which place an upper limit on the inductance. The recommended range of inductor values for small surface mount inductors as a function of peak current is shown in Figure 4. The values in this range are a good compromise between the tradeoffs discussed above. For applications 3631fb 10 LTC3631 APPLICATIONS INFORMATION SWITCHING FREQUENCY (kHz) 700 VOUT = 5V ISET OPEN 600 where board area is not a limiting factor, inductors with larger cores can be used, which extends the recommended range of Figure 4 to larger values. L = 47μH 500 L = 100μH Inductor Core Selection 400 300 L = 220μH 200 L = 470μH 100 0 5 10 15 20 25 30 35 VIN INPUT VOLTAGE (V) 40 45 3631 F02 Figure 2. Switching Frequency for VOUT = 5V 500 SWITCHING FREQUENCY (kHz) 450 L = 47μH 400 VOUT = 3.3V ISET OPEN 350 300 L = 100μH 250 200 150 L = 220μH 100 L = 470μH 50 0 5 10 15 20 25 30 35 VIN INPUT VOLTAGE (V) 40 45 3631 F03 Figure 3. Switching Frequency for VOUT = 3.3V INDUCTOR VALUE (μH) 10000 Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of the more expensive ferrite cores. Actual core loss is independent of core size for a fixed inductor value but is very dependent of the inductance selected. As the inductance increases, core losses decrease. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core losses and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequently output voltage ripple. Do not allow the core to saturate! Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate energy but generally cost more than powdered iron core inductors with similar characteristics. The choice of which style inductor to use mainly depends on the price vs size requirements and any radiated field/EMI requirements. New designs for surface mount inductors are available from Coiltronics, Coilcraft, TDK, Toko, Sumida and Vishay. 1000 CIN and COUT Selection The input capacitor, CIN, is needed to filter the trapezoidal current at the source of the top high side MOSFET. To prevent large ripple voltage, a low ESR input capacitor sized for the maximum RMS current should be used. Approximate RMS current is given by: 100 10 10 100 PEAK INDUCTOR CURRENT (mA) 1000 3631 F04 Figure 4. Recommended Inductor Values for Maximum Efficiency IRMS = IOUT(MAX ) • VOUT VIN • −1 VIN VOUT 3631fb 11 LTC3631 APPLICATIONS INFORMATION This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based only on 2000 hours of life which makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet size or height requirements in the design. electrolytic capacitors have significantly higher ESR but can be used in cost-sensitive applications provided that consideration is given to ripple current ratings and longterm reliability. Ceramic capacitors have excellent low ESR characteristics but can have high voltage coefficient and audible piezoelectric effects. The high quality factor (Q) of ceramic capacitors in series with trace inductance can also lead to significant ringing. The output capacitor, COUT, filters the inductor’s ripple current and stores energy to satisfy the load current when the LTC3631 is in sleep. The output voltage ripple during a burst cycle is dominated by the output capacitor equivalent series resistance (ESR) and can be estimated by the following equation: Higher value, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. However, care must be taken when these capacitors are used at the input and output. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN large enough to damage the part. VOUT < ΔVOUT ≤ IPEAK • ESR 160 where the lower limit of VOUT/160 is due to the 5mV feedback comparator hysteresis. The value of the output capacitor must be large enough to accept the energy stored in the inductor without a large change in output voltage. Setting this voltage step equal to 1% of the output voltage, the output capacitor must be: COUT I 2 PEAK > 50 • L • VOUT Typically, a capacitor that satisfies the ESR requirement is adequate to filter the inductor ripple. To avoid overheating, the output capacitor must also be sized to handle the ripple current generated by the inductor. The worst-case ripple current in the output capacitor is given by IRMS = IPEAK/2. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic, and ceramic capacitors are all available in surface mount packages. Special polymer capacitors offer very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important only to use types that have been surge tested for use in switching power supplies. Aluminum Using Ceramic Input and Output Capacitors For applications with inductive source impedance, such as a long wire, a series RC network may be required in parallel with CIN to dampen the ringing of the input supply. Figure 5 shows this circuit and the typical values required to dampen the ringing. LTC3631 LIN VIN R= LIN CIN 4 • CIN 3631 F05 CIN Figure 5. Series RC to Reduce VIN Ringing Output Voltage Programming For the adjustable version, the output voltage is set by an external resistive divider according to the following equation: R1 VOUT = 0.8V • 1+ R2 3631fb 12 LTC3631 APPLICATIONS INFORMATION The resistive divider allows the VFB pin to sense a fraction of the output voltage as shown in Figure 6. Output voltage can range from 0.8V to VIN. VOUT R1 VFB LTC3631 R2 GND 3631 F06 Figure 6. Setting the Output Voltage The RUN pin can alternatively be configured as a precise undervoltage lockout (UVLO) on the VIN supply with a resistive divider from VIN to ground. The RUN pin comparator nominally provides 10% hysteresis when used in this method; however, additional hysteresis may be added with the use of the HYST pin. The HYST pin is an open-drain output that is pulled to ground whenever the RUN comparator is not tripped. A simple resistive divider can be used as shown in Figure 8 to meet specific VIN voltage requirements. VIN To minimize the no-load supply current, resistor values in the megohm range should be used; however, large resistor values should be used with caution. The feedback divider is the only load current when in shutdown. If PCB leakage current to the output node or switch node exceeds the load current, the output voltage will be pulled up. In normal operation, this is generally a minor concern since the load current is much greater than the leakage. The increase in supply current due to the feedback resistors can be calculated from: V V IVIN = OUT • OUT R1+ R2 VIN The LTC3631 has a low power shutdown mode controlled by the RUN pin. Pulling the RUN pin below 0.7V puts the LTC3631 into a low quiescent current shutdown mode (IQ ~ 3μA). When the RUN pin is greater than 1.2V, the controller is enabled. Figure 7 shows examples of configurations for driving the RUN pin from logic. VIN LTC3631 RUN RUN R2 LTC3631 HYST R3 3631 F08 Figure 8. Adjustable Undervoltage Lockout Specific values for these UVLO thresholds can be computed from the following equations: R1 Rising VIN UVLO Threshold = 1.21V • 1+ R2 Run Pin with Programmable Hysteresis VSUPPLY R1 4.7M LTC3631 RUN 3631 F07 Figure 7. RUN Pin Interface to Logic R1 Falling VIN UVLO Threshold = 1.10V • 1+ R2 + R3 The minimum value of these thresholds is limited to the internal VIN UVLO thresholds that are shown in the Electrical Characteristics table. The current that flows through this divider will directly add to the shutdown, sleep and active current of the LTC3631, and care should be taken to minimize the impact of this current on the overall efficiency of the application circuit. Resistor values in the megohm range may be required to keep the impact on quiescent shutdown and sleep currents low. Be aware that the HYST pin cannot be allowed to exceed its absolute maximum rating of 6V. To keep the voltage on the HYST pin from exceeding 6V, the following relation should be satisfied: R3 VIN(MAX) • < 6V R1+ R2 + R3 3631fb 13 LTC3631 APPLICATIONS INFORMATION The RUN pin may also be directly tied to the VIN supply for applications that do not require the programmable undervoltage lockout feature. In this configuration, switching is enabled when VIN surpasses the internal undervoltage lockout threshold. Soft-Start The internal 0.75ms soft-start is implemented by ramping both the effective reference voltage from 0V to 0.8V and the peak current limit set by the ISET pin (50mA to 225mA). To increase the duration of the reference voltage soft-start, place a capacitor from the SS pin to ground. An internal 5μA pull-up current will charge this capacitor, resulting in a soft-start ramp time given by: tSS = CSS • 0.8 V 5μA Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 + L3 + ...) When the LTC3631 detects a fault condition (input supply undervoltage or overvoltage) or when the RUN pin falls below 1.1V, the SS pin is quickly pulled to ground and the internal soft-start timer is reset. This ensures an orderly restart when using an external soft-start capacitor. The duration of the 0.75ms internal peak current soft-start may be increased by placing a capacitor from the ISET pin to ground. The peak current soft-start will ramp from 50mA to the final peak current value determined by a resistor from ISET to ground. A 1μA current is sourced out of the ISET pin. With only a capacitor connected between ISET and ground, the peak current ramps linearly from 50mA to 225mA, and the peak current soft-start time can be expressed as: tSS(ISET) = CISET • Unlike the SS pin, the ISET pin does not get pulled to ground during an abnormal event; however, if the ISET pin is floating (programmed to 225mA peak current), the SS and ISET pins may be tied together and connected to a capacitor to ground. For this special case, both the peak current and the reference voltage will soft-start on power-up and after fault conditions. The ramp time for this combination is CSS(ISET) • (0.8V/6μA). 0.8 V 1μA A linear ramp of peak current appears as a quadratic waveform on the output voltage. For the case where the peak current is reduced by placing a resistor from ISET to ground, the peak current offset ramps as a decaying exponential with a time constant of RISET • CISET. For this case, the peak current soft-start time is approximately 3 • RISET • CISET. where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, two main sources usually account for most of the losses: VIN operating current and I2R losses. The VIN operating current dominates the efficiency loss at very low load currents whereas the I2R loss dominates the efficiency loss at medium to high load currents. 1. The VIN operating current comprises two components: The DC supply current as given in the electrical characteristics and the internal MOSFET gate charge currents. The gate charge current results from switching the gate capacitance of the internal power MOSFET switches. Each time the gate is switched from high to low to high again, a packet of charge, dQ, moves from VIN to ground. The resulting dQ/dt is the current out of VIN that is typically larger than the DC bias current. 2. I2R losses are calculated from the resistances of the internal switches, RSW, and external inductor RL. When switching, the average output current flowing through the inductor is “chopped” between the high side PMOS switch and the low side NMOS switch. Thus, the series resistance looking back into the switch pin is a function of the top and bottom switch RDS(ON) values and the duty cycle (DC = VOUT/VIN) as follows: RSW = (RDS(ON)TOP)DC + (RDS(ON)BOT)(1 – DC) 3631fb 14 LTC3631 APPLICATIONS INFORMATION The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain the I2R losses, simply add RSW to RL and multiply the result by the square of the average output current: I2R Loss = IO2(RSW + RL) Other losses, including CIN and COUT ESR dissipative losses and inductor core losses, generally account for less than 2% of the total power loss. Thermal Considerations The LTC3631 does not dissipate much heat due to its high efficiency and low peak current level. Even in worst-case conditions (high ambient temperature, maximum peak current and high duty cycle), the junction temperature will exceed ambient temperature by only a few degrees. Design Example As a design example, consider using the LTC3631 in an application with the following specifications: VIN = 24V, VOUT = 3.3V, IOUT = 100mA, f = 250kHz. Furthermore, assume for this example that switching should start when VIN is greater than 12V and should stop when VIN is less than 8V. First, calculate the inductor value that gives the required switching frequency: 3.3V 3.3V L= • 1– 47μH 250kHz • 225mA 24V Next, verify that this value meets the LMIN requirement. For this input voltage and peak current, the minimum inductor value is: L MIN = 24V • 100ns ≅ 10μH 225mA Therefore, the minimum inductor requirement is satisfied, and the 47μH inductor value may be used. Next, CIN and COUT are selected. For this design, CIN should be sized for a current rating of at least: Due to the low peak current of the LTC3631, decoupling the VIN supply with a 1μF capacitor is adequate for most applications. COUT will be selected based on the ESR that is required to satisfy the output voltage ripple requirement. For a 50mV output ripple, the value of the output capacitor ESR can be calculated from: ΔVOUT = 50mV ≤ 225mA • ESR A capacitor with a 200mΩ ESR satisfies this requirement. A 10μF ceramic capacitor has significantly less ESR than 200mΩ. The output voltage can now be programmed by choosing the values of R1 and R2. Choose R2 = 240k and calculate R1 as: V R1= OUT – 1 • R2 = 750k 0.8V The undervoltage lockout requirement on VIN can be satisfied with a resistive divider from VIN to the RUN and HYST pins. Choose R1 = 2M and calculate R2 and R3 as follows: 1.21V • R1= 224k R2 = – 1.21V V IN(RISING) 1.1V • R1– R2 = 90.8k R3 = – 1.1V V IN(FALLING) Choose standard values for R2 = 226k and R3 = 91k. The ISET pin should be left open in this example to select maximum peak current (225mA). Figure 9 shows a complete schematic for this design example. 47μH VIN 24V VIN 1μF 2M SW LTC3631 RUN 226k 91k ISET SS 750k VFB HYST GND 240k VOUT 3.3V 100mA 10μF 3631 F09 3.3V 24V IRMS = 100mA • • – 1 ≅ 35mA RMS 24V 3.3V Figure 9. 24V to 3.3V, 100mA Regulator at 250kHz 3631fb 15 LTC3631 APPLICATIONS INFORMATION Example Layout PC Board Layout Checklist L1 When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3631. Check the following in your layout: VIN VIN LTC3631 R1 RUN CIN 1. Large switched currents flow in the power switches and input capacitor. The loop formed by these components should be as small as possible. A ground plane is recommended to minimize ground impedance. CSS HYST VFB SS ISET 4. Flood all unused area on all layers with copper. Flooding with copper will reduce the temperature rise of power components. You can connect the copper areas to any DC net (VIN, VOUT, GND or any other DC rail in your system). COUT GND RSET R2 3631 F09a 2. Connect the (+) terminal of the input capacitor, CIN, as close as possible to the VIN pin. This capacitor provides the AC current into the internal power MOSFETs. 3. Keep the switching node, SW, away from all sensitive small signal nodes. The rapid transitions on the switching node can couple to high impedance nodes, in particular VFB, and create increased output ripple. VOUT SW L1 VIN CIN VOUT COUT R1 R2 RSET CSS GND VIAS TO GROUND PLANE VIAS TO INPUT SUPPLY (VIN) OUTLINE OF LOCAL GROUND PLANE 3631fb 16 LTC3631 TYPICAL APPLICATIONS L1 100μH VIN 5V TO 45V VIN CIN 4.7μF SW LTC3631 RUN HYST ISET R1 1.47M VFB VOUT 5V COUT 100μF R2 280k SS GND CSS 47nF 3631 F10a CIN: TDK C5750X7R2A475MT COUT: AVX 1812D107MAT L1: TDK SLF7045T-101MR50-PF Figure 10. High Efficiency 5V Regulator 3.3V, 100mA Regulator with Peak Current Soft-Start, Small Size VIN 4.5V TO 24V L1 22μH VIN CIN 1μF SW R1 294k LTC3631 RUN ISET SS CSS 22nF VFB VOUT 3.3V COUT 100mA 10μF Soft-Start Waveforms OUTPUT VOLTAGE 1V/DIV R2 93.1k HYST GND 3642 TA03a INDUCTOR CURRENT 50mA/DIV CIN: TDK C3216X7R1E105KT COUT: AVX 08056D106KAT2A L1: MURATA LQH43CN220K03 3631 TA03b 500μs/DIV Positive-to-Negative Converter 100 L1 100μH VIN CIN 1μF SW LTC3631 RUN ISET R1 1M COUT 10μF VFB SS HYST GND R2 71.5k CIN: TDK C3225X7R1H105KT COUT: MURATA GRM32DR71C106KA01 L1: TYCO/COEV DQ6545-101M VOUT –12V 3631 TA04a MAXIMUM LOAD CURRENT (mA) VIN 4.5V TO 33V Maximum Load Current vs Input Voltage ISET OPEN VOUT = –3V 90 VOUT = –5V 80 70 VOUT = –12V 60 50 40 30 20 5 10 15 20 25 30 35 VIN VOLTAGE (V) 40 45 3631 TA04b 3631fb 17 LTC3631 TYPICAL APPLICATIONS Small Size, Limited Peak Current, 20mA Regulator VIN 7V TO 45V L1 470μH VIN CIN 1μF R3 470k SW RUN R4 100k R5 33k R1 470k LTC3631 VFB COUT 10μF VOUT 5V 20mA R2 88.7k HYST SS ISET GND 3631 TA05a CIN: TDK C3225X7R1H105KT COUT: AVX 08056D106KAT2A L1: MURATA LQH43CN471K03 High Efficiency 15V, 20mA Regulator 100 L1 1000μH VIN CIN 1μF SW LTC3631 RUN ISET SS VFB HYST GND CIN: AVX 18125C105KAT2A COUT: TDK C3216X7R1E475KT L1: TDK SLF7045T-102MR14 R1 3M R2 169k 3631 TA07a COUT 4.7μF VOUT 15V 20mA EFFICIENCY (%) VIN 15V TO 45V Efficiency vs Load Current 95 VIN = 24V 90 VIN = 36V 85 80 VIN = 45V 75 70 65 60 0.1 1 10 LOAD CURRENT (mA) 100 3631 TA07b 3631fb 18 LTC3631 PACKAGE DESCRIPTION DD Package 8-Lead Plastic DFN (3mm × 3mm) (Reference LTC DWG # 05-08-1698) 0.675 p0.05 3.5 p0.05 1.65 p0.05 2.15 p0.05 (2 SIDES) PACKAGE OUTLINE 0.25 p 0.05 0.50 BSC 2.38 p0.05 (2 SIDES) RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS 3.00 p0.10 (4 SIDES) R = 0.115 TYP 5 0.38 p 0.10 8 1.65 p 0.10 (2 SIDES) PIN 1 TOP MARK (NOTE 6) (DD) DFN 1203 0.200 REF 0.75 p0.05 4 0.25 p 0.05 1 0.50 BSC 0.00 – 0.05 2.38 p0.10 (2 SIDES) BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON TOP AND BOTTOM OF PACKAGE 3631fb 19 LTC3631 PACKAGE DESCRIPTION MS8E Package 8-Lead Plastic MSOP, Exposed Die Pad (Reference LTC DWG # 05-08-1662 Rev E) BOTTOM VIEW OF EXPOSED PAD OPTION 2.06 p 0.102 (.081 p .004) 1 0.889 p 0.127 (.035 p .005) 2.794 p 0.102 (.110 p .004) 0.29 REF 1.83 p 0.102 (.072 p .004) 0.05 REF 5.23 (.206) MIN DETAIL “B” CORNER TAIL IS PART OF DETAIL “B” THE LEADFRAME FEATURE. FOR REFERENCE ONLY NO MEASUREMENT PURPOSE 2.083 p 0.102 3.20 – 3.45 (.082 p .004) (.126 – .136) 8 0.42 p 0.038 (.0165 p .0015) TYP 3.00 p 0.102 (.118 p .004) (NOTE 3) 0.65 (.0256) BSC 8 7 6 5 0.52 (.0205) REF RECOMMENDED SOLDER PAD LAYOUT 0.254 (.010) 3.00 p 0.102 (.118 p .004) (NOTE 4) 4.90 p 0.152 (.193 p .006) DETAIL “A” 0o – 6o TYP GAUGE PLANE 1 0.53 p 0.152 (.021 p .006) DETAIL “A” 2 3 4 1.10 (.043) MAX 0.86 (.034) REF 0.18 (.007) SEATING PLANE 0.22 – 0.38 (.009 – .015) TYP 0.65 (.0256) BSC 0.1016 p 0.0508 (.004 p .002) MSOP (MS8E) 0908 REV E NOTE: 1. DIMENSIONS IN MILLIMETER/(INCH) 2. DRAWING NOT TO SCALE 3. DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 4. DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.152mm (.006") PER SIDE 5. LEAD COPLANARITY (BOTTOM OF LEADS AFTER FORMING) SHALL BE 0.102mm (.004") MAX 3631fb 20 LTC3631 REVISION HISTORY (Revision history begins at Rev B) REV DATE DESCRIPTION B 05/10 Updated Absolute Maximum Ratings and Order Information Sections PAGE NUMBER 2 Updated Note 2 3 Updated Graphs G09, G18 and G19 Updated GND pin text in Pin Functions 4, 5, 6 6 Text added to “Output Voltage Programming” section 12 “Example Layout” Art added 16 Updated Typical Application 22 Updated Related Parts 22 3631fb Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 21 LTC3631 TYPICAL APPLICATION 5V, 100mA Regulator for Automotive Applications VBATT 4.5V TO 45V AND SURVIVES TRANSIENTS UP TO 60V L1 100μH VIN CIN 2.2μF SW LTC3631 RUN ISET SS VFB HYST GND CIN: TDK C3225X7R2A225M COUT: KEMET C1210C106K4RAC L1: COILTRONICS DRA73-101-R R1 470k COUT 10μF VOUT* 5V 100mA R2 88.7k 3631 TA06a *VOUT = VBATT FOR VBATT < 5V RELATED PARTS PART NUMBER LTC3632 DESCRIPTION 50V, 20mA Synchronous Micropower Step-Down DC/DC Converter COMMENTS VIN: 4.5V to 50V (60VMAX), VOUT(MIN) = 0.8V, IQ = 12μA, ISD = 3μA, 3mm × 3mm DFN, MS8E LTC3642 45V, 50mA Synchronous Micropower Step-Down DC/DC Converter VIN: 4.5V to 45V (60VMAX), VOUT(MIN) = 0.8V, IQ = 12μA, ISD = 3μA, 3mm × 3mm DFN8, MS8E VIN: 3V to 18V, VOUT(MIN) = 1.2V, IQ = 10μA, ISD = 6μA, MS8E LTC1474 18V, 250mA (IOUT), High Efficiency Step-Down DC/DC Converter VIN: 3.2V to 34V, VOUT(MIN) = 1.25V, IQ = 12μA, ISD < 1μA, LT1934/LT1934-1 36V, 250mA (IOUT), Micropower Step-Down DC/DC Converter with Burst Mode Operation ThinSOT™ Package LT1939 25V, 2A, 2.5MHz High Efficiency DC/DC Converter and LDO VIN: 3.6V to 25V, VOUT(MIN) = 0.8V, IQ = 2.5mA, ISD < 10μA, 3mm × 3mm DFN10 Controller LT1976/LT1977 60V, 1.2A (IOUT), 200kHz/500kHz, High Efficiency Step-Down DC/DC VIN: 3.3V to 60V, VOUT(MIN) = 1.2V, IQ = 100μA, ISD < 1μA, Converter with Burst Mode Operation TSSOP16E VIN: 3.3V to 60V, VOUT(MIN) = 1.25V, IQ = 100μA, ISD < 1μA, LT3437 60V, 400mA (IOUT), Micropower Step-Down DC/DC Converter with Burst Mode Operation 3mm × 3mm DFN10, TSSOP16E VIN: 4V to 40V, VOUT(MIN) = 1.2V, IQ = 26μA, ISD < 1μA, LT3470 40V, 250mA (IOUT), High Efficiency Step-Down DC/DC Converter with Burst Mode Operation 2mm × 3mm DFN8, ThinSOT VIN: 3.6V to 38V, VOUT(MIN) = 0.78V, IQ = 70μA, ISD < 1μA, LT3685 36V with Transient Protection to 60V, 2A (IOUT), 2.4MHz, High Efficiency Step-Down DC/DC Converter 3mm × 3mm DFN10, MSOP10E 3631fb 22 Linear Technology Corporation LT 0510 REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2009