LTC3606B 800mA Synchronous Step-Down DC/DC with Average Input Current Limit DESCRIPTION FEATURES n n n n n n n n n n n n n n Programmable Average Input Current Limit: ±5% Accuracy Step-Down Output: Up to 96% Efficiency Low Noise Pulse-Skipping Operation at Light Loads Input Voltage Range: 2.5V to 5.5V Output Voltage Range: 0.6V to 5V 2.25MHz Constant-Frequency Operation Power Good Output Voltage Monitor Low Dropout Operation: 100% Duty Cycle Internal Soft-Start Current Mode Operation for Excellent Line and Load Transient Response ±2% Output Voltage Accuracy Short-Circuit Protected Shutdown Current ≤ 1μA Available in Small Thermally Enhanced 8-Lead 3mm × 3mm DFN Package APPLICATIONS n n n n The LTC®3606B is an 800mA monolithic synchronous buck regulator using a constant frequency current mode architecture. The input supply voltage range is 2.5V to 5.5V, making it ideal for Li-Ion and USB powered applications. 100% duty cycle capability provides low dropout operation, extending the run time in battery-operated systems. Low output voltages are supported with the 0.6V feedback reference voltage. The LTC3606B can supply 800mA output current. The LTC3606B’s programmable average input current limit is ideal for USB applications and for point-of-load power supplies because the LTC3606B’s limited input current will still allow its output to deliver high peak load currents without collapsing the input supply. The operating frequency is internally set at 2.25MHz allowing the use of small surface mount inductors. Internal soft-start reduces in-rush current during start-up. The LTC3606B is available in an 8-Lead 3mm × 3mm DFN package. L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks are the property of their respective owners. Protected by U.S.Patents, including 5481178, 6127815, 6304066, 6498466, 6580258, 6611131. High Peak Load Current Applications USB Powered Devices Supercapacitor Charging Radio Transmitters and Other Handheld Devices TYPICAL APPLICATION Monolithic Buck Regulator with Input Current Limit 1.5μH VIN 3.4V TO 5.5V VIN CIN 10μF RUN + PGOOD VFB RLIM PGOOD GND 1000pF VOUT 3.4V AT 800mA SW LTC3606B 499k 116k 1210k GSM Pulse Load 2.2mF s2 SuperCap VIN AC-COUPLED 1V/DIV IOUT 500mA/DIV 255k 3606B TA01 ILIM = 475mA VOUT 200mV/DIV IIN 500mA/DIV 1ms/DIV 3606B TA01b VIN = 5V, 500mA COMPLIANT ILOAD = 0A to 2.2A 3606bfa 1 LTC3606B ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) TOP VIEW Input Supply Voltage (VIN) ........................... –0.3V to 6V VFB ................................................... –0.3V to VIN + 0.3V RUN, RLIM ....................................... –0.3V to VIN + 0.3V SW ................................................... –0.3V to VIN + 0.3V PGOOD............................................. –0.3V to VIN + 0.3V P-Channel SW Source Current (DC) (Note 2) ..............1A N-Channel SW Source Current (DC) (Note 2) .............1A Peak SW Source and Sink Current (Note 2) ............. 2.7A Operating Junction Temperature Range (Notes 3, 6, 8) ........................................ –40°C to 125°C Storage Temperature Range .................. –65°C to 125°C Reflow Peak Body Temperature ............................ 260°C GND 1 RLIM 2 GND 3 SW 4 8 VFB 9 GND 7 RUN 6 PGOOD 5 VIN DD PACKAGE 8-LEAD (3mm s 3mm) PLASTIC DFN TJMAX = 125°C, θJA = 40°C/W EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3606BEDD#PBF LTC3606BEDD#TRPBF LFMB 8-Lead (3mm × 3mm) Plastic DFN –40°C to 85°C LTC3606BIDD#PBF LTC3606BIDD#TRPBF LFMB 8-Lead (3mm × 3mm) Plastic DFN –40°C to 125°C Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3606bfa 2 LTC3606B ELECTRICAL CHARACTERISTICS The l denotes the specifications which apply over the full operating junction temperature range, otherwise specifications are at TA = 25°C, VIN = 5V, unless otherwise noted. SYMBOL PARAMETER VIN VIN Operating Voltage Range CONDITIONS MIN VUV VIN Undervoltage Lockout IFB Feedback Pin Input Current VFBREG Feedback Voltage LTC3606BE, –40°C < TJ < 85°C (Note 7) LTC3606BI, –40°C < TJ < 125°C (Note 7) ΔVLINEREG VFB Line Regulation ΔVLOADREG IS l TYP MAX UNITS 5.5 V 2.5 V ±30 nA 0.600 0.600 0.612 0.618 V V VIN = 2.5V to 5.5V (Note 7) 0.01 0.25 %/V VFB Load Regulation ILOAD = 0mA to 800mA (Note 7) 0.5 Supply Current Active Mode (Note 4) Shutdown VFB = 0.95 × VFBREG VRUN = 0V, VIN = 5.5V 420 650 1 μA μA fOSC Oscillator Frequency VFB = VFBREG 1.8 2.25 2.7 MHz ILIM(PEAK) Peak Switch Current Limit VIN = 5V, VFB < VFBREG , Duty Cycle <35% 1800 2400 IINLIM Input Average Current Limit RLIM = 116k RLIM = 116k, LTC3606BE RLIM = 116k, LTC3606BI 450 437 427 475 475 475 RDS(ON) Main Switch On-Resistance (Note 5) VIN = 5V, ISW = 100mA Synchronous Switch On-Resistance (Note 5) VIN = 5V, ISW = 100mA 0.27 0.25 ISW(LKG) Switch Leakage Current VIN = 5V, VRUN = 0V 0.01 1 μA tSOFTSTART Soft-Start Time VFB from 0.06V to 0.54V 0.3 0.95 1.3 ms VRUN RUN Threshold High 0.4 1 1.2 V IRUN RUN Leakage Current 0V ≤ VRUN ≤ 5V 0.01 1 μA PGOOD Power Good Threshold Entering Window VFB Ramping Up VFB Ramping Down Leaving Window VFB Ramping Up VFB Ramping Down PGOOD Blanking Power Good Blanking Time RPGOOD Power Good Pull-Down On-Resistance IPGOOD PGOOD Leakage Current VIN Low to High 2.5 l 2.1 l l l l l l l 0.588 0.582 l –5 5 Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: Guaranteed by long term current density limitations. Note 3: The LTC3606BE is guaranteed to meet performance specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating junction temperature range are assured by design, characterization and correlation with statistical process controls. The LTC3606BI is guaranteed to meet specified performance over the full –40°C to 125°C operating junction temperature range. Note 4: Dynamic supply current is higher due to the internal gate charge being delivered at the switching frequency. 500 513 523 15 mA mA mA Ω Ω % % 11 –11 90 8 VPGOOD = 5V mA –7 7 9 –9 PGOOD Rising and Falling, VIN = 5V % % % μs 30 Ω ±1 μA Note 5: The switch on-resistance is guaranteed by correlation to wafer level measurements. Note 6: This IC includes overtemperature protection that is intended to protect the device during momentary overload conditions. Junction temperature will exceed 125°C when overtemperature protection is active. Continuous operation above the specified maximum operating junction temperature may impair device reliability. Note 7: The converter is tested in a proprietary test mode that connects the output of the error amplifier to the SW pin, which is connected to an external servo loop. Note 8: TJ is calculated from the ambient temperature TA and the power dissipation as follows: TJ = TA + (PD)(θJA°C/W) 3606bfa 3 LTC3606B TYPICAL PERFORMANCE CHARACTERISTICS TA = 25°C, VIN = 5V, unless otherwise noted. Supply Current vs Temperature Pulse-Skipping Mode Operation 550 SW 2V/DIV 500 Efficiency vs Input Voltage 100 RUN = VIN ILOAD = 0A 90 80 IL 100mA/DIV 450 400 EFFICIENCY (%) SUPPLY CURRENT (μA) VIN = 5.5V VOUT 50mV/DIV ACCOUPLED VIN = 2.7V 350 70 60 50 30 300 IOUT = 10mA IOUT = 1mA IOUT = 0.1mA IOUT = 100mA IOUT = 400mA IOUT = 800mA 40 20 3606B G01 5μs/DIV VIN = 5V VOUT = 3.3V ILOAD = 5mA 250 10 200 –50 –25 0 25 50 75 TEMPERATURE (°C) 100 VOUT = 3.3V 0 3.5 125 4 4.5 VIN (V) 5 3606B G03 3606B G02 Oscillator Frequency vs Temperature 1.5 2.5 1.0 2.4 Switch Leakage vs Input Voltage 1000 0.5 0 –0.5 –1.0 2.3 2.2 2.1 2.0 VIN = 2.7V VIN = 3.6V VIN = 4.2V VIN = 5V 1.9 –25 0 25 50 75 TEMPERATURE (°C) 100 1.8 –50 125 –25 0 25 50 75 TEMPERATURE (°C) 3606B G04 0.5 MAIN PFET RDS(ON) (Ω) RDS(ON) (mΩ) 400 MAIN SWITCH 300 4.5 0 2.5 125 3 5.5 MAIN SWITCH 0.6 0.5 0.2 0.4 0.1 0.3 0 0.2 3606B G07 –0.1 –50 –25 25 50 75 0 TEMPERATURE (°C) 4 VIN (V) 4.5 5 5.5 Efficiency vs Load Current 0.7 100 90 VOUT = 3.3V 80 70 60 50 40 30 20 10 SYNCHRONOUS SWITCH 5 3.5 3606B G06 0.3 SYNCHRONOUS SWITCH 4 VIN (V) SYNCHRONOUS SWITCH 200 SYNCHRONOUS NFET RDS(ON) (Ω) VIN = 2.7V VIN = 3.6V VIN = 5V 0.4 500 3.5 400 Switch On-Resistance vs Temperature 600 3 MAIN SWITCH 3606B G05 Switch On-Resistance vs Input Voltage 200 2.5 100 600 EFFICIENCY (%) –1.5 –50 LEAKAGE CURRENT (pA) 800 FREQUENCY (MHz) VFB ERROR (%) Regulated Voltage vs Temperature 5.5 100 0.1 125 3606B G09 0 0.0001 VIN = 3.6V VIN = 4.2V VIN = 5V 0.001 0.01 0.1 OUTPUT CURRENT (A) 1 3606B G11 3606bfa 4 LTC3606B TYPICAL PERFORMANCE CHARACTERISTICS Efficiency vs Load Current 100 90 TA = 25°C, VIN = 5V, unless otherwise noted. Load Regulation Line Regulation 0.6 3.0 VOUT = 1.2V 2.5 0.4 80 70 60 50 40 30 10 0 0.0001 0.001 0.01 0.1 OUTPUT CURRENT (A) 1.5 1.0 0.5 0.2 0 –0.2 0 VIN = 2.7V VIN = 3.6V VIN = 4.2V VIN = 5V 20 VOUT ERROR (%) 2.0 VOUT ERROR (%) EFFICIENCY (%) VOUT = 1.8V ILOAD = 100mA –1.0 1 –0.4 VOUT = 1.8V VOUT = 2.5V VOUT = 3.3V –0.5 0 –0.6 2.5 100 200 300 400 500 600 700 800 LOAD CURRENT (mA) 3606B G13 3.5 4.0 VIN (V) 4.5 1.2 1.0 VOUT 2V/DIV 0.8 RLIM 1V/DIV IL 250mA/DIV IIN 500mA/DIV VRLIM (V) RUN 2V/DIV VOUT 1V/DIV 5.5 VRLIM vs Input Current Start-Up from Shutdown RUN 2V/DIV 5.0 3606B G16 3606B G15 Start-Up from Shutdown ILIM = 475mA RLIM = 116k 0.6 0.4 3606B G17 200μs/DIV VIN = 5V, VOUT = 3.3V RLOAD = 7Ω CLOAD = 4.7μF 3.0 2ms/DIV 3606B G18 VIN = 5V, VOUT = 3.4V RL = NO LOAD, CL = 4.4mF CLIM = 2200pF, ILIM = 500mA 0.2 0 0 100 200 300 400 IIN (mA) 500 600 3606B G18b Average Input Current Limit vs Temperature Load Step Load Step 8 VIN = 5V 6 ILIM = 475mA VOUT 200mV/DIV AC-COUPLED IINLIM ERROR (%) 4 VOUT 200mV/DIV AC-COUPLED 2 0 –2 –4 IL 1A/DIV ILOAD 1A/DIV ILOAD 1A/DIV 20μs/DIV –6 –8 –50 IL 1A/DIV –25 0 25 50 75 TEMPERATURE (°C) 100 125 VIN = 5V, VOUT = 3.3V ILOAD = 0A TO 800mA COUT = 100μF, CF = 20pF 3606B G20 20μs/DIV 3606B G21 VIN = 5V, VOUT = 1.8V ILOAD = 80mA TO 800mA COUT = 100μF, CF = 20pF 3606B G19 3606bfa 5 LTC3606B PIN FUNCTIONS GND (Pins 1, 3, Exposed Pad Pin 9): Ground. Connect to the (–) terminal of COUT, and the (–) terminal of CIN. The Exposed Pad must be soldered to PCB. PGOOD (Pin 6): Open-Drain Logic Output. PGOOD is pulled to ground if the voltage on the VFB pin is not within power good threshold. RLIM (Pin 2): Average Input Current Limit Program Pin. Place a resistor and capacitor in parallel from this pin to ground. RUN (Pin 7): Regulator Enable. Forcing this pin to VIN enables regulator, while forcing it to GND causes regulator to shut down. SW (Pin 4): Regulator Switch Node Connection to the Inductor. This pin swings from VIN to GND. VFB (Pin 8): Regulator Output Feedback. Receives the feedback voltage from the external resistive divider across the regulator output. Nominal voltage for this pin is 0.6V. VIN (Pin 5): Main Power Supply. Must be closely decoupled to GND. FUNCTIONAL DIAGRAM RUN 7 0.6V REF OSC + – OSC 2 RLIM 1V MIN CLAMP 5 VIN SLOPE COMP VFB – – + 8 EA ITH – – 0.6V VSLEEP Q RS LATCH SOFT-START + – + – 6 + ICOMP + S PGOOD SLEEP R + Q SWITCHING LOGIC AND BLANKING CIRCUIT 0.654V ICOMP – ANTI SHOOTTHRU 4 SW VFB 0.546V + IRCMP – SHUTDOWN 9 GND 3606B FD 3606bfa 6 LTC3606B OPERATION The LTC3606B uses a constant-frequency, current mode architecture. The operating frequency is set at 2.25MHz. The output voltage is set by an external resistor divider returned to the VFB pins. An error amplifier compares the divided output voltage with a reference voltage of 0.6V and regulates the peak inductor current accordingly. The LTC3606B continuously monitors the input current via the voltage drop across the RDS(ON) of the internal P-channel MOSFET. When the input current exceeds the programmed input current limit set by an external resistor, RLIM , the regulator’s input current is limited. The regulator output voltage will drop to meet output current demand and to maintain constant input current. Main Control Loop During normal operation, the top power switch (P-channel MOSFET) is turned on at the beginning of a clock cycle when the VFB voltage is below the reference voltage. The current into the inductor and the load increases until the peak inductor current (controlled by ITH) is reached. The RS latch turns off the synchronous switch and energy stored in the inductor is discharged through the bottom switch (N-channel MOSFET) into the load until the next clock cycle begins, or until the inductor current begins to reverse (sensed by the IRCMP comparator). The peak inductor current is controlled by the internally compensated ITH voltage, which is the output of the error amplifier. This amplifier regulates the VFB pin to the internal 0.6V reference by adjusting the peak inductor current accordingly. When the input current limit is engaged, the peak inductor current will be lowered, which then reduces the switching duty cycle and VOUT. This allows the input voltage to stay regulated when its programmed current output capability is met. Light Load Operation The LTC3606B will automatically transition from continuous operation to the pulse-skipping operation when the load current is low. The inductor current is not fixed during the pulse-skipping mode which allows the LTC3606B to switch at constant-frequency down to very low currents, where it will begin skipping pulses to maintain output regulation. This mode of operation exhibits low output ripple as well as low audio noise and reduced RF interference while providing reasonable low current efficiency. Dropout Operation When the input supply voltage decreases toward the output voltage the duty cycle increases to 100%, which is the dropout condition. In dropout, the PMOS switch is turned on continuously with the output voltage being equal to the input voltage minus the voltage drops across the internal P-channel MOSFET and the inductor. An important design consideration is that the RDS(ON) of the P-channel switch increases with decreasing input supply voltage (see the Typical Performance Characteristics section). Therefore, the user should calculate the worstcase power dissipation when the LTC3606B is used at 100% duty cycle with low input voltage (see Thermal Considerations in the Applications Information section). Soft-Start In order to minimize the inrush current on the input bypass capacitor, the LTC3606B slowly ramps up the output voltage during start-up. Whenever the RUN pin is pulled high, the corresponding output will ramp from zero to full-scale over a time period of approximately 750μs. This prevents the LTC3606B from having to quickly charge the output capacitor and thus supplying an excessive amount of instantaneous current. When the output is loaded heavily, for example, with millifarad of capacitance, it may take longer than 750μs to charge the output to regulation. If the output is still low after the soft-start time, the LTC3606B will try to quickly charge the output capacitor. In this case, the input current limit (after it engages) can prevent excessive amount of instantaneous current that is required to quickly charge the output. See the Start-Up from Shutdown curve (CL = 4.4mF)in the Typical Performance Characteristics section. After input current limit is engaged, the output slowly ramps up to regulation while limited by its 500mA of input current. 3606bfa 7 LTC3606B OPERATION Short-Circuit Protection Programming Input Current Limit When either regulator output is shorted to ground, the corresponding internal N-channel switch is forced on for a longer time period for each cycle in order to allow the inductor to discharge, thus preventing inductor current runaway. This technique has the effect of decreasing switching frequency. Once the short is removed, normal operation resumes and the regulator output will return to its nominal voltage. Selection of one external RLIM resistor will program the input current limit. The current limit can be programmed from 200mA up to IPEAK current. As the input current increases, RLIM voltage will follow. When RLIM reaches the internal comparator threshold of 1V, the power PFET on-time will be shortened, thereby, limiting the input current. Input Current Limit Internal current sense circuitry measures the inductor current through the voltage drop across the power PFET switch and forces the same voltage across the small sense PFET. The voltage across the small sense PFET generates a current representing 1/55,000th of the inductor current during the on-cycle. The current out of RLIM pin is the representation of the inductor current, which can be expressed in the following equation. IRLIM = IOUT • D1 • K1 where D1 = VOUT1/VIN is the duty cycle. K1 is the ratio RDS(ON) (power PFET)/RDS(ON)(sense PFET). The ratio of the power PFET to the sense PFET is trimmed to within 2%. Given that both PFETs are carefully laid out and matched, their temperature and voltage coefficient effects will be similar and their terms be canceled out in the equation. In that case, the constant K1 will only be dependent on area scaling, which is trimmed to within 2%. Thus, the IRLIM current will track the input current very well over varying temperature and VIN. The RLIM pin can be grounded to disable input current limit function. Use the following equation to select the RLIM resistance that corresponds to the input current limit. RLIM = 55k / IDC IDC is the input current (at VIN) to be limited. The following are some RLIM values with the corresponding current limit. RLIM IDC 91.6k 600mA 110k 500mA 137.5k 400mA Selection of CLIM Capacitance Since IRLIM current is a function of the inductor current, its dependency on the duty cycle cannot be ignored. Thus, a CLIM capacitor is needed to integrate the IRLIM current and smooth out transient currents. The LTC3606B is stable with any size capacitance >100pF at the RLIM pin. Each application input current limit will call for different CLIM value to optimize its response time. Using a large CLIM capacitor requires longer time for the RLIM pin voltage to charge. For example, consider the application 500mA input current limit, 5V input and 1A, 2.5V output with a 50% duty cycle. When an instantaneous 1A output pulse is applied, the current out of the RLIM pin becomes 1A/55k = 18.2μA during the 50% on-time or 9.1μA full duty cycle. With a CLIM capacitor of 1μF, RLIM of 116k, and using I = CdV/dt, it will take 110ms for CLIM to charge from 0V to 1V. This is the time after which the LTC3606B will start input current limiting. Any current within this time must be considered in each application to determine if it is tolerable. 3606bfa 8 LTC3606B OPERATION Figure 1a shows VIN (IIN) current below input current limit with a CLIM capacitor of 0.1μF. When the load pulse is applied, under the specified condition, ILIM current is 1.1A/55k • 0.66 = 13.2μA, where 0.66 is the duty cycle. It will take a little more than 7.5ms to charge the CLIM capacitor from 0V to 1V, after which the LTC3606B begins to limit input current. The IIN current is not limited during this 7.5ms time and is more than 725mA. This current transient may cause the input supply to temporarily droop if the supply current compliance is exceeded, but recovers after the input current limit engages. The output will continue to deliver the required current load while the output voltage droops to allow the input voltage to remain regulated during input current limit. and the output must deliver the required current load. This may cause the input voltage to droop if the current compliance is exceeded. Depending on how long this time is, the VIN supply decoupling capacitor can provide some of this current before VIN droops too much. In applications with a bigger VIN supply decoupling capacitor and where VIN supply is allow to droop closer to dropout, the CLIM capacitor can be increased slightly. This will delay the start of input current limit and artificially regulated VOUT before input current limit is engaged. In this case, within the 577μs load pulse, the VOUT voltage will stay artificially regulated for 92μs out of the total 577μs before the input current limit activates. This approach may be used if a faster recovery on the output is desired. For applications with short load pulse duration, a smaller CLIM capacitor may be the better choice as in the example shown in Figure 1b. In this example, a 577μs, 0A to 2A output pulse is applied once every 4.7ms. A CLIM capacitor of 2.2nF requires 92μs for VRLIM to charge from 0V to 1V. During this 92μs, the input current limit is not yet engaged Selecting a very small CLIM will speed up response time but it can put the device within threshold of interfering with normal operation and input current limit in every few switching cycles. This may be undesirable in terms of noise. Use 2πRC >> 100/clock frequency (2.25MHz) as a starting point, R being RLIM, C being CLIM. VOUT 2V/DIV VOUT 200mV/DIV IIN 500mA/DIV VIN AC-COUPLED 1V/DIV VRLIM 1V/DIV IOUT 500mA/DIV IIN 500mA/DIV IL 1A/DIV 50ms/DIV 3606B F01a VIN = 5V, 500mA COMPLIANT RLIM = 116k, CLIM = 0.1μF ILOAD = 0A to 1.1A, COUT = 2.2mF, VOUT = 3.3V ILIM = 475mA Figure 1a. Input Current Limit Within 100ms Load Pulses 1ms/DIV 3606B F01b VIN = 5V, 500mA COMPLIANT RLIM = 116k, CLIM = 2200pF ILOAD = 0A to 2A, COUT = 2.2mF, VOUT = 3.3V ILIM = 475mA Figure 1b. Input Current Limit Within 577μs, 2A Repeating Load Pulses 3606bfa 9 LTC3606B APPLICATIONS INFORMATION A general LTC3606B application circuit is shown in Figure 2. External component selection is driven by the load requirement, and begins with the selection of the inductor L. Once the inductor is chosen, CIN and COUT can be selected. Inductor Selection Although the inductor does not influence the operating frequency, the inductor value has a direct effect on ripple current. The inductor ripple current ΔIL decreases with higher inductance and increases with higher VIN or VOUT : V V IL = OUT • 1 OUT (1) fO • L VIN Accepting larger values of ΔIL allows the use of low inductances, but results in higher output voltage ripple, greater core losses, and lower output current capability. A reasonable starting point for setting ripple current is 40% of the maximum output load current. So, for a 800mA regulator, ΔIL = 320mA (40% of 800mA). The inductor value will also have an effect on Burst Mode operation. The transition to low current operation begins when the peak inductor current falls below a level set by the internal burst clamp. Lower inductor values result in higher ripple current which causes the transition to occur at lower load currents. This causes a dip in efficiency in the upper range of low current operation. Furthermore, lower inductance values will cause the bursts to occur with increased frequency. L1 VIN 2.5V TO 5.5V VIN RPGD CIN VOUT SW LTC3606B CF RUN COUT PGOOD VFB RLIM PGOOD GND RLIM R2 R1 CLIM 3606B F02 Figure 2. LTC3606B General Schematic Inductor Core Selection Different core materials and shapes will change the size/ current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. The choice of which style inductor to use often depends more on the price versus size requirements, and any radiated field/EMI requirements, than on what the LTC3606B requires to operate. Table 1 shows some typical surface mount inductors that work well in LTC3606B applications. Table 1. Representative Surface Mount Inductors MANUFACTURER PART NUMBER MAX DC VALUE CURRENT DCR HEIGHT Coilcraft LPS4012-152ML LPS4012-222ML LPS4012-332ML LPS4012-472ML LPS4018-222ML LPS4018-332ML LPS4018-472ML 1.5μH 2.2μH 3.3μH 4.7μH 2.2μH 3.3μH 4.7μH 2200mA 1750mA 1450mA 1450mA 2300mA 2000mA 1800mA 0.070Ω 0.100Ω 0.100Ω 0.170Ω 0.070Ω 0.080Ω 0.125Ω 1.2mm 1.2mm 1.2mm 1.2mm 1.8mm 1.8mm 1.8mm FDK FDKMIPF2520D FDKMIPF2520D FDKMIPF2520D 4.7μH 3.3μH 2.2μH 1100mA 1200mA 1300mA 0.11Ω 0.1Ω 0.08Ω 1mm 1mm 1mm LQH32CN4R7M23 4.7μH 450mA 0.2Ω 2mm ELT5KT4R7M 4.7μH 950mA 0.2Ω 1.2mm CDRH2D18/LD CDH38D11SNP3R3M CDH38D11SNP2R2M 4.7μH 3.3μH 630mA 1560mA 0.086Ω 0.115Ω 2mm 1.2mm 2.2μH 1900mA 0.082Ω 1.2mm 2.2μH 2.2μH 3.3μH 2.2μH 4.7μH 510mA 530mA 410mA 1100mA 750mA 0.13Ω 0.33Ω 0.27Ω 0.1Ω 0.19Ω 1.6mm 1.25mm 1.6mm 1mm 1mm 4.7μH 700mA 0.28Ω 1mm 3.3μH 870mA 0.17Ω 1mm 2.2μH 1000mA 0.12Ω 1mm 2.2μH 1500mA 0.076Ω 1.2mm 3.3μH 1700mA 0.095Ω 1.2mm 2.2μH 2300mA 0.059Ω 1.4mm Murata Panasonic Sumida Taiyo Yuden CB2016T2R2M CB2012T2R2M CB2016T3R3M NR30102R2M NR30104R7M TDK VLF3010AT4R7MR70 VLF3010AT3R3MR87 VLF3010AT2R2M1R0 VLF4012AT-2R2 M1R5 VLF5012ST-3R3 M1R7 VLF5014ST-2R2 M2R3 3606bfa 10 LTC3606B APPLICATIONS INFORMATION Input Capacitor (CIN) Selection In continuous mode, the input current of the converter is a square wave with a duty cycle of approximately VOUT / VIN . To prevent large voltage transients, a low equivalent series resistance (ESR) input capacitor sized for the maximum RMS current must be used. The maximum RMS capacitor current is given by: IRMS IMAX VOUT (VIN VOUT ) VIN Where the maximum average output current IMAX equals the peak current minus half the peak-to-peak ripple current, IMAX = ILIM – ΔIL /2. This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case is commonly used to design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or choose a capacitor rated at a higher temperature than required. Several capacitors may also be paralleled to meet the size or height requirements of the design. An additional 0.1μF to 1μF ceramic capacitor is also recommended on VIN for high frequency decoupling when not using an all-ceramic capacitor solution. Output Capacitor (COUT) Selection The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. The output ripple ΔVOUT is determined by: 1 VOUT IL ESR+ 8fOCOUT where fO = operating frequency, COUT = output capacitance and ΔIL = ripple current in the inductor. For a fixed output voltage, the output ripple is highest at maximum input voltage since ΔIL increases with input voltage. If tantalum capacitors are used, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalum. These are specially constructed and tested for low ESR so they give the lowest ESR for a given volume. Other capacitor types include Sanyo POSCAP, Kemet T510 and T495 series, and Sprague 593D and 595D series. Consult the manufacturer for other specific recommendations. Using Ceramic Input and Output Capacitors Higher values, lower cost ceramic capacitors are now becoming available in smaller case sizes. Their high ripple current, high voltage rating and low ESR make them ideal for switching regulator applications. Because the LTC3606B control loop does not depend on the output capacitor’s ESR for stable operation, ceramic capacitors can be used freely to achieve very low output ripple and small circuit size. However, care must be taken when ceramic capacitors are used at the input. When a ceramic capacitor is used at the input and the power is supplied by a wall adapter through long wires, a load step at the output can induce ringing at the input, VIN. At best, this ringing can couple to the output and be mistaken as loop instability. At worst, a sudden inrush of current through the long wires can potentially cause a voltage spike at VIN, large enough to damage the part. For more information, see Application Note 88. When choosing the input and output ceramic capacitors, choose the X5R or X7R dielectric formulations. These dielectrics have the best temperature and voltage characteristics of all the ceramics for a given value and size. 3606bfa 11 LTC3606B APPLICATIONS INFORMATION Setting the Output Voltage The LTC3606B regulates the VFB pin to 0.6V during regulation. Thus, the output voltage is set by a resistive divider, Figure 2, according to the following formula: VOUT = 0.6V 1+ R2 R1 (2) The output voltage settling behavior is related to the stability of the closed-loop system and will demonstrate the actual overall supply performance. For a detailed explanation of optimizing the compensation components, including a review of control loop theory, refer to Application Note 76. To improve the frequency response of the main control loop, a feedback capacitor (CF) may also be used. Great care should be taken to route the VFB line away from noise sources, such as the inductor or the SW line. In some applications, a more severe transient can be caused by switching in loads with large (>1μF) input capacitors. The discharged input capacitors are effectively put in parallel with COUT, causing a rapid drop in VOUT. No regulator can deliver enough current to prevent this problem if the switch connecting the load has low resistance and is driven quickly. The solution is to limit the turn-on speed of the load switch driver. A Hot Swap™ controller is designed specifically for this purpose and usually incorporates current limiting, short-circuit protection, and soft-starting. Checking Transient Response Efficiency Considerations The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ΔILOAD • ESR, where ESR is the effective series resistance of COUT. ΔILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Percent efficiency can be expressed as: The initial output voltage step may not be within the bandwidth of the feedback loop, so the standard second order overshoot/DC ratio cannot be used to determine the phase margin. In addition, feedback capacitors (CF) can be added to improve the high frequency response, as shown in Figure 2. Capacitor CF provides phase lead by creating a high frequency zero with R2 which improves the phase margin. Although all dissipative elements in the circuit produce losses, four sources usually account for the losses in LTC3606B circuits: 1) VIN quiescent current, 2) switching losses, 3) I2R losses, 4) other system losses. Keeping the current small (< 10μA) in these resistors maximizes efficiency, but making it too small may allow stray capacitance to cause noise problems or reduce the phase margin of the error amp loop. % Efficiency = 100% – (L1 + L2 + L3 + ...) where L1, L2, etc., are the individual losses as a percentage of input power. 1. The VIN current is the DC supply current given in the Electrical Characteristics which excludes MOSFET driver and control currents. VIN current results in a small (<0.1%) loss that increases with VIN, even at no load. 3606bfa 12 LTC3606B APPLICATIONS INFORMATION 2. The switching current is the sum of the MOSFET driver and control currents. The MOSFET driver current results from switching the gate capacitance of the power MOSFETs. Each time a MOSFET gate is switched from low to high to low again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is a current out of VIN that is typically much larger than the DC bias current. In continuous mode, IGATECHG = fO(QT + QB), where QT and QB are the gate charges of the internal top and bottom MOSFET switches. The gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. Thermal Considerations 3. I2R losses are calculated from the DC resistances of the internal switches, RSW , and external inductor, RL. In continuous mode, the average output current flows through inductor L, but is “chopped” between the internal top and bottom switches. Thus, the series resistance looking into the SW pin is a function of both top and bottom MOSFET RDS(ON) and the duty cycle (DC) as follows: where PD is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature, TJ, is given by: RSW = (RDS(ON)TOP) • (DC) + (RDS(ON)BOT) • (1– DC) The RDS(ON) for both the top and bottom MOSFETs can be obtained from the Typical Performance Characteristics curves. Thus, to obtain I2R losses: I2R losses = IOUT 2 • (RSW + RL) 4. Other “hidden” losses, such as copper trace and internal battery resistances, can account for additional efficiency degradations in portable systems. It is very important to include these “system” level losses in the design of a system. The internal battery and fuse resistance losses can be minimized by making sure that CIN has adequate charge storage and very low ESR at the switching frequency. Other losses, including diode conduction losses during dead-time, and inductor core losses, generally account for less than 2% total additional loss. In a majority of applications, the LTC3606B does not dissipate much heat due to its high efficiency. In the unlikely event that the junction temperature somehow reaches approximately 150°C, both power switches will be turned off and the SW node will become high impedance. The goal of the following thermal analysis is to determine whether the power dissipated causes enough temperature rise to exceed the maximum junction temperature (125°C) of the part. The temperature rise is given by: TRISE = PD • θJA TJ = TRISE + TAMBIENT As a worst-case example, consider the case when the LTC3606B is in dropout at an input voltage of 2.7V with a load current of 800mA and an ambient temperature of 70°C. From the Typical Performance Characteristics graph of Switch Resistance, the RDS(ON) of the switch is 0.33Ω. Therefore, the power dissipated is: PD = IOUT 2 • RDS(ON) = 212mV Given that the thermal resistance of a properly soldered DFN package is approximately 40°C/W, the junction temperature of an LTC3606B device operating in a 70°C ambient temperature is approximately: TJ = (0.212W • 40°C/W) + 70°C = 78.5°C which is well below the absolute maximum junction temperature of 125°C. 3606bfa 13 LTC3606B APPLICATIONS INFORMATION should be routed away from noisy components and traces, such as the SW line (Pin 4), and their trace length should be minimized. PC Board Layout Considerations When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC3606B. These items are also illustrated graphically in the layout diagrams of Figures 3a and 3b. Check the following in your layout: 4. Keep sensitive components away from the SW pin, if possible. The input capacitor CIN, CLIM and the resistors R1, R2, and RLIM should be routed away from the SW traces and the inductors. 1. Does the capacitor CIN connect to the power VIN (Pin 5) and GND (Pin 9) as closely as possible? This capacitor provides the AC current of the internal power MOSFETs and their drivers. 5. A ground plane is preferred, but if not available, keep the signal and power grounds segregated with small signal components returning to the GND pin at a single point. These ground traces should not share the high current path of CIN or COUT. 2. Are the respective COUT and L closely connected? The (–) plate of COUT returns current to GND and the (–) plate of CIN. 6. Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power components. These copper areas should be connected to VIN or GND. 3. The resistor divider, R1 and R2, must be connected between the (+) plate of COUT and a ground sense line terminated near GND (Pin 9). The feedback signal VFB L1 VIN 2.5V TO 5.5V VIN CIN SW RPGD VOUT CF LTC3606B RUN PGOOD GND RLIM VFB R2 RLIM COUT R1 CLIM 3606B F03a BOLD LINES INDICATE HIGH CURRENT PATHS Figure 3a. LTC3606B Layout Diagram (See Board Layout Checklist) 3606bfa 14 LTC3606B APPLICATIONS INFORMATION VIA TO VOUT SENSE GND GND VFB RLIM RUN GND PGOOD SW VIN VIN SW GND VOUT Figure 3b. LTC3606B Suggested Layout 3606bfa 15 LTC3606B APPLICATIONS INFORMATION of CIN = 10μF should suffice, if the source impedance is very low. Design Example As a design example, consider using the LTC3606B in a USB-GSM application, with VIN = 5V, IINMAX = 500mA, with the output charging a SuperCap of 4.4mF. The load requires 800mA in active mode and 1mA in standby mode. The output voltage VOUT = 3.4V. The feedback resistors program the output voltage. To maintain high efficiency at light loads, the current in these resistors should be kept small. Choosing 10μA with the 0.6V feedback voltage makes R1~60k. A close standard 1% resistor is 59k. Using Equation (2). First, calculate the inductor value for about 40% ripple current (320mA in this example) at maximum VIN. Using a derivation of Equation (1): L1= R2 = 3.4V 3.4V • 1 =1.51μH 2.25MHz • (320mA) 5V VOUT 1 • R1= 276k, 280k for 1% 0.6 A feedforward capacitor is not used since the 4.4mF SuperCap will inhibit any fast output voltage transients. Figure 4 shows the complete schematic for this example, along with the efficiency curve and transient response. Input current limit is set at 475mA average current, RLIM = 116k, CLIM = 2200pF. See Programming Input Current Limit section for selecting RLIM and Selection of CLIM Capacitance section for CLIM. For the inductor, use the closest standard value of 1.5μH. The 4.4mF supercaps are used to deliver the required 2A pulses to power the RF power amplifiers, while the LTC3606B recharges the supercap after the pulse ends, see Figure 4c. As for the input capacitor, a typical value L1 1.5μH VIN USB INPUT 5V CIN 10μF RPGD 499k LTC3606B RUN R2 280k PGOOD VFB RLIM PGOOD GND CLIM 2200pF VOUT 3.4V AT 800mA SW VIN + COUT 2.2mF s2 SuperCap R1 59k RLIM 116k ILIM = 475mA CIN: AVX 08056D106KAT2A COUT: VISHAY 592D228X96R3X2T20H L1: COILCRAFT LPS4012-152ML 3606B F04 Figure 4a. Design Example Circuit 3606bfa 16 LTC3606B APPLICATIONS INFORMATION 100 90 10 VOUT = 3.4V 1 70 60 0.1 50 40 30 POWER LOSS (W) EFFICIENCY (%) 80 0.01 20 VIN = 3.6V VIN = 4.2V VIN = 5V 10 0 0.0001 0.001 0.01 0.1 OUTPUT CURRENT (A) 0.001 1 3606B F04b Figure 4b. Efficiency vs Output Current VOUT 200mV/DIV VIN 1V/DIV AC-COUPLED IOUT 500mA/DIV IIN 500mA/DIV 1ms/DIV VIN = 5V, 500mA COMPLIANT RLIM = 116kΩ, CLIM = 2200pF ILOAD = 0A TO 2A, COUT = 4.4mF, VOUT = 3.4V ILIM = 475mA 3606B F04c Figure 4c. Transient Response 3606bfa 17 LTC3606B PACKAGE DESCRIPTION DD Package 8-Lead Plastic DFN (3mm × 3mm) (Reference LTC DWG # 05-08-1698) 0.70 p0.05 3.5 p0.05 1.65 p0.05 2.10 p0.05 (2 SIDES) PACKAGE OUTLINE 0.25 p 0.05 0.50 BSC 2.38 p0.05 RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 3.00 p0.10 (4 SIDES) R = 0.125 TYP 5 0.40 p 0.10 8 1.65 p 0.10 (2 SIDES) PIN 1 TOP MARK (NOTE 6) (DD8) DFN 0509 REV C 0.200 REF 0.75 p0.05 4 0.25 p 0.05 1 0.50 BSC 2.38 p0.10 0.00 – 0.05 BOTTOM VIEW—EXPOSED PAD NOTE: 1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON TOP AND BOTTOM OF PACKAGE 3606bfa 18 LTC3606B REVISION HISTORY REV DATE DESCRIPTION PAGE NUMBER A 3/10 Changes to Electrical Characteristics 3 3606bfa Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 19 LTC3606B TYPICAL APPLICATIONS 800mA Buck Converter, ILIM = 500mA L1 1.5μH VIN USB INPUT 5V VIN CIN 10μF RPGD 499k LTC3606B RUN PGOOD VFB RLIM PGOOD GND CLIM 1000pF VOUT 3.4V AT 800mA SW R2 1210k + COUT 2.2mF s2 SuperCap R1 255k RLIM 110k L1: COILCRAFT LPS4012-152ML CIN: AVX 08056D106KAT2A COUT: VISHAY 592D228X96R3X2T20H 3606B TA02 800mA Buck Converter, ILIM = 475mA or Disabled L1 1.5μH VIN USB INPUT 5V VIN CIN 10μF RPGD 499k LTC3606B RUN PGOOD VFB RLIM PGOOD GND ILIM DISABLE RLIM 116k VOUT 3.4V AT 800mA SW CLIM 2200pF CIN: AVX 08056D106KAT2A COUT: VISHAY 592D228X96R3X2T20H R2 1210k + COUT 2.2mF ×2 SuperCap R1 255k L1: COILCRAFT LPS4012-152ML 3606B TA03 RELATED PARTS PART NUMBER LTC3619/LTC3619B LTC3127 LTC3125 LTC3417A/ LTC3417A-2 LTC3407A/ LTC3407A-2 LTC3548/LTC3548-1/ LTC3548-2 LTC3546 DESCRIPTION Dual 400mA and 800mA IOUT, 2.25MHz, Synchronous Step-Down DC/DC Converter 1.2A IOUT, 1.6MHz, Synchronous Buck-Boost DC/DC Converter with Adjustable Input Current Limit 1.2A IOUT, 1.6MHz, Synchronous Boost DC/DC Converter with Adjustable Input Current Limit Dual 1.5A/1A, 4MHz, Synchronous Step-Down DC/DC Converter Dual 600mA/600mA, 1.5MHz, Synchronous Step-Down DC/DC Converter Dual 400mA and 800mA IOUT, 2.25MHz, Synchronous Step-Down DC/DC Converter Dual 3A/1A, 4MHz, Synchronous Step-Down DC/DC Converter COMMENTS 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V, IQ = 50μA, ISD < 1μA, MS10E, 3mm × 3mm DFN-10 94% Efficiency, VIN(MIN) = 1.8V, VIN(MAX) = 5.5V, VOUT(MAX) = 5.25V, IQ = 18μA, ISD < 1μA, 3mm × 3mm DFN-MSOP10E 94% Efficiency, VIN(MIN) = 1.8V, VIN(MAX) = 5.5V, VOUT(MAX) = 5.25V, IQ = 15μA, ISD < 1μA, 2mm × 3mm DFN-8 95% Efficiency, VIN(MIN) = 2.3V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.8V, IQ = 125μA, ISD = <1μA, TSSOP-16E, 3mm × 5mm DFN-16 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V, IQ = 40μA, ISD = <1μA, MS10E, 3mm × 3mm DFN-10 95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V, IQ = 40μA, ISD = <1μA, MS10E, 3mm × 3mm DFN-10 95% Efficiency, VIN(MIN) = 2.3V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V, IQ = 160μA, ISD = <1μA, 4mm × 5mm QFN-28 3606bfa 20 Linear Technology Corporation LT 0310 REV A • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2009