LINER LTC3606BIDD

LTC3606B
800mA Synchronous
Step-Down DC/DC with
Average Input Current Limit
DESCRIPTION
FEATURES
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Programmable Average Input Current Limit:
±5% Accuracy
Step-Down Output: Up to 96% Efficiency
Low Noise Pulse-Skipping Operation at Light Loads
Input Voltage Range: 2.5V to 5.5V
Output Voltage Range: 0.6V to 5V
2.25MHz Constant-Frequency Operation
Power Good Output Voltage Monitor
Low Dropout Operation: 100% Duty Cycle
Internal Soft-Start
Current Mode Operation for Excellent Line and Load
Transient Response
±2% Output Voltage Accuracy
Short-Circuit Protected
Shutdown Current ≤ 1μA
Available in Small Thermally Enhanced 8-Lead
3mm × 3mm DFN Package
APPLICATIONS
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The LTC®3606B is an 800mA monolithic synchronous
buck regulator using a constant frequency current mode
architecture.
The input supply voltage range is 2.5V to 5.5V, making it
ideal for Li-Ion and USB powered applications. 100% duty
cycle capability provides low dropout operation, extending
the run time in battery-operated systems. Low output
voltages are supported with the 0.6V feedback reference
voltage. The LTC3606B can supply 800mA output current.
The LTC3606B’s programmable average input current
limit is ideal for USB applications and for point-of-load
power supplies because the LTC3606B’s limited input
current will still allow its output to deliver high peak load
currents without collapsing the input supply. The operating
frequency is internally set at 2.25MHz allowing the use of
small surface mount inductors. Internal soft-start reduces
in-rush current during start-up. The LTC3606B is available
in an 8-Lead 3mm × 3mm DFN package.
L, LT, LTC, LTM, Linear Technology, the Linear logo and Burst Mode are registered trademarks
and Hot Swap is a trademark of Linear Technology Corporation. All other trademarks are the
property of their respective owners. Protected by U.S.Patents, including 5481178, 6127815,
6304066, 6498466, 6580258, 6611131.
High Peak Load Current Applications
USB Powered Devices
Supercapacitor Charging
Radio Transmitters and Other Handheld Devices
TYPICAL APPLICATION
Monolithic Buck Regulator with Input Current Limit
1.5μH
VIN 3.4V
TO 5.5V
VIN
CIN
10μF
RUN
+
PGOOD
VFB
RLIM
PGOOD
GND
1000pF
VOUT
3.4V AT
800mA
SW
LTC3606B
499k
116k
1210k
GSM Pulse Load
2.2mF
s2
SuperCap
VIN
AC-COUPLED
1V/DIV
IOUT
500mA/DIV
255k
3606B TA01
ILIM = 475mA
VOUT
200mV/DIV
IIN
500mA/DIV
1ms/DIV
3606B TA01b
VIN = 5V, 500mA COMPLIANT
ILOAD = 0A to 2.2A
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LTC3606B
ABSOLUTE MAXIMUM RATINGS
PIN CONFIGURATION
(Note 1)
TOP VIEW
Input Supply Voltage (VIN) ........................... –0.3V to 6V
VFB ................................................... –0.3V to VIN + 0.3V
RUN, RLIM ....................................... –0.3V to VIN + 0.3V
SW ................................................... –0.3V to VIN + 0.3V
PGOOD............................................. –0.3V to VIN + 0.3V
P-Channel SW Source Current (DC) (Note 2) ..............1A
N-Channel SW Source Current (DC) (Note 2) .............1A
Peak SW Source and Sink Current (Note 2) ............. 2.7A
Operating Junction Temperature Range
(Notes 3, 6, 8) ........................................ –40°C to 125°C
Storage Temperature Range .................. –65°C to 125°C
Reflow Peak Body Temperature ............................ 260°C
GND
1
RLIM
2
GND
3
SW
4
8 VFB
9
GND
7 RUN
6 PGOOD
5 VIN
DD PACKAGE
8-LEAD (3mm s 3mm) PLASTIC DFN
TJMAX = 125°C, θJA = 40°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH
TAPE AND REEL
PART MARKING*
PACKAGE DESCRIPTION
TEMPERATURE RANGE
LTC3606BEDD#PBF
LTC3606BEDD#TRPBF
LFMB
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 85°C
LTC3606BIDD#PBF
LTC3606BIDD#TRPBF
LFMB
8-Lead (3mm × 3mm) Plastic DFN
–40°C to 125°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container.
Consult LTC Marketing for information on non-standard lead based finish parts.
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/
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LTC3606B
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
junction temperature range, otherwise specifications are at TA = 25°C, VIN = 5V, unless otherwise noted.
SYMBOL
PARAMETER
VIN
VIN Operating Voltage Range
CONDITIONS
MIN
VUV
VIN Undervoltage Lockout
IFB
Feedback Pin Input Current
VFBREG
Feedback Voltage
LTC3606BE, –40°C < TJ < 85°C (Note 7)
LTC3606BI, –40°C < TJ < 125°C (Note 7)
ΔVLINEREG
VFB Line Regulation
ΔVLOADREG
IS
l
TYP
MAX
UNITS
5.5
V
2.5
V
±30
nA
0.600
0.600
0.612
0.618
V
V
VIN = 2.5V to 5.5V (Note 7)
0.01
0.25
%/V
VFB Load Regulation
ILOAD = 0mA to 800mA (Note 7)
0.5
Supply Current
Active Mode (Note 4)
Shutdown
VFB = 0.95 × VFBREG
VRUN = 0V, VIN = 5.5V
420
650
1
μA
μA
fOSC
Oscillator Frequency
VFB = VFBREG
1.8
2.25
2.7
MHz
ILIM(PEAK)
Peak Switch Current Limit
VIN = 5V, VFB < VFBREG , Duty Cycle <35%
1800
2400
IINLIM
Input Average Current Limit
RLIM = 116k
RLIM = 116k, LTC3606BE
RLIM = 116k, LTC3606BI
450
437
427
475
475
475
RDS(ON)
Main Switch On-Resistance (Note 5)
VIN = 5V, ISW = 100mA
Synchronous Switch On-Resistance (Note 5) VIN = 5V, ISW = 100mA
0.27
0.25
ISW(LKG)
Switch Leakage Current
VIN = 5V, VRUN = 0V
0.01
1
μA
tSOFTSTART
Soft-Start Time
VFB from 0.06V to 0.54V
0.3
0.95
1.3
ms
VRUN
RUN Threshold High
0.4
1
1.2
V
IRUN
RUN Leakage Current
0V ≤ VRUN ≤ 5V
0.01
1
μA
PGOOD
Power Good Threshold
Entering Window
VFB Ramping Up
VFB Ramping Down
Leaving Window
VFB Ramping Up
VFB Ramping Down
PGOOD Blanking Power Good Blanking Time
RPGOOD
Power Good Pull-Down On-Resistance
IPGOOD
PGOOD Leakage Current
VIN Low to High
2.5
l
2.1
l
l
l
l
l
l
l
0.588
0.582
l
–5
5
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: Guaranteed by long term current density limitations.
Note 3: The LTC3606BE is guaranteed to meet performance specifications
from 0°C to 85°C. Specifications over the –40°C to 85°C operating
junction temperature range are assured by design, characterization and
correlation with statistical process controls. The LTC3606BI is guaranteed
to meet specified performance over the full –40°C to 125°C operating
junction temperature range.
Note 4: Dynamic supply current is higher due to the internal gate charge
being delivered at the switching frequency.
500
513
523
15
mA
mA
mA
Ω
Ω
%
%
11
–11
90
8
VPGOOD = 5V
mA
–7
7
9
–9
PGOOD Rising and Falling, VIN = 5V
%
%
%
μs
30
Ω
±1
μA
Note 5: The switch on-resistance is guaranteed by correlation to wafer
level measurements.
Note 6: This IC includes overtemperature protection that is intended
to protect the device during momentary overload conditions. Junction
temperature will exceed 125°C when overtemperature protection is active.
Continuous operation above the specified maximum operating junction
temperature may impair device reliability.
Note 7: The converter is tested in a proprietary test mode that connects
the output of the error amplifier to the SW pin, which is connected to an
external servo loop.
Note 8: TJ is calculated from the ambient temperature TA and the power
dissipation as follows: TJ = TA + (PD)(θJA°C/W)
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LTC3606B
TYPICAL PERFORMANCE CHARACTERISTICS
TA = 25°C, VIN = 5V, unless otherwise noted.
Supply Current vs Temperature
Pulse-Skipping Mode Operation
550
SW
2V/DIV
500
Efficiency vs Input Voltage
100
RUN = VIN
ILOAD = 0A
90
80
IL
100mA/DIV
450
400
EFFICIENCY (%)
SUPPLY CURRENT (μA)
VIN = 5.5V
VOUT
50mV/DIV
ACCOUPLED
VIN = 2.7V
350
70
60
50
30
300
IOUT = 10mA
IOUT = 1mA
IOUT = 0.1mA
IOUT = 100mA
IOUT = 400mA
IOUT = 800mA
40
20
3606B G01
5μs/DIV
VIN = 5V
VOUT = 3.3V
ILOAD = 5mA
250
10
200
–50
–25
0
25
50
75
TEMPERATURE (°C)
100
VOUT = 3.3V
0
3.5
125
4
4.5
VIN (V)
5
3606B G03
3606B G02
Oscillator Frequency
vs Temperature
1.5
2.5
1.0
2.4
Switch Leakage vs Input Voltage
1000
0.5
0
–0.5
–1.0
2.3
2.2
2.1
2.0
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
VIN = 5V
1.9
–25
0
25
50
75
TEMPERATURE (°C)
100
1.8
–50
125
–25
0
25
50
75
TEMPERATURE (°C)
3606B G04
0.5
MAIN PFET RDS(ON) (Ω)
RDS(ON) (mΩ)
400
MAIN SWITCH
300
4.5
0
2.5
125
3
5.5
MAIN SWITCH
0.6
0.5
0.2
0.4
0.1
0.3
0
0.2
3606B G07
–0.1
–50
–25
25
50
75
0
TEMPERATURE (°C)
4
VIN (V)
4.5
5
5.5
Efficiency vs Load Current
0.7
100
90
VOUT = 3.3V
80
70
60
50
40
30
20
10
SYNCHRONOUS SWITCH
5
3.5
3606B G06
0.3
SYNCHRONOUS SWITCH
4
VIN (V)
SYNCHRONOUS SWITCH
200
SYNCHRONOUS NFET RDS(ON) (Ω)
VIN = 2.7V
VIN = 3.6V
VIN = 5V
0.4
500
3.5
400
Switch On-Resistance
vs Temperature
600
3
MAIN SWITCH
3606B G05
Switch On-Resistance
vs Input Voltage
200
2.5
100
600
EFFICIENCY (%)
–1.5
–50
LEAKAGE CURRENT (pA)
800
FREQUENCY (MHz)
VFB ERROR (%)
Regulated Voltage vs Temperature
5.5
100
0.1
125
3606B G09
0
0.0001
VIN = 3.6V
VIN = 4.2V
VIN = 5V
0.001
0.01
0.1
OUTPUT CURRENT (A)
1
3606B G11
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LTC3606B
TYPICAL PERFORMANCE CHARACTERISTICS
Efficiency vs Load Current
100
90
TA = 25°C, VIN = 5V, unless otherwise noted.
Load Regulation
Line Regulation
0.6
3.0
VOUT = 1.2V
2.5
0.4
80
70
60
50
40
30
10
0
0.0001
0.001
0.01
0.1
OUTPUT CURRENT (A)
1.5
1.0
0.5
0.2
0
–0.2
0
VIN = 2.7V
VIN = 3.6V
VIN = 4.2V
VIN = 5V
20
VOUT ERROR (%)
2.0
VOUT ERROR (%)
EFFICIENCY (%)
VOUT = 1.8V
ILOAD = 100mA
–1.0
1
–0.4
VOUT = 1.8V
VOUT = 2.5V
VOUT = 3.3V
–0.5
0
–0.6
2.5
100 200 300 400 500 600 700 800
LOAD CURRENT (mA)
3606B G13
3.5
4.0
VIN (V)
4.5
1.2
1.0
VOUT
2V/DIV
0.8
RLIM
1V/DIV
IL
250mA/DIV
IIN
500mA/DIV
VRLIM (V)
RUN
2V/DIV
VOUT
1V/DIV
5.5
VRLIM vs Input Current
Start-Up from Shutdown
RUN
2V/DIV
5.0
3606B G16
3606B G15
Start-Up from Shutdown
ILIM = 475mA
RLIM = 116k
0.6
0.4
3606B G17
200μs/DIV
VIN = 5V, VOUT = 3.3V
RLOAD = 7Ω
CLOAD = 4.7μF
3.0
2ms/DIV
3606B G18
VIN = 5V, VOUT = 3.4V
RL = NO LOAD, CL = 4.4mF
CLIM = 2200pF, ILIM = 500mA
0.2
0
0
100
200
300
400
IIN (mA)
500
600
3606B G18b
Average Input Current Limit
vs Temperature
Load Step
Load Step
8
VIN = 5V
6 ILIM = 475mA
VOUT
200mV/DIV
AC-COUPLED
IINLIM ERROR (%)
4
VOUT
200mV/DIV
AC-COUPLED
2
0
–2
–4
IL
1A/DIV
ILOAD
1A/DIV
ILOAD
1A/DIV
20μs/DIV
–6
–8
–50
IL
1A/DIV
–25
0
25
50
75
TEMPERATURE (°C)
100
125
VIN = 5V, VOUT = 3.3V
ILOAD = 0A TO 800mA
COUT = 100μF, CF = 20pF
3606B G20
20μs/DIV
3606B G21
VIN = 5V, VOUT = 1.8V
ILOAD = 80mA TO 800mA
COUT = 100μF, CF = 20pF
3606B G19
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LTC3606B
PIN FUNCTIONS
GND (Pins 1, 3, Exposed Pad Pin 9): Ground. Connect to
the (–) terminal of COUT, and the (–) terminal of CIN. The
Exposed Pad must be soldered to PCB.
PGOOD (Pin 6): Open-Drain Logic Output. PGOOD is pulled
to ground if the voltage on the VFB pin is not within power
good threshold.
RLIM (Pin 2): Average Input Current Limit Program Pin.
Place a resistor and capacitor in parallel from this pin to
ground.
RUN (Pin 7): Regulator Enable. Forcing this pin to VIN
enables regulator, while forcing it to GND causes regulator
to shut down.
SW (Pin 4): Regulator Switch Node Connection to the
Inductor. This pin swings from VIN to GND.
VFB (Pin 8): Regulator Output Feedback. Receives the
feedback voltage from the external resistive divider
across the regulator output. Nominal voltage for this pin
is 0.6V.
VIN (Pin 5): Main Power Supply. Must be closely decoupled to GND.
FUNCTIONAL DIAGRAM
RUN
7
0.6V REF
OSC
+
–
OSC
2 RLIM
1V
MIN
CLAMP
5 VIN
SLOPE
COMP
VFB
–
–
+
8
EA
ITH
–
–
0.6V
VSLEEP
Q
RS
LATCH
SOFT-START
+
–
+
–
6
+
ICOMP
+
S
PGOOD
SLEEP
R
+
Q
SWITCHING
LOGIC
AND
BLANKING
CIRCUIT
0.654V
ICOMP
–
ANTI
SHOOTTHRU
4 SW
VFB
0.546V
+
IRCMP
–
SHUTDOWN
9 GND
3606B FD
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LTC3606B
OPERATION
The LTC3606B uses a constant-frequency, current mode
architecture. The operating frequency is set at 2.25MHz.
The output voltage is set by an external resistor divider
returned to the VFB pins. An error amplifier compares the
divided output voltage with a reference voltage of 0.6V and
regulates the peak inductor current accordingly.
The LTC3606B continuously monitors the input current
via the voltage drop across the RDS(ON) of the internal
P-channel MOSFET. When the input current exceeds the
programmed input current limit set by an external resistor,
RLIM , the regulator’s input current is limited. The regulator
output voltage will drop to meet output current demand
and to maintain constant input current.
Main Control Loop
During normal operation, the top power switch (P-channel
MOSFET) is turned on at the beginning of a clock cycle
when the VFB voltage is below the reference voltage. The
current into the inductor and the load increases until the
peak inductor current (controlled by ITH) is reached. The
RS latch turns off the synchronous switch and energy
stored in the inductor is discharged through the bottom
switch (N-channel MOSFET) into the load until the next
clock cycle begins, or until the inductor current begins to
reverse (sensed by the IRCMP comparator).
The peak inductor current is controlled by the internally
compensated ITH voltage, which is the output of the error
amplifier. This amplifier regulates the VFB pin to the internal
0.6V reference by adjusting the peak inductor current
accordingly.
When the input current limit is engaged, the peak inductor
current will be lowered, which then reduces the switching
duty cycle and VOUT. This allows the input voltage to stay
regulated when its programmed current output capability
is met.
Light Load Operation
The LTC3606B will automatically transition from continuous
operation to the pulse-skipping operation when the load
current is low. The inductor current is not fixed during the
pulse-skipping mode which allows the LTC3606B to switch
at constant-frequency down to very low currents, where it
will begin skipping pulses to maintain output regulation.
This mode of operation exhibits low output ripple as well
as low audio noise and reduced RF interference while
providing reasonable low current efficiency.
Dropout Operation
When the input supply voltage decreases toward the
output voltage the duty cycle increases to 100%, which
is the dropout condition. In dropout, the PMOS switch is
turned on continuously with the output voltage being equal
to the input voltage minus the voltage drops across the
internal P-channel MOSFET and the inductor.
An important design consideration is that the RDS(ON)
of the P-channel switch increases with decreasing input
supply voltage (see the Typical Performance Characteristics
section). Therefore, the user should calculate the worstcase power dissipation when the LTC3606B is used at
100% duty cycle with low input voltage (see Thermal
Considerations in the Applications Information section).
Soft-Start
In order to minimize the inrush current on the input bypass
capacitor, the LTC3606B slowly ramps up the output
voltage during start-up. Whenever the RUN pin is pulled
high, the corresponding output will ramp from zero to
full-scale over a time period of approximately 750μs. This
prevents the LTC3606B from having to quickly charge the
output capacitor and thus supplying an excessive amount
of instantaneous current.
When the output is loaded heavily, for example, with
millifarad of capacitance, it may take longer than 750μs to
charge the output to regulation. If the output is still low
after the soft-start time, the LTC3606B will try to quickly
charge the output capacitor. In this case, the input current
limit (after it engages) can prevent excessive amount of
instantaneous current that is required to quickly charge
the output. See the Start-Up from Shutdown curve
(CL = 4.4mF)in the Typical Performance Characteristics
section. After input current limit is engaged, the output
slowly ramps up to regulation while limited by its 500mA
of input current.
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LTC3606B
OPERATION
Short-Circuit Protection
Programming Input Current Limit
When either regulator output is shorted to ground, the
corresponding internal N-channel switch is forced on for
a longer time period for each cycle in order to allow the
inductor to discharge, thus preventing inductor current
runaway. This technique has the effect of decreasing
switching frequency. Once the short is removed, normal
operation resumes and the regulator output will return to
its nominal voltage.
Selection of one external RLIM resistor will program the input
current limit. The current limit can be programmed from
200mA up to IPEAK current. As the input current increases,
RLIM voltage will follow. When RLIM reaches the internal
comparator threshold of 1V, the power PFET on-time will
be shortened, thereby, limiting the input current.
Input Current Limit
Internal current sense circuitry measures the inductor
current through the voltage drop across the power PFET
switch and forces the same voltage across the small sense
PFET. The voltage across the small sense PFET generates
a current representing 1/55,000th of the inductor current
during the on-cycle. The current out of RLIM pin is the
representation of the inductor current, which can be
expressed in the following equation.
IRLIM = IOUT • D1 • K1
where D1 = VOUT1/VIN is the duty cycle.
K1 is the ratio RDS(ON) (power PFET)/RDS(ON)(sense PFET).
The ratio of the power PFET to the sense PFET is trimmed
to within 2%.
Given that both PFETs are carefully laid out and matched,
their temperature and voltage coefficient effects will be
similar and their terms be canceled out in the equation. In
that case, the constant K1 will only be dependent on area
scaling, which is trimmed to within 2%. Thus, the IRLIM
current will track the input current very well over varying
temperature and VIN.
The RLIM pin can be grounded to disable input current
limit function.
Use the following equation to select the RLIM resistance
that corresponds to the input current limit.
RLIM = 55k / IDC
IDC is the input current (at VIN) to be limited. The following are
some RLIM values with the corresponding current limit.
RLIM
IDC
91.6k
600mA
110k
500mA
137.5k
400mA
Selection of CLIM Capacitance
Since IRLIM current is a function of the inductor current,
its dependency on the duty cycle cannot be ignored. Thus,
a CLIM capacitor is needed to integrate the IRLIM current
and smooth out transient currents. The LTC3606B is stable
with any size capacitance >100pF at the RLIM pin.
Each application input current limit will call for different
CLIM value to optimize its response time. Using a large CLIM
capacitor requires longer time for the RLIM pin voltage to
charge. For example, consider the application 500mA input
current limit, 5V input and 1A, 2.5V output with a 50% duty
cycle. When an instantaneous 1A output pulse is applied,
the current out of the RLIM pin becomes 1A/55k = 18.2μA
during the 50% on-time or 9.1μA full duty cycle. With a
CLIM capacitor of 1μF, RLIM of 116k, and using I = CdV/dt,
it will take 110ms for CLIM to charge from 0V to 1V. This is
the time after which the LTC3606B will start input current
limiting. Any current within this time must be considered
in each application to determine if it is tolerable.
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LTC3606B
OPERATION
Figure 1a shows VIN (IIN) current below input current
limit with a CLIM capacitor of 0.1μF. When the load pulse
is applied, under the specified condition, ILIM current is
1.1A/55k • 0.66 = 13.2μA, where 0.66 is the duty cycle.
It will take a little more than 7.5ms to charge the CLIM
capacitor from 0V to 1V, after which the LTC3606B begins
to limit input current. The IIN current is not limited during
this 7.5ms time and is more than 725mA. This current
transient may cause the input supply to temporarily
droop if the supply current compliance is exceeded, but
recovers after the input current limit engages. The output
will continue to deliver the required current load while the
output voltage droops to allow the input voltage to remain
regulated during input current limit.
and the output must deliver the required current load.
This may cause the input voltage to droop if the current
compliance is exceeded. Depending on how long this time
is, the VIN supply decoupling capacitor can provide some
of this current before VIN droops too much. In applications
with a bigger VIN supply decoupling capacitor and where
VIN supply is allow to droop closer to dropout, the CLIM
capacitor can be increased slightly. This will delay the
start of input current limit and artificially regulated VOUT
before input current limit is engaged. In this case, within
the 577μs load pulse, the VOUT voltage will stay artificially
regulated for 92μs out of the total 577μs before the input
current limit activates. This approach may be used if a
faster recovery on the output is desired.
For applications with short load pulse duration, a smaller
CLIM capacitor may be the better choice as in the example
shown in Figure 1b. In this example, a 577μs, 0A to 2A
output pulse is applied once every 4.7ms. A CLIM capacitor
of 2.2nF requires 92μs for VRLIM to charge from 0V to 1V.
During this 92μs, the input current limit is not yet engaged
Selecting a very small CLIM will speed up response time
but it can put the device within threshold of interfering
with normal operation and input current limit in every
few switching cycles. This may be undesirable in terms
of noise. Use 2πRC >> 100/clock frequency (2.25MHz) as
a starting point, R being RLIM, C being CLIM.
VOUT
2V/DIV
VOUT
200mV/DIV
IIN
500mA/DIV
VIN
AC-COUPLED
1V/DIV
VRLIM
1V/DIV
IOUT
500mA/DIV
IIN
500mA/DIV
IL
1A/DIV
50ms/DIV
3606B F01a
VIN = 5V, 500mA COMPLIANT
RLIM = 116k, CLIM = 0.1μF
ILOAD = 0A to 1.1A, COUT = 2.2mF, VOUT = 3.3V
ILIM = 475mA
Figure 1a. Input Current Limit Within 100ms Load Pulses
1ms/DIV
3606B F01b
VIN = 5V, 500mA COMPLIANT
RLIM = 116k, CLIM = 2200pF
ILOAD = 0A to 2A, COUT = 2.2mF, VOUT = 3.3V
ILIM = 475mA
Figure 1b. Input Current Limit Within
577μs, 2A Repeating Load Pulses
3606bfa
9
LTC3606B
APPLICATIONS INFORMATION
A general LTC3606B application circuit is shown in Figure 2.
External component selection is driven by the load requirement, and begins with the selection of the inductor L. Once
the inductor is chosen, CIN and COUT can be selected.
Inductor Selection
Although the inductor does not influence the operating frequency, the inductor value has a direct effect on
ripple current. The inductor ripple current ΔIL decreases
with higher inductance and increases with higher VIN or
VOUT :
V V
IL = OUT • 1 OUT (1)
fO • L VIN Accepting larger values of ΔIL allows the use of low
inductances, but results in higher output voltage ripple,
greater core losses, and lower output current capability.
A reasonable starting point for setting ripple current is
40% of the maximum output load current. So, for a 800mA
regulator, ΔIL = 320mA (40% of 800mA).
The inductor value will also have an effect on Burst Mode
operation. The transition to low current operation begins
when the peak inductor current falls below a level set by
the internal burst clamp. Lower inductor values result in
higher ripple current which causes the transition to occur
at lower load currents. This causes a dip in efficiency in
the upper range of low current operation. Furthermore,
lower inductance values will cause the bursts to occur
with increased frequency.
L1
VIN
2.5V TO 5.5V
VIN
RPGD
CIN
VOUT
SW
LTC3606B
CF
RUN
COUT
PGOOD
VFB
RLIM
PGOOD
GND
RLIM
R2
R1
CLIM
3606B F02
Figure 2. LTC3606B General Schematic
Inductor Core Selection
Different core materials and shapes will change the size/
current and price/current relationship of an inductor. Toroid
or shielded pot cores in ferrite or permalloy materials are
small and do not radiate much energy, but generally cost
more than powdered iron core inductors with similar
electrical characteristics. The choice of which style
inductor to use often depends more on the price versus
size requirements, and any radiated field/EMI requirements,
than on what the LTC3606B requires to operate. Table 1
shows some typical surface mount inductors that work
well in LTC3606B applications.
Table 1. Representative Surface Mount Inductors
MANUFACTURER
PART NUMBER
MAX DC
VALUE CURRENT
DCR
HEIGHT
Coilcraft
LPS4012-152ML
LPS4012-222ML
LPS4012-332ML
LPS4012-472ML
LPS4018-222ML
LPS4018-332ML
LPS4018-472ML
1.5μH
2.2μH
3.3μH
4.7μH
2.2μH
3.3μH
4.7μH
2200mA
1750mA
1450mA
1450mA
2300mA
2000mA
1800mA
0.070Ω
0.100Ω
0.100Ω
0.170Ω
0.070Ω
0.080Ω
0.125Ω
1.2mm
1.2mm
1.2mm
1.2mm
1.8mm
1.8mm
1.8mm
FDK
FDKMIPF2520D
FDKMIPF2520D
FDKMIPF2520D
4.7μH
3.3μH
2.2μH
1100mA
1200mA
1300mA
0.11Ω
0.1Ω
0.08Ω
1mm
1mm
1mm
LQH32CN4R7M23 4.7μH
450mA
0.2Ω
2mm
ELT5KT4R7M
4.7μH
950mA
0.2Ω
1.2mm
CDRH2D18/LD
CDH38D11SNP3R3M
CDH38D11SNP2R2M
4.7μH
3.3μH
630mA
1560mA
0.086Ω
0.115Ω
2mm
1.2mm
2.2μH
1900mA
0.082Ω
1.2mm
2.2μH
2.2μH
3.3μH
2.2μH
4.7μH
510mA
530mA
410mA
1100mA
750mA
0.13Ω
0.33Ω
0.27Ω
0.1Ω
0.19Ω
1.6mm
1.25mm
1.6mm
1mm
1mm
4.7μH
700mA
0.28Ω
1mm
3.3μH
870mA
0.17Ω
1mm
2.2μH
1000mA
0.12Ω
1mm
2.2μH
1500mA
0.076Ω
1.2mm
3.3μH
1700mA
0.095Ω
1.2mm
2.2μH
2300mA
0.059Ω
1.4mm
Murata
Panasonic
Sumida
Taiyo Yuden CB2016T2R2M
CB2012T2R2M
CB2016T3R3M
NR30102R2M
NR30104R7M
TDK
VLF3010AT4R7MR70
VLF3010AT3R3MR87
VLF3010AT2R2M1R0
VLF4012AT-2R2
M1R5
VLF5012ST-3R3
M1R7
VLF5014ST-2R2
M2R3
3606bfa
10
LTC3606B
APPLICATIONS INFORMATION
Input Capacitor (CIN) Selection
In continuous mode, the input current of the converter is a
square wave with a duty cycle of approximately VOUT / VIN .
To prevent large voltage transients, a low equivalent series
resistance (ESR) input capacitor sized for the maximum
RMS current must be used. The maximum RMS capacitor
current is given by:
IRMS IMAX
VOUT (VIN VOUT )
VIN
Where the maximum average output current IMAX equals
the peak current minus half the peak-to-peak ripple current, IMAX = ILIM – ΔIL /2. This formula has a maximum at
VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case
is commonly used to design because even significant
deviations do not offer much relief. Note that capacitor
manufacturer’s ripple current ratings are often based on
only 2000 hours lifetime. This makes it advisable to further
derate the capacitor, or choose a capacitor rated at a higher
temperature than required. Several capacitors may also be
paralleled to meet the size or height requirements of the
design. An additional 0.1μF to 1μF ceramic capacitor is
also recommended on VIN for high frequency decoupling
when not using an all-ceramic capacitor solution.
Output Capacitor (COUT) Selection
The selection of COUT is driven by the required effective
series resistance (ESR). Typically, once the ESR requirement for COUT has been met, the RMS current rating
generally far exceeds the IRIPPLE(P-P) requirement. The
output ripple ΔVOUT is determined by:
1 VOUT IL ESR+
8fOCOUT where fO = operating frequency, COUT = output capacitance
and ΔIL = ripple current in the inductor. For a fixed output
voltage, the output ripple is highest at maximum input
voltage since ΔIL increases with input voltage.
If tantalum capacitors are used, it is critical that the
capacitors are surge tested for use in switching power
supplies. An excellent choice is the AVX TPS series of
surface mount tantalum. These are specially constructed
and tested for low ESR so they give the lowest ESR for a
given volume. Other capacitor types include Sanyo POSCAP,
Kemet T510 and T495 series, and Sprague 593D and
595D series. Consult the manufacturer for other specific
recommendations.
Using Ceramic Input and Output Capacitors
Higher values, lower cost ceramic capacitors are now
becoming available in smaller case sizes. Their high ripple
current, high voltage rating and low ESR make them
ideal for switching regulator applications. Because the
LTC3606B control loop does not depend on the output
capacitor’s ESR for stable operation, ceramic capacitors
can be used freely to achieve very low output ripple and
small circuit size.
However, care must be taken when ceramic capacitors are
used at the input. When a ceramic capacitor is used at the
input and the power is supplied by a wall adapter through
long wires, a load step at the output can induce ringing at
the input, VIN. At best, this ringing can couple to the output
and be mistaken as loop instability. At worst, a sudden
inrush of current through the long wires can potentially
cause a voltage spike at VIN, large enough to damage the
part. For more information, see Application Note 88.
When choosing the input and output ceramic capacitors,
choose the X5R or X7R dielectric formulations. These
dielectrics have the best temperature and voltage
characteristics of all the ceramics for a given value and
size.
3606bfa
11
LTC3606B
APPLICATIONS INFORMATION
Setting the Output Voltage
The LTC3606B regulates the VFB pin to 0.6V during
regulation. Thus, the output voltage is set by a resistive
divider, Figure 2, according to the following formula:
VOUT = 0.6V 1+
R2
R1
(2)
The output voltage settling behavior is related to the
stability of the closed-loop system and will demonstrate
the actual overall supply performance. For a detailed
explanation of optimizing the compensation components,
including a review of control loop theory, refer to
Application Note 76.
To improve the frequency response of the main control
loop, a feedback capacitor (CF) may also be used. Great
care should be taken to route the VFB line away from noise
sources, such as the inductor or the SW line.
In some applications, a more severe transient can be caused
by switching in loads with large (>1μF) input capacitors. The
discharged input capacitors are effectively put in parallel
with COUT, causing a rapid drop in VOUT. No regulator can
deliver enough current to prevent this problem if the switch
connecting the load has low resistance and is driven quickly.
The solution is to limit the turn-on speed of the load switch
driver. A Hot Swap™ controller is designed specifically for
this purpose and usually incorporates current limiting,
short-circuit protection, and soft-starting.
Checking Transient Response
Efficiency Considerations
The regulator loop response can be checked by looking
at the load transient response. Switching regulators take
several cycles to respond to a step in load current. When
a load step occurs, VOUT immediately shifts by an amount
equal to ΔILOAD • ESR, where ESR is the effective series
resistance of COUT. ΔILOAD also begins to charge or discharge COUT generating a feedback error signal used by the
regulator to return VOUT to its steady-state value. During
this recovery time, VOUT can be monitored for overshoot
or ringing that would indicate a stability problem.
The percent efficiency of a switching regulator is equal to
the output power divided by the input power times 100%.
It is often useful to analyze individual losses to determine
what is limiting the efficiency and which change would
produce the most improvement. Percent efficiency can
be expressed as:
The initial output voltage step may not be within the
bandwidth of the feedback loop, so the standard second
order overshoot/DC ratio cannot be used to determine
the phase margin. In addition, feedback capacitors (CF)
can be added to improve the high frequency response, as
shown in Figure 2. Capacitor CF provides phase lead by
creating a high frequency zero with R2 which improves
the phase margin.
Although all dissipative elements in the circuit produce
losses, four sources usually account for the losses in
LTC3606B circuits: 1) VIN quiescent current, 2) switching
losses, 3) I2R losses, 4) other system losses.
Keeping the current small (< 10μA) in these resistors
maximizes efficiency, but making it too small may allow
stray capacitance to cause noise problems or reduce the
phase margin of the error amp loop.
% Efficiency = 100% – (L1 + L2 + L3 + ...)
where L1, L2, etc., are the individual losses as a percentage
of input power.
1. The VIN current is the DC supply current given in the
Electrical Characteristics which excludes MOSFET
driver and control currents. VIN current results in a
small (<0.1%) loss that increases with VIN, even at
no load.
3606bfa
12
LTC3606B
APPLICATIONS INFORMATION
2. The switching current is the sum of the MOSFET driver
and control currents. The MOSFET driver current results
from switching the gate capacitance of the power
MOSFETs. Each time a MOSFET gate is switched from
low to high to low again, a packet of charge dQ moves
from VIN to ground. The resulting dQ/dt is a current
out of VIN that is typically much larger than the DC bias
current. In continuous mode, IGATECHG = fO(QT + QB),
where QT and QB are the gate charges of the internal
top and bottom MOSFET switches. The gate charge
losses are proportional to VIN and thus their effects
will be more pronounced at higher supply voltages.
Thermal Considerations
3. I2R losses are calculated from the DC resistances of
the internal switches, RSW , and external inductor, RL.
In continuous mode, the average output current flows
through inductor L, but is “chopped” between the internal
top and bottom switches. Thus, the series resistance
looking into the SW pin is a function of both top and
bottom MOSFET RDS(ON) and the duty cycle (DC) as
follows:
where PD is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to
the ambient temperature. The junction temperature, TJ,
is given by:
RSW = (RDS(ON)TOP) • (DC) + (RDS(ON)BOT) • (1– DC)
The RDS(ON) for both the top and bottom MOSFETs can be
obtained from the Typical Performance Characteristics
curves. Thus, to obtain I2R losses:
I2R losses = IOUT 2 • (RSW + RL)
4. Other “hidden” losses, such as copper trace and
internal battery resistances, can account for additional
efficiency degradations in portable systems. It is very
important to include these “system” level losses in
the design of a system. The internal battery and fuse
resistance losses can be minimized by making sure that
CIN has adequate charge storage and very low ESR at
the switching frequency. Other losses, including diode
conduction losses during dead-time, and inductor
core losses, generally account for less than 2% total
additional loss.
In a majority of applications, the LTC3606B does not
dissipate much heat due to its high efficiency. In the
unlikely event that the junction temperature somehow
reaches approximately 150°C, both power switches will be
turned off and the SW node will become high impedance.
The goal of the following thermal analysis is to determine
whether the power dissipated causes enough temperature
rise to exceed the maximum junction temperature (125°C)
of the part. The temperature rise is given by:
TRISE = PD • θJA
TJ = TRISE + TAMBIENT
As a worst-case example, consider the case when the
LTC3606B is in dropout at an input voltage of 2.7V with
a load current of 800mA and an ambient temperature of
70°C. From the Typical Performance Characteristics graph
of Switch Resistance, the RDS(ON) of the switch is 0.33Ω.
Therefore, the power dissipated is:
PD = IOUT 2 • RDS(ON) = 212mV
Given that the thermal resistance of a properly soldered
DFN package is approximately 40°C/W, the junction
temperature of an LTC3606B device operating in a 70°C
ambient temperature is approximately:
TJ = (0.212W • 40°C/W) + 70°C = 78.5°C
which is well below the absolute maximum junction
temperature of 125°C.
3606bfa
13
LTC3606B
APPLICATIONS INFORMATION
should be routed away from noisy components and
traces, such as the SW line (Pin 4), and their trace
length should be minimized.
PC Board Layout Considerations
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of
the LTC3606B. These items are also illustrated graphically
in the layout diagrams of Figures 3a and 3b. Check the
following in your layout:
4. Keep sensitive components away from the SW pin, if
possible. The input capacitor CIN, CLIM and the resistors
R1, R2, and RLIM should be routed away from the SW
traces and the inductors.
1. Does the capacitor CIN connect to the power VIN (Pin 5)
and GND (Pin 9) as closely as possible? This capacitor
provides the AC current of the internal power MOSFETs
and their drivers.
5. A ground plane is preferred, but if not available, keep
the signal and power grounds segregated with small
signal components returning to the GND pin at a single
point. These ground traces should not share the high
current path of CIN or COUT.
2. Are the respective COUT and L closely connected? The
(–) plate of COUT returns current to GND and the (–)
plate of CIN.
6. Flood all unused areas on all layers with copper.
Flooding with copper will reduce the temperature rise
of power components. These copper areas should be
connected to VIN or GND.
3. The resistor divider, R1 and R2, must be connected
between the (+) plate of COUT and a ground sense line
terminated near GND (Pin 9). The feedback signal VFB
L1
VIN
2.5V TO 5.5V
VIN
CIN
SW
RPGD
VOUT
CF
LTC3606B
RUN
PGOOD
GND
RLIM
VFB
R2
RLIM
COUT
R1
CLIM
3606B F03a
BOLD LINES INDICATE HIGH CURRENT PATHS
Figure 3a. LTC3606B Layout Diagram (See Board Layout Checklist)
3606bfa
14
LTC3606B
APPLICATIONS INFORMATION
VIA TO
VOUT SENSE
GND
GND
VFB
RLIM
RUN
GND
PGOOD
SW
VIN
VIN
SW
GND
VOUT
Figure 3b. LTC3606B Suggested Layout
3606bfa
15
LTC3606B
APPLICATIONS INFORMATION
of CIN = 10μF should suffice, if the source impedance is
very low.
Design Example
As a design example, consider using the LTC3606B in a
USB-GSM application, with VIN = 5V, IINMAX = 500mA,
with the output charging a SuperCap of 4.4mF. The load
requires 800mA in active mode and 1mA in standby mode.
The output voltage VOUT = 3.4V.
The feedback resistors program the output voltage. To
maintain high efficiency at light loads, the current in these
resistors should be kept small. Choosing 10μA with the
0.6V feedback voltage makes R1~60k. A close standard
1% resistor is 59k. Using Equation (2).
First, calculate the inductor value for about 40% ripple
current (320mA in this example) at maximum VIN. Using
a derivation of Equation (1):
L1=
R2 =
3.4V
3.4V
• 1
=1.51μH
2.25MHz • (320mA)
5V
VOUT
1 • R1= 276k, 280k for 1%
0.6
A feedforward capacitor is not used since the 4.4mF
SuperCap will inhibit any fast output voltage transients.
Figure 4 shows the complete schematic for this example,
along with the efficiency curve and transient response.
Input current limit is set at 475mA average current, RLIM
= 116k, CLIM = 2200pF. See Programming Input Current
Limit section for selecting RLIM and Selection of CLIM
Capacitance section for CLIM.
For the inductor, use the closest standard value of 1.5μH.
The 4.4mF supercaps are used to deliver the required
2A pulses to power the RF power amplifiers, while the
LTC3606B recharges the supercap after the pulse ends,
see Figure 4c. As for the input capacitor, a typical value
L1
1.5μH
VIN
USB INPUT 5V
CIN
10μF
RPGD
499k
LTC3606B
RUN
R2
280k
PGOOD
VFB
RLIM
PGOOD
GND
CLIM
2200pF
VOUT
3.4V AT
800mA
SW
VIN
+
COUT
2.2mF
s2
SuperCap
R1
59k
RLIM
116k
ILIM = 475mA
CIN: AVX 08056D106KAT2A
COUT: VISHAY 592D228X96R3X2T20H
L1: COILCRAFT LPS4012-152ML
3606B F04
Figure 4a. Design Example Circuit
3606bfa
16
LTC3606B
APPLICATIONS INFORMATION
100
90
10
VOUT = 3.4V
1
70
60
0.1
50
40
30
POWER LOSS (W)
EFFICIENCY (%)
80
0.01
20
VIN = 3.6V
VIN = 4.2V
VIN = 5V
10
0
0.0001
0.001
0.01
0.1
OUTPUT CURRENT (A)
0.001
1
3606B F04b
Figure 4b. Efficiency vs Output Current
VOUT
200mV/DIV
VIN
1V/DIV
AC-COUPLED
IOUT
500mA/DIV
IIN
500mA/DIV
1ms/DIV
VIN = 5V, 500mA COMPLIANT
RLIM = 116kΩ, CLIM = 2200pF
ILOAD = 0A TO 2A, COUT = 4.4mF, VOUT = 3.4V
ILIM = 475mA
3606B F04c
Figure 4c. Transient Response
3606bfa
17
LTC3606B
PACKAGE DESCRIPTION
DD Package
8-Lead Plastic DFN (3mm × 3mm)
(Reference LTC DWG # 05-08-1698)
0.70 p0.05
3.5 p0.05
1.65 p0.05
2.10 p0.05 (2 SIDES)
PACKAGE
OUTLINE
0.25 p 0.05
0.50
BSC
2.38 p0.05
RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS
APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED
3.00 p0.10
(4 SIDES)
R = 0.125
TYP
5
0.40 p 0.10
8
1.65 p 0.10
(2 SIDES)
PIN 1
TOP MARK
(NOTE 6)
(DD8) DFN 0509 REV C
0.200 REF
0.75 p0.05
4
0.25 p 0.05
1
0.50 BSC
2.38 p0.10
0.00 – 0.05
BOTTOM VIEW—EXPOSED PAD
NOTE:
1. DRAWING TO BE MADE A JEDEC PACKAGE OUTLINE M0-229 VARIATION OF (WEED-1)
2. DRAWING NOT TO SCALE
3. ALL DIMENSIONS ARE IN MILLIMETERS
4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE
MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE
5. EXPOSED PAD SHALL BE SOLDER PLATED
6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION
ON TOP AND BOTTOM OF PACKAGE
3606bfa
18
LTC3606B
REVISION HISTORY
REV
DATE
DESCRIPTION
PAGE NUMBER
A
3/10
Changes to Electrical Characteristics
3
3606bfa
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
19
LTC3606B
TYPICAL APPLICATIONS
800mA Buck Converter, ILIM = 500mA
L1
1.5μH
VIN
USB INPUT 5V
VIN
CIN
10μF
RPGD
499k
LTC3606B
RUN
PGOOD
VFB
RLIM
PGOOD
GND
CLIM
1000pF
VOUT
3.4V AT
800mA
SW
R2
1210k
+
COUT
2.2mF
s2
SuperCap
R1
255k
RLIM
110k
L1: COILCRAFT LPS4012-152ML
CIN: AVX 08056D106KAT2A
COUT: VISHAY 592D228X96R3X2T20H
3606B TA02
800mA Buck Converter, ILIM = 475mA or Disabled
L1
1.5μH
VIN
USB INPUT 5V
VIN
CIN
10μF
RPGD
499k
LTC3606B
RUN
PGOOD
VFB
RLIM
PGOOD
GND
ILIM
DISABLE
RLIM
116k
VOUT
3.4V AT
800mA
SW
CLIM
2200pF
CIN: AVX 08056D106KAT2A
COUT: VISHAY 592D228X96R3X2T20H
R2
1210k
+
COUT
2.2mF
×2
SuperCap
R1
255k
L1: COILCRAFT LPS4012-152ML
3606B TA03
RELATED PARTS
PART NUMBER
LTC3619/LTC3619B
LTC3127
LTC3125
LTC3417A/
LTC3417A-2
LTC3407A/
LTC3407A-2
LTC3548/LTC3548-1/
LTC3548-2
LTC3546
DESCRIPTION
Dual 400mA and 800mA IOUT, 2.25MHz,
Synchronous Step-Down DC/DC Converter
1.2A IOUT, 1.6MHz, Synchronous Buck-Boost DC/DC
Converter with Adjustable Input Current Limit
1.2A IOUT, 1.6MHz, Synchronous Boost DC/DC
Converter with Adjustable Input Current Limit
Dual 1.5A/1A, 4MHz, Synchronous Step-Down
DC/DC Converter
Dual 600mA/600mA, 1.5MHz, Synchronous
Step-Down DC/DC Converter
Dual 400mA and 800mA IOUT, 2.25MHz,
Synchronous Step-Down DC/DC Converter
Dual 3A/1A, 4MHz, Synchronous Step-Down
DC/DC Converter
COMMENTS
95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V,
IQ = 50μA, ISD < 1μA, MS10E, 3mm × 3mm DFN-10
94% Efficiency, VIN(MIN) = 1.8V, VIN(MAX) = 5.5V, VOUT(MAX) = 5.25V,
IQ = 18μA, ISD < 1μA, 3mm × 3mm DFN-MSOP10E
94% Efficiency, VIN(MIN) = 1.8V, VIN(MAX) = 5.5V, VOUT(MAX) = 5.25V,
IQ = 15μA, ISD < 1μA, 2mm × 3mm DFN-8
95% Efficiency, VIN(MIN) = 2.3V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.8V,
IQ = 125μA, ISD = <1μA, TSSOP-16E, 3mm × 5mm DFN-16
95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V,
IQ = 40μA, ISD = <1μA, MS10E, 3mm × 3mm DFN-10
95% Efficiency, VIN(MIN) = 2.5V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V,
IQ = 40μA, ISD = <1μA, MS10E, 3mm × 3mm DFN-10
95% Efficiency, VIN(MIN) = 2.3V, VIN(MAX) = 5.5V, VOUT(MIN) = 0.6V,
IQ = 160μA, ISD = <1μA, 4mm × 5mm QFN-28
3606bfa
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