a Micropower, Step-Up/Step-Down SW Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V ADP1111 FEATURES Operates from 2 V to 30 V Input Voltage Range 72 kHz Frequency Operation Utilizes Surface Mount Inductors Very Few External Components Required Operates in Step-Up/Step-Down or Inverting Mode Low Battery Detector User Adjustable Current Limit Internal 1 A Power Switch Fixed or Adjustable Output Voltage 8-Pin DIP or SO-8 Package APPLICATIONS 3 V to 5 V, 5 V to 12 V Step-Up Converters 9 V to 5 V, 12 V to 5 V Step-Down Converters Laptop and Palmtop Computers Cellular Telephones Flash Memory VPP Generators Remote Controls Peripherals and Add-On Cards Battery Backup Supplies Uninterruptible Supplies Portable Instruments FUNCTIONAL BLOCK DIAGRAMS SET ADP1111 A2 VIN The fixed outputs are 3.3 V, 5 V and 12 V; and an adjustable version is also available. The ADP1111 can deliver 100 mA at 5 V from a 3 V input in step-up mode, or it can deliver 200 mA at 5 V from a 12 V input in step-down mode. OSCILLATOR A1 COMPARATOR GND DRIVER SW2 FB SET ADP1111-5 ADP1111-12 A2 VIN GAIN BLOCK/ ERROR AMP A0 ILIM SW1 R1 The ADP1111 is part of a family of step-up/step-down switching regulators that operates from an input voltage supply of 2 V to 12 V in step-up mode and up to 30 V in step-down mode. The ADP1111 can be programmed to operate in step-up/stepdown or inverting applications with only 3 external components. ILIM SW1 1.25V REFERENCE 1.25V REFERENCE GENERAL DESCRIPTION A0 GAIN BLOCK/ ERROR AMP GND A1 OSCILLATOR COMPARATOR R2 220k DRIVER SW2 SENSE Maximum switch current can be programmed with a single resistor, and an open collector gain block can be arranged in multiple configuration for low battery detection, as a post linear regulator, undervoltage lockout, or as an error amplifier. If input voltages are lower than 2 V, see the ADP1110. REV. REV. 0 A Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. One One Way, P.O. P.O.Box Box9106, 9106,Norwood, Norwood, MA MA 02062-9106, 02062-9106, U.S.A. U.S.A. One Technology Technology Way, Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: www.analog.com Tel: 781.329.4700 www.analog.com Tel:781.329.4700 617/329-4700 World Wide Web Site: http://www.analog.com Fax:781.461.3113 617/326-8703 ©1996–2009 © Analog Inc., 1996 Fax: Analog Devices, All rights reserved. Fax: 781.461.3113 ©2009 Analog Devices, Inc.Inc. AllDevices, rights reserved. ADP1111–SPECIFICATIONS (0ⴗC ≤ T ≤ +70ⴗC, V A IN = 3 V unless otherwise noted) Parameter Conditions VS QUIESCENT CURRENT Switch Off IQ INPUT VOLTAGE Step-Up Mode Step-Down Mode VIN COMPARATOR TRIP POINT VOLTAGE ADP11111 VOUT Min Typ Max Units 300 500 μA 12.6 30.0 V V 2.0 1.20 1.25 1.30 V 3.13 4.75 11.40 3.30 5.00 12.00 3.47 5.25 12.60 V V V OUTPUT SENSE VOLTAGE ADP1111-3.3 ADP1111-52 ADP1111-122 COMPARATOR HYSTERESIS ADP1111 8 12.5 mV OUTPUT HYSTERESIS ADP1111-3.3 ADP1111-5 ADP1111-12 21 32 75 50 50 120 mV mV mV OSCILLATOR FREQUENCY fOSC 54 72 88 kHz DUTY CYCLE Full Load DC 43 50 65 % SWITCH ON TIME ILIM Tied to VIN tON 5 7 9 μs SW SATURATION VOLTAGE STEP-UP MODE TA = +25°C VIN = 3.0 V, ISW = 650 mA VIN = 5.0 V, ISW = 1 A VIN = 12 V, ISW = 650 mA VSAT 0.5 0.8 1.1 0.65 1.0 1.5 V V V STEP-DOWN MODE FEEDBACK PIN BIAS CURRENT ADP1111 VFB = 0 V IFB 160 300 nA SET PIN BIAS CURRENT VSET = VREF ISET 270 400 nA GAIN BLOCK OUTPUT LOW ISINK = 300 μA VSET = 1.00 V VOL 0.15 0.4 V 0.02 0.4 0.075 %/V %/V REFERENCE LINE REGULATION 5 V ≤ VIN ≤ 30 V 2 V ≤ VIN ≤ 5 V GAIN BLOCK GAIN RL = 100 kΩ3 AV CURRENT LIMIT TA = +25°C 220 Ω from ILIM to VIN ILIM CURRENT LIMIT TEMPERATURE COEFFICIENT SWITCH OFF LEAKAGE CURRENT MAXIMUM EXCURSION BELOW GND 1000 6000 V/V 400 mA –0.3 %/°C TA = +25°C Measured at SW1 Pin VSW1 = 12 V 1 10 μA TA = +25°C ISW1 ≤ 10 μA, Switch Off –400 –350 mV NOTES 1 This specification guarantees that both the high and low trip points of the comparator fall within the 1.20 V to 1.30 V range. 2 The output voltage waveform will exhibit a sawtooth shape due to the comparator hysteresis. The output voltage on the fixed output versions will always be within the specified range. 3 100 kΩ resistor connected between a 5 V source and the AO pin. 4 All limits at temperature extremes are guaranteed via correlation using standard statistical methods. Specifications subject to change without notice. –2– REV. REV. A0 ADP1111 ABSOLUTE MAXIMUM RATINGS Parameter Supply Voltage SW1 Pin Voltage SW2 Pin Voltage Feedback Pin Voltage (ADP1111) Switch Current Maximum Power Dissipation Operating Temperature Range ADP1111A Storage Temperature Range Lead Temperature (Soldering, 10 sec) PIN DESCRIPTIONS Rating 36 V 50 V −0.5 V to VIN 5.5 V 1.5 A 500 mW Mnemonic ILIM VIN SW1 0°C to 70°C −65°C to +150°C 300°C SW2 TYPICAL APPLICATION GND AO SUMIDA CD54-220K 22µH MBRS120T3 3V INPUT ILIM VIN 5V 100mA SET SW1 10µF (OPTIONAL) ADP1111AR-5 33µF FB/SENSE SENSE GND SW2 Figure 1. 3 V to 5 V Step-Up Converter Function For normal conditions this pin is connected to VIN. When lower current is required, a resistor should be connected between ILIM and VIN. Limiting the switch current to 400 mA is achieved by connecting a 220 Ω resistor. Input Voltage. Collector Node of Power Transistor. For step-down con-figuration, connect to VIN. For step-up configuration, connect to an inductor/diode. Emitter Node of Power Transistor. For step-down configuration, connect to inductor/diode. For step-up configuration, connect to ground. Do not allow this pin to go more than a diode drop below ground. Ground. Auxiliary Gain (GB) Output. The open collector can sink 300 μA. It can be left open if unused. Gain Amplifier Input. The amplifier’s positive input is connected to SET pin and its negative input is connected to the 1.25 V reference. It can be left open if unused. On the ADP1111 (adjustable) version this pin is connected to the comparator input. On the ADP1111-3.3, ADP1111-5 and ADP1111-12, the pin goes directly to the internal application resistor that sets output voltage. PIN CONFIGURATIONS ESD CAUTION 8-Lead Plastic DIP (N-8) 8 FB (SENSE)* ILIM 1 VIN 2 ADP1111 7 SET TOP VIEW SW1 3 (Not to Scale) 6 A0 SW2 4 *FIXED VERSIONS REV.A 3 5 GND 8-Lead Plastic SOIC (SO-8) 8 FB (SENSE)* ILIM 1 VIN 2 ADP1111 7 SET TOP VIEW SW1 3 (Not to Scale) 6 A0 SW2 4 *FIXED VERSIONS 5 GND ADP1111–Typical Characteristics 76 1.4 75 OSCILLATOR FREQUENCY – kHz SATURATION VOLTAGE – V 1.2 1.0 0.8 VIN = 3V VIN = 5V 0.6 VIN = 2V 0.4 0.2 73 OSCILLATOR FREQUENCY 72 71 70 69 68 67 0 0.1 0.2 0.4 0.6 0.8 ISWITCH CURRENT – A 1.0 2 1.2 Figure 2. Saturation Voltage vs. ISWITCH Current in Step-Up Mode 4 6 8 10 12 15 18 INPUT VOLTAGE – V 2.0 1.9 1.8 1.7 1.5 SWITCH CURRENT – A 1.4 1.2 21 24 27 30 Figure 5. Oscillator Frequency vs. Input Voltage 1.6 ON VOLTAGE – V 74 VIN = 12V 1.0 0.8 0.6 STEP-DOWN WITH VIN = 12V 1.3 1.1 0.9 STEP-UP WITH 2V < VIN < 5V 0.7 0.4 0.5 0.2 0.3 0.1 0 0.1 0.2 0.4 0.6 ISWITCH CURRENT – A 0.8 1 0.9 10 100 1000 RLIM – Ω Figure 6. Maximum Switch Current vs. RLIM Figure 3. Switch ON Voltage vs. ISWITCH Current In Step-Down Mode 80 1400 78 OSCILLATOR FREQUENCY – kHz QUIESCENT CURRENT – μA 1200 1000 QUIESCENT CURRENT 800 600 400 200 0 1.5 76 74 72 OSCILLATOR FREQUENCY 70 68 66 64 62 3 6 9 12 15 18 21 24 27 60 –40 30 INPUT VOLTAGE – V Figure 4. Quiescent Current vs. Input Voltage 0 25 TEMPERATURE – ⴗC 70 85 Figure 7. Oscillator Frequency vs. Temperature –4– REV.A0 REV. ADP1111 7.5 1.10 7.4 1.05 7.2 ON VOLTAGE – V ON TIME – μs 7.3 ON TIME 7.1 7.0 6.9 VIN = 12V @ ISW = 0.65A 1.00 0.95 0.90 6.8 0.85 6.7 6.6 –40 0 25 TEMPERATURE – ⴗC 70 0.80 –40 85 Figure 8. Switch ON Time vs. Temperature 0 25 TEMPERATURE – ⴗC 70 85 Figure 11. Switch ON Voltage vs. Temperature in StepDown Mode 58 500 450 QUIESCENT CURRENT – μA 56 DUTY CYCLE – % DUTY CYCLE 54 52 50 400 350 QUIESCENT CURRENT 300 250 200 150 100 48 50 46 –40 0 25 TEMPERATURE – ⴗC 70 0 –40 85 Figure 9. Duty Cycle vs. Temperature 70 85 250 0.5 200 VIN = 3V @ ISW = 0.65A BIAS CURRENT – μA SATURATION VOLTAGE – V 25 TEMPERATURE – ⴗC Figure 12. Quiescent Current vs. Temperature 0.6 0.4 0.3 0.2 0 –40 BIAS CURRENT 150 100 50 0.1 0 25 TEMPERATURE – ⴗC 70 0 –40 85 Figure 10. Saturation Voltage vs. Temperature in Step-Up Mode REV. REV. A 0 0 0 25 TEMPERATURE – ⴗC 70 85 Figure 13. Feedback Bias Current vs. Temperature –5– ADP1111 350 The ADP1111 provides external connections for both the collector and emitter of its internal power switch that permit both step-up and step-down modes of operation. For the stepup mode, the emitter (Pin SW2) is connected to GND, and the collector (Pin SW1) drives the inductor. For step-down mode, the emitter drives the inductor while the collector is connected to VIN. BIAS CURRENT – μA 300 BIAS CURRENT 250 200 150 The output voltage of the ADP1111 is set with two external resistors. Three fixed-voltage models are also available: ADP1111–3.3 (+3.3 V), ADP1111–5 (+5 V) and ADP1111–12 (+12 V). The fixed-voltage models are identical to the ADP1111, except that laser-trimmed voltage-setting resistors are included on the chip. On the fixed-voltage models of the ADP1111, simply connect the feedback pin (Pin 8) directly to the output voltage. 100 50 0 –40 0 25 TEMPERATURE – ⴗC 70 85 Figure 14. Set Pin Bias Current vs. Temperature COMPONENT SELECTION General Notes on Inductor Selection THEORY OF OPERATION When the ADP1111 internal power switch turns on, current begins to flow in the inductor. Energy is stored in the inductor core while the switch is on, and this stored energy is transferred to the load when the switch turns off. Since both the collector and the emitter of the switch transistor are accessible on the ADP1111, the output voltage can be higher, lower, or of opposite polarity than the input voltage. The ADP1111 is a flexible, low-power, switch-mode power supply (SMPS) controller. The regulated output voltage can be greater than the input voltage (boost or step-up mode) or less than the input (buck or step-down mode). This device uses a gated-oscillator technique to provide very high performance with low quiescent current. A functional block diagram of the ADP1111 is shown on the first page of this data sheet. The internal 1.25 V reference is connected to one input of the comparator, while the other input is externally connected (via the FB pin) to a feedback network connected to the regulated output. When the voltage at the FB pin falls below 1.25 V, the 72 kHz oscillator turns on. A driver amplifier provides base drive to the internal power switch, and the switching action raises the output voltage. When the voltage at the FB pin exceeds 1.25 V, the oscillator is shut off. While the oscillator is off, the ADP1111 quiescent current is only 300 μA. The comparator includes a small amount of hysteresis, which ensures loop stability without requiring external components for frequency compensation. To specify an inductor for the ADP1111, the proper values of inductance, saturation current and dc resistance must be determined. This process is not difficult, and specific equations for each circuit configuration are provided in this data sheet. In general terms, however, the inductance value must be low enough to store the required amount of energy (when both input voltage and switch ON time are at a minimum) but high enough that the inductor will not saturate when both VIN and switch ON time are at their maximum values. The inductor must also store enough energy to supply the load, without saturating. Finally, the dc resistance of the inductor should be low so that excessive power will not be wasted by heating the windings. For most ADP1111 applications, an inductor of 15 μH to 100 μH with a saturation current rating of 300 mA to 1 A and dc resistance <0.4 Ω is suitable. Ferrite-core inductors that meet these specifications are available in small, surfacemount packages. The maximum current in the internal power switch can be set by connecting a resistor between VIN and the ILIM pin. When the maximum current is exceeded, the switch is turned OFF. The current limit circuitry has a time delay of about 1 μs. If an external resistor is not used, connect ILIM to VIN. Further information on ILIM is included in the “APPLICATIONS” section of this data sheet. To minimize Electro-Magnetic Interference (EMI), a toroid or pot-core type inductor is recommended. Rod-core inductors are a lower-cost alternative if EMI is not a problem. The ADP1111 internal oscillator provides 7 μs ON and 7 μs OFF times that are ideal for applications where the ratio between VIN and VOUT is roughly a factor of two (such as converting +3 V to + 5 V). However, wider range conversions (such as generating +12 V from a +5 V supply) can easily be accomplished. CALCULATING THE INDUCTOR VALUE Selecting the proper inductor value is a simple three step process: 1. Define the operating parameters: minimum input voltage, maximum input voltage, output voltage and output current. An uncommitted gain block on the ADP1111 can be connected as a low-battery detector. The inverting input of the gain block is internally connected to the 1.25 V reference. The noninverting input is available at the SET pin. A resistor divider, connected between VIN and GND with the junction connected to the SET pin, causes the AO output to go LOW when the low battery set point is exceeded. The AO output is an open collector NPN transistor that can sink 300 μA. 2. Select the appropriate conversion topology (step-up, stepdown, or inverting). 3. Calculate the inductor value using the equations in the following sections. –6– REV. REV. A 0 ADP1111 INDUCTOR SELECTION–STEP-UP CONVERTER In a step-up or boost converter (Figure 18), the inductor must store enough power to make up the difference between the input voltage and the output voltage. The power that must be stored is calculated from the equation: P L = (V OUT +V D −V IN(MIN ) ) • ( IOUT ) (Equation 1) where VD is the diode forward voltage (0.5 V for a 1N5818 Schottky). Because energy is only stored in the inductor while the ADP1111 switch is ON, the energy stored in the inductor on each switching cycle must be equal to or greater than: PL (Equation 2) f OSC in order for the ADP1111 to regulate the output voltage. When the internal power switch turns ON, current flow in the inductor increases at the rate of: I L (t ) = −R't ⎞ V IN ⎛ 1− e L ⎟ ⎜ R' ⎝ ⎠ (Equation 3) where L is in Henrys and R' is the sum of the switch equivalent resistance (typically 0.8 Ω at +25°C) and the dc resistance of the inductor. In most applications, the voltage drop across the switch is small compared to VIN so a simpler equation can be used: I L (t ) = V IN t L (Equation 4) Replacing ‘t’ in the above equation with the ON time of the ADP1111 (7 μs, typical) will define the peak current for a given inductor value and input voltage. At this point, the inductor energy can be calculated as follows: 1 E L = L • I 2 PEAK 2 (Equation 5) As previously mentioned, EL must be greater than PL/fOSC so that the ADP1111 can deliver the necessary power to the load. For best efficiency, peak current should be limited to 1 A or less. Higher switch currents will reduce efficiency because of increased saturation voltage in the switch. High peak current also increases output ripple. As a general rule, keep peak current as low as possible to minimize losses in the switch, inductor and diode. In practice, the inductor value is easily selected using the equations above. For example, consider a supply that will generate 12 V at 40 mA from a 9 V battery, assuming a 6 V end-of-life voltage. The inductor power required is, from Equation 1: P L = (12V + 0.5V − 6V ) • (40 mA) = 260 mW Substituting a standard inductor value of 68 μH with 0.2 Ω dc resistance will produce a peak switch current of: I PEAK = Once the peak current is known, the inductor energy can be calculated from Equation 5: EL = When selecting an inductor, the peak current must not exceed the maximum switch current of 1.5 A. If the equations shown above result in peak currents > 1.5 A, the ADP1110 should be considered. Since this device has a 70% duty cycle, more energy is stored in the inductor on each cycle. This results is greater output power. The peak current must be evaluated for both minimum and maximum values of input voltage. If the switch current is high when VIN is at its minimum, the 1.5 A limit may be exceeded at the maximum value of VIN. In this case, the ADP1111’s current limit feature can be used to limit switch current. Simply select a resistor (using Figure 6) that will limit the maximum switch current to the IPEAK value calculated for the minimum value of VIN. This will improve efficiency by producing a constant IPEAK as VIN increases. See the “Limiting the Switch Current” section of this data sheet for more information. Note that the switch current limit feature does not protect the circuit if the output is shorted to ground. In this case, current is only limited by the dc resistance of the inductor and the forward voltage of the diode. INDUCTOR SELECTION–STEP-DOWN CONVERTER The step-down mode of operation is shown in Figure 19. Unlike the step-up mode, the ADP1111’s power switch does not saturate when operating in the step-down mode; therefore, switch current should be limited to 650 mA in this mode. If the input voltage will vary over a wide range, the ILIM pin can be used to limit the maximum switch current. Higher switch current is possible by adding an external switching transistor as shown in Figure 21. The first step in selecting the step-down inductor is to calculate the peak switch current as follows: I PEAK = 2 IOUT ⎛ VOUT + VD ⎞ DC ⎜⎝V IN −VSW +VD ⎟⎠ where DC = duty cycle (0.5 for the ADP1111) VSW = voltage drop across the switch Since the required inductor power is fairly low in this example, the peak current can also be low. Assuming a peak current of 500 mA as a starting point, Equation 4 can be rearranged to recommend an inductor value: VD = diode drop (0.5 V for a 1N5818) IOUT = output current VOUT = the output voltage V IN 6V L= t= 7 μs = 84 μH I L(MAX ) 500 mA REV. A REV. 0 1 (68 μH ) • (587 mA)2 =11.7 μJ 2 Since the inductor energy of 11.7 μJ is greater than the PL/fOSC requirement of 3.6 μJ, the 68 μH inductor will work in this application. By substituting other inductor values into the same equations, the optimum inductor value can be selected. On each switching cycle, the inductor must supply: P L 260 mW = = 3.6 μJ f OSC 72 kHz −1.0 Ω • 7 μs ⎞ 6V ⎛ 1− e 68 μH ⎟ = 587 mA ⎜ 1.0 Ω ⎝ ⎠ VIN = the minimum input voltage –7– (Equation 6) ADP1111 As previously mentioned, the switch voltage is higher in stepdown mode than in step-up mode. VSW is a function of switch current and is therefore a function of VIN, L, time and VOUT. For most applications, a VSW value of 1.5 V is recommended. During each switching cycle, the inductor must supply the following energy: P L 275 mW = = 3.8 μJ f OSC 72 kHz The inductor value can now be calculated: L= VIN (MIN ) −V SW −V OUT I PEAK • tON Using a standard inductor value of 56 μH with 0.2 Ω dc resistance will produce a peak switch current of: (Equation 7) where tON = switch ON time (7 μs). I PEAK = If the input voltage will vary (such as an application that must operate from a 9 V, 12 V or 15 V source), an RLIM resistor should be selected from Figure 6. The RLIM resistor will keep switch current constant as the input voltage rises. Note that there are separate RLIM values for step-up and step-down modes of operation. Once the peak current is known, the inductor energy can be calculated from (Equation 9): EL = For example, assume that +5 V at 300 mA is required from a +12 V to +24 V source. Deriving the peak current from Equation 6 yields: I PEAK = The input voltage only varies between 4.5 V and 5.5 V in this application. Therefore, the peak current will not change enough to require an RLIM resistor and the ILIM pin can be connected directly to VIN. Care should be taken, of course, to ensure that the peak current does not exceed 650 mA. Then, the peak current can be inserted into Equation 7 to calculate the inductor value: 12 −1.5 − 5 • 7 μs = 64 μH 600 mA CAPACITOR SELECTION For optimum performance, the ADP1111’s output capacitor must be selected carefully. Choosing an inappropriate capacitor can result in low efficiency and/or high output ripple. Since 64 μH is not a standard value, the next lower standard value of 56 μH would be specified. To avoid exceeding the maximum switch current when the input voltage is at +24 V, an RLIM resistor should be specified. Using the step-down curve of Figure 6, a value of 560 Ω will limit the switch current to 600 mA. Ordinary aluminum electrolytic capacitors are inexpensive but often have poor Equivalent Series Resistance (ESR) and Equivalent Series Inductance (ESL). Low ESR aluminum capacitors, specifically designed for switch mode converter applications, are also available, and these are a better choice than general purpose devices. Even better performance can be achieved with tantalum capacitors, although their cost is higher. Very low values of ESR can be achieved by using OS-CON capacitors (Sanyo Corporation, San Diego, CA). These devices are fairly small, available with tape-and-reel packaging and have very low ESR. INDUCTOR SELECTION–POSITIVE-TO-NEGATIVE CONVERTER The configuration for a positive-to-negative converter using the ADP1111 is shown in Figure 22. As with the step-up converter, all of the output power for the inverting circuit must be supplied by the inductor. The required inductor power is derived from the formula: PL = (V OUT ) ( + V D • I OUT ) The effects of capacitor selection on output ripple are demonstrated in Figures 15, 16 and 17. These figures show the output of the same ADP1111 converter that was evaluated with three different output capacitors. In each case, the peak switch current is 500 mA, and the capacitor value is 100 μF. Figure 15 shows a Panasonic HF-series 16-volt radial cap. When the switch turns off, the output voltage jumps by about 90 mV and then decays as the inductor discharges into the capacitor. The rise in voltage indicates an ESR of about 0.18 Ω. In Figure 16, the aluminum electrolytic has been replaced by a Sprague 293D series, a 6 V tantalum device. In this case the output jumps about 30 mV, which indicates an ESR of 0.06 Ω. Figure 17 shows an OS-CON 16–volt capacitor in the same circuit, and ESR is only 0.02 Ω. (Equation 8) The ADP1111 power switch does not saturate in positive-tonegative mode. The voltage drop across the switch can be modeled as a 0.75 V base-emitter diode in series with a 0.65 Ω resistor. When the switch turns on, inductor current will rise at a rate determined by: I L (t ) = VL R' −R't ⎞ ⎛ ⎜1 − e L ⎟ ⎝ ⎠ 1 (56 μH ) • (445 mA)2 = 5.54 μJ 2 Since the inductor energy of 5.54 μJ is greater than the PL/fOSC requirement of 3.82 μJ, the 56 μH inductor will work in this application. 2 • 300 mA ⎛ 5 + 0.5 ⎞ ⎜12 −1.5 + 0.5⎟ = 600 mA 0.5 ⎝ ⎠ L= −0.85 Ω • 7 μs ⎞ 4.5V − 0.75V ⎛ 1− e 56 μH ⎟ = 445 mA ⎜ 0.65 Ω + 0.2 Ω ⎝ ⎠ (Equation 9) where: R' = 0.65 Ω + RL(DC) VL = VIN – 0.75 V For example, assume that a –5 V output at 50 mA is to be generated from a +4.5 V to +5.5 V source. The power in the inductor is calculated from Equation 8: P L = (|−5V|+0.5V|) • (50 mA) = 275 mW –8– REV. REV.A0 ADP1111 For most circuits, the 1N5818 is a suitable companion to the ADP1111. This diode has a VF of 0.5 V at 1 A, 4 μA to 10 μA leakage, and fast turn-on and turn-off times. A surface mount version, the MBRS130T3, is also available. For switch currents of 100 mA or less, a Shottky diode such as the BAT85 provides a VF of 0.8 V at 100 mA and leakage less than 1 μA. A similar device, the BAT54, is available in a SOT23 package. Even lower leakage, in the 1 nA to 5 nA range, can be obtained with a 1N4148 signal diode. Figure 15. Aluminum Electrolytic General purpose rectifiers, such as the 1N4001, are not suitable for ADP1111 circuits. These devices, which have turn-on times of 10 μs or more, are far too slow for switching power supply applications. Using such a diode “just to get started” will result in wasted time and effort. Even if an ADP1111 circuit appears to function with a 1N4001, the resulting performance will not be indicative of the circuit performance when the correct diode is used. CIRCUIT OPERATION, STEP-UP (BOOST) MODE In boost mode, the ADP1111 produces an output voltage that is higher than the input voltage. For example, +12 V can be generated from a +5 V logic power supply or +5 V can be derived from two alkaline cells (+3 V). Figure 16. Tantalum Electrolytic Figure 18 shows an ADP1111 configured for step-up operation. The collector of the internal power switch is connected to the output side of the inductor, while the emitter is connected to GND. When the switch turns on, pin SW1 is pulled near ground. This action forces a voltage across L1 equal to VIN – VCE(SAT), and current begins to flow through L1. This current reaches a final value (ignoring second-order effects) of: I PEAK ≅ VIN − VCE (SAT ) • 7μs L where 7 μs is the ADP1111 switch’s “on” time. L1 VIN D1 1N5818 VOUT R3 (OPTIONAL) 1 2 ILIM VIN ADP1111 R2 SW1 3 GND SW2 5 4 + FB 8 C1 R1 Figure 17. OS-CON Capacitor If low output ripple is important, the user should consider the ADP3000. Because this device switches at 400 kHz, lower peak current can be used. Also, the higher switching frequency simplifies the design of the output filter. Consult the ADP3000 data sheet for additional details. DIODE SELECTION In specifying a diode, consideration must be given to speed, forward voltage drop and reverse leakage current. When the ADP1111 switch turns off, the diode must turn on rapidly if high efficiency is to be maintained. Shottky rectifiers, as well as fast signal diodes such as the 1N4148, are appropriate. The forward voltage of the diode represents power that is not delivered to the load, so VF must also be minimized. Again, Schottky diodes are recommended. Leakage current is especially important in low-current applications where the leakage can be a significant percentage of the total quiescent current. REV. REV. 0A Figure 18. Step-Up Mode Operation When the switch turns off, the magnetic field collapses. The polarity across the inductor changes, current begins to flow through D1 into the load, and the output voltage is driven above the input voltage. The output voltage is fed back to the ADP1111 via resistors R1 and R2. When the voltage at pin FB falls below 1.25 V, SW1 turns “on” again, and the cycle repeats. The output voltage is therefore set by the formula: ⎛ R2 ⎞ VOUT = 1.25 V • ⎜1 + R1 ⎟⎠ ⎝ The circuit of Figure 18 shows a direct current path from VIN to VOUT, via the inductor and D1. Therefore, the boost converter is not protected if the output is short circuited to ground. –9– ADP1111 VIN CIRCUIT OPERATION, STEP DOWN (BUCK) MODE) The ADP1111’s step down mode is used to produce an output voltage that is lower than the input voltage. For example, the output of four NiCd cells (+4.8 V) can be converted to a +3 V logic supply. C2 1 VIN − VCE − VOUT • 7μs L RLIM 100Ω 2 1 ILIM VIN 3 FB 8 NC 5 VOUT D1 1N5818 + R2 CL R1 NC Figure 19. Step-Down Mode Operation When the switch turns off, the magnetic field collapses. The polarity across the inductor changes, and the switch side of the inductor is driven below ground. Schottky diode D1 then turns on, and current flows into the load. Notice that the Absolute Maximum Rating for the ADP1111’s SW2 pin is 0.5 V below ground. To avoid exceeding this limit, D1 must be a Schottky diode. If a silicon diode is used for D1, Pin SW2 can go to –0.8 V, which will cause potentially damaging power dissipation within the ADP1111. The output voltage of the buck regulator is fed back to the ADP1111’s FB pin by resistors R1 and R2. When the voltage at pin FB falls below 1.25 V, the internal power switch turns “on” again, and the cycle repeats. The output voltage is set by the formula: VOUT D2 L1 SW2 4 GND 5 D1 C1 R2 + R1 D1, D2 = 1N5818 SCHOTTKY DIODES Figure 20. Step-Down Model, VOUT > 6.2 V INCREASING OUTPUT CURRENT IN THE STEP-DOWN REGULATOR L1 SW2 4 AO SET GND 7 VOUT FB 8 SW1 ADP1111 6 3 If the input voltage to the ADP1111 varies over a wide range, a current limiting resistor at Pin 1 may be required. If a particular circuit requires high peak inductor current with minimum input supply voltage, the peak current may exceed the switch maximum rating and/or saturate the inductor when the supply voltage is at the maximum value. See the “Limiting the Switch Current” section of this data sheet for specific recommendations. VIN + 2 ADP1111 where 7 μs is the ADP1111 switch’s “on” time. C2 R3 ILIM VIN SW1 A typical configuration for step down operation of the ADP1111 is shown in Figure 19. In this case, the collector of the internal power switch is connected to VIN and the emitter drives the inductor. When the switch turns on, SW2 is pulled up towards VIN. This forces a voltage across L1 equal to VIN – VCE – VOUT and causes current to flow in L1. This current reaches a final value of: I PEAK ≅ + Unlike the boost configuration, the ADP1111’s internal power switch is not saturated when operating in step-down mode. A conservative value for the voltage across the switch in step-down mode is 1.5 V. This results in high power dissipation within the ADP1111 when high peak current is required. To increase the output current, an external PNP switch can be added (Figure 21). In this circuit, the ADP1111 provides base drive to Q1 through R3, while R4 ensures that Q1 turns off rapidly. Because the ADP1111’s internal current limiting function will not work in this circuit, R5 is provided for this purpose. With the value shown, R5 limits current to 2 A. In addition to reducing power dissipation on the ADP1111, this circuit also reduces the switch voltage. When selecting an inductor value for the circuit of Figure 21, the switch voltage can be calculated from the formula: V SW = V R5 + V Q1(SAT) ≅ 0.6 V + 0.4 V ≅ 1 V ⎛ R2 ⎞ = 1.25 V • ⎜1 + R1 ⎟⎠ ⎝ INPUT + CINPUT R5 0.3Ω R4 220Ω 1 ILIM 2 VIN SW1 3 R3 330Ω ADP1111 Q1 MJE210 L1 OUTPUT R2 FB 8 AO SET GND SW2 When operating the ADP1111 in step-down mode, the output voltage is impressed across the internal power switch’s emitterbase junction when the switch is off. To protect the switch, the output voltage should be limited to 6.2 V or less. If a higher output voltage is required, a Schottky diode should be placed in series with SW2 as shown in Figure 20. 6 7 NC NC 5 4 D1 1N5821 R1 + CL Figure 21. High Current Step-Down Operation –10– REV. A 0 REV. ADP1111 Table I. Component Selection for Typical Converters Input Voltage Output Voltage Output Current (mA) Circuit Figure Inductor Value Inductor Part No. Capacitor Value 2 to 3.1 2 to 3.1 2 to 3.1 2 to 3.1 5 5 6.5 to 11 12 to 20 20 to 30 5 12 5 5 12 12 12 12 5 5 5 –5 –5 90 mA 10 mA 30 mA 10 mA 90 MA 30 mA 50 mA 300 mA 300 mA 7 mA 250 mA 4 4 4 4 4 4 5 5 5 6 6 15 μH 47 μH 15 μH 47 μH 33 μH 47 μH 15 μH 56 μH 120 μH 56 μH 120 μH CD75-150K CTX50-1 CD75-150K CTX50-1 CD75-330K CTX50-1 33 μF 10 μF 22 μF 10 μF 22 μF 15 μF 47 μF 47 μF 47 μF 47 μF 100 μF CTX50-4 CTX100-4 CTX50-4 CTX100-4 Notes * ** ** ** ** NOTES CD = Sumida. CTX = Coiltronics. **Add 47 Ω from ILIM to VIN. **Add 220 Ω from ILIM to VIN. POSITIVE-TO-NEGATIVE CONVERSION The ADP1111 can convert a positive input voltage to a negative output voltage as shown in Figure 22. This circuit is essentially identical to the step-down application of Figure 19, except that the “output” side of the inductor is connected to power ground. When the ADP1111’s internal power switch turns off, current flowing in the inductor forces the output (–VOUT) to a negative potential. The ADP1111 will continue to turn the switch on until its FB pin is 1.25 V above its GND pin, so the output voltage is determined by the formula: also reduces the circuit’s output voltage sensitivity to temperature, which otherwise would be dominated by the –2 mV VBE contribution of Q1. The output voltage for this circuit is determined by the formula: VOUT = 1.25 V • Unlike the positive step-up converter, the negative-to-positive converter’s output voltage can be either higher or lower than the input voltage. ⎛ R2 ⎞ VOUT = 1.25 V • ⎜1 + R1 ⎟⎠ ⎝ L1 + + RLIM C2 1 2 ILIM VIN L1 Q1 1 2 ILIM VIN MJE210 D2 2N3906 POSITIVE OUTPUT CL SW1 3 FB 8 AO SET GND SW2 OUTPUT R2 AO SET GND 6 7 5 FB 8 D1 1N5818 R1 + NC NEGATIVE INPUT CL NEGATIVE OUTPUT Figure 22. Positive-to-Negative Converter The design criteria for the step-down application also apply to the positive-to-negative converter. The output voltage should be limited to |6.2 V| unless a diode is inserted in series with the SW2 pin (see Figure 20.) Also, D1 must again be a Schottky diode to prevent excessive power dissipation in the ADP1111. NEGATIVE-TO-POSITIVE CONVERSION The circuit of Figure 23 converts a negative input voltage to a positive output voltage. Operation of this circuit configuration is similar to the step-up topology of Figure 18, except the current through feedback resistor R2 is level-shifted below ground by a PNP transistor. The voltage across R2 is VOUT –VBEQ1. However, diode D2 level-shifts the base of Q1 about 0.6 V below ground thereby cancelling the VBE of Q1. The addition of D2 REV. REV. A 0 + ADP1111 3 SW1 SW2 4 ADP1111 NC D1 1N5818 R2 RLIM INPUT CINPUT R2 R1 6 7 NC NC 5 4 10kΩ R1 Figure 23. ADP1111 Negative-to-Positive Converter LIMITING THE SWITCH CURRENT The ADP1111’s RLIM pin permits the switch current to be limited with a single resistor. This current limiting action occurs on a pulse by pulse basis. This feature allows the input voltage to vary over a wide range without saturating the inductor or exceeding the maximum switch rating. For example, a particular design may require peak switch current of 800 mA with a 2.0 V input. If VIN rises to 4 V, however, the switch current will exceed 1.6 A. The ADP1111 limits switch current to 1.5 A and thereby protects the switch, but the output ripple will increase. Selecting the proper resistor will limit the switch current to 800 mA, even if VIN increases. The relationship between RLIM and maximum switch current is shown in Figure 6. The ILIM feature is also valuable for controlling inductor current when the ADP1111 goes into continuous-conduction mode. –11– ADP1111 RLIM (EXTERNAL) This occurs in the step-up mode when the following condition is met: VIN ILIM VIN VOUT + VDIODE 1 < VIN − VSW 1 − DC R1 80Ω (INTERNAL) Q3 where DC is the ADP1111’s duty cycle. When this relationship exists, the inductor current does not go all the way to zero during the time that the switch is OFF. When the switch turns on for the next cycle, the inductor current begins to ramp up from the residual level. If the switch ON time remains constant, the inductor current will increase to a high level (see Figure 24). This increases output ripple and can require a larger inductor and capacitor. By controlling switch current with the ILIM resistor, output ripple current can be maintained at the design values. Figure 25 illustrates the action of the ILIM circuit. IQ1 ADP1111 DRIVER 72kHz OSC 200 SW1 Q2 Q1 POWER SWITCH SW2 Figure 26. ADP1111 Current Limit Operation The delay through the current limiting circuit is approximately 1 μs. If the switch ON time is reduced to less than 3 μs, accuracy of the current trip-point is reduced. Attempting to program a switch ON time of 1 μs or less will produce spurious responses in the switch ON time; however, the ADP1111 will still provide a properly regulated output voltage. PROGRAMMING THE GAIN BLOCK The gain block of the ADP1111 can be used as a low-battery detector, error amplifier or linear post regulator. The gain block consists of an op amp with PNP inputs and an open-collector NPN output. The inverting input is internally connected to the ADP1111’s 1.25 V reference, while the noninverting input is available at the SET pin. The NPN output transistor will sink about 300 μA. 200mA/div Figure 24. Figure 27a shows the gain block configured as a low-battery monitor. Resistors R1 and R2 should be set to high values to reduce quiescent current, but not so high that bias current in the SET input causes large errors. A value of 33 kΩ for R2 is a good compromise. The value for R1 is then calculated from the formula: R1 = V LOBATT − 1.25 V 1.25 V R2 where VLOBATT is the desired low battery trip point. Since the gain block output is an open-collector NPN, a pull-up resistor should be connected to the positive logic power supply. 200mA/div 5V RL 47k VIN Figure 25. The internal structure of the ILIM circuit is shown in Figure 26. Q1 is the ADP1111’s internal power switch that is paralleled by sense transistor Q2. The relative sizes of Q1 and Q2 are scaled so that IQ2 is 0.5% of IQ1. Current flows to Q2 through an internal 80 Ω resistor and through the RLIM resistor. These two resistors parallel the base-emitter junction of the oscillatordisable transistor, Q3. When the voltage across R1 and RLIM exceeds 0.6 V, Q3 turns on and terminates the output pulse. If only the 80 Ω internal resistor is used (i.e. the ILIM pin is connected directly to VIN), the maximum switch current will be 1.5 A. Figure 6 gives RLIM values for lower current-limit values. R1 VBAT 1.25V REF ADP1111 AO SET 33k R2 TO PROCESSOR GND VLB–1.25V R1= ––––––––– 35.1μA VLB = BATTERY TRIP POINT Figure 27a. Setting the Low Battery Detector Trip Point –12– REV. A 0 REV. ADP1111 The circuit of Figure 27b may produce multiple pulses when approaching the trip point due to noise coupled into the SET input. To prevent multiple interrupts to the digital logic, hysteresis can be added to the circuit (Figure 27). Resistor RHYS, with a value of 1 MΩ to 10 MΩ, provides the hysteresis. The addition of RHYS will change the trip point slightly, so the new value for R1 will be: 9 V to 5 V Step-Down Converter This circuit uses a 9 V battery to generate a +5 V output. The circuit will work down to 6.5 V, supplying 50 mA at this lower limit. Switch current is limited to around 500 mA by the 100 Ω resistor. INPUT 9V V LOBATT − 1.25 V R1 = ⎛ 1.25 V ⎞ ⎛ V L − 1.25 V ⎞ ⎜ R2 ⎟ − ⎜ R + R ⎟ ⎝ ⎠ ⎝ L HYS ⎠ RLIM 100Ω R1 NC RL 47k AO TO PROCESSOR GND R2 D1 1N5818 + CL 22μF NC 20 V to 5 V Step-Down Converter SET 33k OUTPUT (9VIN TO 5V @ 150mA, 6.5VIN TO 5V @ 50mA) 15μH Figure 29. 9 V to 5 V Step-Down Converter ADP1111 1.25V REF L1 CTX15-4 SENSE 8 AO SET GND 6 7 5 5V VBAT 3 SW1 SW2 4 ADP1111-5 where VL is the logic power supply voltage, RL is the pull-up resistor, and RHYS creates the hysteresis. VIN 2 1 ILIM VIN 1.6M This circuit is similar to Figure 29, except it supplies much higher output current and operates over a much wider range of input voltage. As in the previous examples, switch current is limited to 500 mA. RHYS 12V TO 28V INPUT RLIM 100Ω Figure 27b. APPLICATION CIRCUITS All Surface Mount 3 V to 5 V Step-Up Converter INPUT +3V R3* (OPTIONAL) L1 20μH CTX20-4 MBRS120T3 SENSE 8 AO SET GND 6 7 5 SW1 3 ADP1111-5 SENSE 8 AO SET GND SW2 6 7 NC NC 5 NC L1 CTX68-4 OUTPUT (+5V @ 300mA) 68μH D1 1N5818 + CL 47μF NC Figure 30. 20 V to 5 V Step-Down Converter +5 V to –5 V Converter OUTPUT (5V @ 100mA) 2 1 ILIM VIN 3 SW1 SW2 4 ADP1111-5 This is the most basic application (along with the basic stepdown configuration to follow) of the ADP1111. It takes full advantage of surface mount packaging for all the devices used in the design. The circuit can provide +5 V at 100 mA of output current and can be operated off of battery power for use in portable equipment. D1 2 1 ILIM VIN + This circuit is essentially identical to Figure 22, except it uses a fixed-output version of the ADP1111 to simplify the design somewhat. 5V TO 12V TO 25V 28V INPUT INPUT CL 33μF RLIM LIM 100Ω 1 4 2 3 ILIM LIM VIN IN SW1 SW2 4 ADP1111-5 SENSE 8 AO SET GND 6 7 5 Figure 28. All Surface Mount +3 V to +5 V Step-Up Converter NC L1 CTX33-2 33μH D1 1N5818 + CLL 33μF NC Figure 31. +5 V to –5 V Converter REV. 0 A –13– –5V @ 75mA ADP1111 Voltage-Controlled Positive-to-Negative Converter High Power, Low Quiescent Current Step-Down Converter By including an op amp in the feedback path, a simple positiveto-negative converter can be made to give an output that is a linear multiple of a controlling voltage, Vc. The op amp, an OP196, rail-to-rail input and output amplifier, sums the currents from the output and controlling voltage and drives the FB pin either high or low, thereby controlling the on-board oscillator. The 0.22 Ω resistor limits the short-circuit current to about 3 A and, along with the BAT54 Schottky diode, helps limit the peak switch current over varying input voltages. The external power switch features an active pull-up to speed up the turn-off time of the switch. Although an IRF9530 was used in the evaluation, almost any device that can handle at least 3 A of peak current at a VDS of at least 50 V is suitable for use in this application, provided that adequate attention is paid to power dissipation. The circuit can deliver 2 W of output power with a +6-volt input from a control voltage range of 0 V to 5 V. By making use of the fact that the feedback pin directly controls the internal oscillator, this circuit achieves a shutdown-like state by forcing the feedback pin above the 1.25 V comparator threshold. The logic level at the 1N4148 diode anode needs to be at least 2 V for reliable standby operation. The external switch driver circuit features an active pull-up device, a 2N3904 transistor, to ensure that the power MOSFET turns off quickly. Almost any power MOSFET will do as the switch as long as the device can withstand the 18 volt VGS and is reasonably robust. The 0.22 Ω resistor limits the short-circuit current to about 3 A and, along with the BAT54 Schottky diode, helps to limit the peak switch current over varying input voltages. +8V TO +18V IRF9540 S D RLIM INPUT 0.22 +5V TO +12V INPUT BAT54 RLIM IRF9530 0.22Ω BAT54 2kΩ 2 1 VIN ILIM 2N3904 1N4148 6 7 5 1kΩ 4 6 7 2 NC 200kΩ +3 V to –22 V LCD Bias Generator This circuit uses an adjustable-output version of the ADP1111 to generate a +22.5 V reference output that is level-shifted to give an output of –22 V. If operation from a +5 volt supply is desired, change R1 to 47 ohms. The circuit will deliver 7 mA with a 3 volt supply and 40 mA with a 5 volt supply. 2xAA CELLS 1 2 VIN NC 4 1N4148 LI = COILTRONICS CTX20-4 Figure 34. High Power, Low Quiescent Current Step-Down Converter NOTES 1. All inductors referenced are Coiltronics CTX-series except where noted. 2. If the source of power is more than an inch or so from the converter, the input to the converter should be bypassed with approximately 10 μF of capacitance. This capacitor should be a good quality tantalum or aluminum electrolytic. OUTPUT 25μH ILIM NC 5 D1 1N4148 L1 +3V 7 121kΩ VC (0V TO +5V) Figure 32. Voltage Controlled Positive-to-Negative Converter RLIM 100Ω 1N4148 OPERATE/STANDBY 2V ≤ VIN ≤ 5 OUTPUT 39kΩ 6 CL 220μF 40.2kΩ –VOUT = –5.13 *VC 2W MAXIMUM OUTPUT 1N5231 4 NC CTX20-4 CL + 47μF 35V 51Ω + D1 IN5821 +5V 500mA SW1 3 FB 8 AO SET GND SW2 L1 20μH 3 2N3904 ADP1111 D1 IN5821 VIN FB 8 AO SET GND SW2 1 ILIM 51Ω SW1 3 ADP1111 2 VIN G 2k LI 20μH SW1 3 732kΩ CL 0.1μF ADP1111 FB 8 + AO SET GND SW2 4.7 μF 6 7 5 4 42.2kΩ 1N5818 NC NC L1 = CTX25-4 1N5818 + 22μF –22V OUTPUT 7mA @ 2V INPUT Figure 33. 3 V to –22 V LCD Bias Generator –14– REV. REV.A0 ADP1111 OUTLINE DIMENSIONS 0.400 (10.16) 0.365 (9.27) 0.355 (9.02) 1 5 4 0.280 (7.11) 0.250 (6.35) 0.240 (6.10) 0.100 (2.54) BSC 0.015 (0.38) MIN SEATING PLANE 0.022 (0.56) 0.018 (0.46) 0.014 (0.36) 5.00 (0.1968) 4.80 (0.1890) 0.060 (1.52) MAX 0.210 (5.33) MAX 0.150 (3.81) 0.130 (3.30) 0.115 (2.92) 0.325 (8.26) 0.310 (7.87) 0.300 (7.62) 0.195 (4.95) 0.130 (3.30) 0.115 (2.92) 0.015 (0.38) GAUGE PLANE 0.005 (0.13) MIN 8 4.00 (0.1574) 3.80 (0.1497) 0.014 (0.36) 0.010 (0.25) 0.008 (0.20) 0.25 (0.0098) 0.10 (0.0040) COPLANARITY 0.10 SEATING PLANE 070606-A COMPLIANT TO JEDEC STANDARDS MS-001 CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. CORNER LEADS MAY BE CONFIGURED AS WHOLE OR HALF LEADS. 5 4 1.27 (0.0500) BSC 0.430 (10.92) MAX 0.070 (1.78) 0.060 (1.52) 0.045 (1.14) 1 6.20 (0.2441) 5.80 (0.2284) 1.75 (0.0688) 1.35 (0.0532) 0.51 (0.0201) 0.31 (0.0122) 0.50 (0.0196) 0.25 (0.0099) 45° 8° 0° 0.25 (0.0098) 0.17 (0.0067) 1.27 (0.0500) 0.40 (0.0157) COMPLIANT TO JEDEC STANDARDS MS-012-A A CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. Figure 36. 8-Lead Standard Small Outline Package (SOIC_N) Narrow Body (R-8) Dimensions shown in millimeters and (inches) Figure 35. 8-Lead Plastic Dual In-Line Package (PDIP) Narrow Body (N-8) Dimensions shown in inches and (millimeters) ORDERING GUIDE Model1 ADP1111ANZ-12 ADP1111ANZ-3.3 ADP1111ANZ-5 ADP1111ARZ ADP1111ARZ-12 ADP1111ARZ-12-REEL ADP1111ARZ-3.3 ADP1111ARZ-5 ADP1111ARZ-5-REEL ADP1111ARZ-REEL 1 Output Voltage 12 V 3.3 V 5V ADJ 12 V 12 V 3.3 V 5V 5V ADJ Temperature Range −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C −40°C to +85°C Package Description 8-Lead Plastic Dual In-Line Package [PDIP] 8-Lead Plastic Dual In-Line Package [PDIP] 8-Lead Plastic Dual In-Line Package [PDIP] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] Z = RoHS Compliant Part. ©1996–2009 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D08365-0-11/09(A) REV. A ‐15‐ Package Option N-8 N-8 N-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8 012407-A 8