www.fairchildsemi.com Application Note AN4137 Design Guidelines for Off-line Flyback Converters Using Fairchild Power Switch (FPS) Abstract The step-by-step design procedure described in this paper helps engineers to design SMPS easily. In order to make the design process more efficient, a software design tool, FPS design assistant that contains all the equations described in this paper is also provided. The design procedure is verified through experimental prototype converter. This paper presents practical design guidelines for off-line flyback converters employing FPS (Fairchild Power Switch). Switched mode power supply (SMPS) design is inherently a time consuming job requiring many trade-offs and iterations with a large number of design variables. L P(n) D R(n) Bridge rectifier diode V DC + CDC R sn Vsn + C sn Np N S(n) CO(n) - VO(n) CP(n) Dsn D R1 FPS L P1 Drain 1 N S1 AC line GND 2 FB 4 Vcc 3 Ra Da CO1 Rd Rbias H11A817A CB Ca Na V O1 CP1 H11A817A R1 RF KA431 CF R2 Figure 1. Basic Off-line Flyback Converter Using FPS 1. Introduction Figure 1 shows the schematic of the basic off-line flyback converter using FPS, which also serves as the reference circuit for the design process described in this paper. Because the MOSFET and PWM controller together with various additional circuits are integrated into a single package, the design of SMPS is much easier than the discrete MOSFET and PWM controller solution. This paper provides a step-by-step design procedure for a FPS based off-line flyback converter, which includes designing the transformer and output filter, selecting the components and closing the feedback loop. The design procedure described herein is general enough to be applied to various applications. The design procedure presented in this paper is also implemented in a software design tool (FPS design assistant) to enable the engineer finish their SMPS design in a short time. In the appendix, a step-by-step design example using the software tool is provided. An experimental flyback converter from the design example has been built and tested to show the validity of the design procedure. Rev. 1.3.0 ©2003 Fairchild Semiconductor Corporation AN4137 APPLICATION NOTE 2. Step-by-step Design Procedure 1. Determine the system specifications (Vlinemin, Vlinemax, fL, Po, Eff) In this section, a design procedure is presented using the schematic of figure 1 as a reference. In general, most FPS devices have the same pin configuration from pin 1 to pin 4, as shown in figure 1. Figure 2 illustrates the design flow chart. The detailed design procedures are as follows: (1) STEP-1 : Define the system specifications 2. Determine DC link capacitor (CDC) and DC link voltage range - Line voltage range (Vlinemin and Vlinemax). - Line frequency (fL). - Maximum output power (Po). 3. Determine the maximum duty ratio (Dmax) - Estimated efficiency (Eff) : It is required to estimate the power conversion efficiency to calculate the maximum input power. If no reference data is available, set Eff = 0.7~0.75 for low voltage output applications and Eff = 0.8~0.85 for high voltage output applications. 4. Determine the transformer primary side inductance (Lm) 5. Choose proper FPS considering input power and Idspeak With the estimated efficiency, the maximum input power is given by 6. Determine the proper core and the minimum primary turns (Npmin) P Pin = ------oE ff 7. Determine the number of turns for each output For multiple output SMPS, the load occupying factor for each output is defined as 8. Determine the wire diameter for each winding Is the winding window area (Aw) enough ? (1) Po (n ) KL ( n ) = -----------Po Y (2) where Po(n) is the maximum output power for the n-th output. For single output SMPS, KL(1)=1. N (2) STEP-2 : Determine DC link capacitor (CDC) and the DC link voltage range. Y Is it possible to change the core ? N It is typical to select the DC link capacitor as 2-3uF per watt of input power for universal input range (85-265Vrms) and 1uF per watt of input power for European input range (195V265Vrms). With the DC link capacitor chosen, the minimum link voltage is obtained as 9. Choose the proper rectifier diode for each output V DC min = 2 ⋅ ( V line 10. Determine the output capacitor 11. Design the RCD snubber 12. Feedback loop design (3) where Dch is the DC link capacitor charging duty ratio defined as shown in figure 3, which is typically about 0.2 and Pin, Vlinemin and fL are specified in step-1. The maximum DC link voltage is given as V DC Design finished P in ⋅ ( 1 – D ch ) ) – -----------------------------------C DC ⋅ f L min 2 max = 2V line max (4) where Vlinemax is specified in step-1. Figure 2. Flow chart of design procedure 2 ©2002 Fairchild Semiconductor Corporation APPLICATION NOTE AN4137 Minimum DC link voltage DC link voltage T1 Dch = T1 / T2 = 0.2 T2 Figure 3. DC Link Voltage Waveform In the case of a CCM flyback converter, the design process is straight forward since the input-to-output voltage gain depends only on the duty cycle. Meanwhile, the input-to-output voltage gain of a DCM flyback converter depends not only on the duty cycle but also on the load condition, which causes the circuit design to be somewhat complicated. However, it is generally accepted that a DCM flyback converter is designed to operate at the boundary of DCM and CCM with minimum input voltage and maximum load as shown in Fig. 4. This minimizes MOSFET conduction losses. Therefore, under these circumstances, we can use the same voltage gain equation as the CCM flyback converter with maximum load and minimum input voltage. (3) STEP-3 : Determine the maximum duty ratio (Dmax). A Flyback converter has two kinds of operation modes ; continuous conduction mode (CCM) and discontinuous conduction mode (DCM). CCM and DCM have their own advantages and disadvantages, respectively. In general, DCM provides better switching conditions for the rectifier diodes, since the diodes are operating at zero current just before becoming reverse biased. The transformer size can be reduced using DCM because the average energy storage is low compared to CCM. However, DCM inherently causes high RMS current, which increases the conduction loss of the MOSFET and the current stress on the output capacitors. Therefore DCM is usually recommended for high voltage and low current output applications. Meanwhile, CCM is preferred for low voltage and high current output applications. - VDC + VRO - + F PS D rain + GND V ds - V RO Minimum input voltage and full load condition MOSFET Drain Current Rectifier Diode Current D As input voltage increases or load current decreases MOSFET Drain Current Rectifier Diode Current D Figure 4. Current waveforms of DCM flyback converter ©2002 Fairchild Semiconductor Corporation V DC 0V Figure 5. The output voltage reflected to the primary When the MOSFET in the FPS is turned off, the input voltage (VDC) together with the output voltage reflected to the primary (VRO) are imposed on the MOSFET as shown in figure 5. After determining Dmax, VRO and the maximum nominal MOSFET voltage (Vdsnom) are obtained as D max min V RO = ----------------------- ⋅ V DC 1 – D max V ds nom = V DC max + VRO (5) (6) where VDCmin and VDCmax are specified in equations (3) and (4) respectively. As can be seen in equation (5) and (6), the voltage stress on MOSFET can be reduced, by decreasing Dmax. However, this increases the voltage stresses on the rectifier diodes in the secondary side. Therefore, it is desirable to set Dmax as large as possible if there is enough margin in the MOSFET voltage rating. The maximum duty ratio 3 AN4137 APPLICATION NOTE (Dmax) should be determined so that Vdsnom would be 65~70% of the MOSFET voltage rating considering the voltage spike caused by the leakage inductance. In the case of 650V rated MOSFET, it is typical to set Dmax to be 0.45~0.5 for an universal input range application. Because the current mode controlled flyback converter operating in CCM causes sub-harmonic oscillation with duty ratio larger than 0.5, set Dmax to be smaller than 0.5 for CCM. (4) STEP-4 : Determine the transformer primary side inductance (Lm). The operation changes between CCM and DCM as the load condition and input voltage vary. For both operation modes, the worst case in designing the inductance of the transformer primary side (Lm) is full load and minimum input voltage condition. Therefore, Lm is obtained in this condition as V DC CCM –1 1 1 - = --------------------------- – --------- 2L m f s P in VRO (12) where Pin, VRO and Lm are specified in equations (1), (5) and (7), respectively, and fs is the FPS switching frequency. If the result of equation (12) has a negative value, the converter is always in CCM under the full load condition regardless of the input voltage variation. ∆I I ds peak I EDC K RF = ∆I 2I EDC CCM operation : KRF < 1 2 min ( VDC ⋅ D max ) L m = --------------------------------------------2Pin f s KRF (7) I ds peak where VDCmin is specified in equation (3), Dmax is specified in step-3, Pin is specified in step-1, fs is the switching frequency of the FPS device and KRF is the ripple factor in full load and minimum input voltage condition, defined as shown in figure 6. For DCM operation, KRF = 1 and for CCM operation KRF < 1. The ripple factor is closely related with the transformer size and the RMS value of the MOSFET current. Even though the conduction loss in the MOSFET can be reduced through reducing the ripple factor, too small a ripple factor forces an increase in transformer size. When designing the flyback converter to operate in CCM, it is reasonable to set KRF = 0.25-0.5 for the universal input range and KRF = 0.4-0.8 for the European input range. Once Lm is determined, the maximum peak current and RMS current of the MOSFET in normal operation are obtained as I ds I ds rms where and peak = ∆I = I EDC + ----2 2 ∆ I 2 D max 3 ( IEDC ) + ----- ------------2 3 P in I EDC = ------------------------------------min V DC ⋅ D max V min D DC max ∆ I = ----------------------------------- Lm fs (8) (9) (10) (11) where Pin, VDCmin and Lm are specified in equations (1), (3), and (7) respectively, Dmax is specified in step-3 and fs is the FPS switching frequency. The flyback converter designed for CCM at the minimum input voltage and full load condition may enter into DCM as the input voltage increases. The maximum input voltage guaranteeing CCM in the full load condition is obtained as 4 ∆I I EDC K RF = ∆I 2I EDC DCM operation : KRF =1 Figure 6. MOSFET Drain Current and Ripple Factor (KRF) (5) STEP-5 : Choose the proper FPS considering input power and peak drain current. With the resulting maximum peak drain current of the MOSFET (Idspeak) from equation (8), choose the proper FPS of which the pulse-by-pulse current limit level (Iover) is higher than Idspeak. Since FPS has ± 12% tolerance of Iover, there should be some margin in choosing the proper FPS device.The FPS lineup with proper power rating is also included in the software design tool. (6) STEP-6 : Determine the proper core and the minimum primary turns. Actually, the initial selection of the core is bound to be crude since there are too many variables. One way to select the proper core is to refer to the manufacture's core selection guide. If there is no proper reference, use the table 1 as a starting point. The core recommended in table 1 is typical for the universal input range, 67kHz switching frequency and single output application. When the input voltage range is 195-265 Vac or the switching frequency is higher than 67kHz, a smaller core can be used. For an application with multiple outputs, usually a larger core should be used than recommended in the table. ©2002 Fairchild Semiconductor Corporation APPLICATION NOTE AN4137 With the chosen core, the minimum number of turns for the transformer primary side to avoid the core saturation is given by NP min L m Iover 6 = ------------------- × 10 B sat A e (turns) (13) where Lm is specified in equation (7), Iover is the FPS pulseby-pulse current limit level, Ae is the cross-sectional area of the core as shown in figure 7 and Bsat is the saturation flux density in tesla. Figure 8 shows the typical characteristics of ferrite core from TDK (PC40). Since the saturation flux density (Bsat) decreases as the temperature goes high, the high temperature characteristics should be considered. If there is no reference data, use Bsat =0.3~0.35 T. Since the MOSFET drain current exceeds Idspeak and reaches Iover in a transition or fault condition, Iover is used in equation (13) instead of Idspeak to prevent core saturation during transition. Output Power EI core EE core EPC core 0-10W EI12.5 EI16 EI19 EE8 EE10 EE13 EE16 EPC10 EPC13 EPC17 10-20W EI22 EE19 EPC19 EE22 EPC25 EER25.5 EE25 EPC30 EER28 20-30W EER core EI25 30-50W EI28 EI30 50-70W EI35 EE30 EER28L 70-100W EI40 EE35 EER35 100-150W EI50 EE40 EER40 EER42 150-200W EI60 EE50 EE60 EER49 Table 1. Core quick selection table (For universal input range, fs=67kHz and single output) Aw (7) STEP-7 : Determine the number of turns for each output Figure 9 shows the simplified diagram of the transformer. First, determine the turns ratio (n) between the primary side and the feedback controlled secondary side as a reference. NP V R0 = ------------------------n = --------Ns1 V o1 + V F1 Ae Figure 7. Window Area and Cross Sectional Area M agnetization Curves (typical) M aterial :PC40 25 ℃ 500 60 ℃ 100 ℃ 400 Flux density B (mT) 120 ℃ (14) where Np and Ns1 are the number of turns for primary side and reference output, respectively, Vo1 is the output voltage and VF1 is the diode (DR1) forward voltage drop of the reference output. Then, determine the proper integer for Ns1 so that the resulting Np is larger than Npmin obtained from equation (13). The number of turns for the other output (n-th output) is determined as Vo (n ) + VF ( n) ⋅ Ns1 N s ( n ) = --------------------------------V o1 + V F1 ( turns ) ( 15 ) 300 The number of turns for Vcc winding is determined as 200 V cc * + V Fa - ⋅ N s1 N a = --------------------------V o1 + VF1 100 0 0 800 M agnetic field H (A/m ) 1600 Figure 8. Typical B-H characteristics of ferrite core (TDK/PC40) ©2002 Fairchild Semiconductor Corporation ( turns ) ( 16 ) where Vcc* is the nominal value of the supply voltage of the FPS device, and VFa is the forward voltage drop of Da as defined in figure 9. Since Vcc increases as the output load increases, it is proper to set Vcc* as Vcc start voltage (refer to the data sheet) to avoid the over voltage protection condition during normal operation. 5 AN4137 APPLICATION NOTE severe eddy current losses as well as to make winding easier. For high current output, it is better to use parallel windings with multiple strands of thinner wire to minimize skin effect. + V F(n) - D R(n) Np V RO Check if the winding window area of the core, Aw (refer to figure 7) is enough to accommodate the wires. The required winding window area (Awr) is given by + V O(n) N S(n) A w r = Ac ⁄ K F - + Da + V cc * where Ac is the actual conductor area and KF is the fill factor. Typically the fill factor is 0.2~0.25 for single output application and 0.15~0.2 for multiple outputs application. If the required window (Awr) is larger than the actual window area (Aw), go back to the step-6 and change the core to a bigger one. Sometimes it is impossible to change the core due to cost or size constraints. If the converter is designed for CCM and the winding window (Aw) is slightly insufficient, go back to step-4 and reduce Lm by increasing the ripple factor (KRF). Then, the minimum number of turns for the primary (Npmin) of the equation (13) will decrease, which results in the reduced required winding window area (Awr). + V F1 - - V Fa + + V O1 D R1 N S1 Na - - (19) Figure 9. Simplified diagram of the transformer With the determined turns of the primary side, the gap length of the core is obtained as (9) STEP-9 : Choose the rectifier diode in the secondary side based on the voltage and current ratings. The maximum reverse voltage and the rms current of the rectifier diode (DR(n)) of the n-th output are obtained as 2 NP 1 G = 40 πA e -------------------- – ------ 1000L A m L ( mm ) ( 17 ) where AL is the AL-value with no gap in nH/turns2, Ae is the cross sectional area of the core as shown in figure 7, Lm is specified in equation (7) and Np is the number of turns for the primary side of the transformer (8) STEP-8 : Determine the wire diameter for each winding based on the rms current of each output. The rms current of the n-th secondary winding is obtained as I sec ( n ) rms = I ds rms rms V RO ⋅ K L ( n ) 1 – D max ----------------------- ⋅ -------------------------------------( Vo (n ) + VF ( n) ) D max ( 18 ) are specified in equations (5) and (9), where VRO and Ids Vo(n) is the output voltage of the n-th output, VF(n) is the diode (DR(n)) forward voltage drop, Dmax is specified in step3 and KL(n) is the load occupying factor for n-th output defined in equation (2). The current density is typically 5A/mm2 when the wire is long (>1m). When the wire is short with a small number of turns, a current density of 6-10 A/mm2 is also acceptable. Avoid using wire with a diameter larger than 1 mm to avoid 6 max V DC ⋅ ( Vo ( n) + VF ( n ) ) V D ( n ) = Vo ( n ) + --------------------------------------------------------------V RO ( 20 ) V RO K L ( n ) 1 – Dmax ----------------------- ⋅ -------------------------------------( Vo( n) + VF ( n) ) D max ( 21 ) ID ( n) rms = I ds rms where KL(n), VDCmax, VRO, Idsrms are specified in equations (2), (4), (5) and (9) respectively, Dmax is specified in step-3, Vo(n) is the output voltage of the n-th output and VF(n) is the diode (DR(n)) forward voltage. The typical voltage and current margins for the rectifier diode are as follows V RRM > 1.3 ⋅ V D ( n ) IF > 1.5 ⋅ I D ( n ) rms (22) (23) where VRRM is the maximum reverse voltage and IF is the average forward current of the diode. A quick selection guide for Fairchild Semiconductor rectifier diodes is given in table 2. In this table trr is the maximum reverse recovery time. ©2002 Fairchild Semiconductor Corporation APPLICATION NOTE AN4137 (10) STEP-10 : Determine the output capacitor considering the voltage and current ripple. Schottky Barrier Diode Products VRRM IF trr Package SB330 30 V 3A - TO-210AD SB530 30 V 5A - TO-210AD MBR1035 35 V 10 A - TO-220AC MBR1635 35 V 16 A - TO-220AC SB340 40 V 3A - TO-210AD SB540 40 V 5A - TO-210AD SB350 50 V 3A - TO-210AD SB550 50 V 5A - TO-210AD SB360 60 V 3A - TO-210AD SB560 60 V 5A - TO-210AD MBR1060 60 V 10 A - TO-220AC MBR1660 60 V 16 A - TO-220AC Ultra Fast Recovery diode Products VRRM IF trr Package EGP10B 100 V 1A 50 ns DO-41 UF4002 100 V 1A 50 ns DO-41 EGP20B 100 V 2A 50 ns DO-15 EGP30B 100 V 3A 50 ns DO-210AD The ripple current of the n-th output capacitor (Co(n)) is obtained as I cap ( n ) rms = ( ID ( n ) rms 2 ) – Io ( n) 2 (24) where Io(n) is the load current of the n-th output and ID(n)rms is specified in equation (21). The ripple current should be smaller than the ripple current specification of the capacitor. The voltage ripple on the n-th output is given by I D I peak V R K ( V o ( n ) + VF ( n ) ) o ( n ) max ds RO C ( n ) L ( n ) ∆ V o ( n ) = ------------------------ + ---------------------------------------------------------- (25) C o ( n ) fs where Co(n) is the capacitance, Rc(n) is the effective series resistance (ESR) of the n-th output capacitor, KL(n), VRO and Idspeak are specified in equations (2), (5) and (8) respectively, Dmax is specified in step-3, Io(n) and Vo(n) are the load current and output voltage of the n-th output, respectively and VF(n) is the diode (DR(n)) forward voltage. Sometimes it is impossible to meet the ripple specification with a single output capacitor due to the high ESR of the electrolytic capacitor. Then, additional LC filter stages (post filter) can be used. When using the post filters, be careful not to place the corner frequency too low. Too low a corner frequency may make the system unstable or limit the control bandwidth. It is typical to set the corner frequency of the post filter at around 1/10~1/5 of the switching frequency. FES16BT 100 V 16 A 35 ns TO-220AC EGP10C 150 V 1A 50 ns DO-41 EGP20C 150 V 2A 50 ns DO-15 EGP30C 150 V 3A 50 ns DO-210AD FES16CT 150 V 16 A 35 ns TO-220AC EGP10D 200 V 1A 50 ns DO-41 UF4003 200 V 1A 50 ns DO-41 (11) STEP-11 : Design the RCD snubber. EGP20D 200 V 2A 50 ns DO-15 When the power MOSFET is turned off, there is a high voltage spike on the drain due to the transformer leakage inductance. This excessive voltage on the MOSFET may lead to an avalanche breakdown and eventually failure of FPS. Therefore, it is necessary to use an additional network to clamp the voltage. EGP30D 200 V 3A 50 ns DO-210AD FES16DT 200 V 16 A 35 ns TO-220AC EGP10F 300 V 1A 50 ns DO-41 EGP20F 300 V 2A 50 ns DO-15 EGP30F 300 V 3A 50 ns DO-210AD EGP10G 400 V 1A 50 ns DO-41 UF4004 400 V 1A 50 ns DO-41 EGP20G 400 V 2A 50 ns DO-15 EGP30G 400 V 3A 50 ns DO-210AD UF4005 600 V 1A 75 ns DO-41 EGP10J 600 V 1A 50 ns DO-41 EGP20J 600 V 2A 50 ns DO-15 EGP30J 600 V 3A 50 ns DO-210AD UF4006 800 V 1A 75 ns TO-41 UF4007 1000 V 1A 75 ns TO-41 Table 2. Fairchild Diode quick selection table ©2002 Fairchild Semiconductor Corporation The RCD snubber circuit and MOSFET drain voltage waveform are shown in figure 10 and 11, respectively. The RCD snubber network absorbs the current in the leakage inductance by turning on the snubber diode (Dsn) once the MOSFET drain voltage exceeds the voltage of node X as depicted in figure 10. In the analysis of snubber network, it is assumed that the snubber capacitor is large enough that its voltage does not change significantly during one switching cycle. The first step in designing the snubber circuit is to determine the snubber capacitor voltage at the minimum input voltage and full load condition (Vsn). Once Vsn is determined, the power dissipated in the snubber network at the minimum input voltage and full load condition is obtained as 7 AN4137 APPLICATION NOTE From equation (28), the maximum voltage stress on the internal MOSFET is given by 2 V ( V sn ) peak 2 sn 1 - = --- fs L lK ( I ds ) --------------------------P sn = ---------------2 R sn Vsn – V RO (26) peak where Ids is specified in equation (8), fs is the FPS switching frequency, Llk is the leakage inductance, Vsn is the snubber capacitor voltage at the minimum input voltage and full load condition, VRO is the reflected output voltage and Rsn is the snubber resistor. Vsn should be larger than VRO and it is typical to set Vsn to be 2~2.5 times of VRO. Too small a Vsn results in a severe loss in the snubber network as shown in equation (26). The leakage inductance is measured at the switching frequency on the primary winding with all other windings shorted. Then, the snubber resistor with proper rated wattage should be chosen based on the power loss. The maximum ripple of the snubber capacitor voltage is obtained as ∆ V sn V sn1 = ----------------------C sn R sn f s V ds (31) VDC + CDC Rsn Csn X - VX Vsn + Np VRO + Dsn FPS - Llk Drain GND + Vds - (28) where fs is the FPS switching frequency, Llk is the primary side leakage inductance, VRO is the reflected output voltage, Rsn is the snubber resistor and Ids2 is the peak drain current at the maximum input voltage and full load condition. When the converter operates in CCM at the maximum input voltage and full load condition (refer to equation (12)), the Ids2 of equation (28) is obtained as max max P in ⋅ V DC + V RO V DC ⋅ VRO I ds2 = ------------------------------------------------------------------ + ----------------------------------------------------------------------------max V ⋅ VRO 2L m f s ⋅ V DCmax + V RO DC (29) Figure 10. Circuit diagram of the snubber network Voltage Margin > 10% of BVdss BVdss Effect of stray inductance (5-10V) Vsn2 When the converter operates in DCM at the maximum input voltage and full load condition (refer to equation (12)), the Ids2 of equation (28) is obtained as 2 ⋅ P in ---------------fs ⋅ Lm + V sn2 2 V RO + ( VRO ) + 2R sn L lk fs ( I ds2 ) V sn2 = ------------------------------------------------------------------------------------------2 I ds2 = max Check if Vdsmax is below 90% of the rated voltage of the MOSFET (BVdss) as shown in figure 11. The voltage rating of the snubber diode should be higher than BVdss. Usually, an ultra fast diode with 1A current rating is used for the snubber network. In the snubber design in this section, neither the lossy discharge of the inductor nor stray capacitance is considered. In the actual converter, the loss in the snubber network is less than the designed value due to this effects. The snubber capacitor voltage (Vsn) of equation (26) is for the minimum input voltage and full load condition. When the converter is designed to operate in CCM, the peak drain current together with the snubber capacitor voltage decrease as the input voltage increases. The snubber capacitor voltage under maximum input voltage and full load condition is obtained as 2 = V DC where VDCmax is specified in equation (4). (27) where fs is the FPS switching frequency. In general, 5~10% ripple is reasonable. max VDC max (30) where Pin, VDCmax, VRO and Lm are specified in equations (1), (4), (5) and (7), respectively, and fs is the FPS switching frequency. VRO 0V Figure 11. MOSFET drain voltage and snubber capacitor voltage ©2003 Fairchild Semiconductor Corporation 8 APPLICATION NOTE AN4137 2 (12) STEP-12 : Design the feed back loop. Since most FPS devices employ current mode control as shown in figure 12, the feedback loop can be simply implemented with a one-pole and one-zero compensation circuit. In the feedback circuit analysis, it is assumed that the current transfer ratio (CTR) of the opto coupler is 100%. The current control factor of FPS, K is defined as I pk Iover K = --------- = ----------------V FB VFBsat (32) where Ipk is the peak drain current and VFB is the feedback voltage, respectively for a given operating condition, Iover is the current limit of the FPS and VFBsat is the feedback saturation voltage, which is typically 2.5V. In order to express the small signal AC transfer functions, the small signal variations of feedback voltage (vFB) and controlled output voltage (vo1) are introduced as vˆFB and vˆo1. vo1' FPS vFB RB CB RD vo1 ibias Rbias iD 1:1 CF RF R1 KA431 R2 RL ( 1 – D ) 1 (1 + D) - and w p = ------------------w z = -------------------- , w rz = ---------------------------------------2 R c1 C o1 R L C o1 DL m ( N s1 ⁄ N p ) where Lm is specified in equation (7), D is the duty cycle of the FPS, Co1 is the reference output capacitor and RC1 is the ESR of Co1. When the converter has more than one output, the low frequency control-to-output transfer function is proportional to the parallel combination of all load resistance, adjusted by the square of the turns ratio. Therefore, the effective load resistance is used in equation (33) instead of the actual load resistance of Vo1. Notice that there is a right half plane (RHP) zero (wrz) in the control-to-output transfer function of equation (33). Because the RHP zero reduces the phase by 90 degrees, the crossover frequency should be placed below the RHP zero. Figure 13 shows the variation of a CCM flyback converter control-to-output transfer function for different input voltages. This figure shows the system poles and zeros together with the DC gain change for different input voltages. The gain is highest at the high input voltage condition and the RHP zero is lowest at the low input voltage condition. Figure 14 shows the variation of a CCM flyback converter control-to-output transfer function for different loads. This figure shows that the low frequency gain does not change for different loads and the RHP zero is lowest at the full load condition. For DCM operation, the control-to-output transfer function of the flyback converter using current mode control is given by Vo1 ( 1 + s ⁄ w z ) vˆ o1 G vc = -------- = ---------- ⋅ ---------------------------VFB ( 1 + s ⁄ w p ) vˆ FB Ipk MOSFET current where (34) 1 wz = -------------------- , w p = 2 ⁄ R L C o1 R c1 C o1 Figure 12. Control Block Diagram For CCM operation, the control-to-output transfer function of the flyback converter using current mode control is given by G vc vˆ o1 = -------vˆ FB K ⋅ R L V DC ( N p ⁄ N s1 ) ( 1 + s ⁄ w z ) ( 1 – s ⁄ w rz ) = ----------------------------------------------------- ⋅ ---------------------------------------------------------1 + s ⁄ wp 2V RO + v DC ( 33 ) where VDC is the DC input voltage, RL is the effective total load resistance of the controlled output, defined as Vo12/Po, Np and Ns1 are specified in step-7, VRO is specified in equation (5), Vo1 is the reference output voltage, Po is specified in step-1 and K is specified in equation (32). The pole and zeros of equation (33) are defined as Vo1 is the reference output voltage, VFB is the feedback voltage for a given condition, RL is the effective total resistance of the controlled output, Co1 is the controlled output capacitance and Rc1 is the ESR of Co1. Figure 15 shows the variation of the control-to-output transfer function of a flyback converter in DCM for different loads. Contrary to the flyback converter in CCM, there is no RHP zero and the DC gain does not change as the input voltage varies. As can be seen, the overall gain except for the DC gain is highest at the full load condition. The feedback compensation network transfer function of figure 12 is obtained as ©2003 Fairchild Semiconductor Corporation 9 AN4137 APPLICATION NOTE ˆ w i 1 + s ⁄ w zc v FB -------- = - ----- ⋅ --------------------------ˆ s 1 + 1 ⁄ w pc v o1 (35) RB 1 1 - , w zc = --------------------------------- , w pc = --------------where w i = ---------------------R1 RD CF ( R F + R 1 )C F RB CB and RB is the internal feedback bias resistor of FPS, which is typically 2.8kΩ and R1, RD, RF, CF and CB are shown in figure 12. 40 dB fp 20 dB fp High input voltage 0 dB Low input voltage fz -20 dB frz fz frz When the input voltage and the load current vary over a wide range, it is not easy to determine the worst case for the feedback loop design. The gain together with zeros and poles vary according to the operating condition. Moreover, even though the converter is designed to operate in CCM or at the boundary of DCM and CCM in the minimum input voltage and full load condition, the converter enters into DCM changing the system transfer functions as the load current decreases and/or input voltage increases. One simple and practical way to this problem is designing the feedback loop for low input voltage and full load condition with enough phase and gain margin. When the converter operates in CCM, the RHP zero is lowest in low input voltage and full load condition. The gain increases only about 6dB as the operating condition is changed from the lowest input voltage to the highest input voltage condition under universal input condition. When the operating mode changes from CCM to DCM, the RHP zero disappears making the system stable. Therefore, by designing the feedback loop with more than 45 degrees phase margin in low input voltage and full load condition, the stability over all the operating ranges can be guaranteed. -40 dB The procedure to design the feedback loop is as follows 1Hz 10Hz 100Hz 1kHz 10kHz 100kHz Figure 13. CCM flyback converter control-to output transfer function variation for different input voltages 40 dB fp Light load 20 dB fp 0 dB Heavy load -20 dB fz frz f rz -40 dB 1Hz 10Hz 100Hz 1kHz 10kHz 100kHz Figure 14. CCM flyback converter control-to output transfer function variation for different loads (a) Determine the crossover frequency (fc). For CCM mode flyback, set fc below 1/3 of right half plane (RHP) zero to minimize the effect of the RHP zero. For DCM mode fc can be placed at a higher frequency, since there is no RHP zero. (b) When an additional LC filter is employed, the crossover frequency should be placed below 1/3 of the corner frequency of the LC filter, since it introduces a -180 degrees phase drop. Never place the crossover frequency beyond the corner frequency of the LC filter. If the crossover frequency is too close to the corner frequency, the controller should be designed to have a phase margin greater than 90 degrees when ignoring the effect of the post filter. (c) Determine the DC gain of the compensator (wi/wzc) to cancel the control-to-output gain at fc. (d) Place a compensator zero (fzc) around fc/3. (e) Place a compensator pole (fpc) above 3fc. 40 dB Loop gain T 40 dB fp fp 20 dB fzc 20 dB Heavy load 0 dB 0 dB -20 dB Control to output fz Light load Compensator fpc fp fc frz -20 dB fz -40 dB 1Hz fz 10Hz 100Hz 1kHz 10kHz 100kHz Figure 15. DCM flyback converter control-to output transfer function variation for different loads -40 dB 1Hz 10Hz 100Hz 1kHz 10kHz 100kHz Figure 16. Compensator design ©2003 Fairchild Semiconductor Corporation 10 APPLICATION NOTE AN4137 When determining the feedback circuit component, there are some restrictions as follows. (a) The voltage divider network of R1 and R2 should be designed to provide 2.5V to the reference pin of the KA431. The relationship between R1 and R2 is given as 2.5 ⋅ R 1 R 2 = ----------------------Vo1 – 2.5 ing the startup. While, too large a capacitor may increase the startup time. (b) Vcc resistor (Ra) : The typical value for Ra is 5-20Ω. In the case of multiple outputs flyback converter, the voltage of the lightly loaded output such as Vcc varies as the load currents of other outputs change due to the imperfect coupling of the transformer. Ra reduces the sensitivity of Vcc to other outputs and improves the regulations of Vcc. (36) where Vo1 is the reference output voltage. (b) The capacitor connected to feedback pin (CB) is related to the shutdown delay time in an overload condition by T delay = ( V SD – 2.5 ) ⋅ C B ⁄ I delay (37 where VSD is the shutdown feedback voltage and Idelay is the shutdown delay current. These values are given in the data sheet. In general, a 10 ~ 50 ms delay time is typical for most applications. Because CB also determines the high frequency pole (wpc) of the compensator transfer function as shown in equation (36), too large a CB can limit the control bandwidth by placing wpc at too low a frequency. Typical value for CB is 10-50nF. (c) The resistors Rbias and RD used together with the optocoupler H11A817A and the shunt regulator KA431 should be designed to provide proper operating current for the KA431 and to guarantee the full swing of the feedback voltage for the FPS device chosen. In general, the minimum cathode voltage and current for the KA431 are 2.5V and 1mA, respectively. Therefore, Rbias and RD should be designed to satisfy the following conditions. V o1 – V OP – 2.5 ----------------------------------------- > I FB RD V OP ------------- > 1mA R bias (38) (39) where Vo1 is the reference output voltage, VOP is opto-diode forward voltage drop, which is typically 1V and IFB is the feedback current of FPS, which is typically 1mA. For example, Rbias< 1kΩ and RD < 1.5kΩ for Vo1=5V. Miscellaneous (a) Vcc capacitor (Ca) : The typical value for Ca is 10-50uF, which is enough for most application. A smaller capacitor than this may result in an under voltage lockout of FPS dur©2003 Fairchild Semiconductor Corporation 11 AN4137 APPLICATION NOTE - Summary of symbols Aw Ae Bsat Co(n) Dmax Eff fL fs Idspeak Idsrms Ids2 Iover Isec(n)rms ID(n)rms Icap(n)rms Io(n) KL(n) KRF Lm Llk Losssn Npmin Np Ns1 Ns(n) Po Pin Rc(n) Rsn RL Vlinemin Vlinemax VDCmin VDCmax Vdsnom Vo1 VF1 Vcc* VFa VD(n) ∆Vo(n) VRO Vsn Vsn2 ∆Vsn Vdsmax : Winding window area of the core in mm2 : Cross sectional area of the core in mm2 : Saturation flux density in tesla. : Output capacitor of the n-th output : Maximum duty cycle ratio : Estimated efficiency : Line frequency : Switching frequency of FPS : Maximum peak current of MOSFET : RMS current of MOSFET : Maximum peak drain current at the maximum input voltage condition. : FPS current limit level. : RMS current of the secondary winding for the n-th output : Maximum rms current of the rectifier diode for the n-th output : RMS Ripple current of the output capacitor for the n-th output : Output load current for the n-th output : Load occupying factor for the n-th output : Current ripple factor : Transformer primary side inductance : Transformer primary side leakage inductance : Maximum power loss of the snubber network in normal operation : The minimum number of turns for the transformer primary side to avoid saturation : Number of turns for primary side : Number of turns for the reference output : Number of turns for the n-th output : Maximum output power : Maximum input power : Effective series resistance (ESR) of the n-th output capacitor. : Snubber resistor : Effective total output load resistor of the controlled output : Minimum line voltage : Maximum line voltage : Minimum DC link voltage : Maximum DC line voltage : Maximum nominal MOSFET voltage : Output voltage of the reference output. : Diode forward voltage drop of the reference output. : Nominal voltage for Vcc : Diode forward voltage drop of Vcc winding : Maximum voltage of the rectifier diode for n-th output : Output voltage ripple for the n-th output : Output voltage reflected to the primary : Snubber capacitor voltage under minimum input voltage and full load condition : Snubber capacitor voltage under maximum input voltage and full load condition : Maximum Snubber capacitor voltage ripple : Maximum voltage stress of the MOSFET ©2003 Fairchild Semiconductor Corporation 12 APPLICATION NOTE AN4137 Appendix : Design example using FPS Design Assistant Application Set-top Box Output Input voltage Output voltage (Max Current) Power Device FSDM07652R 47W ± 5% ± 5% ± 5% ±5% ±5% 3.3V (2A) 5V (2A) 12V (1.5A) 18V (0.5A) 33V (0.1A) 85V-265VAC 1. Define the system specifications Minimum Line voltage (Vlinemin) 85 V.rms Maximum Line voltage (Vlinemax) 265 V.rms Line frequency (fL) Ripple spec 60 Hz Vo(n) 1st output for feedback 2nd output 3rd output 4th output 5th output 6th output Maximum output power (Po) = Estimated efficiency (Eff) 3.3 5 12 18 33 V V V V V V 46.9 W 70 % Io(n) 2.00 2.00 1.50 0.50 0.10 Po(n) A A A A A A 7 10 18 9 3 0 W W W W W W KL(n) 14 21 38 19 7 0 % % % % % % 67.0 W Maximum input power (Pin) = ☞ The estimated efficiency (Eff) is set to be 0.7 considering the low voltage outputs (3.3V and 5V) 2. Determine DC link capacitor and DC link voltage range DC link capacitor (CDC) 150 uF min min Minimum DC link voltage (VDC ) = 92 V (VDCmax)= 375 V Maximum DC link voltage ☞ Since the input power is 67 W, the DC link capacitor is set to be 150uF by 2uF/Watt. 3. Determine Maximum duty ratio (Dmax) Maximum duty ratio (Dmax) (Vdsnom) Max nominal MOSFET voltage = Output voltage reflected to primary (VRO)= 0.48 460 V 85 V ☞ Dmax is set to be 0.48 so that Vdsnom would be about 70% of BVdss (650V× ×0.7=455V) 4. Determine transformer primary inductance (Lm) Switching frequency of FPS (fs) 66 kHz 0.33 Ripple factor (KRF) 671 uH Primary side inductance (Lm) = Maximum peak drain current (Idspeak) = RMS drain current rms (Ids ) = Maximum DC link voltage in CCM (VDCCCM) ©2002 Fairchild Semiconductor Corporation K RF = 1 ( DCM ) K RF < 1 (CCM ) ∆I I EDC 2.01 A 1.07 A ## K RF = ∆I 2 I EDC 375 V 13 AN4137 APPLICATION NOTE 5. Choose the proper FPS considering the input power and current limit 2.50 A Typical current limit of FPS (Iover) Minimum Iover considering tolerance of 12% 2.20 A > 2.01 A ->O.K. ☞ Since the maximum peak drain current (Idspeak) is 2.0A, FSDM07652R is chosen, whose current limit level (Iover) is 2.5A. The current limit tolerance (12%) is considered. 6. Determine the proper core and the minimum primary turns Saturation flux density (Bsat) 0.35 T Cross sectional area of core (Ae) Minimum primary turns 109.4 mm (Npmin)= 2 43.8 T ☞ Ferrite core EER3530 is chosen (Ae=109.4 Aw=210mm2), which is a little bit larger than the core recommended in table 1 to provide enough winding window area. mm2, 7. Determine the number of turns for each output Vo(n) VF(n) # of turns 1.2 V 6.9 => 7T 0.5 V 2 => 2T 0.5 V 2.9 => 3T 1.2 V 6.9 => 7T 1.2 V 10 T 10.1 => 1.2 V 18.0 => 18 T 0V 0.0 => 0T Primary turns (Np)= 45 T --->enough turns 2130 nH/T2 0.34631 mm Vcc (Use Vcc start voltage) 1st output for feedback 2nd output 3rd output 4th output 5th output 6th output VF : Forward voltage drop of rectifier diode 12 3.3 5 12 18 33 0 Ungapped AL value (AL) Gap length (G) ; center pole gap = V V V V V V V ☞ In general, the optimum turn ratio between 5V and 3.3V is 3/2, considering the diode forward voltage drop. 8. Determine the wire diameter for each winding Primary winding Vcc winding 1st output winding 2nd output winding 3rd output winding 4th output winding 5th output winding 6th output winding Copper area (Ac) = Fill factor (KF) Required window area (Awr) Diameter 0.5 0.3 0.4 0.4 0.4 0.4 0.4 Parallel 1T mm mm 2T mm 4T mm 4T mm 3T mm 2T mm 1T mm T 2 19.70 mm 0.15 2 rms ID(n)rms 1.1 0.1 3.5 3.7 2.8 0.9 0.2 ##### A A A A A A A A (A/mm2) 5.44 0.71 6.97 7.30 7.30 3.76 1.55 ##### 2 131.33 mm ☞ Since the windings for 3.3V and 5V are short with small number of turns, relatively large current densities (> 5A/mm2) are allowed. The fill factor is set to be 0.15 due to multiple outputs. 14 ©2002 Fairchild Semiconductor Corporation APPLICATION NOTE AN4137 9. Choose the rectifier diode in the secondary side ID(n)rms 0.10 3.50 3.67 2.75 0.95 0.19 ##### VD(n) Vcc diode 1st output diode 2nd output diode 3rd output diode 4th output diode 5th output diode 6th output diode 70 20 29 70 103 184 0 V V V V V V V A A A A A A A Vcc winding UF4003 (200V /1A, VF=1V) Ultra Fast Recovery Diode 1st output (3.3V) SB540 (40V/5A, VF=0.55V) × 2 Schottky Barrier Diode 2nd output (5V) SB560 (60V/5A, VF=0.67V) × 2 Schottky Barrier Diode 3rd output (12V) EGP30D (200V/3A, VF=0.95V) Ultra Fast Recovery Diode 4st output (18V) EGP20D (200V/2A, VF=0.95V) Ultra Fast Recovery Diode 5st output (30V) UF4004 (400V /1A, VF=1V) Ultra Fast Recovery Diode 10. Determine the output capacitor RC(n) Co(n) 1st output capacitor 2nd output capacitor 3rd output capacitor 4th output capacitor 5th output capacitor 6th output capacitor 2000 2000 330 470 47 uF uF uF uF uF uF 100 100 300 300 480 Icap(n) mΩ 2.9 mΩ 3.1 mΩ 2.3 mΩ 0.8 mΩ 0.2 mΩ ##### A A A A A A ΔVo(n) 0.64 0.67 1.53 0.52 0.18 ##### V V V V V V Since the voltage ripples for 3.3V, 5V and 12V exceed the ripple spec of ± 5%, additional LC filter stage should be used for these three outputs. 220uF capacitor together with 2.2uH inductor are used for the post filter. To attenuate the voltage ripple caused by switching, the corner frequency of the post filter (fo) is set at about one decade below the switching frequency. fo = 1 2π L p1C p1 = 10 6 2π 2.2 × 220 11. Design RCD snubber Primary side leakage inductance (Llk) = 7.2kHz 4.5 uH Maximum Voltage of snubber capacitor (Vsn) Maximum snubber capacitor voltage ripple Snubber resistor (Rsn)= Snubber capacitor (Csn)= Power loss in snubber resistor (Psn)= 190 5 33.1 9.2 1.1 max (Ids2) = Peak drain current at VDC 1.75 A Max Voltage of Csn at VDCmax (Vsn2)= Max Voltage stress of MOSFET max (Vds )= V % ㏀ 0.185 nF W (In Normal Operation) 172 V 547 V ☞ The snubber capacitor and snubber resistor are chosen as 10nF and 33kΩ Ω, respectively. The maximum voltage stress on the MOSFET is designed to be 84% of 650V BVdss voltage of the FSDM07652R. The actual Vdsmax would be lower than this. ©2002 Fairchild Semiconductor Corporation 15 AN4137 APPLICATION NOTE 12. Design Feedback control loop Control-to-output Control-to-output Control-to-output Control-to-output DC gain = zero (wz) = RHP zero (wrz)= pole (wp)= 2 5000 rad/s => fz= 694765 rad/s => frz= 2153 rad/s => fp= Voltage divider resistor (R1) 5.6 ㏀ Voltage divider resistor (R2)= Opto coupler diode resistor (RD) KA431 Bias resistor (Rbias) Feeback pin capacitor (CB) = Feedback Capacitor (CF) = Feedback resistor (RF) = 18 1 1.2 33 47 1.2 Feedback integrator gain (wi) = Compensator zero (wzc)= Compensator pole (wpc)= ㏀ ㏀ ㏀ nF nF ㏀ 796 Hz 110,631 Hz 343 Hz vo1' FPS vFB CB vo1 RD ibias Rbias iD 1:1 B CF R1 RF KA431 R2 11398 rad/s => fi= 3129 rad/s => fzc= 10101 rad/s => fpc= 1,815 Hz 498 Hz 1,608 Hz 60 Gain (dB) 40 20 0 10 100 -20 -40 0 10 Phase (degree) -30 -60 100 16 3.64105 25 3.63 40 3.60099 63 3.53167 100 3.36222 160 2.96516 250 2.20575 400 1000 0.89557 630 -0.6501 1000 -2.0185 1600 -2.9016 2500 -3.3275 4000 -3.5257 frequency (Hz) 6300 -3.5983 10000 -3.6107 16000 -3.5697 1000 25000 -3.4485 40000 -3.1334 63000 -2.448 100000 -1.0745 41 37 33 29 25 21 18 15 13 11 9 6 3 -1 -5 -9 # # # # 44.7 16 -2 -88.7 Control-to-output 40.9 25 -2 -88 36.8 Compensator 40 -4 -86.8 32.8 T (Closed 63loop -6 gain)-85 28.7 100 -9 -82.2 24.4 160 ## -77.9 20.3 250 ## -72.2 15.9 400 ## 100000 -65.2 10000 12.1 630 ## -59.7 8.75 1000 ## -58.3 5.74 1600 ## -62.1 2.74 2500 ## -68.5 -0.8 4000 -8 -75.2 -4.5 6300 -7 -80.2 -8.4 10000 -8 -83.7 -12 16000 ## -86 10000 100000 -16 25000 ## -87.5 -20 40000 ## -88.4 -23 63000 ## -89 -26 1E+05 ## -89.4 # # # # # # # # # # # # # # # # # # # # -90 -120 -150 -180 frequency (Hz) ☞ The control bandwidth is 4kHz. Since the crossover frequency is too close to the corner frequency of the post filter (fo=7.2 kHz), the controller is designed to have enough phase margin when ignoring the effect of the post filter. 16 ©2002 Fairchild Semiconductor Corporation APPLICATION NOTE AN4137 Design Summary • For the FPS, FSDM07652R is chosen. This device has a fixed switching frequency of 66kHz. Startup and soft-start circuits are implemented inside the device. • To limit the current, 10 ohms resistors (Ra and Rdamp) are used in series with Da and DR5. These damping resistors improve the regulations of the very lightly loaded outputs. Figure 17 shows the final schematic of the flyback converter designed by FPS Design Assistant. VO5 33V Rdamp 10 DR5 NS5 Co5 UF4004 47uF/ 50V VO4 18V DR4 NS4 EGP20D 470uF/25V Rsn CDC 10nF 1kV Csn Dsn Cp3 Co3 220uF/ 25V 330uF/25V Np Lp2 DR2 150uF/400V 2.2 uH EGP30D NS3 33k 2W VO3 12V Lp3 DR3 GBLA06 Co4 VO2 5V 2.2 uH SB560 NS2 Cp2 Co2 UF4007 220uF/ 10V 1000uF× 2 /10V 6 Vstr 1.5nF/275Vac CL2 Lp1 1 Vcc GND FB Line Filter (33mH) 4 2 3 Da VO1 3.3V 2.2 uH SB540 Cp1 Co1 NS1 220uF/ 10V 1000uF× 2 /10V Ca 33uF/35V CL1 DR1 UF4003 R a 10 FPS (DM07652R) CL2 Drain Na 1k Rd Rbias 0.47uF/275Vac 1k H11A817A 5.6k R1 RL1 CB 1.5M NTC 5D-13 H11A817A 1.2k 47nF RF CF 33nF Fuse KA431 18k AC line R2 Figure 17. The final schematic of the flyback converter ©2003 Fairchild Semiconductor Corporation 17 AN4137 APPLICATION NOTE Experimental Verification In order to show the validity of the design procedure presented in this paper, the converter of the design example has been built and tested. All the circuit components are used as designed in the design example and the measured transformer characteristics are shown in table 3. Figure 18 shows the FPS drain current and DC link voltage waveforms at the minimum input voltage and full load condition. As can be seen, the maximum peak drain current (Idspeak) is 2A and the minimum DC link voltage (VDCmin) is about 90V. The designed values are 2.01A and 92V, respectively. Figure 19 shows the FPS drain current and voltage waveforms at the minimum input voltage and full load condition. As designed, the maximum duty ratio (Dmax) is about 0.5 and the maximum peak drain current (Idspeak) is 2A. Figure 20 shows the FPS drain current and voltage waveforms at the maximum input voltage and full load condition. The maximum voltage stress on the MOSFET is about 520V, which is lower than the designed value (547V). This is because of the lossy discharge of the inductor or the stray capacitance. Another reason is that the power conversion efficiency at the maximum input voltage is higher than the estimated efficiency used in step-1. Figure 18. Waveforms of drain current and DC link voltage at 85Vac and full load condition (time:2ms/div) As calculated in design step-4, the converter operates at the boundary between CCM and DCM under the maximum input voltage and full load condition (The maximum DC link voltage guaranteeing CCM at full load was obtained as 375V in design step-4). Figure 21 shows the current and voltage waveforms of the first output (3.3V) rectifier diode. The maximum reverse voltage of this diode was calculated as 20V in step-9 and the measured value is 23V. Table 4 shows the line regulation of each output. 3.3V and 5V output shows ±3% and ±4% regulations, respectively. Figure 22 shows the measured efficiency at the full load condition for different input voltages. The minimum efficiency is about 73% at the minimum input voltage condition better than the 70% target efficiency specified in step-1. Core EER3530 (ISU ceramics) Primary side inductance 682 uH @ 70kHz Leakage inductance Resistance Figure 19. Waveforms of drain current and voltage at 85Vac and full load condition (time : 5us/div) 4.5 uH @70kHz with all other windings shorted. 0.76 Ω Table 3. The measured transformer characteristics Figure 20. Waveforms of drain current and voltage at 265Vac and full load condition (time : 5us/div) 18 ©2002 Fairchild Semiconductor Corporation APPLICATION NOTE AN4137 Efficiency 0.81 0.80 0.79 0.78 0.77 0.76 0.75 0.74 0.73 0.72 85 115 145 175 205 235 265 Input voltage (Vac) Figure 22. Measured efficiency Figure 21. Current and voltage waveforms of the first output (3.3V) rectifier diode at 265Vac and full load condition (time : 5us/div) Input voltage Vo1 (3.3V) Vo2 (5V) Vo3 (12V) Vo4 (18V) Vo5 (33V) 85Vac 3.21 V 5.18 V 12.88 V 19.7 V 35.7 V -2.7 % 3.6 % 7.3 % 9.4 % 8.2 % 3.21 V 5.14 V 12.77 V 19.4 V 34.6 V -2.7 % 2.8 % 6.4 % 7.8 % 4.8 % 3.21 V 5.11 V 12.67 V 19.2 V 34.1 V -2.7 % 2.2 % 5.6 % 6.7 % 3.2 % 3.21 V 5.09 V 12.57 V 19.1 V 33.8 V -2.7 % 1.8 % 4.8 % 5.9 % 2.4 % 3.21 V 5.08 V 12.52 V 19.0 V 33.6 V -2.7 % 1.6 % 4.3 % 5.4 % 1.9 % 3.21 V 5.07 V 12.48 V 18.9 V 33.5 V -2.7 % 1.4 % 4.0 % 5.1 % 1.6 % 3.21 V 5.06 V 12.47 V 18.9 V 33.5 V -2.7 % 1.2 % 3.9 % 5.1 % 1.4 % 115Vac 145Vac 175Vac 205Vac 235Vac 265Vac Table 4. Line regulation of each output at full load condition ©2002 Fairchild Semiconductor Corporation 19 AN4137 APPLICATION NOTE by Hang-Seok Choi / Ph. D FPS Application Group / Fairchild Semiconductor Phone : +82-32-680-1383 Facsimile : +82-32-680-1317 E-mail : [email protected] DISCLAIMER FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO ANY PRODUCTS HEREIN TO IMPROVE RELIABILITY, FUNCTION OR DESIGN. FAIRCHILD DOES NOT ASSUME ANY LIABILITY ARISING OUT OF THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN; NEITHER DOES IT CONVEY ANY LICENSE UNDER ITS PATENT RIGHTS, NOR THE RIGHTS OF OTHERS. LIFE SUPPORT POLICY FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR CORPROATION. As used herein: 1. Life support devices or systems are devices or systems which, (a) are intended for surgical implant into the body, or (b) support or sustain life, or (c) whose failure to perform when properly used in accordance with instructions for use provided in the labeling, can be reasonably expected to result in significant injury to the user. 2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause the failure of the life support device or system, or to affect its safety or effectiveness. www.fairchildsemi.com 1/13/04 0.0m 002 Stock#ANxxxxxxxxx 2003 Fairchild Semiconductor Corporation

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