Design Guidelines for Off-line Flyback

www.fairchildsemi.com
Application Note AN4137
Design Guidelines for Off-line Flyback Converters
Using Fairchild Power Switch (FPS)
Abstract
The step-by-step design procedure described in this paper
helps engineers to design SMPS easily. In order to make the
design process more efficient, a software design tool, FPS
design assistant that contains all the equations described in
this paper is also provided. The design procedure is verified
through experimental prototype converter.
This paper presents practical design guidelines for off-line
flyback converters employing FPS (Fairchild Power
Switch). Switched mode power supply (SMPS) design is
inherently a time consuming job requiring many trade-offs
and iterations with a large number of design variables.
L P(n)
D R(n)
Bridge
rectifier
diode
V DC
+
CDC
R sn
Vsn
+
C sn
Np
N S(n)
CO(n)
-
VO(n)
CP(n)
Dsn
D R1
FPS
L P1
Drain 1
N S1
AC line
GND 2
FB
4
Vcc
3
Ra
Da
CO1
Rd
Rbias
H11A817A
CB
Ca
Na
V O1
CP1
H11A817A
R1
RF
KA431
CF
R2
Figure 1. Basic Off-line Flyback Converter Using FPS
1. Introduction
Figure 1 shows the schematic of the basic off-line flyback
converter using FPS, which also serves as the reference
circuit for the design process described in this paper.
Because the MOSFET and PWM controller together with
various additional circuits are integrated into a single
package, the design of SMPS is much easier than the discrete
MOSFET and PWM controller solution. This paper provides
a step-by-step design procedure for a FPS based off-line
flyback converter, which includes designing the transformer
and output filter, selecting the components and closing the
feedback loop. The design procedure described herein is
general enough to be applied to various applications. The
design procedure presented in this paper is also implemented in a software design tool (FPS design assistant) to
enable the engineer finish their SMPS design in a short time.
In the appendix, a step-by-step design example using the
software tool is provided. An experimental flyback converter
from the design example has been built and tested to show
the validity of the design procedure.
Rev. 1.3.0
©2003 Fairchild Semiconductor Corporation
AN4137
APPLICATION NOTE
2. Step-by-step Design Procedure
1. Determine the system specifications
(Vlinemin, Vlinemax, fL, Po, Eff)
In this section, a design procedure is presented using the
schematic of figure 1 as a reference. In general, most FPS
devices have the same pin configuration from pin 1 to pin 4,
as shown in figure 1. Figure 2 illustrates the design flow
chart. The detailed design procedures are as follows:
(1) STEP-1 : Define the system specifications
2. Determine DC link capacitor (CDC)
and DC link voltage range
- Line voltage range (Vlinemin and Vlinemax).
- Line frequency (fL).
- Maximum output power (Po).
3. Determine the maximum duty
ratio (Dmax)
- Estimated efficiency (Eff) : It is required to estimate the
power conversion efficiency to calculate the maximum input
power. If no reference data is available, set Eff = 0.7~0.75 for
low voltage output applications and Eff = 0.8~0.85 for high
voltage output applications.
4. Determine the transformer primary side
inductance (Lm)
5. Choose proper FPS considering input
power and Idspeak
With the estimated efficiency, the maximum input power is
given by
6. Determine the proper core and the
minimum primary turns (Npmin)
P
Pin = ------oE ff
7. Determine the number of turns for each
output
For multiple output SMPS, the load occupying factor for
each output is defined as
8. Determine the wire diameter for each
winding
Is the winding window
area (Aw) enough ?
(1)
Po (n )
KL ( n ) = -----------Po
Y
(2)
where Po(n) is the maximum output power for the n-th output. For single output SMPS, KL(1)=1.
N
(2) STEP-2 : Determine DC link capacitor (CDC) and the
DC link voltage range.
Y
Is it possible to change the core ?
N
It is typical to select the DC link capacitor as 2-3uF per watt
of input power for universal input range (85-265Vrms) and
1uF per watt of input power for European input range (195V265Vrms). With the DC link capacitor chosen, the minimum
link voltage is obtained as
9. Choose the proper rectifier diode for each
output
V DC
min
=
2 ⋅ ( V line
10. Determine the output capacitor
11. Design the RCD snubber
12. Feedback loop design
(3)
where Dch is the DC link capacitor charging duty ratio
defined as shown in figure 3, which is typically about 0.2
and Pin, Vlinemin and fL are specified in step-1.
The maximum DC link voltage is given as
V DC
Design finished
P in ⋅ ( 1 – D ch )
) – -----------------------------------C DC ⋅ f L
min 2
max
=
2V line
max
(4)
where Vlinemax is specified in step-1.
Figure 2. Flow chart of design procedure
2
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4137
Minimum DC link voltage
DC link voltage
T1
Dch = T1 / T2
= 0.2
T2
Figure 3. DC Link Voltage Waveform
In the case of a CCM flyback converter, the design process is
straight forward since the input-to-output voltage gain
depends only on the duty cycle. Meanwhile, the input-to-output voltage gain of a DCM flyback converter depends not
only on the duty cycle but also on the load condition, which
causes the circuit design to be somewhat complicated. However, it is generally accepted that a DCM flyback converter is
designed to operate at the boundary of DCM and CCM with
minimum input voltage and maximum load as shown in Fig.
4. This minimizes MOSFET conduction losses. Therefore,
under these circumstances, we can use the same voltage gain
equation as the CCM flyback converter with maximum load
and minimum input voltage.
(3) STEP-3 : Determine the maximum duty ratio (Dmax).
A Flyback converter has two kinds of operation modes ;
continuous conduction mode (CCM) and discontinuous conduction mode (DCM). CCM and DCM have their own
advantages and disadvantages, respectively. In general,
DCM provides better switching conditions for the rectifier
diodes, since the diodes are operating at zero current just
before becoming reverse biased. The transformer size can be
reduced using DCM because the average energy storage is
low compared to CCM. However, DCM inherently causes
high RMS current, which increases the conduction loss of
the MOSFET and the current stress on the output capacitors.
Therefore DCM is usually recommended for high voltage
and low current output applications. Meanwhile, CCM is
preferred for low voltage and high current output applications.
-
VDC
+
VRO
-
+
F PS
D rain
+
GND
V ds
-
V RO
Minimum input voltage
and full load condition
MOSFET
Drain
Current
Rectifier
Diode
Current
D
As input voltage increases or
load current decreases
MOSFET
Drain
Current
Rectifier
Diode
Current
D
Figure 4. Current waveforms of DCM flyback converter
©2002 Fairchild Semiconductor Corporation
V DC
0V
Figure 5. The output voltage reflected to the primary
When the MOSFET in the FPS is turned off, the input voltage (VDC) together with the output voltage reflected to the
primary (VRO) are imposed on the MOSFET as shown in figure 5. After determining Dmax, VRO and the maximum nominal MOSFET voltage (Vdsnom) are obtained as
D max
min
V RO = ----------------------- ⋅ V DC
1 – D max
V ds
nom
= V DC
max
+ VRO
(5)
(6)
where VDCmin and VDCmax are specified in equations (3) and
(4) respectively. As can be seen in equation (5) and (6), the
voltage stress on MOSFET can be reduced, by decreasing
Dmax. However, this increases the voltage stresses on the rectifier diodes in the secondary side. Therefore, it is desirable
to set Dmax as large as possible if there is enough margin in
the MOSFET voltage rating. The maximum duty ratio
3
AN4137
APPLICATION NOTE
(Dmax) should be determined so that Vdsnom would be
65~70% of the MOSFET voltage rating considering the voltage spike caused by the leakage inductance. In the case of
650V rated MOSFET, it is typical to set Dmax to be 0.45~0.5
for an universal input range application. Because the current
mode controlled flyback converter operating in CCM causes
sub-harmonic oscillation with duty ratio larger than 0.5, set
Dmax to be smaller than 0.5 for CCM.
(4) STEP-4 : Determine the transformer primary side
inductance (Lm).
The operation changes between CCM and DCM as the load
condition and input voltage vary. For both operation modes,
the worst case in designing the inductance of the transformer
primary side (Lm) is full load and minimum input voltage
condition. Therefore, Lm is obtained in this condition as
V DC
CCM
–1

1
1 -
=  --------------------------- – ---------
 2L m f s P in VRO
(12)
where Pin, VRO and Lm are specified in equations (1), (5) and
(7), respectively, and fs is the FPS switching frequency.
If the result of equation (12) has a negative value, the converter is always in CCM under the full load condition regardless of the input voltage variation.
∆I
I ds peak
I EDC
K RF =
∆I
2I EDC
CCM operation : KRF < 1
2
min
( VDC
⋅ D max )
L m = --------------------------------------------2Pin f s KRF
(7)
I ds peak
where VDCmin is specified in equation (3), Dmax is specified
in step-3, Pin is specified in step-1, fs is the switching frequency of the FPS device and KRF is the ripple factor in full
load and minimum input voltage condition, defined as
shown in figure 6. For DCM operation, KRF = 1 and for
CCM operation KRF < 1. The ripple factor is closely related
with the transformer size and the RMS value of the MOSFET current. Even though the conduction loss in the MOSFET can be reduced through reducing the ripple factor, too
small a ripple factor forces an increase in transformer size.
When designing the flyback converter to operate in CCM, it
is reasonable to set KRF = 0.25-0.5 for the universal input
range and KRF = 0.4-0.8 for the European input range.
Once Lm is determined, the maximum peak current and RMS
current of the MOSFET in normal operation are obtained as
I ds
I ds
rms
where
and
peak
=
∆I
= I EDC + ----2
2
∆ I 2 D max
3 ( IEDC ) +  ----- ------------2
3
P in
I EDC = ------------------------------------min
V DC
⋅ D max
V
min
D
DC
max
∆ I = -----------------------------------
Lm fs
(8)
(9)
(10)
(11)
where Pin, VDCmin and Lm are specified in equations (1), (3),
and (7) respectively, Dmax is specified in step-3 and fs is the
FPS switching frequency.
The flyback converter designed for CCM at the minimum
input voltage and full load condition may enter into DCM as
the input voltage increases. The maximum input voltage
guaranteeing CCM in the full load condition is obtained as
4
∆I
I EDC
K RF =
∆I
2I EDC
DCM operation : KRF =1
Figure 6. MOSFET Drain Current and Ripple Factor (KRF)
(5) STEP-5 : Choose the proper FPS considering input
power and peak drain current.
With the resulting maximum peak drain current of the MOSFET (Idspeak) from equation (8), choose the proper FPS of
which the pulse-by-pulse current limit level (Iover) is higher
than Idspeak. Since FPS has ± 12% tolerance of Iover, there
should be some margin in choosing the proper FPS
device.The FPS lineup with proper power rating is also
included in the software design tool.
(6) STEP-6 : Determine the proper core and the minimum
primary turns.
Actually, the initial selection of the core is bound to be crude
since there are too many variables. One way to select the
proper core is to refer to the manufacture's core selection
guide. If there is no proper reference, use the table 1 as a
starting point. The core recommended in table 1 is typical for
the universal input range, 67kHz switching frequency and
single output application. When the input voltage range is
195-265 Vac or the switching frequency is higher than
67kHz, a smaller core can be used. For an application with
multiple outputs, usually a larger core should be used than
recommended in the table.
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4137
With the chosen core, the minimum number of turns for the
transformer primary side to avoid the core saturation is given
by
NP
min
L m Iover
6
= ------------------- × 10
B sat A e
(turns)
(13)
where Lm is specified in equation (7), Iover is the FPS pulseby-pulse current limit level, Ae is the cross-sectional area of
the core as shown in figure 7 and Bsat is the saturation flux
density in tesla. Figure 8 shows the typical characteristics of
ferrite core from TDK (PC40). Since the saturation flux density (Bsat) decreases as the temperature goes high, the high
temperature characteristics should be considered.
If there is no reference data, use Bsat =0.3~0.35 T. Since the
MOSFET drain current exceeds Idspeak and reaches Iover in a
transition or fault condition, Iover is used in equation (13)
instead of Idspeak to prevent core saturation during transition.
Output
Power
EI core
EE core
EPC core
0-10W
EI12.5
EI16
EI19
EE8
EE10
EE13
EE16
EPC10
EPC13
EPC17
10-20W
EI22
EE19
EPC19
EE22
EPC25
EER25.5
EE25
EPC30
EER28
20-30W
EER core
EI25
30-50W
EI28
EI30
50-70W
EI35
EE30
EER28L
70-100W
EI40
EE35
EER35
100-150W EI50
EE40
EER40
EER42
150-200W EI60
EE50
EE60
EER49
Table 1. Core quick selection table (For universal input
range, fs=67kHz and single output)
Aw
(7) STEP-7 : Determine the number of turns for each
output
Figure 9 shows the simplified diagram of the transformer.
First, determine the turns ratio (n) between the primary side
and the feedback controlled secondary side as a reference.
NP
V R0
= ------------------------n = --------Ns1
V o1 + V F1
Ae
Figure 7. Window Area and Cross Sectional Area
M agnetization Curves (typical)
M aterial :PC40
25 ℃
500
60 ℃
100 ℃
400
Flux density B (mT)
120 ℃
(14)
where Np and Ns1 are the number of turns for primary side
and reference output, respectively, Vo1 is the output voltage
and VF1 is the diode (DR1) forward voltage drop of the reference output.
Then, determine the proper integer for Ns1 so that the resulting Np is larger than Npmin obtained from equation (13).
The number of turns for the other output (n-th output) is
determined as
Vo (n ) + VF ( n)
⋅ Ns1
N s ( n ) = --------------------------------V o1 + V F1
( turns )
( 15 )
300
The number of turns for Vcc winding is determined as
200
V cc * + V Fa
- ⋅ N s1
N a = --------------------------V o1 + VF1
100
0
0
800
M agnetic field H (A/m )
1600
Figure 8. Typical B-H characteristics of ferrite core
(TDK/PC40)
©2002 Fairchild Semiconductor Corporation
( turns )
( 16 )
where Vcc* is the nominal value of the supply voltage of the
FPS device, and VFa is the forward voltage drop of Da as
defined in figure 9. Since Vcc increases as the output load
increases, it is proper to set Vcc* as Vcc start voltage (refer to
the data sheet) to avoid the over voltage protection condition
during normal operation.
5
AN4137
APPLICATION NOTE
severe eddy current losses as well as to make winding easier.
For high current output, it is better to use parallel windings
with multiple strands of thinner wire to minimize skin effect.
+ V F(n) -
D R(n)
Np
V RO
Check if the winding window area of the core, Aw (refer to
figure 7) is enough to accommodate the wires. The required
winding window area (Awr) is given by
+
V O(n)
N S(n)
A w r = Ac ⁄ K F
-
+
Da
+
V cc *
where Ac is the actual conductor area and KF is the fill factor.
Typically the fill factor is 0.2~0.25 for single output application and 0.15~0.2 for multiple outputs application.
If the required window (Awr) is larger than the actual window
area (Aw), go back to the step-6 and change the core to a bigger one. Sometimes it is impossible to change the core due to
cost or size constraints. If the converter is designed for CCM
and the winding window (Aw) is slightly insufficient, go back
to step-4 and reduce Lm by increasing the ripple factor (KRF).
Then, the minimum number of turns for the primary (Npmin)
of the equation (13) will decrease, which results in the
reduced required winding window area (Awr).
+ V F1 -
- V Fa +
+
V O1
D R1
N S1
Na
-
-
(19)
Figure 9. Simplified diagram of the transformer
With the determined turns of the primary side, the gap length
of the core is obtained as
(9) STEP-9 : Choose the rectifier diode in the secondary
side based on the voltage and current ratings.
The maximum reverse voltage and the rms current of the rectifier diode (DR(n)) of the n-th output are obtained as
2
 NP
1
G = 40 πA e  -------------------- – ------
1000L
A

m
L
( mm )
( 17 )
where AL is the AL-value with no gap in nH/turns2, Ae is the
cross sectional area of the core as shown in figure 7, Lm is
specified in equation (7) and Np is the number of turns for
the primary side of the transformer
(8) STEP-8 : Determine the wire diameter for each
winding based on the rms current of each output.
The rms current of the n-th secondary winding is obtained as
I sec ( n )
rms
= I ds
rms
rms
V RO ⋅ K L ( n )
1 – D max
----------------------- ⋅ -------------------------------------( Vo (n ) + VF ( n) )
D max
( 18 )
are specified in equations (5) and (9),
where VRO and Ids
Vo(n) is the output voltage of the n-th output, VF(n) is the
diode (DR(n)) forward voltage drop, Dmax is specified in step3 and KL(n) is the load occupying factor for n-th output
defined in equation (2).
The current density is typically 5A/mm2 when the wire is
long (>1m). When the wire is short with a small number of
turns, a current density of 6-10 A/mm2 is also acceptable.
Avoid using wire with a diameter larger than 1 mm to avoid
6
max
V DC
⋅ ( Vo ( n) + VF ( n ) )
V D ( n ) = Vo ( n ) + --------------------------------------------------------------V RO
( 20 )
V RO K L ( n )
1 – Dmax
----------------------- ⋅ -------------------------------------( Vo( n) + VF ( n) )
D max
( 21 )
ID ( n)
rms
= I ds
rms
where KL(n), VDCmax, VRO, Idsrms are specified in equations
(2), (4), (5) and (9) respectively, Dmax is specified in step-3,
Vo(n) is the output voltage of the n-th output and VF(n) is the
diode (DR(n)) forward voltage. The typical voltage and
current margins for the rectifier diode are as follows
V RRM > 1.3 ⋅ V D ( n )
IF > 1.5 ⋅ I D ( n )
rms
(22)
(23)
where VRRM is the maximum reverse voltage and IF is the
average forward current of the diode.
A quick selection guide for Fairchild Semiconductor rectifier
diodes is given in table 2. In this table trr is the maximum
reverse recovery time.
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4137
(10) STEP-10 : Determine the output capacitor
considering the voltage and current ripple.
Schottky Barrier Diode
Products
VRRM
IF
trr
Package
SB330
30 V
3A
-
TO-210AD
SB530
30 V
5A
-
TO-210AD
MBR1035
35 V
10 A
-
TO-220AC
MBR1635
35 V
16 A
-
TO-220AC
SB340
40 V
3A
-
TO-210AD
SB540
40 V
5A
-
TO-210AD
SB350
50 V
3A
-
TO-210AD
SB550
50 V
5A
-
TO-210AD
SB360
60 V
3A
-
TO-210AD
SB560
60 V
5A
-
TO-210AD
MBR1060
60 V
10 A
-
TO-220AC
MBR1660
60 V
16 A
-
TO-220AC
Ultra Fast Recovery diode
Products
VRRM
IF
trr
Package
EGP10B
100 V
1A
50 ns
DO-41
UF4002
100 V
1A
50 ns
DO-41
EGP20B
100 V
2A
50 ns
DO-15
EGP30B
100 V
3A
50 ns
DO-210AD
The ripple current of the n-th output capacitor (Co(n)) is
obtained as
I cap ( n )
rms
=
( ID ( n )
rms 2
) – Io ( n)
2
(24)
where Io(n) is the load current of the n-th output and ID(n)rms
is specified in equation (21). The ripple current should be
smaller than the ripple current specification of the capacitor.
The voltage ripple on the n-th output is given by
I
D
I
peak
V R
K
( V o ( n ) + VF ( n ) )
o ( n ) max
ds
RO C ( n ) L ( n )
∆ V o ( n ) = ------------------------ + ---------------------------------------------------------- (25)
C o ( n ) fs
where Co(n) is the capacitance, Rc(n) is the effective series
resistance (ESR) of the n-th output capacitor, KL(n), VRO and
Idspeak are specified in equations (2), (5) and (8) respectively,
Dmax is specified in step-3, Io(n) and Vo(n) are the load current
and output voltage of the n-th output, respectively and VF(n)
is the diode (DR(n)) forward voltage.
Sometimes it is impossible to meet the ripple specification
with a single output capacitor due to the high ESR of the
electrolytic capacitor. Then, additional LC filter stages (post
filter) can be used. When using the post filters, be careful not
to place the corner frequency too low. Too low a corner frequency may make the system unstable or limit the control
bandwidth. It is typical to set the corner frequency of the
post filter at around 1/10~1/5 of the switching frequency.
FES16BT
100 V
16 A
35 ns
TO-220AC
EGP10C
150 V
1A
50 ns
DO-41
EGP20C
150 V
2A
50 ns
DO-15
EGP30C
150 V
3A
50 ns
DO-210AD
FES16CT
150 V
16 A
35 ns
TO-220AC
EGP10D
200 V
1A
50 ns
DO-41
UF4003
200 V
1A
50 ns
DO-41
(11) STEP-11 : Design the RCD snubber.
EGP20D
200 V
2A
50 ns
DO-15
When the power MOSFET is turned off, there is a high voltage spike on the drain due to the transformer leakage inductance. This excessive voltage on the MOSFET may lead to
an avalanche breakdown and eventually failure of FPS.
Therefore, it is necessary to use an additional network to
clamp the voltage.
EGP30D
200 V
3A
50 ns
DO-210AD
FES16DT
200 V
16 A
35 ns
TO-220AC
EGP10F
300 V
1A
50 ns
DO-41
EGP20F
300 V
2A
50 ns
DO-15
EGP30F
300 V
3A
50 ns
DO-210AD
EGP10G
400 V
1A
50 ns
DO-41
UF4004
400 V
1A
50 ns
DO-41
EGP20G
400 V
2A
50 ns
DO-15
EGP30G
400 V
3A
50 ns
DO-210AD
UF4005
600 V
1A
75 ns
DO-41
EGP10J
600 V
1A
50 ns
DO-41
EGP20J
600 V
2A
50 ns
DO-15
EGP30J
600 V
3A
50 ns
DO-210AD
UF4006
800 V
1A
75 ns
TO-41
UF4007
1000 V
1A
75 ns
TO-41
Table 2. Fairchild Diode quick selection table
©2002 Fairchild Semiconductor Corporation
The RCD snubber circuit and MOSFET drain voltage waveform are shown in figure 10 and 11, respectively. The RCD
snubber network absorbs the current in the leakage inductance by turning on the snubber diode (Dsn) once the MOSFET drain voltage exceeds the voltage of node X as depicted
in figure 10. In the analysis of snubber network, it is
assumed that the snubber capacitor is large enough that its
voltage does not change significantly during one switching
cycle.
The first step in designing the snubber circuit is to determine
the snubber capacitor voltage at the minimum input voltage
and full load condition (Vsn). Once Vsn is determined, the
power dissipated in the snubber network at the minimum
input voltage and full load condition is obtained as
7
AN4137
APPLICATION NOTE
From equation (28), the maximum voltage stress on the internal MOSFET is given by
2
V
( V sn )
peak 2
sn
1
- = --- fs L lK ( I ds
) --------------------------P sn = ---------------2
R sn
Vsn – V RO
(26)
peak
where Ids
is specified in equation (8), fs is the FPS
switching frequency, Llk is the leakage inductance, Vsn is the
snubber capacitor voltage at the minimum input voltage and
full load condition, VRO is the reflected output voltage and
Rsn is the snubber resistor. Vsn should be larger than VRO
and it is typical to set Vsn to be 2~2.5 times of VRO. Too
small a Vsn results in a severe loss in the snubber network as
shown in equation (26). The leakage inductance is measured
at the switching frequency on the primary winding with all
other windings shorted.
Then, the snubber resistor with proper rated wattage should
be chosen based on the power loss. The maximum ripple of
the snubber capacitor voltage is obtained as
∆ V sn
V sn1
= ----------------------C sn R sn f s
V ds
(31)
VDC
+
CDC
Rsn
Csn
X
-
VX
Vsn
+
Np
VRO
+
Dsn
FPS
-
Llk
Drain
GND
+
Vds
-
(28)
where fs is the FPS switching frequency, Llk is the primary
side leakage inductance, VRO is the reflected output voltage,
Rsn is the snubber resistor and Ids2 is the peak drain current at
the maximum input voltage and full load condition. When
the converter operates in CCM at the maximum input voltage
and full load condition (refer to equation (12)), the Ids2 of
equation (28) is obtained as
max
max
P in ⋅  V DC
+ V RO
V DC
⋅ VRO
I ds2 = ------------------------------------------------------------------ + ----------------------------------------------------------------------------max
V
⋅ VRO
2L m f s ⋅  V DCmax + V RO
DC
(29)
Figure 10. Circuit diagram of the snubber network
Voltage Margin > 10% of BVdss
BVdss
Effect of stray inductance (5-10V)
Vsn2
When the converter operates in DCM at the maximum input
voltage and full load condition (refer to equation (12)), the
Ids2 of equation (28) is obtained as
2 ⋅ P in
---------------fs ⋅ Lm
+ V sn2
2
V RO + ( VRO ) + 2R sn L lk fs ( I ds2 )
V sn2 = ------------------------------------------------------------------------------------------2
I ds2 =
max
Check if Vdsmax is below 90% of the rated voltage of the
MOSFET (BVdss) as shown in figure 11. The voltage rating
of the snubber diode should be higher than BVdss. Usually,
an ultra fast diode with 1A current rating is used for the
snubber network.
In the snubber design in this section, neither the lossy discharge of the inductor nor stray capacitance is considered. In
the actual converter, the loss in the snubber network is less
than the designed value due to this effects.
The snubber capacitor voltage (Vsn) of equation (26) is for
the minimum input voltage and full load condition. When the
converter is designed to operate in CCM, the peak drain current together with the snubber capacitor voltage decrease as
the input voltage increases. The snubber capacitor voltage
under maximum input voltage and full load condition is
obtained as
2
= V DC
where VDCmax is specified in equation (4).
(27)
where fs is the FPS switching frequency. In general, 5~10%
ripple is reasonable.
max
VDC max
(30)
where Pin, VDCmax, VRO and Lm are specified in equations
(1), (4), (5) and (7), respectively, and fs is the FPS switching
frequency.
VRO
0V
Figure 11. MOSFET drain voltage and snubber capacitor
voltage
©2003 Fairchild Semiconductor Corporation
8
APPLICATION NOTE
AN4137
2
(12) STEP-12 : Design the feed back loop.
Since most FPS devices employ current mode control as
shown in figure 12, the feedback loop can be simply implemented with a one-pole and one-zero compensation circuit.
In the feedback circuit analysis, it is assumed that the current
transfer ratio (CTR) of the opto coupler is 100%.
The current control factor of FPS, K is defined as
I pk
Iover
K = --------- = ----------------V FB
VFBsat
(32)
where Ipk is the peak drain current and VFB is the feedback
voltage, respectively for a given operating condition, Iover is
the current limit of the FPS and VFBsat is the feedback saturation voltage, which is typically 2.5V.
In order to express the small signal AC transfer functions,
the small signal variations of feedback voltage (vFB) and
controlled output voltage (vo1) are introduced as vˆFB and vˆo1.
vo1'
FPS
vFB
RB
CB
RD
vo1
ibias
Rbias
iD
1:1
CF
RF
R1
KA431
R2
RL ( 1 – D )
1
(1 + D)
- and w p = ------------------w z = -------------------- , w rz = ---------------------------------------2
R c1 C o1
R L C o1
DL m ( N s1 ⁄ N p )
where Lm is specified in equation (7), D is the duty cycle of
the FPS, Co1 is the reference output capacitor and RC1 is the
ESR of Co1.
When the converter has more than one output, the low frequency control-to-output transfer function is proportional to
the parallel combination of all load resistance, adjusted by
the square of the turns ratio. Therefore, the effective load
resistance is used in equation (33) instead of the actual load
resistance of Vo1.
Notice that there is a right half plane (RHP) zero (wrz) in the
control-to-output transfer function of equation (33). Because
the RHP zero reduces the phase by 90 degrees, the crossover
frequency should be placed below the RHP zero.
Figure 13 shows the variation of a CCM flyback converter
control-to-output transfer function for different input voltages. This figure shows the system poles and zeros together
with the DC gain change for different input voltages. The
gain is highest at the high input voltage condition and the
RHP zero is lowest at the low input voltage condition.
Figure 14 shows the variation of a CCM flyback converter
control-to-output transfer function for different loads. This
figure shows that the low frequency gain does not change for
different loads and the RHP zero is lowest at the full load
condition.
For DCM operation, the control-to-output transfer function
of the flyback converter using current mode control is given
by
Vo1 ( 1 + s ⁄ w z )
vˆ o1
G vc = -------- = ---------- ⋅ ---------------------------VFB ( 1 + s ⁄ w p )
vˆ FB
Ipk
MOSFET
current
where
(34)
1
wz = -------------------- , w p = 2 ⁄ R L C o1
R c1 C o1
Figure 12. Control Block Diagram
For CCM operation, the control-to-output transfer function
of the flyback converter using current mode control is given
by
G vc
vˆ o1
= -------vˆ
FB
K ⋅ R L V DC ( N p ⁄ N s1 ) ( 1 + s ⁄ w z ) ( 1 – s ⁄ w rz )
= ----------------------------------------------------- ⋅ ---------------------------------------------------------1 + s ⁄ wp
2V RO + v DC
( 33 )
where VDC is the DC input voltage, RL is the effective total
load resistance of the controlled output, defined as Vo12/Po,
Np and Ns1 are specified in step-7, VRO is specified in equation (5), Vo1 is the reference output voltage, Po is specified in
step-1 and K is specified in equation (32). The pole and zeros
of equation (33) are defined as
Vo1 is the reference output voltage, VFB is the feedback voltage for a given condition, RL is the effective total resistance
of the controlled output, Co1 is the controlled output capacitance and Rc1 is the ESR of Co1.
Figure 15 shows the variation of the control-to-output transfer function of a flyback converter in DCM for different
loads. Contrary to the flyback converter in CCM, there is no
RHP zero and the DC gain does not change as the input voltage varies. As can be seen, the overall gain except for the DC
gain is highest at the full load condition.
The feedback compensation network transfer function of figure 12 is obtained as
©2003 Fairchild Semiconductor Corporation
9
AN4137
APPLICATION NOTE
ˆ
w i 1 + s ⁄ w zc
v FB
-------- = - ----- ⋅ --------------------------ˆ
s 1 + 1 ⁄ w pc
v o1
(35)
RB
1
1
- , w zc = --------------------------------- , w pc = --------------where w i = ---------------------R1 RD CF
( R F + R 1 )C F
RB CB
and RB is the internal feedback bias resistor of FPS, which is
typically 2.8kΩ and R1, RD, RF, CF and CB are shown in figure 12.
40 dB
fp
20 dB
fp
High input voltage
0 dB
Low input voltage
fz
-20 dB
frz
fz
frz
When the input voltage and the load current vary over a wide
range, it is not easy to determine the worst case for the feedback loop design. The gain together with zeros and poles
vary according to the operating condition. Moreover, even
though the converter is designed to operate in CCM or at the
boundary of DCM and CCM in the minimum input voltage
and full load condition, the converter enters into DCM
changing the system transfer functions as the load current
decreases and/or input voltage increases.
One simple and practical way to this problem is designing
the feedback loop for low input voltage and full load condition with enough phase and gain margin. When the converter
operates in CCM, the RHP zero is lowest in low input voltage and full load condition. The gain increases only about
6dB as the operating condition is changed from the lowest
input voltage to the highest input voltage condition under
universal input condition. When the operating mode changes
from CCM to DCM, the RHP zero disappears making the
system stable. Therefore, by designing the feedback loop
with more than 45 degrees phase margin in low input voltage
and full load condition, the stability over all the operating
ranges can be guaranteed.
-40 dB
The procedure to design the feedback loop is as follows
1Hz
10Hz
100Hz
1kHz
10kHz
100kHz
Figure 13. CCM flyback converter control-to output transfer function variation for different input voltages
40 dB
fp
Light load
20 dB
fp
0 dB
Heavy load
-20 dB
fz
frz
f rz
-40 dB
1Hz
10Hz
100Hz
1kHz
10kHz
100kHz
Figure 14. CCM flyback converter control-to output transfer function variation for different loads
(a) Determine the crossover frequency (fc). For CCM mode
flyback, set fc below 1/3 of right half plane (RHP) zero to
minimize the effect of the RHP zero. For DCM mode fc can
be placed at a higher frequency, since there is no RHP zero.
(b) When an additional LC filter is employed, the crossover
frequency should be placed below 1/3 of the corner frequency of the LC filter, since it introduces a -180 degrees
phase drop. Never place the crossover frequency beyond the
corner frequency of the LC filter. If the crossover frequency
is too close to the corner frequency, the controller should be
designed to have a phase margin greater than 90 degrees
when ignoring the effect of the post filter.
(c) Determine the DC gain of the compensator (wi/wzc) to
cancel the control-to-output gain at fc.
(d) Place a compensator zero (fzc) around fc/3.
(e) Place a compensator pole (fpc) above 3fc.
40 dB
Loop gain T
40 dB
fp
fp
20 dB
fzc
20 dB
Heavy load
0 dB
0 dB
-20 dB
Control to output
fz
Light load
Compensator
fpc
fp
fc
frz
-20 dB
fz
-40 dB
1Hz
fz
10Hz
100Hz
1kHz
10kHz
100kHz
Figure 15. DCM flyback converter control-to output transfer function variation for different loads
-40 dB
1Hz
10Hz
100Hz
1kHz
10kHz
100kHz
Figure 16. Compensator design
©2003 Fairchild Semiconductor Corporation
10
APPLICATION NOTE
AN4137
When determining the feedback circuit component, there are
some restrictions as follows.
(a) The voltage divider network of R1 and R2 should be
designed to provide 2.5V to the reference pin of the KA431.
The relationship between R1 and R2 is given as
2.5 ⋅ R 1
R 2 = ----------------------Vo1 – 2.5
ing the startup. While, too large a capacitor may increase the
startup time.
(b) Vcc resistor (Ra) : The typical value for Ra is 5-20Ω. In
the case of multiple outputs flyback converter, the voltage of
the lightly loaded output such as Vcc varies as the load currents of other outputs change due to the imperfect coupling
of the transformer. Ra reduces the sensitivity of Vcc to other
outputs and improves the regulations of Vcc.
(36)
where Vo1 is the reference output voltage.
(b) The capacitor connected to feedback pin (CB) is related to
the shutdown delay time in an overload condition by
T delay = ( V SD – 2.5 ) ⋅ C B ⁄ I delay
(37
where VSD is the shutdown feedback voltage and Idelay is the
shutdown delay current. These values are given in the data
sheet. In general, a 10 ~ 50 ms delay time is typical for most
applications. Because CB also determines the high frequency
pole (wpc) of the compensator transfer function as shown in
equation (36), too large a CB can limit the control bandwidth
by placing wpc at too low a frequency. Typical value for CB is
10-50nF.
(c) The resistors Rbias and RD used together with the optocoupler H11A817A and the shunt regulator KA431 should
be designed to provide proper operating current for the
KA431 and to guarantee the full swing of the feedback voltage for the FPS device chosen. In general, the minimum
cathode voltage and current for the KA431 are 2.5V and
1mA, respectively. Therefore, Rbias and RD should be
designed to satisfy the following conditions.
V o1 – V OP – 2.5
----------------------------------------- > I FB
RD
V OP
------------- > 1mA
R bias
(38)
(39)
where Vo1 is the reference output voltage, VOP is opto-diode
forward voltage drop, which is typically 1V and IFB is the
feedback current of FPS, which is typically 1mA. For example, Rbias< 1kΩ and RD < 1.5kΩ for Vo1=5V.
Miscellaneous
(a) Vcc capacitor (Ca) : The typical value for Ca is 10-50uF,
which is enough for most application. A smaller capacitor
than this may result in an under voltage lockout of FPS dur©2003 Fairchild Semiconductor Corporation
11
AN4137
APPLICATION NOTE
- Summary of symbols Aw
Ae
Bsat
Co(n)
Dmax
Eff
fL
fs
Idspeak
Idsrms
Ids2
Iover
Isec(n)rms
ID(n)rms
Icap(n)rms
Io(n)
KL(n)
KRF
Lm
Llk
Losssn
Npmin
Np
Ns1
Ns(n)
Po
Pin
Rc(n)
Rsn
RL
Vlinemin
Vlinemax
VDCmin
VDCmax
Vdsnom
Vo1
VF1
Vcc*
VFa
VD(n)
∆Vo(n)
VRO
Vsn
Vsn2
∆Vsn
Vdsmax
: Winding window area of the core in mm2
: Cross sectional area of the core in mm2
: Saturation flux density in tesla.
: Output capacitor of the n-th output
: Maximum duty cycle ratio
: Estimated efficiency
: Line frequency
: Switching frequency of FPS
: Maximum peak current of MOSFET
: RMS current of MOSFET
: Maximum peak drain current at the maximum input voltage condition.
: FPS current limit level.
: RMS current of the secondary winding for the n-th output
: Maximum rms current of the rectifier diode for the n-th output
: RMS Ripple current of the output capacitor for the n-th output
: Output load current for the n-th output
: Load occupying factor for the n-th output
: Current ripple factor
: Transformer primary side inductance
: Transformer primary side leakage inductance
: Maximum power loss of the snubber network in normal operation
: The minimum number of turns for the transformer primary side to avoid saturation
: Number of turns for primary side
: Number of turns for the reference output
: Number of turns for the n-th output
: Maximum output power
: Maximum input power
: Effective series resistance (ESR) of the n-th output capacitor.
: Snubber resistor
: Effective total output load resistor of the controlled output
: Minimum line voltage
: Maximum line voltage
: Minimum DC link voltage
: Maximum DC line voltage
: Maximum nominal MOSFET voltage
: Output voltage of the reference output.
: Diode forward voltage drop of the reference output.
: Nominal voltage for Vcc
: Diode forward voltage drop of Vcc winding
: Maximum voltage of the rectifier diode for n-th output
: Output voltage ripple for the n-th output
: Output voltage reflected to the primary
: Snubber capacitor voltage under minimum input voltage and full load condition
: Snubber capacitor voltage under maximum input voltage and full load condition
: Maximum Snubber capacitor voltage ripple
: Maximum voltage stress of the MOSFET
©2003 Fairchild Semiconductor Corporation
12
APPLICATION NOTE
AN4137
Appendix : Design example using FPS Design Assistant
Application
Set-top Box
Output
Input voltage Output voltage (Max Current)
Power
Device
FSDM07652R
47W
± 5%
± 5%
± 5%
±5%
±5%
3.3V (2A)
5V (2A)
12V (1.5A)
18V (0.5A)
33V (0.1A)
85V-265VAC
1. Define the system specifications
Minimum Line voltage (Vlinemin)
85 V.rms
Maximum Line voltage (Vlinemax)
265 V.rms
Line frequency (fL)
Ripple
spec
60 Hz
Vo(n)
1st output for feedback
2nd output
3rd output
4th output
5th output
6th output
Maximum output power (Po) =
Estimated efficiency (Eff)
3.3
5
12
18
33
V
V
V
V
V
V
46.9 W
70 %
Io(n)
2.00
2.00
1.50
0.50
0.10
Po(n)
A
A
A
A
A
A
7
10
18
9
3
0
W
W
W
W
W
W
KL(n)
14
21
38
19
7
0
%
%
%
%
%
%
67.0 W
Maximum input power (Pin) =
☞ The estimated efficiency (Eff) is set to be 0.7 considering the low voltage outputs (3.3V and 5V)
2. Determine DC link capacitor and DC link voltage range
DC link capacitor (CDC)
150 uF
min
min
Minimum DC link voltage (VDC ) =
92 V
(VDCmax)=
375 V
Maximum DC link voltage
☞ Since the input power is 67 W, the DC link capacitor is set to be 150uF by 2uF/Watt.
3. Determine Maximum duty ratio (Dmax)
Maximum duty ratio (Dmax)
(Vdsnom)
Max nominal MOSFET voltage
=
Output voltage reflected to primary (VRO)=
0.48
460 V
85 V
☞ Dmax is set to be 0.48 so that Vdsnom would be about 70% of BVdss (650V×
×0.7=455V)
4. Determine transformer primary inductance (Lm)
Switching frequency of FPS (fs)
66 kHz
0.33
Ripple factor (KRF)
671 uH
Primary side inductance (Lm) =
Maximum peak drain current (Idspeak) =
RMS drain current
rms
(Ids )
=
Maximum DC link voltage in CCM (VDCCCM)
©2002 Fairchild Semiconductor Corporation
K RF = 1 ( DCM )
K RF < 1 (CCM )
∆I
I EDC
2.01 A
1.07 A
##
K RF =
∆I
2 I EDC
375 V
13
AN4137
APPLICATION NOTE
5. Choose the proper FPS considering the input power and current limit
2.50 A
Typical current limit of FPS (Iover)
Minimum Iover considering tolerance of 12%
2.20 A
> 2.01
A
->O.K.
☞ Since the maximum peak drain current (Idspeak) is 2.0A, FSDM07652R is chosen, whose current limit
level (Iover) is 2.5A. The current limit tolerance (12%) is considered.
6. Determine the proper core and the minimum primary turns
Saturation flux density (Bsat)
0.35 T
Cross sectional area of core (Ae)
Minimum primary turns
109.4 mm
(Npmin)=
2
43.8 T
☞ Ferrite core EER3530 is chosen (Ae=109.4
Aw=210mm2), which is a little bit larger than the core
recommended in table 1 to provide enough winding window area.
mm2,
7. Determine the number of turns for each output
Vo(n)
VF(n)
# of turns
1.2 V
6.9 =>
7T
0.5 V
2 =>
2T
0.5 V
2.9 =>
3T
1.2 V
6.9 =>
7T
1.2 V
10 T
10.1 =>
1.2 V
18.0 =>
18 T
0V
0.0 =>
0T
Primary turns (Np)=
45 T
--->enough turns
2130 nH/T2
0.34631 mm
Vcc (Use Vcc start voltage)
1st output for feedback
2nd output
3rd output
4th output
5th output
6th output
VF : Forward voltage drop of rectifier diode
12
3.3
5
12
18
33
0
Ungapped AL value (AL)
Gap length (G) ; center pole gap =
V
V
V
V
V
V
V
☞ In general, the optimum turn ratio between 5V and 3.3V is 3/2, considering the diode forward voltage
drop.
8. Determine the wire diameter for each winding
Primary winding
Vcc winding
1st output winding
2nd output winding
3rd output winding
4th output winding
5th output winding
6th output winding
Copper area (Ac) =
Fill factor (KF)
Required window area (Awr)
Diameter
0.5
0.3
0.4
0.4
0.4
0.4
0.4
Parallel
1T
mm
mm
2T
mm
4T
mm
4T
mm
3T
mm
2T
mm
1T
mm
T
2
19.70 mm
0.15
2
rms
ID(n)rms
1.1
0.1
3.5
3.7
2.8
0.9
0.2
#####
A
A
A
A
A
A
A
A
(A/mm2)
5.44
0.71
6.97
7.30
7.30
3.76
1.55
#####
2
131.33 mm
☞ Since the windings for 3.3V and 5V are short with small number of turns, relatively large current
densities (> 5A/mm2) are allowed. The fill factor is set to be 0.15 due to multiple outputs.
14
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4137
9. Choose the rectifier diode in the secondary side
ID(n)rms
0.10
3.50
3.67
2.75
0.95
0.19
#####
VD(n)
Vcc diode
1st output diode
2nd output diode
3rd output diode
4th output diode
5th output diode
6th output diode
70
20
29
70
103
184
0
V
V
V
V
V
V
V
A
A
A
A
A
A
A
Vcc winding
UF4003 (200V /1A, VF=1V)
Ultra Fast Recovery Diode
1st output (3.3V)
SB540 (40V/5A, VF=0.55V) × 2
Schottky Barrier Diode
2nd output (5V)
SB560 (60V/5A, VF=0.67V) × 2
Schottky Barrier Diode
3rd output (12V)
EGP30D (200V/3A, VF=0.95V)
Ultra Fast Recovery Diode
4st output (18V)
EGP20D (200V/2A, VF=0.95V)
Ultra Fast Recovery Diode
5st output (30V)
UF4004 (400V /1A, VF=1V)
Ultra Fast Recovery Diode
10. Determine the output capacitor
RC(n)
Co(n)
1st output capacitor
2nd output capacitor
3rd output capacitor
4th output capacitor
5th output capacitor
6th output capacitor
2000
2000
330
470
47
uF
uF
uF
uF
uF
uF
100
100
300
300
480
Icap(n)
mΩ
2.9
mΩ
3.1
mΩ
2.3
mΩ
0.8
mΩ
0.2
mΩ #####
A
A
A
A
A
A
ΔVo(n)
0.64
0.67
1.53
0.52
0.18
#####
V
V
V
V
V
V
Since the voltage ripples for 3.3V, 5V and 12V exceed the ripple spec of ± 5%, additional LC filter stage
should be used for these three outputs. 220uF capacitor together with 2.2uH inductor are used for the post
filter. To attenuate the voltage ripple caused by switching, the corner frequency of the post filter (fo) is set
at about one decade below the switching frequency.
fo =
1
2π L p1C p1
=
10 6
2π 2.2 × 220
11. Design RCD snubber
Primary side leakage inductance (Llk)
= 7.2kHz
4.5 uH
Maximum Voltage of snubber capacitor (Vsn)
Maximum snubber capacitor voltage ripple
Snubber resistor (Rsn)=
Snubber capacitor (Csn)=
Power loss in snubber resistor (Psn)=
190
5
33.1
9.2
1.1
max
(Ids2) =
Peak drain current at VDC
1.75 A
Max Voltage of Csn at
VDCmax
(Vsn2)=
Max Voltage stress of MOSFET
max
(Vds )=
V
%
㏀
0.185
nF
W (In Normal Operation)
172 V
547 V
☞ The snubber capacitor and snubber resistor are chosen as 10nF and 33kΩ
Ω, respectively. The maximum
voltage stress on the MOSFET is designed to be 84% of 650V BVdss voltage of the FSDM07652R. The
actual Vdsmax would be lower than this.
©2002 Fairchild Semiconductor Corporation
15
AN4137
APPLICATION NOTE
12. Design Feedback control loop
Control-to-output
Control-to-output
Control-to-output
Control-to-output
DC gain =
zero (wz) =
RHP zero (wrz)=
pole (wp)=
2
5000 rad/s => fz=
694765 rad/s => frz=
2153 rad/s => fp=
Voltage divider resistor (R1)
5.6 ㏀
Voltage divider resistor (R2)=
Opto coupler diode resistor (RD)
KA431 Bias resistor (Rbias)
Feeback pin capacitor (CB) =
Feedback Capacitor (CF) =
Feedback resistor (RF) =
18
1
1.2
33
47
1.2
Feedback integrator gain (wi) =
Compensator zero (wzc)=
Compensator pole (wpc)=
㏀
㏀
㏀
nF
nF
㏀
796 Hz
110,631 Hz
343 Hz
vo1'
FPS
vFB
CB
vo1
RD
ibias
Rbias
iD
1:1
B
CF
R1
RF
KA431
R2
11398 rad/s => fi=
3129 rad/s => fzc=
10101 rad/s => fpc=
1,815 Hz
498 Hz
1,608 Hz
60
Gain (dB)
40
20
0
10
100
-20
-40
0
10
Phase (degree)
-30
-60
100
16 3.64105
25
3.63
40 3.60099
63 3.53167
100 3.36222
160 2.96516
250 2.20575
400 1000
0.89557
630 -0.6501
1000 -2.0185
1600 -2.9016
2500 -3.3275
4000
-3.5257
frequency
(Hz)
6300 -3.5983
10000 -3.6107
16000 -3.5697
1000
25000 -3.4485
40000 -3.1334
63000
-2.448
100000 -1.0745
41
37
33
29
25
21
18
15
13
11
9
6
3
-1
-5
-9
#
#
#
#
44.7
16 -2 -88.7
Control-to-output
40.9
25 -2
-88
36.8 Compensator
40 -4 -86.8
32.8 T (Closed
63loop
-6 gain)-85
28.7
100 -9 -82.2
24.4
160 ## -77.9
20.3
250 ## -72.2
15.9
400 ## 100000
-65.2
10000
12.1
630 ## -59.7
8.75
1000 ## -58.3
5.74
1600 ## -62.1
2.74
2500 ## -68.5
-0.8
4000 -8 -75.2
-4.5
6300 -7 -80.2
-8.4
10000 -8 -83.7
-12
16000 ##
-86
10000
100000
-16
25000 ## -87.5
-20
40000 ## -88.4
-23
63000 ##
-89
-26
1E+05 ## -89.4
#
#
#
#
#
#
#
#
#
#
#
#
#
#
#
#
#
#
#
#
-90
-120
-150
-180
frequency (Hz)
☞ The control bandwidth is 4kHz. Since the crossover frequency is too close to the corner frequency of the
post filter (fo=7.2 kHz), the controller is designed to have enough phase margin when ignoring the effect of
the post filter.
16
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4137
Design Summary
• For the FPS, FSDM07652R is chosen. This device has a fixed switching frequency of 66kHz. Startup and soft-start circuits
are implemented inside the device.
• To limit the current, 10 ohms resistors (Ra and Rdamp) are used in series with Da and DR5. These damping resistors improve
the regulations of the very lightly loaded outputs.
Figure 17 shows the final schematic of the flyback converter designed by FPS Design Assistant.
VO5 33V
Rdamp 10
DR5
NS5
Co5
UF4004
47uF/ 50V
VO4 18V
DR4
NS4
EGP20D
470uF/25V
Rsn
CDC
10nF
1kV
Csn
Dsn
Cp3
Co3
220uF/ 25V
330uF/25V
Np
Lp2
DR2
150uF/400V
2.2 uH
EGP30D
NS3
33k
2W
VO3 12V
Lp3
DR3
GBLA06
Co4
VO2 5V
2.2 uH
SB560
NS2
Cp2
Co2
UF4007
220uF/ 10V
1000uF× 2 /10V
6
Vstr
1.5nF/275Vac
CL2
Lp1
1
Vcc
GND
FB
Line Filter
(33mH)
4
2
3
Da
VO1 3.3V
2.2 uH
SB540
Cp1
Co1
NS1
220uF/ 10V
1000uF× 2 /10V
Ca
33uF/35V
CL1
DR1
UF4003
R a 10
FPS
(DM07652R)
CL2
Drain
Na
1k
Rd
Rbias
0.47uF/275Vac
1k
H11A817A
5.6k
R1
RL1
CB
1.5M
NTC
5D-13
H11A817A
1.2k
47nF
RF
CF
33nF
Fuse
KA431
18k
AC line
R2
Figure 17. The final schematic of the flyback converter
©2003 Fairchild Semiconductor Corporation
17
AN4137
APPLICATION NOTE
Experimental Verification
In order to show the validity of the design procedure presented in this paper, the converter of the design example has
been built and tested. All the circuit components are used as
designed in the design example and the measured transformer characteristics are shown in table 3.
Figure 18 shows the FPS drain current and DC link voltage
waveforms at the minimum input voltage and full load condition. As can be seen, the maximum peak drain current
(Idspeak) is 2A and the minimum DC link voltage (VDCmin) is
about 90V. The designed values are 2.01A and 92V, respectively.
Figure 19 shows the FPS drain current and voltage waveforms at the minimum input voltage and full load condition.
As designed, the maximum duty ratio (Dmax) is about 0.5
and the maximum peak drain current (Idspeak) is 2A.
Figure 20 shows the FPS drain current and voltage waveforms at the maximum input voltage and full load condition.
The maximum voltage stress on the MOSFET is about 520V,
which is lower than the designed value (547V). This is
because of the lossy discharge of the inductor or the stray
capacitance. Another reason is that the power conversion
efficiency at the maximum input voltage is higher than the
estimated efficiency used in step-1.
Figure 18. Waveforms of drain current and DC link
voltage at 85Vac and full load condition (time:2ms/div)
As calculated in design step-4, the converter operates at the
boundary between CCM and DCM under the maximum
input voltage and full load condition (The maximum DC link
voltage guaranteeing CCM at full load was obtained as 375V
in design step-4).
Figure 21 shows the current and voltage waveforms of the
first output (3.3V) rectifier diode. The maximum reverse
voltage of this diode was calculated as 20V in step-9 and the
measured value is 23V.
Table 4 shows the line regulation of each output. 3.3V and
5V output shows ±3% and ±4% regulations, respectively.
Figure 22 shows the measured efficiency at the full load condition for different input voltages. The minimum efficiency
is about 73% at the minimum input voltage condition better
than the 70% target efficiency specified in step-1.
Core
EER3530 (ISU ceramics)
Primary side
inductance
682 uH @ 70kHz
Leakage
inductance
Resistance
Figure 19. Waveforms of drain current and voltage
at 85Vac and full load condition (time : 5us/div)
4.5 uH @70kHz with all other
windings shorted.
0.76 Ω
Table 3. The measured transformer characteristics
Figure 20. Waveforms of drain current and voltage
at 265Vac and full load condition (time : 5us/div)
18
©2002 Fairchild Semiconductor Corporation
APPLICATION NOTE
AN4137
Efficiency
0.81
0.80
0.79
0.78
0.77
0.76
0.75
0.74
0.73
0.72
85
115
145
175
205
235
265
Input voltage (Vac)
Figure 22. Measured efficiency
Figure 21. Current and voltage waveforms of the first
output (3.3V) rectifier diode at 265Vac and full load
condition (time : 5us/div)
Input
voltage
Vo1
(3.3V)
Vo2
(5V)
Vo3
(12V)
Vo4
(18V)
Vo5
(33V)
85Vac
3.21 V
5.18 V
12.88 V
19.7 V
35.7 V
-2.7 %
3.6 %
7.3 %
9.4 %
8.2 %
3.21 V
5.14 V
12.77 V
19.4 V
34.6 V
-2.7 %
2.8 %
6.4 %
7.8 %
4.8 %
3.21 V
5.11 V
12.67 V
19.2 V
34.1 V
-2.7 %
2.2 %
5.6 %
6.7 %
3.2 %
3.21 V
5.09 V
12.57 V
19.1 V
33.8 V
-2.7 %
1.8 %
4.8 %
5.9 %
2.4 %
3.21 V
5.08 V
12.52 V
19.0 V
33.6 V
-2.7 %
1.6 %
4.3 %
5.4 %
1.9 %
3.21 V
5.07 V
12.48 V
18.9 V
33.5 V
-2.7 %
1.4 %
4.0 %
5.1 %
1.6 %
3.21 V
5.06 V
12.47 V
18.9 V
33.5 V
-2.7 %
1.2 %
3.9 %
5.1 %
1.4 %
115Vac
145Vac
175Vac
205Vac
235Vac
265Vac
Table 4. Line regulation of each output at full load
condition
©2002 Fairchild Semiconductor Corporation
19
AN4137
APPLICATION NOTE
by Hang-Seok Choi / Ph. D
FPS Application Group / Fairchild Semiconductor
Phone : +82-32-680-1383 Facsimile : +82-32-680-1317
E-mail : [email protected]
DISCLAIMER
FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO ANY
PRODUCTS HEREIN TO IMPROVE RELIABILITY, FUNCTION OR DESIGN. FAIRCHILD DOES NOT ASSUME ANY
LIABILITY ARISING OUT OF THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN; NEITHER
DOES IT CONVEY ANY LICENSE UNDER ITS PATENT RIGHTS, NOR THE RIGHTS OF OTHERS.
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which, (a) are intended for surgical implant into the body,
or (b) support or sustain life, or (c) whose failure to perform
when properly used in accordance with instructions for use
provided in the labeling, can be reasonably expected to
result in significant injury to the user.
2. A critical component is any component of a life support
device or system whose failure to perform can be
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