INTERSIL LMH6505MM

LMH6505
Wideband, Low Power, Linear-in-dB, Variable Gain
Amplifier
General Description
Features
The LMH6505 is a wideband DC coupled voltage controlled
gain stage followed by a high speed current feedback operational amplifier which can directly drive a low impedance load.
The gain adjustment range is 80 dB for up to 10 MHz which
is accomplished by varying the gain control input voltage,
VG.
Maximum gain is set by external components, and the gain
can be reduced all the way to cutoff. Power consumption is
110 mW with a speed of 150 MHz and a gain control bandwidth (BW) of 100 MHz. Output referred DC offset voltage is
less than 55 mV over the entire gain control voltage range.
Device-to-device gain matching is within ±0.5 dB at maximum
gain. Furthermore, gain is tested and guaranteed over a wide
range. The output current feedback op amp allows high frequency large signals (Slew Rate = 1500 V/μs) and can also
drive a heavy load current (60 mA) guaranteed. Near ideal
input characteristics (i.e. low input bias current, low offset, low
pin 3 resistance) enable the device to be easily configured as
an inverting amplifier as well.
To provide ease of use when working with a single supply, the
VG range is set to be from 0V to +2V relative to the ground pin
potential (pin 4). VG input impedance is high in order to ease
drive requirement. In single supply operation, the ground pin
is tied to a "virtual" half supply.
The LMH6505’s gain control is linear in dB for a large portion
of the total gain control range from 0 dB down to −85 dB at
25°C, as shown below. This makes the device suitable for
AGC applications. For linear gain control applications, see the
LMH6503 datasheet.
The LMH6505 is available in either the 8-Pin SOIC or the 8Pin MSOP package. The combination of minimal external
components and small outline packages allows the LMH6505
to be used in space-constrained applications.
VS = ±5V, TA = 25°C, RF = 1 kΩ, RG = 100Ω, RL = 100Ω,
AV = AVMAX = 9.4 V/V, Typical values unless specified.
150 MHz
■ −3 dB BW
100 MHz
■ Gain control BW
80 dB
■ Adjustment range (<10 MHz)
±0.50 dB
■ Gain matching (limit)
7V to 12V
■ Supply voltage range
1500 V/μs
■ Slew rate (inverting)
11 mA
■ Supply current (no load)
±60 mA
■ Linear output current
±2.4V
■ Output voltage swing
4.4 nV/√Hz
■ Input noise voltage
2.6 pA/√Hz
■ Input noise current
−45 dBc
■ THD (20 MHz, RL = 100Ω, VO = 2 VPP)
Applications
■
■
■
■
Variable attenuator
AGC
Voltage controlled filter
Video imaging processing
Typical Application
20171002
20171011
AVMAX = 9.4 V/V
Gain vs. VG
© 2007 National Semiconductor Corporation
201710
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LMH6505 Wideband, Low Power, Linear-in-dB, Variable Gain Amplifier
December 14, 2007
LMH6505
Junction Temperature
Soldering Information:
Infrared or Convection (20 sec)
Wave Soldering (10 sec)
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
ESD Tolerance (Note 6)
Human Body Model
Machine Model
Input Current
Output Current (Note 3)
Supply Voltages (V+ - V−)
Voltage at Input/ Output pins
Storage Temperature Range
Operating Ratings
2000V
200V
±10 mA
120 mA
12.6V
V+ +0.8V, V− −0.8V
−65°C to 150°C
Electrical Characteristics
235°C
260°C
(Note 1)
Supply Voltages (V+ - V−)
Temperature Range (Note 5)
Thermal Resistance:
8 -Pin SOIC
8-Pin MSOP
150°C
7V to 12V
−40°C to +85°C
(θJC)
60
65
(θJA)
165
235
(Note 2)
Unless otherwise specified, all limits are guaranteed for TJ = 25°C, VS = ±5V, AVMAX = 9.4 V/V, RF = 1 kΩ, RG = 100Ω, VIN = ±0.1V,
RL = 100Ω, VG = +2V. Boldface limits apply at the temperature extremes.
Symbol
Parameter
Conditions
Min
(Note 9)
Typ
(Note 8)
Max
(Note 9)
Units
Frequency Domain Response
BW
GF
−3 dB Bandwidth
Gain Flatness
VOUT < 1 VPP
150
VOUT < 4 VPP, AVMAX = 100
38
VOUT < 1 VPP
40
MHz
MHz
0.9V ≤ VG ≤ 2V, ±0.2 dB
Att Range Flat Band (Relative to Max Gain)
Attenuation Range (Note 15)
±0.2 dB Flatness, f < 30 MHz
26
±0.1 dB Flatness, f < 30 MHz
9.5
BW
Control
Gain control Bandwidth
VG = 1V (Note 4)
100
MHz
CT (dB)
Feed-through
VG = 0V, 30 MHz
(Output/Input)
−51
dB
GR
Gain Adjustment Range
f < 10 MHz
80
f < 30 MHz
71
0.5V Step
2.1
ns
10
%
dB
dB
Time Domain Response
tr, tf
Rise and Fall Time
OS %
Overshoot
SR
Slew Rate (Note 7)
Non Inverting
900
Inverting
1500
2VPP, 20 MHz
−47
V/μs
Distortion & Noise Performance
HD2
2nd Harmonic Distortion
HD3
3rd
Harmonic Distortion
–61
THD
Total Harmonic Distortion
−45
En tot
Total Equivalent Input Noise
f > 1 MHz, RSOURCE = 50Ω
IN
Input Noise Current
DG
Differential Gain
DP
Differential Phase
dBc
4.4
nV/
f > 1 MHz
2.6
pA/
f = 4.43 MHz, RL = 100Ω
0.30
%
0.15
deg
DC & Miscellaneous Performance
GACCU
G Match
K
Gain Accuracy
(See Application Information)
VG = 2.0V
0
±0.50
+0.1/−0.53
+4.3/−3.9
Gain Matching
(See Application Information)
VG = 2.0V
—
±0.50
0.8V < VG < 2V
—
+4.2/−4.0
0.940
0.990
1.04
0.8V < VG < 2V
Gain Multiplier
(See Application Information)
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0.890
0.830
2
dB
dB
V/V
VIN NL
Parameter
Input Voltage Range
VIN L
Conditions
Min
(Note 9)
Typ
(Note 8)
RG = 100Ω
±0.60
±0.50
±0.74
V
±6.0
±5.0
±7.4
mA
RG Open
Max
(Note 9)
Units
±3
I RG_MAX
RG Current
Pin 3
IBIAS
Bias Current
Pin 2 (Note 10)
−0.6
TC IBIAS
Bias Current Drift
Pin 2 (Note 11)
1.28
nA/°C
RIN
Input Resistance
Pin 2
7
MΩ
CIN
Input Capacitance
Pin 2
2.8
pF
IVG
VG Bias Current
Pin 1, VG = 2V (Note 10)
0.9
µA
TC IVG
VG Bias Drift
Pin 1 (Note 11)
10
pA/°C
R VG
VG Input Resistance
Pin 1
25
MΩ
C VG
VG Input Capacitance
Pin 1
2.8
pF
VOUT L
Output Voltage Range
RL = 100Ω
VOUT NL
±2.1
±1.9
RL = Open
−2.5
−2.6
µA
±2.4
V
±3.1
0.12
Ω
±80
mA
ROUT
Output Impedance
DC
IOUT
Output Current
VOUT = ±4V from Rails
VO OFFSET
Output Offset Voltage
0V < VG < 2V
+PSRR
+Power Supply Rejection Ratio
(Note 12)
Input Referred, 1V change, VG =
2.2V
–65
–72
−PSRR
−Power Supply Rejection Ratio
(Note 12)
Input Referred, 1V change, VG =
2.2V
–65
–75
IS
Supply Current
No Load
9.5
7.5
11
±60
±40
±10
±55
±70
mV
dB
dB
14
16
mA
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is
intended to be functional, but specific performance is not guaranteed. For guaranteed specifications, see the Electrical Characteristics.
Note 2: Electrical Table values apply only for factory testing conditions at the temperature indicated. Factory testing conditions result in very limited self-heating
of the device such that TJ = TA. No guarantee of parametric performance is indicated in the Electrical Tables under conditions of internal self-heating where TJ >
TA.
Note 3: The maximum output current (IOUT) is determined by device power dissipation limitations or value specified, whichever is lower.
Note 4: Gain control frequency response schematic:
20171016
Note 5: The maximum power dissipation is a function of TJ(MAX), θJA. The maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) –
TA)/ θJA. All numbers apply for packages soldered directly onto a PC Board.
Note 6: Human Body Model, applicable std. MIL-STD-883, Method 3015.7. Machine Model, applicable std. JESD22-A115-A (ESD MM std. of JEDEC)
Field-Induced Charge-Device Model, applicable std. JESD22-C101-C (ESD FICDM std. of JEDEC).
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LMH6505
Symbol
LMH6505
Note 7: Slew rate is the average of the rising and falling slew rates.
Note 8: Typical values represent the most likely parametric norm as determined at the time of characterization. Actual typical values may vary over time and will
also depend on the application and configuration. The typical values are not tested and are not guaranteed on shipped production material.
Note 9: Limits are 100% production tested at 25°C. Limits over the operating temperature range are guaranteed through correlations using the Statistical Quality
Control (SQC) method.
Note 10: Positive current corresponds to current flowing into the device.
Note 11: Drift is determined by dividing the change in parameter distribution at temperature extremes by the total temperature change.
Note 12: +PSRR definition: [|ΔVOUT/ΔV+| / AV], −PSRR definition: [|ΔVOUT/ΔV−| / AV] with 0.1V input voltage. ΔVOUT is the change in output voltage with offset
shift subtracted out.
Note 13: Gain/Phase normalized to low frequency value at 25°C.
Note 14: Gain/Phase normalized to low frequency value at each setting.
Note 15: Flat Band Attenuation (Relative To Max Gain) Range Definition: Specified as the attenuation range from maximum which allows gain flatness specified
(either ±0.2 dB or ±0.1 dB), relative to AVMAX gain. For example, for f < 30 MHz, here are the Flat Band Attenuation ranges:
±0.2 dB: 19.7 dB down to -6.3 dB = 26 dB range
±0.1 dB: 19.7 dB down to 10.2 dB = 9.5 dB range
Connection Diagram
8-Pin SOIC/MSOP
20171001
Top View
Ordering Information
Package
8-Pin SOIC
8-Pin MSOP
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Part Number
LMH6505MA
LMH6505MAX
LMH6505MM
LMH6505MMX
Package Marking
Transport Media
95 Units/Rail
LMH6505MA
2.5k Units Tape and Reel
1k Units Tape and Reel
AZ2A
3.5k Units Tape and Reel
4
NSC Drawing
M08A
MUA08A
LMH6505
Typical Performance Characteristics
Unless otherwise specified: VS = ±5V, TA = 25°C,
VG = VGMAX, RF = 1 kΩ, RG = 100Ω, VIN = 0.1V, input terminated in 50Ω. RL = 100Ω, Typical values.
Frequency Response Over Temperature
Frequency Response for Various VG
20171003
20171004
Frequency Response (AVMAX = 2)
Inverting Frequency Response
20171044
20171046
Frequency Response for Various VG (AVMAX = 100) (Large
Signal)
Frequency Response for Various Amplitudes
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20171045
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LMH6505
Gain Control Frequency Response
IS vs. VS
20171033
20171021
IS vs. VS
Input Bias Current vs. VS
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20171020
PSRR
AVMAX vs. Supply Voltage
20171034
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20171023
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Gain Variation Over entire Temp Range vs. VG
20171012
20171041
IRG vs. VIN
Gain vs. VG
20171011
20171018
Output Offset Voltage vs. VG (Typical Unit #1)
Output Offset Voltage vs. VG (Typical Unit #2)
20171030
20171025
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LMH6505
Feed through Isolation for Various AVMAX
LMH6505
Output Offset Voltage vs. VG (Typical Unit #3)
Distribution of Output Offset Voltage
20171061
20171028
Output Noise Density vs. Frequency
Output Noise Density vs. Frequency
20171008
20171038
Output Noise Density vs. Frequency
Input Referred Noise Density vs. Frequency
20171037
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20171036
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Output Voltage vs. Output Current (Sourcing)
20171065
20171031
Distortion vs. Frequency
HD vs. POUT
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20171043
THD vs. POUT
THD vs. POUT
20171010
20171009
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LMH6505
Output Voltage vs. Output Current (Sinking)
LMH6505
THD vs. Gain
THD vs. Gain
20171039
20171040
Differential Gain & Phase
VG Bias Current vs. VG
20171035
20171014
Step Response Plot
Step Response Plot
20171015
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20171017
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LMH6505
Gain vs. VG Step
20171032
SETTING THE LMH6505 MAXIMUM GAIN
Application Information
GENERAL DESCRIPTION
The key features of the LMH6505 are:
• Low power
• Broad voltage controlled gain and attenuation range (from
AVMAX down to complete cutoff)
• Bandwidth independent, resistor programmable gain
range (RG)
• Broad signal and gain control bandwidths
• Frequency response may be adjusted with RF
• High impedance signal and gain control inputs
The LMH6505 combines a closed loop input buffer (“X1”
Block in Figure 1), a voltage controlled variable gain cell
(“MULT” Block) and an output amplifier (“CFA” Block). The
input buffer is a transconductance stage whose gain is set by
the gain setting resistor, RG. The output amplifier is a current
feedback op amp and is configured as a transimpedance
stage whose gain is set by, and is equal to, the feedback resistor, RF. The maximum gain, AVMAX, of the LMH6505 is
defined by the ratio: K · RF/RG where “K” is the gain multiplier
with a nominal value of 0.940. As the gain control input (VG)
changes over its 0 to 2V range, the gain is adjusted over a
range of about 80 dB relative to the maximum set gain.
(1)
Although the LMH6505 is specified at AVMAX = 9.4 V/V, the
recommended AVMAX varies between 2 and 100. Higher gains
are possible but usually impractical due to output offsets,
noise and distortion. When varying AVMAX several tradeoffs
are made:
RG: determines the input voltage range
RF: determines overall bandwidth
The amount of current which the input buffer can source/sink
into RG is limited and is given in the IRG_MAX specification. This
sets the maximum input voltage:
(2)
As the IRG_MAX limit is approached with increasing the input
voltage or with the lowering of RG, the device's harmonic distortion will increase. Changes in RF will have a dramatic effect
on the small signal bandwidth. The output amplifier of the
LMH6505 is a current feedback amplifier (CFA) and its bandwidth is determined by RF. As with any CFA, doubling the
feedback resistor will roughly cut the bandwidth of the device
in half. For more about CFAs, see the basic tutorial, OA-20,
“Current Feedback Myths Debunked,” or a more rigorous
analysis, OA-13, “Current Feedback Amplifier Loop Gain
Analysis and Performance Enhancements.”
OTHER CONFIGURATIONS
1) Single Supply Operation
The LMH6505 can be configured for use in a single supply
environment. Doing so requires the following:
a) Bias pin 4 and RG to a “virtual half supply” somewhere
close to the middle of V+ and V− range. The other end of
RG is tied to pin 3. The “virtual half supply” needs to be
capable of sinking and sourcing the expected current flow
through RG.
b) Ensure that VG can be adjusted from 0V to 2V above the
“virtual half supply”.
c) Bias the input (pin 2) to make sure that it stays within the
range of 2V above V− to 2V below V+. See the Input Voltage Range specification in the Electrical Characteristics
table. This can be accomplished by either DC biasing the
20171047
FIGURE 1. LMH6505 Typical Application and Block
Diagram
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LMH6505
input and AC coupling the input signal, or alternatively, by
direct coupling if the output of the driving stage is also
biased to half supply.
Arranged this way, the LMH6505 will respond to the current
flowing through RG. The gain control relationship will be similar to the split supply arrangement with VG measured with
reference to pin 4. Keep in mind that the circuit described
above will also center the output voltage to the “virtual half
supply voltage.”
2) Arbitrarily Referenced Input Signal
Having a wide input voltage range on the input (pin 2)
(±3V typical), the LMH6505 can be configured to control the
gain on signals which are not referenced to ground (e.g. Half
Supply biased circuits). This node will be called the “reference
node”. In such cases, the other end of RG which is the side
not tied to pin 3 can be tied to this reference node so that
RG will “look at” the difference between the signal and this
reference only. Keep in mind that the reference node needs
to source and sink the current flowing through RG.
20171051
FIGURE 2. LMH6505 Gain Accuracy & Gain Matching
Defined
GAIN ACCURACY
Gain accuracy is defined as the actual gain compared against
the theoretical gain at a certain VG, the results of which are
expressed in dB. (See Figure 2).
Theoretical gain is given by:
GAIN PARTITIONING
If high levels of gain are needed, gain partitioning should be
considered:
(3)
Where K = 0.940 (nominal) N = 1.01V & VC = 79 mV at room
temperature
For a VG range, the value specified in the tables represents
the worst case accuracy over the entire range. The "Typical"
value would be the difference between the "Typical Gain" and
the "Theoretical Gain." The "Max" value would be the worst
case difference between the actual gain and the "Theoretical
Gain" for the entire population.
20171052
GAIN MATCHING
As Figure 2 shows, gain matching is the limit on gain variation
at a certain VG, expressed in dB, and is specified as "±Max"
only. There is no "Typical." For a VG range, the value specified
represents the worst case matching over the entire range.
The "Max" value would be the worst case difference between
the actual gain and the typical gain for the entire population.
FIGURE 3. Gain Partitioning
The maximum gain range for this circuit is given by the following equation:
(4)
The LMH6624 is a low noise wideband voltage feedback amplifier. Setting R2 at 909Ω and R1 at 100Ω produces a gain of
20 dB. Setting RF at 1000Ω as recommended and RG at
50Ω, produces a gain of about 26 dB in the LMH6505. The
total gain of this circuit is therefore approximately 46 dB. It is
important to understand that when partitioning to obtain high
levels of gain, very small signal levels will drive the amplifiers
to full scale output. For example, with 46 dB of gain, a 20 mV
signal at the input will drive the output of the LMH6624 to
200 mV and the output of the LMH6505 to 4V. Accordingly,
the designer must carefully consider the contributions of each
stage to the overall characteristics. Through gain partitioning
the designer is provided with an opportunity to optimize the
frequency response, noise, distortion, settling time, and loading effects of each amplifier to achieve improved overall performance.
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AVOIDING OVERDRIVE OF THE LMH6505 GAIN
CONTROL INPUT
There is an additional requirement for the LMH6505 Gain
Control Input (VG): VG must not exceed +2.3V (with ±5V supplies). The gain control circuitry may saturate and the gain
may actually be reduced. In applications where VG is being
driven from a DAC, this can easily be addressed in the software. If there is a linear loop driving VG, such as an AGC loop,
other methods of limiting the input voltage should be implemented. One simple solution is to place a 2.2:1 resistive
divider on the VG input. If the device driving this divider is operating off of ±5V supplies as well, its output will not exceed
5V and through the divider VG can not exceed 2.3V.
LMH6505 GAIN CONTROL FUNCTION
In the plot, Gain vs. VG, we can see the gain as a function of
the control voltage. The “Gain (V/V)” plot, sometimes referred
to as the S-curve, is the linear (V/V) gain. This is a hyperbolic
tangent relationship and is given by Equation 3. The “Gain
(dB)” plots the gain in dB and is linear over a wide range of
gains. Because of this, the LMH6505 gain control is referred
to as “linear-in-dB.”
For applications where the LMH6505 will be used at the heart
of a closed loop AGC circuit, the S-curve control characteristic
provides a broad linear (in dB) control range with soft limiting
at the highest gains where large changes in control voltage
result in small changes in gain. For applications requiring a
fully linear (in dB) control characteristic, use the LMH6505 at
half gain and below (VG ≤ 1V).
IMPROVING THE LMH6505 LARGE SIGNAL
PERFORMANCE
Figure 5 illustrates an inverting gain scheme for the
LMH6505.
GAIN STABILITY
The LMH6505 architecture allows complete attenuation of the
output signal from full gain to complete cutoff. This is achieved
by having the gain control signal VG “throttle” the signal which
gets through to the final stage and which results in the output
signal. As a consequence, the RG pin's (pin 3) average current
(DC current) influences the operating point of this “throttle”
circuit and affects the LMH6505's gain slightly. Figure 4 below, shows this effect as a function of the gain set by VG.
20171054
FIGURE 5. Inverting Amplifier
The input signal is applied through the RG resistor. The VIN
pin should be grounded through a 25Ω resistor. The maximum gain range of this configuration is given in the following
equation:
(5)
The inverting slew rate of the LMH6505 is much higher than
that of the non-inverting slew rate. This ≈ 2X performance
improvement comes about because in the non-inverting configuration the slew rate of the overall amplifier is limited by the
input buffer. In the inverting circuit, the input buffer remains at
a fixed voltage and does not affect slew rate.
20171066
FIGURE 4. LMH6505 Gain Variation over RG DC Current
Capability vs. Gain
TRANSMISSION LINE MATCHING
One method for matching the characteristic impedance of a
transmission line is to place the appropriate resistor at the
input or output of the amplifier. Figure 6 shows a typical circuit
configuration for matching transmission lines.
This plot shows the expected gain variation for the maximum
RG DC current capability (±4.5 mA). For example, with gain
(AV) set to −60 dB, if the RG pin DC current is increased to 4.5
mA sourcing, one would expect to see the gain increase by
about 3 dB (to −57 dB). Conversely, 4.5 mA DC sinking cur13
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LMH6505
rent through RG would increase gain by 1.75 dB (to −58.25
dB). As you can see from Figure 4 above, the effect is most
pronounced with reduced gain and is limited to less than 3.75
dB variation maximum.
If the application is expected to experience RG DC current
variation and the LMH6505 gain variation is beyond acceptable limits, please refer to the LMH6502 (Differential Linear
in dB variable gain amplifier) datasheet instead at http://
www.national.com/ds/LM/LMH6502.pdf.
LMH6505 GAIN CONTROL RANGE AND MINIMUM GAIN
Before discussing Gain Control Range, it is important to understand the issues which limit it. The minimum gain of the
LMH6505 is theoretically zero, but in practical circuits it is
limited by the amount of feedthrough, here defined as the gain
when VG = 0V. Capacitive coupling through the board and
package, as well as coupling through the supplies, will determine the amount of feedthrough. Even at DC, the input signal
will not be completely rejected. At high frequencies
feedthrough will get worse because of its capacitive nature.
At frequencies below 10 MHz, the feed through will be less
than −60 dB and therefore, it can be said that with
AVMAX = 20 dB, the gain control range is 80 dB.
LMH6505
20171056
FIGURE 6. Transmission Line Matching
The resistors RS, RI, RO, and RT are equal to the characteristic
impedance, ZO, of the transmission line or cable. Use CO to
match the output transmission line over a greater frequency
range. It compensates for the increase of the op amp’s output
impedance with frequency.
ADJUSTING OFFSETS AND DC LEVEL SHIFTING
Offsets can be broken into two parts: an input-referred term
and an output-referred term. These errors can be trimmed
using the circuit in Figure 7. First set VG to 0V and adjust the
trim pot R4 to null the offset voltage at the output. This will
eliminate the output stage offsets. Next set VG to 2V and adjust the trim pot R1 to null the offset voltage at the output. This
will eliminate the input stage offsets.
MINIMIZING PARASITIC EFFECTS ON SMALL SIGNAL
BANDWIDTH
The best way to minimize parasitic effects is to use surface
mount components and to minimize lead lengths and component distance from the LMH6505. For designs utilizing
through-hole components, specifically axial resistors, resistor
self-capacitance should be considered. For example, the average magnitude of parasitic capacitance of RN55D 1% metal
film resistors is about 0.15 pF with variations of as much as
0.1 pF between lots. Given the LMH6505’s extended bandwidth, these small parasitic reactance variations can cause
measurable frequency response variations in the highest octave. We therefore recommend the use of surface mount
resistors to minimize these parasitic reactance effects.
RECOMMENDATIONS
Here are some recommendations to avoid problems and to
get the best performance:
• Do not place a capacitor across RF. However, an
appropriately chosen series RC combination can be used
to shape the frequency response.
• Keep traces connecting RF separated and as short as
possible.
• Place a small resistor (20-50Ω) between the output and
CL.
• Cut away the ground plane, if any, under RG.
• Keep decoupling capacitors as close as possible to the
LMH6505.
• Connect pin 2 through a minimum resistance of 25Ω.
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20171057
FIGURE 7. Offset Adjust Circuit
DIGITAL GAIN CONTROL
Digitally variable gain control can be easily realized by driving
the LMH6505 gain control input with a digital-to-analog converter (DAC). Figure 8 illustrates such an application. This
circuit employs National Semiconductor’s eight-bit DAC0830,
the LMC8101 MOS input op amp (Rail-to-Rail Input/Output),
and the LMH6505 VGA. With VREF set to 2V, the circuit provides up to 80 dB of gain control in 256 steps with up to 0.05%
full scale resolution. The maximum gain of this circuit is 20
dB.
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20171058
FIGURE 8. Digital Gain Control
USING THE LMH6505 IN AGC APPLICATIONS
In AGC applications, the control loop forces the LMH6505 to
have a fixed output amplitude. The input amplitude will vary
over a wide range and this can be the issue that limits dynamic
range. At high input amplitudes, the distortion due to the input
buffer driving RG may exceed that which is produced by the
output amplifier driving the load. In the plot, THD vs. Gain,
total harmonic distortion (THD) is plotted over a gain range of
nearly 35 dB for a fixed output amplitude of 0.25 VPP in the
specified configuration, RF = 1 kΩ, RG = 100Ω. When the gain
is adjusted to −15 dB (i.e. 35 dB down from AVMAX), the input
AUTOMATIC GAIN CONTROL (AGC)
Fast Response AGC Loop
The AGC circuit shown in Figure 9 will correct a 6 dB input
amplitude step in 100 ns. The circuit includes a two op amp
precision rectifier amplitude detector (U1 and U2), and an integrator (U3) to provide high loop gain at low frequencies. The
output amplitude is set by R9. The following are some suggestions for building fast AGC loops: Precision rectifiers work
best with large output signals. Accuracy is improved by blocking DC offsets, as shown in Figure 9.
20171059
FIGURE 9. Automatic Gain Control Circuit
15
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LMH6505
amplitude would be 1.41 VPP and we can see the distortion is
at its worst at this gain. If the output amplitude of the AGC
were to be raised above 0.25 VPP, the input amplitudes for
gains 40 dB down from AVMAX would be even higher and the
distortion would degrade further. It is for this reason that we
recommend lower output amplitudes if wide gain ranges are
desired. Using a post-amp like the LMH6714/LMH6720/
LMH6722 family or the LMH6702 would be the best way to
preserve dynamic range and yield output amplitudes much
higher than 100 mVPP. Another way of addressing distortion
performance and its limitations on dynamic range, would be
to raise the value of RG. Just like any other high-speed amplifier, by increasing the load resistance, and therefore decreasing the demanded load current, the distortion performance will be improved in most cases. With an increased
RG, RF will also have to be increased to keep the same
AVMAX and this will decrease the overall bandwidth. It may be
possible to insert a series RC combination across RF in order
to counteract the negative effect on BW when a large RF is
used.
LMH6505
Signal frequencies must not reach the gain control port of the
LMH6505, or the output signal will be distorted (modulated by
itself). A fast settling AGC needs additional filtering beyond
the integrator stage to block signal frequencies. This is provided in Figure 9 by a simple R-C filter (R10 and C3); better
distortion performance can be achieved with a more complex
filter. These filters should be scaled with the input signal frequency. Loops with slower response time, which means
longer integration time constants, may not need the R10 –
C3 filter.
Checking the loop stability can be done by monitoring the
VG voltage while applying a step change in input signal amplitude. Changing the input signal amplitude can be easily
done with an arbitrary waveform generator.
The LMH6505 is fully stable when driving a 100Ω load. With
reduced load (e.g. 1k.) there is a possibility of instability at
very high frequencies beyond 400 MHz especially with a capacitive load. When the LMH6505 is connected to a light load
as such, it is recommended to add a snubber network to the
output (e.g. 100Ω and 39 pF in series tied between the
LMH6505 output and ground). CL can also be isolated from
the output by placing a small resistor in series with the output
(pin 6).
Component parasitics also influence high frequency results.
Therefore it is recommended to use metal film resistors such
as RN55D or leadless components such as surface mount
devices. High profile sockets are not recommended.
National Semiconductor suggests the following evaluation
board as a guide for high frequency layout and as an aid in
device testing and characterization:
CIRCUIT LAYOUT CONSIDERATIONS & EVALUATION
BOARDS
A good high frequency PCB layout including ground plane
construction and power supply bypassing close to the package is critical to achieving full performance. The amplifier is
sensitive to stray capacitance to ground at the I- input (pin 7)
so it is best to keep the node trace area small. Shunt capacitance across the feedback resistor should not be used to
compensate for this effect. Capacitance to ground should be
minimized by removing the ground plane from under the body
of RG. Parasitic or load capacitance directly on the output (pin
6) degrades phase margin leading to frequency response
peaking.
www.national.com
Device
Package
LMH6505
SOIC
Evaluation Board
Part Number
LMH730066
The evaluation board can be shipped when a device sample
request is placed with National Semiconductor. Evaluation
board documentation can be found in the LMH6505 product
folder at www.National.com.
16
LMH6505
Physical Dimensions inches (millimeters) unless otherwise noted
8-Pin SOIC
NS Package Number M08A
8-Pin MSOP
NS Package Number MUA08A
17
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LMH6505 Wideband, Low Power, Linear-in-dB, Variable Gain Amplifier
Notes
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