Application Note ICE1HS01G

Application Note, V1.0, 12 August 2009
Application Note
ANPS0031 -ICE1HS01G
Half Bridge LLC Resonant Converter Design
using ICE1HS01G
Power Management & Supply
N e v e r
s t o p
t h i n k i n g .
Published by
Infineon Technologies AG
81726 Munich, Germany
© 2007 Infineon Technologies AG
All Rights Reserved.
Legal Disclaimer
The information given in this document shall in no event be regarded as a guarantee of
conditions or characteristics. With respect to any examples or hints given herein, any typical
values stated herein and/or any information regarding the application of the device,
Infineon Technologies hereby disclaims any and all warranties and liabilities of any kind,
including without limitation, warranties of non-infringement of intellectual property rights
of any third party.
Information
For further information on technology, delivery terms and conditions and prices, please
contact the nearest Infineon Technologies Office (www.infineon.com).
Warnings
Due to technical requirements, components may contain dangerous substances. For information
on the types in question, please contact the nearest Infineon Technologies Office.
Infineon Technologies components may be used in life-support devices or systems only with
the express written approval of Infineon Technologies, if a failure of such components can
reasonably be expected to cause the failure of that life-support device or system or to affect
the safety or effectiveness of that device or system. Life support devices or systems are
intended to be implanted in the human body or to support and/or maintain and sustain
and/or protect human life. If they fail, it is reasonable to assume that the health of the user
or other persons may be endangered.
Title: ICE1HS01G Application Note ANPS0031
Revision History:
Previous Version:
Page
25 MAY 2009
V1.0
none
Subjects (major changes since last revision)
Half Bridge LLC Resonant Converter Design using ICE1HS01G
License to Infineon Technologies Asia Pacific Pte Ltd
Mao Mingping
[email protected]
He Yi
[email protected]
Jeoh Meng kiat
[email protected]
We Listen to Your Comments
Any information within this document that you feel is wrong, unclear or missing at all?
Your feedback will help us to continuously improve the quality of this document.
Please send your proposal (including a reference to this document) to:
[email protected]
AN-PS0031
Half Bridge LLC Resonant Converter Design using ICE1HS01G
Table of Contents
1
Introduction..........................................................................................................5
2
Overview of Half Bridge LLC Resonant Converter ..........................................5
3
IC Description ......................................................................................................6
3.1
Main Features .............................................................................................................................6
3.2
Pin Configuration .......................................................................................................................6
3.3
Pin Functions..............................................................................................................................6
4
Application Information ......................................................................................7
4.1
Minimum switching frequency ..................................................................................................7
4.2
IC power supply and soft start ..................................................................................................8
4.3
Over Current Protection.............................................................................................................9
4.4
Feedback...................................................................................................................................10
4.5
Input voltage sense ..................................................................................................................10
4.6
Blanking time in case of over load protection........................................................................11
4.7
Auto restart time in case of over load protection ..................................................................12
5
Design Example.................................................................................................14
5.1
Design Specifications ..............................................................................................................14
5.2
Define System Specifications..................................................................................................14
5.3
Define the Required Voltage Gain of the Resonant Network ................................................14
5.4
Calculate the Transformer Turns Ratio...................................................................................17
5.5
Calculate the Effective Load Resistance ................................................................................17
5.6
Determine the Resonant Network ...........................................................................................17
5.7
Transformer Design..................................................................................................................19
5.8
Primary current, Resonant Cap Voltage and OCP level ........................................................19
5.9
Output Rectifier ........................................................................................................................20
6
Experiment Verification ....................................................................................20
6.1
200W 24V HB LLC Resonant Converter using ICE1HS01G ...................................................20
6.2
Schematic of 200W Half Bridge LLC Resonant Converter ....................................................21
6.3
PCB Bottom Layer ....................................................................................................................22
6.4
Transformer Construction .......................................................................................................22
6.5
Test Results ..............................................................................................................................23
7
References .........................................................................................................25
Application Note
4
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
1
Introduction
This application note describes how to design half bridge LLC resonant converter using ICE1HS01G, which
is an 8-pin LLC controller developed by Infineon Technologies. ICE1HS01G is specially designed for
applications of switch mode power supplies used in LCD / PDP TV, AC/DC adapter and Audio system.
In this application note, an overview of half bridge LLC resonant converter will be given at first, followed by
the introduction of ICE1HS01G functions and operations. A typical application example, 200W HB LLC
resonant converter demoboard using ICE1HS01G, will be given in the last past of this document.
2
Overview of Half Bridge LLC Resonant Converter
The increasing requirements of lighter, smaller and more efficient electronic products demand the power
supply designers to develop DC/DC converter with high power density and efficiency.
The conventional PWM power converters are widely used in low and medium power applications. However,
due to the known limitations exhibited by PWM converters, such as drop in efficiency and deterioration of
EMI problem at high-switching frequency and high-input voltage, the efficiency and power density can not be
further improved easily. For this reason, the resonant converter is a good alternative because of its softswitching characteristic. The resonant DC/DC converter can considerably reduce the switching loss and
obtain low EMI emission, which has facilitated its adoption in a diverse range of applications [1,2].
A lot of advantages of the LLC resonant converter, such as zero-voltage switching (ZVS) capability of
MOSFETs, load insensitive characteristic at normal operation point, output voltage regulation even at zero
load condition and low EMI emission, have been investigated and verified in many literatures [3-5]. These
features can fully meet the power supply’s demands in many modern applications such as LCD/PDP TV,
AC/DC adapter, audio system, etc.
Figure 1 Typical application of 8-pin half bridge LLC controller ICE1HS01G
Figure 1 shows a typical application of ICE1HS01G in half bridge LLC resonant converter. The driver module
can be implemented by either a pulse transformer or a high voltage driver IC. The mains input voltage is
normally around 380Vdc delivered by the frontend PFC pre-regulator. The MOSFETs Q1 and Q2 are driven
complementarily to generate a square waveform at the input of the resonant tank. The elements of the
resonant tank are the resonant inductance Lr, the magnetising inductance Lm of the transformer and the
resonant capacitor Cs. Lr is often realized with the leakage inductance of the transformer. During operation,
the primary MOSFETs Q1 and Q2 are turned-on under ZVS condition, and the secondary rectifier diodes DO1
and DO2 are turned-on and turned-off under ZCS condition. Hence high switching frequency and high power
density can be achieved. In addition, MOSFETs Q1 and Q2 and rectifier diodes DO1 and DO2 have low
voltage stresses clamped by the input and output voltages, respectively. Hence, the devices with lower
voltage rating can be used, and consequently lower conduction loss and lower cost can be further achieved.
Application Note
5
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
3
IC Description
ICE1HS01G is an 8-pin controller IC; nevertheless, it includes all necessary protection features for HB LLC
resonant converter. ICE1HS01G allows the designer to choose suitable operation frequency range by
programming the oscillator with an external resistor. And the programmed soft-start function to limit both the
inrush current and the overshoot of output voltage is also provided. In addition, ICE1HS01G performs all
necessary protection functions in HB LLC resonant converters. All of these make ICE1HS01G an
outstanding product for HB LLC resonant converter in the market.
3.1
•
•
•
•
•
•
•
•
3.2
Main Features
Maximum 600kHz switching frequency
Adjustable minimum switching frequency with high accuracy
50% duty cycle
Mains input under voltage protection with adjustable hysteresis
Two levels of overcurrent protection: frequency shift and latch off
Open-loop/over load protection with adjustable blanking time
Built-in digital and nonlinear softstart
Adjustable restart time during over load protection
Pin Configuration
Figure 2 Pin configuration (top view)
3.3
Pin Functions
FMIN (Minimum Switching Frequency)
An external resistor RFMIN is connected between this pin and the ground. The voltage of this pin is constant
during operation and thus the resistance determines the current flowing out of this pin. The minimum
switching frequency is determined by this current. The maximum switching frequency during normal
operation and the maximum switching frequency during soft start are both related to the current flowing out
of FMIN pin.
CS (Current Sense)
The current sense signal is fed to this pin. Inside the IC, two comparators are provided. If the voltage on CS
pin is higher than the first threshold (0.8V typically), IC will increase the switching frequency to limit the
maximum output power of the converter. If the voltage on this pin exceeds the second threshold(1.6V
typically), IC will be latched off immediately.
FB (Feedback)
This pin is connected to the collector of the external optocoupler. Internally, during normal operation, this pin
is connected to reference voltage source with a pull-up resistor (RFB). The IC uses the voltage on this pin to
adjust the switching frequency within the range of maximum and minimum frequency set by FMIN pin. If FB
Application Note
6
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
voltage is higher than VFBH for a certain internallyn fixed blanking time (20ms), an extended timer will be
started. If over load/open loop protection exists longer than the extended blanking time, IC will enter autorestart mode. Another off timer starts from the instant IC stops switching till IC starts another soft start. This
off timer is determined by the resistors and capacitor connected to VINS pin. More details regarding this
function are provided in section 4.6 and 4.7.
VINS (Mains Input Voltage Sense)
The mains input voltage is fed to this pin via a resistive voltage divider. If the voltage on VINS pin is higher
than the threshold VINSon (1.25V typically), IC will start to operate with softstart when VCC increases beyond
turn on threshold (12V typically). During operation, if the voltage on this pin falls below the threshold VINSon,
IC will stop switching until the voltage at this pin increases again. When IC goes into over load protection
mode, IC will stop switching and try to restart after a period of time. This period is adjustable by the RC
network connected between VINS pin and ground. More details regarding this function are provided in
section 4.7.
GND (Ground)
IC common ground.
LG (Low Side Gate Drive)
Low side power MOSFET driver.
HG (High Side Gate Drive)
High side power MOSFET driver.
VCC (IC Power Supply)
Supply voltage of this IC, VCC pin should be connected to an external auxiliary supply.
4
Application Information
4.1
Minimum switching frequency
The minimum switching frequency is a very important factor. ICE1HS01G allows the minimum switching
frequency easily programmed by connecting an external resistor RFMIN between FMIN pin and ground.
The IC internal circuit provides a regulated 1.5V voltage at FMIN pin. The resistor RFMIN , connected from
FMIN pin to GND, determines the current(IFMIN) flowing out from FMIN pin. A certain current proportional to
IFMIN is defined as the minimum charging current(Ichg_min), which in turn defines the minimum switching
frequency. The maximum switching frequency during normal operation and the switching frequency variation
range during soft start and over current protection are all related to this current flowing out of FMIN pin,
which will be discussed in the following section.
The relationship between minimum switching frequency and RFMIN is shown in Figure 3.
Application Note
7
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
Figure 3 Minimum switching frequency VS RFMIN
4.2
IC power supply and soft start
The controller ICE1HS01G is targetting at applications with auxiliary power supply. In most cases, a frontend PFC pre-regulator with a PFC controller is used in the same system.
The controller ICE1HS01G starts to operate when the supply voltage VCC reaches the on-threshold,
typically 12V. The minimum operating voltage after turn-on, VCCoff, is typically 11V. The maximum supply
voltage VCCmax is 18V. It is suggested that IC is supplied with a regulated dc power supply for stable
operation. At the same time, a small bypass filter capacitor 100nF is suggested to be put between VCC and
GND pins, as closely as possible.
After IC supply voltage is higher than 12V, and if the voltage on VINS pin is higher than 1.25V, IC will start
switching with soft start. The soft start function is built inside the IC with a digital manner. During softstart, the
switching frequency of the MOSFET is controlled internally by changing the current ISS instead of by the
feedback voltage. The charging current ISS during soft start, which determines the switching frequency, is
reduced step by step as shown in product datasheet [1]. The maximum duration of softstart is 32ms with
1ms for each step. Figure 4 illustrates the actual switching frequency VS start time when RFMIN=22kohm.
During softstart, the frequency starts from 250kHz, and step by step drops to normal operation point.
Application Note
8
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
Figure 4 Switching frequency during softstart @ RFmin=22kohm
600
550
500
450
400
350
300
250
200
150
100
50
0
0
5
10
15
20
25
30
35
40
45
50
55
60
65
RFMIN [kohm]
st
Figure 5 Soft start 1 step switching frequency VS RFmin
The soft start 1st step switching frequency, maximum frequency during softstart, is also closely related to the
minimum switching frequency fixed by external RFmin resistance. Figure 5 illustrates the relationship between
st
the 1 step frequency and RFmin.
During this 32ms soft start, the overload protection is disabled.
4.3
Over Current Protection
Current sense pin in ICE1HS01G is only for protection purpose. ICE1HS01G features two-level over current
protection. In case of over-load condition, the lower OCP level,0.8V,will be triggerred, the switching
frequency will be increased according to the duration and power of the over load. The higher OCP level,
1.6V,is used to protect the converter if transformer winding is shorted. When Vcs reaches 1.6V, the IC will
be latched immediately.
Application Note
9
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
If Vcs is higher than 0.8V, IC will boost up the switching frequency. If Vcs is lower than 0.75V, IC will resume
to normal operation gradually. If Vcs is always higher than 0.8V for 1.5ms, the frequency will rise to its
maximum level, and vice versa.
To sum up, ICE1HS01G will increase the switching frequency to limit the resonant current in case of
temporary over-load and will also decrease the switching frequency to its normal value after over-load
condition goes away.
4.4
Feedback
The output load information is fed into the controller through feedback voltage VFB. Inside the IC, the
feedback (FB) pin is connected to the 5V voltage source through a pull-up resistor RFB. Outside the IC, this
pin is connected to the collector of opto-coupler. Normally, a ceramic capacitor CFB can be put between this
pin and ground for signal smoothing purpose, also CFB is used to determine the extended blanking time for
over load protection, which will be discussed in section 4.6
If the output load is increased, and consequently VFB is higher, ICE1HS01G will reduce the switching
frequency to regulate the output voltage and vice versa. The regulation of switching frequency is achieved by
changing the charging current IFB. The relationship between IFB and VFB can be found in product datasheet
[1]. The effective range of feedback voltage VFB is from 0.8V to 3.8V. Figure 6 graphs the relationship
between the actual switching frequency and feedback voltage VFB when RFMIN=22kohm.
Figure 6 Switching frequency VS feedback @ RFmin=22kohm
At very light load condition with high input voltage, the designed maximum frequency may not be high
enough to regulate the output voltage. In order to avoid this case, the feedback signal VFB is continuously
monitored. When VFB drops below VFB_off (typical 0.2V), the switching signal will be disabled after a fixed
blanking time TFB (typical 200ns). VFB will then rise as Vout starts to decrease due to no switching signal.
Once VFB exceeds the threshold VFB_on (typical 0.3V), IC resumes to normal operation.
4.5
Input voltage sense
The working range of mains input voltage needs to be specified for LLC resonant converter. It is important
for the controller to have input voltage sensing function and protection features, which allows the IC to stop
switching when the input voltage drops below the specified range and restart with soft start when the input
voltage resumes to its normal level. The mains input voltage sensing circuit is shown in product datasheet [1].
Thanks to the internal current source Ihys connected between VINS pin and Ground, an adjustable hysteresis
between the on and off threshold of mains input voltage can be created as
Application Note
10
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
Vhys = RINS1 ⋅ I hys
The mains input voltage is divided by RINS1 and RINS2. If the on and off threshold for mains input voltage is
Vmainon and Vmainoff, the resistors RINS1 and RINS2 can be selected as
RINS1 =
4.6
Vmainon − Vmainoff
,
I hys
RINS 2 = RINS 1 ⋅
VVINSon
Vmainoff − VVINSon
Blanking time in case of over load protection
In case of output over load or open control loop fault, the FB voltage will increase to its maximum level. If FB
voltage is higher than VFBH and this condition last longer than a fixed blanking time of TOLP (20ms), the IC will
start the extended blanking timer. The extended blanking timer is realized by charging and discharging the
filter capacitor CFB via the internal pull up resistor RFB and switch QFB. The circuit for extended blanking time
is shown in Figure 7.
Iref
I
ICE1HS01G
IFB
Vdd
1.0V
S
RFB
FB
TOLP
20ms
CFB1
4.5V
OPTO
Q
R
CFB
CFB2
EnA
S
QFB
TOLP_R
1.2ms
Q
R
CFB3
EnA
UP Reset
OLP
CLK
0.5V
S
Q
AR
R
AR_R
EnA
CFB4
R
. 0.2V
0.3V
EnA
Q
Gate_off
S
CFB5
Figure 7 Circuit connected to FB pin
The FB voltage waveform during an OLP period is shown in Figure 8. After FB voltage has been higher than
VFBH (4.5V typically) for the fixed blanking time t1 shown in Figure 8, IC will use internal switch QFB to
discharge VFB to VFBL (0.5V typically). After the switch QFB is released, CFB will be charged up by Vdd through
RFB. The time needed for CFB being charged from VFBL to VFBH can be calculated as:
 V − VFBH
tchg _ olp = − ln dd
 Vdd − VFBL

 ⋅ RFB ⋅ CFB

The time needed for CFB being discharged from VFBH to VFBL can be calculated as:
V
t dischg _ olp = ln FBH
 VFBL
where

 ⋅ RQFB ⋅ C FB

RQFB is switch QFB‘s on resitance. If CFB is 680pF, tchg _ olp is about 30us, t dischg _ olp is about 1.4us.
Application Note
11
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
After VFB reaches VFBH, an internal counter will increase by 1 and the capacitor is discharged to VFBL by QFB
again. The charging and discharging process of CFB will be repeated for NOLP_E times (NOLP_E=512) if the fault
condition still exists. After the last time of NOLP_E, the FB voltage is pulled down to VFBL, IC will stop switching
when FB voltage rises to VFBH again. This is called over load/ open loop proteciton. During the charging and
discharging period, the IC will operate with frequency determined by Ichg_min and ICS.
Figure 8 FB voltage waveform during over load protection
If the converter returns to normal operation during the extended blanking time period as mentioned above,
FB voltage can not go up to VFBH again. Therefore, after FB voltage is discharged to zero voltage, if it can not
go up to VFBH within TOLP_R (1.28ms typically), IC will reset all the fault timer to zero and return to normal
operation.
4.7
Auto restart time in case of over load protection
After IC enters into OLP, both switches will be stopped. However, the IC remains active and will try to start
with soft start after an adjustable period. This period is realized by charging and discharging the capacitor
CINS ,connected to VINS pin, for NOLP_R times (NOLP_R=2048), hence, this period can be adjusted by CINS,
RINS1 and RINS2. The circuit implementation of the adjustable off time is shown in Figure 9, and Figure 10
shows the voltage waveform of VINS pin in this case.
Figure 9 Circuit connected to VINS pin
Application Note
12
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
VINS_H
VINS_L
t2
t1
t3 Time
Figure 10 VINS voltage waveform during restart time
As shown in Figure 10, the voltage across CINS is discharged to VINS_L (0.5V typically) when IC goes into OLP
at time t1. After that, an internal constant current source IINST is turned on to charge CINS. Once the voltage at
VINS pin is charged to VINS_H (4.5V typically), the current source will be turned off and CINS is discharged by
another switch Q3 to VINS_L again. The charging and discharging of CINS comprise one cycle. This cylce time
is also influenced by the bus voltage. The charging and discharging time of CINS can be approximated as
tch arg ing
R

 VBUS ⋅ eq + I INST ⋅ Req − VINSH
RINS1
= − Req ⋅ C INS ⋅ ln
Req
+ I INST ⋅ Req − VINSL
 VBUS ⋅
RINS1

t dich arg ing
R

 VBUS ⋅ eq 2 − VINSL
RINS1
= − Req 2 ⋅ C INS ⋅ ln
Req 2
− VINSH
 VBUS ⋅
RINS1













where Req is the equivalent resistance for parallelling of RINS1 and RINS2,
Req = RINS1 // RINS 2
Req2 is the equivalent resistance for parallelling of RINS1, RINS2 and RQ3 (900ohm typically).
Req 2 = RINS1 // RINS 2 // RQ 3
IC will repeat the charging and discharging process for NOLP_R times (NOLP_R=2048). After that, IC will turn off
the switches for both charging and discharging. In addition, the current source for hysteresis will be turned
on and another blanking time of TBL_VINS, the time between t2 and t3 as shown in Fiugre 10 will be added so
that VINS pin fully recovers and represents the bus voltage information. IC will start the soft start after the
additional blanking time in case VVINS is higher than the VVINSon.
Application Note
13
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
5
Design Example
A 24V 200W LLC demoboard using ICE1HS01G is now available. In order to simplify and speed up this IC’s
feature evaluation, no PFC stage is implemented, so 280Vac input is recommended to feed this 200W
demboard, thus around 380Vdc across bulk capacitor can be obtained.
5.1
Design Specifications
The LLC stage is usually used to follow a PFC stage, thus the nominal input voltage for LLC stage can be
specified as:
Vin _ nom = 400Vdc
Output voltage is specified as 24V:
Vout = 24V
Output current is specified as 8A:
I out = 8A
The required hold-up time:
T h= 20ms
The PFC output capacitor:
Co = 220uF
5.2
Define System Specifications
The estimated efficiency:
η = 0.93
The input power will then be:
Pin =
Vout ⋅ I out
η
= 206.45W
During the hold up time, the input voltage for LLC stage drops gradually to a lower level, and the output
voltage of LLC stage is still required to be regulated. The required minimum input DC voltage can be
estimated as:
2
Vin _ min = Vin _ nom −
The resonant frequency is selected as:
5.3
2 ⋅ Pin ⋅ Th
= 349.95V
Co
f r = 100kHz
Define the Required Voltage Gain of the Resonant Network
The integrated magnetic solution is often be used for LLC resonant design. The resonant inductance Lr is
often combined with the transformer into a single magnetic part. The transfomer’s physical model is shown in
Figure 11, where the topological analogy with the inductive part of the LLC resonant tank circuit is apparent.
Figure 12 Transformer primary referred model
Figure 11 LLC integrated transformer model
Application Note
14
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
The transformer’s all primary referred model is shown as Figure 12, where
neq is the equivalent turn ratio. All
the elements related to leakage flux are located on the primary side.
If Lp is defined as the primary inductance mearured at primary side with secondary winding open, Lp will be
the sum of Lr and Lm. Considering the influence of secondary leakage inductance, the equivalent turn ratio
can be estimated as [4]:
neq =
n
=
Lp
Lp − Lr
n
m
m −1
where n is the transformer’s physical turn ratio;
m is the ratio between primary inductance L p and resonant inductance Lr : m = L p Lr
The equivalent circuit for LLC resonant network is ploted as Figure 13.
Figure 13 Equivalent circuit for LLC resonant network
where Cr is the resonant capacitor;
Lr is the resonant inductance, which is made by the transformer primary leakage inductance and the
reflected secondary leakage inductance;
Lm is the magnetizing inductane;
Reff _ 2 is the effective load resistance considering the influence of secondary leakage inductance:
Reff _ 2 =
8
π
2
⋅ neq ⋅
2
Vout
I out
The input RMS voltage Vin _ ac across the resonant network can be calculated as:
Vin _ ac =
where
2
Vin _ dc
π
Vin _ dc is the DC input voltage for LLC stage, which is usually powered by frontend PFC stage.
The output RMS voltage
Vout _ ac across the effective load resistance is:
Vout _ ac =
Thus the voltage gain can be calculated as: G =
2 2
π
Vout _ ac
Vin _ ac
Vout ⋅ neq
=
2neqVout
Vin _ dc
As shown in Figure 13, the resonant network and the load acts as a voltage divider, the expression of the
voltage gain can be obtained as:
Application Note
15
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
G=
2neqVout
Vin _ dc
=



 
− 1 + j 
 

 f2
 2
f
 p
2
f 
 ⋅ (m − 1)
f r 

f   f2
 ⋅  2 − 1 ⋅ (m − 1) ⋅ Qe
fr   fr

=
(m ⋅ F
F 2 (m − 1)
2
− 1) + F 2 (F 2 − 1) (m − 1) Qe
2
2
2
2
with the following parameter definitions:
right-side resonant frequency:
fr =
1
2π LrCr
left-side resonant frequency:
fp =
1
2π LpCr
inductance ratio:
m=
normalized frequency:
F=
quality factor:
Qe =
Considering neq =
n
=
Lp
Lp − Lr
Lp
Lr
f
fr
1
Lr
Lr 8
2 V
⋅
=
⋅ 2 ⋅ neq ⋅ out .
Cr Reff _ 2
Cr π
I out
n
, another voltage gain G ′ directly related to the physical turn ratio
m
m −1
n can be expressed as:
2
G′ =
2nVout
=
Vin _ dc
 f2
 2
f
 p
 f 
  ⋅ m( m − 1)
 fr 
=
  f   f2

− 1 + j   ⋅  2 − 1 ⋅ (m − 1) ⋅ Qe
  fr  f
 r


(m ⋅ F
F 2 ⋅ m( m − 1)
2
− 1) + F 2 ⋅ (F 2 − 1) (m − 1) Qe
2
2
2
2
For LLC topology, when operating close to the resonant point, the frequency’s load-insensitivity can be
achieved, thus the optimal operation point is put close to the resonant frequency point. Hence, the switching
frequency is recommended to be put at the resonant point when LLC operates with nominal input voltage:
f s _ nom = f r
Based on above G ′ expression, the voltage gain at f s _ nom and Vin _ nom can be given as:
Gnom =
m
m −1
Hence the gain at f s _ nom is determined by choosing the inductance ratio m .
Too small m means smaller L p and bigger Lr , which will result in poor coupling of the transformer and
deteriorate the efficiency due to increased circulating current. Typically m is set between 3 to 7.
If the chosen m value is:
Application Note
m=5
16
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
Then the voltage gain at f s _ nom and Vin _ nom can be calculated as:
m
= 1.12
m −1
Gnom =
Accordingly the maximum required voltage gain is at Vin _ min and can be given as:
Gmax =
5.4
Vin _ nom
Vin _ min
⋅ Gnom = 1.28
Calculate the Transformer Turns Ratio
Assuming the secondary rectifier diode voltage drop V f is:
V f = 0.6V
The transformer turns ratio will be:
5.5
Vin _ nom
n=
2 ⋅ (Vout + V f )
⋅ Gnom = 9.09 ≈ 9
Calculate the Effective Load Resistance
Reff =
The effective load resistance is:
8
π
2
⋅ n2 ⋅
Vout
= 196.97Ω
I out
If considering the transformer’s secondary leakage inductance, the effective load resistance is:
Reff _ 2 =
5.6
8
π
2
2
⋅ neq ⋅
Reff
Vout
=
= 157.57Ω
I out Gnom 2
Determine the Resonant Network
The m value is chosen same as mentioned above:
m =5.
The gain equation can be recalculated as:
G′ =
2nVout
=
Vin
F 2 ⋅ m( m − 1)
(m ⋅ F
2
− 1) + F 2 ⋅ (F 2 − 1) (m − 1) Qe
2
2
2
2
=
1
 5F 2 − 1 
1
2 
 + 0.8 ⋅ Qe ⋅  F − 
 2
F

 F ⋅ 20 
2
Then the frequency response of the voltage gain G ′ vs F and Qe is plotted as following:
Application Note
17
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
2.5
Qe=0.2
Qe=0.3
2.25
2
1.75
Qe=0.4
1.5
Qe=0.5
Gpk=1.47
Qe=0.6
Gmax=1.28
1.25
Gnom=1.12
Qe=0.7
Qe=0.8
1
0.75
0.5
0.2
0.4
0.6
0.8
1
1.2
1.4
F
Normalized frequency
Figure 14 Voltage gain vs normalized frequency
As discussed above, the maximum required voltage gain Gmax is calculated to meet the hold up time
requirement. In addition, in order to ensure stable ZVS operation and meet the output voltage regulation
requirement when LLC operates at the lowest allowable input voltage, the peak gain G PK at full load
condition should be somehow higher than Gmax .
Considering 15% margin:
G pk = 1.15 ⋅ Gmax = 1.47
According to Figure 14, the voltage gain when Qe = 0.5 can meet this peak gain requirement.
Accordingly, the resonant capacitor can be selected as:
Cr =
1
= 20.2nF
2π ⋅ Qe ⋅ f r ⋅ Reff _ 2
Also the resonant inductance will then be: Lr =
The primary inductance is:
1
= 125.39uH
(2π ⋅ f r )2 ⋅ C r
L p = m ⋅ Lr = 626.97uH
When using the integrated magnetic solution, the resonant inductance Lr is implemented by the leakage
inductance, hence the value of Lr is not easy to control. The resonant network parameters based on above
calculation often needs to be changed according to the measured inductance values and the standard
capacitance value.
Application Note
18
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
5.7
Transformer Design
The worst case for LLC transformer’s operation is that when LLC operates at minimum switching frequency
when input DC voltage drops to the lowest allowable voltage and full load.
The maximum required voltage gain occurs at Vin _ min and can be given as:
Gmax =
Vin _ nom
Vin _ min
⋅ Gnom = 1.28
Base on Figure 14, it can be roughly estimated that
G = 1.28 when F = 0.78 and Qe = 0.5
Accordingly the minimum switching frequency can be obtained as:
f min = F ⋅ f r = 78kHz
According to Faraday’s Law, the minimum number of primary turns for LLC transformer can be estimated as:
N p _ min ==
neq ⋅ (Vout + V f
)=
2 f min ⋅ ∆B ⋅ Ae
n ⋅ (Vout + V f )
m ( m − 1) ⋅ 2 f min ⋅ ∆B ⋅ Ae
The transformer core is selected as PC47 EER32, its effective cross sectional area is:
Ae = 79.6mm 2
To avoid saturation, the flux density swing (peak to peak) is chosen as:
∆B = 0.45T
The minimum number of primary turns can be calculated as:
N p _ min =
n ⋅ (Vout + V f )
m ( m − 1) ⋅ 2 f min ⋅ ∆B ⋅ Ae
=
9 ⋅ ( 24V + 0.6V )
= 35.44
1.12 ⋅ 2 ⋅ 78kHz ⋅ 0.45T ⋅ 79.6mm 2
The actual number of primary turns can be selected as: N p = 36
Accordingly the number of secondary turns is: N s = N p n = 4
5.8
Primary current, Resonant Cap Voltage and OCP level
The primary RMS current flowing through resonant capacitor can be obtained as:
I r _ rms
2

n (Vout + V f )
 π ⋅ I out  
 = 1.3A
= ⋅ 
 +
η  2 2 ⋅ n   4 2 ⋅ f r ⋅ M ⋅ (L p − Lr ) 
2
1
Hence, the primary peak current is: I r _ pk =
2 ⋅ I r _ rms = 1.84A
The over current protection level is set to about 50% margin of this peak current:
I ocp = (1 + 50%) ⋅ I r _ pk = 2.76A
The maximum voltage across resonant capacitor occurs at nominal input dc voltage and OCP level:
VCr _ max =
Application Note
Vin _ nom
2
+
I ocp
2π ⋅ f r ⋅ C r
19
= 436.36V
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
Thus, a 630V film capacitor with low ESR is recommened to be used as resonant capacitor.
5.9
Output Rectifier
For the output winding with center tap, the output diode voltage stress is
Vd = (Vout + V f ) ⋅ 2 = 49.2 V
The RMS current flowing through the output rectifier diode can be estimated as:
Id =
6
π
4
⋅ I out = 6.28A
Experiment Verification
A 200W half bridge LLC resonant converter demoboard with ICE1HS01G is implemented as shown in Figure
15. Also the full load efficiency of LLC stage has reached 94.35% as shown in Figure 22. The detailed
schematic circuit is shown in Figure 16 and Figure 17.
The specification of this 200W LLC demoboard is listed as following table 1.
Table 1 200W Demoboard Specification
6.1
Normal Input AC voltage
280Vac
Normal DC bulk voltage
400Vdc
Mains under voltage protection point
300Vdc
Auxiliary power supply for IC VCC
15Vdc
Normal output full load
24V/8A
Switching frequency
100kHz @ 24V/8A and 400Vdc input
200W 24V HB LLC Resonant Converter using ICE1HS01G
Figure 15 200W half bridge LLC resonant converter demoboard using ICE1HS01G
Application Note
20
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
Schematic of 200W Half Bridge LLC Resonant Converter
BR1
KBU8G (8A / 400V)
2
CY3 2n2/Y1
+
VCC
HG
LG
GND
C1
220u/450V
3
Q1
IPA50R299CP
QHG
QHS
QLG
QLS
TR1
L1 Short
ER42/ER35/ER32
R26N.C. R25N.C. R24N.C.
2n2/Y1
CX1
S10k/275
VR1
FUSE1
5A/250V
N1
HG
LG
D9
1N4148
D10
1N4148
ICE1HS01G
R10
2M
FMIN
CS
FB
75
R14
24k+1k
D14 VF30100SG
R23
2k2
6
R27
N.C.
C13
D11
N.C.
R28
N.C.
N.C.
R11
2M
8
3
220pF / 630V
R15
VINS
470uF/35V
GND
5
GND
IC1
+ C19
+
ZD5
15V
R22
5k6
C15
1uF/35V
D12
N.C.
+
R17
30k / 1%
R18
1k1 / 1%
C11
47uF N.C.
C10 68nF
R20 27k
C8 220nF
3
4
1
L3
3.3mH/4.6A
VCC
C2
22nF/630V
C9
R16
150
220nF/275Vac
L
2
R1 N.C.
C12
100nF
+
C14 N.C.
IPA50R299CP
3
2
C5
47uF/50V
+
C18
C16
C17
1000uF/35V 1000uF/35V 1000uF/35V
10
9
+
CX2
100nF/275Vac
24V
+
CY2
Q2
2n2/Y1
L2 1.2u/7.5A
1
1
RT1
S237/5
CY1
D13 VF30100SG
11
IC2
SFH615A-2
C7
R12
1M
680pF5%
C6
R13
22k
4
1
3
2
IC3
1
R21
1k1
C20 N.C.
TL431
R19
3k6 / 1%
2
4
6.2
22nF5%
Figure 16 Schematics of 200W half-bridge LLC resonant converter
VCC
Q4
Q6
D2
1N4148
BC557
BC546
C3 TR2
100nF
1
2
8
R5
10k
2
2
LG
1
1
ZD1
15V
D3
1N4148
D4
1N4148
C4
100nF
3
Q7
BC557
ZD2
15V
QHS
6
QLG
D7
1N4148
R8
10
R9
10
ZD3
15V
1
Q8
BC557
3
GND
D6
1N4148
7
5
R4
150
R7
10
1
Q5
BC557
3
2
2
1
D1
1N4148
3
R3
10k
1
3
R2
150
HG
R6
10
2
3
D5
1N4148
BC546
2
QHG
Q3
D8
1N4148
ZD4
15V
QLS
Figure 17 Schematics of driver circuit
The AC line input side comprises the input fuse FUSE1 as overcurrent protection. The X2 Capacitors CX1,
CX2 and Choke L3 and Y1 capacitors CY1 and CY2 forms a main filter to minimize the feedback of RFI into
the main supply. RT1 is placed in series with input to limit the initial peak inrush current. After the bridge
rectifier BR1, together with a smoothing capacitor C1, a voltage of 300VDC to 400 VDC is provided,
depending on mains input voltage, to simulate the real operation condition with front end PFC pre_regulator.
As shown in Figure 17, a cost-effective pulse transformer TR2 is used to transmit the driver signal to
MOSFETs for isolation purpose. The totem pole driver circuits (optional), including a NPN and a PNP
transistor is used to drive the pulse transformer. In the secondary side of the driver circuit, R7, R9, D6, D8,
Q5, and Q8 are used to accelerate the turn-off speed of MOSFET. If the sink impedance of the pulse
transformer is enough, these circuits can also be saved. In this case, a simplified driver circuit is shown in
Figure 18.
Application Note
21
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
QLG
11R
3R3
HG
3R3
5
MBR160
1
6
2
7
15V
QLS
QHG
8
LG
15V
1N4148
11R
3R3
1u0
3R3
15V
1N4148
MBR160
15V
QHS
Figure 18 A simplified driver circuit
6.3
PCB Bottom Layer
Figure 19 Solder side copper – View from component side
6.4
Transformer Construction
•
•
•
•
•
Bobbin: Split type EC32, Horizontal version from TDK
Core: PC47 ER32 from TDK
Primary inductance: 616µH±5%, Gapped between Pin 1 and Pin 3
Leakage inductance: 136µH±5%, measured between Pin 1 and Pin 3 by shorting (Pin 7 & 8)
Measured at frequency of 40kHz
Application Note
22
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
Figure 20 LLC resonant transformer electrical diagram
Figure 21 LLC resonant transformer winding position
Figure 22 LLC resonant transformer complete – top view
Table 2 LLC resonant transformer winding characteristics
6.5
¾
Test Results
Efficiency Mesurements
Table 3 shows the output voltage measurements at the nominal input bulk voltage 400Vdc, with different load
conditions. The bulk voltage 400Vdc is directly supplied from Chroma programmable DC power supply.
Hense, there is no current flowing through the bridge rectifier, and the measured efficiency is actually the LLC
stage’s efficiency.
Table 3 Efficiency measurements @ Vbulk=400Vdc
24Vout
Pinput_main
Pinput_IC and
24Vout [V]
current [A]
power [W]
Driver [W]
Poutput [W]
Efficiency [%]
23.92
7.980
201.8
15*0.034=0.51
190.882
94.35
Application Note
23
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
23.92
6.977
175.9
15*0.034=0.51
166.890
94.60
23.93
5.989
150.9
15*0.034=0.51
143.317
94.65
23.93
4.986
125.6
15*0.034=0.51
119.315
94.61
23.93
3.984
100.5
15*0.034=0.51
95.337
94.38
23.93
2.996
75.9
15*0.034=0.51
71.694
93.83
23.93
1.993
51.1
15*0.035=0.525
47.692
92.38
23.93
0.990
26.5
15*0.035=0.525
23.691
87.66
Figure 23 LLC stage efficiency
The power losses due to IC and driver circuit are both included. In addition, the efficiency values were
measured after 30 minutes of warm-up at full load.
¾
Resonant stage operating waveforms
a. full load 8A output
b. light load 0.5A output
Figure 24 Resonant stage operating waveforms
Figure 24 shows the resonant waveforms during steady state operation of this LLC resonant circuit at
nominal dc input voltage 400Vdc and full load 8A and light load 0.5A condition. Channel 1 shows the Mosfet
drive signal Vgs. The half bridge square voltage, which driving the resonant circuit, is shown in channel 2. It
Application Note
24
12 August 2009
Half Bridge LLC Resonant Converter Design using ICE1HS01G
can be found that the zero voltage switching is achieved. The primary resonant current is shown by channel 4,
it is almost sinusoidal because the operating point is close to the resonant frequency f r .
¾
Secondary side operating waveforms
a. full load 8A output
b. light load 0.5A output
Figure 25 Secondary rectifier diode voltage stress and flowing current
Figure 25 shows the voltage across the secondary rectifier diode, the voltage stress equals to 51V. The
current flowing through rectifier diode is also shown by channel 4. This current shape is almost a sine wave,
and its average value equals to one half the output current.
7
References
[1]
ICE1HS01G datasheet, Infineon Technologies AG, 2008
[2]
RW ERICKSON, D MAKSIMOVIC: ‘Fundamentals of power electronics’ (Kluwer Academic
Publishers, 2001), pp. 705–755
[3]
B Yang: ‘Topology investigation for front end DC/DC power conversion for distributed power system’,
PhD thesis, Virginia Polytechnic Institute and State University, 2003
[4]
S.De Simone.: ‘Design-oriented steady state analysis of LLC resonant converters based on FHA’,
SPEEDAM 2006, 2006.
[5]
Mingping Mao, Dimitar Tchobanov, Dong Li, Martin Maerz, Tobias Gerber, Gerald Deboy, Leo
Lorenz.: ‘Analysis and design of a 1MHz LLC Resonant Converter with Coreless transformer driver’.
PCIM Conference, Shanghai. 2007
[6]
M Mao, D Tchobanov, D Li, M Maerz.: ‘Design optimization of a 1MHz half bridge CLL resonant
converter’. IET Power Electronics, 2008, Vol.1, pp. 100-108.
Application Note
25
12 August 2009
Similar pages