SEMTECH SC480

SC480
Complete DDR1/2/3
Memory Power Supply
POWER MANAGEMENT
Description
Features
The SC480 is a combination switching regulator and
linear source/sink regulator intended for DDR1/2/3
memory systems. The purpose of the switching regulator
is to generate the supply voltage, VDDQ, for the memory
system. It is a pseudo-fixed frequency constant ontime controller designed for high efficiency, superior DC
accuracy, and fast transient response. The purpose of the
linear source/sink regulator is to generate the memory
termination voltage, VTT, with the ability to source and
sink a 3A peak current.
‹ Constant On-Time Controller for Fast Dynamic
Response on VDDQ
‹ DDR1/DDR2/DDR3 Compatible
‹ VDDQ = Fixed 1.8V or 2.5V, or Adjustable from
1.5V to 3.0V
‹ 1% Internal Reference (2% System Accuracy)
‹ Resistor Programmable On-Time for VDDQ
‹ VCCA/VDDP Range = 4.5V to 5.5V
‹ VIN Range = 2.5V to 25V
‹ VDDQ DC Current Sense Using Low-Side RDS(ON)
Sensing or External RSENSE in Series with Low-Side
FET
‹ Cycle-by-Cycle Current Limit for VDDQ
‹ Digital Soft-Start for VDDQ
‹ Analog Soft-Start for VTT/REF
‹ Smart Over-Voltage VDDQ Protection Against SourceCurrent Loads
‹ Combined EN and PSAVE Pin for VDDQ
‹ Over-Voltage/Under-Voltage Fault Protection
‹ Power Good Output
‹ Separate VCCA and VDDP Supplies
‹ VTT/REF Range = 0.75V – 1.5V
‹ VTT Source/Sink 3A Peak
‹ Internal Resistor Divider for VTT/REF
‹ VTT is High Impedance in S3
‹ VDDQ, VTT, REF Are Actively Discharged in S4/S5
‹ 24 Lead MLPQ (4x4 mm) Lead-Free Package
‹ Fully WEEE and RoHS Compliant
For the VDDQ regulator, the switching frequency is constant
until a step in load or line voltage occurs at which time the
pulse density, i.e., frequency, will increase or decrease to
counter the transient change in output or input voltage.
After the transient, the frequency will return to steady-state
operation. At lighter loads, the selectable Power-Save
Mode enables the PWM converter to reduce its switching
frequency and improve efficiency. The integrated gate
drivers feature adaptive shoot-through protection and
soft-switching. Additional features include cycle-by-cycle
current limiting, digital soft-start, over-voltage and undervoltage protection and a power good flag.
For the VTT regulator, the output voltage tracks REF,
which is ½ VDDQ to provide an accurate termination
voltage. The VTT output is generated from a 1.2V to VDDQ
input by a linear source/sink regulator which is designed
for high DC accuracy, fast transient response, and low
external component count. All three outputs (VDDQ, VTT
and REF) are actively discharged when VDDQ is disabled,
reducing external component count and cost. The SC480
is available in a 24-pin MLPQ (4x4 mm) package.
Applications
Notebook Computers
CPU I/O Supplies
Handheld Terminals and PDAs
LCD Monitors
Network Power Supplies
¡
¡
¡
¡
¡
Typical Application Circuit
5V
D1
20
21
19
DL
LX
22
C9
1uF
NC
PGD
PAD
+
C6
18
17
16
RILIM
15
14
13
PGOOD
PAD
R4
12
C10
1uF
EN/PSV
VDDP
VCCA
11
REF
DH
VDDP
VTTEN
TON
7
R6
10R
ILIM
REF
10
6
PGND1
SC480
FB
5
C7
1nF
VSSA
VDDQ
L1
Q2
PGND1
U1
VTTS
9
4
BST
24
VTT
3
VTTSNS
PGND2
VDDQS
2
NC
1
R1
1Meg
VTTIN
C1
1uF
8
C5
10uF
C4
10uF
23
Q1
VTT
VBAT
VBAT
C3
2x10uF
C2
0.1uF
VDDQ
R7
10R
5V
VDDQ
C8
0.1uF
EN/PSV
C11
1uF
VTT_EN
November 3, 2006
1
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SC480
POWER
POWER MANAGEMENT
MANAGEMENT
Absolute Maximum Ratings
Exceeding the specifications below may result in permanent damage to the device or device malfunction. Operation outside of the parameters specified in the
Electrical Characteristics section is not implied. Exposure to absolute maximum rated conditions for extended periods of time may affect device reliability.
Parameter
Symbol
Maximum
Units
TON to VSSA
-0.3 to +25.0
V
DH, BST to PGND1
-0.3 to +31.0
V
BST, DH to LX
-0.3 to +6.0
V
LX to PGND1
-2.0 to +25.0
V
DL, ILIM, VDDP to PGND1
-0.3 to +6.0
V
VDDP to DL
-0.3 to +6.0
V
VTTIN to PGND2; VTT to PGND2; VTTIN to VTT
-0.3 to +6.0
V
EN/PSV, FB, PGD, REF, VCCA, VDDQS, VTTEN, VTTS to VSSA
-0.3 to +6.0
V
VCCA to EN/PSV, FB, REF, VDDQS, VTT, VTTEN, VTTIN, VTTS
-0.3 to +6.0
V
PGND1 to PGND2; PGND1 to VSSA; PGND2 to VSSA
-0.3 to +0.3
V
Thermal Resistance Junction to Ambient(1)
θJA
29
°C/W
Operating Junction Temperature Range
TJ
-40 to +150
°C
Storage Temperature Range
TSTG
-65 to +150
°C
Peak IR Reflow Temperature, 10s - 40s
TPKG
260
°C
ESD Protection Level(2)
VESD
2
kV
Notes:
1) Calculated from package in still air, mounted to 3” x 4.5”, 4 layer FR4 PCB with thermal vias under the exposed pad per JESD51 standards.
2) Tested according to JEDEC standard JESD22-A114-B.
Electrical Characteristics
TEST CONDITIONS: VIN = 15V, VCCA = VDDP = VTTEN = EN/PSV = 5V, VDDQ = VTTIN = 1.8V, RTON = 1MΩ. TAMB = -40 TO +85C.
25°C
Parameter
-40°C to 85°C
Conditions
Units
Min
Typ
Max
Min
Max
Input Supplies
VCCA Operating Current
S0 State (VTT on);
EN/PSV = VCCA;
FB > Regulation Point, IVDDQ = 0A
1500
2500
μA
VCCA Operating Current
S3 State (VTT off);
EN/PSV = VCCA;
FB > Regulation Point, IVDDQ = 0A
800
1400
μA
5.5
V
150
μA
VCCA Operating Voltage
VDDP Operating Current
5
4.5
FB > Regulation Point, IVDDQ = 0A
70
RTON = 1MΩ
15
IVTT = 0A
1
5
μA
VCCA + VDDP + TON
Shutdown Current
EN/PSV = VTTEN = 0V
5
11
μA
VTTIN Shutdown Current
EN/PSV = VTTEN = 0V
1
TON Operating Current
VTTIN Operating Current
© 2006 Semtech Corp.
2
μA
μA
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SC480
POWER MANAGEMENT
Electrical Characteristics (Cont.)
TEST CONDITIONS: VIN = 15V, VCCA = VDDP = VTTEN = EN/PSV = 5V, VDDQ = VTTIN = 1.8V, RTON = 1MΩ. TAMB = -40 TO +85C.
25°C
Parameter
-40°C to 85°C
Conditions
Units
Min
Typ
Max
Min
Max
VDDQ Controller
FB Error Comparator Threshold(1)
VDDQS Regulation Threshold
On-Time
With Adjustable Resistor Divider
1.500
1.485
1.515
V
FB = AGND
2.5
2.475
2.525
V
FB = VCCA
1.8
1.782
1.818
V
RTON = 1MΩ, VDDQ = 1.8V
460
368
552
RTON = 500kΩ, VDDQ = 1.8V
265
212
318
Minimum Off-Time
VDDQS Input Resistance
VDDQS Shutdown
Discharge Resistance
425
550
ns
ns
FB < 0.3V
80
kΩ
FB > 0.3V
91
kΩ
EN/PSV = GND
16
Ω
FB Leakage Current
-1.0
VDDQ Smart Psave Threshold
1.0
8
μA
%
VTT Controller
REF Source Current
10
REF Sink Resistance
REF Output Accuracy
Shutdown Discharge Resistance
(EN/PSV = GND)
VTT Output Accuracy
(with respect to REF)
mA
50
kΩ
IREF = 0 to 10mA
900
VTT
0.25
Ω
REF
8
Ω
-2A < IVTT < 2A(9)
0
VTTS Leakage Current
882
918
mV
-40
+40
mV
-1.0
1.0
μA
9
11
μA
-10
10
mV
Current Sensing
ILIM Current
Current Comparator Offset
Zero-Crossing Threshold
DL High
10
PGND1 - ILIM
PGND1 - LX, EN/PSV = 5V
5
mV
PGND1 - LX, RLIM = 5kΩ
50
35
65
PGND1 - LX, RLIM = 10kΩ
100
80
120
PGND1 - LX, RLIM = 20kΩ
200
170
230
PGND1 - LX
-125
-160
-90
mV
With Respect to FB Regulation Point
-30
-35
-25
%
VDDQ Fault Protection
Current Limit (Positive)(2)
Current Limit (Negative)
Output Under-Voltage Fault
© 2006 Semtech Corp.
3
mV
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SC480
POWER
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Electrical Characteristics (Cont.)
TEST CONDITIONS: VIN = 15V, VCCA = VDDP = VTTEN = EN/PSV = 5V, VDDQ = VTTIN = 1.8V, RTON = 1MΩ. TAMB = -40 TO +85C.
25°C
Parameter
-40°C to 85°C
Conditions
Units
Min
Typ
Max
Min
Max
Under-Voltage Fault Delay
FB Forced Below UV VTH
8
clks(3)
Under-Voltage Blank Time
From EN High
440
clks(3)
Output Over-Voltage Fault
With Respect to FB Regulation Point
+16
Over-Voltage Fault Delay
FB Above Over-Voltage Threshold
5
PGD Low Output Voltage
Sink 1mA
0.1
V
FB in Regulation, PGD = 5V
1
μA
-8
%
PGD Leakage Current
PGD UV Threshold
+12
+20
%
μs
With Respect to FB Regulation Point
-10
-12
PGD Fault Delay
FB Forced Outside PGD Window
5
VCCA Under-Voltage (UVLO)
Falling Edge (Hysteresis 100 mV)
4
3.70
4.35
V
UV Lower Threshold
VTT w/rt REF
-12
-16
-8
%
OV Upper Threshold
VTT w/rt REF
+12
+8
+16
%
VTT Outside OV/UV Window
50
μs
VTT Fault Protection
Fault Shutdown Delay
Thermal Shutdown(4)(5)
160
μs
150
170
°C
Inputs/Outputs
EN/PSV Low/Low (Disabled)
1.2
VTTEN Low (VTT Disabled)
0.6
Logic Input Low Voltage
Logic Input High Voltage
Logic Input High Voltage
EN/PSV Input Resistance
V
EN/PSV Low/High
(Enabled, Psave Disabled)
1.2
VTTEN High (VTT Enabled)
2.4
EN/PSV High/High
(Enabled, Psave Enabled)
3.1
2.4
V
V
Sourcing
1.5
MΩ
Sinking
1.0
MΩ
VTTEN Leakage Current
-1
+1
μA
Soft-Start
VDDQ Soft-Start Ramp Time
EN/PSV High to PGD High
VTT Soft-Start Ramp Rate(6)
© 2006 Semtech Corp.
4
440
clks(3)
5.5
mV/μs
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SC480
POWER MANAGEMENT
Electrical Characteristics (Cont.)
TEST CONDITIONS: VIN = 15V, VCCA = VDDP = VTTEN = EN/PSV = 5V, VDDQ = VTTIN = 1.8V, RTON = 1MΩ. TAMB = -40 TO +85C.
25°C
Parameter
-40°C to 85°C
Conditions
Units
Min
Typ
Max
Min
Max
FB Input Thresholds
FB Logic Input Low
VDDQ Set for 2.5V (DDR1)
0.3
V
FB Logic Input High
VDDQ Set for 1.8V (DDR2)
VCCA
- 0.7
V
DH or DL Rising
30
ns
DL Low
0.8
Ω
VDL = 2.5V
3.1
A
DL High
2
Ω
VDL = 2.5V
1.3
A
DH Pull-Down Resistance
DH Low, BST - LX = 5V
2
Ω
DH Pull-Up Resistance
DH High, BST - LX = 5V
2
Ω
DH Sink/Source Current
VDH = 2.5V
1.3
A
VTT Pull-Up Resistance
VTTS < REF
0.25
Ω
VTT Pull-Down Resistance
VTTS > REF
0.25
Ω
Gate Drives
Shoot-Thru Protection Delay(4)(7)
DL Pull-Down Resistance
DL Sink Current
DL Pull-Up Resistance
DL Source Current
(8)
VTT Peak Sink/Source Current(9)
3.6
2.0
A
Notes:
1)
2)
3)
4)
5)
6)
The VDDQ DC regulation level is higher than the FB error comparator threshold by 50% of the ripple voltage.
Using a current sense resistor, this measurement relates to PGND1 minus the source of the low-side MOSFET.
clks = switching cycles, consisting of one high side and one low side gate pulse.
Guaranteed by design.
Thermal shutdown latches both outputs (VTT and VDDQ) off, requiring VCCA or EN/PSV cycling to reset.
VTT soft-start ramp rate is limited to 5.5mV/μs typical. If the VDDQ/2 ramp rate is slower than 5.5mV/μsec, the VTT soft-start ramp will follow the VDDQ/2
ramp.
7) See Shoot-Through Delay Timing Diagram on Page 6.
8) Semtech’s SmartDriver™ FET drive first pulls DH high with a pull-up resistance of 10Ω (typ.) until LX = 1.5V (typ.). At this point, an additional pull-up device is
activated, reducing the resistance to 2Ω (typical). This creates a softer turn-on with minimal power loss, eliminating the need for an external gate or boost
resistor.
9) Provided operation below TJ(MAX) is maintained. VTT output current is also limited by internal MOSFET resistance which is typically 0.25Ω at 25°C and which
increases with temperature, and by available source voltage (typically VDDQ/2).
© 2006 Semtech Corp.
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SC480
POWER
POWER MANAGEMENT
MANAGEMENT
Shoot-Through Delay Timing Diagram
LX
DH
DL
DL
tplhDL
© 2006 Semtech Corp.
tplhDH
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SC480
POWER MANAGEMENT
Pin Configuration
Ordering Information
Device(2)
Package(1)
SC480IMLTRT
MLPQ-24
SC480EVB
Evaluation Board
DL
19
LX
20
21
22
DH
BST
VTTIN
23
24
VTT
SC480 MLP24 Pin Out
PGND2
1
18
PGND1
Notes:
VTTS
2
17
PGND1
1) Only available in tape and reel packaging. A reel contains 3000 devices.
2) This product is fully WEEE and RoHS compliant.
VSSA
3
16
ILIM
TON
4
15
VDDP
REF
5
14
VDDP
VCCA
6
13
PGD
12
11
EN/PSV
NC
10
VTTEN
9
FB
8
VDDQS
NC
7
T
Pin Description
Pin #
Pin Name
1
PGND2
2
VTTS
Sense pin for VTT. Connect to VTT at the load.
3
VSSA
Ground reference for analog circuitry. Connect to thermal pad.
4
TON
This pin is used to sense VBAT through a pull-up resistor, RTON, which sets the top MOSFET
on-time. Bypass this pin with a 1nF capacitor to VSSA.
5
REF
Reference output. An internal resistor divider from VDDQS sets this voltage to 50% VDDQ (nominal). Bypass this pin with a series 10Ω/1μF to VSSA.
6
VCCA
7
NC
8
VDDQS
9
FB
10
VTTEN
Enable pin for VTT. Pull this pin low to disable VTT (REF remains present as long as VDDQ is
present).
11
EN/PSV
Enable/Power Save input pin. Tie to ground to disable VDDQ. Tie to +5V to enable VDDQ and
activate PSAVE mode. Float to enable VDDQ and activate continous conduction mode. If floated,
bypass to VSSA with a 10nF capacitor.
12
NC
13
PGD
14,15
VDDP
© 2006 Semtech Corp.
Pin Function
Power ground for VTT output. Connect to thermal pad and ground plane.
Analog supply voltage input. Use a 10Ω/1μF RC filter from +5V to VSSA.
No Connect.
Sense input for VDDQ. Used to set the on-time for the top MOSFET and also to set REF/VTT.
Feedback select input for VDDQ. See FB Configuration Table.
No Connect.
Power good output for VDDQ. PGD is low if VDDQ is outside the power good thresholds. This
pin is an open drain NMOS output and requires an external pull-up resistor.
+5V supply voltage input for the VDDQ gate drivers.
7
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Pin Description (Cont.)
Current limit input pin. Connect to drain of low-side MOSFET for RDS(on) sensing or the source
for resistor sensing through a threshold sensing resistor.
16
ILIM
17,18
PGND1
19
DL
Gate drive output for the low side MOSFET switch.
20
LX
Phase node - the junction between the top and bottom FETs and the output inductor.
21
DH
Gate drive output for the high side MOSFET switch.
22
BST
Boost capacitor connection for the high side gate drive.
23
VTTIN
Input supply for the high side switch for VTT regulator. Decouple with a 1μF capacitor to
PGND2.
24
VTT
Output of the linear regulator. Decouple with two (minimum) 10μF ceramic capacitors to
PGND2, locating them directly across pins 24 and 1.
T
THERMAL
PAD
Power ground for VDDQ switching circuits. Connect to thermal pad and ground plane.
Pad for heatsinking purposes. Connect to ground plane using multiple vias. Not connected
internally.
Enable Control Logic
Enable Pin Status
Output Status
EN/PSV (1)
VTTEN
VDDQ(3)
0
0
OFF, Discharged
0
1
OFF, Discharged
1
0
ON
OFF, High Impedance
ON
1
1
ON
ON
ON
VTT(2)
OFF, Discharged
(2)(3)
OFF, Discharged
(2)(3)
REF(2)
(2)
(2)
OFF, Discharged
OFF, Discharged
(2)
(2)
Notes:
1) EN/PSV = 1 = EN/PSV high or floating.
2) Typical discharge resistances: VTT = 0.25Ω. REF = 8Ω.
3) VDDQ is discharged via external series resistance which must be added to SC480 internal discharge resistance to calculate discharge times.
This is separate from any external load on VDDQ.
FB Configuration Table
The FB pin can be configured for fixed or adjustable output voltage as shown.
FB
VDDQ(V)
VREF & VTT (V)
Note
GND
2.5
VDDQS/2
DDR1
VCCA
1.8
VDDQS/2
DDR2
FB Resistors
Adjustable
VDDQS/2
1.5V < VDDQ < 3.0V
© 2006 Semtech Corp.
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SC480
POWER MANAGEMENT
Block Diagram
VTTIN
VTTEN
VCCA
OTSD
VDDQS
DRVH
POR/SS
+12%
VTT
NOVLP
DRVL
-12%
PGND2
REF
DSCHG
DRVH
FEDLY
VCCA
+12%
DRVL
VTTS
VTTPGD
VTTRUN
EN/ PSV
OTSD
TON
TON/ TOFF
POR/SS
DSCHG
-12%
VDDQS
VDDQS
BST
DSCHG
HI
1.5V
REF
-10%
+16%
-30%
VMON
CONTROL
OV
LX
SD
DL
LX
SHOOT
THRU
VDDP
LO
1.5V
DH
DL
PGND1
FB
PWM
SENSE
-10%
UV
-30%
ILIM
SD
PGD
OV
+16%
FAULTMON
VSSA
Figure 1
© 2006 Semtech Corp.
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SC480
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Application Information
+5V Bias Supplies
The SC480 requires an external +5V bias supply in addition
to the battery. If stand-alone capability is required, the
+5V supply can be generated with an external linear
regulator. To minimize crosstalk, the controller has seven
supply pins: VDDP (2 pins), PGND1 (2 pins), PGND2,
VCCA and AGND.
The controller requires its own AGND plane which should
be tied by a single trace to the negative terminal of the
output capacitor. All external components referenced
to AGND in the schematic should then be connected to
the AGND plane. The supply decoupling capacitor should
be tied between VCCA and AGND. A single 10Ω resistor
should be used to decouple the VCCA supply from the
main VDDP supply. PGND can then be a separate plane
which is not used for routing analog traces. All PGND
connections should connect directly to this plane with
special attention given to avoiding indirect connections
between AGND and PGND which will create ground loops.
As mentioned above, the AGND plane must be connected
to the PGND plane at the negative terminal of the output
capacitor. The VDDP input provides power to the upper
and lower gate drivers. A decoupling capacitor for the
VDDP supply and PGND is recommended. No series
resistor between VDDP and the 5 volt bias is required.
Pseudo-Fixed Frequency Constant On-Time
PWM Controller
The PWM control method is a constant-on-time, pseudofixed frequency PWM controller, see Figure 1. The ripple
voltage seen across the output capacitor’s ESR provides
the PWM ramp signal, eliminating the need for a current
sense resistor. The on-time is determined by a one-shot
whose period is proportional to output voltage, and
inversely proportional to input voltage. A separate oneshot sets the minimum off-time (typically 425ns).
On-Time One-Shot (TON)
The on-time one-shot comparator has two inputs. One
input looks at the output voltage, while the other input
samples the input voltage and converts it to a proportional
current. This current charges an internal on-time capacitor.
The TON time is the time required for this capacitor to
charge from zero volts to VOUT, thereby making the ontime of the high-side switch directly proportional to output
voltage and inversely proportional to input voltage. This
implementation results in a nearly constant switching
frequency without the need of a clock generator.
© 2006 Semtech Corp.
10
TON
§V
·
3.3x10 12 x (RTON 37x10 3 ) x ¨ OUT ¸ 50ns
¨ V ¸
© IN ¹
RTON is a resistor connected between the input supply and
the TON pin.
VDDQ/VTT Enable & Power-Save
The EN/PSV pin controls the VDDQ supply and the REF
output (1/2 of VDDQ). VTTEN enables the VTT supply. The
VTT and VDDQ supplies may be enabled independently.
When EN/PSV is tied to VCCA the VDDQ controller is
enabled in power-save mode. When the EN/PSV pin is
floated, an internal resistor divider activates the VDDQ
controller with power-save disabled. If PSAVE is enabled,
the SC480 PSAVE comparator looks for inductor current
to cross zero on eight consecutive cycles. Once observed,
the controller enters power-save and turns off the lowside MOSFET when the current crosses zero. To improve
the efficiency and add hysteresis, the on-time is increased
by 20% in power-save. The efficiency improvement at light
loads more than offsets the disadvantage of slightly higher
output ripple. If the inductor current does not cross zero
on any switching cycle, the controller immediately exits
power-save. Since the controller counts zero crossings,
the converter can sink current as long as the current does
not cross zero on eight consecutive cycles. This allows the
output voltage to recover quickly in response to negative
load steps even when power-save is enabled.
VDDQ Voltage Selection
VDDQ voltage is set using the FB pin. Grounding FB sets
VDDQ to fixed 2.5V. Connecting FB to +5V sets VDDQ to
fixed 1.8V. VDDQ can also be adjusted from 1.5 to 3.0V
using external resistors, see Figure 2. The voltage at FB is
then compared to the internal 1.5V reference.
To VDDQ output capacitor
C
R2
To SC480 FB (pin 9)
R3
Figure 2
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Application Information
Referencing Figure 2, the equation for setting the output
voltage is:
V OUT
(1 R2
R3
+5V
) x 1 .5
+VIN
+
D1
C1
Q1
C2
BST
DH
LX
ILIM
VDDP
DL
PGND
Current Limit Circuit
Current limiting of the SC480 can be accomplished in two
ways. The on-state resistance of the low-side MOSFETs
can be used as the current sensing element, or a sense
resistor in the low-side source can be used if greater accuracy is desired. RDSON sensing is more efficient and
less expensive. In both cases, the RILIM resistor between
the ILIM pin and LX sets the over-current threshold. This
resistor RILIM is connected to a 10μA current source within
the SC480 which is turned on when the low-side MOSFET
turns on. When the voltage drop across the sense resistor
or low-side MOSFET equals the voltage across the RILIMresistor, current limit will activate. The high-side MOSFET will
not be allowed to turn on until the voltage drop across the
sense element (resistor or MOSFET) falls below the voltage
across the RILIM resistor.
L1
Vout
R1
D2
SC480
+
Q2
C3
Figure 4
The schematic of RDSON sensing circuit is shown in Figure
4 with RILIM = R1 and RDSON of Q2.
Similarly, for resistor sensing, the current through the lower
MOSFET and the source sense resistor develops a voltage
that opposes the voltage developed across RILIM. When the
voltage developed across the RSENSE resistor reaches voltage
drop across RILIM, an over-current exists and the high-side
MOSFET will not be allowed to turn on. The over-current
equation when using an external sense resistor is:
The current sensing circuit actually regulates the inductor
valley current, see Figure 3. This means that if the current
limit is set to 10A, the peak current through the inductor
would be 10A plus the peak ripple current, and the average
current through the inductor would be 10A plus 1/2 the
peak-to-peak ripple current.
IL OC Valley 10ƫA x
RILIM
RSENSE
INDUCTOR CURRENT
Schematic of resistor sensing circuit is shown in Figure 5
with RILIM = R1 and RSENSE = R4.
I PEAK
+VIN
+5V
I LOAD
+
D1
I LIMIT
C2
Q1
BST
DH
LX
ILIM
VDDP
DL
PGND
L1
Vout
D2
SC480
TIME
Valley Current - Limit Threshold Point
C1
+
C3
Q2
R1
R4
Figure 3
Figure 5
© 2006 Semtech Corp.
11
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SC480
POWER
POWER MANAGEMENT
MANAGEMENT
Application Information (Cont.)
Power Good Output
The VDDQ controller has a power good (PGD) output.
Power good is an open-drain output and requires a pullup resistor. When the output voltage is +16%/-10% from
its nominal voltage, PGD gets pulled low. It is held low
until the output voltage returns to within +16%/-10% of
nominal. PGD is also held low during start-up and will not
be allowed to transition high until soft-start is over and the
output reaches 90% of its set voltage. There is a 5μs delay
built into the PGD circuit to prevent false transitions.
Output Over-Voltage Protection
When the VDDQ output exceeds 16% of its set voltage,
the low-side MOSFET is latched on. It stays latched and
the SMPS stays off until the EN/PSV input is toggled or
VCCA is recycled. There is a 5μs delay built into the OV
protection circuit to prevent false transitions. During a
VDDQ OV shutdown, VTT is alive until VDDQ falls to typically
0.4V, at which point VTT is tri-stated.
When VTT exceeds 12% above its set voltage, the VTT
regulator will tristate. There is a 50μs delay to prevent false
OV trips due to transients or noise. The VDDQ regulator
continues to operate after VTT OV shutdown. The VTT OV
condition is removed by toggling VTTEN or EN/PSV, or by
recycling VCCA.
Smart Over-Voltage Protection
In some applications, the active loads on VDDQ can
actually leak current into VDDQ. If PSAVE mode is enabled
at very light loading, this leak can cause VDDQ to slowly
rise and reach the OV threshold, causing a hard shutdown.
To prevent this, the SC480 uses Smart OVP to prevent
this. When VDDQ exceeds 8% above nominal, DL drives
high to turn on the low-side MOSFET, which starts to draw
current from VDDQ via the inductor. When VDDQ drops
to the FB trip point, a normal TON switching cycle begins.
This prevents a hard OV shutdown.
Output Under-Voltage Protection
When VDDQ falls 30% below its set point for eight clock
cycles, the VDDQ output is shut off; the DL/DH drives are
pulled low to tristate the MOSFETS, and the SMPS stays
off until the Enable input is toggled or VCCA is recycled.
When VTT is 12% below its set voltage the VTT output is
tristated. There is a 50μs delay for VTT built into the UV
protection circuits to prevent false transitions.
© 2006 Semtech Corp.
12
POR, UVLO and Soft-Start
An internal power-on reset (POR) occurs when VCCA exceeds 3V, resetting the fault latch and soft-start counter,
and preparing the PWM for switching. VCCA under-voltage
lockout (UVLO), circuitry inhibits switching and tristates the
drivers until VCCA rises above 4.2V. At this time the circuit
will come out of UVLO and begin switching and the softstart circuit will progressively limit the output current over
a pre-determined time period. The ramp occurs in four
steps: 25%, 50%, 75% and 100%, thereby limiting the slew
rate of the output voltage. There is 100mV of hysteresis
built into the UVLO circuit and when VCCA falls to 4.1V the
output drivers are shutdown and tristated.
MOSFET Gate Drivers
The DH and DL drivers are optimized for moderate, highside, and larger low-side power MOSFETs. An adaptive
dead-time circuit monitors the DL output and prevents the
high-side MOSFET from turning on until DL is fully off, and
conversely, monitors the DH output and prevents the lowside MOSFET from turning on until DH is fully off. (Note:
be sure there is low resistance and low inductance between the
DH and DL outputs to the gate of each MOSFET.)
Design Procedure
Prior to designing a switch mode supply for a notebook computer, the input voltage, load current, switching frequency
and inductor ripple current must be specified.
Input Voltage Range
The maximum input voltage (VINMAX) is determined by the
highest AC adaptor voltage. The minimum input voltage
(VINMIN) is determined by the lowest battery voltage after
accounting for voltage drops due to connectors, fuses and
battery selector switches.
Maximum Load Current
There are two values of load current to consider: continuous load current and peak load current. Continuous load
current has more to do with thermal stresses and therefore drives the selection of input capacitors, MOSFETs
and commutation diodes. Peak load current determines
instantaneous component stresses and filtering requirements such as, inductor saturation, output capacitors and
design of the current limit circuit.
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SC480
POWER MANAGEMENT
Application Information (Cont.)
Switching Frequency
Switching frequency determines the trade-off between
size and effi ciency. Higher frequency increases switching losses in the MOSFETs, since losses are a function of
F*VIN2. Knowing the maximum input voltage and budget
for MOSFET switches usually dictates the final design.
The best way for checking stability is to apply a zero to
full load transient and observe the output voltage ripple
envelope for overshoot and ringing. Over one cycle of
ringing after the initial step is a sign that the ESR should
be increased.
+5V
Inductor Ripple Current
Low inductor values result in smaller size, but create higher ripple current and are less efficient because of the high
AC current flowing in the inductor. Higher inductor values
do reduce the ripple current and are more efficient, but
are larger and more costly. The selection of the ripple current is based on the maximum output current and tends
to be between 20% to 50% of the maximum load current.
Again, cost, size and efficiency all play a part in the selection process.
+
D1
BST
DH
LX
ILIM
VDDP
DL
PGND
SC480
14
13
12
11
10
9
8
C2
C1
Q1
L1
0.5V - 5.5V
R1
D2
Q2
R2
+
C4
10pF
C3
R3
FBK
Stability Considerations
Unstable operation shows up in two related but distinctly
different ways: double pulsing and fast-feedback loop instability. Double-pulsing occurs due to noise on the output
or because the ESR is too low, causing insufficient voltage
ramp in the output signal. This causes the error amplifier to
trigger prematurely after the 400ns minimum off-time has
expired. Double-pulsing will result in higher ripple voltage at
the output, but in most cases is harmless. In some cases,
however, double-pulsing can indicate the presence of loop
instability, which is caused by insufficient ESR. One simple
way to solve this problem is to add some trace resistance
in the high current output path. A side effect of doing this
is output voltage droop with load. Another way to eliminate
doubling-pulsing is to add a 10pF capacitor across the
upper feedback resistor divider network. This is shown in
Figure 6, by capacitor C4 in the schematic. This capacitance
should be left out until confirmation that double-pulsing exists. Adding this capacitance will add a zero in the transfer
function and should eliminate the problem. It is best to
leave a spot on the PCB in case it is needed.
Figure 6
SC480 ESR Requirements
The constant on-time control used in the SC480 regulates
the ripple voltage at the output capacitor. This signal
consists of a term generated by the output ESR of the
capacitor and a term based on the increase in voltage
across the capacitor due to charging and discharging
during the switching cycle. The minimum ESR is set to
generate the required ripple voltage for regulation. For most
applications the minimum ESR ripple voltage is dominated
by PCB layout and the properties of SP or POSCAP type
output capacitors. For applications using ceramic output
capacitors, the absolute minimum ESR must be considered.
If the ESR is low enough the ripple voltage is dominated
by the charging of the output capacitor. This ripple voltage
lags the on-time due to the LC poles and can cause double
pulsing if the phase delay exceeds the off-time of the
converter. Referring to Figure 5 on Page 11, the equation
for the minimum ESR as a function of output capacitance
and switching frequency and duty cycle is:
Loop instability can cause oscillations at the output as a
response to line or load transients. These oscillations can
trip the over-voltage protection latch or cause the output
voltage to fall below the tolerance limit.
© 2006 Semtech Corp.
+VIN
§
§ Fs - 200000 · ·
¨ 1 3x¨
¸ ¸
Fs
§ VOUT · ¨
©
¹ ¸
ESR ! ¨
¸x
© 1.5V ¹ ¨ 2 x Ư x Cout x Fs x 1 D 2 ¸
¨
¸
©
¹
13
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SC480
POWER
POWER MANAGEMENT
MANAGEMENT
Application Information (Cont.)
Dropout Performance
The output voltage adjust range for continuous-conduction
operation is limited by the fixed 400nS (typical) Minimum
Off-time One-shot. For best dropout performance, use the
slowest on-time setting of 200KHz. When working with
low input voltages, the duty-factor limit must be calculated
using worst-case values for on and off times. The IC dutyfactor limitation is given by:
DUTY
TON (MIN)
TON (MIN) TOFF (MAX)
Be sure to include inductor resistance and MOSFET onstate voltage drops when performing worst-case dropout
duty-factor calculations.
SC480 System DC Accuracy (VDDQ Controller)
Three IC parameters affect VDDQ accuracy: the internal
1.5V reference, the error comparator offset voltage, and
the switching frequency variation with line and load.
The internal 1%, 1.5V reference contains two error
components, a 0.5% DC error and a 0.5% supply and
temperature error. The error comparator offset is trimmed
so that it trips when the feedback pin is nominally 1.5
volts +/-1% at room temperature. The comparator offset
trim compensates for any DC error in the reference. Thus,
the percentage error is the sum of the reference variation
over supply and temperature and the offset in the error
comparator, or 1.5% total.
The on-time pulse in the SC480 is calculated to give a
pseudo-fixed frequency. Nevertheless, some frequency
variation with line and load can be expected. This variation
changes the output ripple voltage. Because constant ontime converters regulate to the valley of the output ripple,
½ of the output ripple appears as a DC regulation error.
For example, If the output ripple is 50mV with VIN = 6
volts, then the measured DC output will be 25mV above
the comparator trip point. If the ripple increases to 80mV
with VIN = 25 volts, then the measured DC output will
be 40mV above the comparator trip. The best way to
minimize this effect is to minimize the output ripple.
To compensate for valley regulation it is often desirable
to use passive droop. Take the feedback directly from the
output side of the inductor, incorporating a small amount
© 2006 Semtech Corp.
14
of trace resistance between the inductor and output
capacitor. This trace resistance should be optimized
so that at full load the output droops to near the lower
regulation limit. Passive droop minimizes the required
output capacitance because the voltage excursions due
to load steps are reduced.
Board components and layout also influence DC accuracy.
The use of 1% feedback resistors contributes up to 1%
error. If tighter DC accuracy is required use 0.1% feedback
resistors.
The output inductor value may change with current. This
will change the output ripple and thus the DC output
voltage (it will not change the frequency).
Switching frequency variation with load can be minimized
by choosing lower RDSON MOSFETs. High RDSON MOSFETS
will cause the switching frequency to increase as the load
current increases. This will reduce the ripple and thus
the DC output voltage. This inherent droop should be
considered when deciding if passive droop is required, or
if passive droop is desired in order to further reduce the
output capacitance.
Output DC Accuracy (VTT Output)
The VTT accuracy compared to VDDQ is determined by
two parameters: the REF output accuracy, and the VTT
output accuracy with respect to REF. The REF output
is generated internally from the VDDQS (sense input),
and tracks VDDQS with 2% accuracy. This REF output
becomes the reference for the VTT regulator. The VTT
regulator then tracks REF within +/-40mV (typically zero).
The total VTT/VDDQ tracking accuracy is then:
VTT error
VDDQS
x r 0.02 r 40mV
2
DDR Reference Buffer
The reference buffer is capable of sourcing 10mA. The
reference buffer has a class A output stage and therefore
will not sink significant current; there is an internal 50 kΩ
(typical) pulldown to ground. If higher current sinking is
required, an external pulldown resistor should be added.
Make sure that the ground side of this pulldown is tied to
the VTT ground plane near the PGND2 pin.
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SC480
POWER MANAGEMENT
Application Information (Cont.)
For stability, place a 10Ω/1μF series combination from
REF to VSSA. If REF load capacitance exceeds 1μF, place
at least 10Ω in series with the load capacitance to prevent
instability. It is possible to use only one 10Ω resistor, by
connecting the load capacitors in parallel with the 1μF,
and connecting the load REF to the capacitor side of the
10Ω resistor. (See the Typical Application Circuit on Page
1.) Note that this resistor creates an error term when REF
has a DC load. In most applications this is not a concern
since the DC load on REF is negligible.
t ON_VBAT(MIN)
ª
«3.3 x 10 12 x R tON 37 x 10 3
«¬
x
VOUT º
» 50 x 10 9 s
VBAT(MIN) »
¼
ª
«3.3 x 10 12 x R tON 37 x 10 3
¬«
x
VOUT º
» 50 x 10 9 s
VBAT(MAX) »
¼
and,
t ON_VBAT(MAX)
From these values of tON we can calculate the nominal
switching frequency as follows:
VOUT
f SW_VBAT (MIN)
Design Procedure
Prior to designing a switching output and making component selections, it is necessary to determine the input
voltage range and output voltage specifications. To demonstrate the procedure, the output for the schematic in
Figure 7 on page 18 will be designed.
§¨ V
·¸
xt
© BAT(MIN) ON_VBAT(MIN) ¹
Hz
and,
f SW_VBAT (MAX)
VOUT
Hz
§¨ V
·¸
x
t
ON_VBAT(MAX) ¹
© BAT(MAX)
The maximum input voltage (VBAT(MAX)) is determined by the
highest AC adaptor voltage. The minimum input voltage
(VBAT(MIN)) is determined by the lowest battery voltage after accounting for voltage drops due to connectors, fuses
and battery selector switches. For the purposes of this
design example we will use a VBAT range of 8V to 20V to
design VDDQ.
tON is generated by a one-shot comparator that samples
VBAT via RtON, converting this to a current. This current is
used to charge an internal 3.3pF capacitor to VOUT. The
equations above reflect this along with any internal components or delays that influence tON. For our example we
select RtON = 1MΩ:
Four parameters are needed for the design:
tON_VBAT(MIN) = 820ns and, tON_VBAT(MAX) = 358ns
1. Nominal output voltage, VOUT. We will use 1.8V with
internal feedback resistors (FB pin tied to VCCA).
2. Static (or DC) tolerance, TOLST (we will use +/-2%).
3. Transient tolerance, TOLTR and size of transient (we
will use +/-8% for a 10A to 5A load release for this
demonstration).
4. Maximum output current, IOUT (we will design for 10A).
fSW_VBAT(MIN) = 274kHz and fSW_VBAT(MAX) = 251kHz
Now that we know tON we can calculate suitable values for
the inductor. To do this we select an acceptable inductor
ripple current. The calculations below assume 50% of IOUT
which will give us a starting place.
Switching frequency determines the trade-off between
size and efficiency. Increased frequency increases
the switching losses in the MOSFETs, and losses are a
function of VBAT2. Knowing the maximum input voltage
and budget for MOSFET switches usually dictates where
the design ends up. The default RtON values of 1MΩ and
715kΩ are suggested only as a starting point.
VBAT(MIN) VOUT x
L VBAT (MAX)
VBAT(MAX) VOUT x
and,
§¨ 0.5 x I
·
OUT ¸¹
©
H
t ON_VBAT(MAX)
§¨ 0.5 x I
·
OUT ¸¹
©
H
For our example:
The first thing to do is to calculate the on-time, tON, at
VBAT(MIN) and VBAT(MAX), since this depends only upon VBAT, VOUT
and RtON.
© 2006 Semtech Corp.
t ON_VBAT (MIN)
L VBAT(MIN)
LVBAT(MIN) = 1.02μH and LVBAT(MAX) = 1.30μH,
15
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SC480
POWER
POWER MANAGEMENT
MANAGEMENT
Application Information (Cont.)
We will select an inductor value of 1.5μH to reduce the
ripple current, which can be calculated as follows:
I RIPPLE_VBAT( MIN )
V OUT ¸· x
¨§ V
© BAT (MIN )
¹
t ON_ VBAT ( MIN )
L
where ERRTR is the transient output tolerance. For this
case, ITRANS is the load transient of 5A (10A - 5A).
For our example:
A P P
ERRTR = 144mV and ERRDC = 18mV, therefore,
and,
I RIPPLE_VBAT( MAX )
V OUT ¸· x
¨§ V
© BAT (MAX )
¹
RESR_TR(MAX) = 17.6mΩ for a full 5A load transient.
t ON_ VBAT ( MAX )
AP P
L
We will select a value of 6mΩ maximum for our design,
which would be achieved by using two 12mΩ output capacitors in parallel. Now that we know the output ESR we
can calculate the output ripple voltage:
For our example:
IRIPPLE_VBAT(MIN) = 3.39AP-P and IRIPPLE_VBAT(MAX) = 4.34AP-P
From this we can calculate the minimum inductor current
rating for normal operation:
I RIPPLE_VBAT ( MAX )
I OUT ( MAX ) A (MIN)
2
I INDUCTOR ( MIN )
V RIPPLE_VBAT ( MIN )
R ESR x I RIPPLE_VBAT ( MIN ) VP P
V RIPPLE_VBAT ( MAX )
R ESR x IRIPPLE_VBAT ( MAX ) VP P
and,
For our example:
For our example:
IINDUCTOR(MIN) = 12.2A(MIN)
Next we will calculate the maximum output capacitor
equivalent series resistance (ESR). This is determined by
calculating the remaining static and transient tolerance
allowances. Then the maximum ESR is the smaller of the
calculated static ESR (RESR_ST(MAX)) and transient ESR
(RESR_TR(MAX)):
R ESR _ ST ( MAX )
ERR ST ERR DC x 2
Ohms
I RIPPLE _ V BAT ( MAX )
Where ERRST is the static output tolerance and ERRDC is
the DC error. The DC error will be 1% plus the tolerance
of the internal feedback. (Use 2% for external feedback,
which is 1% plus another 1% for the external resistors.)
For our example:
RESR_ST(MAX) = 8.3mΩ
© 2006 Semtech Corp.
Note that in order for the device to regulate in a controlled
manner, the ripple content at the feedback pin, VFB, should
be approximately 15mVP-P at minimum VBAT, and worst case
no smaller than 10mVP-P. Note that the voltage ripple at
FB is smaller than the voltage ripple at the output capacitor, due to the resistor divider. Also, when using internal
feedback (FB pin tied to 5V or GND), the FB resistor divider is actually inside the IC. If VRIPPLE_VBAT(MIN) as seen at
the FB point is less than 15mVP-P - whether internal or external FB is used - the above component values should be
revisited in order to improve this. For our example, since
the internal divider reduces the ripple signal by a factor of
(1.5V/1.8V), the internal FB ripple values are then 17mV
and 22mV, which is above the 15mV minimum.
When using external feedback, and with VDDQ greater
than 1.5V, a small capacitor, CTOP, can be used in parallel
with the top feedback resistor, RTOP, in order to ensure that
ripple at VFB is large enough. CTOP should not be greater
than 100pF. The value of CTOP can be calculated as follows, where RBOT is the bottom feedback resistor. Firstly
calculating the value of ZTOP required:
ERRST = 36mV and, ERRDC = 18mV, therefore,
R ESR_ TR ( MAX)
VRIPPLE_VBAT(MAX) = 20mVP-P and VRIPPLE_VBAT(MIN) = 26mVP-P
¨§ ERRTR ERRDC ¸·
©
¹
Ohms
IRIPPLE_VBAT( MAX ) ·
§
¨I
¸
¨ TRANS
¸
2
©
¹
16
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SC480
POWER MANAGEMENT
Application Information (Cont.)
Z TOP
RBOT
x
0.015
at its highest. The capacitance required for smaller transient steps my be calculated by substituting the desired
current for the Ifinal term. In this case Ifinal is set for 5A.
For our example:
V RIPPLE_VBAT (MIN ) 0.015 Ohms
Secondly calculating the value of CTOP required to achieve
this:
COUT(MIN) = 392μF.
§ 1
1 ·¸
¨
¸
¨Z
R
TOP ¹
© TOP
F
2 x Ư x f SW _ VBAT ( MIN )
C TOP
We will select 440μF, using two 220μF, 12mΩ capacitors
in parallel.
Next we calculate the RMS input ripple current, which is
largest at the minimum battery voltage:
Since our example uses internal feedback ,this method
cannot be used, however the voltage seen at the internal
FB point is already greater than 15mV.
Next we need to calculate the minimum output capacitance required to ensure that the output voltage does not
exceed the transient maximum limit, POSLIMTR, starting
from the actual static maximum, VOUT_ST_POS, when a load
release occurs:
V OUT_ST_POS
Where TOLTR is the transient tolerance. For our example:
POSLIMTR = 1.944V,
I VALLEY
The minimum output capacitance is calculated as follows:
2
C OUT(MIN)
Lx
A
R ILIM
F
IOUT IRIPPLE_VBAT( MIN )
2
A
IVALLEY x 1.2 x
R DS (ON) x 1.4
10 x 10 6
Ohms
For our example:
This calculation assumes the condition of a full-load to noload step transient occurring when the inductor current is
© 2006 Semtech Corp.
A RMS
The ripple at low battery voltage is used because we want
to make sure that current limit does not occur under normal operating conditions.
and,
Iinit 2 Ifinal 2
POSLIMTR2 VOUT_ST_POS2
VBAT _ MIN
Finally, we calculate the current limit resistor value. As described in the current limit section, the current limit looks
at the “valley current”, which is the average output current minus half the ripple current.
V OUT x TOL TR V
I OUT ( MAX ) IOUT
Input capacitors should be selected with sufficient ripple
current rating for this RMS current, for example a 10μF,
1210 size, 25V ceramic capacitor can handle approximately 3ARMS. Refer to manufacturer’s data sheets and
derate appropriately.
V OUT ERR DC V
IRIPPLE_VBAT ( MAX )
x
IIN(RMS) = 4.17ARMS
VOUT_ST_POS = 1.818V,
Iinit
V OUT x VBAT ( MIN ) VOUT
For our example:
For our example:
POSLIM TR
I IN ( RMS )
IVALLEY = 8.31A, RDS(ON) = 4mΩ, giving RILIM = 5.62kΩ
17
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SC480
POWER
POWER MANAGEMENT
MANAGEMENT
Application Information (Cont.)
Thermal Considerations
The junction temperature of the device may be calculated
as follows:
TJ = TAMB + θJA
where TJ is the junction temperature, TAMB is the ambient
temperature, PD is the total SC480 device dissipation,
The SC480 device dissipation can be determined using:
PD = VCCA • ICCA + VDDP • IDDP + VTT • |ITT|
The first two terms are losses for the analog and gate drive
circuits and generally do not present a thermal problem.
Typical ICCA (VCCA operating current) is roughly 1.5mA,
which creates 7.5mW loss from the 5V VCCA supply. The
VDDP supply current is used to drive the MOSFETs and
can be much higher, on the order of 30mA, which can
create up to 150mW of dissipation.
The last term, VTT * |ITT|, is the most significant term
from a thermal standpoint. The VTT regulator is a linear
device and will dissipate power proportional to the VTT
current and the voltage drop across the regulator. If VTT
= VDDQ/2, then the voltage drop across the regulator
is always VDDQ2, regardless of whether the regulator is
sinking or sourcing current. In either case the power lost
in the VTT regulator is VTT * |ITT|. The average or longterm value for ITT should be used.
The thermal resistance of the MLPQ package is affected
by PCB layout and the available ground planes and vias
which conduct heat away. A typical value is 29°C/watt.
Example:
ICCA = 1.5mA
IDDP = 25mA
VCCA = VDDP = 5V
VTT = 1.25V
ITT = 0.75A (average)
Ambient = 45 degrees C
Thermal resistance = 29
PD = 5V • 0.0015 A + 5V • 0.025A + 0.9V • |0.75|A
PD = 0.808W
TJ = TAMB + PD • TJA = 45 + 0.808W • 29°C/W = 68.4°C
© 2006 Semtech Corp.
18
Layout Guidelines
One (or more) ground planes are recommended to
minimize the effect of switching noise and copper losses,
and maximize heat dissipation. The IC ground reference,
VSSA, should be connected to PGND1 and PGND2 as a
star connection at the thermal pad, which in connects
using 4 vias to the ground plane. All components that are
:referenced to VSSA should connect to it directly on the
chip side, and not through the ground plane.
VDDQ: The feedback trace must be kept far away from
noise sources such as switching nodes, inductors and
gate drives. Route the feedback trace in a quiet layer if
possible, from the output capacitor back to the chip. Chip
supply decoupling capacitors (VCCA, VDDP) should be
located next to the pins (VCCA/VSSA, VDDP/PGND1) and
connected directly to them on the same side.
VTT: Because of the high bandwidth of the VTT regulator,
proper component placement and routing is essential to
prevent unwanted high-frequency oscillations which can
be caused by parasitic inductance and noise. The input
capacitors should be located at the VTT input pins (VTTIN
and PGND2), as close as possible to the chip to minimize
parasitics. Output capacitors should be directly located at
the VTT output pins (VTT and PGND2). The routing of the
feedback signal VTTS is critical. The trace from VTTS (pin
2) should be connected directly to the output capacitor
that is farthest from VTT (pin24); route this signal away
from noise sources such as the VDDQ power train or highspeed digital signals.
The switcher power section should connect directly to the
ground plane(s) using multiple vias as required for current
handling (including the chip power ground connections).
Power components should be placed to minimize loops
and reduce losses. Make all the connections on one side
of the PCB using wide copper filled areas if possible. Do
not use “minimum” land patterns for power components.
Minimize trace lengths between the gate drivers and the
gates of the MOSFETs to reduce parasitic impedances
(and MOSFET switching losses); the low-side MOSFET is
most critical. Maintain a length to width ratio of <20:1 for
gate drive signals. Use multiple vias as required by current
handling requirement (and to reduce parasitics) if routed
on more than one layer. Current sense connections must
always be made using Kelvin connections to ensure an
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SC480
POWER MANAGEMENT
Application Information (Cont.)
accurate signal. The layout can be generally considered
in three parts; the control section referenced to VSSA, the
VTT output, and the switcher power section.
Finally, connecting the control and switcher power sections
should be accomplished as follows:
1. Route VDDQ feedback trace in a “quiet” layer, away
from noise sources.
2. Route DL, DH and LX (low side FET gate drive, high
side FET gate drive and phase node) to the chip using
wide traces with multiple vias if using more than one
layer. These connections are to be as short as possible
for loop minimization, with a length to width ratio less
than 20:1 to minimize impedance. DL is the most
critical gate drive, with power ground as its return
path. LX is the noisiest node in the circuit, switching
between VBAT and ground at high frequencies, thus
should be kept as short as practical. DH has LX as its
return path.
3. BST is also a noisy node and should be kept as short
as possible.
4. Connect PGND1 pins on the chip directly to the VDDP
decoupling capacitor and then drop vias directly to
theground plane. Locate the current limit resistor at
the chip with a kelvin connection to the phase node.
Looking at the control section first, locate all components
referenced to VSSA on the schematic and place these
components at the chip. Connect VSSA using a wide
(>0.020”) trace. Very little current flows in the chip ground
therefore large areas of copper are not needed. Connect
the VSSA pin directly to the thermal pad under the device
as the only connection from PGND1 and PGND2 from
VSSA.
Decoupling capacitors for VCCA/VSSA and VDDP/PGND1
should be placed is as close as possible to the chip. The
feedback components connected to FB, along with the
VDDQ sense components, should also be located at the
chip. The feedback trace from the VDDQ output should
route from the top of the output capacitors, in a quiet
layer back to the FB components.
Next, looking at the switcher power section, there are a
few key guidelines to follow:
1. There should be a very small input loop, well
decoupled.
2. The phase node should be a large copper pour, but
still compact since this is the noisiest node.
3. Input power ground and output power ground should
not connect directly, but through the ground planes
instead.
© 2006 Semtech Corp.
19
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SC480
POWER
POWER MANAGEMENT
MANAGEMENT
Application Information (Cont.)
C1
D1
5V
0.1uF
Q1
IRF7811
MBR0530
L1
1.5uH
VDDQ
C4
1uF
C15
1uF
20
19
DL
DH
LX
21
22
BST
VTTIN
REF
VDDP
VCCA
PGD
NC
C14
1uF
6
EN/PSV
C13
1nF
10R
PAD
18
17
16
C9*
+
220uF/12m 220uF/12m
13
PAD
C11
0.1uF
*Sanyo 4TPL220MC
15
14
+
C10*
5V
C12
1uF
R4
10K
PGOOD
12
R5
5V
5
VDDP
11
REF
10R
SC480
VTTEN
R3
ILIM
TON
10
4
PGND1
U1
VSSA
7
VBAT
1MEG
VTTS
VDDQ
R1
5.62K
PGND1
FB
3
R2
PGND2
9
2
23
24
1
VDDQS
C7
10uF
0805
8
C6
10uF
0805
C8
0.1uF
VTT
C5
10uF
0805
Vishay IHLP-5050
Q2
IRF7832
NC
VTT
VBAT
C3
10uF/25V
1210
C2
10uF/25V
1210
EN/PSV
VDDQ
VTT_EN
C16
0.1uF
1.8V fixed: connect to 5V
2.5V fixed: connect to VSSA
Adjustable 1.5V-3.0V: connect to divider network
Figure 7 - Reference Design
© 2006 Semtech Corp.
20
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SC480
POWER MANAGEMENT
Typical Characteristics
1.8V Efficiency vs. Output Current
Powersave Mode
100%
1.8V Efficiency vs. Output Current
Continuous Conduction Mode
100%
VBAT = 10
VBAT = 10
90%
90%
VBAT = 20
Efficiency (%)
Efficiency (%)
VBAT = 20
80%
70%
60%
80%
70%
60%
50%
50%
0
2
4
6
8
0
10
2
4
2.5V Efficiency vs. Output Current
Powersave Mode
100%
100%
90%
VBAT = 20
Efficiency (%)
VBAT = 20
Efficiency (%)
10
VBAT = 10
90%
80%
70%
80%
70%
60%
60%
50%
50%
0
2
4
6
8
0
10
2
4
6
8
10
IOUT (A)
IOUT (A)
1.5V Efficiency vs. Output Current
Powersave Mode
1.5V Efficiency vs. Output Current
Continuous Conduction Mode
100%
VBAT = 10
VBAT = 10
90%
90%
VBAT = 20
VBAT = 20
Efficiency (%)
Efficiency (%)
8
2.5V Efficiency vs. Output Current
Continuous Conduction Mode
VBAT = 10
100%
6
IOUT (A)
IOUT (A)
80%
70%
60%
80%
70%
60%
50%
50%
0
2
4
6
8
10
0
IOUT (A)
© 2006 Semtech Corp.
21
2
4
IOUT (A)
6
8
10
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SC480
POWER
POWER MANAGEMENT
MANAGEMENT
Typical Characteristics (Cont.)
Load Transient Response, 0 to 5A, Psave Mode
Load Transient Response, 0 to 5A,
Continuous Conduction Mode
Load Transient Response, 5 to 0A, Psave Mode
Load Transient Response, 5 to 0A,
Continuous Conduction Mode
Load Transient Response, 5 to 10A
Load Transient Response, 10 to 5A
© 2006 Semtech Corp.
22
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SC480
POWER MANAGEMENT
Typical Characteristics (Cont.)
VTT Load Transient Response,
1A Sink/Source, Psave Mode
VTT Load Transient Response, 1A Sink/Source,
Continuous Conduction Mode
Startup (PSV), EN/PSV Going High
Startup (PSV), EN/PSV Going Low, VDDQ = 5A
© 2006 Semtech Corp.
23
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SC480
POWER
POWER MANAGEMENT
MANAGEMENT
Outline Drawing - MLPQ 24 (4x4mm)
A
D
B
DIM
PIN 1
INDICATOR
(LASER MARK)
A
A1
A2
b
D
D1
E
E1
e
L
N
aaa
bbb
E
A2
A
DIMENSIONS
INCHES
MILLIMETERS
MIN NOM MAX MIN NOM MAX
.031 .035 .040
.000 .001 .002
- (.008) .007 .010 .012
.151 .157 .163
.100 .106 .110
.151 .157 .163
.100 .106 .110
.020 BSC
.011 .016 .020
24
.004
.004
0.80 0.90 1.00
0.00 0.02 0.05
- (0.20) 0.18 0.25 0.30
3.85 4.00 4.15
2.55 2.70 2.80
3.85 4.00 4.15
2.55 2.70 2.80
0.50 BSC
0.30 0.40 0.50
24
0.10
0.10
SEATING
PLANE
aaa C
A1
C
D1
LxN
E/2
E1
2
1
N
bxN
bbb
e
C A B
D/2
NOTES:
1. CONTROLLING DIMENSIONS ARE IN MILLIMETERS (ANGLES IN DEGREES).
2. COPLANARITY APPLIES TO THE EXPOSED PAD AS WELL AS THE TERMINALS.
© 2006 Semtech Corp.
24
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SC480
POWER MANAGEMENT
Land Pattern - MLPQ 24 (4x4mm)
K
DIMENSIONS
(C)
G
H
Z
DIM
C
G
H
K
P
X
Y
Z
INCHES
(.155)
.122
.106
.106
.021
.010
.033
.189
MILLIMETERS
(3.95)
3.10
2.70
2.70
0.50
0.25
0.85
4.80
X
P
NOTES:
1. THIS LAND PATTERN IS FOR REFERENCE PURPOSES ONLY.
CONSULT YOUR MANUFACTURING GROUP TO ENSURE YOUR
COMPANY'S MANUFACTURING GUIDELINES ARE MET.
Contact Information
Semtech Corporation
Power Management Products Division
200 Flynn Road, Camarillo, CA 93012
Phone: (805) 498-2111 Fax: (805) 498-3804
www.semtech.com
© 2006 Semtech Corp.
25
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