EXAR SP6134HCU-L

SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
January 2009
Rev. 2.0.0
GENERAL DESCRIPTION
APPLICATIONS
The SP6134H is a 600kHz constant frequency,
voltage mode, synchronous PWM step down
controller optimized for high efficiency.
• Wireless Base Station
• Automotive
• Industrial
The SP6134H is adequately suited for split
plane applications utilizing a low power 5V rail
to power the controller circuitry, minimizing
power dissipation. Its wide input voltage range
of 3V to 15V allows for conversions from the
standard 3.3V, 5V, 9.6V and 12V power rails
to an output voltage adjustable down to 0.8V.
Developed around a wide bandwidth internal
amplifier, the SP6134H can accommodate type
II and type III compensation schemes.
• Power Supply
FEATURES
• 3V to 28V Step Down Achieved Using
Dual Input
• On-Board 1.5Ω sink (2Ω source) NFET
Drivers
• Up to 15A Output Capability
Protection features include a programmable
UVLO, thermal shutdown and output short
circuit protection.
• UVLO Detects Both VCC and VIN
• Short-Circuit Protection with AutoRestart
The SP6134H is part of a larger family of step
down
controllers
operating
at
various
switching frequencies up to 1300kHz and input
voltages up to 28V. Refer to Exar’s SP6134,
SP6132, SP6132H, SP6137 and SP6139 for
complete details.
• Supports Type II or III Compensation
• Programmable Soft Start
• Fast Transient Response
• High Efficiency: Greater than 94%
The SP6134H is available in lead free, RoHS
compliant,
space
saving
10-pin
MSOP
package.
• Non-synchronous Start-Up
• Small 10-Pin MSOP Package
• U.S. Patent #6,922,041
TYPICAL APPLICATION DIAGRAM
VIN
3.5V - 15V
C1
22μF
16V
8 7 6 5
C2
22μF
16V
VCC = 5V @ 30mA
FDS6676S
14.5A, 6mΩ
QT
4
RLF
3.0,5%
GND
C1, C2
Ceramic
1210
X5R
1 2 3
CBST
1μF
U1
DBST
MBR0530
SP6134H
1
R3
221k, 1%
CVCC
10μF
6.3V
GND 3
2
3
4
0.8V
5
UVIN
R4
100k, 1%
CVCC
Ceramic
8050
X5R
VCC
BST 10
R5
Bead
GH 9
GL
GND
SWN 8
VFB
SS 7
COMP
UVIN 6
CZ2
RZ2
RZ3
4.64k, 1%
8 7 6 5
C3
47μF
6.3V
QB
4
SS
CSS
47nF
1 2 3
R1
68.1k, 1%
GND2
C3, C4
Ceramic
1210
X5R
56pF
fs=600Khz
CZ3
220pF
FDS6676S
14.5A, 6.0mΩ
820pF 40.2k, 1%
CP1
CF1
100pF
C4
47μF
6.3V
VOUT≤VIN
0.8V - 3.3V
0 - 10 A
V
R2
21.5k, 1%
=(R1/R2 +1)V
OUT
FB
Fig. 1: SP6134H Application Diagram
Exar Corporation
48720 Kato Road, Fremont CA 94538, USA
www.exar.com
Tel. +1 510 668-7000 – Fax. +1 510 668-7001
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
ABSOLUTE MAXIMUM RATINGS
These are stress ratings only and functional operation of
the device at these ratings or any other above those
indicated in the operation sections of the specifications
below is not implied. Exposure to absolute maximum
rating conditions for extended periods of time may affect
reliability.
GH, GL peak output current <10us..............................2A
Storage Temperature ..............................-65°C to 150°C
Power Dissipation.................................Internally Limited
ESD Rating BST Pin (HBM - Human Body Model)...... 1.5kV
ESD Rating All Other Pins (HBM) ............................... 2kV
Thermal Resistance θJA ....................................41.9°C/W
Operating Voltage Range ................................ 3V to 28V
Vcc .......................................................................... 7V
BST ...................................................................... 33V
BST-SWN ......................................................-0.3 to 7V
SWN ............................................................-1V to 30V
GH................................................... -0.3V to BST+0.3V
GH-SWN.................................................................. 7V
All other pins ..................................... -0.3V to VCC+0.3V
ELECTRICAL SPECIFICATIONS
Specifications with standard type are for an Operating Junction Temperature of TJ = 25°C only; limits applying over the full
Operating Junction Temperature range are denoted by a “•”. Minimum and Maximum limits are guaranteed through test,
design, or statistical correlation. Typical values represent the most likely parametric norm at TJ = 25°C, and are provided for
reference purposes only. Unless otherwise indicated, Vcc = 4.5V to 5.5V, BST= Vcc, SWN=GND=0V, UVIN=3V, CVcc = 10µF,
CCOMP=0.1uF, CGH=CGL=3.3nF, CSS=50nF, TA= –40°C to 85°C (SP6134HEU) or TA= –0°C to 70°C (SP6134HCU).
Parameter
Min.
VCC Supply Current
BST Supply Current
VCC UVLO Start Threshold
4.00
Typ.
Max.
Units
1.5
3
mA
0.2
0.4
mA
•
4.25
4.50
V
•
mV
VCC UVLO Hysteresis
100
200
300
UVIN Start Threshold
2.30
2.50
2.65
V
UVIN Hysteresis
260
300
390
mV
1
uA
UVIN Input Current
Error Amplifier Reference
0.792
0.800
0.808
V
Error Amplifier Reference
Over Line and Temperature
0.788
0.800
0.812
V
Error Amplifier
Transconductance
6
Conditions
VFB =0.9V (No switching)
VFB =0.9V (No switching)
•
UVIN=3.0V
2X Gain Config., Measure VFB, VCC=5V,
T=25°C
•
mS
Error Amplifier Gain
60
dB
No Load
COMP Sink Current
150
uA
VFB=0.9V, COMP=0.9V
COMP Source Current
150
uA
VFB Input Bias Current
50
200
nA
Internal Pole
4
COMP Clamp
2.5
V
COMP Clamp Temp. Coefficient
-2
mV/°C
Ramp Amplitude
0.92
RAMP Offset
1.10
-2
GH Minimum Pulse Width
90
Maximum Controllable Duty
Ratio
Maximum Duty Ratio
Internal Oscillator Frequency
SS Charge Current:
© 2008 Exar Corporation
92
1.28
V
VFB=0.7V, TA = 25°C
•
TA = 25°C, RAMP COMP until GH starts
switching
V
mV/°C
180
97
100
510
ns
•
%
•
%
600
VFB=0.8V
MHz
1.1
RAMP Offset Temp. Coefficient
VFB=0.7V, COMP=2.2V
•
690
10
kHz
Maximum Duty Ratio Measured just
before pulse skipping begins
Valid for 20 Cycles
•
uA
2/15
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
Parameter
SS Discharge Current:
Short Circuit Threshold Voltage
Hiccup Timeout
Min.
Typ.
Max.
1
0.20
0.25
0.30
Units
Conditions
mA
•
V
•
Fault Present, SS = 0.2V
Measured VREF (0.8V) - VFB
100
ms
VFB = 0.5V
Number of Allowable Clock
Cycles at 100% Duty Cycle
20
Cycles
VFB = 0.7V
Minimum GL Pulse After 20
Cycles
0.5
Cycle
VFB = 0.7V
Thermal Shutdown Temperature
145
°C
Thermal Hysteresis
10
GH & GL Rise Times
35
°C
50
ns
•
Measured 10% to 90%
GH & GL Fall Times
30
40
ns
•
Measured 90% to 10%
GL to GH Non Overlap Time
45
70
ns
•
GH & GL Measured at 2.0V
SWN to GL Non Overlap Time
25
40
ns
•
Measured SWN = 100mV to GL = 2.0V
GH & GL Pull Down Resistance
50
Ω
Driver Pull Down Resistance
1.5
Ω
Driver Pull Up Resistance
2.5
Ω
BLOCK DIAGRAM
Fig. 2:SP6134H Block Diagram
© 2008 Exar Corporation
3/15
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
PIN ASSIGNEMENT
Fig. 3: SP6134H Pin Assignment
PIN DESCRIPTION
Name
Pin Number
Description
VCC
1
Bias Supply Input. Connect to external 5V supply. Used to power internal circuits and
low side gate driver.
GL
2
High current driver output for the low side NFET switch. It is always low if GH is high or
during a fault. Resistor pull down ensure low state at low voltage.
GND
3
Ground Pin. The control circuitry of the IC and lower power driver are referenced to this
pin. Return separately from other ground traces to the (-) terminal of COUT.
VFB
4
Feedback Voltage and Short Circuit Detection pin. It is the inverting input of the Error
Amplifier and serves as the output voltage feedback point for the Buck Converter. The
output voltage is sensed and can be adjusted through an external resistor divider.
Whenever VFB drops 0.25V below the positive reference, a short circuit fault is detected
and the IC enters hiccup mode.
COMP
5
Output of the Error Amplifier. It is internally connected to the inverting input of the
PWM comparator. An optimal filter combination is chosen and connected to this pin and
either ground or VFB to stabilize the voltage mode loop.
UVIN
6
UVLO input for VIN voltage. Connect a resistor divider between VIN and UVIN to set
minimum operating voltage.
SS
7
Soft Start. Connect an external capacitor between SS and GND to set the soft start rate
based on the 10μA source current. The SS pin is held low via a 1mA (min) current
during all fault conditions.
SWN
8
Lower supply rail for the GH high-side gate driver. Connect this pin to the switching
node at the junction between the two external power MOSFET transistors.
GH
9
High current driver output for the high side NFET switch. It is always low if GL is high or
during a fault. Resistor pull down ensure low state at low voltage.
BST
10
© 2008 Exar Corporation
High side driver supply pin. Connect BST to the external boost diode and capacitor as
shown in the Typical Application Circuit on page 1. High side driver is connected
between BST pin and SWN pin.
4/15
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
ORDERING INFORMATION
Part Number
Temperature
Range
Marking
Package
Packing
Quantity
Note 1
SP6134HCU-L
0°C≤TA≤+70°C
SP6134HCU
CXXX
10 Pin MSOP
YWW
Bulk
Lead free
SP6134HCU-L/TR
0°C≤TA≤+70°C
SP6134HCU
CXXX
10 Pin MSOP
YWW
Tape & Reel
Lead free
SP6134HEU-L
-40°C≤TA≤+85°C
SP6134HEU
EXXX
10 Pin MSOP
YWW
Bulk
Lead free
SP6134HEU-L/TR
-40°C≤TA≤+85°C
SP6134HEU
EXXX
10 Pin MSOP
YWW
Tape & Reel
Lead free
SP6134HVLEDEB
Note 2
SP6134H LED Evaluation Board
“YY” = Year – “WW” = Work Week – “XXX” = Lot Number
© 2008 Exar Corporation
5/15
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
THEORY OF OPERATION
SOFT START
“Soft Start” is achieved when a power
converter ramps up the output voltage while
controlling the magnitude of the input supply
source current. In a modern step down
converter, ramping up the positive terminal of
the error amplifier controls soft start. As a
result, excess source current can be defined as
the current required to charge the output
capacitor.
GENERAL OVERVIEW
The SP6134H is a fixed frequency, voltage
mode, synchronous PWM controller optimized
for high efficiency. The part has been designed
to be especially attractive for split plane
applications utilizing 5V to power the controller
and 3V to 28V for step down conversion. The
heart of the SP6134H is a wide bandwidth
transconductance
amplifier
designed
to
accommodate
Type
II
and
Type
III
compensation schemes. A precision 0.8V
reference present on the positive terminal of
the error amplifier permits the programming
of the output voltage down to 0.8V via the VFB
pin. The output of the error amplifier, COMP,
compared to a 1.1V peak-to-peak ramp is
responsible for trailing edge PWM control. This
voltage ramp and PWM control logic are
governed by the internal oscillator that
accurately sets the PWM frequency to 300kHz.
The SP6134H contains two unique control
features that are very powerful in distributed
applications. First, Non-synchronous driver
control is enabled during start up to prohibit
the low side NFET from pulling down the
output until the high side NFET has attempted
to turn on. Second, a 100% duty cycle timeout
ensures that the low side NFET is periodically
enhanced during extended periods at 100%
duty cycle. This guarantees the synchronized
refreshing of the BST capacitor during very
large duty ratios. The SP6134H also contains a
number of valuable protection features. A
programmable input (VIN) UVLO allows a user
to set the exact value at which the conversion
voltage is at a safe point to begin down
conversion, and an internal VCC UVLO ensures
that the controller itself has enough voltage to
properly operate. Other protection features
include thermal shutdown and short-circuit
detection. In the event that either a thermal,
short-circuit, or UVLO fault is detected, the
SP6134H is forced into an idle state where the
output drivers are held off for a finite period
before a re-start is attempted.
© 2008 Exar Corporation
IVIN = COUT * DVOUT / DTSoft-start
The SP6134H provides the user with the
option to program the soft start rate by tying
a capacitor from the SS pin to GND. The
selection of this capacitor is based on the
10uA pull up current present at the SS pin and
the 0.8V reference voltage. Therefore, the
excess source can be redefined as:
IVIN = COUT * DVOUT *10μA / (CSS * 0.8V)
UNDER VOLTAGE LOCK OUT (UVLO)
The SP6134H contains two separate UVLO
comparators to monitor the bias (VCC) and
conversion (VIN) voltages independently. The
VCC UVLO threshold is internally set to 4.25V,
whereas
the
VIN
UVLO
threshold
is
programmable through the UVIN pin. When
the UVIN pin is greater than 2.5V, the
SP6134H is permitted to start up pending the
removal of all other faults. Both the VCC and
VIN UVLO comparators have been designed
with hysteresis to prevent noise from resetting
a fault.
THERMAL AND SHORT-CIRCUIT PROTECTION
Because the SP6134H is designed to drive
large NFETs running at high current, there is a
chance that either the controller or power
converter will become too hot. Therefore, an
internal thermal shutdown (145°C) has been
included to prevent the IC from malfunctioning
at extreme temperatures.
A short-circuit detection comparator has also
been included in the SP6134H to protect
against the accidental short or sever build up
of current at the output of the power
converter.
This
comparator
constantly
6/15
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
monitors the positive and negative terminals
of the error amplifier, and if the VFB pin ever
falls more than 250mV (typical) below the
positive reference, a short-circuit fault is set.
Because the SS pin overrides the internal 0.8V
reference during soft start, the SP6134H is
capable of detecting short-circuit faults
throughout the duration of soft start as well as
in regular operation.
capability. The voltage loop also includes two
other very important features. One is an Nonsynchronous start up mode. Basically, the GL
driver can not turn on unless the GH driver
has attempted to turn on or the SS pin has
exceeded 1.7V. This feature prevents the
controller from “dragging down” the output
voltage during startup or in fault modes. The
second feature is a 100% duty cycle timeout
that ensures synchronized refreshing of the
BST capacitor at very high duty ratios. In the
event that the GH driver is on for 20
continuous clock cycles, a reset is given to the
PWM flip flop half way through the 21st cycle.
This forces GL to rise for the remainder of the
cycle, in turn refreshing the BST capacitor.
HANDLING OF FAULTS
Upon the detection of power (UVLO), thermal,
or short-circuit faults, the SP6134H is forced
into an idle state where the SS and COMP pins
are pulled low and the gate drivers are held
off. In the event of UVLO fault, the SP6134H
remains in this idle state until the UVLO fault
is removed. Upon the detection of a thermal or
short-circuit fault, an internal 200ms (typical)
timer is activated. In the event of a shortcircuit
fault,
a
restart
is
attempted
immediately after the 200ms timeout expires.
Whereas, when a thermal fault is detected the
200ms delay continuously recycles and a
restart cannot be attempted until the thermal
fault is removed and the timer expires.
GATE DRIVERS
The SP6134H contains a pair of powerful 2Ω
SOURCE and 1.5Ω SINK drivers. These state
of the art drivers are designed to drive
external NFETs capable of handling up to 30A.
Rise, fall, and non-overlap times have all been
minimized to achieve maximum efficiency. All
drive pins GH, GL & SWN are monitored
continuously to ensure that only one external
NFET is ever on at any given time.
ERROR AMPLIFIER AND VOLTAGE LOOP
As stated before, the heart of the SP6134H
voltage error loop is a high performance, wide
bandwidth
transconductance
amplifier.
Because of the amplifier’s current limited
(±150μA) transconductance, there are many
ways to compensate the voltage loop or to
control the COMP pin externally. A simple,
single pole, single zero compensation can be a
RC to ground. However Exar recommends a
Type II or Type III compensation which
eliminates the gm of the amplifier from the
control loop equations.
The amplifier has
enough bandwidth (45° at 4 MHz) and enough
gain (60dB) to run Type III compensation
schemes with adequate gain and phase
margins at cross over frequencies greater than
50kHz.
The common mode output of the error
amplifier is 0.9V to 2.2V. Therefore, the PWM
voltage ramp has been set between 1.1V and
2.2V to ensure proper 0% to 100% duty cycle
© 2008 Exar Corporation
7/15
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
requirements. The core must be large enough
not to saturate at the peak inductor current
and provide low core loss at the high switching
frequency. Low cost powdered iron cores have
a gradual saturation characteristic but can
introduce considerable ac core loss, especially
when the inductor value is relatively low and
the ripple current is high. Ferrite materials, on
the other hand, are more expensive and have
an abrupt saturation characteristic with the
inductance dropping sharply when the peak
design current is exceeded. Nevertheless, they
are preferred at high switching frequencies
because they present very low core loss and
the design only needs to prevent saturation.
In general, ferrite or molypermalloy materials
are better choice for all but the most cost
sensitive applications. The power dissipated in
the inductor is equal to the sum of the core
and copper losses. To minimize copper losses,
the winding resistance needs to be minimized,
but this usually comes at the expense of a
larger inductor. Core losses have a more
significant contribution at low output current
where the copper losses are at a minimum,
and can typically be neglected at higher output
currents where the copper losses dominate.
Core loss information is usually available from
the magnetic vendor.
APPLICATIONS INFORMATION
INDUCTOR SELECTION
There are many factors to consider in selecting
the inductor including cost, efficiency, size and
EMI. In a typical SP6134H circuit, the inductor
is chosen primarily for value, saturation
current and DC resistance. Increasing the
inductor value will decrease output voltage
ripple, but degrade transient response. Low
inductor values provide the smallest size, but
cause large ripple currents, poor efficiency and
more output capacitance to smooth out the
larger ripple current. The inductor must also
be able to handle the peak current at the
switching frequency without saturating, and
the copper resistance in the winding should be
kept as low as possible to minimize resistive
power loss. A good compromise between size,
loss and cost is to set the inductor ripple
current to be within 20% to 40% of the
maximum output current.
The switching frequency and the inductor
operating point determine the inductor value
as follows:
L=
VOUT (VIN (max ) − VOUT )
The copper loss in the inductor can be
calculated using the following equation:
VIN (max ) FS K r I OUT (max )
PL (Cu ) = I 2 L ( RMS ) RWINDING
where:
where IL(RMS) is the RMS inductor current
that can be calculated as follows:
Fs = switching frequency Kr = ratio of the ac
inductor ripple current to the maximum output
current
I L ( RMS ) − I OUT (max )
The peak to peak inductor ripple current is:
L=
VOUT (VIN (max ) − VOUT )
VIN (max ) FS L
I PEAK = I OUT (max ) +
2
OUTPUT CAPACITOR SELECTION
The
required
ESR
(Equivalent
Series
Resistance)
and
capacitance
drive
the
selection of the type and quantity of the
output capacitors. The ESR must be small
enough that both the resistive voltage
deviation due to a step change in the load
current and the output ripple voltage do not
I PP
2
Once the required inductor value is selected,
the proper selection of core material is based
on peak inductor current and efficiency
© 2008 Exar Corporation
1 ⎛ I PP ⎞⎟
1+ ⎜
3 ⎜⎝ I OUT (max ) ⎟⎠
8/15
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
exceed the tolerance limits expected on the
output voltage. During an output load
transient, the output capacitor must supply all
the additional current demanded by the load
until the SP6134HCU adjusts the inductor
current to the new value.
INPUT CAPACITOR SELECTION
The input capacitor should be selected for
ripple current rating, capacitance and voltage
rating. The input capacitor must meet the
ripple current requirement imposed by the
switching current. In continuous conduction
mode, the source current of the high-side
MOSFET is approximately a square wave of
duty cycle VOUT/VIN. Most of this current is
supplied by the input bypass capacitors. The
RMS value of input capacitor current is
determined at the maximum output current
and under the assumption that the peak to
peak inductor ripple current is low, it is given
by:
Therefore the capacitance must be large
enough so that the output voltage is help up
while the inductor current ramps up or down
to the value corresponding to the new load
current. Additionally, the ESR in the output
capacitor causes a step in the output voltage
equal to the current. Because of the fast
transient response and inherent 100% and 0%
duty cycle capability provided by the
SP6134HCU when exposed to output load
transient, the output capacitor is typically
chosen for ESR, not for capacitance value.
I CIN (rms ) = I OUT (max ) D(1 − D )
The worse case occurs when the duty cycle D
is 50% and gives an RMS current value equal
to IOUT/2.
The output capacitor’s ESR, combined with the
inductor ripple current, is typically the main
contributor to output voltage ripple. The
maximum allowable ESR required to maintain
a specified output voltage ripple can be
calculated by:
RESR ≤
Select input capacitors with adequate ripple
current rating to ensure reliable operation. The
power dissipated in the input capacitor is:
PCIN = I 2 CIN (rms ) RESR (CIN )
ΔVOUT
I PP
This can become a significant part of power
losses in a converter and hurt the overall
energy transfer efficiency. The input voltage
ripple primarily depends on the input capacitor
ESR and capacitance. Ignoring the inductor
ripple current, the input voltage ripple can be
determined by:
ΔVOUT = Peak to Peak Output Voltage Ripple
IPP = Peak to Peak Inductor Ripple Current
The total output ripple is a combination of the
ΔVIN = I OUT (max ) RESR (CIN ) +
ESR and the output capacitance value and can
be calculated as follows:
Where:
FS = Switching Frequency
D = Duty Cycle
COUT = Output Capacitance Value
© 2008 Exar Corporation
FS C IN VIN
2
The capacitor type suitable for the output
capacitors can also be used for the input
capacitors. However, exercise extra caution
when tantalum capacitors are considered.
Tantalum
capacitors
are
known
for
catastrophic failure when exposed to surge
current, and input capacitors are prone to
such surge current when power supplies are
connected “live” to low impedance power
sources.
⎛ I (1 − D ) ⎞
⎟⎟ + (I PP RESR )2
= ⎜⎜ PP
⎝ COUT FS ⎠
2
ΔVOUT
I OUT (max )VOUT (VIN − VOUT )
9/15
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
lowering the RDS(ON) of the MOSFETs always
improves efficiency even though it gives rise
to higher switching losses due to increased
Crss.
MOSFET SELECTION
The losses associated with MOSFETs can be
divided into conduction and switching losses.
Conduction losses are related to the on
resistance of MOSFETs, and increase with the
load current. Switching losses occur on each
on/off
transition
when
the
MOSFETs
experience both high current and voltage.
Since the bottom MOSFET switches current
from/to a paralleled diode (either its own body
diode or a Schottky diode), the voltage across
the MOSFET is no more than 1V during
switching transition. As a result, its switching
losses are negligible. The switching losses are
difficult to quantify due to all the variables
affecting turn on/ off time. However, the
following equation provides an approximation
on the switching losses associated with the top
MOSFET driven by SP6134H.
Top and bottom MOSFETs experience unequal
conduction losses if their on time is unequal.
For applications running at large or small duty
cycle, it makes sense to use different top and
bottom
MOSFETs.
Alternatively,
parallel
multiple MOSFETs to conduct large duty
factor.
RDS(ON) varies greatly with the gate driver
voltage. The MOSFET vendors often specify
RDS(ON) on multiple gate to source voltages
(VGS), as well as provide typical curve of
RDS(ON) versus VGS. For 5V input, use the
RDS(ON) specified at 4.5V VGS. At the time of
this publication, vendors, such as Fairchild,
Siliconix and International Rectifier, have
started to specify RDS(ON) at VGS less than
3V. This has provided necessary data for
designs in which these MOSFETs are driven
with 3.3V and made it possible to use
SP6134H in 3.3V only applications.
PSH (max ) = 12C rssVIN (max ) I OUT (max ) FS
where
Crss = reverse transfer capacitance of the top
MOSFET
Thermal calculation must be conducted to
ensure the MOSFET can handle the maximum
load current. The junction temperature of the
MOSFET, determined as follows, must stay
below the maximum rating.
Switching losses need to be taken into account
for high switching frequency, since they are
directly proportional to switching frequency.
The conduction losses associated with top and
bottom MOSFETs are determined by:
TJ (max ) = TA(max ) +
2
PCH (max ) = RDS (ON ) I OUT (max ) D
PCL (max ) = RDS (ON ) I OUT (max ) (1 − D )
PMOSFET (max )
RθJA
2
where
where
PCH(max) = conduction losses of the high side
MOSFET
TA(max) = maximum ambient temperature
PMOSFET(max) = maximum power dissipation
of the MOSFET
PCL(max) = conduction losses of the low side
MOSFET
RDS(ON) = drain to source on resistance.
RΘJA = junction
resistance.
The total power losses of the top MOSFET are
the sum of switching and conduction losses.
For synchronous buck converters of efficiency
over 90%, allow no more than 4% power
losses for high or low side MOSFETs. For input
voltages of 3.3V and 5V, conduction losses
often dominate switching losses. Therefore,
RΘJA of the device depends greatly on the
board layout, as well as device package.
Significant thermal improvement can be
achieved in the maximum power dissipation
through the proper design of copper mounting
pads on the circuit board. For example, in a
SO-8 package, placing two 0.04 square inches
© 2008 Exar Corporation
10/15
to
ambient
thermal
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
copper pad directly under the package,
without occupying additional board space, can
increase
the
maximum
power
from
approximately 1 to 1.2W. For DPAK package,
enlarging the tap mounting pad to 1 square
inches reduces the RΘJA from 96°C/W to
40°C/W.
LOOP COMPENSATION DESIGN
The open loop gain of the whole system can
be divided into the gain of the error amplifier,
PWM modulator, buck converter output stage,
and feedback resistor divider. In order to
crossover at the selected frequency FCO, the
gain of the error amplifier has to compensate
for the attenuation caused by the rest of the
loop at this frequency. The goal of loop
compensation is to manipulate loop frequency
response such that its gain crosses over 0db
at a slope of -20db/dec. The first step of
compensation design is to pick the loop
crossover frequency. High crossover frequency
is desirable for fast transient response, but
often jeopardizes the system stability.
Crossover frequency should be higher than the
ESR zero but less than 1/5 of the switching
frequency. The ESR zero is contributed by the
ESR associated with the output capacitors and
can be determined by:
SCHOTTKY DIODE SELECTION
When paralleled with the bottom MOSFET, an
optional Schottky diode can improve efficiency
and reduce noises. Without this Schottky
diode, the body diode of the bottom MOSFET
conducts the current during the non-overlap
time when both MOSFETs are turned off.
Unfortunately, the body diode has high
forward voltage and reverse recovery problem.
The reverse recovery of the body diode causes
additional switching noises when the diode
turns off. The Schottky diode alleviates these
noises and additionally improves efficiency
thanks to its low forward voltage. The reverse
voltage across the diode is equal to input
voltage, and the diode must be able to handle
the peak current equal to the maximum load
current.
f Z ( ESR ) =
2πC OUT R ESR
The next step is to calculated the complex
conjugate poles contributed by the LC output
filter,
The power dissipation of the Schottky diode is
determined by
f P ( LC ) =
PDiode = 2VF I OUT TNOL FS
where
1
2π LC
When the output capacitors are of a Ceramic
Type, the SP6134HCU Evaluation Board
requires a Type III compensation circuit to
give a phase boost of 180° in order to
counteract the effects of an under damped
resonance of the output filter at the double
pole frequency.
TNOL = non-overlap time between GH and GL.
VF = forward voltage of the Schottky diode.
© 2008 Exar Corporation
1
11/15
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
SP6134H Voltage Mode Control Loop with Loop Dynamic
Definitions:
Resr = Output Capacitor Equivalent Series Resistance
Rdc = Output Inductor DC Resistance
Vramp_pp = SP6134H internal RAMP Amplitude Peak to Peak Voltage
Conditions:
Cz2 >> Cp1 and R1 >> Rz3
Output Load Resistance >> Resr and Rdc
Bode Plot of Type III Error Amplifier Compensation.
© 2008 Exar Corporation
12/15
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
TYPICAL APPLICATION DIAGRAM
VIN
3.5V - 15V
C1
22μF
16V
8 7 6 5
C2
22μF
16V
VCC = 5V @ 30mA
FDS6676S
14.5A, 6mΩ
QT
4
RLF
3.0,5%
GND
C1, C2
Ceramic
1210
X5R
1 2 3
CBST
1μF
U1
DBST
MBR0530
SP6134H
1
R3
221k, 1%
CVCC
10μF
6.3V
GND 3
2
3
4
0.8V
5
UVIN
R4
100k, 1%
CVCC
Ceramic
8050
X5R
VCC
BST 10
R5
Bead
GH 9
GL
GND
SWN 8
VFB
SS 7
COMP
CZ2
SS
UVIN 6
RZ2
C3
47μF
6.3V
QB
4
CSS
47nF
1 2 3
CZ3
220pF
R1
68.1k, 1%
GND2
C3, C4
Ceramic
1210
X5R
56pF
fs=600Khz
C4
47μF
6.3V
FDS6676S
14.5A, 6.0mΩ
820pF 40.2k, 1%
CP1
CF1
100pF
VOUT≤VIN
0.8V - 3.3V
0 - 10 A
RZ3
4.64k, 1%
8 7 6 5
V
R2
21.5k, 1%
=(R1/R2 +1)V
FB
OUT
Note: Components highlighted in bold are those used on the SP6134H Evaluation Board.
Table 1. Input and Output Stage Components Selection Charts.
© 2008 Exar Corporation
13/15
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
PACKAGE SPECIFICATION
© 2008 Exar Corporation
MSOP-10
14/15
Rev. 2.0.0
SP6134H
High Voltage, 600kHz Synchronous PWM Step Down
Controller
REVISION HISTORY – TO BE DELETED PRIOR TO PUBLICATION Revision
2.0.0
Date
1/20/2009
Description
Complete re-formatting
Changes from PCN #09-0120-01
FOR FURTHER ASSISTANCE
Email:
[email protected]
Exar Technical Documentation:
http://www.exar.com/TechDoc/default.aspx?
EXAR CORPORATION
HEADQUARTERS AND SALES OFFICES
48720 Kato Road
Fremont, CA 94538 – USA
Tel.: +1 (510) 668-7000
Fax: +1 (510) 668-7030
www.exar.com
NOTICE
EXAR Corporation reserves the right to make changes to the products contained in this publication in order to improve
design, performance or reliability. EXAR Corporation assumes no responsibility for the use of any circuits described herein,
conveys no license under any patent or other right, and makes no representation that the circuits are free of patent
infringement. Charts and schedules contained here in are only for illustration purposes and may vary depending upon a
user’s specific application. While the information in this publication has been carefully checked; no responsibility, however,
is assumed for inaccuracies.
EXAR Corporation does not recommend the use of any of its products in life support applications where the failure
malfunction of the product can reasonably be expected to cause failure of the life support system or to significantly affect
safety or effectiveness. Products are not authorized for use in such applications unless EXAR Corporation receives,
writing, assurances to its satisfaction that: (a) the risk of injury or damage has been minimized; (b) the user assumes
such risks; (c) potential liability of EXAR Corporation is adequately protected under the circumstances.
or
its
in
all
Reproduction, in part or whole, without the prior written consent of EXAR Corporation is prohibited.
© 2008 Exar Corporation
15/15
Rev. 2.0.0