STMICROELECTRONICS PM6680

PM6680
No Rsense dual step-down controller with adjustable voltages
for notebook system power
Features
■
6 V to 28 V input voltage range
■
Adjustable output voltages
■
5V always voltage available deliver 100 mA
peak current
■
1.237 V ± 1% reference voltage available
■
Lossless current sensing using low side
MOSFETs RDS(on)
■
Negative current limit
■
Soft-start internally fixed at 2ms
■
Soft output discharge
■
Latched OVP and UVP
■
Selectable pulse skipping at light loads
■
Selectable minimum frequency (33 kHz) in
pulse skip mode
■
4mW maximum quiescent power
■
Independent power good signals
■
Output voltage ripple compensation
Applications
■
Notebook computers
■
Tablet PC or slates
■
Mobile system power supply
■
3-4 cells Li+ battery powered devices
Table 1.
VFQFPN-32 5X5
Description
PM6680 is a dual step-down controller
specifically designed to provide extremely high
efficiency conversion, with lossless current
sensing technique. The constant on-time
architecture assures fast load transient response
and the embedded voltage feed-forward provides
nearly constant switching frequency operation. An
embedded integrator control loop compensates
the DC voltage error due to the output ripple.
Pulse skipping technique increases efficiency at
very light load. Moreover a minimum switching
frequency of 33kHz is selectable to avoid audio
noise issues. The PM6680 provides a selectable
switching frequency, allowing three different
values of switching frequencies for the two
switching sections. The output voltages OUT1
and OUT2 can be adjusted from 0.9 V to 5.5 V
and from 0.9 V to 3.3 V respectively.
Device summary
Order codes
Package
Packaging
PM6680
VFQFPN-32 5mm x 5mm (Exposed pad)
Tube
PM6680TR
VFQFPN-32 5mm x 5mm (Exposed pad)
Tape and reel
August 2007
Rev 6
1/49
www.st.com
49
Contents
PM6680
Contents
1
Simplified application schematic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 4
2
Electrical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
3
2.1
Maximum rating . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
2.2
Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5
Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
3.1
Connections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
3.2
Functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
4
Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
5
Typical operating characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
6
Block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
7
Device description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
2/49
7.1
Constant On time PWM control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
7.2
Constant on time architecture . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
7.3
Output ripple compensation and loop stability . . . . . . . . . . . . . . . . . . . . . 22
7.4
Pulse skip mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
7.5
No-audible skip mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
7.6
Current limit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
7.7
Soft start and soft end . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
7.8
Gate drivers . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
7.9
Reference voltage and bandgap . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
7.10
Internal linear regulator . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
7.11
Power up sequencing and operative modes . . . . . . . . . . . . . . . . . . . . . . . 28
7.12
Monitoring and protections . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
7.12.1
Power good signals . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
7.12.2
Thermal protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
7.12.3
Overvoltage protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
7.12.4
Undervoltage protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
PM6680
Contents
7.13
Design guidelines . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
7.13.1
Switching frequency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
7.13.2
Inductor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
7.13.3
Output capacitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
7.13.4
Input capacitors selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 32
7.13.5
Power MOSFETs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
7.13.6
Closing the integrator loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
7.13.7
Other parts design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
7.13.8
Design example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
8
Package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
9
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
3/49
List of figures
PM6680
List of figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.
Figure 7.
Figure 8.
Figure 9.
Figure 10.
Figure 11.
Figure 12.
Figure 13.
Figure 14.
Figure 15.
Figure 16.
Figure 17.
Figure 18.
Figure 19.
Figure 20.
Figure 21.
Figure 22.
Figure 23.
Figure 24.
Figure 25.
Figure 26.
Figure 27.
Figure 28.
Figure 29.
Figure 30.
Figure 31.
Figure 32.
Figure 33.
Figure 34.
Figure 35.
Figure 36.
Figure 37.
Figure 38.
Figure 39.
Figure 40.
Figure 41.
Figure 42.
4/49
Simplified application schematic . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Pin connection (Through top view) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
1.5V output efficiency vs load current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
1.05V output efficiency vs load current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
PWM no load input battery vs input voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
Skip no load battery current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 15
No-audible skip no load battery current vs input voltage . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Stand-by mode input battery current vs input voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Shutdown mode input battery current vs input voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
1.5V switching frequency vs load current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
1.05V switching frequency vs load current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
LDO5 vs output current. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
1.5V voltage regulation vs load current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
1.05V voltage regulation vs load current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
Voltage reference vs load current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
OUT1, OUT2 and LDO5 power-up . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
1.5V load transient . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
1.05V load transient . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17
1.5V soft start (0.25Ω load). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
1.05V soft start (0.175Ω load). . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
1.5V soft end (No load) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
1.05V soft end (No load) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
1.5V soft end (1Ω Load) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
1.05V soft end (1Ω Load) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
1.5V no-audible skip mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
1.05V no-audible skip mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Functional block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
Constant ON time PWM control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Constant on-time block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
Circuitry for output ripple compensation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
PWM and pulse skip mode inductor current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
No audible skip mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
RDSON sensing technique . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 26
Current waveforms in current limit conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
Soft start waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Circuitry for output ripple compensation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
Virtual ESR network . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 38
VIN pin filter . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
Inductor current waveforms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
Bootstrap circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
Current paths, ground connection and driver traces layout . . . . . . . . . . . . . . . . . . . . . . . . 44
Package dimensions. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47
PM6680
List of tables
List of tables
Table 1.
Table 2.
Table 3.
Table 4.
Table 5.
Table 6.
Table 7.
Table 8.
Table 9.
Table 10.
Table 11.
Table 12.
Table 13.
Table 14.
Table 15.
Table 16.
Table 17.
Table 18.
Table 19.
Device summary . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Absolute maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Thermal data. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Pin functions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 9
Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
FSEL pin selection: typical switching frequency . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
V5SW multifunction pin. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Operatives modes. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
Protections and operatives modes . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
Inductor manufacturer. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Output capacitor manufacturer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Input capacitor manufacturer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
High side MOSFET manufacturer. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
Low side MOSFET manufacturer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Dual MOSFET manufacturer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Shottky diode manufacturer . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
VFQFPN 5x5x1.0 32L Pitch 0.50 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
Exposed pad variations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
Document revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
5/49
OUT1-
1
OUT1+
1
SGND
FB1
VIN
+
PGND
SGND
PGOOD2
PGOOD1
V+
V+
PGND
5
27
26
16
30
29
17
20
15
21
22
23
SGND
SGND
SGND
BOOT2
SHDN
PGOOD2
PGOOD1
SGND2
COMP1
OUT1
V5SW
CSENSE1
LGATE1
PHASE1
HGATE1
BOOT1
SGND
PM6680
EN2
4
EN1
25
V+
31
VCC
V+
32
+
18
LDO5
VREF
19
LGATE2
PHASE2
HGATE2
BOOT2
SGND
FB2
FB1
NC
COMP2
OUT2
SGND
PGND
CSENSE2
VIN
SKIP
6/49
24
7
28 FB1
6
2
8
1
14
12
13
11
10
SGND
9
SGND
PGND
SGND
PGND
PGND
+
VIN
SGND
1
1
O
O
Figure 1.
FSEL
1
3
BOOT1
Simplified application schematic
PM6680
Simplified application schematic
Simplified application schematic
PM6680
Electrical data
2
Electrical data
2.1
Maximum rating
Table 2. Absolute maximum ratings
Parameter
Value
Unit
V5SW, LDO5 to PGND
-0.3 to 6
V
VIN to PGND
-0.3 to 36
V
HGATEx and BOOTx, to PHASEx
-0.3 to 6
V
(1)
V
PHASEx to PGND
-0.6
CSENSEx , to PGND
CSENSEx to BOOTx
LGATEx to PGND
-0.3
FBx, COMPx, SKIP, , FSEL,VREF to SGND1,SGND2
PGND to SGND1,SGND2
SHDN,PGOODx, OUTx, VCC, ENx to SGND1,SGND2
Power dissipation at TA = 25ºC
Maximum withstanding voltage range test condition:
CDF-AEC-Q100-002- “Human Body Model”
acceptance criteria: “Normal Performance”
to36
-0.6 to 42
V
-6 to 0.3
V
(2)
to LDO5 +0.3
V
-0.3 to Vcc +0.3
V
-0.3 to 0.3
V
-0.3 to 6
V
2.8
W
VIN
±1000
Other pins
±2000
V
1. PHASE to PGND up to -2.5 V for t<10 ns
2. LGATEx to PGND up to -1 V for t<40 ns
2.2
Thermal data
Table 3.
Symbol
Thermal data
Parameter
Value
Unit
35
°C/W
RthJA
Thermal resistance junction to ambient
TSTG
Storage temperature range
-40 to 150
°C
Junction operating temperature range
-10 to 125
°C
TJ
7/49
Pin settings
PM6680
3
Pin settings
3.1
Connections
Figure 2.
Pin connection (Through top view)
PM6680
8/49
PM6680
3.2
Pin settings
Functions
Table 4.
N°
Pin functions
Pin
Function
1
SGND1
Signal ground. Reference for internal logic circuitry. It must be connected to
the signal ground plan of the power supply. The signal ground plan and the
power ground plan must be connected together in one point near the PGND
pin.
2
COMP2
DC voltage error compensation pin for the switching section 2
3
FSEL
Frequency selection pin. It provides a selectable switching frequency,
allowing three different values of switching frequencies for the switching
sections.
EN2
Enable input for the switching section 2.
• The section 2 is enabled applying a voltage greater than 2.4V to this pin.
• The section 2 is disabled applying a voltage lower than 0.8V.
When the section is disabled the High Side gate driver goes low and Low
Side gate driver goes high. If both EN1 and EN2 pins are low and SHDN pin
is high the device enters in standby mode.
5
SHDN
Shutdown control input.
• The device switch off if the SHDN voltage is lower than the device off
threshold (Shutdown mode)
• The device switch on if the SHDN voltage is greater than the device on
threshold.
The SHDN pin can be connected to the battery through a voltage divider to
program an undervoltage lockout. In shutdown mode, the gate drivers of the
two switching sections are in high impedance (high-Z).
6
NC
Not connected.
7
FB2
Feedback input for the switching section 2 This pin is connected to a
resistive voltage-divider from OUT2 to PGND to adjust the output voltage
from 0.9V to 3.3V.
8
OUT2
Output voltage sense for the switching section 2.This pin must be directly
connected to the output voltage of the switching section.
9
BOOT2
Bootstrap capacitor connection for the switching section 2. It supplies the
high-side gate driver.
10
HGATE2
High-side gate driver output for section 2. This is the floating gate driver
output.
11
PHASE2
Switch node connection and return path for the high side driver for the
section 2.It is also used as negative current sense input.
4
12
CSENSE2
13
LGATE2
14
PGND
15
LGATE1
Positive current sense input for the switching section 2. This pin must be
connected through a resistor to the drain of the synchronous rectifier
(RDSON sensing) to obtain a positive current limit threshold for the power
supply controller.
Low-side gate driver output for the section 2.
Power ground. This pin must be connected to the power ground plan of the
power supply.
Low-side gate driver output for the section 1.
9/49
Pin settings
PM6680
Table 4.
N°
Pin
Function
16
SGND2
Signal ground for analog circuitry. It must be connected to the signal ground
plan of the power supply.
17
V5SW
Internal 5V regulator bypass connection.
• If V5SW is connected to OUT5 (or to an external 5V supply) and V5SW is
greater than 4.9V, the LDO5 regulator shuts down and the LDO5 pin is
directly connected to OUT5 through a 3Ω (max) switch.
If V5SW is connected to GND, the LDO5 linear regulator is always on.
18
LDO5
5V internal regulator output. It can provide up to 100mA peak current. LDO5
pin supplies embedded low side gate drivers and an external load.
19
VIN
Device supply voltage input and battery voltage sense. A bypass filter (4Ω
and 4.7µF) between the battery and this pin is recommended.
Positive current sense input for the switching section 1. This pin must be
connected through a resistor to the drain of the synchronous rectifier
(RDSON sensing) to obtain a positive current limit threshold for the power
supply controller.
20
CSENSE1
21
PHASE1
Switch node connection and return path for the high side driver for the
section 1.It is also used as negative current sense input.
22
HGATE1
High-side gate driver output for section 1. This is the floating gate driver
output.
23
BOOT1
Bootstrap capacitor connection for the switching section 1. It supplies the
high-side gate driver.
SKIP
Pulse skipping mode control input.
• If the pin is connected to LDO5 the PWM mode is enabled.
• If the pin is connected to GND, the pulse skip mode is enabled.
• If the pin is connected to VREF the pulse skip mode is enabled but the
switching frequency is kept higher than 33KHz (No-audible pulse skip
mode).
25
EN1
Enable input for the switching section 1.
• The section 1 is enabled applying a voltage greater than 2.4V to this pin.
• The section 1 is disabled applying a voltage lower than 0.8V.
When the section is disabled the High Side gate driver goes low and Low
Side gate driver goes high.
26
PGOOD1
Power Good output signal for the section 1. This pin is an open drain output
and when the output of the switching section 1 is out of +/- 10% of its
nominal value.It is pulled down.
27
PGOOD2
Power Good output signal for the section 2. This pin is an open drain output
and when the output of the switching section 2 is out of +/- 10% of its
nominal value.It is pulled down.
28
FB1
24
10/49
Pin functions (continued)
Feedback input for the switching section 1. This pin is connected to a
resistive voltage-divider from OUT1 to PGND to adjust the output voltage
from 0.9V to 5.5V.
PM6680
Pin settings
Table 4.
Pin functions (continued)
N°
Pin
Function
29
OUT1
30
COMP1
31
VCC
Device Supply Voltage pin. It supplies all the internal analog circuitry except
the gate drivers (see LDO5). Connect this pin to LDO5.
32
VREF
Internal 1.237V high accuracy voltage reference. It can deliver 50uA.
Bypass to SGND with a 100nF capacitor to reduce noise.
Output voltage sense for the switching section 1.This pin must be directly
connected to the output voltage of the switching section.
DC voltage error compensation pin for the switching section 1.
11/49
Electrical characteristics
4
PM6680
Electrical characteristics
VIN = 12V, TA = 0°C to 85°C, unless otherwise specified
Table 5.
Electrical characteristics
Symbol
Parameter
Test condition
Min
Typ
Max
Unit
5.5
28
V
4.5
5.5
V
4.9
V
Supply section
VIN
Input voltage range
Vcc
IC supply voltage
Vout = Vref, LDO5 in
regulation
Turn-on voltage threshold
VV5SW
4.8
Turn-off voltage threshold
4.6
4.75
V
Hysteresis
20
50
mV
VV5SW
Maximum operating range
Rdson
LDO5 internal bootstrap
switch resistance
V5SW > 4.9
OUTx,OUTx discharge-mode
on-resistance
OUTx,OUTx discharge-mode
synchronous rectifier
Turn-on level
0.2
5.5
V
1.8
3
Ω
18
25
Ω
0.36
0.6
V
4
mW
Pin
Operating power consumption
FBx > VREF, Vref in
regulation, V5WS to 5V
Ish
Operating current sunk by VIN
SHDN connected to GND,
14
18
µA
Isb
Operating current sunk by VIN
ENx to GND, V5SW to GND
190
250
µA
Shutdown section
VSHDN
Device On threshold
1.2
1.5
1.7
V
Device Off threshold
0.8
0.85
0.9
V
3.5
ms
110
µA
Soft start section
Soft start ramp time
2
Current limit and zero crossing comparator
ICSENSE
12/49
Input bias current limit
90
100
Comparator offset
VCSENSE-VPGND
-6
6
mV
Zero crossing comparator offset
VPGND - VPHASE
-1
11
mV
Fixed negative current limit
threshold
VPGND - VPHASE
-120
mV
PM6680
Table 5.
Electrical characteristics
Electrical characteristics (continued)
Symbol
Parameter
Test condition
Min
Typ
Max
Unit
OUT1=1.5V
550
650
750
OUT2=1.05V
230
270
315
OUT1=1.5V
375
445
515
OUT2=1.05V
175
210
245
OUT1=1.5V
285
340
395
OUT2=1.05V
125
150
175
350
500
ns
1.236
1.249
V
4
mV
0.95
mV
909
mV
On time pulse width
FSEL to
GND
Ton
On time duration
FSEL to
VREF
FSEL to
LDO5
ns
OFF time
TOFFMIN
Minimum off time
Voltage reference
VREF
Voltage accuracy
4V < VLDO5 < 5.5V
Load regulation
-100µA< IREF < 100µA
Undervoltage lockout fault
threshold
Falling edge of REF
1.224
-4
Integrator
FB
Voltage accuracy
FB
Input bias current
-909
900
0.1
Normal mode
250
Pulse skip mode
60
µA
Over voltage clamp
COMP
Under voltage clamp
mV
-150
Line regulation
Both SMPS, 6V<Vin<28V (1)
0.004
%/V
LDO5 linear regulator
LDO5 linear output voltage
6V<VIN<28V,
0 < ILDO5 < 50mA
LDO5 line regulation
6V< VIN < 28V,
ILDO5 = 20mA ,
ILDO5
LDO5 Current limit
VLDO5 > UVLO
ULVO
Under Voltage Lockout of LDO5
VLDO5
4.9
5.0
5.1
V
0.004
%/V
270
330
400
mA
3.94
4
4.13
V
13/49
Electrical characteristics
Table 5.
PM6680
Electrical characteristics (continued)
Symbol
Parameter
Test condition
Min
Typ
Max
Unit
HGATEx high state(pullup)
2.0
3
Ω
HGATEx low state
(pulldown)
1.6
2.7
Ω
LGATEx high state(pullup)
1.4
2.1
Ω
LGATEx low state
(pulldown)
0.8
1.2
Ω
112
116
120
%
65
68
71
%
Upper threshold
(VFB-VREF)
107
110
113
%
Lower threshold
(VFB-VREF)
88
91
94
%
1
uA
250
mV
High and low gate drivers
HGATE driver on-resistance
LGATE driver on-resistance
PGOOD pins UVP/OVP protections
OVP
Over voltage threshold
UVP
Under voltage threshold
Both SMPS sections with
respect to VREF.
PGOOD1,2
IPGOOD1,2
PGOOD leakage current
VPGOOD1,2 Output low voltage
VPGOOD1,2 forced to 5.5V
ISink = 4mA
150
Thermal shutdown
TSDN
Shutdown temperature
150
°C
Power management pins
SMPS disabled level
0.8
EN1,2
V
SMPS enabled level
2.4
Low level (2)
FSEL
Frequency selection range
Middle level
High level
SKIP
(2)
1.0
Pulse skip mode
PWM mode
(2)
1.0
Ultrasonic mode
(2)
VLDO5-0.8
1. by design
2. by demoboard test
VLDO5-1.5
V
VLDO5-0.8
(2)
Input leakage current
14/49
0.5
(2)
0.5
VLDO5-1.5
VEN1,2= 0 to 5V
1
VSKIP= 0 to 5V
1
VSHDN= 0 to 5V
1
VFSEL= 0 to 5V
1
V
µA
PM6680
5
Typical operating characteristics
Typical operating characteristics
FSEL = GND (200/300KHz), SKIP = GND (skip mode), V5SW = V5SW=EXT5V
(external 5V power supply connected), input voltage VIN = 12V, SHDN, EN1 and EN2 high,
OUT1 = 1.5V, OUT2 = 1.05V, no load unless specified)
Figure 3.
1.5V output efficiency vs
load current
Figure 4.
Figure 5.
PWM no load input battery vs Figure 6.
input voltage
1.05V output efficiency vs
load current
Skip no load battery current
15/49
Typical operating characteristics
PM6680
Figure 7.
No-audible skip no load
battery current vs input
voltage
Figure 8.
Stand-by mode input battery
current vs input voltage
Figure 9.
Shutdown mode input battery Figure 10. 1.5V switching frequency vs
current vs input voltage
load current
Figure 11. 1.05V switching frequency vs Figure 12. LDO5 vs output current
load current
16/49
PM6680
Typical operating characteristics
Figure 13. 1.5V voltage regulation
vs load current
Figure 14. 1.05V voltage regulation
vs load current
Figure 15. Voltage reference vs
load current
Figure 16. OUT1, OUT2 and LDO5
power-up
Figure 17. 1.5V load transient
Figure 18. 1.05V load transient
17/49
Typical operating characteristics
18/49
PM6680
Figure 19. 1.5V soft start (0.25Ω load)
Figure 20. 1.05V soft start (0.175Ω load)
Figure 21. 1.5V soft end (No load)
Figure 22. 1.05V soft end (No load)
Figure 23. 1.5V soft end (1Ω Load)
Figure 24. 1.05V soft end (1Ω Load)
PM6680
Typical operating characteristics
Figure 25. 1.5V no-audible skip mode
Figure 26. 1.05V no-audible skip mode
19/49
Block diagram
6
PM6680
Block diagram
Figure 27. Functional block diagram
VIN
VCC
REFERENCE
GENERATOR
VREF
5V
LINEAR
REGULATOR
+
-
4V
UVLO
LDO5
VREF
LDO5 ENABLE
4.8V
NC
+
UVLO
FB2
V5SW
OUT2
FB1
VCC
SKIP
FSEL
FREQUENCY
SELECTOR
OUT1
BOOT1
BOOT2
LEVEL
SHIFTER
HGATE2
PHASE2
OUT2
OUT1
SMPS
SMPS
CONTROLLER
CONTROLLER
LEVEL
SHIFTER
HGATE1
PHASE1
CSENSE1
CSENSE2
COMP1
COMP2
LDO5
LDO5
LGATE2
LGATE1
PGOOD2
PGOOD1
SHDN
LDO5 ENABLE
STARTUP
CONTROLLER
EN1
EN2
UVLO
20/49
TERMIC
FAULT
TERMIC
CONTROLLER
PM6680
7
Device description
Device description
The PM6680 is a dual step-down controller dedicated to provide logic voltages for notebook
computers.
It is based on a Constant On Time control architecture. This type of control offers a very fast
load transient response with a minimum external component count. A typical application
circuit is shown in Figure 27.
The PM6680 regulates two adjustable output voltages: OUT1 and OUT2. The switching
frequency of the two sections can be adjusted to 200/300kHz, 300/400kHz or 400/500kHz
respectively. In order to maximize the efficiency at light load condition, a pulse skipping
mode can be selected.
The PM6680 includes also a 5V linear regulator (LDO5) that can power the switching
drivers. If the output OUT1 regulates 5V, in order to maximize the efficiency in higher
consumption status, the linear regulator can be turned off and their outputs can be supplied
directly from the switching outputs. The PM6680 provides protection versus overvoltage,
undervoltage and over temperature as well as power good signals for monitoring purposes.
An external 1.237V reference is available.
7.1
Constant On time PWM control
If the SKIP pin is tied to 5V, the device works in PWM mode. Each power section has an
independent on time control.The PM6680 employees a pseudo-fixed switching frequency,
Constant On Time (COT) controller as core of the switched mode section. Each power
section has an independent COT control.
The COT controller is based on a relatively simple algorithm and uses the ripple voltage due
to the output capacitor's ESR to trigger the fixed on-time one-shot generator. In this way, the
output capacitor's ESR acts as a current sense resistor providing the appropriate ramp
signal to the PWM comparator. On-time one-shot duration is directly proportional to the
output voltage, sensed at the OUT1/OUT2 pins, and inversely proportional to the input
voltage, sensed at the VIN pin, as follows:
Equation 1
V OUT
T ON = K ⋅ -------------V IN
This leads to a nearly constant switching frequency, regardless of input and output voltages.
When the output voltage goes lower than the regulated voltage Vreg, the on-time one shot
generator directly drives the high side MOSFET for a fixed on time allowing the inductor
current to increase; after the on time, an off time phase, in which the low side MOSFET is
turned on, follows. Figure 28 shows the inductor current and the output voltage waveforms
in PWM mode.
21/49
Device description
PM6680
Figure 28. Constant ON time PWM control
The duty cycle of the buck converter in steady state is:
Equation 2
V OUT
D = -------------V IN
The PWM control works at a nearly fixed frequency fSW:
Equation 3
V OUT
-------------V IN
f SW = ------------------------------- = 1 ⁄ K on
V OUT
K on × -------------V IN
As mentioned the steady state switching frequency is theoretically independent from battery
voltage and from output voltage.
Actually the frequency depends on parasitic voltage drops that are present during the
charging path(high side switch resistance, inductor resistance(DCR)) and discharging
path(low side switch resistance, DCR).
As a result the switching frequency increases as a function of the load current.
Standard switching frequency values can be selected for both sections by pin FSEL as
shown in the following table:
Table 6.
22/49
FSEL pin selection: typical switching frequency
Fsw@OUT1=1.5V(kHz)
Fsw@OUT2=1.05V(kHz)
FSEL = GND
200
325
FSEL = VREF
290
425
FSEL = LDO5
390
590
PM6680
7.2
Device description
Constant on time architecture
Figure 29 shows the simplified block diagram of a Constant On Time controller. A minimum
off-time constrain (300ns typ.) is introduced to allow inductor valley current sensing on
synchronous switch. A minimum on-time (70ns) is also introduced to assure the start-up
switching sequence.
PM6680 has a one-shot generator for each power section that turns on the high side
MOSFET when the following conditions are satisfied simultaneously: the PWM comparator
is high, the synchronous rectifier current is below the current limit threshold, and the
minimum off-time has timed out.
Once the on-time has timed out, the high side switch is turned off, while the synchronous
switch is turned on according to the anti-cross conduction circuitry management.
When the negative input voltage at the PWM comparator (Figure 29), which is a scaleddown replica of the output voltage (see the external R1/R2 divider in Figure 29), reaches the
valley limit (determined by internal reference Vr = 0.9V), the low-side MOSFET is turned off
according to the anti-cross conduction logic once again, and a new cycle begins.
Figure 29. Constant on-time block diagram
ToffToff-min
BOOT
CSENSE
+
Positive
S
-
Current
Limit
Vr
COMP
R
+
shifte
r
Q
+
- PWM
gm
-
+
HS
driver
Level
Q
+
Ton
Comparato
r
0.5V
0.25V
HGATE
PHASE
-
LDO5
Vr
FB
ZeroZero-cross.
OUT
+
-
VIN
SKIP
S
LS
driver
Q
PGND
R
LDO5
0.5V
+
-
LGATE
Comp.
bandga
p
VREF
1.236V
Vr
23/49
Device description
7.3
PM6680
Output ripple compensation and loop stability
In a classic Constant On Time control, the system regulates the valley value of the output
voltage and not the average value, as shown in Figure 28 In this condition, the output
voltage ripple is source of a DC static error.
To compensate this error, an integrator network can be introduced in the control loop, by
connecting the output voltage to the COMP1/COMP2 (for the OUT1 and OUT2 sections
respectively) pin through a capacitor CINT as in Figure 30
Figure 30. Circuitry for output ripple compensation
COMP PIN
VOLTAGE
∆V
Vr
t
OUTPUT
VOLTAGE
CFILT
CINT
∆V
t
L
D
COMP
I=gm(V1-Vr)
Vr
+
-
PWM
Comparator
gm
VCINT
RINT
+
Vr
V1
OUT
ROUT
COUT
R2
FB
R1
The integrator amplifier generates a current, proportional to the DC errors between the FB
voltage and Vr, which decreases the output voltage in order to compensate the total static
error, including the voltage drop on PCB traces. In addition, CINT provides an AC path for the
output ripple. In steady state, the voltage on COMP1/COMP2 pin is the sum of the reference
voltage Vr and the output ripple (see Figure 30). In fact when the voltage on the COMP pin
reaches Vr, a fixed Ton begins and the output increases.
For example, we consider Vout = 5V with an output ripple of ∆V = 50mV. Considering
CINT >> CFILT, the CINT DC voltage drop VCINT is about 5V-Vr+25mV = 4.125V. CINT assures
an AC path for the output voltage ripple. Then the COMP pin ripple is a replica of the output
ripple, with a DC value of Vr + 25mV = 925mV.
For more details about the output ripple compensation network, see the Section 7.13.6:
Closing the integrator loop on page 37 in the Design guidelines.
In steady state the FB pin voltage is about Vr and the regulated output voltage depends on
the external divider:
Equation 4
R
OUT = Vr × ⎛⎝ 1 + ------2-⎞⎠
R1
24/49
PM6680
7.4
Device description
Pulse skip mode
If the SKIP pin is tied to ground, the device works in skip mode.
At light loads a zero-crossing comparator truncates the low-side switch on-time when the
inductor current becomes negative. In this condition the section works in discontinuous
conduction mode. The threshold between continuous and discontinuous conduction mode
is:
Equation 5
ILOAD(SKIP) =
VIN − VOUT
× TON
2×L
For higher loads the inductor current doesn't cross the zero and the device works in the
same way as in PWM mode and the frequency is fixed to the nominal value.
Figure 31. PWM and pulse skip mode inductor current
PWM mode
Pulse skip mode
Low side on
Inductor current
Load current
0
Ton1 Toff
Ton2
Ton1=Ton2
Toff
Time
Low side off
Figure 31 shows inductor current waveforms in PWM and SKIP mode. In order to keep
average inductor current equal to load current, in SKIP mode some switching cycles are
skipped. When the output ripple reaches the regulated voltage Vreg, a new cycle begins.
The off cycle duration and the switching frequency depend on the load condition.
As a result of the control technique, losses are reduced at light loads, improving the system
efficiency.
7.5
No-audible skip mode
If SKIP pin is tied to VREF, a no-audible skip mode with a minimum switching frequency of
33 KHz is enabled. At light load condition, If there is not a new switching cycle within a 30us
(typ.) period, a no-audible skip mode cycle begins.
25/49
Device description
PM6680
Figure 32. No audible skip mode
Inductor current
No audible skip mode
∼30us
Time
0
Low side
The low side switch is turned on until the output voltage crosses about Vreg+1%. Then the
high side MOSFET is turned on for a fixed on time period. Afterwards the low side switch is
enabled until the inductor current reaches the zero-crossing threshold. This keeps the
switching frequency higher than 33KHz. As a consequence of the control, the regulated
voltage can be slightly higher than Vreg (up to 1%).
If, due to the load, the frequency is higher than 33KHz, the device works like in skip mode.
No-audible skip mode reduces audio frequency noise that may occur in pulse skip mode at
very light loads, keeping the efficiency higher than in PWM mode.
7.6
Current limit
The current-limit circuit employs a "valley" current-sensing algorithm. During the conduction
time of the low side MOSFET the current flowing through it is sensed. The current-sensing
element is the low side MOSFET on-resistance (Figure 33).
Figure 33. RDSON sensing technique
HGATE
HS
PHASE
Rcsense
CSENSE
LGATE
LS
RDSon
26/49
PM6680
Device description
An internal 100µA current source is connected to CSENSE pin and determines a voltage
drop on RCSENSE. If the voltage across the sensing element is greater than this voltage
drop, the controller doesn't initiate a new cycle. A new cycle starts only when the sensed
current goes below the current limit.
Since the current limit circuit is a valley current limit, the actual peak current limit is greater
than the current limit threshold by an amount equal to the inductor ripple current. Moreover
the maximum output current is equal to the valley current limit plus half of the inductor ripple
current:
Equation 6
ILOAD (max) = ILvalley +
∆IL
2
The output current limit depends on the current ripple, as shown in Figure 34:
Figure 34. Current waveforms in current limit conditions
Current
DC current limit = maximum load
Maximum load current
is influenced by the
inductor current ripple
Inductor current
Valley current threshold
Time
Being fixed the valley threshold, the greater the current ripple is, greater the DC output
current is The valley current limit can be set with resistor RCSENSE:
Equation 7
R CSENSE =
R DSon × ILvalley
ICSENSE
Where ICSENSE = 100uA, RDSon is the drain-source on resistance of the low side switch.
Consider the temperature effect and the worst case value in RDSon calculation.
The accuracy of the valley current threshold detection depends on the offset of the internal
comparator (∆VOFF) and on the accuracy of the current generator(∆VCSENSE):
Equation 8
∆ILvalley
ILvalley
=
⎤ ∆R CSENSE ∆R SNS
∆I CSENSE ⎡
∆VOFF
+⎢
× 100 ⎥ +
+
ICSENSE
R SNS
⎣ R CSENSE × ICSENSE
⎦ R CSENSE
27/49
Device description
PM6680
Where RSNS is the sensing element (RDSon)
PM6680 provides also a fixed negative peak current limit to prevent an excessive reverse
inductor current when the switching section sinks current from the load in PWM mode. This
negative current limit threshold is measured between PHASE and SGND pins, comparing
the magnitude drop on the PHASE node during the conduction time of the low side
MOSFET with an internal fixed voltage of 120mV.
If the current is sensed on the low side MOSFET, the negative valley-current limit INEG (if the
device works in PWM mode) is given by:
Equation 9
INEG =
7.7
120mV
RDSon
Soft start and soft end
Each switching section is enabled separately by asserting high EN1/EN2 pins respectively.
In order to realize the soft start, at the startup the overcurrent threshold is set 25% of the
nominal value and the undervoltage protection (see related sections) is disabled. The
controller starts charging the output capacitor working in current limit. The overcurrent
threshold is increased from 25% to 100% of the nominal value with steps of 25% every
700?s (typ.). After 2.8ms (typ.) the undervoltage protection is enabled. The soft start time is
not programmable. A minimum capacitor CINT is required to ensure a soft start without any
overshoot on the output:
Equation 10
CINT ≥
6uA
ILvalley
4
∆I
+ L
2
× C out
Figure 35. Soft start waveforms
Switching output
Current limit threshold
EN1/EN2
28/49
Time
PM6680
Device description
When a switching section is turned off (EN1/EN2 pins low), the controller enters in soft end
mode.The output capacitor is discharged through an internal 18Ω P-MOSFET switch; when
the output voltage reaches 0.3V, the low-side MOSFET turns on, keeping the output to
ground. The soft end time also depends on load condition.
7.8
Gate drivers
The integrated high-current drivers allow to use different power MOSFETs. The high side
driver MOSFET uses a bootstrap circuit which is indirectly supplied by LDO5 output. The
BOOT and PHASE pins work respectively as supply and return rails for the HS driver.
The low side driver uses the internal LDO5 output for the supply rail and PGND pin as return
rail.
An important feature of the gate drivers is the adaptive anti-cross conduction protection,
which prevents high side and low side MOSFETs from being on at the same time. When the
high side MOSFET is turned off the voltage at the phase node begins to fall. The low side
MOSFET is turned on when the voltage at the phase node reaches an internal threshold.
When the low side MOSFET is turned off, the high side remains off until the LGATE pin
voltage goes approximately under 1V.
The power dissipation of the drivers is a function of the total gate charge of the external
power MOSFETs and the switching frequency, as shown in the following equation:
Equation 11
Pdriver = Vdriver × Q g × fsw
Where Vdriver is the 5V driver supply.
7.9
Reference voltage and bandgap
The 1.237V (typ.) internal bandgap voltage is accurate to ±1% over the temperature range.
It is externally available (VREF pin) and can supply up to 100µA and can be used as a
voltage threshold for the multifunction pins FSEL and SKIP to select the appropriate working
mode. Bypass VREF to ground with a 100nF minimum capacitor.
If VREF goes below 0.87V (typ.) , the system detects a fault condition and all the circuitry is
turned off. A toggle on the input voltage (power on reset) or a toggle on SHDN pin is
necessary to restart the device.
An internal divider of the the bandgap provides a voltage reference Vr of 0.9V. This voltage
is used as reference for the linear and the switching regulators outputs. The overvoltage
protection, the undervoltage protection and the power good signals are referred to Vr.
29/49
Device description
7.10
PM6680
Internal linear regulator
The PM6680 has an internal linear regulator providing 5V (LDO5) at ±2% accuracy. High
side drivers, low side drivers and most of internal circuitry are supplied by LDO5 output
through VCC pin (an external RC filter may be applied between LDO5 and VCC). The linear
regulator can provide an average output current of 50mA and a peak output current of
100mA. Bypass LDO5 output with a minimum 1µF ceramic capacitor and a 4,7µF tantalum
capacitor ( ESR 2Ω). If the 5V output goes below 4V, the system detects a fault condition
and all the circuitry is turned off. A power on reset or a toggle on SHDN pin is necessary to
restart the device.
V5SW pin allows to keep the 5V linear regulator always active or to enable the internal
bootstrap-switch over function: if the 5V switching output is connected to V5SW, when the
voltage on V5SW pin is above 4.8V, an internal 3.0 Ω max p-channel MOSFET switch
connects V5SW pin to LDO5 pin and simultaneously LDO5 shuts down. This configuration
allows to achieve higher efficiency. V5SW can be connected also to an external 5V supply.
LDO5 regulator turns off and LDO5 is supplied externally. If V5SW is connected to ground,
the internal 5V regulator is always on and supplies LDO5 output.
Table 7.
V5SW multifunction pin
V5SW
GND
Description
The 5V linear regulator is always turned on and supplies LDO5 output.
Switching 5V The 5V linear regulator is turned off when the voltage on V5SW is above 4.8V and
output
LDO5 output is supplied by the switching 5V output.
External 5V
supply
7.11
The 5V linear regulator is turned off when the voltage on V5SW is above 4.8V and
LDO5 output is supplied by the external 5V.
Power up sequencing and operative modes
Let's consider SHDN, EN1 and EN2 low at the beginning. The battery voltage is applied as
input voltage. The device is in shutdown mode.
When the SHDN pin voltage is above the shutdown device on threshold (1.5V typ.), the
controller begins the power-up sequence. All the latched faults are cleared. LDO5
undervoltage control is blanked for 4 ms and the internal regulator LDO5 turns on. If the
LDO5 output is above the UVLO threshold after this time, the device enters in standby
mode. The switching outputs are kept to ground by turning on the low side MOSFETs.
When EN1 and EN2 pins are forced high the switching sections begin their soft start
sequence.
Table 8.
Mode
Run
Standby
Shutdown
30/49
Operatives modes
Conditions
SHDN is high,
EN1/EN2 pins are high
Description
Switching regulators are enabled; internal linear
regulators outputs are enabled.
Internal Linear regulators active (LDO5 is always on).
Both EN1/EN2 pins are low
In Standby mode LGATE1/LGATE2 pins are forced
and SHDN pin is high
high while HGATE1/HGATE2 pins are forced low.
SHDN is low
All circuits off.
PM6680
Device description
7.12
Monitoring and protections
7.12.1
Power good signals
The PM6680 provides three independent power good signals: one for each switching
section (PGOOD1/PGOOD2).
PGOOD1/PGOOD2 signals are low if the output voltage is out of ±10% of the designed set
point or during the soft-start, the soft end and when the device works in standby and
shutdown mode.
7.12.2
Thermal protection
The PM6680 has a thermal protection to preserve the device from overheating. The thermal
shutdown occurs when the die temperature goes above +150°C. In this case all internal
circuitry is turned off and the power sections are turned off after the discharge mode.
A power on reset or a toggle on the SHDN pin is necessary to restart the device.
7.12.3
Overvoltage protection
When the switching output voltage is about 115% of its nominal value, a latched
overvoltage protection occurs. In this case, the synchronous rectifier immediately turns on
while the high-side MOSFET turns off. The output capacitor is rapidly discharged and the
load is preserved from being damaged. The overvoltage protection is also active during the
soft start. Once an overvoltage protection has been detected, a toggle on SHDN, EN1/EN2
pins or a power on reset is necessary to exit from the latched state.
7.12.4
Undervoltage protection
When the switching output voltage is below 70% of its nominal value, a latched undervoltage
protection occurs. In this case the switching section is immediately disabled and both
switches are open. The controller enters in soft end mode and the output is eventually kept
to ground, turning low side MOSFET on. The undervoltage circuit protection is enabled only
at the end of the soft-start. Once an overvoltage protection has been detected, a toggle on
SHDN, EN1/EN2 pin or a power on reset is necessary to clear the undervoltage fault and
starts with a new soft-start phase.
Table 9.
Protections and operatives modes
Mode
Conditions
Description
Overvoltage
protection
LGATE1/LGATE2 pin is forced high, LDO5 remains
OUT1/OUT2 > 115% of the
active. Exit by a power on reset or toggling SHDN or
nominal value
EN1/EN2
Undervoltage
protection
OUT1/OUT2 < 70% of the
nominal value
LGATE1/LGATE2 is forced high after the soft end
mode, LDO5 remains active. Exit by a power on reset
or toggling SHDN or EN1/EN2
TJ > +150ºC
All circuitry off. Exit by a POR on VIN or toggling
SHDN.
Thermal
shutdown
31/49
Device description
7.13
PM6680
Design guidelines
The design of a switching section starts from two parameters:
7.13.1
●
Input voltage range: in notebook applications it varies from the minimum battery
voltage, VINmin to the AC adapter voltage, VINmax.
●
Maximum load current: it is the maximum required output current, ILOAD(max).
Switching frequency
It's possible to set 3 different working frequency ranges for the two sections with FSEL pin
(Table 7 on page 30).
Switching frequency mainly influences two parameters:
7.13.2
●
Inductor size: for a given saturation current and RMS current, greater frequency allows
to use lower inductor values, which means smaller size.
●
Efficiency: switching losses are proportional to frequency. High frequency generally
involves low efficiency.
Inductor selection
Once that switching frequency is defined, inductor selection depends on the desired
inductor ripple current and load transient performance.
Low inductance means great ripple current and could generate great output noise. On the
other hand, low inductor values involve fast load transient response.
A good compromise between the transient response time, the efficiency, the cost and the
size is to choose the inductor value in order to maintain the inductor ripple current ∆IL
between 20% and 50% of the maximum output current ILOAD(max). The maximum ∆IL
occurs at the maximum input voltage. With this considerations, the inductor value can be
calculated with the following relationship:
Equation 12
L=
VIN − VOUT VOUT
×
fsw × ∆IL
VIN
where fSW is the switching frequency, VIN is the input voltage, VOUT is the output voltage and
∆IL is the selected inductor ripple current.
In order to prevent overtemperature working conditions, inductor must be able to provide an
RMS current greater than the maximum RMS inductor current ILRMS:
Equation 13
ILRMS = (ILOAD (max)) 2 +
Where ∆IL(max) is the maximum ripple current:
32/49
(∆IL (max)) 2
12
PM6680
Device description
Equation 14
∆IL (max) =
VIN max − VOUT
V
× OUT
fsw × L
VIN max
If hard saturation inductors are used, the inductor saturation current should be much greater
than the maximum inductor peak current Ipeak:
Equation 15
Ipeak = ILOAD (max) +
∆IL (max)
2
Using soft saturation inductors it's possible to choose inductors with saturation current limit
nearly to Ipeak.
Below there is a list of some inductor manufacturers.
Table 10.
7.13.3
Inductor manufacturer
Manufacturer
Series
Inductor value (uH)
RMS current (A)
Saturation current (A)
COILCRAFT
SER1360
1 to 8
6 to 9.5
7 to 31
COILCRAFT
MLC
0.7 to 4.5
13.6 to 17.3
11.5 to 26
TDK
RLF12560
1 to 10
7.5 to 14.4
7.5 to 18.5
Output capacitor
The selection of the output capacitor is based on the ESR value Rout and the voltage rating
rather than on the capacitor value Cout.
The output capacitor has to satisfy the output voltage ripple requirements. Lower inductor
value can reduce the size of the choke but increases the inductor current ripple ∆IL.
Since the voltage ripple VRIPPLEout is given by:
Equation 16
VRIPPLEout = R out × ∆IL
A low ESR capacitor is required to reduce the output voltage ripple. Switching sections can
work correctly even with 20mV output ripple.
However, to reduce jitter noise between the two switching sections it's preferable to work
with an output voltage ripple greater than 30mV. If lower output ripple is required, a further
compensation network is needed (see Section 7.13.6: Closing the integrator loop on
page 37).
Finally the output capacitor choice deeply impacts on the load transient response (see Load
transient response paragraph). Below there is a list of some capacitor manufacturers.
33/49
Device description
Table 11.
PM6680
Output capacitor manufacturer
Manufacturer
Series
Capacitor value (uF)
Rated voltage (V)
ESR max (mΩ)
SANYO
POSCAP TPB,TPD, TPE
100 to 470
2.5 to 6.3
12 to 65
PANASONIC
SPCAP UD, UE
100 to 470
2 to 6.3
7 to 18
7.13.4
Input capacitors selection
In a buck topology converter the current that flows into the input capacitor is a pulsed current
with zero average value. The input RMS current of the two switching sections can be roughly
estimated as follows:
Equation 17
ICinRMS = D1 × I12 × (1 − D1 ) + D 2 × I22 × (1 − D 2 )
Where D1, D2 are the duty cycles and I1, I2 are the maximum load currents of the two
sections.
Input capacitor should be chosen with an RMS rated current higher than the maximum RMS
current given by both sections.
Tantalum capacitors are good in term of low ESR and small size, but they occasionally can
burn out if subjected to very high current during the charge. Ceramic capacitors have
usually a higher RMS current rating with smaller size and they remain the best choice.
Below there is a list of some ceramic capacitor manufacturers.
Table 12.
34/49
Input capacitor manufacturer
Manufacturer
Series
Capacitor value (uF)
Rated voltage (V)
TAYIO YUDEN
UMK325BJ106KM-T
10
50
TAYIO YUDEN
GMK325BJ106MN
10
35
TDK
C3225X5R1E106M
10
25
PM6680
7.13.5
Device description
Power MOSFETs
Logic-level MOSFETs are recommended, since low side and high side gate drivers are
powered by LDO5. Their breakdown voltage VBRDSS must be higher than VINmax.
In notebook applications, power management efficiency is a high level requirement. The
power dissipation on the power switches becomes an important factor in switching
selections. Losses of high-side and low-side MOSFETs depend on their working conditions.
The power dissipation of the high-side MOSFET is given by:
Equation 18
PDHighSide = Pconduction + Pswitching
Maximum conduction losses are approximately:
Equation 19
Pconduction = R DSon ×
VOUT
× ILOAD (max) 2
VIN min
where RDSon is the drain-source on resistance of the high side MOSFET. Switching losses
are approximately:
Equation 20
Pswitching =
∆IL
∆I
) × t on × fsw VIN × (ILOAD (max) + L ) × t off × fsw
2
2
+
2
2
VIN × (ILOAD (max) −
where ton and toff are the switching times of the turn off and turn off phases of the MOSFET.
As general rule, high side MOSFETs with low gate charge are recommended, in order to
minimize driver losses. Below there is a list of possible choices for the high side MOSFET.
Table 13.
High side MOSFET manufacturer
Manufacturer
Type
Gate charge (nC)
Rated reverse voltage (V)
ST
STS12NH3LL
10
30
ST
STS17NH3LL
18
30
The power dissipation of the low side MOSFET is given by:
Equation 21
PDLowSide = Pconduction
Maximum conduction losses occur at the maximum input voltage:
35/49
Device description
PM6680
Equation 22
⎛
V
Pconduction = RDSon × ⎜⎜1 − OUT
VIN max
⎝
⎞
⎟ × ILOAD (max)2
⎟
⎠
Choose a synchronous rectifier with low RDSon. When high side MOSFET turns on, the fast
variation of the phase node voltage can bring up even the low side gate through its gatedrain capacitance CRSS, causing cross-conduction problems. Choose a low side MOSFET
that minimizes the ratio CRSS/CGS (CGS = CISS - CRSS).
Below there is a list of some possible low side MOSFETs.
Table 14.
Low side MOSFET manufacturer
Manufacturer
Type
RDSon (mΩ)
CRSS
C GS
Rated reverse voltage (V)
ST
STS17NF3LL
5.5
0.047
30
ST
STS25NH3LL
3.5
0.011
30
Dual n-channel MOSFETS can be used in applications with a maximum output current of
about 3 A. Below there is a list of some MOSFET manufacturers.
Table 15.
Dual MOSFET manufacturer
Manufacturer
Type
RDSon (mΩ)
Gate charge (nC)
Rated reverse
voltage (V)
ST
STS8DNH3LL
25
10
30
ST
STS4DNF60L
65
32
60
A rectifier across the low side MOSFET is recommended. The rectifier works as a voltage
clamp across the synchronous rectifier and reduces the negative inductor swing during the
dead time between turning the high-side MOSFET off and the synchronous rectifier on. It
can increase the efficiency of the switching section, since it reduces the low side switch
losses. A shottky diode is suitable for its low forward voltage drop (0.3V). The diode reverse
voltage must be greater than the maximum input voltage VINmax. A minimum recovery
reverse charge is preferable. Below there is a list of some shottky diode manufacturers.
Table 16.
36/49
Shottky diode manufacturer
Manufacturer
Series
Forward
voltage (V)
Rated reverse
voltage (V)
Reverse current
(uA)
ST
STPS1L30M
0.34
30
0.00039
ST
STPS1L20M
0.37
20
0.000075
PM6680
7.13.6
Device description
Closing the integrator loop
The design of external feedback network depends on the output voltage ripple. If the ripple
is higher than approximately 30mV, the feedback network (Figure 35) is usually enough to
keep the loop stable.
Figure 36. Circuitry for output ripple compensation
COMP PIN
VOLTAGE
?V
Vr
Vr
t
OUTPUT
VOLTAGE
I=gm(V1-Vr)
COMP
PW M
Comparator
CFILT
gm
+
CINT
?V
+
VCINT
Vr
RINT
t
L
OUT
R2
ROUT
D
V1
COUT
FB
R1
The stability of the system depends firstly on the output capacitor zero frequency. The
following condition should be satisfied:
Equation 23
fsw > k × fZout =
k
2π × C out × R out
where k is a design parameter greater than 3 and Rout is the ESR of the output capacitor. It
determinates the minimum integrator capacitor value CINT:
Equation 24
CINT >
gm
Vr
×
⎛ fsw
⎞ VOUT
2π × ⎜
− fZout ⎟
⎝ k
⎠
where gm = 50us is the integrator transconductance.
In order to ensure stability it must be also verified that:
Equation 25
C INT >
gm
Vr
×
2π × fZout VOUT
37/49
Device description
PM6680
In order to reduce ground noise due to load transient on the other section, it is
recommended to add a resistor RINT and a capacitor Cfilt that, together with CINT, realize a
low pass filter (see Figure 36). The cutoff frequency fCUT must be much greater (10 or more
times) than the switching frequency of the section:
Equation 26
R INT =
2π × fCUT
1
C
× C filt
× INT
C INT + C filt
Due to the capacitive divider (CINT, Cfilt), the ripple voltage at the COMP pin is given by:
Equation 27
VRIPPLEINT = VRIPPLEout ×
CINT
= VRIPPLEout × q
CINT + Cfilt
Where VRIPPLEout is the output ripple and q is the attenuation factor of the output ripple.
If the ripple is very small (lower than approximately 30mV), a further compensation network,
named virtual ESR network, is needed. This additional part generates a triangular ripple
that is added to the ESR output voltage ripple at the input of the integrator network. The
complete control schematic is represented in Figure 37.
Figure 37. Virtual ESR network
COMP PIN
VOLTAGE
T NODE
VOLTAGE
? V1
? V1
Vr
OUTPUT
VOLTAGE
t
?V
t
CFILT
Vr
COMP
+
- PWM
t
T
RINT
Vr
C
R
OUT
L
ROUT
D
38/49
COUT
R2
R1
Comparator
gm
CINT
-
R1
FB
+
V1
PM6680
Device description
The T node voltage is the sum of the output voltage and the triangular waveform generated
by the virtual ESR network. In fact the virtual ESR network behaves like a further equivalent
ESR RESR.
A good trade-off is to design the network in order to achieve an RESR given by:
Equation 28
RESR =
VRIPPLE
− Rout
∆IL
where ∆IL is the inductor current ripple and VRIPPLE is the overall ripple of the T node
voltage. It should be chosen higher than approximately 30mV.
The new closed loop gain depends on CINT. In order to ensure stability it must be verified
that:
Equation 29
C INT >
gm
Vr
×
2π × fZ VOUT
Where:
Equation 30
fZ =
1
2π × C out × R TOT
where RTOT is the sum of the ESR of the output capacitor Rout and the equivalent ESR
given by the virtual ESR network RESR.
Moreover CINT must meet the following condition:
Equation 31
fsw > k × fZ =
k
2π × C out × R TOT
Where k is a free design parameter greater than 3 and determines the minimum integrator
capacitor value CINT:
Equation 32
CINT >
gm
Vr
×
V
f
⎞
⎛
OUT
2π × ⎜⎜ sw − fZ ⎟⎟
k
⎠
⎝
C must be selected as shown:
Equation 33
C > 5 × C INT
39/49
Device description
PM6680
R must be chosen in order to have enough ripple voltage on integrator input:
Equation 34
R=
L
RESR × C
R1 can be selected as follows:
Equation 35
⎛
1
R × ⎜⎜
⎝ C × π × fZ
R1 =
1
R−
C × π × fZ
⎞
⎟⎟
⎠
Example:
OUT1 = 1.5V, fSW = 290KHz, L = 2.5uH, Cout = 330uF with Rout
≈ 12mΩ.
We design RESR = 12mW. We choose CINT = 1nF by equations 30, 33 and Cfilt = 47pF,
RINT = 1KΩ by eq.27, 28. C = 5.6nF by Eq.34. Then R = 36KΩ (eq.35) and R1 = 3KΩ
(eq.36).
7.13.7
Other parts design
●
VIN filter
A VIN pin low pass filter is suggested to reduce switching noise. The low pass filter is
shown in the next figure:
Figure 38. VIN pin filter
R
Input
voltage
VIN
C
100pF
Typical components values are: R = 3.9Ω and C = 4.7uF.
●
40/49
VCC filter
A VCC low pass filter helps to reject switching commutations noise:
PM6680
Device description
Figure 39. Inductor current waveforms
LDO5
R
VC C
C
Typical components values are: R=47Ω and C = 1uF.
●
VREF capacitor
A 10nF to 100nF ceramic capacitor on VREF pin must be added to ensure noise
rejection.
●
LDO5 output capacitors
Bypass the output of each linear regulator with 1uF ceramic capacitor closer to the
LDO pin and a 4.7uF tantalum capacitor (ESR = 2Ω). In most applicative conditions a
4.7uF ceramic output capacitor can be enough to ensure stability.
●
Bootstrap circuit
The external bootstrap circuit is represented in the next figure:
Figure 40. Bootstrap circuit
D
RBOOT
L
CBOOT
LDO5
BOOT
PHASE
The bootstrap circuit capacitor value CBOOT must provide the total gate charge to the high
side MOSFET during turn on phase. A typical value is 100nF.
The bootstrap diode D must charge the capacitor during the off time phases. The maximum
rated voltage must be higher than VINmax.
A resistor RBOOT on the BOOT pin could be added in order to reduce noise when the phase
node rises up, working like a gate resistor for the turn on phase of the high side MOSFET.
41/49
Device description
7.13.8
PM6680
Design example
The following design example considers an input voltage from 7V to 16V. The two switching
outputs are OUT1=1.5V and OUT2=1.05V and must deliver a maximum current of 5A. The
selected switching frequencies are about 290kHz for OUT1 section and about 425kHz for
OUT2 section (see Table 7 on page 30).
1.
Inductor selection
OUT1: ILOAD = 5A, 35% ripple current.
Equation 36
L=
1.5V ⋅ (16V − 1.5V)
≈ 2.5µH
290KHz ⋅ 16V ⋅ 0.35 ⋅ 5
We choose standard value L = 2.5 uH.
∆IL(max) = 1.8A @VIN =12V
ILRMS = 5.03A
IPEAK = 5A + 0.9A = 5.9A
OUT2: ILOAD = 5A, 30% ripple current.
Equation 37
L=
1.05V ⋅ (16V − 1.05V)
≈ 1.6µH
425KHz ⋅ 16V ⋅ 0.3 ⋅ 5
∆IL(max) = 1.6A @VIN = 12V.
ILRMS = 5.02A
IPEAK = 5A + 0.8A = 5.8A
2.
Output capacitor selection
We would like to have an output ripple smaller than 25mV.
OUT1: POSCAP 6TPB330M
OUT2: POSCAP 6TPB330M
3.
Power MOSFETS
OUT1: High side: STS12NH3LL
Low side: STS12NH3LL
OUT2:High side: STS12NH3LL
Low side: STS12NH3LL
4.
Current limit
OUT1:
Equation 38
ILvalley (min) = ILOAD (max) −
42/49
∆IL (min)
= 4.12A
2
PM6680
Device description
Equation 39
RCSENSE ≡
4.12A
⋅ 16.25mΩ ≈ 670Ω
100µA
(Let's assume the maximum temperature Tmax = 75°C in RDSon calculation)
OUT2:
Equation 40
ILvalley (min) = ILOAD (max) −
∆IL (min)
= 4.2A
2
Equation 41
RCSENSE ≡
4 .2 A
⋅ 16.25mΩ ≈ 680Ω
100µA
(Let's assume Tmax = 75°C in RDSon calculation)
5.
Input capacitor
Maximum input capacitor RMS current is about 2.8A. Then ICinRMS > 2.8A.
We put three 10uF ceramic capacitors with Irms = 1.5A.
6.
Synchronous rectifier
OUT1: Shottky diode STPS1L30M
OUT2: Shottky diode STPS1L30M
7.
Integrator loop
(Refer to Figure 37)
OUT1: The ripple is smaller than 40mV, then the virtual ESR network is required.
CINT = 1nF; Cfilt = 47pF; RINT =1kΩ
C = 5.6nF; R= 36kΩ; R1 = 3kΩ
OUT2: The ripple is smaller than 40mV, then the virtual ESR network is required.
CINT = 1nF; Cfilt = 110pF; RINT = 1kΩ
C = 5.6nF; R = 22kΩ; R1 = 3.3kΩ
8.
Output feedback divider
(Refer to Figure 30 on page 24)
OUT1: R1 = 10kΩ; R2 = 6.8kΩ
OUT2: R1 = 11kΩ; R2 = 1.8kΩ
9.
Layout guidelines
The layout is very important in terms of efficiency, stability and noise of the system. It is
possible to refer to the PM6680 demoboard for a complete layout example.
For good PC board layout follows these guidelines:
●
Place on the top side all the power components (inductors, input and output capacitors,
MOSFETs and diodes). Refer them to a power ground plan, PGND. If possible, reserve
a layer to PGND plan. The PGND plan is the same for both the switching sections.
●
AC current paths layout is very critical (see Figure 41 on page 44). The first priority is to
minimize their length. Trace the LS MOSFET connection to PGND plan as short as
43/49
Device description
●
●
PM6680
possible. Place the synchronous diode D near the LS MOSFET. Connect the LS
MOSFET drain to the switching node with a short trace.
Place input capacitors near HS MOSFET drain. It is recommended to use the same
input voltage plan for both the switching sections, in order to put together all input
capacitors.
Place all the sensitive analog signals (feedbacks, voltage reference, current sense
paths) on the bottom side of the board or in an inner layer. Isolate them from the power
top side with a signal ground layer, SGND. Connect the SGND and PGND plans only in
one point (a multiple vias connection is preferable to a 0 ohm resistor connection) near
the PGND device pin. Place the device on the top or on the bottom size and connect
the exposed pad and the SGND pins to the SGND plan (see Figure 41).
Figure 41. Current paths, ground connection and driver traces layout
●
●
●
●
44/49
As general rule, make the high side and low side drivers traces wide and short. The
high side driver is powered by the bootstrap circuit. It's very important to place
capacitor CBOOT and diode DBOOT as near as possible to the HGATE pin (for
example on the layer opposite to the device). Route HGATE and PHASE traces as near
as possible in order to minimize the area between them.
The Low side gate driver is powered by the 5V linear regulator output. Placing PGND
and LGATE pins near the low side MOSFETs reduces the length of the traces and the
crosstalk noise between the two sections.
The linear regulator output LDO5 is referred to SGND as long as the reference voltage
Vref. Place their output filtering capacitors as near as possible to the device.
Place input filtering capacitors near VCC and VIN pins.
It would be better if the feedback networks connected to COMP, FB and OUT pins are
"referred" to SGND in the same point as reference voltage Vref. To avoid capacitive
PM6680
Device description
●
coupling place these traces as far as possible from the gate drivers and phase
(switching) paths.
Place the current sense traces on the bottom side. If low side MOSFET RDSon sensing
is enabled, use a dedicated connection between the switching node and the current
limit resistor RCSENSE.
45/49
Package mechanical data
8
PM6680
Package mechanical data
In order to meet environmental requirements, ST offers these devices in ECOPACK®
packages. These packages have a Lead-free second level interconnect. The category of
second Level Interconnect is marked on the package and on the inner box label, in
compliance with JEDEC Standard JESD97. The maximum ratings related to soldering
conditions are also marked on the inner box label. ECOPACK is an ST trademark.
ECOPACK specifications are available at: www.st.com.
Table 17.
VFQFPN 5x5x1.0 32L Pitch 0.50
Databook (mm)
Dim.
Min
Typ
Max
A
0.8
0.9
1
A1
0
0.02
0.05
A3
0.2
b
0.18
0.25
D
4.85
5
D2
0.3
5.15
See exposed pad variations
E
4.85
E2
(2)
5
5.15
See exposed pad variations
e
(2)
0.5
L
0.3
0.4
0.5
ddd
Table 18.
0.05
Exposed pad variations
(1)(2)D2
E2
Min
Typ
Max
Min
Typ
Max
2.90
3.10
3.20
2.90
3.10
3.20
1. VFQFPN stands for Thermally Enhanced Very thin Fine pitch Quad Flat Package No lead. Very thin:
A = 1.00mm Max.
2. Dimensions D2 & E2 are not in accordance with JEDEC.
46/49
PM6680
Package mechanical data
Figure 42. Package dimensions
47/49
Revision history
9
PM6680
Revision history
Table 19.
48/49
Document revision history
Date
Revision
Changes
17-Mar-2006
1
Initial release
10-May-2006
2
Few updates
29-Jun-2006
3
Mechanical data updated
28-Jul-2006
4
Application schematic updated Figure 27 on page 16
25-Oct-2006
5
Changes electrical characteristics, added COMP value skip mode,
Order code table updated
28-Aug-2007
6
Updated: Current sensing option and Absolute Maximum Ratings
PM6680
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