LTC1771 10µA Quiescent Current High Efficiency Step-Down DC/DC Controller U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO The LTC®1771 is a high efficiency current mode stepdown DC/DC controller that draws as little as 10µA DC supply current to regulate the output at no load while maintaining high efficiency for loads up to several amps. Very Low Standby Current: 10µA Available in Space-Saving 8-Lead MSOP Package Output Currents: Up to 5A Wide VIN Range: 2.8V to 20V Operation VOUT Range: 1.23V to 18V High Efficiency: Over 93% Possible ±2% Output Accuracy Very Low Dropout Operation: 100% Duty Cycle Current Mode Operation for Excellent Line and Load Transient Response Defeatable Burst ModeTM Operation Short-Circuit Protected Optional Programmable Soft-Start Micropower Shutdown: IQ = 2µA The LTC1771 drives an external P-channel power MOSFET using a current mode, constant off-time architecture. An external sense resistor is used to program the operating current level. Current mode control provides short-circuit protection, excellent transient response and controlled start-up behavior. Burst Mode operation enables the LTC1771 to maintain high efficiency down to extremely low currents. Shutdown mode further reduces the supply current to a mere 2µA. For low noise applications, Burst Mode operation can be easily disabled with the MODE pin. U APPLICATIO S ■ ■ ■ ■ ■ ■ Wide input supply range of 2.8V to 18V (20V maximum) and 100% duty cycle operation for low dropout make the LTC1771 ideal for a wide variety of battery-powered applications where maximizing battery life is important. Cellular Telephones and Wireless Modems 1- to 4-Cell Lithium-Ion-Powered Applications Portable Instruments Battery-Powered Equipment Battery Chargers Scanners The LTC1771’s availability in both 8-lead MSOP and SO packages provides for a minimum area solution. , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. U TYPICAL APPLICATIO VIN 4.5V TO 18V RSENSE 0.05Ω LTC1771 Efficiency + 100 22µF 25V VIN = 5V VIN RUN/SS CSS 0.01µF SENSE PGATE M1 Si6447DQ ITH LTC1771 RC 10k CC 22OpF VFB MODE GND VIN L1 15µH UPS5817 R2 1.64M 1% + COUT 150µF 6.3V VOUT 3.3V 2A EFFICIENCY (%) 90 80 VIN = 10V VIN = 15V 70 60 50 R1 1M 1% CFF 5pF Figure 1. High Efficiency Step-Down Converter 1771 F01 VOUT = 3.3V RSENSE = 0.05Ω 40 10 0.1 1 100 1000 LOAD CURRENT (mA) 10000 1771 F01b 1 LTC1771 W W W AXI U U ABSOLUTE RATI GS (Note 1) Input Supply Voltage (VIN)........................ – 0.3V to 20V Peak Driver Output Current < 10µs (PGATE) ............. 1A RUN/SS Voltage ......................... – 0.3V to (VIN + 0.3V)* MODE Voltage .......................................... – 0.3V to 20V ITH, VFB Voltage .......................................... – 0.3V to 5V SENSE Voltage (VIN > 12V)..(VIN – 12V) to (VIN + 0.3V)* SENSE Voltage (VIN ≤ 12V) ........ – 0.3V to (VIN + 0.3V)* Junction Temperature (Note 2) ............................ 125°C Operating Temperature Range (Note 3) LTC1771E ......................................... – 40°C to 85°C LTC1771I ......................................... – 40°C to 85°C Storage Temperature Range ................. – 65°C to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C *RUN/SS and SENSE cannot exceed 20V. U U W PACKAGE/ORDER I FOR ATIO ORDER PART NUMBER TOP VIEW RUN/SS ITH VFB GND 1 2 3 4 8 7 6 5 LTC1771EMS8 MODE SENSE VIN PGATE MS8 PACKAGE 8-LEAD PLASTIC MSOP MS8 PART MARKING TJMAX = 125°C, θJA = 200°C/ W LTKD ORDER PART NUMBER TOP VIEW RUN/SS 1 8 MODE ITH 2 7 SENSE VFB 3 6 VIN GND 4 5 PGATE LTC1771ES8 LTC1771IS8 S8 PART MARKING S8 PACKAGE 8-LEAD PLASTIC SO 1771 1771I TJMAX = 125°C, θJA = 110°C/ W Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = 10V, VRUN = open unless otherwise specified. SYMBOL PARAMETER VFB Feedback Voltage (Note 5) ● IFB Feedback Current (Note 5) ● ISUPPLY No-Load Supply Current VIN = 10V, ILOAD = 0 (Note 6) ∆VLINEREG Reference Voltage Line Regulation VIN = 5V to 15V (Note 5) ● 0.003 0.03 ∆VLOADREG Output Voltage Load Regulation ITH = 0.5V to 2V, Burst Disabled (Note 5) ● 0.25 1 % IQ Input DC Supply Current Active Mode (PGATE = 0V) Sleep Mode (Note 6) Shutdown Short Circuit (Note 4) VIN = 2.8V to 18V VIN = 2.8V to 18V, VFB = 1.5V VIN = 2.8V to 18V, VRUN = 0V VIN = 2.8V to 18V, VFB = 0V 150 9 2 175 235 15 6 275 µA µA µA µA ∆VSENSE(MAX) Maximum Current Sense Threshold VFB = VREF – 20mV 140 180 mV ∆VSENSE(MIN) Minimum Current Sense Threshold VFB = VREF + 20mV, Burst Disabled – 25 mV ∆VSENSE(SLEEP) Sleep Current Sense Threshold ITH = 1V 50 mV t OFF Switch Off Time VFB at Regulated Value VFB = 0V VMODE Mode Pin Threshold VMODE Rising 2 CONDITIONS MIN TYP MAX 1.205 1.230 1.255 V 1 10 nA µA 10 ● ● 110 UNITS %/V 3 3.5 70 4 µs µs 0.5 1.3 2 V LTC1771 ELECTRICAL CHARACTERISTICS The ● denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25°C. VIN = 10V, VRUN = open unless otherwise specified. SYMBOL PARAMETER CONDITIONS VRUN/SS RUN/SS Pin Threshold VRUN/SS Rising IRUN Source Current VRUN = 0V, VIN = 2.8V to 18V PGATE t r, tf PGATE Transition Time (Note 7) Rise Time Fall Time CLOAD = 2000pF CLOAD = 2000pF Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC1771S8: TJ = TA + (PD)(110°C/W) LTC1771MS8: TJ = TA + (PD)(150°C/W) Note 3: The LTC1771E is guaranteed to meet performance specifications from 0°C to 70°C. Specifications over the – 40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. The LTC1771I is guaranteed and tested over the – 40°C to 85°C operating temperature range. MIN TYP 0.5 1.0 2 V 0.3 1 3 µA 70 70 140 140 ns ns ● MAX UNITS Note 4: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. See Applications Information. Note 5: The LTC1771 is tested in a feedback loop that servos VFB to the balance point for the error amplifier (VITH = 1.23V). Note 6: No-load supply current consists of sleep mode current (9µA typical) plus a small switching component necessary to overcome Schottky diode leakage and feedback resistor current. Note 7: tr and tf are measured at 10% to 90% levels. U W TYPICAL PERFOR A CE CHARACTERISTICS Efficiency vs Input Voltage Efficiency vs Load Current 100 100 FIGURE 1 CIRCUIT 90 Line Regulation 0.4 Burst Mode OPERATION ENABLED FIGURE 1 CIRCUIT 0.3 ILOAD = 50mA 80 ILOAD = 1mA 0.2 70 60 ∆VOUT (%) EFFICIENCY (%) ILOAD = 1A EFFICIENCY (%) 80 90 Burst Mode OPERATION DISABLED 50 40 0.1 ILOAD = 100mA 0 – 0.1 ILOAD = 1A 30 70 – 0.2 20 10 60 2 4 6 8 10 12 14 16 INPUT VOLTAGE (V) 18 20 1771 G01 0 0.1 VIN = 10V FIGURE 1 CIRCUIT 1 10 100 LOAD CURRENT (mA) 1000 – 0.3 – 0.4 0 5 10 15 20 INPUT VOLTAGE (V) 1771 G02 1771 G03 3 LTC1771 U W TYPICAL PERFOR A CE CHARACTERISTICS Burst Mode OPERATION DISABLED VIN = 15V –0.2 Burst Mode OPERATION ENABLED –0.4 VIN = 5V –0.6 –0.8 150 TA = –50°C 100 50 0 –1.0 0.5 0 1.0 2.0 1.5 TA = 25°C 2 4 LOAD CURRENT (A) 2 0 SOFT-START CURRENT (µA) TA = 25°C 2 4 6 8 10 12 14 16 18 20 INPUT VOLTAGE (V) Current Sense Voltage vs Temperature 200 TA = –50°C 4 2 1771 G06 5 8 SHUTDOWN QUIESCENT CURRENT (µA) 4 Run/SS Current vs Input Voltage TA = –50°C TA = –50°C 6 1771 G05 Shutdown Quiescent Current vs Input Voltage 4 3 TA = 25°C 2 TA = 100°C 1 VIN = 10V MAXIMUM 150 100 BURST THRESHOLD 50 0 MINIMUM TA = 100°C 0 0 2 0 4 6 8 10 12 14 16 18 20 INPUT VOLTAGE (V) 0 2 4 6 8 10 12 14 16 18 20 INPUT VOLTAGE (V) Reference Voltage vs Temperature 1.25 –50 –50 –25 0 25 50 TEMPERATURE (°C) 75 100 1771 G09 1771 G08 1771 G07 REFERENCE VOLTAGE (V) TA = 25°C 8 6 8 10 12 14 16 18 20 INPUT VOLTAGE (V) 1771 G04 6 TA = 100°C 10 0 0 CURRENT SENSE VOLTAGE (mV) ∆VOUT (%) 0 TA = 100°C SLEEP QUIESCENT CURRENT (µA) 0.2 12 200 FIGURE 1 CIRCUIT ACTIVE MODE QUIESCENT CURRENT (µA) 0.4 Sleep Quiescent Current vs Input Voltage Active Mode Quiescent Current vs Input Voltage Load Regulation Load Step Transient Response Burst Mode Operation VIN = 10V 1.24 1.23 VOUT 100mV/DIV VOUT 50mV/DIV INDUCTOR CURRENT 1A/DIV INDUCTOR CURRENT 0.5A/DIV 1.22 1.21 –50 –25 0 25 50 TEMPERATURE (°C) 75 100 1771 G10 4 50µs/DIV VIN = 10V VOUT = 3.3V ILOAD = 100mA TO 2A FIGURE 1 CIRCUIT 1771 G11 10µs/DIV VIN = 10V VOUT = 3.3V ILOAD = 100mA FIGURE 1 CIRCUIT 1771 G12 LTC1771 U U U PI FU CTIO S RUN/SS (Pin 1): The voltage level on this pin controls shutdown/run mode (ground = shutdown, open/high = run). Connecting an external capacitor to this pin provides soft-start. PGATE (Pin 5): High Current Gate Driver for External P-Channel MOSFET Switch. Voltage swing is from ground to VIN. ITH (Pin 2): Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. Nominal voltage range for this pin is 0V to 3V. SENSE (Pin 7): Current Sense Input for Monitoring Switch Current. Maximum switch current and Burst Mode threshold is programmed with an external resistor between SENSE and VIN. VIN (Pin 6): Main Input Voltage Supply Pin. VFB (Pin 3): Feedback of Output Voltage for Comparison to Internal 1.23V Reference. An external resistive divider across the output is returned to this pin. MODE (Pin 8): Burst Mode Enable/Disable Pin. Connecting this pin to VIN (or above 2V) enables Burst Mode operation, while connecting this pin to ground disables Burst Mode operation. Do not leave floating. GND (Pin 4): Ground Pin. W FUNCTIONAL BLOCK DIAGRA U VIN U VIN 6 1µA CSS READY RUN/SS 1 MODE 8 + VIN CIN 1.23V REFERENCE 22k RSENSE (BURST ENABLE) 10% CURRENT EA ON C SENSE + ON + – 1.23V SOFT-START – 7 10% CURRENT VOUT SLEEP * 250k ITH 2 2V RC CC 1V READY GND 4 1V + CURRENT LIMITING PGATE B 5 MODE *OPTIONAL FOR FOLDBACK BLANKING VIN – ON TRIGGER D1 L 1-SHOT 3.5µs STRETCH VOUT VFB R2 + 3 COUT R1 1771 BD 5 LTC1771 U OPERATIO (Refer to Functional Block Diagram) Main Control Loop The LTC1771 uses a constant off-time, current mode step-down architecture. During normal operation, the P-channel MOSFET is turned on at the beginning of each cycle and turned off when the current comparator C triggers the 1-shot timer. The external MOSFET switch stays off for the 3.5µs 1-shot duration and then turns back on again to begin a new cycle. The peak inductor current at which C triggers the 1-shot is controlled by the voltage on Pin 3 (ITH), the output of the error amplifier EA. An external resistive divider connected between VOUT and ground allows EA to receive an output feedback voltage VFB. When the load current increases, it causes a slight decrease in VFB relative to the 1.23V reference, which in turn causes the ITH voltage to increase until the average inductor current matches the new load current. The main control loop is shut down by pulling Pin 1 (RUN/SS) low. Releasing RUN/SS allows an internal 1µA current source to charge soft-start capacitor CSS. When CSS reaches 1V, the main control loop is enabled with the ITH voltage clamped at approximately 40% of its maximum value. As CSS continues to charge, ITH is gradually released allowing normal operation to resume. Burst Mode Operation The LTC1771 provides outstanding low current efficiency and ultralow no-load supply current by using Burst Mode operation when the MODE pin is pulled above 2V. During Burst Mode operation, short burst cycles of normal switching are followed by a longer idle period with the switch off and the load current is supplied by the output capacitor. During this idle period, only the minimum required circuitry—1.23V reference and error amp—are left on, and the supply current is reduced to 9µA. At no load, the output capacitor is still discharged very slowly by leakage current in the Schottky diode and feedback resistor current resulting in very low frequency burst cycles that add a few more microamps to the supply current. 6 Burst Mode operation is provided by clamping the minimum ITH voltage at 1V which represents about 25% of maximum load current. If the load falls below this level, i.e. the ITH voltage tries to fall below 1V, the burst comparator B switches state signaling the LTC1771 to enter sleep mode. During this time, EA is reduced to 10% of its normal operating current and the external compensation capacitor is disconnected and clamped to 1V so that the EA can drive its output with the lower available current. As the load discharges the output capacitor, the internal ITH voltage increases. When it exceeds 1V the burst comparator exits sleep mode, reconnects the external compensation components to the error amplifier output, and returns EA to full power along with the other necessary circuitry. This scheme (patent pending) allows the EA to be reduced to such a low operating current during sleep mode without adding unacceptable delay to wake up the LTC1771 due to the compensation capacitor on ITH required for stability in normal operation. Burst Mode operation can be disabled by pulling the MODE pin to ground. In this mode of operation, the burst comparator B is disabled and the ITH voltage allowed to go all the way to 0V. The load can now be reduced to about 1% of maximum load before the loop skips cycles to maintain regulation. This mode provides a low noise output spectrum, useful for reducing both audio and RF interference, at the expense of reduced efficiency at light loads. Off-Time The off-time duration is 3.5µs when the feedback voltage is close to the reference voltage; however, as the feedback voltage drops, the off-time lengthens and reaches a maximum value of about 70µs when VFB is zero. This ensures that the inductor current has enough time to decay when the reverse voltage across the inductor is low such as during short circuit, thus protecting the MOSFET and inductor. LTC1771 U W U U APPLICATIO S I FOR ATIO The basic LTC1771 application circuit is shown in Figure 1 on the first page. External component selection is driven by the load requirement and begins with the selection of RSENSE. Once RSENSE is known, L can be chosen. Next, the MOSFET and D1 are selected. The inductor is chosen based largely on the desired amount of ripple current and for Burst Mode operation. Finally CIN is selected for its ability to handle the required RMS input current and COUT is chosen with low enough ESR to meet the output voltage ripple and transient specifications. RSENSE is chosen based on the required output current. The LTC1771 current comparator has a maximum threshold of 140mV/RSENSE. The current comparator threshold sets the peak inductor current, yielding a maximum average output current IMAX equal to the peak less half the peak-to-peak ripple current ∆IL. For best performance when Burst Mode operation is enabled, choose ∆IL equal to 35% of peak current. Allowing a margin for variations in the LTC1771 and external components gives the following equation for choosing RSENSE: RSENSE = 100mV/IMAX At higher supply voltages, the peak currents may be slightly higher due to overshoot from current comparator delay and can be predicted from the second term in the following equation: IPEAK 1/ 2 Inductor Value Selection Once RSENSE is known, the inductor value can be determined. The inductance value has a direct effect on ripple current. The ripple current decreases with higher inductance and increases with higher VOUT. The ripple current during continuous mode operation is set by the off-time and inductance to be: V +V ∆IL(CONT) = t OFF OUT D L Kool Mµ is a registered trademark of Magnetics, Inc. ∆IL(BURST) ≈ 35% of IPEAK ≈ 0.05/RSENSE For best efficiency when Burst Mode operation is enabled, choose: ∆IL(CONT) ≤ ∆IL(BURST) so that the inductor current is continuous during the burst periods. This sets a minimum inductor value of: LMIN = (75µH)(VOUT + VD)(RSENSE) RSENSE Selection V –V 0.14 ≅ + 0.5 IN OUT RSENSE L(µH) where tOFF = 3.5µs. However, the ripple current at low loads during Burst Mode operation is: When burst is disabled, ripple currents less than ∆IL(BURST) can be achieved by choosing L > LMIN. Lower ripple current reduces output voltage ripple and core losses, but too low of ripple current will adversely effect efficiency. Inductor Core Selection Once the value of L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool Mµ® cores. Actual core loss is independent of core size for a fixed inductor value, but is very dependent on inductance selected. As inductance increases, core losses go down. Unfortunately, increased inductance requires more turns of wire and therefore copper losses will increase. Ferrite designs have very low core loss and are preferred at high switching frequencies, so design goals can concentrate on copper loss and preventing saturation. Ferrite core material saturates “hard,” which means that inductance collapses abruptly when the peak design current is exceeded. This results in an abrupt increase in inductor ripple current and consequent increase in voltage ripple. Do not allow the core to saturate! Molypermalloy (from Magnetics, Inc.) is a very good, low loss core material for toroids, but it is more expensive than ferrite. A reasonable compromise from the same manufacturer is Kool Mµ. Toroids are space efficient, especially when you can use several layers of wire. Because they generally lack a bobbin, mounting is more difficult. However, designs for surface mount are available that do not increase the height significantly. 7 LTC1771 U W U U APPLICATIO S I FOR ATIO Power MOSFET Selection An external P-channel power MOSFET must be selected for use with the LTC1771. The main selection criteria for the power MOSFET are the threshold voltage VGS(TH) and the “on” resistance RDS(ON), reverse transfer capacitance and total gate charge. Since the LTC1771 can operate down to input voltages as low as 2.8V, a sublogic level threshold MOSFET (RDS(ON) guaranteed at VGS = 2.5V) is required for applications that work close to this voltage. When these MOSFETs are used, make sure that the input supply to the LTC1771 is less than the absolute maximum VGS rating (typically 12V), as the MOSFET gate will see the full supply voltage. The required RDS(ON) of the MOSFET is governed by its allowable power dissipation. For applications that may operate the LTC1771 in dropout, i.e. 100% duty cycle, at its worst case the required RDS(ON) is given by: RDS(ON) = ( ) 2 (1+ δP ) where PP is the allowable power dissipation and δP is the temperature dependency of RDS(ON). (1 + δP) is generally given for a MOSFET in the form of a normalized RDS(ON) vs temperature curve, but = 0.005/°C can be used as an approximation for low voltage MOSFETs. In applications where the maximum duty cycle is less than 100% and the LTC1771 is in continuous mode, the RDS(ON) is governed by: RDS(ON) = DC = PP (DC)IOUT2 (1+ δP ) VOUT + VD VIN + VD where DC is the maximum operating duty cycle of the LTC1771. Catch Diode Selection The catch diode carries load current during the off-time. The average diode current is therefore dependent on the P-channel switch duty cycle. At high input voltages the 8 To maximize both low and high current efficiencies, a fast switching diode with low forward drop and low reverse leakage should be used. Low reverse leakage current is critical to maximize low current efficiency since the leakage can potentially exceed the magnitude of the LTC1771 supply current. Low forward drop is critical for high current efficiency since loss is proportional to forward drop. The effect of reverse leakage and forward drop on no- load supply current and efficiency for various Schottky diodes is shown in Table 1. As can be seen, these are conflicting parameters and the user must weigh the importance of each spec in choosing the best diode for the application. Table 1. Effect of Catch Diode on Performance PP IOUT (MAX) diode conducts most of the time. As VIN approaches VOUT the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the output is short-circuited. Under this condition, the diode must safely handle IPEAK at close to 100% duty cycle. DIODE LEAKAGE NO-LOAD EFFICIENCY (VR = 3.3V) VF @ 1A SUPPLY CURRENT AT 10V/1A MBR0540 0.25µA 0.50V 10.4µA 86.3% UPS5817 2.8µA 0.41V 11.8µA 88.2% MBR0520 3.7µA 0.36V 12.2µA 88.4% MBRS120T3 4.4µA 0.43V 12.2µA 87.9% MBRM120LT3 8.3µA 0.32V 14.0µA 89.4% MBRS320 19.7µA 0.29V 20.0µA 89.8% CIN and COUT Selection At higher load currents, when the inductor current is continuous, the source current of the P-channel MOSFET is a square wave of duty cycle VOUT/VIN. To prevent large voltage transients, a low ESR input capacitor sized for the maximum RMS current must be used. The maximum capacitor current is given by: CIN required IRMS = [ ]1/ 2 IMAX VOUT (VIN − VOUT ) VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s LTC1771 U W U U APPLICATIO S I FOR ATIO ripple current ratings are often based on 2000 hours of life. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Do not underspecify this component. An additional 0.1µF ceramic capacitor is also helpful on VIN for high frequency decoupling. The selection of COUT is driven by the required effective series resistance (ESR). Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering. The output ripple (∆VOUT) in continuous mode is approximated by: 1 ∆VOUT ≈ IRIPPLE ESR + 8 fCOUT where f is the operating frequency, COUT is the output capacitance and IRIPPLE is the ripple current in the inductor. For output ripple less than 100mV, assure COUT required ESR is <2RSENSE. The first condition relates to the ripple current into the ESR of the output capacitance while the second term guarantees that the output capacitance does not significantly discharge during the operating frequency period due to ripple current. The choice of using smaller output capacitance increases the ripple voltage due to the discharging term but can be compensated for by using capacitors of very low ESR to maintain the ripple voltage at or below 50mV. The ITH pin OPTI-LOOPTM compensation components can be optimized to provide stable, high performance transient response regardless of the output capacitors selected. When running into dropout, extra input and output capacitance may be necessary for optimal performance due to the drop in frequency as the duty cycle approaches 100%. Compare Figure 1 to the low dropout regulators shown in the Typical Applications section for recommended CIN, COUT, CFF and CC values for low dropout regulators vs regulators not requiring low dropout. Manufacturers such as Nichicon, United Chemicon and Sanyo should be considered for high performance throughhole capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR for its OPTI-LOOP is a trademark of Linear Technology Corporation. size of any aluminum electrolytic at a somewhat higher price. Typically once the ESR requirement is satisfied, the RMS current rating generally far exceeds the IRIPPLE(P-P) requirement. In surface mount applications multiple capacitors may have to be paralleled to meet the ESR or RMS current handling requirements of the application. Aluminum electrolytics and dry tantalum capacitors are both available in surface mount configurations. In case of tantalum, it is critical that the capacitors are surge tested for use in switching power supplies. An excellent choice is the AVX TPS, AVX TPSV and KEMET T510 series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Other capacitor types include Sanyo OS-CON, Sanyo POSCAP, Nichicon PL series and Panasonic SP. Efficiency Considerations The efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting efficiency and which change would produce the most improvement. Efficiency can be expressed as: Efficiency = 100% – (L1 + L2 +L3 + ...) where L1, L2, etc. are the individual losses as a percentage of input power. Although all dissipative elements in the circuit produce losses, four main sources usually account for most of the losses in the LTC1771 circuits: the LTC1771 DC bias current, MOSFET gate charge current, I2R losses and catch diode losses. 1. The DC bias current is 9µA at no load and increases proportionally with load up to a constant 150µA during continuous mode. This bias current is so small that this loss is negligible at loads above a milliamp but at no load accounts for nearly all of the loss. 2. The MOSFET gate charge current results from switching the gate capacitance of the power MOSFET switch. Each time the gate is switched from high to low to high again, a packet of charge dQ moves from VIN to ground. The resulting dQ/dt is the current out of VIN which is typically much larger than the DC bias current. In 9 LTC1771 U W U U APPLICATIO S I FOR ATIO continuous mode, IGATECHG = fQP where QP is the gate charge of the internal switch. Both the DC bias and gate charge losses are proportional to VIN and thus their effects will be more pronounced at higher supply voltages. 3. I2R losses are predicted from the internal switch, inductor and current sense resistor. In continuous mode the average output current flows through L but is “chopped” between the P-channel MOSFET in series with RSENSE and the output diode. The MOSFET RDS(ON) plus RSENSE multiplied by the duty cycle can be summed with the resistance of L to obtain I2R losses. 4. The catch diode loss is proportional to the forward drop as the diode conducts current during the off-time and is more pronounced at high supply voltages where the off-time is long. However, as discussed in the Catch Diode section, diodes with lower forward drops often have higher leakage currents, so although efficiency is improved, the no-load supply current will increase. The diode loss is calculated by multiplying the forward voltage drop times the diode duty cycle multiplied by the load current. Other losses including CIN and COUT ESR dissipative losses, and inductor core losses, generally account for less than 2% total additional loss. Output Voltage Programming The output voltage is programmed with an external divider from VOUT to VFB (Pin 1) as shown in Figure 2. The regulated voltage is determined by: VOUT VOUT VOUT R1 + R2 VIN A 5pF feedforward capacitor across R2 is recommended to minimize output voltage ripple in Burst Mode operation. Run/Soft-Start Function The RUN/SS pin is a dual purpose pin that provides the soft- start function and a means to shut down the LTC1771. Soft-start reduces the input surge current from VIN by gradually increasing the internal current limit. Power supply sequencing can also be accomplished using this pin. An internal 1µA current source charges up an external capacitor CSS. When the voltage on the RUN/SS reaches 1V, the LTC1771 begins operating. As the voltage on the RUN/SS continues to ramp from 1V to 2.2V, the internal current limit is also ramped at a proportional linear rate. The current limits begins near 40% maximum load at VRUN/SS = 1V and ends at maximum load at VRUN/SS = 2.2V. The output current thus ramps up slowly, reducing the starting surge current required from the input power supply. If the RUN/SS has been pulled all the way to ground, there will be a delay before the current limit starts increasing and is given by: tDELAY ≈ CSS/ICHG Foldback Current Limiting CFF 5pF R2 VFB R1 GND 1771 F02 Figure 2. LTC1771 Adjustable Configuaration 10 ∆IVIN = where ICHG ≅ 1µA. Pulling the RUN/SS pin below 0.5V puts the LTC1771 into a low quiescent current shutdown (IQ < 2µA). R2 VOUT = 1.23 1 + R1 LTC1771 To minimize no-load supply current, resistor values in the megohm range should be used. The increase in supply current due to the feedback resistors can be calculated from: As described in the Catch Diode Selection, the worst-case dissipation for diode occurs with a short-circuit output, when the diode conducts the current limit value almost continuously. In most applications this will not cause excessive heating, even for extended fault intervals. However, when heat sinking is at a premium or higher forward voltage drop diodes are being used, foldback current LTC1771 U W U U APPLICATIO S I FOR ATIO limiting should be added to reduce the current in proportion to the severity of the fault. Foldback current limiting is implemented by adding two diodes in series between the output and the ITH pin as shown in the Functional Diagram. In a hard short (VOUT = 0V) the current will be reduced to approximately 25% of the maximum output current. Minimum On-Time Considerations Minimum on-time tON(MIN) is the smallest amount of time that the LTC1771 is capable of turning the top MOSFET on and off again. It is determined by internal timing delays and the amount of gate charge required to turn on the P-channel MOSFET. Low duty cycle applications may approach this minimum on-time limit and care should be taken to ensure that: V + VD tON = tOFF OUT > tON(MIN) VIN − VOUT Mode Pin Burst Mode operation is disabled by pulling MODE (Pin 8) below 0.5V. Disabling Burst Mode operation provides a low noise output spectrum, useful for reducing both audio and RF interference. It does this by keeping the frequency constant (for fixed VIN) down to much lower load current (1% to 2% of IMAX) and reducing the amount of output voltage and current ripple at light loads. When Burst Mode operation is disabled, efficiency is reduced at light loads and no load supply current increases to 175µA. Low Supply Operation Although the LTC1771 can function down to 2.8V, the maximum allowable output current is reduced when VIN decreases below 3.2V. Figure 4 shows the amount of change as the supply is reduced below 3.2V, where 100% of maximum load equals 0.1/RSENSE. To ensure adequate output current at VIN < 3.2V, simply lower RSENSE by the same percentage as the current reduction in Figure 4. where tOFF = 3.5µs and tON(MIN) is generally about 0.4µs for the LTC1771. 120 MAXIMUM LOAD (%) As the on-time approaches tON(MIN), the LTC1771 will remain in Burst Mode operation for an increasingly larger portion of the load range (see Figure 3) and at or below tON(MIN) will remain in Burst Mode operation 100% of the time. The output voltage will continue to be regulated, but the ripple current and ripple voltage will increase. 140 100 80 60 40 20 100 0 2.5 % OF MAXIMUM LOAD 80 3.0 3.5 4.0 4.5 INPUT VOLTAGE (V) 5.0 1771 F04 Figure 4. Maximum Load vs Input Voltage 60 40 PC Board Layout Checklist 20 0 0 0.5 1.5 1.0 ON-TIME (µs) 2.0 2.5 1771 F03 Figure 3. Burst Threshold vs On-Time When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1771. These items are also illustrated graphically in the layout diagram of Figure 5. Check the following in your layout: 1. Is the Schottky diode closely connected to the drain of the external MOSFET and the input cap ground? 11 LTC1771 U W U U APPLICATIO S I FOR ATIO 2. Is the 0.1µF input decoupling capacitor closely connected between VIN (Pin 6) and ground (Pin 4)? This capacitor carries the high frequency peak currents. 3. Does the VFB pin connect directly to the feedback resistors? The resistive divider R1 and R2 must be connected between the (+) plate of COUT and signal ground. Locate the feedback resistors right next to the LTC1771. The VFB line should not be routed close to any nodes with high slew rates. 4. Is the 1000pF decoupling capacitor for the current sense resistor connected as close as possible to Pins 6 and 7? Ensure accurate current sensing with Kelvin connections to the sense resistor. 5. Is the (+) plate of CIN closely connected to the sense resistor ? This capacitor provides the AC current to the MOSFET. 6. Are the signal and power grounds segregated? The signal ground consists of the (–) plate of COUT, Pin 4 of the LTC1771 and the resistive divider. The power ground consists of the Schottky diode anode and the (–) plate of CIN which should have as short lead lengths as possible. 7. Keep the switching node (SW) and the gate node (PGATE) away from sensitive small signal nodes, especially the voltage sensing feedback pin (VFB), and minimize their PC trace area. CSS 1 CITH R1 RITH 2 3 4 R2 CFF RUN/SS MODE ITH SENSE LTC1771 VFB GND VIN PGATE 8 MODE 7 6 5 0.1µF CIN D1 + COUT L VOUT BOLD LINES INDICATE HIGH CURRENT PATHS Figure 5. LTC1771 Layout Diagram 12 As a design example, assume VIN = 10V (nominal), VIN = 15V(MAX), VOUT = 3.3V, and IMAX = 2A. With this information, we can easily calculate all the important components. RSENSE = 100mV/2A = 0.05Ω To optimize low current efficiency, MODE pin is tied to VIN to enable Burst Mode operation, thus the minimum inductance necessary is: LMIN = 70µH(3.3V + 0.5)(0.05Ω) = 13.3µH 15µH is chosen for the application. 3.3V + 0.5V ∆IL = 3.5µs = 0.89A 15µH For the feedback resistors, choose R1 = 1M to minimize supply current. R2 can then be calculated to be: R2 = (VOUT/1.23 – 1) • R1 = 1.68M Assume that the MOSFET dissipation is to be limited to PP = 0.25W. If TA = 70°C and the thermal resistance of the MOSFET is 83°C/W, then the junction temperatures will be 91°C and δP = 0.33. The required RDS(ON) for the MOSFET can now be calculated: 0.25W 3.3V + 0.5V 2A 10 V + 0.5V = 0.130Ω P - Channel RDS(ON) = ( ) (1.33) 2 Since the gate of the MOSFET will see the full input voltage, a MOSFET must be selected whose VGS(MAX) > 15V. A P-channel MOSFET that meets both the VGS(MAX) and RDS(ON) requirement is the Si6447DQ. Q1 + Design Example 1771 F05 The most stringent requirement for the Schottky diode occurs when VOUT = 0V (i.e., short circuit) at maximum VIN. In this case the worst-case dissipation rises to: VIN PD = ISC(AVG) (VD) VIN + VD LTC1771 U W U U APPLICATIO S I FOR ATIO With a 0.05Ω sense resistor ISC(AVG) = 2A will result, increasing the 0.5V Schottky diode dissipation to 1W. output ripple. The output voltage ripple due to ESR is approximately: CIN is chosen for a RMS current rating of at least 1A at temperature. COUT is chosen with an ESR of 0.05Ω for low VORIPPLE ≈ (RESR)(∆IL) = 0.05Ω (0.89AP-P) = 45mVP-P U TYPICAL APPLICATIO S 3.3V to 2.5V/1A Regulator with Burst Mode Operation Enabled 0.01µF 1 220pF 10k 2 3 4 RUN/SS MODE ITH SENSE LTC1771 8 7 6 VFB 1000pF VIN GND PGATE 5 RSENSE 0.1Ω Si3443DV 1M 1% 1.02M 1% + L1 22µH + 5pF D1 CIN 33µF 16V COUT 150µF 6.3V VIN 3.3V TO 12V VOUT 2.5V 1A 1771 TA01 CIN: AVX TPSC336M016R0300 COUT: SANYO POSCAP 6TPB150M D1: MICROSEMI UPS5817 L1: SUMIDA CDRH6D38-220 Low Dropout 5V/2A Regulator with Burst Mode Operation Disabled 0.01µF 1 330pF 10k 2 3 4 1M 1% RUN/SS MODE ITH SENSE LTC1771 VFB 8 7 6 1000pF VIN GND PGATE 3.09M 1% 15pF 5 RSENSE 0.05Ω Si6447DQ L1 22µH + + D1 CIN 22µF 25V ×2 COUT 150µF 6.3V VIN 5.5V TO 18V VOUT 5V 2A 1771 TA04 CIN: AVX TPSD226M025R0200 COUT: SANYO POSCAP 6TPB150M D1: MICROSEMI UPS5817 L1: SUMIDA CR75-220 13 LTC1771 U TYPICAL APPLICATIO S Low Dropout Single Cell Lithium-Ion to 3V 0.01µF 1 330pF 10k 2 RUN/SS MODE ITH SENSE 8 7 LTC1771 3 1000pF 6 VFB 4 MODE VIN GND 5 PGATE Si3443DV 1M 1% + RSENSE 0.05Ω L1 15µH 1.43M 1% CIN 47µF 10V + 15pF Li-Ion 3.4V TO 4.2V VOUT 3V 2A COUT 220µF 4V D1 1771 TA02 CIN: TAIYO YUDEN LMK550BJ476MM COUT: SANYO POSCAP 4TPB220M D1: MICROSEMI UPS5817 L1: SUMIDA CR75-150 12V/1A Zeta Converter VIN VIN (V) ILOAD(MAX) (A) 0.01µF 1 3.01M 1% 220pF Q1 10k 2 280k 1% 3 402k 1% 4 RUN/SS MODE ITH SENSE LTC1771 VFB 4.5 5 10 15 18 MODE 7 1000pF 6 0.7 0.9 1.8 2.4 2.6 VIN GND 1M 1% 8 PGATE 5 Si6459DQ 8.66M 1% • L1A 47µH CIN: AVX TPSD226M025R0200 COUT: AVX TPSV107M020R0085 C1: AVX TPSD336M020R0200 D1: MOTOROLA MBRS140T3 L1A, L1B: COILTRONICS VP4-0075, B H ELECTRONICS Q10549 Q1: MOTOROLA MMBT2N2222LT1 CIN 22µF 25V ×2 • + 5pF + RSENSE 0.025Ω C1 33µF 20V ×2 L1B 47µH D1 + COUT 100µF 20V VIN 5V TO 18V VOUT 12V 1A 1771 TA05 2.5V/1A Regulator with Foldback Current Limit 0.01µF 1 220pF 10k 2 3 4 1M 1% RUN/SS MODE ITH SENSE LTC1771 VFB 8 MODE 7 6 1000pF VIN GND PGATE 1.02M 1% 5 + RSENSE 0.1Ω 4 3 2 CIN: AVX TPSC336M016R0300 COUT: SANYO POSCAP 6TPB150M L1: SUMIDA CDRH6D38-220 U1: INTERNATIONAL RECTIFIER FETKY TM IRF7422D2 1 ITH U1 5pF 5 6 7 VIN 2.8V TO 12V CIN 33µF 16V 8 1N4148 ×2 L1 22µH + COUT 150µF 6.3V 1771 TA06 14 VOUT 2.5V 1A LTC1771 U TYPICAL APPLICATIONS 4-NiCd/NiMH Battery Charger 0.01µF 1 220pF 10k 2 3 4 RUN/SS MODE ITH SENSE LTC1771 VFB 8 MODE 7 6 1000pF VIN GND PGATE 5 RSENSE 0.1Ω Si6447DQ 1M 1% 4.69M 1% + L1 47µH + 5pF D1 VIN 8V TO 18V CIN 22µF 25V D2 VOUT 4-NiCd 1A COUT 100µF 10V 1771 TA07 CIN: AVX TPSD226M025R0200 COUT: SANYO POSCAP 10TPB100M D1, D2: MICROSEMI UPS5817 L1: COILTRONICS UP2B-470, GOWANDA SMP3316-472M U PACKAGE DESCRIPTIO Dimension in inches (millimeters) unless otherwise noted. MS8 Package 8-Lead Plastic MSOP (LTC DWG # 05-08-1660) 0.040 ± 0.006 (1.02 ± 0.15) 0.007 (0.18) 0.034 ± 0.004 (0.86 ± 0.102) 0.118 ± 0.004* (3.00 ± 0.102) 8 7 6 5 0° – 6° TYP 0.021 ± 0.006 (0.53 ± 0.015) SEATING PLANE 0.012 (0.30) 0.0256 REF (0.65) BSC 0.006 ± 0.004 (0.15 ± 0.102) 0.118 ± 0.004** (3.00 ± 0.102) 0.193 ± 0.006 (4.90 ± 0.15) MSOP (MS8) 1098 1 * DIMENSION DOES NOT INCLUDE MOLD FLASH, PROTRUSIONS OR GATE BURRS. MOLD FLASH, PROTRUSIONS OR GATE BURRS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE ** DIMENSION DOES NOT INCLUDE INTERLEAD FLASH OR PROTRUSIONS. INTERLEAD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE 2 3 4 S8 Package 8-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.189 – 0.197* (4.801 – 5.004) 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0.053 – 0.069 (1.346 – 1.752) 0°– 8° TYP 0.016 – 0.050 (0.406 – 1.270) 0.014 – 0.019 (0.355 – 0.483) TYP *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 8 7 6 5 0.004 – 0.010 (0.101 – 0.254) 0.050 (1.270) BSC 0.150 – 0.157** (3.810 – 3.988) 0.228 – 0.244 (5.791 – 6.197) SO8 1298 1 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 2 3 4 15 LTC1771 U TYPICAL APPLICATIO 5V/1A Zeta Converter VIN (V) ILOAD(MAX) (A) 2.8 0.8 3.3 1.1 5 1.7 7.5 2.3 10 2.7 12 2.9 0.01µF 1 220pF 10k 2 3 4 1M 1% RUN/SS MODE ITH SENSE LTC1771 VFB 8 MODE 7 6 1000pF VIN GND PGATE 5 Si3443DV 3.09M 1% • + 5pF • L1A 22µH CIN: AVX TPSD336M020R0200 COUT: SANYO POSCAP 10TPB100M C1: AVX TPSD107M010R065 D1: MICROSEMI UPS5817 L1A, L1B: COILTRONICS CTX10-4, BH ELECTRONICS S10-1013 C1 100µF 10V + RSENSE 0.025Ω CIN 33µF 20V ×2 D1 L1B 22µH + COUT 100µF 10V VIN 2.8V TO 12V VOUT 5V 1A 1771 TA03 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1147 Series High Efficiency Step-Down Switching Regulator Controllers 100% DC, 3.5V ≤ VIN ≤ 16V LTC1174/LTC1174-3.3/LTC1174-5 High Efficiency Step-Down and Inverting DC/DC Converters Selectable IPEAK = 300mA or 600mA LTC1265 1.2A High Efficiency Step-Down DC/DC Converter Burst Mode Operation, Internal MOSFET LTC1474/LTC1475 Low Quiscent Current Step-Down Regulators Monolithic, IQ = 10µA, 400mA, MS8 LTC1574/LTC1574-3.3/LTC1574-5 High Efficiency Step-Down DC/DC Converters with Internal Schottky Diode LTC1174 with Internal Schottky Diode LTC1622 Low Input Voltage Step-Down DC/DC Controller Constant Frequency, 2V to 10V VIN, MS8 LTC1624 High Efficiency SO-8 N-Channel Switching Regulator Controller 95% DC, 3.5V to 36V VIN LT®1761 Series 100mA, Low Noise, LDO Micropower Regulators in SOT-23 20µA Quiescent Current, 20µVRMS Noise LT1763 Series 500mA, Low Noise, LDO Micropower Regulators 30µA Quiescent Current, 20µVRMS Noise LTC1772 Constant Frequency Step-Down DC/DC Controller SOT-23, 2.2V to 9.8V VIN LTC1877 High Efficiency Monolithic Step-Down Regulator 550kHz, MS8, VIN Up to 10V, IQ = 10µA, IOUT to 600mA LTC1878 High Efficiency Monolithic Step-Down Regulator 550kHz, MS8, VIN Up to 6V, IQ = 10µA, IOUT to 600mA 16 Linear Technology Corporation 1771f LT/TP 1000 4K • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 2000