ETC 522964H

LTC1265/LTC1265-3.3/LTC1265-5
1.2A, High Efficiency
Step-Down DC/DC Converter
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FEATURES
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DESCRIPTIO
The LTC®1265 is a monolithic step-down current mode
DC/DC converter featuring Burst Mode TM operation at low
output current. The LTC1265 incorporates a 0.3Ω switch
(VIN =10V) allowing up to 1.2A of output current.
High Efficiency: Up to 95%
Current Mode Operation for Excellent Line and Load
Transient Response
Internal 0.3Ω Power Switch (VIN = 10V)
Short-Circuit Protection
Low Dropout Operation: 100% Duty Cycle
Low-Battery Detector
Low 160µA Standby Current at Light Loads
Active-High Micropower Shutdown: IQ < 15µA
Peak Inductor Current Independent of Inductor Value
Available in 14-pin SO Package
Under no load condition, the converter draws only 160µA.
In shutdown it typically draws a mere 5µA making this
converter ideal for current sensitive applications. In dropout the internal P-channel MOSFET switch is turned on
continuously maximizing the life of the battery source. The
LTC1265 incorporates automatic power saving Burst Mode
operation to reduce gate charge losses when the load
currents drop below the level required for continuous
operation.
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APPLICATIO S
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5V to 3.3V Conversion
Distributed Power Systems
Step-Down Converters
Inverting Converters
Memory Backup Supply
Portable Instruments
Battery-Powered Equipment
Cellular Telephones
The inductor current is user-programmable via an external
current sense resistor. Operation up to 700kHz permits
the use of small surface mount inductors and capacitors.
, LTC and LT are registered trademarks of Linear Technology Corporation.
Burst Mode is a trademark of Linear Technology Corporation.
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TYPICAL APPLICATIO
+
LTC1265-5 Efficiency
CIN†††
68µF
20V
0.1µF
SW
SHDN
1k
3900pF
130pF
L1*
33µH
VIN
PWR VIN
D1†
LTC1265-5
ITH
CT
100
RSENSE**
0.1Ω
PGND
VOUT
5V
1A
+
COUT††
220µF
10V
SENSE+
1000pF
SENSE
SGND
95
90
VIN = 12V
85
80
–
* COILTRONICS CTX33-4
** IRC LRC2010-01-R100-J
† MBRS130LT3
†† AVX TPSE227K010
††† AVX TPSE686K020
LTC1265-FO1
VIN = 6V
VIN = 9V
EFFICIENCY (%)
VIN
5.4V TO
12V
75
70
0.01
L = 33µH
VOUT = 5V
RSENSE = 0.1Ω
CT = 130pF
0.10
LOAD CURRENT (A)
1.00
LTC1265 TA01
Figure 1. High Efficiency Step-Down Converter
1
LTC1265/LTC1265-3.3/LTC1265-5
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ABSOLUTE
RATI GS
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PACKAGE/ORDER I FOR ATIO
(Voltages Refer to GND Pin) (Note 1)
ORDER PART
NUMBER
TOP VIEW
Input Supply Voltage (Pins 1, 2, 13) ..........– 0.3V to 13V
DC Switch Current (Pin 14) .................................... 1.2A
Peak Switch Current (Pin 14) ................................. 1.6A
Switch Voltage (Pin 14) .................................. VIN – 13.0
Operating Temperature Range
LTC1265C ............................................... 0° to 70°C
LTC1265I ........................................ – 40°C to 85°C
Junction Temperature (Note 2) ............................. 125°C
Storage Temperature Range ....................– 65° to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
PWR VIN 1
VIN 2
14 SW
13 PWR VIN
LBOUT 3
12 PGND
LBIN 4
11 SGND
CT 5
10 SHDN
ITH 6
9
N/C (VFB*)
SENSE– 7
8
SENSE+
LTC1265CS
LTC1265CS-5
LTC1265CS-3.3
LTC1265IS
S PACKAGE
14-LEAD PLASTIC SO
*ADJUSTABLE OUTPUT VERSION
TJMAX = 125°C, θJA = 110°C/W
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VSHDN = 0V, unless otherwise specified.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
IFB
Feedback Current into Pin 9
LTC1265
VFB
Feedback Voltage
LTC1265C
VIN = 9V, LTC1265I
VOUT
Regulator Output Voltage
LTC1265-3.3: ILOAD = 800mA
LTC1265-5: ILOAD = 800mA
∆VOUT
Output Voltage Line Regulation
VIN = 6.5V to 10V, ILOAD = 800mA
Output Voltage Load Regulation
0.2
1
µA
●
●
1.22
1.20
1.25
1.25
1.28
1.30
V
V
●
●
3.22
4.9
3.3
5
3.40
5.2
V
V
–40
0
40
mV
LTC1265-3.3: 10mA < ILOAD < 800mA
LTC1265-5: 10mA < ILOAD < 800mA
40
60
65
100
mV
mV
Burst Mode Operation Output Ripple
ILOAD = 0mA
50
IQ
Input DC Supply Current (Note 3)
Active Mode: 3.5V < VIN < 10V
Sleep Mode: 3.5V < VIN < 10V
Sleep Mode: 5V < VIN < 10V (LTC1265-5)
Shutdown: VSHDN = VIN, 3.5V < VIN < 10V
1.8
160
160
5
2.4
230
230
15
mA
µA
µA
µA
VLBTRIP
Low-Battery Trip Point
1.15
1.25
1.35
V
ILBIN
Current into Pin 4
0.5
µA
ILBOUT
Current Sunk by Pin 3
VLBOUT = 0.4V, VLBIN = 0V
VLBOUT = 5V, VLBIN = 10V
0.5
1.0
1.5
1.0
mA
µA
V8 – V 7
Current Sense Threshold Voltage
LTC1265: VSENSE– = 5V, V9 = VOUT/4 + 25mV (Forced)
VSENSE– = 5V, V9 = VOUT/4 – 25mV (Forced)
LTC1265-3.3: VSENSE– = VOUT + 100mV (Forced)
VSENSE– = VOUT – 100mV (Forced)
LTC1265-5: VSENSE– = VOUT + 100mV (Forced)
VSENSE– = VOUT – 100mV (Forced)
RON
ON Resistance of Switch
LTC1265C
LTC1265I
I5
CT Pin Discharge Current
VOUT in Regulation, VSENSE– = VOUT
VOUT = 0V
tOFF
Switch Off Time (Note 4)
CT = 390pF, ILOAD = 800mA (LTC1265C)
CT = 390pF, ILOAD = 800mA (LTC1265I)
2
UNITS
mVP-P
25
150
25
150
25
150
180
mV
mV
mV
mV
mV
mV
0.3
0.3
0.60
0.70
Ω
Ω
40
60
2
100
10
µA
µA
4
3.5
5
5
6
7
µs
µs
130
130
130
●
●
●
MAX
180
180
LTC1265/LTC1265-3.3/LTC1265-5
ELECTRICAL CHARACTERISTICS
The ● denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VSHDN = 0V, unless otherwise specified.
SYMBOL
PARAMETER
CONDITIONS
MIN
VIH
Shutdown Pin High
Min Voltage at Pin 10 for Device to be in Shutdown
1.2
VIL
Shutdown Pin Low
Max Voltage at Pin 10 for Device to be Active
0.6
V
I10
Shutdown Pin Input Current
VSHDN = 8V
0.5
µA
Note 1: Absolute Maximum Ratings are those values beyond which the life
of the device may be impaired.
Note 2: TJ is calculated from the ambient temperature TA and power
dissipation PD according to the following formulas:
LTC1265CS, LTC1265CS-3.3, LTC1265CS-5:
TJ = TA + (PD • 110°C/W)
TYP
MAX
UNITS
V
Note 3: Dynamic supply current is higher due to the gate charge being
delivered at the switching frequency.
Note 4: In applications where RSENSE is placed at ground potential, the
off time increases by approximately 40%.
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TYPICAL PERFOR A CE CHARACTERISTICS
Efficiency vs Input Voltage
(VOUT = 5V)
Efficiency vs Load Current
100
100
100
VIN = 9V
VIN = 12V
80
LTC1265-3.3
VOUT = 3.3V
RSENSE = 0.1Ω
CT = 130pF
COIL = CTX33-4
75
70
0.01
ILOAD = 250mA
94
92
EFFICIENCY (%)
EFFICIENCY (%)
ILOAD = 800mA
90
88
LTC1265-5
RSENSE = 0.1Ω
CT = 130pF
COIL = CTX33-4
84
82
1.00
4
5
6
80
7
9 10 11
8
INPUT VOLTAGE (V)
25°C
70°C
0.4
300
0.8
270
0.7
240
TJ = 125°C
0.5
TJ = 70°C
0.4
0.3
TJ = 25°C
0.2
TJ = 0°C
0.2
2
3
5 6 7 8
(VIN – VOUT) VOLTAGE (V)
4
9
10
1265 G04
7
9 10 11
8
INPUT VOLTAGE (V)
12
13
VIN = 12V
210
180
150
120
90
30
0
1
6
60
0.1
0
5
Switch Leakage Current
0.9
0.6
0.6
0
4
13
Switch Resistance
RDS(ON) (Ω)
NORMALIZED FREQUENCY
12
LTC1265 G03
1.2
0.8
ILOAD = 800mA
1265 G02
Operating Frequency
vs (VIN – VOUT)
0°C
88
82
1265 G01
1.0
ILOAD = 250mA
90
84
80
0.10
LOAD CURRENT (A)
94
92
86
86
LEAKAGE CURRENT (nA)
EFFICIENCY (%)
90
85
96
96
VIN = 5V
LTC1265-3.3
RSENSE = 0.1Ω
CT = 130pF
COIL = CTX33-4
98
98
95
Efficiency vs Input Voltage
(VOUT = 3.3V)
3
4
5
6 7 8 9 10 11 12 13
INPUT VOLTAGE (V)
1265 G05
0
0
20
60
40
TEMPERATURE (°C)
80
100
1265 G06
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LTC1265/LTC1265-3.3/LTC1265-5
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TYPICAL PERFOR A CE CHARACTERISTICS
DC Supply Current
DOES NOT INCLUDE
GATE CHARGE
1.8
ACTIVE MODE
1.5
1.2
0.9
0.6
0.3
SLEEP MODE
10
4
8
6
INPUT VOLTAGE (V)
2
SHUTDOWN = 3V
TA = 25C
5.0
6
5
4
3
2
1
0
0
Gate Charge Losses
5.5
7
SUPPLY CURRENT (µA)
SUPPLY CURRENT (mA)
Supply Current in Shutdown
8
SWITCHING CURRENT (mA)
2.1
12
14
0
VIN = 12V
4.5
4.0
3.5
VIN = 9V
3.0
2.5
2.0
VIN = 6V
1.5
1.0
0.5
0
3
4
5
6 7 8 9 10 11 12 13
INPUT VOLTAGE (V)
1265 G07
1265 G08
0
200
400
600
FREQUENCY (kHz)
800
1000
1265 G09
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PI FU CTIO S
PWR VIN (Pins 1, 13): Supply for the Power MOSFET and
its Driver. Must decouple this pin properly to ground. Must
always tie Pins 1 and 13 together.
SENSE+ (Pin 8): The (+) Pin to the Current Comparator. A
built-in offset between Pins 7 and 8 in conjunction with
RSENSE sets the current trip threshold.
VIN (Pin 2): Main Supply for All the Control Circuitry in the
LTC1265.
N/C,VFB (Pin 9): For the LTC1265 adjustable version, this
pin serves as the feedback pin from an external resistive
divider used to set the output voltage. On the LTC1265-3.3
and LTC1265-5 versions, this pin is not used.
LBOUT (Pin 3): Open-Drain Output of the Low-Battery
Comparator. This pin will sink current when Pin 4 (LBIN)
goes below 1.25V. During shutdown, this pin is high
impedance.
LBIN (Pin 4): The (–) Input of the Low-Battery Comparator.
The (+) input is connected to a reference voltage of 1.25V.
CT (Pin 5): External capacitor CT from Pin 5 to ground sets
the switch off time. The operating frequency is dependent
on the input voltage and CT.
ITH (Pin 6): Feedback Amplifier Decoupling Point. The
current comparator threshold is proportional to Pin 6
voltage.
SENSE – (Pin 7): Connect to the (–) input of the current
comparator. For LTC1265-3.3 and LTC1265-5, it also
connects to an internal resistive divider which sets the
output voltage.
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SHDN (Pin 10): Pulling this pin HIGH keeps the internal
switch off and puts the LTC1265 in micropower shutdown. Do not float this pin.
SGND (Pin 11): Small-Signal Ground. Must be routed
separately from other grounds to the (–) terminal of COUT.
PGND (Pin 12): Switch Driver Ground. Connects to the
(–) terminal of CIN. Anode of the Schottky diode must be
connected close to this pin.
SW (Pin 14): Drain of the P-Channel MOSFET Switch.
Cathode of the Schottky diode must be connected close to
this pin.
LTC1265/LTC1265-3.3/LTC1265-5
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(Pin 9 connection shown for LTC1265-3.3 and LTC1265-5; change create LTC1265)
1, 13 PWR VIN
SENSE +
SENSE –
8
7
14 SW
12 PGND
–
9
V
VFB
ADJUSTABLE
VERSION
+
–
SLEEP
R
25mV TO 150mV
C
Q
5pF
+
VOS
+
S
–
VTH2
–
S
13k
–
G
100k
+
VTH1
ITH 6
T
2 VIN
+
3 LB0UT
+
OFF-TIME
CONTROL
VFB
11 SGND
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OPERATIO
REFERENCE
A3
SENSE–
10
SHDN
–
5 CT
4 LBIN
1265 FD
(Refer to Functional Diagram)
The LTC1265 uses a constant off-time architecture to
switch its internal P-channel power MOSFET. The off time
is set by an external timing capacitor at CT (Pin 5). The
operating frequency is then determined by the off time and
the difference between VIN and VOUT.
the voltage across the shunt reaches the comparator’s
threshold value, its output signal will change state, setting
the flip flop and turning the internal P-channel MOSFET off.
The timing capacitor connected to Pin 5 is now allowed to
discharge at a rate determined by the off-time controller.
The output voltage is set by an internal resistive divider
(LTC1265-3.3 and LTC1265-5) connected to SENSE –
(Pin 7) or an external divider returned to V FB (Pin 9 for
LTC1265). A voltage comparator V, and a gain block G,
compare the divided output voltage with a reference
voltage of 1.25V.
When the voltage on the timing capacitor has discharged
past VTH1, comparator T trips, sets the flip flop and causes
the switch to turn on. Also, the timing capacitor is recharged. The inductor current will again ramp up until the
current comparator C trips. The cycle then repeats.
To optimize efficiency, the LTC1265 automatically switches
between continuous and Burst Mode operation. The voltage comparator is the primary control element when the
device is in Burst Mode operation, while the gain block
controls the output voltage in continuous mode.
When the load is heavy, the LTC1265 is in continuous
operation. During the switch ON time, current comparator
C monitors the voltage between Pins 7 and 8 connected
across an external shunt in series with the inductor. When
When the load current increases, the output voltage decreases slightly. This causes the output of the gain stage
(Pin 6) to increase the current comparator threshold, thus
tracking the load current.
When the load is relatively light, the LTC1265 automatically goes into Burst Mode operation. The current loop is
interrupted when the output voltage exceeds the desired
regulated value. The hysteretic voltage comparator V trips
when VOUT is above the desired output voltage, shutting
off the switch and causing the capacitor to discharge. This
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LTC1265/LTC1265-3.3/LTC1265-5
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OPERATIO (Refer to Functional Diagram)
capacitor discharges past VTH1 until its voltage drops
below VTH2. Comparator S then trips and a sleep signal is
generated. The circuit now enters into sleep mode with the
power MOSFET turned off. In sleep mode, the LTC1265 is
in standby and the load current is supplied by the output
capacitor. All unused circuitry is shut off, reducing quiescent current from 2mA to 160µA. When the output capacitor discharges by the amount of the hysteresis of the
comparator V, the P-channel switch turns on again and the
process repeats itself. During Burst Mode operation the
peak inductor current is set at 25mV/RSENSE.
To avoid the operation of the current loop interfering with
Burst Mode operation, a built-in offset VOS is incorporated
in the gain stage. This prevents the current from increasing until the output voltage has dropped below a minimum
threshold.
Using constant off-time architecture, the operating frequency is a function of the voltage. To minimize the
frequency variation as dropout is approached, the off-time
controller increases the discharge current as VIN drops
below VOUT + 2V. In dropout the P-channel MOSFET is
turned on continuously (100% duty cycle) providing low
dropout operation with VOUT ≅ VIN.
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APPLICATIO S I FOR ATIO
The basic LTC1265 application circuit is shown in
Figure 1. External component selection is driven by the
load requirement, and begins with the selection of RSENSE.
Once RSENSE is known, CT and L can be chosen. Next, the
Schottky diode D1 is selected followed by CIN and COUT.
RSENSE Selection for Output Current
RSENSE is chosen based on the required output current.
With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator
determines the peak inductor current. Depending on the
load current condition, the threshold of the comparator
lies between 25mV/RSENSE and 150mV/RSENSE. The maximum output current of the LTC1265 is:
25mV (Amps)
IOUT(MAX) = 150mV –
RSENSE
2 • RSENSE
= 137.5mV (Amps)
RSENSE
Solving for RSENSE and allowing a margin of variations in
the LTC1265 and extended component values yields:
RSENSE = 100mV (Ω)
IOUT(MAX)
The LTC1265 is rated with a capability to supply a maximum
of 1.2A of output current. Therefore, the minimum value of
RSENSE that can be used is 0.083Ω. A graph for selecting
RSENSE versus maximum output is given in Figure 2.
IOUT(MAX) = 150mV – IRIPPLE (Amps)
RSENSE
2
0.5
where IRIPPLE is the peak-to-peak inductor ripple current.
To account for light and heavy load conditions, the IOUT(MAX)
is then given by:
RSENSE (Ω)
At a relatively light load, the LTC1265 is in Burst Mode
operation. In this mode the peak inductor current is set at
25mV/RSENSE. To fully benefit from Burst Mode operation,
the inductor current should be continuous during burst
periods. Hence, the peak-to-peak inductor ripple current
must not exceed 25mV/RSENSE.
0.4
0.3
0.2
0.1
0
0
0.2
0.6
0.8
0.4
MAXIMUM OUTPUT CURRENT (A)
1
1265 G10
Figure 2. Selecting RSENSE
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LTC1265/LTC1265-3.3/LTC1265-5
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APPLICATIO S I FOR ATIO
Under short-circuit condition, the peak inductor current is
determined by:
ISC(PK) = 150mV (Amps)
RSENSE
2V, the LTC1265 reduces tOFF by increasing the discharge
current in CT. This prevents audible operation prior to
dropout. (See shelving effect shown in the Operating
Frequency curve under Typical Performance Characteristics.)
In this condition, the LTC1265 automatically extends the
off time of the P-channel MOSFET to allow the inductor
current to decay far enough to prevent any current buildup. The resulting ripple current causes the average shortcircuit current to be approximately IOUT(MAX).
To maintain continuous inductor current at light load, the
inductor must be chosen to provide no more than 25mV/
RSENSE of peak-to-peak ripple current. This results in the
following expression for L:
CT and L Selection for Operating Frequency
Using an inductance smaller than the above value will
result in the inductor current being discontinuous. A
consequence of this is that the LTC1265 will delay entering
Burst Mode operation and efficiency will be degraded at
low currents.
The LTC1265 uses a constant off-time architecture with
tOFF determined by an external capacitor CT. Each time the
P-channel MOSFET turns on, the voltage on CT is reset to
approximately 3.3V. During the off time, CT is discharged
by a current that is proportional to VOUT. The voltage on CT
is analogous to the current in inductor L, which likewise,
decays at a rate proportional to VOUT. Thus the inductor
value must track the timing capacitor value.
The value of CT is calculated from the desired continuous
mode operating frequency:
CT =
1
1.3(104)f
)
VIN – VOUT
VIN + VD
)
(Farads)
where VD is the drop across the Schottky diode.
As the operating frequency is increased, the gate charge
losses will reduce efficiency. The complete expression for
operating frequency is given by:
)
)
tOFF = 1.3(104)CT
) )
f≈
VIN – VOUT
(Hz)
tOFF VIN + VD
1
where:
VREG
(sec)
VOUT
VREG is the desired output voltage (i.e. 5V, 3.3V). VOUT is
the measured output voltage. Thus VREG/VOUT = 1
in regulation.
Note that as VIN decreases, the frequency decreases.
When the input-to-output voltage differential drops below
L ≥ 5.2(105)RSENSE (CT)VREG
Inductor Core Selection
With the value of L selected, the type of inductor must be
chosen. Basically, there are two kinds of losses in an
inductor; core and copper losses.
Core losses are dependent on the peak-to-peak ripple
current and core material. However it is independent of
the physical size of the core. By increasing the inductance, the peak-to-peak inductor ripple current will decrease, therefore reducing core loss. Utilizing low core
loss material, such as molypermalloy or Kool Mµ® will
allow user to concentrate on reducing copper loss and
preventing saturation.
Although higher inductance reduces core loss, it increases copper loss as it requires more windings. When
space is not at a premium, larger wire can be used to
reduce the wire resistance. This also prevents excessive
heat dissipation.
CATCH DIODE SELECTION
Losses in the catch diode depend on forward drop and
switching times. Therefore Schottky diodes are a good
choice for low drop and fast switching times.
The catch diode carries load current during the off time.
The average diode current is therefore dependent on the
Kool Mµ is a registered trademark of Magnetics, Inc.
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LTC1265/LTC1265-3.3/LTC1265-5
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APPLICATIO S I FOR ATIO
P-channel switch duty cycle. At high input voltages, the
diode conducts most of the time. As VIN approaches VOUT,
the diode conducts only a small fraction of the time. The
most stressful condition for the diode is when the output
is short circuited. Under this condition, the diode must
safely handle ISC(PK) at close to 100% duty cycle. Most
LTC1265 circuits will be well served by either a 1N5818 or
a MBRS130LT3 Schottky diode. An MBRS0520 is a good
choice for IOUT(MAX) ≤ 500mA.
CIN
In continuous mode, the input current of the converter is
a square wave of duty cycle VOUT/ VIN. To prevent large
voltage transients, a low ESR input capacitor must be
used. In addition, the capacitor must handle a high RMS
current. The CIN RMS current is given by:
1
IOUT [VOUT (VIN – VOUT)] /2
(ARMS)
IRMS ≈
VIN
This formula has a maximum at VIN = 2VOUT, where IRMS
= IOUT/2. This simple worst case is commonly used for
design because even significant deviations do not offer
much relief. Note that capacitor manufacturer’s ripple
current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature
than required. Do not underspecify this component. An
additional 0.1µF ceramic capacitor is also required on
PWR VIN for high frequency decoupling.
COUT
The selection of COUT is based upon the effective series
resistance (ESR) for proper operation of the LTC1265. The
required ESR of COUT is:
ESRCOUT < 50mV/IRIPPLE
where IRIPPLE is the ripple current of the inductor. For the
case where the IRIPPLE is 25mV/RSENSE, the required ESR
of COUT is:
ESRCOUT < 2(RSENSE)
To avoid overheating, the output capacitor must be sized
to handle the ripple current generated by the inductor. The
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worst-case RMS ripple current in the output capacitor is
given by:
IRMS ≈
150mV
(ARMS)
2(RSENSE)
Generally, once the ESR requirement for COUT has been
met, the RMS current rating far exceeds the IRIPPLE(P-P)
requirement.
ESR is a direct function of the volume of the capacitor.
Manufacturers such as Nichicon, AVX and Sprague should
be considered for high performance capacitors. The
OS-CON semiconductor dielectric capacitor available
from Sanyo has the lowest ESR for its size at a somewhat
higher price.
In surface mount applications, multiple capacitors may
have to be paralleled to meet the capacitance, ESR or RMS
current handling requirement of the application. Aluminum electrolyte and dry tantalum capacitors are both
available in surface mount configurations. In the case of
tantalum, it is critical that the capacitors are both available
in surface mount configuration and are surge tested for
use in switching power supplies. An excellent choice is the
AVX TPS series of surface mount tantalums, available in
case heights ranging from 2mm to 4mm. Consult the
manufacturer for other specific recommendations.
When the capacitance of COUT is made too small, the
output ripple at low frequencies will be large enough to trip
the voltage comparator. This causes Burst Mode operation to be activated when the LTC1265 would normally be
in continuous operation. The effect will be most pronounced with low value of RSENSE and can be improved at
higher frequencies with lower values of L.
Low-Battery Detection
The low-battery comparator senses the input voltage
through an external resistive divider. This divided voltage
connects to the (–) input of a voltage comparator (Pin 4)
which is compared with a 1.25V reference voltage. Neglecting Pin 4 bias current, the following expression is
used for setting the trip limit:
)
R4
VLB_TRIP = 1.25 1 +
R3
)
LTC1265/LTC1265-3.3/LTC1265-5
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APPLICATIO S I FOR ATIO
The output, Pin 3, is an N-channel open drain that goes low
when the battery voltage is below the threshold set by R3
and R4. In shutdown, the comparator is disabled and Pin
3 is in a high impedance state.
VIN
R4
LTC1265
4
3
–
+
R3
1.25V REFERENCE
LTC1265 F03
Figure 3. Low-Battery Comparator
LTC1265 ADJUSTABLE APPLICATIONS
The LTC1265 develops a 1.25V reference voltage between
the feedback (Pin 9) terminal and signal ground (see
Figure 4). By selecting resistor R1, a constant current is
caused to flow through R1 and R2 to set overall output
voltage. The regulated output voltage is determined by:
)
R2
VOUT = 1.25 1 +
R1
)
For most applications a 30k resistor is suggested for R1.
To prevent stray pickup, a 100pF capacitor is suggested
across R1 located close to the LTC1265.
Absolute Maximum Ratings and Latchup Prevention
The absolute maximum ratings specify that SW (Pin 14)
can never exceed VIN (Pins 1, 2, 13) by more than 0.3V.
Normally this situation should never occur. It could,
however, if the output is held up while the VIN supply is
pulled down. A condition where this could potentially
occur is when a battery is supplying power to an LTC1265
regulator and also to one or more loads in parallel with the
the regulator’s VIN. If the battery is disconnected while the
LTC1265 regulator is supplying a light load and one of the
parallel circuits has a heavy load, the input capacitor of the
LTC1265 regulator could be pulled down faster than the
output capacitor, causing the absolute maximum ratings
to be exceeded. The result is often a latchup which can be
destructive if VIN is reapplied quickly. Battery disconnect
is possible as a result of mechanical stress, bad battery
contacts or use of a lithium-ion battery with a built-in
internal disconnect. The user needs to assess his/her
application to determine whether this situation could
occur. If so, additional protection is necessary.
Prevention against latchup can be accomplished by
simply connecting a Schottky diode across the SW and
VIN pins as shown in Figure 5. The diode will normally be
reverse biased unless VIN is pulled below VOUT at which
time the diode will clamp the (VOUT – VIN) potential to less
than the 0.6V required for latchup. Note that a low leakage
Schottky should be used to minimize the effect on noload supply current. Schottky diodes such as MBR0530,
BAS85 and BAT84 work well. Another more serious
effect of the protection diode leakage is that at no load
with nothing to provide a sink for this leakage current, the
VOUT
LATCHUP
PROTECTION
SCHOTTKY
R2
LTC1265 VFB
SGND
9
100pF
R1
11
PWR
VIN
VOUT
SW
+
LTC1265
LTC1265 F04
1265 F05
Figure 4. LTC1265 Adjustable Configuration
Figure 5. Preventing Absolute Maximum
Ratings from Being Exceeded
9
LTC1265/LTC1265-3.3/LTC1265-5
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APPLICATIONS INFORMATION
output voltage can potentially float above the maximum
allowable tolerance. To prevent this from occuring, a
resistor must be connected between VOUT and ground
with a value low enough to sink the maximum possible
leakage current.
THERMAL CONSIDERATIONS
In a majority of applications, the LTC1265 does not
dissipate much heat due to its high efficiency. However, in
applications where the switching regulator is running at
high duty cycles or the part is in dropout with the switch
turned on continuously (DC), the user will need to do some
thermal analysis. The goal of the thermal analysis is to
determine whether the power dissipated by the regulator
exceeds the maximum junction temperature of the part.
The temperature rise is given by:
TR = P(θJA)
where P is the power dissipated by the regulator and θJA
is the thermal resistance from the junction of the die to the
ambient temperature.
The junction temperature is simply given by:
T J = TR + T A
As an example, consider the LTC1265 is in dropout at an
input voltage of 4V with a load current of 0.5A. From the
Typical Performance Characteristics graph of Switch Resistance, the ON resistance of the P-channel is 0.55Ω.
Therefore power dissipated by the part is:
P = I2(RDSON) = 0.1375W
For the SO package, the θJA is 110°C/W.
Therefore the junction temperature of the regulator when
it is operating in ambient temperature of 25°C is:
TJ = 0.1375(110) + 25 = 40.1°C
Remembering that the above junction temperature is
obtained from a RDSON at 25°C, we need to recalculate the
junction temperature based on a higher RDSON since it
increases with temperature. However, we can safely assume that the actual junction temperature will not exceed
the absolute maximum junction temperature of 125°C.
10
Now consider the case of a 1A regulator with VIN = 4V and
TA = 65°C. Starting with the same 0.55Ω assumption for
RDSON, the TJ calculation will yield 125°C. But from the
graph, this will increase the RDSON to 0.76Ω, which when
used in the above calculation yields an actual TJ > 148°C.
Therefore the LTC1265 would be unsuitable for a 4V input,
1A output regulator operating at TA = 65°C.
Board Layout Checklist
When laying out the printed circuit board, the following
checklist should be used to ensure proper operation of the
LTC1265. These items are also illustrated graphically in
the layout diagram of Figure 6. Check the following in your
layout:
1. Are the signal and power grounds segregated? The
LTC1265 signal ground (Pin 11) must return to the (–)
plate of COUT. The power ground (Pin 12) returns to the
anode of the Schottky diode, and the (–) plate of CIN,
whose leads should be as short as possible.
2. Does the (+) plate of the CIN connect to the power VIN
(Pins 1,13) as close as possible? This capacitor provides the AC current to the internal P-channel MOSFET
and its driver.
3. Is the input decoupling capacitor (0.1µF) connected
closely between power VIN (Pins 1,13) and power
ground (Pin 12)? This capacitor carries the high frequency peak currents.
4. Is the Schottky diode closely connected between the
power ground (Pin 12) and switch (Pin 14)?
5. Does the LTC1265 SENSE – (Pin 7) connect to a point
close to RSENSE and the (+) plate of COUT? In adjustable
applications, the resistive divider, R1 and R2, must be
connected between the (+) plate of COUT and signal
ground.
6. Are the SENSE – and SENSE + leads routed together with
minimum PC trace spacing? The 1000pF capacitor
between Pins 7 and 8 should be as close as possible to
the LTC1265.
7. Is SHDN (Pin 10) actively pulled to ground during
normal operation? The SHDN pin is high impedance
and must not be allowed to float.
LTC1265/LTC1265-3.3/LTC1265-5
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APPLICATIO S I FOR ATIO
1
PWR VIN
VIN
2
VIN
SW
PWR VIN
14
D1
13
+
LTC1265
1k
3900pF
1000pF
4
5
6
7
LBOUT
PGND
LBIN
SGND
CT
SHDN
ITH
N/C (VFB)
SENSE–
CIN
12
0.1µF
10
R1
SHDN
9
SENSE+
L
11
+
3
8
COUT
RSENSE
R2
VOUT
1000pF
OUTPUT DIVIDER REQUIRED
WITH ADJUSTABLE VERSION ONLY
LTC1265 F06
BOLD LINES INDICATE
HIGH PATH CURRENTS
Figure 6. LTC1265 Layout Diagram (See Board Layout Checklist)
In continuous mode the voltage on the CT pin is a sawtooth
with approximately 0.9VP-P swing. This voltage should
never dip below 2V as shown in Figure 7a.
When the load currents are low (ILOAD < IBURST) Burst
Mode operation occurs. The voltage on CT pin now falls to
ground for periods of time as shown in Figure 7b. During
this time the LTC1265 is in sleep mode with quiescent
current reduced to 160µA.
The inductor current should also be monitored. If the
circuit is poorly decoupled, the peak inductor current will
be haphazard as in Figure 8a. A well decoupled LTC1265
has a clean inductor current as in Figure 8b.
3.3V
2.4V
0V
TIME
(a) CONTINUOUS MODE OPERATION
LTC1265 F07a
SLEEP MODE
VOLTAGE AT CT (PIN 5)
Since efficiency is critical to LTC1265 applications, it is
very important to verify that the circuit is functioning
correctly in both continuous and Burst Mode operation. As
the LTC1265 is highly tolerant of poor layout, the output
voltage will still be regulated. Therefore, monitoring the
output voltage will not tell you whether you have a good or
bad layout. The waveform to monitor is the voltage on the
timing capacitor Pin 5.
VOLTAGE AT CT (PIN 5)
Troubleshooting Hints
3.3V
2.4V
0V
TIME
(b) Burst Mode OPERATION
LTC1265 F07b
Figure 7. CT Waveforms
11
LTC1265/LTC1265-3.3/LTC1265-5
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APPLICATIO S I FOR ATIO
(a) POORLY DECOUPLED LTC1265
(b) WELL DECOUPLED LTC1265
Figure 8. Inductor Waveforms
Design Example
VIN
5V
As a design example, assume VIN = 5V, VOUT = 3.3V, IMAX
= 0.8A and f = 250kHz. With this information we can easily
calculate all the important components.
+
CIN
VIN
PWR VIN
0.1µF
From (1),
LTC1265-3.3
1k
ITH
RSENSE = 100mV/0.8 = 0.125Ω
From (2) and assuming VD = 0.4V,
3900pF 100pF
22µH
D1
+
COUT
PGND
SENSE +
CT
CT ≅ 100pF
1000pF
SENSE –
Using (3), the value of the inductor is:
LTC1265 F09
SGND
L ≥ 5.2(105)(0.125)(100pF)3.3V = 22µH
For the catch diode, a MBRS130LT3 or 1N5818 will be
sufficient in this application.
Figure 9. Design Example Circuit
CIN will require an RMS current rating of at least 0.4A at
temperature, and COUT will require an ESR of (from 5):
100
ESRCOUT < 0.25Ω
The inductor ripple current is given by:
IRIPPLE =
)
)
VOUT + VD
tOFF = 0.22A
L
At light loads the peak inductor current is at:
IPEAK = 25mV/0.125 = 0.2A
Therefore, at load current less than 0.1A the LTC1265 will
be in Burst Mode operation. Figure 9 shows the complete
circuit and Figure 10 shows the efficiency curve with the
above calculated component values.
12
EFFICIENCY (% )
95
L = DALE LPT4545-220 (22µH)
VOUT = 3.3V
CT = 100pF
90
85
80
75
70
0.01
VOUT
3.3V
0.8A
0.125Ω
SW
SHDN
0.1
LOAD CURRENT (mA)
1.0
1265 G11
Figure 10. Design Example Efficiency Curve
LTC1265/LTC1265-3.3/LTC1265-5
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TYPICAL APPLICATIO S
High Efficiency 5V to 3.3V Converter
VIN
5V
2
4
3
270pF
5
3900pF
14
SW
LBIN
LTC1265-3.3
LBOUT
PGND
CT
SGND
6
7
*AVX TPSD107K010
**AVX TPSE227K010
†
COILCRAFT D03316-473
††
DALE WSL2010-0.1-1%
SENSE –
L1†
47µH
RSENSE††
0.1Ω
+
MBRS130LT1
12
VOUT
3.3V
COUT** 1A
220µF
10V
11
10
SHDN
9
NC
ITHR
CIN*
100µF
10V
0.1µF
PWR VIN
SHDN
1k
+
1, 13
VIN
8
SENSE +
1000pF
LTC1265 TA02
Positive-to-Negative (–5V) Converter
VIN
3.5V TO 7.5V
MANUFACTURER
PART NO.
COILCRAFT
COILTRONICS
DALE
SUMIDA
DO3316-473
CTX50-4
LPT4545-500LA
CD74-470
2
4
3
220pF
5
††
IRC LRC2010-01-R100-J
D1= MBRS130LT3
VIN (V)
IOUT(MAX) (mA)
3.5
4.0
5.0
6.0
7.0
7.5
360
430
540
630
720
740
2200pF
1k
6
7
1, 13
L1†
50µH
VIN
LBIN
LTC1265-5
LBOUT
PGND
CT
SGND
ITHR
SHDN
SENSE –
SENSE +
1000pF
CIN*
22µF
25V
×2
+
SHDN
TP0610L
PWR VIN
14
SW
0.1µF
D1
12
VOUT
–5V
11
10
100k
+
*AVX TPSD226K025
**AVX TPSD107K010
†
L1 SELECTION
8
COUT**
100µF
10V
RSENSE††
0.1Ω
LTC1265 TA03
13
LTC1265/LTC1265-3.3/LTC1265-5
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TYPICAL APPLICATIO S
5V Buck-Boost Converter
3.5
4.0
5.0
6.0
7.0
7.5
240
275
365
490
610
665
VIN
3.5V TO 7.5V
2
4
5
L1A
SW
LTC1265
3
L1B
PWR VIN
LBIN
75pF
LBOUT
PGND
CT
SGND
2
3
TOP VIEW
4
1•
3300pF
SHDN
1k
6
VFB
ITHR
L1A
L1B
7
SENSE –
*SANYO OS-CON CAPACITOR
**IRC LRC2010-01-R162-J
†L1A, L2A SELECTION
+
1, 13
VIN
SENSE +
CIN*
100µF
16V
0.1µF
14
PART NO.
COILTRONICS
DALE
CTX33-4
LPT4545-330LA
L1A††
33µH
12
4
11
10
VOUT
5V
2
1
1N5818
•
L1B††
33µH
75k
+
3
SHDN
COUT*
100µF
10V
9
8
0.01µF
MANUFACTURER
33µF
10V*
•
IOUT(MAX) (mA)
+
VIN (V)
100pF
RSENSE**
0.162Ω
25k
LTC1265 F09
9V to 12V and – 12V Outputs
MBRS130LT3
2
4
3
75pF
5
L1A
L1B
3
3300pF
L1A
*AVX TPSE686K020
**AVX TPSE336K025
†
IRC LRC2010-01-R162-J
††
L1A,L2A SELECTION
MANUFACTURER
PART NO.
COILTRONICS
DALE
CTX50-4
LPT4545-500LA
14
PWR VIN
SW
LBIN
LTC1265
LBOUT
PGND
CT
SGND
2
TOP VIEW
4
1•
L1B
VIN
SHDN
1k
6
7
VFB
ITHR
SENSE –
+
1, 13
SENSE +
0.01µF
0.1µF
14
CIN*
68µF
20V
33µF**
25V
1N914
L1A††
50µH
COUT*
68µF
20V
VOUT
12V
•
40
60
80
100
115
130
150
165
180
VOUT
–12V
VIN
4V TO 12V
+
4.0
5.0
6.0
7.0
8.0
9.0
10.0
11.0
12.0
IOUT(MAX) (mA)
+
VIN (V)
2
1
MBRS130LT3
12
SI19430DY 4
11
10
•
SHDN
301k
L1B††
50µH
+
3
9
8
RSENSE*
0.162Ω
100pF
34k
LTC1265 TA05
COUT*
68µF
20V
LTC1265/LTC1265-3.3/LTC1265-5
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TYPICAL APPLICATIO S
2.5mm Max Height 5V-to-3.3V (500mA)
2
VIN
4
SW
LTC1265-3.3
51pF
5
3300pF
PWR VIN
LBIN
3
LBOUT
PGND
CT
SGND
SHDN
1k
6
N/C
ITHR
7
+
1, 13
SENSE –
SENSE +
0.1µF
14
CIN*
15µF
10V × 2
*AVX TAJB156K010
**AVX TAJB226K06
†
IRC LRC2010-01-R200-J
††
SUMIDA CLS62-180
MBRS0520LT1
12
11
10
L1††
18µH
SHDN
9
+
VIN
3.5V TO 12.5V
8
1000pF
COUT**
22µF
6.3V × 2
RSENSE†
0.20Ω
LTC1265 TA06
VOUT
3.3V
500mA
Logic Selectable 0V/3.3V/5V 700mA Regulator
*DALE 593D68X0020E2W
**DALE 593D107X0010D2W
†
IRC LRC2010-01-R15-J
††
L1 SELECTION
0V: VOUT = 5V
5V: VOUT = 3.3V
2
4
3
75pF
5
3300pF
VIN
SW
LBIN
LTC1265
LBOUT
PGND
CT
SGND
SHDN
1k
6
7
VFB
ITHR
SENSE –
+
1, 13
PWR VIN
SENSE +
1000pF
0.1µF
14
CIN*
68µF
20V
PART NO.
COILCRAFT
COILTRONICS
DALE
SUMIDA
DO3316-333
CTX33-4
LPT4545-330LA
CD74-330
†††
VSHDN = 0V: VOUT = 3.3V/5V
= 5V: VOUT = 0V
MBRS130LT3
12
11
10
VSHDN
†††
100pF
L1††
33µH
45.3k
9
+
VIN
3.5V TO 12.5V
MANUFACTURER
8
RSENSE†
0.15Ω
56.2k
75k
LTC1265 TA07
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
COUT**
100µF
10V
VOUT
0V/3.3V/5V
700mA
15
LTC1265/LTC1265-3.3/LTC1265-5
U
TYPICAL APPLICATIO S
4-NiCad Battery Charger
*DALE 593D226X0025D2W
**DALE 593D107X0016E2W
†
DALE WSL2010-0.10-1%
††
L1 SELECTION
2
VIN
4
51Ω
SW
LTC1265
5
FAST CHARGE: = 0V
TRICKLE CHARGE: > 2V
PWR VIN
LBIN
3
LBOUT
PGND
CT
SGND
SHDN
270pF
6
1k
VN2222L
VFB
ITHR
7
SENSE –
+
1, 13
SENSE +
0.1µF
CIN*
22µF, 25V
14
PART NO.
COILCRAFT
COILTRONICS
SUMIDA
DO3316-104
CTX100-4P
CD105-101
MBRS130LT3
12
11
10
CHARGER
ON/OFF
100pF
L1††
100µH
30k
9
8
1000pF
3300pF
MANUFACTURER
+
VIN
8V TO 12.5V
COUT**
100µF
10V
138k
RSENSE†
0.10Ω
MBRS130LT3
LTC1265 TA08
U
PACKAGE DESCRIPTIO
VOUT
4 NICAD
1A FAST CHARGE
0.1A TRICKLE CHARGE
Dimension in inches (millimeters) unless otherwise noted.
S Package
14-Lead Plastic Small Outline (Narrow 0.150)
(LTC DWG # 05-08-1610)
0.337 – 0.344*
(8.560 – 8.738)
0.010 – 0.020
× 45°
(0.254 – 0.508)
0.008 – 0.010
(0.203 – 0.254)
0.053 – 0.069
(1.346 – 1.752)
0.004 – 0.010
(0.101 – 0.254)
14
13
12
11
10
9
8
0° – 8° TYP
0.016 – 0.050
(0.406 – 1.270)
0.014 – 0.019
(0.355 – 0.483)
TYP
0.228 – 0.244
(5.791 – 6.197)
0.050
(1.270)
BSC
0.150 – 0.157**
(3.810 – 3.988)
S14 1298
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
1
2
3
4
5
6
7
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LTC1143
Dual Step-Down Switching Regulator Controller
Dual Version of LTC1147
LTC1147
Step-Down Switching Regulator Controller
Nonsynchronous, 8-Pin, VIN ≤ 16V
LTC1148HV
Step-Down Switching Regulator Controller
Synchronous, VIN ≤ 20V
LTC1174
Step-Down Switching Regulator with Internal 0.5A Switch
VIN ≤ 18.5V, Comparator/Low Battery Detector
LTC1474/LTC1475 Low Quiescent Current Step-Down Regulators
Monolithic, IQ = 40µA, 400mA, MS8
LTC1574
Step-Down Switching Regulator with Internal 0.5A Switch
and Schottky Diode
VIN ≤ 18.5V, Comparator
LTC1622
Low Input Voltage Step-Down DC/DC Controller
Constant Frequency, 2V to 10V VIN, MS8
LTC1627
Monolithic Synchronous Step-Down Switching Regulator
Constant Frequency, IOUT to 500mA, 2.65V to 8.5V VIN
LTC1772
Constant Frequency Step-Down DC/DC Controller
SOT-23, 2.2V to 9.8V VIN
16
Linear Technology Corporation
126535fb LT/LCG 0800 2K REV B • PRINTED IN USA
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com
 LINEAR TECHNOLOGY CORPORATION 1995