LTC1265/LTC1265-3.3/LTC1265-5 1.2A, High Efficiency Step-Down DC/DC Converter U FEATURES ■ ■ ■ ■ ■ ■ ■ ■ ■ ■ DESCRIPTIO The LTC®1265 is a monolithic step-down current mode DC/DC converter featuring Burst Mode TM operation at low output current. The LTC1265 incorporates a 0.3Ω switch (VIN =10V) allowing up to 1.2A of output current. High Efficiency: Up to 95% Current Mode Operation for Excellent Line and Load Transient Response Internal 0.3Ω Power Switch (VIN = 10V) Short-Circuit Protection Low Dropout Operation: 100% Duty Cycle Low-Battery Detector Low 160µA Standby Current at Light Loads Active-High Micropower Shutdown: IQ < 15µA Peak Inductor Current Independent of Inductor Value Available in 14-pin SO Package Under no load condition, the converter draws only 160µA. In shutdown it typically draws a mere 5µA making this converter ideal for current sensitive applications. In dropout the internal P-channel MOSFET switch is turned on continuously maximizing the life of the battery source. The LTC1265 incorporates automatic power saving Burst Mode operation to reduce gate charge losses when the load currents drop below the level required for continuous operation. U APPLICATIO S ■ ■ ■ ■ ■ ■ ■ ■ 5V to 3.3V Conversion Distributed Power Systems Step-Down Converters Inverting Converters Memory Backup Supply Portable Instruments Battery-Powered Equipment Cellular Telephones The inductor current is user-programmable via an external current sense resistor. Operation up to 700kHz permits the use of small surface mount inductors and capacitors. , LTC and LT are registered trademarks of Linear Technology Corporation. Burst Mode is a trademark of Linear Technology Corporation. U TYPICAL APPLICATIO + LTC1265-5 Efficiency CIN††† 68µF 20V 0.1µF SW SHDN 1k 3900pF 130pF L1* 33µH VIN PWR VIN D1† LTC1265-5 ITH CT 100 RSENSE** 0.1Ω PGND VOUT 5V 1A + COUT†† 220µF 10V SENSE+ 1000pF SENSE SGND 95 90 VIN = 12V 85 80 – * COILTRONICS CTX33-4 ** IRC LRC2010-01-R100-J † MBRS130LT3 †† AVX TPSE227K010 ††† AVX TPSE686K020 LTC1265-FO1 VIN = 6V VIN = 9V EFFICIENCY (%) VIN 5.4V TO 12V 75 70 0.01 L = 33µH VOUT = 5V RSENSE = 0.1Ω CT = 130pF 0.10 LOAD CURRENT (A) 1.00 LTC1265 TA01 Figure 1. High Efficiency Step-Down Converter 1 LTC1265/LTC1265-3.3/LTC1265-5 W W W AXI U U ABSOLUTE RATI GS U U W PACKAGE/ORDER I FOR ATIO (Voltages Refer to GND Pin) (Note 1) ORDER PART NUMBER TOP VIEW Input Supply Voltage (Pins 1, 2, 13) ..........– 0.3V to 13V DC Switch Current (Pin 14) .................................... 1.2A Peak Switch Current (Pin 14) ................................. 1.6A Switch Voltage (Pin 14) .................................. VIN – 13.0 Operating Temperature Range LTC1265C ............................................... 0° to 70°C LTC1265I ........................................ – 40°C to 85°C Junction Temperature (Note 2) ............................. 125°C Storage Temperature Range ....................– 65° to 150°C Lead Temperature (Soldering, 10 sec).................. 300°C PWR VIN 1 VIN 2 14 SW 13 PWR VIN LBOUT 3 12 PGND LBIN 4 11 SGND CT 5 10 SHDN ITH 6 9 N/C (VFB*) SENSE– 7 8 SENSE+ LTC1265CS LTC1265CS-5 LTC1265CS-3.3 LTC1265IS S PACKAGE 14-LEAD PLASTIC SO *ADJUSTABLE OUTPUT VERSION TJMAX = 125°C, θJA = 110°C/W Consult factory for Military grade parts. ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VSHDN = 0V, unless otherwise specified. SYMBOL PARAMETER CONDITIONS MIN TYP IFB Feedback Current into Pin 9 LTC1265 VFB Feedback Voltage LTC1265C VIN = 9V, LTC1265I VOUT Regulator Output Voltage LTC1265-3.3: ILOAD = 800mA LTC1265-5: ILOAD = 800mA ∆VOUT Output Voltage Line Regulation VIN = 6.5V to 10V, ILOAD = 800mA Output Voltage Load Regulation 0.2 1 µA ● ● 1.22 1.20 1.25 1.25 1.28 1.30 V V ● ● 3.22 4.9 3.3 5 3.40 5.2 V V –40 0 40 mV LTC1265-3.3: 10mA < ILOAD < 800mA LTC1265-5: 10mA < ILOAD < 800mA 40 60 65 100 mV mV Burst Mode Operation Output Ripple ILOAD = 0mA 50 IQ Input DC Supply Current (Note 3) Active Mode: 3.5V < VIN < 10V Sleep Mode: 3.5V < VIN < 10V Sleep Mode: 5V < VIN < 10V (LTC1265-5) Shutdown: VSHDN = VIN, 3.5V < VIN < 10V 1.8 160 160 5 2.4 230 230 15 mA µA µA µA VLBTRIP Low-Battery Trip Point 1.15 1.25 1.35 V ILBIN Current into Pin 4 0.5 µA ILBOUT Current Sunk by Pin 3 VLBOUT = 0.4V, VLBIN = 0V VLBOUT = 5V, VLBIN = 10V 0.5 1.0 1.5 1.0 mA µA V8 – V 7 Current Sense Threshold Voltage LTC1265: VSENSE– = 5V, V9 = VOUT/4 + 25mV (Forced) VSENSE– = 5V, V9 = VOUT/4 – 25mV (Forced) LTC1265-3.3: VSENSE– = VOUT + 100mV (Forced) VSENSE– = VOUT – 100mV (Forced) LTC1265-5: VSENSE– = VOUT + 100mV (Forced) VSENSE– = VOUT – 100mV (Forced) RON ON Resistance of Switch LTC1265C LTC1265I I5 CT Pin Discharge Current VOUT in Regulation, VSENSE– = VOUT VOUT = 0V tOFF Switch Off Time (Note 4) CT = 390pF, ILOAD = 800mA (LTC1265C) CT = 390pF, ILOAD = 800mA (LTC1265I) 2 UNITS mVP-P 25 150 25 150 25 150 180 mV mV mV mV mV mV 0.3 0.3 0.60 0.70 Ω Ω 40 60 2 100 10 µA µA 4 3.5 5 5 6 7 µs µs 130 130 130 ● ● ● MAX 180 180 LTC1265/LTC1265-3.3/LTC1265-5 ELECTRICAL CHARACTERISTICS The ● denotes the specifications which apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VIN = 10V, VSHDN = 0V, unless otherwise specified. SYMBOL PARAMETER CONDITIONS MIN VIH Shutdown Pin High Min Voltage at Pin 10 for Device to be in Shutdown 1.2 VIL Shutdown Pin Low Max Voltage at Pin 10 for Device to be Active 0.6 V I10 Shutdown Pin Input Current VSHDN = 8V 0.5 µA Note 1: Absolute Maximum Ratings are those values beyond which the life of the device may be impaired. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formulas: LTC1265CS, LTC1265CS-3.3, LTC1265CS-5: TJ = TA + (PD • 110°C/W) TYP MAX UNITS V Note 3: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. Note 4: In applications where RSENSE is placed at ground potential, the off time increases by approximately 40%. U W TYPICAL PERFOR A CE CHARACTERISTICS Efficiency vs Input Voltage (VOUT = 5V) Efficiency vs Load Current 100 100 100 VIN = 9V VIN = 12V 80 LTC1265-3.3 VOUT = 3.3V RSENSE = 0.1Ω CT = 130pF COIL = CTX33-4 75 70 0.01 ILOAD = 250mA 94 92 EFFICIENCY (%) EFFICIENCY (%) ILOAD = 800mA 90 88 LTC1265-5 RSENSE = 0.1Ω CT = 130pF COIL = CTX33-4 84 82 1.00 4 5 6 80 7 9 10 11 8 INPUT VOLTAGE (V) 25°C 70°C 0.4 300 0.8 270 0.7 240 TJ = 125°C 0.5 TJ = 70°C 0.4 0.3 TJ = 25°C 0.2 TJ = 0°C 0.2 2 3 5 6 7 8 (VIN – VOUT) VOLTAGE (V) 4 9 10 1265 G04 7 9 10 11 8 INPUT VOLTAGE (V) 12 13 VIN = 12V 210 180 150 120 90 30 0 1 6 60 0.1 0 5 Switch Leakage Current 0.9 0.6 0.6 0 4 13 Switch Resistance RDS(ON) (Ω) NORMALIZED FREQUENCY 12 LTC1265 G03 1.2 0.8 ILOAD = 800mA 1265 G02 Operating Frequency vs (VIN – VOUT) 0°C 88 82 1265 G01 1.0 ILOAD = 250mA 90 84 80 0.10 LOAD CURRENT (A) 94 92 86 86 LEAKAGE CURRENT (nA) EFFICIENCY (%) 90 85 96 96 VIN = 5V LTC1265-3.3 RSENSE = 0.1Ω CT = 130pF COIL = CTX33-4 98 98 95 Efficiency vs Input Voltage (VOUT = 3.3V) 3 4 5 6 7 8 9 10 11 12 13 INPUT VOLTAGE (V) 1265 G05 0 0 20 60 40 TEMPERATURE (°C) 80 100 1265 G06 3 LTC1265/LTC1265-3.3/LTC1265-5 U W TYPICAL PERFOR A CE CHARACTERISTICS DC Supply Current DOES NOT INCLUDE GATE CHARGE 1.8 ACTIVE MODE 1.5 1.2 0.9 0.6 0.3 SLEEP MODE 10 4 8 6 INPUT VOLTAGE (V) 2 SHUTDOWN = 3V TA = 25C 5.0 6 5 4 3 2 1 0 0 Gate Charge Losses 5.5 7 SUPPLY CURRENT (µA) SUPPLY CURRENT (mA) Supply Current in Shutdown 8 SWITCHING CURRENT (mA) 2.1 12 14 0 VIN = 12V 4.5 4.0 3.5 VIN = 9V 3.0 2.5 2.0 VIN = 6V 1.5 1.0 0.5 0 3 4 5 6 7 8 9 10 11 12 13 INPUT VOLTAGE (V) 1265 G07 1265 G08 0 200 400 600 FREQUENCY (kHz) 800 1000 1265 G09 U U U PI FU CTIO S PWR VIN (Pins 1, 13): Supply for the Power MOSFET and its Driver. Must decouple this pin properly to ground. Must always tie Pins 1 and 13 together. SENSE+ (Pin 8): The (+) Pin to the Current Comparator. A built-in offset between Pins 7 and 8 in conjunction with RSENSE sets the current trip threshold. VIN (Pin 2): Main Supply for All the Control Circuitry in the LTC1265. N/C,VFB (Pin 9): For the LTC1265 adjustable version, this pin serves as the feedback pin from an external resistive divider used to set the output voltage. On the LTC1265-3.3 and LTC1265-5 versions, this pin is not used. LBOUT (Pin 3): Open-Drain Output of the Low-Battery Comparator. This pin will sink current when Pin 4 (LBIN) goes below 1.25V. During shutdown, this pin is high impedance. LBIN (Pin 4): The (–) Input of the Low-Battery Comparator. The (+) input is connected to a reference voltage of 1.25V. CT (Pin 5): External capacitor CT from Pin 5 to ground sets the switch off time. The operating frequency is dependent on the input voltage and CT. ITH (Pin 6): Feedback Amplifier Decoupling Point. The current comparator threshold is proportional to Pin 6 voltage. SENSE – (Pin 7): Connect to the (–) input of the current comparator. For LTC1265-3.3 and LTC1265-5, it also connects to an internal resistive divider which sets the output voltage. 4 SHDN (Pin 10): Pulling this pin HIGH keeps the internal switch off and puts the LTC1265 in micropower shutdown. Do not float this pin. SGND (Pin 11): Small-Signal Ground. Must be routed separately from other grounds to the (–) terminal of COUT. PGND (Pin 12): Switch Driver Ground. Connects to the (–) terminal of CIN. Anode of the Schottky diode must be connected close to this pin. SW (Pin 14): Drain of the P-Channel MOSFET Switch. Cathode of the Schottky diode must be connected close to this pin. LTC1265/LTC1265-3.3/LTC1265-5 W FU CTIO AL DIAGRA U U (Pin 9 connection shown for LTC1265-3.3 and LTC1265-5; change create LTC1265) 1, 13 PWR VIN SENSE + SENSE – 8 7 14 SW 12 PGND – 9 V VFB ADJUSTABLE VERSION + – SLEEP R 25mV TO 150mV C Q 5pF + VOS + S – VTH2 – S 13k – G 100k + VTH1 ITH 6 T 2 VIN + 3 LB0UT + OFF-TIME CONTROL VFB 11 SGND U OPERATIO REFERENCE A3 SENSE– 10 SHDN – 5 CT 4 LBIN 1265 FD (Refer to Functional Diagram) The LTC1265 uses a constant off-time architecture to switch its internal P-channel power MOSFET. The off time is set by an external timing capacitor at CT (Pin 5). The operating frequency is then determined by the off time and the difference between VIN and VOUT. the voltage across the shunt reaches the comparator’s threshold value, its output signal will change state, setting the flip flop and turning the internal P-channel MOSFET off. The timing capacitor connected to Pin 5 is now allowed to discharge at a rate determined by the off-time controller. The output voltage is set by an internal resistive divider (LTC1265-3.3 and LTC1265-5) connected to SENSE – (Pin 7) or an external divider returned to V FB (Pin 9 for LTC1265). A voltage comparator V, and a gain block G, compare the divided output voltage with a reference voltage of 1.25V. When the voltage on the timing capacitor has discharged past VTH1, comparator T trips, sets the flip flop and causes the switch to turn on. Also, the timing capacitor is recharged. The inductor current will again ramp up until the current comparator C trips. The cycle then repeats. To optimize efficiency, the LTC1265 automatically switches between continuous and Burst Mode operation. The voltage comparator is the primary control element when the device is in Burst Mode operation, while the gain block controls the output voltage in continuous mode. When the load is heavy, the LTC1265 is in continuous operation. During the switch ON time, current comparator C monitors the voltage between Pins 7 and 8 connected across an external shunt in series with the inductor. When When the load current increases, the output voltage decreases slightly. This causes the output of the gain stage (Pin 6) to increase the current comparator threshold, thus tracking the load current. When the load is relatively light, the LTC1265 automatically goes into Burst Mode operation. The current loop is interrupted when the output voltage exceeds the desired regulated value. The hysteretic voltage comparator V trips when VOUT is above the desired output voltage, shutting off the switch and causing the capacitor to discharge. This 5 LTC1265/LTC1265-3.3/LTC1265-5 U OPERATIO (Refer to Functional Diagram) capacitor discharges past VTH1 until its voltage drops below VTH2. Comparator S then trips and a sleep signal is generated. The circuit now enters into sleep mode with the power MOSFET turned off. In sleep mode, the LTC1265 is in standby and the load current is supplied by the output capacitor. All unused circuitry is shut off, reducing quiescent current from 2mA to 160µA. When the output capacitor discharges by the amount of the hysteresis of the comparator V, the P-channel switch turns on again and the process repeats itself. During Burst Mode operation the peak inductor current is set at 25mV/RSENSE. To avoid the operation of the current loop interfering with Burst Mode operation, a built-in offset VOS is incorporated in the gain stage. This prevents the current from increasing until the output voltage has dropped below a minimum threshold. Using constant off-time architecture, the operating frequency is a function of the voltage. To minimize the frequency variation as dropout is approached, the off-time controller increases the discharge current as VIN drops below VOUT + 2V. In dropout the P-channel MOSFET is turned on continuously (100% duty cycle) providing low dropout operation with VOUT ≅ VIN. U W U U APPLICATIO S I FOR ATIO The basic LTC1265 application circuit is shown in Figure 1. External component selection is driven by the load requirement, and begins with the selection of RSENSE. Once RSENSE is known, CT and L can be chosen. Next, the Schottky diode D1 is selected followed by CIN and COUT. RSENSE Selection for Output Current RSENSE is chosen based on the required output current. With the current comparator monitoring the voltage developed across RSENSE, the threshold of the comparator determines the peak inductor current. Depending on the load current condition, the threshold of the comparator lies between 25mV/RSENSE and 150mV/RSENSE. The maximum output current of the LTC1265 is: 25mV (Amps) IOUT(MAX) = 150mV – RSENSE 2 • RSENSE = 137.5mV (Amps) RSENSE Solving for RSENSE and allowing a margin of variations in the LTC1265 and extended component values yields: RSENSE = 100mV (Ω) IOUT(MAX) The LTC1265 is rated with a capability to supply a maximum of 1.2A of output current. Therefore, the minimum value of RSENSE that can be used is 0.083Ω. A graph for selecting RSENSE versus maximum output is given in Figure 2. IOUT(MAX) = 150mV – IRIPPLE (Amps) RSENSE 2 0.5 where IRIPPLE is the peak-to-peak inductor ripple current. To account for light and heavy load conditions, the IOUT(MAX) is then given by: RSENSE (Ω) At a relatively light load, the LTC1265 is in Burst Mode operation. In this mode the peak inductor current is set at 25mV/RSENSE. To fully benefit from Burst Mode operation, the inductor current should be continuous during burst periods. Hence, the peak-to-peak inductor ripple current must not exceed 25mV/RSENSE. 0.4 0.3 0.2 0.1 0 0 0.2 0.6 0.8 0.4 MAXIMUM OUTPUT CURRENT (A) 1 1265 G10 Figure 2. Selecting RSENSE 6 LTC1265/LTC1265-3.3/LTC1265-5 U W U U APPLICATIO S I FOR ATIO Under short-circuit condition, the peak inductor current is determined by: ISC(PK) = 150mV (Amps) RSENSE 2V, the LTC1265 reduces tOFF by increasing the discharge current in CT. This prevents audible operation prior to dropout. (See shelving effect shown in the Operating Frequency curve under Typical Performance Characteristics.) In this condition, the LTC1265 automatically extends the off time of the P-channel MOSFET to allow the inductor current to decay far enough to prevent any current buildup. The resulting ripple current causes the average shortcircuit current to be approximately IOUT(MAX). To maintain continuous inductor current at light load, the inductor must be chosen to provide no more than 25mV/ RSENSE of peak-to-peak ripple current. This results in the following expression for L: CT and L Selection for Operating Frequency Using an inductance smaller than the above value will result in the inductor current being discontinuous. A consequence of this is that the LTC1265 will delay entering Burst Mode operation and efficiency will be degraded at low currents. The LTC1265 uses a constant off-time architecture with tOFF determined by an external capacitor CT. Each time the P-channel MOSFET turns on, the voltage on CT is reset to approximately 3.3V. During the off time, CT is discharged by a current that is proportional to VOUT. The voltage on CT is analogous to the current in inductor L, which likewise, decays at a rate proportional to VOUT. Thus the inductor value must track the timing capacitor value. The value of CT is calculated from the desired continuous mode operating frequency: CT = 1 1.3(104)f ) VIN – VOUT VIN + VD ) (Farads) where VD is the drop across the Schottky diode. As the operating frequency is increased, the gate charge losses will reduce efficiency. The complete expression for operating frequency is given by: ) ) tOFF = 1.3(104)CT ) ) f≈ VIN – VOUT (Hz) tOFF VIN + VD 1 where: VREG (sec) VOUT VREG is the desired output voltage (i.e. 5V, 3.3V). VOUT is the measured output voltage. Thus VREG/VOUT = 1 in regulation. Note that as VIN decreases, the frequency decreases. When the input-to-output voltage differential drops below L ≥ 5.2(105)RSENSE (CT)VREG Inductor Core Selection With the value of L selected, the type of inductor must be chosen. Basically, there are two kinds of losses in an inductor; core and copper losses. Core losses are dependent on the peak-to-peak ripple current and core material. However it is independent of the physical size of the core. By increasing the inductance, the peak-to-peak inductor ripple current will decrease, therefore reducing core loss. Utilizing low core loss material, such as molypermalloy or Kool Mµ® will allow user to concentrate on reducing copper loss and preventing saturation. Although higher inductance reduces core loss, it increases copper loss as it requires more windings. When space is not at a premium, larger wire can be used to reduce the wire resistance. This also prevents excessive heat dissipation. CATCH DIODE SELECTION Losses in the catch diode depend on forward drop and switching times. Therefore Schottky diodes are a good choice for low drop and fast switching times. The catch diode carries load current during the off time. The average diode current is therefore dependent on the Kool Mµ is a registered trademark of Magnetics, Inc. 7 LTC1265/LTC1265-3.3/LTC1265-5 U W U U APPLICATIO S I FOR ATIO P-channel switch duty cycle. At high input voltages, the diode conducts most of the time. As VIN approaches VOUT, the diode conducts only a small fraction of the time. The most stressful condition for the diode is when the output is short circuited. Under this condition, the diode must safely handle ISC(PK) at close to 100% duty cycle. Most LTC1265 circuits will be well served by either a 1N5818 or a MBRS130LT3 Schottky diode. An MBRS0520 is a good choice for IOUT(MAX) ≤ 500mA. CIN In continuous mode, the input current of the converter is a square wave of duty cycle VOUT/ VIN. To prevent large voltage transients, a low ESR input capacitor must be used. In addition, the capacitor must handle a high RMS current. The CIN RMS current is given by: 1 IOUT [VOUT (VIN – VOUT)] /2 (ARMS) IRMS ≈ VIN This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT/2. This simple worst case is commonly used for design because even significant deviations do not offer much relief. Note that capacitor manufacturer’s ripple current ratings are often based on only 2000 hours lifetime. This makes it advisable to further derate the capacitor, or to choose a capacitor rated at a higher temperature than required. Do not underspecify this component. An additional 0.1µF ceramic capacitor is also required on PWR VIN for high frequency decoupling. COUT The selection of COUT is based upon the effective series resistance (ESR) for proper operation of the LTC1265. The required ESR of COUT is: ESRCOUT < 50mV/IRIPPLE where IRIPPLE is the ripple current of the inductor. For the case where the IRIPPLE is 25mV/RSENSE, the required ESR of COUT is: ESRCOUT < 2(RSENSE) To avoid overheating, the output capacitor must be sized to handle the ripple current generated by the inductor. The 8 worst-case RMS ripple current in the output capacitor is given by: IRMS ≈ 150mV (ARMS) 2(RSENSE) Generally, once the ESR requirement for COUT has been met, the RMS current rating far exceeds the IRIPPLE(P-P) requirement. ESR is a direct function of the volume of the capacitor. Manufacturers such as Nichicon, AVX and Sprague should be considered for high performance capacitors. The OS-CON semiconductor dielectric capacitor available from Sanyo has the lowest ESR for its size at a somewhat higher price. In surface mount applications, multiple capacitors may have to be paralleled to meet the capacitance, ESR or RMS current handling requirement of the application. Aluminum electrolyte and dry tantalum capacitors are both available in surface mount configurations. In the case of tantalum, it is critical that the capacitors are both available in surface mount configuration and are surge tested for use in switching power supplies. An excellent choice is the AVX TPS series of surface mount tantalums, available in case heights ranging from 2mm to 4mm. Consult the manufacturer for other specific recommendations. When the capacitance of COUT is made too small, the output ripple at low frequencies will be large enough to trip the voltage comparator. This causes Burst Mode operation to be activated when the LTC1265 would normally be in continuous operation. The effect will be most pronounced with low value of RSENSE and can be improved at higher frequencies with lower values of L. Low-Battery Detection The low-battery comparator senses the input voltage through an external resistive divider. This divided voltage connects to the (–) input of a voltage comparator (Pin 4) which is compared with a 1.25V reference voltage. Neglecting Pin 4 bias current, the following expression is used for setting the trip limit: ) R4 VLB_TRIP = 1.25 1 + R3 ) LTC1265/LTC1265-3.3/LTC1265-5 U W U U APPLICATIO S I FOR ATIO The output, Pin 3, is an N-channel open drain that goes low when the battery voltage is below the threshold set by R3 and R4. In shutdown, the comparator is disabled and Pin 3 is in a high impedance state. VIN R4 LTC1265 4 3 – + R3 1.25V REFERENCE LTC1265 F03 Figure 3. Low-Battery Comparator LTC1265 ADJUSTABLE APPLICATIONS The LTC1265 develops a 1.25V reference voltage between the feedback (Pin 9) terminal and signal ground (see Figure 4). By selecting resistor R1, a constant current is caused to flow through R1 and R2 to set overall output voltage. The regulated output voltage is determined by: ) R2 VOUT = 1.25 1 + R1 ) For most applications a 30k resistor is suggested for R1. To prevent stray pickup, a 100pF capacitor is suggested across R1 located close to the LTC1265. Absolute Maximum Ratings and Latchup Prevention The absolute maximum ratings specify that SW (Pin 14) can never exceed VIN (Pins 1, 2, 13) by more than 0.3V. Normally this situation should never occur. It could, however, if the output is held up while the VIN supply is pulled down. A condition where this could potentially occur is when a battery is supplying power to an LTC1265 regulator and also to one or more loads in parallel with the the regulator’s VIN. If the battery is disconnected while the LTC1265 regulator is supplying a light load and one of the parallel circuits has a heavy load, the input capacitor of the LTC1265 regulator could be pulled down faster than the output capacitor, causing the absolute maximum ratings to be exceeded. The result is often a latchup which can be destructive if VIN is reapplied quickly. Battery disconnect is possible as a result of mechanical stress, bad battery contacts or use of a lithium-ion battery with a built-in internal disconnect. The user needs to assess his/her application to determine whether this situation could occur. If so, additional protection is necessary. Prevention against latchup can be accomplished by simply connecting a Schottky diode across the SW and VIN pins as shown in Figure 5. The diode will normally be reverse biased unless VIN is pulled below VOUT at which time the diode will clamp the (VOUT – VIN) potential to less than the 0.6V required for latchup. Note that a low leakage Schottky should be used to minimize the effect on noload supply current. Schottky diodes such as MBR0530, BAS85 and BAT84 work well. Another more serious effect of the protection diode leakage is that at no load with nothing to provide a sink for this leakage current, the VOUT LATCHUP PROTECTION SCHOTTKY R2 LTC1265 VFB SGND 9 100pF R1 11 PWR VIN VOUT SW + LTC1265 LTC1265 F04 1265 F05 Figure 4. LTC1265 Adjustable Configuration Figure 5. Preventing Absolute Maximum Ratings from Being Exceeded 9 LTC1265/LTC1265-3.3/LTC1265-5 U W U U APPLICATIONS INFORMATION output voltage can potentially float above the maximum allowable tolerance. To prevent this from occuring, a resistor must be connected between VOUT and ground with a value low enough to sink the maximum possible leakage current. THERMAL CONSIDERATIONS In a majority of applications, the LTC1265 does not dissipate much heat due to its high efficiency. However, in applications where the switching regulator is running at high duty cycles or the part is in dropout with the switch turned on continuously (DC), the user will need to do some thermal analysis. The goal of the thermal analysis is to determine whether the power dissipated by the regulator exceeds the maximum junction temperature of the part. The temperature rise is given by: TR = P(θJA) where P is the power dissipated by the regulator and θJA is the thermal resistance from the junction of the die to the ambient temperature. The junction temperature is simply given by: T J = TR + T A As an example, consider the LTC1265 is in dropout at an input voltage of 4V with a load current of 0.5A. From the Typical Performance Characteristics graph of Switch Resistance, the ON resistance of the P-channel is 0.55Ω. Therefore power dissipated by the part is: P = I2(RDSON) = 0.1375W For the SO package, the θJA is 110°C/W. Therefore the junction temperature of the regulator when it is operating in ambient temperature of 25°C is: TJ = 0.1375(110) + 25 = 40.1°C Remembering that the above junction temperature is obtained from a RDSON at 25°C, we need to recalculate the junction temperature based on a higher RDSON since it increases with temperature. However, we can safely assume that the actual junction temperature will not exceed the absolute maximum junction temperature of 125°C. 10 Now consider the case of a 1A regulator with VIN = 4V and TA = 65°C. Starting with the same 0.55Ω assumption for RDSON, the TJ calculation will yield 125°C. But from the graph, this will increase the RDSON to 0.76Ω, which when used in the above calculation yields an actual TJ > 148°C. Therefore the LTC1265 would be unsuitable for a 4V input, 1A output regulator operating at TA = 65°C. Board Layout Checklist When laying out the printed circuit board, the following checklist should be used to ensure proper operation of the LTC1265. These items are also illustrated graphically in the layout diagram of Figure 6. Check the following in your layout: 1. Are the signal and power grounds segregated? The LTC1265 signal ground (Pin 11) must return to the (–) plate of COUT. The power ground (Pin 12) returns to the anode of the Schottky diode, and the (–) plate of CIN, whose leads should be as short as possible. 2. Does the (+) plate of the CIN connect to the power VIN (Pins 1,13) as close as possible? This capacitor provides the AC current to the internal P-channel MOSFET and its driver. 3. Is the input decoupling capacitor (0.1µF) connected closely between power VIN (Pins 1,13) and power ground (Pin 12)? This capacitor carries the high frequency peak currents. 4. Is the Schottky diode closely connected between the power ground (Pin 12) and switch (Pin 14)? 5. Does the LTC1265 SENSE – (Pin 7) connect to a point close to RSENSE and the (+) plate of COUT? In adjustable applications, the resistive divider, R1 and R2, must be connected between the (+) plate of COUT and signal ground. 6. Are the SENSE – and SENSE + leads routed together with minimum PC trace spacing? The 1000pF capacitor between Pins 7 and 8 should be as close as possible to the LTC1265. 7. Is SHDN (Pin 10) actively pulled to ground during normal operation? The SHDN pin is high impedance and must not be allowed to float. LTC1265/LTC1265-3.3/LTC1265-5 U W U U APPLICATIO S I FOR ATIO 1 PWR VIN VIN 2 VIN SW PWR VIN 14 D1 13 + LTC1265 1k 3900pF 1000pF 4 5 6 7 LBOUT PGND LBIN SGND CT SHDN ITH N/C (VFB) SENSE– CIN 12 0.1µF 10 R1 SHDN 9 SENSE+ L 11 + 3 8 COUT RSENSE R2 VOUT 1000pF OUTPUT DIVIDER REQUIRED WITH ADJUSTABLE VERSION ONLY LTC1265 F06 BOLD LINES INDICATE HIGH PATH CURRENTS Figure 6. LTC1265 Layout Diagram (See Board Layout Checklist) In continuous mode the voltage on the CT pin is a sawtooth with approximately 0.9VP-P swing. This voltage should never dip below 2V as shown in Figure 7a. When the load currents are low (ILOAD < IBURST) Burst Mode operation occurs. The voltage on CT pin now falls to ground for periods of time as shown in Figure 7b. During this time the LTC1265 is in sleep mode with quiescent current reduced to 160µA. The inductor current should also be monitored. If the circuit is poorly decoupled, the peak inductor current will be haphazard as in Figure 8a. A well decoupled LTC1265 has a clean inductor current as in Figure 8b. 3.3V 2.4V 0V TIME (a) CONTINUOUS MODE OPERATION LTC1265 F07a SLEEP MODE VOLTAGE AT CT (PIN 5) Since efficiency is critical to LTC1265 applications, it is very important to verify that the circuit is functioning correctly in both continuous and Burst Mode operation. As the LTC1265 is highly tolerant of poor layout, the output voltage will still be regulated. Therefore, monitoring the output voltage will not tell you whether you have a good or bad layout. The waveform to monitor is the voltage on the timing capacitor Pin 5. VOLTAGE AT CT (PIN 5) Troubleshooting Hints 3.3V 2.4V 0V TIME (b) Burst Mode OPERATION LTC1265 F07b Figure 7. CT Waveforms 11 LTC1265/LTC1265-3.3/LTC1265-5 U W U U APPLICATIO S I FOR ATIO (a) POORLY DECOUPLED LTC1265 (b) WELL DECOUPLED LTC1265 Figure 8. Inductor Waveforms Design Example VIN 5V As a design example, assume VIN = 5V, VOUT = 3.3V, IMAX = 0.8A and f = 250kHz. With this information we can easily calculate all the important components. + CIN VIN PWR VIN 0.1µF From (1), LTC1265-3.3 1k ITH RSENSE = 100mV/0.8 = 0.125Ω From (2) and assuming VD = 0.4V, 3900pF 100pF 22µH D1 + COUT PGND SENSE + CT CT ≅ 100pF 1000pF SENSE – Using (3), the value of the inductor is: LTC1265 F09 SGND L ≥ 5.2(105)(0.125)(100pF)3.3V = 22µH For the catch diode, a MBRS130LT3 or 1N5818 will be sufficient in this application. Figure 9. Design Example Circuit CIN will require an RMS current rating of at least 0.4A at temperature, and COUT will require an ESR of (from 5): 100 ESRCOUT < 0.25Ω The inductor ripple current is given by: IRIPPLE = ) ) VOUT + VD tOFF = 0.22A L At light loads the peak inductor current is at: IPEAK = 25mV/0.125 = 0.2A Therefore, at load current less than 0.1A the LTC1265 will be in Burst Mode operation. Figure 9 shows the complete circuit and Figure 10 shows the efficiency curve with the above calculated component values. 12 EFFICIENCY (% ) 95 L = DALE LPT4545-220 (22µH) VOUT = 3.3V CT = 100pF 90 85 80 75 70 0.01 VOUT 3.3V 0.8A 0.125Ω SW SHDN 0.1 LOAD CURRENT (mA) 1.0 1265 G11 Figure 10. Design Example Efficiency Curve LTC1265/LTC1265-3.3/LTC1265-5 U TYPICAL APPLICATIO S High Efficiency 5V to 3.3V Converter VIN 5V 2 4 3 270pF 5 3900pF 14 SW LBIN LTC1265-3.3 LBOUT PGND CT SGND 6 7 *AVX TPSD107K010 **AVX TPSE227K010 † COILCRAFT D03316-473 †† DALE WSL2010-0.1-1% SENSE – L1† 47µH RSENSE†† 0.1Ω + MBRS130LT1 12 VOUT 3.3V COUT** 1A 220µF 10V 11 10 SHDN 9 NC ITHR CIN* 100µF 10V 0.1µF PWR VIN SHDN 1k + 1, 13 VIN 8 SENSE + 1000pF LTC1265 TA02 Positive-to-Negative (–5V) Converter VIN 3.5V TO 7.5V MANUFACTURER PART NO. COILCRAFT COILTRONICS DALE SUMIDA DO3316-473 CTX50-4 LPT4545-500LA CD74-470 2 4 3 220pF 5 †† IRC LRC2010-01-R100-J D1= MBRS130LT3 VIN (V) IOUT(MAX) (mA) 3.5 4.0 5.0 6.0 7.0 7.5 360 430 540 630 720 740 2200pF 1k 6 7 1, 13 L1† 50µH VIN LBIN LTC1265-5 LBOUT PGND CT SGND ITHR SHDN SENSE – SENSE + 1000pF CIN* 22µF 25V ×2 + SHDN TP0610L PWR VIN 14 SW 0.1µF D1 12 VOUT –5V 11 10 100k + *AVX TPSD226K025 **AVX TPSD107K010 † L1 SELECTION 8 COUT** 100µF 10V RSENSE†† 0.1Ω LTC1265 TA03 13 LTC1265/LTC1265-3.3/LTC1265-5 U TYPICAL APPLICATIO S 5V Buck-Boost Converter 3.5 4.0 5.0 6.0 7.0 7.5 240 275 365 490 610 665 VIN 3.5V TO 7.5V 2 4 5 L1A SW LTC1265 3 L1B PWR VIN LBIN 75pF LBOUT PGND CT SGND 2 3 TOP VIEW 4 1• 3300pF SHDN 1k 6 VFB ITHR L1A L1B 7 SENSE – *SANYO OS-CON CAPACITOR **IRC LRC2010-01-R162-J †L1A, L2A SELECTION + 1, 13 VIN SENSE + CIN* 100µF 16V 0.1µF 14 PART NO. COILTRONICS DALE CTX33-4 LPT4545-330LA L1A†† 33µH 12 4 11 10 VOUT 5V 2 1 1N5818 • L1B†† 33µH 75k + 3 SHDN COUT* 100µF 10V 9 8 0.01µF MANUFACTURER 33µF 10V* • IOUT(MAX) (mA) + VIN (V) 100pF RSENSE** 0.162Ω 25k LTC1265 F09 9V to 12V and – 12V Outputs MBRS130LT3 2 4 3 75pF 5 L1A L1B 3 3300pF L1A *AVX TPSE686K020 **AVX TPSE336K025 † IRC LRC2010-01-R162-J †† L1A,L2A SELECTION MANUFACTURER PART NO. COILTRONICS DALE CTX50-4 LPT4545-500LA 14 PWR VIN SW LBIN LTC1265 LBOUT PGND CT SGND 2 TOP VIEW 4 1• L1B VIN SHDN 1k 6 7 VFB ITHR SENSE – + 1, 13 SENSE + 0.01µF 0.1µF 14 CIN* 68µF 20V 33µF** 25V 1N914 L1A†† 50µH COUT* 68µF 20V VOUT 12V • 40 60 80 100 115 130 150 165 180 VOUT –12V VIN 4V TO 12V + 4.0 5.0 6.0 7.0 8.0 9.0 10.0 11.0 12.0 IOUT(MAX) (mA) + VIN (V) 2 1 MBRS130LT3 12 SI19430DY 4 11 10 • SHDN 301k L1B†† 50µH + 3 9 8 RSENSE* 0.162Ω 100pF 34k LTC1265 TA05 COUT* 68µF 20V LTC1265/LTC1265-3.3/LTC1265-5 U TYPICAL APPLICATIO S 2.5mm Max Height 5V-to-3.3V (500mA) 2 VIN 4 SW LTC1265-3.3 51pF 5 3300pF PWR VIN LBIN 3 LBOUT PGND CT SGND SHDN 1k 6 N/C ITHR 7 + 1, 13 SENSE – SENSE + 0.1µF 14 CIN* 15µF 10V × 2 *AVX TAJB156K010 **AVX TAJB226K06 † IRC LRC2010-01-R200-J †† SUMIDA CLS62-180 MBRS0520LT1 12 11 10 L1†† 18µH SHDN 9 + VIN 3.5V TO 12.5V 8 1000pF COUT** 22µF 6.3V × 2 RSENSE† 0.20Ω LTC1265 TA06 VOUT 3.3V 500mA Logic Selectable 0V/3.3V/5V 700mA Regulator *DALE 593D68X0020E2W **DALE 593D107X0010D2W † IRC LRC2010-01-R15-J †† L1 SELECTION 0V: VOUT = 5V 5V: VOUT = 3.3V 2 4 3 75pF 5 3300pF VIN SW LBIN LTC1265 LBOUT PGND CT SGND SHDN 1k 6 7 VFB ITHR SENSE – + 1, 13 PWR VIN SENSE + 1000pF 0.1µF 14 CIN* 68µF 20V PART NO. COILCRAFT COILTRONICS DALE SUMIDA DO3316-333 CTX33-4 LPT4545-330LA CD74-330 ††† VSHDN = 0V: VOUT = 3.3V/5V = 5V: VOUT = 0V MBRS130LT3 12 11 10 VSHDN ††† 100pF L1†† 33µH 45.3k 9 + VIN 3.5V TO 12.5V MANUFACTURER 8 RSENSE† 0.15Ω 56.2k 75k LTC1265 TA07 Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. COUT** 100µF 10V VOUT 0V/3.3V/5V 700mA 15 LTC1265/LTC1265-3.3/LTC1265-5 U TYPICAL APPLICATIO S 4-NiCad Battery Charger *DALE 593D226X0025D2W **DALE 593D107X0016E2W † DALE WSL2010-0.10-1% †† L1 SELECTION 2 VIN 4 51Ω SW LTC1265 5 FAST CHARGE: = 0V TRICKLE CHARGE: > 2V PWR VIN LBIN 3 LBOUT PGND CT SGND SHDN 270pF 6 1k VN2222L VFB ITHR 7 SENSE – + 1, 13 SENSE + 0.1µF CIN* 22µF, 25V 14 PART NO. COILCRAFT COILTRONICS SUMIDA DO3316-104 CTX100-4P CD105-101 MBRS130LT3 12 11 10 CHARGER ON/OFF 100pF L1†† 100µH 30k 9 8 1000pF 3300pF MANUFACTURER + VIN 8V TO 12.5V COUT** 100µF 10V 138k RSENSE† 0.10Ω MBRS130LT3 LTC1265 TA08 U PACKAGE DESCRIPTIO VOUT 4 NICAD 1A FAST CHARGE 0.1A TRICKLE CHARGE Dimension in inches (millimeters) unless otherwise noted. S Package 14-Lead Plastic Small Outline (Narrow 0.150) (LTC DWG # 05-08-1610) 0.337 – 0.344* (8.560 – 8.738) 0.010 – 0.020 × 45° (0.254 – 0.508) 0.008 – 0.010 (0.203 – 0.254) 0.053 – 0.069 (1.346 – 1.752) 0.004 – 0.010 (0.101 – 0.254) 14 13 12 11 10 9 8 0° – 8° TYP 0.016 – 0.050 (0.406 – 1.270) 0.014 – 0.019 (0.355 – 0.483) TYP 0.228 – 0.244 (5.791 – 6.197) 0.050 (1.270) BSC 0.150 – 0.157** (3.810 – 3.988) S14 1298 *DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE **DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE 1 2 3 4 5 6 7 RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC1143 Dual Step-Down Switching Regulator Controller Dual Version of LTC1147 LTC1147 Step-Down Switching Regulator Controller Nonsynchronous, 8-Pin, VIN ≤ 16V LTC1148HV Step-Down Switching Regulator Controller Synchronous, VIN ≤ 20V LTC1174 Step-Down Switching Regulator with Internal 0.5A Switch VIN ≤ 18.5V, Comparator/Low Battery Detector LTC1474/LTC1475 Low Quiescent Current Step-Down Regulators Monolithic, IQ = 40µA, 400mA, MS8 LTC1574 Step-Down Switching Regulator with Internal 0.5A Switch and Schottky Diode VIN ≤ 18.5V, Comparator LTC1622 Low Input Voltage Step-Down DC/DC Controller Constant Frequency, 2V to 10V VIN, MS8 LTC1627 Monolithic Synchronous Step-Down Switching Regulator Constant Frequency, IOUT to 500mA, 2.65V to 8.5V VIN LTC1772 Constant Frequency Step-Down DC/DC Controller SOT-23, 2.2V to 9.8V VIN 16 Linear Technology Corporation 126535fb LT/LCG 0800 2K REV B • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408)432-1900 ● FAX: (408) 434-0507 ● www.linear-tech.com LINEAR TECHNOLOGY CORPORATION 1995