LTC3558 Linear USB Battery Charger with Buck and Buck-Boost Regulators FEATURES DESCRIPTION Battery Charger n Standalone USB Charger n Up to 950mA Charge Current Programmable via Single Resistor n HPWR Input Selects 20% or 100% of Programmed Charge Current n NTC Input for Temperature Qualified Charging n Internal Timer Termination n Bad Battery Detection Switching Regulators (Buck and Buck-Boost) n Up to 400mA Output Current per Regulator n 2.25MHz Constant-Frequency Operation n Power Saving Burst Mode® Operation n Low Profile, 20-Lead, 3mm × 3mm QFN Package The LTC®3558 is a USB battery charger with dual high efficiency switching regulators. The device is ideally suited to power single-cell Li-Ion/Polymer based handheld applications needing multiple supply rails. Battery charge current is programmed via the PROG pin and the HPWR pin with capability up to 950mA of current at the BAT pin. The CHRG pin allows battery status to be monitored continuously during the charging process. An internal timer controls charger termination. The part includes monolithic synchronous buck and buckboost regulators that can provide up to 400mA of output current each and operate at efficiencies greater than 90% over the entire Li-Ion/Polymer battery range. The buckboost regulator can regulate its programmed output voltage at its rated deliverable current over the entire Li-Ion range without drop out, increasing battery runtime. APPLICATIONS n n The LTC3558 is offered in a low profile (0.75mm), thermally enhanced, 20-lead (3mm × 3mm) QFN package. SD/Flash-Based MP3 Players Low Power Handheld Applications , LT, LTC, LTM and Burst Mode are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. TYPICAL APPLICATION USB Charger Plus Buck Regulator and Buck-Boost Regulator USB (4.3V TO 5.5V) BAT VCC 1μF + PVIN1 PVIN2 1.74k PROG Demo Board SINGLE Li-lon CELL (2.7V TO 4.2V) 10μF 1.2V AT 400mA NTC SW1 LTC3558 CHRG 4.7μH FB1 SWAB2 SUSP 324k 10pF 10μF 649k 2.2μH DIGITAL CONTROL HPWR 3.3V AT 400mA SWCD2 VOUT2 MODE EN1 324k 121k 22μF 33pF EN2 FB2 105k GND EXPOSED PAD 15k 330pF 10pF VC2 3558 TA01 3558f 1 LTC3558 ABSOLUTE MAXIMUM RATINGS PIN CONFIGURATION (Note 1) HPWR NTC PROG CHRG VCC TOP VIEW 20 19 18 17 16 15 EN2 GND 1 14 VC2 13 FB2 BAT 2 21 MODE 3 12 SUSP FB1 4 11 VOUT2 7 8 9 10 PVIN2 SWAB2 SWCD2 6 SW1 EN1 5 PVIN1 VCC (Transient); t < 1ms and Duty Cycle < 1%....................... –0.3V to 7V VCC (Static) .................................................. –0.3V to 6V BAT, CHRG ................................................... –0.3V to 6V PROG, SUSP .................................–0.3V to (VCC + 0.3V) HPWR, NTC................... –0.3V to Max (VCC, BAT) + 0.3V PROG Pin Current ...............................................1.25mA BAT Pin Current ..........................................................1A PVIN1, PVIN2 ..................................–0.3V to (BAT + 0.3V) EN1, EN2, MODE, VOUT2 .............................. –0.3V to 6V FB1, SW1 ......................... –0.3V to (PVIN1 + 0.3V) or 6V FB2, VC2, SWAB2 ............. –0.3V to (PVIN2 + 0.3V) or 6V SWCD2 ............................–0.3V to (VOUT2 + 0.3V) or 6V ISW1 ...............................................................600mA DC ISWAB2, ISWCD2, IVOUT2 ...................................750mA DC Junction Temperature (Note 2) ............................. 125°C Operating Temperature Range (Note 3).... –40°C to 85°C Storage Temperature.............................. –65°C to 125°C UD PACKAGE 20-LEAD (3mm × 3mm) PLASTIC QFN TJMAX = 125°C, θJA = 68°C/W EXPOSED PAD (PIN 21) IS GND, MUST BE SOLDERED TO PCB ORDER INFORMATION LEAD FREE FINISH TAPE AND REEL PART MARKING PACKAGE DESCRIPTION TEMPERATURE RANGE LTC3558EUD#PBF LTC3558EUD#TRPBF LDCD 20-Lead (3mm × 3mm) Plastic QFN –40°C to 85°C Consult LTC Marketing for parts specified with wider operating temperature ranges. Consult LTC Marketing for information on non-standard lead based finish parts. For more information on lead free part marking, go to: http://www.linear.com/leadfree/ For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/ 3558f 2 LTC3558 ELECTRICAL CHARACTERISTICS The l denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 5V, BAT = PVIN1 = PVIN2 = 3.6V, RPROG = 1.74k, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Battery Charger VCC Input Supply Voltage IVCC Battery Charger Quiescent Current (Note 4) BAT Regulated Output Voltage VFLOAT l 4.3 Standby Mode, Charge Terminated Suspend Mode, VSUSP = 5V 4.179 4.165 440 84 5.5 V 285 8.5 4.200 4.200 460 92 –3.5 –2.5 –1.5 400 17 4.221 4.235 500 100 –7 –4 –3 μA μA V V mA mA μA μA μA –50 4 –100 4.125 μA V IBAT Constant-Current Mode Charge Current Battery Drain Current VUVLO Undervoltage Lockout Threshold 0°C ≤ TA ≤ 85°C l HPWR = 1 HPWR = 0 Standby Mode, Charger Terminated, EN1 = EN2 = 0 Shutdown, VCC < VUVLO, BAT = 4.2V, EN1 = EN2 = 0 Suspend Mode, SUSP = 5V, BAT = 4.2V, EN1 = EN2 = 0 VCC = 0V, EN1 = EN2 = 1, MODE = 1, FB1 = FB2 = 0.85V, VOUT2 = 3.6V BAT = 3.5V, VCC Rising ΔVUVLO Undervoltage Lockout Hysteresis BAT = 3.5V VDUVLO BAT = 4.05V, (VCC – BAT) Falling VPROG Differential Undervoltage Lockout Threshold Differential Undervoltage Lockout Hysteresis PROG Pin Servo Voltage hPROG Ratio of IBAT to PROG Pin Current ITRKL Trickle Charge Current BAT < VTRKL 36 VTRKL Trickle Charge Threshold Voltage BAT Rising 2.8 ΔVTRKL Trickle Charge Hysteresis Voltage ΔVRECHRG Recharge Battery Threshold Voltage Threshold Voltage Relative to VFLOAT tRECHRG Recharge Comparator Filter Time BAT Falling tTERM Safety Timer Termination Period BAT = VFLOAT tBADBAT Bad Battery Termination Time BAT < VTRKL hC/10 End-of-Charge Indication Current Ratio (Note 5) tC/10 End-of-Charge Comparator Filter Time IBAT Falling 2.2 ms RON(CHG) Battery Charger Power FET OnResistance (Between VCC and BAT) Junction Temperature in Constant Temperature Mode IBAT = 190mA 500 mΩ 105 °C VDIS Cold Temperature Fault Threshold Voltage Hot Temperature Fault Threshold Voltage NTC Disable Threshold Voltage INTC NTC Leakage Current Rising NTC Voltage Hysteresis Falling NTC Voltage Hysteresis Falling NTC Voltage Hysteresis VNTC = VCC = 5V ICHG ΔVDUVLO TLIM 3.85 200 30 50 BAT = 4.05V 130 HPWR = 1 HPWR = 0 BAT < VTRKL 1.000 0.200 0.100 800 mV 70 mV V V V mA/mA 46 56 2.9 3 100 –75 –95 mV mA V mV –115 1.7 mV ms 3.5 4 4.5 Hour 0.4 0.5 0.6 Hour 0.085 0.1 0.11 mA/mA NTC VCOLD VHOT 75 33.4 l 0.7 –1 76.5 1.6 34.9 1.6 1.7 50 78 36.4 2.7 1 %VCC %VCC %VCC %VCC %VCC mV μA 3558f 3 LTC3558 ELECTRICAL CHARACTERISTICS The l denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 5V, BAT = PVIN1 = PVIN2 = 3.6V, RPROG = 1.74k, unless otherwise noted. SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNITS Logic (HPWR, SUSP, CHRG, EN1, EN2, MODE) VIL Input Low Voltage HPWR, SUSP, MODE, EN1, EN2 Pins VIH Input High Voltage HPWR, SUSP, MODE, EN1, EN2 Pins RDN Logic Pin Pull-Down Resistance HPWR, SUSP Pins VCHRG CHRG Pin Output Low Voltage ICHRG = 5mA ICHRG CHRG Pin Input Current BAT = 4.5V, VCHRG = 5V 0.4 1.2 l 1.9 V V 4 6.3 MΩ 100 250 mV 0 1 μA 4.2 V 400 50 2 8 Buck Switching Regulator l PVIN1 Input Supply Voltage IPVIN1 fOSC Pulse Skip Input Current Burst Mode Current Shutdown Current Supply Current in UVLO PVIN1 Falling PVIN1 Rising Switching Frequency ILIMSW1 Peak PMOS Current Limit VFB1 Feedback Voltage MODE = 0 IFB1 FB Input Current FB1 = 0.85V DMAX1 Maximum Duty Cycle FB1 = 0V RPMOS1 RDS(ON) of PMOS ISW1 = 100mA 0.65 Ω RNMOS1 RDS(ON) of NMOS ISW1 = –100mA 0.75 Ω RSW1(PD) SW Pull-Down in Shutdown 13 kΩ PVIN1 UVLO FB1 = 0.85V, MODE = 0 (Note 6) FB1 = 0.85V, MODE = 1 (Note 6) EN1 = 0 PVIN1 = PVIN2 = 2V 2.7 1.91 220 35 0 4 2.45 2.55 2.25 2.70 2.59 μA μA μA μA V V MHz 550 800 1050 mA 780 800 820 mV 50 nA l l l MODE = 0 l 2.30 –50 l 100 % Buck-Boost Switching Regulator l PVIN2 Input Supply Voltage IPVIN2 VOUT2(LOW) PWM Input Current Burst Mode Input Current Shutdown Current Supply Current in UVLO PVIN2 Falling PVIN2 Rising Minimum Regulated Buck-Boost VOUT VOUT2(HIGH) Maximum Regulated Buck-Boost VOUT ILIMF2 Forward Current Limit (Switch A) MODE = 0 IPEAK2(BURST) Forward Current Limit (Switch A) ILIMR2 Reverse Current Limit (Switch D) IZERO2(BURST) Reverse Current Limit (Switch D) PVIN2 UVLO 2.7 MODE = 0, IOUT = 0A, FB2 = 0.85V (Note 6) MODE = 1, IOUT = 0A, FB2 = 0.85V (Note 6) EN2 = 0, IOUT = 0A PVIN1 = PVIN2 = 2V l l 2.30 V 400 30 1 8 μA μA μA μA V V V 2.70 2.75 5.45 5.60 l 580 700 820 mA MODE = 1 l 180 250 320 mA MODE = 0 l 325 450 575 mA MODE = 1 l –35 0 35 mA 2.7V < PVIN2 < 4.2V 2.75V < VOUT2 < 5.5V VFB2 Maximum Deliverable Output Current in Burst Mode Operation Feedback Servo Voltage IFB2 FB2 Input Current FB2 = 0.85V –50 fOSC Switching Frequency MODE = 0 1.91 IMAX2(BURST) 220 20 0 4 2.45 2.55 2.65 4.2 V 50 l 780 mA 800 2.25 820 mV 50 nA 2.59 MHz 3558f 4 LTC3558 ELECTRICAL CHARACTERISTICS The l denotes specifications that apply over the full operating temperature range, otherwise specifications are at TA = 25°C. VCC = 5V, BAT = PVIN1 = PVIN2 = 3.6V, RPROG = 1.74k, unless otherwise noted. SYMBOL PARAMETER CONDITIONS RDSP(ON) PMOS RDS(ON) VOUT = 3.6V RDSN(ON) NMOS RDS(ON) ILEAK(P) PMOS Switch Leakage Switches A, D –1 1 μA ILEAK(N) NMOS Switch Leakage Switches B, C –1 1 μA DCBUCK(MAX) Maximum Buck Duty Cycle MODE = 0 DCBOOST(MAX) Maximum Boost Duty Cycle MODE = 0 MIN l TYP MAX UNITS 0.6 Ω 0.6 Ω 100 % 75 % tSS2 Soft-Start Time 0.5 ms ROUT(PD) VOUT Pull-Down in Shutdown 10 kΩ Note 1: Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. Exposure to any Absolute Maximum Rating condition for extended periods may affect device reliability and lifetime. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD according to the following formula: TJ = TA + (PD • θJA) Note 3: The LTC3558E is guaranteed to meet specifications from 0°C to 85°C. Specifications over the –40°C to 85°C operating temperature range are assured by design, characterization and correlation with statistical process controls. Note 4: VCC supply current does not include current through the PROG pin or any current delivered to the BAT pin. Total input current is equal to this specification plus 1.00125 • IBAT where IBAT is the charge current. Note 5: IC/10 is expressed as a fraction of measured full charge current with indicated PROG resistor. Note 6: Dynamic supply current is higher due to the gate charge being delivered at the switching frequency. 3558f 5 LTC3558 TYPICAL PERFORMANCE CHARACTERISTICS Suspend State Supply and BAT Currents vs Temperature 4.24 4.205 VCC = 5V 4.200 4.23 IVCC 8 Battery Regulation (Float) Voltage vs Battery Charge Current, Constant-Voltage Charging Battery Regulation (Float) Voltage vs Temperature 10 9 TA = 25°C, unless otherwise noted. 4.195 4.22 4.190 5 4 VFLOAT (V) 6 VCC = 5V BAT = 4.2V SUSP = 5V EN1 = EN2 = 0V 4.185 4.21 VBAT (V) CURRENT (μA) 7 4.180 4.20 4.175 4.19 4.170 4.18 4.165 3 2 IBAT 1 4.17 0 –55 4.16 –55 –35 –35 25 5 –15 45 TEMPERATURE (°C) 65 85 4.155 45 25 5 TEMPERATURE (°C) 65 –15 3558 G01 4.150 500 VCC = 5V 495 HPWR = 5V 490 RPROG = 1.74k 485 EN1 = EN2 = 0V Battery Charge Current vs Ambient Temperature in Thermal Regulation 500 HPWR = 5V VCC = 5V RPROG = 1.74k 450 450 480 350 475 300 300 460 250 200 450 100 445 50 0 4.3 4.4 4.5 4.6 4.7 4.8 4.9 5.0 5.1 5.2 5.3 5.4 5.5 VCC (V) HPWR = 0V 50 2 2.5 3.5 3 VBAT (V) 4 3.0 1.2 1.0 0.8 3.7 3.6 65 85 3558 G07 VPROG (V) 3.8 25 5 45 –15 TEMPERATURE (°C) VCC = 5V HPWR = 5V RPROG = 1.74k EN1 = EN2 = 0V EN1 = EN2 = 0V 2.0 IBAT (μA) VCC (V) PROG Voltage vs Battery Charge Current BAT = 4.2V FALLING 5 25 45 65 85 105 125 TEMPERATURE (°C) 3558 G06 2.5 RISING 4.0 3.5 –55 –35 0 –55 –35 –15 4.5 Battery Drain Current in Undervoltage Lockout vs Temperature BAT = 3.5V 3.9 VCC = 5V HPWR = 5V RPROG = 1.74k EN1 = EN2 = 0 3558 G05 Battery Charger Undervoltage Lockout Threshold vs Temperature 4.1 200 100 3558 G04 4.2 250 150 150 455 440 IBAT (mA) 400 350 IBAT (mA) 400 465 0 100 200 300 400 500 600 700 800 900 1000 IBAT (mA) 3558 G03 Battery Charge Current vs Battery Voltage 500 IBAT (mA) 85 3558 G02 Battery Charge Current vs Supply Voltage 470 VCC = 5V HPWR = 5V RPROG = 845Ω EN1 = EN2 = 0V 4.160 BAT = 3.6V 1.5 0.6 1.0 0.4 0.5 0.2 0 –55 –35 0 25 5 45 –15 TEMPERATURE (°C) 65 85 3558 G08 0 50 100 150 200 250 300 350 400 450 500 IBAT (mA) 3558 G09 3558f 6 LTC3558 TYPICAL PERFORMANCE CHARACTERISTICS Recharge Threshold vs Temperature 115 111 Battery Charger FET On-Resistance vs Temperature SUSP/HPWR Pin Rising Thresholds vs Temperature 1.2 700 VCC = 5V VCC = 4V IBAT = 200mA EN1 = EN2 = 0V 650 107 1.0 95 91 THRESHOLD (V) 99 VCC = 5V 1.1 600 103 RDS(ON) (mΩ) VRECHARGE (mV) TA = 25°C, unless otherwise noted. 550 500 450 0.9 0.8 0.7 87 83 79 75 –55 –35 25 5 –15 45 TEMPERATURE (°C) 65 400 0.6 350 0.5 300 –55 85 –35 –15 5 25 45 65 PERCENT ERROR (%) 1.5 50 ICHRG (mA) VOLTAGE (mV) VCC = 5V BAT = 3.8V 60 100 Timer Accuracy vs Supply Voltage 2.0 70 VCC = 5V ICHRG = 5mA 60 40 30 40 20 20 10 0 –55 –35 25 5 45 –15 TEMPERATURE (°C) 65 1 2 4 3 CHRG (V) 3 2 1 0 –1 65 85 3558 G16 4.5 4.7 4.9 VCC (V) 5.1 5.3 5.5 Buck and Buck-Boost Regulator Switching Frequency vs Temperature VCC = 5V RPROG = 0.845k HPWR = 5V 2.325 FREQUENCY (MHz) BAT (V) 4 CHRG (V) PERCENT ERROR (%) 5 4.3 3558 G15 2.425 1000 800 600 400 200 0 5.0 4.5 4.0 3.5 3.0 5.0 4.0 3.0 2.0 1.0 0 IBAT (mA) 6 –15 5 25 45 TEMPERATURE (°C) 6 5 Complete Charge Cycle 2400mAh Battery VCC = 5V –35 0 3558 G14 Timer Accuracy vs Temperature –2 –55 0.5 –1.0 0 3558 G13 7 1.0 –0.5 0 85 85 3558 G12 CHRG Pin I-V Curve 80 65 3558 G11 CHRG Pin Output Low Voltage vs Temperature 120 45 25 5 TEMPERATURE (°C) –15 TEMPERATURE (°C) 3558 G10 140 0.4 –55 –35 85 VCC = 0V, MODE = 0 BAT = PVIN1 = PVIN2 BAT = 4.2V 2.225 BAT = 2.7V BAT = 3.6V 2.125 2.025 1.925 1.825 0 1 2 4 3 TIME (HOUR) 5 6 3558 G17 1.725 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 3558 G18 3558f 7 LTC3558 TYPICAL PERFORMANCE CHARACTERISTICS Buck and Buck-Boost Regulator Undervoltage Thresholds vs Temperature 2.700 Buck and Buck-Boost Regulator Enable Thresholds vs Temperature 1200 BAT = PVIN1 = PVIN2 50 BAT = PVIN1 = PVIN2 = 3.6V 1100 2.650 2.600 RISING 900 2.550 2.500 FALLING 800 2.450 RISING 700 2.400 FALLING 600 2.350 2.250 –55 –35 –15 400 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) PVIN1 = 4.2V 35 PVIN1 = 2.7V 30 20 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 5 25 45 65 85 105 125 TEMPERATURE (°C) 3558 G20 3558 G19 Buck Regulator Input Current vs Temperature, Pulse Skip Mode 400 40 25 500 2.300 FB1 = 0.85V 45 1000 VEN (V) INPUT VOLTAGE (V) Buck Regulator Input Current vs Temperature, Burst Mode Operation INPUT CURRENT (μA) 2.750 TA = 25°C, unless otherwise noted. 3558 G21 Buck Regulator PMOS RDS(0N) vs Temperature Buck Regulator NMOS RDS(0N) vs Temperature 1300 1300 1200 1200 1100 1100 1000 1000 FB1 = 0.85V PVIN1 = 4.2V 250 PVIN1 = 2.7V 200 150 100 –55 –35 –15 900 PVIN1 = 2.7V 800 PVIN1 = 4.2V 700 600 500 Buck Regulator Efficiency vs ILOAD 1.23 70 1.22 60 1.21 1.230 Burst Mode OPERATION 1.20 1.220 PULSE SKIP MODE 1.19 1.210 1.200 1.190 30 1.18 1.180 20 1.17 1.170 1.16 1.160 VOUT = 1.2V PVIN1 = 2.7V PVIN1 = 4.2V 0 0.1 1 10 ILOAD (mA) 100 1000 1.15 1 10 100 1000 ILOAD (mA) 3558 G25 ILOAD = 200mA 1.240 VOUT (V) 80 VOUT (V) EFFICIENCY (%) Buck Regulator Line Regulation 1.250 PVIN1 = 3.6V 1.24 VOUT = 1.2V 40 5 25 45 65 85 105 125 TEMPERATURE (°C) 3558 G24 Buck Regulator Load Regulation Burst Mode OPERATION 10 8 400 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 1.25 PULSE SKIP MODE PVIN1 = 4.2V 700 3558 G23 100 50 800 500 3558 G22 90 PVIN1 = 2.7V 900 600 400 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) RDS(ON) (mΩ) 300 RDS(ON) (mΩ) INPUT CURRENT (μA) 350 3558 G26 1.150 2.700 3.000 3.600 3.300 PVIN1 (V) 3.900 4.200 3558 G27 3558f LTC3558 TYPICAL PERFORMANCE CHARACTERISTICS Buck Regulator Pulse Skip Mode Operation Buck Regulator Start-Up Transient VOUT 500mV/DIV INDUCTOR CURRENT IL = 200mA/ DIV EN 2V/DIV Buck Regulator Burst Mode Operation VOUT 20mV/ DIV (AC) VOUT 20mV/ DIV (AC) SW 2V/DIV SW 2V/DIV INDUCTOR CURRENT IL = 50mA/ DIV INDUCTOR CURRENT IL = 60mA/ DIV 3558 G28 PVIN1 = 3.8V LOAD = 10mA Buck Regulator Transient Response, Pulse Skip Mode 200ns/DIV 3558 G29 Buck Regulator Transient Response, Burst Mode Operation INDUCTOR CURRENT IL = 200mA/ DIV INDUCTOR CURRENT IL = 200mA/ DIV VOUT 50mV/ DIV (AC) VOUT 50mV/ DIV (AC) LOAD STEP 5mA TO 290mA 50μs/DIV 3558 G31 2μs/DIV 3558 G30 Buck-Boost Regulator Input Current vs Temperature 30 Burst Mode OPERATION FB2 = 0.85V 25 LOAD STEP 5mA TO 290mA PVIN1 = 3.8V PVIN1 = 3.8V LOAD = 60mA INPUT CURRENT (μA) PVIN1 = 3.8V 50μs/DIV PULSE SKIP MODE LOAD = 6Ω TA = 25°C, unless otherwise noted. PVIN1 = 3.8V 50μs/DIV PVIN2 = 4.2V 20 PVIN2 = 2.7V 15 10 3558 G32 5 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 3558 G33 Buck-Boost Regulator Input Current vs Temperature 500 1100 PVIN2 = 4.2V 250 PVIN2 = 2.7V 600 1000 PVIN2 = 2.7V 900 RDS(ON) (mΩ) 350 550 500 450 400 PVIN2 = 4.2V 3558 G34 600 250 300 200 –55 –35 –15 200 –55 –35 –15 5 25 45 65 85 105 125 TEMPERATURE (°C) 3558 G35 PVIN2 = 2.7V 700 400 300 5 25 45 65 85 105 125 TEMPERATURE (°C) 800 500 350 150 100 –55 –35 –15 1200 750 650 400 200 800 700 RDS(ON) (mΩ) INPUT CURRENT (μA) PWM MODE 450 FB2 = 0.85V 300 Buck-Boost Regulator NMOS RDS(ON) vs Temperature Buck-Boost Regulator PMOS RDS(ON) vs Temperature PVIN2 = 4.2V 5 25 45 65 85 105 125 TEMPERATURE (°C) 3558 G36 3558f 9 LTC3558 TYPICAL PERFORMANCE CHARACTERISTICS Buck-Boost Efficiency vs Load Current Buck-Boost Regulator Efficiency vs Input Voltage 100 VOUT = 3.3V 100 ILOAD = 10mA 90 80 85 70 80 ILOAD = 100mA 75 ILOAD = 400mA 70 VOUT = 3.3V 90 ILOAD = 1mA EFFICIENCY (%) EFFICIENCY (%) 95 65 3.6V 4.2V 50 40 2.7V 20 Burst Mode OPERATION PWM MODE 55 50 2.700 3.000 3.600 3.300 PVIN2 (V) 3.900 PVIN2, Burst Mode OPERATION PVIN2, PWM MODE 3.6V 4.2V 10 0 0.10 4.200 2.7V 60 30 60 TA = 25°C, unless otherwise noted. 1 10 ILOAD (mA) 100 3558 G38 3558 G37 Buck-Boost Regulator Load Regulation 3.36 3.35 Buck-Boost Regulator Line Regulation 3.36 PVIN2 = 3.6V 3.35 3.34 3.34 3.33 3.33 3.32 3.32 Burst Mode OPERATION 3.31 PWM MODE 3.30 3.29 VOUT (V) VOUT (V) 1000 3.31 3.30 3.29 3.28 3.28 3.27 3.27 3.26 3.26 3.25 3.25 3.24 0.10 1 10 ILOAD (mA) 100 1000 PWM MODE ILOAD = 100mA Burst Mode OPERATION ILOAD = 10mA 3.24 2.700 3.000 3.600 3.300 PVIN2 (V) 4.200 3.900 3558 G40 3558 G39 Buck-Boost Regulator Start-Up Transient, Burst Mode Operation Buck-Boost Regulator Start-Up Transient, PWM Mode PVIN2 = 3.6V RLOAD = 16Ω PVIN2 = 3.6V RLOAD = 332Ω VOUT 1V/DIV VOUT 1V/DIV INDUCTOR CURRENT IL = 200mA/DIV INDUCTOR CURRENT IL = 200mA/DIV EN2 1V/DIV EN2 1V/DIV 100μs/DIV 3558 G41 100μs/DIV 3558 G42 3558f 10 LTC3558 PIN FUNCTIONS GND (Pin 1): Ground. Connect to Exposed Pad (Pin 21). BAT (Pin 2): Charge Current Output. Provides charge current to the battery and regulates final float voltage to 4.2V. VC2 (Pin 14): Output of the Error Amplifier and Voltage Compensation Node for the Buck-Boost Regulator. External Type I or Type III compensation (to FB2) connects to this pin. MODE (Pin 3): MODE Pin for Switching Regulators. When held high, both regulators operate in Burst Mode Operation. When held low, the buck regulator operates in pulse skip mode and the buck-boost regulator operates in PWM mode. This pin is a high impedance input; do not float. EN2 (Pin 15): Enable Input Pin for the Buck-Boost Regulator. This pin is a high impedance input; do not float. Active high. FB1 (Pin 4): Buck Regulator Feedback Voltage Pin. Receives feedback by a resistor divider connected across the output. EN1 (Pin 5): Enable Input Pin for the Buck Regulator. This pin is a high impedance input; do not float. Active high. SW1 (Pin 6): Buck Regulator Switching Node. External inductor connects to this node. PVIN1 (Pin 7): Input Supply Pin for Buck Regulator. Connect to BAT and PVIN2. A single 10μF input decoupling capacitor to GND is required. PVIN2 (Pin 8): Input Supply Pin for Buck-Boost Regulator. Connect to BAT and PVIN1. A single 10μF input decoupling capacitor to GND is required. SWAB2 (Pin 9): Switch Node for Buck-Boost Regulator Connected to the Internal Power Switches A and B. External inductor connects between this node and SWCD2. SWCD2 (Pin 10): Switch Node for Buck-Boost Regulator Connected to the Internal Power Switches C and D. External inductor connects between this node and SWAB2. VOUT2 (Pin 11): Regulated Output Voltage for Buck-Boost Regulator. SUSP (Pin 12): Suspend Battery Charging Operation. A voltage greater than 1.2V on this pin puts the battery charger in suspend mode, disables the charger and resets the termination timer. A weak pull-down current is internally applied to this pin to ensure it is low at power-up when the input is not being driven externally. FB2 (Pin 13): Buck-Boost Regulator Feedback Voltage Pin. Receives feedback by a resistor divider connected across the output. HPWR (Pin 16): High Current Battery Charging Enabled. A voltage greater than 1.2V at this pin programs the BAT pin current at 100% of the maximum programmed charge current. A voltage less than 0.4V sets the BAT pin current to 20% of the maximum programmed charge current. When used with a 1.74k PROG resistor, this pin can toggle between low power and high power modes per USB specification. A weak pull-down current is internally applied to this pin to ensure it is low at power-up when the input is not being driven externally. NTC (Pin 17): Input to the NTC Thermistor Monitoring Circuit. The NTC pin connects to a negative temperature coefficient thermistor which is typically co-packaged with the battery pack to determine if the battery is too hot or too cold to charge. If the battery temperature is out of range, charging is paused until the battery temperature re-enters the valid range. A low drift bias resistor is required from VCC to NTC and a thermistor is required from NTC to ground. To disable the NTC function, the NTC pin should be tied to ground. PROG (Pin 18): Charge Current Program and Charge Current Monitor Pin. Charge current is programmed by connecting a resistor from PROG to ground. When charging in constant-current mode, the PROG pin servos to 1V if the HPWR pin is pulled high, or 200mV if the HPWR pin is pulled low. The voltage on this pin always represents the BAT pin current through the following formula: IBAT = PROG • 800 RPROG CHRG (Pin 19): Open-Drain Charge Status Output. The CHRG pin indicates the status of the battery charger. Four possible states are represented by CHRG charging, not charging (i.e., the charge current is less than one-tenth 3558f 11 LTC3558 PIN FUNCTIONS of the full-scale charge current), unresponsive battery (i.e., the battery voltage remains below 2.9V after one half hour of charging) and battery temperature out of range. CHRG requires a pull-up resistor and/or LED to provide indication. VCC (Pin 20): Battery Charger Input. A 1μF decoupling capacitor to GND is recommended. Exposed Pad (Pin 21): Ground. The Exposed Pad must be soldered to PCB ground to provide electrical contact and rated thermal performance. BLOCK DIAGRAM 20 VCC BAT BODY MAXER VCC 1x 800x BAT – 16 12 + 19 CHRG HPWR TA CA LOGIC SUSP TDIE PROG NTCA 17 5 15 4 18 NTC NTC REF 3 2 PVIN1 MODE BATTERY CHARGER PVIN2 PVIN1 7 EN1 UNDERVOLTAGE LOCKOUT EN2 EN – FB1 OT DIE TEMPERATURE 0.8V TDIE + MODE CLK MP CONTROL LOGIC Gm SW1 6 MN BUCK REGULATOR BANDGAP OSCILLATOR 2.25MHz 13 PVIN2 VREF = 0.8V CLK VOUT2 11 BUCK-BOOST REGULATOR EN MODE CLK – FB2 0.8V 14 8 ERROR + AMP VC2 A D SWAB2 CONTROL LOGIC SWCD2 9 10 B C VC2 GND EXPOSED PAD 1 21 3558 BD 3558f 12 LTC3558 OPERATION The LTC3558 is a linear battery charger with a monolithic synchronous buck regulator and a monolithic synchronous buck-boost regulator. The buck regulator is internally compensated and needs no external compensation components. The battery charger employs a constant-current, constantvoltage charging algorithm and is capable of charging a single Li-Ion battery at charging currents up to 950mA. The user can program the maximum charging current available at the BAT pin via a single PROG resistor. The actual BAT pin current is set by the status of the HPWR pin. For proper operation, the BAT, PVIN1 and PVIN2 pins must be tied together, as shown in Figure 1. Current being delivered at the BAT pin is 500mA. Both switching regulators are enabled. The sum of the average input currents drawn by both switching regulators is 200mA. This makes the effective battery charging current only 300mA. If the HPWR pin were tied LO, the BAT pin current would be 100mA. With the switching regulator conditions unchanged, this would cause the battery to discharge at 100mA. 500mA USB (5V) 300mA BAT VCC + 200mA PVIN1 PROG SUSP HIGH HIGH HIGH LOW + PVIN2 LTC3558 RPROG 10μF SWAB2 2.2μH HPWR EN1 EN2 MODE SINGLE Li-lon CELL 3.6V SWCD2 VOUT2 SW1 VOUT1 3558 F01 Figure 1. For Proper Operation, the BAT, PVIN1 and PVIN2 Pins Must Be Tied Together APPLICATIONS INFORMATION Battery Charger Introduction Input Current vs Charge Current The LTC3558 has a linear battery charger designed to charge single-cell lithium-ion batteries. The charger uses a constant-current/constant-voltage charge algorithm with a charge current programmable up to 950mA. Additional features include automatic recharge, an internal termination timer, low-battery trickle charge conditioning, bad-battery detection, and a thermistor sensor input for out of temperature charge pausing. The battery charger regulates the total current delivered to the BAT pin; this is the charge current. To calculate the total input current (i.e., the total current drawn from the VCC pin), it is necessary to sum the battery charge current, charger quiescent current and PROG pin current. Furthermore, the battery charger is capable of operating from a USB power source. In this application, charge current can be programmed to a maximum of 100mA or 500mA per USB power specifications. Undervoltage Lockout (UVLO) The undervoltage lockout circuit monitors the input voltage (VCC) and disables the battery charger until VCC rises above VUVLO (typically 4V). 200mV of hysteresis prevents oscillations around the trip point. In addition, a differential undervoltage lockout circuit disables the battery charger 3558f 13 LTC3558 APPLICATIONS INFORMATION when VCC falls to within VDUVLO (typically 50mV) of the BAT voltage. Suspend Mode The battery charger can also be disabled by pulling the SUSP pin above 1.2V. In suspend mode, the battery drain current is reduced to 1.5μA and the input current is reduced to 8.5μA. Charge Cycle Overview When a battery charge cycle begins, the battery charger first determines if the battery is deeply discharged. If the battery voltage is below VTRKL, typically 2.9V, an automatic trickle charge feature sets the battery charge current to 10% of the full-scale value. Once the battery voltage is above 2.9V, the battery charger begins charging in constant-current mode. When the battery voltage approaches the 4.2V required to maintain a full charge, otherwise known as the float voltage, the charge current begins to decrease as the battery charger switches into constant-voltage mode. Trickle Charge and Defective Battery Detection Any time the battery voltage is below VTRKL, the charger goes into trickle charge mode and reduces the charge current to 10% of the full-scale current. If the battery voltage remains below VTRKL for more than 1/2 hour, the charger latches the bad-battery state, automatically terminates, and indicates via the CHRG pin that the battery was unresponsive. If for any reason the battery voltage rises above VTRKL, the charger will resume charging. Since the charger has latched the bad-battery state, if the battery voltage then falls below VTRKL again but without rising past VRECHRG first, the charger will immediately assume that the battery is defective. To reset the charger (i.e., when the dead battery is replaced with a new battery), simply remove the input voltage and reapply it or put the part in and out of suspend mode. Charge Termination The battery charger has a built-in safety timer that sets the total charge time for 4 hours. Once the battery voltage rises above VRECHRG (typically 4.105V) and the charger enters constant-voltage mode, the 4-hour timer is started. After the safety timer expires, charging of the battery will discontinue and no more current will be delivered. Automatic Recharge After the battery charger terminates, it will remain off, drawing only microamperes of current from the battery. If the portable product remains in this state long enough, the battery will eventually self discharge. To ensure that the battery is always topped off, a charge cycle will automatically begin when the battery voltage falls below VRECHRG (typically 4.105V). In the event that the safety timer is running when the battery voltage falls below VRECHRG, it will reset back to zero. To prevent brief excursions below VRECHRG from resetting the safety timer, the battery voltage must be below VRECHRG for more than 1.7ms. The charge cycle and safety timer will also restart if the VCC UVLO or DUVLO cycles low and then high (e.g., VCC is removed and then replaced) or the charger enters and then exits suspend mode. Programming Charge Current The PROG pin serves both as a charge current program pin, and as a charge current monitor pin. By design, the PROG pin current is 1/800th of the battery charge current. Therefore, connecting a resistor from PROG to ground programs the charge current while measuring the PROG pin voltage allows the user to calculate the charge current. Full-scale charge current is defined as 100% of the constant-current mode charge current programmed by the PROG resistor. In constant-current mode, the PROG pin servos to 1V if HPWR is high, which corresponds to charging at the full-scale charge current, or 200mV if HPWR is low, which corresponds to charging at 20% of the fullscale charge current. Thus, the full-scale charge current and desired program resistor for a given full-scale charge current are calculated using the following equations: ICHG = 800 V RPROG RPROG = 800 V ICHG 3558f 14 LTC3558 APPLICATIONS INFORMATION In any mode, the actual battery current can be determined by monitoring the PROG pin voltage and using the following equation: IBAT = PROG • 800 RPROG Thermal Regulation To prevent thermal damage to the IC or surrounding components, an internal thermal feedback loop will automatically decrease the programmed charge current if the die temperature rises to approximately 115°C. Thermal regulation protects the battery charger from excessive temperature due to high power operation or high ambient thermal conditions and allows the user to push the limits of the power handling capability with a given circuit board design without risk of damaging the LTC3558 or external components. The benefit of the LTC3558 battery charger thermal regulation loop is that charge current can be set according to actual conditions rather than worst-case conditions with the assurance that the battery charger will automatically reduce the current in worst-case conditions. Charge Status Indication The CHRG pin indicates the status of the battery charger. Four possible states are represented by CHRG charging, not charging, unresponsive battery and battery temperature out of range. The signal at the CHRG pin can be easily recognized as one of the above four states by either a human or a microprocessor. The CHRG pin, which is an open-drain output, can drive an indicator LED through a current limiting resistor for human interfacing, or simply a pull-up resistor for microprocessor interfacing. To make the CHRG pin easily recognized by both humans and microprocessors, the pin is either a low for charging, a high for not charging, or it is switched at high frequency (35kHz) to indicate the two possible faults: unresponsive battery and battery temperature out of range. When charging begins, CHRG is pulled low and remains low for the duration of a normal charge cycle. When the charge current has dropped to below 10% of the full-scale current, the CHRG pin is released (high impedance). If a fault occurs after the CHRG pin is released, the pin remains high impedance. However, if a fault occurs before the CHRG pin is released, the pin is switched at 35kHz. While switching, its duty cycle is modulated between a high and low value at a very low frequency. The low and high duty cycles are disparate enough to make an LED appear to be on or off thus giving the appearance of “blinking”. Each of the two faults has its own unique “blink” rate for human recognition as well as two unique duty cycles for microprocessor recognition. Table 1 illustrates the four possible states of the CHRG pin when the battery charger is active. Table 1. CHRG Output Pin STATUS FREQUENCY MODULATION (BLINK) FREQUENCY Charging 0Hz 0 Hz (Lo-Z) 100% IBAT < C/10 0Hz 0 Hz (Hi-Z) 0% NTC Fault 35kHz 1.5Hz at 50% 6.25%, 93.75% Bad Battery 35kHz 6.1Hz at 50% 12.5%, 87.5% DUTY CYCLE An NTC fault is represented by a 35kHz pulse train whose duty cycle alternates between 6.25% and 93.75% at a 1.5Hz rate. A human will easily recognize the 1.5Hz rate as a “slow” blinking which indicates the out of range battery temperature while a microprocessor will be able to decode either the 6.25% or 93.75% duty cycles as an NTC fault. If a battery is found to be unresponsive to charging (i.e., its voltage remains below VTRKL for over 1/2 hour), the CHRG pin gives the battery fault indication. For this fault, a human would easily recognize the frantic 6.1Hz “fast” blinking of the LED while a microprocessor would be able to decode either the 12.5% or 87.5% duty cycles as a bad battery fault. Although very improbable, it is possible that a duty cycle reading could be taken at the bright-dim transition (low duty cycle to high duty cycle). When this happens the duty cycle reading will be precisely 50%. If the duty cycle reading is 50%, system software should disqualify it and take a new duty cycle reading. 3558f 15 LTC3558 APPLICATIONS INFORMATION NTC Thermistor The battery temperature is measured by placing a negative temperature coefficient (NTC) thermistor close to the battery pack. The NTC circuitry is shown in Figure 3. To use this feature, connect the NTC thermistor, RNTC, between the NTC pin and ground, and a bias resistor, RNOM, from VCC to NTC. RNOM should be a 1% resistor with a value equal to the value of the chosen NTC thermistor at 25°C (R25). A 100k thermistor is recommended since thermistor current is not measured by the battery charger and its current will have to be considered for compliance with USB specifications. The battery charger will pause charging when the resistance of the NTC thermistor drops to 0.54 times the value of R25 or approximately 54k (for a Vishay “Curve 1” thermistor, this corresponds to approximately 40°C). If the battery charger is in constant-voltage mode, the safety timer will pause until the thermistor indicates a return to a valid temperature. As the temperature drops, the resistance of the NTC thermistor rises. The battery charger is also designed to pause charging when the value of the NTC thermistor increases to 3.25 times the value of R25. For a Vishay “Curve 1” thermistor, this resistance, 325k, corresponds to approximately 0°C. The hot and cold comparators each have approximately 3°C of hysteresis to prevent oscillation about the trip point. Grounding the NTC pin disables all NTC functionality. DUVLO, UVLO AND SUSPEND DISABLE MODE NO POWER ON IF SUSP < 0.4V AND VCC > 4V AND VCC > BAT + 130mV? CHRG HIGH IMPEDANCE YES FAULT NTC FAULT STANDBY MODE BATTERY CHARGING SUSPENDED CHRG PULSES NO CHARGE CURRENT CHRG HIGH IMPEDANCE NO FAULT BAT b 2.9V TRICKLE CHARGE MODE 1/10 FULL CHARGE CURRENT CHRG STRONG PULL-DOWN 30 MINUTE TIMER BEGINS 2.9V < BAT < 4.105V BAT > 2.9V CONSTANT CURRENT MODE FULL CHARGE CURRENT CHRG STRONG PULL-DOWN 4-HOUR TIMEOUT 30 MINUTE TIMEOUT DEFECTIVE BATTERY NO CHARGE CURRENT CHRG PULSES CONSTANT VOLTAGE MODE 4-HOUR TERMINATION TIMER BEGINS BAT DROPS BELOW 4.105V 4-HOUR TERMINATION TIMER RESETS 3558 F02 Figure 2. State Diagram of Battery Charger Operation 3558f 16 LTC3558 APPLICATIONS INFORMATION Alternate NTC Thermistors and Biasing The battery charger provides temperature qualified charging if a grounded thermistor and a bias resistor are connected to the NTC pin. By using a bias resistor whose value is equal to the room temperature resistance of the thermistor (R25) the upper and lower temperatures are pre-programmed to approximately 40°C and 0°C, respectively (assuming a Vishay “Curve 1” thermistor). The upper and lower temperature thresholds can be adjusted by either a modification of the bias resistor value or by adding a second adjustment resistor to the circuit. If only the bias resistor is adjusted, then either the upper or the lower threshold can be modified but not both. The other trip point will be determined by the characteristics of the thermistor. Using the bias resistor in addition to an adjustment resistor, both the upper and the lower temperature trip points can be independently programmed with the constraint that the difference between the upper and lower temperature thresholds cannot decrease. Examples of each technique are given below. NTC thermistors have temperature characteristics which are indicated on resistance-temperature conversion tables. The Vishay-Dale thermistor NTHS0603N011-N1003F, used in the following examples, has a nominal value of 100k and follows the Vishay “Curve 1” resistance-temperature characteristic. In the explanation below, the following notation is used. R25 = Value of the thermistor at 25°C RNTC|COLD = Value of thermistor at the cold trip point RNTC|HOT = Value of the thermistor at the hot trip point rCOLD = Ratio of RNTC|COLD to R25 rHOT = Ratio of RNTC|HOT to R25 RNOM = Primary thermistor bias resistor (see Figure 3) R1 = Optional temperature range adjustment resistor (see Figure 4) The trip points for the battery charger’s temperature qualification are internally programmed at 0.349 • VCC for the hot threshold and 0.765 • VCC for the cold threshold. Therefore, the hot trip point is set when: RNTCHOT | RNOM + RNTCHOT | • VCC = 0.349 • VCC and the cold trip point is set when: RNTC|COLD RNOM + RNTC|COLD • VCC = 0.765 • VCC Solving these equations for RNTC|COLD and RNTC|HOT results in the following: RNTC|HOT = 0.536 • RNOM and RNTC|COLD = 3.25 • RNOM By setting RNOM equal to R25, the above equations result in rHOT = 0.536 and rCOLD = 3.25. Referencing these ratios to the Vishay Resistance-Temperature Curve 1 chart gives a hot trip point of about 40°C and a cold trip point of about 0°C. The difference between the hot and cold trip points is approximately 40°C. By using a bias resistor, RNOM, different in value from R25, the hot and cold trip points can be moved in either direction. The temperature span will change somewhat due to the nonlinear behavior of the thermistor. The following equations can be used to easily calculate a new value for the bias resistor: r RNOM = HOT • R25 0.536 RNOM = rCOLD • R25 3.25 where rHOT and rCOLD are the resistance ratios at the desired hot and cold trip points. Note that these equations are linked. Therefore, only one of the two trip points can be chosen, the other is determined by the default ratios designed in the IC. Consider an example where a 60°C hot trip point is desired. From the Vishay Curve 1 R-T characteristics, rHOT is 0.2488 at 60°C. Using the above equation, RNOM should be set 3558f 17 LTC3558 APPLICATIONS INFORMATION to 46.4k. With this value of RNOM, the cold trip point is about 16°C. Notice that the span is now 44°C rather than the previous 40°C. The upper and lower temperature trip points can be independently programmed by using an additional bias resistor as shown in Figure 4. The following formulas can be used to compute the values of RNOM and R1: RNOM = rCOLD – rHOT • R25 2.714 0.765 • VCC (NTC RISING) 17 R1 = 0.536 • 105k – 0.4368 • 100k = 12.6k 20 – + NTC 3.266 – 0.4368 • 100k = 104.2k 2.714 the nearest 1% value is 105k. NTC BLOCK VCC RNOM 100k RNOM = the nearest 1% value is 12.7k. The final solution is shown in Figure 4 and results in an upper trip point of 45°C and a lower trip point of 0°C. R1 = 0.536 • RNOM – rHOT • R25 20 For example, to set the trip points to 0°C and 45°C with a Vishay Curve 1 thermistor choose: VCC 0.765 • VCC (NTC RISING) RNOM 105k TOO_COLD 17 – + NTC TOO_COLD R1 12.7k RNTC 100k – – 0.349 • VCC (NTC FALLING) + RNTC 100k TOO_HOT 0.349 • VCC (NTC FALLING) + + + NTC_ENABLE NTC_ENABLE 0.017 • VCC (NTC FALLING) TOO_HOT – 0.017 • VCC (NTC FALLING) 3558 F03 Figure 3. Typical NTC Thermistor Circuit – 3558 F04 Figure 4. NTC Thermistor Circuit with Additional Bias Resistor 3558f 18 LTC3558 APPLICATIONS INFORMATION USB and Wall Adapter Power Although the battery charger is designed to draw power from a USB port to charge Li-Ion batteries, a wall adapter can also be used. Figure 5 shows an example of how to combine wall adapter and USB power inputs. A P-channel MOSFET, MP1, is used to prevent back conduction into the USB port when a wall adapter is present and Schottky diode, D1, is used to prevent USB power loss through the 1k pull-down resistor. Typically, a wall adapter can supply significantly more current than the 500mA-limited USB port. Therefore, an N-channel MOSFET, MN1, and an extra program resistor are used to increase the maximum charge current to 950mA when the wall adapter is present. 5V WALL ADAPTER 950mA ICHG USB POWER 500mA ICHG BAT BATTERY CHARGER VCC MP1 PROG MN1 1.65k TA = 105°C – PDθ JA TA = 105°C – ( VCC – VBAT ) • IBAT • θ JA Example: Consider an LTC3558 operating from a USB port providing 500mA to a 3.5V Li-Ion battery. The ambient temperature above which the LTC3558 will begin to reduce the 500mA charge current is approximately: TA = 105°C – ( 5V – 3.5V ) • ( 500mA ) • 68°C / W TA = 105°C – 0.75W • 68°C / W = 105°C – 51°C TA = 54°C IBAT D1 current. It is not necessary to perform any worst-case power dissipation scenarios because the LTC3558 will automatically reduce the charge current to maintain the die temperature at approximately 105°C. However, the approximate ambient temperature at which the thermal feedback begins to protect the IC is: + Li-Ion BATTERY The LTC3558 can be used above 70°C, but the charge current will be reduced from 500mA. The approximate current at a given ambient temperature can be calculated: IBAT = 1.74k 1k 3558 F05 Figure 5. Combining Wall Adapter and USB Power Power Dissipation The conditions that cause the LTC3558 to reduce charge current through thermal feedback can be approximated by considering the power dissipated in the IC. For high charge currents, the LTC3558 power dissipation is approximately: PD = ( VCC – VBAT ) • IBAT where PD is the power dissipated, VCC is the input supply voltage, VBAT is the battery voltage, and IBAT is the charge 105°C – TA ( VCC – VBAT ) • θJA Using the previous example with an ambient temperature of 88°C, the charge current will be reduced to approximately: IBAT = 105°C – 88°C 17°C = (5V – 3.5V ) • 68°C / W 102°C / A IBAT = 167mA Furthermore, the voltage at the PROG pin will change proportionally with the charge current as discussed in the Programming Charge Current section. It is important to remember that LTC3558 applications do not need to be designed for worst-case thermal conditions since the IC will automatically reduce power dissipation when the junction temperature reaches approximately 105°C. 3558f 19 LTC3558 APPLICATIONS INFORMATION Battery Charger Stability Considerations The LTC3558 battery charger contains two control loops: the constant-voltage and constant-current loops. The constantvoltage loop is stable without any compensation when a battery is connected with low impedance leads. Excessive lead length, however, may add enough series inductance to require a bypass capacitor of at least 1.5μF from BAT to GND. Furthermore, a 4.7μF capacitor with a 0.2Ω to 1Ω series resistor from BAT to GND is required to keep ripple voltage low when the battery is disconnected. High value capacitors with very low ESR (especially ceramic) reduce the constant-voltage loop phase margin, possibly resulting in instability. Ceramic capacitors up to 22μF may be used in parallel with a battery, but larger ceramics should be decoupled with 0.2Ω to 1Ω of series resistance. In constant-current mode, the PROG pin is in the feedback loop, not the battery. Because of the additional pole created by the PROG pin capacitance, capacitance on this pin must be kept to a minimum. With no additional capacitance on the PROG pin, the charger is stable with program resistor values as high as 25K. However, additional capacitance on this node reduces the maximum allowed program resistor. The pole frequency at the PROG pin should be kept above 100kHz. Therefore, if the PROG pin is loaded with a capacitance, CPROG, the following equation should be used to calculate the maximum resistance value for RPROG: RPROG ≤ 1 Average, rather than instantaneous, battery current may be of interest to the user. For example, if a switching power supply operating in low-current mode is connected in parallel with the battery, the average current being pulled out of the BAT pin is typically of more interest than the instantaneous current pulses. In such a case, a simple RC filter can be used on the PROG pin to measure the average battery current as shown in Figure 6. A 10k resistor has been added between the PROG pin and the filter capacitor to ensure stability. USB Inrush Limiting When a USB cable is plugged into a portable product, the inductance of the cable and the high-Q ceramic input capacitor form an L-C resonant circuit. If there is not much impedance in the cable, it is possible for the voltage at the input of the product to reach as high as twice the USB voltage (~10V) before it settles out. In fact, due to the high voltage coefficient of many ceramic capacitors (a nonlinearity), the voltage may even exceed twice the USB voltage. To prevent excessive voltage from damaging the LTC3558 during a hot insertion, the soft connect circuit in Figure 7 can be employed. In the circuit of Figure 7, capacitor C1 holds MP1 off when the cable is first connected. Eventually C1 begins to charge up to the USB input voltage applying increasing gate support to MP1. The long time constant of R1 and C1 prevents the current from building up in the cable too fast thus dampening out any resonant overshoot. 5 2π • 10 • CPROG MP1 Si2333 VCC LTC3558 10k PROG GND RPROG CFILTER CHARGE CURRENT MONITOR CIRCUITRY 5V USB INPUT C1 100nF USB CABLE R1 40k C2 10μF LTC3558 GND 3558 F06 3558 F07 Figure 6. Isolated Capacitive Load on PROG Pin and Filtering Figure 7. USB Soft Connect Circuit 3558f 20 LTC3558 APPLICATIONS INFORMATION Buck Switching Regulator General Information Buck Switching Regulator Output Voltage Programming The LTC3558 contains a 2.25MHz constant-frequency current mode buck switching regulator that can provide up to 400mA. The switcher can be programmed for a minimum output voltage of 0.8V and can be used to power a microcontroller core, microcontroller I/O, memory or other logic circuitry. The regulator supports 100% duty cycle operation (dropout mode) when the input voltage drops very close to the output voltage and is also capable of operating in Burst Mode operation for highest efficiencies at light loads (Burst Mode operation is pin selectable). The buck switching regulator also includes soft-start to limit inrush current when powering on, short-circuit current protection, and switch node slew limiting circuitry to reduce radiated EMI. The buck switching regulator can be programmed for output voltages greater than 0.8V. The output voltage for the buck switching regulator is programmed using a resistor divider from the switching regulator output connected to its feedback pin (FB1), as shown in Figure 8, such that: VOUT = 0.8(1 + R1/R2) Typical values for R1 are in the range of 40k to 1M. The capacitor CFB cancels the pole created by feedback resistors and the input capacitance of the FB pin and also helps to improve transient response for output voltages much greater than 0.8V. A variety of capacitor sizes can be used for CFB but a value of 10pF is recommended for most applications. Experimentation with capacitor sizes between 2pF and 22pF may yield improved transient response if so desired by the user. A MODE pin sets the buck switching regulator in Burst Mode operation or pulse skip operating mode. The regulator is enabled individually through its enable pin. The buck regulator input supply (PVIN1) should be connected to the battery pin (BAT) and PVIN2. This allows the undervoltage lockout circuit on the BAT pin to disable the buck regulators when the BAT voltage drops below 2.45V. Do not drive the buck switching regulator from a voltage other than BAT. A 10μF decoupling capacitor from the PVIN1 pin to GND is recommended. Buck Switching Regulator Operating Modes The buck switching regulator includes two possible operating modes to meet the noise/power needs of a variety of applications. In pulse skip mode, an internal latch is set at the start of every cycle, which turns on the main P-channel MOSFET PVIN EN MP PWM CONTROL MODE SW MN L VOUT CO CFB R1 FB 0.8V GND R2 3558 F08 Figure 8. Buck Converter Application Circuit 3558f 21 LTC3558 APPLICATIONS INFORMATION switch. During each cycle, a current comparator compares the peak inductor current to the output of an error amplifier. The output of the current comparator resets the internal latch, which causes the main P-channel MOSFET switch to turn off and the N-channel MOSFET synchronous rectifier to turn on. The N-channel MOSFET synchronous rectifier turns off at the end of the 2.25MHz cycle or if the current through the N-channel MOSFET synchronous rectifier drops to zero. Using this method of operation, the error amplifier adjusts the peak inductor current to deliver the required output power. All necessary compensation is internal to the buck switching regulator requiring only a single ceramic output capacitor for stability. At light loads in pulse skip mode, the inductor current may reach zero on each pulse which will turn off the N-channel MOSFET synchronous rectifier. In this case, the switch node (SW1) goes high impedance and the switch node voltage will “ring”. This is discontinuous operation, and is normal behavior for a switching regulator. At very light loads in pulse skip mode, the buck switching regulator will automatically skip pulses as needed to maintain output regulation. At high duty cycle (VOUT > PVIN1 /2) in pulse skip mode, it is possible for the inductor current to reverse causing the buck converter to switch continuously. Regulation and low noise operation are maintained but the input supply current will increase to a couple mA due to the continuous gate switching. During Burst Mode operation, the buck switching regulator automatically switches between fixed frequency PWM operation and hysteretic control as a function of the load current. At light loads the buck switching regulator controls the inductor current directly and use a hysteretic control loop to minimize both noise and switching losses. During Burst Mode operation, the output capacitor is charged to a voltage slightly higher than the regulation point. The buck switching regulator then goes into sleep mode, during which the output capacitor provides the load current. In sleep mode, most of the switching regulator’s circuitry is powered down, helping conserve battery power. When the output voltage drops below a pre-determined value, the buck switching regulator circuitry is powered on and another burst cycle begins. The sleep time decreases as the load current increases. Beyond a certain load current point (about 1/4 rated output load current) the buck switching regulator will switch to a low noise constant-frequency PWM mode of operation, much the same as pulse skip operation at high loads. For applications that can tolerate some output ripple at low output currents, Burst Mode operation provides better efficiency than pulse skip at light loads. The buck switching regulator allows mode transition onthe-fly, providing seamless transition between modes even under load. This allows the user to switch back and forth between modes to reduce output ripple or increase low current efficiency as needed. Burst Mode operation is set by driving the MODE pin high, while pulse skip mode is achieved by driving the MODE pin low. Buck Switching Regulator in Shutdown The buck switching regulator is in shutdown when not enabled for operation. In shutdown, all circuitry in the buck switching regulator is disconnected from the regulator input supply, leaving only a few nanoamps of leakage pulled to ground through a 13k resistor on the switch (SW1) pin when in shutdown. Buck Switching Regulator Dropout Operation It is possible for the buck switching regulator’s input voltage to approach its programmed output voltage (e.g., a battery voltage of 3.4V with a programmed output voltage of 3.3V). When this happens, the PMOS switch duty cycle increases until it is turned on continuously at 100%. In this dropout condition, the respective output voltage equals the regulator’s input voltage minus the voltage drops across the internal P-channel MOSFET and the inductor. 3558f 22 LTC3558 APPLICATIONS INFORMATION Buck Switching Regulator Soft-Start Operation Buck Switching Regulator Inductor Selection Soft-start is accomplished by gradually increasing the peak inductor current for each switching regulator over a 500μs period. This allows an output to rise slowly, helping minimize the battery in-rush current required to charge up the regulator’s output capacitor. A soft-start cycle occurs when the buck switcher first turns on, or after a fault condition has occurred (thermal shutdown or UVLO). A soft-start cycle is not triggered by changing operating modes using the MODE pin. This allows seamless output operation when transitioning between operating modes. The buck switching regulator is designed to work with inductors in the range of 2.2μH to 10μH, but for most applications a 4.7μH inductor is suggested. Larger value inductors reduce ripple current which improves output ripple voltage. Lower value inductors result in higher ripple current which improves transient response time. To maximize efficiency, choose an inductor with a low DC resistance. For a 1.2V output efficiency is reduced about 2% for every 100mΩ series resistance at 400mA load current, and about 2% for every 300mΩ series resistance at 100mA load current. Choose an inductor with a DC current rating at least 1.5 times larger than the maximum load current to ensure that the inductor does not saturate during normal operation. If output short-circuit is a possible condition the inductor should be rated to handle the maximum peak current specified for the buck regulators. Buck Switching Regulator Switching Slew Rate Control The buck switching regulator contains circuitry to limit the slew rate of the switch node (SW1). This circuitry is designed to transition the switch node over a period of a couple of nanoseconds, significantly reducing radiated EMI and conducted supply noise while maintaining high efficiency. Buck Switching Regulator Low Supply Operation An undervoltage lockout (UVLO) circuit on PVIN1 shuts down the step-down switching regulators when BAT drops below 2.45V. This UVLO prevents the buck switching regulator from operating at low supply voltages where loss of regulation or other undesirable operation may occur. Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and don’t radiate much energy, but generally cost more than powdered iron core inductors with similar electrical characteristics. Inductors that are very thin or have a very small volume typically have much higher DCR losses, and will not give the best efficiency. The choice of which style inductor to use often depends more on the price vs size, performance, and any radiated EMI requirements than on what the buck regulator requires to operate. The inductor value also has an effect on Burst Mode operation. Lower inductor values will cause Burst Mode switching frequency to increase. 3558f 23 LTC3558 APPLICATIONS INFORMATION Table 2 shows several inductors that work well with the LTC3558 buck switching regulator. These inductors offer a good compromise in current rating, DCR and physical size. Consult each manufacturer for detailed information on their entire selection of inductors. Buck Switching Regulator Input/Output Capacitor Selection Low ESR (equivalent series resistance) ceramic capacitors should be used at switching regulator outputs as well as the switching regulator input supply. Ceramic capacitor dielectrics are a compromise between high dielectric constant and stability versus temperature and versus DC bias voltage. The X5R/X7R dielectrics offer the best compromise with high dielectric constant and acceptable performance over temperature and under bias. Do not use Y5V dielectrics. A 10μF output capacitor is sufficient for most applications. For good transient response and stability the output capacitor should retain at least 4μF of capacitance over operating temperature and bias voltage. The buck switching regulator input supply should be bypassed with a 10μF capacitor. Consult manufacturer for detailed information on their selection and specifications of ceramic capacitors. Many manufacturers now offer very thin (< 1mm tall) ceramic capacitors ideal for use in height-restricted designs. Table 3 shows a list of several ceramic capacitor manufacturers. Table 3: Recommended Ceramic Capacitor Manufacturers AVX (803) 448-9411 www.avxcorp.com Murata (714) 852-2001 www.murata.com Taiyo Yuden (408) 537-4150 www.t-yuden.com TDK (888) 835-6646 www.tdk.com Table 2. Recommended Inductors for Buck Switching Regulators INDUCTOR TYPE L (μH) MAX IDC (A) MAX DCR (mΩ) SIZE IN mm (L × W × H) MANUFACTURER DE2818C DE2812C 4.7 4.7 1.25 1.15 72* 130* 3 × 2.8 × 1.8 3 × 2.8 × 1.2 Toko www.toko.com CDRH3D16 4.7 0.9 110 4 × 4 × 1.8 Sumida www.sumida.com SD3118 SD3112 4.7 4.7 1.3 0.8 162 246 3.1 × 3.1 × 1.8 3.1 × 3.1 × 1.2 Cooper www.cooperet.com LPS3015 4.7 1.1 200 3 × 3 × 1.5 Coilcraft www.coilcraft.com *Typical DCR 3558f 24 LTC3558 APPLICATIONS INFORMATION Buck-Boost Switching Regulator The LTC3558 contains a 2.25MHz constant-frequency, voltage mode, buck-boost switching regulator. The regulator provides up to 400mA of output load current. The buck-boost switching regulator can be programmed for a minimum output voltage of 2.75V and can be used to power a microcontroller core, microcontroller I/O, memory, disk drive, or other logic circuitry. To suit a variety of applications, different mode functions allow the user to trade off noise for efficiency. Two modes are available to control the operation of the buck-boost regulator. At moderate to heavy loads, the constant-frequency PWM mode provides the least noise switching solution. At lighter loads, Burst Mode operation may be selected. Regulation is maintained by an error amplifier that compares the divided output voltage with a reference and adjusts the compensation voltage accordingly until the FB2 voltage has stabilized at 0.8V. The buck-boost switching regulator also includes soft-start to limit inrush current and voltage overshoot when powering on, short-circuit current protection, and switch node slew limiting circuitry for reduced radiated EMI. Buck-Boost Regulator PWM Operating Mode In PWM mode, the voltage seen at the feedback node is compared to a 0.8V reference. From the feedback voltage, an error amplifier generates an error signal seen at the VC2 pin. This error signal controls PWM waveforms that modulate switches A (input PMOS), B (input NMOS), C (output NMOS), and D (output PMOS). Switches A and B operate synchronously, as do switches C and D. If the input voltage is significantly greater than the programmed output voltage, then the regulator will operate in buck mode. In this case, switches A and B will be modulated, with switch D always on (and switch C always off), to stepdown the input voltage to the programmed output. If the input voltage is significantly less than the programmed output voltage, then the converter will operate in boost mode. In this case, switches C and D are modulated, with switch A always on (and switch B always off), to step up the input voltage to the programmed output. If the input voltage is close to the programmed output voltage, then the converter will operate in four-switch mode. While operating in four-switch mode, switches turn on as per the following sequence: switches A and D → switches A and C → switches B and D → switches A and D. Buck-Boost Regulator Burst Mode Operation In Burst Mode operation, the switching regulator uses a hysteretic feedback voltage algorithm to control the output voltage. By limiting FET switching and using a hysteretic control loop switching losses are greatly reduced. In this mode, output current is limited to 50mA. While in Burst Mode operation, the output capacitor is charged to a voltage slightly higher than the regulation point. The buck-boost converter then goes into a SLEEP state, during which the output capacitor provides the load current. The output capacitor is charged by charging the inductor until the input current reaches 250mA typical, and then discharging the inductor until the reverse current reaches 0mA typical. This process of bursting current is repeated until the feedback voltage has charged to the reference voltage plus 6mV (806mV typical). In the SLEEP state, most of the regulator’s circuitry is powered down, helping to conserve battery power. When the feedback voltage drops below the reference voltage minus 6mV (794mV typical), the switching regulator circuitry is powered on and another burst cycle begins. The duration for which the regulator operates in SLEEP depends on the load current and output capacitor value. The SLEEP time decreases as the load current increases. The maximum deliverable load current in Burst Mode operation is 50mA typical. The buck-boost regulator may not enter SLEEP if the load current is greater than 50mA. If the load current increases beyond this point while in Burst Mode operation, the output may lose regulation. Burst Mode operation provides a significant improvement in efficiency at light loads at the expense of higher output ripple when compared to PWM mode. For many noise-sensitive systems, Burst Mode operation might be undesirable at certain times (i.e., during a transmit or receive cycle of a wireless device), but highly desirable at others (i.e., when the device is in low power standby mode). 3558f 25 LTC3558 APPLICATIONS INFORMATION Buck-Boost Switching Regulator Output Voltage Programming The output filter zero is given by: f FILTER _ ZERO = The buck-boost switching regulator can be programmed for output voltages greater than 2.75V and less than 5.45V. To program the output voltage, a resistor divider is connected between VOUT2 and the feedback node (FB2) as shown in Figure 9. The output voltage is given by VOUT2 = 0.8(1 + R1/R2). 2 • π • RESR • COUT Hz where RESR is the capacitor equivalent series resistance. A troublesome feature in boost mode is the right-half plane zero (RHP), and is given by: f RHPZ = LTC3558 VOUT2 R1 PVIN22 Hz 2 • π • IOUT • L • VOUT2 The loop gain is typically rolled off before the RHP zero frequency. FB2 R2 A simple Type I compensation network, as shown in Figure 10, can be incorporated to stabilize the loop, but at the cost of reduced bandwidth and slower transient response. To ensure proper phase margin, the loop requires to be crossed over a decade before the LC double pole. 3558 F09 Figure 9. Programming the Buck-Boost Output Voltage Requires a Resistor Divider Connected Between VOUT2 and FB2 The unity-gain frequency of the error amplifier with the Type I compensation is given by: Closing the Feedback Loop The LTC3558 incorporates voltage mode PWM control. The control to output gain varies with operation region (buck, boost, buck-boost), but is usually no greater than 20. The output filter exhibits a double pole response given by: f FILTER _ POLE = 1 f UG = 1 Hz 2 • π • R1• CP1 1 Hz 2 • π • L • COUT where COUT is the output filter capacitor. VOUT2 + ERROR AMP 0.8V R1 FB2 – VC2 CP1 R2 3558 F10 Figure 10. Error Amplifier with Type I Compensation 3558f 26 LTC3558 APPLICATIONS INFORMATION Most applications demand an improved transient response to allow a smaller output filter capacitor. To achieve a higher bandwidth, Type III compensation is required. Two zeros are required to compensate for the double-pole response. Type III compensation also reduces any VOUT2 overshoot seen during a start-up condition. A Type III compensation circuit is shown in Figure 11 and yields the following transfer function: VC2 1 = VOUT 2 R1 (C1 + C2) • (1 + sR2C2) [1 + s (R1 + R3)C3 ] s ⎣⎡1 + sR2(C1|| C2)⎤⎦ (1 + sR3C3) A Type III compensation network attempts to introduce a phase bump at a higher frequency than the LC double pole. This allows the system to cross unity gain after the LC double pole, and achieve a higher bandwidth. While attempting to cross over after the LC double pole, the system must still cross over before the boost right-half plane zero. If unity gain is not reached sufficiently before the right-half plane zero, then the –180° of phase lag from the LC double pole combined with the –90° of phase lag from the right-half plane zero will result in negating the phase bump of the compensator. at the filter double pole. If they are placed at too low of a frequency, they will introduce too much gain to the system and the crossover frequency will be too high. The two high frequency poles should be placed such that the system crosses unity gain during the phase bump introduced by the zeros and before the boost right-half plane zero and such that the compensator bandwidth is less than the bandwidth of the error amp (typically 900kHz). If the gain of the compensation network is ever greater than the gain of the error amplifier, then the error amplifier no longer acts as an ideal op amp, and another pole will be introduced at the same point. Recommended Type III compensation components for a 3.3V output are: R1: 324kΩ RFB: 105kΩ C1: 10pF R2: 15kΩ C2: 330pF R3: 121kΩ C3: 33pF COUT : 22μF The compensator zeros should be placed either before or only slightly after the LC double pole such that their positive phase contributions offset the –180° that occurs LOUT : 2.2μH VOUT2 + ERROR AMP R3 0.8V R1 C3 FB2 – VC2 RFB C2 R2 C1 3558 F11 Figure 11. Error Amplifier with Type III Compensation 3558f 27 LTC3558 APPLICATIONS INFORMATION Input Current Limit Buck-Boost Switching Regulator Inductor Selection The input current limit comparator will shut the input PMOS switch off once current exceeds 700mA typical. Before the switch current limit, the average current limit amp (620mA typical) will source current into the feedback pin to drop the output voltage. The input current limit also protects against a short-circuit condition at the VOUT2 pin. The buck-boost switching regulator is designed to work with inductors in the range of 1μH to 5μH. For most applications, a 2.2μH inductor will suffice. Larger value inductors reduce ripple current which improves output ripple voltage. Lower value inductors result in higher ripple current and improved transient response time. To maximize efficiency, choose an inductor with a low DC resistance and a DC current rating at least 1.5 times larger than the maximum load current to ensure that the inductor does not saturate during normal operation. If output short-circuit is a possible condition, the inductor current should be rated to handle up to the peak current specified for the buck-boost regulator. Reverse Current Limit The reverse current limit comparator will shut the output PMOS switch off once current returning from the output exceeds 450mA typical. Output Overvoltage Protection If the feedback node were inadvertently shorted to ground, then the output would increase indefinitely with the maximum current that could be sourced from the input supply. The buck-boost regulator protects against this by shutting off the input PMOS if the output voltage exceeds a 5.75V maximum. Buck-Boost Regulator Soft-Start Operation Soft-start is accomplished by gradually increasing the reference voltage over a 500μs typical period. A softstart cycle occurs whenever the buck-boost is enabled, or after a fault condition has occurred (thermal shutdown or UVLO). A soft-start cycle is not triggered by changing operating modes. This allows seamless output operation when transitioning between Burst Mode operation and PWM mode operation. The inductor value also affects Burst Mode operation. Lower inductor values will cause Burst Mode switching frequencies to increase. Different core materials and shapes will change the size/current and price/current relationship of an inductor. Toroid or shielded pot cores in ferrite or permalloy materials are small and do not radiate much energy, but cost more than powdered iron core inductors with similar electrical characteristics. Inductors that are very thin or have a very small volume typically have much higher core and DCR losses and will not give the best efficiency. Table 4 shows some inductors that work well with the buck-boost regulator. These inductors offer a good compromise in current rating, DCR and physical size. Consult each manufacturer for detailed information on their entire selection of inductors. Table 4. Recommended Inductors for the Buck-Boost Switching Regulator. INDUCTOR TYPE L (μH) MAX IDC (A) MAX DCR (mΩ) SIZE IN mm (L × W × H) DB3018C D312C DE2812C DE2812C 2.4 2.2 2 2.7 1.31 1.14 1.4 1.2 80 140 81 87 3.8 × 3.8 × 1.4 3.6 × 3.6 × 1.2 3 × 3.2 × 1.2 3 × 3.2 × 1.2 Toko www.toko.com CDRH3D16 2.2 1.2 72 4 × 4 × 1.8 Sumida www.sumida.com SD12 2.2 1.8 74 5.2 × 5.2 × 1.2 Cooper www.cooperet.com MANUFACTURER *Typical DCR 3558f 28 LTC3558 APPLICATIONS INFORMATION Buck-Boost Switching Regulator Input/Output Capacitor Selection Low ESR (equivalent series resistance) ceramic capacitors should be used at both the buck-boost regulator input (PVIN2) and the output (VOUT2). It is recommended that the input be bypassed with a 10μF capacitor. The output should be bypassed with at least a 10μF capacitor if using Type I compensation and 22μF if using Type III compensation. The same selection criteria apply for the buck-boost regulator input and output capacitors as described in the Buck Switching Regulator Input/Output Capacitor Selection section. PCB Layout Considerations In order to deliver maximum charge current under all conditions, it is critical that the backside of the LTC3558 be soldered to the PC board ground. The LTC3558 has dual switching regulators. As with all switching regulators, care must be taken while laying out a PC board and placing components. The input decoupling capacitors, the output capacitor and the inductors must all be placed as close to the pins as possible and on the same side of the board as the LTC3558. All connections must also be made on the same layer. Place a local unbroken ground plane below these components. Avoid routing noisy high frequency lines such as those that connect to switch pins over or parallel to lines that drive high impedance inputs. 3558f 29 LTC3558 TYPICAL APPLICATIONS UP TO 500mA USB (4.3V TO 5.5V) OR AC ADAPTER VCC 110k BAT PVIN1 PVIN2 10μF NTC + 1 10μF 4.7μF 28.7K 100k (NTC) NTH50603NO1 510Ω SINGLE Li-lon CELL (2.7V TO 4.2V) LTC3558 1.8V AT 400mA 4.7μH SW1 CHRG 806k 1.74k PROG FB1 SUSP SWAB2 EN1 EN2 10μF 649k 2.2μH HPWR DIGITAL CONTROL 10pF 3.3V AT 400mA SWCD2 VOUT2 619k MODE 10μF FB2 GND2 (EXPOSED GND PAD) 200k 15k 150pF VC2 3558 TA02 Figure 12. Li-Ion to 3.3V at 400mA, 1.8V at 400mA and USB-Compatible Battery Charger As shown in Figure 12, the LTC3558 can be operated with no battery connected to the BAT pin. A 1Ω resistor in series with a 4.7μF capacitor at the BAT pin ensures battery charger stability. 10μF VCC decoupling capacitors are required for proper operation of the DC/DC converters. A three-resistor bias network for NTC sets hot and cold trip points at approximately 55°C and 0°C. The battery can be charged with up to 950mA of charge current when powered from a 5V wall adaptor, as shown in Figure 13. CHRG has a LED to provide a user with a visual indication of battery charge status. The buck-boost regulator starts up only after VOUT1 is up to approximately 0.7V. This provides a sequencing function which may be desirable in applications where a microprocessor needs to be powered up before peripherals. A Type III compensation network improves the transient response of the buck-boost switching regulator. 3558f 30 LTC3558 PACKAGE DESCRIPTION UD Package 20-Lead Plastic QFN (3mm × 3mm) (Reference LTC DWG # 05-08-1720 Rev A) 0.70 ±0.05 3.50 ± 0.05 (4 SIDES) 1.65 ± 0.05 2.10 ± 0.05 PACKAGE OUTLINE 0.20 ±0.05 0.40 BSC RECOMMENDED SOLDER PAD PITCH AND DIMENSIONS APPLY SOLDER MASK TO AREAS THAT ARE NOT SOLDERED 3.00 ± 0.10 (4 SIDES) BOTTOM VIEW—EXPOSED PAD R = 0.115 TYP 0.75 ± 0.05 R = 0.05 TYP PIN 1 TOP MARK (NOTE 6) PIN 1 NOTCH R = 0.20 TYP OR 0.25 × 45° CHAMFER 19 20 0.40 ± 0.10 1 2 1.65 ± 0.10 (4-SIDES) (UD20) QFN 0306 REV A 0.200 REF 0.00 – 0.05 NOTE: 1. DRAWING IS NOT A JEDEC PACKAGE OUTLINE 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.15mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 0.20 ± 0.05 0.40 BSC 3558f Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights. 31 LTC3558 TYPICAL APPLICATIONS UP TO 950mA 5V WALL ADAPTER 100k 510Ω VCC BAT PVIN1 NTC PVIN2 1μF 10μF 1.2V AT 400mA 4.7μH LTC3558 100k (NTC) SW1 324k CHRG FB1 PROG SWAB2 887Ω DIGITAL CONTROL SUSP HPWR MODE EN1 SINGLE Li-lon CELL (2.7V TO 4.2V) + 10pF 10μF 649k 2.2μH 3.3V AT 400mA SWCD2 VOUT2 121k 324k 22μF 33pF EN2 FB2 105k 15k 330pF 10pF VC2 3558 TA03 GND2 (EXPOSED PAD) GND Figure 13. Battery Charger Can Charge a Battery with Up to 950mA When Powered From a Wall Adapter RELATED PARTS PART NUMBER DESCRIPTION COMMENTS LTC3550 Dual Input USB/AC Adapter Li-Ion Battery Charger with Adjustable Output 600mA Buck Converter Synchronous Buck Converter, Efficiency: 93%, Adjustable Output at 600mA, Charge Current: 950mA Programmable, USB Compatible, Automatic Input Power Detection and Selection LTC3552 Standalone Linear Li-Ion Battery Charger with Adjustable Output Dual Synchronous Buck Converter Synchronous Buck Converter, Efficiency: >90%, Adjustable Outputs at 800mA and 400mA, Charge Current Programmable Up to 950mA, USB Compatible, 5mm × 3mm DFN-16 Package LTC3552-1 Standalone Linear Li-Ion Battery Charger with Dual Synchronous Buck Converter Synchronous Buck Converter, Efficiency: >90%, Outputs 1.8V at 800mA and 1.575 at 400mA, Charge Current Programmable up to 950mA, USB Compatible LTC3455 Dual DC/DC Converter with USB Power Manager and Li-Ion Battery Charger Seamless Transition Between Input Power Sources: Li-Ion Battery, USB and 5V Wall Adapter, Two High Efficiency DC/DC Converters: Up to 96%, Full Featured Li-Ion Battery Charger with Accurate USB Current Limiting (500mA/100mA) Pin-Selectable Burst Mode Operation, Hot SwapTM Output for SDIO and Memory Cards, 4mm × 4mm QFN-24 Package LTC3456 2-Cell, Multi-Output DC/DC Converter with Seamless Transition Between 2-Cell Battery, USB and AC Wall Adapter Input Power Sources, USB Power Manager Main Output: Fixed 3.3V Output, Core Output: Adjustable from 0.8V to VBATT(MIN), Hot Swap Output for Memory Cards, Power Supply Sequencing: Main and Hot Swap Accurate USB Current Limiting, High Frequency Operation: 1MHz, High Efficiency: Up to 92%, 4mm × 4mm QFN-24 Package LTC3559 USB Charger with Dual Buck Regulators Adjustable, Synchronous Buck Converters, Efficiency >90%, Outputs: Down to 0.8V at 400mA Each, Charge Current Programmable Up to 950mA, USB-Compatible, 3mm × 3mm QFN-16 Package LTC4080 500mA Standalone Charger with 300mA Synchronous Buck Charges Single-Cell Li-Ion Batteries, Timer Termination + C/10, Thermal Regulation, Buck Output: 0.8V to VBAT, Buck Input VIN: 2.7V to 5.5V, 3mm × 3mm DFN-10 Package Hot Swap is a trademark of Linear Technology Corporation. 3558f 32 Linear Technology Corporation LT 0408 • PRINTED IN USA 1630 McCarthy Blvd., Milpitas, CA 95035-7417 (408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com © LINEAR TECHNOLOGY CORPORATION 2008