ONSEMI ADP3198

8-Bit Programmable 2- to 4-Phase
Synchronous Buck Controller
ADP3198
FEATURES
12
SHUNT
REGULATOR
13
OSCILLATOR
UVLO
SHUTDOWN
+
GND 18
CMP
EN
+
1
DAC
+ 150mV
PWRGD
+
+
DAC
– 500mV
–
–
+
CMP
–
+
CMP
–
CURRENT
MEASUREMENT
AND LIMIT
7
This device uses a multimode PWM architecture to drive the
logic-level outputs at a programmable switching frequency that
can be optimized for VR size and efficiency. The phase relationship of the output signals can be programmed to provide 2-, 3-,
or 4-phase operation, allowing for the construction of up to
four complementary buck switching stages.
PWM1
RESET
29
PWM2
28 PWM3
RESET
2/3/4-PHASE
DRIVER LOGIC 27 PWM4
RESET
CURRENT
LIMIT
+
–
25
SW1
24
SW2
23
SW3
22
SW4
17
CSCOMP
15
CSREF
16
CSSUM
21
IMON
4
FB
14
LLSET
6
SS
IREF 20
COMP
5
FBRTN
–
The ADP31981 is a highly efficient, multiphase, synchronous buck
switching regulator controller optimized for converting a 12 V
main supply into the core supply voltage required by high performance Intel processors. It uses an internal 8-bit DAC to read
a voltage identification (VID) code directly from the processor,
which is used to set the output voltage between 0.5 V and 1.6 V.
30
THERMAL
THROTTLING
CONTROL
PRECISION
REFERENCE
GENERAL DESCRIPTION
OD
CROWBAR
ILIMIT 11
DELAY
+
CMP
DELAY
2
TTSENSE 10
VRHOT 9
VRFAN 8
–
CSREF
–
CURRENT BALANCING
CIRCUIT
–
850mV
19
SET EN
RESET
3
BOOT
VOLTAGE
AND
SOFT START
CONTROL
VID DAC
VIDSEL 40
ADP3198
32
33
34
35
36
37
38
39
VID7 VID6 VID5 VID4 VID3 VID2 VID1 VID0
06094-001
Desktop PC power supplies for next generation
Intel® processors
VRM modules
RT RAMPADJ
31
+
–
APPLICATIONS
VCC
+
Selectable 2-, 3-, or 4-phase operation at up to
1 MHz per phase
±11 mV worst-case differential sensing error over
temperature
Logic-level PWM outputs for interface to external high
power drivers
Enhanced PWM flex mode for excellent load transient
performance
Active current balancing between all output phases
Built-in power-good/crowbar blanking supports on-the-fly
VID code changes
Digitally programmable 0.5 V to 1.6 V output supports both
VR10.x and VR11 specifications
Programmable short-circuit protection with programmable
latch-off delay
FUNCTIONAL BLOCK DIAGRAM
Figure 1.
The ADP3198 has a built-in shunt regulator that allows the part
to be connected to the 12 V system supply through a series resistor.
The ADP3198 is specified over the extended commercial
temperature range of 0°C to 85°C and is available in a
40-lead LFCSP.
The ADP3198 also includes programmable no load offset and
slope functions to adjust the output voltage as a function of the
load current, optimally positioning it for a system transient. The
ADP3198 also provides accurate and reliable short-circuit
protection, adjustable current limiting, and a delayed powergood output that accommodates on-the-fly output voltage
changes requested by the CPU.
1
Protected by U.S. Patent Number 6,683,441; other patents pending.
©2008 SCILLC. All rights reserved.
January 2008 – Rev. 2
Publication Order Number:
ADP3198/D
ADP3198
TABLE OF CONTENTS
Features...............................................................................................1
Power-Good Monitoring ...........................................................13
Applications .......................................................................................1
Output Crowbar..........................................................................14
General Description..........................................................................1
Output Enable and UVLO.........................................................14
Functional Block Diagram...............................................................1
Thermal Monitoring...................................................................14
Revision History................................................................................2
Application Information ................................................................19
Specifications .....................................................................................3
Setting the Clock Frequency .....................................................19
Test Circuits .......................................................................................5
Soft Start Delay Time .................................................................19
Absolute Maximum Ratings ............................................................6
Current-Limit Latch-Off Delay Times.....................................19
ESD Caution ..................................................................................6
Inductor Selection.......................................................................19
Pin Configuration and Function Descriptions .............................7
Current Sense Amplifier ............................................................20
Typical Performance Characteristics..............................................9
Inductor DCR Temperature Correction..................................21
Theory of Operation.......................................................................10
Output Offset...............................................................................22
Start-Up Sequence ......................................................................10
COUT Selection..............................................................................22
Phase Detection Sequence .........................................................10
Power MOSFETs .........................................................................24
Master Clock Frequency ............................................................11
Ramp Resistor Selection ............................................................25
Output Voltage Differential Sensing ........................................11
COMP Pin Ramp ........................................................................25
Output Current Sensing .............................................................11
Current-Limit Setpoint ..............................................................25
Active Impedance Control Mode .............................................11
Feedback Loop Compensation Design ....................................25
Current Control Mode and Thermal Balance.........................11
CIN Selection and Input Current di/dt Reduction ..................27
Voltage Control Mode ................................................................12
Thermal Monitor Design...........................................................27
Current Reference .......................................................................12
Shunt Resistor Design ................................................................28
Enhanced PWM Mode...............................................................12
Tuning the ADP3198..................................................................28
Delay Timer .................................................................................12
Layout and Component Placement..........................................29
Soft Start .......................................................................................12
Outline Dimensions........................................................................31
Current-Limit, Short-Circuit, and Latch-Off Protection ......13
Ordering Guide ...........................................................................31
Dynamic VID ..............................................................................13
REVISION HISTORY
01/08 - Rev 2: Conversion to ON Semiconductor
8/06—Rev. 0 to Rev. A.
6/06—Revision 0: Initial Version
Rev. 2 | Page 2 of 31 | www.onsemi.com
ADP3198
SPECIFICATIONS
VCC = 5 V, FBRTN = GND, TA = 0°C to 85°C, unless otherwise noted.1
Table 1.
Parameter
REFERENCE CURRENT
Reference Bias Voltage
Reference Bias Current
ERROR AMPLIFIER
Output Voltage Range2
Accuracy
Symbol
Conditions
Min
Typ
Max
Unit
VIREF
IIREF
RIREF = 100 kΩ
14.25
1.5
15
15.75
V
μA
4.4
+11
V
mV
1.111
−82
+1
16.5
200
V
mV
LSB
μA
μA
μA
MHz
V/μs
mV
nA
ms
VCOMP
VFB
VFB(BOOT)
Load Line Positioning Accuracy
Differential Nonlinearity
Input Bias Current
FBRTN Current
Output Current
Gain Bandwidth Product
Slew Rate
LLSET Input Voltage Range
LLSET Input Bias Current
BOOT Voltage Hold Time
VID INPUTS
Input Low Voltage
Input High Voltage
Input Current
VID Transition Delay Time2
No CPU Detection Turn-Off Delay Time2
OSCILLATOR
Frequency Range2
Frequency Variation
Output Voltage
RAMPADJ Output Voltage
RAMPADJ Input Current Range
CURRENT SENSE AMPLIFIER
Offset Voltage
Input Bias Current
Gain Bandwidth Product
Slew Rate
Input Common-Mode Range
Output Voltage Range
Output Current
Current Limit Latch-Off Delay Time
IMON Output
CURRENT BALANCE AMPLIFIER
Common-Mode Range
Input Resistance
Input Current
Input Current Matching
CURRENT LIMIT COMPARATOR
ILIMIT Bias Current
IFB
IFBRTN
ICOMP
GBW(ERR)
VLLSET
ILLSET
tBOOT
Relative to nominal DAC output, referenced to
FBRTN, LLSET = CSREF (see Figure 2)
In startup
CSREF − LLSET = 80 mV
IFB = IIREF
FB forced to VOUT – 3%
COMP = FB
COMP = FB
Relative to CSREF
0
−11
1.089
−78
−1
13.5
15
65
500
20
25
−350
−10
CDELAY = 10 nF
VIL(VID)
VIH(VID)
IIN(VID)
1.1
−80
+350
+10
2
VID(X), VIDSEL
VID(X), VIDSEL
0.8
VID code change to FB change
VID code change to PWM going low
400
5
0.4
V
V
μA
ns
μs
4
240
MHz
kHz
kHz
kHz
V
mV
μA
−1
fOSC
fPHASE
VRT
VRAMPADJ
IRAMPADJ
VOS(CSA)
IBIAS(CSSUM)
GBW(CSA)
TA = 25°C, RT = 243 kΩ, 4-phase
TA = 25°C, RT = 113 kΩ, 4-phase
TA = 25°C, RT = 51 kΩ, 4-phase
RT = 243 kΩ to GND
RAMPADJ − FB
CSSUM − CSREF (see Figure 3)
CSSUM = CSCOMP
CCSCOMP = 10 pF
CSSUM and CSREF
0.25
156
1.9
−50
1
−2
−10
2.1
+50
50
+2
+10
+6
mV
nA
MHz
V/μs
V
V
μA
ms
%
17
12
+200
26
20
+5
mV
kΩ
μA
%
10
11
μA
10
10
0
0.05
ICSCOMP
tOC(DELAY)
IMON
CDELAY = 10 nF
10 × (CSREF − CSCOMP) > 50 mV
−6
VSW(X)CM
RSW(X)
ISW(X)
ΔISW(X)
SW(X) = 0 V
SW(X) = 0 V
SW(X) = 0 V
−600
10
8
−5
IILIMIT
IILIMIT = 2/3 × IIREF
9
Rev. 2 | Page 3 of 31 | www.onsemi.com
200
400
800
2.0
3.5
3.5
500
8
ADP3198
Parameter
ILIMIT Voltage
Maximum Output Voltage
Current-Limit Threshold Voltage
Current-Limit Setting Ratio
DELAY TIMER
Normal Mode Output Current
Output Current in Current Limit
Threshold Voltage
SOFT START
Output Current
ENABLE INPUT
Threshold Voltage
Hysteresis
Input Current
Delay Time
OD OUTPUT
Symbol
VILIMIT
Conditions
RILIMIT = 121 kΩ (VILIMIT = (IILIMIT × RILIMIT))
VCL
VCSREF − VCSCOMP, RILIMIT = 121 kΩ
VCL/VILIMIT
IDELAY
IDELAY(CL)
VDELAY(TH)
ISS
VTH(EN)
VHYS(EN)
IIN(EN)
tDELAY(EN)
Output Low Voltage
VOL(OD)
Output High Voltage
VOH(OD)
Min
1.09
3
80
Typ
1.21
Max
1.33
100
82.6
125
IDELAY = IIREF
IDELAY(CL) = 0.25 × IIREF
12
3.0
1.6
15
3.75
1.7
18
4.5
1.8
μA
μA
V
During startup, ISS = IIREF
12
15
18
μA
800
80
850
100
−1
2
900
125
mV
mV
μA
ms
160
500
4
5
V
60
kΩ
EN > 950 mV, CDELAY = 10 nF
OD Pull Down Resistor
THERMAL THROTTLING CONTROL
TTSENSE Voltage Range
TTSENSE Bias Current
TTSENSE VRFAN Threshold Voltage
TTSENSE VRHOT Threshold Voltage
TTSENSE Hysteresis
VRFAN Output Low Voltage
VRHOT Output Low Voltage
POWER-GOOD COMPARATOR
Undervoltage Threshold
Overvoltage Threshold
Output Low Voltage
Power-Good Delay Time
During Soft Start2
VID Code Changing
VID Code Static
Crowbar Trip Point
Crowbar Reset Point
Crowbar Delay Time
VID Code Changing
VID Code Static
PWM OUTPUTS
Output Low Voltage
Output High Voltage
SUPPLY
VCC2
DC Supply Current
UVLO Turn-On Current
UVLO Threshold Voltage
UVLO Turn-Off Voltage
Internally limited
VOL(VRFAN)
VOL(VRHOT)
IVRFAN(SINK) = −4 mA
IVRHOT(SINK) = −4 mA
VPWRGD(UV)
VPWRGD(OV)
VOL(PWRGD)
Relative to nominal DAC output
Relative to nominal DAC output
IPWRGD(SINK) = −4 mA
0
−133
1.06
765
−450
250
CDELAY = 10 nF
−123
1.105
810
50
150
150
−500
300
150
VCROWBAR
Relative to nominal DAC output
250
2
250
200
300
395
450
tCROWBAR
Relative to FBRTN
Overvoltage to PWM going low
100
250
400
100
VOL(PWM)
VOH(PWM)
VCC
IVCC
IPWM(SINK) = −400 μA
IPWM(SOURCE) = 400 μA
VSYSTEM = 12 V, RSHUNT = 340 Ω (see Figure 2)
1
All limits at temperature extremes are guaranteed via correlation using standard statistical quality control (SQC).
2
Guaranteed by design or bench characterization, not tested in production.
Rev. 2 | Page 4 of 31 | www.onsemi.com
300
300
V
μA
V
mV
mV
mV
mV
−550
350
300
mV
mV
mV
350
ms
μs
ns
mV
505
mV
μs
ns
160
5
500
mV
V
4.65
5
5.55
25
11
V
mA
mA
V
V
VSYSTEM = 13.2 V, RSHUNT = 340 Ω
VCC rising
VCC falling
mV
4.0
6.5
VUVLO
5
−113
1.15
855
Unit
V
V
mV
mV/V
9
4.1
ADP3198
TEST CIRCUITS
12V
680Ω
8-BIT CODE
+
1μF
680Ω
680Ω
680Ω
COMP
VIDSEL
VID0
VID1
VID2
VID3
VID4
VID5
VID6
VID7
VCC
EN
PWRGD
FBRTN
FB
COMP
SS
DELAY
VRFAN
VRHOT
TTSENSE
ADP3198
10kΩ
PWM1
PWM2
PWM3
PWM4
NC
SW1
SW2
SW3
SW4
NC
LLSET
14
ΔV
1V
06094-002
100nF
12V
ADP3198
680Ω
VCC
31
CSCOMP
17
39kΩ
100nF
CSSUM
16
1kΩ
CSREF
GND
18
VOS =
CSCOMP – 1V
40
06094-003
15
1V
VID
DAC
GND
ΔVFB = FBΔV = 80mV – FBΔV = 0mV
Figure 2. Closed-Loop Output Voltage Accuracy
680Ω
+
18
20kΩ
NC = NO CONNECT
–
CSREF
15
100kΩ
250kΩ
FB
3
ILIMIT
RT
RAMPADJ
LLSET
CSREF
CSSUM
CSCOMP
GND
OD
IREF
10nF
4
Figure 3. Current Sense Amplifier VOS
Rev. 2 | Page 5 of 31 | www.onsemi.com
Figure 4. Positioning Voltage
06094-004
1kΩ
10nF
1
VCC
31
100nF
40
1.25V
ADP3198
12V
ADP3198
ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter
VCC
FBRTN
PWM3 to PWM4, RAMPADJ
SW1 to SW4
<200 ns
All Other Inputs and Outputs
Storage Temperature Range
Operating Ambient Temperature Range
Operating Junction Temperature
Thermal Impedance (θJA)
Lead Temperature
Soldering (10 sec)
Infrared (15 sec)
Rating
−0.3 V to +6 V
−0.3 V to +0.3 V
−0.3 V to VCC + 0.3 V
−5 V to +25 V
−10 V to +25 V
−0.3 V to VCC + 0.3 V
−65°C to +150°C
0°C to 85°C
125°C
39°C/W
300°C
260°C
Stresses above those listed under Absolute Maximum Ratings
may cause permanent damage to the device. This is a stress
rating only; functional operation of the device at these or any
other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute
maximum rating conditions for extended periods may affect
device reliability.
Absolute maximum ratings apply individually only, not in
combination. Unless otherwise specified, all other voltages
referenced to GND.
ESD CAUTION
Rev. 2 | Page 6 of 31 | www.onsemi.com
ADP3198
40
39
38
37
36
35
34
33
32
31
VIDSEL
VID0
VID1
VID2
VID3
VID4
VID5
VID6
VID7
VCC
PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
PIN 1
INDICATOR
ADP3198
TOP VIEW
(Not to Scale)
30
29
28
27
26
25
24
23
22
21
PWM1
PWM2
PWM3
PWM4
NC
SW1
SW2
SW3
SW4
IMON
NOTES
1. NC = NO CONNECT.
2. THE EXPOSED EPAD ON BOTTOM SIDE OF PACKAGE IS AN
ELECTRICAL CONNECTION AND SHOULD BE SOLDERED TO GROUND.
06094-005
ILIMIT
RT
RAMPADJ
LLSET
CSREF
CSSUM
CSCOMP
GND
OD
IREF
11
12
13
14
15
16
17
18
19
20
EN 1
PWRGD 2
FBRTN 3
FB 4
COMP 5
SS 6
DELAY 7
VRFAN 8
VRHOT 9
TTSENSE 10
Figure 5. Pin Configuration
Table 3. Pin Function Descriptions
Pin No.
1
2
3
4
Mnemonic
EN
PWRGD
FBRTN
FB
5
6
COMP
SS
7
DELAY
8
VRFAN
9
VRHOT
10
TTSENSE
11
12
ILIMIT
RT
13
14
RAMPADJ
LLSET
15
CSREF
16
CSSUM
17
CSCOMP
18
19
GND
OD
Description
Power Supply Enable Input. Pulling this pin to GND disables the PWM outputs and pulls the PWRGD output low.
Power-Good Output. Open-drain output that signals when the output voltage is outside of the proper operating range.
Feedback Return. VID DAC and error amplifier reference for remote sensing of the output voltage.
Feedback Input. Error amplifier input for remote sensing of the output voltage. An external resistor between this
pin and the output voltage sets the no load offset point.
Error Amplifier Output and Compensation Point.
Soft Start Delay Setting Input. An external capacitor connected between this pin and GND sets the soft start
ramp-up time.
Delay Timer Setting Input. An external capacitor connected between this pin and GND sets the overcurrent latchoff delay time, boot voltage hold time, EN delay time, and PWRGD delay time.
VR Fan Activation Output. Open-drain output that signals when the temperature at the monitoring point
connected to TTSENSE exceeds the programmed VRFAN temperature threshold.
VR Hot Output. Open-drain output that signals when the temperature at the monitoring point connected to
TTSENSE exceeds the programmed VRHOT temperature threshold.
VR Hot Thermal Throttling Sense Input. An NTC thermistor between this pin and GND is used to remotely sense
the temperature at the desired thermal monitoring point.
Current-Limit Set Point. An external resistor from this pin to GND sets the current-limit threshold of the converter.
Frequency Setting Resistor Input. An external resistor connected between this pin and GND sets the oscillator
frequency of the device.
PWM Ramp Current Input. An external resistor from the converter input voltage to this pin sets the internal PWM ramp.
Output Load Line Programming Input. This pin can be directly connected to CSCOMP, or it can be connected to
the center point of a resistor divider between CSCOMP and CSREF. Connecting LLSET to CSREF disables positioning.
Current Sense Reference Voltage Input. The voltage on this pin is used as the reference for the current sense
amplifier and the power-good and crowbar functions. This pin should be connected to the common point of the
output inductors.
Current Sense Summing Node. External resistors from each switch node to this pin sum the average inductor
currents together to measure the total output current.
Current Sense Compensation Point. A resistor and capacitor from this pin to CSSUM determines the gain of the
current sense amplifier and the positioning loop response time.
Ground. All internal biasing and the logic output signals of the device are referenced to this ground.
Output Disable Logic Output. This pin is actively pulled low when the EN input is low or when VCC is below its
UVLO threshold to signal to the Driver IC that the driver high-side and low-side outputs should go low.
Rev. 2 | Page 7 of 31 | www.onsemi.com
ADP3198
Pin No.
20
Mnemonic
IREF
21
22 to 25
IMON
SW4 to SW1
26
27 to 30
NC
PWM4 to
PWM1
31
VCC
32 to 39
VID7 to VID0
40
VIDSEL
Description
Current Reference Input. An external resistor from this pin to ground sets the reference current for IFB, IDELAY, ISS,
IILIMIT, and ITTSENSE.
Analog Output. Represents the total load current.
Current Balance Inputs. Inputs for measuring the current level in each phase. The SW pins of unused phases
should be left open.
No Connection.
Logic-Level PWM Outputs. Each output is connected to the input of an external MOSFET driver such as the
ADP3110A. Connecting the PWM4, and PWM3 outputs to VCC causes that phase to turn off, allowing the
ADP3198 to operate as a 2-, 3-, or 4-phase controller.
Supply Voltage for the Device. A 340 Ω resistor should be placed between the 12 V system supply and the VCC
pin. The internal shunt regulator maintains VCC = 5 V.
Voltage Identification DAC Inputs. These eight pins are pulled down to GND, providing a Logic 0 if left open. When in
normal operation mode, the DAC output programs the FB regulation voltage from 0.5 V to 1.6 V (see Table 4).
VID DAC Selection Pin. The logic state of this pin determines whether the internal VID DAC decodes VID0 to VID7
as extended VR10 or VR11 inputs.
Rev. 2 | Page 8 of 31 | www.onsemi.com
ADP3198
TYPICAL PERFORMANCE CHARACTERISTICS
7000
6000
4000
MASTER CLOCK
3000
2000
PHASE 1
IN 4 PHASE DESIGN
1000
0
13
20
30
43
68
75
82 130 180 270 395 430 680 850
RT (kΩ)
06094-018
FREQUENCY (kHz)
5000
Figure 6. Master Clock Frequency vs. RT
Rev. 2 | Page 9 of 31 | www.onsemi.com
ADP3198
THEORY OF OPERATION
The multimode control of the ADP3198 ensures a stable,
high performance topology for the following:
•
Balancing currents and thermals between phases
•
High speed response at the lowest possible switching
frequency and output decoupling
•
Minimizing thermal switching losses by using lower
frequency operation
•
Tight load line regulation and accuracy
•
High current output due to 4-phase operation
•
Reduced output ripple due to multiphase cancellation
•
PC board layout noise immunity
•
Ease of use and design due to independent component
selection
•
Flexibility in operation for tailoring design to low cost or
high performance
START-UP SEQUENCE
The ADP3198 follows the VR11 start-up sequence shown in
Figure 7. After both the EN and UVLO conditions are met,
the DELAY pin goes through one cycle (TD1). The first four
clock cycles of TD2 are blanked from the PWM outputs and
used for phase detection as explained in the Phase Detection
Sequence section. Then, the soft start ramp is enabled (TD2),
and the output comes up to the boot voltage of 1.1 V. The boot
hold time is determined by the DELAY pin as it goes through a
second cycle (TD3). During TD3, the processor VID pins settle
to the required VID code. When TD3 is over, the ADP3198 soft
starts either up or down to the final VID voltage (TD4). After
TD4 is completed and the PWRGD masking time (equal to VID
on-the-fly masking) is completed, a third ramp on the DELAY
pin sets the PWRGD blanking (TD5).
5V
SUPPLY
VTT I/O
(ADP3198 EN)
UVLO
THRESHOLD
0.85V
VDELAY(TH)
(1.7V)
DELAY
1V
SS
VBOOT
(1.1V)
VVID
TD3
VCC_CORE
VBOOT
(1.1V)
TD1
VVID
TD4
TD2
VR READY
(ADP3198 PWRGD)
50μs
CPU
VID INPUTS
VID INVALID
TD5
VID VALID
06094-006
The ADP3198 combines a multimode, fixed frequency,
PWM control with multiphase logic outputs for use in 2-, 3-,
and 4-phase synchronous buck CPU core supply power
converters. The internal VID DAC is designed to interface
with the Intel 8-bit VRD/VRM 11-compatible and 7-bit
VRD/VRM 10×-compatible CPUs. Multiphase operation is
important for producing the high currents and low voltages
demanded by today’s microprocessors. Handling the high
currents in a single-phase converter places high thermal
demands on the components in the system, such as the
inductors and MOSFETs.
Figure 7. System Start-Up Sequence
PHASE DETECTION SEQUENCE
During startup, the number of operational phases and their
phase relationship is determined by the internal circuitry that
monitors the PWM outputs. Normally, the ADP3198 operates
as a 4-phase PWM controller. Connecting the PWM4 pin to
VCC programs 3-phase operation and connecting the PWM4
and PWM3 pins to VCC programs 2-phase operation.
Prior to soft start, while EN is low, the PWM3 and PWM4 pins
sink approximately 100 μA. An internal comparator checks each
pin’s voltage vs. a threshold of 3 V. If the pin is tied to VCC, it is
above the threshold. Otherwise, an internal current sink pulls
the pin to GND, which is below the threshold. PWM1 and
PWM2 are low during the phase detection interval that occurs
during the first four clock cycles of TD2. After this time, if the
remaining PWM outputs are not pulled to VCC, the 100 μA
current sink is removed, and they function as normal PWM
outputs. If they are pulled to VCC, the 100 μA current source is
removed, and the outputs are put into a high impedance state.
The PWM outputs are logic-level devices intended for driving
external gate drivers such as the ADP3110A. Because each
phase is monitored independently, operation approaching 100%
duty cycle is possible. In addition, more than one output can be
on at the same time to allow overlapping phases.
Rev. 2 | Page 10 of 31 | www.onsemi.com
ADP3198
MASTER CLOCK FREQUENCY
The clock frequency of the ADP3198 is set with an external
resistor connected from the RT pin to ground. The frequency
follows the graph in Figure 6. To determine the frequency per
phase, the clock is divided by the number of phases in use. If all
phases are in use, divide by 4. If PWM4 is tied to VCC, divide
the master clock by 3 for the frequency of the remaining phases.
If PWM3 and PWM4 are tied to VCC, divide by 2.
OUTPUT VOLTAGE DIFFERENTIAL SENSING
The ADP3198 combines differential sensing with a high
accuracy VID DAC and reference, and a low offset error amplifier. This maintains a worst-case specification of ±9.5 mV
differential sensing error over its full operating output voltage
and temperature range. The output voltage is sensed between
the FB pin and FBRTN pin. FB should be connected through
a resistor to the regulation point, usually the remote sense pin
of the microprocessor. FBRTN should be connected directly
to the remote sense ground point. The internal VID DAC
and precision reference are referenced to FBRTN, which has a
minimal current of 65 μA to allow accurate remote sensing. The
internal error amplifier compares the output of the DAC to the
FB pin to regulate the output voltage.
OUTPUT CURRENT SENSING
The ADP3198 provides a dedicated current-sense amplifier
(CSA) to monitor the total output current for proper voltage
positioning vs. load current and for current-limit detection.
Sensing the load current at the output gives the total average
current being delivered to the load, which is an inherently more
accurate method than peak current detection or sampling the
current across a sense element such as the low-side MOSFET.
This amplifier can be configured several ways, depending on
the objectives of the system, as follows:
•
Output inductor DCR sensing without a thermistor for
lowest cost
•
Output inductor DCR sensing with a thermistor for
improved accuracy with tracking of inductor temperature
•
Sense resistors for highest accuracy measurements
The positive input of the CSA is connected to the CSREF pin,
which is connected to the output voltage. The inputs to the
amplifier are summed together through resistors from the
sensing element, such as the switch node side of the output
inductors, to the inverting input CSSUM. The feedback resistor
between CSCOMP and CSSUM sets the gain of the amplifier
and a filter capacitor is placed in parallel with this resistor. The
gain of the amplifier is programmable by adjusting the feedback
resistor.
An additional resistor divider connected between CSREF and
CSCOMP (with the midpoint connected to LLSET) can be used
to set the load line required by the microprocessor. The current
information is then given as CSREF − LLSET. This difference
signal is used internally to offset the VID DAC for voltage
positioning. The difference between CSREF and CSCOMP is
then used as a differential input for the current-limit comparator.
This allows the load line to be set independently of the currentlimit threshold. In the event that the current-limit threshold
and load line are not independent, the resistor divider between
CSREF and CSCOMP can be removed and the CSCOMP pin
can be directly connected to LLSET. To disable voltage positioning entirely (that is, no load line), connect LLSET to CSREF.
To provide the best accuracy for sensing current, the CSA is
designed to have a low offset input voltage. Also, the sensing gain
is determined by external resistors to make it extremely accurate.
ACTIVE IMPEDANCE CONTROL MODE
For controlling the dynamic output voltage droop as a function
of output current, a signal proportional to the total output current
at the LLSET pin can be scaled to equal the regulator droop
impedance multiplied by the output current. This droop voltage
is then used to set the input control voltage to the system. The
droop voltage is subtracted from the DAC reference input
voltage to tell the error amplifier where the output voltage
should be. This allows enhanced feed-forward response.
CURRENT CONTROL MODE AND THERMAL
BALANCE
The ADP3198 has individual inputs (SW1 to SW4) for each
phase that are used for monitoring the current of each phase.
This information is combined with an internal ramp to create
a current balancing feedback system that has been optimized for
initial current balance accuracy and dynamic thermal balancing
during operation. This current balance information is independent
of the average output current information used for positioning
as described in the Output Current Sensing section.
The magnitude of the internal ramp can be set to optimize the
transient response of the system. It also monitors the supply
voltage for feed-forward control for changes in the supply. A
resistor connected from the power input voltage to the
RAMPADJ pin determines the slope of the internal PWM ramp.
External resistors can be placed in series with individual phases
to create an intentional current imbalance if desired, such as
when one phase has better cooling and can support higher
currents. Resistor RSW1 through Resistor RSW4 (see Figure 10) can
be used for adjusting thermal balance in this 4-phase example.
It is best to have the ability to add these resistors during the
initial design, therefore, ensure that placeholders are provided
in the layout.
To increase the current in any given phase, enlarge RSW for that
phase (make RSW = 0 for the hottest phase and do not change it
during balancing). Increasing RSW to only 500 Ω makes a
Rev. 2 | Page 11 of 31 | www.onsemi.com
ADP3198
substantial increase in phase current. Increase each RSW value
by small amounts to achieve balance, starting with the coolest
phase first.
VOLTAGE CONTROL MODE
A high gain, high bandwidth, voltage mode error amplifier is
used for the voltage mode control loop. The control input
voltage to the positive input is set via the VID logic according to
the voltages listed in Table 4.
This voltage is also offset by the droop voltage for active
positioning of the output voltage as a function of current,
commonly known as active voltage positioning. The output
of the amplifier is the COMP pin, which sets the termination
voltage for the internal PWM ramps.
The negative input (FB) is tied to the output sense location with
Resistor RB and is used for sensing and controlling the output
voltage at this point. A current source (equal to IREF) from the
FB pin flowing through RB is used for setting the no load offset
voltage from the VID voltage. The no load voltage is negative
with respect to the VID DAC. The main loop compensation is
incorporated into the feedback network between FB and COMP.
CURRENT REFERENCE
The IREF pin is used to set an internal current reference. This
reference current sets IFB, IDELAY, ISS, ILIMIT, and ITTSENSE. A resistor
to ground programs the current based on the 1.5 V output.
1.5 V
IREF =
R IREF
Typically, RIREF is set to 100 kΩ to program IREF = 15 μA. The
following currents are then equal to
IFB = IREF = 15 μA
IDELAY = IREF = 15 μA
ISS = IREF = 15 μA
Figure 7) is initiated. A current flows out of the DELAY pin to
charge CDLY. This current is equal to IREF, which is normally
15 μA. A comparator monitors the DELAY voltage with a
threshold of 1.7 V. The delay time is therefore set by the IREF
current charging a capacitor from 0 V to 1.7 V. This DELAY pin
is used for multiple delay timings (TD1, TD3, and TD5) during
the start-up sequence. In addition, DELAY is used for timing
the current-limit latch off, as explained in the Current-Limit,
Short-Circuit, and Latch-Off Protection section.
SOFT START
The soft start times for the output voltage are set with a
capacitor from the SS pin to ground. After TD1 and the phase
detection cycle are complete, the SS time (TD2 in Figure 7)
starts. The SS pin is disconnected from GND, and the capacitor
is charged up to the 1.1 V boot voltage by the SS amplifier,
which has an output current equal to IREF (normally 15 μA).
The voltage at the FB pin follows the ramping voltage on the
SS pin, limiting the inrush current during startup. The soft start
time depends on the value of the boot voltage and CSS.
Once the SS voltage is within 100 mV of the boot voltage, the
boot voltage delay time (TD3 in Figure 7) is started. The end of
the boot voltage delay time signals the beginning of the second
soft start time (TD4 in Figure 7). The SS voltage now changes
from the boot voltage to the programmed VID DAC voltage
(either higher or lower) using the SS amplifier with the output
current equal to IREF. The voltage of the FB pin follows the
ramping voltage of the SS pin, limiting the inrush current
during the transition from the boot voltage to the final DAC
voltage. The second soft start time depends on the boot voltage,
the programmed VID DAC voltage, and CSS.
If EN is taken low or if VCC drops below UVLO, DELAY and
SS are reset to ground to be ready for another soft start cycle.
Figure 8 shows typical start-up waveforms for the ADP3198.
ILIMIT = 2/3 (IREF) = 10 μA
Enhanced PWM mode is intended to improve the transient
response of the ADP3198 to a load setup. In previous
generations of controllers, when a load step up occurred, the
controller had to wait until the next turn-on of the PWM signal
to respond to the load change. Enhanced PWM mode allows
the controller to immediately respond when a load step up
occurs. This allows the phases to respond more quickly when a
load increase takes place.
DELAY TIMER
The delay times for the start-up timing sequence are set with
a capacitor from the DELAY pin to ground. In UVLO, or when
EN is logic low, the DELAY pin is held at ground. After the
UVLO and EN signals are asserted, the first delay time (TD1 in
1
2
3
4
CH1 1V
CH3 1V
CH2 1V
CH4 10V
M 1ms
T 40.4%
A CH1
700mV
06094-007
ENHANCED PWM MODE
Figure 8. Typical Start-Up Waveforms (Channel 1: CSREF,
Channel 2: DELAY, Channel 3: SS, and Channel 4: Phase 1 Switch Node)
Rev. 2 | Page 12 of 31 | www.onsemi.com
ADP3198
CURRENT-LIMIT, SHORT-CIRCUIT, AND LATCHOFF PROTECTION
If the limit is reached and TD5 in Figure 7 has completed, a
latch-off delay time starts, and the controller shuts down if the
fault is not removed. The current-limit delay time shares the
DELAY pin timing capacitor with the start-up sequence timing.
However, during current limit, the DELAY pin current is
reduced to IREF/4. A comparator monitors the DELAY voltage
and shuts off the controller when the voltage reaches 1.7 V.
Therefore, the current-limit latch-off delay time is set by the
current of IREF/4 charging the delay capacitor from 0 V to 1.7 V.
This delay is four times longer than the delay time during the
start-up sequence.
The current-limit delay time starts only after the TD5 is
complete. If there is a current limit during startup, the
ADP3198 goes through TD1 to TD5, and then starts the latchoff time. Because the controller continues to cycle the phases
during the latch-off delay time, the controller returns to normal
operation and the DELAY capacitor is reset to GND if the short
is removed before the 1.7 V threshold is reached.
The latch-off function can be reset by either removing and
reapplying the supply voltage to the ADP3198, or by toggling
the EN pin low for a short time. To disable the short-circuit
latch-off function, an external resistor should be placed in
parallel with CDLY. This prevents the DELAY capacitor from
charging up to the 1.7 V threshold. The addition of this resistor
causes a slight increase in the delay times.
During startup, when the output voltage is below 200 mV,
a secondary current limit is active. This is necessary because
the voltage swing of CSCOMP cannot go below ground. This
secondary current limit controls the internal COMP voltage
to the PWM comparators to 1.5 V. This limits the voltage drop
across the low-side MOSFETs through the current balance
circuitry. An inherent per-phase current limit protects
individual phases if one or more phases stop functioning
because of a faulty component. This limit is based on the
maximum normal mode COMP voltage. Typical overcurrent
latch-off waveforms are shown in Figure 9.
1
2
3
4
CH1 1V
CH3 2V
CH2 1V
CH4 10V
M 2ms
T 61.8%
A CH1
680mV
06094-008
The ADP3198 compares a programmable current-limit set
point to the voltage from the output of the current-sense
amplifier. The level of current limit is set with the resistor
from the ILIMIT pin to ground. During operation, the current
from ILIMIT is equal to 2/3 of IREF, giving 10 μA normally.
This current through the external resistor sets the ILIMIT
voltage, which is internally scaled to give a current limit
threshold of 82.6 mV/V. If the difference in voltage between
CSREF and CSCOMP rises above the current-limit threshold,
the internal current-limit amplifier controls the internal COMP
voltage to maintain the average output current at the limit.
Figure 9. Overcurrent Latch-Off Waveforms (Channel 1: CSREF,
Channel 2: DELAY, Channel 3: COMP, and Channel 4: Phase 1 Switch Node)
DYNAMIC VID
The ADP3198 has the ability to dynamically change the VID
inputs while the controller is running. This allows the output
voltage to change while the supply is running and supplying
current to the load. This is commonly referred to as VID onthe-fly (OTF). A VID OTF can occur under light or heavy load
conditions. The processor signals the controller by changing the
VID inputs in multiple steps from the start code to the finish
code. This change can be positive or negative.
When a VID input changes state, the ADP3198 detects the
change and ignores the DAC inputs for a minimum of 400 ns.
This time prevents a false code due to logic skew while the eight
VID inputs are changing. Additionally, the first VID change
initiates the PWRGD and crowbar blanking functions for a
minimum of 100 μs to prevent a false PWRGD or crowbar
event. Each VID change resets the internal timer.
POWER-GOOD MONITORING
The power-good comparator monitors the output voltage via
the CSREF pin. The PWRGD pin is an open-drain output whose
high level, when connected to a pull-up resistor, indicates that the
output voltage is within the nominal limits specified based on the
VID voltage setting. PWRGD goes low if the output voltage is
outside of this specified range, if the VID DAC inputs are in no
CPU mode, or if the EN pin is pulled low. PWRGD is blanked
during a VID OTF event for a period of 200 μs to prevent false
signals during the time the output is changing.
The PWRGD circuitry also incorporates an initial turn-on delay
time (TD5), based on the DELAY timer. Prior to the SS voltage
reaching the programmed VID DAC voltage and the PWRGD
masking-time finishing, the PWRGD pin is held low. Once the SS
pin is within 100 mV of the programmed DAC voltage, the
capacitor on the DELAY pin begins to charge. A comparator
monitors the DELAY voltage and enables PWRGD when the
voltage reaches 1.7 V. The PWRGD delay time is set, therefore, by
a current of IREF, charging a capacitor from 0 V to 1.7 V.
Rev. 2 | Page 13 of 31 | www.onsemi.com
ADP3198
OUTPUT CROWBAR
Grounding OD disables the drivers such that both DRVH and
DRVL are grounded. This feature is important in preventing the
discharge of the output capacitors when the controller is shut
off. If the driver outputs are not disabled, a negative voltage can
be generated during output due to the high current discharge of
the output capacitors through the inductors.
To protect the load and output components of the supply, the
PWM outputs are driven low, which turns on the low-side
MOSFETs when the output voltage exceeds the upper crowbar
threshold. This crowbar action stops once the output voltage
falls below the release threshold of approximately 375 mV.
Turning on the low-side MOSFETs pulls down the output as
the reverse current builds up in the inductors. If the output
overvoltage is due to a short in the high-side MOSFET, this
action current limits the input supply or blows its fuse,
protecting the microprocessor from being destroyed.
THERMAL MONITORING
OUTPUT ENABLE AND UVLO
A fixed current of 8 × IREF (normally giving 123 μA) is sourced
out of the TTSENSE pin and into the thermistor. The current
source is internally limited to 5 V. An internal circuit compares
the TTSENSE voltage to a 1.105 V and a 0.81 V threshold, and
outputs an open-drain signal at the VRFAN and VRHOT
outputs, respectively. Once the voltage on the TTSENSE pin
drops below its respective threshold, the open-drain outputs
assert high to signal the system that an overtemperature event
has occurred. Because the TTSENSE voltage changes slowly
with respect to time, 50 mV of hysteresis is built into these comparators. The thermal monitoring circuitry does not depend on
EN and is active when UVLO is above its threshold. When UVLO
is below its threshold, VRFAN and VRHOT are forced low.
The ADP3198 includes a thermal-monitoring circuit to detect
when a point on the VR has exceeded two different userdefined temperatures. The thermal-monitoring circuit requires
an NTC thermistor to be placed between TTSENSE and GND.
For the ADP3198 to begin switching, the input supply (VCC) to
the controller must be higher than the UVLO threshold and the
EN pin must be higher than its 0.85 V threshold. This initiates a
system start-up sequence. If either UVLO or EN is less than
their respective thresholds, the ADP3198 is disabled. This holds
the PWM outputs at ground, shorts the DELAY capacitor to
ground, and forces PWRGD and OD signals low.
In the application circuit (see Figure 10), the OD pin should be
connected to the OD inputs of the ADP3110A drivers.
Table 4.VR11 and VR10.x VID Codes for the ADP3198
OUTPUT
OFF
OFF
1.60000
1.59375
1.58750
1.58125
1.57500
1.56875
1.56250
1.55625
1.55000
1.54375
1.53750
1.53125
1.52500
1.51875
1.51250
1.50625
1.50000
1.49375
1.48750
1.48125
1.47500
1.46875
1.46250
1.45625
VID7
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
VID6
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
VR11 DAC CODES: VIDSEL = HIGH
VID5 VID4 VID3 VID2
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
1
0
0
1
1
0
VID1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
VID0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
VID4
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
Rev. 2 | Page 14 of 31 | www.onsemi.com
VR10.x DAC CODES: VIDSEL = LOW
VID3 VID2 VID1 VID0 VID5
N/A
N/A
1
0
1
0
1
1
0
1
0
1
1
0
1
1
0
1
0
1
1
0
1
0
1
1
1
1
0
1
1
1
1
1
0
0
0
1
1
0
0
0
1
1
0
0
1
1
1
0
0
1
1
1
0
1
0
1
1
0
1
0
1
1
0
1
1
1
1
0
1
1
1
1
1
0
0
1
1
1
0
0
1
1
1
0
1
1
1
1
0
1
1
1
1
1
0
1
1
1
1
0
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
VID6
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
ADP3198
OUTPUT
1.45000
1.44375
1.43750
1.43125
1.42500
1.41875
1.41250
1.40625
1.40000
1.39375
1.38750
1.38125
1.37500
1.36875
1.36250
1.35625
1.35000
1.34375
1.33750
1.33125
1.32500
1.31875
1.31250
1.30625
1.30000
1.29375
1.28750
1.28125
1.27500
1.26875
1.26250
1.25625
1.25000
1.24375
1.23750
1.23125
1.22500
1.21875
1.21250
1.20625
1.20000
1.19375
1.18750
1.18125
1.17500
1.16875
1.16250
1.15625
1.15000
1.14375
1.13750
1.13125
1.12500
1.11875
1.11250
1.10625
1.10000
1.09375
VID7
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
VID6
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
VR11 DAC CODES: VIDSEL = HIGH
VID5 VID4 VID3 VID2
0
1
1
0
0
1
1
0
0
1
1
1
0
1
1
1
0
1
1
1
0
1
1
1
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
1
1
0
1
1
1
0
1
1
1
0
1
1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
1
1
0
1
1
1
0
1
1
1
0
1
1
1
1
0
1
1
1
0
1
1
1
0
1
1
1
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
VID1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
VID0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
VID4
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
Rev. 2 | Page 15 of 31 | www.onsemi.com
VR10.x DAC CODES: VIDSEL = LOW
VID3 VID2 VID1 VID0 VID5
0
0
0
0
1
0
0
0
0
1
0
0
0
1
0
0
0
0
1
0
0
0
0
1
1
0
0
0
1
1
0
0
1
0
0
0
0
1
0
0
0
0
1
0
1
0
0
1
0
1
0
0
1
1
0
0
0
1
1
0
0
0
1
1
1
0
0
1
1
1
0
1
0
0
0
0
1
0
0
0
0
1
0
0
1
0
1
0
0
1
0
1
0
1
0
0
1
0
1
0
0
1
0
1
1
0
1
0
1
1
0
1
1
0
0
0
1
1
0
0
0
1
1
0
1
0
1
1
0
1
0
1
1
1
0
0
1
1
1
0
0
1
1
1
1
0
1
1
1
1
1
0
0
0
0
1
0
0
0
0
1
0
0
0
1
1
0
0
0
1
1
0
0
1
0
1
0
0
1
0
1
0
0
1
1
1
0
0
1
1
1
0
1
0
0
1
0
1
0
0
1
0
1
0
1
1
0
1
0
1
1
0
1
1
0
1
0
1
1
0
1
0
1
1
1
1
0
1
1
1
1
1
0
0
0
1
1
0
0
0
1
1
0
0
1
1
1
0
0
1
1
1
0
1
0
1
1
0
1
0
1
1
0
1
1
1
1
0
1
1
1
1
1
0
0
1
1
1
0
0
1
1
1
0
1
1
1
1
0
1
VID6
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
ADP3198
OUTPUT
OFF
OFF
OFF
OFF
1.08750
1.08125
1.07500
1.06875
1.06250
1.05625
1.05000
1.04375
1.03750
1.03125
1.02500
1.01875
1.01250
1.00625
1.00000
0.99375
0.98750
0.98125
0.97500
0.96875
0.96250
0.95625
0.95000
0.94375
0.93750
0.93125
0.92500
0.91875
0.91250
0.90625
0.90000
0.89375
0.88750
0.88125
0.87500
0.86875
0.86250
0.85625
0.85000
0.84375
0.83750
0.83125
0.82500
0.81875
0.81250
0.80625
0.80000
0.79375
0.78750
0.78125
0.77500
0.76875
0.76250
0.75625
VID7
VID6
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
VR11 DAC CODES: VIDSEL = HIGH
VID5 VID4 VID3 VID2
N/A
N/A
N/A
N/A
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
1
0
1
1
1
0
1
1
1
0
1
1
1
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
1
1
0
1
1
1
0
1
1
1
0
1
1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
1
1
0
1
1
1
0
1
1
1
0
1
1
1
1
0
1
1
1
0
1
1
1
0
1
1
1
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
1
0
0
0
1
0
VID1
VID0
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
VID4
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
Rev. 2 | Page 16 of 31 | www.onsemi.com
VR10.x DAC CODES: VIDSEL = LOW
VID3 VID2 VID1 VID0 VID5
1
1
1
1
0
1
1
1
1
0
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
0
0
0
0
1
0
0
0
1
0
0
0
0
1
0
0
0
0
1
1
0
0
0
1
1
0
0
1
0
0
0
0
1
0
0
0
0
1
0
1
0
0
1
0
1
0
0
1
1
0
0
0
1
1
0
0
0
1
1
1
0
0
1
1
1
0
1
0
0
0
0
1
0
0
0
0
1
0
0
1
0
1
0
0
1
0
1
0
1
0
0
1
0
1
0
0
1
0
1
1
0
1
0
1
1
0
1
1
0
0
0
1
1
0
0
0
1
1
0
1
0
1
1
0
1
0
1
1
1
0
0
1
1
1
0
0
1
1
1
1
0
1
1
1
1
1
0
0
0
0
1
0
0
0
0
1
0
0
0
1
1
0
0
0
1
1
0
0
1
0
1
0
0
1
0
1
0
0
1
1
1
0
0
1
1
1
0
1
0
0
1
0
1
0
0
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
VID6
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
ADP3198
OUTPUT
0.75000
0.74375
0.73750
0.73125
0.72500
0.71875
0.71250
0.70625
0.70000
0.69375
0.68750
0.68125
0.67500
0.66875
0.66250
0.65625
0.65000
0.64375
0.63750
0.63125
0.62500
0.61875
0.61250
0.60625
0.60000
0.59375
0.58750
0.58125
0.57500
0.56875
0.56250
0.55625
0.55000
0.54375
0.53750
0.53125
0.52500
0.51875
0.51250
0.50625
0.50000
OFF
OFF
VID7
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
VID6
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
VR11 DAC CODES: VIDSEL = HIGH
VID5 VID4 VID3 VID2
0
0
1
0
0
0
1
0
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
1
0
1
1
1
0
1
1
1
0
1
1
1
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
0
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
1
1
0
1
1
1
0
1
1
1
0
1
1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
1
1
1
1
1
1
VID1
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
0
0
1
1
1
VID0
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
1
0
0
1
VID4
1
1
Rev. 2 | Page 17 of 31 | www.onsemi.com
VR10.x DAC CODES: VIDSEL = LOW
VID3 VID2 VID1 VID0 VID5
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
N/A
1
1
1
1
1
1
1
1
1
1
VID6
0
1
Figure 10. Typical 4-Phase Application Circuit
Rev. 2 | Page 18 of 31 | www.onsemi.com
R3
1Ω
CDLY
18nF
RA
CA
560pF 13.7kΩ
CFB
15pF
RTH1
100kΩ, 5%
NTC
CSS
18nF
C5
1nF
C6
0.1μF
1
RT
130kΩ
1%
RLIM
205kΩ
1%
EN
PWRGD
FBRTN
FB
COMP
SS
DELAY
VRFAN
VRHOT
TTSENSE
40
C4
1μF
FROM CPU
+
560Ω
C3
100μF
(C3 OPTIONAL)
R2
267kΩ
1%
12V
C2
+
560Ω
U1
ADP3198
C7
1nF
CCS1
2nF
5% NPO
RIREF
100kΩ
PWM1
PWM2
PWM3
PWM4
NC
SW1
SW2
SW3
SW4
IMON
1 FOR A DESCRIPTION OF OPTIONAL R
SW RESISTORS, SEE THE THEORY OF OPERATION SECTION.
2 CONNECT NEAR EACH INDUCTOR.
C8
1nF
RB
1.21kΩ
CB
680pF
POWER GOOD
VRFAN
PROCHOT
VTT I/O
1μF
1kΩ
C1
+
2700μF/16V/3.3A × 2
SANYO MV-WX SERIES
VIDSEL
VID0
VID1
VID2
VID3
VID4
VID5
VID6
VID7
VCC
ILIMIT
RT
RAMPADJ
LLSET
CSREF
CSSUM
CSCOMP
GND
OD
IREF
VIN
RTN
L1
370nH
18A
CCS2
2.2nF
5% NPO
RPH2
RPH4
93.1kΩ 93.1kΩ
1%
1%
RSW21
RCS1
RCS2
RPH3
35.7kΩ 82.5kΩ 93.1kΩ
1%
RSW41
RSW3 1
RSW11
RPH1
93.1kΩ
1%
C22
4.7μF
D5
1N4148
C18
4.7μF
D4
1N4148
C14
4.7μF
D3
1N4148
C10
4.7μF
D2
1N4148
OD
3
PGND 6
DRVL 5
VCC
4
SW 7
DRVH 8
OD
IN
2
3
BST
1
C23
10nF
C21
18nF
DRVL 5
PGND 6
SW 7
DRVH 8
U5
ADP3110A
R7
2.2Ω
VCC
IN
4
BST
2
C19
10nF
C17
18nF
DRVL 5
PGND 6
SW 7
DRVH 8
U4
ADP3110A
R6
2.2Ω
VCC
OD
IN
BST
1
4
3
2
1
C13
18nF
R5
2.2Ω
C15
10nF
DRVL 5
4 VCC
U3
ADP3110A
PGND 6
SW 7
DRVH 8
3 OD
2 IN
1 BST
C11
10nF
C9
18nF
U2
ADP3110A
R4
2.2Ω
Q15
NTD110N02
Q13
NTD40N03
Q11
NTD110N02
Q9
NTD40N03
Q7
NTD110N02
Q5
NTD40N03
Q3
NTD110N02
Q1
NTD40N03
C12
4.7μF
C16
4.7μF
C20
4.7μF
C24
4.7μF
Q16
NTD110N02
Q14
NTD40N03
L5
320nH/1.4mΩ
Q12
NTD110N02
Q10
NTD40N03
L4
320nH/1.4mΩ
Q8
NTD110N02
Q6
NTD40N03
L3
320nH/1.4mΩ
Q4
NTD110N02
+
C25
RTH2
100kΩ, 5%
NTC
10Ω2
10Ω2
10Ω2
10Ω2
+
C32
Q2
NTD40N03
560μF/4V × 8
L2
320nH/1.4mΩ SANYO SEPC SERIES
5mΩ EACH
VCC(CORE) RTN
VCC(CORE)
0.5V TO 1.6V
115A TDC, 130A PK
VSS(SENSE)
VCC(SENSE)
22μF × 18
MLCC
IN SOCKET
06094-009
VIN
12V
ADP3198
ADP3198
APPLICATION INFORMATION
The design parameters for a typical Intel VRD 11 compliant
CPU application are as follows:
•
Input voltage (VIN) = 12 V
•
VID setting voltage (VVID) = 1.300 V
•
Duty cycle (D) = 0.108
Assuming a desired TD2 time of 3 ms, CSS is 41 nF. The closest
standard value for CSS is 39 nF. Although CSS also controls the
time delay for TD4 (determined by the final VID voltage), the
minimum specification for TD4 is 0 ns. This means that as long
as the TD2 time requirement is met, TD4 is within the
specification.
•
Nominal output voltage at no load (VONL) = 1.285 V
CURRENT-LIMIT LATCH-OFF DELAY TIMES
•
Nominal output voltage at 115 A load (VOFL) = 1.170 V
•
Static output voltage drop based on a 1.0 mΩ load line (RO)
from no load to full load (VD) = VONL − VOFL =
1.285 V − 1.170 V = 115 mV
•
Maximum output current (IO) = 130 A
The start-up and current-limit delay times are determined by
the capacitor connected to the DELAY pin. The first step is to
set CDLY for the TD1, TD3, and TD5 delay times (see Figure 7).
The DELAY ramp (IDELAY) is generated using a 15 μA internal
current source. The value for CDLY can be approximated using
•
Maximum output current step (ΔIO) = 100 A
•
Maximum output current slew rate (SR) = 200 A/μs
•
Number of phases (n) = 4
•
Switching frequency per phase (fSW) = 330 kHz
C DLY = I DELAY ×
SETTING THE CLOCK FREQUENCY
The ADP3198 uses a fixed frequency control architecture. The
frequency is set by an external timing resistor (RT). The clock
frequency and the number of phases determine the switching
frequency per phase, which relates directly to switching losses
as well as the sizes of the inductors, the input capacitors, and
output capacitors. With n = 4 for four phases, a clock frequency
of 1.32 MHz sets the switching frequency (fSW) of each phase to
330 kHz, which represents a practical trade-off between the
switching losses and the sizes of the output filter components.
Figure 6 shows that to achieve a 1.32 MHz oscillator frequency,
the correct value for RT is 130 kΩ. Alternatively, the value for RT
can be calculated using
RT =
1
n × f SW × 6 pF
(1)
where 6 pF is the internal IC component values. For good initial
accuracy and frequency stability, a 1% resistor is recommended.
SOFT START DELAY TIME
The value of CSS sets the soft start time. The ramp is generated
with a 15 μA internal current source. The value for CSS can be
found using
C SS = 15 μA ×
TD2
V BOOT
(2)
TD( x )
VDELAY (TH )
(3)
where TD(x) is the desired delay time for TD1, TD3, and TD5.
The DELAY threshold voltage (VDELAY(TH)) is given as 1.7 V. In
this example, 2 ms is chosen for all three delay times, which
meets Intel specifications. Solving for CDLY gives a value of
17.6 nF. The closest standard value for CDLY is 18 nF.
When the ADP3198 enters current limit, the internal current
source changes from 15 μA to 3.75 μA. This makes the latch-off
delay time four times longer than the start-up delay time.
Longer latch-off delay times can be achieved by placing a
resistor in parallel with CDLY.
INDUCTOR SELECTION
The choice of inductance for the inductor determines the ripple
current in the inductor. Less inductance leads to more ripple
current, which increases the output ripple voltage and conduction
losses in the MOSFETs. However, using smaller inductors
allows the converter to meet a specified peak-to-peak transient
deviation with less total output capacitance. Conversely, a higher
inductance means lower ripple current and reduced conduction
losses, but more output capacitance is required to meet the
same peak-to-peak transient deviation.
In any multiphase converter, a practical value for the peak-topeak inductor ripple current is less than 50% of the maximum
dc current in the same inductor. Equation 4 shows the
relationship between the inductance, oscillator frequency, and
peak-to-peak ripple current in the inductor.
IR =
VVID × (1 − D )
where TD2 is the desired soft start time, and VBOOT is internally
set to 1.1 V.
Rev. 2 | Page 19 of 31 | www.onsemi.com
f SW × L
(4)
ADP3198
Equation 5 can be used to determine the minimum inductance
based on a given output ripple voltage.
L≥
VVID × R O × (1 − (n × D ))
(5)
f SW × V RIPPLE
Solving Equation 5 for an 8 mV p-p output ripple voltage yields
L≥
1.3 V × 1.0 mΩ × (1 − 0.432 )
330 kHz × 8 mV
= 280 nH
If the resulting ripple voltage is less than what is designed for,
the inductor can be made smaller until the ripple value is met.
This allows optimal transient response and minimum output
decoupling.
The smallest possible inductor should be used to minimize
the number of output capacitors. For this example, choosing a
320 nH inductor is a good starting point and gives a calculated
ripple current of 11 A. The inductor should not saturate at the
peak current of 35.5 A and should be able to handle the sum of
the power dissipation caused by the average current of 30 A in
the winding and core loss.
The best choice for a core geometry is a closed-loop type such
as a potentiometer core (PQ, U, or E core) or toroid. A good
compromise between price and performance is a core with
a toroidal shape.
Many useful magnetics design references are available for
quickly designing a power inductor, such as
•
Intusoft Magnetic Designer Software
•
Designing Magnetic Components for High Frequency
DC to DC Converters, by William T. McLyman,
Kg Magnetics, Inc., ISBN 1883107008
Selecting a Standard Inductor
The following power inductor manufacturers can provide design
consultation and deliver power inductors optimized for high
power applications upon request.
•
Coilcraft®
•
Coiltronics®
•
Sumida Corporation®
CURRENT SENSE AMPLIFIER
Another important factor in the inductor design is the dc
resistance (DCR), which is used for measuring the phase currents.
A large DCR can cause excessive power losses, though too small
a value can lead to increased measurement error. A good rule is
to have the DCR (RL) be about 1 to 1½ times the droop resistance
(RO). This example uses an inductor with a DCR of 1.4 mΩ.
Designing an Inductor
Once the inductance and DCR are known, the next step is to
either design an inductor or to find a standard inductor that
comes as close as possible to meeting the overall design goals.
It is also important to have the inductance and DCR tolerance
specified to control the accuracy of the system. Reasonable
tolerances most manufacturers can meet are 15% inductance
and 7% DCR at room temperature. The first decision in
designing the inductor is choosing the core material. Several
possibilities for providing low core loss at high frequencies
include the powder cores (from Micrometals, Inc., for example,
or Kool Mu® from Magnetics®) and the gapped soft ferrite cores
(for example, 3F3 or 3F4 from Philips). Low frequency
powdered iron cores should be avoided due to their high core
loss, especially when the inductor value is relatively low and the
ripple current is high.
Most designs require the regulator output voltage, measured at
the CPU pins, to drop when the output current increases. The
specified voltage drop corresponds to a dc output resistance (RO),
also referred to as a load line. The ADP3198 has the flexibility of
adjusting RO, independent of current-limit or compensation
components, and it can also support CPUs that do not require
a load line.
For designs requiring a load line, the impedance gain of the
CS amplifier (RCSA) must be to be greater than or equal to the load
line. All designs, whether they have a load line or not, should
keep RCSA ≥ 1 mΩ.
The output current is measured by summing the voltage across
each inductor and passing the signal through a low-pass filter.
This summer filter is the CS amplifier configured with resistors
RPH(X) (summers), and RCS and CCS (filter). The impedance gain
of the regulator is set by the following equations, where RL is the
DCR of the output inductors:
RCSA =
CCS =
R CS
R PH ( x )
× RL
L
R L × RCS
Rev. 2 | Page 20 of 31 | www.onsemi.com
(6)
(7)
ADP3198
The following procedure and equations yield values to use for
RCS1, RCS2, and RTH (the thermistor value at 25°C) for a given
RCS value.
The user has the flexibility to choose either RCS or RPH(X).
However, it is best to select RCS equal to 100 kΩ, and then solve
for RPH(X) by rearranging Equation 6. Here, RCSA = RO = 1 mΩ
because this is equal to the design load line.
RPH ( x ) =
RPH ( x ) =
1.
Select an NTC based on type and value. Because the value
is unknown, use a thermistor with a value close to RCS. The
NTC should also have an initial tolerance of better than 5%.
2.
Based on the type of NTC, find its relative resistance
value at two temperatures. The temperatures that work
well are 50°C and 90°C. These resistance values are called
A (RTH(50°C))/RTH(25°C)) and B (RTH(90°C))/RTH(25°C)). The relative
value of the NTC is always 1 at 25°C.
3.
Find the relative value of RCS required for each of these
temperatures. This is based on the percentage change
needed, which in this example is initially 0.39%/°C. These
temperatures are called r1 (1/(1 + TC × (T1 − 25°C)))
and r2 (1/(1 + TC × (T2 − 25°C))), where TC = 0.0039 for
copper, T1 = 50°C, and T2 = 90°C. From this, r1 = 0.9112 and
r2 = 0.7978.
4.
Compute the relative values for RCS1, RCS2, and RTH using
RL
× RCS
RCSA
1.4 mΩ
1.0 mΩ
× 100 kΩ = 140 kΩ
Next, use Equation 7 to solve for CCS.
CCS =
320 nH
1.4 mΩ × 100 kΩ
= 2.28 nF
It is best to have a dual location for CCS in the layout so that
standard values can be used in parallel to get as close to the
desired value. For best accuracy, CCS should be a 5% or 10%
NPO capacitor. This example uses a 5% combination for CCS
of two 1 nF capacitors in parallel. Recalculating RCS and RPH(X)
using this capacitor combination yields 114 kΩ and 160 kΩ.
The closest standard 1% value for RPH(X) is 158 kΩ.
rCS2 =
INDUCTOR DCR TEMPERATURE CORRECTION
rCS1 =
When the inductor DCR is used as the sense element and
copper wire is used as the source of the DCR, the user needs to
compensate for temperature changes of the inductor’s winding.
Fortunately, copper has a well known temperature coefficient
(TC) of 0.39%/°C.
rTH =
TO
SWITCH
NODES
RTH
RPH1
ADP3198
RCS1
CSCOMP
RPH2
TO
VOUT
SENSE
CCS1
17
CCS2
KEEP THIS PATH
AS SHORT AS POSSIBLE
AND WELL AWAY FROM
SWITCH NODE LINES
CSREF
06094-010
16
Figure 11. Temperature Compensation Circuit Values
1
1
1
−
1 − rCS2 rCS1
(9)
(10)
RTH ( ACTUAL )
RTH (CALCULATED )
(11)
Calculate values for RCS1 and RCS2 using Equation 12 and 13.
RCS1 = RCS × k × rCS1
(12)
RCS2 = RCS × ((1 − k ) + (k × rCS2 ))
(13)
In this example, RCS is calculated to be 114 kΩ. Look for an
available 100 kΩ thermistor, 0603 size. One such thermistor
is the Vishay NTHS0603N01N1003JR NTC thermistor with
A = 0.3602 and B = 0.09174. From these values, rCS1 = 0.3795,
rCS2 = 0.7195, and rTH = 1.075.
RCS2
18
CSSUM
k=
5.
RPH3
(1 − A)
1
A
−
1 − rCS2 r1 − rCS2
Calculate RTH = rTH × RCS, then select the closest value of
thermistor available. Also, compute a scaling factor (k)
based on the ratio of the actual thermistor value used
relative to the computed one.
If RCS is designed to have an opposite and equal percentage
change in resistance to that of the wire, it cancels the temperature variation of the inductor DCR. Due to the nonlinear nature
of NTC thermistors, Resistor RCS1 and Resistor RCS2 are needed.
See Figure 11 to linearize the NTC and produce the desired
temperature tracking.
PLACE AS CLOSE AS POSSIBLE
TO NEAREST INDUCTOR
OR LOW-SIDE MOSFET
( A − B ) × r1 × r2 − A × (1 − B ) × r2 + B × (1 − A ) × r1
(8)
A × (1 − B ) × r1 − B × (1 − A ) × r2 − ( A − B )
Solving for RTH yields 122.55 kΩ, so 100 kΩ is chosen, making
k = 0.816. Next, find RCS1 and RCS2 to be 35.3 kΩ and 87.9 kΩ.
Finally, choose the closest 1% resistor values, which yields a
choice of 35.7 kΩ and 88.7 kΩ.
Rev. 2 | Page 21 of 31 | www.onsemi.com
ADP3198
By combining Equation 16 with Equation 14 and selecting
minimum values for the resistors, the following equations result:
Load Line Setting
For load line values greater than 1 mΩ, RCSA can be set equal
to RO, and the LLSET pin can be directly connected to the
CSCOMP pin. When the load line value needs to be less than
1 mΩ, two additional resistors are required. Figure 12 shows
the placement of these resistors.
ADP3198
CSSUM
CSREF
16
RLL2
OPTIONAL LOAD LINE
SELECT SWITCH
15
Figure 12. Load Line Setting Resistors
RLL 2
× RCSA
RLL1 + RLL 2
(14)
The resistor values for RLL1 and RLL2 are limited by two factors.
The minimum value is based upon the loading of the
CSCOMP pin. This pin’s drive capability is 500 μA and the
majority of this should be allocated to the CSA feedback. If
the current through RLL1 and RLL2 is limited to 10% of this
(50 μA), the following limit can be placed for the
minimum value for RLL1 and RLL2:
RLL1 + RLL 2 ≥
I LIM × RCSA
50 × 10 −6
(15)
Here, ILIM is the current-limit current, which is the
maximum signal level that the CSA responds to.
•
The maximum value is based upon minimizing induced dc
offset errors based on the bias current of the LLSET pin. To
keep the induced dc error less than 1 mV, which makes this
error statistically negligible, place the following limit of the
parallel combination of RLL1 and RLL2:
RLL1 × RLL2
1 × 10 −3
≤
= 8.33 kΩ
RLL1 + RLL2 120 × 10 −9
For this design, RCSA = RO = 1 mΩ. As a result, connect LLSET
directly to CSCOMP; the RLL1 and RLL2 resistors are not needed.
OUTPUT OFFSET
The two resistors RLL1 and RLL2 set up a divider between the
CSCOMP pin and CSREF pin. This resistor divider is input into
the LLSET pin to set the load line slope RO of the VR according
to the following equation:
•
(18)
Another useful feature for some VR applications is the ability to
select different load lines. Figure 12 shows an optional
MOSFET switch that allows this feature. Here, design for
RCSA = RO(MAX) (selected with QLL on) and then use Equation 14
to set RO = RO(MIN) (selected with QLL off).
17
QLL
RO =
⎛R
⎞
RLL1 = ⎜⎜ CSA − 1⎟⎟ × RLL 2
R
⎝ O
⎠
18
RLL1
LLSET
(17)
Therefore, both RLL1 and RLL2 need to be in parallel and less than
8.33 kΩ.
06094-011
CSCOMP
I LIM × RO
50 μA
R LL 2 =
(16)
It is best to select the resistor values to minimize their values to
reduce the noise and parasitic susceptibility of the feedback path.
The Intel specification requires that at no load the nominal output
voltage of the regulator be offset to a value lower than the
nominal voltage corresponding to the VID code. The offset is
set by a constant current source flowing out of the FB pin (IFB) and
flowing through RB. The value of RB can be found using
Equation 19.
RB =
RB =
VVID − VONL
I FB
1.3 V − 1.285 V
15 μA
= 1.00 kΩ
(19)
The closest standard 1% resistor value is 1.00 kΩ.
COUT SELECTION
The required output decoupling for the regulator is typically
recommended by Intel for various processors and platforms.
Use some simple design guidelines to determine the requirements. These guidelines are based on having both bulk
capacitors and ceramic capacitors in the system.
First, select the total amount of ceramic capacitance. This is
based on the number and type of capacitor to be used. The best
location for ceramic capacitors is inside the socket with 12 to
18, 1206 size being the physical limit. Other capacitors can be
placed along the outer edge of the socket as well.
To determine the minimum amount of ceramic capacitance
required, start with a worst-case load step occurring right after
a switching cycle has stopped. The ceramic capacitance then
delivers the charge to the load while the load is ramping up and
until the VR has responded with the next switching cycle.
Rev. 2 | Page 22 of 31 | www.onsemi.com
ADP3198
Equation 20 gives the designer a rough approximation for
determining the minimum ceramic capacitance. Due to the
complexity of the PCB parasitics and bulk capacitors, the actual
amount of ceramic capacitance required can vary.
C Z ( MIN ) ≥
1 ⎡ 1 ⎛1
⎞ Δ IO ⎤
×⎢
× ⎜ − D⎟ −
⎥
RO ⎣ f SW ⎝ n
⎠ 2 SR ⎦
(20)
The typical ceramic capacitors consist of multiple 10 μF or
22 μF capacitors. For this example, Equation 20 yields 180.8 μF,
so eighteen, 10 μF ceramic capacitors suffice.
Next, there is an upper limit imposed on the total amount of
bulk capacitance (CX) when the user considers the VID on-thefly voltage stepping of the output (voltage step VV in time tV
with error of VERR).
A lower limit is based on meeting the capacitance for load
release for a given maximum load step (ΔIO) and a maximum
allowable overshoot. The total amount of load release voltage
is given as ΔVO = ΔIO × RO + ΔVrl, where ΔVrl is the maximum
allowable overshoot voltage.
(21)
C X ( MAX ) ≤
(22)
⎛V
where K = −1n ⎜⎜ ERR
⎝ VV
⎛
⎞
⎜
⎟
⎜
⎟
320 nH × 100 A
C X ( MIN ) ≤ ⎜
− 180 μF ⎟ = 3.92 mF
⎛
50 mV ⎞
⎜
⎟
⎟ × 1.3 V
⎜⎜ 4 × ⎜⎜ 1.0 mΩ +
⎟⎟
⎟
100 A ⎠
⎝
⎝
⎠
C X ( MAX ) ≤
320 nH × 450 mV
4 × 5.22 × (1.0 mΩ )2 × 1.3 V
×
2
⎛
⎞
⎛ 230 μs × 1.3 V × 4 × 5.2 × 1.0 mΩ ⎞
⎜
⎟
⎜
⎟
− 1⎟ − 180 μF = 43.0 mF
⎜ 1+ ⎜
⎟
×
450
mV
320
nH
⎜
⎟
⎝
⎠
⎝
⎠
where K = 5.2.
⎛
⎞
⎜
⎟
⎜
⎟
L × Δ IO
C X ( MIN ) ≥ ⎜
− CZ ⎟
⎜ n × ⎛⎜ R + ΔVrl ⎞⎟ × V
⎟
⎜ O ΔI ⎟ VID
⎜
⎟
O ⎠
⎝
⎝
⎠
⎛
⎛ V
VV ⎜
nKRO
L
×
×
1 + ⎜⎜ tV VID ×
⎜
2 2
L
nK RO VVID ⎜
⎝ VV
⎝
This example uses 18, 10 μF 1206 MLC capacitors (CZ = 180 μF).
The VID on-the-fly step change is 450 mV in 230 μs with a
settling error of 2.5 mV. The maximum allowable load release
overshoot for this example is 50 mV, therefore, solving for the
bulk capacitance yields
2
⎞
⎞
⎟
⎟ − 1⎟ − C Z
⎟
⎟
⎠
⎠
⎞
⎟.
⎟
⎠
Using 10, 560 μF Al-Poly capacitors with a typical ESR of 6 mΩ
each yields CX = 5.6 mF with an RX = 0.6 mΩ.
One last check should be made to ensure that the ESL of the
bulk capacitors (LX) is low enough to limit the high frequency
ringing during a load change.
This is tested using
LX ≤ CZ × RO 2 × Q 2
L X ≤ 180 μF × (1 mΩ )2 ×
4
= 240 pH
3
(23)
where Q2 is limited to 4/3 to ensure a critically damped system.
To meet the conditions of these equations and transient
response, the ESR of the bulk capacitor bank (RX) should be less
than two times the droop resistance (RO). If the CX(MIN) is larger
than CX(MAX), the system cannot meet the VID on-the-fly specification and can require the use of a smaller inductor or more
phases (and may have to increase the switching frequency to
keep the output ripple the same).
In this example, LX is approximately 240 pH for the 10, Al-Poly
capacitors, which satisfies this limitation. If the LX of the chosen
bulk capacitor bank is too large, the number of ceramic
capacitors needs to be increased, or lower ESL bulks need to be
used if there is excessive undershoot during a load transient.
For this multimode control technique, all ceramic designs can
be used providing the conditions of Equation 20 through
Equation 23 are satisfied.
Rev. 2 | Page 23 of 31 | www.onsemi.com
ADP3198
POWER MOSFETS
For this example, the N-channel power MOSFETs have been
selected for one high-side switch and two low-side switches per
phase. The main selection parameters for the power MOSFETs
are VGS(TH), QG, CISS, CRSS, and RDS(ON). The minimum gate drive
voltage (the supply voltage to the ADP3110A) dictates whether
standard threshold or logic-level threshold MOSFETs must be
used. With VGATE ~10 V, logic-level threshold MOSFETs
(VGS(TH) < 2.5 V) are recommended.
The maximum output current (IO) determines the RDS(ON)
requirement for the low-side (synchronous) MOSFETs. With
the ADP3198, currents are balanced between phases, thus, the
current in each low-side MOSFET is the output current divided
by the total number of MOSFETs (nSF). With conduction losses
being dominant, Equation 24 shows the total power that is
dissipated in each synchronous MOSFET in terms of the ripple
current per phase (IR) and average total output current (IO):
PSF
⎡⎛ I
= (1 − D ) × ⎢⎜⎜ O
⎢⎣⎝ n SF
2
⎞
1 ⎛ n IR
⎟ +
×⎜
⎟
12 ⎜⎝ n SF
⎠
⎞
⎟
⎟
⎠
2
⎤
⎥ × R DS ( SF )
⎥⎦
(24)
Knowing the maximum output current being designed for and
the maximum allowed power dissipation, the user can find the
required RDS(ON) for the MOSFET. For D-PAK MOSFETs up to
an ambient temperature of 50°C, a safe limit for PSF is 1 W to
1.5 W at 120°C junction temperature. Thus, for this example
(119 A maximum), RDS(SF) (per MOSFET) < 7.5 mΩ. This RDS(SF)
is also at a junction temperature of about 120°C. As a result,
users need to account for this when making this selection. This
example uses two lower-side MOSFETs at 4.8 mΩ, each at 120°C.
value for the switching loss per main MOSFET, where nMF is the
total number of main MOSFETs.
V ×I
n
PS ( MF ) = 2 × f SW × CC O × RG × MF × C ISS
(25)
n MF
n
where RG is the total gate resistance (2 Ω for the ADP3110A and
about 1 Ω for typical high speed switching MOSFETs, making
RG = 3 Ω), and CISS is the input capacitance of the main MOSFET.
Adding more main MOSFETs (nMF) does not help the switching
loss per MOSFET because the additional gate capacitance slows
switching. Use lower gate capacitance devices to reduce
switching loss.
The conduction loss of the main MOSFET is given by the
following, where RDS(MF) is the on resistance of the MOSFET:
PC ( MF )
⎡⎛ I
= D × ⎢⎜⎜ O
⎢⎣⎝ n MF
2
⎞
1 ⎛ n × IR
⎟ + ×⎜
⎜ n
⎟
12
⎝ MF
⎠
⎞
⎟
⎟
⎠
2
⎤
⎥ × R DS ( MF )
⎥⎦
(26)
Typically, for main MOSFETs, the highest speed (low CISS)
device is preferred, but these usually have higher on resistance.
Select a device that meets the total power dissipation (about
1.5 W for a single D-PAK) when combining the switching and
conduction losses.
For this example, an NTD40N03L is selected as the main MOSFET
(eight total; nMF = 8), with CISS = 584 pF (maximum) and
RDS(MF) = 19 mΩ (maximum at TJ = 120°C). An NTD110N02L
is selected as the synchronous MOSFET (eight total; nSF = 8),
with CISS = 2710 pF (maximum) and RDS(SF) = 4.8 mΩ
(maximum at TJ = 120°C). The synchronous MOSFET CISS is
less than 3000 pF, satisfying this requirement.
Another important factor for the synchronous MOSFET is the
input capacitance and feedback capacitance. The ratio of the
feedback to input needs to be small (less than 10% is recommended) to prevent accidental turn-on of the synchronous
MOSFETs when the switch node goes high.
Solving for the power dissipation per MOSFET at IO = 119 A and
IR = 11 A yields 958 mW for each synchronous MOSFET and
872 mW for each main MOSFET. A guideline to follow is to limit
the MOSFET power dissipation to 1 W. The values calculated in
Equation 25 and Equation 26 comply with this guideline.
Also, the time to switch the synchronous MOSFETs off should
not exceed the nonoverlap dead time of the MOSFET driver
(40 ns typical for the ADP3110A). The output impedance of
the driver is approximately 2 Ω, and the typical MOSFET input
gate resistances are about 1 Ω to 2 Ω. Therefore, a total gate
capacitance of less than 6000 pF should be adhered to. Because
two MOSFETs are in parallel, the input capacitance for each
synchronous MOSFET should be limited to 3000 pF.
Finally, consider the power dissipation in the driver for each
phase. This is best expressed as QG for the MOSFETs and is
given by Equation 27, where QGMF is the total gate charge for
each main MOSFET and QGSF is the total gate charge for each
synchronous MOSFET.
The high-side (main) MOSFET has to be able to handle two
main power dissipation components: conduction and switching
losses. The switching loss is related to the amount of time it
takes for the main MOSFET to turn on and off, and to the
current and voltage that are being switched. Basing the switching
speed on the rise and fall time of the gate driver impedance and
MOSFET input capacitance, Equation 25 provides an approximate
⎡f
⎤
PDRV = ⎢ SW × (n MF × QGMF + nSF × Q GSF ) + I CC ⎥ × VCC (27)
⎢⎣ 2 × n
⎥⎦
Also shown is the standby dissipation factor (ICC × VCC) of the
driver. For the ADP3110A, the maximum dissipation should be less
than 400 mW. In this example, with ICC = 7 mA, QGMF = 5.8 nC,
and QGSF = 48 nC, there is 297 mW in each driver, which is
below the 400 mW dissipation limit. See the ADP3110A data
sheet for more details.
Rev. 2 | Page 24 of 31 | www.onsemi.com
ADP3198
RAMP RESISTOR SELECTION
CURRENT-LIMIT SETPOINT
The ramp resistor (RR) is used for setting the size of the internal
PWM ramp. The value of this resistor is chosen to provide the best
combination of thermal balance, stability, and transient response.
Equation 28 is used for determining the optimum value.
To select the current-limit setpoint, first find the resistor value
for RLIM. The current-limit threshold for the ADP3198 is set
with a constant current source flowing out of the ILIMIT pin,
which sets up a voltage (VLIM) across RLIM with a gain of
82.6 mV/V (ALIM). Thus, increasing RLIM now increases the
current limit. RLIM can be found using
VCL
I × RCSA
R LIM =
= LIM
× R REF
(31)
A LIM × I ILIMIT
82.6 mV
AR × L
RR =
3 × A D × R DS × C R
(28)
0.2 × 320 nH
RR =
3 × 5 × 2.4 mΩ × 5 pF
= 356 kΩ
Here, ILIM is the peak average current limit for the supply output.
The peak average current is the dc current limit plus the output
ripple current. In this example, choosing a dc current limit of
159 A and having a ripple current of 11 A gives an ILIM of 170 A.
This results in an RLIM = 205.8 kΩ, for which 205 kΩ is chosen
as the nearest 1% value.
where:
AR is the internal ramp amplifier gain.
AD is the current balancing amplifier gain.
RDS is the total low-side MOSFET on resistance.
CR is the internal ramp capacitor value.
The internal ramp voltage magnitude can be calculated by using
A R × (1 − D ) × VVID
VR =
R R × C R × f SW
(29)
VR =
0.2 × (1 − 0.108 ) × 1.3 V
357 kΩ × 5 pF × 330 kHz
= 394 mV
The size of the internal ramp can be made larger or smaller.
If it is made larger, stability and noise rejection improves, but
transient degrades. Likewise, if the ramp is made smaller,
transient response improves at the sacrifice of noise rejection
and stability.
The factor of 3 in the denominator of Equation 28 sets a ramp
size that gives an optimal balance for good stability, transient
response, and thermal balance.
COMP PIN RAMP
A ramp signal on the COMP pin is due to the droop voltage
and output voltage ramps. This ramp amplitude adds to the
internal ramp to produce the following overall ramp signal
at the PWM input:
VRT =
VR
⎛
2 × (1 − n × D )
⎜1 −
⎜ n× f ×C × R
X
SW
O
⎝
⎞
⎟
⎟
⎠
The per-phase initial duty cycle limit and peak current during a
load step are determined by
VCOMP ( MAX ) − V BIAS
(32)
D MAX = D ×
V RT
I PHMAX ≅
D MAX (VIN − VVID )
×
f SW
L
(33)
For the ADP3198, the maximum COMP voltage (VCOMP(MAX))
is 4.0 V and the COMP pin bias voltage (VBIAS) is 1.1 V. In this
example, the maximum duty cycle is 0.61 and the peak current
is 62 A.
The limit of the peak per-phase current described earlier during
the secondary current limit is determined by
VCOMP (CLAMPED ) − V BIAS
(34)
I PHLIM ≅
A D × R DS ( MAX )
For the ADP3198, the current balancing amplifier gain (AD) is 5
and the clamped COMP pin voltage is 2 V. Using an RDS(MAX) of
2.8 mΩ (low-side on resistance at 150°C) results in a per-phase
peak current limit of 64 A. This current level can be reached only
with an absolute short at the output, and the current-limit latch-off
function shuts down the regulator before overheating can occur.
FEEDBACK LOOP COMPENSATION DESIGN
(30)
In this example, the overall ramp signal is 0.46 V. However,
if the ramp size is smaller than 0.5 V, increase the ramp size
to be at least 0.5 V by decreasing the ramp resistor for noise
immunity. Because there is only 0.46 V initially, a ramp resistor
value of 332 kΩ is chosen for this example, yielding an overall
ramp of 0.51 V.
Optimized compensation of the ADP3198 allows the best
possible response of the regulator output to a load change. The
basis for determining the optimum compensation is to make the
regulator and output decoupling appear as an output impedance
that is entirely resistive over the widest possible frequency
range, including dc, and equal to the droop resistance (RO).
With the resistive output impedance, the output voltage droops
in proportion to the load current at any load current slew rate.
This ensures optimal positioning and minimizes the output
decoupling.
Rev. 2 | Page 25 of 31 | www.onsemi.com
ADP3198
Because of the multimode feedback structure of the ADP3198,
the feedback compensation must be set to make the converter
output impedance work in parallel with the output decoupling
to make the load look entirely resistive. Compensation is
needed for several poles and zeros created by the output
inductor and the decoupling capacitors (output filter).
Tuning the ADP3198 section).
A type-three compensator on the voltage feedback is adequate
for proper compensation of the output filter. Equation 35 to
Equation 39 are intended to yield an optimal starting point for
the design; some adjustments may be necessary to account for
PCB and component parasitic effects (see the
First, compute the time constants for all the poles and zeros in the system using Equation 35 to Equation 39.
R E = n × RO + A D × R DS +
R L × V RT
VVID
R E = 4 × 1 mΩ + 5 × 2.4 mΩ +
TA = C X × (RO − R ' ) +
+
2 × L × (1 − n × D ) × V RT
n × C X × R O × VVID
1.4 mΩ × 0.51 V
1.3 V
+
2 × 320 nH × (1 − 0.432 ) × 0.51 V
4 × 5.6 mF × 1 mΩ × 1.3 V
= 22.9 mΩ
240 pH 1 mΩ − 0.5 mΩ
L X RO − R '
×
= 5.6 mF × (1 mΩ − 0.5 mΩ ) +
×
= 3.00 μs
1 mΩ
0.6 mΩ
RO
RX
TB = (R X + R ' − R O ) × C X = (0.6 mΩ + 0.5 mΩ − 1 mΩ ) × 5.6 mF = 560 ns
⎛
A × RDS
VRT × ⎜ L − D
⎜
2 × f SW
⎝
TC =
VVID × RE
TD =
C X × (RO − R ' ) + C Z × RO
=
5.6 mF × 180 μF × (1 mΩ )2
(36)
(37)
⎞
⎛
5 × 2.4 mΩ ⎞
⎟ 0.51 V × ⎜ 320 nH −
⎟
⎟
⎜
2 × 330 kHz ⎟⎠
⎠=
⎝
= 5.17 μs
1.3 V × 22.9 mΩ
C X × C Z × RO2
(35)
5.6 mF × (1 mΩ − 0.5 mΩ ) + 180 μF × 1 mΩ
(38)
= 338 ns
(39)
where:
R' is the PCB resistance from the bulk capacitors to the ceramics.
RDS is the total low-side MOSFET on resistance per phase.
In this example, AD is 5, VRT equals 0.51 V, R' is approximately 0.5 mΩ (assuming a 4-layer, 1 ounce motherboard), and LX is 240 pH for
the 10 Al-Poly capacitors.
The compensation values can then be solved using
n × RO × TA 4 × 1 mΩ × 3.00 μs
CA =
=
= 524 pF
22.9 mΩ × 1.00 kΩ
R E × RB
C FB =
(40)
RA =
TC 5.17 μs
=
= 9.87 kΩ
C A 524 pF
(41)
CB =
TB
560 ns
=
= 560 pF
R B 1.00 kΩ
(42)
TD
338 ns
=
= 34.2 pF
R A 9.87 kΩ
(43)
These are the starting values prior to tuning the design that
account for layout and other parasitic effects (see the Tuning
the ADP3198 section). The final values selected after tuning are
CA = 560 pF
RA = 10.0 kΩ
CB = 560 pF
CFB = 27 pF
Rev. 2 | Page 26 of 31 | www.onsemi.com
ADP3198
The capacitor manufacturer’s ripple-current ratings are often
based on only 2000 hours of life. As a result, it advisable to
further derate the capacitor or to choose a capacitor rated at a
higher temperature than required. Several capacitors can be
placed in parallel to meet size or height requirements in the
design. In this example, the input capacitor bank is formed by
two 2700 μF, 16 V aluminum electrolytic capacitors and eight
4.7 μF ceramic capacitors.
Figure 13 and Figure 14 show the typical transient response
using these compensation values.
1
M 10μs
A CH1
–36mV
06094-012
CH1 50mV
To reduce the input current di/dt to a level below the recommended maximum of 0.1 A/μs, an additional small inductor
(L > 370 nH at 18 A) should be inserted between the converter
and the supply bus. This inductor also acts as a filter between
the converter and the primary power source.
THERMAL MONITOR DESIGN
Figure 13. Typical Transient Response for Design Example Load Step
A thermistor is used on the TTSENSE input of the ADP3198
for monitoring the temperature of the VR. A constant current
of 123 μA is sourced out of this pin and runs through a
thermistor network such as the one shown in Figure 15.
ADP3198
1
OPTIONAL
TEMPERATURE
ADJUST RESISTOR
A CH1
–36mV
Figure 14. Typical Transient Response for Design Example Load Release
CIN SELECTION AND INPUT CURRENT
di/dt REDUCTION
In continuous inductor current mode, the source current of the
high-side MOSFET is approximately a square wave with a duty
ratio equal to n × VOUT/VIN and an amplitude of one-nth the
maximum output current. To prevent large voltage transients,
a low ESR input capacitor, sized for the maximum rms current,
must be used. The maximum rms capacitor current is given by
I CRMS = D × I O ×
1
−1
N×D
(44)
I CRMS = 0.108 × 119 A ×
1
− 1 = 14.7 A
4 × 0.108
VRFAN
9
VRHOT
10
TTSENSE
0.1μF
RTTSENSE
06094-014
M 10μs
06094-013
CH1 50mV
PLACE
THERMISTOR
NEAR CLOSEST
PHASE
8
Figure 15. VR Thermal Monitor Circuit
A voltage is generated from this current through the thermistor
and sensed inside the IC. When the voltage reaches 1.105 V,
the VRFAN output gets set. When the voltage reaches 0.81 V,
the VRHOT gets set. This corresponds to RTTSENSE values of
8.98 kΩ for VRFAN and 6.58 kΩ for VRHOT.
These values correspond to a thermistor temperature of ~100°C
and ~110°C when using the same type of 100 kΩ NTC thermistor
used in the current sense amplifier.
An additional fixed resistor in parallel with the thermistor allows
tuning of the trip point temperatures to match the hottest temperature in the VR, when the thermistor itself is directly sensing
a proportionately lower temperature. Setting this resistor value
is best accomplished with a variable resistor during thermal
validation and then fixing this value for the final design.
Additionally, a 0.1 μF capacitor should be used for filtering noise.
Rev. 2 | Page 27 of 31 | www.onsemi.com
ADP3198
SHUNT RESISTOR DESIGN
TUNING THE ADP3198
The ADP3198 uses a shunt to generate 5 V from the 12 V
supply range. A trade-off can be made between the power
dissipated in the shunt resistor and the UVLO threshold.
Figure 16 shows the typical resistor value needed to realize
certain UVLO voltages. It also gives the maximum power
dissipated in the shunt resistor for these UVLO voltages.
1.
Build a circuit based on the compensation values
computed from the design spreadsheet.
2.
Hook up the dc load to the circuit, turn it on, and verify its
operation. Also, check for jitter at no load and full load.
550
0.50
500
0.45
400
RSHUNT
0.35
350
0.30
300
0.25
250
0.20
200
0.15
150
7.0
7.5
8.0
8.5
9.0
9.5
10.0
10.5
3.
Measure the output voltage at no load (VNL). Verify that it
is within tolerance.
4.
Measure the output voltage at full load cold (VFLCOLD). Let
the board sit for ~10 minutes at full load, and then measure
the output (VFLHOT). If there is a change of more than a few mV,
adjust RCS1 and RCS2 using Equation 46 and Equation 48.
PSHUNT (W)
0.40
PSHUNT
0.10
11.0
VIN (UVLO)
RCS2 ( NEW ) = R CS2 (OLD ) ×
06094-019
RSHUNT (Ω)
450
DC Load Line Setting
PMAX
(V
=
IN ( MAX ) − VCC ( MIN )
)
Repeat Step 4 until the cold and hot voltage measurements
remain the same.
6.
Measure the output voltage from no load to full load using
5 A steps. Compute the load line slope for each change, and
then average to get the overall load line slope (ROMEAS).
7.
If ROMEAS is off from RO by more than 0.05 mΩ, use
Equation 47 to adjust the RPH values.
2
(45)
R SHUNT
R PH ( NEW ) = R PH (OLD ) ×
where:
VIN(MAX) is the maximum voltage from the 12 V input supply (if
the 12 V input supply is 12 V ± 5%, VIN(MAX) = 12.6 V; if the 12 V
input supply is 12 V ± 10%, VIN(MAX) = 13.2 V).
VCC(MIN) is the minimum VCC voltage of the ADP3198. This is
specified as 4.75 V.
RSHUNT is the shunt resistor value.
ROMEAS
RCS1( NEW ) =
8.
Repeat Step 6 and Step 7 to check the load line. Repeat
adjustments if necessary.
9.
When the dc load line adjustment is complete, do not
change RPH, RCS1, RCS2, or RTH for the remainder of the
procedure.
(
1
RCS1(OLD ) + RTH (25° C )
) (
RCS1(OLD ) × RTH (25° C ) + RCS1(OLD ) − RCS2 ( NEW ) × RCS1(OLD ) − RTH (25° C )
11. Remove the dc load from the circuit and hook up the
dynamic load.
12. Hook up the scope to the output voltage and set it to dc
coupling with the time scale at 100 μs/div.
(47)
RO
10. Measure the output ripple at no load and full load with
a scope, and make sure it is within specifications.
The CECC standard specification for power rating in surface
mount resistors is: 0603 = 0.1 W, 0805 = 0.125 W, 1206 = 0.25 W.
AC Load Line Setting
(46)
5.
Figure 16. Typical Shunt Resistor Value and Power Dissipation
for Different UVLO Voltage
The maximum power dissipated is calculated using Equation 45.
V NL − VFLCOLD
V NL − VFLHOT
)− R
1
(48)
TH ( 25° C )
13. Set the dynamic load for a transient step of about 40 A at
1 kHz with 50% duty cycle.
14. Measure the output waveform (use dc offset on scope to see
the waveform). Try to use a vertical scale of 100 mV/div or
finer. This waveform should look similar to Figure 17.
Rev. 2 | Page 28 of 31 | www.onsemi.com
ADP3198
19. If both overshoots are larger than desired, try making
the adjustments using the following suggestions:
VACDRP
VDCDRP
Make the ramp resistor larger by 25% (RRAMP)
•
For VTRAN1, increase CB or increase the switching
frequency
•
For VTRAN2, increase RA and decrease CA by 25%
06094-015
If these adjustments do not change the response, the design
is limited by the output decoupling. Check the output
response every time a change is made, and check the switching nodes to ensure that the response is still stable.
Figure 17. AC Load Line Waveform
15. Use the horizontal cursors to measure VACDRP and VDCDRP as
shown in Figure 17. Do not measure the undershoot or
overshoot that happens immediately after this step.
16. If VACDRP and VDCDRP are different by more than a few
millivolts, use Equation 49 to adjust CCS. Users may need to
parallel different values to get the right one because limited
standard capacitor values are available. It is a good idea to
have locations for two capacitors in the layout for this.
C CS ( NEW ) = C CS (OLD ) ×
•
20. For load release (see Figure 19), if VTRANREL is larger
than the allowed overshoot, there is not enough output
capacitance. Either more capacitance is needed, or the
inductor values need to be made smaller. When changing
inductors, start the design again using a spreadsheet and
this tuning procedure.
VTRANREL
V ACDRP
(49)
V DCDRP
VDROOP
06094-017
17. Repeat Step 11 to Step 13 and repeat the adjustments, if
necessary. Once complete, do not change CCS for the
remainder of the procedure. Set the dynamic load step to
maximum step size. Do not use a step size larger than
needed. Verify that the output waveform is square, which
means that VACDRP and VDCDRP are equal.
Figure 19. Transient Setting Waveform
Initial Transient Setting
18. With the dynamic load still set at the maximum step size,
expand the scope time scale to either 2 μs/div or 5 μs/div.
The waveform can have two overshoots and one minor
undershoot (see Figure 18). Here, VDROOP is the final
desired value.
Because the ADP3198 turns off all of the phases (switches
inductors to ground), no ripple voltage is present during load
release. Therefore, the user does not have to add headroom for
ripple. This allows load release VTRANREL to be larger than VTRAN1
by the amount of ripple, and still meet specifications.
If VTRAN1 and VTRANREL are less than the desired final droop, this
implies that capacitors can be removed. When removing capacitors, also check the output ripple voltage to make sure it is still
within specifications.
VDROOP
LAYOUT AND COMPONENT PLACEMENT
The following guidelines are recommended for optimal
performance of a switching regulator in a PC system.
VTRAN2
06094-016
VTRAN1
Figure 18. Transient Setting Waveform
Rev. 2 | Page 29 of 31 | www.onsemi.com
ADP3198
General Recommendations
For good results, a PCB with at least four layers is recommended.
This provides the needed versatility for control circuitry
interconnections with optimal placement, power planes for
ground, input and output power, and wide interconnection
traces in the remainder of the power delivery current paths.
Keep in mind that each square unit of 1 ounce copper trace
has a resistance of ~0.53 mΩ at room temperature.
Whenever high currents must be routed between PCB layers,
use vias liberally to create several parallel current paths, so the
resistance and inductance introduced by these current paths is
minimized and the via current rating is not exceeded.
If critical signal lines (including the output voltage sense lines of
the ADP3198) must cross through power circuitry, it is best to
interpose a signal ground plane between those signal lines and
the traces of the power circuitry. This serves as a shield to
minimize noise injection into the signals at the expense of
making signal ground a bit noisier.
An analog ground plane should be used around and under the
ADP3198 as a reference for the components associated with the
controller. This plane should be tied to the nearest output
decoupling capacitor ground and should not be tied to any other
power circuitry to prevent power currents from flowing into it.
The components around the ADP3198 should be located close
to the controller with short traces. The most important traces to
keep short and away from other traces are the FB pin and CSSUM
pin. The output capacitors should be connected as close as
possible to the load (or connector), for example, a microprocessor core, that receives the power. If the load is distributed, the
capacitors should also be distributed and generally be in
proportion to where the load tends to be more dynamic.
Avoid crossing any signal lines over the switching power path loop
(described in the Power Circuitry Recommendations section).
Power Circuitry Recommendations
The switching power path should be routed on the PCB to
encompass the shortest possible length to minimize radiated
switching noise energy (EMI) and conduction losses in the
board. Failure to take proper precautions often results in EMI
problems for the entire PC system and noise-related operational
problems in the power converter control circuitry. The
switching power path is the loop formed by the current path
through the input capacitors and the power MOSFETs, including
all interconnecting PCB traces and planes. Using short and wide
interconnection traces is especially critical in this path for two
reasons: it minimizes the inductance in the switching loop,
which can cause high energy ringing; and it accommodates the
high current demand with minimal voltage loss.
When a power dissipating component, for example, a power
MOSFET, is soldered to a PCB, it is recommended to liberally
use the vias, both directly on the mounting pad and immediately
surrounding it. Two important reasons for this are improved
current rating through the vias and improved thermal performance from vias extended to the opposite side of the PCB, where a
plane can more readily transfer the heat to the air. Make a
mirror image of any pad being used to heatsink the MOSFETs
on the opposite side of the PCB to achieve the best thermal
dissipation in the air around the board. To further improve
thermal performance, use the largest possible pad area.
The output power path should also be routed to encompass a
short distance. The output power path is formed by the current
path through the inductor, the output capacitors, and the load.
For best EMI containment, a solid power ground plane should
be used as one of the inner layers extending fully under all the
power components.
Signal Circuitry Recommendations
The output voltage is sensed and regulated between the FB pin
and the FBRTN pin, which connect to the signal ground at the
load. To avoid differential mode noise pickup in the sensed
signal, the loop area should be small. Thus, the FB trace and
FBRTN trace should be routed adjacent to each other on top
of the power ground plane back to the controller.
The feedback traces from the switch nodes should be connected
as close as possible to the inductor. The CSREF signal should be
connected to the output voltage at the nearest inductor to the
controller.
Rev. 2 | Page 30 of 31 | www.onsemi.com
ADP3198
OUTLINE DIMENSIONS
6.00
BSC SQ
0.60 MAX
0.60 MAX
PIN 1
INDICATOR
TOP
VIEW
0.50
BSC
5.75
BCS SQ
0.50
0.40
0.30
12° MAX
1.00
0.85
0.80
PIN 1
INDICATOR
31
30
40
1
4.25
4.10 SQ
3.95
EXPOSED
PAD
(BOTTOM VIEW)
10
11
21
20
0.25 MIN
4.50
REF
0.80 MAX
0.65 TYP
0.05 MAX
0.02 NOM
0.30
0.23
0.18
SEATING
PLANE
0.20 REF
COPLANARITY
0.08
COMPLIANT TO JEDEC STANDARDS MO-220-VJJD-2
Figure 20. 40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
6 mm × 6 mm Body, Very Thin Quad
(CP-40)
Dimensions shown in millimeters
ORDERING GUIDE
Model
ADP3198JCPZ-RL1
1
Temperature Range
0°C to 85°C
Package Description
40-Lead Lead Frame Chip Scale Package [LFCSP_VQ]
Package Option
CP-40
Ordering Quantity
2,500
Z = Pb-free part.
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ON Semiconductor and
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