6-Bit, Programmable 2-, 3-, 4-Phase Synchronous Buck Controller ADP3168 FEATURES FUNCTIONAL BLOCK DIAGRAM Selectable 2-, 3-, or 4-phase operation at up to 1 MHz per phase ±10 mV worst-case differential sensing error over temperature Logic-level PWM outputs for interface to external high power drivers Active current balancing between all output phases Built-in Power Good/crowbar blanking supports On-the-fly VID code changes 6-bit digitally programmable 0.8375 V to 1.6 V output Programmable short-circuit protection with programmable latch-off delay VCC RAMPADJ RT 28 14 13 ADP3168 11 EN UVLO SHUTDOWN AND BIAS OSCILLATOR SET CMP 19 GND EN RESET 27 PWM1 DAC +150mV CMP RESET CURRENTBALANCING CIRCUIT CSREF 26 PWM2 2-, 3-, 4-PHASE DRIVER LOGIC CMP RESET 25 PWM3 DAC –250mV 24 CMP RESET PWM4 APPLICATIONS Desktop PC power supplies for: Next generation Intel® processors VRM modules DELAY 10 CROWBAR PWRGD CURRENT LIMIT 23 SW1 22 SW2 21 15 SW3 ILIMIT The ADP3168 is a highly efficient, multiphase, synchronous buck switching regulator controller optimized for converting a 12 V main supply into the core supply voltage required by high performance Intel processors. It uses an internal 6-bit DAC to read a voltage identification (VID) code directly from the processor, which is used to set the output voltage between 0.8375 V and 1.6 V, and uses a multimode PWM architecture to drive the logic-level outputs at a programmable switching frequency that can be optimized for VR size and efficiency. The phase relationship of the output signals can be programmed to provide 2-, 3-, or 4-phase operation, allowing for the construction of up to four complementary buck switching stages. The ADP3168 also includes programmable no-load offset and slope functions to adjust the output voltage as a function of the load current so that it is always optimally positioned for a system transient. The ADP3168 also provides accurate and reliable short-circuit protection, adjustable current limiting, and a delayed Power Good output that accommodates on-the-fly output voltage changes requested by the CPU. 20 EN SW4 17 CURRENTLIMIT CIRCUIT CSSUM 16 CSREF 12 DELAY 18 CSCOMP SOFTSTART 8 9 FB COMP PRECISION REFERENCE VID DAC 7 1 2 3 4 5 6 FBRTN VID4 VID3 VID2 VID1 VID0 VID5 Figure 1. The device is specified over the commercial temperature range of 0°C to 85°C and is available in a 28-lead TSSOP package. Rev. B Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners. 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All rights reserved. 03258-001 GENERAL DESCRIPTION ADP3168 TABLE OF CONTENTS Specifications..................................................................................... 3 Soft Start and Current Limit Latch-Off Delay Times............ 13 Absolute Maximum Ratings............................................................ 5 Inductor Selection ...................................................................... 13 ESD Caution.................................................................................. 5 Designing an Inductor............................................................... 15 Pin Configuration and Function Descriptions............................. 6 Selecting a Standard Inductor................................................... 15 Typical Performance Characteristics and Test Circuits............... 7 Output Droop Resistance.......................................................... 15 Theory of Operation ........................................................................ 9 Inductor DCR Temperature Correction ................................. 16 Number of Phases......................................................................... 9 Output Offset .............................................................................. 16 Master Clock Frequency.............................................................. 9 COUT Selection ............................................................................. 17 Output Voltage Differential Sensing .......................................... 9 Power MOSFETs......................................................................... 18 Output Current Sensing .............................................................. 9 Ramp Resistor Selection............................................................ 19 Active Impedance Control Mode............................................. 10 COMP Pin Ramp ....................................................................... 19 Current-Control Mode and Thermal Balance........................ 10 Current-Limit Set Point............................................................. 19 Voltage Control Mode................................................................ 10 Feedback Loop Compensation Design.................................... 19 Soft Start ...................................................................................... 10 CIN Selection and Input Current di/dt Reduction.................. 21 Current-Limit, Short-Circuit, and Latch-Off Protection...... 11 Tuning Procedure for the ADP3168 ........................................ 21 Dynamic VID.............................................................................. 12 Layout and Component Placement.............................................. 23 Power-Good Monitoring........................................................... 12 General Recommendations....................................................... 23 Output Crowbar ......................................................................... 12 Power Circuitry .......................................................................... 23 Output Enable and UVLO ........................................................ 12 Signal Circuitry........................................................................... 23 Application Information................................................................ 13 Outline Dimensions ....................................................................... 24 Setting the Clock Frequency ..................................................... 13 Ordering Guide .......................................................................... 24 REVISION HISTORY 11/04—Rev. A to Rev. B Changes to Specifications ................................................................ 3 Updated Outline Dimensions ....................................................... 24 Changes to Ordering Guide .......................................................... 24 4/03—Data Sheet Changed from Rev. 0 to Rev. A. Changes to Specifications ................................................................ 2 Rev. B | Page 2 of 24 ADP3168 SPECIFICATIONS All limits at temperature extremes are guaranteed via correlation using standard statistical quality control (SQC). VCC = 12 V, FBRTN = GND, TA = 0°C to 85°C, unless otherwise noted. Table 1. Parameter ERROR AMPLIFIER Output Voltage Range Accuracy Symbol VCOMP VFB Line Regulation Input Bias Current FBRTN Current Output Current Gain Bandwidth Product Slew Rate VID INPUTS Input Low Voltage Input High Voltage Input Current, Input Voltage Low Input Current, Input Voltage High Pull-Up Resistance Internal Pull-Up Voltage VID Transition Delay Time1 No CPU Detection Turn-Off Delay Time OSCILLATOR Frequency Range1 Frequency Variation ∆VFB IFB IFBRTN IO(ERR) GBW(ERR) Output Voltage RAMPADJ Output Voltage RAMPADJ Input Current Range CURRENT SENSE AMPLIFIER Offset Voltage Input Bias Current Gain Bandwidth Product Slew Rate Input Common-Mode Range Positioning Accuracy Output Voltage Range Output Current CURRENT-BALANCE CIRCUIT Common-Mode Range Input Resistance Input Current Input Current Matching VRT VRAMPADJ IRAMPADJ 1 VIL(VID) VIH(VID) IIL(VID) IIH(VID) RVID Conditions Relative to nominal DAC output, Referenced to FBRTN, CSSUM = CSCOMP; see Figure 10 VCC = 10 V to 14 V VOS(CSA) IBIAS(CSA) GBW(CSA) ∆VFB FB forced to VOUT − 3% COMP = FB CCOMP = 10 pF 0.05 15.5 90 500 20 25 Max Unit 3.5 +10 V mV 17 120 0.4 0.8 VID(X) = 0 V VID(X) = 1.25 V TA = 25°C, RT = 250 kΩ, 4-phase TA = 25°C, RT = 115 kΩ, 4-phase TA = 25°C, RT = 75 kΩ, 4-phase RT = 100 kΩ to GND RAMPADJ − FB CSSUM − CSREF; see Figure 5 CCSCOMP = 10 pF CSSUM and CSREF See Figure 6 ICSCOMP = ±100 µA 35 0.825 400 400 0.25 155 1.9 −50 0 −20 15 60 1.00 200 400 600 2.0 −1.5 −50 −30 25 115 4 245 2.1 +50 100 +1.5 +50 10 10 0 −77 0.05 ICSCOMP VSW(X)CM RSW(X) ISW(X) ∆ISW(X) Typ 0.5 −10 14 VID code change to FB change VID code change to 11111 to PWM going low fOSC fPHASE Min −80 3 −83 3.3 500 SW(X) = 0 V SW(X) = 0 V SW(X) = 0 V Guaranteed by design, not tested in production. Rev. B | Page 3 of 24 −600 20 4 −5 30 7 +200 40 10 +5 % µA µA µA MHz V/µs V V µA µA kΩ V ns ns MHz kHz kHz kHz V mV µA mV nA MHz V/µs V mV V µA mV kΩ µA % ADP3168 Parameter CURRENT-LIMIT COMPARATOR ILIMIT Output Voltage Normal Mode Shutdown Mode Output Current, Normal Mode Current-Limit Threshold Voltage Current-Limit Setting Ratio DELAY Normal Mode Voltage DELAY Overcurrent Threshold Latch-Off Delay Time SOFT START Output Current, Soft-Start Mode Soft-Start Delay Time ENABLE INPUT Input Low Voltage Input High Voltage Input Current, Input Voltage Low Input Current, Input Voltage High POWER-GOOD COMPARATOR Undervoltage Threshold Overvoltage Threshold Output Low Voltage Power-Good Delay Time VID Code Changing VID Code Static Crowbar Trip Point Crowbar Reset Point Crowbar Delay Time VID Code Changing VID Code Static PWM OUTPUTS Output Voltage Low Output Voltage High SUPPLY DC Supply Current UVLO Threshold Voltage UVLO Hysteresis Symbol Conditions Min Typ Max Unit VILIMIT(NM) VILIMIT(SD) IILIMIT(NM) VCL EN > 1.7 V, RILIMIT = 250 kΩ EN > 0.8 V, IILIMIT = −100 µA EN > 1.7 V, RILIMIT = 250 kΩ VCSREF − VCSCOMP, RILIMIT = 250 kΩ VCL/IILIMIT 2.9 3 3.1 400 V mV µA mV mV/µA V V µs VDELAY(NM) VDELAY(OC) tDELAY IDELAY(SS) tDELAY(SS) 2.9 1.7 RDELAY = 250 kΩ, CDELAY = 4.7 nF During startup, DELAY < 2.8 V RDELAY = 250 kΩ, CDELAY= 4.7 nF VID code = 011111 VIL(EN) VIH(EN) IIL(EN) IIH(EN) EN = 0 V EN = 1.25 V VPWRGD(UV) VPWRGD(OV) VOL(PWRGD) Relative to nominal DAC output Relative to nominal DAC output IPWRGD(SINK) = 4 mA VCROWBAR tCROWBAR 105 Relative to nominal DAC output Relative to FBRTN Overvoltage to PWM going low 15 IPWM(SINK) = 400 µA IPWM(SOURCE) = 400 µA VUVLO VCC rising Rev. B | Page 4 of 24 20 350 145 3.1 1.9 25 µA µs 0.4 10 +1 25 V V µA µA −200 90 −250 150 225 −325 200 400 mV mV mV 100 250 200 150 550 200 650 µs ns mV mV 0.8 −1 90 450 100 VOL(PWM) VOH(PWM) 12 125 10.4 3 1.8 600 250 400 µs ns 160 5.0 500 4.0 mV V 6.5 0.7 5 6.9 0.9 8 7.3 1.1 mA V V ADP3168 ABSOLUTE MAXIMUM RATINGS Table 2. Parameter VCC FBRTN VID0 to VID5, EN, DELAY, ILIMIT, CSCOMP, RT, PWM1 to PWM4, COMP SW1-SW4 All Other Inputs and Outputs Operating Ambient Temperature Range Operating Junction Temperature Storage Temperature Range Junction to Air Thermal Resistance (θJA) Lead Temperature (Soldering, 10 sec) Vapor Phase (60 sec) Infrared (15 sec) Rating −0.3 V to +15 V −0.3 V to +0.3 V −0.3 V to +5.5 V −5 V to +25 V −0.3 V to VCC + 0.3 V 0°C to 85°C Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those listed in the operational sections of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. Absolute maximum ratings apply individually only, not in combination. Unless otherwise specified, all other voltages are referenced to GND. 125°C −65°C to +150°C 100°C/W 300°C 215°C 220°C ESD CAUTION ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality. Rev. B | Page 5 of 24 ADP3168 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS VID4 1 28 VCC VID3 2 27 PWM1 26 PWM2 VID2 3 ADP3168 25 PWM3 VID0 5 24 PWM4 VID5 6 23 SW1 FBRTN 7 FB 8 TOP VIEW (Not to Scale) 22 SW2 21 SW3 COMP 9 20 SW4 PWRGD 10 19 GND EN 11 18 CSCOMP DELAY 12 17 CSSUM RT 13 16 CSREF RAMPADJ 14 15 ILIMIT 03258-B-002 VID1 4 Figure 2. Pin Configuration Table 3. Pin Function Descriptions Pin No. 1 to 6 Mnemonic VID4 to VID0, VID5 7 8 FBRTN FB 9 10 COMP PWRGD 11 12 EN DELAY 13 RT 14 RAMPADJ 15 ILIMIT 16 CSREF 17 CSSUM 18 CSCOMP 19 20 to 23 GND SW4 to SW1 24 to 27 PWM4 to PWM1 28 VCC Function Voltage Identification DAC Inputs. These six pins are pulled up to an internal reference, providing a Logic 1 if left open. When in normal operation mode, the DAC output programs the FB regulation voltage from 0.8375 V to 1.6 V. Leaving VID4 through VID0 open results in the ADP3168 going into a no CPU mode, shutting off its PWM outputs. Feedback Return. VID DAC and error amplifier reference for remote sensing of the output voltage. Feedback Input. Error amplifier input for remote sensing of the output voltage. An external resistor between this pin and the output voltage sets the no -load offset point. Error Amplifier Output and Compensation Point. Power Good Output. Open-drain output that pulls to GND when the output voltage is outside the proper operating range. Power Supply Enable Input. Pulling this pin to GND disables the PWM outputs. Soft-Start Delay and Current Limit Latch-Off Delay Setting Input. An external resistor and capacitor connected between this pin and GND set the soft-start ramp-up time and the overcurrent latch-off delay time. Frequency Setting Resistor Input. An external resistor connected between this pin and GND sets the oscillator frequency of the device. PWM Ramp Current Input. An external resistor from the converter input voltage to this pin sets the internal PWM ramp. Current Limit Set Point/Enable Output. An external resistor from this pin to GND sets the current limit threshold of the converter. This pin is actively pulled low when the ADP3168 EN input is low or when VCC is below its UVLO threshold to signal to the driver IC that the driver high-side and low-side outputs should go low. Current Sense Reference Voltage Input. The voltage on this pin is used as the reference for the currentsense amplifier and the Power Good and crowbar functions. This pin should be connected to the common point of the output inductors. Current-Sense Summing Node. External resistors from each switch node to this pin sum the average inductor currents to measure the total output current. Current Sense Compensation Point. A resistor and a capacitor from this pin to CSSUM determine the slope of the load line and the positioning loop response time. Ground. All internal biasing and the logic output signals of the device are referenced to this ground. Current Balance Inputs. Inputs for measuring the current level in each phase. The SW pins of unused phases should be left open. Logic-Level PWM Outputs. Each output is connected to the input of an external MOSFET driver, such as the ADP3413 or ADP3418. Connecting the PWM3 and/or PWM4 outputs to GND causes that phase to turn off, allowing the ADP3168 to operate as a 2-, 3 -, or 4 -phase controller. Supply Voltage for the Device. Rev. B | Page 6 of 24 ADP3168 TYPICAL PERFORMANCE CHARACTERISTICS AND TEST CIRCUITS 5.3 TA = 25°C 4-PHASE OPERATION 3.5 5.2 SUPPLY CURRENT (mA) 3.0 2.5 2.0 1.5 1.0 5.1 5.0 4.9 4.8 4.7 0.5 100 150 200 250 300 RT VALUE (kΩ) 4.6 0 0.5 1.0 1.5 1 18 39kΩ CSCOMP CSSUM VOS = CSCOMP – 1V 40 GND 03258-B-005 CSREF 1.0V 19 4.0 VCC 28 2 VID3 PWM1 27 3 VID2 PWM2 26 4 VID1 PWM3 25 5 VID0 PWM4 24 6 VID5 SW1 23 7 FBRTN SW2 22 8 FB SW3 21 9 COMP SW4 20 10 PWRGD GND 19 11 EN 12 DELAY 12V + 1µF 100nF 20kΩ 100nF 1kΩ Figure 5. Test Circuit 1, Current Sense Amplifier VOS 1.25V ADP3168 12V 3.5 6-BIT CODE 1kΩ 16 VID4 VCC 100nF 17 3.0 ADP3168 ADP3168 28 2.5 MASTER CLOCK FREQUENCY (MHz) Figure 4. Supply Current vs. Master Clock Frequency Figure 3. Master Clock Frequency vs. RT 12V 2.0 03258-B-004 50 28 VCC 4.7nF 250kΩ 13 RT 14 8 FB RAMPADJ CSCOMP 18 CSSUM 17 CSREF 16 ILIMIT 15 250kΩ 10kΩ 9 COMP Figure 7. Test Circuit 3, Closed-Loop Output Voltage Accuracy 200kΩ 18 200kΩ CSCOMP 100nF 17 CSSUM ∆V 16 CSREF 1.0V 19 GND ∆VFB = FB∆V = 80mV – FB∆V = 0mV Figure 6. Test Circuit, Positioning Voltage Rev. B | Page 7 of 24 03258-B-007 0 03258-B-003 SEE EQUATION 1 FOR FREQUENCIES NOT ON THIS GRAPH 0 03258-B-006 MASTER CLOCK FREQUENCY (MHz) 4.0 ADP3168 Table 4. Output Voltage vs. VID Code (X = Don’t Care) VID4 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 VID3 1 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 VID2 1 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 1 0 0 0 0 VID1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 0 0 0 0 1 1 1 1 VID0 1 0 0 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 0 0 1 1 0 0 1 1 0 0 VID5 X 0 0 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 VOUT(NOM) No CPU 0.8375 V 0.850 V 0.8625 V 0.875 V 0.8875 V 0.900 V 0.9125 V 0.925 V 0.9375 V 0.950 V 0.9625 V 0.975 V 0.9875 V 1.000 V 1.0125 V 1.025 V 1.0375 V 1.050 V 1.0625 V 1.075 V 1.0875 V 1.100 V 1.1125 V 1.125 V 1.1375 V 1.150 V 1.1625 V 1.175 V 1.1875 V 1.200 V 1.2125 V VID4 VID3 VID2 VID1 VID0 VID5 VOUT(NOM) 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 1 1 1 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 0 1 1 1 1 1 1 1 1 0 0 0 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 0 0 0 0 1 1 1 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 0 1 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 0 1 1.225 V 1.2375 V 1.250 V 1.2625 V 1.275 V 1.2875 V 1.300 V 1.3125 V 1.325 V 1.3375 V 1.350 V 1.3625 V 1.375 V 1.3875 V 1.400 V 1.4125 V 1.425 V 1.4375 V 1.450 V 1.4625 V 1.475 V 1.4875 V 1.500 V 1.5125 V 1.525 V 1.5375 V 1.550 V 1.5625 V 1.575 V 1.5875 V 1.600 V Rev. B | Page 8 of 24 ADP3168 THEORY OF OPERATION The ADP3168 combines a multimode, fixed frequency PWM control with multiphase logic outputs for use in 2-, 3-, and 4-phase synchronous buck CPU core supply power converters. The internal 6-bit VID DAC conforms to Intel’s VRD/VRM 10 specifications. Multiphase operation is important for producing the high currents and low voltages demanded by today’s microprocessors. Handling the high currents in a single-phase converter would place high thermal demands on system components such as inductors and MOSFETs. The multimode control of the ADP3168 ensures a stable, high performance topology for • Balancing currents and thermals between phases • High speed response at the lowest possible switching frequency and output decoupling • Minimizing thermal switching losses due to lower frequency operation • Tight load-line regulation and accuracy • High current output resulting from having up to a 4-phase operation • Reduced output ripple due to multiphase cancellation • PC board layout noise immunity • Ease of use and design due to independent component selection • Flexibility in operation for tailoring design to low cost or high performance NUMBER OF PHASES The number of operational phases and their phase relationship is determined by the internal circuitry that monitors the PWM outputs. Normally, the ADP3168 operates as a 4-phase PWM controller. Grounding the PWM4 pin programs 3-phase operation; grounding the PWM3 and PWM4 pins programs 2-phase operation. When the ADP3168 is enabled, the controller outputs a voltage on PWM3 and PWM4 of approximately 550 mV. An internal comparator checks each pin’s voltage vs. a threshold of 400 mV. If the pin is grounded, the voltage is below the threshold and the phase is disabled. The output resistance of the PWM pin is approximately 5 kΩ during this detection time. Any external pull-down resistance connected to the PWM pin should be at least 25 kΩ to ensure proper operation. The phase detection is made during the first two clock cycles of the internal oscillator. After this time, if the PWM output is not grounded, the 5 kΩ resistance is removed and switches between 0 V and 5 V. If the PWM output was grounded, it remains off. The PWM outputs become logic-level devices once normal operation starts. The detection is normal and is intended for driving external gate drivers such as the ADP3418. Because each phase is monitored independently, operation approaching 100% duty cycle is possible. Also, more than one output can be on at any given time for overlapping phases. MASTER CLOCK FREQUENCY The clock frequency of the ADP3168 is set with an external resistor connected from the RT pin to ground. The frequency follows the graph in Figure 3. To determine the frequency per phase, the clock is divided by the number of phases in use. If PWM4 is grounded, divide the master clock by 3 for the frequency of the remaining phases. If PWM3 and PWM4 are grounded, divide by 2. If all phases are in use, divide by 4. OUTPUT VOLTAGE DIFFERENTIAL SENSING The ADP3168 combines differential sensing with a high accuracy VID DAC and reference and a low offset error amplifier to maintain a worst-case specification of ±10 mV differential sensing error with a VID input of 1.6000 V over its full operating output voltage and temperature range. The output voltage is sensed between the FB and FBRTN pins. FB should be connected through a resistor to the regulation point, usually the remote sense pin of the microprocessor. FBRTN should be connected directly to the remote sense ground point. The internal VID DAC and precision reference are referenced to FBRTN, which has a minimal current of 90 µA to allow accurate remote sensing. The internal error amplifier compares the output of the DAC to the FB pin to regulate the output voltage. OUTPUT CURRENT SENSING The ADP3168 provides a dedicated current sense amplifier (CSA) to monitor the total output current for proper voltage positioning vs. load current and for current limit detection. Sensing the load current at the output gives the total average current being delivered to the load, which is an inherently more accurate method than peak current detection or sampling the current across a sense element such as the low-side MOSFET. This amplifier can be configured several ways, depending on the objectives of the system: • • • Rev. B | Page 9 of 24 Output inductor ESR sensing without thermistor for lowest cost Output inductor ESR sensing with thermistor for improved accuracy with tracking of inductor temperature Sense resistors for most accurate measurements ADP3168 The positive input of the CSA is connected to the CSREF pin, which is connected to the output voltage. The inputs to the amplifier are summed together through resistors from the sensing element (such as the switch node side of the output inductors) to the inverting input, CSSUM. The feedback resistor between CSCOMP and CSSUM sets the gain of the amplifier, and a filter capacitor is placed in parallel with this resistor. The gain of the amplifier is programmable by adjusting the feedback resistor to set the load line required by the microprocessor. The current information is then given as the difference of CSREF − CSCOMP. This difference signal is used internally to offset the VID DAC for voltage positioning and as a differential input for the current-limit comparator. To provide the best accuracy for the current sensing, the CSA was designed to have a low offset input voltage. Also, the sensing gain is determined by external resistors so that it can be made extremely accurate. ACTIVE IMPEDANCE CONTROL MODE For controlling the dynamic output voltage droop as a function of output current, a signal proportional to the total output current at the CSCOMP pin can be scaled to be equal to the droop impedance of the regulator times the output current. This droop voltage is then used to set the input control voltage to the system. The droop voltage is subtracted from the DAC reference input voltage directly to tell the error amplifier where the output voltage should be. This differs from previous implementations and allows enhanced feed-forward response. Resistors RSW1 through RSW4 (see the typical applica-tion circuit in Figure 11) can be used for adjusting thermal balance. It is best to add these resistors during the initial design, so make sure placeholders are provided in the layout. To increase the current in any given phase, make RSW for that phase larger. (Make RSW = 0 for the hottest phase and do not change during balancing.) Increasing RSW to only 500 Ω makes a substantial increase in phase current. Increase each RSW value by small amounts to achieve balance, starting with the coolest phase first. VOLTAGE CONTROL MODE A high gain bandwidth voltage mode error amplifier is used for the voltage-mode control loop. The control input voltage to the positive input is set via the VID 6-bit logic code, according to the voltages listed in Table 4. This voltage is also offset by the droop voltage for active positioning of the output voltage as a function of current, commonly known as active voltage positioning. The output of the amplifier is the COMP pin, which sets the termination voltage for the internal PWM ramps. The negative input (FB) is tied to the output sense location with a resistor, RB, and is used for sensing and controlling the output voltage at this point. A current source from the FB pin flowing through RB is used for setting the no-load offset voltage from the VID voltage. The no-load voltage is negative with respect to the VID DAC. The main loop compensation is incorporated into the feedback network between FB and COMP. SOFT START CURRENT-CONTROL MODE AND THERMAL BALANCE The ADP3168 has individual inputs that are used for monitoring the current in each phase. This information is combined with an internal ramp to create a current-balancing feedback system that has been optimized for initial current balance accuracy and dynamic thermal balancing during operation. This current-balance information is independent of the average output current information used for positioning described previously. The magnitude of the internal ramp can be set to optimize the transient response of the system. It also monitors the supply voltage for feed-forward control for changes in the supply. A resistor connected from the power input voltage to the RAMPADJ pin determines the slope of the internal PWM ramp. Detailed information about programming the ramp is given in the Application Information section. External resistors can be placed in series with individual phases, for example, to create an intentional current imbalance so one phase may have better cooling and can support higher currents. The power-on ramp-up time of the output voltage is set with a capacitor and a resistor in parallel from the DELAY pin to ground. The RC time constant also determines the current-limit latch-off time, as explained in the following section. In UVLO or when EN is a logic low, the DELAY pin is held at ground. After the UVLO threshold is reached and EN is a logic high, the DELAY capacitor is charged up with an internal 20 µA current source. The output voltage follows the ramping voltage on the DELAY pin, limiting the inrush current. The soft-start time depends on the values of VID DAC and CDLY, with a secondary effect from RDLY. Refer to the Application Information section for detailed information on setting CDLY. When the PWRGD threshold is reached, the soft-start cycle is stopped and the DELAY pin is pulled up to 3 V. This ensures that the output voltage is at the VID voltage when the PWRGD signals to the system that the output voltage is good. If EN is taken low or VCC drops below UVLO, the DELAY capacitor is reset to ground to be ready for another soft-start cycle. Figure 8 shows a typical start-up sequence for the ADP3168. Rev. B | Page 10 of 24 ADP3168 03258-B-008 The latch-off function can be reset either by removing and reapplying VCC to the ADP3168, or by pulling the EN pin low for a short time. To disable the short-circuit latch-off function, the external resistor to ground should be left open, and a high value (>1 MΩ) resistor should be connected from DELAY to VCC. This prevents the DELAY capacitor from discharging, so the 1.8 V threshold is never reached. The resistor has an impact on the soft-start time because the current through it adds to the internal 20 µA current source. Figure 8. Start-Up Waveforms, Circuit of Figure 12. Channel 1—PWRGD, Channel 2—VOUT, Channel 3—High-Side MOSFET VGS, Channel 4—Low-Side MOSFET VGS The ADP3168 compares a programmable current-limit set point to the voltage from the output of the current-sense amplifier. The level of current limit is set with the resistor from the ILIMIT pin to ground. During normal operation, the voltage on ILIMIT is 3 V. The current through the external resistor is internally scaled to give a current-limit threshold of 10.4 mV/µA. If the difference in voltage between CSREF and CSCOMP rises above the current-limit threshold, the internal current-limit amplifier controls the internal COMP voltage to maintain the average output current at the limit. After the limit is reached, the 3 V pull-up on the DELAY pin is disconnected, and the external delay capacitor is discharged through the external resistor. A comparator monitors the DELAY voltage and shuts off the controller when the voltage drops below 1.8 V. The current-limit latch-off delay time is therefore set by the RC time constant discharging from 3 V to 1.8 V. The Application Information section discusses the selection of CDLY and RDLY. 03258-B-009 CURRENT-LIMIT, SHORT-CIRCUIT, AND LATCH-OFF PROTECTION Figure 9. Overcurrent Latch-Off Waveforms, Circuit of Figure 11. Channel 1— PWRGD, Channel 2—VOUT, Channel 3—CSCOMP Pin of ADP3168, Channel 4—High-Side MOSFET VGS During startup, when the output voltage is below 200 mV, a secondary current limit is active. This is necessary because the voltage swing of CSCOMP cannot go below ground. This secondary current limit controls the internal COMP voltage to the PWM comparators to 2 V. This limits the voltage drop across the low-side MOSFETs through the current-balance circuitry. There is also an inherent per-phase current limit that protects individual phases in the case where one or more phases stop functioning because of a faulty component. This limit is based on the maximum normal mode COMP voltage. Because the controller continues to cycle the phases during the latch-off delay time, if the short is removed before the 1.8 V threshold is reached, the controller returns to normal operation. The recovery characteristic depends on the state of PWRGD. If the output voltage is within the PWRGD window, the controller resumes normal operation. However, if a short circuit has caused the output voltage to drop below the PWRGD threshold, a soft-start cycle is initiated. Rev. B | Page 11 of 24 ADP3168 DYNAMIC VID OUTPUT CROWBAR The ADP3168 incorporates the ability to dynamically change the VID input while the controller is running. This allows the output voltage to change while the supply is running and supplying current to the load. This is commonly referred to as VID on-the-fly (OTF). A VID OTF can occur under either light load or heavy load conditions. The processor signals the controller by changing the VID inputs in multiple steps from the start code to the finish code. This change can be either positive or negative. As part of the protection for the load and output components of the supply, the PWM outputs are driven low (turning on the low-side MOSFETs) when the output voltage exceeds the upper Power-Good threshold. This crowbar action stops once the output voltage has fallen below the release threshold of approximately 450 mV. When a VID input changes state, the ADP3168 detects the change and ignores the DAC inputs for a minimum of 400 ns. This prevents a false code due to logic skew while the six VID inputs are changing. Additionally, the first VID change initiates the PWRGD and CROWBAR blanking functions for a minimum of 250 µs to prevent a false PWRGD or CROWBAR event. Each VID change resets the internal timer. Figure 10 shows VID on-the-fly performance when the output voltage is stepping up and the output current is switching between minimum and maximum values, which is the worst-case situation. Turning on the low-side MOSFETs pulls down the output as the reverse current builds up in the inductors. If the output overvoltage is due to a short of the high-side MOSFET, this action current limits the input supply or blows its fuse, protecting the microprocessor from destruction. OUTPUT ENABLE AND UVLO The input supply (VCC) to the controller must be higher than the UVLO threshold, and the EN pin must be higher than its logic threshold for the ADP3168 to begin switching. If UVLO is less than the threshold or the EN pin is a logic low, the ADP3168 is disabled. This holds the PWM outputs at ground, shorts the DELAY capacitor to ground, and holds the ILIMIT pin at ground. 03258-B-010 In the application circuit, the ILIMIT pin should be connected to the OD pins of the ADP3418 drivers. Because ILIMIT is grounded, this disables the drivers so that both DRVH and DRVL are grounded. This feature is important to prevent discharging of the output capacitors when the controller is shut off. If the driver outputs were not disabled, a negative voltage could be generated on the output due to the high current discharge of the output capacitors through the inductors. Figure 10. VID On-the-Fly Waveforms, Circuit of Figure 12. VID Change = 5 mV, 5 µs per Step, 50 Steps, IOUT Change =5 A to 65 A POWER-GOOD MONITORING The Power-Good comparator monitors the output voltage via the CSREF pin. The PWRGD pin is an open-drain output whose high level (when connected to a pull-up resistor) indicates that the output voltage is within the nominal limits specified in Table 1 based on the VID voltage setting. PWRGD goes low if the output voltage is outside of this specified range. PWRGD is blanked during a VID OTF event for a period of 250 µs to prevent false signals during the time the output is changing. Rev. B | Page 12 of 24 ADP3168 APPLICATION INFORMATION The design parameters for a typical Intel VRD 10 compliant CPU application are as follows: • Input voltage (VIN) = 12 V • VID setting voltage (VVID) = 1.500 V • Duty cycle (D) = 0.125 • Nominal output voltage at no load (VONL) = 1.480 V • Nominal output voltage at 65 A load (VOFL) = 1.3955 V • Static output voltage drop based on a 1.3 mΩ load line (RO) from no load to full load • (VD) = VONL − VOFL = 1.480 V − 1.3955 V = 84.5 mV • Maximum output current (IO) = 65 A • Maximum output current step (∆IO) = 60 A • Number of phases (n) = 3 • Switching frequency per phase (fSW) = 267 kHz SETTING THE CLOCK FREQUENCY The ADP3168 uses a fixed-frequency control architecture. The frequency is set by an external timing resistor (RT). The clock frequency and the number of phases determine the switching frequency per phase, which relates directly to switching losses and the sizes of the inductors and input and output capacitors. With n = 3 for three phases, a clock frequency of 800 kHz sets the switching frequency, fSW, of each phase to 267 kHz, which represents a practical trade-off between the switching losses and the sizes of the output filter components. Figure 3 shows that to achieve an 800 kHz oscillator frequency, the correct value for RT is 249 kΩ. Alternatively, the value for RT can be calculated using RT = 1 (n × f SW × 5.83 pF) − 1 1.5M Ω (1) The closest standard value for CDLY is 39 nF. Once CDLY has been chosen, RDLY can be calculated for the current-limit latch-off time using RDLY = The choice of inductance for the inductor determines the ripple current in the inductor. Less inductance leads to more ripple current, which increases the output ripple voltage and conduction losses in the MOSFETs but allows using smaller inductors and, for a specified peak-to-peak transient deviation, less total output capacitance. Conversely, a higher inductance means lower ripple current and reduced conduction losses but requires larger inductors and more output capacitance for the same peak-to-peak transient deviation. In any multiphase converter, a practical value for the peak-to-peak inductor ripple current is less than 50% of the maximum dc current in the same inductor. Equation 4 shows the relationship between the inductance, oscillator frequency, and peak-to-peak ripple current in the inductor. Equation 5 can be used to determine the minimum inductance based on a given output ripple voltage. where tSS is the desired soft-start time. Assuming an RDLY of 390 kΩ and a desired a soft-start time of 3 ms, CDLY is 36 nF. L≥ VVID × (1 − D ) f SW × L VVID × RO × (1 − (n × D )) f SW × V RIPPLE (4) (5) Solving Equation 5 for a 10 mV p-p output ripple voltage yields: Because the soft-start and current limit latch-off delay functions share the DELAY pin, these two parameters must be considered together. The first step is to set CDLY for the soft-start ramp. This ramp is generated with a 20 µA internal current source. The value of RDLY has a second-order impact on the soft-start time because it sinks part of the current source to ground. However, as long as RDLY is kept greater than 200 kΩ, this effect is minor. The value for CDLY can be approximated using (2) (3) INDUCTOR SELECTION IR = SOFT START AND CURRENT LIMIT LATCH-OFF DELAY TIMES ⎞ t ss ⎟× ⎟ V VID ⎠ C DLY If the result for RDLY is less than 200 kΩ, a smaller soft-start time should be considered by recalculating the equation for CDLY, or a longer latch-off time should be used. In no case should RDLY be less than 200 k . In this example, a delay time of 8 ms gives RDLY = 402 kΩ. The closest standard 5% value is 390 kΩ. where 5.83 pF and 1.5 MΩ are internal IC component values. For good initial accuracy and frequency stability, a 1% resistor is recommended. ⎛ VVID C DLY = ⎜ 20 µA − ⎜ 2 × RDLY ⎝ 1.96 × t DELAY L≥ 1.5 V × 1.3 m Ω × (1 − 0.375 ) = 456 nH 267 kHz × 10 mV If the resulting ripple voltage is less than that designed for, the inductor can be made smaller until the ripple value is met. This allows optimal transient response and minimum output decoupling. The smallest possible inductor should be used to minimize the number of output capacitors. Choosing a 600 nH inductor is a good starting point and gives a calculated ripple current of 8.2 A. The inductor should not saturate at the peak current of 25.8 A and should be able to handle the sum of the power dissipation caused by the average current of 22.7 A in the winding and core loss. Rev. B | Page 13 of 24 ADP3168 increased measurement error. A good rule is to have the DCR be about 1 to 1½ times the droop resistance (RO). Our example uses an inductor with a DCR of 1.6 mΩ. Another important factor in the inductor design is the DCR, which is used for measuring the phase currents. A large DCR causes excessive power losses, while too small a value leads to L1 1.6µH VIN 12V VIN RTN 470µF/16V × 6 NICHICON PW SERIES + + C1 C6 D1 1N4148WS C9 4.7µF U2 C8 ADP3418 100nF D2 1N4148WS Q1 IPD12N03L DRVH 8 1 BST 2 IN 3 OD PGND 6 4 VCC DRVL 5 L2 600nH/1.6mΩ SW 7 820µF/2.5V × 8 FUJITSU RE SERIES 8mΩ ESR (EACH) C10 4.7nF R1 2.2Ω C7 4.7µF + + C21 C28 VCC(CORE) 0.8375V–1.6V 65A AVG, 74A PK VCC(CORE) RTN Q3 IPD06N03L Q2 IPD06N03L D3 1N4148WS C13 4.7µF C12 U3 ADP3418 100nF Q4 IPD12N03L DRVH 8 1 BST 2 IN 3 OD PGND 6 4 VCC DRVL 5 10µF × 23MLCC AROUND SOCKET L3 600nH/1.6mΩ SW 7 C14 4.7nF C11 4.7µF R2 2.2Ω Q6 IPD06N03L Q5 IPD06N03L D4 1N4148WS U4 ADP3418 C17 4.7µF C16 100nF Q7 IPD12N03L DRVH 8 1 BST 2 IN 3 OD PGND 6 4 VCC DRVL 5 L4 600nH/1.6mΩ SW 7 C18 4.7nF R3 2.2Ω C15 4.7µF Q9 Q8 IPD06N03L IPD06N03L R4 10Ω C19 1µF RTH 100kΩ, 5% + C20 33µF U1 ADP3168 RR 383kΩ 1 VID4 VCC 28 2 VID3 PWM1 27 3 VID2 PWM2 26 4 VID1 PWM3 25 5 VID0 PWM4 24 6 VID5 SW1 23 FROM CPU RSW11 RSW21 CB 1.5nF RB 1.33kΩ CA RA 390pF 16.9kΩ ENABLE CDLY 39nF FBRTN SW2 22 8 FB SW3 21 RSW31 CFB 33pF RDLY 390kΩ RT 249kΩ 9 COMP SW4 20 10 PWRGD GND 19 11 EN 12 DELAY CSSUM 17 13 RT CSREF 16 14 RAMPADJ CSCOMP 18 RPH3 124kΩ CCS2 RCS1 1.5nF 35.7kΩ RCS2 73.2kΩ RPH1 124kΩ RPH2 124kΩ CCS1 2.2nF ILIMIT 15 RLIM 200kΩ NOTE: 1 FOR A DESCRIPTION OF OPTIONAL R SW RESISTORS, SEE THE THEORY OF OPERATION SECTION. Figure 11. 65 A Intel Pentium 4-CPU Supply Circuit, VRD 10 Design Rev. B | Page 14 of 24 03258-011 POWER GOOD 7 ADP3168 DESIGNING AN INDUCTOR OUTPUT DROOP RESISTANCE Once the inductance and DCR are known, the next step is to either design an inductor or find a standard inductor that comes as close as possible to meeting the overall design goals. It is also important to have the inductance and DCR tolerance specified to control the accuracy of the system. 15% inductance and 8% DCR (at room temperature) are reasonable tolerances that most manufacturers can meet. The design requires that the regulator output voltage measured at the CPU pins drops when the output current increases. The specified voltage drop corresponds to a dc output resistance (RO). The first decision in designing the inductor is to choose the core material. There are several possibilities for providing low core loss at high frequencies. Two examples are the powder cores (e.g., Kool-Mµ® from Magnetics, Inc. or Micrometals) and the gapped soft ferrite cores (e.g., 3F3 or 3F4 from Philips). Low frequency powdered iron cores should be avoided due to their high core loss, especially when the inductor value is relatively low and the ripple current is high. The best choice for a core geometry is a closed-loop type such as a pot core, PQ, U, or E core or toroid. A good compromise between price and performance is a core with a toroidal shape. The output current is measured by summing together the voltage across each inductor and passing the signal through a low-pass filter. This summer filter is the CS amplifier configured with resistors RPH(X) (summers), and RCS and CCS (filter). The output resistance of the regulator is set by the following equations, where RL is the DCR of the output inductors: RO = CCS = Magnetic Designer Software Intusoft (www.intusoft.com) • Designing Magnetic Components for High-Frequency DC-DC Converters, by William T. McLyman, Kg Magnetics, Inc., ISBN 1883107008 SELECTING A STANDARD INDUCTOR The companies listed below can provide design consultation and deliver power inductors optimized for high power applications upon request. Power Inductor Manufacturers • Coilcraft (847)639-6400 www.coilcraft.com • Coiltronics (561)752-5000 www.coiltronics.com • Sumida Electric Company (510) 668-0660 www.sumida.com • Vishay Intertechnology (402) 563-6866 www.vishay.com L RL × RCS (6) (7) One has the flexibility of choosing either RCS or RPH(X). It is best to select RCS equal to 100 kΩ, and then solve for RPH(X) by rearranging Equation 6. RPH (x ) = There are many useful references for quickly designing a power inductor, such as the following: • RCS × RL RPH ( x ) RPH ( x ) = RL × RCS RO 1.6 mΩ × 100 kΩ = 123 kΩ 1.3 mΩ Next, use Equation 6 to solve for CCS. CCS = 600 nH = 3.75 nF 1.6 mΩ × 100 kΩ It is best to have a dual location for CCS in the layout so standard values can be used in parallel to get as close to the value desired. For this example, choosing CCS to be 1.5 nF and 2.2 nF in parallel is a good choice. For best accuracy, CCS should be a 5% or 10% NPO capacitor. The closest standard 1% value for RPH(X) is 124 kΩ. Rev. B | Page 15 of 24 ADP3168 INDUCTOR DCR TEMPERATURE CORRECTION 4. With the inductor’s DCR being used as the sense element and copper wire being the source of the DCR, one needs to compensate for temperature changes of the inductor’s winding. Fortunately, copper has a well-known temperature coefficient (TC) of 0.39%/°C. TO SWITCH NODES RTH RPH1 RPH2 RTH = 5. TO VOUT SENSE 6. RPH3 RCS1 17 KEEP THIS PATH AS SHORT AS POSSIBLE AND WELL AWAY FROM SWITCH NODE LINES 16 03258-B-012 CSREF Figure 12. Temperature Compensation Circuit Values The following procedure and expressions yield values to use for RCS1, RCS2, and RTH (the thermistor value at 25°C) for a given RCS value. 1. 2. 3. 1 1 1 − 1 − RCS2 RCS1 RTH ( ACTUAL ) (9) RTH (CALCULATED ) Finally, calculate values for RCS1 and RCS2 using Equation 10: RCS1 = RCS × k × γ CS1 (10) RCS2 = RCS × ((1 − k ) + (k × γ CS 2 )) 18 CCS 1.8nF CSSUM RCS2 (8) Calculate RTH = RTH × RCS, then select the closest value of thermistor available. Also compute a scaling factor k based on the ratio of the actual thermistor value used relative to the computed one: k= ADP3168 CSCOMP (1 − A) 1 A − 1 − RCS2 γ1 − RCS 2 RCS1 = If RCS is designed to have an opposite and equal percentage change in resistance to that of the wire, it cancels the temperature variation of the inductor’s DCR. Due to the nonlinear nature of NTC thermistors, resistors RCS1 and RCS2 are needed (see Figure 12) to linearize the NTC and produce the desired temperature tracking. PLACE AS CLOSE AS POSSIBLE TO NEAREST INDUCTOR OR LOW-SIDE MOSFET Compute the relative values for RCS1, RCS2, and RTH using: ( A − B ) × γ1 × γ 2 × − A × (1 − B ) × γ 2 + B × (1 − A ) × γ1 RCS 2 = A × (1 − B ) × γ1 − B × (1 − A ) × γ 2 − ( A − B ) Select an NTC based on type and value. Because there is no value yet, start with a thermistor with a value close to RCS. The NTC should also have an initial tolerance of better than 5%. Based on the type of NTC, find its relative resistance value at two temperatures. The temperatures that work well are 50°C and 90°C. We will call these resistance values A (RTH(50°C)/RTH(25°C)) and B (RTH(90°C)/RTH(25°C)). Note that the NTC’s relative value is always 1 at 25°C. Find the relative value of RCS required for each of these temperatures. This is based on the percentage change needed, which in this example is initially 0.39%/°C. These are called r1 (1/(1 + TC × (T1 − 25))) and r2 (1/(1 + TC × (T2 − 25))), where TC = 0.0039, T1 = 50°C, and T2 = 90°C. For this example, RCS has been chosen to be 100 kΩ , so we start with a thermistor value of 100 kΩ. Looking through available 0603 size thermistors, we find a Vishay NTHS0603N01N1003JR NTC thermistor with A = 0.3602 and B = 0.09174. From these we compute RCS1 = 0.3796, RCS2 = 0.7195, and RTH = 1.0751. Solving for RTH yields 107.51 kΩ, so we choose 100 kΩ, making k = 0.9302. Finally, we find RCS1 and RCS2 to be 35.3 kΩ and 73.9 kΩ. Choosing the closest 1% resistor values yields a choice of 35.7 kΩ and 73.2 kΩ. OUTPUT OFFSET Intel’s specification requires that at no load the nominal output voltage of the regulator be offset to a lower value than the nominal voltage corresponding to the VID code. The offset is set by a constant current source flowing out of the FB pin (IFB) and flowing through RB. The value of RB can be found using Equation 11: RB = VVID − VONL I FB (11) RB = 1.5 V − 1.480 V = 1.33 kΩ 15 µA The closest standard 1% resistor value is 1.33 kΩ. Rev. B | Page 16 of 24 ADP3168 COUT SELECTION The required output decoupling for the regulator is typically recommended by Intel for various processors and platforms. One can also use some simple design guidelines to determine what is required. These guidelines are based on having both bulk and ceramic capacitors in the system. For our example, 23 10 µF 1206 MLC capacitors (CZ = 230 µF) were used. The VID on-the-fly step change is 250 mV in 150 µs with a setting error of 2.5 mV. Solving for the bulk capacitance yields ⎞ ⎛ 600 nH × 60 A C x (MIN ) ≤ ⎜⎜ − 230 µF ⎟⎟ = 5.92 mF ⎠ ⎝ 3 × 1.3 m Ω × 1.5 V 600 nH × 250 mV C x (MAX ) ≤ × 2 3 × 4.6 2 × (1.3 mΩ ) × 1.5 V The first thing is to select the total amount of ceramic capacitance. This is based on the number and type of capacitor to be used. The best location for ceramics is inside the socket, with 12 to 18 of size 1206 being the physical limit. Others can be placed along the outer edge of the socket as well. 2 ⎛ ⎞ ⎛ 150 µs × 1.5 V × 3 × 4.6 × 1.3 mΩ ⎞ ⎜ ⎟ ⎟ ⎜ 1 − 1 + ⎜ ⎟ − 230 µF ⎟ ⎜ 250 mV 600 nH × ⎜ ⎟ ⎠ ⎝ ⎝ ⎠ = 23.9 mF where k = 4.6 Combined ceramic values of 200 µF to 300 µF are recommended, usually made up of multiple 10 µF or 22 µF capacitors. Select the number of ceramics and then find the total ceramic capacitance (CZ). Next, there is an upper limit imposed on the total amount of bulk capacitance (CX) when one considers the VID on-the-fly voltage stepping of the output (voltage step VV in time tV with error of VERR) and a lower limit based on meeting the critical capacitance for load release for a given maximum load step ∆IO: ⎞ ⎛ L × Δ IO C x ( MIN ) ≥ ⎜ − Cz ⎟ ⎟ ⎜ n× R ×V VID O ⎠ ⎝ Using eight 820 µF A1-Polys with a typical ESR of 8 mΩ each yields CX = 6.56 mF with an RX = 1.0 mΩ. One last check should be made to ensure that the ESL of the bulk capacitors (LX) is low enough to limit the initial high frequency transient spike. This is tested using (12) Lx ≤ C z × R Lx ≤ 230 µF × (1.3 m Ω ) = 389 pH 2 Cx ( MAX ) ≤ L V × V 2 2 nK RO VVID 2 ⎛ ⎞ ⎛ V ⎜ ⎟ nKRO ⎞ ⎟ − 1 ⎟ − C z (13) × ⎜ 1 + ⎜⎜ tv VID × ⎟ L ⎠ ⎜ ⎟ ⎝ VV ⎝ ⎠ ⎛V where K = 1n ⎜⎜ ERR ⎝ VV (14) In this example, LX is 375 pH for the eight A1-Polys capacitors, which satisfies this limitation. If the LX of the chosen bulk capacitor bank is too large, the number of capacitors must be increased. One should note that for this multimode control technique, all ceramic designs can be used as long as the conditions of Equations 11, 12, and 13 are satisfied. ⎞ ⎟ ⎟ ⎠ To meet the conditions of these expressions and transient response, the ESR of the bulk capacitor bank (RX) should be less than two times the droop resistance, RO. If the CX(MIN) is larger than CX(MAX), the system does not meet the VID on-the-fly specification and may require the use of a smaller inductor or more phases (and may have to increase the switching frequency to keep the output ripple the same). Rev. B | Page 17 of 24 ADP3168 POWER MOSFETS For this example, the N-channel power MOSFETs have been selected for one high-side switch and two low-side switches per phase. The main selection parameters for the power MOSFETs are VGS(TH), QG, CISS, CRSS, and RDS(ON). The minimum gate drive voltage (the supply voltage to the ADP3418) dictates whether standard threshold or logic-level threshold MOSFETs must be used. With VGATE ~10 V, logic-level threshold MOSFETs (VGS(TH) <2.5 V) are recommended. The maximum output current IO determines the RDS(ON) requirement for the low-side (synchronous) MOSFETs. The ADP3168, balances currents between phases, thus the current in each lowside MOSFET is the output current divided by the total number of MOSFETs (nSF). With conduction losses being dominant, the following expression shows the total power being dissipated in each synchronous MOSFET in terms of the ripple current per phase (IR) and average total output current (IO): ⎡⎛ I PSF = (1 − D ) × ⎢⎜⎜ O ⎢⎝ n SF ⎣ 2 ⎞ 1 ⎛ n IR ⎟ + ×⎜ ⎟ 12 ⎜⎝ n SF ⎠ ⎞ ⎟ ⎟ ⎠ 2 ⎤ ⎥ × RDS (SF ) (15) ⎥ ⎦ Knowing the maximum output current being designed for and the maximum allowed power dissipation, one can find the required RDS(ON) for the MOSFET. For D-PAK MOSFETs up to an ambient temperature of 50°C, a safe limit for PSF is 1 W to 1.5 W at 120°C junction temperature. Thus, for this example (65 A maximum), we find RDS(SF) (per MOSFET) < 8.7 mΩ. This RDS(SF) is also at a junction temperature of about 120°C, so we need to make sure we account for this when making this selection. For this example, we selected two lower-side MOSFETs at 7 mΩ each at room temperature, which gives 8.4 mΩ at high temperature. Another important factor for the synchronous MOSFET is the input capacitance and feedback capacitance. The ratio of the feedback to input needs to be small (less than 10% is recommended) to prevent accidental turn-on of the synchronous MOSFETs when the switch node goes high. Also, the time to switch the synchronous MOSFETs off should not exceed the nonoverlap dead time of the MOSFET driver (40 ns typical for the ADP3418). The output impedance of the driver is about 2 Ω and the typical MOSFET input gate resistances are about 1 Ω to 2 Ω, so a total gate capacitance of less than 6000 pF should be adhered to. Because there are two MOSFETs in parallel, the input capacitance for each synchronous MOSFET should be limited to 3000 pF. The high-side (main) MOSFET must be able to handle two main power dissipation components: conduction and switching losses. The switching loss relates to the amount of time it takes for the main MOSFET to turn on and off, and to the current and voltage that are being switched. Basing the switching speed on the rise and fall time of the gate driver impedance and MOSFET input capacitance, the following expression provides an approximate value for the switching loss per main MOSFET, where nMF is the total number of main MOSFETs: PS (MF ) = 2 × f SW × VCC × I O n × RG × MF × C ISS n n MF (16) Here, RG is the total gate resistance (2 Ω for the ADP3418 and about 1 Ω for typical high speed switching MOSFETs, making RG = 3 Ω) and CISS is the input capacitance of the main MOSFET. Note that adding more main MOSFETs (nMF) does not really help the switching loss per MOSFET because the additional gate capacitance slows switching. The best thing to reduce switching loss is to use lower gate capacitance devices. The conduction loss of the main MOSFET is given by the following, where RDS(MF) is the ON resistance of the MOSFET: ⎡⎛ I PC (MF ) = D × ⎢⎜⎜ O ⎢⎝ nMF ⎣ 2 ⎞ 1 ⎛ n × IR ⎟ + ×⎜ ⎟ 12 ⎜⎝ nMF ⎠ ⎞ ⎟ ⎟ ⎠ 2 ⎤ ⎥ × RDS (MF ) (17) ⎥ ⎦ Typically, for main MOSFETs, the highest speed (low CISS) device is preferred, but these usually have higher ON resistance. Select a device that meets the total power dissipation (about 1.5 W for a single D-PAK) when combining the switching and conduction losses. For this example, an Infineon IPD12N03L was selected as the main MOSFET (three total; nMF = 3), with a CISS = 1460 pF (max) and RDS(MF) = 14 mΩ (max at TJ = 120°C), and an Infineon IPD06N03L was selected as the synchronous MOSFET (six total; nSF = 6), with CISS = 2370 pF (max) and RDS(SF) = 8.4 mΩ (max at TJ = 120°C). The synchronous MOSFET CISS is less than 3000 pF, satisfying that requirement. Solving for the power dissipation per MOSFET at IO = 65 A and IR = 8.2 A yields 863 mW for each synchronous MOSFET and 1.44 W for each main MOSFET. These numbers work well considering there is usually more PCB area available for each main MOSFET vs. each synchronous MOSFET. One last thing to consider is the power dissipation in the driver for each phase. This is best described in terms of the QG for the MOSFETs and is given by the following, where QGMF is the total gate charge for each main MOSFET and QGSF is the total gate charge for each synchronous MOSFET: ⎡ f ⎤ PDRV = ⎢ SW × (n MF × QGMF + nSF × QGSF ) + I CC ⎥ × VCC (18) ⎣2× n ⎦ Also shown is the standby dissipation factor (ICC × VCC) for the driver. For the ADP3418, the maximum dissipation should be less than 400 mW. For our example, with ICC = 7 mA, QGMF = 22.8 nC, and QGSF = 34.3 nC, we find 260 mW in each driver, which is below the 400 mW dissipation limit. See the ADP3418 data sheet for more details. Rev. B | Page 18 of 24 ADP3168 RAMP RESISTOR SELECTION The ramp resistor (RR) is used for setting the size of the internal PWM ramp. The value of this resistor is chosen to provide the best combination of thermal balance, stability, and transient response. This expression determines the optimum value: RR = AR × L 3 × AD × RDS × C R RR = 0.2 × 600 nH = 381 k Ω 3 × 5 × 4.2 mΩ × 5 pF (19) where AR is the internal ramp amplifier gain, AD is the current balancing amplifier gain, RDS is the total low-side MOSFET ON resistance, and CR is the internal ramp capacitor value. The closest standard 1% resistor value is 383 kΩ. The internal ramp voltage magnitude can be calculated using VR = AR × (1 − D ) × VVID RR × C R × f SW 0.2 × (1 − 0.125 ) × 1.5 V VR = 383 kΩ × 5 pF × 267 kHz For values of RLIM greater than 500 kΩ, the current limit may be lower than expected, so some adjustment of RLIM may be needed. Here, ILIM is the average current limit for the output of the supply. For our example, choosing 120 A for ILIM, we find RLIM to be 200 kΩ, for which we chose 200 kΩ as the nearest 1% value. The per-phase current limit described earlier has its limit determined by the following: I PHLIM ≅ VCOMP (MAX ) − VR − VBIAS AD × RDS (MAX ) + IR 2 (23) For the ADP3168, the maximum COMP voltage (VCOMP(MAX)) is 3.3 V, the COMP pin bias voltage (VBIAS) is 1.2 V, and the current balancing amplifier gain (AD) is 5. Using VR of 0.63 V and RDS(MAX) of 4.2 mΩ (low-side ON resistance at 150°C), we find a per-phase limit of 66 A. (20) The size of the internal ramp can be made larger or smaller. If it is made larger, stability and transient response improve, but thermal balance degrades. Likewise, if the ramp is made smaller, thermal balance improves at the sacrifice of transient response and stability. The factor of three in the denominator of Equation 19 sets a ramp size that gives an optimal balance for good stability, transient response, and thermal balance. This limit can be adjusted by changing the ramp voltage VR, but make sure not to set the per-phase limit lower than the average per-phase current (ILIM/n). There is also a per-phase initial duty cycle limit determined by DMAX = D × VCOMP (MAX ) − VBIAS VRT (24) For this example, the maximum duty cycle is found to be 0.42. COMP PIN RAMP FEEDBACK LOOP COMPENSATION DESIGN There is a ramp signal on the COMP pin due to the droop voltage and output voltage ramps. This ramp amplitude adds to the internal ramp to produce the following overall ramp signal at the PWM input. Optimized compensation of the ADP3168 allows the best possible response of the regulator’s output to a load change. The basis for determining the optimum compensation is to make the regulator and output decoupling appear as an output impedance that is entirely resistive over the widest possible frequency range, including dc, and equal to the droop resistance (RO). With the resistive output impedance, the output voltage droops in proportion with the load current at any load current slew rate; this ensures the optimal positioning and allows the minimization of the output decoupling. VRT = VR ⎛ 2 × (1 − n × D ) ⎜1 − ⎜ n× f ×C × R SW X O ⎝ ⎞ ⎟ ⎟ ⎠ (21) For this example, the overall ramp signal is found to be 0.63 V. CURRENT-LIMIT SET POINT To select the current-limit set point, first find the resistor value for RLIM. The current limit threshold for the ADP3168 is set with a 3 V source (VLIM) across RLIM with a gain of 10.4 mV/µA (ALIM). RLIM can be found using the following: RLIM = ALIM × VLIM I LIM × RO (22) With the multimode feedback structure of the ADP3168, the feedback compensation must be set to make the converter’s output impedance, working in parallel with the output decoupling, meet this goal. There are several poles and zeros created by the output inductor and decoupling capacitors (output filter) that need to be compensated for. A type-three compensator on the voltage feedback is adequate for proper compensation of the output filter. The expressions given in Equations 25 to 29 are intended to yield an optimal starting point for the design; some adjustments may be necessary to account for PCB and component parasitic effects (see the Tuning Procedure for the ADP3168). Rev. B | Page 19 of 24 ADP3168 The first step is to compute the time constants for all of the poles and zeros in the system: RE = n × RO × AD × RDS + RL × VRT 2 × L × (1 − n × D ) × VRT + VDID n × C X × RO × VVID RE = 3 × 1.3 mΩ + 5 × 4.2 mΩ + TA = C X × (RO − R ') + 1.6 mΩ × 0.63 V 2 × 600 nH × (1 − 0.375) × 0.63 V + 37.9 mΩ 1. 5 V 3 × 6.56 mF × 1.3 mΩ × 1.5 V L X RO − R ' 375 pH 1.3 mΩ − 0.6 mΩ × = 6.56 mF × (1.3 mΩ − 0.6 mΩ ) + × = 4.79 µs RO RX 1.3 mΩ 1.0 mΩ TB = (R X + R '− RO ) × C X = (1.0 mΩ + 0.6 mΩ − 1.3 mΩ ) × 6.56 mF = 1.97 µs ⎛ A × RDS VRT × ⎜⎜ L − D 2 × f SW ⎝ TC = VVID × RE TD = (25) (26) (27) ⎞ ⎛ 5 × 4.2 mΩ ⎞ ⎟ 0.63 V × ⎜ 600 nH = ⎟ ⎟ ⎜ 2 × 267 kHz ⎟⎠ ⎝ ⎠= = 6.2 µs 1.5 V × 37.9 mΩ (28) 2 C X × C Z × RO2 6.56 mF × 230 µF × (1.3 mΩ ) = = 521 ns C X × (RO − R') + C Z × RO 6.56 mF × (1.3 mΩ − 0.6 mΩ ) + 230 µF × 1.3 mΩ (29) where, for the ADP3168, R' is the PCB resistance from the bulk capacitors to the ceramics and where RDS is the total low side MOSFET ON resistance per phase. For this example, AD is 5, VRT equals 0.63 V, R' is approximately 0.6 mΩ (assuming a 4-layer motherboard), and LX is 375 pH for the eight Al-Poly capacitors. Figure 13 shows the typical transient response using the compensation values. The compensation values can be solved using the following: CA = n × RO × TA RE × RB (30) RA = TC 6.2 µs = = 16.7 kΩ C A 371 pF (31) CB = TB 1.97 µs = = 1.48 nF RB 1.33 kΩ (32) C FB = TD 521 ns = = 31.2 pF R A 16.7 kΩ 03258-B-013 3 × 1.3 mΩ × 4.79 µs = 371 pF CA = 37.9 mΩ × 1.33 kΩ (33) Choosing the closest standard values for the components yields C A = 390 pF, R A = 16.9 kΩ, C B = 1.5 nF, C FB = 33 pF Rev. B | Page 20 of 24 Figure 13. Typical Transient Response for Design Example ADP3168 CIN SELECTION AND INPUT CURRENT DI/DT REDUCTION TUNING PROCEDURE FOR THE ADP3168 1. In continuous inductor current mode, the source current of the high-side MOSFET is approximately a square wave with a duty ratio equal to n × VOUT/VIN and an amplitude of one-nth of the maximum output current. To prevent large voltage transients, a low ESR input capacitor sized for the maximum rms current must be used. The maximum rms capacitor current is given by DC Loadline Setting 3. 1 −1 N ×D 4. (34) I CRMS 1 = 0.125 × 65 A × − 1 = 10.5 A 3 × 0.125 5. Note that the capacitor manufacturer’s ripple current ratings are often based on only 2,000 hours of life. This makes it advisable to further derate the capacitor or choose a capacitor rated at a higher temperature than required. Several capacitors may be placed in parallel to meet size or height requirements in the design. In this example, the input capacitor bank is formed by three 2,200 µF, 16 V Nichicon capacitors with a ripple current rating of 3.5 A each. 6. 7. RPH ( NEW ) = RPH (OLD ) × To reduce the input current di/dt to a level below the recommended maximum of 0.1 A/µs, an additional small inductor (L > 1 µH @ 15 A) should be inserted between the converter and the supply bus. That inductor also acts as a filter between the converter and the primary power source. VNL − VFLCOLD VNL − VFLHOT (37) RCS1(OLD ) × RTH ( 25°C ) + (RCS 1(OLD ) − RCS 2( NEW ) ) × (RCS 1(OLD ) − RTH (25°C ) ) 100 90 80 70 60 50 40 30 20 10 0 0 10 20 (36) Repeat Steps 6 and 7 to check loadline and repeat adjustments if necessary. 9. Once complete with dc loadline adjustment, do not change RPH, RCS1, RCS2, or RTH for rest of procedure. 10. Measure output ripple at no-load and full-load with scope and make sure it is within specifications. 1 RCS1(OLD ) + RTH (25°C ) RCS 2( NEW ) = ROMEAS RO 8. (35) EFFICIENCY (%) RCS2( NEW ) = RSC 2(OLD ) × Measure output voltage at no-load (VNL). Verify that it is within tolerance. Measure output voltage at full-load cold (VFLCOLD). Let board set for ~10 minutes at full-load and measure output (VFLHOT). If there is a change of more than a couple of millivolts, adjust RCS1 and RCS2 using Equations 35 and 37. Repeat Step 4 until cold and hot voltage measurements remain the same. Measure output voltage from no-load to full-load using 5 A steps. Compute the loadline slope for each change and then average to get overall loadline slope (ROMEAS). If ROMEAS is off from RO by more than 0.05 mΩ, use the following to adjust the RPH values: 30 40 50 60 OUTPUT CURRENT (A) Figure 14. Efficiency of the Circuit of Figure 11 vs. Output Current Rev. B | Page 21 of 24 03258-B-014 I CRMS = D × I O × 2. Build circuit based on compensation values computed from design spreadsheet. Hook up dc load to circuit, turn on, and verify operation. Also check for jitter at no-load and full-load. − 1 RTH (25°C ) ADP3168 AC Loadline Setting VACDRP VDROOP VTRAN1 Figure 16. Transient Setting Waveform 20. If both overshoots are larger than desired, try making the adjustments described below. (Note: If these adjustments do not change the response, you are limited by the output decoupling.) Check the output response each time you make a change as well as the switching nodes (to make sure the response is still stable). VDCDRP Figure 15. AC Loadline Waveform 16. If the VACDRP and VDCDRP are different by more than a few millivolts, use Equation 38 to adjust CCS. Parallel different values to get the right one because there are limited standard capacitor values available. (Make sure that there are locations for two capacitors in the layout for this.) V ACDRP V DCDRP a. Make ramp resistor larger by 25% (RRAMP). b. For VTRAN1, increase CB or increase switching frequency. c. For VTRAN2, increase RA and decrease CA by 25%. 21. For load release (see Figure 17), if VTRANREL is larger than VTRAN1 (see Figure 16), there is not enough output capacitance. You will either need more capacitance or have to make the inductor values smaller. (If you change inductors, you will need to start the design over using the spreadsheet and this tuning procedure.) (38) 17. Repeat Steps 11 to 13, making adjustments if necessary. Once complete, do not change CCS again in the procedure. 18. Set dynamic load step to maximum step size (do not use a step size larger than needed) and verify that the output waveform is square (which means VACDRP and VDCDRP are equal).Make sure load step slew rate and turn-on are set for a slew rate of ~150 A/µs to 250 A/µs (for example, a load step of 50 A should take 200 ns to 300 ns) with no overshoot. Some dynamic loads have an excessive turn-on overshoot if a minimum current is not set properly. (This is an issue if using a VTT tool.) Initial Transient Setting 19. With dynamic load still set at maximum step size, expand scope time scale to see 2 µs/div to 5 µs/div. The waveform may have two overshoots and one minor undershoot (see Figure 16). Here, VDROOP is the final desired value. VTRANREL VDROOP B-017 C CS ( NEW ) = C CS(OLD ) × VTRAN2 B-016 11. Remove dc load from circuit and hook up dynamic load. 12. Hook up scope to output voltage and set to dc coupling with time scale at 100 µs/div. 13. Set dynamic load for a transient step of about 40 A at 1 kHz with 50% duty cycle. 14. Measure output waveform (may have to use dc offset on scope to see waveform). Try to use vertical scale of 100 mV/div or finer. 15. This waveform should look something like Figure 15. Use the horizontal cursors to measure VACDRP and VDCDRP as shown. Do not measure the undershoot or overshoot that happens immediately after the step. Figure 17. Transient Setting Waveform Because the ADP3168 turns off all of the phases (switches inductors to ground), there is no ripple voltage present during load release. Thus, you do not have to add headroom for ripple, allowing your load release VTRANREL to be larger than VTRAN1 by the amount of ripple and still meet specifications. If VTRAN1 and VTRANREL are less than the desired final droop, this implies that capacitors can be removed. When removing capacitors, check the output ripple voltage as well to make sure it is still within specifications. Rev. B | Page 22 of 24 ADP3168 LAYOUT AND COMPONENT PLACEMENT The following guidelines are recommended for optimal performance of a switching regulator in a PC system. Key layout issues are illustrated in Figure 18. 12V CONNECTOR SWITCH NODE PLANES The output capacitors should be connected as close as possible to the load (or connector) that receives the power (e.g., a microprocessor core). If the load is distributed, the capacitors should also be distributed and generally in proportion to where the load tends to be more dynamic. Avoid crossing signal lines over the switching power path loop, as described next. INPUT POWER PLANE POWER CIRCUITRY The switching power path should be routed on the PCB to encompass the shortest possible length in order to minimize radiated switching noise energy (i.e., EMI) and conduction losses in the board. Failure to take proper precautions often results in EMI problems for the entire PC system as well as noise-related operational problems in the power converter control circuitry. The switching power path is the loop formed by the current path through the input capacitors and the power MOSFETs including all interconnecting PCB traces and planes. Using short and wide interconnection traces is critical in this path because it minimizes the inductance in the switching loop, which can cause high energy ringing, and it accommodates the high current demand with minimal voltage loss. THERMISTOR KEEP-OUT AREA OUTPUT POWER PLANE KEEP-OUT AREA KEEP-OUT AREA KEEP-OUT AREA 03258-B-018 CPU SOCKET Figure 18. Layout Recommendations GENERAL RECOMMENDATIONS For good results, a PCB with at least four layers is recommended. This should allow the needed versatility for control circuitry interconnections with optimal placement, power planes for ground, input, and output power, and wide interconnection traces in the rest of the power delivery current paths. Keep in mind that each square unit of 1 ounce copper trace has a resistance of ~0.53 mΩ at room temperature. Whenever high currents are routed between PCB layers, vias should be used liberally to create several parallel current paths so that the resistance and inductance introduced by the current paths is minimized and the via current rating is not exceeded. If critical signal lines (including the output voltage sense lines of the ADP3168) must cross through power circuitry, it is best if a signal ground plane can be interposed between those signal lines and the traces of the power circuitry. This creates a shield to minimize noise injection into the signals at the expense of making signal ground a bit noisier. Whenever a power dissipating component (e.g., a power MOSFET) is soldered to a PCB, the liberal use of vias, both directly on the mounting pad and immediately surrounding it, is recommended. Two important reasons for this are improved current rating through the vias and improved thermal performance from vias extended to the opposite side of the PCB, where a plane can more readily transfer the heat to the air. Make a mirror image of any pad being used to heat sink the MOSFETs on the opposite side of the PCB to achieve the best thermal dissipation to the air around the board. To further improve thermal performance, use the largest possible pad area. The output power path should also be routed to encompass a short distance. The output power path is formed by the current path through the inductor, the output capacitors, and the load. For best EMI containment, a solid power ground plane should be used as one of the inner layers extending fully under all the power components. SIGNAL CIRCUITRY An analog ground plane should be used around and under the ADP3168 as a reference for the components associated with the controller. This plane should be tied to the nearest output decoupling capacitor ground and should not be tied to any other power circuitry to prevent power currents from flowing in it. The output voltage is sensed and regulated between the FB pin and the FBRTN pin, which connects to the signal ground at the load. To avoid differential mode noise pickup in the sensed signal, the loop area should be small. Thus the FB and FBRTN traces should be routed adjacent to each other on top of the power ground plane back to the controller. The components around the ADP3168 should be located close to the controller with short traces. The most important traces to keep short and away from other traces are the FB and CSSUM pins. See Figure 18 for details on layout for the CSSUM node. Connect the feedback traces from the switch nodes as close as possible to the inductor. The CSREF signal should be connected to the output voltage at the nearest inductor to the controller. Rev. B | Page 23 of 24 ADP3168 OUTLINE DIMENSIONS 9.80 9.70 9.60 28 15 4.50 4.40 4.30 6.40 BSC 1 14 PIN 1 0.65 BSC 0.15 0.05 COPLANARITY 0.10 0.30 0.19 1.20 MAX SEATING PLANE 0.20 0.09 8° 0° 0.75 0.60 0.45 COMPLIANT TO JEDEC STANDARDS MO-153AE Figure 19. 28-Lead Thin Shrink Small Outline Package [TSSOP] (RU-28) Dimensions shown in millimeters ORDERING GUIDE Model ADP3168JRU-REEL7 ADP3168JRU-REEL ADP3168JRUZ-REEL1 1 Temperature Range 0°C to 85°C 0°C to 85°C 0°C to 85°C Package Options RU-28 (TSSOP-28) RU-28 (TSSOP-28) RU-28 (TSSOP-28) Z = Pb-free part. © 2004 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. C03258-0-12/04(B) Rev. B | Page 24 of 24 Quantity per Reel 1000 2500 2500