ISL8102EVAL1: Two Phase Buck Converter with Integrated High Current 5-12V Drivers ® Application Note October 31, 2005 AN1212.0 Author: Nasser A. Ismail Introduction The progress in many parts of modern power systems such as DDR/Chipset core voltage regulators, high current low voltage DC/DC converters, FPGA/ASIC DC/DC converters and many other general purpose applications keeps challenging the power management IC makers to come up with innovative products and new solutions to meet the increase in power, reduction in size and increase in the DC/DC converter’s efficiency targets. The interleaved multiphase synchronous buck topology proves again to be the topology of choice for such high current low voltage applications. The ISL8102 is a space-saving, cost-effective solution for such applications. The ISL8102 is a two-phase PWM control IC with integrated high current MOSFET drivers. The integration of 5-12V high current MOSFET drivers into the controller IC marks a departure from the separate PWM controller and driver configuration of previous multi-phase product families. By reducing the number of external parts, this integration allows for a cost and space saving power management solution. Output voltage can be programmed using the on-chip DAC or an external precision reference. A two bit code programs the DAC reference to one of 4 possible values (0.6V, 0.9V, 1.2V and 1.5V). A unity gain, differential amplifier is provided for remote voltage sensing, compensating for any potential difference between remote and local grounds. The output voltage can also be offset through the use of single external resistor. An optional droop function is also implemented and can be disabled for applications having less stringent output voltage variation requirements or experiencing less severe step loads. A unique feature of the ISL8102 is the combined use of both DCR and rDS(ON) current sensing. Load line voltage positioning and overcurrent protection are accomplished through continuous inductor DCR current sensing, while rDS(ON) current sensing is used for accurate channel-current balance. Using both methods of current sampling utilizes the best advantages of each technique. Protection features of this controller IC include a set of sophisticated overvoltage and overcurrent protection. Overvoltage results in the converter turning the lower MOSFETs ON to clamp the rising output voltage and protect the load. An OVP output is also provided to drive an optional crowbar device. The overcurrent protection level is set through a single external resistor. Other protection features include protection against an open circuit on the remote sensing inputs. Combined, these features provide advanced protection for the output load. 1 The ISL8102EVAL1 evaluation board embodies a 55-60A regulator solution targeted at supplying power to the designated load. The physical board design is optimized for 2 phase operation and ships out configured to provide one of the following four output voltages (0.6V, 0.9V, 1.2V and 1.5V) depending on choice of the REF1, REF0 combination set by DIP switch U2, but can be easily modified to provide any output voltage values in the range of 0.6-2.3V by means of resistor divider composed of R90, R81. For further details on the ISL8102, consult the data sheet [1]. The Intersil multiphase family controller and driver portfolio continues to expand with new selections to better fit our customer’s needs. Refer to our web site for updated information: www.intersil.com. ISL8102EVAL Board Design The evaluation kit consists of the ISL8102EVAL1 evaluation board, the ISL8102 data sheet, and this application note. The evaluation board is optimized for two phase operation without droop, the nominal output voltage is 1.5V (with DIP switch U2 set to 11 position) and the maximum output current is 60A. The evaluation board provides convenient test points, a DIP switch for DAC (REF) voltage selection from four possible values (0.6V, 0.9V, 1.2V and 1.5V), footprint for a resistor divider for output voltage adjustment up to 2.3V, and an onboard transient load generator to facilitate the evaluation process. An on board LED is present to indicate the status of the PGOOD signal. The board is configured for down conversion from 5-12V to the REF setting. The printed circuit board is implemented in 6-layer, 2-ounce copper. Layout plots and part lists are provided at the end of the application note for this design. Quick Start Evaluation The ISL8102EVAL1 is designed for quick evaluation after following only a few simple steps. All that is required is two bench power supplies, Oscilloscope and Load. To begin evaluating the ISL8102EVAL1 follow the steps below. 1. Before doing anything to the evaluation board, make sure the “Enable” switch (S1) and the “Transient Load Generator” switch (S2) are both in the ON position corresponding to the converter being disabled and the transient load generator being turned off. 2. Connect a 12V, 10A Lab power supply between J7 and J8, this Power supply provides VIN and PVCC (with original board configuration). CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc. Copyright © Intersil Americas Inc. 2005. All Rights Reserved All other trademarks mentioned are the property of their respective owners. Application Note 1212 3. Connect a 5V, 1A Lab power supply between J23 and GND, this Power supply provides VCC bias (and PVCC bias if the board is configured for 5V PVCC). 4. Set the “REF Selection” DIP switch (U2) to 11 corresponding to DAC = REF = 1.5V. Figure 1 details the typical default configuration for U2 when the board is received. In this default setting, the evaluation board is set for a reference voltage of 1.5V. 2 1 Logic 1 0 REF0 REF1 ON FIGURE 1. TYPICAL U2 DEFAULT SETTING 11 (1.500V) 5. Connect a load (either resistive or electronic) between VOUT terminal (J1, J2) and GND terminal (J3, J4). 6. Move the “Enable” switch (S1) to the OFF position releasing the IC ENLL pin to rise to begin regulation. After step 6, the ISL8102EVAL1 should be regulating the output voltage, at the “VOUT+” and “VOUT-” test points (P20, P21) and J5 to the REF voltage. The “PGOOD Indicator” LED (D1) should be green to indicate the regulator is operating correctly. ISL8102EVAL1 Board Features Input Power Connections The ISL8102EVAL1 allows for the use of standard bench power supplies for powering up the board. Two femalebanana jacks are provided for connection of bench power supplies. Connect the +5V terminal to P23, +12V terminal to J7, and the common ground to terminal J8. Voltage sequencing is not required when powering the evaluation board. Once power is applied to the board, the PGOOD LED indicator will begin to illuminate red. With S1 in the ON position, the ENABLE input of the ISL8102 is held low and the startup sequence is inhibited. Output Power Connections The ISL8102EVAL1 output can be exercised using either resistive or electronic loads. Copper alloy terminal lugs provide connection points for loading. Tie the positive load connection to VOUT, terminals J1 and J2, and the negative to ground, terminals J3 and J4. A shielded scope probe test point, J5, allows for inspection of the output voltage, VOUT. REF and VOUT Setup The REF DIP switch would be preset to 11 (1.500V). Also 1.2V, 0.9V and 0.6V outputs can be selected using different codes on the DIP switch. If an output voltage different than the 4 possible REF values is desired, the output resistor divider composed of R90 (initially 0Ω) and R81 (initially 2 open) can be used (consult Data sheet and the section entitled Adjusting the Output Voltage at the end of this application note for resistor value calculations). Note that the ISL8102 is usable for output voltages up to 2.3V when the REF voltage is set to 1.5V. See Table 1 below for the maximum possible voltages with different REF setting. TABLE 1. MAXIMUM OUTPUT VOLTAGE WITH DIFFERENT REF SETTING WITH THE USE OF A RESISTOR DIVIDER ON VSEN REF1 REF0 REF = DAC VOUT MAX 0 0 0.6V 1.4V 0 1 0.9V 1.7V 1 0 1.2V 2.0V 1 1 1.5V 2.3V PVCC Power Options One unique feature of the ISL8102 is the variable gate drive bias for the integrated drivers. The gate drive voltage for the internal drivers can be any voltage from +5V to +12V by simply connecting the desired voltage to the PVCC pin of the controller. To accommodate the flexibility of the drivers, the ISL8102EVAL1 has been designed to support a multitude of options for the PVCC voltage. Switching between the different PVCC voltages available on the evaluation board is as simple as populating and depopulating certain resistors. The eval board has three on board voltages available: +5V, +12V, and +8V (from an on board linear regulator). Refer to Table 2 for what resistors to populate for each voltage option. TABLE 2. GATE DRIVE VOLTAGE OPTIONS AND RESISTOR SETTINGS UGATE VOLTAGE LGATE VOLTAGE R48 R68 R71 R72 12.0V 12.0V OPEN OPEN OPEN 0Ω 8.0V 8.0V OPEN OPEN 0Ω OPEN 5.0V 5.0V 0Ω OPEN OPEN OPEN 12.0V 5.0V 0Ω 0Ω OPEN OPEN Enabling the Controller In order to enable the controller, the board must be powered, a REF (DAC) code must be set, and the PVCC and VCC voltages must be set. If these steps have been properly followed, the regulator is enabled by toggling the “ENABLE” switch (S1) to the OFF position. When S1 is switched OFF, the voltage on the ENLL pin of the ISL8102 will rise above the ENLL threshold of 0.66V and the controller will begin its digital soft start sequence. The output voltage ramps up to the programmed setting, at which time the PGOOD indicator will switch from red to green. AN1212.0 October 31, 2005 Application Note 1212 On-Board Load Transient Generator Most bench-top electronic loads are not capable of producing the current slew rates required to emulate most modern loads. For this reason, a discrete transient load generator is provided on the evaluation board, see Figure 2. The generator produces a load pulse of 500µs in duration with a period of 27ms. The pulse magnitude is approximately 25A with rise and fall slew rates of approximately 50A/µs as configured. The short load current pulse and long duty cycle is required to limit the power dissipation in the load resistors (R38-R42) and MOSFETs (Q20, Q21). To engage the load generator simply place switch S2, in the “OFF” position. If the DAC code is changed from 11 (1.500V), the transient generator dynamics must be adjusted relative to the new output voltage level. Place a scope probe in J10 to measure the voltage across the load resistors and the dV/dt across them as well. Adjust the load resistors, R38-R40, to achieve the correct load current level. Change resistors R34-R37 to increase or decrease the dV/dt as required to match the desired dI/dt profile. HO LO HI HS VSS U2 R33 ON 402Ω OFF C57 22µF Q19 2N7002 S3 BAV99LT1 S4 R35 R39 OPEN R40 OPEN 249Ω R37 562kΩ R36 Q21 SUD50N03 BAV99LT1 R38 OPEN S2 562Ω R34 VOUT Q20 SUD50N03 R41 0.12Ω R42 0.12Ω C L ISL8102 DCR INDUCTOR I VOUT COUT L FIGURE 3. DCR STATIC CURRENT SENSE CIRCUIT In order to not affect the rest of the regulator, the time constant of this R-C circuit is very large, so it can only be used to measure static current, and not transient currents. To calculate the current through each inductor measure the voltage across the “DCR SENSE” points on the ISL8102EVAL1 and then divide that number by the DCR of the inductor. This should give you an accurate reading of the current through each channel during static loads. The ISL8102 uses lower MOSFET rDS(ON) current sensing to measure the current through each channel and balance them accordingly. If the lower MOSFETs on the ISL8102EVAL1 are changed, the current balance resistors, R18 and R20, should also be changed to adjust for the change in rDS(ON). Refer to the Current Balancing Component Selection section the ISL8102 data sheet to choose new current sense resistors. R18 adjusts the current in channel 1, R20 adjusts the currents in channel 2. These resistors can also be changed to adjust for any current imbalance due to layout, which is also explained in the ISL8102 data sheet. Load Line (Droop) Regulation The ISL8102 has an optional Droop function, the ISL8102EVAL1 board design is optimized for no Droop case. For Droop option selection follow the following Table 3. 249Ω TABLE 3. SELECTION OF DROOP OPTION J10 VLOAD FIGURE 2. LOAD TRANSIENT GENERATOR Inductor DCR Static Current Sense Points A unique feature of the ISL8102EVAL1 is the ability to measure the voltage drop across the DCR of each channel’s inductor by multimeter. This is accomplished with the use of a capacitor and resistor series circuit which is placed in parallel across each inductor as illustrated in Figure 3. When 3 R VIN Current Balance Resistors HIP2100 R30 46.4kΩ DCR + SENSE - Modifying the ISL8102EVAL1 Design LI VDD C55 1µF HB VCC12 current, IL, flows through the inductor, the voltage drop developed across the DCR will be sensed by the R-C circuit, and an equivalent voltage will be developed across the capacitor C. DROOP R46 R45 Disabled (Droop connected IREF) 0Ω OPEN Enabled (Droop connected ICOMP) OPEN 0Ω If Droop is implemented, the compensation network will need to be recalculated for the Droop case for optimal loop response and stability. AN1212.0 October 31, 2005 Application Note 1212 To create an output voltage change proportional to the total current in all the active channels (droop), the ISL8102 uses an inductor DCR sensing R-C network. This network, shown in Figure 4, is designed not only to precisely control the load line of the regulator, but also to thermally compensate for any changes in DCR that may skew the load line as a result of increases in temperature. L2 PHASE1 IOUT VOUT R15 L3 PHASE2 R12 DNP R16 ISUM R9 DNP R14 R10 0Ω C7 10nF ICOMP RCOMP IREF OPTIONAL NTC RESISTOR NETWORK ISL8102 * R15, R16 ARE EQUAL FIGURE 4. DCR SENSING CONFIGURATION WITH OPTIONAL FOOTPRINTS FOR NTC LOAD-LINE COMPENSATION This sensing technique works off the principle that if the R-C time constant of C7*RCOMP (RCOMP = R14) is equal to the L/DCR time constant of the inductor, the load line impedance will be equal to RCOMP*DCR/R15 (R15 and R16 are equal). If the load line impedance needs to be changed, all that is required is adjusting the values of R15-R16 as explained in the Load Line Regulation Component Selection (DCR Current Sensing) section of the ISL8102 data sheet. If the Inductor is changed though, the resistance of time constant matching network will need to be changed. The NTC resistor network must first be adjusted so that the new L/DCR time constant is precisely matched. Refer to the ISL8102 data sheet to design the entire R-C sense network. An Optional NTC resistor network consisting of 3 resistors (R9, R10, and R14) and a single NTC thermistor (R12), which is placed close the output inductors. This network is designed to compensate for any change in DCR that occurs due to temperature when the Droop option is used and a tight load-line regulation is required, and keep the time constant of the R-C network equal to that of the inductor L/DCR time constant. 4 Overcurrent Protection Level The ISL8102 utilizes a single resistor to set the maximum current level for the IC’s overcurrent protection circuitry. Please refer to the Overcurrent Protection section of the ISL8102 data sheet, and adjust resistor R11 accordingly to set the desired overcurrent trip level. Output Voltage Offset The ISL8102 allows a designer to accurately offset the output voltage both negatively and positively. All that is required is a single resistor between the OFS and VCC pins, or the OFS pin and GND. The ISL8102EVAL1 has both of these resistor options available on the board. To positively offset the output voltage populate resistor R5. To negatively offset the output voltage populate resistor R7. Please refer to the Output Voltage Offset Programming section of the ISL8102 data sheet to accurately calculate these resistor values. Switching Frequency The switching frequency of the ISL8102EVAL1 board is set to an optimal value of 400kHz giving the best efficiency and performance for the given design with R13 = 60.4kΩ. However, the switching frequency can be adjusted anywhere from 80kHz to 1.5MHz per phase. In practice many factors affect the choice of switching frequency among which are efficiency, and gate drive losses (which depend on the MOSFET choice and Gate Driver Voltage). Since the ISL8102 has integrated MOSFET drivers, the driver losses must be taken into account when the switching frequency is chosen. To change the switching frequency refer to the Switching Frequency section of the ISL8102 data sheet and adjust the value of frequency set resistor, R13, accordingly. MOSFET Gate Drive Voltage (PVCC) The gate drive bias voltage of the integrated drivers in the ISL8102 can be any voltage between +5V and +12V. This bias voltage is set by connecting the desired voltage to the PVCC pins of the IC. Please refer to the PVCC Power Options section to set the desired gate drive voltage. Number of Active Phases The ISL8102 has the option of 1 or 2-phase operation. The ISL8102EVAL1 is designed to change the number of active phases by simply populating or depopulating one or two resistors, R26 and R29. Refer to Table 4 for which resistors to populate for 1 or 2-phase operation. TABLE 4. SETTINGS FOR NUMBER OF ACTIVE PHASES # OF ACTIVE PHASES R26 R29 2 0Ω OPEN 1 OPEN 0Ω AN1212.0 October 31, 2005 ISL8102 Performance GND> ICORE, 0A ICC12 0A 0V 1ms/DIV ENLL 2.0V/DIV 10.0A/DIV 50.0A/DIV VOUT 0V 1.0V/DIV PHASE2 PHASE2 GND> VOUT VOUT GND> ENLL ENLL 2.00V/DIV The typical start-up waveforms for the ISL8102EVAL1 are shown in Figure 5. The waveforms represented in this image show the soft-start sequence of the regulator DAC set to 1.50V. Before the soft-start interval begins, VCC and PVCC are above POR and the DAC is set to 11. With these two conditions met, throwing the ENABLE switch into the OFF position causes the voltage on the ENLL pin to rise above the ISL8102’s enable threshold, beginning the soft-start sequence. For a delay time of 0.6ms, VOUT does not move due to the manner in which soft-start is implemented within the controller. After this delay (which is approximately equal to 240 switching cycles), VOUT begins to ramp linearly toward the DAC voltage. With the converter running at 400kHz, this ramp takes approximately 4.3ms, during which time the input current, ICC12, also ramps slowly due to the controlled building of the output voltage. PHASE1 PHASE1 1ms/DIV 1ms/DIV GND> T1 T0 T2 FIGURE 6. ISL8102EVAL1 START-UP INTO A PARTIALLY CHARGED OUTPUT (VDAC = 1.200V) A second scenario can be encountered with a pre-charged output: output being pre-charged above the DAC-set point, as shown in Figure 7. In this situation, the ISL8102 behaves in a way similar to that of Figure 6, keeping the MOSFETs off until the end of the SS ramp. However, once the end of the ramp has been reached, at time T1, the output drivers are enabled for operation, and the output is quickly drained down to set-point level. 500mV/DIV 10.00V/DIV 10.00V/DIV Soft-Start Interval 500mV/DIV 10.00V/DIV 10.00V/DIV Application Note 1212 PHASE1 GND> PHASE2 GND> VOUT FIGURE 5. SOFT-START INTERVAL WAVEFORMS Special consideration is given to start-up into a pre-charged output (where the output is not 0V at the time the SS cycle is initiated). Under such circumstances, the ISL8102 keeps off both sets of output MOSFETs until the internal ramp starts to exceed the output voltage sensed at the FB pin. This special scenario is detailed in Figure 6. The circuit is enabled at time T0. As the internal ramp exceeds the magnitude of the output voltage at time T1, the MOSFETs drivers are enabled and the output voltage ramps up in a seamless fashion from the preexistent level to the DAC-set level, reached at time T2. 5 GND> ENLL 2.00V/DIV Once VOUT reaches the DAC set point, the internal pulldown on the PGOOD pin is released. This allows a resistor from PGOOD to VCC to pull PGOOD high and the PGOOD LED indicator changes from red to green. 1ms/DIV GND> T0 T1 FIGURE 7. ISL8102EVAL1 START-UP INTO AN OVERCHARGED OUTPUT (VDAC = 1.200V) An OV condition during start-up will take precedence over this normal start-up behavior, but will allow reversal back to normal behavior as soon as the condition is removed or brought under control. AN1212.0 October 31, 2005 Application Note 1212 Transient Response The rising edge transient response of the ISL8102EVAL1, is shown in Figure 8. In order to obtain the load current waveform shown, the bench-top load is turned off, while the on-board transient generator is pulsing a 25A step for 500µs. When the load step occurs, the output capacitors provide the initial output current, causing VOUT to drop suddenly due to the ESR and ESL voltage drops in the capacitors. The controller immediately responds to this drop by increasing the PWM duty cycles to as much as 66%. The duty cycles then decrease to stabilize VOUT. 20mV/DIV 500mV/DIV V(ILOAD) GND> 5µs/DIV FIGURE 9. ISL8102EVAL1 FALLING EDGE TRANSIENT RESPONSE Figure 10 shows both the rising and falling edges. VOUT 500mV/DIV 20mV/DIV VOUT VOUT 20mV/DIV The ISL8102EVAL1 is designed without droop for a maximum output load current of 60A. The Load step is approximately 25A and the output voltage variation during the transient is kept below 100mV peak to peak. This load step have a maximum slew rate of approximately 50A/µs on both the rising and falling edges. The on-board load transient generator is designed to provide the specified load step, different load steps and current slew rates can be accommodated with the on Board Transient load generator. V(ILOAD) GND> 500mV/DIV V(ILOAD) 5µs/DIV GND> 100µs/DIV FIGURE 10. ISL8102EVAL1 TRANSIENT RESPONSE Overcurrent Protection FIGURE 8. ISL8102EVAL1 RISING RDGE TRANSIENT LOAD RESPONSE At the end of the 500µs load pulse, the load current returns to 0A. The transient response to this falling edge of the load is shown in Figure 9. When the falling load step occurs, the output capacitors must absorb the inductor current which can not fall at the same rate of the load step. This causes VOUT to rise suddenly due to the ESR and ESL voltage drops in the capacitors. The controller immediately responds to this rise by decreasing the PWM duty cycles to zero, and then increasing them accordingly to regulate VCORE to the programmed 1.5V level. 6 The ISL8102 is designed to stop all regulation and protect the sensitive Load if an overcurrent event occurs. This is done by continuously monitoring the total output current and comparing it to an overcurrent trip level set by the OCSET resistor, R11. If the output current ever exceeds the trip level as shown in Figure 11 (at time T1), the ISL8102 immediately turns the upper and lower MOSFETs off, causing VOUT to fall to 0V. The controller holds the UGATE and LGATE signals in this state for a period of 4096 switching cycles, which at 400kHz is 10.25ms. The controller then re-initializes the soft-start cycle (at time T2). If the load that caused the overcurrent trip remains, another overcurrent trip will occur before the soft-start cycle completes. The controller will continue to try to cycle soft-start indefinitely until the load current is reduced, or the controller is disabled. This operation is shown in Figure 11. AN1212.0 October 31, 2005 Application Note 1212 Overvoltage Protection 20.00V/DIV UGATE2 GND> PGOOD COMP GND> GND> VOUT 2ms/DIV T1 T2 500mV/DIV 1.00V/DIV 5.00V/DIV GND> FIGURE 11. ISL8102EVAL1 OVERCURRENT PROTECTION To protect from an overvoltage event during normal operation, the ISL8102 continually monitors the output voltage. If the output voltage exceeds a specific limit (set internally), the controller commands the LGATE signals high, turning on the lower MOSFETs to keep the output voltage below a level that might cause damage to the Load. As shown in the overvoltage event in Figure 12, turning on the lower MOSFETs not only keeps the output voltage from rising, it also sinks a large amount of current, causing the input voltage to the power stage to drop. If this causes the input power supply voltage to fall below the POR level of the ISL8102, as seen at the end of the waveform in Figure 13, the controller responds by using the pre-POR overvoltage protection explained in the previous section. This allows the ISL8102 to always keep the output load safe from high voltage spikes during an entire overvoltage event. LGATE2 Vin = PVCC GND> GND> 1ms/DIV 1.0V/DIV GND> GND> 1.0V/DIV 1.0V/DIV VOUT 1.0V/DIV VOUT GND> 10.00V/DIV LGATE2 GND> Vin = PVCC 5.00V/DIV Prior to PVCC and VCC exceeding their POR levels, the ISL8102 is designed to protect the load from any overvoltage events that may occur (such an overvoltage may occur if for example one of the upper MOSFETs was shorted at assembly due to manufacturing defects). This is accomplished by means of an internal 10kΩ resistor tied from PHASE to LGATE, which turns on the lower MOSFET to control the output voltage until the input power supply current limits itself and cuts off. In Figure 12, an artificial prePOR overvoltage event has been created by shorting the positive 12V input plane to the PHASE plane. This same 12V input is connected to PVCC pins of the ISL8102. Figure 12 illustrates how the controller protects the load from a high output voltage spike, when the 12V input turns on, by tying LGATE to PHASE. 2.00V/DIV Pre-POR Overvoltage Protection GND> 1ms/DIV FIGURE 13. ISL8102EVAL1 PRE-POR OVERVOLTAGE PROTECTION Efficiency The efficiency of the ISL8102EVAL1 board, loaded from 0A to 60A, at both PVCC = 5V and 12V are plotted in Figure 14 for VOUT = 1.5V. Measurements were performed at room temperature and taken at thermal equilibrium without any air flow. The efficiency peaks just below 88% at 35A for the PVCC = 12V case and then levels off steadily to approximately 86% at 60A, while for the PVCC = 5V, efficiency peaks at around 89% at 17A and then falls down to approximately 84% at 60A. > T0 T1 T2 FIGURE 12. ISL8102EVAL1 PRE-POR OUTPUT OVERVOLTAGE PROTECTION (START-UP WITH SHORTED UPPER FET) 7 AN1212.0 October 31, 2005 Application Note 1212 The offset pin (OFS) allows for small-range (less than 100mV), positive or negative, offsetting of the output voltage. The board is shipped with R90 equal to 0Ω and R82 is not populated to provide an output voltage equal to the internal DAC setting. Should an output voltage setting outside the normal range provided via the internal DAC be required, a separate resistor divider connected from the load output terminals to VSEN pin as shown in Figure 16 is needed. 90 88 86 EFFICIENCY ( % ) 84 PVCC = 5V PVCC = 12V 82 VOUT = 1.5V Fsw = 400kHz 80 78 76 COMP 70 0 5 10 15 20 25 30 35 40 45 50 55 60 R1 LOAD CURRENT ( A ) 5 R4 6 VDIFF R3 C4 The efficiency for VOUT = 1.8V is plotted in Figure 15. The efficiency peaks just below 89% at 35A for the PVCC = 12V case and then levels off steadily to approximately 87% at 60A, while for the PVCC = 5V, efficiency peaks at around 90% at 20A and then falls down to approximately 86% at 60A. The use of air flow could improve the efficiency across the load range and keeps the components cooler leading to better reliability and longer component lives. ISL8102 C1 FB FIGURE 14. EFFICIENCY vs LOAD CURRENT 4 8 7 VSEN C2 72 RGND 74 VOUT+ RL = R81 RH = R90 VOUT- FIGURE 16. ADJUSTING VOUT OUTSIDE THE REF (DAC) RANGE 92 Use the following relationships to calculate the value of the resistors based on the known parameters. 90 88 EFFICIENCY ( % ) 86 R L + R H ≤ 500Ω PVCC = 5V 84 82 Choose R L = 300Ω PVCC = 12V 80 78 VOUT = 1.8V 76 Fsw = 400kHz Choose a value of VREF that meets the following condition VREF ≥ VOUT – 0.8V 74 72 Choose a value for RL (for example 300Ω) 70 0 5 10 15 20 25 30 35 40 45 50 55 60 LOAD CURRENT ( A ) FIGURE 15. EFFICIENCY vs LOAD CURRENT Calculate the value of the resistor RH R L ⋅ ( VOUT – REF ) R H = ----------------------------------------------------REF Modifications Example: Adjusting the Output Voltage VOUT = 2.1V The output voltage can be adjusted by changing the 2 bit inputs (REF1, REF0) of internal DAC (externally connected with a resistor to REF). Please consult the data sheet for the available voltage ranges and the required settings. VREF ≥ 2.1V – 0.8V ≥ 1.3V VREF = 1.5V R L = 300Ω 300Ω ⋅ ( 2.1V – 1.5V ) R H = -------------------------------------------------------- = 120Ω 1.5V 8 AN1212.0 October 31, 2005 Application Note 1212 Down-Converting From a Different Input Voltage Summary The ISL8102EVAL1 is powered from bench supplies, the input labeled ‘+12V’ can be adjusted down as desired. If experimenting with a lower voltage, be mindful of a few aspects: The ISL8102EVAL1 evaluation board showcases a highly integrated approach to providing control in a wide variety of applications. The sophisticated feature set and high-current MOSFET drivers of the ISL8102 yield a highly efficient power conversion solution with a reduced number of external components in a compact footprint. The following pages provide a board schematic, bill of materials and layout drawings to support implementation of this solution. Refer to the ISL8102 data sheet for detailed layout instructions. • The duty cycle of the controller is limited to 66%; the circuit will not be capable of properly regulating the output voltage should the input be reduced to a level low enough to induce duty cycle saturation. • The input-RMS current will likely increase as the input voltage is decreased; maximum will occur at duty cycles around 25% for the two-phase and to 50% for singlephase. • As the evaluation board (as shipped) was not optimized for high duty cycle operation, closely monitor the board temperatures and increase the output current only as allowed by the board thermal behavior. References Intersil documents are available on the web at www.intersil.com. [1] ISL8102 Data Sheet, Intersil Corporation, File No. FN9247. • The reduced input voltage will decrease the amount of loop gain the modulator provides in the feedback loop, as a result, expect a more sluggish transient response when operating the board at reduced down-conversion voltage. • The Evaluation Board (as shipped) have the +12V is connected as the input to be down-converted and provides gate drive bias (PVCC). Since PVCC can assume any value between +5 and +12V, the Input can be reduced only to 5V. If a lower input voltage is desired, the PVCC voltage should be provided by a separate supply whose value does not drop below +5V. The VCC bias supply can be used in this case (Consult the section entitled PVCC Power Options for more details on how this can be accomplished). 9 AN1212.0 October 31, 2005 ISL8102EVAL1 Schematic 10 Application Note 1212 AN1212.0 October 31, 2005 ISL8102EVAL1 Schematic 11 Application Note 1212 AN1212.0 October 31, 2005 ISL8102EVAL1 Schematic 12 Application Note 1212 AN1212.0 October 31, 2005 ISL8102EVAL1 Schematic 13 Application Note 1212 AN1212.0 October 31, 2005 Application Note 1212 Bill of Materials for ISL8102EVAL1 REFERENCE DESIGNATOR PART NUMBER DESCRIPTION CASE/ FOOTPRINT MANUF. OR VENDOR QTY C34, 37, 40, 41, 44, 4SEPC560M 47, 48 Capacitor, TH, 8x13mm, 560µF, 4V, 20%, 7mΩ 8x13 mm Sanyo 7 DNP (C29, 30, 33) 4SEPC560M Capacitor, TH, 8x13mm, 560µF, 4V, 20%, 7mΩ 8x13 mm Sanyo 0 C57 C3225X5R1A226M Ceramic Capacitor, X5R, 10V, 22µF 1210 TDK 1 C56 GRM188R71H102KA ECJ-1VB1H102K C0603X7R500-102KNE Ceramic Capacitor, X7R, 0603, 50V, 10%, 1000pF 0603 Murata Panasonic Venkel 1 C10 C0603X7R500-332KNE 0603B332K500BT CAPACITOR, SMD, 0603, 3300pF, 50V, 10%, X7R 0603 Venkel BC Components 1 C4 0.027µF Ceramic Cap CAPACITOR, SMD, 0603, 0.027µF, 50V, 10%, X7R 0603 Any 1 C1 ECU-V1H101JCG C0805COG500-101JNE CAPACITOR, SMD, 0805, 100pF, 50V, 5%, NPO 0805 Panasonic Venkel 1 C2, C7 C0805C103K5RACTU 08055C103KAT2A ECJ-2VB1H103K C0805X7R500-103KNE CAPACITOR, SMD, 0805, 0.01µF, 50V, 10%, X7R 0805 Kemet AVX Panasonic Venkel 2 C8, C9, C64 GRM40X7R104K050AD C0805 1-4K5RAC7800 C0805X74500-104KNE CAPACITOR, SMD, 0805, 0.1µF, 50V, 10%, X7R 0805 Murata Kemet Venkel 3 C65 C0805X7R160-105KNE C0805C105K4RAC CAPACITOR, SMD, 0805, 1µF, 16V, 10%, X7R 0805 Venkel Kemet 1 C6, C12 08053D105KAT2A C0805X7R250-105KNE CAPACITOR, SMD, 0805, 1.0µF, 25V, 10%, X5R 0805 AVX Venkel 2 C5 ECJ-2VB1H223K C0805X7R500-223KNE CAPACITOR, SMD, 805, 0.022µF, 50V, 10%, X7R 0805 Panasonic Venkel 1 C14, C16, C55, C67, C68. ECJ-3YB1C105K C1206X7R160-105KNE CAPACITOR, SMD, 1206, 1µF, 16V, 10%, X7R 1206 Panasonic Venkel 5 C24, C25 C1206X7R100-106KNE CAPACITOR, SMD, 1206, 10µF, 10V, 10%, X7R 1206 Venkel Any 2 C38, C39, C42, C43, C45, C46, C49C54, C80-C87 22µF Ceramic CAPACITOR, SMD, 1206, 22µF, 6.3V, 20%, X5R 1206 Any 20 CAPACITOR, SMD, 1210, 100µF, 6.3V, 20%, X5R 1210 TDK Panasonic AVX 4 1210 TDK Murata 2 C27, C28, C31, C32 C3225X5R0J107M ECJ-4YB0J107M 12106D107MAT C21, C22 C3225X5R1C226M CAPACITOR, SMD, 1210, 22µF, 16V, 20%, X5R GRM32ER61C226ME20L C20, C88-C90 16MBZ1800M10X23 CAP, RADIAL, 10x23, 1800µF, 16V, 20%, ALUM. ELEC 10x23 Rubycon Panasonic 4 DNP (L4) 1008PS-153K COIL RF INDUC, SMD, 2.74mm, 15µH, 10%, .6A Coilcraft 0 L2, L3 IHLP-5050FD-01-R47M COIL-PWR INDUCTOR, SMD, 13mm, 0.47µH, 20%, 55A, SHIELDED Vishay 2 J5, J10 131-4353-00 CONN-GEN, SCOPE PROBE TEST PT Tektronix 2 P6, P11, P12, P21, 1514-2 P23-P25, P27, P41 CONN-GEN, TERMINAL POST, TH, 0.09 Keystone 9 J8 CONN-PLUG, BANA-INSUL-SDRLESS, BLK, 4.23mm Mouser 1 164-6218 14 AN1212.0 October 31, 2005 Application Note 1212 Bill of Materials for ISL8102EVAL1 (Continued) REFERENCE DESIGNATOR J7 PART NUMBER 164-6219 DESCRIPTION CASE/ FOOTPRINT MANUF. OR VENDOR QTY CONN-PLUG, BANA-INSUL-SDRLESS, RED, 4.23mm, RA Mouser 1 5002 P1-P5, P7-P10, P13-P15, P17, P18, P20, P22, P26, P28, P29, P42 CONN-GEN, MINI TEST POINT, VERTICAL, WHITE Mouser 20 S7 BAS40-06LT1-T DIODE-SCHOTTKY BARRIER, SMD, SOT-23, 3P, 40V On Semiconductor 1 S3, S4 BAV99TA-T DIODE-SWITCHING, SMD, SOT23, 70V, 0.2A Zetex Inc 2 D2 MBR0540T1-T DIODE-RECTIFIER, SMD, SOD-123, 2P, 40V, 0.5A On Semiconductor 1 DNP (D4, D5) MBR0540T1-T DIODE-RECTIFIER, SMD, SOD-123, 2P, 40V, 0.5A On Semiconductor 0 D1 SSL-LXA3025IGC-TR LED, SMD, 3x2.5mm, 4P, RED/GREEN, 12/20MCD, 2V Lumex 1 L1 T50-8/90-8T-16AWG-1UH CORE, RADIAL, TH, 1.0µH, T50-8/90, 8TURNS, 16AWG Any 1 U3 HIP2100IB IC-HI FREQ BRIDGE DRIVER, 8P, SOIC, 100V Intersil 1 U1 ISL8102IRZ IC-2 PHASE PWM CONTROLLER, 32P, QFN, 5X5, Pb-Free Intersil 1 Q19, Q22 2N7002-T TRANSISTOR, N-CHANNEL, 3LD, SOT-23, 60V, 115mA Any 2 Q23 CZT3019 TRANSISTOR, NPN, 4P, SOT-223, 120V, 1A Central Semiconductor 1 Q1, Q2, Q5, Q6. HAT2168H MOSFET, 30V, 8.8mΩ LFPAK Renesas 4 Q4, Q3, Q7, Q8. HAT2165H MOSFET, 30V, 3.4mΩ LFPAK Renesas 4 Q20, Q21 SUD50N03-07 TRANSIST-MOS, N-CHANNEL, SMD, TO-252, 30V, 20A DPAK Vishay 2 R17 2.4kΩ Resistor, SMD, 0, 1/16W, 5% 0603 Any 1 R94, R95 10Ω Resistor, 10Ω, 1/16W, 5% 0603 Any 2 R6 20Ω Resistor, 20Ω, 1/16W, 5% 0603 Any 1 R10, R26, R46, R90, R91, R93, R104, R105 0Ω Shorting resistor 0603 Any 7 R1, R3 1kΩ Resistor, 1kΩ, 1/16W, 1% 0603 Any 2 R27, R28, R49 10kΩ Resistor, 10kΩ, 1/16W, 1% 0603 Any 3 R31, R50, R53 10.7kΩ Resistor, 10.7kΩ, 1/16W, 1% 0603 Any 3 R32 1.87kΩ Resistor, 1.87kΩ, 1/16W, 1% 0603 Any 1 R43, R44 2.43kΩ Resistor, 2.43kΩ, 1/16W, 1% 0603 Any 1 R34, R36 249Ω Resistor, 249Ω, 1/16W, 1% 0603 Any 2 R33 402Ω Resistor, 402Ω, 1/16W, 1% 0603 Any 1 R30 46.4kΩ Resistor, 46.6kΩ, 1/16W, 1% 0603 Any 1 R14 48.7kΩ Resistor, 48.7kΩ, 1/16W, 1% 0603 Any 1 R100 4.99kΩ Resistor, 4.99kΩ, 1/16W, 1% 0603 Any 1 R4 51.1Ω Resistor, 51.1Ω, 1/16W, 1% 0603 Any 1 R35, R37 562Ω Resistor, 562Ω, 1/16W, 1% 0603 Any 2 R13 60.4kΩ Resistor, 60.4kΩ, 1/16W, 1% 0603 Any 1 15 AN1212.0 October 31, 2005 Application Note 1212 Bill of Materials for ISL8102EVAL1 (Continued) REFERENCE DESIGNATOR PART NUMBER DESCRIPTION CASE/ FOOTPRINT MANUF. OR VENDOR QTY R15, R16 62kΩ Resistor, 62kΩ, 1/16W, 1% 0603 Any 2 R18, R20 750Ω Resistor, 750Ω, 1/16W, 1% 0603 Any 2 R11 887Ω Resistor, 887Ω, 1/16W, 1% 0603 Any 1 R69 10Ω Resistor, 10Ω, 1/10W, 1% 0805 Any 1 R8, R21, R22, R101, R102 0Ω Resistor, 0Ω, 1/10W, 5% 0805 Any 5 R67 301Ω Resistor, 301Ω, 1/10W, 1% 0805 Any 1 R70 909Ω Resistor, 909Ω, 1/16W, 1% 0603 Any 1 R72 0Ω Resistor, 0Ω, 1W, 5% 2512 Any 1 R38, R39 0.12Ω Resistor, 0Ω, 1W, 5% 2512 Any 1 U2 218-2LPST SWITCH, SMD, 2P, SLIDE, 150M HALFPITCHGOLD CTS 1 S1, S2 GT11MSCKE SWITCH-TOGGLE, SMD, ULTRAMINI, 1P, SPST MINI C&K 2 J1-J4 KPA8CTP MTG HDWR, CBL.TERMINAL-LUG&SCREW, 6-14AWG BERG/FCI 4 C3, C15, C17, C70, DNP C71, C91 C92, C169 0 R2, R5, R7, R9, R12, R19, R23R25, R29, R40R42, R45, R48, R51, R52, R54R57, R68, R71, R81, R92 DNP 0 DNP (D3) LT1009CLP IC-2.5V ADJ. SHUNT REGULATOR, TH, 3P, TO-92 TO-92 TI DNP (U5) LT1616ES6 IC-SWITCHING REGULATOR, 6P, SOT23, 0.6A SOT23 Linear Tech 16 0 AN1212.0 October 31, 2005 Application Note 1212 ISL8102EVAL1 Layout TOP SILK SCREEN TOP LAYER (1st) 17 AN1212.0 October 31, 2005 Application Note 1212 ISL8102EVAL1 Layout (Continued) GROUND LAYER (2nd) GND/SIGNAL LAYER (3rd) 18 AN1212.0 October 31, 2005 Application Note 1212 ISL8102EVAL1 Layout (Continued) POWER LAYER (4th) GND LAYER (5th) 19 AN1212.0 October 31, 2005 Application Note 1212 ISL8102EVAL1 Layout (Continued) POWER/SIGNAL LAYER (6th) BOTTOM SILK SCREEN Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without notice. Accordingly, the reader is cautioned to verify that the Application Note or Technical Brief is current before proceeding. For information regarding Intersil Corporation and its products, see www.intersil.com 20 AN1212.0 October 31, 2005