Application Note AN-52 HiperPFS Family

Application Note AN-52
HiperPFS Family
™
Design Guide
Introduction
Electronic equipment and power supplies without any active or
passive power factor correction (PFC) circuits have an input
current waveform which has a poor power factor and high
harmonic distortion. Poor power factor and high distortion of
current drawn by equipment is a key contributor of losses in the
power distribution network. In order to enforce a limit on the
amount of harmonic distortion and power factor degradation
caused by electrical and electronic equipment, IEC/EN standard
61000 stipulates limits on the harmonic distortion of different
classes of equipment. It is difficult to meet these limits without
the use of an active or passive power factor correction circuit.
Active power factor correction using a boost PFC is one of the
most cost effective methods of power factor correction and is
the chosen topology of the HiperPFS product family.
The HiperPFS is a family of highly integrated ICs that provides
active power factor correction for switching power supplies. The
HiperPFS enables the design of power factor corrected, rectifier
front ends for switching power supplies rated up to a load of
425 W for universal input applications and up to 900 W for power
supplies with a 230 VAC input. The resulting input current
waveform has very low distortion, and the voltage and current
waveforms achieve near unity power factor, enabling the power
supply to meet the requirements of EN61000-3. Several
innovative features of the HiperPFS result in a circuit that is easy
to design, and reduces design effort and cycle time. The
HiperPFS family also improves design reliability via features such
as input undervoltage protection, built-in overload protection and
hysteretic thermal shutdown protection.
Each member of the family has a high-voltage power MOSFET
and a highly integrated controller in a compact package that
enables design of low profile power supplies. The PFC engine
uses a novel constant amp-seconds control to determine the on
time and a constant volt second control to determine the off-time.
The resulting switching waveform has inherent frequency
modulation which simplifies input filter design by spreading the
EMI energy over a wider frequency range. The integrated
soft-start feature ensures that the start-up is graceful without any
abnormal input current and any significant output voltage
overshoot. In addition, the ICs have integrated functions that
provide system-level protection. The soft-shutdown feature
provides a graceful shutdown of the converter during a line
brown-out. The open feedback pin detection feature ensures
that the power factor correction stage operates only when the
feedback circuit is correctly configured. EcoSmart technology
continuous frequency slide, results in an automatic reduction in
operating frequency as the load level reduces which ensures that
operating efficiency remains high down to extremely light load
TM
www.powerint.com levels. This simplifies meeting energy efficiency standards such
as the European Code of Conduct, 80 PLUS and ENERGY STAR.
This application note describes the operation of the HiperPFS.
Supporting information is provided to facilitate design optimization
and selection of required components together with guidelines
for system design. Design steps are described using the PIXls
spreadsheet to facilitate design of a PFC circuit using the
HiperPFS.
For additional information regarding active power factor correction
in general, see Application Note AN-53 Active Power Factor
Correction – Basics.
Need for Power Factor Correction
For a sinusoidal AC supply with a linear load, Power Factor (PF) is
a measure of the ratio of real power and apparent power supplied
by the AC source. Real power is measured in watts and
represents the energy consumed by the load to do useful work.
Reactive power is the power that flows back and forth between
the source and the load and is a result of the reactive nature of
components on the load side. Apparent power is the vector sum
of the real and reactive power. When the reactive power is high,
the AC source must supply a large apparent power to support
the operation of the load which results in higher RMS current.
High reactive power not only demands a higher source capacity
to support the load, but also results in higher transmission
losses. For a pure sinusoidal voltage and current waveform, PF
is the cosine of the phase angle between the voltage and current
waveforms. The value of PF therefore can vary from 0 to 1 and
can be leading or lagging. In situations where the power factor is
lagging, PF improvement is achieved by connecting capacitor
banks across the source. The resulting current from the source
is in phase with the source voltage and PF correction is achieved.
PF is therefore a figure of merit indicating how effectively energy
is transmitted between the source and the load.
PF =
PAVERAGE
V RMS # I RMS
(1)
AC-DC switching power supplies with a rectifier front end draw a
current waveform which is not sinusoidal. The waveform contains
harmonics at frequencies higher than the fundamental frequency.
This often introduces some distortion in the supply voltage
waveform. The AC source waveform in real life is often distorted
due to the increased number of non-linear loads that are connected
on the grid. The AC power being transmitted to the switching
power supply load now consists of power transmission at the
harmonic frequencies in addition to power being supplied at the
fundamental frequency.
December 2011
Application Note
AN-52
The average power is now calculated using the equation:
3
PAVERAGE = V0 I 0 + / V N I N cos ^{N - iN h
2
N=1
(2)
N represents the harmonic number and the term. Cos(jN-qN) is
a displacement term which accounts for the phase displacement between the harmonic voltage and respective harmonic
current. The equation indicates that only when the voltage and
current waveform contain the same harmonic, power is transmitted
at the harmonic frequency. If the AC supply voltage is purely
sinusoidal and free of harmonic distortion at any other frequency,
the presence of harmonics in the current waveform alone, only
increases the RMS value of the current waveform and leaves the
average power unchanged. Increase in the RMS value of
current therefore only results in added transmission line losses.
The equivalent circuit of a typical non-PFC switching power
supply front end is shown in Figure 1. Figure 2 shows the
resulting non-sinusoidal input current waveform. Several
passive and active techniques to improve the current waveform
and achieve a near sinusoidal wave shape are described in
AN-53. A VF-CCM (Variable Frequency – Continuous
Conduction Mode) boost PFC circuit designed using the
HiperPFS, when used as a power supply front end, results in a
sinusoidal low distortion input current waveform compliant to
most requirements including EN61000-3.
Figure 3 shows the input current waveform of a typical VF-CCM
boost PFC stage designed using the HiperPFS. Near unity
power factor is achieved while simultaneously maintaining
overall high efficiency across load range.
Figure 4 shows the schematic of a typical boost PFC designed
using the HiperPFS.
BR1
PI-6303-020711
8
IBR
6
VIN
4
CIN
RLOAD
Current (A)
+
VOUT
0
-2
-4
PI-5679-012011
PI-6281-011411
Figure 1. Full Wave Rectifier.
2
-6
-8
0
25
50
Time (ms)
Voltage (V)
Figure 3. Input Current Waveform at 115 VAC for a Typical 425 W PFC
using HiperPFS (2 A, 1 ms / div).
0
50
Time (ms)
100
Figure 2. Full Wave Rectifier Stage Waveforms. Top: Input Voltage. Middle: Output Voltage. Bottom: Input Current.
2
Rev. B 12/11
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AN-52
Application Note
D2
D1
385 V
L1
VCC
*CSN1
R4
F1
AC E
IN
LCM
CX1
N
R1
LDM
CY1
CY2
R3
D
CX2
TH1
C2
D3
BR1
L
+
R2
VCC
CONTROL
HiperPFS
S
CIN
V
CSN2
R10
FB
R9
D4
G
Q1
R8
R5
C1
DC
OUT
R6
C3
CFB
CV
CVCC
C4
Q2
R7
PI-6223-020711
*Optional – required for universal input designs >350 W and 230 VAC input designs >700 W
Figure 4. Typical PFC Frontend Designed using HiperPFS (385 V Output).
Scope
This application note is intended for engineers designing an
active PFC circuit using the HiperPFS family of devices. It
provides guidelines to enable an engineer to quickly select key
components and also complete a suitable inductor design. To
simplify the task this application note refers directly to the PIXls
design spreadsheet that is part of the PI Expert™ design
software suite. The basic configuration used in a PFC designed
using the HiperPFS is shown in Figure 4, which also serves as
the reference circuit for component identifications used in the
description throughout this application note.
In addition to this application note the reader may also find the
HiperPFS Reference Design Kit (RDK-236) containing an
engineering prototype board, engineering report and device
samples useful as an example of a working power supply.
Further details on downloading PI Expert, obtaining a RDK and
updates to this document can be found at www.powerint.com.
Theory of Operation
The HiperPFS employs a novel control method that creates an
input current waveform that follows the shape of the input
voltage waveform by varying the on time and the off time of the
power switch.
More specifically, the control technique sets constant voltseconds for the off time. The off-time is controlled such that:
^ VOUT - V IN h # t OFF = K 1 (3)
Since the volt-seconds during the on time must equal the
volt-seconds during the off time, to maintain flux equilibrium in
the PFC choke, the on time is controlled such that:
V IN # t ON = K 1 (4)
The controller also sets a constant value of charge during each
on cycle of the power MOSFET. The charge per cycle is varied
gradually over many switching cycles in response to load
changes so can be regarded as substantially constant for a half
line cycle. With this constant charge (or Amp-Second) control,
the following relationship is therefore also true:
(5)
i IN # t ON = K 2
Substituting tON from (4) into (5) gives:
i IN = V IN # K 2
K1
(6)
The relationship of (6) demonstrates that by controlling a
constant amp-second on time and constant volt-second off
time, the input current IIN is proportional to the input voltage VIN,
therefore providing the fundamental requirement of power factor
correction.
This control produces a continuous mode power switch current
waveform that varies both in frequency and peak current value
across a line half cycle to produce an input current proportional
to the input voltage.
3
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Rev. B 12/11
Application Note
AN-52
PFC Control Engine
Figure 5 illustrates at a high level the PFC control engine of the
HiperPFS device. It comprises of two main sections; an
on-time controller and an off-time controller which both
interface to a latch to generate the control gate-drive signal to
the power MOSFET.
of the off-time control comparator ‘sets’ the latch to terminate
the off-time and begin the on-time of that particular cycle.
The on-time controller comprises of a current source proportional
to the switch current, a resettable integrator and a comparator.
The scaled switch current (from the sense-FET) is used to
charge an integrating capacitor up to a level defined by the
output voltage regulator (error-voltage). The output of the
on-time control comparator ‘resets’ the latch to terminate the
on-time and begin the off-time of that particular cycle. In a
given input line frequency cycle, the error-voltage from the
output voltage regulating error-amplifier is quasi-static providing
the constant amp-second relationship described in equation (5).
VE
VOFF
(VOUT-VIN)dt
Latch
SET
Gate
Drive (Q)
Timing
Supervisor
Figure 6. Idealized Control Waveform.
VOLTAGE MONITOR (V)
BIAS POWER (VCC)
INPUT
LINE INTERFACE
Peak
Detector
MON IVPK
IS dt
Latch
RESET
Maximum
ON-time
Minimum
OFF-time
The off-time controller comprises of a current source
proportional to the difference between the output and input
voltage (VOUT-VIN), a resettable integrator and a comparator. In
this circuit block the integrating capacitor voltage level is
controlled by a fixed reference providing an off-time that has
constant volt-second relationship described in (3). The output
PI-5335-111610
The idealized pertinent control signals of the on-time and
off-time controllers are shown below.
DRAIN (D)
+
INTERNAL
SUPPLY
VCC+
-
OTP
Input UV
(IUV-/IUV+)
6V
Input Voltage
Emulation
SOFT
START
“Off-time derived with
constant Volt-Sec
VO-VIN
+
CINT
-
7 kHz
Filter
VOFF is a function of the error-voltage (VE) and is used to reduce the average
operating frequency as a function of output power for increased efficiency
(PFS704-716).
(VOFF = 0.8 V for PFS723-729).
Frequency
I
Internal
Reference
VREF
FEEDBACK (FB)
Transconductance
Error-Amplifier
FBOV
FBUV/
FBOFF
+
-
OFF
VOFF
V
E
VE
1 kHz
Filter
+
-
+
-
Slide
V
VPK
Comparator
Latch
Input UV
FBOV/UV
MON IS
Sense
FET
VCC
Power
MOSFET
+
Comparator
OTP
Driver
+
IS
LEB
-
MON is the switch current
sense scale factor which
is function of peak line
voltage derived from IVIN
Fast OV
Comparator
-
TIMER
SUPERVISOR
The internal derived error-voltage (VE)
regulates the output voltage
+
OCP
-
IOCP
CINT
UV Comparator
SIGNAL GROUND (G)
SOURCE (S)
PI-5333-113010
Figure 5. HiperPFS Block Diagram.
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Rev. B 12/11
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AN-52
Application Note
Off-time Control
Frequency (kHz)
80
70
60
50
40
Full Load
50% Load
75% Load
30
20
120
100
0
1
2
3
4
5
6
80
60
Full Load 90 VAC
Full Load 115 VAC
Full Load 150 VAC
Full Load 130 VAC
Full Load 170 VAC
Full Load 200 VAC
Full Load 230 VAC
Full Load 245 VAC
40
20
10
0
PI-6305-011711
90
Frequency (kHz)
PI-6304-011111
100
7
8
9
0
10
Time Period (ms)
0
1
2
3
4
5
6
7
8
9
10
Time Period (ms)
Figure 7. Frequency Variation with Load.
Figure 8. Frequency Variation with Input Voltage Change (100% Load at Output).
As the peak input voltage approaches the output voltage the
difference between these two quantities approaches zero and
the required time to satisfy volt-second balance increases. This
relationship is apparent upon closer examination of the off-time
controller shown in Figure 6.
sheet, other parameters will be automatically selected based on
a typical design. References to spreadsheet cell locations are
provided in square brackets [cell reference].
•
Enter the voltage range for the design [B3]
Universal should be selected if the application requires a
wide operating voltage range. High-line should be selected
for applications that only are required to work with 220 V /
240 V nominal AC supply.
Enter AC input voltage range VACMIN, VACMAX and minimum
line frequency fL [B4, B5, B10]
Enter nominal output voltage VO [B8]
Enter maximum continuous output power [B9]
Enter efficiency estimate at VACMIN [B12]
• 0.93 for universal input voltage (85 - 265 VAC) or single 100
/ 115 VAC (85 -132 VAC) and 0.96 for a single 230 VAC
(180 - 265 VAC) design. Correct the number after measuring the efficiency of the prototype board at full load and
VACMIN.
Enter maximum operating temperature [B11]
• If left unpopulated the default value is 40 ºC
Enter maximum output ripple [B15]
• If no specific information is provided a default value of 20 V
is used
Enter hold-up time required [B16]
• 16 ms or 20 ms are standard requirements for most high
performance designs
• Use 10 ms if long hold-up time is not a requirement
• If this cell is left blank then the output capacitance value is
calculated for a hold-up time of 20 ms
Enter minimum PFC output voltage at the end of the hold-up
interval [B17]
• 310 V is typically adequate for most second stage converters
to maintain regulation.
• Use 310 V if no specific information is available
Select a suitable HiperPFS device and enter directly [B23]
• Select the device in Table 1 according to output power and
line input voltage. If this filed is left as “Auto” a suitable
device will be selected automatically.
•
The off-time is derived through a transconductance amplifier
which produces a current that is proportional to the difference
between VOUT and VIN. As VOUT-VIN approaches zero, which is the
case when the peak of the input voltage is close to the set
output voltage regulation threshold, the off-time current source
reduces.
•
•
•
As a natural consequence the rate in which the integrating
capacitor is charged toward the off-time control reference is
decreased causing the off-time to increase. Theoretically for
this type of control, the resultant off-time when VOUT is equal to
VIN is infinity.
•
The HiperPFS includes a timer supervisor function to limit both
the minimum and maximum switch on and off times. Specifically,
limiting the maximum off-time prevents the controller from
causing audible noise in the boost choke if the required switching
frequency falls below ~20 kHz for a load in excess of 20% of the
device peak power rating. The specified maximum off-time in
the HiperPFS is limited to between 30 µs to 40 µs.
•
The operating frequency of the PFC changes as a function of
load level and also as a function of line voltage. The frequency
plots shown in Figure 7 and Figure 8 can be used to estimate
the operating frequency depending on input voltage and load.
•
•
•
Designing with HiperPFS
Quick Start
Readers willing to start immediately can use the following
information to quickly design the inductor and select the
components for a first prototype. Only the information
described below needs to be entered into the PIXls spread-
•
5
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Rev. B 12/11
Application Note
AN-52
Step-by-Step Design Procedure Introduction
Output Power Table
Maximum Continuous
Output Power Rating at
90 VAC
Peak Output Power
Rating at 90 VAC
PFS704EG
110 W
120 W
PFS706EG
140 W
150 W
PFS708EG
190 W
205 W
PFS710EG
240 W
260 W
PFS712EG
300 W
320 W
PFS714EG
350 W
385 W
Product
PFS716EG
The design flow allows for design of a PFC stage for a rated
continuous output power requirement. Operating conditions of
a power supply and test results of evaluation may require
design changes for performance optimization and this design
flow should be used to iterate a design as necessary.
The power table (Table 1) provides guidance for selection of a
suitable HiperPFS device for a design. Selection should be
based on the operating conditions such as maximum operating
temperature and the desired operating voltage range. This
approach ensures that the selected device can deliver the
required output power at minimum input line voltage.
388 W
425 W
Product
Maximum Continuous
Output Power Rating at
180 VAC
Peak Output Power
Rating at 180 VAC
PFS723EG
255 W
280 W
PFS724EG
315 W
350 W
PFS725EG
435 W
480 W
PFS726EG
540 W
600 W
Nominal Input Voltage (VAC)
VACMIN
VACMAX
750 W
100 / 115
90
132
180
265
90
265
PFS727EG
675 W
Step 1. Enter Application Variables VACMIN, VACMAX, fL ,
VO, PO, h, KP, VOUTRIPPLE, tHOLDUP, VHOLDUP, IINRUSH
Determine the input voltage range from Table 2.
PFS728EG
810 W
900 W
230 (High-Line Only)
PFS729EG
900 W
1000 W
Universal
Table 1. Output Power Table.
Notes:
1. Maximum practical continuous power at 90 VAC in an open-frame design with
adequate heat sinking, measured at 50 °C ambient.
2. Maximum practical continuous power at 180 VAC in an open-frame design with
adequate heat sinking, measured at 50 °C ambient.
Table 2. Standard Worldwide Input Line Voltage Ranges.
Brownout Voltage VBROWNOUT (V)
This is the voltage at which the power supply will shutdown due
to input undervoltage. During the initial design stage the
desired value should be adjusted by changing VAC(MIN) and will
determine the correct value of the V pin resistors to program
VBROWNOUT. Once a prototype has been constructed the actual
measured value of the lowest line voltage at which the converter
still operates should be entered.
Table 1 Output Power Table
•
Enter core type (if desired) from drop down menu [B51]
A suggested core size will be selected automatically if none
is entered
•
If any warnings are generated, make changes to the design by
following instructions in spreadsheet column F
• Build the inductor
• Select key components
• See Steps 7 through 12.
• Build prototype and iterate design as necessary, entering
measured values into spreadsheet where estimates were
used (e.g. efficiency, VMIN).
Enter Applications Variables
Input Voltage Range
VACMIN
VACMAX
VBROWNIN
VBROWNOUT
VO
PO
fL
TA Max
n
KP
VO_MIN
VO_RIPPLE_MAX
tHOLDUP
VHOLDUP_MIN
I_INRUSH
Forced Air Cooling
Universal
Universal
90
265
77.76
70.4
385
347
0.677
16
Yes
50
40
0.93
0.677
365.75
20
16
310
40
Yes
Nominal Output Voltage, VO (V)
Enter the nominal output voltage of the PFC. The HiperPFS is
designed for continuous operation at an output voltage of
385 VDC. Higher output voltages are not recommended due to
increased voltage stress on the internal MOSFET. A lower
output voltage may be used for applications that require
operation only at 100 / 115 VAC nominal, providing the output
V
V
V
V
W
Hz
°C
V
V
ms
V
A
Select Universal or High_Line option
Minimum AC input voltage
Maximum AC input voltage
Expected Minimum Brown-in Voltage
Specify brownout voltage.
Nominal Output voltage
Nominal Output power
Line frequency
Maximum ambient temperature
Enter the efficiency estimate for the boost converter at VACMIN
Ripple to peak inductor current ratio at the peak of VACMIN
Minimum Output voltage
Maximum Output voltage ripple
Holdup time
Minimum Voltage Output can drop to during holdup
Maximum allowable inrush current
Enter "Yes" for Forced air cooling. Otherwise enter "No"
Figure 9. Application Variable Section of HiperPFS Design Spreadsheet.
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Rev. B 12/11
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AN-52
Application Note
voltage is greater than the peak of the maximum AC input
voltage.
Nominal Output Power, PO (W)
Enter the maximum continuous output power of the PFC. If the
PFC is required to meet a peak power specification, the peak
power should be entered in this cell.
Line Frequency, FL
50 Hz for universal or single 100 VAC, 60 Hz for single 115 VAC
input. 50 Hz for single 230 VAC input. These values represent
typical line frequencies rather than minimum. For most
applications this gives adequate overall design margin. For
absolute worst case or based on the product specification,
reduce these numbers by 6% (47 Hz or 56 Hz) as necessary.
Ambient Temperature, TA(MAX)
A default value of 40 ºC is assumed for maximum ambient
temperature. If the ambient temperature inside the power
supply enclosure is higher than this value, the expected internal
ambient temperature should be entered in this cell. This value
will be used for calculation of the estimated thermal resistance
of the heat sinks for the HiperPFS and the PFC output diode.
Power Supply Efficiency, η
Enter the estimated efficiency of the PFC stage at the lowest
input voltage specified as VACMIN. Start with a value of 0.93 for
universal input designs and 0.96 for 230 VAC only designs.
Once a prototype has been constructed, the measured
efficiency should be entered and further inductor design
iteration performed if required.
Ripple to Peak Current Ratio, KP
The HiperPFS is a (Continuous Conduction Mode) CCM PFC.
The factor KP is a ratio of the ripple component in the inductor
current as compared to the peak inductor current at the peak of
the sine wave as shown in Figure 10. For a CCM mode
operation the value of KP is required to be less than 1.
KP ≡ KRP =
IDRAIN
IR
IP
IR
IP
IR
Hold-Up Time, tHOLDUP
Enter the hold-up time specification required to be met, this
being the time the PFC stage output capacitor must supply to
load during a line dropout during which the output voltage falls
to VHOLDUP(MIN).
Voltage at the end of Hold-Up Time, VHOLDUP(MIN)
In order to sustain operation, the load on the PFC will often
require that the PFC output voltage remain above a certain
minimum value at the end of hold-up time. Enter the minimum
voltage expected at the output of the PFC at the end of hold-up
time. A value of 310 V will be acceptable for most designs that
use a second stage converter based on the HiperTFS.
Both VOUTRIPPLE(MAX), tHOLDUP and VHOLDUP(MIN) are used to calculate
the output capacitance.
Maximum Inrush Current Permissible, IINRUSH
Enter total maximum permissible inrush current in this cell. This
information is used by the spreadsheet to determine the
minimum resistance of the inrush limiting thermistor required.
Forced Air Cooling - Option
Enter “YES” if forced air cooling is used otherwise enter “NO”.
In a forced air cooled design, higher current density can be
used in the inductor winding which enables use of thinner wires
reducing cost, size and weight of the inductor.
Select the Correct HiperPFS Device
Refer to the HiperPFS power table and select a device based
on the output power of the design. If the continuous power
exceeds the value given in the power table then the next largest
device should be selected. If the “Auto” option is selected, the
spreadsheet will automatically select the most suitable device
based on application variables entered in Step 1. Close attention
should be paid to the notes associated with the power table.
For some designs, it may be necessary to use the next higher
device depending on the specifics of the application.
IP
(b) Borderline Continuous/Discontinuous, KP = 1
PI-6340-011711
Figure 10. Continuous Mode Current Waveform, KP≤1.
Output Voltage Ripple, VOUTRIPPLE(MAX)
Enter maximum permissible output ripple voltage. A default
value of 20 V is used if this field is left blank which is typical for
most applications.
Step 2 – Enter HiperPFS Variables: HiperPFS Device, Line
Sense Resistor RV, TJ (MAX), Thermal Resistance of
HiperPFS Assembly
(a) Continuous, KP < 1
IDRAIN
A lower value of KP results in a higher inductance however a
higher inductance results in lower THD of input current and a
higher PF. As a start, for high-line only designs, a starting value
of 0.5 or 0.675 should be used for ferrite core and powder core
designs respectively. For universal input designs, a value of
0.25 or 0.675 should be used for ferrite core and powder core
designs respectively. If further improvement in PF is required
after evaluation of the PFC performance, one option to improve
PF is to lower the value of KP and iterate the inductor design.
Select the Line Sense Resistor Value, RV
The line sense resistor programs the feed forward gain for the
PFC controller and also sets the brown-in and brown-out
thresholds. The recommended value of the line sense resistor
7
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Rev. B 12/11
Application Note
PFS Parameters
PFS Part Number
IOCP min
IOCP typ
IOCP max
RDSON
RV
C_VCC
C_V
C_FB
FS_PK
FS_AVG
IP
PFS_IRMS
PCOND_LOSS_PFS
PSW_LOSS_PFS
PFS_TOTAL
TJ Max
Rth-JS
HEAT SINK Theta-CA
AN-52
Auto
PFS714
9
9.95
10.9
0.46
4
1
100
10
84.93
69.13
8.84
4.02
7.43
2.85
10.28
100
3
2.83
A
A
A
ohms
Mohms
uF
nF
nF
kHz
kHz
A
A
W
W
W
°C
°C/W
°C/W
Selected PFS device
Minimum Current limit
Typical current limit
Maximum current limit
Typical RDSon at 100 'C
Line sense resistor
Supply decoupling capacitor
V pin decoupling capacitor
Feedback pin decoupling capacitor
Estimated peak frequency of operation
Estimated average frequency of operation
MOSFET peak current
PFS MOSFET RMS current
Estimated PFS conduction losses
Estimated PFS switching losses
Total Estimated PFS losses
Maximum steady-state junction temperature
Maximum thermal resistance (Junction to heat sink)
Maximum thermal resistance of heat sink
Figure 11. PFS Parameters Section of the HiperPFS Spreadsheet.
for Universal input applications is 4 MΩ and the recommended
for a high-line only application is 9 MΩ. These values may be
adjusted to change the brown-in and brown-out voltage levels
by entering the desired value in MΩ in this cell. A warning will
be generated if a change is being made to the value of this
resistance beyond permissible limits.
Select the Operating Junction Temperature Maximum for
the HiperPFS, TJ(MAX)
The default value of the operating junction temperature for the
HiperPFS is set to 100 ºC. This ensures that there is sufficient
margin to ensure reliability and safe operation including during
brown-out testing. A different value may be used if required
however higher value are generally not recommended.
Select the Thermal Resistance-Junction to Heat Sink
for the HiperPFS, Rth-JS
The default value of the thermal resistance is set to 3 ºC/W
assuming use of a heat spreader and thermally conductive
insulator between the heat spreader and the heat sink. If no
better information is available, the value of 3 ºC/W should be
used. Additional information regarding thermal design and its
effects on thermal resistance are explained in subsequent
sections of the application note. The spreadsheet calculates
the estimated value of the thermal resistance (surface –
ambient) for the heat sink that may be necessary for the
HiperPFS part selected, based on the estimated losses in the
device.
Step 3 – Choose Core and Winding Based on Output
Power and Enter AE, LE, AL , BW, MLT
Core effective cross-sectional area, AE: (mm2)
Core effective path length, LE: (mm).
Core ungapped effective inductance factor, AL: (nH/turn2).
Bobbin width, BW: (mm)
Mean length per turn: (mm)
Core Type
By default, if the “Core Type” cell is left empty, the spreadsheet
will select the smallest commonly available core suitable for the
continuous (average) output power specified. The entire list of
cores available can be shown by selecting the drop down list in
the tool bar of the PIXls design software. The gray override cells
can be used to enter the core and bobbin parameter directly by
the user. This is useful if a core is selected that is not on the list
or the specific core or bobbin information differs from that
recalled by the spreadsheet.
Sendust, ferrite and iron powder are three choices that are
available for selection. Sendust cores are a cost effective option.
Ferrite core inductors have the lowest core loss; however it is
often difficult to achieve a high inductance without using a large
core size. Iron powder cores offer the cheapest alternative
however, core loss can be significant.
Core Material
When the Sendust core type is selected, four different permeability
options are available for selection using a drop down menu.
Core material with 60 µ, 75 µ, 90 µ or 125 µ, can be selected.
High permeability materials will yield a higher 0-bias inductance
for the same number of turns. Inductors made with high
permeability cores will typically reduce in inductance with flux
bias. The increase in inductance with drop in current helps the
PFC to maintain CCM operation at light load levels and is
beneficial in maintaining a high PF and low input current THD
(ITHD) even at light load levels.
Only the -52 material is supported for iron powder cores and
PC44 material is supported for ferrite cores. Other material
types may also be used, however the loss calculations will not
be accurate.
Core Geometry
A choice of EE or toroidal core geometry is available for Sendust
core type. For ferrite, either EE or PQ core geometry is available
as a standard choice. Only toroid geometry is available as a
choice for iron powder cores. Parameter values for the selected
core are updated automatically. If the core parameters are
different as compared to the ones shown in the spreadsheet,
they can be entered in the gray cells in the list and the
spreadsheet will update the calculation results.
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AN-52
Inductor Construction Parameters
Core Type
Core Material
Core Geometry
Core
AE
LE
AL
VE
HT
MLT
BW
NL
LG
ILRMS
Wire type
AWG
Filar
OD
AC Resistance Ratio
J
BM_TARGET
BM
BP
LPFC_CORE_LOSS
LPFC_COPPER_LOSS
LPFC_TOTAL LOSS
Application Note
Sendust
Sendust
Auto
125u
Auto
77324
(OD=36.7)
TOROID
77324
(OD=36.7)
67.8
89.8
119
6088
11.35
43.4
N/A
108
N/A
4.77
LITZ
40
125
40
125
0.079
1.09
7.84
N/A
2591
6898
2.26
2.59
4.85
Enter "Sendust", "Pow Iron" or "Ferrite"
Select from 60u, 75u, 90u or 125 u for Sendust cores. Fixed at PC44 or
equivalent for Ferrite cores. Fixed at 52 material for Pow Iron cores.
Select from Toroid or EE for Sendust cores and from EE, or PQ for Ferrite
cores
Core part number
Core cross sectional area
Core mean path length
Core AL value
Core volume
Core height/Height of window
Mean length per turn
Bobbin width
Inductor turns
mm
Gap length (Ferrite cores only)
A
Inductor RMS current
Select between "Litz" or "Regular" for double coated magnet wire
AWG Inductor wire gauge
Inductor wire number of parallel strands
mm
Outer diameter of single strand of wire
Ratio of AC resistance to the DC resistance (using Dowell curves)
A/mm^2 Estimated current density of wires. It is recommended that 6 < J < 8
Gauss Target flux density at VACMIN (Ferrite cores only)
Gauss Maximum operating flux density
Gauss Peak Flux density (Estimated at VBROWNOUT)
W
Estimated Inductor core Loss
W
Estimated Inductor copper losses
W
Total estimated Inductor Losses
mm^2
mm
nH/t^2
mm^3
mm
cm
mm
Figure 12. Inductor Core and Construction Variables Section of Spreadsheet.
careful consideration of core material saturation characteristics
at high temperature.
Wire Type
By default the LITZ wire option is selected. Litz wires offer the
lowest proximity and skin effect losses and hence provide the
highest efficiency for most designs. Regular (double coated)
magnet wires with a single or multi strand construction may
also be used by entering REGULAR.
Wire Gauge AWG
If the Litz wire option is selected, PIXls will select a Litz wire with
the appropriate number of strands and wire gauge and the
result will be updated. If a different Litz wire is preferred, the
wire gauge (AWG) and number of strands should be entered in
the cells.
Number of Strands
For a Litz wire selection, the number of strands is selected
automatically. This choice can be changed by entering the
number of strands in the cell [B67]. While a multi-filar construction
reduces AC losses, it can also lead to higher proximity losses.
The copper loss calculation corrects for proximity loss based on
the wire gauge (AWG) selected.
Wire Diameter
If the wire diameter is different from the wire diameter displayed,
cell [B68] can be changed to show the actual diameter of wire.
Target Flux Density
For ferrite cores, the target flux density is set to 3000 G (0.3 T).
If a higher flux density is to be selected, cell [B71] should be
changed. Choice of a higher flux density should be made after
Basic Inductor Calculation
LPFC
LPFC (0 Bias)
LPFC_RMS
150.47
1383.32
4.77
uH
uH
A
The spreadsheet will calculate the number of turns required and
the inductance value. The expected inductance with no
DC-bias is calculated. This is the expected inductance value
when measured using an LCR meter.
Step 4 – Iterate Inductor Design and Generate Initial Design
Iterate the design making sure that no warnings are displayed.
Any parameters outside the recommended range of values can
be corrected by following the guidance given in the right hand
column.
Once all warnings have been cleared, the inductor design
parameters can be used to either wind a prototype inductor or
sent to a vendor for samples.
The Key Inductor Electrical Parameters:
Primary Inductance, LPFC(0-BIAS) (mH)
This is the target nominal primary inductance of the transformer.
Primary Inductance Tolerance, LP(TOLERANCE) (%)
This is the assumed primary inductance tolerance. A value of
12% is used by default, however if specific information is known
from the magnetics vendor, then this may be entered in the gray
override cell.
Value of PFC inductor at peak of VACMIN and Full Load
Value of PFC inductor at No load. This is the value measured with LCR meter
Inductor RMS current (calculated at VACMIN and Full Load)
Figure 13. Induction Calculation Section of HiperPFS Design Spreadsheet.
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Rev. B 12/11
Application Note
AN-52
Gapped Core Effective Inductance, ALG (nH/t2)
Used by the transformer vendor to specify the core (gap). This
information is only required for ferrite core designs.
Maximum Operating Flux Density, BM (Gauss)
For ferrite cores, a maximum value of 3000 Gauss (0.3 T) during
normal operation is recommended to limit the maximum flux
density under start-up and output overload conditions.
Peak Flux Density, BP (Gauss)
For ferrite cores, a maximum value of 4200 Gauss (0.42 T) is
recommended to limit the maximum flux density under start-up
and output overload conditions. This calculation assumes
worst-case current limit specification and inductance value. In
high ambient temperature applications, such as sealed adapters,
this value may need to be reduced to 3600 Gauss. To determine
the correct value, verify that core saturation does not occur at
maximum ambient during start-up.
Step 5 – Selection of Output Diode
For a CCM PFC, use of ultrafast recovery diodes is recommended
to reduce losses. Special diodes optimized for PFC application
are available from various manufacturers and are also suitable
for HiperPFS designs. These diodes have both soft recovery
characteristics and low QRR that reduce EMI under the hard
switching of CCM operation while simultaneously reducing
forward conduction and switching losses.
The spreadsheet automatically selects a diode with forward
current rating suitable for the specified output power. The peak
current through the PFC output diode is same as the PFSMOSFET peak current. A different diode can be selected from
the drop down list. If any warnings are generated, selection of a
different diode is necessary.
The diode parameters such as tRR, VF etc. can be modified
based on actual operating conditions and data sheet information.
Change of diode VF or tRR parameter affects diode losses. If a
different diode is used, these numbers should be updated in
the diode parameter cells [B91] and [B92].
Typically the diodes will be required to have a forward continuous
current rating of at least 1.2 A to 1.5 A for every 100 W of output
power.
Output Diode
Part Number
Type
Manufacturer
VRRM
IF
TRR
VF
PCOND_DIODE
PSW_DIODE
P_DIODE
TJ Max
Rth-JS
HEAT SINK Theta-CA
Auto
STTH8S06
ULTRAFAST
ST
600
8
33
1.1
0.99
2.15
3.14
125
3
23.57
V
A
ns
V
W
W
W
°C
°C/W
°C/W
Diode
Part
Number
Manufacturer
Voltage
Rating
(V)
Current
Rating
(A)
Type
LQA03TC600
Special
Power Integrations
600
3
LQA05TC600
Special
Power Integrations
600
5
LQA08TC600
Special
Power Integrations
600
8
LXA04T600
Special
Power Integrations
600
4
LXA06T600
Special
Power Integrations
600
6
LXA08T600
Special
Power Integrations
600
8
STTH1R06
Ultrafast
ST
600
2
STTH2R06
Ultrafast
ST
600
2
STTH3R06
Ultrafast
ST
600
3
STTH506
Ultrafast
ST
600
5
STTH506D
Ultrafast
ST
600
5
STTH5R06
Ultrafast
ST
600
5
STTH806
Ultrafast
ST
600
8
STTH806TTI
Ultrafast
ST
600
8
STTH8R06
Ultrafast
ST
600
8
STTH8S06
Ultrafast
ST
600
8
STTH12R06
Ultrafast
ST
600
12
CSD01060
SiC
CREE
600
1
CSD02060
SiC
CREE
600
2
CSD04060
SiC
CREE
600
4
CSD06060A
SiC
CREE
600
6
CSD08060A
SiC
CREE
600
8
CSD10060A
SiC
CREE
600
10
HFA04TB60
Ultrafast
Vishay
600
4
HFA08TB60
Ultrafast
Vishay
600
4
HFA16TA60C
Ultrafast
Vishay
600
16
Table 3. Diodes Suitable for PFC Application.
Diode Junction Temperature, TJMAX
A default value of 125 ºC as the maximum junction operating
temperature is used for the diode. If a lower stress level is
desirable, this value should be entered in the cell [B96]. The
diode thermal resistance depends on the package selected. If
the thermal resistance of the package is different as compared
to the one used by the spreadsheet, enter the value in cell [B97].
PFC Diode Part Number
Diode Type - Special - Diodes specially catered for PFC applications, SiC Silicon Carbide type, UF - Ultrafast recovery type
Diode Manufacturer
Diode rated reverse voltage
Diode rated forward current
Diode Reverse recovery time
Diode rated forward voltage drop
Estimated Diode conduction losses
Estimated Diode switching losses
Total estimated Diode losses
Maximum Operating temperature
Maximum thermal resistance (Junction to heat sink)
Maximum thermal resistance of heat sink
Figure 14. Output Diode Section Of HiperPFS Design Spreadsheet.
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AN-52
Application Note
The spreadsheet calculates the maximum permissible heat sink
thermal resistance that may be usable for the diode selected
and based on the output power of the PFC. Heat sink
temperature should be measured on a prototype unit and
necessary changes made in order to optimize the design.
Input Bridge Rectifier
The average input rectifier current, peak inverse voltage and
total power dissipation is calculated in the spreadsheet. The
estimated bridge rectifier power dissipation is dependent on the
forward voltage drop of the diodes in the bridge rectifier. If the
forward voltage drop is different from the one shown in cell
[E119], enter the diode forward voltage drop in the cell [B119].
Bridge rectifier with at least a 600 V PIV rating is recommended
for a universal input design. For designs with only a 115 V /
100 V nominal input voltages, bridge rectifiers with PIV rating as
low as 400 V can be used.
Step 6 – Selection of PFC Output Capacitor
Selection of the output capacitor depends on the expected
hold-up time and the permissible output ripple. The spreadsheet automatically selects the most suitable capacitor
depending on the hold-up time and output ripple specified in
the applications variable section.
The low frequency ESR value is the specified ESR at 100 Hz /
120 Hz and the high frequency ESR value is the ESR at 20 kHz.
If the low frequency and high frequency ESR values are different
for the capacitor selected, enter the appropriate values in cell
[B106] and [B107].
Power is dissipated in the output capacitor due to the low
frequency and high frequency ripple currents. The spreadsheet
estimates these losses which can be used for thermal calculations
to estimate temperature rise of the capacitor.
Step 7 – Selection of Other Circuit Components
Fuse Current Rating and I2t Rating
Fuse continuous current rating and the required I2t rating are
calculated in the spreadsheet. These values can be used to
select a suitable fuse.
Auto
270
11.4
18.4
0.68
0.27
0.64
1.83
0.27
0.91
4.15
1.18
0.9
Input Thermistor RT
The cold state resistance of the thermistor required to guarantee
the maximum inrush current specification in the applications
variables section is calculated. A thermistor with a continuous
current rating exceeding the calculated input RMS current and
the calculated value of RT should be used.
Output Capacitor Pre-Charge Diode
A diode is used to bypass the PFC inductor at start in order to
route the charging current for the output capacitor away from
the inductor. In the absence of this diode, the capacitor
charging current flows through the inductor and the resonance
between the inductor and the output capacitor causes the
output voltage to be as high as twice the input voltage which
can result in failure of the capacitor and other circuit
components.
The spreadsheet provides recommended values or parameter
values for selection of the balance circuit components to
complete the design.
Output Capacitor
CO
VO_RIPPLE_EXPECTED
T_HOLDUP_EXPECTED
ESR_LF
ESR_HF
IC_RMS_LF
IC_RMS_HF
CO_LF_LOSS
Critical Parameters
CO_HF_LOSS
IRMS
Total
CO LOSS
IO_AVG
Input Filter Capacitor
A filter capacitor with low ESR and low ESL is recommended to
be placed after the bridge rectifier. This capacitor helps to
reduce EMI. The required value is estimated by the spreadsheet.
uF
V
ms
ohms
ohms
A
A
W
W
A
W
A
Minimum value of Output capacitance
Expected ripple voltage on Output with selected Output capacitor
Expected holdup time with selected Output capacitor
18.96
6.3
0.9
3.92
375
6.72
A^2s
A
V
A
V
W
1
7.54
1N5407
uF
ohms
Minimum I^2t rating for fuse
Minimum Current rating of fuse
Input bridge Diode forward Diode drop
Input average current at 70 VAC.
Peak inverse voltage of input bridge
Estimated Bridge Diode conduction loss
Input capacitor. Use metallized polypropylene or film foil type with high ripple
current rating
Input Thermistor value
Recommended precharge Diode
Low Frequency Capacitor RMS current
High Frequency Capacitor RMS current
Estimated Low Frequency ESR loss in Output capacitor
Estimated
Highcurrent
frequency ESR loss in Output capacitor
AC input RMS
Total
estimated
Output
average losses
currentin Output Capacitor
Figure 15. Output Capacitor Section of HiperPFS Design Spreadsheet.
Input Bridge and Fuse
I^2t Rating
Fuse Current rating
VF
IAVG
PIV_INPUT BRIDGE
PCOND_LOSS_BRIDGE
CIN
RT
D_Precharge
Figure 16. Input and Bridge Section of HiperPFS Design Spreadsheet.
Critical Parameters
IRMS
IO_AVG
4.15
0.9
A
A
AC input RMS current
Output average current
Figure 17. Critical Parameters Section of HiperPFS Design Spreadsheet.
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Rev. B 12/11
Application Note
AN-52
The spreadsheet selects either the 1N4007 or the 1N5407 diode
to ensure that the diode surge current rating is not exceeded
during the pre-charge of the output capacitor.
bandwidth control loop. Figure 18 shows the minimum
recommended circuit with the reference designators matching
those in Figure 19.
Feedback Circuit Components
A feedback circuit arrangement comprising of a voltage divider
network, compensation elements and a set of general purpose
transistors is recommended for use with the HiperPFS. This
simple arrangement ensures that loop stability is achieved while
simultaneously ensuring that the circuit responds to any
transient conditions rapidly, without the delays of a low
The feedback circuit components are calculated in the feedback
components section. Component values closest to the values
shown should be used. If component values used are different
from the values shown in the spreadsheet, the output voltage of
the PFC will be different and some adjustment of resistor values
will be required.
D2
D1
385 V
L1
VCC
*CSN1
R4
F1
AC E
IN
LCM
CX1
N
R1
LDM
CY1
CY2
R3
D
CX2
TH1
C2
D3
BR1
L
+
R2
S
CSN2
VCC
CONTROL
HiperPFS
CIN
V
R10
FB
R9
D4
G
Q1
R8
R5
C1
DC
OUT
R6
C3
CFB
*Optional – required for universal input designs >350 W and 230 VAC input designs >700 W
CV
CVCC
C4
Q2
R7
PI-6223-020711
Figure 18. Minimum Recommended Circuit for HiperPFS.
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AN-52
Feedback Components
R2
R3
R4
C2
R5
R6
R7
C3
R8
R9
R10
C4
D3
D4
Q1
Q2
Application Note
1.54
1.54
698
100
2.2
2.2
57.6
470
160
2.32
10
10
1N4148
1N4001
2N4401
2N4403
Mohms
Mohms
kohms
nF
kohms
kohms
kohms
pF
kohms
kohms
kohms
uF
Feedback network, first high voltage divider resistor
Feedback network, second high voltage divider resistor
Feedback network, third high voltage divider resistor
Feedback network, loop speedup capacitor
Feedback component, NPN transistor bias resistor
Feedback component, PNP transistor bias resistor
Feedback network, lower divider resistor
Feedback component- noise suppression capacitor
Feedback network - pole setting resistor
Feedback network - zero setting resistor
Feedback pin filter resistor
Feedback network - compensation capacitor
Feedback network reverse blocking Diode
Feedback network - capacitor failure detection Diode
Feedback network - speedup circuit NPN transistor
Feedback network - speedup circuit PNP transistor
Figure 19. Feedback and Components Section of HiperPFS Design Spreadsheet.
Loss Budget (Estimated at VACMIN)
PFS Losses
Boost diode Losses
Input Bridge losses
Inductor losses
Output Capacitor Loss
Total losses
Efficiency
10.28
3.14
6.72
4.85
1.18
26.18
0.93
W
W
W
W
W
W
Total estimated losses in PFS
Total estimated losses in Output Diode
Total estimated losses in input bridge module
Total estimated losses in PFC choke
Total estimated losses in Output capacitor
Overall loss estimate
Estimated efficiency at VACMIN. Verify efficiency at other line voltages
Figure 20. Loss Budget Section of HiperPFS Design Spreadsheet.
Step 8 – Verify Loss Budget
B+
Power dissipation in the key components of the circuit is
calculated and listed in the spreadsheet.
R2
VCC
The estimated losses are expected to be close to the actual
losses. In real life the power dissipated in the components may
be slightly different depending on parasitic elements and
material properties of components used. The loss budget
helps to verify the estimated efficiency. If the estimated
efficiency based on the loss budget is different to the efficiency
at VACMIN used in the applications variables section, the
efficiency specification should be changed and the design
should be iterated. The loss budget provides insight into the
power dissipated in the key elements and helps with the thermal
design of the power supply.
R4
C2
D3
R3
D
V
VCC
CONTROL
HiperPFS
S
CSN2
R10
FB
R9
D4
G
Q1
R8
R5
R6
C3
CFB
CV
CVCC
C4
Q2
R7
Designing with HiperPFS
Feedback Circuit Design – Analysis and Calculations
PI-6339-011411
The typical feedback network tied to the FEEDBACK pin is
shown in the Figure 21.
Figure 21. Typical External Feedback Network (Including Transient Response
Loop Speed-Up).
Resistors R2 to R7 comprise of the main output voltage divider
network. The sum of resistors R2, R3, R4 and R5 is the upper
divider resistor and the lower feedback resistor is comprised of the
sum of resistors R6 and R7. Capacitor C2 is a soft-finish
capacitor that reduces output voltage overshoot at start-up.
Resistor R10 and capacitor CFB form a low pass filter to filter any
switching noise from coupling into the FEEDBACK pin. Resistor
R9 and capacitor C4 form a loop compensation network which
introduces a low frequency zero required to tailor the loop
response to ensure low cross-over frequency and sufficient
loop phase margin. Resistor R8 isolates the fast portion
(resistor voltage divider network comprising of resistors R2 to
R7) and the slow feedback loop compensator circuit (resistor
R9 and capacitor C4). Transistors Q1 and Q2, biased with
resistors R5 and R6 respectively, detect output voltage transient
conditions and provide the FEEDBACK pin with “fast” information
to increase the loop response of the system dynamically. Diode
D4 is included to ensure a safe shutdown under the single point
fault condition of capacitor C4 shorted. In this event the
FEEDBACK pin would be forced below the FBUV threshold
through diode D4 and subsequently turn the HiperPFS off.
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Rev. B 12/11
Application Note
AN-52
Only a standard recovery diode should be used for D4. Use of
ultrafast or fast recovery diode is not recommended due to the
difference in behavior in the presence of high frequency noise
that maybe coupled into the FEEDBACK pin.
Steady-State Analysis
In steady-state operation transistors Q1 and Q2 must be in the
“off” state to prevent any distortion of the input current
waveform. The FEEDBACK pin is the high input impedance
inverting terminal of a transconductance amplifier; the internal
non-inverting terminal is tied to a precision reference voltage.
The steady-state current into the FEEDBACK pin is assumed to
be zero in this analysis, thus the current through R10 and R8 is
also zero. In steady-state the mid-point voltage of the voltage
divider network is at internal reference voltage potential. The
output voltage can be computed as a function of the resistor
values in the divider network and the internal FEEDBACK pin
reference voltage as:
+ R3 + R4 + R5 +
B+ = c R 2 1 m V FB
R6 + R7
Using typical values for the resistor divider network; R2+R3=
3.1 MW, R4=732 kW, R5=2.2 kW, R6=2.2 kW, and R7=57.6 kW
and the internal reference voltage: VFB = 6 V, we find that the
output voltage (B+) is:
B+ = ` 3100 + 732 + 2.21 + 1 j # 6 = 385. 4 V
2.21 + 57.6
The steady-state current through the divider network sourced
from the output voltage is given as:
I B+ = c
385.4 V
m = 99 nA
^ 3100 + 732 + 2.21 + 2.21 + 57.6h kX
The voltage across the bias resistors R5 and R6 is:
V R5/R6 = 99 nA # 2.21 kX . 219 mV
At this bias level, transistors Q1 and Q2 are in the “off” state
confirming the original assumption for steady-state operation. It
is important to ensure that the voltage across R5 and R6 remain
safely below the VBE(ON) thresholds of Q1 and Q2 in steady-state
operation, which must also include any line frequency ripple
which may be present across the B+ output capacitor.
VC
Control
RC
R
C
PI-6287-011111
Figure 22. Outer Voltage Loop – Small Signal Model.
Loop Compensation – Compensator Design
The simplified low frequency small signal model of the outer loop
is shown in Figure 22. In order to ensure that the input current
has low harmonic distortion while simultaneously maintaining
good output voltage regulation, the loop gain crossover
frequency of the outer loop of a boost PFC is considerably
lower than the line frequency. For frequencies significantly
lower than the switching frequency, the model shown in Figure
22 provides accurate results. The resistor R represents the load
resistance and the capacitance C represents the load capacitance.
Resistance RC is the ESR of the output capacitor C. With a
source resistance equal to the load resistance at all times for a
current mode controlled power source, the dominant pole of
the loop shown in Figure 23 is at 2/RC.
With a control to output response of a single pole for the outer
loop with a very low pole frequency, which is usually less than
one or two hertz, compensation of the converter is achieved by
using an amplifier in the outer voltage loop that provides
sufficient overall gain to improve regulation of output voltage
while simultaneously increasing the loop gain cross over
frequency to achieve ripple reduction.
The inner current loop generally has a high bandwidth to ensure
a low distortion of the current waveform. Although the bandwidth of the inner current loop is higher than the frequency of
the input, supply, it can be significantly lower than the switching
frequency and yet achieve a low current waveform distortion.
Diode D4 is not required for normal operation and is only
required to protect the circuit in case of failure of capacitor C4
resulting in a short-circuit across capacitor C4, or a short
resulting from manufacturing defect. Diode D3 prevents delay
in start-up when the PFC is remotely turned-on by turning the
VCC supply on. If start-up delay after turn-on of VCC is not a
concern, diode D3 is not required and collector of transistor Q1
can be connected directly to VCC.
14
Rev. B 12/11
www.powerint.com
AN-52
Application Note
The small signal control to output voltage loop gain varies with
the square of the input voltage and hence is line voltage
dependent. With a feed-forward of line voltage using the V pin
signal, the loop gain can be made independent of line voltage
and improves line regulation.
A simplified voltage feedback loop for the PFC using HiperPFS
is shown in Figure 23.
The ESR of the output capacitor is not included in this analysis
only because the zero contributed by the ESR of the output
capacitor is at frequencies significantly beyond the loop gain
crossover frequency.
The block diagram in Figure 24 shows the voltage feedback
loop as a block diagram.
VOUT
R1
IBOOST
CO
R3
RL
RZ
R2
VFB
RF
GEA = 100
fPOLE = 1 kHz
VE
CF
CZ
PI-6282-112310
Figure 23. Simplified Representation of Voltage Feedback Loop.
VIN
d
PFS
Modulator
H3(s)
Power Stage
G(s)
VE
Error Amplifier
H2(s)
VREF
VOUT
VFB
Resistor
Divider
H1(s)
PI-6283-112310
Figure 24. Simplified Block Diagram of the Outer Voltage Loop.
15
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Rev. B 12/11
AN-52
The total loop gain of the outer voltage loop is:
R2
1
#
m#
R 1 + R 2 c 1 + s/~F m
1 + s/~Z # G #
1
c
m
EA
c
s/~1 kHz + 1 m
1 + s/~P1
~F =
1
RF # CF
1
~Z =
RZ # CZ
1
~ P1 =
^R3 + RZh # CZ
2.5
2
1.5
0.5
0
0.28
0.35
0.48
0.60
0.75
0.90
1.00
High-Line Input Family
(PFS723-729)
Universal Input Family
(PFS704-716)
25 30 35 40 45 50 55 60 65 70 75 80 85 90 95 100
% of Peak Load Rating
Figure 25. Modulator Gain.
80
60
for a constant power load
40
vt O
1
ti BOOST = s # C O
dBμV
20
vt O
R L /2
tiBOOST =
1 + s/ c 2 m
CO # RL
0
-20
-40
-60
-80
1
10
1
10
fI
100
1000
100
1000
180
135
90
45
dBμV
The PFS Modulator gain is dependent on a number of parameters
and it can be determined from Figure 25. The resulting system
has a loop gain which is a single pole response. As stated
above, the control to output gain of the power stage has a pole
which is at a very low frequency and typically occurs below
2 Hz. The zero contributed by RZ and CZ restores the phase lag
created by the pole associated with R3, RZ and CZ. The zero is
placed in the 4 Hz to 10 Hz region which is at a frequency
above the pole. The high frequency pole contributed by RF, CF
ensures that the system does not respond to noise and has a
gain roll off at higher frequencies. The resulting loop gain and
phase plot are shown in Figure 26.
Scaling Factors
0.28 PFS704
0.35 PFS706
0.48 PFS708
0.60 PFS710
0.75 PFS712
0.90 PFS714
1.00 PFS716
1
t
t
m # c vO m
H 3 ^ s h # G ^ s h = c i BOOST
ti BOOST
vt E
for a resistive load
PFS723
PFS724
PFS725
PFS726
PFS727
PFS728
PFS729
PI-6306-011411
H1 ^ s h # H2 ^ s h = c
Modulator Gain H3(s)
T^ s h = H 1 ^ s h # H 2 ^ s h # H 3 ^ s h # G^ s h
3
PI-6280-011811
Application Note
0
-45
-90
-135
-180
fI
Figure 26. Theoretical Loop Gain for the PFC.
16
Rev. B 12/11
www.powerint.com
AN-52
Application Note
Value of capacitor CZ can be increased to reduce phase lag at
low frequency and restore the gain slope to a near single pole
response.
returning signal together with the difference in their amplitudes
can be plotted as a Bode plot to evaluate the stability of the
converter. The test set-up shown in Figure 27 can be used to
obtain a bode plot
Loop Gain Measurement – Test Set-Up and Load Step
Response Measurement
Measurement of control loop response of a switching power
converter such as the boost PFC designed using the HiperPFS,
requires use of a specialized test set-up. This test set-up uses
a network analyzer to inject a signal into the feedback loop and
measure the amplitude of the signal appearing in the output of
the power supply. The phase shift in the injected signal and the
Loop Gain Measurement – Procedure
• The PFC stage is supplied from a adjustable DC source for
this test
• Connect the circuit as shown in the picture. Open the top end
of the feedback divider network and insert a 100 Ω, 2 W
resistor in series as shown. The signal injected in the loop for
gain-phase measurement will be injected across this resistor.
A
100 Ω
2W
B
D
CONTROL
HiperPFS
High-Voltage
Probes
×100
COAX
to CH2
FB
COAX
to CH1
S
PFC Circuit
using PFS714
Non-Linear
Feedback Circuit
COAX
CPU
Monitor
GPIB
Signal Out
➪➪
CH1 CH2
Venable Model 5060A
(Frequency Response Analyzer)
Venable Injection Box
Model 200-300
PI-6214-020711
Figure 27. Loop Gain Measurement Set-Up.
17
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Rev. B 12/11
Application Note
•
•
AN-52
Nodes A and B (two ends of the injection resistor) are
connected to Channel 1 and Channel 2 of the frequency
response analyzer using high-voltage x100 attenuator probes.
GND leads of both probes are connected to output return as
shown.
The signal to be injected is isolated using the Bode box
injection transformer model – 200-000 from Venable Industries.
Test Procedure:
1. Adjust the input voltage to 150 VDC and confirm that the
PFC output voltage is within regulation limits
2. Adjust the output load to full rated load and reduce the input
voltage to 100 VDC for universal input power supplies and
200 VDC for 230 V only power supplies.
3. Inject a signal from the frequency response analyzer.
4. The injected signal should be seen in the output voltage
ripple of the PFC.
5. Plot the Gain Phase Plot by sweeping the injected signal
frequency from 1 ~ 2 Hz to 90Hz
Loop Gain Measurement – Test Result
Figure 28 shows a typical gain phase plot obtained using the
measurement method described above. The gain-phase plot
shows a healthy phase margin in excess of 60º. It can be seen
from the gain-phase plot that system has a single-pole role off
and the gain goes up with increasing load level.
60
Gain
Blue Trace – 100% Load
Yellow Trace – ≈ 50% Load
Green Trace – ≈ 10% Load
PI-6307-011411
Taking measurements using the procedure provided is often
difficult. Stability can be evaluated using load step response as
shown in Figure 48a. It is necessary to connect the feedback
divider network directly to the output capacitor. The loop
shorting capacitor CSN2 should always be connected directly
between the cathode of the output diode D9 and the SOURCE
pin of the HiperPFS device.
-60
2
Frequency
1k
Figure 28. Gain-Phase Plot Example for a 347 W PFC using HiperPFS PFS714EG.
VCC Decoupling Requirements
A low-ESR decoupling capacitor of at least 1 mF is recommended
to be connected across the VCC and G pins of the HiperPFS.
The VCC pin supplies power to not only the internal control
circuit of the HiperPFS but also the MOSFET driver which draws
pulsating current with each switching transition of the highvoltage MOSFET. Ceramic capacitors are most suitable for this
application and use of surface mount capacitors is recommended
to minimize stray inductance. It is recommended that the VCC
be decoupled using an electrolytic capacitor of at least 10 mF in
addition to the low ESR 1 mF capacitor placed physically close
to the HiperPFS with short trace length between the capacitor
leads and the HiperPFS pins. Note that when using ceramic
capacitors verify the voltage coefficient of the dielectric for the
selected capacitor. Some dielectrics can have a capacitance
-80% of nominal when the rated voltage is applied.
V Pin and FEEDBACK Pin Decoupling Capacitor
Requirements
Switching noise can be easily coupled into the FEEDBACK pin
and the V pin of the HiperPFS. To ensure reliable operation, it is
required that a ceramic dielectric capacitor of 10 nF to 20 nF be
connected from the FEEDBACK pin to the GROUND pin of the
HiperPFS. The V pin needs to be decoupled using a 100 nF
capacitor for the universal input parts and a 47 nF capacitor for
the high-line only parts. The capacitor should be connected
from the V pin to the G pin of the HiperPFS.
Line Sense Network
The line sense resistor is connected from the bridge rectifier
output to the V pin of the HiperPFS. The voltage drop across
this resistor is in excess of 350 V peak when operating at the
higher end of the input voltage range for universal Input or
high-line only applications. It is therefore necessary to divide
this resistor into two or more resistors to distribute the voltage
stress between multiple resistors. The line sense resistors are
required to be 4 MΩ for most universal input applications and
9 MΩ for most 230 V only (180 VAC - 264 VAC) applications.
The resistors should be physically located close to the V pin of
the HiperPFS to prevent noise injection due to the high dv/dt
switching waveforms of the main power loop.
Inrush Limiting
Without the use of any limiting mechanism, the input current of
a boost PFC at start can be significantly large and even exceed
100 A. This current is due to the charging of the output capacitor.
The current is only limited by the parasitic impedance of the
components in the circuit such as the ESR of the output capacitor
and the impedance of the common mode and differential mode
filters. This current can often damage components in the circuit
and result in fuse failure. Power supply specifications typically
limit the magnitude of the inrush current and make it necessary
to use a suitable limiting mechanism to keep this current below
safe limits.
A simple and inexpensive method to limit inrush current is to
use a NTC thermistor in series with the line. The resistance of
the thermistor limits the inrush current at start and the rapid
drop in its resistance thereafter ensures that the effect on
system efficiency is minimal. There are two locations in which
the thermistor can be used. The location TH1 shown in Figure 30
is the preferred location although in this location since the input
current is high when the input voltage is low, it results in some
efficiency degradation especially at lower input voltages. When
used in location TH2, there is a reduced efficiency penalty,
however there is an increase in the MOSFET drain source
18
Rev. B 12/11
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AN-52
Application Note
voltage at switch-off and hence this is not particularly desirable.
For most high performance PFC circuits when the thermistor is
used in location TH1, it is common practice to use a relay to
bypass the thermistor after start-up. This has the advantage
that the highest efficiency is achieved and also it ensures that
the thermistor is cold and hence will limit the inrush current if
the input is switched off and switched on again.
PCB Design Guidelines
The boost PFC circuit is a combination of a high-voltage switch
mode converter, a control circuit consisting of the line sense
network, the output voltage feedback divider network and
compensation elements. The line sense network and the feedback network use large resistance values in order to minimize
power dissipation in the feedback and line sense network.
A third arrangement uses a thermistor in series with the bypass
diode D2 (Figure 30). While this provides some inrush current
limiting, it is not as effective as putting the thermistor in location
TH1, since some inrush current flows through the inductor and
the PFC output diode. Placement of the thermistor in this path is
therefore not recommended.
Care should be taken to place the feedback circuit and the line
sense network away from the high-voltage and high current
nodes to minimize any interference. Any noise injected in the
feedback network or the line sense network will typically
manifest as degradation of power factor. Excessive noise
injection can lead to waveform instability or dissymmetry.
Power Circuit
Control Circuit
PI-6345-011811
Figure 29. Example Layout of Power and Control Circuit for a PFC Stage.
19
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Rev. B 12/11
Application Note
AN-52
The EMI filter components should be clustered together to
improve filter effectiveness. The placement of the EMI filter
components on the circuit board should be such that the input
circuit is located away from the drain node, the output diode of
the PFC and the PFC inductor.
L
A filter or decoupling capacitor should be placed at the output
of the bridge rectifier. This capacitor together with the X
capacitance in the EMI filter and the differential inductance of
the source, works as a filter to reduce the switching frequency
current ripple in the input current. This capacitor also helps to
minimize the loop area of the switching frequency current loop
thereby reducing EMI.
LDM1**
F1
TH2
L1
LCM
E
MOV1
Location 1
CX2
CX1
D1
BR1
CY1
Location 2
CSN1
D2
D
CIN
CY2
HiperPFS
N
V
VCC
CONTROL
S
FB
Compensation
and Feedback
Circuit
CSN2
COUT
RLOAD
G
TH1
*Alternate location for thermistor
**Not required for all designs
PI-6208-011811
PI-6308-1011411
Figure 30. PFC Schematic Showing Locations Where an Inrush Limiter may be Placed.
Upper: Line Current, 2 A / div.
Middle: Output Voltage, 100 V / div.
Lower: Line Voltage, 200 V / div.
Upper: Line Current, 5 A / div.
Middle: Output Voltage, 100 V / div.
Lower: Line Voltage, 200 V / div.
Figure 31. Examples of Waveform Distortion due to Violation of PCB Design Rules.
20
Rev. B 12/11
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AN-52
The connection between the HiperPFS drain node, output
diode anode terminal and the PFC inductor should be kept as
small as possible.
Application Note
terminal of the HiperPFS. This ensures that the loop area of the
loop carrying high frequency currents at the transition of
switch-off of the MOSFET and helps to reduce radiated EMI
due to high frequency pulsating nature of the diode current.
A low loss ceramic dielectric capacitor should be connected
between the cathode of the PFC output diode and the source
PI-6344-011811
Figure 32. Low Loop Area Routing of High-Frequency Loops.
21
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Rev. B 12/11
Application Note
AN-52
During placement of components on the board, it is best to
place the V pin, FEEDBACK pin and VCC pin decoupling
capacitors close to the HiperPFS before the other components
are placed and routed.
V Pin Decoupling
Capacitor
To minimize effect of trace impedance affecting regulation,
output feedback should be taken directly from the output
capacitor positive terminal. The upper end of the line sense
resistors should be connected to the high frequency filter
capacitor connected at the output of the bridge rectifier.
FEEDBACK Pin
Decoupling Capacitor
VCC Pin Decoupling
Capacitor
V Pin Sense
Resistors
Shield
Trace
PI-6343-011911
Figure 33. Placement of Decoupling Capacitor and Control Circuit Components.
22
Rev. B 12/11
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AN-52
Application Note
PCB Design Example
PFC Output
Capacitor
PFC
Output
Second Stage
Converter
HiperPFS
PFC
Inductor
Auxiliary Supply for PFC –
from Standby Power Supply
Input
Capacitor
(CIN)
Bridge
Rectifier
Thermistor
Shorting Relay
EMI Filter
AC Input
PI-6238-012611
Figure 34. PCB Layout Example for System Power Supply Consisting of a PFC and a Second Stage Converter.
Common Layout Problems to Avoid
A poor layout will often result in performance issues that may
be time consuming to analyze and occur especially at the end
of a development cycle when PCB design changes are difficult
to make. Figures 35-38 should be useful in quickly identifying
the root cause and correct the layout. Figures 35-38
schematically show common layout mistakes and the reason
they should be avoided
23
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Rev. B 12/11
Application Note
AN-52
✘ Poor Choice of Location for Termination of Feedback Divider Network
D2
D1
+
R2
L1
VCC
*CSN1
R4
D3
BR1
F1
L
AC E
IN
LCM
CX1
R3
D
CX2
CY2
N
R1
LDM
CY1
C2
S
CIN
CSN2
VCC
CONTROL
HiperPFS
TH1
V
R10
FB
R5
R8
R9
D4
G
Q1
C1
DC
OUT
R6
C3
CFB
CVCC
CV
C4
Q2
R7
PI-6223a-011711
*Optional – required for universal input designs >350 W and 230 V input designs >700 W
Note: Feedback divider network should be connected as close to the output capacitor as possible. Connecting the feedback divider network close to the output
ensures the best possible load regulation.
Figure 35. Feedback Divider Network Location.
✘ Poor Choice of Location for Termination of Line-Sense Resistor
D2
D1
L1
VCC
*CSN1
R1
R4
F1
AC E
IN
LCM
CX1
N
R3
LDM
CY1
CY2
D
CX2
TH1
C2
D3
BR1
L
+
R2
VCC
CONTROL
HiperPFS
CIN
V
S
CSN2
R10
FB
R9
D4
G
Q1
R8
R5
C1
DC
OUT
R6
C3
CFB
*Optional – required for universal input designs >350 W and 230 V input designs >700 W
CV
CVCC
C4
Q2
R7
PI-6223b-011711
Note: Line sense resistor network should only be connected directly to the filter capacitor after the bridge rectifier. If connected as shown in the picture above, it
could result in noise injection into the V pin signal that can result in power limited operation or incorrect brown-out threshold.
Figure 36. Location of Line Sense Network Resistor Termination.
24
Rev. B 12/11
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AN-52
Application Note
✘ Indirect Short Between “G” and “S” Pin on the Circuit Board
D2
D1
L1
VCC
*CSN1
R4
F1
AC E
IN
LCM
CY2
N
R1
LDM
CY1
CX1
C2
D3
BR1
L
+
R2
R3
D
CX2
S
CIN
VCC
CONTROL
HiperPFS
TH1
V
CSN2
R10
FB
R8
R9
D4
G
Q1
R5
C1
DC
OUT
R6
C3
CFB
CVCC
CV
C4
Q2
R7
PI-6223c-011711
*Optional – required for universal input designs >350 W and 230 V input designs >700 W
Note: The lower resistor of the feedback network and the decoupling capacitors for the HiperPFS pins should only be returned to the G pin which is the internal
controller ground. This pin is internally connected to the SOURCE pin. The G pin and the S pin should not be connected externally. If a connection is made as
shown in the figure, it will typically result in erratic input current and poor load regulation.
Figure 37. Signal Ground for Feedback Network.
✘ Poor Grounding of CVCC
D2
D1
L1
VCC
*CSN1
R4
F1
AC E
IN
LCM
CX1
N
R1
LDM
CY1
CY2
R3
D
CX2
TH1
C2
D3
BR1
L
+
R2
VCC
CONTROL
HiperPFS
CIN
V
S
CSN2
R10
FB
R9
D4
G
Q1
R8
R5
C1
DC
OUT
R6
C3
CFB
*Optional – required for universal input designs >350 W and 230 V input designs >700 W
CV
CVCC
C4
Q2
R7
PI-6223d-011711
Note: The VCC decoupling capacitor should be terminated close to the G pin. If the trace connecting this capacitor to the G pin is shared by the feedback circuit,
it will result in common impedance coupling thereby introducing switching noise on the feedback signal resulting in waveform distortion. Inappropriate
decoupling can cause failure of the brown-out feature. See Figure 31 for example of waveform distortion due to this configuration.
Figure 38. Grounding for VCC Decoupling Capacitor.
25
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Rev. B 12/11
Application Note
AN-52
Figure 39 shows the parts of the boost converter stage that
have high frequency switching currents. Parts of these loops
are shared between the two loops shown. The segments that
are shared have a small AC component of current. The segments
that are highlighted have a large AC component of current.
These segments can be a source of significant amount of
electromagnetic interference if their length is large. PCB design
effort should be directed at keeping the length of these segments
small and also keeping the loop area of the two loops shown in
Figure 45 as small as practical.
The segments that carry pulsating currents are highlighted in
Figure 39.
D2
D1
L1
VCC
*CSN1
R4
F1
AC E
IN
LCM
CX1
N
R1
LDM
CY1
CY2
R3
D
CX2
TH1
C2
D3
BR1
L
+
R2
VCC
CONTROL
HiperPFS
CIN
V
S
CSN2
R10
FB
R9
D4
G
Q1
R8
R5
C1
DC
OUT
R6
C3
CFB
*Optional – required for universal input designs >350 W and 230 V input designs >700 W
CV
CVCC
C4
Q2
R7
PI-6223e-011711
Figure 39. Parts of PFC Circuit Carrying Pulsating Currents.
26
Rev. B 12/11
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AN-52
Thermal Design
The eSIP package enables design of compact and high density
PFC design due to its low profile. A heat spreader is recommended
for use with the HiperPFS parts to enhance thermal performance,
especially in universal input designs over 150 W and 230 V only
designs over 300 W. The rear surface of a HiperPFS eSIP part
is the MOSFET Drain connection. This being a high-voltage
switching node, it is essential to isolate the package from the
heat sink. A low thermal resistance silicone rubber insulator
®
such as the Bergquist – Kapton-K10 insulator is recommended
to be placed between the heat spreader and the heat sink.
A heat spreader is a rectangular piece of aluminum or copper.
Application Note
While the physical size may depend on the actual size of heat
sink and mounting arrangement, a larger size yields lower
thermal resistance. The assembly shown in Figure 40 uses a
Kapton K-10 insulator and a 0.76 mm thick aluminum heat
spreader of 16 mm width and 21 mm height. This assembly
provides a junction to heat sink combined thermal resistance of
approximately 3.1 ºC/W for the heat sink shown.
It is important to ensure a good quality surface finish for the
aluminum heat spreader to achieve low thermal resistance and
efficient heat transfer between the heat spreader and the
HiperPFS.
1.SCREW
2. SHOULDER WASHER
3. EDGE CLIP
4.HiperPFS
5. THERMALLY CONDUCTIVE
SILICONE GREASE
6. FIBER WASHER
7. CUSTOM ALUMINUM
HEATSPREADER
8. KAPTON SILPAD INSULATOR
TO-247
9. HEAT SINK
10. FLAT WASHER
11. LOCK WASHER
12.NUT
Figure 40. Heat Sink Assembly Example for Designs >150 W (Universal Input) and >300 W (230 V Input).
Figure 41. Heat Sink Assembly – Side View.
27
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Rev. B 12/11
Application Note
AN-52
Figure 42. Heat Sink Assembly – Front View.
1.SCREW
2. EDGE CLIP
3.HiperPFS
4. KAPTON SILPAD
INSULATOR
5. HEAT SINK
6. FLAT WASHER
7. LOCK WASHER
8.NUT
Figure 43. Heat Sink Assembly Example – Low Power Designs (< 150 W (Universal Input) and < 300 W (230 V Input)).
28
Rev. B 12/11
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AN-52
Application Note
Figure 44. Heat Sink Assembly – Low Power Designs.
It is necessary to apply suitable thermally conductive silicone
grease to the rear surface of the eSIP package to ensure low
thermal impedance between the heat spreader and the eSIP.
method. Thermocouples also suffer from measurement
inaccuracy due to noise pickup from adjacent high-voltage
switching circuitry.
Thermal impedance of the assembly also depends on the
clamping force with which the eSIP assembly is pressed
together. Metallic clips such as the one shown in Figure 43
provide a mounting force of 25 N. Beyond 20 N to 30 N there is
no significant change in the total thermal resistance of the
assembly.
A third method that is often inexpensive is to use RTD
(Resistance Temperature Detector). The RTD devices are
available in packages that have a flat surface that can be attached
to the surface of the device being measured using a highly
conductive epoxy such as Arctic Silver®. The resistance of the
RTD changes as a function of temperature and a simple
ohm-meter or multi-meter with a resistance range can be used
for this measurement. The much larger currents used to
measure the resistance makes this method less noise sensitive.
The heat sink should be connected to the source terminal of the
HiperPFS on the circuit board. This ensures that the heat sink
is a quiet node electrically and hence not a source of EMI.
When using a heat spreader that is electrically connected to the
high-voltage drain node and a heat sink which is electrically
connected to the source terminal, a voltage as high as the
output voltage exists between the heat spreader and the heat
sink. As such care needs to be taken to ensure that there is
sufficient creepage and clearance between surfaces that have
high-voltage between them. One way to achieve the required
clearance is to use a shoulder washer as shown in Figure 40.
In order to evaluate the thermal performance of the assembly, it
is necessary to make temperature measurements. Temperature
can often be measured by using infrared thermometers.
Infrared thermometers being non-contact type, greatly simplify
thermal measurements however care should be taken when
using this method. The emissivity of the surface being measured
is required to be programmed into these instruments without
which there is an error in the measured temperature.
Temperature can also be measured by using thermocouples
glued to the part being measured using a thermally conductive
epoxy. There can be some measurement inaccuracy when
using this method due to the physical size of the thermocouple
junction. Often a reading inaccuracy as high as 10 ºC is noticed
when temperature of flat surfaces is measured using this
Design Considerations for Controlling EMI
The boost converter designed using the HiperPFS is a switching
converter. Without the use of an EMI filter at input, this converter
will not meet conducted and radiated EMI limits imposed by
regulatory agencies. A single stage common mode filter with X
capacitors connected across the line will be generally adequate
for meeting conducted EMI limits for most designs. The
differential inductance of the common mode filter together with
the X capacitors forms a low pass filter that attenuates
switching frequency components in the input current. If this
attenuation is insufficient, in some design a small differential
inductance will often be necessary.
The Drain node of the HiperPFS and the PFC inductor should
be routed away from the input EMI filter. The effectiveness of
the EMI filter is compromised if the switching noise is coupled
into the input wires directly thereby bypassing the EMI filter.
Power Integrations application note AN-15 TOPSwitch Power
Supply Design Techniques for EMI and Safety and application
note AN-53 Active Power Factor Correction - Basics, offer
detailed explanation of causes of conducted and radiated EMI
and guidelines on selection of components for EMI filter.
Addition of X capacitors in the EMI filter will attenuate the switching
frequency components and will reduced low frequency
29
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Rev. B 12/11
Application Note
AN-52
conducted EMI. Excess X capacitance will degrade PF at light
load levels. Power supply specification such as the 80 PLUS
Bronze require that at 230 VAC and 50% load, the input power
factor be better than 0.9. This limits the amount of X capacitance
that can be used in the EMI filter.
L1
D1
+
BR1
A HiperPFS part mounted on a floating heat sink will often
cause unstable PFC operation and also cause radiated and
conducted EMI failures. The heat sink should be connected to
the source terminal of the HiperPFS and as close to the source
terminal as possible.
CIN
AC
IN
Q1
C1
DC
OUT
Loop 3
Loop 1
Loop 2
PI-6222-020711
The windings of the PFC inductor can be a source of V-field and
H-field interference. The PFC inductor should be located away
from the EMI filter components.
It is most economical to minimize EMI at source. Filtering
conducted and radiated EMI often requires use of expensive
components such as X capacitors and inductors. EMI can be
reduced by ensuring shortest possible loop areas of the high
frequency loops or parts of circuit that carry voltages or
currents with sharp rise and fall times.
There are two such loops in the boost PFC designed using the
HiperPFS as shown in Figure 45. The first loop is completed
between input capacitor CIN, Inductor L1 and MOSFET Q1. The
second loop is completed between input capacitor CIN, inductor
L1, output diode D1 and output capacitor C1. The loop area of
these loops should be minimized. The most significant sources
of EMI are the portions of these loops that experience pulsating
currents. The Drain node of the MOSFET has a high-voltage
switching waveform with respect to the source terminal which is
also the circuit common. This makes the copper trace on the
circuit board connected to the drain node a source of V-field.
The trace length of the traces connecting components to this
node should be kept short.
A third loop exists as shown in Figure 45 when the MOSFET
turns on, the reverse recovery current of output diode D1 flows
through the MOSFET. This current is a short duration pulse that
can result in excess EMI if the loop area of this loop is not kept
very small
One way to reduce the loop area of the second loop is to place
a low ESR capacitor from the PFC output diode cathode
L
terminal directly to the MOSFET source terminal. Use of these
capacitors is advised. Generally a 10 nF 1 kV capacitor will
suffice. Use of multilayer surface mount capacitors can
drastically reduce the parasitic lead inductance however these
capacitors have been found to be very fragile. They often
develop cracks easily during PCB assembly or handling or even
due to unequal expansion and contraction of the board due to
thermal cycling. Capacitors that are not coated by the
manufacturer should be avoided as these may suffer from
surface arc-over when a high-voltage is first applied that
initiates a degradation process which finally leads to a
catastrophic failure. Use of high-voltage surface mount
capacitors vs. leaded type should be carefully considered for
application as a loop shortening capacitor in the output of a
boost PFC designed using HiperPFS.
Use of a low ESR and high ripple current capacitor is
recommended for PFC output filter. Use of a low ESR capacitor
ensures low switching ripple in the output of the PFC thereby
reducing EMI coupled from the output stage of the PFC.
Figure 46 shows a typical EMI filter example. Generally a single
common mode inductor will suffice for most applications.
Inductor LDM and capacitor CX3 form a differential low pass filter.
Splitting the inductor LDM and inserting one of the inductors in
series with the inrush limiter shown has the added benefit of
reducing the common mode EMI further. The common mode
filter inductance will typically be over 8 mH and values as high
as 15 mH to 22 mH are not uncommon.
Common
Mode
F1
Differential
Mode
LDM
CY1*
LCM
CX1
TH1
CY2*
Discharge
AC E
IN
N
Figure 45. Critical Loops of Current Resulting in Radiated and Conducted EMI.
BR1
CY3
CX2*
CX3
CY4
*Optional components – not required on all designs
PI-6242-012811
Figure 46. Typical Common Mode and Differential Mode Filter.
30
Rev. B 12/11
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AN-52
Application Note
Although use of a high X capacitance helps in reducing the
differential noise, excess X capacitance degrades the input
power factor at light load and hence is counterproductive.
HiperPFS has variable frequency architecture. Frequency of
operation can vary between devices which can result in slight
variation between EMI signatures of multiple units. When
optimizing EMI performance of a prototype designed using the
HiperPFS, it is recommended that during development the
conducted EMI measurement be made from 140 kHz to 30 MHz.
If a high level of conducted noise is seen in the 140 kHz to the
150 kHz region, the EMI filter should be adjusted to offer sufficient
attenuation for this noise as in some units the operating
frequency may be higher shifting this noise signature to the
150 kHz - 160 kHz region where conducted EMI compliance is
required.
•
•
When designing the EMI filter, care should to be taken to ensure
that the EMI filter has a damped response and does not result
in under damped ringing superimposed on the current waveform.
The effect of this can be seen both in EMI measurements and
also unexpected dips in measured efficiency and PF, typically at
~50% load.
Design for Safety Compliance
Power supplies are required to have capability of withstanding
surge voltages which typically are a result of events such as
lightning strikes. It is expected that such events do not lead to
failure of any components or loss of functionality. Standards
such as IEC61000-4-5 defines surge voltage and current
waveforms as well as source impedance, which emulate typical
worst case transients for testing of protection mechanisms for
line connected power circuits and data line connected
equipment.
•
•
•
•
•
Components of the EMI filter and the capacitors used in the
power supply input stage as well as the PFC output, help in
limiting the voltage and current stress that the components of
the power supply are subjected to during these events.
MOVs will often be required to be added at the input of the
power supply. These MOVs are placed after the input fuse and
help in clamping the voltage at the input of the power supply
when a surge event occurs.
The following checklist can be used to ensure that the design is
compliant to the applicable requirements:
• Define the target market for PFC converter.
• Determine the equipment class to determine Common-Mode
(CM) and Differential-Mode (DM) surge levels.
• Design boost converter front-end: ensure that EMI filter has at
least one CM inductor stage to provide adequate leakage
inductance for spike suppression.
• If DM surge >1000 V, then you will likely need to include an
MOV across the AC line at the front-end of the EMI filter
• Select a MOV for North America 115 VAC or universal input
with adequate stand-off voltage during normal operation as
well as adequate rated surge current and energy capacity.
• An example of selecting an MOV: Assume that you have a
North America application within a Class 3 equipment
•
installation for which you need to select a MOV for differentialmode protection, connected across the AC line. The DM
Spike Energy will be less than 6.9 J. A device rated for
150 VAC continuous operation would provide adequate
stand-off voltage for 115 VAC nominal applications. Littelfuse
part number V150LA5 provides 25 J and 2500 A surge
capability with adequate margin to minimize degraded
performance due to accumulated strikes over the life of the
MOV. For a universal input design, the V320LA10 provides 48
J and 2500 A surge capability.
Conduct both Common-Mode and Differential-Mode surge
tests on the converter and observe voltages across key
components and currents where necessary to validate SOA
operation of components. Verify all voltage and current
extremes are within the rated specification of each X and Y
capacitor. If not, specify a component with a higher rating.
Verify surge transient current rating of the diode bridge used.
Verify rise time of voltage across the film capacitor connected
after the bridge rectifier and ensure that it is within specified
maximum dv/dt for the component selected. If not, specify a
larger capacitor, a capacitor with higher dv/dt rating, or
increase inductance in series with AC line in order to reduce
inrush surge currents.
Verify MOSFET switch BV rating is greater than surge voltage
on switching node. If not, you may need to increase bulk
capacitor size for greater current sink, or clamp drain node
with TVS diode.
Ensure that bulk capacitor surge voltage rating is not exceeded during testing. If surge voltage rating is exceeded, you
may need to increase capacitance.
Ensure that the bulk capacitor surge voltage does not exceed
BV ratings of the second stage DC-DC converter.
Select an AC line fuse which has an I2t rating that will accommodate power-on inrush current at maximum line voltage and
which is rated for continuous AC line current and will not
interrupt due to surge I2t. Do not oversize the fuse more than
necessary to withstand transient currents so as to ensure that
the fuse will interrupt line current in the event of a line-to-line
MOV failure.
Thermistor selection/location: If large differential mode surge
levels are required, it is recommended that the thermistor not
be located in series with the output bulk capacitor, as this
location causes the drain node to rise up to hundreds of volts
above the bulk capacitor (refer Figure 30). A thermistor
located in series with the AC line will increase design robustness during line surges.
When making measurements on a power supply during a line
surge or safety test, care should be taken to ensure that the test
equipment is galvanically isolated. If alternate paths for the
surge energy are created as a result of connection of test
probes, the test result will be incorrect. Care must be taken to
use voltage probes that are rated for measurement of highvoltages in excess of the voltages likely to be encountered
during the test.
Do’s and Don’ts
Location of VCC Decoupling Capacitor
Ensure that the VCC decoupling capacitor is located close to the
IC. If the VCC decoupling capacitor is located further away from
31
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Rev. B 12/11
Application Note
AN-52
the IC and the trace is tapped at additional points for connection
to other circuit sections, it will result in common impedance
coupling and result in unexpected waveform distortion.
Grounding of Feedback Circuit Components
Feedback pin decoupling capacitor and V pin decoupling
capacitors should be placed next to the HiperPFS with minimum
trace length for the interconnecting traces. This maximizes the
effectiveness of these capacitors in filtering any noise coupled
into the FEEDBACK pin or the V pin.
Effects of Floating Heat Sink
The heat sink on which the HiperPFS is assembled, should
never be floated. The heat sink should always be connected
electrically to the source terminal of the PFS. The drain node is
a high-voltage node. Since this node is a switching node, a
small displacement current due to the capacitance between the
heat sink and the PFS flows through the capacitance between
the heat sink and the PFS. A floating heat sink thus becomes a
V-field antenna, coupling noise into the sensitive FEEDBACK pin
and the V pin which can make operation unstable.
This recommendation should be followed even if any additional
power devices are assembled on the same heat sink.
This typically manifests as poor PF or waveform dissymmetry or
occasional glitch in the waveform.
Measurement Techniques
Drain Source – Voltage Measurement
When measuring Drain-Source voltage of the MOSFET, a
high-voltage probe should be used. When the probe tip is
removed, a silver ring in the vicinity of the probe tip can be
seen. This ring is at ground potential and the ultimate ground
connection is at this point. A stiff wire can be wrapped around
the ground ring and then the probe can be connected to the
drain and source terminals of the HiperPFS with the shortest
possible wire length.
Inductor – Drain Current Measurement
A DC current probe is typically the most useful tool to measure
the inductor current and input current of the PFC. When
measuring the Inductor current, the probe should be inserted
between the bridge rectifier and the inductor. The bridge
Short Wire
Loop
Current
Probe
VRECT
L5
D2
JP4
DC Input
After Bridge
Rectifier
RTN
Do Not Open
Jumper J4
D
S
C21
C14
To DC
Output
rectifier end of the inductor is a relatively quiet node as compared
to the end of the inductor which is connected to the MOSFET
drain and the output diode. This minimizes common mode
noise coupling into the probe thereby avoiding distortion in the
measurement.
Most of the required information regarding the drain current at
the turn-on and turn-off of the MOSFET can be obtained from
the inductor current waveform. It is therefore recommended
that the current probe should not be inserted in the connection
between the drain node and the output diode-inductor junction.
Inserting the current probe between the drain node and the
output diode-PFC inductor connection requires that a small
loop of wire be inserted in this location as shown in Figure 47.
This loop contributes some stray inductance which causes
increased voltage spike across the MOSFET at turn-off.
No-Load Power and PF Measurement
The HiperPFS features an EcoSmart mode which results in
burst mode of operation when the PFC is unloaded. The timing
between the bursts is dependent on the size of the output
capacitor and hence can vary between designs. It is often
difficult to accurately measure input power to the circuit when
the input power is pulsating at a low frequency. Many power
meters feature special modes that allow for a long integration
interval which can be used under such conditions. The power
meter modes should be carefully programmed to achieve an
error free measurement. The EPRI (Electric Power Research
Institute) recommends inserting a LISN between the AC source
used and the equipment under test. A 1 mF X capacitor is also
required to be connected at the output of the LISN. The power
analyzer should be connected close to the equipment under
test so that voltage drops in the interconnecting wires do not
result in measurement errors. This arrangement ensures that
the impedance mismatch between the AC source and the PFC
does not result in unwanted behavior such as source voltage
distortion or oscillation which typically results in inaccuracy of
no-load input power measurement or PF measurement at light
loads.
Tips for Design and Performance Improvement
PF Improvement with Higher Inductance
The size of the PFC inductor is a trade-off between cost and
required performance in terms of PF and THD of input current.
For some designs it is a requirement that the input PF be above
a certain value at 50% load. If the inductor value selected is
failing to provide the required PF, the PF can be generally
improved by increasing the inductance value.
PF at light load can often degrade because of excessive X
capacitance in the EMI filter. Before increasing the size of the
inductor, value of the X capacitance should be checked. If the
conducted EMI measurement shows sufficient margin, it may
be possible to improve PF at light load by reducing the size of
the X capacitance.
Scope Probes
PI-6215-011711
Figure 47. Current Probe and Scope Probe Jack Insertion Locations.
Selection of Suitable Diodes
Diode choice can have significant effect on overall performance.
It is often difficult to gauge diode performance on the basis of
32
Rev. B 12/11
www.powerint.com
AN-52
Application Note
the data presented in the data sheet of the diode. Generally
ultrafast diodes with soft recovery characteristics are best
suited for PFC application.
Change of diode can have a very significant effect on efficiency.
Diode choice should be carefully evaluated based on efficiency
measurement. For high performance applications, use of a
Silicon Carbide (SiC) Schottky diode may be considered if the
increased cost is acceptable. Efficiency improvement of the
order of 0.5% is possible when Silicon Carbide Schottky diodes
are used instead of ultrafast recovery diodes.
A number of ultrafast recovery diodes are available however
their data sheets often lack sufficient information to ensure that
they are suitable. Reverse recovery parameters are often
specified at current levels significantly lower than the rated
current. Such diodes should be treated with caution and their
performance carefully verified before use.
Several manufacturers provide diodes that are specially
designed for PFC applications including Qspeed diodes from
Power Integrations. These diodes have soft recovery
characteristics with low QRR and can provide efficiency
performance between that of ultrafast and SiC diodes.
Powder Core Inductor – Potential Benefits
The variable frequency architecture of the HiperPFS provides
significant improvement in light load efficiency as compared to
traditional fixed frequency architectures. The higher the value of
Improving Efficiency
Efficiency improvement can be achieved by:
• Reducing the temperature of the parts by improving the
thermal design
• Using Litz wire instead of magnet wire for the inductor
• Using a core material with lower core loss. Iron powder
inductors will generally result in a lower efficiency.
• Using a high performance PFC output diode
• Using a low ESR output capacitor
• Using the next higher HiperPFS part
Bring-up Procedure – Feedback Circuit Check
A defective feedback circuit will result in inadequate feedback
voltage and could lead to catastrophic failure of the HiperPFS or
the output capacitor. If the defect in the feedback circuit results
in a floating FEEDBACK pin or a low voltage on the FEEDBACK
pin, that will prevent the HiperPFS from switching and will not
result in any failure.
(b)
PI-6309-121010
(a)
the inductance, the lower is the load level at which the PFC will
become discontinuous. The PF and waveform THD performance
is superior when the converter remains in CCM mode of operation.
It is often difficult to achieve a high inductance value using
ferrite cores. Powder cores such as Sendust, Kool-mu and
MPP yield a permeability that drops with DC bias. These cores
can be effectively used to design inductors that offer higher
inductance at light load and the inductance drops with DC bias
resulting in a lower inductance at higher load levels. This can
significantly improve PF at light load levels.
Upper: Line Current, 5 A / div.
Middle: Output Voltage AC Coupled, 20 V / div.
Lower: Line Voltage, 200 V / div.
Upper: Line Current, 5 A / div.
Middle: Output Voltage AC Coupled, 20 V / div.
Lower: Line Voltage, 200 V / div.
Figure 48. Waveforms – Effect on Non-Linear Amplifier on PFC Response (a). Effect of Linear Amplifier on PFC Response (b).
33
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Rev. B 12/11
Application Note
One easy way to determine if the feedback divider network is
correctly wired is to connect a bench power supply to the
output of the PFC and raise its voltage gradually. If the external
voltage source is raised to a level such that at a voltage equal to
the rated PFC output voltage, the FEEDBACK pin voltage is
around 6 V, which indicates that the FEEDBACK pin resistor
divider is correctly configured. This simple test can often
prevent damage at start if there are any mistakes in the
feedback circuit divider network.
Verification of Non-Linear Amplifier Using Dynamic Load
Response
The non-linear amplifier which is a part of the feedback network
is made using a set of NPN and PNP transistors. The transistors
are biased in a way that they do not conduct during normal
operation. When the output experiences undershoot or
overshoot, due to dynamic load changes, these transistors
conduct momentarily to rapidly correct the feedback pin voltage
in order to ensure a rapid response. The examples shown in
Figure 48b shows the difference in performance if the PNP
transistor is removed from the circuit. A step load response is
an easy way to verify the operation of the transistors.
AN-52
•
•
•
Maximum drain voltage – Verify that peak VDS does not exceed
530 V at lowest input voltage and maximum overload output
power. Maximum overload output power occurs when the
output is overloaded to a level just above the highest rated
load or before the power supply output voltage starts falling
out of regulation.
Maximum drain current – At maximum ambient temperature,
minimum input voltage and maximum output load, verify drain
current waveforms at start-up for any signs of inductor
saturation and excessive leading edge current spikes. HiperPFS
has a leading edge blanking time of 220 ns to prevent premature
termination of the ON-cycle. Verify that the leading edge
current spike is below the allowed current limit for the drain
current waveform at the end of the 220 ns blanking period.
Thermal check – At maximum output power, minimum input
voltage and maximum ambient temperature; verify that
temperature specifications are not exceeded for the HiperPFS,
PFC inductor, output diodes and output capacitors. Enough
thermal margin should be allowed for the part-to-part variation
of the RDS(ON) of HiperPFS, as specified in the data sheet. A
maximum package temperature of 110 °C is recommended to
allow for these variations.
Quick Design Checklist
As with any power supply design, all HiperPFS designs should
be verified on the bench to make sure that component
specifications are not exceeded under worst-case conditions.
The following minimum set of tests is strongly recommended:
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www.powerint.com
AN-52
Application Note
Appendix A – Application Example
A High Efficiency, 347 W, 380 VDC Universal Input PFC
The circuit shown in Figure 11 is designed using a PFS714EG
device from the HiperPFS family of integrated PFC controllers.
This design is rated for a continuous output power of 347 W
and provides a regulated output voltage of 380 VDC nominal
maintaining a high input power factor and overall efficiency from
light load to full load.
Fuse F1 provides protection to the circuit and isolates it from the
AC supply in case of a fault. Diode bridge BR1 rectifies the AC
input. Capacitors C3, C4, C5, C6 and C19 together with
inductors L1, L2, L3 and L4 form the EMI filter reducing the
common mode and differential mode noise. Resistors R1, R3
and CAPZero, IC U2 are required to discharge the EMI filter
capacitors once the circuit is disconnected. CAPZero
eliminates static losses in R1 and R2 by only connecting these
components across the input when AC is removed.
The boost converter stage consists of inductor L5, diode rectifier
D2 and the HiperPFS IC U1. This converter stage works as a
boost converter and controls the input current of the power
supply while simultaneously regulating the output DC voltage.
Diode D1 prevents a resonant build up of output voltage at startup by bypassing inductor L5 while simultaneously charging
output capacitor C15. Thermistor RT1 limits the inrush input
current of the circuit at start-up and prevents saturation of L5.
In most high-performance designs, a relay will be used to
bypass the thermistor after start-up to improve power supply
efficiency. Therefore efficiency measurement, that represents
the high performance configuration, the thermistors should be
shorted. Capacitors C14 and C21 are used for reducing the
loop length and area of the output circuit to reduce EMI and
overshoot of voltage across the drain and source of the
MOSFET inside U1 at each switching instant.
The PFS714EG IC requires a regulated supply of 12 V for
operation and must not exceed 13.4 V. Resistors R6, R16, R17,
Zener diode VR1, and transistor Q3 form a shunt regulator that
prevents the supply voltage to IC U1 from exceeding 12 V.
Capacitors C8, C18 and C20 filter the supply voltage and
provide decoupling to ensure reliable operation of IC U1. Diode
D5 prevents destruction of U1 if the auxiliary input is inadvertently
connected reverse polarity.
The rectified AC input voltage of the power supply is sensed by
IC U1 using resistors R4, R5 and R19. The capacitor C12 filters
any noise on this signal.
Divider network comprising of resistors R9, R10, R11, R12, R13,
and R14 are used to scale the output voltage and provide
feedback to IC U1. The circuit comprising of diode D4,
transistor Q1, Q2 and the resistors R12 and R13 form a nonlinear feedback circuit which improves the load transient
response by improving the response time of the PFC circuit.
Resistor R7, R8, R15, and capacitors C13 and C17 are required
for shaping the loop response of the feedback network. The
combination of resistor R8 and capacitor C13 provide a low
frequency zero and the resistor R15 and capacitor C13 form a
low frequency pole.
D1
1N5408
L2
100 µH
F1
6.3 A
L
380 VDC
BR1
RT1 GBU806
10 Ω 600 V
L5
1.38 mH
t
O
D1
CAPZero
U2
CAP006DG
C3
680 nF
275 VAC
C4
680 pF
250 VAC
L1
14 mH
C19
1 µF
310 V
RV1
320 VAC
E
D2
R2
220 kΩ
R18
10 Ω
2W
C6
100 nF
275 VAC
R4
1.5 MΩ
1%
R9
1.5 MΩ
1%
R19
1.5 MΩ
1%
R11
732 kΩ
1%
R5
1 MΩ
1%
R10
1.6 MΩ
1%
C16
100 nF
200 V
C5
680 pF
250 VAC
N
L3
100 µH
L4
Ferrite Bead
R6
100 Ω
R16
100 Ω
*Optional Component
D5
DL4001
D
VR1
BZX84C12LT1G
C8
47 µF
50 V
C14
10 nF
1 kV
C15
270 µF DC
450 V OUT
Q1
MMBT4401
R17
3.01 kΩ
1%
Auxiliary
Power
Supply
C21
10 nF
1 kV
D4
1N4148
Q3
MMBT4401LT1G
+
C20
100 nF
50 V
S
C12
100 nF
50 V
VCC
CONTROL
HiperPFS
U1
PFS714EG
C7
1 µF
400 V
V
FB
G
C11
10 nF
50 V
R15
160 kΩ
R7
2 kΩ
D3
BAV116
130 V
R8
3.01 kΩ
1%
C18
1 µF
25 V
C13
4.7 µF
25 V
Q2
MMBT4403
R1
220 kΩ
+
D2
STTH8S06D
R13
2.21 kΩ
1%
R14
57.6 kΩ
1%
R12
2.21 kΩ
1%
C17
470 pF
100 V
PI-6197-111110
Figure 49. 347 W PFC using PFS714EG.
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Application Note
AN-52
Appendix B – Failure Mode Analysis Summary
Device Level Failure Mode Analysis
Figure 50 is the device level failure mode analysis including the
system effects of an open-circuit for each pin as well as
adjacent pin-pin shorts. In each case a safe failure is expected.
Feedback Network Failure Mode Analysis
Table 4 is the feedback network level failure mode analysis
including the system effects of an open and short-circuit
conditions for each component. In each case a safe failure is
expected. [Refer Figure 18]
Figure 50. Device Lev el Failure Mode Analysis.
Open
Short
R2/R3
Lost feedback signal;
Q2 pulls FEEDBACK pin below FBUV -> IC OFF
Reduced voltage divider; output voltage reduced
R4
Lost feedback signal;
Q2 pulls FEEDBACK pin below FBUV -> IC OFF
Reduced voltage divider; output voltage reduced
R5
Lost feedback signal;
Q2 pulls FEEDBACK pin below FBUV -> IC OFF
Reduced voltage divider; output voltage reduced;
fast UV threshold increases
R6
Lost feedback signal; feedback above FBOV -> IC OFF
Reduced voltage divider; output voltage reduced;
fast UV threshold increases
R7
Lost feedback signal; feedback above FBOV -> IC OFF
Q2 pulls FEEDBACK pin below FBOV -> IC OFF
R8
Lost feedback signal;
internal IFB pulls feedback above FBOV -> IC OFF
Fast loop disabled; lost of loop speed-up circuit
R9
No loop compensation; unstable operation; poor
power factor
Poor loop compensation; unstable operation; poor
power factor
R10
Lost feedback signal;
Internal IFB pulls feedback above FBOV -> IC OFF
Increased noise susceptibility; unstable operation; poor
power factor
C2
Loss of soft-start; increased output overshoot at start-up
Reduced voltage divider; B+ reduced
C4
No loop compensation, unstable operation; poor
power factor
FEEDBACK pin below FBUV -> IC OFF
Increased noise susceptibility; unstable operation; poor
power factor
Feedback pulled below FBUV -> IC OFF
No affect to circuit operation
Loss of fast OV/UV loop
Q1-E
Loss of fast OV loop
Short to Q1-B: Feedback above FBOV -> IC OFF
Short to Q1-C: Feedback above FBOV -> IC OFF
Short to Q1-C: Feedback above FBOV -> IC OFF
CFB
C4
Q1-B
Loss of fast OV loop
Q1-C
Loss of fast OV loop
Q2-E
Loss of fast OV loop
Short to Q2-B: No loop compensation
Short to Q2-E: Feedback below FBUV -> IC OFF
Q2-B
Loss of fast UV loop
Short to Q2-C: FEEDBACK pin below FBUV -> IC OFF
Q2-C
Loss of fast UV loop
Table 4. Feedback Network Failure Mode Analysis.
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AN-52
Application Note
Appendix C – Troubleshooting Matrix
Problem Symptoms
No voltage at the output of the PFC
Input fuse blows at start-up / HiperPFS blows
at start-up
Output voltage goes up above 400 V and
decreases slowly
Output voltage is permanently over 105% of
the nominal output rating
PFC output voltage is below the nominal
voltage permanently
Possible Causes of the Problem
Solutions to the Problem
No AC input voltage at bridge rectifier input
terminals.
1. Check voltage at the input of the bridge
rectifier.
2. Check components and connections in the
EMI filter stage including the thermistor.
Input fuse is open.
1. Verify input fuse is functional.
Components in the power stage are
incorrectly assembled or damaged.
1. Verify polarities of the components used
(Ex. – Bridge rectifier polarity).
2. Verify interconnections between
components. Check connection between
PFC output diode and HiperPFS Drain
terminal.
HiperPFS Drain terminal is shorted to the
Source terminal on the circuit board.
1. Check for solder bridging and short-circuits
on PCB.
2. Verify if HiperPFS is damaged with Drain
and Source short.
3. Verify PFC output diode D1 polarity and
make sure the part is functional.
Failed insulation between HiperPFS and heat
sink.
1. Verify heat sink assembly.
2. Check for damaged insulator.
3. Violation of creepage or clearance distance
in heat sink assembly causes arcing,
resulting in failure of HiperPFS.
Insufficient fuse current rating.
1. Verify fuse rating using PIXls.
2. Replace fuse with sufficient current rating
and I2t rating.
Overload at output.
1. Check the load current.
Load is disconnected intermittently when the
PFC is on.
1. Turn off the AC input and verify the
connection to the load.
NPN transistor Q1 is not functional.
1. Check transistor Q1 connection.
2. Check if diode D3 is open or its polarity is
reversed.
3. Check if capacitor C2 is open.
Incorrect values of feed back circuit
components.
1. Verify values of components used in the
feedback circuit.
Defective or missing capacitors in feed back
network.
1. Replace the defective capacitors and install
any missing components.
Input voltage peak is higher than the VOUT
nominal rating.
1. Verify the PFC input voltage.
Transistor Q2 conducts during normal
operation.
1. Verify values of components used in the
feed back circuit.
2. Confirm transistor Q2 is not defective or
connected incorrectly.
Voltage divider network has a shorted resistor
or resistors with incorrectly values.
1. Check short connections of R2/R3/R4/R5/
R6/C2.
2. Check line-sensing network resistors R1.
VCC of the HiperPFS part is below specified
limits.
1. Verify voltage across VCC and GROUND
pins is greater than 10.2 V.
37
www.powerint.com
Rev. B 12/11
Application Note
Problem Symptoms
AN-52
Possible Causes of the Problem
HiperPFS pins are incorrectly connected on
the circuit board.
1. Confirm that none of the HiperPFS pins are
open or are accidentally connected to the
adjacent pins on the circuit board.
Output capacitor polarity is reversed.
1. Verify polarity of the output capacitor.
FEEDBACK pin is floating.
1. Check feedback circuit components R2/
R3/R4/R5/R6/R7/R8/R10 and ensure that
correct component values are used.
Low-voltage on FEEDBACK pin.
1. Check feedback circuit components R7, C4
and CVCC for likely short.
2. Check PNP transistor Q2 base, emitter and
collector terminals for accidental short
between the pins.
High-voltage on FEEDBACK pin.
1. Check for short between collector, base
and emitter of NPN transistor Q1
V pin current is less than IUV-
1. Verify the values of line sensing resistor R1
and its connections.
PNP transistor Q2 is not functional.
1. Check Q2 connections.
2. Check function of Q2.
3. Check bias of Q2 refer to AN-52 for more
information on waveforms.
Transistors Q1 and Q2 are both not functional.
1. Check if resistor R8 is shorted or of
incorrect value.
2. Check values of resistors R5 and R6.
Loss of loop-compensation.
1. Check the values of components R8, R9
and capacitor C4.
Feedback compensation is incorrect.
1. Check for component values of
components R8, R9, R10, C4 and CFB.
Noise injection in the feedback circuit.
1. Verify layout according to data sheet
recommendation.
2. Verify the placement of decoupling
capacitors.
3. Check the PCB layout of ground loops.
Noise injection in the V pin signal.
1. Verify layout according to data sheet
recommendation.
2. Verify the placement of V pin decoupling
capacitor.
3. Check the layout of ground loops in PCB.
Noise injection due to floating heat sinks.
1. Connect heat sinks close to the source
terminal with the shortest possible trace.
Undersized PFC inductor used.
1. Verify the inductor L1 value used.
Undersized bridge rectifier decoupling
capacitor.
1. Choose the correct value and low ESR/ESL
capacitor to limit the high frequency ripple
across this capacitor (CIN).
Incorrect EMI filter stage components used.
1. Inadequate differential mode filter or filter
resonance is resulting in instability. Verify
input current waveform for signs of
resonance.
Excessive voltage ripple on VCC pin.
1. Verify VCC pin decoupling capacitor and
filter capacitor value and circuit board
layout.
2. Verify the operation of the linear regulator if
used.
3. Reduce high frequency switch noise on VCC.
Excess noise on VCC pin.
1. Inadequate VCC pin decoupling capacitor
and filter capacitor.
2. Verify the operation of the linear regulator if
used.
3. Reduce high frequency switch noise on VCC.
4. Check for layout errors which results in
poor grounding of decoupling capacitor.
PFC does not start-up.
VOUT voltage undershoot >40 V during load
transient.
Excessive output voltage overshoot and
undershoot during startup or during a load
transient.
Unstable operation and/or poor input power
factor.
HiperPFS part fails to brown-in.
Solutions to the Problem
38
Rev. B 12/11
www.powerint.com
AN-52
Application Note
Problem Symptoms
Huge resonant voltage ringing on VDS during
start up, which results in HiperPFS failure.
Brown-in / Brown-out at incorrect voltage
levels.
Possible Causes of the Problem
Solutions to the Problem
By pass diode D2 is not functional or it is
missing.
1. Replace bypass diode D2 with a diode with
appropriate current rating.
Incorrect line-sense resistor value.
1. Verify line sensing network resistor R1 value
(recommend 4 MW for universal input parts,
9 MW for 230 VAC only parts. 1% tolerance
parts are recommended).
Note: For reference diagram, refer to Figure 4.
39
www.powerint.com
Rev. B 12/11
Revision
Notes
Date
A
Initial Release
02/11
B
Updated Equations on page 14
12/11
For the latest updates, visit our website: www.powerint.com
Power Integrations reserves the right to make changes to its products at any time to improve reliability or manufacturability. Power
Integrations does not assume any liability arising from the use of any device or circuit described herein. POWER INTEGRATIONS MAKES
NO WARRANTY HEREIN AND SPECIFICALLY DISCLAIMS ALL WARRANTIES INCLUDING, WITHOUT LIMITATION, THE IMPLIED
WARRANTIES OF MERCHANTABILITY, FITNESS FOR A PARTICULAR PURPOSE, AND NON-INFRINGEMENT OF THIRD PARTY RIGHTS.
Patent Information
The products and applications illustrated herein (including transformer construction and circuits external to the products) may be covered
by one or more U.S. and foreign patents, or potentially by pending U.S. and foreign patent applications assigned to Power Integrations. A
complete list of Power Integrations patents may be found at www.powerint.com. Power Integrations grants its customers a license under
certain patent rights as set forth at http://www.powerint.com/ip.htm.
Life Support Policy
POWER INTEGRATIONS PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES OR
SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF POWER INTEGRATIONS. As used herein:
1. A Life support device or system is one which, (i) is intended for surgical implant into the body, or (ii) supports or sustains life, and (iii) whose failure to perform, when properly used in accordance with instructions for use, can be reasonably expected to result in significant
injury or death to the user.
2. A critical component is any component of a life support device or system whose failure to perform can be reasonably expected to cause
the failure of the life support device or system, or to affect its safety or effectiveness.
The PI logo, TOPSwitch, TinySwitch, LinkSwitch, DPA-Switch, PeakSwitch, CAPZero, SENZero, LinkZero, HiperPFS, HiperTFS, HiperLCS,
Qspeed, EcoSmart, Clampless, E-Shield, Filterfuse, StakFET, PI Expert and PI FACTS are trademarks of Power Integrations, Inc. Other
trademarks are property of their respective companies. ©2011, Power Integrations, Inc.
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