MIC4606 - Micrel

MIC4606
85V Full-Bridge MOSFET Drivers with
Adaptive Dead Time and Shoot-Through
Protection
Features
General Description
The MIC4606 is an 85V full-bridge MOSFET driver that
features adaptive dead time and shoot-through protection.
The adaptive dead time circuitry actively monitors both
sides of the full-bridge to minimize the time between highside and low-side MOSFET transitions, thus maximizing
power efficiency. Antishoot-through circuitry prevents
erroneous inputs and noise from turning both MOSFETs of
each side of the bridge on at the same time.
The MIC4606 also offers a wide 5.5V to 16V operating
supply range to maximize system efficiency. The low 5.5V
operating voltage allows longer run times in batterypowered applications. Additionally, the MIC4606’s
adjustable gate drive sets the gate drive voltage to VDD for
optimal MOSFET RDS(ON), which minimizes power loss due
to the MOSFET’s RDS(ON).
The MC4606-1 features four independent inputs while the
MIC4606-2 utilizes two PWM inputs, one for each side of
the H-bridge. The MIC4606-1 and MIC4606-2 are
available in a 16 pin 4 ₓ 4 QFN package with an operating
temperature range of -40°C to 125°C.
Datasheets and support documentation are available on
Micrel’s web site at: www.micrel.com.
• 5.5V to 16V gate drive supply voltage range
• Advanced adaptive dead time
• Intelligent shoot-through protection
− MIC4606-1: 4 Independent TTL inputs
− MIC4606-2: 2 PWM inputs
• Enable input for on/off control
• On-chip bootstrap diodes
• Fast 35ns propagation times
• Drives 1000pF load with 20ns rise and fall times
• Low power consumption: 235µA total quiescent current
• Separate high- and low-side undervoltage protection
• –40°C to +125°C junction temperature range
Applications
•
•
•
•
•
Full-bridge motor drives
Power inverters
High Voltage step-down regulators
Distributed power systems
Stepper motors
Typical Application
85V Motor Drive Configuration
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
September 8, 2014
Revision 1.0
Micrel, Inc.
MIC4606
Ordering Information
Input
Version
Junction Temperature Range
Package(1)
MIC4606-1YML
TTL
Dual Inputs
–40° to +125°C
16-Pin 4ₓ4 QFN
MIC4606-2YML
TTL
Single PWM Inputs
–40° to +125°C
16-Pin 4ₓ4 QFN
Part Number
Note:
1. QFN is a GREEN, RoHS-compliant package. Lead finish is NiPdAu. Mold compound is Halogen Free.
Pin Configuration
MIC4606-1
MIC4606-2
16-Pin 4mm x 4mm QFN (ML)
(Top View)
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MIC4606
Pin Description
Pin Number
MIC4606-1
Pin Name
MIC4606-2
Pin Name
1
NC
NC
No connect.
Pin Name
2
AHB
AHB
Phase A high-side bootstrap supply. An external bootstrap capacitor is required.
Connect the bootstrap capacitor between this pin and AHS. An on-chip bootstrap
diode is connected from VDD to AHB.
3
AHO
AHO
Phase A high-side drive output. Connect to the external high-side power MOSFET
gate.
4
AHS
AHS
Phase A high-side drive reference connection. Connect to the external high-side
power MOSFET source terminal. Connect a bootstrap capacitor between this pin and
AHB.
5
ALO
ALO
Phase A low-side drive output. Connect to the external low-side power MOSFET
gate.
6
VDD
VDD
Input supply for gate drivers. Decouple this pin to VSS with a >1.0μF capacitor.
7
VSS
VSS
Driver reference supply input. Connect to the power ground of the external circuitry.
8
BLO
BLO
Phase B low-side drive output. Connect to the external low-side power MOSFET
gate.
9
BHS
BHS
Phase B high-side drive reference connection. Connect to the external high-side
power MOSFET source terminal. Connect a bootstrap capacitor between this pin and
BHB.
10
BHO
BHO
Phase B high-side drive output. Connect to the external high-side power MOSFET
gate.
11
BHB
BHB
Phase B high-side bootstrap supply. An external bootstrap capacitor is required.
Connect the bootstrap capacitor between this pin and BHS. An on-chip bootstrap
diode is connected from VDD to BHB.
12
EN
EN
Enable input. A logic high on the enable pin results in normal operation. A logic low
disables all outputs and places the driver into a low current shutdown mode. Do not
leave this pin floating.
13
BHI
—
Phase B high-side drive input.
13
—
BPWM
14
BLI
—
Phase B low-side drive input.
14
—
NC
No connect.
15
ALI
—
Phase A low-side drive input.
15
—
NC
No connect.
16
AHI
—
Phase A high-side drive input.
16
—
APWM
EP
ePad
ePad
September 8, 2014
Phase B PWM input for single input signal drive.
Phase A PWM input for single input signal drive.
Exposed thermal pad. Connect to VSS. A connection to the ground plane is
necessary for optimum thermal performance.
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Micrel, Inc.
MIC4606
Absolute Maximum Ratings(2)(6)
Operating Ratings(3)
Supply Voltage (VDD, VxHB – VxHS) ................... −0.3V to 18V
Input Voltages (VxLI, VxHI, VEN) .............. −0.3V to VDD + 0.3V
Voltage on xLO (VxLO) .......................... −0.3V to VDD + 0.3V
Voltage on xHO (VxHO) .................. VHS − 0.3V to VHB + 0.3V
Voltage on xHS (continuous) ............................. −1V to 90V
Voltage on xHB ............................................................ 108V
Average Current in VDD to HB Diode ....................... 100mA
Storage Temperature (Ts) ......................... −60°C to +150°C
ESD Rating(4)
HBM ......................................................................... 1kV
MM ......................................................................... 200V
Supply Voltage (VDD) [decreasing VDD] ........... 5.25V to 16V
Supply Voltage (VDD) [increasing VDD] .............. 5.5V to 16V
Enable Voltage (VEN) ............................................ 0V to VDD
Voltage on xHS .................................................. −1V to 85V
Voltage on xHS (100ns repetitive transient) ...... −5V to 90V
HS Slew Rate ............................................................ 50V/ns
Voltage on xHB ..................................................... VHS + VDD
and/or........................................... VDD - 1V to VDD +85V
Junction Temperature (TJ) ........................ –40°C to +125°C
Junction Thermal Resistance
4mm ₓ 4mm QFN-16-pin (θJA) ........................... 51°C/W
Electrical Characteristics(5)(6)
VDD = VxHB = 12V; VEN = 5V; VSS = VxHS = 0V; No load on xLO or xHO; TA = 25°C, unless noted.
Bold values indicate –40°C≤ TJ ≤ +125°C.
Symbol
Parameter
Typ.
Max.
Units
xLI = xHI = 0V
200
350
µA
EN = 0V with xHS = floating;
2.5
5
EN = 0V, xLI, xHI = 12V or 0V
40
100
0.35
0.5
mA
Condition
Min.
Supply Current
IDD
VDD Quiescent Current
IDDSH
VDD Shutdown Current
IDDO
VDD Operating Current
fS = 20kHz
IHB
Total xHB Quiescent Current
xLI = xHI = 0V or xLI = 0V and xHI =5V
35
75
µA
IHBO
Total xHB Operating Current
fS = 20kHz
30
400
μA
IHBS
xHB to VSS Quiescent Current
VxHS = VxHB = 90V
0. 5
5
µA
IHBSO
xHB to VSS Operating Current
fS = 20kHz
3
10
µA
0.8
V
Input (TTL: xLI, xHI, EN)
µA
(6)
VIL
Low-Level Input Voltage
VIH
High-Level Input Voltage
VHYS
Input Voltage Hysteresis
RI
Input Pull-Down Resistance
2.2
V
0.1
V
xHI/xLI inputs
100
300
500
kΩ
xPWM inputs
50
150
250
kΩ
4.0
4.4
4.9
V
Under-Voltage Protection
VDDR
VDD Falling Threshold
VDDH
VDD Threshold Hysteresis
VHBR
xHB Falling Threshold
VHBH
xHB Threshold Hysteresis
0.25
4.0
4.4
0.25
V
4.9
V
V
Notes:
2. Exceeding the absolute maximum ratings may damage the device.
3. The device is not guaranteed to function outside its operating ratings.
4. Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5kΩ in series with 100pF.
5. Specification for packaged product only
6. x in front of a pin name refers to either A or B. (e.g. xHI can be either AHI or BHI).
7. VIL (MAX) = maximum positive voltage applied to the input which will be accepted by the device as a logic low.
VIH (MIN) = minimum positive voltage applied to the input which will be accepted by the device as a logic high.
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MIC4606
Electrical Characteristics(5)(6) (Continued)
VDD = VxHB = 12V; VEN = 5V; VSS = VxHS = 0V; No load on xLO or xHO; TA = 25°C, unless noted.
Bold values indicate –40°C≤ TJ ≤ +125°C.
Symbol
Parameter
Condition
Min.
Typ.
Max.
Units
Bootstrap Diode
VDL
Low-Current Forward Voltage
IVDD-xHB = 100µA
0.4
0.70
V
VDH
High-Current Forward Voltage
IVDD-xHB = 50mA
0.7
1.0
V
RD
Dynamic Resistance
IVDD-xHB = 50mA
3
5.0
Ω
LO Gate Driver
VOLL
Low-Level Output Voltage
IxLO = 50mA
0.3
0.6
V
VOHL
High-Level Output Voltage
IxLO = −50mA, VOHL = VDD - VxLO
0.5
1.0
V
IOHL
Peak Sink Current
VxLO = 0V
1
A
IOLL
Peak Source Current
VxLO = 12V
1
A
HO Gate Driver
VOLH
Low-Level Output Voltage
IxHO = 50mA
0.3
0.6
V
VOHH
High-Level Output Voltage
IxHO = −50mA, VOHH = VxHB – VxHO
0.5
1.0
V
IOHH
Peak Sink Current
VxHO = 0V
1
A
IOLH
Peak Source Current
VxHO = 12V
1
A
(8)
Switching Specifications
tLPHL
Lower Turn-Off Propagation Delay
(xLI Falling to xLO Falling)
35
75
ns
tHPHL
Upper Turn-Off Propagation Delay
(xHI Falling to xHO Falling)
35
75
ns
tLPLH
Lower Turn-On Propagation Delay
(xLI Rising to xLO Rising)
35
75
ns
tHPLH
Upper Turn-On Propagation Delay
(xHI Rising to xHO Rising)
35
75
ns
tR/tF
Output Rise/Fall Time
CL = 1000pF
20
ns
tR/tF
Output Rise/Fall Time (3V to 9V)
CL = 0.1µF
0.8
µs
tPW
Minimum Input Pulse Width that Changes the
Output
50
ns
Note:
8. xLI/xHI mode with inputs non-overlapping, assumes xHS low before xLI goes high and xLO low before xHI goes high).
September 8, 2014
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Revision 1.0
Micrel, Inc.
MIC4606
Electrical Characteristics(5)(6) (Continued)
VDD = VxHB = 12V; VEN = 5V; VSS = VHS = 0V; No load on xLO or xHO; TA = 25°C, unless noted.
Bold values indicate –40°C≤ TJ ≤ +125°C.
Symbol
Parameter
Condition
Min.
Switching Specifications
Delay from xPWM High (or xLI Low) to xLO
tLOOFF
Low
Typ.
Max.
Units
35
75
ns
VLOOFF
xLO Output Voltage Threshold for Low-Side
FET to be Considered Off
1.9
tHOON
Delay from xLO off to xHO High
35
75
ns
tHOOFF
Delay from xPWM Low (or xHI Low) to xHO
Low
35
75
ns
VSWTH
Switch Node Voltage Threshold Signaling xHO
is Off
2.2
4
V
tLOON
Delay Between xHO FET neing Considered
Off to xLO Turning On
35
75
ns
tLOONHI
For xHS Low/xLI High, Delay from xPWM/xHI
Low to xLO High
80
150
ns
tSWTO
Force xLO On if VSWTH is Not Detected
250
500
ns
1
100
V
Note:
9. PWM mode (MIC4606-2) or LI/HI mode (MIC4606-1) with overlapping xLI/xHI inputs.
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MIC4606
Timing Diagrams
Likewise, xLI going high forces xLO high after typical delay
of 35ns (tLPLH) and xLO follows low transition of xLI after
typical delay of 35ns (tLPHL).
Non-Overlapping LI/HI Input Mode (MIC4606-1)
In LI/HI input mode, external xLI/xHI inputs are delayed to
the point that xHS is low before xLI is pulled high and
similarly xLO is low before xHI goes high
xHO and xLO output rise and fall times (tR/tF) are typically
20ns driving 1000pF capacitive loads.
xHO goes high with a high signal on xHI after a typical
delay of 35ns (tHPLH). xHI going low drives xHO low also
with typical delay of 35ns (tHPHL).
All propagation delays are measured from the 50% voltage
level and rise/fall times are measured 10% to 90%.
Figure 1. Separate Non-Overlapping LI/HI Input Mode (MIC4606-1)
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MIC4606
Timing Diagrams (Continued)
Overlapping LI/HI Input Mode (MIC4606-1)
When xLI/xHI input high conditions overlap, xLO/xHO
output states are dominated by the first output to be turned
on. That is, if xLI goes high (on), while xHO is high, xHO
stays high until xHI goes low at which point, after a delay
of tHOOFF and when xHS < 2.2V, xLO goes high with a
delay of tLOON. Should xHS never trip the aforementioned
internal comparator reference (2.2V), a falling xHI edge
delayed by a typical 250ns will set “HS latch” allowing xLO
to go high.
If xHS falls very fast, xLO will be held low by a 35ns delay
gated by HI going low. Conversely, xHI going high (on)
when xLO is high has no effect on outputs until xLI is
pulled low (off) and xLO falls to < 1.9V. Delay from xLI
going low to xLO falling is tLOOFF and delay from xLO <
1.9V to xHO being on is tHOON.
Figure 2. Separate Overlapping LI/HI Input Mode (MIC4606-1)
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MIC4606
Timing Diagrams (Continued)
PWM Input Mode (MIC4606-2)
A low going xPWM signal applied to the MIC4606-2
causes xHO to go low, typically 35ns (tHOOFF) after the
xPWM input goes low, at which point the switch node,
xHS, falls (1 − 2).
high. An 80ns delay gated by xPWM going low may
determine the time to xLO going high for fast falling HS
designs. xPWM going high forces xLO low in typically 35ns
(tLOOFF) (5− 6).
When xHS reaches 2.2V (VSWTH), the external high-side
MOSFET is deemed off and xLO goes high, typically within
35ns (tLOON) (3-4). xHS falling below 2.2V sets a latch that
can only be reset by xPWM going high. This design
prevents ringing on xHS from causing an indeterminate
xLO state. Should xHS never trip the aforementioned
internal comparator reference (2.2V), a falling xPWM edge
delayed by 250ns will set “HS latch” allowing xLO to go
When xLO reaches 1.9V (VLOOFF), the low-side MOSFET is
deemed off and xHO is allowed to go high. The delay
between these two points is typically 35ns (tHOON) (7 − 8).
xHO and xLO output rise and fall times (tR/tF) are typically
20ns driving 1000pF capacitive loads.
Note: All propagation delays are measured from the 50% voltage
level and rise/fall times are measured 10% to 90%.
Figure 3. PWM Mode (MIC4606-2)
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MIC4606
Typical Characteristics
100
SHUTDOWN CURRENT (µA)
HS = 0V
250
225
200
T = 125°C
175
150
T = -40°C
HS = 0V
80
T = -40°C
60
40
T = 25°C
20
T = 25°C
6
T = 125°C
0
125
4
700
8
10
12
14
4
16
6
8
10
12
14
16
VDD OPERATING CURRENT (µA)
275
QUIESCENT CURRENT (µA)
VDD Operating Current
vs. Input Voltage
Shutdown Current
vs. Input Voltage
VDD Quiescent Current
vs. Input Voltage
Freq = 20kHz
HS = 0V
VHB = VDD
600
500
T = 125°C
400
T = 25°C
300
T = -40°C
200
100
0
4
INPUT VOLTAGE (V)
6
VHB Operating Current
vs. Input Voltage
T = 25°C
T = -40°C
20
tHPHL
50
tLPHL
tLPLH
35
tHPLH
T = 125°C
0
20
10
12
14
16
4
6
8
10
12
14
VDD = 16V
60
VDD = 12V
40
VDD = 5.5V
20
0
25
50
75
TEMPERATURE (°C)
September 8, 2014
VDD = 5.5V
125
HS = 0V
-50
-25
100
125
0
25
50
75
100
125
TEMPERATURE (°C)
VHB Operating Current
vs. Temperature
80
FREQ = 20kHz
HS = 0V
VHB = VDD
600
VDD = 16V
500
400
VDD = 12V
300
VDD = 5.5V
200
100
-50
-25
0
25
50
75
TEMPERATURE (°C)
10
100
125
VHB OPERATING CURRENT (µA)
VDD OPERATING CURRENT (µA)
HS = 0V
0
175
150
16
700
-25
VDD = 12V
200
VDD Operating Current
vs. Temperature
100
-50
225
INPUT VOLTAGE (V)
Shutdown Current
vs. Temperature
80
VDD = 16V
250
100
INPUT VOLTAGE (V)
SHUTDOWN CURRENT (µA)
QUIESCENT CURRENT (µA)
DELAY (ns)
VHB OPERATING CURRENT (µA)
65
8
16
275
TAMB = 25°C
HS = 0V
60
6
14
300
80
FREQ = 20kHz
HS = 0V
VHB = VDD
4
12
VDD Quiescent Current
vs. Temperature
Propagation Delay
vs. Input Voltage
80
40
10
8
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
FREQ = 20kHz
HS = 0V
VHB = VDD
70
60
50
VHB = 16V
40
30
VHB = 12V
20
VHB = 5.5V
10
0
-50
-25
0
25
50
75
100
125
TEMPERATURE (°C)
Revision 1.0
Micrel, Inc.
MIC4606
Typical Characteristics (Continued)
10
10
5
HS = 0V
ILO , IHO = 50mA
HS = 0V
ILO , IHO = -50mA
4
VDD = 16V
4.8
VDD = 12V
VDD Rising
VDD = 5.5V
6
UVLO (V)
VDD = 5.5V
6
HS = 0V
8
VDD = 12V
VOLL, VOLH (Ω)
VOHL, VOHH (Ω)
8
4
4.6
VHB Rising
4.4
VDD Falling
4.2
VHB Falling
VDD = 16V
2
2
0
0
-50
-25
0
25
50
75
100
125
-50
-25
0
TEMPERATURE (°C)
25
50
75
100
4
125
-50
TEMPERATURE (°C)
UVLO Hysteresis
vs. Temperature
0.5
320
DELAY (ns)
DELAY (ns)
tLPLH
40
tLPHL
tHPHL
30
0
25
50
75
100
-25
160
120
0
25
PWM to LO Low
PWM Low-to-LO High
PWM Low to HO Low
50
75
100
125
-25
0
25
50
75
100
125
TEMPERATURE (°C)
TEMPERATURE (°C)
TEMPERATURE (°C)
VDD Operating Current
vs. Frequency
VHB Operating Current
vs. Frequency
2.5
HS = 0V
VHB = VDD =12V
8
T = -40°C
6
T = 25°C
4
T = 125°C
2
0
0
200
400
600
FREQUENCY (kHz)
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800
1000
VHB OPERATING CURRENT (mA)
10
VDD OPERATING CURRENT (mA)
FORCE LO On
200
-50
-50
125
125
240
0
-25
100
40
20
-50
75
80
tHPLH
0.1
0
50
280
0.4
VDD Hysteresis
MIC4606-2
VDD = VHB = 12V
HS = 0V
360
50
0.2
25
400
MIC4606-1
VDD = VHB = 12V
HS = 0V
0.3
0
Propagation Delay (PWM)
vs. Temperature
60
HS = 0V
VHB Hysteresis
-25
TEMPERATURE (°C)
Propagation Delay
vs. Temperature
0.6
HYSTERESIS (V)
UVLO Thresholds
vs. Temperature
Low Level Output Resistance
vs. Temperature
High Level Output Resistance
vs. Temperature
HS = 0V
VHB = VDD = 12V
2
T = -40°C
1.5
1
T = 25°C
T = 125°C
0.5
0
0
200
400
600
800
1000
FREQUENCY (kHz)
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MIC4606
Typical Characteristics (Continued)
Bootstrap Diode I-V
Characteristics
Bootstrap Diode Reverse Current
100
1000
FORWARD CURRENT (mA)
T = 25°C
100
T = 125°C
T = -40°C
10
1
REVERSE CURRENT (µA)
HS = 0V
HS = 0V
10
1
T = 125°C
0.1
T = 85°C
0.01
0.001
T = 25°C
0.0001
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.8
FORWARD VOLTAGE (V)
September 8, 2014
0.9
1.0
0
10
20
30
40
50
60
70
80
90 100
REVERSE VOLTAGE (V)
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MIC4606
Functional Diagram
The latch is set by the quicker of either the falling edge of
xHS or xLI gated delay of 250ns. The latch is present to
lockout xLO bounce due to ringing on xHS. If xHS never
adequately falls due to the absence of or the presence of
a very weak external pull-down on xHS, the gated delay
of 250ns at xLI will set the latch allowing xLO to transition
high. This in turn allows the xLI startup pulse to charge
the bootstrap capacitor if the load inductor current is very
low and xHS is uncontrolled. The latch is reset by the xLI
falling edge.
For xHO to be high, the xHI must be high and the xLO
must be low. xHO going high is delayed by xLO falling
below 1.9V. The xHI and xLI inputs must not rise at the
same time to prevent a glitch from occurring on the
output. A minimum 50ns delay between both inputs is
recommended.
xLO is turned off very quickly on the xLI falling edge. xLO
going high is delayed by the longer of 35ns delay of xHO
control signal going “off” or the RS latch being set.
There is one external enable pin that controls both
phases.
Figure 4. MIC4606 xPhase Top Level Block Diagram
Figure 5. Input Logic Block in Figure 4.
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MIC4606
A high level applied to xLI pin causes VDD to be applied to
the gate of the external MOSFET. A low level on the xLI
pin grounds the gate of the external MOSFET.
Functional Description
The MIC4606 is a non-inverting, 85V full-bridge MOSFET
driver designed to independently drive all four N-Channel
MOSFETs in the bridge. The MIC4606 offers a wide 5.5V
to 16V operating supply range with either four
independent TTL inputs (MIC4606-1) or two PWM inputs,
one for each phase (MIC4606-2). Refer to Figure 4.
The drivers contain input buffers with hysteresis, three
independent UVLO circuits (two high side and one low
side), and four output drivers. The high-side output
drivers utilize a high-speed level-shifting circuit that is
referenced to its HS pin. Each phase has an internal
diode that is used by the bootstrap circuits to provide the
drive voltages for each of the two high-side outputs.
Startup and UVLO
The UVLO circuits force the driver’s outputs low until the
supply voltage exceeds the UVLO threshold. The lowside UVLO circuit monitors the voltage between the VDD
and VSS pins. The high-side UVLO circuits monitor the
voltage between the xHB and xHS pins. Hysteresis in the
UVLO circuits prevent noise and finite circuit impedance
from causing chatter during turn-on.
Figure 6. Low-Side Driver Block Diagram
High-Side Driver and Bootstrap Circuit
A block diagram of the high-side driver and bootstrap
circuit is shown in Figure 7. This driver is designed to drive
a floating N-channel MOSFET, whose source terminal is
referenced to the HS pin.
Enable Inputs
There is one external enable pin that controls both
phases. A logic high on the enable pin (EN) allows for
startup of both phases and normal operation. Conversely,
when a logic low is applied on the enable pin, both
phases turn-off and the device enters a low current
shutdown mode. All outputs (xHO and xLO) are pulled
low when EN is low. Do not leave the EN pin floating.
Input Stage
All input pins (xLI and xHI) are referenced to the VSS pin.
The MIC4606 has a TTL-compatible input range and can
be used with input signals with amplitude less than the
supply voltage. The threshold level is independent of the
VDD supply voltage and there is no dependence between
IVDD and the input signal amplitude. This feature makes
the MIC4606 an excellent level translator that will drive
high level gate threshold MOSFETs from a low-voltage
PWM IC.
Figure 7. High-Side Driver and Bootstrap Circuit
Block Diagram
Low-Side Driver
A block diagram of the low-side driver is shown in Figure
6. It drives a ground (VSS pin) referenced N-channel
MOSFET.
Low impedances in the driver allow the external MOSFET
to be turned on and off quickly. The rail-to-rail drive
capability of the output ensures high noise immunity and
a low RDS(ON) from the external MOSFET.
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A low-power, high-speed, level-shifting circuit isolates the
low side (VSS pin) referenced circuitry from the high-side
(xHS pin) referenced driver. Power to the high-side driver
and UVLO circuit is supplied by the bootstrap capacitor
(CB) while the voltage level of the xHS pin is shifted high.
Programmable Gate Drive
The MIC4606 offers programmable gate drive, which
means the MOSFET gate drive (gate to source voltage)
equals the VDD voltage. This feature offers designers
flexibility in driving the MOSFETs. Different MOSFETs
require different VGS characteristics for optimum RDSON
performance. Typically, the higher the gate voltage (up to
16V), the lower the RDSON achieved. For example, a
NTMSF4899NF MOSFET can be driven to the ON state
with a gate voltage of 5.5V but RDSON is 5.2mΩ. If driven to
10V, RDSON is 4.1mΩ – a decrease of 20%. In low-current
applications, the losses due to RDSON are minimal, but in
battery-powered high-current motor drive applications such
as power tools, the difference in RDSON can cut into the
efficiency budget, reducing run time.
The bootstrap circuit consists of an internal diode and
external capacitor, CB. In a typical application, such as the
motor driver shown in Figure 8 (only Phase A illustrated),
the AHS pin is at ground potential while the low-side
MOSFET is on. The internal diode allows capacitor CB to
charge up to VDD-VF during this time (where VF is the
forward voltage drop of the internal diode). After the lowside MOSFET is turned off and the AHO pin turns on, the
voltage across capacitor CB is applied to the gate of the
high-side external MOSFET. As the high-side MOSFET
turns on, voltage on the AHS pin rises with the source of
the high-side MOSFET until it reaches VIN. As the AHS
and AHB pins rise, the internal diode is reverse biased,
preventing capacitor CB from discharging.
Figure 9. MOSFET RDSON vs. VGS
Figure 8. MIC4606 Motor Driver Example
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the driver cannot monitor the gate voltage inside the
MOSFET. Figure 10 shows an equivalent circuit of the highside gate drive, including parasitic.
Application Information
Adaptive Dead Time
The door lock/unlock circuit diagram shown in Figure 11 is
used to illustrate the importance of the adaptive dead time
feature of the MIC4606. For each phase, it is important
that both MOSFETs are not conducting at the same time
or VIN will be shorted to ground and current will “shoot
through” the MOSFETs. Excessive shoot-through causes
higher power dissipation in the MOSFETs, voltage spikes
and ringing. The high switching current and voltage ringing
generate conducted and radiated EMI.
Minimizing shoot-through can be done passively, actively
or through a combination of both. Passive shoot-through
protection can be achieved by implementing delays
between the high and low gate drivers to prevent both
MOSFETs from being on at the same time. These delays
can be adjusted for different applications. Although simple,
the disadvantage of this approach is that it requires long
delays to account for process and temperature variations
in the MOSFET and MOSFET driver.
Figure 10. MIC4606 Driving an External MOSFET
The internal gate resistance (RG_FET) and any external
damping resistor (RG) isolate the MOSFET’s gate from the
driver output. There is a delay between when the driver
output goes low and the MOSFET turns off. This turn-off
delay is usually specified in the MOSFET datasheet. This
delay increases when an external damping resistor is
used.
Adaptive dead time monitors voltages on the gate drive
outputs and switch node to determine when to switch the
MOSFETs on and off. This active approach adjusts the
delays to account for some of the variations, but it too has
its disadvantages. High currents and fast switching
voltages in the gate drive and return paths can cause
parasitic ringing to turn the MOSFETs back on even while
the gate driver output is low. Another disadvantage is that
Figure 11. Door Lock/Unlock Circuit
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propagation delay also minimizes phase shift errors in
power supplies with wide bandwidth control loops.
Care must be taken to ensure that the input signal pulse
width is greater than the minimum specified pulse width.
An input signal that is less than the minimum pulse width
may result in no output pulse or an output pulse whose
width is significantly less than the input.
The maximum duty cycle (ratio of high side on-time to
switching period) is determined by the time required for the
CB capacitor to charge during the off-time. Adequate time
must be allowed for the CB capacitor to charge up before
the high-side driver is turned back on.
Figure 12. Adaptive Dead Time Logic Diagram
The MIC4606 uses a combination of active sensing and
passive delay to ensure that both MOSFETs are not on at
the same time. Figure 12 illustrates how the adaptive dead
time circuitry works.
Although the adaptive dead time circuit in the MIC4606
prevents the driver from turning both MOSFETs on at the
same time, other factors outside of the anti shoot-through
circuit’s control can cause shoot-through. Other factors
include ringing on the gate drive node and capacitive
coupling of the switching node voltage on the gate of the
low-side MOSFET.
For the MIC4606-2, a high level on the xPWM pin causes
/HI to go high and /LI to go low. This causes the xLO pin to
go low. The MIC4606 monitors the xLO pin voltage and
prevents the xHO pin from turning on until the voltage on
the xLO pin reaches the VLOOFF threshold. After a short
delay, the MIC4606 drives the xHO pin high. Monitoring
the xLO voltage eliminates any excessive delay due to the
MOSFET drivers turn-off time and the short delay
accounts for the MOSFET turn-off delay as well as letting
the xLO pin voltage settle out. An external resistor
between the xLO output and the MOSFET may affect the
performance of the xLO pin monitoring circuit and is not
recommended.
The scope photo in Figure 13 shows the dead time (<20ns)
between the high and low-side MOSFET transitions as the
low-side driver switches off while the high-side driver
transitions from off to on.
A low on the xPWM pin causes /HI to go low and /LI to go
high. This causes the xHO pin to go low after a short delay
(tHOOFF). Before the xLO pin can go high, the voltage on the
switching node (xHS pin) must have dropped to 2.2V.
Monitoring the switch voltage instead of the xHO pin
voltage eliminates timing variations and excessive delays
due to the high side MOSFET turn-off. The xLO driver
turns on after a short delay (tLOON). Once the xLO driver is
turned on, it is latched on until the xPWM signal goes high.
This prevents any ringing or oscillations on the switch
node or xHS pin from turning off the xLO driver. If the
xPWM pin goes low and the voltage on the xHS pin does
not cross the VSWTH threshold, the xLO pin will be forced
high after a short delay (tSWTO), insuring proper operation.
Figure 13. Adaptive Dead Time LO (low) to HO (high)
1 contains truth tables for the MIC4606-1
(Independent TTL inputs) and Table 2 is for the MIC4606-2
(PWM inputs) that details the “first on” priority as well as
the failsafe delay (tSWTO).
Table
Table 1. MIC4606-1 Truth Table
The internal logic circuits also insure a “first on” priority at
the inputs. If the xHO output is high, the xLI pin is
inhibited. A high signal or noise glitch on the xLI pin has
no effect on the xHO or xLO outputs until the xHI pin goes
low. Similarly, the xLO being high holds xHO low until xLI
and xLO are low.
Fast propagation delay between the input and output drive
waveform is desirable. It improves overcurrent protection
by decreasing the response time between the control
signal and the MOSFET gate drive.
Minimizing
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xLI
xHI
xLO
xHO
Comments
0
0
0
0
Both outputs off.
0
1
0
1
xHO will not go high until xLO falls
below 1.9V.
1
0
1
0
xLO will be delayed an extra 250ns
if xHS never falls below 2.2V.
1
1
X
X
First ON stays on until input of
same goes low.
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Table 2. MIC4606-2 Truth Table
xPWM
xLO
xHO
0
1
0
xLO will be delayed an extra 240ns if
xHS never falls below 2.2V.
1
0
1
xHO will not go high until xLO falls below
1.9V.
An optional external bootstrap diode may be used instead
of the internal diode (Figure 14). An external diode may be
useful if high gate charge MOSFETs are being driven and
the power dissipation of the internal diode is contributing to
excessive die temperatures. The voltage drop of the
external diode must be less than the internal diode for this
option to work. The reverse voltage across the diode will
be equal to the input voltage minus the VDD supply voltage.
The above equations can be used to calculate power
dissipation in the external diode; however, if the external
diode has significant reverse leakage current, the power
dissipated in that diode due to reverse leakage can be
calculated as:
Comments
Power Dissipation Considerations
Power dissipation in the driver can be separated into three
areas:
•
Internal diode dissipation in the bootstrap circuit
•
Internal driver dissipation
•
Quiescent current dissipation used to supply the
internal logic and control functions.
Pdiode REV = I R × VREV × (1 − D )
Eq. 3
where;
IR = reverse current flow at VREV and TJ
VREV = diode reverse voltage
D = duty cycle = tON × fS
Bootstrap Circuit Power Dissipation
Power dissipation of the internal bootstrap diode primarily
comes from the average charging current of the bootstrap
capacitor (CB) multiplied by the forward voltage drop of the
diode. Secondary sources of diode power dissipation are
the reverse leakage current and reverse recovery effects
of the diode.
The on-time is the time the high-side switch is conducting.
In most topologies, the diode is reverse biased during the
switching cycle off-time.
The average current drawn by repeated charging of the
high-side MOSFET is calculated by:
I F ( AVE ) = Q gate × f S
Eq. 1
where;
Qgate = total gate charge at VHB - VHS
fs = gate drive switching frequency
The average power dissipated by the forward voltage drop
of the diode equals:
Pdiode fwd = I F ( AVE ) × V F
Eq. 2
where; VF = diode forward voltage drop
There are two phases in the MIC4606. The power
dissipation for each of the bootstrap diodes must be
calculated and summed to obtain the total bootstrap diode
power dissipation for the package.
Figure 14. Optional Bootstrap Diode
The value of VF should be taken at the peak current
through the diode; however, this current is difficult to
calculate because of differences in source impedances.
The peak current can either be measured or the value of
VF at the average current can be used, which will yield a
good approximation of diode power dissipation.
The reverse leakage current of the internal bootstrap diode
is typically 3µA at a reverse voltage of 85V at 125°C.
Power dissipation due to reverse leakage is typically much
less than 1mW and can be ignored.
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Gate Driver Power Dissipation
Power dissipation in the output driver stage is mainly
caused by charging and discharging the gate to source
and gate to drain capacitance of the external MOSFET.
Figure 15 shows a simplified equivalent circuit of the
MIC4606 driving an external high-side MOSFET.
Figure 16. Typical Gate Charge vs. VGS
The same energy is dissipated by ROFF, RG, and RG_FET
when the driver IC turns the MOSFET off. Assuming RON is
approximately equal to ROFF, the total energy and power
dissipated by the resistive drive elements is:
E DRIVER = QG × VGS
Figure 15. MIC4606 Driving an External High-Side MOSFET
and
PDRIVER = QG × VGS × f S
Dissipation during the External MOSFET Turn-On
Energy from capacitor CB is used to charge up the input
capacitance of the MOSFET (CGD and CGS). The energy
delivered to the MOSFET is dissipated in the three
resistive components, RON, RG and RG_FET. RON is the on
resistance of the upper driver MOSFET in the MIC4606.
RG is the series resistor (if any) between the driver and the
MOSFET. RG_FET is the gate resistance of the MOSFET.
RG_FET is usually listed in the power MOSFET’s
specifications. The ESR of capacitor CB and the resistance
of the connecting etch can be ignored since they are much
less than RON and RG_FET.
Where:
EDRIVER = energy dissipated per switching cycle
PDRIVER = power dissipated per switching cycle
QG = total gate charge at VGS
VGS = gate to source voltage on the MOSFET
fS = switching frequency of the gate drive circuit
The power dissipated in the driver equals the ratio of RON
and ROFF to the external resistive losses in RG and RG_FET.
Letting RON = ROFF, the power dissipated in the driver due
to driving the external MOSFET is:
The effective capacitances of CGD and CGS are difficult to
calculate because they vary non-linearly with ID, VGS, and
VDS. Fortunately, most power MOSFET specifications
include a typical graph of total gate charge versus VGS.
Figure 16 shows a typical gate charge curve for an arbitrary
power MOSFET. This chart shows that for a gate voltage
of 10V, the MOSFET requires about 23.5nC of charge.
The energy dissipated by the resistive components of the
gate drive circuit during turn-on is calculated as:
E = 12 × C ISS × VGS
Pdissdriver = PDRIVER
RON
RON
+ RG + RG _ FET
Eq. 6
There are four MOSFETs driven by the MIC4606. The
power dissipation for each of the drivers must be
calculated and summed to obtain the total driver diode
power dissipation for the package.
2
but
Q = C ×V
so
Eq. 5
In some cases, the high-side FET of one phase may be
pulsed at a frequency, fS, while the low-side FET of the
other phase is kept continuously on. Since the MOSFET
gate is capacitive, there is no driver power if the FET is not
switched. The operation of each of the four drivers must
be considered to accurately calculate power dissipation.
Eq. 4
E = 12 × QG × VGS
Where
CISS = total gate capacitance of the MOSFET
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provides current to the high-side circuit while the highside external MOSFET is on. Ceramic capacitors are
recommended because of their low impedance and small
size. Z5U type ceramic capacitor dielectrics are not
recommended because of the large change in
capacitance over temperature and voltage. A minimum
value of 0.1µF is required for CB (HB to HS capacitors)
and 1µF for the VDD capacitor, regardless of the
MOSFETs being driven. Larger MOSFETs may require
larger capacitance values for proper operation. The
voltage rating of the capacitors depends on the supply
voltage, ambient temperature and the voltage derating
used for reliability. 25V rated X5R or X7R ceramic
capacitors are recommended for most applications. The
minimum capacitance value should be increased if low
voltage capacitors are used because even good quality
dielectric capacitors, such as X5R, will lose 40% to 70%
of their capacitance value at the rated voltage.
Supply Current Power Dissipation
Power is dissipated in the input and control sections of
the MIC4606, even if there is no external load. Current is
still drawn from the VDD and HB pins for the internal
circuitry, the level shifting circuitry, and shoot-through
current in the output drivers. The VDD and HB currents
are proportional to operating frequency and the VDD and
VHB voltages. The typical characteristic graphs show
how supply current varies with switching frequency and
supply voltage.
The power dissipated by the MIC4606 due to supply
current is
Pdiss supply = VDD × I DD + VHB × I HB
Eq. 7
Values for IDD and IHB are found in the EC table and the
typical characteristics graphs.
Total Power Dissipation and Thermal Considerations
Total power dissipation in the MIC4606 is equal to the
power dissipation caused by driving the external
MOSFETs, the supply currents and the internal bootstrap
diodes.
Pdisstotal = Pdiss supply + Pdissdrive + Pdiode
Placement of the decoupling capacitors is critical. The
bypass capacitor for VDD should be placed as close as
possible between the VDD and VSS pins. The bypass
capacitor (CB) for the HB supply pin must be located as
close as possible between the HB and HS pins. The etch
connections must be short, wide, and direct. The use of a
ground plane to minimize connection impedance is
recommended. Refer to the section Grounding, Component
Placement and Circuit Layout for more information.
Eq. 8
The die temperature can be calculated after the total
power dissipation is known.
TJ = T A + Pdisstotal × θ JA
The voltage on the bootstrap capacitor drops each time it
delivers charge to turn on the MOSFET. The voltage drop
depends on the gate charge required by the MOSFET.
Most MOSFET specifications specify gate charge versus
VGS voltage. Based on this information and a
recommended ΔVHB of less than 0.1V, the minimum
value of bootstrap capacitance is calculated as:
Eq. 9
Where:
TA = maximum ambient temperature
TJ = junction temperature (°C)
Pdisstotal = total power dissipation of the MIC4606
θJA = thermal resistance from junction to ambient air
CB ≥
Other Timing Considerations
Make sure the input signal pulse width is greater than the
minimum specified pulse width. An input signal that is
less than the minimum pulse width may result in no
output pulse or an output pulse whose width is
significantly less than the input.
Eq. 10
Where:
QGATE = total gate charge at VHB
∆VHB = voltage drop at the HB pin
If the high-side MOSFET is not switched but held in an on
state, the voltage in the bootstrap capacitor will drop due
to leakage current that flows from the HB pin to ground.
This current is specified in the EC table. In this case, the
value of CB is calculated as:
The maximum duty cycle (ratio of high side on-time to
switching period) is controlled by the minimum pulse
width of the low side and by the time required for the CB
capacitor to charge during the off-time. Adequate time
must be allowed for the CB capacitor to charge up before
the high-side driver is turned on.
CB ≥
Decoupling and Bootstrap Capacitor Selection
Decoupling capacitors are required for both the low side
(VDD) and high side (HB) supply pins. These capacitors
supply the charge necessary to drive the external
MOSFETs and also minimize the voltage ripple on these
pins. The capacitor from HB to HS has two functions: it
provides decoupling for the high-side circuitry and also
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QGATE
∆VHB
I HBS × tON
∆VHB
Eq. 11
Where:
IHBS = maximum HB pin leakage current
tON = maximum high-side FET on-time
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The larger value of CB from Equation 10 or 11 should be
used.
DC Motor Applications
MIC4606 MOSFET drivers are widely used in DC motor
applications. They address both stepper and brushed
motors in full-bridge topologies. As shown in Figure 17
and Figure 18, the driver switches the MOSFETs at
variable duty cycles that modulate the voltage to control
motor speed. The full-bridge topology allows for bidirectional control.
The MIC4606’s 85V operating voltage offers ample
operating voltage margin to protect against voltage
spikes in the motor drive circuitry. It is good practice to
have at least twice the HV voltage of the motor supply.
The MIC4606’s 85V operating voltage allows sufficient
margin for 12V, 24V, and 40V motors.
Figure 18. Full-Bridge DC Motor
The MIC4606 is offered in a small 4mm × 4mm QFN 16
lead package for applications that are space constrained.
The motor trend is to put the motor control circuit inside
the motor casing, which requires small packaging
because of the size of the motor.
The MIC4606 offers low UVLO threshold and
programmable gate drive, which allows for longer
operation time in battery operated motors such as power
hand tools.
Power Inverter
Power inverters are used to supply AC loads from a DC
operated battery system, mainly during power failure. The
battery voltage can be 12VDC, 24VDC, or up to 36VDC,
depending on the power requirements. There two popular
conversion methods, Type I and Type II, that convert the
battery energy to AC line voltage (110VAC or 230VAC).
Figure 19. Type I Inverter Topology
Figure 17. Stepper Motor Driver
As shown in Figure 19, Type I is a dual-stage topology
where line voltage is converted to DC through a
transformer to charge the storage batteries. When a
power failure is detected, the stored DC energy is
converted to AC through another transformer to drive the
AC loads connected to the inverter output. This method is
simplest to design but tends to be bulky and expensive
because it uses two transformers.
Type II is a single-stage topology that uses only one
transformer to charge the bank of batteries to store the
energy. During a power outage, the same transformer is
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used to power the line voltage. The Type II topology
switches at a higher frequency compared to the Type I
topology to maintain a small transformer size.
It is important to note that capacitor CB must be placed
close to the xHB and xHS pins. This capacitor not only
provides all the energy for turn-on but it must also keep
xHB pin noise and ripple low for proper operation of the
high-side drive circuitry.
Both types use a full bridge topology to invert DC to AC.
The MIC4606’s operating voltage offers enough of a
margin to address all of the available banks of batteries
commonly used in inverter applications. The 85V
operating voltage allows designers to increase the bank
of batteries up to 72V, if desired. The MIC4606 can sink
as much as 1A, which is sufficient to drive the MOSFET’s
gate capacitance while switching the MOSFET up to
50kHz. This makes the MIC4606 an ideal solution for
single phase inverter applications.
Grounding, Component Placement and Circuit Layout
Nanosecond switching speeds and ampere peak currents
in and around the MIC4606 driver require proper
placement and trace routing of all components. Improper
placement may cause degraded noise immunity, false
switching, excessive ringing, or circuit latch-up.
Figure 20 shows the critical current paths of the high and
low-side driver when their outputs go high and turn on the
external MOSFETs. It also helps demonstrate the need
for a low impedance ground plane. Charge needed to
turn-on the MOSFET gates comes from the decoupling
capacitors CVDD and CB. Current in the low-side gate
driver flows from CVDD through the internal driver, into the
MOSFET gate, and out the source. The return connection
back to the decoupling capacitor is made through the
ground plane. Any inductance or resistance in the ground
return path causes a voltage spike or ringing to appear
on the source of the MOSFET. This voltage works
against the gate drive voltage and can either slow down
or turn off the MOSFET during the period when it should
be turned on.
Figure 20. Turn-On Current Paths
Figure 21 shows the critical current paths when the driver
outputs go low and turn off the external MOSFETs. Short,
low-impedance connections are important during turn-off
for the same reasons given in the turn-on explanation.
Current flowing through the internal diode replenishes
charge in the bootstrap capacitor, CB.
Current in the high-side driver is sourced from capacitor
CB and flows into the xHB pin and out the xHO pin, into
the gate of the high side MOSFET. The return path for
the current is from the source of the MOSFET and back
to capacitor CB. The high-side circuit return path usually
does not have a low-impedance ground plane so the etch
connections in this critical path should be short and wide
to minimize parasitic inductance. As with the low-side
circuit, impedance between the MOSFET source and the
decoupling capacitor causes negative voltage feedback
that fights the turn-on of the MOSFET.
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Figure 21. Turn-Off Current Paths
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Typical Application Schematic
Bill of Materials
Item
Part Number
Manufacturer
(10)
Description
Qty.
C1
06033D105MAT2A
AVX
1µF Ceramic Capacitor, 25V, X5R, Size 0603
1
C2
C1608X5R1C106M08
0AB
TDK(11)
10µF Ceramic Capacitor, 16V, X5R, Size 0603
1
C3, C12
0805YD105MAT2A
AVX
1µF Ceramic Capacitor, 16V, X5R, Size 0805
2
C4
C3225X7S2A475M200
AB
TDK
4.7µF Ceramic Capacitor, 100V, X7S, Size 1210
1
100µF Aluminum Electrolytic Capacitor, 100V
1
100V, N-Channel MOSFET
4
100kΩ, Tolerance 1%, Size 0603
1
85V Full-Bridge MOSFET Drivers with Adaptive Dead
Time and Shoot-Through Protection
1
C8
B41827A9107M
Q1, Q2, Q3, Q4
AM7414
R1
U1
CRCW06030000FRT1
MIC4606-1YML
(12)
EPCOS
Analog
Power(13)
(14)
Vishay
Micrel, Inc.
(15)
Notes:
10. AVX: www.avx.com.
11. TDK: www.tdk.com.
12. EPCOS: www.epcos.com.
13. Analog Power: www.analogpowerinc.com.
14. Vishay: www.vishay.com.
15. Micrel, Inc.: www.micrel.com.
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PCB Layout Recommendations
Top Layer
Bottom Layer
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Package Information(7)
16-Pin QFN 4mm x 4mm (ML)
Note:
16. Package information is correct as of the publication date. For updates and most current information, go to www.micrel.com.
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MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel, Inc. is a leading global manufacturer of IC solutions for the worldwide high performance linear and power, LAN, and timing & communications
markets. The Company’s products include advanced mixed-signal, analog & power semiconductors; high-performance communication, clock
management, MEMs-based clock oscillators & crystal-less clock generators, Ethernet switches, and physical layer transceiver ICs. Company
customers include leading manufacturers of enterprise, consumer, industrial, mobile, telecommunications, automotive, and computer products.
Corporation headquarters and state-of-the-art wafer fabrication facilities are located in San Jose, CA, with regional sales and support offices and
advanced technology design centers situated throughout the Americas, Europe, and Asia. Additionally, the Company maintains an extensive network
of distributors and reps worldwide.
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this datasheet. This
information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry,
specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual
property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability
whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties
relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright, or other intellectual property right.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical
implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
© 2014 Micrel, Incorporated.
September 8, 2014
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