1MHz PWM Synchronous Buck-Control IC
General Description
The MIC2168A is a high-efficiency, simple to use 1MHz
PWM synchronous buck-control IC housed in a small
MSOP-10 package. The MIC2168A allows compact
DC/DC solutions with a minimal external component count
and cost.
The MIC2168A operates from a 3V to 14.5V input, without
the need of any additional bias voltage. The output voltage
can be precisely regulated down to 0.8V. The adaptive all
N-Channel MOSFET drive scheme allows efficiencies over
95% across a wide load range.
The MIC2168A senses current across the high-side NChannel MOSFET, eliminating the need for an expensive
and lossy current-sense resistor. Current limit accuracy is
maintained by a positive temperature coefficient that tracks
the increasing RDS(ON) of the external MOSFET. Further
cost and space are saved by the internal in-rush-current
limiting and digital soft-start.
The MIC2168A is identical to the MIC2168 with the
exception that the MIC2168A increases the overcurrent
blanking time from 80ns (typ.) to 120ns (typ.)
The MIC2168A is available in a 10-pin MSOP package,
with a wide junction operating range of –40°C to +125°C.
Datasheets and support documentation are available on
Micrel’s web site at: www.micrel.com.
3V to 14.5V input voltage range
Adjustable output voltage down to 0.8V
Up to 95% efficiency
1MHz PWM operation
Adjustable current-limit senses high-side N-Channel
MOSFET current
No external current sense resistor
Adaptive gate drive increases efficiency
Ultra-fast response with hysteretic transient recovery
Overvoltage protection protects the load in fault
Dual mode current limit speeds up recovery time
Hiccup mode short-circuit protection
Internal soft-start
Small-size MSOP 10-pin package
Point-of-load DC/DC conversion
Set-top boxes
Graphic cards
LCD power supplies
Telecom power supplies
Networking power supplies
Cable modems and routers
Typical Application
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
April 22, 2015
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Ordering Information
Part Number
Junction Temperature Range
–40° to +125°C
10-Pin MSOP Pb-Free
Pin Configuration
10-Pin MSOP (MM)
(Top View)
Pin Description
Pin Number
Pin Name
Supply Voltage (Input): 3V to 14.5V.
5V Internal Linear Regulator (Output): VDD is the external MOSFET gate drive supply voltage
and an internal supply bus for the IC. When VIN is <5V, this regulator operates in dropout mode.
Current Sense / Enable (Input): Current-limit comparator noninverting input. The current limit is
sensed across the MOSFET during the ON time. The current can be set by the resistor in series
with the CS pin.
Ground (Return).
Low-Side Drive (Output): High-current driver output for external synchronous MOSFET.
Switch (Return): High-side MOSFET driver return.
High-Side Drive (Output): High-current output-driver for the high-side MOSFET. When VIN is
between 3.0V to 5V, 2.5V threshold-rated MOSFETs should be used. At VIN > 5V, 5V threshold
MOSFETs should be used.
Boost (Input): Provides the drive voltage for the high-side MOSFET driver. The gate-drive voltage
is higher than the source voltage by VIN minus a diode drop.
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Pin Function
Compensation (Input): Dual function pin. Pin for external compensation. If this pin is pulled below
0.2V, with the reference fully up, the device shuts down (50μA typical current draw).
Feedback (Input): Input to error amplifier. Regulates error amplifier to 0.8V.
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Absolute Maximum Ratings(1)
Operating Ratings(2)
Supply Voltage (VIN) .................................................. +15.5V
Bootstrapped Voltage (VBST). ................................... VIN +5V
Junction Temperature (TJ) ..................–40°C ≤ TJ ≤ +125°C
Storage Temperature (TS) ......................... –65°C to +150°C
Supply Voltage (VIN) ...................................... +3V to +14.5V
Output Voltage Range............................. 0.8V to VIN × DMAX
Package Thermal Resistance
10-Pin MSOP (θJA) ........................................... 180°C/W
Electrical Characteristics(3)
TJ = 25°C, VIN = 5V, unless otherwise specified. Bold values indicate –40°C < TJ < +125°C.
Feedback Voltage Reference
(± 1%)
Feedback Voltage Reference
(± 2% over temp)
Feedback Bias Current
Output Voltage Line Regulation
Output Voltage Load Regulation
Output Voltage Total Regulation
3V ≤ VIN ≤ 14.5V; 1A ≤ IOUT ≤ 10A; (VOUT = 2.5V)
Oscillator Section
Oscillator Frequency
Maximum Duty Cycle (DMAX)
Minimum On-Time(4)
Input and VDD Supply
PWM Mode Supply Current
VCS = VIN –0.25V; VFB = 0.7V
(output switching but excluding external MOSFET
gate current.)
Shutdown Quiescent Current
VCOMP Shutdown Threshold
VCOMP Shutdown Blanking
CCOMP = 100nF
Digital Supply Voltage (VDD)
VIN ≥ 6V
Error Amplifier
DC Gain
Soft Start
Soft-Start Current
After timeout of internal timer.
(See “Soft-Start” section.)
Current Sense
CS Overcurrent Trip Point
VCS = VIN –0.25V
Temperature Coefficient
1. Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating
the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(max), the
junction-to-ambient thermal resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive die
temperature, and the regulator will go into thermal shutdown.
2. Devices are ESD sensitive, handling precautions required.
3. Specification for packaged product only.
4. Guaranteed by design.
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Electrical Characteristics (Continued)
TJ = 25°C, VIN = 5V, unless otherwise specified. Bold values indicate –40°C < TJ < +125°C.
Output Fault Correction Thresholds
Upper Threshold, VFB_OVT
(relative to VFB)
Lower Threshold, VFB_UVT
(relative to VFB)
Rise/Fall Time
Into 3000pF at VIN > 5V
Output Driver Impedance
Source, VIN = 5V
Sink, VIN = 5V
Source, VIN = 3V
Sink, VIN = 3V
Gate Drivers
Driver Non-Overlap Time
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Note 4
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Typical Characteristics
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Typical Characteristics (Continued)
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Functional Diagram
MIC2168A Block Diagram
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Functional Description
The MIC2168A is a voltage mode, synchronous stepdown switching regulator controller designed for high
output power without the use of an external sense
resistor. It includes an internal soft-start function which
reduces the power supply input surge current at start-up
by controlling the output voltage rise time, a PWM
generator, a reference voltage, two MOSFET drivers,
and short-circuit current limiting circuitry to form a
complete 1MHz switching regulator.
Theory of Operation
The MIC2168A is a voltage mode step-down regulator.
The block diagram illustrates the voltage control loop.
The output voltage variation due to load or line changes
will be sensed by the inverting input of the
transconductance error amplifier via the feedback
resistors R3, and R2 and compared to a reference
voltage at the non-inverting input. This will cause a small
change in the DC voltage level at the output of the error
amplifier which is the input to the PWM comparator. The
other input to the comparator is a 0 to 1V triangular
waveform. The comparator generates a rectangular
waveform whose width tON is equal to the time from the
start of the clock cycle t0 until t1, the time the triangle
crosses the output waveform of the error amplifier. To
illustrate the control loop, let us assume the output
voltage drops due to sudden load turn-on, this would
cause the inverting input of the error amplifier which is
divided down version of VOUT to be slightly less than the
reference voltage causing the output voltage of the error
amplifier to go high. This will cause the PWM
comparator to increase tON time of the top side
MOSFET, causing the output voltage to go up and
bringing VOUT back in regulation.
The COMP pin on the MIC2168A is used for the
following three functions:
1. Disables the part by grounding this pin
2. External compensation to stabilize the voltage
control loop
3. Soft-start
internal 11-bit counter starts counting which takes
approximately 2ms to complete. During counting, the
COMP voltage is clamped at 0.65V. After this counting
cycle the COMP current source is reduced to 8.5µA and
the COMP pin voltage rises from 0.65V to 0.95V, the
bottom edge of the saw-tooth oscillator. This is the
beginning of 0% duty cycle and it increases slowly
causing the output voltage to rise slowly. The MIC2168A
has two hysteretic comparators that are enabled when
VOUT is within ±3% of steady state. When the output
voltage reaches 97% of programmed output voltage,
then the gm error amplifier is enabled along with the
hysteretic comparator. This point onwards, the voltage
control loop (gm error amplifier) is fully in control and will
regulate the output voltage.
Soft-start time can be calculated approximately by
adding the following four time frames:
t1 = Cap_COMP × 0.18V/8.5μA
t2 = 12 bit counter, approx 2ms
t3 = Cap_COMP × 0.3V/8.5μA
t4 OUT
 × 0.5 ×
Soft-Start Time(Cap_COMP = 100nF) = t1 + t2 + t3 + t4
= 2.1ms + 2ms + 3.5ms + 1.8ms = 10ms
Current Limit
The MIC2168A uses the RDS(ON) of the top power
MOSFET to measure output current. Since it uses the
drain to source resistance of the power MOSFET, it is
not very accurate. This scheme is adequate to protect
the power supply and external components during a
fault condition by cutting back the time the top MOSFET
is on if the feedback voltage is greater than 0.67V. In
case of a hard short when feedback voltage is less than
0.67V, the MIC2168A discharges the COMP capacitor
to 0.65V, resets the digital counter and automatically
shuts off the top gate drive, and the gm error amplifier
and the –3% hysteretic comparators are completely
disabled and the soft-start cycles restarts. This mode of
operation is called the “hiccup mode” and its purpose is
to protect the downstream load in case of a hard short.
The circuit in Figure 1 illustrates the MIC2168A current
limiting circuit.
For better understanding of the soft-start feature, let’s
assume VIN = 12V, and the MIC2168A is allowed to
power-up. The COMP pin has an internal 8.5μA current
source that charges the external compensation
capacitor. As soon as this voltage rises to 180mV (t =
Cap_COMP × 0.18V/8.5μA), the MIC2168A allows the
internal VDD linear regulator to power up and as soon as
it crosses the undervoltage lockout of 2.6V, the chip’s
internal oscillator starts switching. At this point in time,
the COMP pin current source increases to 40μA and an
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Internal VDD Supply
The MIC2168A controller internally generates VDD for
self biasing and to provide power to the gate drives. This
VDD supply is generated through a low-dropout regulator
and generates 5V from VIN supply greater than 5V. For
supply voltage less than 5V, the VDD linear regulator is
approximately 200mV in dropout. Therefore, it is
recommended to short the VDD supply to the input
supply through a 10Ω resistor for input supplies between
2.9V to 5V.
Figure 1. MIC2168A Current Limiting Circuit
The current limiting resistor RCS is calculated by the
following equation:
R CS =
IPP = Inductor Ripple Current =
200μA is the internal sink current to program the
MIC2168A current limit.
MOSFET Gate Drive
The MIC2168A high-side drive circuit is designed to
switch an N-Channel MOSFET. The Functional Diagram
shows a bootstrap circuit, consisting of D1 and CBST,
supplies energy to the high-side drive circuit. Capacitor
CBST is charged while the low-side MOSFET is on and
the voltage on the VSW pin is approximately 0V. When
the high-side MOSFET driver is turned on, energy from
CBST is used to turn the MOSFET on. As the MOSFET
turns on, the voltage on the VSW pin increases to
approximately VIN. Diode D1 is reversed biased and
CBST floats high while continuing to keep the high-side
MOSFET on. When the low-side switch is turned back
on, CBST is recharged through D1. The drive voltage is
derived from the internal 5V VDD bias supply. The
nominal low-side gate drive voltage is 5V and the
nominal high-side gate drive voltage is approximately
4.5V due the voltage drop across D1. An approximate
20ns delay between the high- and low-side driver
transitions is used to prevent current from
simultaneously flowing unimpeded through both
The MOSFET RDS(ON) varies 30% to 40% with
temperature; therefore, it is recommended to add a 50%
margin to the load current (ILOAD) in the above equation
to avoid false current limiting due to increased MOSFET
junction temperature rise. It is also recommended to
connect RCS resistor directly to the drain of the top
MOSFET Q1, and the RSW resistor to the source of Q1
to accurately sense the MOSFETs RDS(ON). To make the
MIC2168A insensitive to board layout and noise, a 1.4Ω
resistor and a 1000pF capacitor is recommended below
the switch node and ground. A 0.1μF capacitor in
parallel with RCS should be connected to filter some of
the switching noise.
MOSFET Selection
The MIC2168A controller works from input voltages of
3V to 13.2V and has an internal 5V regulator to provide
power to turn the external N-Channel power MOSFETs
for high- and low-side switches. For applications where
VIN < 5V, the internal VDD regulator operates in dropout
mode, and it is necessary that the power MOSFETs
used are low threshold and are in full conduction mode
for VGS of 2.5V. For applications when VIN > 5V; logiclevel MOSFETs, whose operation is specified at VGS =
4.5V must be used.
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It is important to note the on-resistance of a MOSFET
increases with increasing temperature. A 75°C rise in
junction temperature will increase the channel
resistance of the MOSFET by 50% to 75% of the
resistance specified at 25°C. This change in resistance
must be accounted for when calculating MOSFET power
dissipation and in calculating the value of current-sense
(CS) resistor.
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Total gate charge is the charge required to turn the
MOSFET on and off under specified operating
conditions (VDS and VGS). The gate charge is supplied
by the MIC2168A gate-drive circuit. At 1MHz switching
frequency and above, the gate charge can be a
significant source of power dissipation in the MIC2168A.
At low output load, this power dissipation is noticeable
as a reduction in efficiency.
The average current required to drive the high-side
IG[high − side](avg) = QG × fS
The voltage ratings for the top and bottom MOSFET are
essentially equal to the input voltage. A safety factor of
20% should be added to the VDS(max) of the MOSFETs
to account for voltage spikes due to circuit parasitics.
The power dissipated in the switching transistor is the
sum of the conduction losses during the on-time
(PCONDUCTION) and the switching losses that occur during
the period of time when the MOSFETs turn on and off
IG[high-side](avg) = average high-side MOSFET gate current.
PAC = PAC(off) + PAC(on)
R SW = on − resistance of the MOSFET switch
QG = total gate charge for the high-side MOSFET taken
from manufacturer’s data sheet for VGS = 5V.
D = duty cycle = 
The low-side MOSFET is turned on and off at VDS = 0
because the freewheeling diode is conducting during
this time. The switching loss for the low-side MOSFET is
usually negligible. Also, the gate-drive current for the
low-side MOSFET is more accurately calculated using
CISS at VDS = 0 instead of gate charge.
Making the assumption the turn-on and turn-off
transition times are equal; the transition times can be
approximated by:
For the low-side MOSFET:
tT =
IG[low − side](avg) = CISS × VGS × fS
Since the current from the gate drive comes from the
input voltage, the power dissipated in the MIC2168A
due to gate drive is:
PGATEDRIVE = VIN IG[high − side](avg) + IG[low − side](avg)
CISS and COSS are measured at VDS = 0
IG = gate-drive current (1A for the MIC2168A)
A convenient figure of merit for switching MOSFETs is
the on resistance times the total gate charge RDS(ON) ×
QG. Lower numbers translate into higher efficiency. Low
gate-charge logic-level MOSFETs are a good choice for
use with the MIC2168A.
The total high-side MOSFET switching loss is:
PAC = (VIN + VD) × IPK × tT × fS
Parameters that are important to MOSFET switch
selection are:
Voltage rating
tT = switching transition time (typically 20ns to 50ns)
VD = freewheeling diode drop, typically 0.5V
Total gate charge
fS it the switching frequency, nominally 1MHz
The low-side MOSFET switching losses are negligible
and can be ignored for these calculations.
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Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance.
The high frequency operation of the MIC2168A requires
the use of ferrite materials for all but the most cost
sensitive applications. Lower cost iron powder cores
may be used but the increase in core loss will reduce
the efficiency of the power supply. This is especially
noticeable at low output power. The winding resistance
decreases efficiency at the higher output current levels.
The winding resistance must be minimized although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of
the core and copper losses. At higher output loads, the
core losses are usually insignificant and can be ignored.
At lower output currents, the core losses can be a
significant contributor. Core loss information is usually
available from the magnetics vendor. Copper loss in the
inductor is calculated by the equation below:
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by:
VOUT × (VIN (max) − VOUT )
VIN (max) × fS × 0.2 × IOUT (max)
The resistance of the copper wire, RWINDING, increases
with temperature. The value of the winding resistance
used should be at the operating temperature.
fS = switching frequency, 1MHz
RWINDING( hot ) = RWINDING( 20°C ) × (1 + 0.0042 × (THOT − T20°C ))
0.2 = ratio of AC ripple current to DC output current
VIN(max) = maximum input voltage
The peak-to-peak inductor current (AC ripple current) is:
THOT = temperature of the wire under operating
T20°C = ambient temperature
VOUT × (VIN (max) − VOUT )
VIN (max) × fS × L
RWINDING(20°C) is room temperature winding resistance
(usually specified by the manufacturer)
Output Capacitor Selection
The output capacitor values are usually determined
capacitors ESR (equivalent series resistance). Voltage
and RMS current capability are two other important
factors selecting the output capacitor. Recommended
electrolytics, and POSCAPS. The output capacitor’s
ESR is usually the main cause of output ripple. The
output capacitor ESR also affects the overall voltage
feedback loop from stability point of view. See
“Feedback Loop Compensation” section for more
The peak inductor current is equal to the average output
current plus one half of the peak-to-peak inductor ripple
IPK = IOUT(max) + 0.5 × IPP
The RMS inductor current is used to calculate the I2 × R
losses in the inductor.
IINDUCTOR (rms) = IOUT (max) × 1 + 
3  IOUT (max) 
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The maximum value of ESR is calculated:
The peak input current is equal to the peak inductor
current, so:
VOUT = peak-to-peak output voltage ripple
IPP = peak-to-peak inductor ripple current
The total output ripple is a combination of the ESR
output capacitance. The total ripple is calculated below:
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor ripple current is low:
ICIN (rms) ≈ IOUT (max) × D × (1 − D)
 I × (1 − D) 
 + (IPP × RESR )2
=  PP
The power dissipated in the input capacitor is:
PDISS(CIN ) = ICIN (rms)2 × RESR(CIN )
Voltage Setting Components
The MIC2168A requires two resistors to set the output
voltage as shown in Figure 2.
D = duty cycle
COUT = output capacitance value
fS = switching frequency
The voltage rating of capacitor should be twice the
voltage for a tantalum and 20% greater for an aluminum
electrolytic. The output capacitor RMS current is
calculated below:
IC OUT(rms) =
Figure 2. Voltage-Divider Configuration
The power dissipated in the output capacitor is:
VREF for the MIC2168A is typically 0.8V
PDISS (COUT ) = ICOUT ( rms ) 2 × RESR (COUT )
The output voltage is determined by the equation:
Input Capacitor Selection
The input capacitor should be selected for ripple current
rating and voltage rating. Tantalum input capacitors may
fail when subjected to high inrush currents, caused by
turning the input supply on. Tantalum input capacitor
voltage rating should be at least 2 times the maximum
input voltage to maximize reliability. Aluminum
electrolytic, OS-CON, and multilayer polymer film
capacitors can handle the higher inrush currents without
voltage derating. The input voltage ripple will primarily
depend on the input capacitor’s ESR.
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R1 
VO = VREF × 1 +
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A typical value of R1 can be between 3kΩ and 10kΩ. If
R1 is too large, it may allow noise to be introduced into
the voltage feedback loop. If R1 is too small, in value, it
will decrease the efficiency of the power supply,
especially at light loads. Once R1 is selected, R2 can be
calculated using:
R2 =
External Schottky Diode
An external freewheeling diode is used to keep the
inductor current flow continuous while both MOSFETs
are turned off. This dead time prevents current from
flowing unimpeded through both MOSFETs and is
typically 15ns. The diode conducts twice during each
switching cycle. Although the average current through
this diode is small, the diode must be able to handle the
peak current.
diode causes less ringing and less power loss.
Depending on the circuit components and operating
conditions, an external Schottky diode will give a /2% to
1% improvement in efficiency.
Feedback Loop Compensation
The MIC2168A controller comes with an internal
transconductance error amplifier used for compensating
the voltage feedback loop by placing a capacitor (C1) in
series with a resistor (R1) and another capacitor C2 in
parallel from the COMP pin to ground. See “Functional
Power Stage
The power stage of a voltage mode controller has an
inductor, L1, with its winding resistance (DCR)
connected to the output capacitor, COUT, with its
electrical series resistance (ESR) as shown in Figure 3.
The reverse voltage requirement of the diode is:
The power dissipated by the Schottky diode is:
Figure 3. The Output LC Filter in a Voltage-Mode
Buck Converter
PDIODE = ID(avg) × VF
The transfer function G(s), for such a system is:
(1 + ESR × s × C
G(S) − 
 DCR × s × C + s × L × C + 1 + ESR × s × C 
VF = forward voltage at the peak diode current
The external Schottky diode, D1, is not necessary for
circuit operation since the low-side MOSFET contains a
parasitic body diode. The external diode will improve
efficiency and decrease high frequency noise. If the
MOSFET body diode is used, it must be rated to handle
the peak and average current. The body diode has a
relatively slow reverse recovery time and a relatively
high forward voltage drop. The power lost in the diode is
proportional to the forward voltage drop of the diode. As
the high-side MOSFET starts to turn on, the body diode
becomes a short circuit for the reverse recovery period,
dissipating additional power. The diode recovery and the
circuit inductance will cause ringing during the high-side
MOSFET turn-on. An external Schottky diode conducts
at a lower forward voltage preventing the body diode in
the MOSFET from turning on. The lower forward voltage
drop dissipates less power than the body diode. The
lack of a reverse recovery mechanism in a Schottky
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Plotting this transfer function with the following assumed
values (L = 2μH, DCR = 0.009Ω, COUT = 1000μF, ESR =
0.025Ω) gives lot of insight as to why one needs to
compensate the loop by adding resistor and capacitors
on the COMP pin. Figures 4 and 5 show the gain curve
and phase curve for the above transfer function.
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From the point of view of compensating the voltage
loop, it is recommended to use higher ESR output
capacitors since they provide a 90° phase gain in the
power path. For comparison purposes, Figure 6, shows
the same phase curve with an ESR value of 0.002Ω.
Figure 4. The Gain Curve for G(s)
Figure 6. The Phase Curve with ESR = 0.002Ω
Figure 5. Phase Curve for G(s)
It can be seen from the transfer function G(s) and the
gain curve that the output inductor and capacitor create
a two pole system with a break frequency at:
fC =
2 × π L × COUT
Therefore, fLC = 3.6kHz.
By looking at the phase curve, it can be seen that the
output capacitor ESR (0.050Ω) cancels one of the two
poles (LCOUT) system by introducing a zero at:
It can be seen from Figure 5 that at 50kHz, the phase is
approximately –90° versus Figure 6 where the number
is –150°. This means that the transconductance error
amplifier has to provide a phase boost of about 45° to
achieve a closed loop phase margin of 45° at a
crossover frequency of 50kHz for Figure 4, versus 105°
for Figure 6. The simple RC and C2 compensation
scheme allows a maximum error amplifier phase boost
of about 90°. Therefore, it is easier to stabilize the
MIC2168A voltage control loop by using high ESR value
output capacitors.
gm Error Amplifier
It is undesirable to have high error amplifier gain at high
frequencies because high-frequency noise spikes would
be picked up and transmitted at large amplitude to the
output, thus, gain should be permitted to fall off at high
frequencies. At low frequency, it is desired to have high
open-loop gain to attenuate the power line ripple. Thus,
the error amplifier gain should be allowed to increase
rapidly at low frequencies.
2 × π × ESR × COUT
Therefore, FZERO = 6.36kHz.
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The transfer function with R1, C1, and C2 for the
internal gm error amplifier can be approximated by the
following equation:
1 + R1 × S × C1
Error Amplifier (s) = g m × 
 s × (C1 + C2) 1 + R1 × C1 × C2 × S  
C1 + C2  
The above equation can be simplified by assuming
Figure 7. Error Amplifier Gain Curve
1 + R1 × S × C1
 s × (C1)(1 + R1 × C2 × S) 
Error Amplifier (s) = g m × 
From the above transfer function, one can see that R1
and C1 introduce a zero and R1 and C2 a pole at the
following frequencies:
2 × π × R1 × C1
2 × π × C2 × R1
FPOLE @origin =
Figure 8. Error Amplifier Phase Curve
2 × π × C1
Figures 7 and 8 show the gain and phase curves for the
above transfer function with R1 = 9.3k, C1 = 1000pF, C2
= 100pF, and gm = .005Ω–1. It can be seen that at
50kHz, the error amplifier exhibits approximately 45° of
phase margin.
April 22, 2015
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Micrel, Inc.
Total Open-Loop Response
The open-loop response for the MIC2168A controller is
easily obtained by adding the power path and the error
amplifier gains together, since they already are in Log
scale. It is desirable to have the gain curve intersect
zero dB at tens of kilohertz, this is commonly called
crossover frequency; the phase margin at crossover
frequency should be at least 45°. Phase margins of 30°
or less cause the power supply to have substantial
ringing when subjected to transients, and have little
tolerance for component or environmental variations.
Figure 9 and Figure 10 show the open-loop gain and
phase margin. It can be seen from Figure 9 that the gain
curve intersects the 0dB at approximately 50kHz, and
from Figure 10 that at 50kHz, the phase shows
approximately 50° of margin.
Design Example
Layout and Checklist:
1. Connect the current limiting (CS) resistor directly to
the drain of top MOSFET Q1.
2. Use a 10Ω resistor from the input supply to the VIN
pin on the MIC2168A. Also, place a 1µF ceramic
capacitor from this pin to GND not through via.
3. The feedback resistors R1 and R2 should be placed
close to the FB pin. The top side of R1 should
connect directly to the output node. Run this trace
away from the switch node (junction of Q1, Q2, and
L1). The bottom side of R1 should connect to the
GND pin on the MIC2168A.
4. The compensation resistor and capacitors should be
placed right next to the COMP pin and the other
side should connect directly to the GND pin on the
MIC2168A rather than going to the plane.
5. Add a snubber circuit (resistor and a capacitor) from
the switch node to GND. A good starting point is
1000pF and 1.4Ω.
6. Add a place holder for a gate resistor on the top
MOSFET gate drive. A gate resistors of 10Ω or less
should be used. No gate resistor should be used on
the low side MOSFET.
7. Low gate charge MOSFETs should be used to
maximize efficiency, such as Si4800, Si4804BDY,
IRF7821, IRF8910, FDS6680A, and FDS6912A to
mention a few.
Figure 9. Open-Loop Gain Margin
8. Add a 1Ω to 4Ω resistor from the SW pin on the
MIC2168A to the switch node on the circuit (junction
of MOSFETs and inductor).
9. Compensation component GND, feedback resistor
ground, chip ground, 1µF VIN ceramic capacitor
ground, and 10µF VDD capacitor ground should all
run together and connect to the output capacitor
ground. See demo board layout, top layer.
10. The 10µF ceramic input capacitor should be placed
between the drain of top MOSFET and source of
bottom MOSFET.
11. The 10µF ceramic capacitor should be placed right
on the VDD pin without any vias.
12. 12. The source of the bottom MOSFET should
connect directly to the input capacitor GND with a
thick trace. The output capacitor and the input
capacitor should connect directly to the GND plane.
Figure 10. Open-Loop Phase Margin
April 22, 2015
13. 13. Place a 0.1µF ceramic capacitor in parallel with
the CS resistor to filter any switching noise.
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Package Information
10-Pin MSOP (MM)
April 22, 2015
Revision 3.0
Micrel, Inc.
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April 22, 2015
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