LED5000 - STMicroelectronics

LED5000
3 A monolithic step-down current source with dimming capability
Datasheet - production data
Applications
 High brightness LED driving
 Street lighting
 Signage
 Halogen bulb replacement
HSOP8
HPSO8
 General lighting
Features
Description
 5.5 V to 48 V operating input voltage range
The LED5000 device is an 850 kHz fixed
switching frequency monolithic step-down DC-DC
converter designed to operate as a precise
constant current source with an adjustable current
capability up to 3 A DC. The embedded PWM
dimming circuitry features LED brightness control.
The regulated output current level is set by
connecting a sensing resistor to the feedback pin.
The 200 mV typical RSENSE voltage drop
enhances performance in terms of efficiency. The
size of the overall application is minimized thanks
to the high switching frequency and its
compatibility with ceramic output capacitors. The
device is fully protected against overheating,
overcurrent and output short-circuit. The
LED5000 is available in an HSOP8 package.
 850 kHz fixed switching frequency
 200 mV typ. current sense voltage drop
 Buck / buck-boost / floating boost topologies
 PWM dimming

± 3% output current accuracy overtemperature
 200 m typical RDSON
 Peak current mode architecture
 Short-circuit protection
 Compliant with ceramic output capacitors
 Inhibit for zero current consumption
 Thermal shutdown
Figure 1. Typical application circuit
LED5000
2
DIM
3
VIN
L
SW
8
DIM
INH
COMP
CIN
U1
BOOT
CFLT
4
FB
GND
6
5
2
7
VIN
1
COUT
1
RS
GND
AM13485v1
September 2014
This is information on a product in full production.
DocID023951 Rev 4
1/55
www.st.com
Contents
LED5000
Contents
1
2
Pin settings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
1.1
Pin connection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
1.2
Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
2.1
Maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
2.2
Thermal data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
2.3
ESD protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
3
Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
4
Functional description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
5
4.1
Power supply and voltage reference . . . . . . . . . . . . . . . . . . . . . . . . . . . . .11
4.2
Voltage monitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .11
4.3
Soft-start . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .11
4.4
Dimming block . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
4.5
Inhibit block . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
4.6
Error amplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
4.7
Thermal shutdown . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Application notes - buck conversion . . . . . . . . . . . . . . . . . . . . . . . . . . 14
5.1
Closing the loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
5.2
GCO(s) control to output transfer function . . . . . . . . . . . . . . . . . . . . . . . . 14
5.3
Error amplifier compensation network . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
5.4
LED small signal model . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
5.5
Total loop gain . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
5.6
Compensation network design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
5.7
Example of system design . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 20
5.8
Dimming operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
Dimming frequency vs. dimming depth . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
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LED5000
Contents
5.9
6
Component selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
5.9.1
Sensing resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
5.9.2
Inductor and output capacitor selection . . . . . . . . . . . . . . . . . . . . . . . . . 25
5.9.3
Input capacitor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
5.10
Layout considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
5.11
Thermal considerations . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
5.12
Short-circuit protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 31
5.13
Application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
Application notes - alternative topologies . . . . . . . . . . . . . . . . . . . . . . 37
6.1
Inverting buck-boost . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
6.2
Positive buck-boost . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
6.3
Floating boost . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
6.4
Compensation network design for alternative topologies . . . . . . . . . . . . . 48
6.4.1
fp < BW . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49
6.4.2
fp > BW . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50
7
Package information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
8
Ordering information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
9
Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54
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List of tables
LED5000
List of tables
Table 1.
Table 2.
Table 3.
Table 4.
Table 5.
Table 6.
Table 7.
Table 8.
Table 9.
Table 10.
Table 11.
Table 12.
Table 13.
4/55
Pin description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
Absolute maximum ratings . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Thermal data. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
ESD protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 7
Electrical characteristics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 8
Uncompensated error amplifier characteristics. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13
Inductor selection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 27
List of ceramic capacitors for the LED5000 . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 28
Component list . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 35
BB and boost parameters . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 49
HSOP8 package mechanical data . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 52
Order code . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 53
Document revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 54
DocID023951 Rev 4
LED5000
List of figures
List of figures
Figure 1.
Figure 2.
Figure 3.
Figure 4.
Figure 5.
Figure 6.
Figure 7.
Figure 8.
Figure 9.
Figure 10.
Figure 11.
Figure 12.
Figure 13.
Figure 14.
Figure 15.
Figure 16.
Figure 17.
Figure 18.
Figure 19.
Figure 20.
Figure 21.
Figure 22.
Figure 23.
Figure 24.
Figure 25.
Figure 26.
Figure 27.
Figure 28.
Figure 29.
Figure 30.
Figure 31.
Figure 32.
Figure 33.
Figure 34.
Figure 35.
Typical application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1
Pin connection (top view) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 6
LED5000 block diagram . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 10
Internal circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 11
Soft-start open . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 12
Block diagram of the loop . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 14
Transconductance embedded error amplifier . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16
Equivalent series resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 18
Load equivalent circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 19
Module plot . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
Phase plot . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 21
dimming operation example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 22
LED rising edge operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
LED rising edge operation (zoom) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 23
dimming signal . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 24
Equivalent circuit. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 25
Layout example . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 29
Switching losses . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 30
constant current protection triggering hiccup mode . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33
Evaluation board application circuit . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 34
PCB layout (component side) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
PCB layout (bottom side) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 36
Inverting buck-boost . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 37
LED current source based on inverting BB topology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 39
Inverting BB dimming operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 40
Inverting BB PCB layout (component side) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
Inverting BB PCB layout (bottom side) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 41
Positive buck-boost. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 42
LED current source based on positive BB+ topology . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 43
Floating boost . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 44
LED current source based on floating boost topology . . . . . . . . . . . . . . . . . . . . . . . . . . . . 46
Floating BB dimming operation. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47
Floating boost PCB layout (component side) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 47
Floating boost PCB layout (bottom side) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 48
HSOP8 package outline . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 51
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Pin settings
LED5000
1
Pin settings
1.1
Pin connection
Figure 2. Pin connection (top view)
HSOP8
1.2
AM13486v1
Pin description
Table 1. Pin description
Type
6/55
Description
1
BOOT
Analog circuitry power supply connection
2
DIM
Dimming control input. Logic low prevents the switching activity, logic high
enables it. A square wave on this pin implements LEDs current PWM
dimming. Connect to VIN if not used (see Section 5.8 on page 22).
3
INH
Inhibit pin. Connect to GND if not used.
4
COMP
5
FB
6
GND
Ground connection
7
VIN
Power input voltage
8
SW
Switching node
-
e.p.
Exposed pad to be connected to GND to increase the package thermal
performance and the device noise immunity.
Analog circuitry
Feedback input. Connect a proper sensing resistor to set the LED current.
DocID023951 Rev 4
LED5000
Maximum ratings
2
Maximum ratings
2.1
Maximum ratings
Table 2. Absolute maximum ratings
Symbol
Value
Unit
VIN
Power supply input voltage
-0.3 to 52
V
VINH
Inhibit input
-0.3 to 7
V
VDIM
Dimming input
-0.3 to (VIN + 0.3)
V
VCOMP
Comp output
-0.3 to 3
V
BOOT
Bootstrap pin
-0.3 to 55
V
-1 to (VIN + 0.3)
V
-0.3 to 3
V
Operating junction temperature range
-40 to 150
°C
TSTG
Storage temperature range
-65 to 150
°C
TLEAD
Lead temperature (soldering 10 sec.)
260
°C
Value
Unit
40
C/W
SW
Switching node
VFB
Feedback voltage
TJ
2.2
Parameter
Thermal data
Table 3. Thermal data
Symbol
Rth JA(1)
Parameter
Thermal resistance junction ambient
1. Device soldered to the STEVAL-ILL056V1 demonstration board.
2.3
ESD protection
Table 4. ESD protection
Symbol
ESD
Test condition
Value
Unit
HBM
4
KV
MM
500
V
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Electrical characteristics
3
LED5000
Electrical characteristics
All tests performed at TJ = 25 °C, VCC = 12 V, VINH = 0 V unless otherwise specified. The
specification is guaranteed from -40 to +125 °C - TJ temperature range by design,
characterization and statistical correlation.
Table 5. Electrical characteristics
Symbol
VIN
RDS(on)
ISW
Parameter
Test condition
Min.
Operating input voltage range
5.5
MOSFET on resistance
ISW = 1 A
Maximum limiting
current
3.7
tHICCUP Hiccup time
fSW
Typ.
Max. Unit
48
V
0.2
0.4

4.5
5.2
A
16
Switching frequency
600
850
ms
1000
kHz
Duty cycle
(1)
90
%
TON MIN
Minimum conduction time of the power
element
(1)
90
ns
TOFF MIN
Minimum conduction time of the external
diode
(1)
75
90
120
ns
194
200
206
mV
DC characteristics
VFB
Voltage feedback
IFB
FB biasing current
Iq
Quiescent current
Iqst-by
Standby quiescent current
50
nA
VDIM > 1.5 V
1.3
2
mA
VDIM > 1.5 V, VIN = 48 V
1.7
2.4
mA
16
34
A
0.5
V
VINH > 1.5 V
12
Inhibit
VINH
IINH
Inhibit levels
Inhibit biasing current
Device ON
VIN = 5.5 V to 48 V
Device OFF
VIN = 5.5 V to 48 V
1.5
VINH = 5 V
0.7
Switching activity
VIN = 5.5 V to 48 V
2.2
V
1.6
2.5
A
Dimming
VDIM
8/55
Dimming levels
Switching activity
prevented
VIN = 5.5 V to 48 V
DocID023951 Rev 4
V
0.5
V
LED5000
Electrical characteristics
Table 5. Electrical characteristics (continued)
Symbol
Parameter
Test condition
Min.
Typ.
Max. Unit
Error amplifier
VOH
High level output voltage
VFB = 0 V
VOL
Low level output voltage
VFB = 400 mV
Io source Source output current
V
150
mV
VCOMP = 1.5 V; VFB = 0 V
16
23
30
A
16
23
30
A
Io sink
Sink output current
VCOMP = 1.5 V; VFB = 0.4 V
Ib
Source bias current
VFB = 250 mV
DC open loop gain
RL = 
Transconductance
ICOMP = TBD;
VCOMP = TBD
gm
3
(1)
50
nA
90
dB
220
S
Thermal shutdown
(1)
TSHDWN Thermal shutdown temperature
THYS
1.
(1)
Thermal shutdown hysteresis
140
150
15
160
C
C
Parameter guaranteed by design.
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Functional description
4
LED5000
Functional description
The LED5000 is based on a “peak current mode” architecture with fixed frequency control.
As a consequence the intersection between the error amplifier output and the sensed
inductor current generates the control signal to drive the power switch.
The main internal blocks shown in the block diagram in Figure 3 are:

A fully integrated sawtooth oscillator with a typical frequency of 850 kHz

A transconductance error amplifier

A high side current sense amplifier to track the inductor current

A pulse width modulator (PWM) comparator and the circuitry necessary to drive the
internal power element

The soft-start circuitry to decrease the inrush current at power-up

The dimming block to implement PWM dimming

The inhibit block for standby operation

The current limitation circuit based on the pulse-by-pulse current and the HICCUP
protection

The bootstrap circuitry to drive the embedded NMOS switch

A circuit to implement the thermal protection function
Figure 3. LED5000 block diagram
%227
9,1
2&3
5()
26&
2&3
5(*8/$725
%2275(*
,B6(16(
026)(7
&21752/
/2*,&
&203
'5,9(5
JP
6:
'5,9(5
273
',00,1*
,1+
',0
,1+
62)767$57
)%
&203
*1'
$0
10/55
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LED5000
4.1
Functional description
Power supply and voltage reference
The internal regulator circuit consists of a startup circuit, an internal voltage pre-regulator,
the bandgap voltage reference and the bias block that provides current to all the blocks. The
starter supplies the startup current to the entire device when the input voltage goes high and
the device is enabled (inhibit pin connected to ground). The pre-regulator block supplies the
bandgap cell with a pre-regulated voltage that has a very low supply voltage noise
sensitivity.
4.2
Voltage monitor
An internal block continuously senses the Vcc, Vref and Vbg. If the monitored voltages are
good, the regulator begins operating. There is also a hysteresis on the Vcc(UVLO).
Figure 4. Internal circuit
AM13488v1
4.3
Soft-start
The startup phase is implemented ramping the reference of the embedded error amplifier in
1 msec typ. time. It minimizes the inrush current and decreases the stress of the power
components at power up.
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Functional description
LED5000
Figure 5. Soft-start open
AM13489v1
During normal operation a new soft-start cycle takes place in case of:

thermal shutdown event

UVLO event
The soft-start is disabled during the dimming operation to maximize the dimming
performance.
4.4
Dimming block
The DIM input features the LED brightness control with the PWM dimming operation (see
Section 5.8 on page 22).
4.5
Inhibit block
The inhibit block features the standby mode accordingly with Table 5: Electrical
characteristics on page 8. The INH pin high level disables the device so the power
consumption is reduced to less than 40 µA. The INH pin is 5 V tolerant.
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4.6
Functional description
Error amplifier
The voltage error amplifier is the core of the loop regulation. It is a transconductance
operational amplifier whose non inverting input is connected to the internal voltage
reference (200 mV), while the inverting input (FB) is connected to the output current sensing
resistor.
Table 6. Uncompensated error amplifier characteristics
Description
Values
Transconductance
220 µS
Low frequency gain
90 dB
The error amplifier output is compared with the inductor current sense information to
perform PWM control.
4.7
Thermal shutdown
The shutdown block generates a signal that disables the power stage if the temperature of
the chip goes higher than a fixed internal threshold (150 ± 10 °C typical). The sensing
element of the chip is close to the PDMOS area, ensuring fast and accurate temperature
detection. A 15 °C typical hysteresis prevents the device from turning ON and OFF
continuously during the protection operation.
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LED5000
5
Application notes - buck conversion
5.1
Closing the loop
Figure 6. Block diagram of the loop
GCO(s)
VIN
PWM control
Current sense
HS
switch
L
VOUT
LC filter
LS
switch
COUT
error
PWM
+
amplifier
VCONTROL
+
comparator
RC
FB
VREF
RS
compensation
network
CC
α LED
AO(s)
AM13490v1
5.2
GCO(s) control to output transfer function
The accurate control to output transfer function for a buck peak current mode converter can
be written as:
Equation 1
s
 1 + ----


R LOAD
1
z
G CO  s  = ------------------  ----------------------------------------------------------------------------------------  ----------------------  F H  s 
R 0  T SW
R CS
s

1 + -----------------------   m C   1 – D  – 0.5   1 + ----- p
L
where RLOAD represents the load resistance (see Section 5.4 on page 18), RCS the
equivalent sensing resistor of the current sense circuitry equal to 0.38, p the single pole
introduced by the LC filter and z the zero given by the ESR of the output capacitor.
FH(s) accounts the sampling effect performed by the PWM comparator on the output of the
error amplifier that introduces a double pole at one half of the switching frequency.
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Equation 2
1
 Z = ------------------------------ESR  C OUT
where ESR is the equivalent series resistor to the output capacitor.
Equation 3
m C   1 – D  – 0.5
1
 P = -------------------------------------- + --------------------------------------------R LOAD  C OUT
L  C OUT  f SW
where:
Equation 4
Se

 m C = 1 + -----Sn

 S = V f
pp SW
 e

V IN – V OUT
 S = -----------------------------  R CS
 n
L
Sn represents the slope of the sensed inductor current, Se the slope of the external ramp
(VPP peak to peak amplitude equal to 1.2 V) that implements the slope compensation to
avoid sub-harmonic oscillations at duty cycle over 50%.
The sampling effect contribution FH(s) is:
Equation 5
1
F H  s  = -----------------------------------------2
s
s
1 + ------------------- + ------2
n  QP 
n
where:
Equation 6
 n =   f SW
and
Equation 7
1
Q P = ---------------------------------------------------------   m C   1 – D  – 0.5 
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5.3
LED5000
Error amplifier compensation network
The external compensation network connected at the output of the error amplifier is
dimensioned to stabilize the system depending on the application conditions.
Figure 7. Transconductance embedded error amplifier
+
E/A
COMP
-
FB
RC
CP
CC
V+
R0
dV
C0
RC
CP
CC
Gm dV
AM13491v1
RC and CC introduce a pole and a zero in the open loop gain. CP does not significantly affect
system stability but it can be useful to reduce the noise at the output of the error amplifier.
The transfer function of the error amplifier and its compensation network is:
Equation 8
A V0   1 + s  R c  C c 
A 0  s  = ---------------------------------------------------------------------------------------------------------------------------------------------------------------------------------------2
s  R0   C0 + Cp   Rc  Cc + s   R0  Cc + R0   C0 + Cp  + Rc  Cc  + 1
Where AV0 = Gm · R0 (RO = output resistor of OTA = 200 * 10 ^ 6 ).
The poles of this transfer function are (if Cc >> C0 + CP):
Equation 9
1
f P LF = ---------------------------------2    R0  Cc
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Equation 10
1
f P HF = ---------------------------------------------------2    Rc   C0 + Cp 
whereas the zero is defined as:
Equation 11
1
F Z = --------------------------------2    Rc  Cc
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5.4
LED5000
LED small signal model
Once the system reaches the working condition the LEDs composing the row are biased
and their equivalent circuit can be considered as a resistor for frequencies << 1 MHz.
The LED manufacturer typically provides the equivalent dynamic resistance of the LED
biased at a different DC current. This parameter is required to study the behavior of the
system in the small signal analysis.
For instance, the equivalent dynamic resistance of Luxeon III Star from Lumiled measured
with different biasing current levels is reported below:
 1.3

 0.9
r LED
I LED = 350mA
I LED = 700mA
In case the LED datasheet does not provide the equivalent resistor value, it can be easily
derived as the tangent to the diode I - V characteristic in the present working point (see
Figure 8).
Figure 8. Equivalent series resistor
[A]
1
working point
0.1
1
2
3
4
[V]
Figure 9 shows the equivalent circuit of the LED constant current generator.
The equivalent loading resistor in the LEDs working point is:
Equation 12
R LOAD = n LED  r LED + R S
where RS is the resistor put in series to the LED string.
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LED5000
Application notes - buck conversion
Figure 9. Load equivalent circuit
L
Dled1
VIN
D
COUT
Dled2
Rs
L
Rd1
VIN
D1
COUT
Rd2
Rs
AM13493v1
As a consequence the LED equivalent circuit gives the LED(s) term correlating the output
voltage with the high impedance FB input:
Equation 13
R SENSE
 LED  n LED  = ---------------------------------------------------------n LED  r LED + R SENSE
5.5
Total loop gain
In summary, the open loop gain can be expressed as:
Equation 14
G  s  = G CO  s   A 0  s   LED  n LED 
5.6
Compensation network design
The maximum bandwidth of the system can be designed up to fSW/6 to guarantee a valid
small signal model.
Equation 15
P
f SW
-----------  BW  BW MAX = --------2
6
where P (Equation 3) is the pole introduced by the power components. The following
calculations are valid in the hypothesis that BW > P which is the typical condition.
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LED5000
With the power components selected in accordance with Section 5.9: Component selection
on page 25 and given the BW specification, the components composing the compensation
network can be calculated as:
Equation 16
RC
R LOAD  T SW
1 + ---------------------------------   m C   1 – D  – 0.5  BW  R
L
CS
= --------------------------------------------------------------------------------------------------  ------------------------Gm  RS
fP
where the term mC is represented in Equation 4 on page 15, RLOAD the equivalent loading
resistor (Equation 12), RS the resistor put in series to the LED string, Gm the error amplifier
transconductance and RCS the equivalent sensing resistor of the current sense circuitry
equal to 0.38 (Table 5: Electrical characteristics on page 8).
Equation 17
K
C C = ---------------------R C  BW
where K represents the leading position of the FZ (Equation 11) with respect to the system
bandwidth. In general, a value of 2 gives enough phase margin to the overall small loop
transfer function.
5.7
Example of system design
Design specification:
VIN = 48 V, VFW_LED = 3.7 V, nLED = 10, rLED = 1.1 , ILED = 1 A, ILED RIPPLE = 2%.
The inductor and capacitor value are dimensioned to meet the ILED RIPPLE specification (see
Section 5.9.2 on page 25 for output capacitor and inductor selection guidelines):
L = 22 H, COUT = 1.0 F mlcc (negligible ESR).
In accordance with Section 5.9.1 on page 25 the sensing resistor value is:
Equation 18
200 mV
R S = -------------------- = 200 m
1A
Assuming a system bandwidth of:
Equation 19
BW = 70 kHz  BW MAX
The ideal values of the components making up the compensation network are:
Equation 20
R C = 43 k
C C = 650 pF
Final component selection is based on commercial values and a small capacitor CP is
added to reduce noise at the error amplifier output. CP slightly decreases the BW and phase
margin.
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Equation 21
R C = 47 k
C C = 680 pF
C C = 12 pF
The gain and phase margin bode diagrams are plotted, respectively, in Figure 10 and
Figure 11.
Figure 10. Module plot
External loop module
100
87
74
61
Module [dB]
48
35
22
9
4
17
30
0.1
1
10
3
1 .10
100
4
1 .10
5
1 .10
Frequency [Hz]
6
1 .10
AM13494v1
Figure 11. Phase plot
External loop gain phase
180
157.5
135
112.5
90
67.5
45
22.5
0
0.1
1
10
100
3
1 .10
4
1 .10
5
1 .10
6
1 .10
AM13495v1
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LED5000
The cut-off frequency and the phase margin are:
Equation 22
f C = 65 kHz
5.8
pm = 66
Dimming operation
The dimming input disables the switching activity, masking the PWM comparator output.
The inductor current dynamic performance when dimming input goes high depends on the
designed system response. The best dimming performance is obtained by maximizing the
bandwidth and phase margin, when possible.
As a general rule, the output capacitor minimization improves dimming performance.
Figure 12. dimming operation example
AM13496v1
In fact, when dimming enables the switching activity, a small capacitor value is fast charged
with low inductor value. As a consequence, the LEDs current rising edge time is improved
and the inductor current oscillation reduced. An oversized output capacitor value requires
extra current for fast charge so generating an inductor current overshoot and oscillations.
The switching activity is prevented as soon as the dimming signal goes low. Nevertheless,
the LED current drops to zero only when the voltage stored in the output capacitor goes
below a minimum voltage determined by the selected LEDs. As a consequence, a big
capacitor value makes the LED current falling time worse than a smaller one.
The LED5000 device embeds dedicated circuitry to improve LED current rising edge time.
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Figure 13. LED rising edge operation
AM13497v1
Figure 14. LED rising edge operation (zoom)
AM13498v1
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LED5000
Dimming frequency vs. dimming depth
As seen in Section 5.8 the LEDs current rising and falling edge time mainly depends on the
system bandwidth (TRISE) and the selected output capacitor value (TRISE and TFALL).
The dimming performance depends on the minimum current pulse shape specification of
the final application. The ideal minimum current pulse has rectangular shape, in any case it
degenerates into a trapezoid or, at worst, into a triangle, depending on the ratio (TRISE +
TFALL)/ TDIM.
Equation 23
rec tan gle
T
+T
RISE
FALL
--------------------------------------------- « 1
T
DIM
trapezoid

T
+T
RISE
FALL
---------------------------------------------  1
T
DIM
triangle

T
+T
RISE
FALL
--------------------------------------------- = 1
T
DIM
The small signal response in Figure 14 is considered as an example.
Equation 24
 T RISE  5s

 T FALL  2s
Assuming the minimum current pulse (TMIN_PULSE) shape specification as:
Equation 25
T RISE + T FALL = 0.75  T MIN_PULSE = 0.75  D MIN  T DIMMING
where TDIMMING represents the dimming period and DMIN the minimum duty cycle which
gives the TMIN_PULSE charge. In the given example TMIN_PULSE = 9 µs.
Figure 15. dimming signal
AM13499v1
Given TMIN_PULSE it is possible to calculate the maximum dimming depth given the dimming
frequency or vice versa.
For example, assuming a 10 KHz dimming frequency the maximum dimming depth is 9% or
given a 5% dimming depth it follows a 5.5 KHz maximum fDIM.
The LED5000 dimming performance is strictly dependent on the system small signal
response. As a consequence, an optimized compensation network (good phase margin and
bandwidth maximized) and minimized COUT value are crucial for best performance. Once
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the external power components and the compensation network are selected, a direct
measurement to determine TRISE, TFALL (see Equation 24) is necessary to certify the
achieved dimming performance.
5.9
Component selection
5.9.1
Sensing resistor
In closed loop operation the LED5000 feedback pin voltage is 200 mV, so the sensing
resistor calculation is expressed as:
Equation 26
200 mV
R S = --------------------I LED
Since the main loop (see Section 5.1 on page 14) regulates the sensing resistor voltage
drop, the average current is regulated into the LEDs. The integration period is at minimum
5 * TSW since the system bandwidth can be dimensioned up to fSW/5 at maximum.
A system loop based on a peak current mode architecture features consistent advantages
in comparison with simpler closed loop regulation schemes like the hysteretic or the
constant ON/OFF control.
The system performs the output current regulation over a period which is at least five times
longer than the switching frequency. The output current regulation neglects the ripple
current contribution and its reliance on external parameters like input voltage and output
voltage variations (line transient and LED forward voltage spread). This performance cannot
be achieved with simpler regulation loops like hysteretic control.
For the same reason, the switching frequency is constant over the application conditions,
that helps to tune the EMI filtering and to guarantee the maximum LED current ripple
specification in the application range. This performance cannot be achieved using constant
ON/OFF time architectures.
Inductor and output capacitor selection
The output capacitor filters the inductor current ripple that, given the application condition,
depends on the inductor value. As a consequence the LED current ripple, that is the main
specification for a switching current source, depends on the inductor and output capacitor
selection.
Figure 16. Equivalent circuit
DCR
L
DCR
L
Dled1
Rd1
ESR
VIN
2
2
ESR
Dledn
D
VIN
D
Rdn
1
1
5.9.2
COUT
COUT
Rs
Rs
AM13500v1
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LED5000
The LED ripple current can be calculated as the inductor ripple current ratio flowing into the
output impedance using the Laplace transform (see Figure 11 on page 21):
Equation 27
8
-----2-  I L   1 + s  ESR  C OUT 

I RIPPLE  s  = ----------------------------------------------------------------------------------------------------------1 + s   R S + ESR + n LED  R LED   C OUT
where the term 8/2 represents the main harmonic of the inductor current ripple (which has
a triangular shape) and IL is the inductor current ripple.
Equation 28
V OUT
n LED  V FW_LED + 200mV
I L = --------------  T OFF = ------------------------------------------------------------------  T OFF
L
L
so L value can be calculated as:
Equation 29
n LED  V FW_LED + 200mV
n LED  V FW_LED + 200mV
n LED  V FW_LED + 200mV
L = ------------------------------------------------------------------  T OFF = ------------------------------------------------------------------   1 – ------------------------------------------------------------------
I L
I L
V IN
where TOFF is the OFF time of the embedded high switch, given by 1 - D.
As a consequence the lower is the inductor value (so higher the current ripple), the higher
would be the COUT value to meet the specification.
A general rule to dimension L value is:
Equation 30
I L
-----------  0.5
I LED
Finally the required output capacitor value can be calculated equalizing the LED current
ripple specification with the module of the Fourier transformer (see Equation 27) calculated
at fSW frequency.
Equation 31
I RIPPLE  s=j    = I RIPPLE_SPEC
Example 1(see Section 5.6 on page 19):
VIN = 48 V, ILED = 700 mA, ILED/ILED = 2%, VFW_LED = 3.7 V, nLED = 10.
A lower inductor value maximizes the inductor current slew rate for better dimming
performance. Equation 30 becomes:
Equation 32
I L
----------- = 0.5
I LED
which is satisfied selecting a 10 H inductor value.
The output capacitor value has to be dimensioned according to Equation 31.
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Finally, given the selected inductor value, a 1 F ceramic capacitor value keeps the LED
current ripple ratio lower than the 2% of the nominal current. An output ceramic capacitor
type (negligible ESR) is suggested to minimize the ripple contribution given a fixed capacitor
value.
Table 7. Inductor selection
Manufacturer
Wurth Elektronik
Coilcraft
5.9.3
Series
Inductor value (µH)
Saturation current (A)
WE-HCI 7040
1 to 4.7
20 to 7
WE-HCI 7050
4.9 to 10
20 to 4.0
XPL 7030
2.2 to 10
29 to 7.2
Input capacitor
The input capacitor must be able to support the maximum input operating voltage and the
maximum RMS input current.
Since step-down converters draw current from the input in pulses, the input current is
squared and the height of each pulse is equal to the output current. The input capacitor has
to absorb all this switching current, whose RMS value can be up to the load current divided
by two (worst case, with duty cycle of 50%). For this reason, the quality of these capacitors
has to be very high to minimize the power dissipation generated by the internal ESR,
thereby improving system reliability and efficiency. The critical parameter is usually the RMS
current rating, which must be higher than the RMS current flowing through the capacitor.
The maximum RMS input current (flowing through the input capacitor) is:
Equation 33
2
2
2D
D
I RMS = I O  D – --------------- + ------2

Where  is the expected system efficiency, D is the duty cycle and IO is the output DC
current. Considering  = 1 this function reaches its maximum value at D = 0.5 and the
equivalent RMS current is equal to IO divided by 2. The maximum and minimum duty cycles
are:
Equation 34
V OUT + V F
D MAX = ------------------------------------V INMIN – V SW
and
Equation 35
V OUT + V F
D MIN = -------------------------------------V INMAX – V SW
Where VF is the freewheeling diode forward voltage and VSW the voltage drop across the
internal PDMOS. Considering the range DMIN to DMAX, it is possible to determine the max.
IRMS going through the input capacitor.
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Capacitors that can be considered are:

Electrolytic capacitors:
These are widely used due to their low price and their availability in a wide range of RMS
current ratings.
The only drawback is that, considering ripple current rating requirements, they are
physically larger than other capacitors.

Ceramic capacitors:
If available for the required value and voltage rating, these capacitors usually have a higher
RMS current rating for a given physical dimension (due to very low ESR).
The drawback is the considerably high cost.

Tantalum capacitors:
Small tantalum capacitors with very low ESR are becoming more available. However, they
can occasionally burn if subjected to very high current during charge.
Therefore, it is suggested to avoid this type of capacitor for the input filter of the device as
they could be stressed by a high surge current when connected to the power supply.
Table 8. List of ceramic capacitors for the LED5000
Manufacturer
Series
Capacitor value (µF)
Rated voltage (V)
Taiyo Yuden
UMK325BJ106MM-T
10
50
Murata
GRM42-2 X7R 475K 50
4.7
50
In case the selected capacitor is ceramic (so neglecting the ESR contribution), the input
voltage ripple can be calculated as:
Equation 36
IO
D
D
V IN PP = -----------------------   1 – ----  D + ----   1 – D 
C IN  f SW 


5.10
Layout considerations
The layout of switching DC-DC converters is very important to minimize noise and
interference. Power-generating portions of the layout are the main cause of noise and so
high switching current loop areas should be kept as small as possible and lead lengths as
short as possible.
High impedance paths (in particular the feedback connections) are susceptible to
interference, so they should be as far as possible from the high current paths. A layout
example is provided in Figure 17.
The input and output loops are minimized to avoid radiation and high frequency resonance
problems. The feedback pin to the sensing resistor path must be designed as short as
possible to avoid pick-up noise. Another important issue is the ground plane of the board.
Since the package has an exposed pad, it is very important to connect it to an extended
ground plane in order to reduce the thermal resistance junction to ambient and increase the
noise immunity during the switching operation.
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In addition, to increase the design noise immunity, different signal and power grounds
should be designed in the layout (see Section 5.13: Application circuit on page 34). The
signal ground serves the small signal components, the device analog ground pin, the
exposed pad and a small filtering capacitor connected to the VCC pin. The power ground
serves the device ground pin and the input filter. The different grounds are connected
underneath the output capacitor. Neglecting the current ripple contribution, the current
flowing through this component is constant during the switching activity and so this is the
cleanest ground point of the buck application circuit.
Figure 17. Layout example
AM13501v1
5.11
Thermal considerations
The dissipated power of the device is tied to three different sources:

Conduction losses due to the RDSON, which are equal to:
Equation 37
2
P ON = R RDSON_HS   I OUT   D
Where D is the duty cycle of the application. Note that the duty cycle is theoretically given by
the ratio between VOUT (nLED VLED + 200 mV) and VIN, but in practice it is substantially
higher than this value to compensate for the losses in the overall application.
For this reason, the conduction losses related to the RDSON increase compared to an ideal
case.
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
LED5000
Switching losses due to turning ON and OFF. These are derived using Equation 38:
Equation 38
 T RISE + T FALL 
P SW = V IN  I OUT  -----------------------------------------  F SW = V IN  I OUT  T SW_EQ  F SW
2
Where TRISE and TFALL represent the switching times of the power element that cause the
switching losses when driving an inductive load (see Figure 18). TSW is the equivalent
switching time.
Figure 18. Switching losses
AM13502v1

Quiescent current losses.
Equation 39
P Q = V IN  I Q
Example 2 (see Section 5.6 on page 19):
VIN = 42 V, VFW_LED = 3.7 V, nLED = 8, ILED = 1500 mA.
The typical output voltage is:
Equation 40
V OUT = n LED  V FW_LED + V FB = 29.4V
RDSON_HS has a typical value of 200  at 25 °C.
For the calculation we can estimate RDSON_HS = 300 m as a consequence of TJ increase
during the operation.
TSW_EQ is approximately 12 ns.
IQ has a typical value of 2.4 mA at VIN = 48 V.
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The overall internal losses are:
Equation 41
2
P TOT = R DSON_HS   I OUT   D + V IN  I OUT  f SW  T SW
EQ
+ V IN  I Q
where TSW_EQ = (TRISE + TFALL )/2 = 12 nS.
Equation 42
2
P TOT = 0.3  1.5  0.7 + 42  1.5  12  10
–9
3
 850  10 + 42  2.4  10
–3
 1.2W
The junction temperature of the device will be:
Equation 43
T J = T A + Rth J – A  P TOT
Where TA is the ambient temperature and RthJ-A is the thermal resistance junction to
ambient. The junction to ambient (RthJ-A) thermal resistance of the device assembled in the
HSOP8 package and mounted on the evaluation is about 40 °C/W.
Assuming the ambient temperature around 40 °C, the estimated junction temperature is:
C
T J = 40 + 1.2W  40  -------  110C
W
5.12
Short-circuit protection
In overcurrent protection mode, when the peak current reaches the current limit threshold,
the device disables the power element and it is able to reduce the conduction time down to
the minimum value (approximately 100 nsec typical) to keep the inductor current limited.
This is the pulse-by-pulse current limitation to implement constant current protection feature.
In overcurrent condition, the duty cycle is strongly reduced and, in most applications, this is
enough to limit the switch current to the current threshold.
The inductor current ripple during ON and OFF phases can be written as:

ON phase
Equation 44
V IN – V OUT –  DCR L + R DSON HS   I
I L TON = -----------------------------------------------------------------------------------------------  T ON 
L

OFF phase
Equation 45
–  V OUT + DCR L  I + V FW DIODE 
I L TON = -------------------------------------------------------------------------------------  T OFF 
L
where DCRL is the series resistance of the inductor and VFWDIODE is the forward voltage
drop across the external rectifying diode.
DocID023951 Rev 4
31/55
55
Application notes - buck conversion
LED5000
The pulse-by-pulse current limitation is effective to implement constant current protection
when:
Equation 46
I L TON = I L TOFF
From Equation 44 and Equation 45 we can gather that the implementation of the constant
current protection becomes more critical the lower is the VOUT and the higher is VIN.
In fact, in short-circuit condition the voltage applied to the inductor during the OFF time
becomes equal to the voltage drop across parasitic components (typically the DCR of the
inductor and the forward voltage of the diode) since VOUT is negligible, while during TON the
voltage applied the inductor is maximized and it is approximately equal to VIN.
In general the worst case scenario is a heavy short-circuit at the output with maximum input
voltage. Equation 44 and Equation 45 in overcurrent conditions can be simplified to:
Equation 47
V IN –  DCR L + R DSON HS   I
V IN
I L TON = ------------------------------------------------------------------------  T ON MIN   ---------  90ns 
L
L
considering TON that has been already reduced to its minimum.
Equation 48
–  DCR L  I + V FW DIODE 
–  DCR L  I + V FW DIODE 
I L TOFF = ----------------------------------------------------------------  T SW – 90ns   ----------------------------------------------------------------  1.18s 
L
L
where TSW = 1/fSW considering the nominal fSW.
At high input voltage IL TON could be higher than IL TOFF and so the inductor current
could escalate. As a consequence, the system typically meets Equation 46 at a current level
higher than the nominal value thanks to the increased voltage drop across stray
components. In most application conditions the pulse-by-pulse current limitation is effective
to limit the inductor current. Whenever the current escalates, a second level current
protection called “hiccup mode” is enabled. The hiccup protection offers an additional
protection against heavy short-circuit condition at very high input voltage even considering
the spread of the minimum conduction time of the power element. In case the hiccup current
level (6.2 A typical) is triggered the switching activity is prevented for 16 msec typ. (see
hiccup time in Table 5: Electrical characteristics on page 8).
32/55
DocID023951 Rev 4
LED5000
Application notes - buck conversion
Figure 19 shows the operation of the constant current protection when a short-circuit is
applied at the output at the maximum input voltage.
Figure 19. constant current protection triggering hiccup mode
AM13503v1
DocID023951 Rev 4
33/55
55
Application notes - buck conversion
5.13
LED5000
Application circuit
Figure 20. Evaluation board application circuit
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The network D3, R4, RS implements an inexpensive overvoltage protection. R4 effect can
be neglected during normal operation since the FB biasing current is negligible (tens of nA,
see Table 5: Electrical characteristics on page 8) but it limits the current flowing in the Zener
diode D3. In case the load is disconnected or in case of open LED:
Equation 49
V OUT = V FB + V ZENER_DIODE
V FB
I ZENER_DIODE = --------------------RS + R1
R1 must be dimensioned to limit the D1 rated power so it is an inexpensive small signal
Zener diode.
The overvoltage limits the output voltage in case of LED disconnection so protecting LEDs
when the string is reconnected with the device enabled. In case the OVP is not
implemented, a large amount of non-controlled current could flow through the LEDs during
the output capacitor discharging phase, thereby damaging the devices.
34/55
DocID023951 Rev 4
LED5000
Application notes - buck conversion
Table 9. Component list
Reference
Part number
Description
Manufacturer
C1, C2
C3225X7S1H106M
10 F 50 V (size 1210)
TDK
C3, C6
100 nF 50 V (size 0805)
C4
470 pF 50 V (size 00603)
C5
22 pF 50 V (size 0603)
C7
C3216X7R1H105K
1 F 50 V (size 1206)
C8
TDK
Not mounted
D1
STPS3L60U
D2
BZX384-C4V7
D3
BZX384-C39
3 A 60 V
ST
Coilcraft
Panasonic
L1
XFL6060-473ME
47 H
ISAT = 1.8 A (10% drop)
IRMS = 3.7 A (40 C rise)
(size 7.3 x 7.3 x 4.1 mm)
RS
ERJ14BSFR27U
0.271%
(size 1206)
R1
10 k 1% (size 0603)
R2
Not mounted
R3
24.9 k 1% (size 0603)
R4
47 k 1% (size 0603)
DocID023951 Rev 4
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55
Application notes - buck conversion
LED5000
Figure 21. PCB layout (component side)
AM13505v1
Figure 22. PCB layout (bottom side)
AM13506v1
36/55
DocID023951 Rev 4
LED5000
6
Application notes - alternative topologies
Application notes - alternative topologies
Thanks to the wide input voltage range, the adjustable external compensation network and
enhanced dimming capability, the LED5000 device is suitable to implement boost and buckboost topologies.
6.1
Inverting buck-boost
The buck-boost topology fits the application with an input voltage range that overlaps the
output voltage, which is the voltage drop across the LEDs and the sensing resistor.
The inverting buck-boost (see Figure 23) requires the same component count as the buck
conversion and it is more efficient than the positive buck-boost. A current generator based
on this topology implies two main application constraints:

the output voltage is negative so the LEDs must be reversed

the device GND floats with the negative output voltage. The device is supplied between
VIN and VOUT (< 0). As a consequence:
Equation 50
V IN_MAX LED5000 = V IN – V OUT
so:
Equation 51
V IN = V IN_MAX LED5000 + V OUT = 48 + V OUT
where VOUT < 0.
Figure 23. Inverting buck-boost
VIN ISW
VREF
AM13507v1
Example 3
VIN RANGE = 12 - 24 V, VFW_LED = 3.7 V, nLED = 5 so VOUT = 18.7 V.
DocID023951 Rev 4
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Application notes - alternative topologies
LED5000
Since the maximum operating voltage of the LED5000 is 48 V, according to Equation 51 the
maximum input voltage of the application is 48 - 18.7 = 29.3 V.
The output voltage is given by:
Equation 52
D IDEAL
V OUT = – V IN  ---------------------------1 – D IDEAL
where the ideal duty cycle DIDEAL for the buck-boost converter is:
Equation 53
– V OUT
D IDEAL = -----------------------------V IN – V OUT
However, due to power losses (mainly switching and conduction losses), the real duty cycle
is always higher than this. The real value (which can be measured in the application) should
be used in the following formulas.
The peak current flowing in the embedded switch is:
Equation 54
I LOAD
I RIPPLE
I LOAD
V IN D
I SW = --------------------------- + -------------------- = --------------------------- + -----------  --------1 – D REAL
2
1 – D REAL 2  L f SW
while its average current level is equal to:
Equation 55
I LOAD
I SW = --------------------------1 – D REAL
This is due to the fact that the current flowing through the internal power switch is delivered
to the output only during the OFF phase.
The switch peak current must be lower than the minimum current limit of the overcurrent
protection (see Table 5: Electrical characteristics on page 8 for details) while the average
current must be lower than the rated DC current of the device.
As a consequence, the maximum output current is:
Equation 56
I LOAD MAX  I SW MAX   1 – D REAL 
where ISW MAX represents the rated current of the device.
The current capability is reduced by the term (1 - DREAL) and so, for example, with a duty
cycle of 0.5, and considering an average current through the switch of 3 A, the maximum
output current deliverable to the load is 1.5 A.
38/55
DocID023951 Rev 4
LED5000
Application notes - alternative topologies
Figure 24 shows the schematic circuit for an LED current source based on inverting buckboost topology. The input voltage ranges from 10 to 26 V and it can drive a string composed
of 5 LEDs with1 A DC.
Figure 24. LED current source based on inverting BB topology
C1 100 nF
1
1
VIN
3
TP1
2
2
3
DIM
8
DIM
T1 jumper
7
TP3
SW
BOOT
VIN
LED5000
INH
Q1 BC807-16
GND
6
EP
TP2
5
VLED-
FB
COMP
EP
4
C9
NM
R3
R4
C4
C5
NM 330 nF
50 V
R5
C7
15 pF
C6
2.2 nF
D2 STPS3L60U
R2
D1 BZX384-C20
C2
C3
10 µF 10 µF
50 V 50 V
L1 22 µH XAL5050
RS1
Small signal
TP4
GND
C8
3.3 µF
50 V
Power plane
VLED+
TP5
AM03647V1
DocID023951 Rev 4
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Application notes - alternative topologies
LED5000
The circuitry Q1, R2, R3, R4 implements a level shifter to convert the dimming signal
voltage levels (referred to GND) to the device rails, since the LED5000 local ground is
referred to the negative output voltage (given by the voltage drop across the LEDs and the
sensing resistor). Figure 25 shows the dimming operation: the light blue trace represents
the DIM pin, the yellow the SW (high level is VIN, low level is -VOUT), the green trace the
inductor current (see Equation 54) and the purple is the output voltage.
Figure 25. Inverting BB dimming operation
AM13509v1
The network D1, R1, RS implements an inexpensive overvoltage protection. R1 effect can
be neglected during normal operation since the FB biasing current is negligible (tens of nA,
see Table 5: Electrical characteristics on page 8) but it limits the current flowing in the Zener
diode D1. In case the load is disconnected or in case of an open LED:
Equation 57
V OUT = V FB + V ZENER_DIODE
V FB
I ZENER_DIODE = --------------------RS + R1
R1 must be dimensioned to limit the D1 rated power so it is an inexpensive small signal
Zener diode.
The overvoltage protection plays an important role for the inverting buck-boost topology. In
fact, in case of open row, the output voltage tends to diverge thus exceeding the input
voltage absolute maximum rate and the device would be damaged (see Equation 50). The
overvoltage protection limits VOUT and thereby it protects the device in case of load
disconnection.
To design the compensation network for the inverting buck-boost topology please refer to
paragraph Chapter 6.4: Compensation network design for alternative topologies on
page 48.
40/55
DocID023951 Rev 4
LED5000
Application notes - alternative topologies
Figure 26. Inverting BB PCB layout (component side)
AM13510v1
Figure 27. Inverting BB PCB layout (bottom side)
AM13511v1
6.2
Positive buck-boost
Positive buck-boost fits those applications that require a buck-boost topology (i.e.: the input
voltage range crosses the output voltage value) and where the inverting buck-boost is not
suitable because of the main constraints for the final application (refer to Section 6.1). As
a consequence the inverting buck-boost is the preferred option because it requires less
components and it has higher efficiency compared to the positive buck-boost topology.
DocID023951 Rev 4
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Application notes - alternative topologies
LED5000
Figure 28. Positive buck-boost
VIN
ISW
VREF
AM13512v1
The positive buck-boost implementation (Figure 28) requires one more diode and an
external power switch than inverting buck-boost. The device is not floating, referred to GND,
and it is supplied with the input voltage of the application (the input voltage in inverting buckboost topology is instead VIN - VOUT, refer to Section 6.1 for details). The LED5000 device
does not see the output voltage during the switching activity so VOUT can be higher than the
maximum input voltage.
The equations for the positive buck-boost are similar to those seen for the inverting.
Equation 58
D IDEAL
V OUT = V IN  ---------------------------1 – D IDEAL
where the ideal duty cycle DIDEAL for the buck-boost converter is:
Equation 59
V OUT
D IDEAL = -----------------------------V IN + V OUT
However, due to power losses (mainly switching and conduction losses), the real duty cycle
is always higher than this. The real value (which can be measured in the application) should
be used in the following formulas.
The peak current flowing in the embedded switch is:
Equation 60
I LOAD
I RIPPLE
I LOAD
V IN D
I SW = --------------------------- + -------------------- = --------------------------- + -----------  --------1 – D REAL
2
1 – D REAL 2  L f SW
while its average current level is equal to:
Equation 61
I LOAD
I SW = --------------------------1 – D REAL
This is due to the fact that the current flowing through the internal power switch is delivered
to the output only during the OFF phase.
42/55
DocID023951 Rev 4
LED5000
Application notes - alternative topologies
The switch peak current must be lower than the minimum current limit of the overcurrent
protection (see Table 5: Electrical characteristics on page 8 for details) while the average
current must be lower than the rated DC current of the device.
As a consequence, the maximum output current is:
Equation 62
I LOAD MAX  I SW MAX   1 – D REAL 
where ISW MAX represents the rated current of the device.
The current capability is reduced by the term (1 - DREAL) and so, for example, with a duty
cycle of 0.5, and considering an average current through the switch of 3 A, the maximum
output current deliverable to the load is 1.5 A.
Figure 29 shows the circuit schematic for an LED current source based on positive buckboost topology. The input voltage ranges from 18 to 30 V and it can drive a string composed
of 7 LEDs with 0.7 A DC (VFW_LED = 3.75 V so VOUT = 26.4 V).
The network D5, R4, RS implements an inexpensive overvoltage protection. R4 effect can
be neglected during the normal operation since the FB biasing current is negligible (tens of
nA, see Table 5: Electrical characteristics on page 8) but it limits the current flowing in the
Zener diode D5. In case the load is disconnected or in case of open LED:
Equation 63
V OUT = V FB + V ZENER_DIODE
V FB
I ZENER_DIODE = --------------------RS + R4
R4 must be dimensioned to limit the D5 rated power so it is an inexpensive small signal
Zener diode.
Figure 29. LED current source based on positive BB+ topology
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DocID023951 Rev 4
43/55
55
Application notes - alternative topologies
LED5000
In case of open row, the positive output voltage tends to diverge, exceeding the D3
maximum reverse voltage and so the diode would be damaged. The overvoltage protection
limits VOUT and it protects the power components in case of load disconnection.
The network D4, R8 implements a level shifter to drive the gate of the transistor Q1. The
voltage at Q1 is:
Equation 64
V Q1 GATE = V SW – V DZ4 = V SW – 15V
Considering the VIN range 18 to 30 V:
Equation 65
V Q1 GATE MIN = V SW – V DZ4 = 18V – 15V = 3V
V Q1 GATE MAX = V SW – V DZ4 = 30V – 16V = 15V
The gate is driven inside the component specification. R8 can be dimensioned to discharge
the gate when VSW is low.
In case the input voltage range of the application is not suitable to implement a level shifter
to drive Q1, a dissipative clamping network (like R5, D6) must be used.
To design the compensation network for the positive buck-boost topology please refer to
Section 6.4: Compensation network design for alternative topologies on page 48.
6.3
Floating boost
The floating boost topology (see Figure 30) serves those applications with an input voltage
range narrower than the output voltage, that is the voltage drop across the LEDs and the
sensing resistor (i.e.: VIN < VOUT). The topology is called floating since the output voltage is
referred to VIN and not GND, but this is typically suitable for a floating load like a string of
LEDs.
Figure 30. Floating boost
VCC
VREF
VIN
LX
GND
AM13514v1
44/55
DocID023951 Rev 4
LED5000
Application notes - alternative topologies
The device is supplied by the output voltage so the maximum voltage drop across the LEDs
string is 48 V. The direct path of the boost conversion (COUT, VDIODE, L) guarantees the
proper startup when the input voltage is:
Equation 66
V IN_START = V OP_MIN + V DIODE = 5.5V + V DIODE
where VOP_MIN is the minimum operating voltage.
The equations for the floating boost are:
Equation 67
V IN
V OUT = ---------------------------1 – D IDEAL
The ideal duty cycle DIDEAL for the boost converter is:
Equation 68
V OUT – V IN
D IDEAL = ----------------------------V OUT
As seen for the buck-boost topologies (Section 6.1 on page 37 and Section 6.2 on page 41),
due to power losses the real duty cycle is always higher than the ideal. The real value (that
can be measured in the application) should be used in the following formulas to estimate the
switch current.
The peak current flowing in the embedded switch is:
Equation 69
I LOAD
I RIPPLE
I LOAD
V IN D
I SW = --------------------------- + -------------------- = --------------------------- + -----------  --------1 – D REAL
2
1 – D REAL 2  L f SW
while its average current level is equal to:
Equation 70
I LOAD
I SW = --------------------------1 – D REAL
This is due to the fact that the current flowing through the internal power switch is delivered
to the output only during the OFF phase.
The switch peak current must be lower than the minimum current limit of the overcurrent
protection (see Table 5: Electrical characteristics on page 8 for details) while the average
current must be lower than the rated DC current of the device.
As a consequence, the maximum output current depends on the application conditions:
Equation 71
I LOAD MAX  I SW MAX   1 – D REAL 
where ISW MAX represents the rated current of the device.
DocID023951 Rev 4
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55
Application notes - alternative topologies
LED5000
The current capability is reduced by the term (1-DREAL) and so, for example, with a duty
cycle of 0.5, and considering an average current through the switch of 3 A, the maximum
output current deliverable to the load is 1.5 A.
Figure 31 shows the circuit schematic for an LED current source based on the floating boost
topology. The input voltage ranges from 12 to 36 V and it can drive a string composed of 11
LEDs with 0.7 A DC (VFW_LED = 3.74 V so VOUT = 41 V).
Figure 31. LED current source based on floating boost topology
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The network D1, R1, RS implements an inexpensive overvoltage protection. R1 effect can
be neglected during the normal operation since the FB biasing current is negligible (tens of
nA, see Table 5: Electrical characteristics on page 8) but it limits the current flowing in the
Zener diode D1. In case the load is disconnected or in case of open LED:
Equation 72
V OUT = V FB + V ZENER_DIODE
V FB
I ZENER_DIODE = --------------------RS + R1
R1 must be dimensioned to limit the D1 rated power so it is an inexpensive small signal
Zener diode.
The circuitry Q1, R3, R4, R5 implements a level shifter to convert the dimming signal
voltage levels (referred to GND) to the device rails, since the LED5000 local ground is
floating. The LED5000 local GND level is:
Equation 73
V LGND = V IN – V OUT
where VLGND represent the local GND value.
46/55
DocID023951 Rev 4
LED5000
Application notes - alternative topologies
Figure 25 on page 40 shows the dimming operation: the light blue trace represents the DIM
pin, the yellow the SW (high level is VIN, low level is VIN - VOUT), the green trace the inductor
current (see Equation 69) and the purple is the output voltage.
Figure 32. Floating BB dimming operation
AM13516v1
To design the compensation network for the boost topology please refer to Section 6.4:
Compensation network design for alternative topologies.
Figure 33. Floating boost PCB layout (component side)
AM13517v1
DocID023951 Rev 4
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55
Application notes - alternative topologies
LED5000
Figure 34. Floating boost PCB layout (bottom side)
AM13518v1
6.4
Compensation network design for alternative topologies
The small signal analysis for the alternative topologies can be written as:
Equation 74
s
s  
 1 – ------------------  1 + ------
 z
 Z_RHP 
R LOAD  1 – D  
G CO  s  = ------------------  ------------------  -------------------------------------------------------------  F H  s 
R CS
K Dx
s
 1 + -----
 p
that shares similar terms with Equation 1 on page 14 which is valid for the buck (see
Equation 1). In addition KDX depends on the topology (different for boost and buck-boost)
and Z_RHP (Equation 75) is a zero in the right half plane:
Equation 75
2
R OUT   1 – D 
 Z_RHP = ----------------------------------------L
The RHP (right half plane) zero has the same 20 dB/dec rising gain magnitude as
a conventional zero, but with 90 degree phase drop instead of lead. This characteristic
cannot be compensated with the error amplifier network so the loop gain is designed to roll
off at lower frequency in order to keep its contribution outside the small signal analysis.
Z_RHP (see Equation 75) depends on the equivalent output resistance, inductor value and
the duty cycle. As a consequence the minimum Z_RHP over the input voltage range
determines the maximum system bandwidth:
Equation 76
1  Z_RHP_MIN f SW
BW  BW MAX = ----  ------------------------------ « --------K
2
6
the system phase margin depends on K.
This paragraph provides the equations to calculate the components of the compensation
network once selected the power components and given the BW specification.
48/55
DocID023951 Rev 4
LED5000
Application notes - alternative topologies
Table 10 summarizes the KD, Km, K parameters useful for the next calculations of the
compensation network.
The DC gain of the total small loop is:
Equation 77
RS 1
A 0 = G m  R EA   1 – D   -----------  ------R CS K D
where Gm is the error amplifier transconductance, REA the equivalent output resistance of
the error amplifier, RCS the internal current sense gain (for these parameters refer to Table 5
on page 8), RS the sensing resistor value, and KD can be calculated from Table 10.
The calculation of the components composing the compensation network depends on the
relative position of the pole fp (see Equation 3 on page 15) and the designed bandwidth BW.
Equation 78
p
KD
1
f p = ----------- = -----------  ------------------------------2
2   C O  R LOAD
Table 10. BB and boost parameters
Boost
KD
Km
Buck-boost
2
R LOAD R LOAD   1 – D 
1
K
1 + ---------------------- + ------------------------------------------------   --------- + ------------------
V OUT
Km  1 – D 
R CS
----------------I LED
1
--------------------------------------------------------------------------------------------------------------------------------------T SW V OUT – V IN
T SW
 0.5 – D   R CS  ------------- + ----------------------------------  R CS  ------------L
V
L
OUT
2
R LOAD  D R LOAD   1 – D 
1
K
1 + --------------------------------- + ------------------------------------------------   --------- + ------------------
K

V OUT
R CS
m 1 – D
----------------I LED
1
----------------------------------------------------------------------------------------------------------------------------------------V
T
T
OUT
SW
SW
 ------------ 0.5 – D   R CS  ------------- + ------------------------------------  R
CS
L
L
V +V
IN
OUT
0.5  R
k
6.4.1
CS
T
SW
 -------------  D   1 – D 
L
T SW
0.5  R CS  -------------  D   1 – D 
L
fp < BW
In case the pole fp is inside the system bandwidth BW, the component values composing the
compensation network can be calculated as:
DocID023951 Rev 4
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55
Application notes - alternative topologies
LED5000
Equation 79
R EA BW
R C = ----------  --------A0
fp
and
Equation 80
K
C C = ------------------------------------2    R C  BW
Where K represents the leading position of the FZ (Equation 11 on page 17) in respect to the
system bandwidth. In general a decade (K = 10) gives enough phase margin to the overall
small loop transfer function.
6.4.2
fp > BW
In case the pole fp is outside the system bandwidth BW, the component values composing
the compensation network can be calculated as:
Equation 81
R EA BW
R C = ----------  --------A0
fp
and
Equation 82
K
C C = -------------------------------2    RC  fp
where K represents the leading position of the FZ (Equation 11 on page 17) in respect to the
pole fp.
50/55
DocID023951 Rev 4
LED5000
7
Package information
Package information
In order to meet environmental requirements, ST offers these devices in different grades of
ECOPACK® packages, depending on their level of environmental compliance. ECOPACK
specifications, grade definitions and product status are available at: www.st.com.
ECOPACK is an ST trademark.
Figure 35. HSOP8 package outline
7195016_D
DocID023951 Rev 4
51/55
55
Package information
LED5000
Table 11. HSOP8 package mechanical data
Dimensions (mm)
Symbol
Min.
Max.
A
1.75
A1
0.15
A2
1.25
b
0.38
0.51
c
0.17
0.25
D
4.80
4.90
5.00
D1
3.10
3.30
3.50
E
5.80
6.00
6.20
E1
3.80
3.90
4.00
E2
2.20
2.40
2.60
e
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Typ.
1.27
h
0.30
0.50
L
0.45
0.80
k
0°
8°
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Ordering information
Ordering information
Table 12. Order code
Order code
Package
Packing
LED5000PHR
HSOP8
Tube
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Revision history
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LED5000
Revision history
Table 13. Document revision history
Date
Revision
31-Jan-2013
1
Initial release.
27-Feb-2014
2
Updated package name in package photo on page 1 (replaced “HPSO8” by “HSOP8”).
Updated Section : Features on page 1 (replaced “200 mW” by “200 mΩ” in “typical
RDSON”).
Updated Table 3 on page 7 (added note 1 below Table 3).
Updated Figure 3 on page 10 (replaced by a new block diagram).
Updated Table 6 on page 13 (changed “transconductance“ from 2200 µS to 220 µS).
Updated Section 5.2 on page 14 [updated text below Equation 1 - added “equal to 0.38”.
Added “where: ESR is the equivalent series resistor to the output capacitor.” below
Equation 2 on page 15. Added “equal to 1.2 V” below Equation 4 on page 15 to “(VPP
peak to peak amplitude)” ].
Updated Section 5.3 (updated text below Equation 8 on page 16 - replaced “Where
Avo = Gm · Ro” by “Where AV0 = Gm · R0 (RO = output resistor of OTA = 200 * 10 ^ 6 )”.
Updated Section 5.4 (updated Equation 12 on page 18 - replaced “RSENSE” by “RS”,
added “where RS is the resistor put in series to the LED string” below Equation 12).
Updated Section 5.6 (updated text below Equation 16 on page 20 - replaced text “RS the
sensing resistor value” by “RS the resistor put in series to the LED string” and “RCS the
current sense gain” by “RCS the equivalent sensing resistor of the current sense circuitry
equal to 0.38”).
Updated Section 5.7 on page 20 (replaced “rLED= 1.1 W” by “rLED= 1.1 “ in “Design
specification”) .
Updated Equation 29 on page 26 (added equations).
Updated Equation 41 on page 31 (replaced “TSW“ by “TSW_EQ“, added text “where:
TSW_EQ = (TRISE + TFALL )/2 = 12 nS” below Equation 41).
Updated text below Equation 43 on page 31 (updated package name - replaced
“HPSO8” by “HSOP8”, replaced “60” of “the estimated junction temperature” by “40”).
Updated Figure 20 on page 34 (replaced by a new application circuit - replaced “R56” by
“R27”).
Updated Table 9 on page 35 [replaced “MSS7341”-473MLD” by
“XFL6060-473ME”, “ISAT = 1.0 A (30% drop)” by “ISAT = 1.8 A (10% drop)” and “IRMS =
1.85 A (40 C rise)” by “IRMS = 3.7 A (40 C rise)”.
of “L1“ component ].
Updated Figure 24 on page 39, Figure 29 on page 43 and Figure 31 on page 46
(replaced by new figures).
Updated Section 7 on page 51 (updated titles, reversed order of Figure 35 and Table 11,
minor modifications).
Updated package name in Table 12 on page 53 (replaced “HPSO8” by “HSOP8”).
Updated cross-references throughout document.
Minor modifications throughout document.
28-Apr-2014
3
Updated Section 6.1: Inverting buck-boost on page 37 (replaced 10 LEDs by 5 LEDs in
text above Figure 24 on page 39).
30-Sep-2014
4
Updated Figure 24 on page 39 (replaced by new figure).
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Changes
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