LT1339 Design Manua

Application Note 73
February 1999
LT1339 Design Manual
Designing the Power Converter
Dale Eagar
START
NO
SANITY
OK
DO YOU
HAVE TO DESIGN
A SWITCHER?
DO YOU
WANT TO DESIGN
A SWITCHER?
YES
YOU’RE
NUTS !
NO
FIND SOMEONE
WHO DOES AND GIVE THEM
THIS APPLICATION NOTE
YES
PROCEED
AN73 • FLOW
AN73-1
Application Note 73
PREFACE
Switching Power Converter: an Early Example
Although the origin of switching power converters is lost
in antiquity, some early machines have been identified that
work on this principle. Among the first was the hydraulic
impulse pump, in use centuries before the use of electricity (see Figure 1). This device allows some of the energy
in the stream of water flowing from point A to point B to be
diverted to pump a smaller amount of water from point B
to point C. Here is how this “water hammer” works (refer
to Figures 1 and 2). First, the control valve SW1 opens at
time t1 and flow commences through the downpipe L1. As
the velocity of water flow in the downpipe increases, so
does the energy stored in the moving mass. (E = 1/2MV2,
where M = mass of the moving water, V = the velocity of
the moving water, and E = energy, usually expressed in
Joules). When the valve SW1 has been open long enough
for the water flow rate to reach a valve that is a significant
percentage of the theoretical maximum flow rate, we shut
the valve SW1. This is t2 in Figure 2. By shutting the control
valve SW1 at time t2, we attempt to arrest the flow through
the downpipe. One thing about energy—it doesn’t like
being abandoned, so the pressure rises until something
gives. Lucky for our downpipe, we have a check valve that
can open and divert the otherwise-trapped energy into C1,
the output damper. Enough H2O moves through the check
valve D1 and into C1 at a pressure head above the load
head H2 to remove all kinetic energy associated with the
down flow in the downpipe. Hence, for every hogshead of
water that flows into the downpipe, a partial hogshead of
water is delivered to the barrel at point C.
CONTROL VALVE
SW1
OUTPUT
DAMPENER
C1
SCREEN
DOWN
PIPE
ENERGY
STORAGE
SITE L1
H2O
A
SUPPLY
HEAD
H1
t2
PRESSURE AT t1
PRESSURE AT t2
AN73-2
H1
H2
FLOWTHROUGH
D1 INTO C1
0
H2O
Figure 1. Hydraulic Impulse Pump (Water Hammer)
0
0
CONTROL
VALVE
SW1
B
JOULES
0
CHECK
VALVE
D1
t1
0
ENERGY STORED
IN L1
C
CLOSED
VELOCITY
H2O FLOW IN
L1
LOAD
HEAD
H2
OPEN
AN73 F02
AN73 F01
t1
t2
Figure 2. Water Hammer Waveforms
Application Note 73
TABLE OF CONTENTS
Introduction and Data Sheet .................................................................................................... 5
Introduction ........................................................................................................................................................... 5
Data Sheet ............................................................................................................................................................. 6
Block Diagram ..................................................................................................................................................... 13
Operation ............................................................................................................................................................. 13
Applications Information ...................................................................................................................................... 15
Typical Applications.............................................................................................................................................. 22
Package Descriptions .......................................................................................................................................... 24
Related Parts ....................................................................................................................................................... 25
Expanded Pin Descriptions .................................................................................................... 26
SYNC (Pin 1) ........................................................................................................................................................ 26
5VREF (Pin 2) ....................................................................................................................................................... 27
CT (Pin 3) ............................................................................................................................................................ 28
SL/ADJ (Pin 4) .................................................................................................................................................... 28
IAVE (Pin 5) .......................................................................................................................................................... 30
SS (Pin 6) ............................................................................................................................................................ 31
VC (Pin 7) ............................................................................................................................................................ 32
SGND (Pin 8) ....................................................................................................................................................... 33
VFB (Pin 9) ........................................................................................................................................................... 33
VREF (Pin 10) ....................................................................................................................................................... 34
SENSE + (Pin 11) ................................................................................................................................................. 34
SENSE – (Pin 12) ................................................................................................................................................. 35
RUN/SHDN (Pin 13) ............................................................................................................................................ 35
PHASE (Pin 14) ................................................................................................................................................... 36
PGND (Pin 15) ..................................................................................................................................................... 36
BG (Pin 16).......................................................................................................................................................... 36
12VIN (Pin 17) ..................................................................................................................................................... 37
TS (Pin 18) .......................................................................................................................................................... 37
TG (Pin 19) .......................................................................................................................................................... 38
VBOOST (Pin 20) ................................................................................................................................................... 38
Buck Regulator Design ........................................................................................................ 39
Overview.............................................................................................................................................................. 39
Graphical Design Example ................................................................................................................................... 40
3.3V Output Graph Set ........................................................................................................................................ 44
5V Output Graph Set ............................................................................................................................................ 47
12V Output Graph Set ......................................................................................................................................... 50
24V Output Graph Set ......................................................................................................................................... 53
AN73-3
Application Note 73
Boost Converters ...............................................................................................................
Overview..............................................................................................................................................................
Characteristics of the Synchronous Switching Boost Converter ..........................................................................
Graphical Design Section ....................................................................................................................................
Conclusion ..........................................................................................................................................................
5V/60A Input to 28V/9A Boost Converter ............................................................................................................
Circuit Collection .................................................................................................................................................
Typical Applications ...........................................................................................................
28V to 5V/20A Buck Converter ............................................................................................................................
Constant-Current Solenoid Driver with 2 × Turn-On Boost ..................................................................................
2.5A SEPIC Converter. The Output Voltage Can Be Lower or Higher Than the Input Voltage ..............................
AN73-4
58
58
58
59
59
59
60
63
63
64
64
Application Note 73
INTRODUCTION
The advent of the switching regulator has greatly reduced
the size, weight and volume of power conversion circuitry,
while improving both the speed of response and efficiency. With the output voltage requirements going ever
lower and currents ever higher, close scrutiny is applied to
the loss mechanisms of the power converter. The loss
mechanisms are divided into three classes: resistance
loss, fixed voltage loss and switching loss. Resistance
losses are caused by the circuit resistances (input capacitor ESR, power switch on-resistance, DC and AC resistance of the inductor, resistance of any current-sense
elements, resistance in the output diode and ESR in the
output capacitors), each multiplied by the squares of their
respective currents. Fixed voltage losses associated with
diode forward drops can be calculated by multiplying
diode forward voltage by diode current. Switching losses
are caused by both the finite turn-on and turn-off times of
the MOSFETs and the stray capacitance on the source of
the top MOSFET. As the trend toward higher output
current progresses, the first thing to do is minimize all
losses caused by resistance (because the power is proportional to I2R). It is easy to minimize resistance because we
have available very low ESR capacitors, low on-resistance
MOSFETs and low series-resistance inductors. We have
controllers that place a very small voltage across the
current sense resistors. We do such a good job of dropping the resistive loss mechanisms that the output diode
forward-voltage drop becomes the greatest loss mechanism. This is how the mandate for synchronous rectification comes about. Synchronous rectification is achieved
by replacing the output diode with a low on-resistance
switch. With synchronous rectification, efficiencies are
higher, and, more importantly, power dissipated in the
switching power supply is lower, often eliminating the
need for heat sinks and/or fans.
AN73-5
Application Note 73
High Power Synchronous
DC/DC Controller
U
DESCRIPTION
FEATURES
■
■
■
■
■
■
■
■
The LT ®1339 is a high power synchronous current mode
switching regulator controller. The IC drives dual
N-channel MOSFETs to create a single IC solution for high
power DC/DC converters in applications up to 60V.
High Voltage: Operation Up to 60V
High Current: Dual N-Channel Synchronous Drive
Handles Up to 10,000pF Gate Capacitance
Programmable Average Load Current Limiting
5V Reference Output with 10mA External
Loading Capability
Programmable Fixed Frequency Synchronizable
Current Mode Operation Up to 150kHz
Undervoltage Lockout with Hysteresis
Programmable Start Inhibit for Power Supply
Sequencing and Protection
Adaptive Nonoverlapping Gate Drive Prevents
Shoot-Through
The LT1339 incorporates programmable average current
limiting, allowing accurate limiting of DC load current
independent of inductor ripple current. The IC also incorporates user-adjustable slope compensation for minimization of magnetics at duty cycles up to 90%.
The LT1339 timing oscillator operating frequency is programmable and can be synchronized up to 150kHz. Minimum off-time operation provides main switch protection.
The IC also incorporates a soft start feature that is gated by
both shutdown and undervoltage lockout conditions.
U
APPLICATIONS
■
■
■
■
■
■
An output phase reversal pin allows flexibility in configuration of converter types, including inverting and negative
topologies.
48V Telecom Power Supplies
Personal Computers and Peripherals
Distributed Power Converters
Industrial Control Systems
Lead-Acid Battery Backup Systems
Automotive and Heavy Equipment
, LTC and LT are registered trademarks of Linear Technology Corporation.
U
TYPICAL APPLICATION
28V to 5V 20A Buck Converter
VBOOST
+
5VREF
+
CAVG
CCT
2200pF 2200pF
CT
SL/ADJ
BG
D2
MBR0520
IAVG
CSS, 1µF
CVC, 1nF
+
PHASE
VC
CREF
0.1µF
SGND
IRL3103D2
×2
100
CIN
1500µF
63V
×3
L1
10µH
90
SENSE –
+
L1 = CTX02-13400-X2
VOUT
COUT 5V AT 20A
2200µF
6.3V
×2
1339 TA03
AN73-6
70
50
RS
0.005Ω
RFB1
3k
80
60
RRUN
100k
RUN/SHDN
SENSE +
VFB
VREF
RFB2
1k
PGND
SS
RVC, 10k
+
D1
MBR0520
TS
12VIN
LT1339
12V
C12VIN
47µF
IRL3803
TG
RCT
10k
C5VREF
1µF
+
CBST
1µF
EFFICIENCY (%)
SYNC
28V to 5V Efficiency
VIN
28V
DBST
IN5819
0
10
5
15
OUTPUT CURRENT (A)
20
1339 TA03a
Application Note 73
U
U
RATI GS
W
W W
W
AXI U
U
ABSOLUTE
PACKAGE/ORDER I FOR ATIO
(Note 1)
Supply Voltages
Power Supply Voltage (12VIN)...............– 0.3V to 20V
Topside Supply Voltage (VBOOST)
VTS – 0.3V to VTS + 20V (VMAX = 75V)
Topside Reference Pin Voltage (TS) ......– 0.3V to 60V
Input Voltages
Sense Amplifier Input Common Mode ...– 0.3V to 60V
RUN/SHDN Pin Voltage ...................... – 0.3V to 12VIN
All Other Inputs .......................................– 0.3V to 7V
Maximum Currents
5V Reference Output Current............................ 65mA
Maximum Temperatures
Operating Ambient Temperature Range
LT1339C............................................. 0°C to 70°C
LT1339I ......................................... – 40°C to 85°C
Storage Temperature Range ................. – 65°C to 150°C
Lead Temperature (Soldering, 10 sec).................. 300°C
ORDER PART
NUMBER
TOP VIEW
SYNC 1
20 VBOOST
5VREF 2
19 TG
CT 3
18 TS
16 BG
IAVG 5
SS 6
15 PGND
VC 7
14 PHASE
13 RUN/SHDN
SGND 8
VFB 9
12 SENSE –
VREF 10
11 SENSE +
N PACKAGE
20-LEAD PDIP
LT1339CN
LT1339CSW
LT1339IN
LT1339ISW
17 12VIN
SL/ADJ 4
SW PACKAGE
20-LEAD PLASTIC SO WIDE
TJMAX = 125°C, θJA = 70°C/W (N)
TJMAX = 125°C, θJA = 85°C/W (SW)
Consult factory for Military grade parts.
ELECTRICAL CHARACTERISTICS
12VIN = VBOOST = 12V, VC = 2V, TS = 0V, VFB = VREF = 1.25V, CTG = CBG = 3000pF, TA = 25°C unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
14
150
20
250
mA
µA
Supply and Protection
I12VIN
IBOOST
DC Active Supply Current (Note 2)
DC Standby Supply Current
VRUN/SHDN < 0.5V
DC Active Supply Current (Note 2)
DC Standby Supply Current
VRUN/SHDN < 0.5V
VRUN/SHDN Shutdown Rising Threshold
●
●
2.2
0
●
1.15
1.25
mA
µA
1.35
25
V
VSSHYST
Shutdown Threshold Hysteresis
ISS
Soft Start Charge Current
●
4
8
14
mV
µA
VUVLO
Undervoltage Lockout Threshold - Falling
Undervoltage Lockout Threshold - Rising
Undervoltage Lockout Hysteresis
●
●
●
8.20
9.75
9.95
200
9.00
9.35
350
V
V
mV
4.75
5.00
5.25
3
5
mV/V
10
20
mA
mA
–2
V/A
5V Reference
VREF5
IREF5
5V Reference Voltage
Line, Load and Temperature
●
5V Reference Line Regulation
10V ≤ 12VIN ≤ 15V
●
5V Reference Load Range - DC
Pulse
5V Reference Load Regulation
ISC
5V Reference Short-Circuit Current
●
●
0 ≤ IREF5 ≤ 20mA
●
– 1.25
45
V
mA
AN73-7
Application Note 73
ELECTRICAL CHARACTERISTICS
12VIN = VBOOST = 12V, VC = 2V, TS = 0V, VFB = VREF = 1.25V, CTG = CBG = 3000pF, TA = 25°C unless otherwise noted.
SYMBOL
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
●
1.242
1.235
1.250
1.250
1.258
1.265
V
V
●
0.1
0.5
1.0
µA
3200
Error Amplifier
VFB
Error Amplifier Reference Voltage
Measured at Feedback Pin
IFB
Feedback Input Current
gm
Error Amplifier Transconductance
●
1200
2000
AV
Error Amplifier Voltage Gain
●
1500
3000
V/V
IVC
Error Amplifier Source Current
Error Amplifier Sink Current
VFB – VREF = 500mV
●
●
200
280
275
400
µA
µA
VVC
Absolute VC Clamp Voltage
Measured at VC Pin
3.5
V
VSENSE
Peak Current Limit Threshold
Average Current Limit Threshold (Note 4)
Measured at Sense Inputs
Measured at Sense Inputs
Average Current Limit Threshold
Measured at IAVG Pin
2.5
V
15
V/V
VIAVG
VFB = VREF
●
●
170
110
190
120
µmho
130
mV
mV
Current Sense Amplifier
AV
Amplifier DC Gain
Measured at IAVG Pin
VOS
Amplifier Input Offset Voltage
2V < VCMSENSE < 60V,
SENSE+ – SENSE– = 5mV
●
IB
Input Bias Current
Sink (VCMSENSE > 5V)
Source (VCMSENSE = 0V)
●
●
fO ≤ 150kHz
●
●
–5
0.1
mV
45
700
75
1200
µA
µA
150
5
kHz
%
2.75
2.75
mA
mA
V
Oscillator
fO
Operating Frequency, Free Run
Frequency Programming Error (Note 3)
ICT
Timing Capacitor Discharge Current
LT1339C
LT1339I
●
●
2.20
2.10
2.50
2.50
VSYNC
SYNC Input Threshold
Rising Edge
●
0.8
2.0
fSYNC
SYNC Frequency Range
fSYNC ≤ 150kHz
●
fO
1.4fO
12VIN ≤ 8V
VRUN < 0.5V
●
●
Output Drivers
VTG,BG
Undervoltage Output Clamp
Standby Mode Output Clamp
VTG
Top Gate On Voltage
Top Gate Off Voltage
●
●
tTGR
Top Gate Rise Time
●
tTGF
Top Gate Fall Time
●
VBG
Bottom Gate On Voltage
Bottom Gate Off Voltage
●
●
tBGR
Bottom Gate Rise Time
tBGF
Bottom Gate Fall Time
The ● denotes specifications which apply over the full operating
temperature range.
Note 1: Absolute maximum ratings are those values beyond which the life
of a device may be impaired.
AN73-8
11.0
0.4
0.7
0.1
V
V
11.9
0.4
12.0
0.7
V
V
130
200
ns
60
140
ns
11.9
0.4
12.0
0.7
V
V
●
70
200
ns
●
60
140
ns
11.0
Note 2: Supply current specification does not include external FET gate
charge currents. Actual supply currents will be higher and vary with
operating frequency, operating voltages and the type of external FETs
used. See Application Information section.
Note 3: Test condition: RCT = 16.9k, CCT = 1000pF.
Note 4: Test Condition: VCMSENSE = 10V.
Application Note 73
U W
TYPICAL PERFORMANCE CHARACTERISTICS
Boost Supply Current vs
Temperature
12VIN Supply Current vs
Temperature
5V REFERENCE SHORT-CIRCUIT CURRENT (mA)
18
17
3.5
I12VIN SUPPLY CURRENT (mA)
BOOST SUPPLY CURRENT (mA)
4.0
3.0
2.5
2.0
1.5
16
15
14
13
12
11
1.0
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
10
–50 –25
125
0
25
50
75
125
55
50
45
40
35
30
–50 –25
180
1.251
160
150
125
5V Reference Voltage vs
Temperature
5.01
5V REFERENCE VOLTAGE (V)
1.252
REFERENCE VOLTAGE (V)
190
100
1339 G03
Reference Voltage vs
Temperature
170
50
25
75
0
TEMPERATURE (°C)
1339 G02
I12VIN Shutdown Current vs
Temperature
I12VIN SHUTDOWN CURRENT (µA)
100
60
TEMPERATURE (°C)
1339 G01
1.250
1.249
1.248
5.00
4.99
1.247
140
130
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
1.246
–50 –25
50
25
75
0
TEMPERATURE (°C)
4.0
3.5
3.0
2.5
2.0
1.5
100
125
1339 G07
50
25
75
0
TEMPERATURE (°C)
2.6
2.4
2.2
2.0
1.8
1.6
1.4
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
1339 G08
100
125
1339 G06
Error Amplifier Maximum Source
Current vs Temperature
ERROR AMPLIFIER SOURCE CURRENT (µA)
Ω
4.5
50
25
75
0
TEMPERATURE (°C)
4.98
–50 –25
125
Error Amplifier Transconductance
vs Temperature
ERROR AMPLIFIER TRANSCONDUCTANCE (m )
Error Amplifier Voltage Gain vs
Temperature
1.0
–50 –25
100
1339 G05
1339 G04
ERROR AMPLIFIER VOLTAGE GAIN (kV/V)
5V Reference Short-Circuit
Current vs Temperature
350
325
300
275
250
225
200
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
1339 G09
AN73-9
Application Note 73
U W
TYPICAL PERFORMANCE CHARACTERISTICS
Soft Start Charge Current
vs Temperature
RUN/SHDN Rising Threshold
vs Temperature
8
7
6
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
26
RUN/SHDN THRESHOLD HYSTERESIS (mV)
1.26
RUN/SHDN RISING THRESHOLD (V)
SOFT START CHARGE CURRENT (µA)
9
RUN/SHDN Threshold Hysteresis
vs Temperature
1.25
1.24
1.23
1.22
1.21
1.20
–50 –25
125
50
25
75
0
TEMPERATURE (°C)
100
1339 G10
60
FALL TIME
40
20
0
1000
5000
7500
2500
BOTTOM GATE CAPACITANCE (pF)
10000
140
UPPER LIMIT
200
RISE TIME
150
100
FALL TIME
130
TYPICAL
120
LOWER LIMIT
110
100
50
90
80
0
1000
5000
7500
2500
TOP GATE CAPACITANCE (pF)
10000
0
1
2
3
4
5
VSENSE(CM) (V)
1339 G14
18
fO = 100kHz
TA = 25°C
CBG = 10000pF
24
22
CBG = 4700pF
20
18
CBG = 3300pF
16
CBG = 1000pF
fO = 100kHz
TA = 25°C
16
CTG = 10000pF
14
12
10
CTG = 4700pF
8
CTG = 3300pF
6
CTG = 1000pF
4
14
2
10
12
13
14
11
12VIN SUPPLY VOLTAGE (V)
15
1339 G16
10
60
1339 G15
Boost Supply Current vs
12VIN Supply Voltage
BOOST SUPPLY CURRENT (mA )
12VIN SUPPLY CURRENT (mA )
FULL OPERATING
TEMPERATURE RANGE
150
30
26
125
1339 G12
250
12VIN Supply Current vs
Supply Voltage
28
100
160
1339 G13
AN73-10
50
25
75
0
TEMPERATURE (°C)
Average Current Limit Threshold
Sense Voltage Tolerance vs
Common Mode Voltage
VSENSE (mV)
RISE TIME
21
20
–50 –25
TA = 25°C
TOP GATE TRANSITION TIMES (ns)
BOTTOM GATE TRANSITION TIMES (ns)
80
22
125
300
TA = 25°C
100
23
Top Gate Transition Times vs
Top Gate Capacitance
160
120
24
1339 G11
Bottom Gate Transition Times vs
Bottom Gate Capacitance
140
25
12
13
14
11
12VIN SUPPLY VOLTAGE (V)
15
1339 G17
Application Note 73
U W
TYPICAL PERFORMANCE CHARACTERISTICS
10.00
1200
9.75
1100
RISING
FALLING
8.75
900
800
700
8.50
600
8.25
500
0
50
25
75
0
25
50
75
100
600
500
UPPER
LIMIT
300
200
100
0
125
1339 G20
600
TYPICAL
LOWER
LIMIT
1.0 (1.25) 1.5
2.0
0.5
RUN/SHDN INPUT VOLTAGE (V)
FULL OPERATING
TEMPERATURE
RANGE
450
TYPICAL
300
LOWER
LIMIT
150
0
2.5
UPPER
LIMIT
0
2
4
6
8
10
RUN/SHDN SUPPLY VOLTAGE (V)
12
1339 G23
1339 G22
Operating Frequency (Normalized)
vs Temperature
Maximum Duty Cycle vs RCT
100
IDISCHG = 2.75mA
80
70
60
50
IDISCHG = 2.1mA
40
30
20
FULL OPERATING
TEMPERATURE
RANGE
10
1
2
4
6
10
20
RCT (kΩ)
40 60 100
1339 G21
OPERATING FREQUENCY (NORMALIZED)
1.01
90
0
100
RUN/SHDN Input Current
vs Pin Voltage
RUN/SHDN INPUT CURRENT (µA)
FULL OPERATING
TEMPERATURE
RANGE
..................................................................
RUN/SHDN INPUT CURRENT (nA )
800
50
25
75
0
TEMPERATURE (°C)
1339 G19
RUN/SHDN Input Current
vs Pin Voltage
0
30
–50 –25
125
TEMPERATURE (°C)
1339 G18
400
45
35
TEMPERATURE (°C)
700
50
40
400
–50 –25
125
100
IB(SINK) (µA)
IB(SOURCE) (µA)
9.00
VCMSENSE = 10V
55
1000
9.25
MAXIMUM DUTY CYCLE (%)
V12VIN (V)
60
VCMSENSE = 0V
9.50
8.00
–50 –25
Sense Amplifier Input Bias
Current (Sink) vs Temperature
Sense Amplifier Input Bias
Current (Source) vs Temperature
UVLO Thresholds vs Temperature
1.00
0.99
0.98
–50 –25
50
25
75
0
TEMPERATURE (°C)
100
125
1339 G24
AN73-11
Application Note 73
U
U
U
PIN FUNCTIONS
SYNC (Pin 1): Oscillator Synchronization Pin with TTLLevel Compatible Input. Input drives internal rising edge
triggered one-shot; sync signal on/off times should be
≥1µs (10% to 90% DC at 100kHz). Does not contain
internal pull-up. Connect to SGND if not used.
5VREF (Pin 2): 5V Output Reference. Allows connection
of external loads up to 10mA DC. (Reference is not
available in shutdown.) Typically bypassed with 1µF
capacitor to SGND.
CT (Pin 3): Oscillator Timing Pin. Connect a capacitor
(CCT) to ground and a pull-up resistor (RCT) to the 5VREF
supply. Typical values are CT = 1000pF and 10k ≤ RCT
≤ 30k.
SL/ADJ (Pin 4): Slope Compensation Adjustment.
Allows increased slope compensation for certain high
duty cycle applications. Resistive loading of the pin
increases effective slope compensation. A resistor
divider from the 5VREF pin can tailor the onset of additional slope compensation to specific regions in each
switch cycle. Pin can be floated or connected to 5VREF if
no additional slope compensation is required. (See
Applications Information section for slope compensation details.)
IAVG (Pin 5): Average Current Limit Integration. Frequency response characteristic is set using the 50kΩ
output impedance and external capacitor to ground.
Averaging roll-off typically set at 1 to 2 orders of magnitude under switching frequency. (Typical capacitor value
~1000pF for fO = 100kHz.) Shorting this pin to SGND will
disable the average current limit function.
SS (Pin 6): Soft Start. Generates ramping threshold for
regulator current limit during start-up and after UVLO
event by sourcing about 8µA into an external capacitor.
VREF (Pin 10): Bandgap Generated Voltage Reference
Decoupling. Connect a capacitor to signal ground. (Typical capacitor value ~0.1µF.)
SENSE + (Pin 11): Current Sense Amplifier Inverting
Input. Connect to most positive (DC) terminal of current
sense resistor.
SENSE – (Pin 12): Current Sense Amplifier Noninverting
Input. Connect to most negative (DC) terminal of current
sense resistor.
RUN/SHDN (Pin 13): Precision Referenced Shutdown.
Can be used as logic level input for shutdown control or
as an analog monitor for input supply undervoltage
protection, etc. IC is enabled when RUN/SHDN pin rising
edge exceeds 1.25V. About 25mV of hysteresis helps
assure stable mode switching. All internal functions are
disabled in shutdown mode. If this function is not
desired, connect RUN/SHDN to 12VIN (typically through
a 100k resistor). See Applications Information section.
PHASE (Pin 14): Output Driver Phase Control. If Pin 14
is not connected (floating), the topside driver operates
the main switch, with the bottom side driver operating
the synchronous switch. Shorting Pin 14 to ground
reverses the roles of the output drivers. PHASE is typically shorted to ground for inverting and boost configurations. Positive buck configuration requires the PHASE
pin to float. See Applications Information section.
PGND (Pin 15): Power Ground. References the bottom
side output switch and internal driver control circuits.
Connect with low impedance trace to VIN decoupling
capacitor negative (ground) terminal.
BG (Pin 16): Bottom Side Output Driver. Connects to gate
of bottom side external power FET.
12VIN (Pin 17): 12V Power Supply Input. Bypass with at
least 1µF to PGND.
VC (Pin 7): Error Amplifier Output. RC load creates
dominant compensation in power supply regulation feedback loop to provide optimum transient response. (See
Applications Information section for compensation details.)
TS (Pin 18): Boost Output Driver Reference. Typically
connects to source of topside external power FET and
inductive switch node.
SGND (Pin 8): Small-Signal Ground. Connect to negative
terminal of COUT.
TG (Pin 19): Topside (Boost) Output Driver. Connects to
gate of topside external power FET.
VFB (Pin 9): Error Amplifier Inverting Input. Used as
voltage feedback input node for regulator loop. Pin
sources about 0.5µA DC bias current to protect from an
open feedback path condition.
VBOOST (Pin 20): Topside Power Supply. Bootstrapped
via 1µF capacitor tied to switch node (Pin 18) and
Schottky diode connected to the 12VIN supply.
AN73-12
Application Note 73
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FUNCTIONAL BLOCK DIAGRA
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VIN
VBOOST
MAIN
SWITCH
CT
PHASE
12VIN
5VREF
TG
TS
NONOVERLAPPING
SWITCH LOGIC
BG
Q
SYNC
SWITCH
S
UVLO
CIRCUIT
R
OSC
SL/ADJ
ONE SHOT
SENSE +
RSENSE
VOUT
+
× 15
SENSE –
+
SYNC
+
–
IC1
CURRENT
SENSE AMP
0.5µA
–
–
VFB
VC
EA
+
VREF
1.25V
5VREF
2.5V
5V
–
8µA
REFERENCE
50k
SOFT START
+
+
RUN/SHDN
AVERAGE
CURRENT
LIMIT
CIRCUIT
ENABLE
1.25V
–
SGND
PGND
SS
IAVG
1339 • BD
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OPERATION
(Refer to Functional Block Diagram)
Basic Control Loop
The LT1339 uses a constant frequency, current mode
synchronous architecture. The timing of the IC is provided
through an internal oscillator circuit, which can be synchronized to an external clock, programmable to operate
at frequencies up to 150kHz. The oscillator creates a
modified sawtooth wave at its timing node (CT) with a slow
charge, rapid discharge characteristic.
During typical positive buck operation, the main switch
MOSFET is enabled at the start of each oscillator cycle. The
main switch stays enabled until the current through the
switched inductor, sensed via the voltage across a series
sense resistor (RSENSE), is sufficient to trip the current
comparator (IC1) and, in turn, reset the RS latch. When the
RS latch resets, the main switch is disabled, and the
synchronous switch MOSFET is enabled. Shoot-through
prevention logic prohibits enabling of the synchronous
switch until the main switch is fully disabled. If the current
comparator threshold is not obtained throughout the
entire oscillator charge period, the RS latch is bypassed
and the main switch is disabled during the oscillator
discharge time. This “minimum off time” assures adequate charging of the bootstrap supply, protects the main
switch, and is typically about 1µs.
AN73-13
Application Note 73
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OPERATION (Refer to Functional Block Diagram)
The current comparator trip threshold is set on the VC pin,
which is the output of a transconductance amplifier, or
error amplifier (EA). The error amplifier integrates the
difference between a feedback voltage (on the VFB pin)
and an internal bandgap generated reference voltage of
1.25V, forming a signal that represents required load
current. If the supplied current is insufficient for a given
load, the output will droop, thus reducing the feedback
voltage. The error amplifier forces current out of the VC
pin, increasing the current comparator threshold. Thus,
the circuit will servo until the provided current is equal to
the required load and the average output voltage is at the
value programmed by the feedback resistors.
Average Current Limit
The output of the sense amplifier is monitored by a single
pole integrator comprised of an external capacitor on the
IAVG pin and an internal impedance of approximately
50kΩ. If this averaged value signal exceeds a level corresponding to 120mV across the external sense resistor, the
current comparator threshold is clamped and cannot
continue to rise in response to the error amplifier. Thus, if
average load current requirements exceed 120mV/RSENSE,
the supply will current limit and the output voltage will fall
out of regulation. The average current limit circuit monitors the sense amplifier output without slope compensation or ripple current contributions, therefore the average
load current limit threshold is unaffected by duty cycle.
Undervoltage Lockout
The LT1339 employs an undervoltage lockout circuit
(UVLO) that monitors the 12V supply rail. This circuit
disables the output drive capability of the LT1339 if
the 12V supply drops below about 9V. Unstable mode
switching is prevented through 350mV of UVLO threshold
hysteresis.
Adaptive Nonoverlapping Output Stage
The FET driver output stage implements adaptive
nonoverlapping control. This circuitry maintains dead
time independent of the type, size or operating conditions
of the switch elements. The control circuit monitors the
AN73-14
output gate drive signals, insuring that the switch gate
(being disabled) is fully discharged before enabling the
other switch driver.
Shutdown
The LT1339 can be put into low current shutdown mode
by pulling the RUN/SHDN pin low, disabling all circuit
functions. The shutdown threshold is a bandgap referred
voltage of 1.25V typical. Use of a precision threshold on
the shutdown circuit enables use of this pin for undervoltage protection of the VIN supply and/or power supply
sequencing.
Soft Start
The LT1339 incorporates a soft start function that operates by slowly increasing the internal current limit. This
limit is controlled by clamping the VC node to a low voltage
that climbs with time as an external capacitor on the SS pin
is charged with about 8µA. This forces a graceful climb of
output current capability, and thus a graceful increase in
output voltage until steady-state regulation is achieved.
The soft start timing capacitor is clamped to ground
during shutdown and during undervoltage lockout, yielding a graceful output recovery from either condition.
5V Internal Reference
Power for the oscillator timing elements and most other
internal LT1339 circuits is derived from an internal 5V
reference, accessible at the 5VREF pin. This supply pin can be
loaded with up to 10mA DC (20mA pulsed) for convenient
biasing of local elements such as control logic, etc.
Slope Compensation
For duty cycles greater than 50%, slope compensation is
required to prevent current mode duty cycle instability in
the regulator control loop. The LT1339 employs internal
slope compensation that is adequate for most applications. However, if additional slope compensation is
desired, it is available through the SL/ADJ pin. Excessive
slope compensation will cause reduction in maximum
load current capability and therefore is not desirable.
Application Note 73
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APPLICATIONS INFORMATION
RSENSE generates a voltage that is proportional to the
inductor current for use by the LT1339 current sense
amplifier. The value of RSENSE is based on the required
load current. The average current limit function has a
typical threshold of 120mV/RSENSE, or:
RSENSE = 120mV/ILIMIT
Operation with VSENSE common mode voltage below 4.5V
may slightly degrade current limit accuracy. See Average
Current Limit Threshold Tolerance vs Common Mode
Voltage curve in the Typical Performance Characteristics
section for more information.
Output Voltage Programming
Output voltage is programmed through a resistor feedback network to VFB (Pin 9) on the LT1339. This pin is the
inverting input of the error amplifier, which is internally
referenced to 1.25V. The divider is ratioed to provide
1.25V at the VFB pin when the output is at its desired value.
The output voltage is thus set following the relation:
VOUT = 1.25(1 + R2/R1)
when an external resistor divider is connected to the
output as shown in Figure 1.
VOUT
R2
LT1339 VFB
SGND
9
R1
8
1339 • F01
Figure 1. Programming LT1339 Output Voltage
If high value feedback resistors are used, the input bias
current of the VFB pin (1µA maximum) could cause a slight
increase in output voltage. A Thevenin resistance at the
VFB pin of <5k is recommended.
the minimum off-time of the PWM controller. This limits
maximum duty cycle (DCMAX) to:
DCMAX = 1 – (tDISCH)(fO)
This relation corresponds to the minimum value of the
timing resistor (RCT), which can be determined according
to the following relation (RCT vs DCMAX graph appears in
the Typical Performance Characteristics section):
RCT(MIN) ≈ [(0.8)(10 –3)(1 – DCMAX)] –1
Values for RCT > 15k yield maximum duty cycles above
90%. Given a timing resistor value, the value of the timing
capacitor (CCT) can then be determined for desired operating frequency (fO) using the relation:
(1/ fO ) − (100) 10−9 
CCT ≈
(RCT / 1.85) +  −3 1.75
(2.5) 10  − (3.375 / RCT )
A plot of Operating Frequency vs RCT and CCT is shown in
Figure 2. Typical 100kHz operational values are CCT =
1000pF and RCT = 16.9k.
160
OSCILLATOR FREQUENCY (kHz)
RSENSE Selection for Output Current
140
CCT = 1.0nF
120
CCT = 1.5nF
100
80
60
CCT = 3.3nF
40
CCT = 2.2nF
20
0
0
5
10
20
25
15
TIMING RESISTOR (kΩ)
30
LT1339 • F02
Figure 2. Oscillator Frequency vs RCT, CCT
Oscillator Components RCT and CCT
Average Current Limit
The LT1339 oscillator creates a modified sawtooth wave
at its timing node (CT) with a slow charge, rapid discharge
characteristic. The rapid discharge time corresponds to
The average current limit function is implemented using
an external capacitor (CAVG) connected from IAVG to SGND
that forms a single pole integrator with the 50kΩ output
AN73-15
Application Note 73
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APPLICATIONS INFORMATION
impedance of the IAVG pin. The integrator corner frequency is typically set 1 to 2 orders of magnitude below the
oscillator frequency and follows the relation:
f–3dB = (3.2)(10– 6)/CAVG
The average current limit function can be disabled by
shorting the IAVG pin directly to SGND.
Soft Start Programming
The current control pin (VC) limits sensed inductor current
to zero at voltages less than a transistor VBE, to full average
current limit at VC = VBE + 1.8V. This generates a 1.8V full
regulation range for average load current. An internal
voltage clamp forces the VC pin to a VBE – 100mV above
the SS pin voltage. This 100mV “dead zone” assures 0%
duty cycle operation at the start of the soft start cycle, or
when the soft start pin is pulled to ground. Given the
typical soft start current of 8µA and a soft start timing
capacitor CSS, the start-up delay time to full available
average current will be:
tSS = (1.5)(105)(CSS)
ing capabilities of that supply, causing the system to lock
up in an undervoltage state. Input supply start-up protection can be achieved by enabling the RUN/SHDN pin using
a resistor divider from the input supply to ground. Setting
the divider output to 1.25V when that supply is almost fully
enabled prevents the LT1339 regulator from drawing large
currents until the input supply is able to provide the
required power.
If additional hysteresis is desired for the enable function,
an external feedback resistor can be used from the LT1339
regulator output. If connection to the regulator output is
not desired, the 5VREF internal supply pin can be used.
Figure 3 shows a resistor connection on a 48V to 5V
converter that yields a 40V VIN start-up threshold for
regulator enable and also provides about 10% input
referred hysteresis.
VIN
48V
300k
390k
Boost Supply
The VBOOST supply is bootstrapped via an external capacitor. This supply provides gate drive to the topside switch
FET. The bootstrap capacitor is charged from 12VIN through
a diode when the switch node is pulled low.
The diode reverse breakdown voltage must be greater than
VIN + 12VIN. The bootstrap capacitor should be at least 100
times greater than the total input capacitance of the
topside FET. A capacitor in the range of 0.1µF to 1µF is
generally adequate for most applications.
Shutdown Function — Input Undervoltage Detect and
Threshold Hysteresis
The LT1339 RUN/SHDN pin uses a bandgap generated
reference threshold of about 1.25V. This precision threshold allows use of the RUN/SHDN pin for both logic-level
shutdown applications and analog monitoring applications such as power supply sequencing.
Because an LT1339 controlled converter is a power transfer device, a voltage that is lower than expected on the
input supply could require currents that exceed the sourc-
AN73-16
VOUT
5V
OPTION 1
OPTION 2
2
13
10k
5VREF
LT1339
RUN/SHDN
1339 • F03
Figure 3. Input Supply Sequencing Programming
The shutdown function can be disabled by connecting the
RUN/SHDN pin to the 12VIN rail. This pin is internally
clamped to 2.5V through a 20k series input resistance and
will therefore draw about 0.5mA when tied directly to 12V.
This additional current can be minimized by making the
connection through an external resistor (100k is typically
used).
Inductor Selection
The inductor for an LT1339 converter is selected based on
output power, operating frequency and efficiency requirements. Generally, the selection of inductor value can be
reduced to desired maximum ripple current in the inductor
(∆I). For a buck converter, the minimum inductor value for
a desired maximum operating ripple current can be determined using the following relation:
Application Note 73
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APPLICATIONS INFORMATION
(VOUT )(VIN − VOUT)
L MIN =
(∆I)(fO)(VIN)
where fO = operating frequency. Given an inductor value
(L), the peak inductor current is the sum of the average
inductor current (IAVG)and half the inductor ripple current
(∆I), or:
(VOUT )(VIN − VOUT)
IPK = IAVG +
(2)(L)(fO)(VIN)
The inductor core type is determined by peak current and
efficiency requirements. The inductor core must withstand peak current without saturating, and series winding
resistance and core losses should be kept as small as is
practical to maximize conversion efficiency.
The LT1339 peak current limit threshold is 40% greater
than the average current limit threshold. Slope compensation effects reduce this margin as duty cycle increases.
This margin must be maintained to prevent peak current
limit from corrupting the programmed value for average
current limit. Programming the peak ripple current to less
than 15% of the desired average current limit value will
assure porper operation of the average current limit
feature through 90% duty cycle (see Slope Compensation
section).
Oscillator Synchronization
The LT1339 oscillator generates a modified sawtooth
waveform at the CT pin between low and high thresholds
of about 0.8V (vl) and 2.5V (vh) respectively. The oscillator
can be synchronized by driving a TTL level pulse into the
SYNC pin. This inputs to a one-shot circuit that reduces the
oscillator high threshold to 2V for about 200ns. The SYNC
input signal should have minimum high/low times of
≥1µs.
Slope Compensation
Current mode switching regulators that operate with a
duty cycle greater than 50% and have continuous inductor
current can exhibit duty cycle instability. While a regulator
will not be damaged and may even continue to function
SYNC
2.5V
(vh)
2V
VCT
(vl)
0.8V
FREE RUN
SYNCHRONIZED
1339 F04
Figure 4. Free Run and Synchronized Oscillator
Waveforms (at CT Pin)
acceptably during this type of subharmonic oscillation, an
irritating high-pitched squeal is usually produced.
The criterion for current mode duty cycle instability is met
when the increasing slope of the inductor ripple current is
less than the decreasing slope, which is the case at duty
cycles greater than 50%. This condition is illustrated in
Figure 5a. The inductor ripple current starts at I1, at the
beginning of each oscillator switch cycle. Current
increases at a rate S1 until the current reaches the control
trip level I2. The controller servo loop then disables the
main switch (and enables the synchronous switch) and
inductor current begins to decrease at a rate S2. If the
current switch point (I 2) is perturbed slightly and
increased by ∆I, the cycle time ends such that the minimum current point is increased by a factor of (1 + S2/S1)
to start the next cycle. On each successive cycle, this error
is multiplied by a factor of S2/S1. Therefore, if S2/S1 is
≥ 1, the system is unstable.
Subharmonic oscillations can be eliminated by augmenting the increasing ripple current slope (S1) in the control
loop. This is accomplished by adding an artificial ramp on
the inductor current waveform internal to the IC (with a
slope SX) as shown in Figure 5b. If the sum of the slopes
S1 + SX is greater than S2, the condition for subharmonic
oscillation no longer exists.
For a buck converter, the required additional current
waveform slope, or “Slope Compensation,” follows the
relation:
V 
SX ≥  IN  2DC − 1
 L 
(
)
AN73-17
Application Note 73
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APPLICATIONS INFORMATION
T1
S1 + SX
I2
S1
I1
0
S2
S1
S2
OSCILLATOR
PERIOD
0
TIME
a
b
1339 • F05
Figure 5. Inductor Current at DC > 50% and
Slope Compensation Adjusted Signal
For duty cycles less than 50% (DC < 0.5), SX is negative
and is not required. For duty cycles greater than 50%, SX
takes on values dependent on S1 and duty cycle. This leads
to a minimum inductance requirement for a given VIN and
duty cycle of:
V 
L MIN =  IN  2DC− 1
 SX 
(
)
The LT1339 contains an internal SX slope compensation
ramp that has an equivalent current referred value of:
 fO 
0.084

 RSENSE 
SXADD =
(2500)( fO )
(REQ )(RSENSE )
Amp/s
where REQ is the effective resistance of the resistor divider.
Actual compensation will be somewhat greater due to
internal curvature correction circuitry that imposes an
exponential increase in the slope compensation waveform, further increasing the effective compensation slope
up to 20% for a given setting.
1.45
Amp/s
where fO is oscillator frequency. This yields a minimum
inductance requirement of:
(VIN)(RSENSE)(2DC− 1)
L MIN ≥
(0.084)(fO)
A down side of slope compensation is that, since the IC
servo loop senses an increase in perceived inductor current, the internal current limit functions are affected such
that the maximum current capability of a regulator is
reduced by the same amount as the effective current
referred slope compensation. The LT1339, however, uses
a current limit scheme that is independent of slope compensation effects (average current limit). This provides
operation at any duty cycle with no reduction in current
sourcing capability, provided ripple current peak amplitude is less than 15% of the current limit value. For
example, if the supply is set up to current limit at 10A, as
long as the peak inductor current is less than 11.5A, duty
cycles up to 90% can be achieved without compromising
the average current limit value.
AN73-18
If an inductor smaller than the minimum required for
internal slope compensation (calculated above as LMIN) is
desired, additional slope compensation is required. The
LT1339 provides this capability through the SL/ADJ pin.
This feature is implemented by referencing this pin via a
resistor divider from the 5VREF pin to ground. The additional slope compensation will be affected at the point in
the oscillator waveform (at pin CT) corresponding to the
voltage set by the resistor divider. Additional slope compensation can be calculated using the relation:
1.40
1.35
PEAK/AVG
∆I
1.30
1.25
1.20
1.15
1.10
0
0.1
0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9
DUTY CYCLE (DC)
LT1339 • F06
Figure 6. Maximum Ripple Current (Normalized)
vs Duty Cycle for Average Current Limit
Design Example:
VIN = 20V
VOUT = 15V (DC = 0.75)
RSENSE = 0.01Ω
fO = 100kHz
L = 5µH
The minimum inductor usable with no additional slope
compensation is:
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(20V)(0.01Ω)(1.5 − 1) = 11.9µH
LMIN ≥
(0.084)(100000)
Since L = 5µH is less than LMIN, additional slope compensation is necessary. The total slope compensation
required is:
 20V 
SX ≥ 
1.5 − 1 = 2  106 

 
 5µH
(
) ()
Amp/s
Subtracting the internally generated slope compensation
and solving for the required effective resistance at SL/ADJ
yields:
REQ ≤
(2500)(fO)
= 21.5k
6

(2)10  (RSENSE) − (0.084)(fO)
Setting the resistor divider reference voltage at 2V assures
that the additional compensation waveform will be
enabled at 75% duty cycle. As shown in Figure 7a, using
2
RSL1
45k
5VREF
SL/ADJ
RSL2
30k
Power MOSFET and Catch Diode Selection
External N-channel MOSFET switches are used with the
LT1339. The positive gate-source drive voltage of the
LT1339 for both switches is roughly equivalent to the
12VIN supply voltage, so standard threshold MOSFETs
can be used.
Selection criteria for the power MOSFETs include the “ON”
resistance (RDS(ON)), reverse transfer capacitance (CRSS),
maximum drain-source voltage (VDSS) and maximum
output current.
The power FETs selected must have a maximum operating
VDSS exceeding the maximum VIN. VGS voltage maximum
must exceed the 12VIN supply voltage.
Once voltage requirements have been determined, RDS(ON)
can be selected based on allowable power dissipation and
required output current.
In an LT1339 buck converter, the average inductor current
is equal to the DC load current. The average currents
through the main and synchronous switches are:
LT1339
4
RSL1 = 45k and RSL2 = 30k sets the desired reference
voltage and has a REQ of 18k, which meets both design
requirements. Figure 7b shows the slope compensation
effective waveforms both with and without the SL/ADJ
external resistors.
1339 • F07a
Figure 7a. External Slope Compensation Resistors
2.5V
IMAIN = (ILOAD)(DC)
ISYNC = (ILOAD)(1 – DC)
The RDS(ON) required for a given conduction loss can be
calculated using the relation:
2V
PLOSS = (ISWITCH)2(RDS(ON))
0.8V
DC = 0.75
(0.084 + 0.139)(fO)
RSENSE
(0.084)(fO)
RSENSE
In high voltage applications (VIN > 20V), the topside switch
is required to slew very large voltages. As VIN increases,
transition losses increase through a square relation, until
it becomes the dominant power loss term in the main
switch. This transition loss takes the form:
PTR ≈ (k)(VIN)2(IMAX)(CRSS)(fO)
where k is a constant inversely related to the gate drive
current, approximated by k = 2 in LT1339 applications.
1339 • F07b
Figure 7b. Slope Compensation Waveforms
AN73-19
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The maximum power loss terms for the switches are thus:
PMAIN = (DC)(IMAX)2(1 + δ)(RDS(ON)) +
2(VIN)2(IMAX)(CRSS)(fO)
PSYNC = (1 – DC)(IMAX)2(1 + δ)(RDS(ON))
The (1 + δ) term in the above relations is the temperature
dependency of RDS(ON), typically given in the form of a
normalized RDS(ON) vs Temperature curve in a MOSFET
data sheet.
In some applications, parasitic FET capacitances couple
the negative going switch node transient onto the bottom
gate drive pin of the LT1339, causing a negative voltage in
excess of the Absolute Maximum Rating to be imposed on
that pin. Connection of a catch Schottky (rated to about 1A
is typically sufficient) from this pin to ground will eliminate
this effect.
CIN and COUT Supply Decoupling Capacitor Selection
The large currents typical of LT1339 applications require
special consideration for the converter input and output
supply decoupling capacitors. Under normal steady state
operation, the source current of the main switch MOSFET
is a square wave of duty cycle VOUT/VIN. Most of this
current is provided by the input bypass capacitor. To
prevent large input voltage transients and avoid bypass
capacitor heating, a low ESR input capacitor sized for the
maximum RMS current must be used. This maximum
capacitor RMS current follows the relation:
(IMAX )(VOUT (VIN – VOUT ))
1/ 2
IRMS ≈
VIN
which peaks at a 50% duty cycle, when IRMS = IMAX/2.
Capacitor ripple current ratings are often based on only
2000 hours (three months) lifetime; it is advisable to
derate either the ESR or temperature rating of the capacitor for increased MTBF of the regulator.
The output capacitor in a buck converter generally has
much less ripple current than the input capacitor. Peak-topeak ripple current is equal to that in the inductor (∆IL),
typically a fraction of the load current. COUT is selected to
reduce output voltage ripple to a desirable value given an
expected output ripple current. Output ripple (∆VOUT) is
approximated by:
AN73-20
∆VOUT ≈ ∆IL{ESR + [(4)(fO) • COUT]–1}
where fO = operating frequency.
Efficiency Considerations and Heat Dissipation
High output power applications have inherent concerns
regarding power dissipation in converter components.
Although high efficiencies are achieved using the LT1339,
the power dissipated in the converter climbs to relatively
high values when the load draws large amounts of power.
Even at 90% efficiency, an application that provides 500W
to the load has conversion loss of 55W.
I2R dissipation through the switches, sense resistor and
inductor series resistance create substantial losses under
high currents. Generally, the dominant I2R loss is evident
in the FET switches. Loss in each switch is proportional to
the conduction time of that switch. For example, in a 48V
to 5V converter the synchronous FET conducts load current for almost 90% of the cycle time and thus, requires
greater consideration for dissipating I2R power.
Gate charge/discharge current creates additional current
drain on the 12V supply. If powered from a high voltage
input through a linear regulator, the losses in that regulator device can become significant. A supply solution
bootstrapped from the output would draw current from a
lower voltage source and reduce this loss component.
Transition losses are significant in the topside switch FET
when high VIN voltages are used. Transition losses can be
estimated as:
PTLOSS ≈ 2(VIN)2(IMAX)(CRSS)(fO)
Since the conduction time in the main switch of a 48V to
5V converter is small, the I2R loss in the main switch FET
is also small. However, since the FET gate must switch up
past the 48V input voltage, transition loss can become a
significant factor. In such a case, it is often prudent to take
the increased I2R loss of a smaller FET in order to reduce
CRSS and thus, the associated transition losses.
Gate Drive Buffers
The LT1339 is designed to drive relatively large capacitive
loads. However, in certain applications, efficiency improvements can be realized by adding an external buffer
stage to drive the gates of the FET switches. When the
Application Note 73
U
W
U
U
APPLICATIONS INFORMATION
switch gates load the driver outputs such that rise/fall
times exceed about 100ns, buffers can sometimes result
in efficiency gains. Buffers also reduce the effect of back
injection into the bottom side driver output due to coupling
of switch node transitions through the switch FET CMILLER.
Paying the Physicists
In high power synchronous buck configurations, certain
physical characteristics of the external MOSFET switches
can impact conversion efficiency. As the input voltage
approaches about 30V, the bottom MOSFETs will begin to
exhibit “phantom turn-on.” This phenomenon is caused
by coupling of the instantaneous voltage step on the
bottom side switch drain through CMILLER to the device
gate, yielding internal localized gate-source voltages above
the turn-on threshold of the FET. This generates a shootthrough blip that ultimately eats away at efficiency numbers. In Figure 8 a negative prebias circuit is added to the
bottom side gate. The addition of this ∼3V of negative
offset to the bottom gate drive provides additional offstate voltage range to prevent phantom turn-on.
TS
3.3V
12VIN
ZTX649
1µF
LT1339
BG
ZTX749
10k
D1N914
PGND
1339 F08
Figure 8. Bottom Side Driver Negative Prebias Circuit
not available for high voltages, so as input voltage continues to increase, they can no longer be used. Because this
necessitates the use of discrete FETs and Schottkys,
interdigitation of a number of smaller devices is required
to minimize parasitic inductances. This technique is also
used in the 48V to 5V, 50A converter shown in the Typical
Applications section.
This type of prebias circuit is used in the 48V to 5V, 50A
converter pictured in the Typical Applications section.
Optimizing Transient Response—Compensation
Component Values
As currents increase beyond the 10A to 15A range, the
bottom side FET body diode experiences hard turn-on
during switch dead time due to local current loop inductance preventing the timely transfer of charge to the
Schottky catch diode. The charge current required to
commutate this body diode creates a high dV/dt Schottky
avalanche when the diode charge is finally exhausted (due
to an effective inductor current discontinuity at the
moment the body diode no longer requires charge). This
generates an increased turn-on power burst in the topside
switch, causing additional conversion efficiency loss. This
effect of this parasitic inductance can be reduced by using
FETKEY TM MOSFETs, which have parallel catch Schottky
diodes internal to their packages. FETKEY MOSFETs are
The dominant compensation point for an LT1339 converter is the VC pin (Pin 7), or error amplifier output. This
pin is connected to a series RC network, RVC and CVC. The
infinite permutations of input/output filtering, capacitor
ESR, input voltage, load current, etc. make for an empirical
method of optimizing loop response for a specific set of
conditions.
Loop response can be observed by injecting a step change
in load current. This can be achieved by using a switchable
load. With the load switching, the transient response of the
output voltage can be observed with an oscilloscope.
Iterating through RC combinations will yield optimized
response. Refer to LTC Application Note 19 in 1990 Linear
Applications Handbook, Volume 1 for more information.
FETKEY is a trademark of International Rectifier Corporation.
AN73-21
AN73-22
C11
0.1µF
+
+
C12
100pF
+
C10
0.1µF
C9
1800pF
5%
NPO
+
+
C14
3300pF
R9
12k
C1: SANYO 63MV680GX
C2: WIMA SMD4036/1.5/63/20/TR
C6: KEMET T510X477M006AS (X8)
L1: GOWANDA 50-318
T1: GOWANDA 50-319
+
R5
2.49k
1%
+
+
10
7
6
5
3
4
2
1
17
C15
0.1µF
8
SGND
VREF
VC
SS
IAVG
CT
11
18
19
20
15
PGND
12
SENSE –
16
BG
14
PHASE
13
RUN/SHDN
9
VFB
SENSE +
TS
TG
5VREF
SL/ADJ
VBOOST
SYNC
D2
MURS120
C2
1.5µF
63V
12VIN
+
U1
LT1339
C5
1µF
C1
680µF
63V
R6, 100Ω
D5
BAT54
R10
10k
1%
R8
301k
1%
R7
100Ω
4
3
2
1
4
3
2
1
VCC1
OUT1
VCC2
GND1
IN2
OUT2
VCC1
IN1
U3, LTC1693-2
GND2
5
6
7
8
5
6
7
8
C7
1µF
VCC2
OUT2
GND2
OUT1
IN2
GND1
IN1
U2, LTC1693-2
+
+
C13
1µF
+
D4
MBR0530T1
C8
1µF
D3
MURS120
Q1
MTD20N06HD
R1
0.04Ω
Q3
MTD20N06HD
13:2
T1
D1
MURS120
3 2 1
8 7 6 5
R2
5.1Ω
+
Q4
Si4420
X2
4
C3
4700pF
25V
3 2 1
8 7 6 5
L1
1.5µH
+
Q2
Si4420
X2
4
C6
470µF
6.3V
X8
1339 TA05
+
C4
0.1µF
R4
1.24k
1%
R3
549Ω
1%
VOUT
1,8V
20A
TYPICAL APPLICATIONS
12V
VIN
48V
48V to 1.8V 2-Transistor Synchronous Forward Converter
Application Note 73
U
C1
1.2µF
100V
CER
68µF
20V
AVX
TSPE
10k
P
+
100k
0.1µF
P
3.9k
GND1
IN1
PHASE
JP3
W2
T1
2
W3
2
7
5
6
18
1
RUN/SHDN
12VIN
20
2.2µF
19
OUT1
VCC1
470Ω
OUT2
IN2
VCC2
LTC1693-1
GND2
JP2
100k
14
13
17
1
8
3
4
12V
5VOUT SHORT JP3, OPEN JP2
3.3VOUT, SHORT JP2, OPEN JP3
BAS21
BAS21
BAS21
13k
MMBD914LT1
C2
1.2µF
100V
CER
COILCRAFT
DO1608-105
36k
+VIN
–VIN
INPUT
36V TO
75V
+VIN
+VIN
BAT54
10Ω
5
10
P
W4, 7T 6 x 26AWG
W5, 10T 2 x 26AWG
W1, 10T 32AWG,
W2, 15T 32AWG
W1, 10T 2 x 26AWG
T2
T2
T1
8 15
W4
W4
4.7nF
7
VFB
BG
4.7k
4.7k
9
2MIL
POLY
FILM
2MIL
POLY
FILM
2.4k
1µF
BAT54
+
OUT1
IN1
T2
P
GND1
OUT2
IN2
GND2
VCC2
LTC1693-1
VCC1
CNY17-3
4
1
3
8
SUD30N04-10
W1
470Ω
BAT54
1nF
C3
330µF
6.3V
85
90
95
100k
470Ω
2
7
5
6
+
C5
330µF
6.3V
0
1
1
2
8
3 4 5 6 7
OUTPUT CURRENT
48VIN
36VIN
8
9
10
4.42k
1%
–VOUT
9.31k
1%
72VIN
5 7
LT1431CS8
REF
BAS21
10Ω
SEC HV
1339 TA06
SHORT JP1
FOR 5VOUT
0.01µF
1k
0.47µF
50V
3.01k
1%
+VOUT
MMFT3904
6
COLL
–VOUT
OUTPUT
5V/10A
+VOUT
2k
3.1V
2 4
0.22µF
1µF
4.7µF
25V
1k
–VOUT
FZT600
+
+VOUT
3
C3, C4, C5:
SANYO OS-CON
C4
330µF
6.3V
4.8µH
PANASONIC ETQP AF4R8H
10Ω
470Ω
16 3.3Ω
T1 PHILIPS EFD20-3F3 CORE
LP = 720µH (AI = 1800)
T2 ER11/5 CORE
AI = 960µH
6
10Ω
SEC HV
SUD30N04-10
1nF
4.7nF
4.7nF
W3
LT1339
W5
W1
0.1µF
W3, 10T 32AWG,
W4, 10T 32AWG
2.2nF
2.2nF
4
12
0.025Ω
1/2W
W1, 18T BIFILAR 31AWG
W3, 6T BIFILAR 31AWG
1µF
4.53k
3
11
10Ω
P
IRF1310NS
MURS120
FMMT718
FMMT718
TS
2.2µF
SGND
470Ω
SENSE +
CT
W2
SL/ADJ
T2
PGND
47Ω
SS
MMBD914LT1
SENSE –
IAVG
VBOOST
SYNC
TG
5VREF
MURS120
VREF
IRF1310NS
VC
10Ω
EFFICIENCY
0.1µF
V+
GND-F
RTOP
+VIN
COMP
GND-S
RMID
48V to 5V Isolated Synchronous Forward DC/DC Converter
Application Note 73
TYPICAL APPLICATIONS
AN73-23
U
Application Note 73
U
TYPICAL APPLICATIONS
5V to 28V DC/DC Synchronous Boost Converter Limits Input Current at 60A (DC)
12V
+
DBST
MBR0530
SYNC
VBOOST
5VREF
+ C5VREF
Q2
FMMT720
TG
TS
12VIN
SL/ADJ
CCT
CAVG
2200pF 2200pF
12L
CVC, 1500pF
VFB
RR1
100k
PHASE
RUN/SHDN
SENSE –
VREF
RFB2, 1.2k
D2
MBR0520
L1
40µH
PGND
SGND
CREF, 0.1µF
Q4
FMMT720
BG
VC
RVC, 7.5k
RFB1, 27k
IRF3205
×4
1µF
SS
+
D1
IR30BQ060
×8
Q3
FMMT619
+C
IAVG LT1339
CSS, 10µF
IRF3205
×2
1µF
CT
1µF
C12VIN
47µF
Q1
FMMT619
+ CBST
RCT
10k
VOUT
28V
COUT
2200µF
35V
×6
+
RSS1
100Ω
RS
0.002Ω
RSS2, 100Ω
SENSE +
CIN
2200µF
6.3V
×4
L1 = 12T 4X12 ON 77439-A7
VIN
5V AT 60A
+
1339 TA04
U
PACKAGE DESCRIPTION
Dimensions in inches (millimeters) unless otherwise noted.
N Package
20-Lead PDIP (Narrow 0.300)
(LTC DWG # 05-08-1510)
0.130 ± 0.005
(3.302 ± 0.127)
0.300 – 0.325
(7.620 – 8.255)
0.020
(0.508)
MIN
0.009 – 0.015
(0.229 – 0.381)
(
+0.035
0.325 –0.015
+0.889
8.255
–0.381
1.040*
(26.416)
MAX
0.045 – 0.065
(1.143 – 1.651)
)
0.065
(1.651)
TYP
0.125
(3.175)
MIN
0.005
(0.127)
MIN
0.100 ± 0.010
(2.540 ± 0.254)
20
19
18
17
16
15
14
13
12
11
1
2
3
4
5
6
7
8
9
10
0.255 ± 0.015*
(6.477 ± 0.381)
0.018 ± 0.003
(0.457 ± 0.076)
N20 1197
*THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED 0.010 INCH (0.254mm)
SW Package
20-Lead Plastic Small Outline (Wide 0.300)
(LTC DWG # 05-08-1620)
0.291 – 0.299**
(7.391 – 7.595)
0.010 – 0.029 × 45°
(0.254 – 0.737)
0.093 – 0.104
(2.362 – 2.642)
0.496 – 0.512*
(12.598 – 13.005)
0.037 – 0.045
(0.940 – 1.143)
20
19
18
17
16
15
14
13
12
11
0° – 8° TYP
0.009 – 0.013
(0.229 – 0.330)
NOTE 1
0.016 – 0.050
(0.406 – 1.270)
0.050
(1.270)
TYP
0.014 – 0.019
(0.356 – 0.482)
TYP
NOTE:
1. PIN 1 IDENT, NOTCH ON TOP AND CAVITIES ON THE BOTTOM OF PACKAGES ARE THE MANUFACTURING OPTIONS.
THE PART MAY BE SUPPLIED WITH OR WITHOUT ANY OF THE OPTIONS
*DIMENSION DOES NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED 0.006" (0.152mm) PER SIDE
**DIMENSION DOES NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED 0.010" (0.254mm) PER SIDE
AN73-24
0.394 – 0.419
(10.007 – 10.643)
NOTE 1
0.004 – 0.012
(0.102 – 0.305)
1
2
3
4
5
6
7
8
9
10
S20 (WIDE) 0396
Application Note 73
U
TYPICAL APPLICATION
48V to 5V 50A DC/DC Converter with Input Supply Start-Up Protection
12V
RCT
10k
5VREF
VBOOST
VIN
48V
+
DBST
IN5819
LT1339
SYNC
50mA
C12VIN
47µF
Q1
+ CBST
CT
+ CCT
TG
CAVG, 2200pF
CSS, 10µF
CVC, 2200pF
D3
MMSZ4684
12VIN
CBG, 1µF
IAVG
SS
PGND
RFB2
1k
RFB1
3k
RR1
22k
RR3
51k
PHASE
SGND
RUN/SHDN
VFB
VREF
D2
MBR0520
Q4
RVC, 4.7k
CREF
0.1µF
D1
Q3
BG
VC
IRFZ44
×2
Q2
SL/ADJ
2200pF
+
1µF
TS
C5VREF
1µF
CIN
1500µF
63V, × 6
RBG
10k
D4
IN914
IRFZ44
×4
L1
40µH
RR2
1.2k
RS
0.002Ω
SENSE –
SENSE +
D1 = IR30BQ060 × 8
Q1, Q3 = FMMT619; Q2, Q4 = FMMT720
L1 = Kool Mµ®, 12T 4X12 ON 77439-A7
Kool Mµ IS A REGISTERED TRADEMARK OF MAGNETICS, INC.
COUT
2200µF
6.3V, × 4
+
VOUT
5V AT 50A
1339 TA01
48V to 5V Efficiency
100
95
EFFICIENCY (%)
90
85
80
75
70
65
60
55
50
0
10
30
40
20
OUTPUT CURRENT (AMPS)
50
LT1339 • TA02
RELATED PARTS
PART NUMBER
DESCRIPTION
COMMENTS
LT1158
Half-Bridge N-Channel MOSFET Driver
Current Limit Protection, 100% of Duty Cycle
LT1160
Half-Bridge N-Channel MOSFET Driver
Up to 60V Input Supply, No Shoot-Through
LT1162
Dual Half-Bridge N-Channel MOSFET Driver
VIN to 60V, Good for Full-Bridge Applications
LT1336
Half-Bridge N-Channel MOSFET Driver
Smooth Operation at High Duty Cycle (95% to 100%)
LTC ® 1530
High Power Step-Down Switching Regulator Controller
Excellent for 5V to 3.xV Up to 50A
LTC1435A
High Efficiency, Low Noise Current Mode Step-Down DC/DC Converter
Drives Synchronous N-Channel MOSFETs
LTC1438
Dual High Efficiency, Low Noise Synchronous Step-Down Controller
Tight 1% Reference
LT1680
High Power DC/DC Current Mode Step-Up Controller
High Side Current Sense, Up to 60V Input
AN73-25
Application Note 73
EXPANDED PIN DESCRIPTIONS
SYNC (PIN 1) OSCILLATOR SYNCHRONIZATION PIN
12V or 15V logic, a resistor divider is recommended, as
shown in Figure 4.
12V OR 15V
LOGIC
This pin allows the user to synchronize the LT1339 to an
external clock.
The synchronized frequency must be faster than the freerunning frequency of the LT1339. This logic-level input
can be driven from TTL or 3.3V or 5V CMOS. Its threshold
is set at 2.2VBE, about 1.5V, and it exhibits no hysteresis.
Its duty factor can be almost anything, but you must
ensure that the pulse is at least 500ns wide, whether
asserted high or low. The internal synchronization event is
triggered on the rising edge of the SYNC pulse; this event
triggers an internal 200ns one-shot. While the output of
the internal one-shot is asserted, the high level trip point
of the CT pin is reduced from 2.5V to 2V (see Figure 3).
SYNC
2.5V
(vh)
2V
15k
OR
20k 1
LT1339
SYNC
10k
SGND
8
AN73 F04
Figure 4. Synchronizing from Higher Voltage Logic Families
Pulling the SYNC pin above 5VREF by a diode drop forward
biases an internal diode which begins to source current
into 5VREF, which has no provision to sink current. This is
not recommended.
SYNCHRONIZING MULTIPLE LT1339s
There are two basic schemes to synchronize multiple
power converters: master/slave synchronization and
multiphase synchronization.
Master/Slave Synchronization
VCT
(vl)
0.8V
FREE RUN
SYNCHRONIZED
AN73 F03
Figure 3. Free Run and Synchronized Oscillator
Waveforms (at Pin CT)
If, during the 200ns at the output of the internal one-shot,
the ramp voltage on the CT pin is above 2V, an early reset
is initiated. Normal reset is initiated at 2.5V. The functions
performed by the reset event are:
1. The CT pin (Pin 3) is discharged by the LT1339 to 0.8V
2. The beginning of a new on cycle ensues (the top gate
goes high in buck mode or the bottom gate goes high
in boost mode, depending on the logic level of the
PHASE pin).
The SYNC pin is used to pull the oscillator frequency up
from the free-running frequency set by RCT and CCT, so the
free-running frequency should be set low enough (20%
lower than the desired running frequency) to ensure
synchronization. If unused, this pin should be tied either to
SGND or to 5VREF, the latter having a penalty of an
additional 1mA of quiescent operating current. Leaving
this pin floating is bad form. If this pin is to be driven from
AN73-26
In master/slave synchronization, one LT1339 (the master)
is set to free-run at the frequency desired for the whole
system and the other LT1339(s) are slaved from the
master. Figure 5 details the master/slave connection. The
free-running frequencies of the slaves are set 20% lower
than the free-running frequency of the master. In such a
system, if the master LT1339 stops switching (shuts
down) the slaves free-run at their natural frequency until
the master resumes switching.
#1
BG
16
SGND
8
20k
#2
1
SYNC
SGND
8
10k
#3
1
SYNC
SGND
8
AN73 F05
Figure 5. Master/Slave Synchronization
Application Note 73
Multiphase Synchronization
Multiphase synchronization is very useful in systems
where high ripple current mandates massive input
capacitors in a buck or forward regulator, or massive
output capacitors in a boost regulator. Refer to Figure 6.
The master oscillator (U1) is set to run at n times the
desired running frequency, where n is the number of
phases desired. Select the subcircuit of U2 that matches
the number of phases you desire (or make more phases by
looking up the 4017 data sheet and choosing additional
[up to ten] phases). Simply connect the sync outputs of
Figure 6 to the SYNC pins of the respective LT1339s. Set
up the LT1339s to free-run 20% slower than the synchronizing frequency, fOSC/n.
5V
16
14
VDD
0
U2
4017
1
>I
5V
13
4
7
GND R
8
2
SYNC 1
SYNC 2
2θ
fOSC/2
4
15
D
7
VDD
U1
LMC555
THRESHOLD OUT
3
14
U2
4017
0
1
2
3
GND
>I
1
13
GND R
8
2. The synchronous transistor (bottom FET of buck, top
FET of boost) is on.
3. A 2.5mA current source is discharging the CT pin.
The LT1339 will stay in this state as long as the CT pin is
held above 0.8V.
When the CT pin is released, the internal discharge current
(2.5mA) will pull it down to 0.8V, the internal flip-flop is
reset and the next on cycle begins:
1. The synchronous transistor is turned off
3. The 2.5mA discharge current is switched off.
5V
16
NC
1. The main transistor (top FET of buck, bottom FET of
boost) is off.
2. The main transistor is turned on.
8
R VCC
TRIG
2
2
3
be done by pulling the CT pin above 2.5V; this sets an
internal flip-flop, which, while set, keeps the LT1339 in the
deadtime phase of operation. The deadtime phase is
maintained until the CT pin is brought down to 0.8V.
During the dead-time phase, the LT1339 enters and
remains in the following state:
3
2
4
SYNC 1
SYNC 2
3θ
fOSC/3
SYNC 3
7
15
RC
Refer to Figure 7. This synchronization scheme has the
advantage of allowing synchronization over the full operating range of the LT1339, but has the disadvantage of
disabling the slope compensation. Due to the internal
structures of the LT1339, the CT pin should not be pulled
above 3.5V; doing so may seriously confuse the internal
logic.
5V
16
C
VDD
fOSC =
1
(1.4)(RC)(C)
0
1
14
U2
4017
2
3
4
>I
13
GND R
8
3
2
4
7
2
SYNC 1
SYNC 2
SYNC 3
4θ
fOSC/4
SYNC 4
10
THIS EXTERNAL
SYNCHRONIZING WAVEFORM
WIDTH SETS
MINIMUM OFF TIME
5VREF
2N3904
LT1339
3
5V
CT
0V
SGND
15
8
AN73 F07
AN73 F06
Figure 7. Wide-Range Synchronization Using the CT Pin
Figure 6. Mulitphase Synchronization
Wide-Range Synchronization
The LT1339 oscillator can be synchronized over a wider
frequency range by acting directly on the CT pin. This can
5VREF (PIN 2) OUTPUT REFERENCE
This is your reference, the reference to power your
external logic, amplifiers and the supply to tailor the
inputs of the LT1339 to your needs—it is 5V ±250mV
over line, load and temperature. You can draw up to 10mA
AN73-27
Application Note 73
to limit maximum duty factor to a number less than 90%,
a better way to do so is described in the SL/ADJ (Pin 4) pin
description.
Figure 10 shows the Oscillator Frequency vs RCT and CCT .
160
OSCILLATOR FREQUENCY (kHz)
from this pin. Sourcing current into this pin is not recommended because it has no pull-down capability other than
the normal operating current of the logic of the LT1339. It
is used as the reference for the oscillator section through
RT. To set up hysteresis for the RUN pin, connect an
appropriate resistor from the RUN pin to the 5VREF pin.
Finally, the 5VREF pin is used with the SL/ADJ pin to set the
maximum duty factor or additional slope compensation
when needed. Internally, the 5VREF pin is used to power
practically all internal functions, with only the RUN/SHDN
comparator and both gate drive circuits powered from
other sources. This pin should be decoupled to ground
(SGND) with a 1µF capacitor having an ESR less than 10Ω.
The decoupling capacitor can be anything from 0.1µF to
many thousands of microfarads.
140
CCT = 1.0nF
120
CCT = 1.5nF
100
80
60
CCT = 3.3nF
40
CCT = 2.2nF
20
0
0
CT (PIN 3) OSCILLATOR TIMING PIN
5
10
20
25
15
TIMING RESISTOR (kΩ)
30
AN73 F10
The free-running (nonsynchronized) frequency of the
LT1339 is set up by the RCT and CCT combination connected to this pin (see Figure 8). The value of the resistor
RCT sets up the minimum off-time of the LT1339. Refer to
Figure 9 for duty factor verses RCT in kΩ. If you would like
2
1µF
LT1339
3
SGND
8
2.5V
AN73 F08
0.8V
Figure 8. Oscillator Pin Connection + Waveform
100
MAXIMUM DUTY CYCLE (%)
70
60
50
IDISCHG = 2.1mA
30
20
10
0
1
2
4
6
10
20
RCT (kΩ)
40 60 100
AN73 F09
Figure 9. Duty Factor vs RCT
AN73-28
Figure 11 details how to use the SL/ADJ pin to limit
maximum duty factor. Limiting the maximum duty factor
effectively limits the voltage input-to-output ratio where
the converter can transform power. If you think of the
converter as a DC variac, limiting the duty factor limits how
far the variac can be turned. What function limiting duty
factor performs depends upon the topology used:
1. In a synchronous buck converter, decreasing the maximum duty factor increases the dropout voltage. This
could be used to set the minimum input voltage at
which a given output voltage can be reached. It also
limits the load-step response time when at the lowest
input voltage.
IDISCHG = 2.75mA
80
40
Limiting Maximum Duty Factor
CT
CCT
90
SL/ADJ (PIN 4) SLOPE COMPENSATION
ADJUSTMENT PIN
This pin is used to limit maximum duty factor and/or
introduce additional slope compensation to the LT1339.
5VREF
RCT
Figure 10. Oscillator Frequency vs RCT, CCT
2. In a synchronous boost converter, decreasing the maximum duty factor decreases the maximum available
output voltage at a given input voltage. This could be
used as an overvoltage protection default. Limiting duty
factor can also affect the load step response time when
operating at the minimum input voltage.
Application Note 73
3. In a forward converter, decreasing the maximum duty
factor is highly desirable because it prevents transformer core saturation, resulting in a robust design.
4. In all topologies, limiting the maximum duty factor can
be used to prevent operation at duty factors at which the
circuit would experience current mode instability, but
this function is better performed by setting the UV
lockout voltage appropriately.
1. By varying VIN, measure the minimum duty factor at
which current mode instability occurs. If the duty factor
is 50% or less, you are not looking at current mode
instability; check loop stability.
Setting Maximum Duty Factor
100
5VREF
90
MAXIMUM DUTY FACTOR
If your system exhibits current mode instability, you will
need to add slope compensation or increase the inductance of the main inductor. For the procedure to calculate
the correct value for the main inductor, refer to the section
entitled “Boost,” or “Buck”, according to which topology
you have, and look up the “Inductor Selection” subsection.
To add slope compensation to your existing design, use
this procedure:
1k
80
SL/ADJ
70
60
R2
2. Construct the subcircuit shown in Figure 12 and connect it to your switcher.
50
5VREF
40
ADD SLOPE COMPENSATION
30
20
POT2
20k
2.49k
10
SW1
POT1
1k
0
100
500
R2 (Ω)
SL/ADJ
1000
AN73 F11
1.5k
+
DVM
SET
THEVENIN
VOLTAGE
–
Figure 11. Using the Slope Comp Pin to Limit Duty Factor
AN73 F12
Adding More Slope Compensation
At duty factors greater than 50%, an instability enters the
current mode switcher world: current mode instability.
This is seen as a two-cycle sequence: first, a cycle with a
long on-time and short off-time, then a cycle with a short
on-time and a long off-time. Current mode instability
always happens at exactly half of the switching frequency
and is completely independent of other forms of
subharmonic oscillation. Current mode instability is independent of the control loop and can be observed when a
voltage source is substituted for the control loop. Current
mode instability occurs whenever the rising main inductor
current ramp is slower than the falling main inductor
current ramp. To make a design that can run at duty factors
approaching unity, slope compensation is a must. The
LT1339 has built-in slope compensation that is adequate
for most buck converter designs, but if you have a boost
converter running at very high duty factors, you may need
more slope compensation.
Figure 12. Slope Compensation Adjustment Setup Circuit
3. Power up and switch SW1 to the “set Thevenin voltage”
position.
4. Select a Thevenin voltage from Table 1 that corresponds to a duty factor that is 10% less than you found
in step 1.
Table 1. Thevinin Voltages for Slope Compensation
DF
VTHEVININ
90%
80%
70%
60%
50%
2.37
2.23
2.09
1.922
1.744
5. Adjust POT1 for that Thevenin voltage, as read on the
DVM.
6. Adjust POT2 for maximum resistance and set SW1 for
“Add slope compensation.”
AN73-29
Application Note 73
7. Decrease the resistance of POT2 to correct the current
mode instability at the minimum input voltage and
maximum load current allowed.
converter is stable in steady-state operation, it will also
be free from current mode instability during load-stepinduced duty factor variations.
8. Calculate from the equations in Figure 13 the resistor
values needed to construct the slope-compensation
network shown in Figure 14.
1
RTHEV =

 

1
1

+

 (VTHEV / 5V)(4k )   (5V – VTHEV ) /(5V)(4k ) 
IAVE (PIN 5) AVERAGE CURRENT LOOP
INTEGRATION CAPACITOR
(
)( )
RSELECT = RPOT 2 0.8
(Margin)
RTOTAL = RTHEV + RSELECT
RTOP =
(RTOTAL )(5)
VTHEV
RBOTTOM =
(RTOTAL )(5)
5 – VTHEV
Figure 13. Calculating the Resistor Values for
a 2-Resistor Slope Compensation Network
2
RTOP
LT1339
3
RBOTTOM
5VREF
SL/ADJ
SGND
8
This pin is used to smooth out the triangular wave shape
of the current waveform and, in the process, to set up the
loop compensation of the current limit loop. In the buck or
boost converter, the current waveform in the inductor of
the power converter is roughly triangular, the average of
which represents the DC output current of the buck
converter (DC input current for a boost converter). This
main inductor current is converted into a voltage by the
current sense resistor. The voltage across the sense
resistor is applied to the SENSE – and SENSE + pins of the
LT1339. Inside, it is amplified by a gain of 15 and is then
offset by a VBE so that 700mV represents zero programmed peak inductor current. This current signal representing inductor current is then given an impedance of
50k and brought out to the IAVE pin. Placing a capacitor
from the IAVE pin to the VC pin (Pin 7), sets up an
integration network that smooths out the triangular ripple,
yielding a DC voltage that represents the average current.
This DC voltage is subtracted from 2.5V and converted to
a current by a transconductance amplifier. If the resultant
current is negative, it is subtracted from the error amplifier’s
output current. The transconductance of the conversion is
0.03A/V. Once the current reaches its threshold, the
average current limit loop comes to life, taking control of
the main control loop. The power converter changes from
a constant voltage source to a constant current source. If
the average current limit feature is not used, this pin
should be grounded.
AN73 F14
Figure 14. Two Resistor Slope Compensation
For the new design, refer to the “Inductor Selection”
subsection of the section describing the topology you are
designing.
The slope compensation in your system need only be
enough to handle the steady-state operation at minimum
VIN with maximum VOUT and maximum load current (UV
lockout can be set to limit minimum input voltage and
IRSENSE can be set to limit maximum current). If the
AN73-30
Using the IAVE Pin to Implement
an Adjustable Current Limit.
Refer to Figure 15. This circuit is a system-level application using the LT1339 where there is a motherboard that
uses one-half of the available power and two daughter
boards that, when present, each use one-quarter of the
total available power. Using this circuit, one can protect
the motherboard and daughterboards with current limits
set appropriately for the actual load.
Application Note 73
5VREF
LT1339
SENSE –
LOAD
CURRENT
200k
SENSE +
× 15
200k
CR2
50k
IAVE
CR3
–
VC
TO CARD
SLOTS
2.5V
+
CARDS GROUND THESE PINS
WHEN PRESENT, INCREASING
CURRENT LIMIT
330pF
EA
AN73 F15
Figure 15. Multiple Level Automatic Current Limit
SS (PIN 6) SOFT-START INTEGRATION CAPACITOR
This pin, with its associated external capacitor, generates
a ramp that limits the maximum voltage possible on the VC
pin. The VC pin is clamped by a PNP transistor to be 1VBE
(~ 0.7V) maximum above the SS pin. This allows the
current output of the power converter to start at zero and
rise to the operating load current at a rate slower than the
loop response. Soft-start ramps minimize or completely
eliminate overshoot at turn-on. This pin is actively pulled
to ground when either the 12VIN pin voltage drops below
the UV lockout point, disabling operation (see the section
on the 12VIN pin for details), or the RUN/SHDN pin is below
its low voltage trip point (see the section on the RUN/
SHDN pin for details). When the voltages on both the
12VIN and the RUN/SHDN pins enable operation, approximately 8µA is sourced out of this pin, charging CSS, the
capacitor connected between ground and this pin. This
current can vary from 4µA to 20µA. For tighter chargecurrent tolerance, an external resistor is recommended
between the 5VREF pin and the SS pin, setting up a much
higher (100µA) charge current. The voltage on this pin is
internally clamped at 3.5V and any resistor pulling up on
this pin should be sized to limit the current to 150µA.
When the LT1339 pulls this pin low, the discharge current
is 10mA. This discharge current becomes available
whenever the voltage at the 12V IN pin is greater than 3V.
When the soft-start pin is fully discharged by the internal
discharge transistor, the peak current programmed is
zero. Grounding the SS pin will stop the top FET drive from
turning on. The bottom FET drive will be on for the
maximum time as set up by the running frequency and
dead-time. Using the internal charge current, the time to
full current is nominally Time (s) = 0.8 • 10 5C. If this pin
is unused, it should be left floating. If your design requires
that the VREF5 output be active while the LT1339 output is
shut down, use the circuit detailed in Figure 16.
LT1339
6
+
H = SHUTDOWN
VN2222
CSS
SS
SGND
8
AN73 F16
NOTE: GROUNDING THE SOFT-START PIN WILL
CAUSE THE OUTPUT TO BE DISCHARGED TO
OR BELOW GROUND
Figure 16. Shutdown While Keeping 5VREF Circuit Alive
AN73-31
Application Note 73
VC (PIN 7) CONTROL LOOP COMPENSATION NODE
Current Sharing Multiple Power Converters
This pin comprises the output of the error amplifier and the
input of the peak current comparator. Normally it is used
to compensate the output voltage control loop. Compensation is performed by adding a pole and a zero to the
control loop function; sometimes an additional pole is
needed—this is provided by CFINAL. These two poles and
one zero are configured by connecting this pin in the
manner indicated in Figure 17. The total voltage loop is
detailed in Figure 18. If your design requires gating on and
off, use the RUN/SHDN pin as your first choice. If, however, you need to keep 5VREF alive during the shutdown
period, the VC pin can be pulled down by an NPN transistor
or MOSFET, as shown in Figure 19. This allows the softstart function to work independently of the gating signal.
If soft-start is desired on each turn-on of your gating
sequence, refer to Figure 16 in the soft-start pin description. When using optoelectronic feedback in a forward
converter, ground the feedback pin and connect the optoisolator to the VC pin according to Figure 20.
As with the synchronization of multiple LT1339 power
converters, current sharing can be achieved by two different strategies: master/slave current sharing and peer-level
sharing controlled by a system master control loop.
AV = 3000
VC
(
1. Master/slave current sharing is performed by the circuit
detailed in Figure 21.
2. Operating peer-level power converters with a master
system loop is detailed in Figure 22.
NC
6
SS
LT1339
7
VC
SGND
2N3904 OR
VN2222
8
AN73 F19
Figure 19. Gating the Power Converter
On and Off Without Soft Start
)
RREF
RFB + RREF
POLE
7
CFINAL
LT1339
F=
RCOMP
CCOMP
SGND
AV = 3000
8
1
(2π)(CCOMP)(1.5)(106)
F=
(
)( )
VC
1
(2π)(RCOMP)(CFINAL)
LT1339
RCOMP
RREF
RFB + RREF 1.5E6
VFB
ZERO
1
F=
(2π)(RCOMP)(CCOMP)
9
OPTO
SGND
8
AN73 F17
AN73 F20
Figure 17. Control Loop Compensation
RFB
7
Figure 20. Optical Feedback Scheme
VFB
–
RREF
gm =
1.5M
–
AV = 3k
15
RSENSE
+
AV =
RREF
RFB + RREF
1.25V
RCOMP
CCOMP
+
CFINAL
1
(2π)(CO)(ESR)
1
FP =
(2π)(CO)(RL)
FZ =
0.8V
CO
RL
ESR
AN73 F18
Figure 18. Total Voltage Loop
AN73-32
Application Note 73
2
7
1k
2N3904
7
2N3906
LT1339
MASTER
VC
VFB
RFB
9
VOUT
SGND
9
8
7
R1 =
VC
LT1339
SLAVE 1
VFB
SGND
RREF
8
R1
VFB (PIN 9) FEEDBACK PIN FOR
OUTPUT VOLTAGE CONTROL
5VREF
1k
NUMBER OF SLAVES
VC
LT1339
SLAVE 2
VFB
9
SGND
This pin is the inverting input to the error amplifier. The
input voltage at the VFB pin is subtracted from the voltage
on the VREF pin and converted into an error current by the
error amplifier’s transconductance of 0.002A/V. The
LT1339, using its loop gain and current-controlled engine,
is constantly forcing this pin to 1.250V. Thus, the output
voltage is set by a resistor divider (See Figure 23) connected to this pin.
8
AN73 F21
VOUT
Figure 21. Master/Slave Current Sharing
R2
LT1339 VFB
VOUT SENSE
CC
RFB
5V
3
RREF
8
REF
V + COMP
1
4
RTOP
COLL
LT1431
7
RMID
2.5V
GNDS
SGND
R1
8
2
RC
GNDF
5
9
VOUT = 1.25 +
(R2)(1.25)
R1 + R2
AN73 F23
Figure 23. Programming LT1339’s Output Voltage
6
Error Amplifier Characteristics
9
7
9
7
9
VC
LT1339
VFB
VC
LT1339
VFB
VC
LT1339
VFB
AN73 F22
Figure 22. Peer Level Current Sharing
SGND (PIN 8) SIGNAL GROUND
This is the “clean” ground pin; it is for the returns of CCT,
the feedback resistor divider, CCOMP, C5VREF, CIAVE, CVREF
and CSOFTSTART. This pin also is the reference for the
RUN/SHDN pin. This pin is electrically connected to the
PGND inside the LT1339; the resistance between these
pins is approximately 10Ω. On the PCB there should be a
trace connecting this pin to the PGND pin. This connecting
trace should be situated so that no switching frequency AC
or power level DC current flows through it. Placing the
LT1339 on a ground plane with PGND and SGND connected to the plane works well.
There is a DC voltage that, when placed on the feedback pin
(VFB ), will cause the current output of the error amplifier
to be zero. This voltage falls between 1.242V and 1.258V
and is called VNULL. Figure 24 details the DC characteristics of the error amplifier. Note the generous width of the
linear region; this is important for noise immunity in high
power systems. It is evident that a 200mVP-P high frequency disturbance on the feedback pin will not take the
error amplifier out of its linear region, and will be integrated in the compensation network. This is the essence
of removing the nasty “R word” (rectification) from the
gm = 0
275
ERROR AMP OUTPUT CURRENT (µA)
7
0
gm = 2000
– 275
gm = 130
0.3
VNULL – 0.14
VNULL
FEEDBACK VOLTAGE (V)
VNULL + 0.14
AN73 F24
Figure 24. Error Amplifier Transconductance
AN73-33
Application Note 73
arena of problems encountered with poorly designed
parts. There is, however, a limit to the ability of the error
amplifier to reject rectification when looking across the
DC-to-daylight spectrum. Rectification happens when the
dV/dT on the feedback pin approaches 50V/µs. The mechanism of this rectification is seen in Figure 25, as Q1 cuts off
when the feedback pin’s dV/dT goes positive faster than I1
can charge C1. If the feedback pin slews at rates approaching this limit, we recommend that you add a small capacitor between the feedback pin and the VREF pin or SGND pin
to slow down the signals at the VFB pin. The feedback pin
has a maximum voltage and current restriction. If you pull
the feedback pin above 5V, you will turn on a parasitic
vertical PNP transistor and inject current into the substrate. Under such conditions the current into the feedback pin should be limited to less than 1mA.
I1
25µA
I2
25µA
BANDGAP
Q2
VFB
Q1
C1
0.5pF
5k
Q3
≈200mV
Q4
VREF
C2
0.5pF
2
6
2N3904
1N914
10k
+
4.3k
2N3906
VREF5
VOUT
SS
LT1339
10
0.1µF
VREF
VOUT
2
SGND
8
10µF
AN73 F26
Figure 26. Voltage Soft Start
Using the VREF Pin to Implement a Noise-Immune
Output Voltage Trim
Although the output voltage can be easily trimmed by
adding a resistor to the VFB pin, the poor noise immunity
of this trim method is less than desirable. A much better
voltage trim is accomplished by connecting an appropriate valued resistor to the VREF pin and adjusting the value
of the decoupling capacitor for a time constant in the tens
to hundreds of milliseconds (refer to Figure 27). The time
constant of noise reduction on the VREF pin has no effect
on loop stability, unlike networks used on the VFB pin. This
function is also useful in applications requiring remote
sense.
AN73 F25
Figure 25. Error Amplifier Input Stage
TRIM
VOLTAGE
0V TO 2.5V
R1
10
VREF
C
LT1339
VREF (PIN 10) ERROR AMPLIFIER VOLTAGE
REFERENCE DECOUPLING
The VREF pin is connected directly to the noninverting
input of the error amplifier. This pin is normally decoupled
with a 0.1µF cap connected from this pin to SGND.
Internally, there is a 5k isolation resistor between this pin
and the bandgap reference. The dynamic output impedance of the bandgap reference is approximately 100Ω.
Setting Up a Voltage Soft-Start Using the VREF Pin
The error amplifier that is connected to this pin has an
input common mode range that extends below 0.6V. This
pin can be used for a voltage soft-start circuit. The circuit
shown in Figure 26 utilizes the error amplifier to program
a soft-start that starts at 50% of nominal output voltage
and ramps up to 100% of the nominal output voltage.
AN73-34
SGND
8
AN73 F27
TRIM RANGE WITH 0V TO 2.5V INPUT =
TIME CONSTANT (tNR) =
( )(
)
VOUT
5k
1.25 R1 + 5k
(5k)(R1)
C(5K + R1)
Figure 27. The Right Way to Trim Output Voltage
SENSE+ (PIN 11) SENSE AMPLIFIER
NONINVERTING INPUT
This pin, in conjunction with the SENSE – pin, forms the
input to a wide common mode range differential amplifier.
This amplifier takes its differential input voltage, adds an
intentional 5mV offset, multiplies the result by 15 and adds
it to a voltage that is one VBE (~ 0.7V) above ground. This
Application Note 73
provides an internal voltage representing the current flowing in the external current sense resistor (RS). This voltage
is used for current mode operation and in the averagecurrent-limiting loop. The input common mode range
extends from – 0.3V to 60V. This common mode range
spans two ranges of operation, – 0.3V to (5V – 1VBE) and
(5V – 1VBE) to 60V. We will investigate these two ranges
of operation because the input characteristics of this
amplifier change somewhat when going from one range to
the other. The input structure of the current sense amplifier is shown in Figure 28.
5VREF
D1
SENSE +
R2
3k
R1
3k
R3
3k
Q3
R4
3k
SENSE –
See the Pin 11 definition.
RUN/SHDN (PIN 13) PRECISION REFERENCED
SHUTDOWN CONTROL PIN
This pin implements the user-programmed undervoltage
(UV) lockout. By placing a resistor divider from VIN at this
pin to ground, any UV lockout voltage from 1.25V through
hundreds of volts can be programmed. Internal hysteresis
of this pin is internally set at 15mV (1.2% of the 1.25V
switching point). Additional hysteresis can be implemented by adding a resistor from this pin to the 5VREF pin
(see Figure 29a).
SENSE AMP
OUT
Q4
Q5
45µA
This pin complements the function of the SENSE + pin
(Pin 12).
Figures 29b and 29c detail digital shutdown with and
without UV lockout.
Q2
45µA
SENSE – (PIN 12) SENSE AMPLIFIER
INVERTING INPUT
R5
45k
AN73 F28
Figure 28. Current Sense Amplifier Equivalent Circuit
0.1µF
VIN
R1
13
R2
SELECT R2 (10k)
RUN/SHDN
R1 = R2
R3
LT1339
2
SENSE –
When the
and
pins are above
(5VREF – 1VBE) D1 is not conducting and the input bias
current is 45µA (the emitter current of the PNPs). When
the voltage on SENSE + goes positive with respect to the
voltage on SENSE –, current in excess of 45µA flows
through R1 and through Q4 into R5. The voltage across
R5 thus becomes a representation of the voltage between
SENSE + and SENSE – multiplied by 15. Keep in mind that
the emitter voltages of Q2 and Q3 are always the same.
This amplifier has a bandwidth of 300kHz.
VSHDN + 3VSU – 5
5
VSHDN + 3VSU – 5
VSU – VSHDN
)
)
VSU = VSTARTUP, VSHDN = VSHUTDOWN
SGND
High Common Mode Voltage
SENSE +
R3 = R2
5VREF
(
(
8
AN73 F29a
Figure 29a RUN/SHDN Pin
VIN
RUV
LOCKOUT
RHY
2
5VREF
LT1339
H = SHUTDOWN
13
VN2222
RUN/SHDN
OPTIONAL
10k
SGND
8
AN73 F29b
The Low Voltage Range
When these pins are pulled below (5VREF – 1VBE), the
direction of input bias current changes and its magnitude
depends upon how far the sense leads are pulled below
(5VREF – 1VBE). When using this amplifier in the low
voltage mode, the input bias current is high, so be sure that
any external resistors connected to these pins are matched.
Figure 29b UV Lockout with Digital Shutdown
LT1339
DIGITAL
INPUT
13
RUN/SHDN
SGND
8
AN73 F29c
Figure 29c Digital Shutdown
AN73-35
Application Note 73
PHASE (PIN 14) OUTPUT DRIVER PHASE CONTROL
This pin configures the LT1339 to function in one of its two
separate modes:
12V
17
1. Synchronous buck converter (pin left floating or tied
to 5V)
LT1339
2. Synchronous boost converter (pin grounded)
PGND
The threshold of this pin is at 1VBE above ground and it has
very good noise immunity when left floating. The input
characteristics of this pin are such that sinking current
from this pin (pulling it low) at about 40µA (approximately
15k to ground) brings the voltage down to the gray area of
its threshold. In this gray area, the LT1339 is in a state
where it is uncertain of the mode in which to operate; in
this condition, the gate-drive level is diminished and
certain failure will result.
Moral: either ground the PHASE pin, float the PHASE pin
or tie it to 5VREF.
PGND (PIN 15) POWER GROUND
This is the “substrate pin” of the LT1339; the pin relative
to which it is desirable to keep the voltages on all other pins
positive. This pin carries the return current for the bottom
gate drive current, the majority of the LT1339’s quiescent
current and the fault current generated when pins are
carelessly allowed to swing more than 1VBE below this pin.
There should be a short trace from this pin to the IC
decoupling capacitor, and this pin should be connected to
the ground plane of the circuit.
BG (PIN 16) BOTTOM-SIDE GATE DRIVE
This pin is the output of the bottom gate driver, a hefty,
fearless drive output. Due to the enormous size of the
output drive transistors, pulling this pin negative with
respect to PGND by 1VBE turns on the large NPN collectorto-substrate diode. This injects current into the substrate.
The bottom gate drive can easily push 10,000pF around,
but efficiency tests have shown that about 1% to 2% of
additional efficiency can be bought by adding a pair of
external buffer transistors (see Figure 30).
AN73-36
1µF
VIN
ZTX649
BG
16
IRFZ44N
×4
1A
ZTX749
15
*EFFICIENCY IMPROVES WHEN
JUMPER IS REMOVED
JP1*
AN73 F30
Figure 30. External Drivers Improve Efficiency
NULLIFYING THE PHANTOM TURN-ON
About 200ns prior to the top gate turning on, the bottom
gate turns off. During that time (the underlap time), the
main inductor current flows through the body diodes of
the bottom FETs. By the time the topside FET turns on,
significant stored charge is packed away in the body
diode(s) of the bottom MOSFET(s). Upon turning on, the
top MOSFET “sees” a short—the unrecovered body
diodes of the bottom MOSFETs. Large currents start to
flow as the stored charge in the body diode is swept out.
Upon releasing the last of its charge, the bottom body
diode snaps off (switch opens in a few nanoseconds). At
this time the drains of the bottom MOSFETs are pulled
positive by tens of volts in a few nanoseconds. Referring
to Figure 31, it can be seen that the bottom MOSFET has
phantom RC networks that act as little voltage dividers
(VG(STEP)S) with time constants τGs. When the dV/dT and
∆V are high enough, the internal VGS reaches one threshold voltage and bad things happen (efficiency drops).
Figure 32 details the removal of the phantom turn-on
mechanism by offsetting the bottom gate voltage to
– 3.3V when the bottom FET is off, effectively placing the
gate-threshold point out of reach of VG(STEP).
Application Note 73
D
VDS
1
RG
0
=
• • •
VGS
VG(STEP)
CM
V
CI + CM D(STEP)
τG = (CM + CI) RG
VG(STEP) =
CM
• • •
CI
τG
0
AN73 F31
S
Figure 31. The Phantoms Lurking Inside Your Bottom Side MOSFETs. At Some dV/dT on the Drain
with a Big Enough Step, This FET will Turn Itself On Even with the Gate Connected to the Source
The power dissipated by the LT1339, when active, is the
sum of the power drawn by the 12VIN pin and the VBOOST
pin, and is calculated below:
13.5V
17
VIN
10.2V
– 3.3V
13.5V
0V
LT1339
BG
ZTX649
PTOTAL = P12VIN + PBOOST
1µF
16
PGND
BOTTOM
FET
1A
ZTX749
3.3V
10k
15
1N4148
*EFFICIENCY IMPROVES WHEN
JUMPER IS REMOVED
JP1*
AN73 F32
Figure 32. Offsetting the Bottom Gate Drive Removes Phantom
Turn-On Losses from Higher Input Voltage Operation
12VIN (PIN 17) 12V POWER SUPPLY INPUT
Although the LT1339 is designed and optimized to run on
input voltages ranging from 10V to 15V, it can run on
voltages from UV lockout to 20V (the absolute maximum
input voltage); however, operation above 18V is not
recommended. This pin should be decoupled to the PGND
pin with short traces, low ESR and capacitor values of at
least 1µF (keep in mind that this pin is providing the
positive drive current for the bottom-side MOSFET driver).
The current flowing into this pin can be divided into three
parts: IACTIVE , IDYNAMIC and ISTANDBY
1. Active Supply Current
Active VRUN > 1.35V (1.25V typ) IACTIVE = 14mA typ,
20mA max
2. Dynamic supply current VRUN > 1.35V (1.25V typ)
IDYNAMIC = (QG (of bottom FETs) • fSW)
where:
P12VIN = (VIN)(QGB • fSW + 20mA)
QGB = the sum of all QGs of all of the bottom FETs
PBOOST = (VIN – 0.6V)(QGT • fSW + 2.2mA)
QGT = the sum of all QGs of the top FETs
TS (PIN 18) BOOST OUTPUT DRIVER REFERENCE
(TOP SOURCE)
This is the negative supply pin for the top driver. Externally
it is connected to the source of the top MOSFET. This pin
should be clamped to ground with a Schottky diode,
preventing it from going below the substrate (PGND pin).
The LT1339 is designed to be robust and will not latch up
if this pin goes to – 3V. Excursions of this pin below
substrate will cause unwanted charge to be injected into
the substrate. When injection occurs, changes can be seen
in the operation of the LT1339 in the form of a shortening
of the underlap time between the bottom gate drive and the
top gate drive. In many high power applications, it is
impossible to keep the source(s) of the top FET(s) from
going volts below PGND. Depending upon which MOSFETs
you use and upon your layout, this can become a real
problem as top switch currents rise above 50A to 100A.
The circuit detailed in Figure 33 uses a common mode
transformer to solve this problem, and should be considered for very high power converters.
3. Standby Supply Current VRUN < 1.15V (1.25V typ)
ISTANDBY = 150µA typ, 220µA max
IACTIVE includes ISTANDBY
AN73-37
Application Note 73
1A
12V
56µF
25V
+
17
1µF
VIN
CENORMOUS
10Ω
DALE 220MBP
LT1339
VBOOST
TG
TS
20
19
18
PGND
15
1µF
•
•
•
3A
SCHOTTKY
ZTX649
+
56µF
25V
2Ω
ZTX749
AN73 F33
Figure 33. Using a Common Mode Transformer to Keep Large Negative
BUS Excursions from Injecting into the Substrate Via the TS Pin
TG (PIN 19) TOPSIDE DRIVER GATE OUTPUT
(TOP GATE)
VBOOST (PIN 20) TOPSIDE DRIVER
POWER SUPPLY PIN.
This pin is the gate driver for the topside MOSFET. It
acquires its current to charge the topside gate positive
from the VBOOST pin, which is sourced by the external
boost capacitor. The negative drive current is obtained
from the TS (Top Source) pin.
This pin is the power supply pin for the top driver. It should
be decoupled to the TS (Top Source) pin with a 1µF low
ESR capacitor. This external capacitor is charged during
the time the bottom transistor is on via an external diode
from 12VIN.
AN73-38
Application Note 73
BUCK REGULATOR DESIGN
OVERVIEW
The buck converter is a converter that has an input voltage
greater than its output voltage. The simplest buck converter is the linear-pass regulator. Its input is a voltage that
varies over the regulator’s input range and its output is a
constant voltage. The power dissipated in a linear-pass
regulator is the product of the differential voltage
(VIN – VOUT) and the output current. The efficiency of a
linear-pass regulator is simply the ratio of output voltage
divided by input voltage (multiplied by 100%). In some
low differential voltage applications, the linear-pass regulator has efficiencies that rival those of switching regulators. It takes a well-designed switcher to surpass the
efficiency of a linear-pass regulator in a 3.3V input/2.5V
output application.
Switching buck converters have current multiplication, in
that the output current exceeds the input current. In an
ideal buck converter, the ratio of output current to input
current is the same as the ratio of input voltage to output
voltage (100% efficiency).
Characteristics of the
Synchronous Switching Buck Converter
The synchronous buck converter has the following characteristics:
1. It has a minimum input voltage that it needs to provide
rated output voltage. For the power converter (the
LT1339 and its associated components), this voltage
is usually limited by the maximum duty factor and the
resistive losses in the top FET, inductor and sense
resistor. In some designs (those with small inductors)
the minimum input voltage will be the voltage at which
the converter experiences current mode instability (an
instability found in clocked, current mode converters
operating at duty factors approaching unity), and
design action should be taken: either increase the
inductance, add slope compensation or use the undervoltage lockout capability of the RUN/SHDN pin to
lock out operation.
2. There is a maximum input voltage, above which something will break.
3. The input current is discontinuous, with high AC RMS
values that require large input capacitors just to
accommodate the high AC current. This also causes
high input ripple voltage; an input filter may be required to keep the switching frequency ripple from
contaminating the power source.
4. The output current is continuous and has a triangular/
sawtooth wave shape. This means small output
capacitors and low output ripple.
5. There is no forward parasitic power path; hence,
shutting down the buck converter turns off the output.
6. There is a parasitic reverse power path at the output:
if the output is pulled below ground, current flows
through the body diode of the bottom MOSFET, through
the inductor, through the sense resistor and out. Were
you to connect a car battery to the output of a synchronous buck converter backwards, current would be
limited by very small resistances. This is not recommended.
7. There is also a parasitic reverse power path at the
input: if the input is pulled below ground, the equivalent circuit simplifies to two diodes to ground. These
diodes—the body diodes of the MOSFETs—are all
that stands between your circuit and disaster. When
reverse polarity input voltage is applied, there is no
built-in mechanism to limit reverse current. The usual
result is irrepairable damage.
8. If the output voltage is pulled above the input voltage,
the body diode of the top MOSFET conducts to supply
power from the output back to the input. If you have a
car battery at the output, don’t short the input.
9. Inductor value is not extremely critical; increasing the
inductor value above the required minimum nets
small returns.
10. Efficiencies range from the low 80%s to the high
90%s.
11. The optimum switching frequency is determined by
tradeoffs in core losses in the inductor, switching
losses and gate-charge losses against smaller inductors and capacitors and faster transient response. For
modern core materials, capacitors and MOSFETs, the
frequency typically falls in the 50kHz to 125kHz
region.
AN73-39
Application Note 73
GRAPHICAL DESIGN EXAMPLE
Although the LT1339 data sheet details the procedure for
designing a buck converter, some of us like to see the
pictures to make our decisions. Here is a graphical design
guide for the LT1339 synchronous buck converter. Everything here is normalized to a 100kHz clock frequency and
1A ILIMIT. Here we will design a 12V input, 3.3V/20A output
converter. Since our maximum output current is 20A, we
will set the current limit at 24A.
above the shaded areas where things get nasty. We could
choose a higher ripple percentage and not get bitten.
Let’s look at some of the other graphs to see what effect
choosing a higher ripple ratio would have.
Now, we pick some value for the ratio of peak-to-peak
ripple current to current limit output current. Any value
between 10% and 80% will do; for this example we’ll use
30%. This results in a ripple current of 24A • 0.3, or 7.2A
Referring to Figure 35, we see that the ripple current ratio
doesn’t make any significant difference in the AC RMS
input current. Further, we see that the input capacitor AC
RMS ripple current is 0.46A for the example above (“1A”
on Figure 35); multiplying this by the rated output current
of 20A yields 9.2A; hence big input caps will be required,
regardless of inductor value. Note that if the minimum
input voltage is 7V, the input AC RMS ripple current goes
up to 10A (1B on Figure 35).
Look at Figure 34. Starting from the right, follow the 30%
isocline to the left until it intercepts the input voltage = 12V
grid line. (This is labeled “1” on Figure 34.) This yields
80µH, which is denormalized by dividing it by ILIMIT
(80µH/24 = 3.33µH). Note that this inductor value is well
What about output capacitor AC RMS current? Refer to
Figure 36 to see the interaction. Here we see that the AC
RMS current is 0.087A (“1” on Figure 36); multiplying this
by 20A yields 1.74ARMS—no problem; maybe 30% is too
low.
400
10
300
20
150
30
100
90
80
1
40
70
50
60
60
50
70
2
40
80
30
20
3.3
4
5
6
THIS REGION IS LIMITED BY
INTERNAL SLOPE
COMPENSATION. ENTRY INTO
THIS REGION WILL CAUSE
CURRENT MODE INSTABILITY
REQUIRING ADDITIONAL
SLOPE COMPENSATION
7
8
9
10
12
15
INPUT VOLTAGE (V)
20
24
30
THIS REGION IS LIMITED BY
IPEAK/IAVE. ENTRY INTO THIS
REGION WILL RESULT IN
INOPERABLE AVERAGE
CURRENT LIMIT FUNCTION.
CURRENT WILL BE LIMITED BY
THE PEAK AT A VALUE LESS
THAN ILIMIT.
Figure 34. 3.3V Output Normalized Minimum Inductor Value
AN73-40
40
50
60
AN73 F34
P-P RIPPLE/ILIMIT (%)
MINIMUM INDUCTANCE (µH)
200
Application Note 73
0.60
1B
0.50
1A
2
0.30
80
10
0.20
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.40
0.15
0.10
3.3
4
5
6
7
8
9
10
12
15
INPUT VOLTAGE (V)
20
24
30
40
50
60
AN73 F35
Figure 35. 3.3V Output Normalized RMS AC Input Capacitor Current
0.300
80
70
60
50
0.200
2
40
0.100
30
1
0.060
20
0.040
10
0.020
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.080
0.010
0.008
0.006
0.004
3.3
4
5
6
7
8
9
10
12
15
INPUT VOLTAGE (V)
20
24
30
40
50
60
AN73 F36
Figure 36. 3.3V Output Normalized Output Capacitor AC RMS Current
AN73-41
Application Note 73
Let’s look at the RMS current in our transistors. Figure 37
details the bottom MOSFET RMS current at 0.85A; multiplying this by 20A yields a modest 17A (“1” on Figure 37).
Referring to Figure 38, the top MOSFET RMS current is
0.525A; multiplying this by 20A yields 10.5A.
HINDSIGHT
Other than the output capacitor’s RMS current from Figure
36 and the shaded areas of Figure 34, the ratio of peak-topeak ripple current to current limit will have little effect on
the buck converter (the actual value of the inductor is quite
noncritical).
The value of inductance we have chosen is 38µH, which
denormalizes to 38/24 or 1.58µH, at 100kHz, which
denormalizes to 100k/60k • 1.58µH or 2.63µH at 60kHz.
Verify that we stay airborne (hitting mountains bends
propellers). We notice from the isoclines that our peak-topeak ripple/ILIMIT varies from 15% at VIN = 4V to 75% at
VIN = 24V. Move on to Figure 35 to verify that the peak input
capacitor current will be at its maximum at VIN = 7V which
denormalizes to 10A (see line labeled “2”). Next, refer to
Figure 36 and draw a line representing the change in P-P
ripple/ILIMIT vs input voltage that we learned from Figure
34 (this is the line labeled “2”).
We can see that our maximum output capacitor AC RMS
current will be 0.25A multiplied by 20A, or 5A.
THIS IS TOO EASY; LETS DO A HARD ONE
4V < VINPUT < 24V, VOUTPUT = 3.3V/20A, operating at
60kHz using the smallest inductance
Start with Figure 34; pick the inductor value bounded by
the darker shaded zone to the left (current mode instability), see the point at the left of the line labeled as 2 on the
figure. Draw a line of constant inductance from 4V to 24V.
1.00
From Figure 37 we learn that our maximum RMS bottom
MOSFET current will be 0.95A • 20A, or 19A when the input
voltage is at 24V.
Figure 38 informs us that our maximum top MOSFET RMS
current will be 0.9A • 20A, or 18A.
80
10
1
0.90
0.80
2
0.70
0.60
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.50
0.40
0.30
0.20
0.15
0.10
3.3
4
5
6
7
8
9
10
12
15
INPUT VOLTAGE (V)
20
24
30
40
50
60
AN73 F37
Figure 37. 3.3V Output Normalized RMS Bottom Transistor Current
AN73-42
Application Note 73
1.5
1.0
0.9
0.7
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.8
0.6
2
0.5
1
0.4
0.3
80
10
0.2
3.3
4
5
6
7
8
9
10
12
15
INPUT VOLTAGE (V)
20
24
30
40
50
60
AN73 F38
Figure 38. 3.3V Output Normalized Top Switch RMS Current
CONCLUSION
The LT1339 has plenty of internal slope compensation for
most buck converters.
The next four sections are graphical design aids for 3.3V,
5V, 12V and 24V converters. The procedure for using
these design aids is the same as described in the preceding examples.
Inductor ripple current has little effect on anything except
stability, average current limit and the inductor’s own core
loss.
AN73-43
Application Note 73
3.3V OUTPUT GRAPH SET
400
10
300
20
150
30
100
90
80
40
70
50
60
P-P RIPPLE/ILIMIT (%)
MINIMUM INDUCTANCE (µH)
200
60
50
70
40
80
30
20
3.3
4
5
6
7
8
9
10
12
15
INPUT VOLTAGE (V)
20
24
30
40
50
60
AN73 F39
Figure 39. 3.3V Output Normalized Minimum Inductor Value
0.60
0.50
0.30
80
10
0.20
0.15
0.10
3.3
4
5
6
7
8
9
10
12
15
INPUT VOLTAGE (V)
20
24
30
40
50
60
AN73 F40
Figure 40. 3.3V Output Normalized RMS AC Input Capacitor Current
AN73-44
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.40
Application Note 73
0.300
80
70
60
50
0.200
40
0.100
30
0.060
20
0.040
10
0.020
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.080
0.010
0.008
0.006
0.004
3.3
4
5
6
7
8
9
10
12
15
INPUT VOLTAGE (V)
20
24
30
40
50
60
AN73 F41
Figure 41. 3.3V Output Normalized Output Capacitor AC RMS Current
80
10
1.00
0.90
0.80
0.70
0.60
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.50
0.40
0.30
0.20
0.15
0.10
3.3
4
5
6
7
8
9
10
12
15
INPUT VOLTAGE (V)
20
24
30
40
50
60
AN73 F42
Figure 42. 3.3V Output Normalized RMS Bottom Transistor Current
AN73-45
Application Note 73
1.5
1.0
0.9
0.7
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.8
0.6
0.5
0.4
0.3
80
10
0.2
3.3
4
5
6
7
8
9
10
12
15
INPUT VOLTAGE (V)
20
24
30
40
50
60
AN73 F43
Figure 43. 3.3V Output Normalized Top Switch RMS Current
AN73-46
Application Note 73
5V OUTPUT GRAPH SET
500
10
400
300
30
150
40
100
90
80
50
P-P RIPPLE/ILIMIT (%)
MINIMUM INDUCTANCE (µH)
20
200
60
70
70
60
80
50
40
30
5
6
7
8
9
10
12
15
20
INPUT VOLTAGE (V)
24
30
40
50
60
AN73 F44
Figure 44. 5V Output Normalized Minimum Inductor Value
0.60
0.50
0.30
80
10
0.20
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.40
0.15
0.10
5
6
7
8
9
10
12
15
20
INPUT VOLTAGE (V)
24
30
40
50
60
AN73 F45
Figure 45. 5V Output Normalized RMS AC Input Capacitor Current
AN73-47
Application Note 73
80
70
60
50
0.200
40
0.100
30
0.060
20
0.040
10
0.020
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.080
0.010
0.008
0.006
0.004
5
6
7
8
9
10
12
15
20
INPUT VOLTAGE (V)
24
30
40
50
60
AN73 F46
Figure 46. 5V Output Normalized Output Capacitor AC RMS Current
80
10
1.00
0.90
0.80
0.70
0.60
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.50
0.40
0.30
0.20
0.15
0.10
5
6
7
8
9
10
12
15
20
INPUT VOLTAGE (V)
24
30
40
50
60
AN73 F47
Figure 47. 5V Output Normalized RMS Bottom Transistor Current
AN73-48
Application Note 73
1.5
1.0
0.9
0.7
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.8
0.6
0.5
0.4
80
10
0.3
0.2
5
6
7
8
9
10
12
15
20
INPUT VOLTAGE (V)
24
30
40
50
60
AN73 F48
Figure 48. 5V Output Normalized Top Switch RMS Current
AN73-49
Application Note 73
12V OUTPUT GRAPH SET
1000
900
800
10
700
600
20
400
30
300
40
200
P-P RIPPLE/ILIMIT (%)
MINIMUM INDUCTANCE (µH)
500
50
60
150
70
80
100
12
13
14 15 16
18
20
24
30
INPUT VOLTAGE (V)
36
40
48 50
56
60
AN73 F49
Figure 49. 12V Output Normalized Minimum Inductor Value
0.60
0.50
80
10
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.40
0.30
0.20
0.15
0.10
12
13
14
15
16
18
20
24
30
INPUT VOLTAGE (V)
36
40
48 50
56
60
AN73 F50
Figure 50. 12V Output Normalized RMS AC Input Capacitor Current
AN73-50
Application Note 73
80
70
60
50
0.200
40
0.100
30
0.060
20
0.040
10
0.020
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.080
0.010
0.008
0.006
0.004
12
13
14
15
16
18
20
24
30
INPUT VOLTAGE (V)
36
40
48 50
56
60
AN73 F51
Figure 51. 12V Output Normalized Output Capacitor AC RMS Current
1.00
80
10
0.90
0.80
0.70
0.60
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.50
0.40
0.30
0.20
0.15
0.10
12
13
14
15
16
18
20
24
30
INPUT VOLTAGE (V)
36
40
48 50
56
60
AN73 F52
Figure 52. 12V Output Normalized RMS Bottom Transistor Current
AN73-51
Application Note 73
1.5
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
1.0
0.9
0.8
0.7
0.6
0.5
80
10
0.4
12
13
14
15
16
18
20
24
30
INPUT VOLTAGE (V)
36
40
48 50
56
60
AN73 F53
Figure 53. 12V Output Normalized Top Switch RMS Current
AN73-52
Application Note 73
24V OUTPUT GRAPH SET
1500
10
1000
900
20
700
600
500
30
400
40
300
P-P RIPPLE/ILIMIT (%)
MINIMUM INDUCTANCE (µH)
800
50
60
70
80
200
150
24 25 26
28
30
32
36
40
INPUT VOLTAGE (V)
45
48
50
56
60
AN73 F54
Figure 54. 24V Output Normalized Minimum Inductor Value
0.60
80
10
0.50
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.40
0.30
0.20
0.15
0.10
24
25
26
28
30
32
36
40
INPUT VOLTAGE (V)
45
48
50
56
60
AN73 F55
Figure 55. 24V Output Normalized RMS AC Input Capacitor Current
AN73-53
Application Note 73
80-70
60
50
0.200
40
0.100
30
0.060
20
0.040
10
0.002
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.080
0.010
0.008
0.006
0.004
24
25
26
28
30
32
36
40
INPUT VOLTAGE (V)
45
48
50
56
60
AN73 F56
Figure 56. 24V Output Normalized Output Capacitor AC RMS Current
80
10
0.80
0.70
0.60
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
0.50
0.40
0.30
0.20
0.15
0.10
24
25
26
28
30
32
36
40
INPUT VOLTAGE (V)
45
48
50
56
60
AN73 F57
Figure 57. 24V Output Normalized RMS Bottom Transistor Current
AN73-54
Application Note 73
P-P RIPPLE/ILIMIT (%)
RMS CURRENT (A)
1.5
1.0
0.9
0.8
0.7
80
10
0.6
24
25
26
28
30
32
36
40
INPUT VOLTAGE (V)
45
48
50
56
60
AN73 F58
Figure 58. 24V Output Normalized Top Switch RMS Current
AN73-55
Application Note 73
HOW TO CONVERT RMS CURRENT IN A CAPACITOR
TO PEAK-TO-PEAK RIPPLE VOLTAGE
Figure 59 is a guide to assist in output capacitor selection
and/or estimating peak-to-peak output ripple voltage.
Divide the output ripple specification (expressed in
mVP-P) by the maximum output capacitor RMS current
obtained from the design graphs to determine the effective resistance needed in the output capacitor. Using the
right-hand scale of Figure 59, follow the isocline that
approximates the needed effective resistance, calculated
above. Looking at the scale to the left, determine the
maximum allowable value for total capacitor ESR. Pick a
capacitor for the output, and plot its capacitance and ESR
on Figure 59. If the point is outside the effective-resistance isocline, you will either need to pick another capacitor or parallel multiple capacitors.
Here are the rules for paralleling capacitors:
1. You must parallel capacitors of the same capacitance
and ESR to correctly use Figure 59.
2. Paralleling n capacitors into a block will move you to
higher ground on the graph:
– The RMS ripple current of the block will be n × the
RMS ripple current of one capacitor.
– The ESR of the block will be the ESR of one capacitor/n.
– The capacitance of the block will be n × the capacitance of one capacitor.
3. Derate the capacitor block for end-of-life considerations (ESR doubles and the capacitance drops by
50%).
4. When using ceramic-type capacitors, the ESR is usually
so low that only the block capacitance need be considered.
5. With aluminum and tantalum capacitors, you will probably be paralleling capacitors to get low enough block
ESR to cross the isocline.
100
100kHz
0.2
0.1
0.05
CAPACITOR ESR (mΩ)
0.02
0.01
0.005
1
0.002
0.001
0.1
15
22
33
47
68
100 150 220 330 470 680 1k 1.5k 2.2k 3.3k 4.7k 6.8k 10k 15k
OUTPUT CAPACITOR (µF)
22k 33k 47k 68k
AN73 F59
Figure 59. RMS Current to Peak-to-Peak Ripple
AN73-56
EFFECTIVE RESISTANCE (Ω)
10
Application Note 73
6. If you are tempted to parallel a 15µF, 0.001Ω ceramic
capacitor with a 1000µF, 0.01Ω aluminum capacitor in
order to get a 1015µF, 0.0009Ω capacitor, you will be
surprised by the result:
– Plot the 15µF, 0.001Ω capacitor on Figure 59.
– Traverse the elevation isocline to its terminus at the
right side of the graph and read its effective resistance (~ 0.22Ω).
– Plot the 1000µF, 0.01Ω capacitor and follow its
isocline to the right side terminus, read its effective resistance (0.04Ω)
– At 100kHz, a 100µF, 0.001Ω ceramic capacitor has
about the same effective resistance as a 1000µF
0.01Ω aluminum electrolytic.
DENORMALIZING FIGURE 59 FOR OTHER
FREQUENCIES
As far as Figure 59 is concerned, a 100µF capacitor at
100kHz will look like a 60µF capacitor at 60kHz. All other
parameters stay the same. To plot a capacitor of xµF
operating at ykHz, first calculate a new capacitance, z,
where z(µF) = y(kHz)/100kHz • x(µF), then plot on Figure
59 using its native ESR value.
– Parallel the two impedances: 0.22 || 0.04 = 0.034;
an effective resistance far more affected by the
aluminum capacitor than the ceramic.
AN73-57
Application Note 73
BOOST CONVERTERS
OVERVIEW
The boost converter is a converter that has an output
voltage greater than its input voltage. Its input is a voltage
that varies over the regulator’s input range and its output
is a constant voltage.
Switching boost converters have current division, in that
the output current will be less than the input current. In an
ideal boost converter, the ratio of output current to input
current is the same as the ratio of input voltage to output
voltage (100% efficiency).
Characteristics of the
Synchronous Switching Boost Converter
The synchronous boost converter has the following characteristics:
1. It has a maximum input voltage that can be tolerated
while providing rated output voltage. For the power
converter (the LT1339 and its associated components), this voltage is usually limited first by the
minimum duty factor and finally by the body diode of
the top MOSFET. Excursions of input voltage above
the output voltage result in uncontrolled rise in output
voltage.
2. There is a minimum input voltage where boost conversion can be performed. Here several factors come
into the picture: undervoltage lockout, current mode
instability, average input current limit (if used) and
peak current limit.
In the boost converter the undervoltage lockout is
used to prevent operation where the performance or
robustness of the design would be compromised.
Because of the negative input impedance of a boost
converter, the input current increases as the input
voltage drops. This increase in input current is accompanied by higher RMS currents throughout the power
converter. Especially significant is the RMS output
ripple current in the output capacitor. Undervoltage
lockout is often used to prevent operation at duty
factors where current mode instability would cause
undesirable (audible) operation. Finally, undervoltage
lockout is used to prevent operation with insufficient
gate drive voltage.
AN73-58
3. The output current is discontinuous, with high AC
RMS values that require large output capacitors just to
accommodate the high AC current. This is also manifested in high output ripple voltage; an output filter
may be required to keep the switching frequency
ripple from disrupting the load.
4. The input current is continuous and has a triangular/
sawtooth wave shape. This means small input capacitors and low input ripple voltage.
5. There is a forward parasitic power path; hence,
shutting down the boost converter leaves a power
path from input through the choke and body diode of
the top MOSFET to the output. There is no provision
for current limiting in this parasitic power path.
Shorting the output of a boost converter can be
disastrous, depending upon the current capability of
the input source. This path exits both when the
converter is operating and when it is in shutdown.
6. The average current limit function in the LT1339
monitors input current rather than output current. The
maximum output current will depend on input voltage.
If the average current limit loop is not used (IAVE, Pin
5 grounded), the output current will be limited by the
peak input current limit. Under this condition the
maximum output current will depend on both the
input voltage and the inductor value, and the amount
of slope compensation used.
7. There is a parasitic reverse power path at the input. If
the input is pulled below ground, current flows through
the body diode of the bottom MOSFET, through the
inductor, through the sense resistor and out. Were
you to connect a car battery to the input of a synchronous boost converter backwards, very small resistances would limit current. This is not recommended.
8. There is a parasitic reverse power path at the output.
If the output is pulled below ground, the equivalent
circuit simplifies to two diodes to ground. These
diodes, the body diodes of the MOSFETs, are all that
stands between your circuit and disaster. When reverse polarity output voltage is applied, there is no
built-in mechanism to limit reverse current.
9. If the output voltage is pulled below the input voltage,
the body diode of the top MOSFET conducts to supply
Application Note 73
power from the input to the output. If you have a car
battery as the input, don’t short the output.
GRAPHICAL DESIGN SECTION
A boost converter can be designed using Figures 60 to 66
to select the inductor value. In boost mode, current mode
stability is more of a problem, as can be seen by the
respectively larger regions in the graphs where current
mode stability is a problem. Once an inductor value and its
ripple current are selected for the boost converter, the
balance of the design is done using the graphs in the buck
design section. Using these graphs with the input and
output voltages in reversed roles will provide the correct
answers. Keep in mind that the input capacitor and output
capacitor have reversed roles also. Their ripple current
charts should be applied accordingly.
10. Inductor value is not extremely critical; increasing the
inductor value above the required minimum nets
small returns. The main effect will be seen at minimum
input voltage, where the inductor value will affect the
stability of the current mode engine. If current mode
instability is a problem, increasing the inductor value
or adding additional slope compensation will be
required.
11. Efficiencies range from the low 80%s to the high
90%s.
12. Several things should be considered when selecting a
switching frequency. Higher switching frequency is
usually accompanied by a smaller inductor, lower
input, and output ripple voltages. In some cases
higher switching frequency can allow the use of smaller
or cheaper output capacitors. However, higher switching frequency is often accompanied by higher wire
loss and core loss in the inductor, higher switching
loss, and more drive power required by the MOSFETs.
For the modern core materials, capacitors and
MOSFETs, the most suitable frequency typically falls
in the 50kHz to 125kHz region.
CONCLUSION
As with the buck regulator design section, the use of the
graph can simplify the design of most boost designs. If for
some reason your application cannot be handled by the
graphical method detailed here, the data sheet has equations that lead you through the design process.
As with any of our products, if these tools do not provide
enough design support, call us. We will help you design a
system that will meet your needs, or we will help you
debug a design that isn’t performing to specification.
5V/60A Input to 28V/9A Boost Converter
IRF3205
×2
VIN
5V
60A
R5
0.002Ω
L1
40µH
Q1
Q2
FMMT720 FMMT619
220µF +
6.3V
×4
1N914
20
1µF
19
18
Q3
FMMT619
100k
VBOOST 12VIN
TG
RUN
TS
FB
5VREF
1µF
16
SL/ADJ
CT
100Ω
12
100Ω
11
10
L1: 12T 4 × AWG12 ON
Kool Mµ® 77437-A7
14
0.1µF
SENSE¯
SENSE +
VC
VREF
IAVE
PHASE
SS
SYNC PGND SGND
1
15
VOUT
28V
RFB
2.7k
1.2k
9
2
4
100k
LT1339
47µF
16V
2200µF
35V
×6
13
BG
Q4
FMMT720
Kool Mµ IS A REGISTERED
TRADEMARK OF MAGNETICS, INC.
+
17
12V
IRF3205
×4
+
12V
2200pF
1µF
3
7.5k
7
5
1.5nF
330pF
6
10µF
8
AN74 TA03
AN73-59
Application Note 73
CIRCUIT COLLECTION
10
500
400
300
200
30
150
40
100
50
P-P RIPPLE/ILIMIT (%)
MINIMUM INDUCTANCE (µH)
20
60
70
80
70
50
40
30
1
1.5
2
2.5
INPUT VOLTAGE (V)
3
3.3
4
5
AN73 F60
Figure 60. 5V Output Normalized Minimum Inductor Value
1500
10
1000
20
500
400
30
300
40
50
200
60
150
70
80
100
70
2
3
4
5
6
INPUT VOLTAGE (V)
7
8
9
10
12
AN73 F61
Figure 61. 12V Output Normalized Minimum Inductor Value
AN73-60
P-P RIPPLE/ILIMIT (%)
MINIMUM INDUCTANCE (µH)
700
Application Note 73
3000
10
2000
20
1000
30
700
40
500
50
400
60
300
70
80
P-P RIPPLE/ILIMIT (%)
MINIMUM INDUCTANCE (µH)
1500
200
150
3
4
5
6
7
8
9
10
INPUT VOLTAGE (V)
15
24
AN73 F62
Figure 62. 24V Output Normalized Minimum Inductor Value
4000
10
3000
1500
20
1000
30
40
700
50
500
60
400
70
80
P-P RIPPLE/ILIMIT (%)
MINIMUM INDUCTANCE (µH)
2000
300
200
150
5
6
7
8
9
10
15
20
24
30
INPUT VOLTAGE (V)
AN73 F63
Figure 63. 30V Output Normalized Minimum Inductor Value
AN73-61
Application Note 73
4000
10
3000
2000
1500
30
1000
40
50
700
60
P-P RIPPLE/ILIMIT (%)
MINIMUM INDUCTANCE (µH)
20
70
80
500
400
300
200
5
6
7
8
9
10
15
INPUT VOLTAGE (V)
20
24
30
36
AN73 F64
Figure 64. 36V Output Normalized Minimum Inductor Value
5000
10
4000
3000
2000
30
1500
40
1000
50
60
700
70
80
500
400
300
5
7.5
10
15
20
INPUT VOLTAGE (V)
24
30
36
42
48
AN73 F65
Figure 65. 48V Output Normalized Minimum Inductor Value
AN73-62
P-P RIPPLE/ILIMIT (%)
MINIMUM INDUCTANCE (µH)
20
Application Note 73
7000
10
5000
3000
20
2000
30
1500
40
50
60
1000
P-P RIPPLE/ILIMIT (%)
MINIMUM INDUCTANCE (µH)
4000
70
80
700
500
400
300
5
7.5
10
15
20
INPUT VOLTAGE (V)
24
30
36
42
48
60
AN73 F66
Figure 66. 60V Output Normalized Minimum Inductor Value
TYPICAL APPLICATIONS
28V to 5V/20A Buck Converter
1000µF 35V × 6
28V
+
+
+
+
+
+
1µF
1N4148
20
56µF
25V
56µF
35V
+
+
7815
17
6
22k
13
+
1.2k
0.1µF
16F
10µF
2.2nF
51k
10k
4
2
14
3
5
7
2.2nF
10
1k
1
0.1µF
0.1µF
3A SCHOTTKYS
VBOOST
VIN
TG
SS
RUN
SLOPE
5VREF
TS
19
IRL3103
3µH
25A
18
0.01µF
0.01µF
+
LT1339
PHASE
BG
RC
IAVE
SENSE +
VC
3.3V CR7
16
11
1µF
5V
2200µF
6.3V
IRL3103
2×
10k
1N4148
VREF
SYNC
SENSE
SGND PGND
2
15
–
12
FB
9
AN73 TA01
20k
10k
SYNC
100k
1k
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
AN73-63
Application Note 73
TYPICAL APPLICATIONS
Constant-Current Solenoid Driver with 2 × Turn-On Boost
+
ON
OFF
10µF
TURN ON
BOOST TIMER
MPS2907
2A
1
2
OFF INPUT
“0” = OFF
3
4
C8
2200pF
C10 1µF
5
C13
330pF
6
7
C11 1000pF
MBRS1100T3
+
100k
R5 10k
C7
1µF
VIN
10V TO 18V
4A
1N4148
R6
10k
8
9
10
SNYC
VBOOST
5VREF
TG
CT
20
19
IAVG
17
12VIN
16
BG
LT1339
SS
PGND
VC
PHASE
Q2
IRL3710S
15
RUN/SHDN
SGND
Q1
IRL3710S
D4
MBR0520LT1
18
TS
SL/ADJ
CIN1
220µF
C6
1µF
C5
1µF
14
R11 30k
13
R12 10k
VFB
SENSE –
12
VREF
SENSE +
11
•
•
•
•
•
•
D6
MBR0520LT1
T1
VP5-0155
COILTRONICS
40µH
4.8A
RS
0.025Ω
1/2W
+
COUT1
220µF
16V
+
COUT2
220µF
16V
LARGE
SOLENOID
AN74 TA02
2.5A SEPIC Converter. The Output Voltage Can Be Lower or Higher Than the Input Voltage
VIN
11V TO 16.5V
17
330pF
7.68k
5
13
2
1k
1µF
4
10k
3
2.2µF
7
6
8
6.8k
+
0.068µF
15
10µF
VCC
+
20
0.02Ω
VBOOST
IAVG
RUN
SENSE +
11
SENSE –
12
T1
40µH
5A
•
5VREF
TG
SL/ADJ
19
470µF
25V
•
0.22Ω SETS OUTPUT
CURRENT LIMIT
NC
LT1339
CT
VREF
VC
10
0.1µF
SS
SGND
BG
PGND
FB
9
TS
+
CLOSE TURNS OFF AND
DISCONNECTS POWER
FROM THE OUTPUT
16
18
+
100µF
20V
19V
25A
1000µF
35V
IRFZ34
RFB SETS OUTPUT
14.2k VOLTAGE
SYNC PHASE
1
14
1k
100Ω
2N3904
0.1µF
AN73 TA04
Linear Technology Corporation
McCarthy Blvd., Milpitas, CA 95035-7487
AN73-64 1630
(408) 432-1900
: (408) 434-0507
: 499-3977
●
FAX
●
TELEX
an73f LT/TP 0299 4K • PRINTED IN USA
 LINEAR TECHNOLOGY CORPORATION 1999