140W Monolithic Switching Regulator Simplifies Constant-Current/Constant-Voltage Regulation Eric Young The LT3956 is a monolithic switching regulator that can generate constant-current/constant-voltage outputs in buck, boost or SEPIC topologies over a wide range of input and output voltages. With input and output voltages of up to 80V, a rugged internal 84V switch and high efficiency operation, the LT3956 can easily produce high power in a small footprint. The LT®3956 combines key amplifier and comparator blocks with a high current/ high voltage switching regulator in a tiny 5mm × 6mm package. See Figure 1 for an example of how little board space is needed to produce a complete constant-current, constant-voltage boost circuit ideal for LED driving, supercap charging or other high power applications that require the added protection of input or output current limiting. WHAT MAKES THE LT3956 TICK? The big mover in the LT3956 is an 84V-rated, 90mΩ low side N-MOSFET switch with an internally programmed current limit of 3.9A (typ). The switching regulator can be powered from a supply as high as 80V because the N-MOSFET switch driver, the PWMOUT pin driver, and most internal loads are powered by an internal LDO linear regulator that converts VIN to 7.15V, provided the VIN supply is high enough. The switch duty cycle and current is controlled by a current-mode pulse-width modulator—an architecture that provides fast transient response, fixed switching frequency operation and an easily stabilized feedback loop at variable inputs and outputs. The switching frequency can be programmed from 100kHz to 1MHz with an external resistor, which allows designers to optimize component 16 | April 2010 : LT Journal of Analog Innovation size and performance parameters, such as min/max duty cycle and efficiency. And at the heart of the LT3956 is a dual input feedback transconductance (gm) amplifier that combines a differential constant current sense with a standard low side voltage feedback. The handoff between these two loops is seamless and predictable. The feedback loop operating closest to its set point is auto-selected to be the loop controlling the flow of charge onto the compensation R-C network attached to the VC pin. The voltage Figure 1. Complete high power, constant current, constant voltage boost circuit level at the VC pin in turn controls the current and duty ratio of the switch. A more thorough description of operation can be found in the LT3956 data sheet. C1 2.2µF ×2 R1 332k R2 100k INTVCC VIN 0.33Ω INTVCC ISP R3 1M LT3956 CTRL 1k Q2 Q3 20k 2k D2 ISN FB 100k VMODE PWM SS RT VC 0.01µF PGND VREF R6 57.6k 28.7k 375kHz C2 2.2µF ×10 SW EN/UVLO R5 1M 750mA D1 22µH VIN 6V TO 60V (80V TRANSIENT) 200Ω 6.8nF Figure 2. This 50W boost LED driver provides wide input range, PWM dimming and LED fault protection and reporting. M1 R4 16.2k LED+ Q1 PWMOUT GND INTVCC INTVCC 4.7µF 1k 18 WHITE LEDs 50W (DERATED IF VIN < 22V) M1: VISHAY SILICONIX Si7113DN D1: DIODES INC PDS5100 L1: COILTRONICS DR125-220 C1,C2: MURATA GRM42-2X7R225 Q1: ZETEX FMMT497 Q2,Q3: ZETEX FMMT589 D2: BAV116W D3: DIODES INC B1100/B D3 design features The big mover in the LT3956 is an 84V-rated, 90mΩ low side N-MOSFET switch with an internally programmed current limit of 3.9A (typ). The switching regulator can be powered from a supply as high as 80V because the N-MOSFET switch driver, the PWMOUT pin driver, and most internal loads are powered by an internal LDO linear regulator that converts VIN to 7.15V. 100 0.8 VLED+ 50V/DIV CURRENT EFFICIENCY (%) 0.6 EFFICIENCY 90 0.4 85 0.2 80 0 10 20 30 VIN (V) 40 50 OUTPUT CURRENT (A) 95 PWM 5V/DIV 60 SW 50V/DIV ILED 2A/DIV ILED 500mA/DIV FB 5V/DIV VIN = 24V 10µs/DIV VIN = 24V 200ns/DIV Figure 3. High 94% efficiency means less than 3W dissipation in the converter shown in Figure 2. Figure 4. Boost PWM dimming waveforms for 60V of LEDs shows microsecond rise and fall times and excellent constant current regulation even over short intervals. Figure 5. LED+ terminal of boost shorted to GND is prevented from damaging switching components by a novel circuit. A RUGGED HIGH POWER BOOST LED DRIVER Analog Dimming solution for the luminary—the LED cathode current can return on a common GND. A scope photo of PWM dimming waveform (Figure 4) shows sharp rise and fall times, less than 200ns, and quick stabilization of the current. Although a low side N-MOSFET disconnect at the cathode is the simpler and more obvious (and a bit faster) implementation for this particular boost circuit using the LT3956, the use of high side PWM disconnect is important to a boost protection strategy to be discussed below. Figure 2 shows a 50W boost LED driver that operates from a 24V input, showing off some of the unique capabilities of this product when applied as an LED driver. This boost circuit tolerates a wide input range—from 6V to 60V. At the low end of this VIN range, the circuit is prevented from operating too close to switch current limit by scaling back the programmed LED current as VIN declines— set by the resistor divider (R5 and R6) on the CTRL pin. Figure 3 shows efficiency and LED current versus VIN. The high efficiency (94%) means passive cooling of the regulator is adequate for all but the most extreme environmental conditions. ANALOG AND PWM LED DIMMING The LT3956 offers two high performance dimming methods: analog dimming via the CTRL pin and the ISP/ISN current sense inputs, and PWM dimming through the PWM input and PWMOUT output. Analog dimming is achieved via the voltage at the CTRL pin. When the CTRL pin is below 1.2V, it programs the current sense threshold from zero to 250mV (typ) with guaranteed accuracy of ±3.5% at 100mV. When CTRL is above 1.2V, the current sense threshold is fixed at 250mV. At CTRL = 100mV (typ), the current sense threshold is set to zero. This built-in offset is important to the feature if the CTRL pin is driven by a resistor divider—a zero programmed current can be reached with a non-zero CTRL voltage. The CTRL pin is high impedance so it can be driven in a wide variety of configurations. PWM Dimming Pulse width modulation (PWM) of LED current is the preferred technique to achieve wide range dimming of the light output. Figure 2 shows a level shift transistor Q1 driving a high side disconnect P-MOSFET M1. This configuration allows PWM dimming with a single wire CONSIDERATIONS FOR PROTECTING THE LED, THE DRIVER, AND THE INPUT POWER SUPPLY LED systems often require load fault detection. Limiting the output voltage in the case of an open LED string has always been a basic requirement and is achieved through a resistor divider (R3 and R4) at the FB input. If the string opens, the switching regulator regulates VFB to a constant 1.25V (typ). In addition to the gm amplifier that provides this constant April 2010 : LT Journal of Analog Innovation | 17 The boost circuit in Figure 2 uses the voltage feedback (FB) input in a unique fashion—protecting the LED+ node from a fault to GND while preserving all the other desirable attributes of the LED driver. A standard boost circuit has a direct path from the supply to the output, and therefore cannot survive a GND fault on its output when the supply current is not limited. There are a number of situations where one might desire to protect the switching regulator from a short to GND of the LED anode—perhaps the luminary is separated from the driver circuit by a connector or by a long wire, and the input supply is a high capacity battery. The LT3956 has a feature to provide this protection. The overvoltage FB (OVFB) comparator is a second comparator on the FB input with a setpoint higher than the VFB regulation voltage. It causes the PWMOUT pin to transition low and switching to stop immediately when the FB input exceeds 1.31V(typ). 18 | April 2010 : LT Journal of Analog Innovation VIN OVLO 53V RISING 51V FALLING VIN EN/UVLO 10k R1 1M INTVCC 4.7µF LT3956 Q1 R2 143k PWM PWMOUT GND BAV116 PWM 332k by the resistor divider from VIN) exceeds 6.5V (INTVCC minus a VBE). When PWM falls below its threshold, PWMOUT goes low as well. Hysteresis of ~2V is provided by PWMOUT. Because of the high PWM threshold (0.85V minimum over temperature), the blocking diode D1 can be added to preserve the PWM dimming capability. Q1: ZETEX FMMT593 Figure 6. VIN overvoltage circuit halts switching and disconnects the load during high input voltage transients. The OVFB comparator can be used in an output GND fault protection scheme (patent pending) for the boost. The key elements are the high side LED disconnect P-MOSFET (M1) and its supporting driving circuit responsive to the PWMOUT signal, and the output GND fault sensing circuit consisting of D2, Q2 and two resistors that provide signal to the FB node. The circuit works by sensing the current flowing in D2 when the output is shorted, and thereby triggering the OVFB comparator. In response to the OVFB comparator, the high side switch M1 is maintained in an offstate and the switching is stopped until the fault condition is removed. Figure 5 shows the current waveform in the M1 switch during and output short circuit event. Additional Considerations for Protecting the LED Some harsh operating environments produce transients on the input power supply that can overdrive a boosted output, if only for a short while, and potentially damage the LEDs with excessive current. To discontinue switching and disconnect the LEDs during such a transient, a simple add-on circuit to the PWM input, shown as a breakout in Figure 6, disconnects the LED string and idles the switcher when VIN exceeds 50V. The circuit works by sourcing current to the PWM input of the LT3956 from the collector of Q1 when VIN is low enough, but cutting off that current when the base of Q1 (set The LT3956 provides solutions to thermal dissipation problems encountered driving LEDs. With high power comes the concern about reduced lifetime of the LED due to continuous operation at high temperatures. Increasing numbers of LED module applications implement thermal sensing for the LED, usually employing an NTC resistor coupled to the LED heat sink with thermal grease. A simple circuit employing the CTRL and VREF pins of the LT3956 and an NTC resistor sensing the LED temperature produces a thermal derating curve for the LED current as shown in Figure 7. A CONSTANT-CURRENT/VOLTAGE REGULATOR SERVES A WIDE RANGE OF APPLICATIONS Driving LEDs makes excellent use of the LT3956 features, but it isn’t the only application that requires constant Figure 7. CTRL and VREF pins provide thermal derating to enhance LED reliability. ISP + CTRL ISN 100k NTC RT1 MURATA NCP18WM104 – VREF 16.9k LT3956 V(ISP-ISN) 300 V(ISP-ISN) THRESHOLD (mV) voltage regulation, the FB input also has two fixed setpoint comparators associated with it. The lower setpoint comparator activates the VMODE open collector pulldown when FB exceeds 1.20V (typ). After the disconnection of the LED and loss of the current regulation signal, the output rises until it reaches the constant voltage regulation setpoint. During this voltage ramp the VMODE pin asserts and holds, indicating that the LED load is open. This signal maintains its state when PWM goes low and the regulator stops switching, allowing for the likelihood that output voltage may fall below the threshold without an occasional refresh provided by switching. The VMODE pin quickly updates when PWM goes high. The VMODE signal can also indicate that the regulation mode is transitioning from constant current to constant voltage, which is the appropriate function for current limited constant voltage applications, such as battery chargers. 250 200 150 100 50 0 25 45 65 85 TEMPERATURE (°C) 105 125 design features voltage at constant current. It can be used for charging batteries and supercapacitors, or driving a current source load such as a thermoelectric cooler, just to name a few examples. It can be used as a voltage regulator with current limited input or output, or a current regulator with a voltage clamp. intermittently, but the available power might be limited based on an overall system budget. The output charging rate of the circuit of Figure 8 is not based on any timer, but rather on the output voltage level as sensed by the CTRL pin. Below a certain output voltage, 22V in this case, the input current is limited so that the switching regulator is maintained within its own current limit. At higher output voltages, the default internal current sensing threshold of 250mV (typ) establishes that the input current cannot exceed 1.2A, and so the output current drops. At very low output voltages less than 1.5V, the network driving the SS pin of LT3956 reduces the switching frequency and the current limit to maintain good control of the charging current. When the load is within 5% of its target voltage, the VMODE pin toggles to indicate the end of constant current mode and entry into constant voltage regulation. Pursuing this line of thinking, Figure 8 shows a SEPIC supercap charger that draws power from a fixed 24V input, and has an input current limit of 1.2A. The SEPIC architecture is chosen for several reasons: it can do both step-up and step-down, and it has inherent isolation of the input from the output. A coupled inductor is chosen over a 2-inductor approach because of the smaller, lowercost circuit. The magnetic coupling effect allows the use of a single coupling capacitor and the switch current levels of the LT3956 make strategic use of the readily available coupled inductor offerings of major magnetics vendors. This circuit is intended for a situation where VIN does not experience much variation during normal operation. The design procedure for this type circuit begins with setting the maximum input current limit with the RSENSE value and the 250mV default threshold. The next design step is to determine the VOUT level below A charging circuit for a large value capacitor (1F or more) might be found in a nonbattery based backup power system. These chargers would draw power from some inductive based DC supply that operates which VIN current is to be reduced through CTRL to maintain less than 2.5A average switching current. Assuming slightly less than 90% efficiency, set the resistor divider R5 and R6 to give CTRL = 1.1V when VOUT = VIN • 0.9 • IIN(MAX ) 2.5A − IIN(MAX ) , at CTRL = 1.1V The values of R5 and R6 should be an order of magnitude higher value than resistor R7. The resistor divider R7 and R8 is set to provide a minimum voltage at CTRL, greater than 125mV, which is needed to set non-zero value for input current. CONCLUSION The LT3956 simplifies power conversion applications needing both constant-current and constant-voltage regulation, especially if they are constrained by board area and/ or bill-of-materials length. Its features are selected to minimize the number of external analog blocks for these types of applications while maintaining flexibility. Careful integration of these components into the switching regulator makes it possible to easily produce applications that would otherwise require a cumbersome combination of numerous externals. n Figure 8. Supercapacitor charger with current limited input provides controlled charging current over a wide output range. SW VIN PGND EN/UVLO FB LT3956 1:1 C1 10µF VMODE CTRL INTVCC VREF SS VC C2 4.7µF RT GND 28.7k 370kHz 10nF L1B C3 10µF R5 1M R6 40.2k INTVCC 10k PWM VOUT 0V TO 28V R3 536k R4 24.9k PWMOUT D1 R8 14k 1M R1 59k R7 2k Q1 R2 30.1k 3 INPUT AND OUTPUT CURRENT (A) ISN ISP 1µF C4 10µF L1A 33µH RSENSE 200mΩ VIN 28V ≤ 1.2A 2.5 2 OUTPUT 1.5 1 INPUT 0.5 0 0 5 10 15 20 VOUT (V) 25 30 L1: WURTH ELEKTRONIK 744871330 D1: ON SEMICONDUCTOR MBRS360T Q1: MMBTA42 C1,C3,C4: TAIYO YUDEN GMK316BJ106 April 2010 : LT Journal of Analog Innovation | 19