V20N1 - APRIL

April 2010
I N
T H I S
I S S U E
our new look 2
dual output step-down
regulator with DCR sensing
in a 5mm × 5mm QFN 9
accurate battery gas
Volume 20 Number 1
Energy Harvester Produces Power
from Local Environment, Eliminating
Batteries in Wireless Sensors
Michael Whitaker
gauges with I2C interface 12
dual buck regulator
operates outside of
AM radio band 20
eight 16-bit VOUT DACs in a
4mm × 5mm QFN 24
Advances in low power technology are making it easier to create
wireless sensor networks in a wide range of applications, from
remote sensing to HVAC monitoring, asset tracking and industrial
automation. The problem is that even wireless sensors require
batteries that must be regularly replaced—a costly and cumbersome
maintenance project. A better wireless power solution would be to
harvest ambient mechanical, thermal or electromagnetic energy from
the sensor’s local environment.
dual-phase converter for
1.2V at 50A with DCR
sensing 28
Typically, harvestable ambient power is on the order of tens of microwatts, so energy harvesting
requires careful power management in order to successfully capture microwatts of ambient power
and store it in a useable energy reservoir. One common form of ambient energy is mechanical
vibration energy, which can be caused by motors running in a factory,
airflow across a fan blade or even by a moving vehicle. A piezoelectric transducer can be used to convert these forms of vibration energy
into electrical energy, which in turn can be used to power circuitry.
To manage the energy harvesting and the energy release to the system,
the LTC®3588-1 piezoelectric energy harvesting power supply (Figure 1)
integrates a low loss internal bridge rectifier with a synchronous stepdown DC/DC converter. It uses an efficient energy harvesting algorithm
to collect and store energy from high impedance piezoelectric elements,
which can have short-circuit currents on the order of tens of microamps.
Piezoelectric energy harvesting power supply
Energy harvesting systems must often support peak load currents
that are much higher than a piezoelectric element can produce, so
the LTC3588-1 accumulates energy that can be released to the load in
short power bursts. Of course, for continuous operation these power
(continued on page 4)
ww w. li n e a r.c om
to our readers
Design Innovation: Our New Look
In this issue...
Bob Dobkin, Co-founder, Vice President, Engineering & CTO
COVER ARTICLE
Energy Harvester Produces Power
from Local Environment, Eliminating
Batteries in Wireless Sensors
Michael Whitaker
1
DESIGN FEATURES
Dual Output Step-Down Regulator Features
Pin Selectable Outputs, DCR Sensing, Reverse
Current Protection and a 5mm × 5mm QFN
Stephanie Dai
9
Tiny, Accurate Battery Gas Gauges with
Easy-to-Use I2C Interface and Optional
Integrated Precision Sense Resistors
Christoph Schwoerer, Axel Klein and Bernhard Engl
12
140W Monolithic Switching Regulator Simplifies
Constant-Current/Constant-Voltage Regulation
Eric Young
16
2MHz Dual Buck Regulator Operates
Outside of AM Radio Band When Delivering
3.3V and 1.8V from 16V Input
Pit-Leong Wong
20
Eight 16-Bit, Low INL, VOUT DACs in a 4mm × 5mm
QFN Package: Unparalleled Density and Flexibility
Leo Chen
24
DESIGN IDEAS
6mm × 6mm DC/DC Controller for
High Current DCR Sensing Applications
Eric Gu, Theo Phillips, Mike Shriver and Kerry Holliday
28
Maximize Cycle Life of Rechargeable
Battery Packs with Multicell Monitor IC
Jon Munson
30
To complement the technical innovation that defines Linear’s products, we have
added color and features to make this publication more readable, useful and
inviting. It’s our goal to continue to provide you with an informative publication,
worth keeping.
The new design is optimized for delivery both electronically and in print for ease
of download or printing on a laser printer. The addition of color makes for more
compelling reading and better understanding of the technical material.
Over the years, we’ve referred to this publication as “LT magazine,” so the new
name is fitting. Since all the material in the publication is new circuitry, we have
added a new subtitle, Journal of Analog Innovation, to emphasize the technical
sophistication.
As before, LT carries in-depth features and design ideas, plus a new section, “highlights from circuits.linear.com” on the back page. What hasn’t changed is what
we’re about. After more than 28 years delivering innovative high performance
analog solutions, we’re still at it—focused on solving the hard-to-do analog problems. We’re confident that the designs discussed in LT will give your end-products
a performance edge.
LT is your publication and we welcome your comments, ideas and suggestions—
both for the products you need and the ways you’d like to receive information. We
hope you like our latest design innovation. n
Dual 500mA µPower LDO Features
Independent 1.8V–20V Inputs and Easy
Sequencing in a 4mm × 3mm DFN
Molly Zhu
33
product briefs
34
back page circuits
36
2 | April 2010 : LT Journal of Analog Innovation
Design innovation. That’s what we’re about—in our
products and here in the newly updatedLinear Technology
magazine. After nearly two decades in a two-color
format, we have updated its design while maintaining
the deep technical focus you’ve come to expect.
Linear in the news
Linear in the News
µMODULE SEMINARS IN EUROPE
ANALOG PRODUCT
OF THE YEAR
The LTC6802 is a highly
integrated multicell battery monitoring IC capable of precisely measuring
the voltages of up to 12
series-connected lithiumion battery cells. With a
maximum measurement
error of less than 0.25%,
the LTC6802 enables battery cells to operate over
their full charge and discharge limits, maximizing
lifetime and useful battery
capacity.
Designers today are challenged with the
increased complexity of power design, a
growing number of system voltage rails
and with designing power management
solutions to drive FPGAs and DSPs. Linear’s
µModule® DC/DC regulators provide
a complete circuit in a tiny package,
simplifying system power design, saving
valuable board space, increasing efficiency,
and reducing development time and risk.
To ease the design challenge, Linear
Technology and its partners are offering
a series of free, half-day seminars across
Europe on the use of its high performance
DC/DC µModule regulators. The seminars,
which include lunch, give an overview
of high end power needs for complex
systems, discuss the power µModule
regulator concept, and include a design lab
with demonstrations of tools and demo
boards. The seminars are held in March,
April and May at various locations in
the UK, Ireland, Sweden, Denmark, Italy,
France, Germany, Austria, Switzerland,
Belgium and the Netherlands. For complete information, visit www.linear.com/
designtools/EasyAnalogSeminars.html.
PARTICIPATION AT IIC-CHINA
Linear had a booth at the IIC-China exhibition in Shenzhen on March 4–5 and in
Chengdu on March 11–12. As the China
electronics market continues to grow in
scale and sophistication, Linear is well
positioned to grow in that market. Linear’s
product emphasis on such markets as
industrial, communications and automotive fits well with the China market, which
is seeing growth in all these segments.
The industrial market demands products
with high precision, quality and reliability—areas in which Linear Technology
excels. The automotive market is being
fueled by the push to develop high
efficiency hybrid and electric vehicles.
In addition, the overall electronic content in vehicles is increasing rapidly,
with analysts estimating that electronic
content in cars will reach 15% of a
new vehicle’s cost by 2012. There is an
explosion underway in the need for
communications infrastructure equipment to keep up with the demand from
the growing use of smart phones and
other broadband wireless devices.
At IIC-China, Linear showcased power
µModule DC/DC regulators, battery stack
monitors for hybrid and electric vehicles,
energy harvesting products that capture ambient energy to power sensors
wirelessly, isolated µModule transceivers + power, communications µModule
products, RF products for communications
infrastructure, digital power management,
LED drivers and power management ICs.
ANALOG PRODUCT OF THE YEAR
At Electronics Weekly’s annual Elektra
Awards ceremony, held in London
in December, Linear Technology’s
LTC6802 Battery Stack Monitor for
Hybrid/Electric Vehicles was named
Semiconductor Product of the Year—
Analogue. The product was selected
based on its performance, design flexibility and suitability for the application.
The LTC6802 is a highly integrated multicell battery monitoring IC capable of
precisely measuring the voltages of up
to 12 series-connected lithium-ion battery cells. Applications for the LTC6802
include electric and hybrid electric
vehicles, scooters, motorcycles, golf
carts, medical equipment and uninterruptible power supply systems. n
April 2010 : LT Journal of Analog Innovation | 3
The LTC3588-1 interfaces with the piezo through its internal
low loss bridge rectifier accessible via the PZ1 and PZ2
pins. The rectified output is stored on the VIN capacitor.
At typical 10µA piezoelectric currents, the voltage drop
associated with the bridge rectifier is on the order of 400mV.
(continued from page 1)
bursts must occur at a low duty cycle,
such that the total output energy during
the burst does not exceed the average
source power integrated over an energy
accumulation cycle. A sensor system
that makes a measurement at regular
intervals, transmits data and powers
down in between is a prime candidate
for an energy harvesting solution.
KEY TO HARVESTING IS
LOW QUIESCENT CURRENT
The energy harvesting process relies on a
low quiescent current energy accumulation
phase. The LTC3588-1 enables this through
an undervoltage lockout (UVLO) mode with
a wide hysteresis window that draws less
than a microamp of quiescent current. The
UVLO mode allows charge to build up on
an input capacitor until an internal buck
converter can efficiently transfer a portion of the stored charge to the output.
Figure 2 shows a profile of the quiescent
current in UVLO, which is monotonic with
VIN so that a current source as low as
700nA could charge the input capacitor
1µF
6V
CSTORAGE
25V
to the UVLO rising threshold and result
in a regulated output. Once in regulation, the LTC3588-1 enters a sleep state in
which both input and output quiescent
currents are minimal. For instance, at
VIN = 4.5V, with the output in regulation, the quiescent current is only 950nA.
The buck converter then turns on and
off as needed to maintain regulation.
Low quiescent current in both the sleep
and UVLO modes allows as much energy
to be accumulated in the input reservoir capacitor as possible, even if the
source current available is very low.
800
700
SW
VOLTAGE (V)
–40°C
500
400
300
PGOOD
D0, D1
0
0
2
3
VIN (V)
4
5
Figure 2. IVIN in UVLO vs VIN
4 | April 2010 : LT Journal of Analog Innovation
6
85°C
1600
25°C
1400
1200
–40°C
1000
6
2
D1 = D0 = 0
2000
8
800
VOUT
600
PGOOD = LOGIC 1
0
200
400
TIME (s)
OUTPUT
VOLTAGE
SELECT
When VIN reaches the UVLO rising threshold, the high efficiency integrated synchronous buck converter turns on and
begins to transfer energy from the input
capacitor to the output capacitor. The
buck regulator uses a hysteretic voltage algorithm to control the output via
internal feedback from the VOUT sense pin.
It charges the output capacitor through
an inductor to a value slightly higher
than the regulation point by ramping the
inductor current up to 250mA through
an internal PMOS switch and then ramping it down to zero current through an
internal NMOS switch. This efficiently
delivers energy to the output capacitor.
2200
10
4
2
GND
2400
VIN
12
100
VOUT
47µF
6V
1800
14
200
10µH
VOUT
VIN2
16
600
LTC3588-1
CAP
CSTORAGE = 22µF, COUT = 47µF
20 NO LOAD, IVIN = 2µA
18
25°C
1
PZ2
VIN
22
85°C
0
PZ1
4.7µF
6V
D1 = D0 = 1
900
IVIN (nA)
ADVANCED CERAMETRICS PFCB-W14
IVIN (nA)
1000
Figure 1. Piezoelectric energy
harvesting power supply
600
Figure 3. A 3.3V regulator start-up profile
400
2
4
6
8
10 12
VIN (V)
14
16
Figure 4. IVIN in sleep vs VIN
18
design features
Figure 8. Block diagram of the LTC3588-1
VIN
20V
INTERNAL RAIL
GENERATION
CAP
PZ1
SW
VIN2
PZ2
If the input voltage falls below the
UVLO falling threshold before the output voltage reaches regulation, the buck
converter shuts off and is not turned on
again until the input voltage rises above
the UVLO rising threshold. During this time
the leakage on the VOUT sense pin is no
greater than 90nA and the output voltage remains near the level it had reached
when the buck was switching. Figure 3
shows a typical start-up waveform of the
LTC3588-1 charged by a 2µA current source.
When the synchronous buck brings the
output voltage into regulation the converter enters a low quiescent current sleep
state that monitors the output voltage with
a sleep comparator. During this operating
mode, load current is provided by the buck
output capacitor. When the output voltage
falls below the regulation point the buck
regulator wakes up and the cycle repeats.
This hysteretic method of providing a regulated output minimizes losses associated
with FET switching and makes it possible
to efficiently regulate at very light loads.
EFFICIENCY (%)
50
40
VOUT = 3.6V
VOUT = 3.3V
VOUT = 2.5V
VOUT = 1.8V
20
10
1µ
10µ
100µ
1m
10m
LOAD CURRENT (A)
Figure 5. Efficiency vs ILOAD
100m
PGOOD
LTC3588-1
The buck delivers up to 100mA of average load current when it is switching.
Four output voltages, 1.8V, 2.5V, 3.3V and
3.6V, are pin selectable and accommodate
powering of microprocessors, sensors and
wireless transmitters. Figure 4 shows the
extremely low quiescent current while in
regulation and in sleep, which allows for
efficient operation at light loads. Although
the quiescent current of the buck regulator while switching is much greater than
the sleep quiescent current, it is still a
small percentage of the load current,
which results in high efficiency over a
wide range of load conditions (Figure 5).
The buck operates only when sufficient
energy has been accumulated in the input
capacitor, and it transfers energy to the
output in short bursts, much shorter than
the time it takes to accumulate energy.
When the buck operating quiescent current is averaged over an entire accumulation/burst period, the average quiescent
current is very low, easily accommodating sources that harvest small amounts
of ambient energy. The extremely low
quiescent current in regulation also allows
the LTC3588-1 to achieve high efficiency at
loads under 100µA as shown in Figure 5.
20
18
60
30
PGOOD
COMPARATOR
12
70
VOUT
2
D1, D0
VIN = 5V
80
0
BANDGAP
REFERENCE
VIN = 18V, LEAKAGE AT PZ1 OR PZ2
16
9
BRIDGE LEAKAGE (nA)
90
GND
SLEEP
RECTIFIED PIEZO VOLTAGE (V)
100
BUCK
CONTROL
UVLO
INCREASING
VIBRATION ENERGY
6
3
14
12
10
8
6
4
2
0
0
10
20
PIEZO CURRENT (µA)
30
Figure 6. Typical piezoelectric load lines for Piezo
Systems T220-A4-503X
0
–55
–10
35
80
125
TEMPERATURE (°C)
170
Figure 7. The internal bridge rectifier outperforms
discrete solutions
April 2010 : LT Journal of Analog Innovation | 5
Though an energy harvesting system can eliminate the need
for batteries, it can also supplement a battery solution.
Figure 9. Piezo energy
harvester with battery
backup
Ambient vibrations can be characterized
in order to select a piezo with optimal
characteristics. The frequency and force
of the vibration as well as the desired
interval between use of the LTC3588-1’s
output capacitor reservoir and the amount
of energy required at each burst can
help to determine the best piezoelectric
element. In this way, a system can be
designed so that it performs its task as
often as the amount of available energy
allows. In some cases, optimization of the
piezoelectric element may not be necessary as just the capability to harvest any
amount of energy may be attractive.
MIDE V21BL
IR05H40CSPTR
1µF
6V
9V
BATTERY
100µF
16V
PZ2
VIN
PGOOD
CAP
LTC3588-1
4.7µF
6V
Piezoelectric elements convert mechanical energy, typically vibration energy, into
electrical energy. Piezoelectric elements
can be made out of PZT (lead zirconate
titanate) ceramics, PVDF (polyvinylidene
fluoride) or other composites. Ceramic
piezoelectric elements exhibit a piezoelectric effect when the crystal structure of
the ceramic is compressed and internal
dipole movement produces a voltage.
Polymer elements comprised of long-chain
molecules produce a voltage when flexed
as molecules repel each other. Ceramics
are often used under direct pressure while
a polymer can be flexed more readily.
A wide range of piezoelectric elements are
available and produce a variety of open
circuit voltages and short-circuit currents.
The open circuit voltage and short-circuit
current form a “load line” for the piezoelectric element that increases with available vibration energy as shown in Figure 6.
The LTC3588-1 can handle up to 20V at its
input, at which point a protective shunt
safeguards against an overvoltage condition on VIN. If ample ambient vibration
causes a piezoelectric element to produce
D1
D0
PGOOD
10µH
SW
VOUT
VIN2
REAPING VIBRATION ENERGY
6 | April 2010 : LT Journal of Analog Innovation
PZ1
PZ1
GND
VOUT
3.3V
47µF
6V
PZ2
more energy than the LTC3588-1 needs, the
shunt consumes the excess power, effectively clamping the piezo on its load line.
The LTC3588-1 interfaces with the piezo
through its internal low loss bridge rectifier accessible via the PZ1 and PZ2 pins.
The rectified output is stored on the
VIN capacitor. At typical 10µA piezoelectric currents, the voltage drop associated
with the bridge rectifier is on the order
of 400mV. The bridge rectifier also suits a
variety of other input sources by featuring less than 1nA of reverse leakage at
125°C (Figure 7), a bandwidth greater
than 1MHz and ability to carry 50mA.
Figure 10. Electric field
energy harvester
COPPER PANEL
(12in × 24in)
1µF
6V
10µF
25V
OPTIONS FOR ENERGY STORAGE
Harvested energy can be stored on the
input capacitor or the output capacitor.
The wide input range takes advantage of
the fact that energy storage on a capacitor is proportional to the square of the
capacitor voltage. After the output voltage
is brought into regulation any excess
energy is stored on the input capacitor
and its voltage increases. When a load
exists at the output, the buck can efficiently transfer energy stored at a high
voltage to the regulated output. While
PANELS ARE PLACED 6"
FROM 2' × 4' FLUORESCENT
LIGHT FIXTURES
PZ1
PZ2
VIN
PGOOD
CAP
LTC3588-1
4.7µF
6V
D1
D0
GND
PGOOD
10µH
SW
VOUT
VIN2
COPPER PANEL
(12in × 24in)
3.3V
10µF
6V
design features
The LTC3588-1 can harvest other sources of energy
besides the ambient vibration energy available from a
piezoelectric element. The integrated bridge rectifier allows
many other AC sources to power the LTC3588-1.
Figure 11. AC line powered
3.6V buck regulator
DANGER! HIGH VOLTAGE!
150k
150k
120VAC
60Hz 150k
150k
1µF
6V
10µF
25V
DANGEROUS AND LETHAL POTENTIALS ARE PRESENT IN OFFLINE CIRCUITS!
PZ1
PZ2
VIN
PGOOD
CAP
LTC3588-1
PGOOD
10µH
VOUT
3.6V
SW
VOUT
VIN2
4.7µF
6V
Before proceeding any further, the reader is warned that caution must be used in
the construction, testing and use of offline circuits. Extreme caution must be
used in working with and making connections to these circuits.
100µF
6V
D1
D0
REPEAT: Offline circuits contain dangerous, AC line-connected high voltage
potentials. Use caution. All testing performed on an offline circuit must be done
with an isolation transformer connected between the offline circuit’s input and
the AC line. Users and constructors of offline circuits must observe this
precaution when connecting test equipment to the circuit to avoid electric
shock.
REPEAT: An isolation transformer must be connected between the circuit input
and the AC line if any test equipment is to be connected.
GND
energy storage at the input utilizes the
high voltage at the input, the load current is limited to the 100mA the buck
converter can supply. If larger transient
loads need to be serviced, the output
capacitor can be sized to support a larger
current for the duration of the transient.
output capacitor until PGOOD went low.
In some cases, using every last joule is
important and the PGOOD pin will remain
high if the output is still within 92% of
the regulation point, even if the input
falls below the lower UVLO threshold
(as might happen if vibrations cease).
A PGOOD output exists that can help
with power management. PGOOD transitions high (referred to VOUT) the first time
the output reaches regulation and stays
high until the output falls to 92% of the
regulation point. PGOOD can be used to
trigger a system load. For example, a current burst could begin when PGOOD goes
high and would continuously deplete the
THE LTC3588-1 EXTENDS
BATTERY LIFE
The battery backup circuit in Figure 9
shows a 9V battery with a series blocking
diode connected to VIN. The piezo charges
VIN through the internal bridge rectifier
and the blocking diode prevents reverse
current from flowing into the battery.
A 9V battery is shown, but any stack of
batteries of a given chemistry can be used
as long as the battery stack voltage does
Though an energy harvesting system can
eliminate the need for batteries, it can also
supplement a battery solution. The system
can be configured such that when ambient
energy is available, the battery is unloaded,
but when the ambient source disappears, the battery engages and serves as
the backup power supply. This approach
300Ω
Figure 12. Solar panel
powering the LTC3588-1
not only improves reliability, but it can
also lead to a more responsive system.
For example, an energy harvesting sensor
node placed on a mobile asset, such as a
tractor trailer, may gather energy when the
trailer is on the road. When the truck is
parked and there is no vibration, a battery
backup still allows polling of the asset.
PZ2
PZ1
IR05H4OCSPTR
+
–
1µF
6V
5V TO 16V
SOLAR PANEL
9V
BATTERY
100µF
25V
4.7µF
6V
PGOOD
VIN
LTC3588-1
PGOOD
10µH
CAP
SW
VIN2
VOUT
+
D0
D1
GND
VOUT
2.5V
10µF
6V
3F
2.7V
NESS SUPER CAPACITOR
ESHSR-0003CO-002R7
April 2010 : LT Journal of Analog Innovation | 7
Not limited to AC sources, DC sources such
as solar panels and thermoelectric couplers
can be used to power the LTC3588-1.
Figure 13. Dual rail power
supply with single piezo
MIDE V25W
PGOOD1
3.6V
PZ1
PZ2
PZ1
PZ2
PGOOD
VIN
VIN
PGOOD
10µH
SW
10µF
6V
LTC3588-1
VOUT
D1
not exceed 18V, the maximum voltage
that can be applied to VIN by an external
low impedance source. When designing a
battery backup system, the piezoelectric
transducer and battery should be chosen
such that the peak piezo voltage exceeds
the battery voltage. This allows the piezo
to “take over” and power the LTC3588-1.
A WEALTH OF ALTERNATIVE
ENERGY SOLUTIONS
The LTC3588-1 can harvest other sources
of energy besides the ambient vibration
energy available from a piezoelectric
element. The integrated bridge rectifier
allows many other AC sources to power the
LTC3588-1. For example, the fluorescent
light energy harvester shown in Figure 10
capacitively harvests the alternating
electric field radiated by an AC powered
fluorescent light tube. Copper panels can
be placed above the light tube on the light
fixture to harness the energy from the
electric field produced by the light tube
and feed that energy to the LTC3588-1
and the integrated bridge rectifier. Such
a harvester can be used throughout
buildings to power HVAC sensor nodes.
D0
1µF
6V
10µF
25V
VIN2
GND
8 | April 2010 : LT Journal of Analog Innovation
CAP
1µF
6V
4.7µF
6V
10µF
25V
CAP
LTC3588-1
D1
D0
Another useful application of the
LTC3588-1 involves powering the IC from
the AC line voltage with current limiting resistors as shown in Figure 11. This
offers a low cost, transformer-free solution for simple plug-in applications.
Appropriate UL guidelines should be
followed when designing circuits connecting directly to the line voltage.
Not limited to AC sources, DC sources
such as solar panels and thermoelectric couplers can be used to power the
LTC3588-1 as shown in Figure 12. Such
sources can connect to one of the PZ1/PZ2
inputs to utilize the reverse current protection that the bridge would provide.
They can also be diode-ORed together
to the VIN pin with external diodes. This
facilitates the use of multiple solar panels
aimed in different directions to catch the
sun at different times during the day.
MULTIPLE OUTPUT RAILS
SHARE A SINGLE PIEZO SOURCE
Many systems require multiple rails to
power different components. A microprocessor may use 1.8V but a wireless transmitter may need 3.6V. Two
LTC3588-1 devices can be connected to
10µH
SW
VOUT
VIN2
4.7µF
6V
PGOOD2
1.8V
10µF
6V
GND
one piezoelectric element and simultaneously provide power to each output as
shown in Figure 13. This setup features
automatic supply sequencing as the
LTC3588-1 with the lower voltage output (i.e., lower UVLO rising threshold)
comes up first. As the piezo provides
input power both VIN rails initially come
up together, but when one output starts
drawing power, only its corresponding
VIN falls as the bridges of each LTC3588-1
provide isolation. Input piezo energy
is then directed to this lower voltage
capacitor until both VIN rails are again
equal. This configuration is expandable
to multiple LTC3588 devices powered
by a single piezo as long as the piezo
can support the sum total of the quiescent currents from each LTC3588-1.
CONCLUSION
The LTC3588-1 provides a unique power
solution for emerging wireless sensor
technologies. With extremely low quiescent current and an efficient energy
harvesting solution it makes distributed
sensor networks easier to deploy. Sensors
can now be used in remote locations
without worrying about battery life. n
design features
Dual Output Step-Down Regulator Features Pin Selectable
Outputs, DCR Sensing, Reverse Current Protection and a
5mm × 5mm QFN
Stephanie Dai
The LTC3865 is a high performance step-down controller with two constant frequency,
current mode, synchronous buck controllers and on-chip drivers. It offers high output
current capability over a wide input range. An important feature offered by the LTC3865 is its
highly accurate programmable output voltage. Internal precision feedback resistors make it
possible to select from nine different output voltages via two VID pins. The internal resistors
reduce the number of external components and assure 1% accuracy for low voltage rails.
The LTC3865 is suitable for applications
with input voltages up to 38V and output
voltages up to 5V. It can be synchronized
to a frequency of up to 750kHz and comes
in a compact 5mm × 5mm QFN package. The part is also capable of inductor DCR sensing, allowing for increased
efficiencies at higher load currents. These
features are ideal for wide input voltage
range, high current applications where
design solution footprint size is restricted.
them floating can result in nine different output voltages from 0.6V to 5V (see
Table 1). Pin programming eliminates
at least four external feedback resistors, making the overall design solution
space conservative and cost effective. If
an application requires an output voltage not supported by pin programming,
one still has the option to use external
resistors to set the output voltage. Since
the LTC3865 integrates precision feedback resistors, it can achieve 1% output
accuracy for outputs from 0.6V to 1.8V and
1.5% percent output accuracy for 2.5V to
5V, with this accuracy maintained over
the temperature range of −40°C to 85°C.
R SENSE AND DCR SENSING
VID11/VID21
VID21/VID22
V OUT1 /V OUT2 (V)
For applications requiring the highest
possible efficiency at high load currents,
a sense resistor would sacrifice several
percentage points of efficiency compared
to DCR sensing. Inductor DCR is a manifestation of the inductor’s copper winding
resistance. In high current applications,
typical inductance values are low, allowing for a high saturation current inductor
with sub 1mΩ DCR values. DCR current
sensing takes advantage of this by sensing
the voltage drop across the low copper
DCR to monitor the inductor current.
This eliminates the sense resistor and its
additional power loss, thus increasing
efficiency as well as lowering solution
size and cost. An example of a DCR sensing application is shown in Figure 1.
Figure 2 shows an efficiency comparison.
INTV CC
INTV CC
5.0
MULTIPHASE OPERATION
INTV CC
Float
3.3
INTV CC
GND
2.5
Float
INTV CC
1.8
Float
Float
0.6 or External Divider
Float
GND
1.5
GND
INTV CC
1.2
The LTC3865 operates both of channels 180° out-of-phase. This reduces the
required input capacitance and power
supply induced noise. With its current
mode architecture, it can be configured
for dual outputs, or for one output
with both power stages tied together.
GND
Float
1.0
GND
GND
1.1
PIN SELECTABLE OUTPUTS
The LTC3865 features pin programmable
output voltages. Tying each channel’s
two VID pins to INTVCC, GND or leaving
Table 1. Programming the Output Voltages
A dual-phase single output application
is easy to configure; just tie the channels’ compensation (ITH), feedback (VFB),
April 2010 : LT Journal of Analog Innovation | 9
The LTC3865 step-down regulator is ideal for
applications requiring inductor DCR sensing for
maximum efficiency under heavy load.
VIN
7V TO 20V
Figure 1. DCR sensing application
1µF
2.2Ω
4.7µF
D3
M1
0.1µF
L1
3.3µH
VIN PGOOD EXTVCC INTVCC
TG1
TG2
BOOST1
SW1
BG1
5.49k
1%
10µF
35V
×2
LTC3865
COUT1
100µF
×2
FREQ
RUN1
TK/SS1
100pF
3.65k
1%
SENSE2+
1000pF
RUN2
SENSE1–
VID11
VID12
VOSENSE1
ITH1
1800pF
4.75k
1%
L2
2.2µH
PGND
SENSE1+
VOUT1
3.3V
5A
M2
0.1µF
BG2
ILIM
1000pF
D4
BOOST2
SW2
MODE/PLLIN
1.37k
1%
22µF
50V
SENSE2–
VID21
VID22
VOSENSE2
ITH2
TK/SS2
SGND
0.1µF
0.1µF
1.58k
1%
VOUT2
1.8V
5A
2200pF
162k
1%
5.49k
1%
100pF
COUT2
100µF
×2
L1, L2: COILTRONICS HCP0703
M1, M2: VISHAY SILICONIX Si4816BDY
COUT1, COUT2: TAIYO YUDEN JMK325BJ107MM
D3, D4: CMDSH-3
The traditional way of protecting an
IC against overvoltage conditions is to use
an overvoltage comparator, which guards
against transient overshoots (>10%) as
well as other more serious conditions that
10 | April 2010 : LT Journal of Analog Innovation
100
10
90
EFFICIENCY
EFFICIENCY (%)
OUTPUT OVERVOLTAGE
PROTECTION WITH A NEGATIVE
REVERSE CURRENT LIMIT
may cause the output voltage to overshoot.
In such cases, the top MOSFET is turned off
and the bottom MOSFET is turned on and
kept on until excessive energy has been
1
80
70
60
POWER LOSS
50
40
0.01
0.1
DCR
8mΩ
0.1
1
LOAD CURRENT (mA)
10
0.01
Figure 2. Efficiency for the circuit in Figure 1
POWER LOSS (mW)
enable (RUN), power good (PGOOD) and
track/soft-start (TRK/SS) pins together. By
doubling the effective switching frequency
and interleaving phases, the single output configuration minimizes the required
input and output capacitance and voltage ripple, and allows for a fast transient
response and increased current capability. An example for a dual-phase single
output application is shown in Figure 3.
discharged from the output capacitor,
bringing the output back to regulation.
One problem with turning on the bottom
MOSFET indefinitely to clear an overvoltage condition is that sometimes excessive
reverse current is required to discharge the
output capacitor. In these cases the bottom
FET experiences extreme current stress. To
avoid this scenario, the LTC3865 adds a
–53mV of reverse current limit. By setting a floor on how much reverse current
is allowed, the LTC3865 limits how long
the bottom FET can be turned on. This
feature is important in applications that
reprogram output voltages on the fly. For
example, if the output voltage is changed
from 1.8V to 1.5V, the reverse current limit
is activated as shown in the Figure 4.
design features
The LTC3865’s VID programmable output voltage
decreases parts count while increasing design flexibility.
10µF
35V
4.7µF
RJK0305DPB
VIN PGOOD EXTVCC INTVCC
TG1
0.1µF
L1
0.47µH
1µF
2.2Ω
D3
BOOST1
SW1
RJK0330DPB
BG1
TG2
LTC3865
MODE/PLLIN
ILIM
100Ω
2mΩ
100Ω
1.21k
1%
RUN1
TK/SS1
BG2
D4
RJK0305DPB
0.1µF
L2
0.47µH
RJK0330DPB
VOUT
500mV/DIV
1.8V TO 1.5V
PGND
100Ω
SENSE2+
1000pF
100Ω
RUN2
SENSE1–
VID11
VID12
VOSENSE1
ITH1
6800pF
COUT1
220µF
BOOST2
SW2
SENSE2–
VID21
VID22
VOSENSE2
ITH2
SGND
2mΩ
IL
5A/DIV
1.2V
30A
TK/SS2
COUT2
220µF
162k
100pF
VIN
7V TO 20V
22µF
50V
FREQ
SENSE1+
1000pF
10µF
35V
0.1µF
COUT1, COUT2: SANYO 4TPE 220µF
L1, L2: VISHAY IHLP4040DZERR47M11
D3, D4: CMDSH-3
RUN1
0Ω
RUN2
Figure 3. Single output application
FREQUENCY SELECTION
AND MODE/PLLIN
To maximize efficiency at light loads,
the LTC3865 can be set for automatic
Burst Mode® operation. Alternately, to
minimize noise at the expense of light
load efficiency, it can be set to operate in
forced continuous conduction mode. For
both relatively high efficiency and low
noise operation, it can be set to operate
with a hybrid of the two, namely pulseskipping mode. Pulse-skipping mode,
like forced continuous mode, exhibits
lower output ripple as well as low audio
noise and reduced RF interference as
compared to Burst Mode operation. It
also improves light load efficiency, but
not as much as Burst Mode operation.
A clock on the MODE/PLLIN pin forces the
controller into forced continuous mode
and synchronizes the internal oscillator
with the clock on this pin. The phaselocked loop integrated at this pin is composed of an internal voltage-controlled
oscillator and a phase detector. This allows
the turn-on of the top MOSFET of controller 1 to be locked to the rising edge of
an external clock signal applied to the
MODE/PLLIN pin. The frequency range for
the LTC3865 is from 250kHz to 750kHz.
If no external synchronization signal is
applied, there is a precision 7.5µA current
flow out of the FREQ pin that can be used
to program the operating frequency of the
LTC3865 from 250kHz to 750kHz through
a single resistor from the pin to SGND.
1ms/DIV
REVERSE CURRENT LIMIT = –53mV
RSENSE = 10mΩ
Figure 4. As VOUT transitions from 1.8V to 1.5V, with
–53mV reverse current limit and 10mΩ sense resistor, the reverse inductor current is limited at around
5.3A.
CONCLUSION
The LTC3865 step-down regulator is
ideal for applications requiring inductor DCR sensing for maximum efficiency
under heavy load. It can regulate two
separate outputs and can be configured
for higher load current capability by
tying its channels together, and/or by
paralleling additional LTC3865 power
stages. The LTC3865’s VID programmable
output voltage decreases parts count
while increasing design flexibility. These
features, along with its additional negative reverse current limit and integrated
PLL features, make the LTC3865 an easy
fit in a wide variety of applications. n
April 2010 : LT Journal of Analog Innovation | 11
Tiny, Accurate Battery Gas Gauges with Easy-to-Use I2C
Interface and Optional Integrated Precision Sense Resistors
Christoph Schwoerer, Axel Klein and Bernhard Engl
Imagine your daughter is catching her first wave on a
surfboard and your video camera shuts down because the
battery—reading half full when you started filming a few
minutes ago—is suddenly empty. The problem is inaccurate
battery gas gauging. Inaccurate fuel gauging is a common
nuisance as many portable devices derive remaining
battery capacity directly from battery voltage. This method
is cheap but inaccurate since the relationship between
battery voltage and capacity has a complex dependency
on temperature, load conditions and usage history.
More accurate battery gauging can be
achieved by monitoring not only the
battery voltage but also by tracking the
charge that goes in and out of the battery. For applications requiring accurate
battery gas gauging, the LTC2941 and
LTC2942 coulomb counters are tiny and
easy-to-use solutions. These feature-rich
devices are small and integrated enough to
fit easily into the latest handheld gadgets.
The LTC2941 is a battery gas gauge device
designed for use with single Li-Ion cells
and other battery types with terminal
voltages between 2.7V and 5.5V. A precision coulomb counter integrates current
through a sense resistor between the
battery’s positive terminal and the load
or charger. The high side sense resistor
avoids splitting the ground path in the
application. The state of charge is continuously updated in an accumulated
charge register (ACR) that can be read out
via an SMBus/I2C interface. The LTC2941
also features programmable high and
low thresholds for accumulated charge.
If a threshold is exceeded, the device
12 | April 2010 : LT Journal of Analog Innovation
ANALOG INTEGRATOR ALLOWS
PRECISE COULOMB COUNTING
Charge is the time integral of current. The LTC2941 and LTC2942 use
a continuous time analog integrator
to determine charge from the voltage drop developed across the sense
resistor RSENSE as shown in Figure 2.
The differential voltage between SENSE+
and SENSE– is applied to an autozeroed
differential integrator to convert the
measured current to charge, as shown
in Figure 2. When the integrator output
ramps to REFHI or REFLO levels, switches
S1, S2, S3 and S4 toggle to reverse the ramp
direction. By observing the condition
of the switches and the ramp direction,
polarity is determined. A programmable
prescaler adjusts integration time to
match the capacity of the battery. At each
underflow or overflow of the prescaler,
the ACR value is incremented or decremented one count. The value of accumulated charge is read via the I2C interface.
communicates an alert using either
the SMBus alert protocol or by setting
a flag in the internal status register.
The LTC2942 adds an ADC to the coulomb
counter functionality of the LTC2941. The
ADC measures battery voltage and chip
temperature and provides programmable
thresholds for these quantities as well.
The LTC2941 and LTC2942 are pin compatible and come in tiny 6-pin 2mm × 3mm
DFN packages. Each consumes only
75µA in normal operation. Figure 1
shows the LTC2942 monitoring the charge
status of a single cell Li-ion battery.
500mA
VIN
5V
BAT
VCC
1µF
Figure 1. Monitoring
the charge status of a
single-cell Li-Ion battery with the LTC2942
The use of an analog integrator distinguishes the LTC coulomb counters from
most other gas gauges available on the
2k
LTC4057-4.2
(CHARGER)
PROG SHDN
GND
2
LOAD
0.1µF
3.3V
2k
VDD
µP
2k
2k
SENSE+
LTC2942
AL/CC
SDA
SENSE–
SCL
GND
RSENSE
100mΩ
+
1-CELL
Li-Ion
design features
Figure 2. Coulomb counter
section of LTC2942
LTC2942
LOAD
1
SENSE+
S1
2
market. It is common to use an ADC to
periodically sample the voltage drop over
the sense resistor and digitally integrate
the sampled values over time. This implementation has two major drawbacks.
First, any current spikes occurring inbetween sampling instants is lost, which
leads to rather poor accuracy—especially in applications with pulsed loads.
Second, the digital integration limits the
precision to the accuracy of the available
time base—typically low if not provided
by additional external components.
In contrast, the coulomb counter of
the LTC2941 and LTC2942 achieves an
accuracy of better than 1% over a wide
range of input signals, battery voltages
and temperatures without such external
components, as shown in Figures 3 and 4.
SENSE–
ACR
+
S4
REFLO
GND
POLARITY
DETECTION
–
TEMPERATURE AND VOLTAGE
MEASUREMENT
gral nonlinearity of the ADC is typically
below 0.5LSB as shown in Figure 6.
The LTC2942 includes a 14-bit No Latency
∆Σ™ analog-to-digital converter with internal clock and voltage reference circuits to
measure battery voltage. The integrated
reference circuit has a temperature coefficient typically less than 20ppm/°C, giving
an ADC gain error of less than 0.3% from
–45°C to 85°C (see Figure 5). The inte-
The ADC is also used to read the output
of the on-chip temperature sensor. The
sensor generates a voltage proportional
to temperature with a slope of 2.5mV/°C,
resulting in a voltage of 750mV at 27°C.
The total temperature error is typically below ±2°C as shown in Figure 7.
3
1.00
0.75
2
1
0
–1
–2
INTEGRATED SENSE RESISTOR
VERSIONS LTC2941-1 AND LTC2942-1
1
10
0.50
0.25
0
–0.25
–0.50
–0.75
VSENSE+ = 2.7V
VSENSE+ = 4.2V
–3
0.1
100
–1.00
–50
VSENSE = –50mV
VSENSE = –10mV
VSENSE (mV)
25
0
50
TEMPERATURE (°C)
Figure 3. Charge error vs VSENSE
Figure 4. Charge error vs temperature
10
–25
75
100
1.0
8
TOTAL UNADJUSTED ERROR (mV)
The accuracy of the charge monitoring
depends not only on the accuracy of the
chosen battery gas gauge but also on
the precision of the sense resistor. The
LTC2941-1 and LTC2942-1 remove the need
for a high precision external resistor by
including an internal, factory trimmed
50mΩ sense resistor. Proprietary internal
circuitry compensates the temperature
coefficient of the integrated metal resistor
to a residual error of only 50ppm/°C which
makes the LTC2941-1 and LTC2942-1 by
far the most precise internal sense resistor battery gas gauges available today.
M
PRESCALER
CHARGE ERROR (%)
6
+
CONTROL
LOGIC
–
+
S3
CHARGE ERROR (%)
BATTERY
–
+
S2
RSENSE
IBAT
REFHI
VCC
TA = 85°C
6
0.5
TA = 85°C
4
2
0
INL (VLSB)
CHARGER
TA = –45°C
–2
–4
–6
0
TA = –40°C
TA = 25°C
–0.5
TA = 25°C
–8
–10
2.5
3.0
3.5
4.0 4.5
5.0
VSENSE– (V)
5.5
6.0
Figure 5. ADC total unadjusted error
–1.0
2.5
3.0
3.5
4.0 4.5 5.0
VSENSE– (V)
5.5
6.0
Figure 6. ADC integrated nonlinearity
April 2010 : LT Journal of Analog Innovation | 13
USB CHARGING
Figure 9 shows a portable application
designed to charge a Li-Ion battery
from a USB connection. The LTC2942-1
monitors the charge status of a singlecell Li-Ion battery in combination
with the LTC4088-1 high efficiency battery charger/USB power manager.
Once a charge cycle is completed, the
LTC4088-1 releases the CHRG pin. The
microcontroller detects this and sets the
accumulated charge register to full either
by writing it via the I2C interface or by
applying a pulse to the charge complete (AL/CC) pin of the LTC2942-1 (if it
is configured as input). Once initialized,
the LTC2942-1 accurately monitors the
3
MONITORING BATTERY STACKS
The LTC2941 and LTC2942 are not
restricted to single cell Li-Ion applications.
They can also monitor the charge state of
a battery stack as shown in Figure 11.
2
TEMPERATURE ERROR (°C)
Conversion of either temperature or
voltage is triggered by setting the control
register via the I2C interface. The LTC2942
also features an optional automatic
mode where a voltage and a temperature
conversion are executed once a second.
At the end of each conversion the corresponding registers are updated before
the converter goes to sleep to minimize
quiescent current. Figure 8 shows a block
diagram of the LTC2942, with the coulomb counter and its ACR, the temperature
sensor, the ADC with the corresponding
data registers and the I2C interface.
1
0
In this configuration, the power consumption of the gas gauge might lead
to an unacceptable imbalance between
the lower and the upper Li-Ion cells
in the stack. This imbalance can be
eliminated by supervising every cell
individually, as shown in Figure 12.
–1
–2
–3
–50
–25
0
25
50
TEMPERATURE (°C)
75
100
Figure 7. Temperature sensor error
charge flowing in and out of the battery,
and the microcontroller can monitor the
state of charge by reading the accumulated charge register via the I2C interface.
By monitoring each cell’s state of
charge, the LTC2942-1 provides enough
information to balance the cells
while charging and discharging.
ENERGY MONITORING
TRACKING BATTERY CAPACITY WITH
TEMPERATURE AND AGING
Real time energy monitoring is increasingly used in non-portable, wall powered
systems such as servers or networking
equipment. The LTC2941 and LTC2942
are just as well suited to monitor energy
flow in any 3.3V or 5V rail application
as they are in battery powered applications. Figure 10 shows an example.
Battery capacity varies with temperature
and aging. There is a wide variety of
approaches and algorithms to track the
battery capacity tailored to specific applications and the chemistry of the battery.
The LTC2942 measures all physical quantities—charge, voltage, temperature and (by
differentiating charge) current—necessary
to model the effects of temperature and
aging on battery capacity. The measured
quantities are easily accessible by reading
the corresponding registers with standard
With a constant supply voltage, the
charge flowing through the sense resistor is proportional to the energy consumed by the load. Thus several LTC2941
devices can help determine exactly
where system energy is consumed.
Figure 8. Block diagram
of the LTC2942
1
SENSE+
VSUPPLY
COULOMB COUNTER
REF
TEMPERATURE
SENSOR
6
2
SENSE–
MUX
CC
CLK
REFERENCE
GENERATOR
OSCILLATOR
REF+
CLK
IN
ACCUMULATED
CHARGE
REGISTER
AL
I2C/
SMBus
AL/CC
SCL
SDA
ADC
DATA AND
CONTROL
REGISTERS
REF–
GND
LTC2942
14 | April 2010 : LT Journal of Analog Innovation
5
3
4
design features
I2C commands. No special instruction
language or programming is required.
3.3uH
The LTC2941 and LTC2942 do not impose
a particular approach or algorithm to
determine battery capacity from the
measured quantities, but instead allow
the system designer to implement algorithms tailored to the special needs of
the system via the host controller. The
microcontroller can then simply adapt
the charge thresholds of LTC2941 and
LTC2942 based on these calculations.
SW
VBUS
USB
10µF
VOUT
LOAD
10µF
VOUTS
GATE
BAT
LT4088-1
0.1µF
3.3V
D0
VDD
D1
SENSE+
2k
2k
8.2k
0.1µF
PROG C/X
+
1 CELL
LI-ION
AL/CC
SDA
µP
CHRG
_
LTC2942-1
D2
CLPROG
SENSE
GND NTC
SCL
GND
2.94k
2k
CONCLUSION
Battery gas gauging is one feature that lags
behind other technological improvements
in many portable electronics. The LTC2941
and LTC2942 integrated coulomb counters
solve this problem with accurate battery
gas gauging that is easy to implement and
fit into the latest portable applications.
For high accuracy in the tightest spots,
the LTC2941-1 and LTC2942-1 versions
integrate factory trimmed and temperature compensated sense resistors for the
ultimate small coulomb counters. This new
family of accurate coulomb counters can
help prevent you from ever again missing
those priceless vacation moments due to
inaccurate battery charge monitoring. n
Figure 11. Using the
LTC2942 in a battery
stack
CHARGER
Figure 9. Battery gas gauge with USB charger
TO WALL
ADAPTER
3.3V OUT
GND
SCL
_
SENSE
GND
SENSE+ SENSE
_
2k
2k
2k
2k
2k
2k
LTC2941
VDD
AL/CC
SDA
0.1µF
SENSE+ SENSE
_
GND SCL
µP
AL/CC
SDA
GND SCL
Figure 10. Monitoring system energy flow using LTC2941s at the loads
+
SENSE+
I2C/SMBus
TO HOST
3.3V
5A LOAD
LTC2941
+
AL/CC
SDA
0.1µF
DC/DC
5V
10A LOAD
5mΩ
5V OUT
10mΩ
+
LTC2942
VIN
LOAD
1 CELL
LI-ION
1 CELL
LI-ION
1 CELL
LI-ION
Figure 12. Individual
cell monitoring of a
2-cell stack
CHARGER
LTC2942-1
SENSE+
I2C/SMBus
TO HOST
RSENSE
+
LOAD
LEVEL–
SHIFT
AL/CC
SDA
SCL
1 CELL
LI-ION
SENSE
GND
_
+
1 CELL
LI-ION
LTC2942-1
SENSE+
I2C/SMBus
TO HOST
AL/CC
SDA
SCL
SENSE
GND
_
+
1 CELL
LI-ION
April 2010 : LT Journal of Analog Innovation | 15
140W Monolithic Switching Regulator Simplifies
Constant-Current/Constant-Voltage Regulation
Eric Young
The LT3956 is a monolithic switching regulator that can
generate constant-current/constant-voltage outputs in buck,
boost or SEPIC topologies over a wide range of input and
output voltages. With input and output voltages of up to 80V,
a rugged internal 84V switch and high efficiency operation, the
LT3956 can easily produce high power in a small footprint.
The LT®3956 combines key amplifier and
comparator blocks with a high current/
high voltage switching regulator in a
tiny 5mm × 6mm package. See Figure 1
for an example of how little board space
is needed to produce a complete constant-current, constant-voltage boost
circuit ideal for LED driving, supercap
charging or other high power applications that require the added protection
of input or output current limiting.
WHAT MAKES THE LT3956 TICK?
The big mover in the LT3956 is an
84V-rated, 90mΩ low side N-MOSFET switch
with an internally programmed current
limit of 3.9A (typ). The switching regulator can be powered from a supply as
high as 80V because the N-MOSFET switch
driver, the PWMOUT pin driver, and most
internal loads are powered by an internal LDO linear regulator that converts
VIN to 7.15V, provided the VIN supply is
high enough. The switch duty cycle and
current is controlled by a current-mode
pulse-width modulator—an architecture
that provides fast transient response, fixed
switching frequency operation and an
easily stabilized feedback loop at variable
inputs and outputs. The switching frequency can be programmed from 100kHz
to 1MHz with an external resistor, which
allows designers to optimize component
16 | April 2010 : LT Journal of Analog Innovation
size and performance parameters, such
as min/max duty cycle and efficiency.
And at the heart of the LT3956 is a dual
input feedback transconductance (gm)
amplifier that combines a differential
constant current sense with a standard
low side voltage feedback. The handoff
between these two loops is seamless and
predictable. The feedback loop operating closest to its set point is auto-selected
to be the loop controlling the flow of
charge onto the compensation R-C network attached to the VC pin. The voltage
Figure 1. Complete high power, constant current,
constant voltage boost circuit
level at the VC pin in turn controls the
current and duty ratio of the switch. A
more thorough description of operation
can be found in the LT3956 data sheet.
C1
2.2µF
!2
R1
332k
R2
100k
INTVCC
VIN
0.33Ω
INTVCC
ISP
R3
1M
LT3956
CTRL
1k
Q2
Q3
20k
2k
D2
ISN
FB
100k
VMODE
PWM
SS
RT
VC
0.01µF
PGND
VREF
R6
57.6k
28.7k
375kHz
C2
2.2µF
!10
SW
EN/UVLO
R5
1M
750mA
D1
22µH
VIN
6V TO 60V
(80V TRANSIENT)
200Ω
6.8nF
Figure 2. This 50W boost LED driver provides wide
input range, PWM dimming and LED fault protection
and reporting.
M1
R4
16.2k
LED+
Q1
PWMOUT
GND INTVCC
INTVCC
4.7µF
1k
18 WHITE
LEDs
50W
(DERATED IF
VIN < 22V)
M1: VISHAY SILICONIX Si7113DN
D1: DIODES INC PDS5100
L1: COILTRONICS DR125-220
C1,C2: MURATA GRM42-2X7R225
Q1: ZETEX FMMT497
Q2,Q3: ZETEX FMMT589
D2: BAV116W
D3: DIODES INC B1100/B
D3
design features
The big mover in the LT3956 is an 84V-rated, 90mΩ low side N-MOSFET
switch with an internally programmed current limit of 3.9A (typ). The switching
regulator can be powered from a supply as high as 80V because the
N-MOSFET switch driver, the PWMOUT pin driver, and most internal loads are
powered by an internal LDO linear regulator that converts VIN to 7.15V.
100
0.8
VLED+
50V/DIV
CURRENT
EFFICIENCY (%)
0.6
EFFICIENCY
90
0.4
85
0.2
80
0
10
20
30
VIN (V)
40
50
OUTPUT CURRENT (A)
95
PWM
5V/DIV
60
SW
50V/DIV
ILED
2A/DIV
ILED
500mA/DIV
FB
5V/DIV
VIN = 24V
10µs/DIV
VIN = 24V
200ns/DIV
Figure 3. High 94% efficiency means less than 3W
dissipation in the converter shown in Figure 2.
Figure 4. Boost PWM dimming waveforms for 60V
of LEDs shows microsecond rise and fall times and
excellent constant current regulation even over
short intervals.
Figure 5. LED+ terminal of boost shorted to GND is
prevented from damaging switching components by
a novel circuit.
A RUGGED HIGH POWER BOOST LED
DRIVER
Analog Dimming
solution for the luminary—the LED cathode current can return on a common
GND. A scope photo of PWM dimming
waveform (Figure 4) shows sharp rise
and fall times, less than 200ns, and quick
stabilization of the current. Although
a low side N-MOSFET disconnect at the
cathode is the simpler and more obvious (and a bit faster) implementation
for this particular boost circuit using the
LT3956, the use of high side PWM disconnect is important to a boost protection strategy to be discussed below.
Figure 2 shows a 50W boost LED driver
that operates from a 24V input, showing off some of the unique capabilities of this product when applied as an
LED driver. This boost circuit tolerates
a wide input range—from 6V to 60V. At
the low end of this VIN range, the circuit
is prevented from operating too close to
switch current limit by scaling back the
programmed LED current as VIN declines—
set by the resistor divider (R5 and R6) on
the CTRL pin. Figure 3 shows efficiency
and LED current versus VIN. The high
efficiency (94%) means passive cooling of
the regulator is adequate for all but the
most extreme environmental conditions.
ANALOG AND PWM LED DIMMING
The LT3956 offers two high performance
dimming methods: analog dimming
via the CTRL pin and the ISP/ISN current
sense inputs, and PWM dimming through
the PWM input and PWMOUT output.
Analog dimming is achieved via the voltage at the CTRL pin. When the CTRL pin
is below 1.2V, it programs the current
sense threshold from zero to 250mV (typ)
with guaranteed accuracy of ±3.5% at
100mV. When CTRL is above 1.2V, the
current sense threshold is fixed at 250mV.
At CTRL = 100mV (typ), the current sense
threshold is set to zero. This built-in offset
is important to the feature if the CTRL pin
is driven by a resistor divider—a zero
programmed current can be reached with
a non-zero CTRL voltage. The CTRL pin
is high impedance so it can be driven
in a wide variety of configurations.
PWM Dimming
Pulse width modulation (PWM) of
LED current is the preferred technique
to achieve wide range dimming of the
light output. Figure 2 shows a level shift
transistor Q1 driving a high side disconnect P-MOSFET M1. This configuration
allows PWM dimming with a single wire
CONSIDERATIONS FOR PROTECTING
THE LED, THE DRIVER, AND THE
INPUT POWER SUPPLY
LED systems often require load fault
detection. Limiting the output voltage in
the case of an open LED string has always
been a basic requirement and is achieved
through a resistor divider (R3 and R4)
at the FB input. If the string opens, the
switching regulator regulates VFB to a
constant 1.25V (typ). In addition to the
gm amplifier that provides this constant
April 2010 : LT Journal of Analog Innovation | 17
The boost circuit in Figure 2 uses the
voltage feedback (FB) input in a unique
fashion—protecting the LED+ node from
a fault to GND while preserving all the
other desirable attributes of the LED driver.
A standard boost circuit has a direct
path from the supply to the output, and
therefore cannot survive a GND fault on
its output when the supply current is not
limited. There are a number of situations
where one might desire to protect the
switching regulator from a short to GND of
the LED anode—perhaps the luminary
is separated from the driver circuit by
a connector or by a long wire, and the
input supply is a high capacity battery.
The LT3956 has a feature to provide this
protection. The overvoltage FB (OVFB)
comparator is a second comparator on
the FB input with a setpoint higher than
the VFB regulation voltage. It causes
the PWMOUT pin to transition low and
switching to stop immediately when
the FB input exceeds 1.31V(typ).
18 | April 2010 : LT Journal of Analog Innovation
VIN
OVLO
53V RISING
51V FALLING
VIN
EN/UVLO
10k
R1
1M
INTVCC
4.7µF
LT3956
Q1
R2
143k
PWM
PWMOUT
GND
BAV116
PWM
332k
by the resistor divider from VIN) exceeds
6.5V (INTVCC minus a VBE). When PWM falls
below its threshold, PWMOUT goes low
as well. Hysteresis of ~2V is provided by
PWMOUT. Because of the high PWM threshold (0.85V minimum over temperature),
the blocking diode D1 can be added to
preserve the PWM dimming capability.
Q1: ZETEX FMMT593
Figure 6. VIN overvoltage circuit halts switching
and disconnects the load during high input voltage
transients.
The OVFB comparator can be used in
an output GND fault protection scheme
(patent pending) for the boost. The key
elements are the high side LED disconnect
P-MOSFET (M1) and its supporting driving
circuit responsive to the PWMOUT signal,
and the output GND fault sensing circuit
consisting of D2, Q2 and two resistors that
provide signal to the FB node. The circuit
works by sensing the current flowing
in D2 when the output is shorted, and
thereby triggering the OVFB comparator. In
response to the OVFB comparator, the high
side switch M1 is maintained in an offstate and the switching is stopped until the
fault condition is removed. Figure 5 shows
the current waveform in the M1 switch
during and output short circuit event.
Additional Considerations for
Protecting the LED
Some harsh operating environments produce transients on the input power supply
that can overdrive a boosted output, if
only for a short while, and potentially
damage the LEDs with excessive current.
To discontinue switching and disconnect the LEDs during such a transient, a
simple add-on circuit to the PWM input,
shown as a breakout in Figure 6, disconnects the LED string and idles the switcher
when VIN exceeds 50V. The circuit works
by sourcing current to the PWM input
of the LT3956 from the collector of Q1
when VIN is low enough, but cutting off
that current when the base of Q1 (set
The LT3956 provides solutions to thermal dissipation problems encountered
driving LEDs. With high power comes
the concern about reduced lifetime of
the LED due to continuous operation at
high temperatures. Increasing numbers
of LED module applications implement
thermal sensing for the LED, usually
employing an NTC resistor coupled to
the LED heat sink with thermal grease.
A simple circuit employing the CTRL and
VREF pins of the LT3956 and an NTC resistor sensing the LED temperature produces a thermal derating curve for the
LED current as shown in Figure 7.
A CONSTANT-CURRENT/VOLTAGE
REGULATOR SERVES A WIDE RANGE
OF APPLICATIONS
Driving LEDs makes excellent use of
the LT3956 features, but it isn’t the
only application that requires constant
Figure 7. CTRL and VREF pins provide thermal derating to enhance LED reliability.
ISP
+
CTRL
ISN
100k
NTC
RT1
MURATA NCP18WM104
–
VREF
16.9k
LT3956
V(ISP-ISN)
300
V(ISP-ISN) THRESHOLD (mV)
voltage regulation, the FB input also has
two fixed setpoint comparators associated
with it. The lower setpoint comparator
activates the VMODE open collector pulldown when FB exceeds 1.20V (typ). After
the disconnection of the LED and loss of
the current regulation signal, the output
rises until it reaches the constant voltage
regulation setpoint. During this voltage
ramp the VMODE pin asserts and holds,
indicating that the LED load is open. This
signal maintains its state when PWM goes
low and the regulator stops switching,
allowing for the likelihood that output
voltage may fall below the threshold
without an occasional refresh provided by
switching. The VMODE pin quickly updates
when PWM goes high. The VMODE signal
can also indicate that the regulation mode
is transitioning from constant current to
constant voltage, which is the appropriate
function for current limited constant voltage applications, such as battery chargers.
250
200
150
100
50
0
25
45
65
85
TEMPERATURE (°C)
105
125
design features
voltage at constant current. It can be
used for charging batteries and supercapacitors, or driving a current source
load such as a thermoelectric cooler,
just to name a few examples. It can be
used as a voltage regulator with current limited input or output, or a current regulator with a voltage clamp.
intermittently, but the available power
might be limited based on an overall
system budget. The output charging rate
of the circuit of Figure 8 is not based on
any timer, but rather on the output voltage
level as sensed by the CTRL pin. Below a
certain output voltage, 22V in this case, the
input current is limited so that the switching regulator is maintained within its
own current limit. At higher output voltages, the default internal current sensing
threshold of 250mV (typ) establishes that
the input current cannot exceed 1.2A, and
so the output current drops. At very low
output voltages less than 1.5V, the network
driving the SS pin of LT3956 reduces the
switching frequency and the current limit
to maintain good control of the charging
current. When the load is within 5% of its
target voltage, the VMODE pin toggles to
indicate the end of constant current mode
and entry into constant voltage regulation.
Pursuing this line of thinking, Figure 8
shows a SEPIC supercap charger that draws
power from a fixed 24V input, and has
an input current limit of 1.2A. The
SEPIC architecture is chosen for several
reasons: it can do both step-up and
step-down, and it has inherent isolation
of the input from the output. A coupled
inductor is chosen over a 2-inductor
approach because of the smaller, lowercost circuit. The magnetic coupling
effect allows the use of a single coupling
capacitor and the switch current levels of the LT3956 make strategic use of
the readily available coupled inductor
offerings of major magnetics vendors.
This circuit is intended for a situation
where VIN does not experience much
variation during normal operation. The
design procedure for this type circuit
begins with setting the maximum input
current limit with the RSENSE value and the
250mV default threshold. The next design
step is to determine the VOUT level below
A charging circuit for a large value capacitor (1F or more) might be found in a nonbattery based backup power system. These
chargers would draw power from some
inductive based DC supply that operates
which VIN current is to be reduced through
CTRL to maintain less than 2.5A average
switching current. Assuming slightly less
than 90% efficiency, set the resistor divider
R5 and R6 to give CTRL = 1.1V when
VOUT =
VIN • 0.9 • IIN(MAX )
2.5A − IIN(MAX )
, at CTRL = 1.1V
The values of R5 and R6 should be an
order of magnitude higher value than
resistor R7. The resistor divider R7 and R8
is set to provide a minimum voltage at
CTRL, greater than 125mV, which is needed
to set non-zero value for input current.
CONCLUSION
The LT3956 simplifies power conversion
applications needing both constant-current
and constant-voltage regulation, especially
if they are constrained by board area and/
or bill-of-materials length. Its features
are selected to minimize the number of
external analog blocks for these types of
applications while maintaining flexibility.
Careful integration of these components
into the switching regulator makes it possible to easily produce applications that
would otherwise require a cumbersome
combination of numerous externals. n
Figure 8. Supercapacitor charger with current limited input provides controlled charging current over a wide output range.
SW
VIN
PGND
EN/UVLO
FB
LT3956
1:1
C1
10µF
VMODE
CTRL
INTVCC
VREF
SS
VC
C2
4.7µF
RT
GND
28.7k
370kHz
10nF
L1B
C3
10µF
R5
1M
R6
40.2k INTVCC
10k
PWM
VOUT
0V TO 28V
R3
536k
R4
24.9k
PWMOUT
D1
R8
14k
1M
R1
59k
R7
2k
Q1
R2
30.1k
3
INPUT AND OUTPUT CURRENT (A)
ISN
ISP
1µF
C4
10µF
L1A
33µH
RSENSE
200mΩ
VIN
28V
≤ 1.2A
2.5
2
OUTPUT
1.5
1
INPUT
0.5
0
0
5
10
15
20
VOUT (V)
25
30
L1: WURTH ELEKTRONIK 744871330
D1: ON SEMICONDUCTOR MBRS360T
Q1: MMBTA42
C1,C3,C4: TAIYO YUDEN GMK316BJ106
April 2010 : LT Journal of Analog Innovation | 19
2MHz Dual Buck Regulator Operates Outside of AM Radio
Band When Delivering 3.3V and 1.8V from 16V Input
Pit-Leong Wong
The number of microprocessor-based control units continues
to grow in both automobiles and industrial systems. Because
the amount of required processing power is also increasing,
even modest processors require a low voltage core supply
in addition to 3.3V or 5V memory, I/O and analog supplies.
In automotive systems, power comes
from the battery, with its voltage typically between 9V and 16V. Including cold
crank and double battery jump-starts, the
minimum input voltage may be as low
as 4V and the maximum up to 36V, with
even higher transient voltages. Likewise,
a 24V industrial supply may be as high
as 32V. With these high input voltages,
linear regulators cannot be used for supply currents greater than 200mA without
overheating the regulator. Instead, high
efficiency switching regulators must be
used to minimize thermal dissipation.
There are challenges in applying switching regulators in these systems. A small
circuit is desired, and certain operating
frequencies may be unacceptable. At
high step-down ratios, switching regulators typically operate at frequencies in
the AM radio band. One solution is to
dynamically move the power converter
switching frequency (and harmonics)
away from the tuned AM frequency, but
moving the switching frequency can
lead to unexpected EMI problems.
A cleaner solution is to simply set the
switching frequency higher than the top
of the AM radio band, which is at 1.8MHz.
This is easier said than done, since most
existing buck converters cannot meet the
low (<100ns over temperature) minimum
20 | April 2010 : LT Journal of Analog Innovation
timer to supervise microprocessors.
The LT3640 is offered in 4mm × 5mm
QFN and 28-lead FE packages.
DUAL BUCK REGULATOR
The LT3640 is a dual channel, constant
frequency, current mode monolithic
buck switching regulator with power-on
reset and watchdog timer. Both channels
are synchronized to a single oscillator
with frequency set by RT. The adjustable frequency ranges from 350kHz
to 2.5MHz. The internal oscillator of
the LT3640 can be synchronized to an
external clock signal on the SYNC pin.
on-time required to produce the step-down
ratio from a 16V input to a 3.3V output.
The LT3640 solves this problem with a
fast non-synchronous high voltage buck
converter and a high efficiency synchronous low voltage buck converter. With a
typical minimum on time of about 60ns
over temperature, the high voltage channel in the LT3640 can deliver 3.3V from
16VIN at 2MHz with comfortable margin.
The synchronous low voltage channel
in the LT3640 can be cascaded from the
3.3V channel to generate the other lower
voltage buses such as 2.5V, 1.8V or 1.2V.
The high voltage channel is a non-synchronous buck with an internal 1.7A NPN top
switch that operates from an input of
4V to 35V. Above 35V, an internal overvoltage lockout circuit suspends switching,
protecting the LT3640 and downstream
circuits from input faults as high as 55V.
The low voltage channel is a synchronous
buck with internal CMOS power switches
The LT3640 also includes a programmable power-on reset timer and watchdog
0.22µF
Figure 1. 2MHz 3.3V/0.8A
and 1.8V/0.8A step-down
regulators
VIN
5V TO 35V CIN
4.7µF
D2
L1,3.3µH
EN/UVLO VIN
SYNC
WDE
PGOOD
SW
BST SW1
D1 80.6k
DA
FB1
49.9k
VOUT1
100k
100k
LT3640
RST1
RST2
WDO
WDI
µP
CIN: TAIYO YUDEN UMK316BJ475KL
COUT1, COUT2: TAIYO YUDEN JMK212BJ226MD
L1: VISHAY IHLP2020BZER3R3
L2: VISHAY IHLP1616BZER1R0
D1, D2: DIODES INC. B240A
D2: CENTRAL SEMICONDUCTOR CMDSH-4E
VOUT1
3.3V/0.8A
COUT1
22µF
VIN2
EN2
L2,1µH
SW2
100k
CWDT
CPOR
1.5nF
RT GNDSS2
1.5nF
32.4k
FB2
SS1
1nF
1nF
49.9k
VOUT2
1.8V/0.8A
COUT2
22µF
design features
The high voltage buck regulator in the LT3640 has
a very fast minimum on time of about 60ns. This
enables the LT3640 to operate at a high step-down
ratio while maintaining high switching frequency.
90
90
VIN2 = 3.3V
85
VIN = 16V
80
VIN = 24V
75
80
75
fSW = 2MHz
3.3V CHANNEL
70
0
0.2
0.4
0.6
0.8
1.0
VOUT1 CURRENT (A)
FAST, HIGH VOLTAGE
BUCK REGULATOR
The high voltage buck regulator in the
LT3640 has a very fast minimum on time
of about 60ns. This enables the LT3640 to
operate at a high step down ratio while
maintaining high switching frequency.
Figure 3 shows the waveform of the
LT3640 operating from input voltage of
35V to regulate a 3.3V output at 2MHz. The
on time of the top power switch is about
60ns, which is also flat over temperature.
85
VIN = 12V
EFFICIENCY (%)
EFFICIENCY (%)
providing high efficiency without the
need of an external Schottky diode and
accepts an input of 2.5V to 5.5V. Typically,
the low voltage channel can operate from
the output of the high voltage channel to
form a cascaded structure as shown in
Figure 1, but it can also operate from a
separate supply source. The low voltage
channel only switches when the high voltage channel output is within regulation.
1.2
fSW = 2MHz
1.8V CHANNEL
70
0
0.2
0.4
0.6
VOUT2 CURRENT (A)
0.8
1
Figure 2. Efficiency of the circuit in Figure 1
Besides the fast minimum on time, the
high voltage buck regulator has fast
switching edges to minimize switching
losses and improve conversion efficiency
at high frequency. Figure 4 shows the efficiency of the high voltage buck regulator
at 2MHz operation for different input voltages. For 5V output voltage, the high voltage buck maintains an efficiency of higher
than 86% for input voltage up to 24V.
OUTPUT SHORT-CIRCUIT
ROBUSTNESS
The LT3640 monitors the catch diode current to guarantee the output short-circuit
robustness for the high voltage buck
converter. The LT3640 waits for the catch
diode current to fall below its limit before
starting a new cycle. The top NPN does
not turn on until the catch diode current
is below its limit. This control scheme
90
85
EFFICIENCY (%)
VSW1
10V/DIV
SW1
10V/DIV
80
75
IL1
0.5A/DIV
fSW = 2MHz
VOUT1 = 5V
100ns/DIV
Figure 3. Fast high voltage buck regulator
70
0
0.2
VIN = 12V
VIN = 16V
VIN = 20V
VIN = 24V
0.4
0.6
0.8
1.0
VOUT1 CURRENT (A)
IL1
0.5A/DIV
1.2
Figure 4. Efficiency of the high voltage buck regulator
1µs/DIV
VIN = 30V
VOUT1 = SHORT
RT SET = 2MHz
Figure 5. Output shorted robustness, high voltage
channel
April 2010 : LT Journal of Analog Innovation | 21
ensures cycle-by-cycle current limit, providing protection against shorted output.
The switching waveform for VIN = 30V and
VOUT = 0V is shown in Figure 5.
LOW VOLTAGE SYNCHRONOUS
BUCK REGULATORS
The low voltage channel is a synchronous
buck with internal CMOS power switches
providing high efficiency without the
need of external Schottky diode. This
channel only switches when the high
voltage channel output is within regulation. The output can be programmed
as low as 0.6V, covering any core voltage in modern microprocessors.
The low voltage buck has a similar scheme
as the high voltage buck of monitoring the current in the bottom NMOS to
guarantee shorted output robustness.
However, when the bottom NMOS current
exceeds its limit, the oscillator frequency
is not affected. The low voltage buck
simply skips one cycle to avoid interference with the high voltage buck.
At light load the low voltage buck also
operates in low ripple Burst Mode operation to minimize output ripple and power
loss. Although the two channels in the
LT3640 share a common oscillator, they
may require different light load operation
frequencies to optimize efficiencies at their
respective loads. In this case, the oscillator always runs at the higher frequency,
with the channel requiring the lower
frequency skipping cycles. Figures 6 and 7
show the light load switching waveforms
of two channels running at same reduced
frequency and at different frequencies,
respectively. Output ripple for both channels remains below 10mVP–P. No-load
quiescent current from the input is only
300µA with both outputs in regulation.
BENEFITS OF CASCADING
As described above, there are clear
advantages of cascading two switchers to
generate I/O and core supplies. The higher
operating frequency reduces circuit size
22 | April 2010 : LT Journal of Analog Innovation
SW1
10V/DIV
SW1
10V/DIV
IL1
0.5A/DIV
IL1
0.5A/DIV
SW2
5V/DIV
SW2
5V/DIV
IL2
0.5A/DIV
IL2
0.5A/DIV
500ns/DIV
VIN = 12V
VOUT1 = 3.3V/25mA
VIN2 = VOUT1
VOUT2 = 1.8V/30mA
2µs/DIV
VIN = 12V
VOUT1 = 3.3V/0mA
VIN2 = VOUT1
VOUT2 = 1.8V/30mA
Figure 6. Two channels running in discontinuous
mode at light load remain synchronized.
Figure 7. At still lighter loads, the two channels
switch at different frequencies to maintain high
efficiency and low output ripple.
and provides faster transient response
for better regulation. The low voltage
technology used for the core supply
switcher further reduces solution size.
Figure 1 against two non-synchronous
bucks operating from VIN, the overall
efficiency is nearly identical. If the core
voltage is reduced from 1.8V to 1.2V, the
LT3640 circuit is actually more efficient.
Although the core supply is generated
via two conversions, with two efficiency
hits, keep in mind that the core supply is often low power, even if it is high
current, so total power loss is minimal.
Also, a buck converter generating the
core voltage directly from the input
does not typically operate in an efficient
region anyway, and it would be larger
and slower. Comparing the circuit in
Figure 8. Power-on reset
and watchdog timing
POWER-ON RESET AND
WATCHDOG TIMER
In high reliability systems, a supervisor monitors the activity of the microprocessor. If the processor appears to
stop, due to either hardware or software
faults, the supervisor resets the microprocessor in an attempt to restore the
system to a functional state. While some
FB
tRST
tUV
RST
a. Power-on reset timing
WDI
WDO
tDLY
t < tWDL tRST
b. Watchdog timing
t < tWDU
tWDL < t < tWDU
tWDU
tRST
design features
modern processors include internal
supervisor functions, it is better practice
to separate the two. Typical supervisor functions are voltage monitors with
power-on resets to qualify supply voltages and watchdog timers to monitor
software and hardware functions.
function for higher reliability. If the falling
edges on the WDI pin are grouped too close
together or too far apart, the WDO pin
is pulled down for a period the same as
the power-on reset timeout period before
the watchdog timer is started again. The
timing diagrams of the power-on reset and
watchdog timer are shown in Figure 8.
The LT3640 includes one power-on reset
timer for each buck regulator and one
common watchdog timer. Power-on
reset and watchdog timers are both
adjustable using external capacitors.
LOW NOISE DATA ACQUISITION
SUPPLY
Figure 9 shows a 4-output supply that
generates low noise 5V and 3.3V rails for
analog circuits, along with 3.3V I/O and
1.5V core supplies for digital circuits.
The high voltage channel of the LT3640
converts the input to a 5.7V intermediate
bus, the low voltage channel bucks to the
1.5V core, and a few LDOs regulate the
5V and 3.3V outputs. The 5.7V bus voltage
gives the 5V regulator suitable headroom
for good PSSR and transient performance.
Once the high voltage buck output voltage reaches 90% of its regulation target,
the high voltage channel reset timer is
started and the RST1 pin is released after
the reset timeout period. The low voltage channel reset timer is started once the
low voltage buck output voltage reaches
92% of its regulation voltage, and releases
RST2 after the reset timeout period.
Diode D2 performs two functions. First,
it lowers the intermediate voltage to a
value below the 5.5V maximum operating
voltage of the synchronous buck regulator. Second, it isolates high frequency
ripple current flowing into the VIN2 and
The watchdog circuit monitors a microprocessor’s activity. As soon as both RST1
and RST2 are released and an additional
delay has expired, the watchdog starts
monitoring the signal at the WDI pin. The
LT3640 implements windowed watchdog
Figure 9. This circuit generates two low
noise analog rails plus I/O and core
supplies for a microprocessor.
BST pins of the LT3640 from the inputs
of the LDOs regulating the analog supplies, resulting in quiet analog rails.
Total current draw from the 5.7V rail
is 820mA, so about 30% more power
is available from this output. RST1 and
RST2 indicate power is good when the
1.5V and 5.7V rails are in regulation.
The watchdog function is not used here,
and is disabled by tying WDE to VIN2.
The associated pins, not shown on this
schematic, should be left floating.
CONCLUSION
The LT3640 is a dual channel buck regulator. The high voltage channel buck is
capable of converting 16V input voltage
to 3.3V output at 2MHz with comfortable
margin. The high voltage buck maintains efficiency above 86% for delivering
5V from up to 24V input at 2MHz switching frequency. The low voltage channel
buck input ranges from 2.5V to 5.5V. The
LT3640 also includes power-on reset and
watchdog timer to monitors a microprocessor’s activity. The high frequency
high efficient buck converters and the
programmable timers make the LT3640
ideal for automotive applications. n
LOW NOISE, LDO
MICROPOWER REGULATORS
IN
OUT
LT1962-5
SHDN BYP
GND
D3
VIN
7V TO 35V
CIN1
10µF
VIN
BST
EN/UVLO
SW
SW1
226k
49.9k
DA
FB1
POR
1nF
CIN1: TAIYO YUDEN UMK325BJ106MM
COUTSW1: TAIYO YUDEN JMK212BJ226MD
CIN2, COUT1, COUT2, COUT3 : TAIYO YUDEN JMK212B7106KG
COUT4 : TAIYO YUDEN JMK212BJ476MG
L1: VISHAY IHLP2525CZER8R2
L2: VISHAY IHLP1616BZER2R2
D1: UPS140
D2: CMMR1-02
D3: PMEG4005
D1 178k
1nF
1nF
82.5k
VIN2
RST1
RST2
WDE
CPOR
EN2
SHDN BYP
GND
RT
SYNC
GND
FB2
VOUT2
3.3V/100mA
COUT2
10µF
10nF
VOUT3
3.3V/300mA
COUT3
10µF
VOUT4
1.5V/800mA
SW2
SS2
SHDN BYP
GND
IN
OUT
LT1763-3.3
CIN2
10µF
75k
10nF
D2
L2
2.2µH
SS1
IN
OUT
LT1761-3.3
COUTSW1
22µF
LT3640
100k
VOUT1
5V/100mA
COUT1
10µF
1µF
L1,8.2µH
49.9k
VOUT3
10nF
COUT4
47µF
49.9k
April 2010 : LT Journal of Analog Innovation | 23
Eight 16-Bit, Low INL, VOUT DACs in a 4mm × 5mm
QFN Package: Unparalleled Density and Flexibility
Leo Chen
While 16-bit VOUT DACs are not uncommon, the
combination of low INL (±4 LSB) and high density
(eight DACs in a 4mm × 5mm QFN package) allows
the LTC2656 to fit an unparalleled range of sockets.
Space-saving features also include a built-in 2ppm/°C
reference, with performance typically reserved for
external references. These characteristics, along with
superior offset and gain error specifications, make the
LTC2656 a powerful device housed in a tiny package.
Although the LTC2656 is a formidable standalone device, it can be
readily combined with other Linear
Technology products to produce
uniquely high performance solutions.
A DIGITALLY CONTROLLED POWER
SUPPLY USING THE LT3080 1.1A
REGULATOR
Another noteworthy product in the
Linear Technology portfolio is the
LT3080. The LT3080 is a 1.1A low dropout
Figure 1. Simple variable output voltage 1.1A supply
The LTC2656 can be combined with the
LT3080 to create a digitally controlled
Figure 2. Digitally controlled version of the power supply in shown in Figure 1 with a 0V–4.096V output range.
For a 0V–2.5V output range use the LTC2656-LI.
5V
RESET SELECT
LT3080
IN
The LTC2656’s 16 bits of resolution yield
finer tuning of the LT3080’s output than
that of a digital potentiometer. Typical
digital potentiometers have resolutions of
only 8 bits; high resolution digital pots are
at 10 bits. The ±4 LSB INL of the LTC2656
is far superior to the INL of typical digital
pots. The LTC2656-L combined with an
LT3080 results in a power-supply with an
adjustable range of 0V–2.5V, while using
the LTC2656-H produces a power supply
with an adjustable range of 0V–4.096V.
linear regulator that can be paralleled to
increase output current or spread heat
on surface mounted boards. Typically
the output voltage of the LT3080 is
adjusted by a resistor at the SET pin of
the part (Figure 1). A fixed current of
10µA flows out of the SET pin through the
resistor and the resulting voltage drop
programs the output of the regulator.
MID-SCALE
VIN
1.2V TO 36V
power supply. If the output of the DAC is
connected to the SET pin of the regulator (Figure 2), the LT3080 acts as a unity
gain buffer. The DAC simply sinks the
10µA from the internal current source
and directly controls the output of the
regulator creating a digitally controlled
power supply. Should more than 1.1A be
needed, multiple LT3080s can be paralleled to provide more output current.
5V
ZERO-SCALE
0.1µF
VCONTROL
+
–
1µF
SET
RSET
VOUT = RSETtµA
0.1µF
OUT
7.5k
REFCOMP REFIN/OUT LDAC PORSEL VCC CLR
VOUT
2.2µF
0.1µF
TO
µC
CS
SCK
SDO
SDI
LTC2656-HI6
REFLO
GND
VOUTA
VOUTB
VOUTC
VOUTD
VOUTE
VOUTF
VOUTG
VOUTH
VIN
4.446V–36V
LT3080
IN
0V–4.096V
VCONTROL
+
–
1µF
SET
NOTE: LT3080 MINIMUM
LOAD CURRENT IS 0.5mA
24 | April 2010 : LT Journal of Analog Innovation
OUT
VOUT
2.2µF
design features
Although the LTC2656 is a formidable standalone device,
it can be readily combined with other Linear Technology
products to produce uniquely high performance solutions.
5V
GAIN = 4
0.1µF
VOUTA
M9 50k
0V–2.5V
SDO
VOUTC
SDI
VOUTD
LTC2656-LI6
PORSEL
VOUTF
CLR
VOUTG
VCONTROL
+
–
4pF
1µF
M1 450k
–
VOUTE
LDAC
VOUTH
REFIN/OUT GND REFLO REFCOMP
0.1µF
450k
M3 150k
VOUTB
SCK
LT3080
IN
VCC
LT1991
VCC
CS
VIN
10.35V TO 36V
12V
P1
450k
P3
150k
P9
50k
OUT
SET
VOUT
0V–10V
2.2µF
OUT
+
NOTE: LT3080 MINIMUM LOAD CURRENT
IS 0.5mA
450k
0.1µF
4pF
VEE
REF
Figure 3. Expanding the power supply
output range to 0V–10V
The two Linear Technology parts complement each other very well. The LT3080 has
a max offset of 2mV in the DFN package,
while the LTC2656 has a max offset of
±2mV. This means that there will only
be a slight degradation in offset performance when combining these two parts.
This circuit can also be easily replicated at
Figure 4. Expanding the power supply
output range to 0V–12V
each of the DAC outputs in order to create
an 8-channel adjustable power supply.
ADDING THE LT1991 TO ADJUST
OUTPUT RANGE
CS
LT1991
VOUTA
VOUTB
SCK
SDO
VOUTC
SDI
VOUTD
PORSEL
LTC2656-LI6
VOUTF
CLR
VOUTG
VOUTH
REFIN/OUT GND REFLO REFCOMP
0.1µF
LT3080
IN
VCC
M9 50k
450k
VCONTROL
+
–
4pF
M3 150k
1µF
M1 450k
VOUTE
LDAC
0.1µF
0V–2.5V
VIN
12.46V TO 36V
15V
GAIN = 4.844
0.1µF
VCC
The LT1991 is a micropower precision gain
selectable amplifier. It combines a precision op amp with eight precision matched
resistors in a small package. Using the
The LTC2656-H combined with an LT3080
provides a nice digitally controlled
1.1A power supply with an output range
from 0V to 4.096V. However, should the
5V
user need a wider output range than what
that particular circuit offers, there is an
LTC part that provides an easy solution.
P1
450k
P3
150k
P9
50k
–
OUT
SET
VOUT
0V–12.11V
2.2µF
OUT
+
NOTE: LT3080 MINIMUM LOAD CURRENT
IS 0.5mA
450k
VEE
4pF
REF
April 2010 : LT Journal of Analog Innovation | 25
LT1991 eliminates the need for any expensive external precision gain setting resistors. Gain can be set by simply changing
how the input pins are connected, which
in turn changes how the internal precision resistors set the gain of the op amp.
12V
LTC6240
VCC
LT1991
–
M9 50k
VCC
VOUTA
SCK
VOUTB
SDO
VOUTC
SDI
VOUTD
LTC2656-LI6
PORSEL
4pF
M1 450k
0.1µF
CS
450k
M3 150k
5V
The limiting factor in output range of
the LTC2656-LT3080 circuit is the output
range of the LTC2656. The LT3080 is fully
capable of going all the way up to 36V,
however the DAC output at the SET pin
is limited to 4.096V which in turn limits
the output of the regulator. This hurdle
is easily overcome by adding an LT1991
between the output of the LTC2656 and the
SET pin of the LT3080. The LT1991 expands
the output range of the DAC, which in
turn sets the output range of the LT3080.
The LT1991 standard grade has 0.08%
gain accuracy and 100µV offset, so there
should be no degradation in performance
of the DAC. With a maximum gain of 13
and a supply range of 40V, the LT1991
can stretch the DAC output range to
match that of the LT3080. Typical output ranges such as 0V–10V (Figure 3) and
0V–12V (Figure 4) are easily achieved.
0.1µF
+
0V–2.5V
VOUTE
LDAC
VOUTF
CLR
VOUTG
P1
450k
P3
150k
P9
50k
–
OUT
+
VOUT
±5V
450k
4pF
VEE
VOUTH
REF
0.1µF
–12V
REFIN/OUT GND REFLO REFCOMP
0.1µF
0.1µF
Figure 5. Using the LT1991 to shift the LTC2656
output range to ±5V
USING THE LT1991 TO GO BIPOLAR
(Figure 5), a ±10V output (Figure 6) and
a variety of other bipolar voltages.
While the LTC2656 does come in two
flavors, neither one is capable of a bipolar
output. This again is a situation where the
LT1991 proves its mettle. Using a combination of gain and offset, the LT1991
can be combined with the LTC2656 to
form circuits that provide a ±5V output
As stated before, the LT1991 has the
advantage of not requiring expensive
external precision gain set resistors.
While some users might balk at using a
multichip solution in order to achieve a
12V
12V
+
0.1µF
LTC6240
–
LT1991
M1 450k
VCC
VOUTA
SCK
VOUTB
SDO
VOUTC
0V–2.5V
VOUTD
SDI
LTC2656-LI6
P1
450k
P3
150k
P9
50k
VOUTE
LDAC
VOUTF
CLR
VOUTG
LT1991
450k
M1 450k
OUT
450k
VEE
4pF
450k
4pF
M3 150k
–
+
VCC
M9 50k
4pF
M3 150k
0.1µF
PORSEL
VCC
M9 50k
5V
CS
0.1µF
±5V
P1
450k
P3
150k
P9
50k
REF
–
+
450k
VEE
VOUTH
REFIN/OUT GND REFLO REFCOMP
0.1µF
0.1µF
–12V
0.1µF
Figure 6. Using the LT1991 to shift the LTC2656 output range to ±10V
26 | April 2010 : LT Journal of Analog Innovation
OUT
0.1µF
–12V
4pF
REF
VOUT
±10V
design features
space equation while likely producing
significant performance improvements.
GOING ALL THE WAY TO GROUND
5V
0.1µF
CS
VCC
VOUTA
VOUTB
SCK
SDO
VOUTC
SDI
VOUTD
PORSEL
D1
BAS70
LTC2656
VOUTE
LDAC
VOUTF
CLR
VOUTG
10k
–5V
The pull-down resistor forces the amplifier’s pull-up stage to turn on. With the
pull-up stage turned on on, the output
loop is correctly closed, putting the amplifier back into regulation. The Schottky
diode prevents the output from being
driven far below ground during power-up
or when the DAC is placed into shutdown.
VOUTH
REFIN/OUT GND REFLO REFCOMP
0.1µF
The LTC2656 has unmatched offset performance, and its ouput can swing within
3mV of ground. For applicatons requiring
the outputs to go completely to the lower
supply rail, some additional circuitry, in
the form of a Schottky diode and pulldown resistor, must be added (Figure 7).
0.1µF
Figure 7. Achieving true rail-to-rail
performance with the LTC2656
bipolar output, it is worth pointing out
that the LTC2656 already saves space by
eliminating the need for an external reference. Also, many bipolar DACs require
an external op amp to convert a current
output to a voltage anyway, so in the
end, the LT1991 adds little to the board
LINKING MULTIPLE LTC2656s TO
THE SAME REFERENCE
outputs, it is possible to drive all of the
LTC2656 from a single internal reference. This is accomplished by tying the
REFCOMP pin low on all but one of the
DACs while also issuing the internal
reference shutdown command through
the digital interface. The one DAC with
REFCOMP not tied to GND becomes the
master reference, the REFIN/REFOUT pin of
this DAC feeds into the REFIN/REFOUT pin
on all the other DACs. All of the DACs on
the board are thus driven from a single
internal reference, avoiding variances
in the respective reference outputs.
It is important to have the correct bypass
capacitors in place at each of the reference
inputs (Figure 8). The master reference
should be treated similarly to a discrete
reference during board layout and design.
CONCLUSION
The LTC2656 offers superior accuracy,
precision and DAC density without the
need for an external reference, thus
reducing overall part count and footprint. It can be easily inserted into a
wide variety of applications to solve
otherwise intractable problems. n
When a design calls for more than
the eight channels available in a single LTC2656, the SPI interface of the
DAC allows for easy expansion.
When the application demands high
precision matching between all analog
Figure 8. Driving multiple LTC2656s with a single internal reference
INDIVIDUAL
DAC CHIP
SELECT LINES
COMMON INPUT
COMMON CLOCK
VCC
VCC
0.1µF
VCC
0.1µF
VOUTA
CS
VOUTB
SCK
SDO
VOUTC
SDI
VOUTD
CS
SCK
LTC2656
VCC
VCC
0.1µF
VCC
VOUTA
CS
VOUTB
SCK
SDO
VOUTC
SDO
VOUTC
SDI
VOUTD
SDI
VOUTD
LTC2656
VOUTA
VOUTB
LTC2656
VOUTE
PORSEL
VOUTE
PORSEL
LDAC
VOUTF
LDAC
VOUTF
LDAC
VOUTF
CLR
VOUTG
CLR
VOUTG
CLR
VOUTG
PORSEL
VOUTH
REFIN/OUT GND REFLO REFCOMP
0.1µF
0.1µF
VOUTH
REFIN/OUT GND REFLO REFCOMP
0.1µF
VOUTE
VOUTH
REFIN/OUT GND REFLO REFCOMP
0.1µF
April 2010 : LT Journal of Analog Innovation | 27
6mm × 6mm DC/DC Controller for High Current
DCR Sensing Applications
Eric Gu, Theo Phillips, Mike Shriver and Kerry Holliday
The LTC3855 is a versatile 2-phase synchronous buck controller IC with on-chip drivers,
remote output voltage sensing and inductor temperature sensing. These features are
ideal for high current applications where cycle-by-cycle current is measured across
the inductor (DCR sensing). Either channel is suitable for inputs up to 38V and outputs
up to 12.5V, further increasing the controller’s versatility. The LTC3855 is based on
the popular LTC3850, described in the October 2007 issue of Linear Technology.
MONITORING THE TEMPERATURE
“lossless” method is less accurate than
using a sense resistor, in large part because
as the inductor heats up, its resistance
increases with a temperature coefficient of resistivity (TCR) of 3930ppm/°C.
Therefore as the temperature rises, the
current limit decreases. When DCR sensing
When current is sensed at the inductor,
either a sense resistor is placed in series
with the inductor, or an R-C network
across the inductor is used to infer the
current information across the inductor’s DC resistance (DCR sensing). This
is used, the current limit for the LTC3855
is determined by the peak sense voltage as
measured across the inductor’s DCR. The
LTC3855 includes a temperature sensing
scheme designed to compensate for the
TCR of copper by effectively raising the
peak sense voltage at high temperature.
Figure 1. A 1.2V, 50A, 2-phase converter. The two channels operate 180º out-of-phase to minimize output ripple and component sizes.
10k
73.2k
22.1k
22.1k
1%
TG1
ITH1
BOOST1
VFB1
PGND1
SGND
EXTVCC
100k
TG2
SW2
20k
1%
PGOOD
2.2Ω
M2
RJK0330DPB
×2
4.7µF
1µF
COUT1
100µF
6.3V
×4
0.1µF
CMDSH-3
M3
RJK0305DPB
×2
M4
RJK0330DPB
×2
L2
0.33µH
3.92k
30.1k
L1, L2: VISHAY IHLP5050FD-01, 0.33µH
COUT1: MURATA GRM31CR60J107ME39L
COUT2: SANYO 2R5TPE330M9
RNTC: MURATA NCP18WB473J03RB
28 | April 2010 : LT Journal of Analog Innovation
VIN
4.5V TO
14V
270µF
16V
3.92k
BOOST2
NC
PGOOD2
PGOOD1
DIFFP
ILIM2
PGND2
ILIM1
BG2
SENSE2–
RUN2
SENSE2+
DIFFOUT
3.92k
0.1µF
+
INTVCC
TK/SS2
DIFFN
20k
1%
30.1k
L1
0.33µH
CMDSH-3
VIN
LTC3855
ITH2
2200pF
0.1µF
BG1
VFB2
220pF
M1
RJK0305DPB
×2
SW1
CLKOUT
PHSASMD
FREQ
ITEMP2
RUN1
ITEMP1
MODE/PLLIN
TK/SS1
0.1µF
22µF
×4
300kHz
SENSE1+
RNTC
47k
(PLACE BETWEEN L1 & L2)
0.1µF
SENSE1–
10k
1%
+
COUT2
330µF
2.5V
×4
VOUT
1.2V
50A
design ideas
80
90
70
IL1
10A/DIV
65
60
IL2
10A/DIV
55
0
20
40
60
80 100
INDUCTOR TEMP (°C)
120
140
85
80
75
ILOAD
20A/DIV
50
45
EFFICIENCY (%)
DC CURRENT LIMIT (A)
75
95
VOUT
(AC COUPLED)
100mV/DIV
COMPENSATED
UNCOMPENSATED
IOUT (MAX)
VIN = 12V
ILOAD = 30A–50A
50µs/DIV
70
MODE = CCM
fSW = 300kHz
VIN = 12V
0
10
40
20
30
LOAD CURRENT (A)
50
60
Figure 2. The converter of Figure 1 can deliver the
current whether hot or cold, with its calculated
worst-case current limit remaining above the target
50A, even well above room temperature.
Figure 3. VOUT is stable in the face of a of 30A to
50A load step for the converter of Figure 1, and the
inductor current sharing is fast and precise.
Figure 4. Efficiency for the converter of Figure 1.
DIFFERENTIAL SENSING
resistance for linearization), the current limit can be maintained above
the nominal operating current, even
at elevated temperatures (Figure 2).
OTHER IMPORTANT FEATURES
At high load current, an offset can
develop between the power ground, where
VOUT is sensed, and the IC’s local ground.
To overcome this load regulation error,
the LTC3855 includes a unity gain differential amplifier for remote output voltage
sensing. Inputs DIFFP and DIFFN are tied
to the point of load, and the difference
between them is expressed with respect
to local ground from the DIFFOUT pin.
Measurement error is limited to the
input offset voltage of the differential
amplifier, which is no more than 2mV.
SINGLE OUTPUT CONVERTER
WITH REMOTE OUTPUT VOLTAGE
SENSING AND INDUCTOR DCR
COMPENSATION
Figure 1 shows a high current DCR application with temperature sensing. The
nominal peak current limit is determined
by the sense voltage (30mV, set by grounding the ILIM pins) across the DC resistance
of the inductor (typically 0.83mΩ), or
36A per phase. This sense voltage can be
raised by biasing the ITEMP pins below
500mV. Since each ITEMP pin sources
10µA, peak sense voltage can be increased
by inserting a resistance of less than
25k from ITEMP to ground. By using an
inexpensive NTC thermistor placed near
the inductors (with series and parallel
The circuit also maintains precise regulation by differentially sensing the output
voltage. The measurement is not contaminated by the difference between power
ground and local ground. As a result,
output voltage typically changes less
than 0.2% from no load to full load.
MULTIPHASE OPERATION
The LTC3855 can be configured for dual
outputs, or for one output with both
power stages tied together, as shown in
Figure 1. In the single output configuration, both channels’ compensation (ITH),
feedback (VFB), enable (RUN) and track/
soft-start (TRK/SS) pins are tied together,
and both PGOOD1 and PGOOD2 will
indicate the power good status of the
output voltage. By doubling the effective
switching frequency, the single output
configuration minimizes the required
input and output capacitance and voltage ripple, and allows for fast transient
response (Figure 3). The LTC3855 provides
inherently fast cycle-by-cycle current
sharing due to its peak current mode
architecture plus tight DC current sharing.
A precise 10µA flows out of the FREQ pin,
allowing the user to set the switching
frequency with a single resistor to ground.
The frequency can be set anywhere from
250kHz to 770kHz. If an external frequency source is available, a phase-locked
loop enables the LTC3855 to sync with
frequencies in the same range. A minimum on-time of 100ns allows low duty
cycle operation even at high frequencies.
If the external sync signal is momentarily
interrupted, the LTC3855 reverts to the
frequency set by the external resistor.
Its internal phase-locked loop filter is
prebiased to this frequency. An internal
switch automatically changes over to the
sync signal when a clock train is detected.
Since the PLL filter barely has to charge or
discharge during this transition, synchronization is achieved in a minimum
number of cycles, without large swings in
switching frequency or output voltage.
The LTC3855 is also useful for designs
using three or more phases. Its CLKOUT pin
can drive the MODE/PLLIN pins of additional regulators. The PHASMD pin tailors the phase delays to interleave
all the switch waveforms.
(continued on page 35)
April 2010 : LT Journal of Analog Innovation | 29
Maximize Cycle Life of Rechargeable Battery Packs
with Multicell Monitor IC
Jon Munson
Rechargeable battery packs prematurely deteriorate in performance if any cells are
allowed to overdischarge. As a pack becomes fully discharged, the ILOAD • RINTERNAL
voltage drop of the weakest cell(s) can overtake the internal VCELL chemical potential
and the cell terminal voltage becomes negative with respect to the normal voltage. In
such a condition, irreversible chemical processes begin altering the internal material
characteristics that originally provided the charge storage capability of the cell,
so subsequent charge cycles of the cell do not retain the original energy content.
Furthermore, once a cell is impaired, it is more likely to suffer reversals in subsequent
usage, exacerbating the problem and rapidly shortening the useful cycle life of the pack.
With nickel-based chemistries, an overdischarge of a set of series-connected
cells does not necessarily lead to a safety
hazard, but it is not uncommon for one or
more cells to suffer a reversal well before
the user is aware of any significant degradation in performance. By then, it is too
late to rehabilitate the pack. In the case
of the more energetic lithium-based cell
chemistries, reversals must be prevented
as a safety measure against overheating or
fire. Monitoring the individual cell voltages is therefore essential to ensure a long
pack life (and safety with lithium cells).
Figure 1. Simple load-disconnect circuit to prevent excessive discharge of a nickel-cell pack.
FDS6675
1k
1.25V
100nF
1k
1.25V
100nF
1k
1.25V
100nF
1k
1.25V
100nF
1k
1.25V
100nF
1k
1.25V
100nF
1k
1.25V
100nF
1k
1.25V
100nF
100nF
V+
C12
C11
C10
C9
C8
C7
C6
C5
C4
C3
C2
C1
V–
7.5k
OV1
OV0
UV1
UV0
HYST
CC1
CC0
SLTB
SLTOK
DC
EOUT
EOUT
SIN
LTST-C190KGKT
2N7002K
CMHD457
1M
SIN
VTEMP1
VTEMP2
100nF
10k
SOUT
SOUT
VREF
EIN
VREG
EIN
LTC6906
V+
OUT
GND GND
1µF
30 | April 2010 : LT Journal of Analog Innovation
1µF
SPST
ENABLE
DIV
100nF
FEATURES OF THE LTC6801
The operating modes and programmable threshold levels are set by pinstrap connections. Nine UV settings (from
0.77V to 2.88V) and nine OV settings
(from 3.7V to 4.5V) are available. The
number of monitored cells can be set
from 4 to 12 and the sampling rate can
be set to one of three different speeds
to optimize the power consumption
versus detection time. Three different
hysteresis settings are also available to
tailor behavior of the alarm recovery.
100k
LTC6801
9.6V
Enter the LTC6801, developed to provide integrated solutions for these
specific problems. The LTC6801 can
detect individual cell overvoltage (OV)
and undervoltage (UV) conditions of up
to twelve series connected cells, with
cascadable interconnections to handle
extended chains of devices, all independent of any microprocessor support.
SET
1M
To support extended configurations of
series-connected cells, fault signaling
is transmitted by passing galvanically
isolated differential clock signals in both
directions in a chain of “stacked” devices,
providing excellent immunity to load
design ideas
Cell reversal is a primary damage mechanism in traditional
nickel-based multicell packs and can occur well before
other noticeable charge-exhaustion symptoms appear.
noise impressed on the battery pack. Any
device in the chain detecting a fault stops
its output clock signal, thus any fault
indication in the entire chain propagates
to the “bottom” device in the stack. The
clock signal originates at the bottom of
the stack by a dedicated IC, such as the
LTC6906, or a host microprocessor if one
is involved, and loops completely through
the chain when conditions are normal.
In many applications, the LTC6801 is used
as a redundant monitor to a more sophisticated acquisition system such as the
LTC6802 (for example, in hybrid automobiles), but it is also ideal as a standalone
solution for lower-cost products like
portable tools and backup power sources.
Since the LTC6801 takes its operating
power directly from the batteries that it
monitors, the range of usable cells per
device varies by chemistry in order to provide the needed voltage to run the part—
from about 10V up to over 50V. This range
supports groupings of 4–12 Li-ion cells
or 8–12 nickel-based cells. Figure 1 shows
how simply an 8-cell nickel pack can be
monitored and protected from the abuse
of overdischarge. Note that only an undervoltage alarm is relevant with the nickel
chemistries, though a pack continuity fault
would still be detected during charging
by the presence of an OV condition.
AVOIDING CELL REVERSALS
Cell reversal is a primary damage
mechanism in traditional nickelbased multicell packs and can actually occur well before other noticeable
charge-exhaustion symptoms set in.
Figure 2. Pack discharge conditions
that promote cell reversals may not
be apparent from load potential.
IDEAL CELLS
5V AT LOAD
3 WEAK CELLS
5V AT LOAD
1 WEAK CELL
8V AT LOAD
0.63V
–0.1V
1.16V
0.62V
1.0V
1.15V
0.63V
1.1
1.16V
0.62V
–0.1V
LOAD
LOAD
1.15V
0.63V
1.1V
1.16V
0.62V
–0.1V
1.16V
0.63V
1.0V
–0.1V
0.62V
1.1V
1.16V
Consider the following scenario. An 8-cell
nickel-cadmium (NiCd) pack is powering
a hand tool such as a drill. The typical
user runs the drill until it slows to perhaps
50% of its original speed, which means
that the nominal 9.6V pack is loading
down to about 5V. Assuming the cells are
perfectly matched as in the left diagram
of Figure 2, this means that each cell has
run down to about 0.6V, which is acceptable for the cells. However, if there is a
mismatch in the cells such that perhaps
five of the cells are still above 1.0V, then
the other three would be below zero
volts and suffer a reverse stress as shown
in the middle diagram of Figure 2.
Even assuming that there is only one weak
cell in the pack (a realistic scenario) as in
the right diagram in Figure 2, the first cell
reversal might well occur while the stack
voltage is still 8V or more, with just a subtle reduction in perceived pack strength.
Because of the inevitable mismatching
that exists in practice, users unknowingly
reverse cells on a regular basis, reducing
LOAD
the capacity and longevity of their battery
packs, so a circuit that makes an early
detection of individual cell exhaustion
offers significant added value to the user.
USING THE LTC6801 SOLUTION
The lowest available UV setting of the
LTC6801 (0.77V) is ideal for detecting
depletion of a nickel-cell pack. Figure 1
shows a MOSFET switch used as a load
disconnect, controlled by the output state
of the LTC6801. Whenever a cell becomes
exhausted and its potential falls below
the threshold, the load is removed so that
cell reversal and its degradation effects
are avoided. It also allows the maximum
safe extraction of energy from the pack
since there are no assumptions made as to
the relative matching of the cells as might
be the case with an overly conservative
single pack-potential threshold function.
A 10kHz clock is generated by the LTC6906
silicon oscillator and the LTC6801 output status signal is detected and used to
control the load disconnect action. Since
April 2010 : LT Journal of Analog Innovation | 31
The LTC6801 simultaneously monitors up to 12 individual
cells in a multicell battery pack, making it possible to
maximize the pack’s capacity and longevity. It can
also be cascaded to support larger battery stacks.
9.6V
Figure 3. Alternative circuit provides an audible
warning of the need to recharge the pack without
interrupting service to the load.
100k
LTC6801
1k
1.25V
100nF
1k
1.25V
100nF
1k
1.25V
100nF
1k
1.25V
100nF
1k
1.25V
100nF
1k
1.25V
100nF
1k
1.25V
100nF
1k
1.25V
100nF
V+
C12
C11
C10
C9
C8
C7
C6
C5
C4
C3
C2
C1
V–
OV1
OV0
UV1
UV0
HYST
CC1
CC0
SLTB
SLTOK
DC
EOUT
EOUT
SIN
SPST
ENABLE
LED1
2N7002K
47k
2N7002K
CMHD457
1M
VREG
100nF
10k
SIN
VTEMP1
VTEMP2
VREF
7.5k
PKB24SPCH3601-B0
SOUT
SOUT
EIN
EIN
LTC6906
V+
100nF
OUT
GND GND
1µF
1µF
100nF
PDZ5.1B
DIV
SET
1M
LED1: LTST-C190KGKT
this example does not involve stacking of
devices, the cascadable clock signals are
simply looped-back rather than passed to
another LTC6801. An LED provides a visual
indication that power is available to the
load. Once the switch opens, the voltage
of the weak cell tends to recover somewhat and the LTC6801 reactivates the load
switch (no hysteresis with 0.77V undervoltage setting). The cycling rate of this
digital load-limiting action depends on
the configuration of the DC pin; in the
fastest response mode (DC = VREG), the
duty cycle of the delivered load power
drops and tapers off, with pulsing becoming noticeable and slower as the weakest
cell safely reaches a complete discharge.
32 | April 2010 : LT Journal of Analog Innovation
In some applications it is not acceptable
to spontaneously interrupt the load when
the weakest cell is nearing full discharge
as depicted in Figure 1. For those situations, the circuit of Figure 3 might be a
good alternative. This circuit does not
force a load intervention, but simply
provides an audible alarm indication that
the battery is near depletion. Here the
LED provides an indication that the alarm
is active and that no cells are exhausted.
An LTC6801 idle mode is invoked whenever the source clock is absent, and
power consumption then drops to a
miniscule 30µA, far less than the typical self-discharge of the pack. In both
figures, the circuits show a switch
that disables the oscillator (and other
peripheral circuitry) in order to place the
circuit into idle mode when not being
used so that battery drain is minimized.
CONCLUSION
The LTC6801 simultaneously monitors up
to 12 individual cells in a multicell battery
pack, making it possible to maximize
the pack’s capacity and longevity. It can
also be cascaded to support larger battery stacks. The device has a high level
of integration, configurability and well
thought out features, including an idle
mode to minimize drain on the pack
during periods of inactivity. This makes
the LTC6801 a compact solution for
improving the performance and reliability of battery powered products. n
design ideas
Dual 500mA µPower LDO Features Independent 1.8V–20V
Inputs and Easy Sequencing in a 4mm × 3mm DFN
Molly Zhu
The LT3029 integrates
two independent 500mA
monolithic LDOs in a
tiny 16-lead MSOP or
4mm × 3mm × 0.75mm DFN
package. Both regulators
have a wide 1.8V to 20V
input voltage range with
a 300mV dropout voltage
at full load. The output
voltage is adjustable down
to the 1.215V reference
voltage. With an external
bypass capacitor, the
output voltage noise is less
than 20µVRMS. A complete
power supply requires only
a minimum 3.3µF ceramic
output capacitor for each
channel to be stable.
Quiescent current is 55µA per channel, dropping below 1µA in shutdown.
Reverse-battery protection, reversecurrent protection, current limit foldback and thermal shutdown are all
Figure 1. Completely independent
channels have separate input and
SHDN pins.
VIN1
5V
IN1
1µF
OUT1
LT3029
10nF
BYP1
ADJ1
VIN2
2.5V
IN2
ON
OFF
SHDN1
OUT2
ON
OFF
SHDN2
BYP2
ADJ2
10nF
GND
The LT3029 includes features that simplify
the design of multivoltage systems. Its two
independent regulators present separate input and shutdown pins. It is also
compatible with the LTC2921, LTC2922 and
LTC2923 power supply tracking controllers, allowing for easy multirail power
supply tracking and sequencing design.
TWO INDEPENDENT REGULATORS
The LT3029’s inputs can be used independently or combined. Figure 1 shows
an application generating two output voltages from two different input
voltages, with independent shutdown control for each channel.
10µF
237k
1%
1µF
integrated into the package, making it
ideal for battery-powered systems.
402k
1%
VOUT1
3.3V
500mA
113k
1%
10µF
VOUT2
1.8V
500mA
237k
1%
DIFFERENT START-UP SLEW RATES
Start-up time is roughly proportional
to the bypass capacitance, regardless
of the input and output voltage. The
output capacitance and the load characteristics also have no influence on
the result. Figure 2 shows the regulator
start-up time versus bypass capacitance.
The LT3029 is compatible with LTC292x
series of power supply sequencing and
tracking controllers. Its ADJ pin should be
connected to LTC292x FB pin. By choosing the right resistors, it can track or
sequence the power supply. Please refer
to the LTC2923 data sheet for details.
(continued on page 35)
Table 1. Comparison between dual channel LDOs
V IN RANGE (V)
I OUT (mA)
DROPOUT VOLTAGE @ IOUT (mV)
INDEPENDENT V IN
I Q /CHANNEL (µA)
DFN PACKAGE SIZE
LT3023
1.8~20
100/100
300/300
N
20
3mm × 3mm
LT3024
1.8~20
100/500
300/300
N
30
4mm × 3mm
LT3027
1.8~20
100/100
300/320
Y
25
3mm × 3mm
LT3028
1.8~20
100/500
300/300
Y
30
5mm × 3mm
LT3029
1.8~20
500/500
300/300
Y
55
4mm × 3mm
April 2010 : LT Journal of Analog Innovation | 33
Product Briefs
PD applications often incorporate the
flexibility and reliability of receiving
power from either a PoE or auxiliary
power source. A PoE based power source
can operate as high as 57V while an
auxiliary power source may come from
a 12V wall adapter. Furthermore, the
switching regulator may need to operate
an additional three diode drops below the
auxiliary input voltage when a PD application requires both 12VDC and 24VAC inputs.
(The diodes are required to rectify the
24VAC and support PoE/Aux diode ORing.)
With the demand for such a wide input
range, the LTC4278-based PD is ideally
suited to handle PoE and low voltage
auxiliary power inputs. The SHDN pin
provides a simple implementation for
PoE or auxiliary power priority when both
power sources are available and ready.
A PoE+ PD requires a high power efficiency delivery, particularly when operating near the PoE+ 25.5W limit. The
LTC4278 serves this application with an
34 | April 2010 : LT Journal of Analog Innovation
OUTPUT VOLTAGE (V)
The LTC4278 joins Linear Technology’s
family of Power-over-Ethernet-Plus (PoE+)
powered device (PD) controller and integrated switching regulators, ideally suited
for the next generation security cameras
and other PoE+ PD applications. The
LTC4278-based PD can receive power from
a PoE source or from 24VAC/12VDC auxiliary
power supplies. This wide input voltage
range distinguishes the LTC4278 from
other PD/switchers on the market today.
1.22
1.21
VOUT = 1.2V
Burst Mode OPERATION
1.20
1.210
LTC3409
VIN = 1.6V
VOUT = 1.2V
PULSE-SKIPPING
1.19
LTC3409A
VIN = 1.6V
1.205
OUTPUT VOLTAGE (V)
POE+ PD/SWITCHER SUPPORTS
SECURITY CAMERA 24VAC / 12V DC
AUXILIARY POWER APPLICATION
VOUT = 1.2V
Burst Mode OPERATION
1.200
VOUT = 1.2V
PULSE-SKIPPING
1.195
1.190
1.185
1.18
0 100 200 300 400 500 600 700 800 900
LOAD CURRENT (mA)
1.100
1
10
100
LOAD CURRENT (mA)
1000
Figure 1. The LTC3409A (right graph) improves regulation over the LTC3409 (left)
onboard synchronous, current-mode,
flyback controller that can implement
Linear Technology’s patented No-Opto
feedback topology. This LTC4278-based
topology is capable of delivering >92%
isolated power supply efficiency, or
>88% efficiency including diode bridges
and Hot Swap™ FET. Furthermore, full
IEEE 802.3 isolation is achieved without the need for an opto-isolator.
The LTC4278 is also fully equipped with
PoE detection, programmable classification
load current and 2-event classification discovery. Inrush current limiting and a true
soft-start function, both provide graceful
ramp-up of line and all output voltages.
Programmable timing, operating frequency, and the optional synchronized timing can be employed to optimize efficiency,
component size, cost and EMI performance. Safety features include overvoltage, undervoltage and thermal shutdown,
and the devices can be configured for
short-circuit protection with auto restart.
LOW V IN BUCK REGULATOR
IMPROVES REGULATION IN BURST
MODE OPERATION
The LTC3409A buck regulator features
improved DC, line and load regulation
in Burst Mode operation as compared
to the LTC3409. These improvements
are the result of reducing offsets in the
regulator as well as increasing the regulator’s DC gain and PSRR. Suggested output capacitance has also been increased
to 44µF for optimal load regulation
and stability. A stress relief coating has
been applied to the die to minimize
offset spreading resulting from encapsulation in the DFN plastic package.
Figure 1 shows the improvement in load
regulation. The LTC3409A output voltage in
Burst Mode operation consistently begins
at a higher voltage at light loads dropping
slightly as load increases, then dropping off at heavy loads. Due to process
variations, the LTC3409’s output voltage
in Burst Mode operation can either start
higher or lower at light loads compared
to the output voltage when in continuous conduction mode at heavy loads.
product briefs
The LTC3108 is a highly integrated
DC/DC converter ideal for harvesting and managing surplus energy from
extremely low input voltage sources
such as TEG (thermoelectric generators),
thermopiles and small solar cells. The
step-up topology operates from input
voltages as low as 20mV. Using a small
step-up transformer, the LTC3108 provides
a complete power management solution
for wireless sensing and data acquisition.
The 2.2V LDO powers an external microprocessor, while the main output is
programmed to one of four fixed voltages
to power a wireless transmitter or sensors. The power good indicator signals
that the main output voltage is within
regulation. A second output can be
(LTC3855 continued from page 29)
The MOSFET drivers and control circuits
are powered by INTVCC, which by default
is powered through an internal low
dropout regulator from the main input
supply, VIN. If lower power dissipation
in the IC is desired, a 5V supply can be
connected to EXTVCC. When a supply is
detected on EXTVCC, the LTC3855 switches
INTVCC over to EXTVCC, with a drop of
just 50mV. The strong gate drivers with
optimized dead time provide high efficiency. The full load efficiency is 86.7%
and the peak efficiency is 89.4% (Figure 4).
The LTC3855 features a RUN and TRACK/SS pin
for each channel. RUN enables the output
and INTVCC, while TRACK/SS acts as a softstart or allows the outputs to track an
external reference. If a multiphase output is desired, all RUN and TRACK/SS pins
are typically tied to one another.
Peak current limiting is used in this
application, with the peak sense voltage set by the three-state ILIM pin. A
high speed rail-to-rail differential current
enabled by the host. A storage capacitor
provides power when the input voltage source is unavailable. Extremely low
quiescent current and high efficiency
design ensure the fastest possible charge
times of the output reservoir capacitor.
POWERFUL SYNCHRONOUS
N-CHANNEL MOSFET DRIVER IN A
2MM × 3MM DFN
The LTC4449 is high speed synchronous MOSFET driver designed to
maximize efficiency and extend the
operating voltage range in a wide variety of DC/DC converter topologies,
from buck to boost to buck-boost.
The LTC4449’s rail-to-rail driver outputs operate over a range of 4V to
6.5V and can sink up to 4.5A and source
up to 3.2A of current, allowing it to
easily drive high gate capacitance and/
or multiple MOSFETs in parallel for high
sense comparator looks across the current sense element (here the inductor’s
DC resistance, implied from the associated R-C network). If a short circuit
occurs, current limit foldback reduces
the peak current to protect the power
components. Foldback is disabled during start-up, for predictable tracking.
CONCLUSION
The LTC3855 is ideal for converters using
inductor DCR sensing to provide high current outputs. Its temperature compensation and remote output voltage sensing
ensure predictable behavior from light
load to high current. From inputs up to
38V it can regulate two separate outputs
from 0.6V to 12.5V, and can be configured
for higher currents by tying its channels
together, or by paralleling additional
LTC3855 power stages. At low duty cycles,
the short minimum on-time ensures constant frequency operation, and peak current limit remains constant even as duty
cycle changes. The LTC3855 incorporates
these features and more into 6mm × 6mm
QFN or 38-lead TSSOP packages. n
current applications. The high side driver
can withstand voltages up to 38V.
Adaptive shoot-through protection circuitry is integrated to prevent
MOSFET cross-conduction current.
With 14ns propagation delays and
4ns to 8ns transition times driving 3nF loads, the LTC4449 minimizes
power loss due to switching losses and
dead time body diode conduction.
The LTC4449 features a three-state
PWM input for power stage control and
shutdown that is compatible with all controllers that employ a three-state output
feature. The LTC4449 also has a separate
supply input for the input logic to match
the signal swing of the controller IC.
Undervoltage lockout detectors monitor
both the driver and logic supplies and disable operation if the voltage is too low. n
(LT3029 continued from page 35)
CONCLUSION
The LT3029 is a dual 500mA/500mA monolithic LDO with a wide input voltage range
and low noise. The two channels are fully
independent, allowing for flexible power
management. It is ideal for battery-powered systems because of its low quiescent
current, small package and integration of battery protection features. n
100
START-UP TIME (ms)
ULTRALOW VOLTAGE STEP-UP
CONVERTER AND POWER MANAGER
FOR ENERGY HARVESTING
10
1
0.1
0.01
10
1k
100
BYPASS CAPACITANCE (pF)
10k
Figure 2. Start-up time vs bypass capacitor value
April 2010 : LT Journal of Analog Innovation | 35
highlights from circuits.linear.com
WIRELESS REMOTE SENSOR APPLICATION POWERED FROM A PELTIER CELL
The LTC3108 is a highly integrated DC/DC converter ideal for harvesting and managing surplus
energy from extremely low input voltage sources such as TEGs (thermoelectric generators),
thermopiles and small solar cells. Using a small step-up transformer, the LTC3108 can boost
from input sources as low as 20mV to provide a complete power management solution for
microprocessors, remote sensors, data acquisition circuitry and low power RF links.
www.linear.com/3108
1nF
+
+
THERMOELECTRIC
GENERATOR
C1
220µF
VSTORE
+
LTC3108
330pF
1000
100
PGOOD
PGD
2.2V
VLDO
SW
µP
2.2µF SENSORS
VOUT
VS2
3.3V
+
VOUT = 3.3V
COUT = 470µF
0.1F
6.3V
VOUT2
C2
20mV TO 500mV
5V
TIME (sec)
T1
1:100
10
RF LINK
470µF
1
1:100 Ratio
1:50 Ratio
1:20 Ratio
VS1 VOUT2_EN
VAUX
GND
3108 TA01a
0
1µF
T1: COILCRAFT LPR6235-752SML
0
50
100 150 200 250 300 350 400
VIN (mV)
3108 TA01b
5V ISOLATED FLYBACK CONVERTER
The LT3574 is a monolithic switching regulator specifically designed for the isolated flyback
topology. The part senses the isolated output voltage directly from the primary side flyback
waveform—no third winding or opto-isolator is required for regulation. This circuit provides
a simple, low component count isolated 5V supply from a 12V to 24V input. The output
voltage is easily set by the resistor at RFB and RREF and the transformer turns ratio.
www.linear.com/3574
B340A
10µF
357k
0.22µF
VIN
3:1
2k
50µH
SHDN/UVLO
PMEG6010
51.1k
LT3574
RFB
t
1.0
VOUT+
5V
0.35A
22µF
5.6µH
t
OUTPUT VOLTAGE ERROR (%)
VIN
12V TO 24V
VOUT–
80.6k
RREF
TC
6.04k
RILIM
SS
VIN = 12V
0
–0.5
SW
VC
28.7k
GND
TEST BIAS
–1.0
59k
10k
10nF
VIN = 24V
0.5
0
100
200
300 400
IOUT (mA)
500
600
4.7µF
1nF
700
3574 TA01b
3574 TA01
910Ω
SINGLE 2.7V SUPPLY 4MHZ 4TH ORDER BUTTERWORTH FILTER
The LTC6247 is a power efficient dual op amp that offers 180MHz gain-bandwidth
product while consuming only 1mA of supply current per amplifier. This simple
implementation of a Butterworth lowpass filter provides a low power, low noise
4MHz lowpass filter. Operating from a single 2.7V supply, the complete circuit
dissipates less than 5mW of power. Both the input and output of the LTC6247
swing rail-to-rail, critical in maximizing dynamic range in low voltage systems.
www.linear.com/6247
1.1k
12pF
5.6pF
VIN
910Ω
2.7k
–
56pF
1/2LTC6247
2.7V
1.1k
+
2.3k
120pF
–
2.7V
1/2LTC6247
+
VOUT
1.2V
L, LT, LT, LTC, LTM, Linear Technology, the Linear logo, Burst Mode, and µModule are registered trademarks and Hot Swap and No Latency ∆∑ are trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners.
Linear Technology Corporation
1630 McCarthy Boulevard, Milpitas, CA 95035
(408) 432-1900
www.linear.com
© 2010 Linear Technology Corporation/Printed in U.S.A./46.5K
Cert no. SW-COC-001530