V17N2 - JUNE

LINEAR TECHNOLOGY
JUNE 2007
IN THIS ISSUE…
Cover Article
SiGe Differential Amplifier Drives
High Speed ADCs at Hundreds of MHz
............................................................1
Kris Lokere and Adam Shou
VOLUME XVII NUMBER 2
SiGe Differential Amplifier
Drives High Speed ADCs
at Hundreds of MHz
by Kris Lokere and Adam Shou
Linear in the News…............................2
Design Features
12-,10-, and 8-Bit DACs with
Integrated 10ppm/°C Reference
in 2mm × 2.1mm SC70.........................5
Kevin Wrenner, Troy Seman
and Mark Thoren
3µA Quiescent Current LDO
Improves Efficiency for Low Power
Circuits in Industrial, Automotive
and Battery-Powered Systems..............8
Sam Rankin
Triple Output LED Driver Delivers
3000:1 Dimming Ratio in Buck,
Boost or Buck-Boost Mode..................10
Bin Zhang
4.5A Monolithic LED Drivers with
3000:1 Dimming are Ideal for a Wide
Range of High Power LED Applications
..........................................................13
Mark W. Marosek
SAR ADCs Feature Speed,
Low Power, Small Package Size
and True Simultaneous Sampling......18
Steve Logan and Atsushi Kawamoto
A Cool Circuit: 48V Ideal Diode-OR
Reduces Heat Dissipation...................22
Dan Eddleman
Highly Integrated USB Power Manager
with Li-Ion Charger and Three
Step-Down Switching Regulators
in 4mm × 4mm QFN............................25
Amit Lele
DESIGN IDEAS
.....................................................29–41
(complete list on page 29)
New Device Cameos............................42
Design Tools.......................................43
Sales Offices......................................44
Introduction
The last few years have seen great place that swath of bandwidth starting
advances in the performance of analog- at DC. For example, with a 100Msps
to-digital converters. Sampling rates ADC, you can digitize signals that are
for 12-, 14- and even 16-bit ADCs bandpass limited between 150MHz
are now well above 100Msps. The and 200MHz. The total bandwidth
LTC®6400 differis still 50MHz,
ential amplifier
which is half the
has been specifisample rate, but
In the high performance
cally designed to
the input frequenreceiver systems of
drive these high
cies at which you
tomorrow’s wireless
performance ADC
operate are much
basestations, the analog
inputs in a way
higher.
that maintains
In modern
signal path that processes
their excellent low
communications
the IF frequency must be
noise and high
receiver systems,
highly linear and low noise.
linearity perforthe practice demance, all while
scribed above is
The LTC6400 fills that need
operating off a low
called IF sampling
in a way that is efficient in
3V or 3.3V supply
or undersampling.
board space and power use.
voltage.
The RF input signal is mixed down
IF Sampling
to an IF frequency
In addition to the higher sample rates, using a downconverting mixer such
the analog input frequency range of as the LT®5557. This IF frequency is
ADCs has been greatly expanded as digitized, and all further processing is
well. Long gone are the days when done digitally. To make this work for
you could only use an ADC with input the high performance receiver systems
frequencies no greater than half the in tomorrow’s wireless basestations,
sample rate. Is Harry Nyquist turning the analog signal path that processes
over in his grave, you ask? Not exactly. the IF frequency must be highly linear
It is still generally a good idea to limit and low noise. The LTC6400 fills that
the total signal bandwidth that gets need in a way that is efficient both in
digitized to one-half of the sample rate. terms of board space and power.
However, nobody says that you have to
continued on page L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology
Corporation. Adaptive Power, Bat-Track, BodeCAD, C-Load, DirectSense, Easy Drive, FilterCAD, Hot Swap, LinearView,
µModule, Micropower SwitcherCAD, Multimode Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational
Filter, PanelProtect, PowerPath, PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, True Color PWM,
UltraFast and VLDO are trademarks of Linear Technology Corporation. Other product names may be trademarks of the
companies that manufacture the products.
L LINEAR IN THE NEWS
Linear in the News…
New Line of Power Management Chips
Linear Opens Expanded
Linear Technology has announced a new line of power Manchester, NH Design Center
management chips that combines unique high performance
power functions in compact formats for use in a wide
range of portable electronic products. The new product
line provides designers with simple, compact and reliable power management integrated circuits (PMICs) that
combine the key power functions for products including
media players, digital cameras, smart phones, personal
navigation devices, PDAs, satellite radios, point-of-sale
terminals, portable medical equipment and other lithium
battery-powered devices.
“Today’s designers are challenged to develop portable
electronic products in increasingly short time-frames that
are both highly compact and efficient in power delivery,”
according to Don Paulus, Vice President and General Manager of Linear Technology’s Power Management Products.
“Our new PMIC family provides a new level of performance
by combining all the key power functions needed for each
application.”
Linear’s new LTC35XX PMIC family was developed
in response to the growing need for power management
solutions for portable electronic products. The first device
in Linear’s new PMIC family, the LTC3555 USB Power
Manager and Triple Step-Down DC/DC Converter, is now
available, with other family members coming soon. The
device incorporates a range of power management functions
including a switching PowerPath™ manager, a stand-alone
battery charger, three monolithic buck regulators and
always-on LDO, controlled via an I2C interface, housed
in a tiny 4mm × 5mm package. The switching PowerPath
control feature seamlessly manages power flow between
an AC/DC wall adapter, USB port, lithium-ion/polymer
battery and system load, while maximizing power available
from the USB and providing up to 1.2A to the system from
the wall adapter. The chip’s “instant-on” feature ensures
system power, even with a dead or missing battery.
Linear Technology announced the expansion of its Manchester, New Hampshire Design Center with the opening
of a new 20,000 square foot design facility. The new Manchester Design Center facility, one of twelve centers focused
on design of high performance analog integrated circuits,
includes design facilities, lab and test development floor.
With the company’s growth, Linear has outgrown its prior
New Hampshire Design Center facility and is relocating to
a new, state-of-the-art design facility.
Lothar Maier, CEO of Linear Technology, stated, “The
Manchester Design Center facility will allow us to grow
our team of analog designers from the rich talent base in
the New Hampshire/Boston area. Our new facility has a
highly favorable location, close to world-class technical
universities. We expect the Manchester Design Center to
further increase its contribution of innovative products
to serve the broad analog market, which will further fuel
the company’s growth.”
Linear Technology’s eleven other design centers are
located in Boston, Massachusetts; Burlington, Vermont;
Colorado Springs, Colorado; Dallas, Texas; Grass Valley,
California; Phoenix, Arizona; Raleigh, North Carolina; Santa
Barbara, California; Singapore; Munich, Germany, and at
the company headquarters in Milpitas, California.
Linear Technology Products
Selected as Ultimate Products
EE Times in April published their latest list of Ultimate
Products, selected by their readers and editors as best-inclass, and highlighted three Linear Technology products
as top 10 products in the Power Products category.
The publication selected Linear’s LTC4263 PSE Controller for Power over Ethernet and stated, “Linear Technology
touts the current-sharing, stand-alone capabilities of
its LTC4263 single-channel IC for Power over Ethernet
(PoE) as unique among Power Sourcing Equipment (PSE)
controllers.”
EE Times also selected Linear’s LTM4601 and LTM4603
µModule™ controllers as Ultimate products, with the headline, “Micromodules simplify point-of-load applications to
48 amps.” The LTM4601 is a 12A DC/DC is a µModule with
PLL, Output Tracking and Margining and the LTM4603
is a 6A DC/DC µModule with PLL, Output Tracking and
Margining. EE Times’ stated, “With capabilities far beyond
the company’s first-generation, high-voltage LTM4600
point-of-load DC/DC supply, the LTM4601 and LTM4603
series from Linear Technology, part of the µModule series
of ‘one-chip’ power supplies, adds margining/tracking,
remote-sense, expanded polyphase/paralleling capacity,
and phase-locked loop synchronization functionality for
advanced 6-48 amp designs. In addition, the new LTM4602
is a modified version of the LTM4600.” L
Linear Technology Magazine • June 2007
DESIGN FEATURES L
LTC6400-20, continued from page Performance
without Precedent
Figure 1 shows the intermodulation
distortion vs input frequency for a
2VP–P output signal. The LTC6400
achieves distortion at the –90dBc level
up to 140MHz, and at the –70dBc level
up to a couple hundred MHz. Previously, this type of performance was
only achievable using much higher
power RF gain blocks (which typically aren’t even differential). Figure 2
shows the equivalent OIP3 (3rd order
output intercept point), which is an RF
figure-of-merit that expresses output
linearity irrespective of signal level.
Besides distortion, the other key
performance requirement of an IF ADC
Driver is low noise contribution. The
LTC6400 is based on a differential op
amp with a very quiet 1nV/√Hz input
noise density. The internal 200Ω differential input resistors inevitably add
some noise of their own, resulting in a
–40
–50
IMD3 (dBc)
–60
–70
SINGLE-ENDED DRIVE
–80
DIFFERENTIAL DRIVE
–90
–100
–110
VOUT = 2VP–P COMPOSITE
1MHz TONE SPACING
0
50
200
100
150
FREQUENCY (MHz)
250
12
Figure 1. The LTC6400 maintains low
intermodulation distortion up to hundreds
of MHz, allowing for high performance IF
sampling applications
OUTPUT IP3 (dBm)
V–
11
ENABLE
The LTC6400 differential amplifier is
manufactured on an advanced complementary bipolar silicon-germanium
(SiGe) process. Because germanium
atoms are larger than silicon atoms,
selectively adding some germanium to
an otherwise silicon process causes
strains within the material’s crystalline
structure. This strain actually results
in beneficial electrical properties, such
as higher carrier mobility and a more
precise control of the base-width, allowing for faster transistors.
Figure 3 shows a block diagram
of the LTC6400. At its core is a very
high speed differential op amp. The
combination of fast transistors and
streamlined circuit topology results
in an op amp with a gain-bandwidth
10
50
+IN
14
40
–IN
15
30
–IN
16
RF
1000Ω
RG
100Ω
IN+
200
100
150
FREQUENCY (MHz)
250
300
Figure 2. The LTC6400 Equivalent Output-IP3
is in excess of 50dBm up to 100MHz, and in
excess of 40dBm up to 250MHz
Linear Technology Magazine • June 2007
+OUT
8
RFILT
50Ω
OUT–
IN–
+OUTF
7
CFILT
2.7pF
1
V+
2
VOCM
–OUTF
6
OUT+
RF
1000Ω
RG
100Ω
ROUT
12.5Ω
–OUT
5
COMMON
MODE CONTROL
5.3pF
50
V–
ROUT
12.5Ω
2k
0
9
RFILT
50Ω
10
0
V+
BIAS CONTROL
+IN
13
20
A Look under the Hood
2.1nV/√Hz total input referred noise
density.
In RF terms, when terminated in a
matched 200Ω system, this translates
to a noise figure of only 6.1dB. Since
the LTC6400 is typically the last stage
before the ADC in a receiver line-up
there are other gain blocks that precede it. To refer a component’s noise
contribution to the actual input of
the entire receiver, you divide it by
the gain that precedes it. Therefore,
300
60
the low 6.1dB noise figure of the
LTC6400 allows for very low noise
receiver designs.
Another way to look at noise is in
terms of SNR (signal-to-noise ratio).
The LTC6400-20 output noise density is 21nV/√Hz (because the gain is
10V/V). If you limit the input signal
bandwidth to a generous 50MHz, this
amounts to 148µVRMS of integrated
noise. This allows for a 74dB SNR
relative to a 2VP-P full-scale signal,
compatible with popular 14-bit ADCs
such as the LTC2249.
The LTC6400 differential
amplifier is manufactured
on an advanced
complementary bipolar
silicon-germanium (SiGe)
process, which allows for
faster transistors. At the
core of the LTC6400 is a
very high speed differential
op amp. The combination
of fast transistors and
streamlined circuit topology
results in an op amp
with a gain-bandwidth
product in excess of 3GHz
relative to a unity-gain
stable transfer function.
3
V+
4
V–
Figure 3. The LTC6400 combines a very high speed
differential op amp with on-chip feedback resistors
L DESIGN FEATURES
25
20
GAIN (dB)
product in excess of 3GHz relative to
a unity-gain stable transfer function.
Furthermore, all feedback resistors
are integrated. In addition to the obvious space savings, integrating the
feedback network results in several
design benefits:
qThe sensitive summing nodes
at the immediate inputs of the
op amp are not exposed to the
vagaries of board layout, which
allows us to carefully control the
amount of parasitic capacitance
seen at that node. Otherwise,
even as little as 100 femtoFarads at this node (for example
due to board traces, package
pins, or bond pads) would cause
unwanted poles in the loop-gain
of the feedback network.
qIf the feedback resistors were offchip, two sets of bond wires (at
the op amp outputs and inputs)
would be in the feedback loop. On
chip resistors eliminate bond wire
or lead inductance associated
with the op amp inputs, and
those at the op amp outputs are
outside of the feedback loop. At
frequencies of 3GHz and above,
even a small 1nH of inductance
exhibits significant impedance
and phase shift, which would
again limit the achievable speed
and performance.
qSince the gain is fixed and higher
than unity, we can internally
de-compensate the op amp to
achieve the maximum possible
open-loop gain for a given closedloop configuration. The more
open-loop gain, the better the
15
10
5
0
10
100
1000
FREQUENCY (MHz)
Figure 4: The op amp inside the LTC6400-20
is internally decompensated, so that even
though the closed loop gain is 10V/V (20dB),
the closed loop –3dB bandwidth is still an
impressive 1.8GHz
feedback action can suppress
non-linear components. In
addition, this compensation
technique preserves a wide –3dB
bandwidth even though the gain
is high, as shown in Figure 4.
Application Example
Figure 5 shows a typical application
of the LTC6400 driving the LTC2208,
a 16-Bit 130Msps ADC. In this case,
the input signal is single-ended, and
applied to the +IN input of the LTC6400
through a DC-blocking capacitor.
(With a little bit of care, the signal
could also be DC-coupled, so long
as the DC voltage is within the input
common mode range of the amplifier.) As can be readily observed from
Figure 3, the input impedance of the
LTC6400-20 is 200Ω differential. The
66.5Ω input resistor changes the total
input impedance to 50Ω, to provide a
match to a 50Ω source impedance.
Alternatively, a 1:4 transformer may
3.3V
1.25V
0.1µF
1000pF
0.1µF
0.1µF
V+
0.1µF
VOCM
+IN
66.5Ω
0.1µF
28.7Ω
+OUT
+OUTF
LTC6400-20
–OUTF
–OUT
–IN
V–
ENABLE
20dB GAIN
10Ω
AIN+
VCM
VDO
LTC2208
10Ω
AIN–
LTC2208 130Msps
16-Bit ADC
Figure 5. The LTC6400 can drive high performance
ADCs with a minimum of external components
3000
be used for matching the amplifier to
a 50Ω load. In other cases, the source
impedance may already be 200Ω and
no additional components would be
necessary. The 29Ω resistor placed
at the –IN input provides a balanced
termination for the internal op amp.
The LTC6400 is powered from the
same 3.3V as the ADC, saving the
need for another power supply rail.
It could do the same with a 3V rail.
Other driver solutions require 5V or
more to drive ADCs to full-scale with
high performance.
The LTC2208 family of ADCs wants
to see its inputs swing centered around
a 1.25V common mode voltage. The
LTC6400 makes this easy: simply
connect the VCM pin of the ADC to
the VOCM pin of the LTC6400, and
the amplifier’s internal commonmode feedback loop ensures that the
outputs swing centered around the
value applied to VOCM. For ADCs that
prefer a 1.5V common-mode voltage,
the interface is the same.
Related Parts
The LTC6401-20 is a lower power
version of the LTC6400-20. The
LTC6401-20 consumes only 45mA
at 3V or 3.3V. Both amplifiers are
pin-compatible and have the same
low noise performance. The LTC6401
maintains excellent linearity up to
140MHz, while consuming only half
the power of the LTC6400.
Conclusion
By combining a new SiGe process
with careful, innovative design, the
LTC6400 offers unprecedented performance at high frequencies, all while
operating at a low 3V or 3.3V supply
voltage. A tiny 9mm2 leadless package, along with a minimal number of
external components, lets you place
the driver right at the ADC inputs,
providing the best performance and
compact board layout. The differential outputs are uniquely optimized
to directly drive state-of-the art high
speed ADCs with high linearity, while
the low input-noise preserves the sensitivity of a high performance receiver
system. L
Linear Technology Magazine • June 2007
DESIGN FEATURES L
12-,10-, and 8-Bit DACs with
Integrated 10ppm/°C Reference
in 2mm × 2.1mm SC70
by Kevin Wrenner, Troy Seman and Mark Thoren
Introduction
Because the output voltage range
of a DAC is directly proportional to
its reference voltage, the accuracy
of the reference directly impacts the
accuracy of the output. Despite the
critical nature of the reference voltage,
it is often overlooked, and simply tied
to a power supply rail. This makes
the DAC output track the power supply—including its inaccuracies and
noise, which may be unspecified and
quite large.
In the LTC2630 family of smallfootprint DACs, a high performance
voltage reference is built in (Figure 1),
eliminating the need for an external
reference. The LTC2630 provides an
unprecedented combination of accuracy, small size, integrated reference
and ease of use, making it ideal for
applications from general-purpose
voltage adjustment in analog signal
conditioning circuits to high accu-
racy industrial controls. An H-grade
version that operates over a –40°C to
+125°C temperature range is available
for demanding industrial, military, or
automotive applications.
Full Scale Defined by
Integrated Reference
or Supply
The LTC2630’s integrated reference
provides a full-scale voltage that is
low drift (±10ppm/°C) and insensitive to supply voltage variations. The
LTC2630-L has a full-scale output
of 2.5V and operates from a single
2.7V to 5.5V supply. The LTC2630-H
has a full-scale output of 4.096V and
operates from a 4.5V to 5.5V supply.
When configured in supply-as-reference mode, the output of the LTC2630
can swing rail-to-rail referenced to the
input supply.
Tiny SC70 Footprint and
Ultralow Power
The LTC2630
offers an unprecedented
combination of accuracy,
small size,
integrated reference
and ease of use,
making it ideal for a
wide range of applications.
The LTC2630 fits the 12-, 10-, or 8bit DAC and internal reference in an
ultracompact 6-lead SC70 package
(2mm × 2.1mm). Power consumption
is low, too. When operating in internal
reference mode, supply current is just
180µA at 3V. Performance of the DAC,
however, is anything but low.
2.7V TO 5.5V (LTC2630-L)
4.5V TO 5.5V (LTC2630-H)
0.1µF
VCC
INTERNAL
REFERENCE
SDI
CONTROL
DECODE LOGIC
SCK
µP
RESISTOR
DIVIDER
24-BIT
SHIFT
REGISTER
DACREF
CS/LD
INPUT
REGISTER
DAC
REGISTER
DAC
VOUT
0V TO VCC, OR
0V TO 2.5V (LTC2630-L)
0V TO 4.096V (LTC2630-H)
GND
Figure 1. The LTC2630 integrates a high performance rail-to-rail amplifier,
10ppm/°C reference, and double-buffered input data path in an SC70 package.
Linear Technology Magazine • June 2007
L DESIGN FEATURES
Outstanding DAC
Performance
1.0
1.0
LTC2630-L12
VCC = 3V
INTERNAL REF.
Excellent Load Regulation Means
Hidden Error is Reduced
The LTC2630’s output buffer is guaranteed to be capable of sourcing and
sinking 5mA at 2.7V and 10mA at
4.5V. Its high gain amplifier holds the
output resistance at only 0.1Ω (0.156Ω
max) despite having a single GND pin.
Figure 4 shows how this minimizes
output voltage error due to DC loading—only 0.1 LSB per mA of load
current (0.16 LSB/mA max) for the
LTC2630-12H and 0.13 LSB per mA
(0.256 LSB/mA max) for the LTC263012L. In comparison, the lowest DC
DNL (LSB)
0.5
0
–0.5
–1.0
0
–0.5
0
1024
2048
3072
–1.0
4095
0
1024
CODE
2048
3072
4095
CODE
Figure 2. Integral and differential nonlinearity in internal reference mode.
The LTC2630’s excellent DNL guarantees its monotonicity.
0.8
OFFSET ERROR (mV)
2
1
0
–1
LTC2630-H
VCC = 5V
INTERNAL REF.
–2
–3
–50 –25
0
FULL SCALE ERROR (%FSR)
3
0.4
0
–0.4
LTC2630-H
VCC = 5V
INTERNAL REF.
–0.8
–50 –25
25 50 75 100 125 150
TEMPERATURE (°C)
0
25 50 75 100 125 150
TEMPERATURE (°C)
Figure 3. Low-drift offset error voltage and full-scale error voltage.
output impedance of any competitor
is 0.5 Ω, easily introducing five times
greater load-induced error.
Easy Operation
The LTC2630 family operates off a
single supply and can drive loads
up to 500pF without any stability
concerns.
Its simple SPI/MICROWIRE-compatible 3-wire interface can be
operated at clock rates of up to 50MHz.
Setup and hold times of only 4ns allow
problem-free operation in optoisolated
and other applications having slow
edge rates. The internal data registers are double-buffered, allowing
simultaneous updating of multiple
devices in a system. All three parts
in the LTC2630 family use the same
24-bit load sequence (32-bit is also
supported). There are six command
codes for selecting internal or supply reference modes, powering down,
writing to the input register, updating
the DAC register and performing a
combined write and update.
Other Features
At power up, the internal reference is
selected by default, and the code is
reset to either midscale (LTC2630-M)
or zero (LTC2630-Z). Internal circuitry
holds the output glitch to less than
5mV if the supply is ramped no faster
than 1V/ms.
The LTC2630 can be placed in a
power-saving mode in which current
10
8
6
4
∆VOUT (mV)
Predictable and Usable
Output Range
Over its rated temperature range, the
LTC2630 has a maximum offset of
±5mV. The low offset enables a starting code voltage closer to 0V than
competing devices. When full scale is
set by the internal reference, the fullscale error voltage is just ±0.8% of the
full-scale range (FSR), and linearity is
guaranteed to the upper code limit.
The invariance of these parameters
over temperature is shown in Figure 3.
Together, low offset and low full-scale
error define a predictable output range
and maximize the number of usable
codes.
INL (LSB)
0.5
Linearity: at 12-Bit Accuracy, DNL
and INL are Guaranteed ±1LSB
The LTC2630 family uses Linear
Technology’s proprietary, inherently
monotonic voltage interpolation architecture, the benefits of which can be
seen in Figure 2. For the LTC2630A12, the DNL is ±0.2 LSB, the INL is
±0.5LSB, and both are guaranteed
to be less than ±1 LSB over the full
operating temperature range of the
part. For the LTC2630-12, DNL and
INL are guaranteed to ±1 LSB and ±2
LSB over temperature, respectively.
At 10 bits (LTC2630-10), DNL and
INL are guaranteed less than ±0.5
LSB and ±1 LSB over temperature,
respectively. At 8 bits (LTC2630-8),
both are guaranteed less than ±0.5
LSB over temperature.
LTC2630-L12
VCC = 3V
INTERNAL REF.
2
0
–2
–4
LTC2630-L
VCC=5V
INTERNAL REF.
CODE = MIDSCALE
–6
–8
–10
–30
–20
–10
0
10
IOUT (mA)
20
30
Figure 4. Load regulation. The high drive
output buffer is guaranteed to source and sink
5mA at 3V, and 10mA at 5V, well inside the
bounds of current limiting. Output resistance
of only 0.1Ω keeps the error contributed by DC
loading to a minimum.
Linear Technology Magazine • June 2007
DESIGN FEATURES L
VLOOP
5.4V TO 80V
LT3010-5
IN
OUT
SHDN SENSE
1µF
ROFFSET
374k
0.1%
+
1µF
GND
FROM
OPTOISOLATED
INPUTS
SDI
SCK
VCC
LTC2630-HZ
VOUT
RGAIN
76.8k
0.1%
CS/LD
+
1k
LTC2054
3.01k
–
10k
Q1
2N3440
1000PF
RS
10Ω
5V
OPTOISOLATORS
SDI
SCK
CS/LD
499Ω
10k
4N28
IOUT
SDI
SCK
CS/LD
Figure 5. Optoisolated 4mA to 20mA process controller. This circuit digitizes an output current for use in an isolated control loop.
draw at 5V is reduced to below 1.8µA
(5µA for H-grade operating at 125°C).
Upon exiting power down mode, the
output settles at midscale to 12-bit
accuracy in 18µs.
Optoisolated 4mA to 20mA
Process Controller
LTC2630 is well-suited to industrial
applications, including control loops.
Figure 5 shows an optically-isolated,
digitally-controlled 4mA to 20mA
transmitter using the LTC2630HZ.
The transmitter circuitry, including
optoisolation, is powered by the loop
voltage, which has a wide 5.4V to 80V
range. The 5V output of the LT30105 sets the 4mA offset current and
the DAC digitally controls the 0mA
to 16mA signal current. The supply
current for the regulator, DAC and
op amp is well below the 4mA budget
Table 1. Available part options. The LTC2630 is offered in twelve
combinations of full-scale voltage, power-on reset, and accuracy.
Full-Scale
Reference
Power-On
Reset Code
Accuracy
(Bits)
VCC
(V)
LTC2630-LM
2.5V
Midscale
12
10
8
2.7–5.5
LTC2630-LZ
2.5V
Zero
12
10
8
2.7–5.5
LTC2630-HM
4.096V
Midscale
12
10
8
4.5–5.5
LTC2630-HZ
4.096V
Zero
12
10
8
4.5–5.5
Linear Technology Magazine • June 2007
at zero scale. RS senses the total loop
current, which includes the quiescent
supply current and additional current
through Q1. Note that at the maximum
loop voltage of 80V, Q1 dissipates 1.6W
when IOUT is 20mA, so it must have
an appropriate heat sink.
The values of ROFFSET and RGAIN
are as close to ideal as possible using
0.1% resistors to meet the 4mA–20mA
design objective. Alternatively, ROFFSET
can be a 365k, 1% resistor in series
with a 20k trim pot and RGAIN can be
a 75.0k, 1% resistor in series with a
5k trim pot. If the application calls for
a high speed serial bus, use 6N139
rather than 4N28 optocouplers.
Conclusion
The LTC2630 is a family of single
voltage output DACs in 6-lead SC70
packages with integrated references.
Each DAC can provide its own accurate full-scale voltage and can operate
rail-to-rail referenced to the input
supply. Twelve options are available
in various combinations of accuracy
(12-, 10-, and 8-bit), full-scale voltage (2.5V or 4.096V), and power on
reset value (zero or midscale); see
Table 1. L
L DESIGN FEATURES
3µA Quiescent Current LDO Improves
Efficiency for Low Power Circuits in
Industrial, Automotive and
Battery-Powered Systems
by Sam Rankin
Introduction
Ultralow Quiescent Current
PNP LDO
Figure 1 shows a typical application
for the LT3009, a 3µA quiescent current low dropout linear regulator in
tiny 2mm × 2mm DFN and 8-lead
SC70 packages. Its ultralow 3µA quiescent current is well controlled—it
does not rise excessively in dropout
as happens with many regulators.
Quiescent current is less than 5% of
1.4
0.8
0.6
0.4
0.2
1µF
2.8M
1%
LT3009
ADJ
VOUT
3.3V
20mA
619k
1%
Figure 1. New 3µA quiescent current low dropout regulator
1000
GND CURRENT (µA)
output current at 20mA IOUT, even in
dropout (Figure 2).
The LT3009 can supply up to 20mA
from input supplies ranging from 1.6V
to 20V to output voltages ranging from
0.6V to 19.5V. Dropout voltage on the
LT3009 is only 280mV while delivering up to 20mA of output current. It
can be put into a low power shutdown
state by pulling the SHDN pin low. In
shutdown state, the already low quiescent current is reduced to the leakage
currents of the internal transistors.
This leakage, typically a few nA at
room temperature, stays below 1µA
over the entire operating temperature
range. Low quiescent current and tiny
package size does not translate into
poor performance in the LT3009. The
LT3009 features industry leading load,
line, and temperature regulation (see
Figures 3, 4 and 5)
VIN = 3.8V
VOUT = 3.3V
100
10
1
0.001
0.01
0.1
1
LOAD (mA)
10
100
Figure 2. GND Pin current vs ILOAD
Aside from the output voltage
setting resistors, the only external
components required are input and
output bypass capacitors. Internal
frequency compensation in the LT3009
stabilizes the output for a wide range
of capacitors. A minimum of 1µF of
0.6
0.612
0.610
0.608
0.5
0.4
0.3
0.2
0.606
0.604
0.602
0.600
0.598
0.596
0.594
0.592
0.1
0
0.590
–40 –20
0
20 40 60 80
TEMPERATURE (°C)
100 120
Figure 3. Load regulation vs temperature
1µF
OUT
GND
LINE REGULATION (mV)
LOAD REGULATION (mV)
1.0
IN
SHDN
ΔIL = 1µA TO 20mA
VOUT = 600mV
VIN = 1.6V
1.2
–0.2
VIN
3.75V TO
20V
ADJ PIN VOLTAGE (mV)
Many electronic systems spend much
of their time in an idle state, waiting
for something to happen. Industrial
remote monitoring systems and keepalive circuits are but two examples.
Many of these systems depend on
battery power, so a high efficiency
power supply is paramount to preserve battery life. Efficiency during
quiescent state is of particular importance since active operation may
draw milliamps while quiescent operation only microamps. Small size and
reverse output and input protection
capabilities are also desirable features
in a power supply. This is a demanding
combination of power supply requirements, but there is an easy way to
satisfy them with one device.
0
–40 –20
0
20 40 60 80
TEMPERATURE (°C)
100 120
Figure 4. Line regulation vs temperature
0.588
–40 –20
0
20 40 60 80
TEMPERATURE (°C)
100 120
Figure 5. Output voltage vs temperature
Linear Technology Magazine • June 2007
DESIGN FEATURES L
output capacitance is required for
stability, and almost any type of output capacitor can be used. Even small
ceramic capacitors with low ESR can
be used without the additional series
resistance commonly required with
other regulators. The combination of
small package size and the ability to
use small ceramic capacitors enable
the LT3009 to fit almost anywhere.
The LT3009 has a number of protection features to safeguard itself
and sensitive load circuits. Should the
input voltage become reversed (due to
a battery inserted backwards or a fault
on the line, for example), current flow
from the IN pin is limited by a 100k
resistance and no negative voltage is
seen at the load. No external protection
diodes are necessary when using the
LT3009. With a reverse voltage from
output to input, the LT3009 acts as
though it has a 500k limiting resistor
in series with two diodes from output
to input to limit reverse current flow.
For dual-supply applications where
the regulator load is returned to a
negative supply, the OUT and ADJ
pins can be pulled below ground (by up
to a 20V input-to-output differential)
while still allowing the device to start
and operate. The LT3009 also includes
protection features found standard on
linear regulators such as current and
thermal limiting.
The Ideal Solution for
Remote Monitoring
The LT3009 provides an optimum
solution for remote monitoring applications. The duty cycle of many of
these applications is very short—they
spend most of their time in shutdown,
waking briefly to take and communicate measurements, then returning
immediately to shutdown. Aside from
LINE POWER
VLINE
12V TO 15V
DCHARGE
IN
1µF
OUT
5V
IN
SUPERCAP
1µF
SHDN
ADJ
GND
4.32M
1%
LT3009
ADJ
GND
1µF
FAULT
GND
TO
MONITORING
CENTER
590k
1%
Figure 6. Typical last-gasp circuit
the typical supply regulation requirements required by sensitive analog
circuitry (tight supply regulation,
quiet supply, load protection, etc.), the
principle supply requirement is low
quiescent power consumption. With
its 3µA quiescent current coupled with
industry leading supply regulation
capability and myriad of protection
features, the LT3009 fits the bill.
A typical remote monitoring application used frequently in utility
meters is a “last-gasp” circuit, shown
in Figure 6. In this application, a 12V
to 15V supply derived from line power
charges a large capacitor (SuperCap)
through a diode and a current limiting resistor. This stored voltage on the
SuperCap provides input voltage for
the LT3009. The LT3009 provides a
quiet, well-regulated 5V supply to the
analog fault detection circuits as well
as a digital communication module
used to send distress signals to the
remote monitoring center. The fault
detection circuitry is typically active for
only a few hundred milliseconds every
15-minute detection cycle. In the event
of a line failure, the ultralow quiescent
current of the LT3009 enables the SuperCap to provide enough power to the
3.3V
1µF
LOAD:
SYSTEM MONITOR,
VOLATILE MEMORY, ETC
619k
1%
Figure 7. Typical keep-alive power supply
Linear Technology Magazine • June 2007
PWR
OUT
SHDN
2.8M
1%
LT3009
LINE
INTERRUPT
DETECT
RLIMIT
NO PROTECTION
DIODES NEEDED!
VIN
12V
SENSE
fault detection and communications
circuitry for several detection cycles.
The 3µA quiescent current of the
LT3009 reduces the required size and
cost of the SuperCap while simultaneously extending the life of the detection
and communications circuits after
line failure. Additionally, with its
output regulation of ±2% over load
line and temperature, the LT3009 can
do double duty as a highly accurate
voltage reference for the fault detection circuits.
An Excellent Choice for
Keep-Alive Power Supplies
Switching power supplies provide
robust local low voltage/high current
power from high voltage rails, but
switching power supplies are overly
complex for the low power keep-alive
circuits that typically run only a few
milliamps of current. There are many
such low current applications in industrial, monitoring, security systems,
smoke detectors, and other always-on
circuits. For many of these applications, the LT3009 provides a relatively
simple and inexpensive solution.
A typical keep-alive application is
shown in Figure 7. A 12V rail powers
a keep-alive circuit for monitoring or
other purposes. Low quiescent current is critical here to reduce battery
drain. A battery backup keeps the
output alive when a fault on the input
occurs. Should a fault on the 12V rail
occur, the battery backup takes over.
The internal protection of the LT3009
limits current flow from the output
back to the input, removing the need
for protection diodes.
continued on page 24
L DESIGN FEATURES
Triple Output LED Driver Delivers
3000:1 Dimming Ratio in Buck,
by Bin Zhang
Boost or Buck-Boost Mode
Introduction
The LT3496 is a triple output DC/DC
converter designed for high performance, True Color PWMTM dimming
in multichannel LED lighting applications. By integrating three independent
driver channels, the LT3496 provides
a space-saving and cost-efficient solution to drive multiple LED strings.
Figure 1 shows a 50W LT3496 3-channel LED driver that occupies 350mm2
and with a sub-1.5mm profile.
SIMPLIFIED TRADITIONAL
LED DRIVER BOARD
SIX WIRES
L1
15µH
LED
DRIVER
ratio in buck, boost, or buck-boost
configurations. The 45V capability of
the internal power switch, 3V–40V
input voltage range, and adjustable
frequency result in reliable operation
over a wide range of supply and output
voltages. Applications for the LT3496
include RGB lighting, billboards and
large displays, automotive and avionic lighting, and constant-current
sources.
Figure 1. A complete LT3496
LED driver fits into 350mm2
The LT3496 features high side current sensing and built-in gate drivers
for PMOS high side LED disconnect
(patent pending). These two features
give the LT3496 its versatility, allowing
it to drive LED’s to high PWM dimming
M1
LED
DRIVER
M2
PVIN
42V
LED
DRIVER
High Side LED Disconnect
with High Side Current
Sensing for System
Versatility, Simplicity
and Reliability
The LT3496’s high side LED disconnect and high side current sensing
enable 3000:1 dimming control in
buck, boost, or buck-boost configurations. No traditional LED driver can
match the simplicity and high PWM
CAP1
CAP2
0.28Ω
TG1
a. Traditional boost LED driver
SIMPLIFIED LT3496
LED DRIVER BOARD
L1
15µH
THREE WIRES
TG2
M2
350mA
LED3
M3
TG3
350mA
C5
C4
0.47µF
0.47µF
D1
0.28Ω
LED2
C1-C3
1µF
×3
350mA
C6
0.47µF
L2
D2 15µH
L3
15µH
D3
M2
M3
VIN
3.3V TO 24V
C7
1µF
b. LT3496-based boost LED driver
Figure 2. An LT3496-based boost LED driver
requires half as many wires as a traditional
boost LED driver
10
M1
7 LEDs
M1
LT3496
0.28Ω
LED1
M3
CAP3
PWM1-3
SHDN
SW1
CAP1-3
LED1-3
VIN
PWM1-3
SHDN
SW2
SW3
LT3496
GND
TG1-3
VC1-3
VREF
CTRL1-3
FADJ
OVP1-3
C8-C10
100nF
C1, C2, C3: MURATA GRM31MR71H105KA88
C4, C5, C6: MURATA GRM21BR71H474KA88
C7, GRM188R71C105KA12
L1, L2, L3: TAIYO YUDEN NP04SZB 150M
M1, M2, M3: ZETEX ZXMP6A13F
D1, D2, D3: DIODES DFLS160
Figure 3. The LT3496 RGB driver for large TFT LCD TVs
Linear Technology Magazine • June 2007
DESIGN FEATURES L
IL
0.5A/DIV
ILED
0.5A/DIV
0.5µs/DIV
Figure 4. 5000:1 dimming waveforms
for the application circuit of Figure 3
dimming performance of LT3496,
especially in buck-boost mode. Implementation of a high side disconnect
switch with traditional LED drivers
is possible, but uses many additional
components, has slow response and
burns extra power.
Because the LED disconnect and
current sensing are on the high side
of each LED string, the low sides of
the LED strings can be tied together
in boost or buck-boost mode to reduce
the number of wires returning to the
LED driver. In a boost configuration,
each of the low side connections can
be returned to ground anywhere, allowing a simple 1-wire LED connection
VIN
8V TO 30V
90
85
80
75
70
65
60
55
50
0
20
40
60
80
100
PWM DUTY CYCLE (%)
Figure 5. Efficiency of the application
circuit of Figure 3
here. If the PWM1 pin is pulled low, M1
is turned off, disconnecting the LED
string of channel 1 and stopping the
current draw from output capacitor
C4. The VC1 pin is also disconnected
from the compensation capacitor C8.
C4 stores the state of the LED voltage
and C8 stores the state of the LED
current until PWM1 is pulled up again.
This leads to a highly linear relationship between pulse width and output
light, a large and accurate dimming
range, and high efficiency. At 120Hz
PWM frequency, the PWM control of
the circuit allows 5000:1 dimming as
shown in Figure 4. Figure 5 shows the
Applications
Buck Mode LED Driver
The LT3496 can be configured as a
buck mode LED driver for applications
where the LED voltage is lower than
the supply voltage. Figure 3 shows an
LT3496 RGB driver for a large TFT
LCD TV.
The three LT3496 channels operate
independently, but function in the
same way. For simplicity, the PWM
operation of channel 1 is described
C1
3.3µF
150mA
150mA
L1
22µH
150mA
L2
22µH
L3
22µH
M1
TG1
TG2
LED1
TG3
0.68Ω
CAP2
D1
C2
0.1µF
C3
1µF
R1
3.9M
OVP1
R2
100k
CAP3
D2
C4
0.1µF
VIN
C5
1µF
R3
3.9M
OVP2
R4
100k
D3
C6
0.1µF
VIN
SW2
LT3496
GND
M3
LED3
0.68Ω
CAP1
SW1
CAP1-3
LED1-3
VIN
PWM1-3
SHDN
M2
LED2
0.68Ω
PWM 1-3
SHDN
95
EFFICIENCY (%)
PWM
5V/DIV
100
for each LED string. Traditional LED
drivers employ a low side LED disconnect approach, in which both the high
side and the low side of each LED
string must connect to the LED driver.
Figure 2a shows simplified traditional
boost LED drivers, where M1–M3
are LED-disconnect NMOS switches.
Figure 2b shows a simplified LT3496
triple boost LED driver, where M1–M3
are LED-disconnect PMOS switches.
The LT3496 solution removes three
wires, increasing system simplicity
and reliability. These advantages will
become increasingly important as
the channels are multiplied in high
performance displays.
C7
1µF
R5
3.9M
OVP3
R6
100k
VIN
SW3
TG1-3
OVP1-3
VC1-3
VREF
CTRL1-3
FADJ
C1: MURATA GRM55DR71H335KA0193
C3, C5, C7: MURATA GRM31MR71H105KA88
C2, C4, C6: GRM21BR71H104KA01
M1, M2, M3: ZETEX ZXMP6A13F
L1, L2, L3: TAIYO YUDEN NP04SZB 220M
D1, D2, D3: DIODES DFLS160
0.1µF
R7
75k
R8
24k
Figure 6. Buck-boost mode LED driver for automotive lighting
Linear Technology Magazine • June 2007
11
L DESIGN FEATURES
efficiency as a function of the PWM
duty cycle.
Buck-Boost Mode LED Driver
In some LED applications, the desired
supply voltage range and LED voltage
range overlap, thus requiring buckboost mode configuration. Figure 6
shows a LT3496 buck-boost mode LED
driver for automotive lighting. The LED
voltage is 9V–12V and the automobile
battery voltage is 8V–30V. R1–R6 set
the overvoltage protection voltage
at 40V to guarantee the voltages of
SW1–SW3, CAP1–CAP3, LED1–LED3,
and TG1–TG3 pins are below the
maximum rating voltage. R7–R8 set
the switching frequency at 1.3MHz to
limit the LT3496 power dissipation and
ensure that a junction temperature of
125°C is not exceeded. Figure 7 shows
the 3000:1 PWM dimming waveforms
at 120Hz PWM frequency.
Conclusion
IL
0.2A/DIV
ILED
0.2A/DIV
0.5µs/DIV
Figure 7. 3000:1 dimming waveforms
for the application circuit of Figure 6
the supply voltage. Figure 8 shows a
LT3496 boost LED driver for automotive lighting. D4, Q1–Q3, and R1–R4
create the battery surge voltage protection circuits to protect the LED string
from being damaged by a battery surge
voltage. The zener breakdown voltage
of D4 is chosen to be lower than the
LED voltage. When the VIN surge voltage increases to be close to the LED
voltage, D4 breaks down and turns on
Q1–Q3. Q1–Q3 pull PWM1–3 low and
M1–M3 are turned off immediately to
disconnect the LED strings from the
LED driver.
Boost LED driver
The LT3496 can be configured as a
boost LED driver for the applications
where the LED voltage is higher than
VIN
8V TO 16 V
Figure 9 shows the 3000:1 PWM
dimming waveforms at 120Hz PWM
frequency.
PWM
5V/DIV
C1
3.3µF
L1
15µH
D4
R1
1k
R2
1k
R3
1k
R4
1k
1Ω
TG2
M1
PWM2
0.1A
6 LEDs
20k
SW2
PWM1
SHDN
PWM3
1k
LT3496
PWM2
1k
PWM1
Q1
Q2
Figure 9. 3000:1 dimming waveforms
for the application circuit of Figure 8
D3
C4
1µF
CAP2
1Ω
CAP3
1Ω
LED3
TG3
M2
M3
825k
OVP1
SW1
SHDN
1k
0.5µs/DIV
LED2
VIN
PWM3
ILED
0.1A/DIV
D2
C3
1µF
CAP1
825k
6 LEDs
IL
0.5A/DIV
L3
15µH
LED1
TG1
PWM
5V/DIV
L2
15µH
D1
C2
1µF
The LT3496 provides a compact, low
cost, high reliability, and high efficiency solution to multichannel LED
lighting. With the capability of operating in buck, boost and buck-boost
mode, the LT3496 LED driver delivers
3000:1 True Color PWMTM dimming
ratio over a wide range of supply and
output voltages. L
GND
0.1A
825k
OVP2
6 LEDs
20k
SW3
CAP1-3
LED1-3
TG1-3
OVP1-3
VC1-3
VREF
FADJ
CTRL1-3
0.1A
OVP3
20k
100nF
Q3
C1: MURATA GRM55DR71H335KA0193
C2, C3, C4: MURATA GRM31MR71H105KA88
M1, M2, M3: ZETEX ZXMP6A13F
L1, L2, L3: TAIYO YUDEN NP04SZB 150M
D1, D2, D3: DIODES DFLS160
Figure 8. Boost mode LED driver with battery surge voltage protection for automotive lighting
12
Linear Technology Magazine • June 2007
DESIGN FEATURES L
4.5A Monolithic LED Drivers with
3000:1 Dimming are Ideal for
a Wide Range of High Power
LED Applications
by Mark W. Marosek
Introduction
The LT3478 and LT3478-1 are monolithic step-up DC/DC converters
specifically designed to drive high
brightness LEDs with a constant current over a wide programmable range.
They are extremely easy to use and
include programmable features for
optimizing performance, reliability,
size and overall solution cost. These
devices can operate in boost, buckmode boost and buck-boost mode
LED driver topologies. Depending on
the topology, they can provide up to
4A of LED current, a level unmatched
by other monolithic LED drivers. The
LT3478 and LT3478-1 are ideal for
high power LED applications, including automotive and avionic lighting,
and are available in a 16-pin thermally enhanced TSSOP package with
either E-grade or I-grade temperature
ratings.
The LT3478 and LT3478-1 operate similarly to conventional current
mode boost converters, but use LED
current (instead of output voltage) as
the main source of feedback for the
control loop. The block diagram in
Figure 2 shows the major functions
of each part. Both parts use high side
LED current sensing to extend operation to buck and buck-boost modes.
The LT3478-1 saves space and cost by
integrating the current sense resistor
and limits maximum LED current to
1.05A. The LT3478 uses an external
sense resistor to allow programming of
maximum LED current up to 4A.
boards and airplane cockpits, require
very high levels of PWM dimming. The
LT3478 and LT3478-1 offer a 3000:1
PWM dimming range (preserving LED
color) in addition to an optional 10:1
analog dimming range.
Current control for dimming is an
important feature, but it is just as
important to avoid overdriving LEDs
beyond their maximum rated current.
The LT3478 and LT3478-1 make it
easy to set the maximum current and
to derate the maximum current relative
to temperature.
Programming the LED
Current for Protection
and Dimming
Maximum LED Current
The LT3478 and LT3478-1 control
maximum LED current using the
voltage at the CTRL1 pin, unless the
device is set to derate the maximum
LED current relative to temperature
(using CTRL2 pin described below).
The voltage at CTRL1 pin can be set
using a simple resistor divider from
LEDs are a desirable lighting solution
in part because of their wide dimming
range via simple current control.
For instance, environments with the
potential for very low ambient light
conditions, such as automotive dashL1
10µH
VIN
8V TO 16V
C1
4.7µF
25V
VIN
VS
L
D1
C2
10µF
25V
SW
SHDN
OUT
100
VREF
R1
45.3k
CTRL2
LT3478-1
700mA
LED
95
EFFICIENCY (%)
OVPSET
R4
54.9k
CTRL1
R2
130k
PWM
SS
CSS
1µF
L1: CDRH104R-100NC
D1: PDS560
Q1: Si2318DS
LEDs: LUXEON III (WHITE)
VC
RT
CC
0.1µF
RT
69.8k
90
85
fOSC = 500kHz
80
3.3V
0V
ILED = 700mA
fOSC = 500kHz
PWM DUTY CYCLE = 100%
100Hz
10
Q1
PWM
DIMMING RATIO = 1000:1
6 LEDs LUXEON III (WHITE)
8
12
VIN (V)
14
16
R3
10k
Figure 1. Automotive TFT LCD backlight, 15W, 6 LEDs at 700mA, boost LED driver
Linear Technology Magazine • June 2007
13
L DESIGN FEATURES
SHDN
VS
11
L
4
SS
5
10µA
9.5mΩ
+
–
+
1.4V
VIN
VOUT
6
OVERVOLTAGE
DETECT
–
57mV
OVPSET
INRUSH
CURRENT
PROTECTION
UVLO
REF
1.24V
3
1, 2
VC
–
SW
16
+
100Ω
RSENSE
0.1Ω
(INTERNAL FOR
LT3478-1)
SOFT-START
RSENSE
(EXTERNAL FOR
LT3478)
LED
7
PWM
DETECT
VREF
10
S
Q
Q1
R
LED
PWM
+
+
+
–
–
+
LED
Σ
1000Ω
1V
PWM
+
13
GM
+
CTRL2
LED
SLOPE
COMP
Q2
–
12
LED
+
1.05V
–
CTRL1
OSC
14
RS
–
TO OVERVOLTAGE
DETECT CIRCUIT
8
15
OVPSET
RT
17
9
EXPOSED PAD
(GND)
VC
Figure 2. LT3478 and LT3478-1 block diagram
13
R2
12
LT3478/LT3478-1
VREF
VOUT
(LT3478)
RSENSE
CTRL2
CTRL1
LED
R1
Figure 3. Programming maximum LED current
LED CURRENT (mA)
1400
TA = 25°C
CTRL2 = VREF
(FOR LT3478 SCALE
BY 0.1Ω/RSENSE)
1050
LT3478-1
700
350
VREF
0
0
0.35
0.70
CTRL1 (V)
1.05
1.40
Figure 4. LED current vs CTRL1 voltage
14
VREF (see Figure 3), from an external
voltage source, or by connecting it
directly to the VREF pin for maximum
current. Figure 4 shows LED current
versus CTRL1 pin voltage.
Temperature-Based Derating of
the Maximum LED Current
To ensure optimum reliability, LED
manufacturers specify curves of
maximum allowed LED current versus
temperature (Figure 5). If the LED
current is not derated relative to temperature, it is possible to permanently
damage the LED.
The LT3478 and LT3478-1 enable
temperature derating via the CTRL2
pin. Simply connect CTRL2 to VREF
via a temperature-dependent resistor divider as shown in Figure 6. As
the temperature rises, the voltage at
CTRL2 falls. When CTRL2 falls below
CTRL1, the voltage at CTRL2 takes
over in setting the maximum LED
current (Figure 7).
900
800
If FORWARD CURRENT (mA)
10
700
LUXEON V EMITTER
CURRENT DERATING
CURVE
600
500
EXAMPLE
LT3478-1
PROGRAMMED LED
CURRENT DERATING CURVE
400
300
200
100
0
0
25
50
75
TA AMBIENT TEMPERATURE (°C)
100
LUXEON V EMITTER
(GREEN, CYAN, BLUE, ROYAL BLUE)
θJA = 20°C/W
Figure 5. LED current derating curve
vs ambient temperature
Linear Technology Magazine • June 2007
DESIGN FEATURES L
R4
13
12
R1
VREF
LT3478/LT3478-1
CTRL2
CTRL1
OPTION A TO D
R3
RY
RNTC
RNTC
A
RX RNTC
B
RY
RNTC
C
D
The temperature at which LED
current begins to decrease and the
rate of decrease are selectable by the
resistor network/values chosen. Table
1 lists several NTC resistor manufacturers. Murata Electronics notably
provides an online simulator to select
the required resistor combinations as
shown in Figure 6 including a catalog
describing the NTC resistor specifications. Figure 5 shows an example of
LT3478-1 programmed LED current
falling versus temperature using the
option C, shown in Figure 6, with R4
= 19.3k, RY = 3.01k and RNTC = 22k
(NCP15XW223J0SRC). A more detailed description of how to determine
these values by hand calculation is
given in the LT3478 and LT3478-1
data sheet.
Analog Dimming
Many LED applications require
accurate brightness control. LED
brightness can be reduced by simply
decreasing the programmed LED
VS
L
CTRL2
Contact
Murata Electronics
North America
www.murata.com
TDK Corporation
www.tdk.com
Digi-Key
www.digikey.com
COUT
SW
current, but reducing the operating
current of the LED changes the color
of the LED. This method is known as
analog dimming and is available in the
LT3478 and LT3478-1 by reducing the
voltage at the CTRL1 pin to as low as
0.1V (10:1 dimming from 1V). If color
preservation is important, then PWM
dimming is a better option.
PWM Dimming
PWM dimming (Figures 8 and 9) yields
high dimming ratios with no current-related LED color change. PWM
dimming is implemented in the LT3478
and LT3478-1 via the PWM pin. When
the PWM pin is active high (TPWM(ON))
or low, the LED current is either at
its maximum or off, respectively. The
LED on time, and hence the average
current, is controlled by the duty cycle
of the PWM pin. Because the LED is
always operating at the same current
(maximum set by CTRL1), and only the
average current changes, dimming is
achieved without changing the color
of the LED.
PWM dimming is not new, but the
ability to achieve high PWM dimming
ratios (requiring extremely low PWM
duty cycles) is challenging. The LT3478
and LT3478-1 use a patented architecture to achieve PWM dimming ratios
exceeding 3000:1 at 100Hz. The application circuit and waveforms shown
1000
900
800
CTRL1
700
600
500
400
CTRL2
300
200
LED CURRENT = MINIMUM
100 OF CTRL1, CTRL2
R3 = OPTION C
0
0
25
50
75
TA AMBIENT TEMPERATURE (°C)
100
Figure 7. CTRL1 and CTRL2 voltages vs
temperature. The voltage at CTRL1 sets the
maximum LED current until the voltage at
CTRL2 falls below that of CTRL1. At that point
(here at 25°C) CTRL2 takes over and derates
the maximum current to rising temperature.
in Figures 10, 11 and 12 show a PWM
dimming ratio that can actually exceed
3000:1 if PWM on time is reduced to
only 3 switching cycles (TPWM(ON) <
3.3µs for fPWM = 100Hz).
The simplified waveforms in Figure 10 and guidelines listed below
explain the relationship between PWM
duty cycle, PWM frequency, PWM dimming ratio and LED current. Strategies
for achieving maximum possible PWM
dimming using the PWM pin fall out
of the relation:
PWM DIMMING RATIO
1
=
MINIMUM PWM DUTY CYCLE
1
=
TPWM(OON)MIN • f PWM
qFor a PWM frequency (fPWM) of
100Hz, a PDR of 3000 implies a
PWM on time of 3.3µs.
qThe lower the PWM frequency,
the greater the PWM dimming
ratio (for a fixed PWM on time).
However, there are limits to how
VOUT
SHDN
VREF
Manufacturer
RX
Figure 6. Programming LED current derating
curve vs temperature (RNTC located on LED’s
circuit board)
VIN
1100
Table 1. NTC resistor
manufacturers/distributors
CTRL1, CTRL2 PIN VOLTAGES (mV)
10
R2
TPWM
TPWM(ON)
(LT3478)
LT3478/
LT3478-1
RSENSE
(= 1/fPWM)
PWM
CTRL1
OVPSET
RT
LED
VC
PWM
PWM DIMMING
CONTROL
Figure 8. PWM dimming control
Linear Technology Magazine • June 2007
INDUCTOR
CURRENT
LED
CURRENT
MAX ILED
Figure 9. PWM dimming waveforms
15
L DESIGN FEATURES
VIN
VS
L
1000
SW
SHDN
100
OUT
VREF
CTRL2
100k
LT3478-1
LED
VIN = 12V
6 LEDS AT 700mA
PWM FREQ=100Hz
fOSC = 1.67MHz
4.7µF
LED CURRENT (mA)
3.3µF
PDS560
2.2µH
12V
700mA
CTRL1
10
1
OVPSET
TA=25°C
CTRL1=0.7V
CTRL2=VREF
130k
PWM
SS
1µF
VC
RT
0.1µF
0
11k
1
10
100
1000
PWM DIMMING RATIO
10000
Figure 11. LED current versus PWM dimming
ratio for the circuit in Figure 10
3.3V
0V
100Hz
PWM
DIMMING RATIO = 3000:1
Q1
voltage limits the maximum output
voltage, given by:
Maximum output voltage = OVPSET • 41
Figure 10. Boost LED driver optimized for high PWM
dimming ratio (3000:1): 15W, 6 LEDs at 700mA
low the PWM frequency can be
operated since the human eye
can see flicker below about 80Hz.
qHigher programmed switching
frequency (fOSC) improves PDR
but reduces efficiency and
increases internal heating. In
general, TPWM(ON)MIN = 3 • 1/fOSC
(approximately 3 switch cycles).
qLeakage currents from the output
capacitor should be minimized.
The LT3478 and LT3478-1 both
turn off any circuitry running
from VOUT when the PWM pin is
low.
qFor an even wider dimming range,
the PWM and analog dimming
features can be combined, where
TDR = PDR • ADR
where
TDR = Total Dimming Ratio
PDR = PWM Dimming Ratio
ADR = Analog Dimming Ratio
A PDR of 3000:1 and an ADR
of 10:1 (CTRL = 0.1V) yields a
TDR of 30,000:1.
Open LED Protection
The output voltage has a programmable maximum to avoid damaging
the LEDs due to a disconnect (open
LED) followed by a reconnect. During
LED disconnect, the converter can go
open loop and drive the output voltage so high that the internal power
switch is damaged. Most LED drivers
have a fixed maximum output voltage
to save the switch, but this may be
too high for the reconnected string
of LEDs. The LT3478 and LT3478-1
provide a programmable overvoltage
protection (OVP) level to limit output
voltage based on the number of series
connected LEDs. The OVPSET pin
OVPSET voltage can be derived
from VREF by it’s own resistor divider
or by adding one resistor to the divider
used to define CTRL1 voltage. OVPSET
program level should not exceed 1V
to ensure the switch voltage does not
exceed 42V.
Robust Operation: Fault
Detection and Soft-Start
For robust performance during hotplugging, startup, or during normal
operation, the LT3478 and LT3478-1
monitor system parameters for any
of the following faults: VIN < 2.8V,
SHDN < 1.4V, inductor inrush current
greater than 6A, and/or output voltage
greater than programmed OVP. On
detection of any of these faults, the
LT3478 and LT3478-1 stop switching
immediately and the soft-start pin
is discharged (Figure 13). When all
faults are removed and the SS pin has
SW
SS
PWM
5V/DIV
FAULTS TRIGGERING
SOFT-START LATCH
WITH SW TURNED OFF
IMMEDIATELY:
IL
1A/DIV
VIN < 2.8V OR
SHDN < 1.4V OR
VOUT > OVP OR
I(INDUCTOR) > 6A
ILED
1A/DIV
1µs/DIV
Figure 12. PWM dimming waveforms
for the circuit in Figure 10
16
0.65V (ACTIVE THRESHOLD)
0.25V (RESET THRESHOLD)
0.15V
SOFT-START LATCH RESET:
SOFT-START
LATCH SET:
SS < 0.25V AND
VIN > 2.8V AND
SHDN > 1.4V AND
VOUT < OVP AND
I(INDUCTOR) < 6A
Figure 13. LT3478/ LT3478-1 fault detection and SS pin timing diagram
Linear Technology Magazine • June 2007
DESIGN FEATURES L
VIN
3.8V TO 6.5V
NiMH 4×
C1
10µF
10V
L1
6.8µH
VIN
ON OFF
VS
D1
L
SHDN
80
ILED = 1A
fOSC = 500kHz
75 PWM DUTY CYCLE = 100%
C2
4.7µF
16V
SW
OUT
CTRL2
R1
100k
LT3478-1
Q2
1A
LED
CTRL1
R4
510Ω
OVPSET
L1: CDRH105R-6R8
D1: B320
Q1: Si2302ADS
Q2: Si2315BDS
LED: LUXEON III (WHITE)
R2
34k
PWM
SS
CSS
1µF
3.3V
0V
VC
70
EFFICIENCY (%)
VREF
65
60
RT
CC
0.1µF
55
RT
69.8k
R5
510Ω
50
SINGLE LED
LUXEON III (WHITE)
3
4
5
VIN (V)
fOSC = 500kHz
1kHz
6
7
Q1
PWM
DIMMING RATIO = 200:1
R3
10k
Figure 14. Portable camera flash: 4W single LED at 1A buck-boost mode LED driver
been discharged to at least 0.25V, an
internal 12µA supply charges the SS
pin with a rate programmed using an
external capacitor CSS. A gradual ramp
up of SS pin voltage is equivalent to a
ramp up of switch current limit until
SS exceeds the VC pin voltage.
Conclusion
The LT3478 and LT3478-1 are ideal for
boost, buck or buck-boost mode LED
applications requiring high LED current operation and high PWM dimming
ratios. The high 4.5A peak switch curPVIN
32V
High Efficiency: Separate
Inductor and IC Supplies,
Programmable fOSC,
60mΩ Switch
The LT3478 and LT3478-1 can use
separate supplies for the IC and the
inductor to optimize efficiency and
switch duty cycle range. Detection of
inductor inrush current uses VS and
L pins independent of the VIN supply
of the IC (Figure 2). This allows VIN to
be supplied from the lowest available
supply (at least 2.8V) in the system to
minimize efficiency lost in the power
switch driver. The inductor can then
be powered from a supply (between
2.8V and 36V) better suited to the
duty cycle and power requirements of
the LED load. The switching frequency
of the power switch can be tailored to
achieve the optimum inductor size and
efficiency performance required for
the system. The 60mΩ switch further
improves efficiency by keeping switch
losses to a minimum for high duty
cycle operation.
rent limit combined with a new patent
pending PWM dimming architecture
allow the LT3478 and LT3478-1 to
provide high PWM dimming ratios for
LED currents up to 4A. L
C1
3.3µF
50V
RSENSE
0.068Ω
1.5A
4 LEDs
R4
365Ω
TYPICAL EFFICIENCY = 90%
FOR CONDITIONS/COMPONENTS SHOWN
(PWM DUTY CYCLE = 100%, TA =25°C)
C3
10µF
25V
Q2
L1
10µH
VIN
3.3V
C2
4.7µF
10V
D1
VIN
VS
L
OUT LED SW
SHDN
L1: CDRH105R-100
D1: PDS560
Q1: 2N7002
Q2: Si2319DS
LEDs: LXK2 (WHITE)
Q1
PWM
R3
10k
VREF
R1
24k
R5
510Ω
LT3478
CTRL2
PWM
CTRL1
DIMMING RATIO = 3000:1
OVPSET
3.3V
R2
100k
SS
CSS
1µF
VC
RT
CC
0.1µF
0V
100Hz
RT
69.8k
fOSC = 500kHz
Figure 15. High powered LED lighting: 24W, 4 LEDs at 1.5A buck-boost mode LED driver
Linear Technology Magazine • June 2007
17
L DESIGN FEATURES
SAR ADCs Feature Speed,
Low Power, Small Package Size
and True Simultaneous Sampling
by Steve Logan and Atsushi Kawamoto
Introduction
When it comes to quickly digitizing
analog signals from a few hertz to a few
megahertz, successive approximation
register (SAR) ADCs are the best choice
for a broad range of applications. Their
fast response and low latency make
SAR ADCs ideal for single channel or
multichannel data acquisition.
Low power SAR ADCs are crucial as
more designs migrate to lower supply
voltages and tighter power budgets.
Solution size is also a key requirement
for designers needing a single snapshot
of the input, as many low power SAR
ADCs are used in portable or multichannel systems in which PCB space
is limited. With designers trying to do
more with less space, a small package
becomes vital.
As package size shrinks, it makes
sense to replace a parallel interface
with a serial interface to reduce the
number of data lines, which in turn
reduces the size of both the SAR
ADC and the microprocessor. Serial
interfaces also reduce the headaches
associated with routing many parallel data lines across a board. Linear
Technology offers multiple families of
fast SAR ADCs that combine speed, low
When it comes to quickly
digitizing analog signals
from a few hertz to a few
megahertz, successive
approximation register
(SAR) ADCs are the best
choice for a broad range
of applications. Their fast
response and low latency
make SAR ADCs ideal
for single channel
or multichannel
data acquisition.
power, small package size and simple
serial interfaces.
6-Channel Simultaneous
Sampling ADCs
Motor control is one of many applications that benefit from simultaneous
sampling SAR ADCs. In motor control
circuits, the phase relationship of measured channels must be preserved,
thus requiring simultaneous sampling
ADCs with multiple sample-and-hold
amplifiers (S/HA’s). Data can be stored
internally to be read out sequentially,
with the phase relationship from the
inputs intact. Without simultaneous
sampling, control algorithms could
incorrectly adjust the motor’s torque
or speed control, leading to vibrations
and additional wear on the motor. Linear Technology has a growing family
of low power simultaneous sampling
ADCs that target motor control, servos, and general purpose AC power
monitoring.
Linear Technology offers four low
power, 6-channel simultaneous sampling ADCs, optimized for two fast
sample rates (250ksps per channel
and 100ksps) as well as two different
resolutions (14 bits and 12 bits). All
are pin- and software-compatible,
making it easy to optimize designs for
resolution, speed and cost. By using
a 5mm × 5mm 32-pin QFN package,
these ADCs achieve a solution size as
much as six times smaller than comparable performance ADCs. A single
3V supply powers both the analog and
digital circuitry, thus reducing power
dissipation eliminating the need for
higher voltage supplies.
Table 1. Simultaneous sampling ADCs from Linear Technology
Part Number
Resolution
Number of
Channels
Sample Rate
per channel
Power
Package
Input Voltage Range
LTC2351-14
14-Bit
6
250ksps
16.5mW
QFN-32 (5mm × 5mm)
±1.25V, 0V to 2.5V
LTC1408
14-Bit
6
100ksps
15mW
QFN-32 (5mm × 5mm)
±1.25V, 0V to 2.5V
LTC1407A
14-Bit
2
1.5Msps
14mW
MSOP-10
0V to 2.5V
LTC1407A-1
14-Bit
2
1.5Msps
14mW
MSOP-10
±1.25V
LTC2351-12
12-Bit
6
250ksps
16.5mW
QFN-32 (5mm × 5mm)
±1.25V, 0V to 2.5V
LTC1408-12
12-Bit
6
100ksps
15mW
QFN-32 (5mm × 5mm)
±1.25V, 0V to 2.5V
LTC1407
12-Bit
2
1.5Msps
14mW
MSOP-10
0V to 2.5V
LTC1407-1
12-Bit
2
1.5Msps
14mW
MSOP-10
±1.25V
18
Linear Technology Magazine • June 2007
DESIGN FEATURES L
Low Power ADCs Optimized
for 250ksps–750ksps
The 14-bit LTC2351-14 is a 1.5Msps,
low power SAR ADC with six simultaneously sampled differential input
channels. It operates from a single 3V
supply and features six independent
sample-and-hold amplifiers and a
single ADC. The single ADC with multiple S/HA’s enables excellent range
match (1mV) between channels and
channel-to-channel skew (200ps). The
six channels can monitor two separate
motors, providing vital information
about motor torque, speed, shaft position, and direction.
The versatile LTC2351-14 also
suits other industrial monitoring applications such as 3-phase voltage
monitoring to ensure line voltage
compliance, 3-phase power monitoring
of current and voltage, power factor
correction, and data acquisition. These
applications may require portability,
and it is here that the LTC2351-14’s
low power and small size are most
desirable. Power consumption is a
mere 16.5mW, which extends battery life. The 3-wire serial interface
means fewer pins than traditional
parallel output devices, allowing the
LTC2351-14 to fit in a 32-pin, 5mm
× 5mm QFN package.
When the LTC2351-14 is not converting, the ADC offers two power
saving modes. Power dissipation can
be reduced to 4.5mW in nap mode with
the internal 2.5V reference remaining
active. Sleep mode further reduces
0.1µF
10µF
4
5
CH0+
CH0–
3V
24
VCC
+
25
VDD
S&H
–
6
7
8
CH1+
CH1–
+
S&H
–
9
10
11
CH2+
CH2–
+
S&H
–
MUX
12 13
14
15
CH3+
CH3–
1.5Msps
14-BIT ADC
+
S&H
14-BIT LATCH 0
14-BIT LATCH 1
14-BIT LATCH 2
14-BIT LATCH 3
14-BIT LATCH 4
14-BIT LATCH 5
OVDD
3V
THREESTATE
SERIAL
OUTPUT
PORT
SD0
OGND
3
1
0.1µF
2
–
16
17
18
CH4+
CH4–
CONV
TIMING
LOGIC
+
SCK
S&H
30
32
–
19
20
21
CH5+
CH5–
+
S&H
–
2.5V
REFERENCE
EXPOSED PAD
33
GND
22
VREF
23
BIP
29
SEL2 SEL1 SEL0
26
27
28
DGND
31
10µF
Figureﾊ1. The LTC2351-14 includes six sample-and-hold amplifiers.
Linear Technology Magazine • June 2007
19
L DESIGN FEATURES
power consumption to 12µW, with
all internal circuitry powered down,
further extending battery life. Upon
waking up from sleep mode, the internal reference settles within 2ms, and
conversions resume thereafter within
a single clock cycle.
Three input-select lines configure
the number of differential inputs
converted. Thus, higher speeds are
possible as the number of channels
converted decreases, from six differential inputs at 250ksps, two differential
inputs at 750ksps, to one differential
input at 1.5Msps. A bipolar/unipolar
input line selects either a ±1.25V bipolar or a 0V to 2.5V unipolar input
range. A 100kHz input signal yields
a SINAD of 75dB and –90dB THD.
The LTC2351-14’s true differential
inputs and 83dB common mode rejection make it ideal for minimizing
common mode noise prevalent in
harsh industrial environments. For
lower resolution applications and
performance-cost optimization, Linear
Technology offers the pin- and software-compatible 12-bit LTC2351-12
ADC. The LTC2351-12 also simultaneously samples up to six differential
channels, draws only 16.5mW of power
and features 72dB SINAD.
Some simultaneous sampling ADCs
are capable of measuring six channels,
but use only two S/HA’s, two ADCs,
and two 3-to-1 multiplexers. In these
competing ADCs, only two channels
are simultaneously sampled. Multiple
ADCs can mean mismatches from one
ADC to the other within the package.
INL could be within the maximum ratings, but bow in one polarity on one
ADC and the opposite polarity on the
second ADC. By integrating six S/HA’s
and a single ADC, the LTC2351-14
does not suffer the anomalies associated with multiple ADCs and is
ideal for applications that require
simultaneously sampling more than
two channels.
Lower Sampling Rate ADCs
with Improved AC Performance
Linear Technology also offers a second pair of 6-channel simultaneous
sampling ADCs optimized for slower
sampling rates. The 14-bit LTC1408
and 12-bit LTC1408-12 are optimized
for output rates up to 100ksps/channel for all six channels, 300ksps for
two channels, and 600ksps for one
channel. The LTC1408 features improved AC performance (79dB SINAD
at 300kHz, with an external reference). Like the LTC2351 family, both
LTC1408 ADCs are low power (15mW),
offered in a small 5mm × 5mm 32-pin
QFN package, and include six sampleand-hold amplifiers. See Table 1 for a
complete listing of these simultaneous
sampling ADCs.
The LTC1408 and LTC2351-14
6-channel SAR ADCs are ideal for
monitoring 3-phase voltages and
currents, as shown in Figure 2. Attenuation networks externally reduce
the voltage to within the selected
bipolar/unipolar input ranges. While
ATTENUATION
NETWORK
ATTENUATION
NETWORK
ATTENUATION
NETWORK
LTC1408
SIGNAL
CONDITIONING
AND FILTERING
SIGNAL
CONDITIONING
AND FILTERING
SIGNAL
CONDITIONING
AND FILTERING
Figureﾊ2. The LTC1408 is ideal for 3-phase power monitoring.
20
three analog inputs measure the
voltage, the other three channels use
signal conditioning and filtering to
convert the currents. The six S/HA’s
keep the phase relationship between
the voltages and currents intact and
data can be read out through the serial interface. These ADCs also include
a digital output supply voltage that
can be set between the analog supply voltage down to 1.8V, making it
possible to interface with 1.8V, 2.5V
or 3V digital logic.
2-Channel
Simultaneous Sampling ADCs
For applications such as encoders and
communications requiring simultaneous sampling on only two channels at
rates greater than 1Msps per channel,
fast SAR ADCs again work very well.
Linear Technology offers a pin- and
software-compatible family of 14-bit
and 12-bit, 2-channel, simultaneous
sampling SAR ADCs.
Like the 6-channel simultaneous
sampling ADCs, the 14-bit, 2-channel
LTC1407A-1 is also optimized for low
power and small package size, further
extending battery life and reducing
total solution area. The LTC1407A-1
is available in a 10-pin MSOP package
and dissipates only 14mW. This small
ADC measures two ±1.25V bipolar
channels simultaneously at 1.5Msps
per channel or a single channel at
3Msps. No competing ADCs of similar
size can meet the speed and input
frequency range of the LTC1407A-1.
The pin- and software-compatible
LTC1407A is a 0V to 2.5V unipolar
14-bit ADC. Both the unipolar and
bipolar LTC1407 ADCs perform
well when measuring differential AC
inputs, making it a good choice for
communications applications. The
LTC1407A-1 and LTC1407A achieve
76.3dB SINAD and –86dB THD with a
750kHz input frequency and an external 3.3V reference. SFDR is 86dB and
intermodulation distortion is –82dB at
the same input frequency.
For applications requiring less resolution, the 12-bit LTC1407-1 (bipolar)
and 12-bit LTC1407 (unipolar) ADCs
are available. All four LTC1407 ADCs
include a 2.5V internal reference, nap
Linear Technology Magazine • June 2007
DESIGN FEATURES L
Table 2. Fast single-channel SAR ADCs from Linear Technology
Part Number
Resolution
Sample Rate
Package
Power
Input Voltage Range
I/O
LTC2355-14
14-Bit
3.5Msps
MSOP-10
18mW
0V to 2.5V
Serial
LTC2356-14
14-Bit
3.5Msps
MSOP-10
18mW
±1.25V
Serial
LTC1403A
14-Bit
2.8Msps
MSOP-10
14mW
0V to 2.5V
Serial
LTC1403A-1
14-Bit
2.8Msps
MSOP-10
14mW
±1.25V
Serial
LTC2355-12
12-Bit
3.5Msps
MSOP-10
18mW
0V to 2.5V
Serial
LTC2356-12
12-Bit
3.5Msps
MSOP-10
18mW
±1.25V
Serial
LTC1403
12-Bit
2.8Msps
MSOP-10
14mW
0V to 2.5V
Serial
LTC1403-1
12-Bit
2.8Msps
MSOP-10
14mW
±1.25V
Serial
Data Acquisition Systems
SAR ADCs also excel in data acquisition applications due to the ability to
multiplex multiple channels with little
or no data latency. Data acquisition
requires the ability to monitor a wide
array of analog signals in industrial
settings, often including temperature,
pressure, voltage, or load currents. For
example, an industrial control design
may use thermocouples to monitor
temperature variations, pressure sensors to measure physical changes, or
chemical sensors to detect various
environmental settings. Data acquisition could mean monitoring a single
channel or hundreds of channels.
Figure 3 shows an example of the
analog signal chain for a multichannel
data acquisition system. After being
routed through a series of multiplexers and signal conditioning circuits,
these signals can be digitized by a fast
single-channel SAR ADC, such as the
LTC2355-14. With a fast SAR ADC,
multiplexers and amplifiers with high
gain bandwidths are used to switch
through the various data inputs. The
LTC1391 is an 8-to-1 multiplexer used
to switch the various analog signals
on the front end of the system. The
LT6241 is a precision amplifier that
has low noise (550nVP–P), 1pA bias
current, 17MHz unity gain bandwidth,
and provides a low impedance connection to the ADC.
Linear Technology Magazine • June 2007
High Speed
Single-Channel SAR ADCs
sures a single differential input and
communicates via an SPI-compatible serial interface. This SAR ADC
operates from a single 3.3V supply,
draws only 18mW at the maximum
conversion rate, and is available
in a tiny 10-pin MSOP package.
The combination of high speed, low
Along with its growing family of simultaneous sampling ADCs, Linear
Technology is also adding to its family
of pin- and software-compatible high
speed single-channel SAR ADCs. The
14-bit, 3.5Msps LTC2356-14 mea-
continued on page 38
8
LTC1391
3V
MULTIPLE
INPUTS
MEASURE
TEMPERATURE,
PRESSURE, VOLTAGE,
LOAD CURRENTS
LTC1391
8
LT6241
3-WIRE
SERIAL
INTERFACE
LTC2355-14
LTC1391
MULTIPLEXING
INPUTS
SIGNAL
CONDITIONING
ADC DIGITIZES
ALL ANALOG INPUTS
Figure 3. Industrial control data acquisition systems
measure numerous signals with a single ADC.
10µF 3.3V
7
LTC2356-14
AIN+
1
+
14-BIT ADC
S&H
AIN–
2
VDD
–
THREESTATE
SERIAL
OUTPUT
PORT
14-BIT LATCH
(3.3mW) and sleep (6µW) power-down
modes. Both families of 6-channel and
2-channel simultaneous sampling
ADCs are detailed in Table 1.
8
SDO
10
CONV
9
SCK
14
3
VREF
2.5V
REFERENCE
10µF
4
GND
5
6
TIMING
LOGIC
11
EXPOSED PAD
Figure 4. The LTC2356-14 single channel ADC is ideal for fast, low power applications.
21
L DESIGN FEATURES
A Cool Circuit: 48V Ideal Diode-OR
by Dan Eddleman
Reduces Heat Dissipation
Introduction
High availability systems commonly
demand redundant power supplies or
backup battery feeds to enhance reliability. Traditionally, Schottky diodes
were used to diode-OR these supplies
at the point of load. However, as load
currents climb, the forward voltage
drop of the ORing diodes becomes a
significant source of power loss. Designers are thus tasked with creating
elaborate thermal layouts and heat
sinks to contend with the diodes’ rising temperatures.
A better solution for a high current,
high availability system is to replace
the Schottky diodes with MOSFETbased ideal diodes. This lowers the
forward voltage drop of the diode-OR,
shrinking thermal layouts and improving system power efficiency. The 4mm
× 3mm LTC4355 simplifies the design
of MOSFET ORing circuits by controlling two N-channel MOSFETs, which
can combine supplies with voltages
between 9V and 80V. The LTC4355
also provides the input voltage monitors, input fuse monitors, and forward
voltage drop monitors frequently required in these systems.
Operation
The LTC4355’s basic operation is
straightforward. It uses a linear amplifier and an internal charge pump
to maintain a 25mV forward voltage
drop across the external N-channel
MOSFETs. The MOSFET sources are
connected to the input supplies and
the drains are joined at the output
(Figure 1). When power is first applied,
load current flows from the input supply with the higher voltage through
the body diode of the MOSFET. The
LTC4355 senses the voltage drop and
enhances the MOSFET. For small
load currents, the voltage across the
MOSFET is limited to 25mV. Larger
load currents cause the LTC4355 to
fully enhance the MOSFET, resulting
in a voltage drop of RDS(ON) • ILOAD. The
22
F1
15A
VIN1 = 12V
M1
HAT2165H
F2
15A
VIN2 = 12V
R2
86.6k
R4
86.6k
M2
HAT2165H
R5
10k
IN1
GATE1 IN2
MON1
SET
MON2
R1
12.7k
TO
LOAD
LTC4355
GND
R3
12.7k
R7
10k
GATE2 OUT
R6
10k
VDSFLT
FUSEFLT1
FUSEFLT2
PWRFLT1
PWRFLT2
GREEN LEDs D1
PANASONIC LN1351C
R8
10k
D3
D2
GND
R9
10k
D5
D4
Figure 1. 12V/15A ideal diode-OR application
linear amplifier provides a smooth
switchover between supplies without
the oscillations, chatter, and reverse
current common to comparator-based
designs. If the higher input supply
abruptly drops more than 25mV below the output voltage, as may occur
during an input short circuit, the
LTC4355 pulls the MOSFET gate low
within about 0.5µs to limit the amount
of reverse current that flows from the
output back to the input.
Fault Monitors
In addition to controlling the MOSFETs,
the LTC4355 also performs several
system health monitoring functions
required in high availability systems. It
detects when a fuse is blown, an input
supply is low, or the forward voltage
across a MOSFET is excessively large.
If a fuse blows open, the FUSEFLT1
or FUSEFLT2 pin pulls low to signal
which fuse has opened. Similarly,
when an input supply is below its minimum voltage, configured by a resistive
divider, the PWRFLT1 or PWRFLT2 pin
pulls low to indicate which supply is
out of regulation. The PWRFLT1 and
PWRFLT2 pins also indicate when the
forward voltage across a MOSFET exceeds a voltage programmed with the
SET pin. Excessive forward voltage is
a sign that a MOSFET may have failed
or is conducting too much current.
The LTC4355 in the DFN-14 package
provides a VDSFLT pin, which also
pulls low under this condition to allow
the system to differentiate between a
supply that is out of regulation and
a MOSFET with too much forward
voltage.
12V/15A Ideal Diode-OR
Figure 1 shows a simple 12V/15A ideal
diode-OR application. An MBR1635
Schottky diode would dissipate 8W in
this circuit. In contrast, the HAT2165
3.4mΩ MOSFET drops 15A • 3.4mΩ
= 51mV and dissipates only 51mV •
15A = 0.765W. The result is a drastic
reduction in PCB area and heat sinking
required to dissipate the power, not
to mention a 4-point improvement in
efficiency.
In this circuit, green LEDs indicate
normal operation, and fault conditions
cause the LEDs to turn off. Resistive
dividers connected between the input
supplies and the MON1 and MON2
pins configure the supply monitor
thresholds near 10V. When a supply
is below its minimum voltage, the
respective PWRFLT1 or PWRFLT2 pin
pulls low, thus turning off the D4 or
D5 LED.
Likewise, the D2 or D3 green LED
turns off to signal when a fuse has
blown open. Under this condition, the
IN1 or IN2 pin is pulled to ground by
an internal 0.5mA pulldown current.
As soon as the LTC4355 senses that
Linear Technology Magazine • June 2007
DESIGN FEATURES L
one of these pins is below 3.5V, it pulls
the FUSEFLT1 or FUSEFLT2 pin low.
Note that this condition also occurs
when an input supply falls below
3.5V. Therefore, it may be necessary
to confirm that PWRFLT1 or PWRFLT2
is high impedance, signaling a valid
input supply voltage, before concluding that a fuse is blown open.
In Figure 1, the LTC4355 detects
that a MOSFET has failed or is conducting excessive current by sensing
the forward voltage drop across the
MOSFET. The faults detected include
a MOSFET that is open on the higher
supply, excessive MOSFET current
due to overcurrent on the load, or a
shorted MOSFET on the lower supply.
When one of these conditions occurs, the LTC4355 pulls the VDSFLT
pin (DFN-14 package only) and the
PWRFLT1 or PWRFLT2 pin low to indicate which supply has the fault. The
forward voltage threshold is configured
at 1.5V by leaving the SET pin open.
Tying the SET pin directly to ground
7A
48V/5.5A High Side and
Low Side Ideal Diode-ORs
Many high availability systems require
diodes on both the high and low side of
the redundant power feeds. Combining the LTC4355 with the LTC4354
provides a complete solution for
these applications. In the 48V/5.5A
circuit of Figure 2, the LTC4355 and
two FDS3672 MOSFETs perform the
high side ORing function while the
LTC4354 and two FDS3672s perform
low side ORing.
7A
48VB
FDS3672
33k
340k
IN1
IN2
GATE1
GATE2
LTC4355
MON2
SET
OUT
FUSEFLT1
FUSEFLT2
PWRFLT1
PWRFLT2
MON1
12.7k
At 5.5A, an MBR10100 Schottky
Diode in a TO-220 package dissipates
over 3W. The current passes through
both a high side and a low side diode,
resulting in a total power dissipation
of over 6W. In contrast, an FDS3672
in a smaller SO-8 package dissipates
0.6W for a total of 1.2W. The ideal diode
solution lowers the total power dissipation by 80%, reducing the necessary
PCB area and heat sinking.
In the circuit in Figure 2, the
LTC4355 and LTC4354 receive power
when either input supply is present.
The LTC4354’s positive supply pin,
VCC, is regulated from the output of
the LTC4355, always within a diode
drop of the higher input voltage
(+48VA or +48VB). At the low side, the
LTC4355’s negative supply pin, GND,
connects to the output of the LTC4354,
always within a diode drop of the more
negative voltage (RTNA or RTNB).
Consequently, both parts remain powered even when one of the supplies is
disconnected or is out of regulation.
FDS3672
48VA
340k
or through a 10kΩ resistor to ground
configures this threshold at 0.25V or
0.5V, respectively. Note that during
startup or when a switchover between
supplies occurs, the VDSFLT pin and
the PWRFLT1 or PWRFLT2 pin may
momentarily indicate that the forward
voltage has exceeded the programmed
threshold during the short interval
when MOSFET gate ramps up and
the body diode conducts.
GND
12.7k
MOC207
LOAD
12k
33k
VCC
LTC4354
DA
DB
GA
FAULT
GB
VSS
1µF
2k
2k
10A
RTNA
RTNB
10A
FDS3672
FDS3672
Figure 2. 48V/5.5A positive supply and negative supply diode-ORing with combined fault outputs.
Linear Technology Magazine • June 2007
23
L DESIGN FEATURES
10A
VRTN_A
10A
VRTN_B
340k
FDS3672
FDS3672
–48V/5.5A
High side and Low Side
Diode-ORs for Telecom
340k
IN1
IN2
GATE1
GATE2
MON1
SET
GND
12.7k
OUT
VDSFLT
FUSEFLT1
FUSEFLT2
PWRFLT1
PWRFLT2
LTC4355
MON2
12.7k
pin spacing sometimes desirable in
higher voltage applications.
LOAD
12k
VCC
LTC4354
DA
–48V_A
–48V_B
7A
7A
DB
2k
GA
FAULT
GB
2k
VSS
1µF
FDS3672
FDS3672
Figure 3. –48V/5.5A positive and negative supply diode-ORing for telecom systems.
Large supply variations and transients
are easily accommodated by the wide
operating voltage ranges of these two
parts, 4.5V to 80V for the LTC4354 and
9V to 80V (100V absolute maximum)
for the LTC4355.
This circuit combines all fault indicators to drive one optoisolator. If an
input supply falls to less than 36V or
the forward voltage drop across one
of the positive-side MOSFETs exceeds
0.25V, the LTC4355’s PWRFLT1 or
PWRFLT2 pin pulls low to signal the
fault. If a positive-side fuse blows
open, the LTC4355 indicates a fault by
pulling the FUSEFLT1 or FUSEFLT2
pin low. Finally, if the forward voltage
across a low side MOSFET exceeds
0.26V, the LTC4354’s FAULT pin
drives an NPN that turns off the same
optoisolator driven by the LTC4355’s
pins.
Because the high side fuses have
lower current ratings than the return
fuses, the high side fuses blow first
under most fault conditions. With the
return fuses intact, system potentials
tend to settle near ground after a fuse
blows open.
The VDSFLT pin is not shown in
this schematic. Since the PWRFLT1
or PWRFLT2 pin pulls low when the
VDSFLT pin pulls low, VDSFLT is
redundant in this application. Furthermore, this schematic is capable of
accommodating not just the smaller
DFN-14 package, but also the larger
SO-16 package. While the SO-16 lacks
a VDSFLT pin, it features the wider
Many –48V telecom systems, including those that conform to the new
AdvancedTCA specification, require
ORing circuits on both the high and
low side of the redundant power feeds.
A few simple modifications convert
the +48V solution in Figure 2 to the
–48V solution in Figure 3. The +48V
supply input becomes the return
feed, VRTN, and the returns in the
+48V system now serve as the –48V
input feeds. The 10A and 7A fuses
have been swapped, placing the 10A
fuse in the high side return path. As
a result, most fault conditions cause
the high side 7A fuse to blow before
the low side 10A fuse. Consequently,
system potentials generally settle near
VRTN after a fuse blows. The minimal
circuit in Figure 3 does not connect
the fault pins. If desired, faults can
be monitored with a circuit similar to
that in Figure 2.
Conclusion
The LTC4355 frees up PCB area by
reducing power dissipation and the
size of associated heat sinks in applications that require supply ORing. Its
wide 9V to 80V supply operating range
and 100V absolute maximum rating
accommodate a broad range of input
supply voltages with ample margin for
supply variations and transients. In
addition, the ability to provide system
health monitoring functions makes it
especially well suited to high-availability applications. Those systems
that require both high side and low
side ORing can combine the LTC4355
with the LTC4354 to form a complete
solution. L
LT3009, continued from page Conclusion
The LT3009 offers ultralow quiescent
current, a shutdown mode, and wide
input and output voltage ranges in tiny
2mm × 2mm DFN and SC70 packages
without sacrificing performance or
24
reliability. A stable output is available
with a wide range of output capacitors,
including small ceramics. Internal
protection circuitry in the LT3009
eliminates the need for external protections diodes, further saving space
and lowering cost. Competing devices
can’t come close to the performance
and advantages that the LT3009 offers in the world of ultralow quiescent
current regulators. L
Linear Technology Magazine • June 2007
DESIGN FEATURES L
Highly Integrated USB Power Manager
with Li-Ion Charger and Three
Step-Down Switching Regulators
in 4mm × 4mm QFN
by Amit Lele
Introduction
Mobile technology has radically
changed the way we acquire, share
and disseminate information. Modern,
feature-rich handheld and portable
devices require several power management circuits, including a battery
charger, multiple step-down switching
regulators and low power LDOs for
watchdog circuitry. If each of these
functions is served by a separate power
supply IC, each IC (and its external
components) occupies valuable board
space, consumes battery-draining
quiescent current and significantly
increases the overall development
and material costs of the device. The
LTC3557 solves this problem by bringing all power management functions
into a single device. It combines a full
featured USB power manager, a Li-ion
battery charger, three high frequency
step-down switching regulators and a
3.3V always-on LDO in a single 4mm
× 4mm QFN package.
Features
The LTC3557 is a highly integrated
power management and battery charger IC for single cell Li-Ion/Polymer
battery applications. Table 1 high-
lights some of the key features of the
LTC3557.
The LTC3557 can derive power from
a current limited input such as USB.
The programmable current limit is set
by a single external resistor (RCLPROG)
on the CLRPOG pin and the logic state
of ILIM0 and ILIM1 pins. Table 2 shows
the different operating modes of the
input current limit.
The 1A (10x) mode is reserved for
use with a higher current input power
supply such as an AC wall adapter.
Alternatively, power can be directly
provided to the system load (VOUT)
via an external PFET connected in
Table 1. Features of the LTC3557
Feature
Benefits
PowerPath Control
Allows seamless transition between input power sources (Li-Ion battery, USB, wall
adapter or high voltage buck regulator) to supply system load.
WALL Input
Provides power from 5V wall adapter directly to system load through an external
low impedance PFET
USB Input
Precision input current limit which communicates with the battery charger to
ensure that input current never violates the USB specification
High Voltage Buck Control with Bat-Track™
Controls external HV buck to expand input voltage range up to 38V. The Bat-Track
feature allows efficient charging of the battery to minimize heat dissipation in the
application
Li-Ion Charger
Uses constant current/constant voltage architecture with thermal regulation for
optimal charging. Preset float voltage accurate to 0.85%.
Temperature qualified charging using NTC
Disables charging of the battery under extreme temperature conditions outside a
programmable range
Internal Safety Timer
Limits maximum charge cycle to 4 hours
CHRG Fault Reporting
Four modes of CHRG pin including ON, OFF, Slow Blink and Fast Blink to report
various operating states
Three High Efficiency Step-Down Switching
Regulators
High frequency switching (2.25MHz) stays out of the AM band and enables use
of tiny inductors. Internally compensated to save valuable board space. Userprogrammable output voltages with external resistor divider. Power on Reset output
for power sequencing.
Always on 3.3V LDO
Ultra low quiescent current 3.3V LDO for real time clock, standby power,
pushbutton control, etc.
Linear Technology Magazine • June 2007
25
L DESIGN FEATURES
series with an AC wall adapter. The
input supply range can be expanded
by using an appropriate high voltage
buck regulator as shown in Figure 1.
The LTC3557 takes over the control
of buck regulator via the VC pin and
sets the VOUT pin voltage at a fixed
offset above the battery voltage. This
Bat-Track feature charges the battery
at the highest efficiency. Absent all
other input power sources, the battery
provides power to the system (VOUT)
through an internal 200mΩ ideal
diode. An optional external <50mΩ
ideal diode can be used to minimize
the voltage drop from BAT to VOUT in
high current applications.
The LTC3557 charger circuitry uses
constant current/constant voltage
architecture to optimize the charging
of the battery. The battery charge
current is set by an external resistor
(RPROG) connected to the PROG pin
as follows:
ICHG ( A) =
A Typical Application
Figure 2 shows a typical application
using the LTC3557. In this configuration, the LTC3557 automatically
switches between the high voltage
buck power supply or the USB/5V
wall adapter. The USB input current
is programmed to nominal value of
476mA using a 2.1k resistor on the
26
SW
FB
C
VC
WALL
ACPR
SYSTEM
LOAD
VOUT
LTC3557/LTC3557-1
OPTIONAL
EXTERNAL
IDEAL DIODE
PMOS
GATE
BAT
+
Li-Ion
BATTERY
Figure 1. High voltage buck control using VC
HVIN
8V TO 38V
4
4.7µF
68nF
150k
5
10
1000 V
RPROG
The LTC3557 includes several safety mechanisms to handle situations
when the available input current is less
than the programmed charge current.
This allows the system designer to set
the charge current based on normal
operating conditions rather than reducing the charging current to account
for worst-case scenarios. These safety
mechanisms are explained in more
detail in the “Getting the Priorities
Right” section below.
The LTC3557 includes three stepdown switching regulators capable of
delivering up to 600mA. Additionally,
an always-on LDO with fixed 3.3V
output voltage can deliver up to 25mA
of load current. This can be used to
power watchdog circuitry or other low
power circuitry.
HIGH VOLTAGE
BUCK
REGULATOR
VIN
LT3480, LT3481
V OR LT3505
VIN
UP TO 38V
TRANSIENTS
TO 60V
40.2k
NC
7
VIN
BOOST
LT3480
RUN/SS
RT
SW
SYNC
BD
PG
FB
GND
0.47µF
3
6
DFLS240L
24
10µF
22µF
100k
8
Si2333DS
3
25
VC WALL ACPR
VBUS
VOUT
BVIN1
2.1k 27
20
100k 18
19
100k
NTC
CLPROG BVIN2
PROG
CHRG
GATE
BAT
VNTC
NTC
LDO3V3
VOUT
23
6
16
1
2
9
10
11
8
SW1
ILIM0
ILIM1
FB1
EN1
EN2
SW3
EN3
MODE
FB3
RST2
SW2
GND
FB2
10µF
2.2µF
510Ω
2.2µF
28
21
Si2333DS
(OPT)
22
4
LTC3557/
LTC3557-1
PMIC
CONTROL
499k
1
9
26
2k
6.8µH
VC
11
USB OR
5V WALL
ADAPTER
OPTIONAL HIGH VOLTAGE
BUCK INPUT
2
5
3.3V
25mA
1µF ALWAYS ON
BAT
Li-Ion
3.3µH
10pF
7
17
+
1.02M
324k
4.7µH
10pF
12
806k
649k
14
15
13
VOUT1
3.3V
10µF 600mA
VOUT3
1.8V
10µF 400mA
RST2
100k
4.7µH
10pF
232k
464k
VOUT2
1.2V
10µF 400mA
29
Figure 2. Typical application circuit for LTC3557
Linear Technology Magazine • June 2007
DESIGN FEATURES L
FROM AC ADAPTER (OR HIGH VOLTAGE BUCK OUTPUT)
26
3
WALL
4.3V
(RISING)
3.2V
(FALLING)
–
+
ACPR
+
–
+
–
FROM
USB
24
VC
OPTIONAL CONTROL
FOR HIGH VOLTAGE BUCK REGS
LT3480, LT3481 OR LT3505
25
75mV (RISING)
25mV (FALLING)
ENABLE
VBUS
VOUT
VOUT
23
SYSTEM
LOAD
USB CURRENT LIMIT
IDEAL
DIODE
CONSTANT CURRENT
CONSTANT VOLTAGE
BATTERY CHARGER
+
–
–
+
GATE
OPTIONAL
EXTERNAL
IDEAL DIODE
PMOS
21
15mV
BAT
BAT
22
+
35571 F01
Li-Ion
Figure 3. Simplified PowerPath block diagram
CLPROG pin. The charge current is
programmed to 500mA using a 2k
resistor on PROG pin. The resistor
network on the NTC pin sets the
battery charging temperature range
from 0°C to 40°C based on R-T curve
1 characteristics for the 100k NTC
thermistor. An LED on the CHRG pin
provides battery charging and status
information.
VOUT1 is set to 3.3V to drive higher
power applications such as I/O or disk
drives. VOUT3 is set to 1.8V to drive medium power applications while VOUT2 is
set to 1.2V to drive a microprocessor
core. The RST2 output can be used to
provide power supply sequencing using the PMIC control pins. The optional
external ideal diode can be used to
provide a lower impedance path from
BAT to VOUT for applications that draw
heavy loads from the battery.
Table 2. Controlled input current limit
ILIM1
ILIM0
IBUS(LIM)
0
0
100mA(1x)
0
1
1A(10x)
1
0
SUSPEND
1
1
500mA(5x)
Linear Technology Magazine • June 2007
Safety Timer and
Automatic Recharge
An internal safety timer shuts off all
charge current to the battery after 4
hours of charging. As long as the load
current at VOUT does not exceed the
current available from the external
power source, the battery remains fully
charged. If the load current at VOUT
exceeds the current available from
the external power source, the extra
current is pulled from the battery. This
VNTC
Getting the Priorities Right
The USB specification has very strict
restrictions on the maximum current
that can be pulled out of the bus. For
this reason the LTC3557 provides
load prioritization on the system load
NTC BLOCK
18
RNOM
100k
NTC
causes the battery to discharge and if
the battery voltage drops below 100mV
of its float voltage (4.2V for LTC3557
or 4.1V for LTC3557-1), an automatic
recharge cycle is initiated.
0.765 • VVNTC
–
TOO_COLD
19
+
RNTC
100k
–
0.349 • VVNTC
TOO_HOT
+
+
NTC_ENABLE
0.017 • VVNTC
–
Figure 4. Temperature qualified charging using NTC
27
L DESIGN FEATURES
(VOUT) as shown in Figure 3. Power
is always prioritized at VOUT and the
battery charge current is automatically
dialed back so that the USB current
limit is never exceeded. This feature
enables the battery charge current to
be programmed to normal operating
conditions rather than worst case
load on VOUT.
The charge current is also automatically dialed back at high temperatures
to prevent the part from overheating.
Additionally, if VOUT starts to drop
due to heavy load, the charge current
is dialed back to maintain VOUT near
VBUS. If the system load exceeds the
programmed USB current limit, the
additional current needed is drawn
from the battery. Power provided directly to VOUT pin via the WALL input
is prioritized over USB power as USB
power is current limited.
Status Symbols
The CHRG pin provides valuable information regarding the status of battery
charging. The CHRG pin is an open
drain output that is pulled low during a
normal charge cycle. When the charge
current reduces to one tenth of the
programmed value of charge current
(C/10) the CHRG pin is let go and is
pulled high by the external pull-up
device to the appropriate rail voltage.
Two Fault modes are also encoded
on to the CHRG output. If the battery
voltage fails to rise above 2.85V even
after charging it for a half hour, it is
deemed to be a bad battery and this
fault is reported at the CHRG pin as
a fast blink (6Hz signal modulated at
35kHz). Temperature qualified charging can be enabled with an external
resistor divider on the VNTC and NTC
pins as shown in Figure 4. This defines
a range of temperatures for charging
the battery and is a function of the
thermal characteristics of the NTC
resistor. When the battery temperature
is outside the defined range, an NTC
fault is indicated at the CHRG pin by a
slow blinking (1.5Hz signal modulated
at 35KHz).
Step-Down
Switching Regulators
The LTC3557 includes three internally compensated 2.25MHz
constant-frequency current-mode
step-down switching regulators providing 600mA, 400mA and 400mA
each. All step-down switching regulators can be programmed for a
minimum output voltage of 0.8V and
can be used to power a microcontroller
core, microcontroller I/O, memory or
other logic circuitry. Figure 5 shows
the step-down switching regulator
application circuit. The full-scale
output voltage for each step-down
switching regulator is programmed
using a resistor divider as shown in
the figure such that
 R1 
VOUTx = 0.8 V • 
+1
 R2 
Typical values of R1 are in the
range of 40kΩ to 1MΩ. The capacitor CFB cancels the pole created by
the feedback resistors and the input
capacitance of the FB pin, and also
helps to improve transient response
for output voltages much greater than
0.8V. A value of 10pF is recommended
for CFB for most applications.
All three of the step-down switching
regulators support 100% duty cycle
operation (low dropout mode) when
the input voltage drops very close to
the output voltage. Each regulator
VIN
EN
MODE
PWM
CONTROL
MP SWx
MN
L
CFB
R1
FBx
GND
0.8V
R2
VOUTx
COUT
can be individually enabled through
its respective enable pin.
A single MODE pin sets the three
voltage regulators in a high efficiency
Burst Mode operation (MODE = 1) or
low ripple pulse-skip mode (MODE =
0). For high enough load currents, in
either mode, the step-down switching regulators automatically switch
into constant frequency PWM mode
operation. The high 2.25MHz switching frequency allows the use of tiny
power inductors and stays out of the
AM Band.
The step-down switching regulators
also include soft-start to limit inrush
current when powering on, shortcircuit current protection and switch
node slew rate limiting circuitry to reduce EMI radiation. It is recommended
that the step-down switching regulator
input supplies (VIN1 and VIN2) be connected to the system supply pin (VOUT).
This allows the undervoltage lockout
circuit on the VOUT pin to disable the
step-down switching regulators from
operating outside the specified voltage range.
Power Sequencing using RST2
The RST2 open drain output responds
to step-down switching regulator 2 and
issues a Power ON reset signal 230ms
after the feedback voltage (FB2) rises to
within 8% of its final value. This output
can be pulled to a desired voltage level
using an external pull-up resistor and
used for sequencing power rails. For
example, it could be used to drive the
enable inputs of the other switching
regulators.
Conclusion
In summary, the LTC3557 provides a
highly integrated solution for handheld
and mobile applications in a compact
4mm × 4mm QFN package. The variety
of input power sources and externally
programmable output voltages make
it ideally suited for a broad range of
applications. The feature rich Li-Ion
charger provides protection against
several real-world fault conditions
while the versatile high frequency stepdown switching regulators provide
high efficiency power. L
Figure 5. Buck converter application circuit
28
Linear Technology Magazine • June 2007
DESIGN IDEAS L
Smart Battery Charger
for Battery Backup
Introduction
The most common power source used
for backup power is a battery. In a
backup power system it is important to
know if the battery is ready and reliable
at all times by constantly monitoring
its health and state of charge. Smart
Batteries are currently the best available industry standard system that
can satisfy these requirements.
Two important features of Smart
Battery Systems (SBS) are that they
are battery chemistry independent and
provide a built-in gas gauge. Because
the charging system no longer carries
the burden of charge monitoring and
applying chemistry-specific charge
algorithms, the charger itself can be
truly generic, accepting any Smart Battery, regardless of type or capacity. A
host system needs to do nothing other
than provide a Smart Battery charger
to guarantee that a healthy battery is
kept at full charge and a bad battery
is detected.
Design Ideas
Smart Battery Charger
for Battery Backup............................29
Mark Gurries
Tiny Comparator Fits Anywhere You
Need Micropower Control Functions
..........................................................31
Alexi Sevastopoulos
3-in-1 Device Replaces Battery Charger,
Overvoltage Protection and PowerPath
Manager for USB/Battery Powered
Devices...............................................33
Andy Bishop
Universal 12-Output
LED Driver Controls 4-RGB LEDs........35
Ted Henderson
12A Monolithic Synchronous Buck
Regulator Accepts Inputs up to 24V
..........................................................36
Stephanie Dai and Theo Phillips
0.25in2 × 1.8mm
Dual Output Converter
for Li-Ion to 3.3V and 1.8V.................39
John Canfield
Sub-µA RMS Current Measurement
for Quartz Crystals............................41
Jim Williams
Linear Technology Magazine • June 2007
This certainly simplifies charger
design. The same charger can be used
without modification in a variety of
products. It also simplifies field and
factory upgrades to different chemistries or higher capacities.
The LTC4100 Smart Battery charger
is primarily targeted at big battery
configurations in power hungry portable products, such as notebook
computers. Many new products do
not require the high voltage capability of the LTC4100, but still need all
the advantages of a SBS system. The
LTC4101 is a special version of the
LTC4100 Smart Battery charger that is
optimized to work with battery voltages
below 5.5V, while retaining the space
saving advantages of the LTC4100.
The LTC4101
Smart Battery Charger
The LTC4101 is a compact Smart
Battery charger optimized for battery
voltages below 5.5V. It shrinks overall
circuit size by reducing the size of
external components. For instance,
it takes advantage of the compact
ceramic capacitors’ space saving
features while avoiding any audible
noise. It also operates at a high 300kHz
switching frequency, which allows the
use of a very small, low cost inductor.
Inductor values can be as low as 4µH
at 4A with 7.5V of input.
The LTC4101 is a Level 2 (slave)
Smart Battery charger that is compliant with both Smart Battery charger
V1.1 and SMBus V1.1 standards.
Input voltage range is 6V to 28V while
the output charge voltage range is from
3V to 5.5V. A 10-bit current DAC and
an 11-bit voltage DAC, with current
accuracy of 5% and voltage accuracy
of 0.8%, respectively, provide precision
charge capabilities. A topside P-channel MOSFET allows 98% maximum
duty cycle, dramatically reducing total
part count and IC pin count while
maintaining efficiency greater than
95% (see Figure 2).
by Mark Gurries
The LTC4101 also offers many
unique features, including a current
limit and voltage limit system that
prevents SMBus data corruption errors from generating harmful charge
values. A patented SMBus accelerator1 increases data rates in high
capacitance traces while preventing
bus noise from corrupting data (see
Figure 3).
Figure 1 shows a typical compact
single battery charger. This circuit
can charge batteries with up to 1A
and switch continuously down to zero
load current. The LTC4101 is capable
of charging currents up to 4A. Other
features include:
qan AC present signal with
precision 3%-accurate user
adjustable trip points
qa safety signal circuit that rejects
false thermistor tripping due to
ground bounce caused by the
sudden presence of high charge
currents
qa DC input FET diode circuit that
prevents battery current from
flowing backwards into the wall
adapter or DC power source
qan ultrafast overvoltage
comparator circuit that prevents
voltage overshoot when the
battery is suddenly removed or
disconnects itself during charge.
qVLIM and ILIM settings that
are used to protect the battery
from excessive voltage or current
conditions that could occur if
there are data corruption errors
in SMBus communication.
qan input current limit sensing
circuit2 that is used to limit
charge current to prevent wall
adapter overload as the system
power increases.
Ceramic Capacitors Reduce
Size and Improve Reliability
One of the biggest space saving changes that has occurred in recent years is
the use of high capacitance and volt29
L DESIGN IDEAS
DCIN
9V to 12V, 2A
1.21k
17
3V TO 5.5V
11
6
CHGEN
10
ACP
7
9
8
15
16
13
1.13k
14
10k
54.9k
0.1µF
6.04k
20
LTC4101
VDD
DCIN
DCDIV
INFET
CHGEN
CLP
ACP
CLN
SMBALERT
TGATE
SCL
BGATE
SDA
PGND
THB
CSP
THA
BAT
ILIM
VSET
VLIM
ITH
IDC
0.068µF
GND
Q1A
Q1: SIA811DJ
Q2: SI5513DC
0.1µF
5
4
0.05Ω
VBAT
PART
< 5.5V
> 5.5V
LTC4101
LTC4100
5.11k
24
SYSTEM LOAD
23
1
3
Q2A
4.7µF
Q2B
24µH
1A
2
0.1Ω 1%
Q1B
4.7µF
21
22
18
19
12
0.03µF
6.04k
100Ω
0.0015µF
0.12µF
0.1µF
SMART
BATTERY
SafetySignal
SMBALERT#
SMBCLK
SMBCLK
SMBDAT
SMBDAT
Figure 1. Charger with 2A input current limiting and 1A of charge power
age (high C/V) ceramic capacitors. In
switching regulator applications, the
low ESR of ceramics allows them to
handle a relatively large ripple current
per microfarad while remaining relatively inexpensive. Battery chargers
can reap the same benefits provided
their feedback loops are stable with
ceramic capacitors.
Ceramics come with their own
unique challenges such as piezoelectric properties that can result in
audible noise if there are AC currents
with audible frequencies present. Such
frequencies can occur in battery chargers at two load extremes: low dropout
and light load.
Battery chargers run up against
wall adapter voltages that are often
100
VIN = 8V
POWER EFFICIENCY (%)
96
VIN = 20V
84
80
76
72
0.0
0.5
1.0
1.5 2.0
IOUT (A)
2.5
3.0
Figure 2. Efficiency at
single-cell Li-ion voltages
30
increases the switching frequency
proportional to the reduction in inductance, the output capacitance
can remain the same. The LTC4101
operates at a switching frequency
of 300kHz, allowing tiny, low profile
inductors to be used.
Conclusion
The LTC4101 Smart Battery retains
all the same compact form factor advantages of the LTC4100 while being
optimized for low voltage battery packs
that can be found in compact products
that require battery backup. L
Notes
1. U.S. patent number 6650174
2. U.S. patent number 5723970
VDD = 5V
5V CBUS = 200pF
High Switching Frequencies
Keep Inductors Small
92
88
just a few volts above the peak battery
voltage. Depending on the design, as
the charger approaches 100% duty
cycle, the switching frequency passes
though the audible range on the way
to DC. Alternatively, conditions where
the charge current falls below the
PWM controller’s ability to maintain
regulation can create discontinuous
switching cycles or cycle-skipping.
Cycle-skipping switching periods
can occur in the audible range. This
typically happens when batteries
momentarily disconnect themselves
during the charge process for termination condition evaluation, thus forcing
the charge current to zero. Ceramic
capacitors translate cycle skipping
or low dropout switching activity into
audible noise. The LTC4101 avoids
this problem by switching continuously under all loads, even 0A.
3.5
Charger system designers are often
driven to reduce inductance values
to take advantage of smaller form factor components. The problem is that
less inductance for a given switching
frequency results in more inductor
ripple current, which increases the
output capacitor size. However, if one
LTC4101
RPULLUP = 15k
0V
1µs/DIV
4101 G09
Figure 3. Built-in SMBus accelerator improves
rise time performance and noise margin
Linear Technology Magazine • June 2007
DESIGN IDEAS L
Tiny Comparator Fits Anywhere You
Need Micropower Control Functions
by Alexi Sevastopoulos
Introduction
It’s rare that an IC offers such a simple
solution to so many common problems
that it instantly becomes a favorite
building block in the system designer’s
toolset. The LT6703 micropower, low
voltage comparator and reference
does just that by squeezing a single
micropower comparator and accurate
reference into a tiny 2mm × 2mm
DFN package. Although only one of
its comparator inputs is accessible
(the other is connected to a 400mV
internal precision voltage reference)
its size makes it easy to fit just about
anywhere even on the most crowded
circuit boards.
The LT6703 is a smaller and simpler version of its sibling, the LT6700
dual comparator and reference. Its
open-collector output enables level
shifting, while its Over-The-Top® capabilities allow the input voltage
range to span from –0.3V to 18V with
respect to ground, regardless of the
supply voltage. The internal bandgap
voltage reference has an output voltage of 400mV ±1.25% over its wide
temperature range (–40 to 125°C). The
LT6703-2 and LT6703-3 differ by the
polarity of the available comparator
input and runs on 6.5µA with a typical
propagation delay of 25µs.
The LT6703-2 has an available
inverting input while the LT6703-3
(Figure 1) has an available non-inverting input. The comparator has
6.5mV of built-in hysteresis to ensure
stable operation. In the LT6703-3,
this hysteresis level can be increased
using positive feedback circuitry. The
threshold voltage, which represents
the combined reference accuracy
and comparator offset, is guaranteed
at ±1.25% at 25ºC. This threshold
accuracy, in addition to the built-in
6.5mV of hysteresis, provides a clean
switching threshold that the user can
rely on even with slow varying inputs.
For extra protection and to help elimiLinear Technology Magazine • June 2007
VS
LT6703-3
+
+IN
OUT
–
VS
400mV
REFERENCE
GND
Figure 1. Block diagram of tiny
2mm × 2mm 1.4V-to-18V comparator
nate false triggering, a supply bypass
capacitor should be added to prevent
power supply glitches from disturbing
the reference voltage.
Features for Versatility
and Ease of Use
Wide Supply Range
The unique supply range of the LT6703
enables it to meet the standards of
many industrial or battery-operated
applications. In industrial applications where voltages above 5.5V are
typically used, the LT6703 has no
problem since its supply stretches up
to 18V. Likewise, in battery-powered
applications the supply reaches as
1.4V ≤ VIN ≤ 18V
(VTH = 3V)
1M
+IN
0.1µF
1M
VS
LT6703-3
+
OUT
–
154k
400mV
REFERENCE
VS
GND
Figure 2. Micropower supply voltage monitor
far down as 1.4V. This ability to run
from a low voltage, combined with a
low 6.5µA supply current, make the
LT6703 ideal for low voltage system
monitoring (shown in Figure 2).
As shown in Figure 2, the LT67033 can be run from a power supply
rail or from a battery. In this system
monitoring application, the output of
the comparator goes low whenever the
supply drops below the 3V threshold
voltage—indicating that the system is
running low on batteries or that there
was a power failure or brown-out.
Although the LT6703 is specified as
having ±10nA of input bias current,
large input resistors are recommended
to reduce overall supply current as
shown in Figure 2. However, if the
two input resistors are increased by
a factor of ten, the input bias current
of the comparator begins to affect the
threshold value. With these larger
input resistors and a supply voltage
of 3V, the current through the input
resistors is 260nA. With an input bias
current of ±10nA, the comparator now
sinks a significant portion of the supply
current required to set the threshold
voltage at the comparator input. As a
result, an increase in supply voltage of
a few hundred millivolts is required in
compensation to reach the 400mV trip
point. However, with the values shown
in Figure 2, the current through the
two input resistors is 2.6µA at the trip
point, which considerably outweighs
the comparator bias current and
thus produces a reliable threshold
voltage.
Over-The-Top Input and
Open-Collector Output
The LT6703 features Over-The-Top
operation, which allows inputs with
amplitudes as high as 18V, regardless of the supply voltage. In other
words, operation at a low supply does
not limit the input level. This feature,
31
L DESIGN IDEAS
12V
1.8V
1.8V
VS
15V
0V
+IN
LT6703-3
OUT
VOUT
15V
VS
10k
1.8V
1.8V
0V
0V
10k
LT6703-3
+IN
GND
ILOAD
VOUT
OUT
RS
0.1Ω
above the 400mV threshold voltage, a
relay is tripped, cutting off the supply.
Current conduction through the load
is prevented as well. The output of the
comparator remains high until the
power supply is cycled back on and the
load current decreases to below 4mA.
When the output of the comparator
is low, the part is capable of sinking
up to 40mA from the supply through
the relay although in this case it will
only sink 6mA.
The 100µF capacitor shown in Figure 6 is responsible for pulling current
through the relay coil. The large value
is important because it allows enough
time for the relay’s internal switch to
close and kick-start the circuit. The
response time between the relay trip
and supply reset is 40µs, regardless
of the capacitor value. Figure 7 shows
a modification to the circuit, allowing
the circuit to restart without cycling
the power supply. The auto-restart
loop monitors the current through
the load. The 1µF capacitor in the
loop ensures that the supply of the
1k
LT6703-2
–IN
GND
along with the part’s wide supply
range, is especially useful in portable
battery-powered applications, allowing a flexibility in input and supply
voltage ranges that cannot be found
in competing devices.
The comparator’s open collector
output also provides great flexibility.
This allows the device to be used as
a level translator since the output
can be pulled up to 18V regardless
of the supply voltage (Figures 3 and
4). In Figure 3, the LT6703-3 takes a
15V pulse input and translates it to
a 1.8V output, all while running on
a 1.8V supply. A simple modification
reverses the translation as shown in
Figure 4.
The use of multiple LT6703’s also
permits logical wire-AND implementation and can drive relatively heavy
loads (up to 40mA) such as relays or
LED indicators.
32
VS
LOAD
0V
Figure 4. Simple level translator for
shifting low voltages to high voltages
The LT6703 can also be used to trigger
an alarm dependent upon the amount
of load current through an external
sense resistor. In Figure 5, an LED is
used on the output as an alarm signal.
If the load current exceeds 4A, the
sense resistor voltage rises above the
400mV threshold, triggering a state
change on the output of the comparator. The internal NPN transistor at the
output of the comparator now allows
current to flow through it to ground,
lighting up the LED and letting the user
know that there is excessive current
being conducted through the load.
In Figure 6, the load is protected
by more than just an LED warning
indicator. Once current through the
load has exceeded the set limit and the
voltage across the sense resistor rises
LED
(ON IF ILOAD > 4A)
15V
Figure 3. Simple level translator for
shifting high voltages to low voltages
Overload Protection
5V
OUT
GND
Figure 5. Low side current sense alarm
comparator does not turn back on
when the output goes high. As the
load current is decreased, the supply
voltage gradually increases. When it
hits 1.4V, the output goes low and
the relay switch closes, turning the
circuit back on.
Conclusion
Linear Technology continues to innovate by crafting the LT6703 series
of precision, micropower comparators
in a tiny 2mm × 2mm DFN package.
These products provide an excellent
solution to many design challenges
for threshold detection applications,
with characteristics accommodating wide temperature spans and
space-critical designs. Its unique
Over-The-Top® feature offers versatility and performance ideal for
portable, battery-powered commercial products as well as industrial
or high-temperature grade system
monitoring applications. The LT6703
excels in all specifications that set
system performance. L
COTO 9001-12-01
COTO 9001-12-01
12V
12V
100µF
100µF
10k
1k
1k
ILOAD
VS
LOAD
+IN
RS
100Ω
LT6703-3
1µF
ILOAD
OUT
GND
Figure 6. Latch-off protection circuit
+IN
RS
100Ω
1k
VS
LOAD
LT6703-3
OUT
GND
Figure 7. Latch-off protection circuit
with load sensing auto-reset
Linear Technology Magazine • June 2007
DESIGN IDEAS L
3-in-1 Device Replaces Battery
Charger, Overvoltage Protection
and PowerPath Manager for
USB/Battery Powered Devices
by Andy Bishop
Introduction
An efficient Li-Ion battery powered
system requires at least three distinct
circuits to control the power path
between the load, the battery and the
power source (see Figure 1). The minimal circuit requirements include:
qa battery charger,
qa power switch to select powering
the load from either the battery or
the wall adapter (when present),
qand a current regulator for the
wall adapter/USB input.
This, of course, assumes that the
load draws power from a communal
power bus, as opposed to attaching
directly and exclusively to the battery.
A direct-to-battery topology might be
simpler, precluding the need for the
power path controller and regulator,
but it is far less efficient and significantly more restrictive. For instance,
if the battery is fully drained, no power
can be delivered to the load, even if
wall adapter power is available.
The LTC4067 Li-Ion charger and
PowerPath™ controller combines the
efficiency, flexibility and robust nature
of a 3-chip solution with the simplicity of a direct-to-battery topology by
replacing three components with a
single device, as shown in ‎Figure 2.
The LTC4067’s advanced topology
battery charger optimizes power utilization while limiting input current to
a programmable level, making it ideal
for USB powered applications.
Working with USB Port
Current Limits
In applications where input current
consumption is constrained, the
LTC4067 is able to satisfy USB power
requirements. Take the example of a
portable device with a disk drive that
draws power from either a battery or
Linear Technology Magazine • June 2007
CURRENT
LIMIT
charge current monitoring, allowing
the application to perform advanced
gas-gauge functions.
USB OR
WALL ADAPTER
U2: LTC4411
IDEAL DIODE
Li-Ion
U3: LTC4053
CHARGER
SYSTEM
SUPPLY
Figure 1. Battery charger current-limit
and ideal diode supply connections with
intermediate voltage bus.
U1: LTC4067
USB OR
WALL
ADAPTER
IN CURRENT
LIMIT
IDEAL DIODE
OUT
BAT
CHARGER
SYSTEM
SUPPLY
Li-Ion
Figure 2. Intermediate voltage bus
supply connections with the LTC4067
the USB. Peak current consumption
may readily exceed USB limits when
the disk is spinning up. In this situation, the LTC4067 optimizes power
management by sharing the load between the battery and the USB, while
limiting the current from the USB port.
When load current decreases, the
LTC4067 automatically switches over
to charge the battery with any excess
USB current that is not consumed
by the load.
The LTC4067’s input current limit
is programmable via a resistor at the
CLPROG pin. Control inputs ILIM0
and ILIM1 are used to set USB high
power, low power or suspend operating modes—or allow for much larger
current limit when powering from a
wall adapter. The LTC4067 also provides instantaneous USB current and
Working with Unregulated
Wall Adapters
With the addition of an external highvoltage PFET, the LTC4067 provides
an automatic overvoltage protection
function that allows the LTC4067 to
automatically disconnect itself in the
event that the wrong wall adapter is
applied.
Figure 3 illustrates an application
where the LTC4067 charges a singlecell Li-Ion battery from a 1A wall
adapter. The overvoltage protection
circuit includes the OVI and OVP pins
of the LTC4067 and an external PFET
in series with the IN pin. The PFET
serves to disconnect the LTC4067
from potentially damaging overvoltage
conditions. When the OVI input senses
a voltage greater than 6V, the OVP
output pulls up to disable the PFET.
When OVI falls below this threshold,
the OVP output falls low, turning on
this PFET. Note that the body diode of
this PFET is connected so that it does
not forward bias when an overvoltage
condition exists. While the overvoltage
condition persists, the input power
path is disabled, but system power is
provided by the battery. A 10nF capacitor placed from OVI to OVP ensures
that the PFET is quickly disabled in the
event that fast edges occur when the
wall adapter is suddenly hot-plugged.
An optional, low power Zener diode is
also recommended in the event that
voltage surges occur after the device
is powered.
In the example of Figure 3, the input
current limit from the wall adapter is
programmed to 1A with a 1k resistor
33
L DESIGN IDEAS
from CLPROG to GND, assuming ILIM0
and ILIM1 are held high, or 200mA if
ILIM0 and ILIM1 are both held low.
The charge current is independently
programmed to 500mA via the 2k resistor from the PROG pin to GND.
An optional second external PFET
connected between OUT and BAT
serves as a high performance ideal
diode to connect the load to the battery
with an extremely low impedance. The
GATE output pin enables this ideal
diode when the wall adapter disconnects or when the load demands more
current than the wall adapter supplies.
Note that this PFET is connected so
that the internal body diode from drain
to source does not forward bias when
the voltage at OUT is greater than the
voltage at BAT.
The LTC4067 allows for instantaneous monitoring of both input current
and charge current for advanced gas
gauge functions by measuring the
voltages at the CLPROG and PROG
pins, respectively. The optional NFET
(Q3) tied in series with the PROG pin
resistor serves to engage a low power
shutdown mode, where total quiescent
current drops to less than 20µA.
Full Featured USB Li-Ion Charger
Figure 4 illustrates an application for
charging a single-cell Li-Ion battery
directly from the USB, conforming
to the USB requirements for low
power (LPWR), high power (HPWR),
or self-powered functions. Here, the
LTC4067 ensures that the load at OUT
sees the USB potential when the USB
port is present. When the USB port is
removed, the load powers from the battery through an internal 200mΩ ideal
diode. Optionally, for more demanding
applications, an external PFET driven
by the GATE pin improves performance
by reducing the series resistance of
the ideal diode.
The 2k resistor at the CLPROG
pin ensures that the maximum current drawn from the USB input port
remains below the maximum allowed
depending on the permitted power
allocation: 500mA for HPWR USB function or 100mA for LPWR USB function.
By driving the ILIM0 pin low and the
ILIM1 pin high, the LTC4067 complies
34
OVI
LOAD
OUT
10nF
SOURCE
OVP
10µF
CHRG
GATE
Q1
WALL
ADAPTER
1µF
Q2
IN
DRAIN
LTC4067
D1
OPTIONAL
10µF
OPTIONAL
NTC
BAT
1-CELL
Li-Ion
CHARGE CURRENT
MONITOR
PROG
RPROG
2k
ILIM0
INPUT
CURRENT
MONITOR
IN
ILIM1
CLPROG
RCLPROG
1k
GND
Q3
ENABLE
Q1, Q2: IRLML6402
D1: MMSZ5234B
Figure 3. Li-Ion charger/controller with overvoltage protection
with the USB SUSPEND specification,
whereby the load at OUT powers from
the battery and the only current drawn
from the USB port is due to the two
series NTC pin resistors.
The 2k resistor at the PROG pin
selects 500mA for the charge current,
automatically charging a single-cell
Li-Ion battery following a constantcurrent/constant-voltage (CC/CV)
algorithm with a built-in timer that
halts charging two hours after the
charger enters constant-voltage mode.
Note that actual charge current depends on the load current, as the
charger shares the USB current with
the load.
During a charge cycle, the CHGB
status pin signals that the battery is
charging in constant-current mode by
pulling to GND through an open-drain
drive output capable of driving an LED
for visual indication of charge status.
When the charge current drops to less
than about 9% of the programmed
charge current and the battery is above
the recharge threshold (4.1V), the
CHGB pin assumes a high impedance
state (although top-off charge current
continues to flow until the internal
charge timer elapses). Bad battery and
battery out-of-temperature conditions
are also flagged with the CHGB pin by
a series of flashing pulses.
If the load demands more current
than allowed by the USB current
limit, the charge current automatically
scales back. As the load demands
more current than available from the
USB port, charge current decreases
to zero, at which point an ideal diode
function from BAT to OUT turns on as
the OUT voltage drops below the BAT
voltage. When the ideal diode engages,
the battery charge cycle pauses, and
the load draws current from both the
USB port and the battery. When the
load current decreases such that the
continued on page 38
SOURCE
CHRG
USB INPUT
IN
OPTIONAL
OUT
10µF
NTC
TO LOAD
10µF
GATE
LTC4067
BAT
ILIM0
IN
RCLPROG
2k
ILIM1
PROG
CLPROG
GND
DRAIN
1-CELL
Li-Ion
RPROG
2k
Figure 4. USB battery charger/controller
Linear Technology Magazine • June 2007
DESIGN IDEAS L
Universal 12-Output LED Driver
By Ted Henderson
Controls 4-RGB LEDs
Introduction
RGB LEDs produce a wide range
of colors, including white, making
them highly versatile. Each RGB LED
requires three drivers, one for each
color LED. Using a multiple output
LED driver for RGB applications can
save solution size and cost versus
single LED drivers. The LTC3207 and
LTC3207-1 each provide 12 individually programmable current sources
(universal drivers). This allows them
to drive up to 12 white LEDs or four
RGB LEDs, as shown in Figure 1. Each
universal LED driver is controlled
by a dedicated 6-bit linear DAC that
covers an LED output current range
of 400µA to 28mA. Any unused universal or camera outputs can simply
be connected to ground and left unprogrammed by the I2C port.
The LTC3207 and LTC3207-1 include all of the functions required to
drive 12 LEDs and one camera LED,
including the following features: a
high power multimode charge pump
with automatic mode reset, a precision
internal current source and voltage
reference to set full scale current,
and 13 precision LED current source
outputs each controlled by a DAC
and an I2C data interface. Only five
small external ceramic capacitors are
required. The LTC3207 and LTC32071 are packaged in a small, low profile
C2
2.2µF
VBAT
C1
2.2µF
C3
2.2µF
4mm × 4mm 24-pin QFN plastic package and can operate over an input
voltage range of 2.9V to 5.5V.
Features
Automatic Blinking and Gradation
Reduce I2C Bus Traffic
The LTC3207 and LTC3207-1 have
incorporated features that greatly reduce I2C bus traffic. The universal LED
outputs can be programmed to blink
at rates up to 2.5 seconds independent
of direct I2C control. Gradation times
from 0.25s to 1s can be programmed
to smoothly ramp the brightness of any
channel from off to the programmed
current and down to zero independent
of the I2C port.
An ENU pin is also available to
directly enable the universal drivers
independent of the I2C port once the
device has been programmed. Each
universal output can be individually
programmed to gradate or blink. Each
universal output can also be controlled
by the ENU pin. Application Note 108
(available on our web site at www.
linear.com) outlines recommended
programming examples for all of these
features.
High Power Charge Pump
Both parts automatically change the
charge pump mode to optimize ef-
C4
4.7µF
RGB
C1P C1M C2P C2M
VBAT
CPO
RGB
LTC3207/LTC3207-1
ULED1-12
I2C
ENABLE DISABLE
2
SCL/SDA
ENU
CAMHL
12
CAM
GND
Figure 1. A four RGB LED driver
Linear Technology Magazine • June 2007
RGB
RGB
ficiency. Initially the part starts in 1x
mode. When a dropout is detected,
indicating that the LED driver voltage
is too low to maintain the programmed
current, the charge pump changes
modes to 1.5x (4.6V). A subsequent
dropout changes the charge pump to
2x mode (5.1V). The charge pump is
automatically reset to 1x mode whenever an I2C write occurs, gradating
down has completed, a blink period
has completed, a camera flash has
completed, or when ENU goes low.
Soft-start at power up and between
charge pump mode changes ensures
low inrush currents. Slew rates on the
flying capacitor pins C1M, C1P, C2M
and C2P are controlled to minimize
conducted and radiated noise. The
charge pump can be forced to operate in 1x, 1.5x or 2x mode via the I2C
port for applications where the charge
pump is used to power external devices
or when the supply voltage is high
enough such that the charge pump
is not required.
Serial Port
The microcontroller-compatible I2C
serial port provides all of the command
and control inputs for the LTC3207
and LTC3207-1. There are 16 data
register, one address register and one
sub-address register. The maximum
clock operating frequency is 400kHz.
These parts are receive-only (slave)
devices. Two I2C addresses are available by using either the LTC3207 or
LTC3207-1.
Conclusion
The small package and high level
of integration of the LTC3207 and
LTC3207-1 make these parts an excellent choice for a wide range of LED
applications. The blinking and gradation features coupled with individual
LED current control and simple LED
disable features make these parts truly
universal, extremely easy to use with
minimal I2C bus traffic. L
35
L DESIGN IDEAS
12A Monolithic Synchronous Buck
Regulator Accepts Inputs up to 24V
by Stephanie Dai and Theo Phillips
Introduction
Flexible Control
The LTC3610 is a high power monolithic synchronous buck regulator
capable of providing up to 12A from
inputs as high as 24V in a complete
solution that takes little space (Figure 1). It integrates the step-down
controller and power MOSFETs into
a single, compact 9mm × 9mm QFN
package. Its high step-down ratio,
wide input and output voltage range
and high current capability present a
single IC solution for many applications previously requiring separate
FETs and controller ICs. Its very low
profile (0.9mm max) allows mounting
on the back of a circuit board, freeing
up valuable front-side board space.
High step-down ratios (Figure 2) are
possible because of the LTC3610’s
constant on-time operation and
valley current control architecture,
which allow a minimum on-time of
less than 100ns. Output voltages
approaching VIN are also possible
(Figure 5). In either case, efficiency is
very high—up to 97% (Figures 4 and
6). Synchronous operation affords high
efficiency at low duty cycles, whereas
a non-synchronous converter would
conduct current through the forward
drop of a Schottky diode most of the
time. Transient response (Figure 3) is
fast because the LTC3610 reacts immediately to a load increase. It does
Figure 1. Who says a lot of space is needed
for a complete high power density stepdown regulator? The LTC3610 is capable of
providing up to 12A from inputs as high as
28V. Its low 0.9mm profile allows it to be
mounted on the back of the board too.
INTVCC
CVCC
4.7µF
6.3V
GND
CF
0.1µF
25V
SW
RF1
1Ω
VIN
VIN
VIN
5V TO 24V
CIN
10µF
35V
3×
C6
10µF
35V
+
(OPTIONAL)
GND
12
13
14
15
16
SGND
SVIN
SGND
SVIN
INTVCC
SW
INTVCC
PGND
PGND
PGND
PGND
PGND
PGND
PGND
PGND
SGND
SW
ION
LTC3610
SW
SGND
SW
FCB
SW
ITH
PVIN
VRNG
PVIN
PGOOD
PVIN
VON
PVIN
SGND
PVIN
RX1
0Ω
47
46
R1
9.5k
1%
45
44
43
41
40
(OPTIONAL)
CON
0.01µF
39
38
DB
CMDSH-3
SW
C2
VOUT
R5
31.84k
VIN
CC1
470pF
36
35
RPG1
100k
34
33
RVON PGOOD
0Ω
RSS1
510k
CB1
0.22µF
(OPTIONAL)
37
17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32
INTVCC
R2
30.1k
1%
RON
182k
1%
42
SW
CIN: TAIYO YUDEN GMK325BJ106MM-B
COUT: SANYO 10TPE220ML
L1: CDEP85NP-R80MC-50
C5: MURATA GRM31CR60J226KE19
EXTVCC
C4
0.01µF
48
SGND
11
SW
SGND
10
VFB
RUN/SS
9
SW
BOOST
8
EXTVCC
SGND
7
(OPTIONAL)
GND
SGND
SW
NC
6
SW
SW
5
PVIN
L1
0.8µH
PVIN
+
PVIN
COUT1
220µF
2×
SGND
PVIN
C5
22µF
6.3V
PGND
PVIN
4
SGND
PVIN
VOUT
2.5V AT
12A
SGND
PGND
PVIN
3
PGND
PVIN
2
PVIN
1
PGND
64 63 62 61 60 59 58 57 56 55 54 53 52 51 50 49
CSS
0.1µF
CC2
100pF
INTVCC
VOUT
VIN
(OPTIONAL)
RUN/SS
Figure 2. This converter runs at 550kHz and delivers 2.5V at 12A from an extremely wide 5V–24V input.
36
Linear Technology Magazine • June 2007
DESIGN IDEAS L
100
VOUT
100mV/DIV
EFFICIENCY (%)
90
IL
5A/DIV
ILOAD
5A/DIV
VIN = 12V
80
70
60
40µs/DIV
LOAD STEP 0A TO 8A
VIN = 12V
VOUT = 2.5V
FCB = 0V
FIGURE 6 CIRCUIT
(the ITH pin) rises, initiating another
cycle. As the load current rises, so
does the average inductor current.
Eventually, the interval between
constant on-time pulses ends before
the inductor current can reach zero,
at which point the inductor continuously conducts current. This point is
determined by duty cycle, inductance
value, and the interval between constant on-time pulses. By using single
on-time pulses of fixed width, this
mode provides well-controlled output
ripple at any supported load. This
process also prevents reverse inductor
current, which minimizes power loss
at light loads.
The on-time is set by the current into
the ION pin and the voltage at the VON
pin according to a simple equation
VIN = 5V
VOUT=2.5V
EXT VCC=5V
50
0.01
0.1
1
10
LOAD CURRENT (A)
100
Figure 3. The LTC3610 responds quickly to an
8A transient (circuit of Figure 2).
Figure 4. Efficiency vs load current for the
circuit of Figure 2
not wait for the beginning of the next
clock cycle to respond, so there is no
clock latency.
The LTC3610 can be programmed
for two kinds of light-load operation:
forced continuous mode or discontinuous mode. Forced continuous
operation offers the lowest possible
noise and output ripple. The top
MOSFET turns on for the programmed
on-time and the bottom MOSFET
turns on for the (remaining) off-time.
Inductor current is allowed to reverse,
even at no load.
In discontinuous mode, the top
MOSFET turns on for a preset ontime. Then (after a brief non-overlap
period) the bottom MOSFET turns on
until the current comparator senses
reverse inductor current. When the
error amplifier senses a small decrease
at the feedback node VFB, its output
CVCC
4.7µF
6.3V
INTVCC
CF
0.1µF
25V
SW
GND
TON =
VVON
IION • 10pF
Tying a resistor RON from VIN to
the ION pin yields an on-time inversely
proportional to VIN.
RF1
1Ω
VIN
11
VIN
VIN
5V TO 24V
CIN
10µF
25V
3×
C6
10µF
35V
+
(OPTIONAL)
GND
12
13
14
15
16
SGND
SVIN
SGND
SVIN
INTVCC
SW
INTVCC
PGND
PGND
PGND
PGND
PGND
PGND
PGND
PGND
SGND
SW
ION
LTC3610
SW
SGND
SW
FCB
SW
ITH
PVIN
VRNG
PVIN
PGOOD
PVIN
VON
PVIN
SGND
PVIN
SGND
RX1
0Ω
CIN: TAIYO YUDEN TMK432BJ106MM
COUT: SANYO 16SVP180MX
L1: SUMIDA CDEP1055R7
48
EXTVCC
C4
0.01µF
47
46
R1
1.58k
1%
45
44
43
41
40
(OPTIONAL)
R2
30.1k
1%
C2
VOUT
RON
3.4M
1%
42
(OPTIONAL)
CON
0.01µF
39
38
VIN
CC1
560pF
R5
20k
37
36
35
34
33
SGND
10
SW
SGND
9
VFB
RUN/SS
8
SW
BOOST
(OPTIONAL)
GND
EXTVCC
SGND
7
SGND
SW
NC
6
SW
SW
5
PVIN
L1
5.7µH
PVIN
+
PVIN
COUT
180µF
16V
SGND
PVIN
C5
22µF
25V
PGND
PVIN
4
SGND
PVIN
VOUT
12V AT
5A
SGND
PGND
PVIN
3
PGND
PVIN
2
PVIN
1
PGND
64 63 62 61 60 59 58 57 56 55 54 53 52 51 50 49
CC2
100pF
RPG1
100k
INTVCC
PGOOD
(OPTIONAL)
RVON
17 18 19 20 21 22 23 24 25 26 27 28 29 30 31 32
RSS1
510k
CB1
0.22µF
INTVCC
DB
CMDSH-3
CSS
0.1µF
CVON
VOUT
(OPTIONAL)
VIN
(OPTIONAL)
RUN/SS
Figure 5. Although the LTC3610 is optimized for high step-down ratios, it can also regulate output voltages beyond the range of many DC/DC buck
converters. For example, this schematic shows a 500kHz regulator delivering 12V at up to 5A, with high efficiency and low output ripple.
Linear Technology Magazine • June 2007
37
L DESIGN IDEAS
100
Adjustable current limit is also builtin. The inductor current of LTC3610 is
determined by measuring the voltage
across the sense resistance between
the PGND and SW pins, where RDS(ON)
of the bottom MOSFET is about 6.5mΩ.
The current limit is set by applying a
voltage to the VRNG pin, which sets the
relative maximum voltage across the
sense resistance. An external resistive
divider from the internal bias, INTVCC,
can be used to set the voltage of the
VRNG pin between 0.5V and 1V resulting in a typical current limit of 16A to
19A. Tying VRNG to SGND defaults the
current limit to 19A.
The LTC3610 also has soft-start
and latch off functions enabled by
the Run/SS pin. Pulling the Run/SS
below 0.8V puts the LTC3610 into
a low quiescent current shut down
state, whereas releasing the pin allows
a 1.2µA current source to charge up
the external soft-start capacitor. When
the voltage on Run/SS reaches 1.5V,
the LTC3610 begins operating with
an initial clamp on ITH of approximately 0.9V. This prevents current
overshoot during start up. As the
soft-start capacitor charges, the ITH
clamp increases, allowing normal
operation at full load current. If the
output voltage falls below 75% of the
LTC4067, continued from page 34
Conclusion
OUT voltage rises above the BAT voltage, the charge cycle restarts where
it left off.
At any time, the user may monitor
both instantaneous charge current
and instantaneous USB current by
observing the PROG pin and CLPROG
pin voltages respectively.
LTC2355/56, continued from page 21
power, and small package makes the
LTC2356-14 ideal for high speed,
portable applications including data
acquisition, communications, and
medical instrumentation.
The LTC2356-14 achieves 72.3dB
SINAD and –82dB SFDR with a 1.4MHz
input frequency. While measuring
±1.25V bipolar inputs differentially,
the LTC2356-14’s 80dB common mode
rejection ratio allows users to eliminate
ground loops and common mode noise.
When the ADC is not converting, power
dissipation can be reduced to 4mW
in nap mode, with the internal 2.5V
reference remaining active, and 13µW
with all analog circuitry powered down
in sleep mode.
38
VIN = 24V
EFFICIENCY (%)
80
60
40
20
VOUT=12V
1
10
100
1000
LOAD CURRENT (mA)
10000
Figure 6. Efficiency vs load current
for the circuit of Figure 4
regulated voltage, then a short-circuit fault is assumed. At this point,
a 1.8µA current discharges capacitor
CSS. If the fault condition persists until
Run/SS drops to 3.5V, the controller’s
overcurrent latch off turns off the
MOSFETS until Run/SS is grounded
and released. If latch off is not desired,
a pull-up current source at Run/SS
defeats this feature.
Conclusion
Few synchronous monolithic DC/DC
converters are versatile enough to use
in low power portable devices such as
notebook and palmtop computers, as
well as high power industrial distributed power systems. The LTC3610’s
broad input and output ranges, efficiency greater than 90% and high
current capability make it a superior
alternative to many solutions requiring
separate power switches. L
The LTC4067 satisfies the needs of voltage sensitive battery operated devices,
replacing as many as three separate devices. With accuracy better than ±0.4%
on the battery float voltage, the LTC4067
is ideally suited for demanding highprecision applications. The LTC4067
offers both a power management
strategy that complies with USB port
specifications as well as providing an
advanced battery charger. The LTC4067
also offers overvoltage protection up to
13V, to protect itself as well as system
devices in the event that an incorrect
wall adapter is attached. L
For applications requiring a unipolar measurement, the LTC2355-14
measures 0V to 2.5V input signals, but
is otherwise identical to the LTC235614. For lower resolution applications,
the LTC2356-12 and LTC2355-12
are pin- and software-compatible 12bit versions of the LTC2356-14 and
LTC2355-14.
The LTC2355-14/LTC2356-14/
LTC2355-12/LTC2356-12 ADCs are
pin- and software-compatible with
the LTC1403 2.8Msps ADC family,
allowing users to easily upgrade their
design for a 25% faster sample rate.
Table 2 details these fast single-channel unipolar and bipolar ADCs.
Summary
With PCB real estate getting tighter and
designers always searching for lower
power ICs, fast data acquisition can
be a challenge. Linear Technology’s
families of simultaneous sampling
ADCs and fast single-channel ADCs
make it possible to optimize solution
size, power and cost. The pin- and software-compatible families of 6-channel,
2-channel and single-channel ADCs
offer flexibility to upgrade from 12bit resolution to 14-bit resolution.
Whatever your motor control, power
monitoring, or data acquisition system
requires, Linear Technology has a fast
SAR ADC to do the job. L
Linear Technology Magazine • June 2007
DESIGN IDEAS L
0.25in2 × 1.8mm Dual Output
Converter for Li-Ion to 3.3V and 1.8V
by John Canfield
Introduction
One quarter inch square. That is all
the area needed for a complete Li-Ion
to dual output, buck and buck-boost
converter. Figure 1 shows a compact
dual output converter made possible
by the LTC3522—a complete, high
efficiency, dual rail power supply solution in a 3mm × 3mm QFN. As shown,
only a few external components are
required, and they can all be low profile
(≤1mm)—perfect for the demanding
space requirements of even the most
compact portable electronic devices.
The LTC3522 combines a monolithic
buck-boost converter and synchronous buck converter in a single, low
profile 0.75m × 3mm × 3mm 16-lead
QFN. Soft-start and feedback loop
compensation circuitry is included in
the IC. An entire application circuit
for a dual converter requires only
the IC, inductors, bypass capacitors
and feedback resistor dividers. Both
converters maintain a low transient
voltage deviation under full load
step, even with small ceramic output
capacitors. These features result in a
simple application circuit as shown in
Figure 2 and a total PCB area of less
than 0.25 square inches as illustrated
by Figure 1. The LTC3522 features
a fixed internal switching frequency
of 1.1MHz that allows for the use of
low profile capacitors and inductors,
resulting in a total application height
of only 1mm.
While requiring only a single inductor, the LTC3522 is capable of high
efficiency fixed frequency operation
with input voltages that are above,
below, or equal to the output voltage.
The buck-boost converter utilizes a
proprietary switching algorithm to
provide seamless transitions between
buck and boost functional modes while
simultaneously maximizing conversion efficiency. The buck-boost output
The LTC3522 combines
a monolithic buck-boost
converter and synchronous
buck converter in a
single, low profile
0.75m × 3mm × 3mm
16-lead QFN.
Circuitry for soft-start and
feedback loop compensation
is integrated into the
IC. An entire application
circuit for a dual converter
requires only the LTC3522
and a minimal number
of external components.
Figure 1. Buck-boost and buck converter
occupy less than 0.25in2 of board space
voltage can be set as low as 2.2V or
as high as 5.25V. With a 3.3V output,
the buck-boost converter is able to
supply a 300mA load current over the
full 2.4V to 5.5V input voltage range.
When powered by a standard Li-Ion
battery with a minimum voltage of 3V,
a 400mA load can be supported.
The LTC3522 buck converter features internally compensated current
mode control that ensures a rapid
transient response over a wide range
of output capacitor values. The buck
converter can supply a load current
of up to 200mA over the entire input
voltage range and its output voltage
can be set as low as 0.6V. The buck
converter transitions smoothly to
100% duty cycle operation to extend
battery life in low dropout operation.
Despite its tiny size, the LTC3522
boasts an efficiency of up to 95% for
100
VOUT2
1.8V
200mA
+
4.7µF
8.2µH
PVIN1 PVIN2
SW2
6.8µF
137k
OFF
BURST
ON
4.7µH
SW1A
SW1B
12pF
FB2
68.1k
BUCK-BOOST
BUCK
80
LTC3522
VOUT1
1M
SHDN2
SHDN1
FB1
PGOOD2
PWM
PWM
432k
PGOOD1
PGND1 GND PGND2
VOUT1
3.3V
300mA
(400mA
4.7µF
VIN > 3V)
EFFICIENCY (%)
Li-Ion
2.4V TO 4.2V
BUCK,
Burst Mode
OPERATION
90
BUCK-BOOST,
Burst Mode
OPERATION
70
60
50
40
30
20
1
100
10
LOAD CURRENT (mA)
1000
BUCK-BOOST L = COILCRAFT MSS6132 – 4.7µH
BUCK L = COILCRAFT MSS6132 – 8.2µH
Figure 2. Li-Ion to 3.3V at 300mA and 1.8V at 200mA
Linear Technology Magazine • June 2007
Figure 3. Efficiency vs load current
39
L DESIGN IDEAS
BUCK VOUT
100mV/DIV
BUCK-BOOST VOUT
100mV/DIV
100µs/DIV
Figure 4. Alternating load step responses
each converter and incorporates a variety of useful features. Both converters
include an internal, closed-loop soft
start to ensure a reliable output voltage
rise time, independent of loading and
output capacitor value. In addition,
each converter includes its own opendrain power-good indicator, which
allows for undervoltage fault detection
and sequenced start-up. Each converter can be independently enabled.
With both converters disabled, the
total supply current is reduced to
under 1µA.
Efficiency
Figure 3 shows the efficiency of each
converter for the circuit of Figure 2.
The buck-boost converter reaches
a peak efficiency of 95%, while the
buck converter peaks at 94%. In PWM
mode, both converters are greater
than 90% efficient at all load currents
above 30mA.
Pin selectable Burst Mode® operation improves efficiency at light load
currents. In Burst Mode, the total quiLi-Ion
2.4V TO 4.2V
VOUT2
1.8V
200mA
+
Supply Sequencing
Many dual supply applications require
that the supply rails power up in a
particular order. A common example
is a microprocessor in which the core
supply voltage must be up and in
regulation before the peripheral supply powering the output pin drivers is
enabled. This ensures that the core
logic is functioning before the outputs
become active, thereby preventing
erratic output fluctuations during
power-up.
The LTC3522 has an independent
power-good output for each converter.
This allows the two output voltages to
C3
4.7µF
L1
8.2µH
499k
escent current is reduced to only 25µA
with both converters enabled. In noise
sensitive applications, both converters
can be forced into low noise, fixed frequency PWM operation by connecting
the PWM pin to VIN. Alternatively, the
PWM pin can be driven dynamically
in the application to provide low noise
performance during critical phases of
operation.
C1
6.8µF
SW2
12pF
L2
4.7µH
PVIN1 PVIN2
137k
FB2
SW1A
SW1B
LTC3522
VOUT1
1M
68.1k
VOUT1
3V
300mA
C2
4.7µF (400mA, VIN > 3V)
FB1
be sequenced in either order without
requiring any additional external
components. Figure 5 shows a sequenced LTC3522 application circuit
that waits for the 1.8V buck output
rail to reach regulation before enabling
the buck-boost converter to power the
3.0V output rail. This is accomplished
by simply connecting the SHDN1
pin to the buck power-good output,
PGOOD2. With the external enable
signal held low, both converters are
disabled. When the external enable
is brought high, the buck converter
is immediately enabled. The buckboost converter remains disabled
until PGOOD2 goes high, indicating
that the buck converter has reached
regulation.
Inter-Channel Performance
While in PWM mode, both converters
operate synchronously from a common
1.1Mhz oscillator. This minimizes the
interaction between the two converters so that load steps on the output
of one converter have little impact on
the opposite output. For example,
Figure 4 shows both output voltages
as a 20mA to 200mA load step is applied to the buck channel and a 0mA
to 300mA load step is applied to the
buck-boost channel. In this case, even
with small 4.7µF output capacitors
on each converter, the interaction
between channels is minimal.
Conclusion
The LTC3522 provides a complete,
sequenced dual rail power supply
solution in a compact footprint. Its
high efficiency and exceptional performance make the LTC3522 well suited
for even the most demanding portable
applications. L
VOUT2
1V/DIV
499k
PGOOD1
PGOOD1
PWM
BURST MODE OPERATION
C1: TDK C3216X5R0J685M
C2, C3: TAIYO YUDEN JMK212BJ106MG
L1: COOPER BUSSMANN SD18-8R2
L2: COOPER BUSSMANN SD18-4R7
SHDN2
PWM
PGOOD2
SHDN1
PGND1 GND1 PGND2
Figure 5. Sequenced power-up application
40
ON
OFF
499k
VOUT1
2V/DIV
PGOOD2
5V/DIV
PGOOD1
5V/DIV
200µs/DIV
Figure 6. Sequenced power-up waveforms
Linear Technology Magazine • June 2007
DESIGN IDEAS L
Sub-µA RMS Current Measurement
by Jim Williams
for Quartz Crystals
Quartz crystal RMS operating current is critical to long-term stability,
temperature coefficient and reliability.
Accurate determination of RMS crystal current, especially in micropower
types, is complicated by the necessity
to minimize introduced parasitics,
particulary capacitance, which corrupt
crystal operation.
Figure 1’s high gain, low noise
amplifier combines with a commercially available closed core current
probe to permit the measurement.
An RMS-to-DC converter supplies
the RMS value. The quartz crystal
test circuit shown in dashed lines
exemplifies a typical measurement
situation. The Tektronix CT-1 current
probe monitors crystal current while
introducing minimal parasitic loading.
The probe’s 50Ω terminated output
is fed to A1. A1 and A2 take a closed
loop gain of 1120; excess gain over a
nominal gain of 1000 corrects for the
CT-1’s 12% low frequency gain error
at 32.768kHz.1 A3 and A4 contribute a
gain of 200, resulting in total amplifier
gain of 224,000. This figure results in
a 1V/µA scale factor at A4 referred to
the gain corrected CT-1’s output. A4’s
LTC1563-2 bandpass filtered output
feeds an LTC1968-based RMS-to-DC
converter (A5), which provides the
circuit’s output. The signal processing
path constitutes an extremely narrow
band amplifier tuned to the crystal’s
A = 224,000
CRYSTAL OSCILLATOR TEST CIRCUIT
TEKTRONIX CT-1
CURRENT PROBE
5mV/mA (5µV/µA)
A = 1120 (CT-1 GAIN ERROR AT 32.768kHz ≈ 12%, SEE TEXT)
EPSON C-100R
f = 32.768kHz
2V
+
2V
2M
A1
LT1028
+
–
C1
LTC1440
1M
+
39pF
1.5k*
1µF
A2
LT1222
–
–
+
1740Ω*
1k
A3
LT1222
–
+
825Ω*
–5V
39Ω
(SEE TEXT)
49.9Ω*
10pF
A = 200
5V
–
A4
LT1222
825Ω*
1.2M
61.9Ω*
49.9Ω*
63.4Ω*
10µF
32.7kHz BANDPASS FILTER
5V
43k
RMS-TO-DC CONVERTER
V+
0.01µF
I1
LPB
21k*
INVB
LPA
84.5k*
5V
10k
V+
LTC1968
E
G
OUT
5.62k*
24.9k*
+
A5
LT1077
OUT
0V–1V = 0µA–1µA
R
10µF
1µF
10k
SB
SA
V–
I2
42.2k*
INVA
20k*
–
5V
LTC1563-2
LP
0.1µF
GND EN
*1% METAL FILM RESISTOR
10µF, 1µF CAPACITORS = WIMA MKS-2
–5V
Figure 1. Op amps A1–A4 furnish gain of > 200,000, permitting sub-µA crystal current measurement. The LTC1563-2 bandpass filter
smooths residual noise while providing unity gain at 32.768kHz. The LTC1968 RMS-to-DC converter supplies RMS calibrated output.
Linear Technology Magazine • June 2007
41
L NEW DEVICE CAMEOS
A
2V/DIV
B
1µA/DIV
C
1µA/DIV
10µs/DIV
Figure 2. The 32.768kHz output of the crystal oscillator (Trace A) and crystal current monitored
at A4 output (Trace B) and the RMS-to-DC converter input (Trace C). Peaks in Trace B’s unfiltered
waveform derive from inherent and parasitic paths shunting the crystal.
frequency. Figure 2 shows typical circuit waveforms. Crystal drive, taken at
C1’s output (trace A), causes a 530nA
RMS crystal current, which is represented at A4’s output (Trace B) and the
RMS-to-DC converter input (Trace C).
Peaking visible in Trace B’s unfiltered
presentation derive from inherent and
parasitic paths shunting crystal.
Typical circuit accuracy is 5%.
Uncertainty terms include the transformer’s tolerances, its approximately
1.5pF loading and resistor/RMS-toDC converter error. Calibrating the
New Device Cameos
High Voltage Dual Input
Li-Ion Battery Charger
The LTC4075HVX is a standalone
linear charger that is capable of
charging a single-cell Li-Ion/Polymer
battery from both wall adapter and
USB inputs. The charger can detect
power at the inputs and automatically
select the appropriate power source
for charging.
No external sense resistor or blocking diode is required for charging due to
the internal MOSFET architecture. The
LTC4075HVX features a maximum
22V rating for both wall adapter and
USB inputs although charging stops if
the selected power source exceeds the
overvoltage limit (typical 6V). Internal
thermal feedback regulates the battery
charge current to maintain a constant
die temperature during high power
operation or high ambient temperature
conditions. The float voltage is fixed
at 4.2V and the charge current is programmed with an external resistor. The
LTC4075HVX terminates the charge
cycle when the charge current drops
below the programmed termination
threshold after the final float voltage
is reached.
Other features include automatic
recharge, undervoltage lockout,
42
charge status outputs, and “power
present” status outputs to indicate the
presence of wall adapter or USB power.
No trickle charge allows full current
from the charger when a load is connected directly to the battery.
Small 1.8A Step-Down
Regulator Switches at
4MHz for Space-Sensitive
Applications
The LTC3568 is a 10-lead DFN, synchronous, step-down, current mode,
DC/DC converter, intended for medium power applications. It operates
within a 2.5V to 5.5V input voltage
range and switches at up to 4MHz,
making it possible to use tiny capacitors and inductors that are under 1mm
in height. The output of the LTC3568
is adjustable from 0.8V to 5V, and its
0.11Ω switches allows up to 1.8A of
output current at high efficiency. By
using the LTC3568 in a small 3mm ×
3mm, 10-lead DFN package, a complete DC/DC converter can consume
less than 0.3 square inches of board
real estate.
Efficiency is extremely important in
battery-powered applications, and the
LTC3568 keeps efficiency high with an
automatic, power saving Burst Mode
circuit reduces error to less than
1%. Calibration involves driving the
transformer with 1µA at 32.7kHz. This
is facilitated by biasing a 100k, 0.1%
resistor with an oscillator set at 0.1V
output. The output voltage should be
verified with an RMS voltmeter having appropriate accuracy. Figure 1 is
calibrated by padding A2’s gain with
a small resistive correction, typically
39Ω.L
Notes
1The validity of this gain error correction at one
sinusoidal frequency—32.768kHz—was investigated with a 7-sample group of Tektronix CT-1s.
Device outputs were collectively within 0.5% of
12% down for a 1.00µA, 32.768kHz sinusoidal
input current. Although this tends to support
the measurement scheme, it is worth noting that
these results are as measured. Tektronix does not
guarantee performance below the specified –3dB,
25kHz low frequency roll-off.
operation, which reduces gate charge
losses at low load currents. With no
load, the part only draws 60µA, and in
shutdown, the device draws less than
1µA, making it ideal for low current
applications.
The LTC3568 uses a current-mode,
constant frequency architecture that
benefits noise sensitive applications.
Burst Mode operation is an efficient
solution for low current applications,
but sometimes noise suppression is
a priority. To reduce noise problems,
a pulse-skipping mode and a forced
continuous mode are available, which
decreases the ripple noise at low
currents. Although not as efficient
as Burst Mode operation at low currents, pulse-skipping mode and forced
continuous mode still provide high efficiency for moderate loads. In dropout,
the internal P-channel MOSFET switch
is turned on continuously, thereby
maximizing the usable battery life.
A Power Good output is available for
power supply monitoring or for Power
On Reset use. Internal overvoltage
and undervoltage comparators pull
the open-drain PGOOD output low
if the output voltages are not within
about ±7.5%.
The LTC3568’s small size, high
efficiency, low component count and
flexibility make it an ideal DC/DC
converter for portable devices. L
Linear Technology Magazine • June 2007
DESIGN TOOLS L
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Linear Technology Magazine • June 2007
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Linear Technology Magazine • June 2007