V20N4 - JANUARY

January 2011
I N
T H I S
TimerBlox: Function-Specific ICs
Quickly and Reliably Solve
Timing Problems
I S S U E
solar battery charger tracks
panel maximum power 10
I2C system monitor
combines temperature,
voltage and current
Andy Crofts
measurements 22
Your design is nearly complete, but a nagging timing requirement
has suddenly cropped up. It might call for a variable frequency
oscillator, a low frequency timer, a pulse-width modulator, a
controlled one-shot pulse generator, or an accurate delay.
Regardless of the requirement, you need a quick, reliable, stable
solution—there is no time to develop code for a microcontroller. You
could build something out of discrete components and a comparator
or two, or maybe the good old 555 timer could do the job, but will
the accuracy be there? Will it take up too much room on the board?
What about time to test and specify the bench-built timer?
isolated data transmission
and power conversion
combo in surface mount
package 30
isolated power supplies
made easy 38
nanopower buck converter
for energy harvesting apps
41
POWER
SUPPLY
Volume 20 Number 4
CONNECTOR
DROPS
WIRING DROPS
CONNECTOR
DROPS
WIRING DROPS
CONNECTOR
DROPS
LOAD
CONNECTOR
DROPS
Figure 1. The simplest model for load regulation over
resistive interconnections.
There is a better way. Linear Technology’s TimerBlox® family of silicon timing devices solves specific timing problems with
minimal effort. TimerBlox devices easily drop into designs with a
fraction of the design effort or space requirements that a microcontroller or discrete-component solution would demand. It only
takes a few resistors to nail down the frequency or time duration you require. That’s it, no coding or testing required. Complete
solutions are tiny, composed of a 2mm × 3mm DFN, or a popular
6-lead SOT-23, plus a couple of resistors and decoupling cap.
A TOOLBOX OF TIMERBLOX DEVICES
All TimerBlox devices use Linear’s silicon oscillator technology,
featuring low component count, vibration-immunity, fast startup, and ease-of-use. Each TimerBlox device is purpose-built to
solve a specific timing problem (see Table 1), so the performance
TimerBlox devices solve timing problems
w w w. li n ea r.com
(continued on page 2)
…continued from the cover
In this issue...
COVER STORY
TimerBlox: Function-Specific ICs Quickly
and Reliably Solve Timing Problems
Andy Crofts
1
(LTC699x, continued from page 1)
DESIGN FEATURES
of each device is specified for its intended application, eliminating the
guesswork involved with configuring and applying do-it-all timers.
Battery Charger’s Unique Input
Regulation Loop Simplifies Solar Panel
Maximum Power Point Tracking
Jay Celani
10
Two High Power Monolithic Switching
Regulators Include Integrated 6A, 42V or
3.3A, 42V Power Switches, Built-in Fault
Protection and Operation up to 2.5MHz
Matthew Topp and Joshua Moore
16
I2C System Monitor Combines Temperature,
Voltage and Current Measurements
for Single-IC System Monitoring
David Schneider
22
Isolated Data Transmission and Power Conversion
Integrated Into a Surface Mount Package
Keith Bennett
30
Isolated Power Supplies Made Easy
John D. Morris
It only takes a few resistors to nail down the frequency
or time duration you require. That’s it, no coding or
testing required. Complete solutions are tiny, composed
of a 2mm × 3mm DFN, or a popular 6-lead SOT-23,
plus a couple of resistors and decoupling cap.
38
DESIGN IDEAS
Nanopower Buck Converter Runs on
720nA, Easily Fits into Energy Harvesting
and Other Low Power Applications
Michael Whitaker
41
product briefs
42
back page circuits
44
Because each TimerBlox device is designed to perform a specific timing function, the most significant design decision is choosing the proper part number.
To further simplify design, five of the six package pins in all TimerBlox devices
share the same name and function—with the remaining pin unique to the
device function. Figure 1 details the function of each pin (SOT-23 shown).
Each Timerblox device offers eight different timing ranges and two modes
of operation (which vary for each device). The operational state is represented by a 4-bit DIVCODE value, which is set by the voltage on the
DIV pin. For the ultimate in simplicity, a resistor divider can be used to
set the DIVCODE. For example, Figure 2 shows how changing the voltage at the DIV pin sets the functionality of the LTC6992 by selecting a
DIVCODE from 0–15. The MSB of DIVCODE is a “mode” bit, in this case selecting the output polarity. The remaining bits choose the frequency range.
Once the proper DIVCODE has been determined, the frequency or timing duration is fine-tuned by a simple calculation for RSET. The set
resistor establishes the frequency of an internal silicon oscillator master clock. The resulting circuit has guaranteed accuracy over the full
2.25V–5.5V supply range and –40°C to 125°C temperature range.
(continued on page 4)
Figure 1. All TimerBlox devices share common pin functions
DEVICE-SPECIFIC FUNCTION PIN
LTC6990: Output Enable
LTC6991: Reset
LTC6992: Modulation Control
LTC6993: Trigger
LTC6994: Logic Input
Scan This
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2 | January 2011 : LT Journal of Analog Innovation
OUTPUT
Sources and Sinks 20mA
SUPPLY VOLTAGE
2.25V TO 5.5V
OUT
GND
RVCO (OPTIONAL)
CONTROL VOLTAGE
Modulates Output Frequency
V+
LTC699x
CONTROL RESISTOR
Allows for VCO Operation
SET
V+
C1
0.1µF
R1
DIV
R2
RSET
SET RESISTOR
Only a Single Resistor is Required
to Set the Master Frequency
RESISTOR DIVIDER SETS VDIV
Voltage at DIV (Divider) Pin Selects
One of 16 States, which Sets the
Output Frequency Range
and Mode Bit
Linear in the news
Linear in the News
APEC SHOW
Linear will have a booth at APEC, the
Applied Power Electronics Conference
and Exposition, held in Fort Worth, Texas,
March 6-10. At the booth (151 and 152),
Linear will showcase a broad range of
power management solutions, including:
•Energy harvesting solutions
•Power µModule® regulators
•Digital power products
•LED drivers
•Linear regulators
•Switching regulators
The booth will be staffed by Linear’s
power experts and technical field
staff. Info at www.apec-conf.org.
AUTOMOTIVE SHOWS
Linear Technology is rapidly growing its
commitment to the automotive electronics market. This parallels the increase in
the electronics content in cars, as well as
new innovations in hybrid and electric
vehicles. Linear’s products for automotive now cover most electronics systems
including navigation and entertainment,
safety systems, security, electronic steering and braking, LED lighting, engine
control and battery management systems for hybrid/electric vehicles.
Linear plans to participate with
booths at the following automotive events in the coming months:
International Automotive Electronics Technology
Expo, Tokyo Bigsight Convention Center, Japan,
January 19–21: At this show, Linear will
showcase its LTC6802 family of battery management ICs for hybrid/electric
vehicles, as well as H-grade power management ICs for automotive applications and power µModule regulators.
Information at www.car-ele.jp/carele/en.
Advanced Automotive Batteries Conference,
Pasadena Convention Center, Pasadena, California,
January 24–28: Linear will showcase the
company’s innovative battery management IC family. including the LTC6802, a
highly integrated multicell battery monitoring IC capable of precisely measuring
the voltages of up to 12 series-connected
battery cells. Using a novel stacking
technique, multiple LTC6802s can be
placed in series without opto-couplers
or isolators. Stacked LTC6802s enable
precision measurement of all battery
cell voltages, independent of battery
string size, within 13ms. Linear will also
show the companion LTC6801 independent multicell battery stack monitor
and other devices in the family. Info at
www.advancedautobat.com/automotivebattery-conference-2011/index.html.
SAE 2011 Hybrid Vehicle Technology Symposium,
Hilton Anaheim, Anaheim, California, February 9–11:
At this automotive-focused conference,
Linear will showcase its growing family of battery stack monitors for hybrid/
electric vehicles. This conference will
focus on technology advances and
platform strategies for hybrid/electric
vehicles, plug-in hybrid electric vehicles
and all-electric vehicles. Info at www.sae.
org/events/training/symposia/hybrid.
EVENTS IN CHINA
IIC China Conference & Exhibition, Shenzhen
Convention & Exhibition Center, Shenzhen, China,
February 24–26: IIC China is attended by
design engineers and technical managers in China. It is China’s largest
showcase of IC application technologies
and high-end components. At Linear’s
booth (2C15), visitors will gain an
overview of the company’s products
across a range of diverse applications.
Info at www.english.iic-china.com.
Electronica & Productronica China 2011, Shanghai
New International Expo Center, Shanghai, China,
March 15–17: This is the 10th anniver-
sary of Electronica & Productronica.
The show focuses on the latest technology breakthroughs in growth-oriented
markets, including telecommunications,
industrial, automotive and IT products, and consumer electronics. Linear
will be in Hall E2, booth 2466. More
info at e-p-china.com/en/home.
At both IIC China and Electronica
& Productronica, Linear will
showcase the following:
•Automotive electronic solutions
•Battery management systems
for hybrid/electric vehicles
•Wireless communications solutions
•Industrial & medical solutions
•Instrumentation
•Energy harvesting/ Nanopower solutions
•High power LED drivers
•High voltage step-down
(buck) regulators
•TimerBlox product family
•Power µModule regulators n
January 2011 : LT Journal of Analog Innovation | 3
The LTC6990 can easily be used as a voltage-controlled
frequency modulator. Although this technique can be
used with other silicon oscillators, they typically are
limited in accuracy and suffer from poor supply rejection.
The LTC6990 does not have these limitations.
Figure 2. LTC6992
Frequency Range and
“POL” Bit vs DIVCODE
(LTC699x, continued from page 2)
While it can be used as a fixed-frequency
oscillator, the LTC6990 can easily be
applied as a frequency modulator. A
second SET-pin resistor, RVCO, allows a
control voltage to vary ISET and change the
VCTRL
0V to 3.3V
9
6
8
7
0V
•VSET is GND-referenced, allowing
for a GND-referenced control voltage that is easy to work with.
¼V+
½V+
¾V+
V+
C1
0.1µF
R1
976k
DIV
R2
102k
Figure 3 shows the LTC6990 configured as
a VCO that translates a 0V to 3.3V control
kHz
V
DIVCODE = 1
(NDIV = 2, Hi-Z = 0)
OE
2V/DIV
VCTRL
2V/DIV
OUT
2V/DIV
20µs/DIV
4 | January 2011 : LT Journal of Analog Innovation
V+
•All TimerBlox devices allow for a
wide 16:1 timing range within each
NDIV setting, but only the LTC6990 uses
a small 2× step through divider settings. That allows for maximum overlap between ranges to accommodate
any 8:1 range of VCO frequencies (or
16:1 with a reduced-accuracy extended
range). And since each TimerBlox
device has eight different timing
ranges, the LTC6990 still maintains a
large 4096:1 total frequency range.
Figure 4. Performance of the voltage-controlled
oscillator shown in Figure 3
f OUT = 400kHz − VCTRL • 109
LTC6990
V+
10
5
•VSET (the SET pin voltage) is regulated
to 1V and is accurate to ±30mV over
all conditions. This allows RVCO to
establish an accurate VCO gain.
OUT
SET
11
4
INCREASING VDIV
40kHz TO 400kHz
RSET
86.6k
12
1
Figure 3. LTC6990 voltage-controlled oscillator
RVCO
232k
13
3
output frequency. Although this technique
can be used with other silicon oscillators,
they typically are limited in accuracy and
suffer from poor supply rejection. The
LTC6990 does not have these limitations
because of three important enhancements:
where NDIV = 1, 2, 4, …, 128
GND
2
0.01
1MHz • 50k ISET
•
NDIV
VSET
OE
14
0.1
The LTC6990 is a resistor-programmable
oscillator featuring 1.5% accuracy and
an output enable function to force the
output low or into a high-impedance
state. The output frequency is determined by the NDIV frequency divider
and RSET (which replaces VSET/ISET):
OUTPUT
ENABLE
15
1
10
0.001
DIVCODE MSB (“POL” BIT) = 1
0
100
VOLTAGE-CONTROLLED
OSCILLATOR CAN BE USED FOR
FIXED FREQUENCY OR FREQUENCY
MODULATION
fOUT =
DIVCODE MSB (“POL” BIT) = 0
1000
fOUT (kHz)
For an even easier design process,
download “The TimerBlox Designer”
from www.linear.com/timerblox—
a free Excel-based tool that generates component values, schematics,
and timing diagrams automatically.
cover story
PWM
CONTROL
TRIGGER
INPUT
LTC6991
LTC6992
OUTPUT
RESET
LTC6993
VCTRL
LTC6994
OUTPUT
ENABLE
LTC6990
DEVICE
FUNCTION
OPTIONS
RANGE
Voltage-Controlled
Silicon Oscillator
Configurable frequency gain and voltage range
488Hz to 2MHz
Low Frequency Oscillator
Period range from 1ms to 9.5 hours
29µHz to 977Hz
LTC6992-1
0%–100% Duty Cycle
LTC6992-2
5%–95% Duty Cycle
LTC6992-3
0%–95% Duty Cycle
LTC6992-4
5%–100% Duty Cycle
LTC6993-1
Rising-Edge Triggered
LTC6993-2
Rising-Edge Re-Triggerable
LTC6993-3
Falling-Edge Triggered
LTC6993-4
Falling-Edge Re-Triggerable
LTC6994-1
1-Edge Delay
Voltage-Controlled PWM
3.8Hz to 1MHz
One-Shot
1µs to 34 sec
Delay
1µs to 34 sec
LTC6994-2
2-Edge Delay
Table 1. TimerBlox family members
voltage into a 40kHz to 400kHz frequency. Due to the LTC6990’s high
modulation bandwidth, the output
responds quickly to control voltage
changes, as can be seen in Figure 4.
LOW FREQUENCY SOLUTIONS
The LTC6991 picks up in frequency where
the LTC6990 leaves off, with an enormous 29µ Hz to 977Hz range (a period
range of 1ms to 9.5 hours). It incorporates a fixed 10-stage frequency divider
and a programmable 21-stage divider.
Since the applications for frequency
modulation are rare at such low frequencies, the emphasis for this part is
on covering as wide a range as possible.
Therefore, the LTC6991 uses large 8×
steps between NDIV settings. The trade-off
is a smaller 2× overlap between ranges.
The output interval relationship is:
t OUT =
NDIV • RSET
• 1.024ms
50k
where NDIV = 1, 8, 64, …, 221
The LTC6991 is designed to handle long
duration timing events. In place of an
output enable, it includes a similar reset
function. The RST pin can truncate the
output pulse or prevent the output from
oscillating at all, but it has no effect
on the timing of the next rising edge.
This function allows the LTC6991 to
initiate an event with a variable duration, perhaps controlled by another
circuit. Otherwise, if RST is inactive, the
LTC6991 produces a square wave.
Figure 5 shows how a simple camera
intervalometer can be constructed from
the LTC6991 and a handful of discrete
January 2011 : LT Journal of Analog Innovation | 5
Figure 5. An LTC6991-based camera intervalometer
3 SEC PULSE WIDTH
RPW
100k
RST
CPW
33µF
LTC6991
GND
RS3
95.3k
8 SEC TO
64 SEC
RS1
1M
OUT
V+
+
V
R1B
1M
R1A
332k
SET
VARIABLE INTERVAL
(8 SEC TO 8.5 MIN)
SHUTTER
OUT
DIV
1µF
RST
“SLOW RANGE”
1.1 MIN TO 8.5 MIN
RS2
2M
R2
130k
Figure 6. An upgraded camera intervalometer—
the LTC6994-1 is added to allow shutter speed
adjustment
IN
RST
OUT
GND
47.5k
RSHUTTER
1M
V
+
1M
SET
DIV
3M
RSHUTTER: SHUTTER SPEED ADJUSTMENT
0Ω FOR 0.25 SEC
1M FOR 4 SEC
components. An intervalometer is used
in time-lapse photography to capture
images at specific intervals. The shutter
rate might range from a few seconds
to a few hours. In this example, the
photographer can choose any interval
between 8 seconds and 8.5 minutes.
An RC delay from OUT to RST allows for
a 3-second shutter pulse before resetting the output. Potentiometer RS1 varies the total resistance at the SET pin
from 95.3k to 762k to adjust the period
100
DIVCODE = 0
90 3 PARTS
LTC6992-1/
LTC6992-4
DUTY CYCLE (%)
80
LTC6992-2/
LTC6992-3
70
60
50
40
LTC6992-2/
LTC6992-4
30
20
10
0
LTC6992-1/LTC6992-3
0
0.2
0.4
0.6
VMOD/VSET (V/V)
0.8
6 | January 2011 : LT Journal of Analog Innovation
681k
RINTERVAL
1M
1
95.3k
2.25V TO 5V
GND
V+
SET
DIV
332k
from 8 seconds to 64 seconds, with
DIVCODE set to 4 by R1A and R2. Closing
the SLOW RANGE switch changes the
DIVCODE to 5, increasing NDIV by 8× to
extend the interval up to 8.5 minutes.
Figure 6 shows how easy it is to add
timing functions on top of each other
using TimerBlox devices. Here the
LTC6994-1 is added to the intervalometer in Figure 5 to create an intervalometer with shutter-speed adjustment.
PULSE-WIDTH MODULATOR
The MOD pin accepts a control voltage with a range of 0.1V to 0.9V that
linearly regulates the output duty cycle.
The 0.1V “pedestal” ensures that
an op-amp or other input driver is
1M
1µF
“SLOW INTERVAL RANGE”
1 MIN TO 8 MIN
2M
130k
RINTERVAL: INTERVAL ADJUSTMENT
0Ω FOR 8 SEC
1M FOR 64 SEC
The LTC6992 TimerBlox oscillator features pulse-width modulation—the
ability to control output duty cycle with
a simple input voltage. The LTC6992
makes quick work of a technique that is
useful for many applications: light dimming, isolated proportional control, and
efficient load control, to name a few.
Figure 7. Measured transfer
function of the LTC6992 family
SHUTTER
OUT
LTC6991
LTC6994-1
able to reach the bottom of the control range. The duty cycle is given by:
DutyCycle =
VMOD
1 V
− 100mV
− ≈ MOD
0.8 • VSET 8
800mV
The output frequency is governed by the
simple relationship shown below. The
total frequency range of the LTC6992
covers 3.8Hz to 1MHz, using 4× divider
steps in the eight NDIV settings.
fOUT =
1MHz • 50k
NDIV • RSET
where NDIV = 1, 4, 16, …, 16384
The LTC6992-1 allows for the full
duty cycle range, covering 0% (for
VMOD ≤ 0.1V) to 100% (for VMOD ≥ 0.9V).
At the extremes, the output stops oscillating, resting at GND (0% duty) or V+
(100% duty). Some applications (such
as coupling a control signal across an
isolation transformer) require continuous
oscillation. For such applications, choose
the LTC6992-2, which limits the output
duty cycle to 5% min and 95% max.
The LTC6992-3 and LTC6992-4 complete
cover story
The LTC6992 makes quick work of producing a
voltage-controlled PWM signal—useful for many
applications: light dimming, isolated proportional
control and efficient load control, to name a few.
the family by limiting the duty cycle at
only one extreme. Figure 7mV shows the
measured response for the LTC6992 family.
Figure 8 shows a typical circuit. With the
frequency divider (NDIV) set to 1 and
RSET = 200k, this PWM circuit is configured
for a 250kHz output frequency. Figure 9
demonstrates the circuit in action for
both the LTC6992-1 and the LTC6992-2.
The high modulation bandwidth allows
the output duty cycle to quickly track
changes in the modulation voltage.
VMOD
0.5V/DIV
250kHz
VMOD
0.1V TO 0.9V
MOD
OUT
LTC6992
LTC6992-1 OUT
1V/DIV
2.25V TO 5.5V
GND
V+
SET
DIV
RSET
200k
C1
0.1µF
TIE DIV TO GND
FOR DIVCODE = 0
LTC6992-2 OUT
1V/DIV
10µs/DIV
Figure 8. An LTC6992 pulse-width modulator
Figure 9. Performance of the PWM shown in Figure 7
ONE-SHOT EVENTS
Of course, not all timing applications
require a stable frequency oscillator
output. Some circuits require an eventtriggered fixed-duration pulse, like that
produced by the LTC6993 monostable
(one-shot) pulse generator, which offers
eight different logic functions and a huge
1µs to 34-second timing range. The oneshot duration tOUT is established by RSET:
NDIV • RSET
• 1µs
50k
where NDIV = 1, 8, 64, …, 221
t OUT =
The LTC6993 is triggered by a rising or
falling transition on its TRIG pin, which
initiates an output pulse with pulse width
tOUT. Some variations include the ability to
“retrigger” the pulse, extending the output
pulse duration with additional trigger
signals. And each version can be configured to produce logic high or low output
pulses using the MSB of the DIVCODE.
Table 2 summarizes the different options.
Figure 10 shows a basic circuit, with the
DIVCODE set to 3 (NDIV = 512, POL = 0) by a
resistor divider and a 97.6k RSET defining a
TRIG
2V/DIV
1ms
1ms
TRIG
OUT
LTC6993
2.25V TO 5.5V
LTC6993-1 OUT
2V/DIV
1ms
V+
GND
0.1µF
SET
R1
1M
DIVCODE = 3
DIV
RSET
97.6k
R2
280k
LTC6993-2 OUT
2V/DIV
200µs/DIV
Figure 10. An LTC6993 monostable pulse generator
(one-shot)
Figure 11. The LTC6993 non-retriggerable and
retriggerable functionality
IN
2V/DIV
IN
OUT
LTC6994
GND
100µs
2.25V TO 5.5V
V+
0.1µF
SET
RSET
619k
DIV
LTC6994-1 OUT
2V/DIV
R1
976k
DIVCODE = 1
R2
102k
LTC6994-2 OUT
2V/DIV
tDELAY = 100µs
Figure 12. LTC6994 delay interval generator
100µs/DIV
Figure 13. LTC6994 single and double-edge delay
functionality
January 2011 : LT Journal of Analog Innovation | 7
The LTC6993 is triggered by a rising or falling transition
on its TRIG pin, which initiates an output pulse
with pulse width tOUT. Some variations include the
ability to “retrigger” the pulse, extending the output
pulse duration with additional trigger signals.
1ms
output pulse width. To demonstrate
the difference between retriggerable and
non-retriggerable functionality, Figure 11
shows the results of using either the
LTC6993-1 or LTC6993-2 in this circuit.
THE LTC6994 FOR PROGRAMMABLE
DELAY AND PULSE QUALIFICATION
The LTC6994 is a programmable delay
or pulse qualifier. It can perform noise
filtering, which distinguishes its function
from a delay line. The LTC6994 is available in two versions, as detailed in Table 3.
The LTC6994-1 delays the rising or falling
edge of the input signal. The LTC6994-2
delays any input transition, rising or falling, and can invert the output signal.
The LTC6994’s programmable delay
(denoted as tDELAY below) can vary
from 1µs to 34 seconds, accurate
to ±3% in most conditions.
tDELAY =
Table 2. LTC6993 options
Table 3. LTC6994 options
DEVICE
INPUT POLARITY
RE-TRIGGER
LTC6993-1
Rising-Edge
No
LTC6993-2
Rising-Edge
Yes
LTC6993-3
Falling-Edge
No
LTC6993-4
Falling-Edge
Yes
delays either the rising or falling transition,
and the LTC6994-2, which delays transitions in both directions. Both versions will
reject narrow pulses, but the LTC6994-2
preserves the original signal’s pulse width.
In addition to this type of noise filtering,
the LTC6994 is useful for delay matching, generating multiple clock phases,
or doubling the clock frequency of the
input signal, as shown in Figure 14.
NDIV • RSET
• 1µs
50kΩ
where NDIV = 1, 8, 64, …, 221
The output will only respond to input
changes that persist longer than the delay
period. This operation is well suited for
pulse qualification, switch debouncing, or
guaranteeing minimum pulse widths. The
basic circuit in Figure 12 is configured for
a 100µs delay. Figure 13 demonstrates the
difference between the LTC6994-1, which
74AC86
FREQ2X
500kHz
fIN
250kHz
IN
OUT
LTC6994
2.25V TO 5.5V
GND
V+
SET
DIV
RSET
49.9k
OUT
250kHz
C1
0.1µF
TIE DIV TO GND
FOR DIVCODE = 0
4µs
IN
More Online
Learn more about TimerBlox devices at
www.linear.com/timerblox. There you can find
data sheets, TimerBlox Designer software, even
an introductory video about the products.
8 | January 2011 : LT Journal of Analog Innovation
1µs
OUT
FREQ2X
Figure 14. 90° phase-shifted (quadrature) signal
generator and frequency doubler
DEVICE
DELAY FUNCTION
LTC6994-1
or
LTC6994-2
or
MOTOR SPEED ALARM
There is no limit to how TimerBlox
devices can be combined to easily produce
esoteric timing functions. For instance,
the design in Figure 15 combines one
shots and delay blocks with a VCO to
produce a high/low motor speed alarm.
The circuit sounds a high frequency
tone if a motor is spinning too fast and
a low frequency tone if too slow.
The input is taken from a motor shaft
encoder or other rotational sensor
and used to trigger a one shot to produce a 1ms pulse per revolution.
The fast alarm threshold can be set
between 10,000 rpm and 1500 rpm which,
in time, is one pulse every 6ms to 40ms.
Re-triggerable one shot, U3, is adjusted
for a time interval equal to the warning threshold value. If it is continually
re-triggered and not allowed to time
out, then the motor is turning too fast.
For time-filtering, a delay timer, U4,
is programmed by the same threshold adjust voltage to delay an output
signal until the motor has exceeded
the threshold speed for 100 revolutions (600ms to 4000ms). The delayed
cover story
The output of the LTC6994 will only respond to input
changes that persist longer than the delay period. This
operation is well suited for pulse qualification, switch
debouncing, or guaranteeing minimum pulse widths.
output signal enables an LTC6990 oscillator to produce a 5kHz warning tone.
Another time filter is created with delay
block U7 which sounds a lower frequency
alarm if the motor remains too slow
for 10 revolutions (500ms to 5000ms).
Two OR gates are used to detect when
the motor has stopped completely.
The slow alarm threshold can be set
between 1200 rpm and 120 rpm or one
pulse every 50ms to 500ms. The delay
timer, U5, pulses its output if allowed
to time out because the motor speed is
too slow. This output re-triggers one
shot U6 and keeps its output high as
long as the speed remains too slow.
design effort to produce accurate and
reliable circuits. Several of the five core
products are available in multiple versions to cover more applications and
reduce the need for external components.
Each part is designed to be as flexible as
possible with a 2.25V to 5.5V supply range,
up to –40°C to 125°C temp range and wide
timing ranges. In addition, the parts are
offered in a small 2mm × 3mm DFN or a
low-profile SOT-23 (ThinSOT™) package
when a leaded package is required. n
CONCLUSION
The Linear Technology TimerBlox family of silicon oscillators fills a designer’s
toolbox with simple and dependable
timing solutions that require minimal
Figure 15. Motor speed alarm
OUT
U3
LTC6993-2
IN
TRIG
GND
V+
SET
DIV
5V
OUT
U4
LTC6994-1
GND
V+
SET
DIV
5V
1M
1M
5V
TRIG
392k
681k
137k
681k
OE
10k
1N4148
10k RPM
U2
LTC6990
GND
5V
ALARM OUTPUT
TOO SLOW = 250Hz
TOO FAST= 5kHz
V+
1M
A2
SET
5V
97.6k
DIV
887k
V+
GND
976k
IN
182k
GND
V+
SET
DIV
DIV
787k
OUT
U5
LTC6994-1
OUT
U6
LTC6993-4
IN
TRIG
5V
5V
OUT
U7
LTC6994-1
GND
V+
GND
V+
SET
DIV
SET
DIV
523k
5V
TOO SLOW
WARNING 10k
THRESHOLD ADJUST
OUT
383k
A1
OUT
U1
LTC6993-1
SET
88.7k
1.5k RPM
TOO FAST
WARNING 10k
THRESHOLD ADJUST
MOTOR
SPEED INPUT
1 PULSE/REV
432k
1N4148
383k
84.5k
1M
5V
1M
383k
86.6k
383k
84.5k
681k
120 RPM
1.2k RPM
U1: MOTOR SPEED INPUT. Output is 1ms, one-shot pulse per revolution.
U2: ALARM OUTPUT. Output of this VCO is alarm oscillator tone.
FAST SPEED SENSOR
U3: RETRIGGERABLE ONE SHOT. Output stops pulsing if rpm > fast threshold.
U4: DELAY TIME FILTER. If rpm too fast for 100 cycles, ouput alarm is sounded.
SLOW SPEED SENSOR
U5: DELAY TIMER. Output never pulses if rpm > slow threshold.
U6: RETRIGGERABLE ONE SHOT. If triggered output goes high and stays high if rpm below threshold.
U7: DELAY TIME FILTER. If rpm too slow for 10 cycles, output alarm is sounded.
A1, A2: Logic to sound alarm if motor too slow or stopped.
January 2011 : LT Journal of Analog Innovation | 9
Battery Charger’s Unique Input Regulation Loop Simplifies
Solar Panel Maximum Power Point Tracking
Jay Celani
Solar panels have great potential as
energy harvesting power sources—they
just need batteries to store the harvested
power and to provide carry-though during dark periods. Solar panels are relatively expensive, so extracting maximum
power from the panels is paramount to
minimizing the panel size. The tricky part
is a balancing of solar panel size with
required power. The characteristics of
solar panels require careful management
of the panel’s output power versus load
to effectively optimize the panel’s output
power for various lighting conditions.
(see Figure 1). Maintaining this peakpower point during operation as lighting
conditions change is called maximum
peak power tracking (MPPT). Complex
algorithms are often used to perform this
function, such as varying the panel’s load
periodically while directly measuring panel
output voltage and output current, calculating panel output power, then forcing
the point of operation that provides the
peak output power as illumination and/or
temperature conditions change. This type
of algorithm generally requires complex
circuitry and microprocessor control.
For a given illumination level, a solar
panel has a specific operating point that
produces the maximum amount of power
There exists, however, an interesting
relationship between the output voltage of a solar panel and the power that
the panel produces. A solar panel output
voltage at the maximum power point
remains relatively constant regardless of
illumination level. It follows that forcing operation of the panel such that the
output voltage is maintained at this peak
power voltage (VMP) yields peak output
power from the panel. A battery-charger
can therefore maintain peak power
In Depth
For an in-depth discussion of the maximum
power point tracking feature of the LT3652,
see “Designing a Solar Cell Battery Charger”
in the December 2009 issue of LT Magazine.
You can find this article and the LT3652
data sheet at www.linear.com/3652.
10 | January 2011 : LT Journal of Analog Innovation
INCREASING ILLUMINATION
2.2
2
24
VMP
22
20
I VS V
1.8
18
P VS V
1.6
16
1.4
14
1.2
12
1
10
0.8
8
0.6
6
0.4
4
0.2
2
0
0
0
2
4
6
8
10
VPANEL (V)
12
14
16
PPANEL (W)
2.4
IPANEL (A)
Advances in battery technology and device performance
have made it possible to produce complex electronics that
run for long periods between charges. Even so, for some
devices, recharging the batteries by plugging into the grid
is not possible. Emergency roadside telephones, navigation
buoys, and remote weather monitoring stations are just a
few applications that have no access to the power grid,
so they must harvest energy from their environment.
Figure 1. Current vs voltage and power vs voltage
for a solar panel at a number of different illumination
levels. The panel output voltage at the maximum
power point (VMP) remains relatively constant
regardless of illumination level.
transfer by exploiting this VMP characteristic instead of implementing complex MPPT circuitry and algorithms.
A FEW FEATURES OF
THE LT3652 BATTERY CHARGER
The LT3652 is a complete monolithic stepdown multi-chemistry battery charger
that operates with input voltages as high
as 32V (40V abs max) and charges battery
stacks with float voltages up to 14.4V.
The LT3652 incorporates an innovative
input regulation circuit, which implements a simple and automatic method
for controlling the charger’s input supply voltage when using poorly regulated sources, such as solar panels. The
LT3652HV, a high voltage version of the
charger, is available to charge battery
stacks with float voltages up to 18V.
design features
The LT3652 is a versatile platform for simple and efficient
solar-powered battery charger solutions, applicable to
a wide variety of battery chemistries and configurations.
The LT3652 is equally at home in conventionally powered
applications, providing small and efficient charging solutions
for a wide variety of battery chemistries and stack voltages.
Input Regulation Loop Maintains
Peak Power Point for Solar Panels
The LT3652 input regulation loop linearly reduces the output battery charge
current if the input supply voltage falls
toward a programmed level. This closedloop regulation circuit servos the charge
current, and thus the load on the input
supply, such that the input supply voltage
is maintained at or above a programmed
level. When powered by a solar panel,
the LT3652 implements MPPT operation by
simply programming the minimum input
voltage level to that panel’s peak power
voltage, VMP. The desired peak-power voltage is programmed via a resistor divider.
current or drop in solar panel illumination levels. In either case the regulation
loop maintains the solar panel output
voltage at the programmed VMP as set
by the resistor divider on VIN_REG.
The input regulation loop is a simple
and elegant method of forcing peak
power operation from a particular solar
panel. The input voltage regulation
loop also allows optimized operation
from other types of poorly regulated
sources, where the input supply can collapse during overcurrent conditions.
Integrated, Full-Featured
Battery Charger
The LT3652 operates at a fixed switching frequency of 1MHz, and provides
a constant-current/constant-voltage
(CC/CV) charge characteristic. The part
is externally resistor-programmable to
provide charge current up to 2A, with
charge-current accuracy of ±5%. The IC is
If during charging, the power required
by the LT3652 exceeds the available
power from the solar panel, the LT3652
input regulation loop servos the charge
current lower. This might occur due to
an increase in desired battery charge
particularly suitable for the voltage ranges
associated with popular and inexpensive
“12V system” solar panels, which typically
have open-circuit voltages around 25V.
The charger employs a 3.3V float voltage
feedback reference, so any desired battery float voltage from 3.3V to 14.4V (or
up to 18V with the LT3652HV) can be
programmed with a resistor divider. The
float-voltage feedback accuracy for the
LT3652 is ±0.5%. The wide LT3652 output
voltage range accommodates many battery
chemistries and configurations, including
up to three Li-ion/polymer cells in series,
up to four LiFePO4 (lithium iron phosphate) cells in series, and sealed lead acid
(SLA) batteries containing up to six cells
in series. The LT3652HV, a high-voltage
version of the charger, is also available.
The LT3652HV operates with input voltages
up to 34V and can charge to float voltages of 18V, accommodating 4-cell Li-ion/
polymer or 5-cell LiFePO4 battery stacks.
CMSH1-40MA
523k
VIN
LT3652
VIN_REG
100k
10k
10k
LED
SHDN
1N4148
10µH
0.05Ω
BAT
NTC
LED
FAULT
VFB
TIMER
Figure 2. A 2A solar panel power
manager for a 2-cell LiFePO4 battery
with 17V peak power tracking
1µF
SENSE
CHRG
CMSH3-40MA
SW
BOOST
549k
22
SYSTEM
LOAD
CMSH3-40MA
10µF
10µF
464k
10k
B = 3380
MURATA NCP18XH103
2-CELL LiFePO4 (2 × 3.6V) BATTERY PACK
+
INPUT REGULATION VOLTAGE (V)
SOLAR PANEL INPUT
(<40V OC VOLTAGE)
TA = 25°C
20
18
100% TO 98% PEAK POWER
16
98% TO 95% PEAK POWER
14
12
10
0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8
CHARGER OUTPUT CURRENT (A)
2
Figure 3. A 17V input voltage regulation threshold
tracks solar panel peak power to greater than 98%
January 2011 : LT Journal of Analog Innovation | 11
The LT3652 incorporates an innovative input regulation
circuit, which implements a simple and automatic method
for controlling the charger’s input supply voltage when
using poorly regulated sources, such as solar panels.
Energy Saving
Low Quiescent Current Shutdown
The LT3652 employs a precision-threshold
shutdown pin, allowing simple implementation of undervoltage lockout functions using a resistor divider. While in
low-current shutdown mode, the LT3652
draws only 15µ A from the input supply.
The IC also supports temperature-qualified
charging by monitoring battery temperature using a single thermistor attached
to the part’s NTC pin. The device has two
binary coded open-collector status pins
that display the operational state of the
LT3652 battery charger, CHRG and FAULT.
These status pins can drive LEDs for visual
signaling of charger status, or be used as
logic-level signals for control systems.
REPLACED BLOCKING DIODE
Si2319
SOLAR PANEL
INPUT
(14.5V TO 40V)
442k
PFET SUBCIRCUIT
TO REPLACE
BLOCKING DIODE
VIN
LT3652
VIN_REG
10k
49.9k
SENSE
SHDN
49.9k
Figure 4. A 2A 3-cell LiFePO4 charger
using a P-channel FET for input
blocking to increase high-current
charging efficiencies
FAULT
SW
1µF
1N4148
10µH
BOOST
0.05Ω
3.3V
BAT
CHRG
VFB
TIMER
NTC
0.68µF
226k
180k
10µF
100k
10k
B = 3380
MURATA NCP18XH103
+
3-CELL LiFePO4 (3 × 3.6V) BATTERY PACK
SIMPLE SOLAR POWERED
BATTERY CHARGER
Figure 2 shows a 2A 2-cell LiFePO4 battery charger with power path management. This circuit provides power to the
system load from the battery when the
solar panel is not adequately illuminated
and directly from the solar panel when
Figure 5. Comparative efficiencies for blocking
Schottky diode vs blocking FET as battery voltage
rises for 15V to 10.8V 3-cell LiFePO4 charger
the power required for the system load
is available. The input voltage regulation loop is programmed for a 17V peak
power input panel. The charger uses
C/10 termination, so the charge circuit is
disabled when the required battery charge
current falls below 200m A. This LT3652
charger also uses two LEDs that provide
status and signal fault conditions. These
binary-coded pins signal battery charging, standby or shutdown modes, battery
temperature faults and bad battery faults.
90
89
88
FET
87
86
85
84
SCHOTTKY
83
82
81
80
12 | January 2011 : LT Journal of Analog Innovation
CMSH3-40MA
10µF
10V
100k
EFFICIENCY (%)
The LT3652 contains a programmable
safety timer used to terminate charging
after a desired time is reached. Simply
attaching a capacitor to the TIMER pin
enables the timer. Shorting the TIMER pin
to ground configures the LT3652 to terminate charging when charge current
falls below 10% of the programmed
maximum (C/10), with C/10 detection
accuracy of ±2.5%. Using the safety timer
for termination allows top-off charging
at currents less than C/10. Once charging is terminated, the LT3652 enters a
low-current (85µ A) standby mode. An
auto-recharge feature starts a new charging cycle if the battery voltage falls 2.5%
below the programmed float voltage. The
LT3652 is packaged in low-profile, 12-lead
3mm × 3mm DFN and MSOP packages.
7.7
8.2
8.7
9.7
9.2
VBAT (V)
10.2
10.7
The input voltage regulation point is
programmed using a resistor divider
from the panel output to the VIN_REG pin.
Maximum output charge current is
reduced as the voltage on the solar panel
output collapses toward 17V, which corresponds to 2.7V on the VIN_REG pin.
This servo loop thus acts to dynamically reduce the power requirements
of the charger system to the maximum
design features
When powered by a solar panel, the LT3652 implements
maximum power point tracking (MPPT) operation by
simply programming the minimum input voltage level to
that panel’s peak power voltage, VMP. The specific desired
peak-power voltage is programmed via a resistor divider.
L1
10µH
SOLAR PANEL INPUT
(<40V OC VOLTAGE)
1µF
CIN
390µF
50V
523k
+
1.15M
VIN
SW
SENSE
VIN_REG
FAULT
10µF
50V
CMSH3-40MA
CMSH1-4
0.05Ω
BOOST
LT3652
BAT
IN
CHRG
SHDN
100k
NTC
100µF
10V
10µF
16V
100k
TIMER
VFB
Figure 6. A solar powered, 2A Li-ion charger with
ideal diode output pass element; the LTC4411 ideal
diode IC prevents reverse conduction during lowlight conditions
OUT
LTC4411
GND
CTL
STAT
316k
1.18M
0.1Ω
+ Li-ION
BATTERY
4.2V
L1: IHLP-2525CZ-01
power that the panel can provide, maintaining solar panel power utilization
close to 100%, as shown in Figure 3.
WANT BETTER EFFICIENCY?
REPLACE THE BLOCKING DIODE
WITH A BLOCKING FET
The LT3652 requires a blocking diode
when used with battery voltages
higher than 4.2V. The voltage drop
across this diode creates a power loss
term that reduces charging efficiency.
This term can be greatly reduced by
replacing the blocking diode with a
P-channel FET, as shown in Figure 4.
Figure 4 shows a 3-cell LiFePO4 2A charger with a float voltage of 10.8V. This
charger has an input voltage regulation
threshold of 14.5V and is enabled by the
SHDN pin when VIN ≥ 13V. Charge cycle
termination is controlled by a 3-hour
timer cycle. The blocking diode normally
used in series with the input supply for
reverse voltage protection is replaced
by a FET. A 10V Zener diode clamp is
used to prevent exceeding the FET maximum VGS . If the specified VIN range does
not exceed the maximum VGS of the
input FET, this clamp is not required.
During the high-current charging period
of a normal charge cycle (ICHG > C/10), the
CHRG status pin is held low. In the charger shown in Figure 4, this CHRG signal
is used to pull the gate of the blocking FET low, enabling a low-impedance
power supply path that eliminates the
blocking diode drop to improve conversion efficiency. Figure 5 shows that the
addition of this blocking FET improves
efficiency by 4% compared to operation with a blocking Schottky diode.
Should the timer be used for termination, the body diode of the FET provides
a conduction path once charge currents
of < C/10 is achieved, and the CHRG pin
becomes high-impedance. If desired, a
blocking Schottky diode can be left in
parallel with the blocking FET to improve
conversion efficiency during the top-off
portion of the timer-controlled charge
cycle. Use of a FETKEY as the blocking
element also increases top-off efficiency.
SCARED OF THE DARK?
USE AN IDEAL DIODE FOR
LOW-LIGHT APPLICATIONS
When the LT3652 is actively charging,
the IC provides an internal load onto the
switching loop to ensure closed-loop
operation during all conditions. This is
accomplished by sinking 2m A into the
BAT pin whenever a charging cycle is
active. In a solar-panel-powered battery
charger, low-light conditions can make the
input solar panel voltage collapse below
the input regulation threshold, causing
January 2011 : LT Journal of Analog Innovation | 13
SOLAR PANEL INPUT
VP(MAX) = 3.8V
L1
4.7µH
C1
2.2µF
VIN VS L
L2
10µH
MBRS230LT1
470pF
SW
FBN
38.3k
SHDN
30.9k
LT3479
VREF
SS
RT
0.1µF
GND
17.8k
1µF
50V
169k
100pF
0.1µF
10V
SENSE
100k
4.32k
10nF
CMSH3-40MA
SW
VIN_REG
10µF
10V
210k
FBP
VC
VIN
CMSH1-4
BOOST
FAULT
0.068Ω
LT3652
CHRG
BAT
SHDN
NTC
100k
TIMER
VFB
1.18M
L1: CDRH-6D28-3R0
L2: IHLP-2525CZ-01
VBAT
4.2V
1.5A
316k
100µF
10V
10µF
16V
0.1Ω
Figure 7. Low-voltage solar panel powers 1.5A single cell Li-ion buck/boost battery charger. The LT3479 boosts the solar panel’s 3.8V output to operate an LT3652
charger. The LT3652’s closed loop operation includes the boost converter, thus regulating the LT3479’s input to the solar panel’s VMP of 3.8V.
output charge current to be reduced to
zero. If the charger remains enabled during this condition (i.e., the panel voltage remains above the UVLO threshold),
the internal battery load results in a net
current drain from the battery. This is
undesirable for obvious reasons, but this
condition can be eliminated by incorporating a unidirectional pass element that prevents current backflow from the battery.
Linear Technology makes a high-efficiency pass element IC, the LTC4411 ideal
diode, which has an effective forward
drop close to zero. The effect on overall
charger efficiency and final float voltage is negligible due to the extremely
low forward drop during conduction.
Figure 6 shows an LT3652 solar-powered
battery charger that employs low-light
reverse protection using an LTC4411
ideal-diode IC. During a low light condition, should the panel voltage collapse
below the input regulation threshold,
the LT3652 reduces battery charge current to zero. In the case where the input
voltage remains above the UVLO threshold, the charger remains enabled but is
held in a zero charge current state. The
LT3652 attempts to sink 2m A into the
14 | January 2011 : LT Journal of Analog Innovation
BAT pin; however, the LTC4411 prevents
reverse conduction from the battery.
NEED TO STEP-UP? NO PROBLEM.
A 2-STAGE BUCK-BOOST
BATTERY CHARGER
The LT3652 can be used for step-up
and step-up/step-down charger applications by incorporating a front-end
step-up DC/DC converter. The frontend converter generates a local highvoltage supply for the LT3652 to use
as an input supply. The LT3652 input
regulation loop functions perfectly when
wrapped around both converters.
Figure 7 shows a low-voltage solar panel
powered 1.5A single-cell Li-ion charger
with a 4.2V float voltage. This charger is
designed to operate from a solar panel
that has a peak power voltage of 3.8V.
An LT3479 switching boost converter
running at 1MHz is used on the front-end
to create an 8V supply, which is used to
power the LT3652. This charger operates
with input voltages as low as the input
regulation threshold of 3.8V, up to 24V, the
maximum input voltage for the LT3479.
When input voltages approach 8V (or
higher), the LT3479 boost converter no
longer regulates, ultimately operating at
0% duty cycle and effectively shorting the
input supply through the pass Schottky
diode to the LT3652. Because the input
regulation loop monitors the input to the
LT3479, when the input voltage collapses
toward the input regulation threshold,
the LT3652 scales back charge current,
reducing the current requirements of
the LT3479 boost converter. The input
voltage servos to the regulation point,
with the boost converter and LT3652
charger combination extracting the peak
power available from the solar panel.
NEED MORE CHARGE CURRENT?
USE MORE LT3652s
Multiple LT3652 chargers can be used in
parallel to produce a charger that exceeds
the charge current capability of a single
LT3652. In the application shown in
Figure 8, three 2A LT3652 charger networks are connected in parallel to yield a
6A 3-cell Li-ion charger with a float voltage
of 12.3V that uses C/10 termination. This
charger is solar power compatible, having
an input regulation threshold of 20V. This
charger also implements an input blocking FET to increase charging efficiencies.
The three LT3652 charger ICs share a
common float voltage feedback network
design features
SOLAR PANEL INPUT
(20V TO 32V)
Si4401DY
649k
10V
100k
CIN
390µF
50V
10k
LED
+
MBRS340
VIN_REG
100k
VIN
SW
1µF
SHDN
CHRG
BAT
FAULT
NTC
VFB
0.05Ω
TIMER
28.7k
280k
10k
B = 3380
and a common input regulation network.
A feedback network with an equivalent
resistance of 250k W is recommended
to compensate for input bias currents
into the LT3652 VFB pin. Since the three
LT3652s share the same feedback network in this charger, and the input bias
currents are also shared through the
network, the network equivalent resistance is reduced to 250k/3, or ~83k W.
Due to tolerances in reference voltages,
one of the ICs will likely power up before
the other during an auto-recharge event.
In this case, the battery auto-recharges
at a maximum of 2A. Should the battery
continue to discharge due to a >2A load,
the second charger engages. Higher
discharge currents will engage the third
charger IC, allowing the charger to produce the full 6A system charge current.
The CHRG pins on all of the LT3652s are
tied together to enable the input blocking
FET, so the FET is low-impedance regardless of which order the ICs auto-restart.
The NTC and status functions are shared
by all three LT3652s, with each IC using a
dedicated NTC thermistor. The open collector status pins of the ICs are shorted
together, so engaging any or all of the
individual chargers lights the CHRG status
indicator. Likewise, an NTC fault in any of
the ICs lights the FAULT status indicator.
The individual LT3652 NTC functions are
slaved to each other via a diode connected from the common FAULT pins to
the common VIN_REG pins of all three ICs.
10µH
SENSE
LT3652
10k
4.7V 1N4148
BOOST
1N4148
10k
LED
4.7µF
x3
1.4M
100k
MBRS340
VIN_REG
VIN
SW
1µF
SHDN
4.7V 1N4148
BOOST
10µH
0.05Ω
SENSE
LT3652
CHRG
BAT
FAULT
NTC
VFB
TIMER
28.7k
10k
B = 3380
MBRS340
VIN_REG
VIN
SW
1µF
SHDN
4.7V 1N4148
BOOST
0.05Ω
SENSE
LT3652
CHRG
BAT
FAULT
NTC
VFB
TIMER
100k
10k
B = 3380
10µH
28.7k
3-CELL Li-ION
(3 × 4.2V)
BATTERY
+
MURATA NCP18XH103
Figure 8. A 6A 3-cell Li-ion battery charger using three LT3652 charger ICs
This diode pulls the VIN_REG pin below
the VIN_REG threshold should any of the
ICs trigger an NTC fault, which disables
all output charge current until the temperature fault condition is relieved.
CONCLUSION
The LT3652 is a versatile platform for
simple and efficient solar-powered battery charger solutions, applicable to a
wide variety of battery chemistries and
configurations. The LT3652 is equally at
home in conventionally powered applications, providing small and efficient
charging solutions for a wide variety of
battery chemistries and stack voltages.
Solar-powered charger solutions maintain
panel utilization close to 100%, reducing solution costs due to minimized panel
area. The compact size of the IC, coupled
with modest external component requirements, allows construction of stand-alone
charger systems that are both tiny and
inexpensive, providing a simple and
efficient solution to realize true gridindependence for portable electronics. n
January 2011 : LT Journal of Analog Innovation | 15
Two High Power Monolithic Switching Regulators Include
Integrated 6A, 42V or 3.3A, 42V Power Switches,
Built-in Fault Protection and Operation up to 2.5MHz
Matthew Topp and Joshua Moore
Power supply designers looking to shrink applications and simplify layout often turn to
monolithic switching regulators. Monolithics simplify power supply layout by including
the power switch on the die—no external FETs or precision sense resistors are needed.
Monolithics can also operate at substantially higher switching frequencies than their
controller-only counterparts, thus reducing the size and number of external passive
components. The benefits of monolithic regulators are clear, but they traditionally have one
major limitation: as the required power level goes up, the likelihood of finding a suitable
monolithic regulator diminishes. Two new high power monolithics, the LT3579 and LT3581,
solve this problem by integrating 6A (42V) and 3.3A (42V) power switches, respectively.
The LT3579 and LT3581 are highly flexible parts and can be configured in
boost, SEPIC, inverting, or flyback configurations. They also offer many unique
performance and fault protection features. When configured as high power
boost converters, these parts can survive
output overloads with only a few additional external components. They can
also be configured to provide hot-plug
and reverse input voltage protection.
In addition, a novel master and slave
power switch design allows high voltage
charge pump circuits to be made with low
power dissipation and few components.
Both parts can be programmed to free-run
from 200kHz to 2.5Mhz or can be synchronized with an outside clock source.
The parts also provide a clock output
pin, enabling the ICs to synchronize
other switching regulators. The LT3579
comes in a 4mm × 5mm QFN and 20-lead
TSSOP package, and the LT3581 comes in a
4mm × 3mm DFN and 16-lead MSE package.
L1
2.2µH
VIN
5V
D1
VOUT
12V
1.7A
M1
COUT1
10µF
100k
200k
CIN
22µF
VIN
SW1 SW2
FAULT
LT3579
130k
FB
CLKOUT
RT
SYNC
VC
86.6k
GND
D2
VIN
GATE
SHDN
COUT
10µF
6.34k
SS
47pF
CIN: 22µF, 16V, X7R, 1210
COUT1, COUT: 10µF, 25V, X7R, 1210
D1: VISHAY SSB43L
D2: CENTRAL SEMI CMDSH-3TR
L1: WÜRTH WE-PD 744771002
M1: SILICONIX SI7123DN
8.06k
0.1µF
2.2nF
Figure 1. This 5V to 12V boost converter can survive the infamous metal file test where a wire attached to the
output is dragged across the jagged surface of a grounded metal file
VOUT
10V/DIV
TO ONE OR MORE
LT3579/LT3581
TO FAULT PIN OF
ONE OR MORE
LT3579/LT3581
TO ONE OR MORE
LT3579/LT3581
VIN
CLKOUT
2V/DIV
RGATE
IL
5A/DIV
GATE
FAULT
LT3579/LT3581
FAULT
5V/DIV
VIN
10µs/DIV
Figure 2. Operating waveforms for Figure 1 circuit
during brutal metal file test
16 | January 2011 : LT Journal of Analog Innovation
Figure 3. Input disconnect schematic
design features
L1
4.7µH
VIN
5V
D1
M1
VOUT
12V
0.5A (VIN = 3V)
0.9A (VIN = 5V)
C1
10µF
100k
51.1k
VIN
SW1 SW2
FAULT
CIN2
4.7µF
CIN1
10µF
130k
6.34k
FB
D3
VIN
GATE
LT3581
SHDN
CLKOUT
RT
SYNC
VC
GND
COUT1
10µF
SS
86.6k
16.9k
0.22µF
2.2nF
47pF
Figure 4. A 3V–5V input to ±12V output converter
C2
4.7µF
L2
3.3µH
L3
3.3µH
VOUT
–12V
0.5A (VIN = 3V)
0.9A (VIN = 5V)
D2
VIN
CIN1, CIN3, CIN4: 10µF, 10V, X5R, 1206
CIN2 : 4.7µF, 10V, X5R, 0805
C1, COUT1, COUT2: 10µF, 25V, X7R, 1210
C2: 4.7µF, 25V, X7R, 1206
D1, D2: DIODES INC SBR2A40P1
D3: CENTRAL SEMI CTLSH1-40M563
CIN3
L1: COILCRAFT XPL7030-472ML
10µF
L2, L3: COOPER DRQ125-3R3
M1: SILICONIX SI7123DN
FAULT PROTECTION FEATURE:
CURRENT OVERLOADS
Most high power boost converters cannot survive an output overload condition
because of the inherent DC pathway that
exists from the input to output through
the inductor and rectifying diode. An
output overload or short causes the
current in this pathway to increase and
run away, thus damaging anything in
this pathway or connected to it. The
12VOUT
500mV/DIV
AC COUPLED
CIN4
10µF
86.6k
LT3579 and LT3581 include features that
protect against such fault events.
Figure 1 shows an LT3579 configured
as a 5V input to 12V output boost converter with output short protection. An
external PFET, diode, and resistor are
all it takes to implement robust output
short protection. In fact, this circuit can
survive the infamous “file test,” where a
wire tied to the output is swiped across
the surface of a metal woodworking
file tied to ground. Figure 2 shows the
IL1
2A/DIV
–12VOUT
500mV/DIV
AC COUPLED
IL2 + IL3
2A/DIV
IL2 + IL3
2A/DIV
100µs/DIV
Figure 5. Load step from 0.25A to 0.75A between
+12V and −12V rail, with 5V input
143k
FB
FAULT
GATE
LT3579-1
FAULT
IL1
2A/DIV
SW1 SW2
10µs/DIV
Figure 6. Transient short between rails with 5V input,
0.9A load before short
SHDN
CLKOUT
RT
SYNC
VC
GND
SS
COUT2
10µF
×2
14.3k
0.22µF
2.2nF
47pF
operating waveforms during this normally
destructive test—the LT3579 survives
this brutal test without any problems.
These parts also protect against several
other types of fault conditions, including
overcurrent conditions, overvoltage on
VIN, and over-temperature inside the part.
In systems where multiple LT3579s/LT3581s
are incorporated to produce multiple
rails, a single PFET and resistor can be
used on the input side to protect all the
rails from a current overload. Figure 3
shows how to set this up. Simply tie the
FAULT pins of all ICs together and connect to a single pull-up resistor. The fault
control scheme is designed so that if one
part goes into fault, it pulls its FAULT pin
low, causing the other parts to go into
fault as well. Switching activity in all
parts stops and all enter into a time-out
period. This time-out period allows the
components in the system to cool down.
Only after the last part exits the time-out
period do all parts attempt to restart. To
January 2011 : LT Journal of Analog Innovation | 17
The LT3579 and LT3581 include features that protect
against a number of fault events including output
overloads or shorts, overcurrent conditions, overvoltage
on VIN, and over-temperature inside the part.
C2
2.2µF
L1
3.3µH
M2
6.34k
•
SW1
GATE
VIN
C1
3.3µF
100k
GATE
LT3579/LT3581
CIN: 0.1µF, 25V, X7R, 0805
C1: 3.3µF, 25V, X7R, 1206
C2: 2.2µF, 50V, X7R, 1206
C3: 10µF, 25V, X7R, 1210
D1: CENTRAL SEMI CTLSH2-40M832
L1, L2: COILCRAFT MSD7342-332MLB
M1, M2: SILICONIX SI7123DN
VIN
Figure 7. Recommended connections for hot plug,
reverse input voltage, and input overvoltage events
isolate a fault to only one part, simply
do not connect the FAULT pins together.
The LT3579 and LT3581 can be easily mixed
within a system while maintaining all
overload and protection features. Figure 4
shows the LT3579-1 configured as an
inverting converter working together with
an LT3581 configured as a boost converter.
Together, these converters generate a
VIN
10V/DIV
43.2k
L2
3.3µH
FB
130k
SHDN
FAULT
C3
10µF
CLKOUT
RT
VC
SYNC
SS
GND
100pF
0.1µF
10k
2.2nF
Figure 8. A 9V–16V input to 12V output SEPIC with hot plug, reverse input voltage, and input
overvoltage protection
regulated ±12V output at up to 0.9A running off a 3V–5V input, with overload and
over-temperature protection. The LT3579-1
is used because it features low input
ripple (see page 19 for more about this
feature of the LT3579-1). Figure 5 shows
the load step response. This system not
only accommodates output shorts and
overloads between each rail to ground,
but it can also tolerate these conditions
between the rails as shown in Figure 6.
FAULT PROTECTION FEATURES: HOT
PLUG, REVERSE INPUT VOLTAGE,
AND INPUT OVERVOLTAGE
The GATE pin, SS pin and related circuitry
can also be used to protect against hot
plug, reverse input voltage, and input
overvoltage events. Figure 7 shows one
way to set this up. Hot plug protection is
VIN
10V/DIV
GND
VIN
10V/DIV
IL1 + IL2
2A/DIV
VOUT
12V
1A (VIN = 9V)
1.1A (VIN = 12V)
1.3A (VIN = 16V)
SW2
LT3581
VIN
CIN
0.1µF
D1
•
VIN
9V TO 16V
M1
IL1 + IL2
2A/DIV
VGATE
20V/DIV
SS
1V/DIV
VOUT
20V/DIV
SS
1V/DIV
VOUT
10V/DIV
IL1 + IL2
2A/DIV
VOUT
10V/DIV
200ms/DIV
Figure 9. Operating waveforms for a hot plug event
18 | January 2011 : LT Journal of Analog Innovation
100ms/DIV
Figure 10. Operating waveforms for a negative VIN
transient
500ms/DIV
Figure 11. Operating waveforms for a VIN
overvoltage transient
design features
Packed with the latest features and some of the highest
power levels of any monolithic converters in the industry,
the LT3579 and LT3581 venture into applications
once reserved for controllers with external FETs.
C2
1µF
L1
3.3µH
L2
3.3µH
VOUT
–12V
625mA
•
•
VIN
5V
D1
CLKOUT
2V/DIV
SW1 SW2
VIN
C1
3.3µF
100k
43.2k
LT3581
FB
SHDN
GATE
FAULT
CLKOUT
RT
VC
SYNC
SS
143k
SW
10V/DIV
C3
4.7µF
47pF
GND
0.1µF
11k
1nF
VOUT
50mV/DIV
AC COUPLED
C1: 3.3µF, 16V, X7R, 1206
C2: 1µF, 25V, X7R, 1206
C3: 4.7µF, 25V, X7R, 1206
D1: DIODES INC. PD3S230H-7
L1, L2: COILCRAFT MSD7342-332MLB
IL1 + IL2
2A/DIV
200ns/DIV
Figure 13. High operating frequency results in low
output ripple, even at maximum load
Figure 12. A 5V input to −12V output inverting DC/DC converter
useful for limiting the surge current when
the input to the power supply is suddenly
stepped from low voltage to normal.
In a boost converter, there is a DC path
from the input to the output capacitors
of the circuit. Since these capacitors are
initially discharged, large surge currents
are possible if this feature is not used.
Figure 8 shows a circuit designed to
handle all these potentially dangerous
conditions. Figure 9 shows the operating waveforms during a hot plug event,
Figure 10 shows the waveforms during a
negative VIN transient, and Figure 11 shows
the result of a VIN overvoltage transient.
The LT3579/LT3581 survives all these
fault conditions and when the fault is
removed, resumes a normal start-up cycle.
parts can synchronize to an external clock.
The CLKOUT pin on the parts is designed
to drive the SYNC pins of other switching regulators. The LT3579 and LT3581
also encode die temperature information
into the duty cycle of the CLKOUT signal,
making thermal measurements simple.
small. The amount of output ripple is
also very low, as shown in Figure 13.
Figure 14 shows a 2.8V to 4.2V input to
5V output boost running at 2MHz using
the LT3579. This circuit is configured
to survive output overloads and can
deliver up to 2A of output current.
Figure 12 shows a 2MHz, 5V input to
−12V output inverting converter with
625m A of output current capability using
the LT3581. Due to the high switching frequency, external components are
USE THE LT3579-1 FOR EVEN MORE
POWER AND SPEED
The LT3579-1 is nearly identical to the
LT3579 with one exception: the CLKOUT pin
has a 50% duty cycle that does not vary
Figure 14. Li-ion battery to 5V output boost running
at 2MHz can deliver 2A of output current.
L1
0.47µH
VIN
2.8V TO 4.2V
VIN
SW1 SW2
SHDN
100k
CIN: 10µF, 16V, X7R, 1206
COUT1, COUT: 22µF, 16V, X7R, 1210
D1: CENTRAL SEMI CTLSH3-30M833
L1: VISHAY IHLP-2020BZ-01-R47
M1: SILICONIX SI7123DN
VOUT
5V
2A
M1
COUT1
22µF
HIGH POWER AND HIGH SPEED
The combination of high current capability and high switching frequency make
the LT3579/LT3581 useful in a wide range
of applications. Not only can the parts
be set for an internal oscillator frequency
between 200kHz and 2.5MHz, but the
D1
45.3k
GATE
LT3579
COUT
22µF
CLKOUT
FAULT
VC
RT
SYNC GND
SS
CIN
10µF
43.2k
10k
FB
47pF
22nF
6.34k
2.2nF
January 2011 : LT Journal of Analog Innovation | 19
L2
4.7µH
Figure 15. Dual phase 8V–16V input to 24V boost converter uses
two LT3579-1s and can deliver up to 5.1A of output current
D2
CPWR1, CPWR2: 10µF, 25V, X7R, 1210
CVIN1, CVIN2: 4.7µF, 25V, X7R, 1206
COUT1M, COUT1S, COUT: 4.7µF, 50V, X5R, 1210
D1, D2: CENTRAL SEMI CTLSH5-40M833
D3: CENTRAL SEMI CTLSH1-40M563
L1, L2: VISHAY IHLP-2525CZ-01-4R7
M1: SILICONIX SI7461DP
COUT1S
4.7µF
×2
CPWR2
10µF
SW1 SW2
CLKOUT
LT3579-1 FB
SLAVE
FAULT
VIN
CVIN2
4.7µF
SHDN
RT
SYNC GND
GATE
VC
SS
0.22µF
86.6k
with die temperature and is 180° out of
phase with its own internal clock whether
the part free runs or is synchronized.
This difference allows for the construction of a dual phase converter in the
boost, SEPIC, or inverting configurations.
VPWR
8V TO 16V
CPWR1
10µF
VIN
3.3V TO VPWR
MASTER AND SLAVE SWITCHES
Both the LT3579 and LT3581 have a novel
master/slave switch configuration. To
implement current mode control, the
MASTER
CLKOUT
2V/DIV
D1
VIN
VOUT1
86.6k
SW1 SW2
CLKOUT
LT3579-1
MASTER FB
CVIN1
4.7µF
SYNC GND
4.99k
86.6k
D3**
VPWR
GATE
VC
47pF
SS
0.22µF
VPWR = 8V
VPWR = 12V
VPWR = 16V
2.4A
3.7A
5.1A
VIN = VPWR
2.2A
3.1A
3.9A
**OPTIONAL FOR OUTPUT SHORT CIRCUIT PROTECTION
master switch (SW1 pin) has a current
comparator to monitor the current. The
slave switch (SW2 pin) has no current
comparator and simply operates in phase
with the master. For most applications,
simply tie SW1 and SW2 pins together to
get a 6A or 3.3A total current limit for
the LT3579 and LT3581, respectively. Since
it may be desirable in some situations
to have a lower current limit with an
easy way to upgrade to a higher current
in the future, these parts can operate
using only the master switch. To do this,
simply float the slave switch pins. As a
result, the LT3579 becomes a 3.4A part
and the LT3581 becomes a 1.9A part.
Figure 16. Output ripple at maximum load for the
dual phase circuit shown in Figure 15
20 | January 2011 : LT Journal of Analog Innovation
EFFICIENCY (%)
100µs/DIV
Figure 17. Transient load response for the dual
phase circuit shown in Figure 15
100
8
90
7
80
6
VIN = 12V
70
60
5
4
VIN = 3.3V
50
3
40
2
30
1
20
0
0.5
1
1.5 2 2.5 3 3.5
LOAD CURRENT (A)
4
POWER LOSS (W)
ILOAD
1A/DIV
500ns/DIV
2.2nF
VIN = 3.3V TO 5V
IL1 + IL2
5A/DIV
IL1 + IL2
5A/DIV
6.98k
*MAX OUTPUT CURRENT
VOUT
1V/DIV
AC COUPLED
VOUT
AC COUPLED
100mV/DIV
COUT
4.7µF
×2
137k
FAULT
SHDN
RT
VOUT
24V
5.1A*
6.34k**
COUT1M
4.7µF
×2
21.5k
499k
A major benefit of out-of-phase operation is an inherent reduction in input
and output ripple. Figure 15 shows an
8V–16V input to 24V output dual-phase
boost converter capable of delivering
up to 5.1A of output current. Each part
operates at 1MHz, but because the outputs operate out of phase the effective
switching frequency of the converter is
2MHz. Figure 16 shows the output ripple
at maximum load, Figure 17 shows the
transient load response, and Figure 18
shows the efficiency. This circuit features output short circuit protection,
which is easily removed if not needed.
M1**
VOUT1
L1
4.7µH
0
4.5
Figure 18. Converter efficiency reaches 93% for the
dual phase circuit shown in Figure 15
design features
The master/slave architecture provides
a clear advantage when creating high
voltage charge pump circuits. It is common practice to create high voltage rails
by building a boost converter and adding
charge pump stages to double or even
triple the boost converter’s output voltage.
At higher power levels, it becomes necessary to dampen the current spikes inherent
in these charge pump circuits. Figure 19
shows a traditional approach, which uses
high power resistors within the charge
pumps. Without these resistors, the current spikes would cause the switching
regulator to false trip, causing erratic
and unstable operation. The problem is
that these resistors add to the component
count and generate additional heat.
Figure 20 shows a better solution in
which the master/slave switch configuration eliminates the need for the
high power resistors. All current spikes
caused by the charge pump stages are
only seen by the slave switch, eliminating the possibility of false tripping.
controllers, resulting in solution sizes
unachievable by controller solutions.
Advanced fault protection features make
it possible to produce compact and rugged solutions without additional ICs.
switching regulators. Both parts feature
a wide input operating voltage range.
The LT3579 can operate from 2.5V to
16V and survive transients to 40V. The
LT3581 can operate from 2.5V to 22V with
transients to 40V. Both parts have built-in
programmable soft-start and automatic
frequency foldback. Single pin feedback enables both positive and negative output voltages. Each part has an
accurate comparator/reference for the
SHDN pin, allowing the pin to be used as
a programmable undervoltage lockout.
A new master and slave switch architecture not only allows adjustment of the
current limit but also significantly eases
the design of high voltage boost plus
charge pump circuits. These new features
are simple to implement, yet stay out the
way if not required. The LT3579, with a 6A,
42V switch comes in a 4mm × 5mm QFN or
20-lead TSSOP package. The LT3581, with a
3.3A, 42V switch comes in a 3mm × 4mm
DFN or 16-lead MSE package. n
CONCLUSION
Packed with the latest features and
some of the highest power levels of any
monolithic converter in the industry, the
LT3579 and LT3581 venture into applications once reserved for controllers.
Monolithic converters can operate at
clock speeds far beyond the ability of
D6
C4
2.2µF
×2
BEST IN CLASS SPECIFICATIONS
C3
2.2µF
×2
D2
L1
10µH
VIN
9V TO 16V
D1
M1**
C1
2.2µF
×3
D9**
VOUT2
100k
VOUT1
536k
CIN
10µF
VIN
SW1 SW2
FAULT
SHDN
RT
SYNC
LT3579
383k
DC/DC
GND
Figure 19. Traditional method for building high power
boost plus charge pump circuits
D8**
8.2V
6.49k**
FB
D7**
C2
2.2µF
×3
VIN
GATE
CLKOUT
VC
GND
SS
86.6k
SW
VOUT1
67V
C5
500mA*
2.2µF
×2
D4
D3
With so many new features, it is easy
to overlook that the LT3579 and LT3581
include all the standard features available in many modern Linear Technology
VIN
C6
2.2µF
×2
D5
VOUT2
100V
330mA*
CIN: 10µF, 25V, X7R, 1210
C1-C6: 2.2µF, 50V, X7R, 1210
D1-D6: DIODES INC SBR2A40P1
D7: CENTRAL SEMI CMDSH-3TR
D8: CENTRAL SEMI CMDZ5237B-LTZ
D9: DIODES INC MBRM360
L1: WÜRTH WE-PD 7447710
M1: SILICONIX SI7461DP
27pF
2.2µF
34k
470pF
*MAX TOTAL
OUTPUT POWER
22W (VIN = 9V)
27W (VIN = 12V)
33W (VIN = 16V)
**OPTIONAL FOR OUTPUT
SHORT CIRCUIT PROTECTION
Figure 20. Master and slave switches of the LT3579/LT3581 allow a cooler running, simpler
method for building boost plus charge pump circuits.
January 2011 : LT Journal of Analog Innovation | 21
I2C System Monitor Combines Temperature, Voltage and
Current Measurements for Single-IC System Monitoring
David Schneider
LTC2990 Features
• Measures Voltage, Current and Temperature
• Measures Two Remote Diode Temperatures
• ±1ºC Accuracy, 0.06ºC Resolution
• ±2ºC Internal Temperature
• 15-Bit ADC Measures Voltage and Current
• 3V to 5.5V Supply Operating Voltage
• I2C Serial Interface with Four
Selectable Addresses
• Internal 10ppm Voltage Reference
• 10-Lead MSOP Package
22 | January 2011 : LT Journal of Analog Innovation
1.0
380
100µA
10µA
0.8
∆VD
0.6
0.4
100
200
300
RSERIES = 0Ω
RSERIES = 30Ω
360
DIODE TEMPERATURE (K)
DIODE VOLTAGE (V)
The limit on the complexity
of large integrated circuits
is dominated by how much
power they can dissipate.
The trend in µprocessors
and FPGAs is toward
packing more features into
smaller ICs, run at ever-lower
voltages. The resulting rise
in power dissipation makes
it increasingly difficult to
monitor and control sources
of heat. Where it was once
suitable to have a single
chassis temperature monitor
to deduce the health of the
system, modern electronic
systems produce many
high power, point sources
of heat that would go undetected with a simple chassis
monitor.
400
340
320
300
280
280
300
TEMPERATURE (K)
320
340
360
380
TEMPERATURE (K)
Figure 1. Diode voltage VD vs temperature T(K) for
different bias currents
Figure 2. Reported uncompensated diode
temperature TD(K) vs temperature T(K) with series
resistance
Even PC processors feature dedicated
secondary fans in order to keep specific die
junction temperatures below an acceptable level. One line of defense against
overheating is to increase fan speeds, while
another is to temporarily disable the heat
source. In telecommunication systems and
other always-on applications, it is not
acceptable to disable the system, so the
only line of defense is to increase cooling.
The LTC2990 measures ambient and
remote temperature, plus voltage and
current, so the measurements are easily combined. Temperature sensors can
be diodes or transistor sensors—remote
sensor diodes are available as substrate
diodes in large microprocessors and
One problem with reactive cooling is that
large HVAC systems have lag—they require
time to reduce the ambient temperature.
Moreover, microprocessors and FPGAs
are embedded in chassis with surrounding thermal mass, which take even longer
to respond to a request for cooling.
Therefore it is important to monitor not
only the temperature, but also the rate of
temperature change in order to apply the
correction before temperatures escalate
to dangerous levels. An integrated power
and temperature monitoring system can
use changes in power consumption to
anticipate changes in temperature.
Figure 3. Single LTC2990 accurately monitors three
voltage rails and microprocessor temperature (via
substrate diode)
12V
5V
3.3V
0.1µF
2-WIRE
I2C
INTERFACE
10.0k
1%
30.1k
1%
10.0k
1%
10.0k
1%
VCC V1
SDA
SCL
ADR0
ADR1
µP
V2
LTC2990
GND
V3
470pF
V4
VOLTAGE, CURRENT AND TEMPERATURE CONFIGURATION:
CONTROL REGISTER: 0x58
REG 4, 5
0.0625°C/LSB
TAMB
V1 (+5)
REG 6, 7
0.61mVLSB
V2(+12)
REG 8, 9
1.22mV/LSB
REG A, B
0.0625°C/LSB
TPROCESSOR
REG E, F
2.5V + 305.18µV/LSB
VCC
design features
It is important to monitor not only the temperature,
but also the rate of temperature change in order to
apply corrections before temperatures escalate to
dangerous levels. An integrated power and temperature
monitoring system can use changes in power
consumption to anticipate changes in temperature.
FPGAs. The I2C serial interface provides
four addresses accommodating up to
four LTC2990s on the same bus.
where KD = 8.62−5, and knowing
ln(ID /IS) is always positive because
ID is always greater than IS, leaves
us with the equation that:
PRINCIPLE OF OPERATION
Measuring the absolute temperature of a
diode is possible due to the relationship
between current, voltage and temperature
described by the classic diode equation:
ID = I S • e(VD
η• VT )
or
I
VD = η • VT • ln D
IS
(1)
where ID is the diode current, VD is the
diode voltage, η is the ideality factor
(typically close to 1.0) and IS (saturation current) is a process dependent
parameter. VT can be broken out to:
VT =
k•T
q
where T is the diode junction temperature
in Kelvin, q is the electron charge and k is
Boltzmann’s constant. VT is approximately
26mV at room temperature (298K) and
scales linearly with Kelvin temperature.
It is this linear temperature relationship
that makes diodes suitable temperature
sensors. The IS term in the equation above
is the extrapolated current through a
diode junction when the diode has zero
volts across the terminals. The IS term
varies from process to process, varies with temperature, and by definition
must always be less than ID. Combining
all of the constants into one term:
KD =
η•k
q
VD = T(KELVIN) • K D • ln
ID
IS
Subtracting we get:
∆VD =
T(KELVIN) • K D • ln
I
I1
− T(KELVIN) • K D •ln 2
IS
IS
Combining like terms, then simplifying the natural log terms yields:
where VD appears to increase with temperature. It is common knowledge that
a silicon diode biased with a current
source has an approximately –2mV/°C
temperature relationship (Figure 1),
which is at odds with the equation. In
fact, the IS term increases with temperature, reducing the ln(ID /IS ) absolute value
yielding an approximately –2mV/deg
composite diode voltage slope.
To obtain a linear voltage proportional
to temperature we cancel the IS variable in the natural logarithm term to
remove the IS dependency from the
equation 1. This is accomplished by
measuring the diode voltage at two
currents I1, and I2, where I1 = 10 • I2),
∆VD = T(KELVIN) • K D • ln(10)
and redefining constant
K'D = K D • ln(10) =
198µV
K
yields
∆VD = K'D • T(KELVIN)
Solving for temperature:
T(KELVIN) =
∆VD
K'D
means that is we take the difference in
voltage across the diode measured at two
currents with a ratio of 10, the resulting
voltage is 198µV per Kelvin of the junction with a zero intercept at 0 Kelvin.
Table 1. Recommended transistors to be used as temperature sensors
MANUFACTURER
PART NUMBER
PACKAGE
Fairchild Semiconductor
MMBT3904
SOT-23
CMPT3904
SOT-23
CET3904E
SOT883L
Diodes, Inc.
MMBT3904
SOT-23
On Semiconductor
MMBT3904LT1
SOT-23
NXP
MMBT3904
SOT-23
Infineon
MMBT3904
SOT-23
Rohm
UMT3904
SC-70
Central Semiconductor
January 2011 : LT Journal of Analog Innovation | 23
Figure 4. High voltage current sensing
RSENSE
1mΩ
1%
ILOAD
0A TO 10A
VIN
5V TO 100V
(SURVIVAL TO 105V)
RIN
20Ω
1%
+IN
–INS
–INF
–
+
V–
V+
VREG
Thus, the equation describes a perfectly
linear, monotonically increasing temperature result provided that the current
ratio is constant, but arbitrary to the
absolute value of the currents. The two
independent diode voltages measured at
I1 and I2 both have negative temperature
dependence (~2mV/°C), but the diode
voltage at the larger bias current has a
slightly smaller negative slope, yielding
a positive composite ∆VD term (Figure 1).
Another way to think of it is that when
the junction is biased at a higher current, it is more probable (by a factor of
ln(I1 /I2) that a thermally generated carrier will have sufficient energy to exceed
the diode junction energy barrier. Using
this method, common diodes and transistors can be used as temperature sensors
over an operating range of –55ºC to 150°C,
typically limited by packaging materials.
One complication with the method
described above is the effect of series
resistance with the sensor diode. At
193µV/°C slope, it does not take much
Figure 5. Liquid level sensor
3.3V
µC
OUT
SDA
SCL LTC2990
ADR0
ADR1
GND
LTC6102HV
200k
1%
4.75k
1%
5V
ROUT
4.99k
1%
0.1µF
0.1µF
0.1µF
2-WIRE
I2C
INTERFACE
VCC
V1
V2
SDA
SCL LTC2990
ADR0
ADR1
GND
MMBT3904
V3
470pF
V4
ALL CAPACITORS ±20%
VOLTAGE, CURRENT AND TEMPERATURE CONFIGURATION:
CONTROL REGISTER: 0x58
REG 4, 5
0.0625°C/LSB
TAMB
REG 6, 7
13.2mVLSB
VLOAD
REG 8, 9
1.223mA/LSB
V2(ILOAD)
REG A, B
0.0625°C/LSB
TREMOTE
REG E, F
2.5V + 305.18µV/LSB
VCC
series resistance to yield artificially high
temperature readings due to the additional
voltage drop (the temperature would
always report falsely high). This series
resistance can be in the form of copper
traces and junction contact resistances.
Moreover, this resistance can have a temperature coefficient (copper is 3930ppm/°C)
yielding a temperature dependent additive
3.3V
SENSOR HI*
0.1µF
VCC
0.1µF
HEATER ENABLE
V1
V2
V3
V4
470pF
SENSOR LO*
SENSOR LO
470pF
TINTERNAL HEATER ENABLE
2 SECOND PULSE
CONTROL REGISTER: 0x5D
REG 4, 5
0.0625°C/LSB
TAMB
REG 6, 7
0.0625°C/LSB
THI
REG A, B
0.0625°C/LSB
TLO
REG E, F
2.5V + 305.18µV/LSB
VCC
24 | January 2011 : LT Journal of Analog Innovation
SENSOR HI
∆T = ~2.0°C pp, SENSOR HI
~0.2°C pp, SENSOR LO
NDS351AN
HEATER: 75Ω 0.125W
*SENSOR MMBT3904, DIODE CONNECTED
term. To combat this, multiple ∆VD measurements are made at multiple operating points, so the series resistance can be
calculated and compensated. The LTC2990
simplifies all of these complications, compensates for them and converts the diode
temperature straight to a digital result,
where it can be read over the I2C interface to a host microcontroller or FPGA.
IDEALITY FACTOR AND
COMPENSATION
The LTC2990 can report temperature in
units of degrees Celsius or Kelvin. Kelvin
temperature is valuable when fine-tuning
scaling calibration factors (η) of various
manufacturers’ devices. Since absolute
temperature is measured by silicon diodes,
the gain or slope of a sensor extrapolates
to absolute zero, or 0 Kelvin. An ideality
factor error of +1%, or 1.01, represents a
temperature error of 273.15 • 0.01 ≈ 2.7°C
at 0°C. At 100°C (398.15K), a 1% error in
ideality factor translates to an error of
approximately 4°C. The LTC2990 is factory
calibrated for an ideality factor of 1.004,
which is typical of the popular MMBT3904
NPN transistor. Transistor sensors are
made of ultra-pure materials, inherently
hermetic, small and inexpensive, making
them very attractive for –55°C to 125°C
applications. The linearity of transistor
sensors eliminates the need for linearization in contrast to thermocouples, RTDs
and thermistors. The semiconductor purity
and wafer-level processing limits deviceto-device variation, making these devices
interchangeable (typically < 0.5°C) for no
additional cost. Several manufacturers
design features
Figure 6. Fan/air filter/temperature alarm
MMBT3904
VCC
V1
where TACT and TMEAS are in Kelvin
degrees. To perform the scaling in Celsius degrees:
TACTUAL (°C) =
(TMEAS (°C) + 273.15) •
ηSENSOR
− 273.15
1.004
TEMPERATURE MONITORING
APPLICATIONS
Figure 3 illustrates a typical application where the LTC2990 is configured to
measure a substrate diode, which monitors the microprocessor temperature and
three system power supply voltages (12V,
5V and 3.3V). To extend the measurement range of the voltage inputs, resistive
voltage dividers are used for the 5V and
12V voltages. For this application the
0.1% accuracy of the LTC2990 introduces
negligible gain error over what is produced by the resistor divider network.
V3
22Ω
0.125W
470pF
V4
FAN
HEATER
TINTERNAL
TEMPERATURE FOR:
GOOD FAN
NDS351AN
HEATER ENABLE
2 SECOND PULSE
CONTROL REGISTER: 0x5D
REG 4, 5
TAMB
REG 6, 7
TR1
REG A, B
TR2
REG E, F
VCC
BAD FAN
accuracy, again dominated by the
1% precision of the external voltage
divider network components.
Figure 4 shows an example of high voltage
monitoring. The LTC6102HV is optimized
for accurate high-side current sensing.
Using a voltage divider for current sensing
would result in large gain errors, > 4%,
and low resolution for the current sensing
function. By attenuating the commonmode voltage with a voltage divider, the
differential voltage is attenuated by the
same factor. By making the sense resistor
larger to increase the gain, the power-loss
scales with the square of sense voltage.
Liquid Level Sensing
Figure 5 illustrates a simple application that uses temperature measurement to indicate liquid level. A heater
is pulsed, and the temperature sensor is
monitored for a corresponding change in
5V
The 14-bit resolution also allows the
use of larger voltage divider networks
while maintaining high resolution.
For example, compare the LTC2990 to
a device with a 10-bit dynamic range
that can measure 12V to 2% accuracy.
The LTC2990 can measure a 192V signal
with the same LSB weight (11.72 mV)
as the 10-bit part and maintain 2%
MMBT3904
V2
SDA
SCL LTC2990
ADR0
ADR1
GND
HEATER ENABLE
T
(K) • ηSENSOR
TACTUAL (K) = MEAS
1.004
FAN
0.1µF
2-WIRE
I2C
INTERFACE
If a target sensor ideality factor differs from 1.004, it can be compensated in the following manner:
22Ω
0.125W
470pF
3.3V
supply suitable transistors—some recommended sources are listed in Table 1.
3.3V
temperature. The measurement indicating the liquid level threshold is actually a
combined thermal conductivity and heat
capacity measurement, which is proportional to the change in temperature.
In operation, the remote temperature is
measured and stored for reference, after
which the heater is switched on, and given
a few seconds to heat the surroundings.
The temperature is again measured and
compared with the first temperature.
If the temperature difference is greater
than a preset threshold, the sensor is
determined to be above the liquid level.
If the sensor is submerged in the liquid,
the relatively larger heat capacity of the
liquid prevents the temperature from
rising quickly. The smaller the discernable temperature change, the less heater
power is required for detection. For this
0.1µF
VCC
µC
0.0625°C/LSB
0.0625°C/LSB
0.0625°C/LSB
2.5V + 305.18µV/LSB
V1
MMBT3904
V3
∆T
470pF
470pF
V4
TDRY
TINTERNAL
CONTROL REGISTER: 0x5D
REG 4, 5
TAMB
REG 6, 7
TWET
REG A, B
TDRY
REG E, F
VCC
MMBT3904
V2
SDA
SCL LTC2990
ADR0
ADR1
GND
FAN: SUNON
KDE0504PFB2
0.0625°C/LSB
0.0625°C/LSB
0.0625°C/LSB
2.5V + 305.18µV/LSB
TWET
DAMP MUSLIN
FAN
5V
FAN ENABLE
WATER
RESERVOIR
NDS351AN
Figure 7. Wet bulb psychrometer
January 2011 : LT Journal of Analog Innovation | 25
Figure 8. Current sensing with inductor
parasitic resistance
INDUCTOR WITH
RPARASITIC
“QUIET” NODE
ILOAD
5V
1µF
BUCK REGULATOR
5V
2.1k
2.1k
1µF
THERMAL
COUPLING
1µF
1µF
VCC
2-WIRE
I2C
INTERFACE
application, filtering the remote temperature sensor can trade time for power.
SDA
SCL
CA0
CA1
V1
V2
V3
LTC2990
GND
Airflow Measurement
Airflow can also be measured by monitoring temperature. Figure 6 illustrates a
method using a heater and a temperature
sensor similar to the liquid level application. In this application the cooling
power of the fan is tested by turning on a
small heater and measuring the temperature rise, or the rate of temperature rise
with the remote sensor. In the absence
of cooling air, both the absolute and
the rate of temperature rise increases.
This method can be used to detect
faulty fans, or dust buildup on air filters. Whatever the cause, the circuit
can signal inadequate cooling conditions. Thermistors are undesirable in
this application because their change
in resistance is not consistent over a
broad operating temperature range.
Temperature monitoring can signal the
alarm for overheating, but simple temperature monitoring cannot predict overheating. By measuring power (voltage and
current) and cooling capacity, one can
predict a problem prior to a catastrophic
failure. This is important, because it
takes time to correct an over-temperature
condition due to the heat capacity of the
system and its immediate evironment.
Humidity Measurement
Humidity can also be measured using
temperature monitoring as shown in
Figure 7. One can implement a humidity sensor in the form of a psychrometer.
A psychrometer uses two temperature
26 | January 2011 : LT Journal of Analog Innovation
V4
MMBT3904
RPARASITIC ~3930ppm/°C
sensors to detect humidity: one of them
is dry and acts as a reference; the other
is dampened and exposed to airflow. The
cooling effectiveness of the water on the
wet sensor is a function of humidity. In a
100% humidity environment, the forced
air on the wet sensor yields no evaporation and thus yields no cooling effect.
Conversely, in an arid environment, the
cooling due to the heat of evaporation can
cool the “wet bulb” temperature sensor
significantly. The dry temperature sensor
reads the same with or without airflow.
The temperature difference function is
non-linear, and commonly implemented
with lookup tables in a host microprocessor. Thus the temperature difference
between the wet and dry temperature
sensor in the presence of air movement
is an indirect measurement of humidity.
CURRENT SENSING WITH
PARASITIC RESISTANCE
The application circuit in Figure 8 uses
the LTC2990 as a current monitor. The
sense resistor in this application is the
parasitic resistance in a buck switching
Figure 9. Example pseudo-code for an FIR filter
//FIR digital filter example (Moving Average Filter)
#define filter_dim 16
int16 FIR_temp[filter_dim]; //memory allocation scales with filter size!
int8 i = 0;
int8 j;
int32 accumulator;
int16 filtered_data;
// Moving Average filter for ltc2990 temperature
// Reduces noise by factor of sqrt(filter_dim), or in this case ~4
if((ltc2990_temperature && 0x1000) == 0x1000)
FIR_temp[i++%filter_dim] = ltc2990_temperature | 0xE000; //sign extend data
else
FIR_temp[i++%filter_dim] = ltc2990_temperature & 0x1FFF; //strip off alarms & data_valid
accumulator=0; //cleared each pass through filter routine
for(j=0; j<filter_dim; j++)
accumulator += FIR_temp[j];
filtered_data = (int16)(accumulator/filter_dim); //could use >>4, where 4 = log2(filter_dim)
design features
Temperature sensors can be diodes or transistor sensors.
Remote sensor diodes are available as substrate
diodes in large microprocessors and FPGAs.
regulator. At the output of the buck
regulator is the switching node, which
typically toggles between VCC and ground.
The average value of this voltage is the
output regulated voltage. The load current
runs through the power supply inductor,
which has a series parasitic resistance.
This parasitic resistance is typically
small and is minimized in the power
supply design to maximize efficiency.
The RC filter across the inductor into the
LTC2990 V1 and V2 pins filters out the
transitions seen on the switching node.
The quiet node is equivalently filtered to
maintain circuit balance due to LTC2990
input common-mode sampling currents.
Knowing RPARASITIC and V1 – V2, the load
current can be calculated. Moreover,
VCC is measured by the LTC2990, so load
voltage and load current are known;
thus load power can be calculated.
Because RPARASITIC is typically copper, it
has a temperature coefficient of resistance
(TCR) of ~3930ppm/°C. By measuring
the inductor temperature, this relatively
large error source can be compensated by
introducing a temperature dependent gain
coefficient inversely proportional to the
resistor TCR. Knowing the load power, the
inductor temperature and ambient temperature from the LTC2990 internal temperature sensor, you can predict the rise
in temperature of the inductor for various
load currents. This can be important to
avoid inductor core saturation at high
Figure 10. Example pseudo-code for an IIR filter
temperatures, which can be a potentially
catastrophic event to the buck regulator.
MEASUREMENT ACCURACY
AND NOISE
The LTC2990 can measure temperatures at
a rate of ~20Hz. This allows the designer
to trade resolution and noise performance
for speed. At 20Hz, the noise is ~1.2°C
peak to peak, or ~0.2°C RMS. For most
board level monitoring applications this
is excellent performance, though there
are applications that require lower noise
levels, which can be obtained by controlling the measurement bandwidth. The
temperature data output is digital, so this
requires the band limiting function to be
in the form of a digital filter. Example
filters and their simulated performance
Figure 11. Simulated IIR filter response
1.8
1.6
#define filter_coefficient 16 //a power of 2 here can speed up filter
int8 j;
int16 filtered_data;
static int32 accumulator = 0; //GLOBAL, only cleared at boot time. Does not change with filter growth!
// implements: filtered_temperature = (filtered_temperature*(filter_coefficient-1) +
// ltc2990_temperature)/filter_coefficient
for(j=0;j<(filter_coefficient-1);j++)
//multiply by repeated add resulting in accumulator = filter_coefficient-1
accumulator += accumulator;
TEMPERATURE DATA (°C)
//IIR digital filter example (higher averaging for limited ram application)
1.4
1.2
1
0.8
0.6
0.4
FILTERED DATA
SIMULATED DATA
0.2
0
0
1
2
3
4 5 6
TIME (s)
7
8
9
10
//add the latest LTC2990 output to the accumulator once
if((ltc2990_temperature && 0x1000) == 0x1000)
accumulator = ltc2990_temperature | 0xE000; //sign extend data
else
accumulator = ltc2990_temperature & 0x1FFF; //strip off alarms & data_valid
accumulator >>= 4; // where 4 = log2(filter_coefficient)
filtered_data = (int16)accumulator;
January 2011 : LT Journal of Analog Innovation | 27
The LTC2990 serves up the results with 14-bit resolution via I2C. Its
small package size, integrated voltage reference and 1µA shutdown
current are ideal for portable electronics applications.
RSENSE
2.5V
ILOAD
5V
VCC
SDA
SCL
ADR0
ADR1
V2
V1
V3
LTC2990
TREMOTE
V4
GND
TINTERNAL
MEASURES: TWO SUPPLY VOLTAGES,
SUPPLY CURRENT, INTERNAL AND
REMOTE TEMPERATURES
Figure 12. Temperature compensated
current sense resistor
for equal over-sampling ratios are
illustrated in Figures 9 through 11.
The LTC2990 measurement resolution is
14-bit for voltages and 15-bit for currents. The monitor contains an internal
reference with 10ppm/°C stability, requiring no external support components.
Ground referenced single-ended voltages
can be measured in a range of zero volts
to VCC + 0.2V, (4.9V max), and differential voltages in a range of ±300mV with
a common mode voltage range of zero
volts to VCC + 0.2V, which is suitable for
current sensing and bridge circuits.
Scaling single-ended, ground referenced
voltages is common practice using
standard voltage dividers with precision resistors. Sensing current with high
accuracy requires more attention to detail.
In the case of current measurements, the
external sense resistor is typically small,
and determined by the full-scale input
voltage of the LTC2990. The full-scale
differential voltage is 0.3V. The external
sense resistance is then a function of
the maximum measurable current, or:
REXTERNAL(MAX ) =
0.3V
.
IMAX
THE FINE POINTS OF CURRENT
SENSE MEASUREMENT
If you wanted to measure a current
range of ±10A, the external shunt resistance would equal 0.3V/10A = 30mΩ. This
resistance is fairly small, and one may be
tempted to implement this resistor using
a thin copper trace on the printed circuit
board. The dimension of this resistor is
determined by the bulk resistance of the
PCB copper, the thickness of the copper
clad sheet, the length and width of the
copper trace. PCB clad material thickness
is specified by weight in units of ounces
per square foot. Typical copper thicknesses are ½, 1, and 2 oz, corresponding
to 0.7, 1.4 and 2.8 mils thickness, respectively. When multi-layer printed circuit
boards are manufactured, via holes are
electroplated. This electroplating process,
also adds copper thickness to the outer
copper layers or the PCB. Even if the thickness of the copper clad on the PCB stock
30mΩ, 0.5%, <50ppm/°C
4-TERMINAL, SURFACE MOUNT
WITH KELVIN CONNECTIONS
OHMITE RW1/RW2 SERIES
VISHAY VCS1625 SERIES
Figure 13. Current sense connections
SENSE
RESISTOR
FOR ~10A
W = 0.097''
L = 6''
1oz Cu
INTERNAL TRACE
FROM
POWER
30mΩ
30mΩ, 1%, 75ppm/°C
2512 SURFACE MOUNT
SENSE RESISTOR
FOR ~10A
FROM
POWER
2512
SENSE RESISTOR
FOR ~10A
FULL-SCALE APP
FROM
POWER
TO
LOAD
VIA
4-MIL TRACES
TO V1 AND V2
4-MIL TRACES
TO V1 AND V2
~4-MIL TRACE SENSE
LINES TO V1 AND V2
WORST
BETTER
BEST
COST: LEAST EXPENSIVE
TOLERANCE: WORST ~±30%
TCR: WORST ~±400ppm/°C
COST: LOW
TOLERANCE: GOOD ~±2%
TCR: GOOD ~75ppm/°C
COST: HIGH
TOLERANCE: EXCELLENT ±0.5%
TCR: EXCELLENT <10ppm/°C
28 | January 2011 : LT Journal of Analog Innovation
design features
RESISTANCE
TOLERANCE (%)
SENSE RESISTOR TYPE
Copper Trace
• R Not calibrated
20
• TCR Not calibrated
2-Terminal Discrete Resistor
• R Not calibrated
2
• TCR Not calibrated
4-Terminal Precision Discrete Resistor
• R Not calibrated
0.5
• TCR Not calibrated
Copper Trace
• R Calibrated & Compensated
0.025
• TCR Calibrated & Compensated
2-Terminal Discrete Resistor
• R Calibrated & Compensated
0.025
• TCR Compensated
4-Terminal Precision Discrete Resistor
• R Calibrated & Compensated
0.025
• TCR Compensated
4-Terminal Precision Discrete Resistor
• R Calibrated & Compensated
0.025
• TCR Calibrated & Compensated
TCR % FOR 50°C
RISE, (ppm)
TOTAL ERROR %,
(BITS PRECISION)
20,
40,
(3970)
(1.3)
0.375,
2.375,
(75)
(5.4)
0.05,
0.55,
(10)
(7.5)
0.5,
0.525,
(3970 ±100)
(7.5)
0.375,
0.4,
(75)
(8.0)
0.05,
0.075,
(10)
(10.4)
0.005,
0.075,
(x ±1)
(11.7)
Table 2. Current sense resistor
precision comparison table
changes with temperature. Assuming that
the current through the sense resistor produces negligible self-heating over a –40°C
to 85°C temperature range, the copper
resistance changes about 50%. If the sense
resistor does heat itself, there is a nonlinear current-to-voltage distortion in the
measurement. For this reason, there are
special sense resistors manufactured with
low TCR values (Figure 13). If the temperature rise in the sense resistor is large due
to large currents, even small TCRs can yield
large measurement errors. The LTC2990
can be used to track the sense resistor
temperature so its TCR can be compensated
for, improving measurement accuracy.
CONCLUSION
material is well controlled, the thickness
of the trace will have a manufacturing
variable due to plating thickness, when
plating the via holes. The copper thickness uncertainty impacts the sense resistor
Figure 14. Temperature compensated
copper trace resistor
value, and hence the resulting differential
volts/amp that the LTC2990 measures.
Copper has a relatively high temperature
coefficient of resistance (TCR), with a value
of ~3930ppm/K. The TCR of copper also
30mΩ
SENSE
RESISTOR
FOR ~10A
FULL-SCALE APP
W = 0.097''
L = 6''
1oz Cu
INTERNAL TRACE
2N3904
CALIBRATE R AND TCR,
COMPENSATE IN SOFTWARE
TO LTC2990
V3 AND V4
TEMPERATURE
MODE
FROM
POWER
TO
LOAD
VIA
4-MIL TRACES
TO V1 AND V2 CURRENT MODE
COMPROMISE
The LTC2990 is able to measure electrical power (via voltage and current) and
temperature and serve up the results with
14-bit resolution via I2C. Combo power
and temperature measurements are commonly used for industrial control and
fault monitoring applications, including
air and fluid flow, liquid level, over/under
temperature, power sharing and limiting,
redundancy management, alarm generation, nonvolatile memory write/erase
protection, and countless others. The small
package size, integrated voltage reference
and 1µ A shutdown current are ideal for
portable electronics applications. Remote
diode sensors are available in extremely
small packages (Central Semiconductor
CET3904E: 1.05mm × 0.65mm) allowing
for fast thermal response times, taking
advantage of the 50ms temperature measurement capabilities of the LTC2990. n
COST: MEDIUM (INCLUDING CALIBRATION)
TOLERANCE: EXCELLENT ~0.025%
TCR: EXCELLENT ~±2000ppm/°C
January 2011 : LT Journal of Analog Innovation | 29
Isolated Data Transmission and Power Conversion
Integrated Into a Surface Mount Package
Keith Bennett
Galvanic isolation is used in a variety of industries, most commonly to provide safety
against potentially lethal voltages. Isolation is also used to eliminate the effects of noise
and common mode voltage differences created by ground loops, or as a level shifter
between dissimilar operating voltages. Typically, building an isolated system requires
a number of passive and active components on either side of the isolation barrier, in
addition to the barrier components themselves. These barrier components can be
optical, magnetic, capacitive, RF or even GMR (giant magnetoresistance) devices.
Barrier components are notoriously difficult to use, adding significant design
time and cost to isolated systems. With
this in mind, Linear Technology has
developed a line of µModule® isolators that reduces the design of isolated
systems to simply plugging in a module—no complex barrier components
are required. In fact, µModule isolators
require no external components at all.
Linear Technology µModule isolators
use magnetically coupled techniques to
provide data and power isolation in a
single package. The result is an easy-touse, low power, robust solution with
excellent field immunity. The LTM2881
isolated RS485/RS422 transceiver and
LTM2882 dual isolated RS232 transceivers use µModule isolator technology
to provide complete transceiver-pluspower solutions in 15mm × 11.25mm
× 2.8mm surface mount packages.
ISOLATED DATA TRANSMISSION
µModule isolators use inductively coupled coils, or coreless transformers, to
pass data across the isolation boundary.
Electrical and mechanical representations are illustrated in Figure 1. Dedicated
integrated circuits perform the data
transmission and receiving functions for
all channels and both data directions.
Three channels of data are encoded,
serialized into a packet, and transmitted
Figure 1. Data isolation equivalent electrical and mechanical configuration
Rx
Tx
ENABLE
DATAIN
DATAOUT
RCVR
ENABLE
Tx
Rx
DATAIN
RCVR
30 | January 2011 : LT Journal of Analog Innovation
DATAOUT
across the isolation boundary in each
direction. Data transmission in one direction is completely independent of data
transmission in the other direction.
The encoding process is initiated when
the data changes (edge triggered) on
any of the three inputs. The three data
inputs consist of one high priority and
two low priority channels. A state change
on the high priority channel preempts
an encoding process initiated by a low
priority channel state change. This scheme
ensures that the data on the high priority channel is transmitted with no jitter
to the output, but in turn produces some
amount of timing uncertainty on the
design features
Linear Technology has developed a line of µModule isolators
that reduces the design of isolated systems to simply plugging
in a module—no complex barrier components are required. In
fact, µModule isolators require no external components at all.
LDO
3.9
VCC2
6
-3 VERSIONS
VIN (V)
3.6
5.5
-5 VERSIONS
3.3
5
3
GND
GND2
Figure 2. Simplified isolated power conversion stage
low priority channels. The high priority
channel is assigned to DI to (Y,Z) and (A,B)
to RO for the LTM2881, and from T1IN to
T1OUT and R1IN to R1OUT for the LTM2882.
All remaining channels for these products
are low priority. Encoding is completed by
sampling all three input data states and
transmitting the data across the coreless
isolation transformer as a series of pulses.
Each transmission is received as a differential signal, checked for errors,
decoded, and clocked to the associated
data output. The receiver is connected
to a center tapped transformer secondary. This winding arrangement provides
common mode rejection since any coupled
signals are cancelled by the opposing winding geometry at the receiver
input. The error checking process determines if the transmitted data is valid;
if not, the outputs are not updated.
The encoding/decoding process supports a
minimum data throughput rate of approximately 21Mbps. Edge events occurring
during a packet transmission initiate a
new capture operation, which is completed
at the end of the current transmit cycle.
Data is refreshed at a rate of approximately 1MHz, ensuring DC correctness of
all data outputs. If four invalid packets are
received in succession then a communication fault is generated. This fault mode
forces a high impedance state on certain
outputs, for instance RO and DOUT for
the LTM2881, and R1OUT and R2OUT for
the LTM2882. This fault condition can be
easily detected in critical applications.
The coreless transformer is fabricated
within the µModule substrate’s inner
layers. Minimum coil layer separation of 60µm is provided by a 2-ply
Bismaleimide-Triazine (BT) high performance resin-based laminate.
ISOLATED POWER TRANSMISSION
Isolated power is generated by more
conventional means. The overall power
converter consists of a full-bridge square
wave oscillator AC-coupled to an isolation transformer primary, rectified with
a full wave voltage doubler connected
2.7
VIN (V)
VCC
4.5
VCC2 ≥ 4.75V
0.4
0.6
0.8
1
1.2 1.4
OUTPUT POWER (W)
1.6
1.8
4
Figure 3. µModule isolator power capability
to the transformer secondary, and
post regulated by a low-dropout linear
regulator (LDO). Figure 2 schematically
shows the power conversion stage.
The overall power topology provides
a simple, flexible, fault tolerant and
relatively efficient design (~65%). Bridge
current is monitored and limited to
protect the power switches and transformer. The primary and secondary are
both AC-coupled, preventing transformer
saturation under any condition. All
components are integrated within the
µModule package; no external decoupling is required for proper operation.
Two input operating voltage ranges,
determined by transformer turns ratio,
are available: 3V to 3.6V (-3 versions) and
4.5V to 5.5V (-5 versions). Input voltage level is internally sensed, setting the
primary current limit to approximately
550m A for -3 part versions and 400m A for
-5 part versions. The converter output
power capability versus input voltage
is shown in Figure 3 for VCC2 ≥ 4.75V.
January 2011 : LT Journal of Analog Innovation | 31
PARAMETER
CONDITIONS
MIN
Rated Dielectric Insulation Voltage
1 Minute
2500
V RMS
Continuous
400
V RMS
560
V PEAK
Partial Discharge
V PR = 1050V PEAK
Common Mode Transient Immunity
MAX
5
30
kV/µs
Ω
f = 1MHz
6
pF
External Tracking (Creepage) Distance
L/BGA
9.53
mm
External Air Gap (Clearance) Distance
BGA
9.38
mm
V IO = 500V
Input to Output (Barrier) Capacitance
Comparitive Tracking Index (CTI)
Highest Allowable Overvoltage
175
Isolation Barrier ESD, HBM
V
t = 10s
4000
Minimum Distance Through Insulation
0.06
V PEAK
mm
(V CC2 , GND2) to GND
±10
kV
Isolated I/O to GND
±8
kV
Primary current limit is not active
under normal operating conditions.
and components, and related topics of interest are listed in Table 2.
The transformer comprises a toroidal ferrite core with one high voltage insulated
winding and a second low voltage insulated winding. The high voltage winding
uses supplementary-rated Teflon-insulated
wire, consisting of two independent
layers with a total insulation thickness
of 76µm. The transformer is completely
encapsulated during the µModule molding
process, providing additional protection.
UL1577 is a component level standard
designed to verify that the dielectric withstands voltages of up to 2500VRMS under
a variety of environmental conditions.
The LTM2881 and LTM2882 devices have
achieved UL approval up to an operating temperature of 100ºC. The dielectric withstand is 100% production
tested by applying a DC test voltage of
±4400V (≈2500 • 1.2 • √2) for one second
at each polarity, with all pins shorted
on each side of the isolation barrier.
ISOLATION BARRIER PERFORMANCE
As detailed above, the isolation barrier
consists of two components: data coils
insulated by BT substrate, and Teflon
insulated windings within the power
transformer. The isolation barrier is
designed to have a minimum dielectric withstand rating of 2500VRMS for
sixty seconds, and continuous working voltage of 400VRMS or 560VPEAK .
Key isolation parameters are listed
in Table 1, adopted from a wide
variety of international standards.
Standards relevant to isolation systems
32 | January 2011 : LT Journal of Analog Innovation
Table 1. Key isolation
specifications of the
LTM2881 and LTM2882
pC
10 11
Input to Output (Insulation) Resistance
10 9
UNITS
IEC 60747-5-2 is the European equivalent
component level standard, requiring a
measurement of partial discharge (PD)
across the isolation barrier at a voltage related to the rated system working
voltage. Tests under a variety of environmental conditions are required for
certification, as well as 100% production
screening. The LTM2881 and LTM2882
have been characterized for PD and easily
meet the requirements of the standard,
with less than 5pC at 1050VPEAK for a
working voltage of 560VPEAK . Certification
to IEC 60747-5-2 is in process.
The rated continuous working voltage is
not explicitly defined in any standard; its
value is dependent on a variety of environmental operating conditions, including creepage/clearance, test requirements
defined within the standards, as well as the
operational lifetime. The operating lifetime
at the rated working voltage is extrapolated from accelerated lifetime test data.
Tests results for the LTM2881 and LTM2882
are shown in Figure 4. Data conforms to
the generally accepted Weibull distribution
for accelerated dielectric breakdown testing.1 The plot shows the minimum lifetime
at 500VRMS is greater than 100 years.
Parameters related to the isolation barrier voltage rating include electrostatic
discharge (ESD), surge immunity, and
electrical fast transients. One of the major
benefits of galvanic isolation is the ability of the isolation barrier to hold off
large voltage potentials, eliminating the
need for other protection devices such as
1G
1M
LIFETIME (HOURS)
Maximum Working Insulation Voltage
TYP
MEAN
1k
MINIMUM
1
0.001
0.1
1
VAC(RMS) (kV)
10
Figure 4. µModule isolator insulation lifetime
design features
Table 2. Standards
relevant to isolation
systems and components
transient voltage suppressors. The key is
configuring the system in a way that the
barrier sees the transient event, which
is typically achieved by proper shielding of system interconnects. The isolation barrier easily withstands transient
events equal to the dielectric withstand
voltage of 2500VRMS or 3500VPEAK, and
can withstand much higher voltages for
short periods of time as evidenced by
the 8kV–10kV barrier ESD ratings, and
operating lifetimes at elevated voltages.
The input to output capacitance, or barrier capacitance, is an important parameter impacting many aspects of overall
performance. In general, the lower the
capacitance the better. The capacitance
is a parasitic element and is the parallel
combination of data coil and transformer
winding capacitance. Typical capacitance is 6pF at 1MHz with the data coils
accounting for 1.2pF each and the power
transformer 3.6pF. In the case of transients, one or more ESD or body diodes
conducts the transient energy to the
barrier capacitance and back to ground.
A smaller barrier capacitor absorbs more
transient voltage, reducing the energy
dissipated in the functional IC, thus
minimizing the potential for damage.
The magnitude of capacitance also
impacts the common mode transient
immunity. This parameter is a measure of
the component’s ability to maintain proper
function in the presence of high slew
rate signals across the isolation barrier.
µModule isolators support a minimum
common mode transient rate of 30kV/µs
(50kV/µs typical), operating through these
STANDARD
DESCRIPTION
UL1577
Standard for Safety, Optical Isolators
IEC 60747-5-2 (VDE 0884-10)
Optoelectronic Devices, Essential Ratings and Characteristics
IEC 60664-1
Insulation Coordination for Equipment within Low-Voltage Systems
IEC 60950-1
Information Technology Equipment—Safety
IEC 61010-1
Measurement, Control, Laboratory Equipment—Safety
IEC 60601-1
Medical Electrical Equipment
IEC 61000-4-2
Electrostatic Discharge Immunity
IEC 61000-4-3
Radio Frequency Field Immunity
IEC 61000-4-4
Electrical Fast Transients
IEC 61000-4-5
Surge Immunity
IEC 61000-4-8
Power Frequency Magnetic Field Immunity
IEC 61000-4-9
Pulse Magnetic Field Immunity
CISPR 22
Radiated Emissions - Information Technology Equipment
IEC 60079-11
Intrinsic Safety
transient events error-free while transmitting data. The barrier capacitance displaces a current equal to the capacitance
times the slew rate into the application
die, potentially disrupting its function
by inducing noise or triggering parasitic
device structures. Under the extreme conditions needed to produce common mode
errors, µModule isolator products do not
exhibit any latch-up, instead showing
only momentary state changes corrected
during the next communication cycle.
ELECTROMAGNETIC COMPATIBILITY
The barrier capacitance also plays a role
in the system’s electromagnetic compatibility, specifically radiated emissions.
Just as transient common mode events
inject current through the parasitic barrier
capacitance, so do the data and power
drive circuits. These drive circuits have fast
edge rates which ultimately create a voltage transient on the isolated ground plane
(GND2). Most isolated printed circuit board
layouts use separate ground planes for
the input (GND) and output (GND2) sides.
This dual ground plane structure forms
a dipole antenna, providing an effective
radiator of the common mode voltage created by the parasitic barrier capacitance.
Another way to look at this problem is
to consider the currents generated in the
parasitic capacitance by the drive circuits.
These currents require a return path.
If the current is not returned through
one of the parallel parasitic capacitors,
it will be returned through the capacitance created by a pair of ground planes,
or potentially through the capacitance
formed between interconnect wires and
ground. The radiated emissions can be
quite high if not properly mitigated.
The most effective technique is to provide a low impedance return path at
the frequencies of interest—namely an
additional capacitor between the isolated
ground planes, or bridge capacitor.
The structures and isolation techniques
used in isolator µModule technology
are insufficient in themselves to radiate enough power to exceed the Class
B limit of CISPR 22. The data coils are
essentially small loop antennas, whose
January 2011 : LT Journal of Analog Innovation | 33
The center tapped receiver coils provide a high level of
rejection to external RF and magnetic fields or common
mode signals, due to the opposing winding turns.
60
CISPR 22 CLASS A LIMIT
50
•Insure that all signal and power traces
have closely coupled return paths to
minimize radiation created by these
localized loops. Avoid transitioning signals from layer to layer as the image current often cannot follow the same path.
FCC CLASS A LIMIT
AMPLITUDE (dBµV/m)
40
CISPR 22 CLASS B LIMIT
30
20
10
PREDICTED
DATA COIL
EMISSIONS
0
−10
Figure 5. Predicted
versus actual total
radiated emissions
LTM2881 LOW-EMI DEMO BOARD EMISSIONS
RBW = 120kHz
VBW = 300kHz
DETECTOR = PEAK
SWEEP TIME = 17s
10
100
FREQUENCY (MHz)
N
∑ (rn λ )
4
WATTS
(1)
n=1
If = current at frequency
rn = radius of nth coil turn in meters
N = Total number of turns
λ = wavelength at frequency
As shown, the emission levels fall far
below the required CISPR 22 and related
FCC limits.
Additional EMI mitigation techniques
include:
•Minimize the size of the isolated ground
plane.
100
1k
C
AN
ST
DI
=
1m
=
10
1
100
E
CE
AN
ST
DI
10
0m
10
m
E
=
0.1
m
5m
1
0.01
0.001
0.1
CURRENT
FLUX DENSITY
1
10
100
1k
10k
MAGNETIC FIELD FREQUENCY (kHz)
0.01
100k
•Use filtering for all off-board interconnection. This can often be done with
ferrite based chip beads and common
mode filters. Care must be exercised
to ensure the filter components do not
compromise the integrity of signals on
the associated data lines. The filtering
provides a high impedance block to
radio frequency signals. Shunt capacitance can also be used to provide a
local low impedance return path.
•Reduce the operating supply voltage, or
use the lower input voltage version (-3)
of the part.
•Low-EMI versions of the LTM2881 and
LTM2882 demo boards, DC1746A and
DC1747A respectively, are available,
leveraging many of these techniques.
Figure 6. Magnetic field immunity
C
AN
ST
34 | January 2011 : LT Journal of Analog Innovation
with discrete capacitors to further reduce
emissions below 300MHz, above which
frequency the parasitic inductance limits
their usefulness. Recommended discrete
capacitors are AC safety rated, Y2 type,
applied two in series to meet safety standard requirements. Murata’s Type GF series
of Y2 capacitors meet this requirement.
DI
Several techniques can be used to ensure
a design with minimal radiated emissions. The first, as mentioned, is the use
of a bridge capacitor, implemented by
overlapping a floating plane of printed
circuit board copper, typically using an
inner layer, over the input and isolated
ground planes. This provides a nearly ideal
capacitor, and two insulation barriers are
created to support the isolation voltage.
This technique can be used in combination
1k
MAXIMUM ALLOWABLE CURRENT (kA)
−30
MAXIMUM ALLOWABLE FLUX DENSITY (kGauss)
2
•Add a combination of low ESL and high
impedance decoupling capacitors to
high current power rails, preventing
conducted noise and parasitic ringing from becoming radiated noise.
−20
emissions level can be predicted by solving
the radiated power loop antenna equation (1). Figure 5 shows the results of this
equation using simulated coil current
spectrum data as well as actual measured
data, which also includes contributions
from the isolated power component.
PRAD = 160π 6 (If )
FCC CLASS B LIMIT
ELECTROMAGNETIC IMMUNITY
By superposition, an inefficient radiator is also an inefficient receiver. The
LTM2881 and LTM2882 have been independently evaluated for radio frequency
and magnetic field immunity. Table 3
summarizes the applicable test standards
with passing field strength levels.
design features
The LTM2881 RS485 transceiver and the LTM2882 dual RS232 transceiver
combine desirable data-plus-power isolation features in a robust small-package
system. Both devices feature superior transient common mode rejection, magnetic
field immunity, ESD and transient barrier withstand, and insulation lifetime.
Passing the test requirements, however,
does not give much insight to the system’s
true immunity level. The external magnetic
field required to corrupt data communication can be calculated by equation (2),
where V represents the differential receiver
threshold voltage. The center tapped
receiver coils provide a high level of rejection to external RF and magnetic fields or
common mode signals, due to the opposing winding turns. Since the data coils are
not completely symmetric, some differential voltage is generated, represented
by the net difference in area between
the center tap coil turns. The maximum
external magnetic field relative to receiver
threshold is plotted in Figure 6, taking into
account the common mode cancelation.
TEST
FREQUENCY
FIELD STRENGTH
80MHz–1GHz
10V/m
1.4MHz–2GHz
3V/m
2GHz–2.7GHz
1V/m
IEC 61000-4-8 Level 4
50Hz and 60Hz
30A/m
IEC 61000-4-8 Level 5
60Hz
100A/m (non IEC method)
IEC 61000-4-9 Level 5
Pulse
1000A/m
IEC 61000-4-3 Annex D
Table 3. Electromagnetic field immunity
IEC 60664-1 deals specifically with insulation systems and is a reference standard to IEC 60950-1, IEC 61010-1, and
IEC 60601-1. These standards define the
required creepage, clearance, etc., based
on the type of equipment installation
and operating environment. The following terms and definitions are used
throughout the various standards:
•Material Group: Classification based
on an insulating components relative
comparative tracking index, a measure
of the surface electrical breakdown
properties of an insulating material.
N = number of turns in receiving coil
rn = radius of nth coil turn in centimeters
•Basic Insulation: Insulation to provide
basic protection against electric shock.
•Pollution Degree: Numeral characterizing the expected pollution of the
microenvironment, degrees include 1, 2
and 3. The level of pollution can result
in a reduction of electric strength or
surface resistivity of the insulation.
Alternatively, the magnetic field due to
an alternating current flowing in a wire
some distance away can be calculated.
Figure 6 also plots this result for distances
of 5mm, 100mm, and 1m away from the
device in question. For example, a current of 1000A running at 1MHz in a wire
5mm away would be required to corrupt data transmission at the receiver.
•Supplementary Insulation: Independent
insulation applied in addition to
basic insulation in order to reduce
the risk of electric shock in the event
of a failure of the basic insulation.
•Overvoltage Category: Numeral defining
a transient overvoltage condition; categories include I, II, III and IV. Operating
category is a function of the highest
RMS operating voltage within the system.
•Double Insulation: Insulation comprising both basic insulation and
supplementary insulation.
•Creepage: Shortest distance along the
surface of a solid insulating material between two conductive parts.
•Reinforced Insulation: Single insulation
system that provides a degree of protection against electric shock equivalent to
double insulation. It may comprise several layers that cannot be tested as basic
insulation and supplementary insulation.
•Clearance: Shortest distance in air
between two conductive parts.
N
 dβ 
2
V = −  ∑ π (rn ) VOLTS
 dt  n=1
(2)
β = magnetic flux density in gauss
SAFETY STANDARDS
Creepage, clearance, tracking index, and
minimum distance through insulation
are safety related parameters used in the
various equipment level standards and
insulation coordination standards to
determine proper component application.
A complete interpretation of IEC 60664-1
and related standards as they apply to
µModule isolators is beyond the scope of
this article. Neverthless, Table 4 summarizes some of the key parameters for
January 2011 : LT Journal of Analog Innovation | 35
Table 4. Isolation categories
PARAMETER
CONDITION
SPECIFICATION
Basic Isolation Group
Material Group, 175 ≤ CTI < 400
IIIa
Rated Mains Voltage ≤ 150V RMS
I–IV
Rated Mains Voltage ≤ 300V RMS
I–III
Rated Mains Voltage ≤ 400V RMS*
I–II
Installation Class
*Range is normally 600V RMS, limited by devices rated working voltage
µModule isolators often listed on component data sheets. It is worth noting that
the creepage and clearance distances of
the LTM2881 and LTM2882 modules greatly
exceed the requirements for the rated
working voltage for all insulation categories, pollution degrees, overvoltage categories, and material groups. For components
classified with a Basic Insulation level
there is no minimum for distance through
insulation. Supplementary and Reinforced
Insulation systems require a distance of
400µm, or they must meet the type (conditioned sample) and routine (production)
test requirements for partial discharge
and/or dielectric withstand. µModule
isolators are considered Basic Insulation
systems at the rated working voltage.
and reactive component energy storage
to prevent the generation of sparks. The
LTM2881 and LTM2882 are suitable for
level of protection “ic” for peak voltages up to 60V. These products meet
all of the requirements for protection
levels “ia”, “ib”, and “ic” except for
distance through solid insulation.
ESD, EFT AND SURGE
Electrical transients are covered in
IEC 61000-4-2 (Electrostatic Discharge
Immunity), IEC 61000-4-4 (Electrical Fast
Transients), and IEC 61000-4-5 (Surge
Immunity). Each of these standards covers
transient events that are similar in nature
and approximate different real world
events such as lighting strikes, conductive
circuit interruption, and device handling.
The differences between the transients
lie in the peak voltages, impulse duration, and repetition rates. In non-isolated
systems the addition of protective components is often required to protect against
transient events. Proper application of
Intrinsic Safety is a standard covering equipment protection in explosive
atmospheres. Component requirements
are more stringent, and include limits
for temperature rise, maximum current,
µModule isolators allows the isolation
barrier to absorb the transient, eliminating the need for additional protection.
Figure 7 shows a properly configured
test set-up to meet the requirements of
EFT testing. Surge testing is performed with
the signal applied directly to the shield.
The transient (burst) generator is capacitively coupled to the shielded I/O lines
of the LTM2881. The coupling device is
depicted as a second shield with length of
0.5 meters per standard. The return side of
the burst generator is tied to the logic side
ground. The shield conducts the transient
to the isolated ground then across the isolation barrier to the generator return. The
parasitic inductance (LPAR) must be minimized or some of the transient is coupled
into the signal lines, decreasing the transient immunity effectiveness. The isolation
barrier has been tested and is compliant
to level 4 (4kV) for both surge and EFT.
Figure 7. EFT test configuration
0.5m
5V
LTM2881-5
VCC
VCC2
VL
PWR
RO
RE
TE
DE
DI
ISOLATION BARRIER
ON
DOUT
GND
36 | January 2011 : LT Journal of Analog Innovation
SLO
CPAR
A
B
CPAR
LPAR
CC 100pF–1000pF
Y
Z
DIN
GND2
BURST
GENERATOR
design features
Electrostatic discharge testing is performed
directly from pin to pin of the device.
Barrier ESD is performed from any logic
side pin to isolated side pin. The LTM2881
and LTM2882 have an ±8kV HBM (human
body model) ESD rating from any logic
side I/O pin to isolated side I/O pin, and
a ±10kV HBM, ±8kV IEC, ESD rating from
any isolated side power pin, VCC2 or GND2,
to logic side power pin, VCC , VL , or GND.
THE LTM2881 AND LTM2882
Linear Technology’s first released µModule
isolators are the LTM2881 RS485/RS422
transceiver plus power, and the LTM2882
dual RS232 transceiver plus power. Both
offer distinct advantages over alternative
solutions including excellent common
mode rejection, integrated high efficiency
isolated power and low EMI. Furthermore,
no external components, including power
decoupling capacitors, are required.
Each has separate logic power supply
inputs for easy interface to low voltage
systems ranging from 1.62V to 5.5V.
Two main power supply options are
offered, 3.0V to 3.6V and 4.5V to 5.5V,
which are completely independent of
the logic supply. Devices are available
in LGA and BGA packages with ambient temperature ranges of 0ºC to 70ºC,
–40ºC to 85ºC, and –55ºC to 105ºC.
The LTM2881, shown in Figure 8, is a
20Mbps transceiver with integrated selectable termination and 250kbps reduced
Figure 8. LTM2881 isolated RS485 µModule transceiver
3.3V OR 5V
Isolated
Power
REG
5V
RO
RS485/RS422
BUS
DI
ON
LTM2881
Figure 9. LTM2882 dual RS232 µModule transceiver
3.3V OR 5V
Power
T1IN
REG
5V
T1OUT
R1OUT
R1IN
T2IN
T2OUT
R2OUT
R2IN
The LTM2882, shown in Figure 9, is a
dual channel 1Mbps transceiver featuring one uncommitted isolated digital channel from logic to isolated
side and ±10kV HBM ESD protection
on the RS232 interface pins.
CONCLUSION
The LTM2881 RS485 transceiver and the
LTM2882 dual RS232 transceiver use Linear
Technology µModule isolator technology
to combine desirable data-plus-power
isolation features in a robust smallpackage system. Both devices feature
superior transient common mode rejection, magnetic field immunity, ESD and
transient barrier withstand, and insulation
lifetime. The core isolation techniques may
be applied to a breadth of applications.
Certification to UL1577 has been completed. The certification process has
been initiated for IEC 60747-5-2, including routine (production) test support for
partial discharge, and certification to
CSA (Canadian) Component Acceptance
Notice #5A in relation to IEC 60950-1.
Isolation terminology and relevant safety
standards have been introduced to aid
in the application of Linear Technology
µModule isolator products across all
industries. Isolation barrier performance is
under constant evaluation (i.e., insulation
lifetime, partial discharge, etc.) to ensure
the highest reliability and performance.
GALVANIC ISOLATION
Isolated
slew rate operating mode. The device
includes one uncommitted isolated digital
channel from isolated side to the logic
side and features ±15kV HBM ESD protection on the RS485 interface pins.
RS232
BUS
Additional product certifications may be
added to further distinguish the products’
exceptional isolation characteristics. n
Notes
1V.Y.
Ushakov, Insulation of High-Voltage Equipment,
Springer-Verlag, Berlin (2004)
LTM2882
GALVANIC ISOLATION
January 2011 : LT Journal of Analog Innovation | 37
Isolated Power Supplies Made Easy
John D. Morris
The LT3748 is a switching regulator controller specifically
designed to simplify the design of isolated power supplies
using a flyback topology. No third winding or opto-isolator
is required as the LT3748 senses the isolated output
voltage directly from the primary-side flyback waveform.
One challenge in designing a flyback
converter is that information relating to
the output voltage on the secondary side
of the transformer must be fed back to the
regulator on the primary side in order to
maintain regulation. Historically, feedback
across the isolation barrier is achieved
using opto-isolators or extra transformer
windings, though both methods present a
number of design problems. Opto-isolator
feedback circuits add components, increasing converter size and cost. They also draw
power, degrading efficiency and complicating thermal design. Opto-isolators also
make it difficult to accurately regulate
the output due to their limited dynamic
response, inherent nonlinearities, typical
variation from unit-to-unit and variation with age. The usual alternative is
to add an extra transformer winding,
but this may introduce other problems,
including bigger, more expensive magnetics or limited dynamic response.
Figure 2. A no-opto-coupler
30W design with an 18V-to-90V
input range (actual size)
38 | January 2011 : LT Journal of Analog Innovation
SW VOLTAGE
SW CURRENT
DIODE CURRENT
By contrast, the LT3748 infers the
isolated output voltage by examining the primary-side flyback pulse
waveform. In this manner, neither an
opto-isolator nor an extra transformer
winding is required to maintain regulation, and the output voltage is easily
programmed with two resistors.
The LT3748 features a boundary mode
control method (also called critical conduction mode), where the part operates at
the boundary between continuous conduction mode and discontinuous conduction
mode, as illustrated in Figure 1. Due to
the boundary control mode operation, the
output voltage can be calculated from the
transformer primary voltage when the secondary current is approximately zero. This
method improves load regulation without external resistors and capacitors and
results in typical line and load regulation
of better than ±5% while allowing for a
Figure 1. Idealized waveforms for an LT3748-based
flyback converter operating in boundary mode
simple and compact solution, as shown
by the 12V, 30W demo board in Figure 2.
OUTPUT POWER
Because the MOSFET power switch is
located outside the LT3748, the maximum
output power is limited primarily by external components—not the LT3748. Output
power limitations can be separated into
three categories: voltage limitations, current limitations and thermal limitations.
The voltage limitations in a flyback design
are primarily the MOSFET switch maximum
drain to source voltage and the output
diode reverse-bias rating. The current
limitation on output power delivery is generally constrained by transformer saturation current in higher power applications,
although the MOSFET switch and output
diode may need to be rated for the desired
Figure 3. The LT3748 is available in an MSOP-16
package with four pins removed for high voltage
operation.
design features
The LT3748 infers the isolated output voltage by examining
the primary-side flyback pulse waveform. In this manner,
neither an opto-isolator nor an extra transformer winding
is required to maintain regulation, and the output
voltage is easily programmed with two resistors.
currents, as well. The thermal limitation in
flyback applications for lower output voltages is dominated by losses in the output
diode, with resistive and leakage losses
in the transformer increasing in significance as the output voltage is increased.
OPTIMIZED FEATURES
The LT3748 is capable of driving the vast
majority of appropriate MOSFETs at frequencies of up to several hundred kilohertz using its built-in gate driver capable
of 1.9A average output current (both rising
and falling) and its internal INTVCC lowdropout regulator. In addition, start-up is
well controlled with programmable softstart and undervoltage lockout. Although
the LT3748 fits in a compact MSOP-16
package, four pins have been removed to
provide sufficient spacing for high voltage operation, as shown in Figure 3.
Figure 4. Efficiency of the LT3748 application in
Figure 5 with and without a third winding
The LT3748 provides excellent output
voltage regulation without the need for
an opto-coupler or third winding, but for
some applications with high input voltages, an additional winding may improve
overall system efficiency, particularly at
lighter loads. The third winding should
be designed to output a voltage above
7.2V but never exceeding 20V. In typical
applications over 15W, overdriving the
INTVCC pin may improve efficiency by several percent at maximum load and more
than 10% at light loads. Figure 4 shows
the efficiency of the circuit in Figure 5 with
and without the third winding connected.
This can occur when an inductive output
short-circuit causes the output voltage to
dip below zero or when the transformer
saturation current is exceeded. Regardless
of the cause, when the voltage at the
SENSE pin exceeds ~130mV—or 30% higher
than the programmed maximum current
limit in the RSENSE resistor—the SS pin is
reset, thus halting switching operation.
Once the soft-start capacitor is recharged
and the soft-start threshold is reached,
switching resumes at the minimum current limit. In output short circuit cases
where the reflected output voltage plus
the forward diode drop is greater than
zero, the LT3748 functions normally and
no external components are stressed.
OVER-CURRENT PROTECTION
HIGH TEMPERATURE OPERATION
The LT3748 has an internal threshold to
detect when current in the RSENSE resistor
exceeds the programmed range to protect
external devices in case of a system fault.
The LT3748 is available in E, I and H
grades, and is designed for excellent performance across a wide temperature range.
Other than the internal INTVCC regulator,
Figure 5. Schematic for the converter pictured in Figure 2. This converter takes an 18V-to-90V input and
produces a 2.5A output at 12V.
VIN
22V TO 75V
95
90
EFFICIENCY (%)
OVERDRIVING INTV CC
WITH A THIRD WINDING
WITH THIRD WINDING
T1
2.5:1:0.83 D1
+
100µF
100V
85
1µF
100V
2.2µF
100V
×2
402k
EN/UVLO
VIN
RFB
32.4k
80
INTVCC
WITHOUT THIRD WINDING
LT3748
75
70
65
VIN = 72V
VOUT = 12V
0
0.5
1
1.5
ILOAD (A)
2
2.5
43Ω
63.4k
GATE
SS
SENSE
VC
0.1µF
3
TC
470pF
GND
47.5k
2200pF
RREF
6.04k
150k
4.7µF
16V
43Ω
220pF
100V
22µF
16V
×3
VOUT+
12V
2.5A
D3
VOUT–
D2
M1
0.015Ω
D1: PDS5100
D2: BAT41KFILM
D3: DFLZ13, 13V
T1: MIDCOM 750311607
M1: Si7450DP
January 2011 : LT Journal of Analog Innovation | 39
Figure 6. IGBT controller supply for hybrid
and electric vehicle applications
10µF
1µF
825k
D1
15V
300mA
6µH
VIN
EN/UVLO
71.5k
RFB
RREF
150k
6.04k
VC
2nF
GND
D3
15V
300mA
10k
C1
IGBT
DRIVER
C2
IGBT
DRIVER
C3
4.7µF
D4
15V
300mA
C1-C4: 22µH 25V X7R ×2
D1-D4: DIODES INC. PDS3100
M1: VISHAY Si7898DP
T1: COILTRONICS VERSA-PAC VP4-0075-R
3-PHASE
MOTOR
IGBT
DRIVER
C4
0V
HIGH OUTPUT VOLTAGE FOR
REMOTE SENSORS
CONCLUSION
The LT3748 simplifies the design of isolated
flyback converters by using a primaryside sensing, boundary mode control
scheme that precludes the need for an
opto-coupler and its related circuitry.
The LT3748 also features a wide input
range, low internal power dissipation, a
1.9A gate driver and user-programmable
protection features that further simplify
design and add to its versatility. n
A flyback topology is often the only
way to produce a high voltage isolated
output for long cable runs or for powering interface equipment. Figure 7
shows a typical application for this style
of application with complementary
300V outputs. For the low power levels
in this application, an off-the-shelf EP13
transformer is more than sufficient and
keeps the total solution size small.
DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY
VIN
7V TO 15V
T1
1:10:10
C1
10µF
C2
1µF
C8
0.22µF
50V
R1
357k
R2
93.1k
Figure 7. A ±300V isolated flyback converter
VIN
EN/UVLO
RFB
C5
SS
TC
VOUT+
300V
8mA
VOUT–
D2
C6
R4
6.04k
GATE
R7
600k
D3
R3
140k
LT3748
D1
R5
10k
RREF
M1
R8
600k
VOUT+
300V
8mA
VOUT–
SENSE
C7
0.1µF
C9
100pF
40 | January 2011 : LT Journal of Analog Innovation
INTVCC 0.0125Ω
4700pF
IGBT CONTROLLER SUPPLY FOR
AUTOMOTIVE APPLICATIONS
IGBT
DRIVER
SENSE
SS
133k
M1
GATE
TC
12V, 3A OUTPUT
FROM 18V–90V INPUT
The LT3748 can easily produce multiple isolated supplies to power IGBTs
that drive synchronous motors from
high battery voltages in electric or
hybrid electric vehicles, as illustrated
in Figure 6. A MOSFET with 150V maximum VDS is selected so that any snubbing
circuitry is optional and the hysteretic
UVLO threshold is set to start switching when VIN equals 10V while allowing
VIN to droop to 8V while switching.
D2
15V
300mA
LT3748
the LT3748 dissipates very little power,
even at high input voltages, so limitations
to thermal performance are almost entirely
in the external components, which can
be correctly sized or cooled as needed.
Figure 5 shows an application that
efficiently converts a wide input range
to a 12V output. Because the LT3748 is
capable of handling up to 100V at its
input, no additional interface circuitry
is required between the line voltage and
the controller. A simple RC snubber is all
that is required to protect the 200V Si7450
MOSFET from excessive voltage across the
full line and load range. Although a third
winding is normally connected to boost
efficiency at lighter loads, all regulation is done on the primary winding—a
transformer without a third winding
would be nearly as efficient at lower
input voltages and high output loads.
320V
T1
1:1:1:1:1
VIN
12V TYP
VC
GND
R6
24.9k
C4
2.2nF
INTVCC
50mΩ
C3
4.7µF
C5, C6: 0.1µF 600V ×2
D1, D2: CENTRAL SEMICONDUCTOR CMR1U-06M LTC
M1: FAIRCHILD FDM3622
T1: WÜRTH ELEKTRONIK 750311486
D3: CENTRAL SEMICONDUCTOR CMMR1U-02
design ideas
Nanopower Buck Converter Runs on 720nA, Easily Fits into
Energy Harvesting and Other Low Power Applications
Michael Whitaker
The LTC3388-1/LTC3388-3 integrated synchronous stepdown regulator provides a regulated output while consuming
a mere 720nA of quiescent current. It accepts inputs up to
20V and can deliver up to 50mA of load current. Eight pinselectable output voltages are offered: 1.2V, 1.5V, 1.8V, and
2.5V on the LTC3388-1 and 2.8V, 3.0V, 3.3V, and 5.0V on the
LTC3388-3. The extremely low quiescent current prolongs
battery life for keep-alive circuits in portable electronics and
makes it a good fit for energy harvesting applications.
HIGH EFFICIENCY AT LIGHT LOADS
The LTC3388-1/LTC3388-3 (Figure 1)
achieves its low quiescent current by
entering a sleep state once the output
is in regulation. In the sleep state, load
current is provided by the output capacitor while the output voltage is monitored. When the output falls below a
fixed hysteresis window, the converter
wakes up and refreshes the capacitor.
SUPPORTS ENERGY HARVESTING
APPLICATIONS
current required to maintain regulation
approaches the DC sleep quiescent current.
ENABLE AND STANDBY
This hysteretic method of providing a
regulated output minimizes losses associated with FET switching and makes it
possible to efficiently regulate at very
light loads. The total quiescent current at
VIN in the sleep state is 720n A when VIN is
4V and increases to only 820n A when
VIN is 20V. At light loads, the time the
buck is active is miniscule relative to the
time it sleeps, so the average quiescent
2.7V TO 20V
1µF
6V
2.2µF
25V
CAP
LTC3388-1/
LTC3388-3
100µH
SW
VOUT
47µF
6V
VOUT
VIN2
4.7µF
6V
Figure 1: Low quiescent
current buck converter
VIN
The LTC3388-1/LTC3388-3 is especially
well suited for energy harvesting applications where only low amounts of
energy are available. Figure 3 shows the
LTC3588-1 piezoelectric energy harvesting
power supply harvesting ambient vibration energy with a piezoelectric transducer and producing a 3.3V output. The
LTC3388-3 is powered from this output
and is configured to provide a −3.3V rail,
producing a low power dual output
supply. The LTC3388-3’s low quiescent
current, combined with the LTC3588-1’s
(continued on page 43)
100
90
EN
PGOOD
STBY
D0, D1
GND
2
OUTPUT
VOLTAGE
SELECT
VOUT = 1.8V, L = 100µH
80
EFFICIENCY (%)
Two pins, EN and STBY, control enable
and standby functions on the LTC3388-1/
LTC3388-3. When EN is low, the buck
is turned off and only 520n A of quiescent current appears at VIN . When EN is
high, the STBY pin places the LTC3388-1/
LTC3388-3 in standby. In this mode the
buck is prevented from switching, resulting in a quiet supply. The PGOOD pin,
which is high when the output is in regulation, remains active while in standby.
PGOOD transitioning low can serve as an
indicator that the output has fallen and
that the LTC3388-1/LTC3388-3 should leave
standby mode to refresh the output.
The extremely low quiescent current of the LTC3388-1/LTC3388-3 allows
for high efficiency at loads as low as
10µ A. This is especially useful for low
power systems that spend a long time
idling and only periodically wake up
to perform a task. Figure 2 shows typical efficiency of the LTC3388-1 for the
1.8V output, which is suitable for powering low power microprocessors.
70
60
50
40
30
20
VIN = 3.0V
VIN = 10V
VIN = 20V
10
0
1µ
10µ
100µ
1m
LOAD CURRENT (A)
10m
Figure 2: LTC3388-1 Efficiency vs load current for
the 1.8V output
January 2011 : LT Journal of Analog Innovation | 41
Product Briefs
10PPM/°C, 3mV DROPOUT VOLTAGE
REFERENCE OPERATES WITH LESS
THAN 1µA
The LT6656 is a precision SOT23 voltage
reference that operates on only 850n A of
supply current. The LT6656 has an initial
error of less than 0.05% and a guaranteed
temperature drift of less than 10ppm/°C.
This combination of precision and
ultralow power is ideal for portable, wireless and remote devices. With an output
drive capability of 5m A, the LT6656 is
suitable for a wide range of applications.
For example, the LT6656 could serve as
both the supply voltage and the precision
reference for a low power, high resolution
ADC. The LT6656 can replace low power
shunt references, offering better efficiency
and regulation in the presence of variable load current and supply voltages.
For many applications, the extremely
low quiescent current allows the LT6656
to remain in an always-on, stable state.
Building on Linear Technology’s extensive
family of precision voltage references, the
LT6656 is optimized for battery-powered
operation. This high performance bipolar device can withstand reverse battery
conditions, accept input voltages up to
18V and operate from voltages as low
as 3mV above the output voltage. When
unpowered, the output is high impedance, avoiding loading on the rest of
the circuit. The LT6656 is fully specified
for operation from –40°C to 85°C, and is
guaranteed functional over the extreme
temperature range of –55°C to 125°C.
“The high precision of the LT6656, combined with only 1µ A of supply current
42 | January 2011 : LT Journal of Analog Innovation
packaged in a tiny SOT23, demonstrates
a significant advance in voltage reference
technology,” stated Brendan Whelan,
design manager for Linear Technology.
400mA SYNCHRONOUS STEPUP DC/DC CONVERTER WITH
MAXIMUM POWER POINT CONTROL
& 250mV START-UP FOR ENERGY
HARVESTING APPLICATIONS
2.6A, 2.5MHz STEP-DOWN DC/DC
CONVERTER PLUS DUAL LINEAR
CONTROLLERS IN 4mm × 5mm QFN
OPERATES FROM 4V–36V INPUTS
The LTC3105 is a high performance,
synchronous boost converter that incorporates maximum power point control
(MPPC) and starts up with inputs as low
as 250mV. The LTC3105 operates over an
extremely wide input range of 0.225V to
5V, making it ideal for harvesting energy
from high impedance alternative power
sources, including photovoltaic cells, thermoelectric generators (TEGs) and fuel cells.
The LT3694 is a 2.6A, 36V step-down
switching regulator with dual linear
controllers, packaged in a 4mm × 5mm
QFN or TSSOP-20E. The LT3694 operates from a VIN range of 4V to 36V with
transient protection to 70V, making it
ideal for load dump and cold-crank
conditions found in automotive applications. Its internal 3.5A switch can deliver
up to 2.6A of continuous output current
at voltages as low as 0.75V. Its switching
frequency is user-programmable from
250kHz to 2.5MHz, enabling the designer to
optimize efficiency while avoiding critical noise-sensitive frequency bands. Each
integrated LDO controller has an accurate
programmable current limit up to 1A to
provide an additional level of reliability.
Combined with NPN transistors, they can
be used to offer dual low noise outputs in
addition to the primary channel. Although
the LDO controllers can be powered by
independent inputs, powering them from
the primary switching output ensures
both high efficiency and low noise.
The combination of its 4mm × 5mm
QFN package (or thermally enhanced
TSSOP-20E) and high switching frequency, which keeps external capacitors and inductors small, provides a
compact, thermally efficient footprint.
The LTC3105’s internal 400m A synchronous switches maximize efficiency while
its Burst Mode® operation offers quiescent
current of only 24µ A, further optimizing converter efficiency over all operating conditions. A user-programmable
MPPC set point maximizes the energy that
can be extracted from any power source
without collapsing its internal voltage.
The LTC3105 is ideally suited to power
wireless sensors and data acquisition
applications. Surplus or ambient energy
can be harvested and then used to generate
system power in lieu of traditional wired
or battery power, which may be expensive
or impractical. Typically, these applications require very low average power,
but require periodic pulses of higher
load current. For example, the LTC3105
can be used in wireless sensor applications where the power load is extremely
low when the sensor is in standby mode,
which is interrupted by periodic high load
design ideas
The LTC3787 is a high power 2-phase single output synchronous step-up DC/DC controller,
which replaces the boost diodes with high efficiency N-channel MOSFETs. This solution
eliminates the heat sink normally required in medium to high power boost converters. The
LTC3787 can produce a 24V at 10A output from a 12V input at up to 97% efficiency.
bursts, when the circuitry is powered up
to take measurements and transmit data.
The LTC3105 offers an auxiliary LDO that
delivers up to 6m A of output current to
power external microcontrollers and sensors while the main output is charging.
Once fully charged, the main output can
deliver voltages as high as 5.25V with up
to 100m A of output current. It can also
regulate VOUT even when VIN is greater than
or equal to VOUT, offering further design
flexibility. In shutdown, the LTC3105 offers
output disconnect, isolating VIN from VOUT,
requiring only 10µ A of quiescent current.
The combination of the LT3105’s
3mm × 3mm DFN package (or MSOP-12)
and very small external components
offers a very compact solution for
energy harvesting applications.
HIGH POWER POLYPHASE
SYNCHRONOUS BOOST
CONTROLLER ELIMINATES HEAT
SINK WITH 97% EFFICIENCY
The LTC3787 is a high power 2-phase
single output synchronous step-up
DC/DC controller, which replaces the boost
diodes with high efficiency N-channel
MOSFETs. This solution eliminates the
heat sink normally required in medium
to high power boost converters. The
LTC3787 can produce a 24V at 10A output
from a 12V input at up to 97% efficiency.
The LTC3787’s 135µ A standby quiescent
current when configured for Burst Mode
operation makes it ideal for high power
automotive audio amplifiers, as well as
industrial and medical applications where
a step-up DC/DC converter must deliver
high power in a small solution size.
The LTC3787 operates from an input
voltage ranging from 4.5V to 38V during
start-up, maintains operation down to
2.5V after start-up and can regulate an
(LTC3388-1/LTC3388-3, continued from page 41)
PIEZO
MIDE V25W
own low quiescent current results in
a complete solution that draws only
1600n A with no load at both outputs.
CONCLUSION
The LTC3388-1/LTC3388-3 monolithic
buck converter’s extremely low quiescent
current makes it ideal for low power
applications. A quiescent current of less
than a microamp prolongs battery life
for keep-alive circuits in portable electronics and enables a new generation
of energy harvesting applications. n
output voltage as high as 60V. The powerful 1.2Ω onboard N-channel MOSFET gate
drivers are capable of slewing large
MOSFET gates quickly. The device’s current mode architecture, clock output
and phase modulation enables easy
paralleling of multiple devices for up to
12-phase operation. The LTC3787 has a
phase-lockable frequency from 75kHz to
850kHz or a selectable fixed frequency
from 50kHz to 900kHz. In applications
where the input voltage exceeds the regulated output voltage, the LTC3787 keeps
the synchronous MOSFET on continuously
so that the output voltage follows the
input voltage with minimal power loss.
In addition, this device features adjustable cycle-by-cycle current limit and can
use a sense resistor or monitor the voltage
drop across the inductor (DCR) for current sensing. Furthermore, the LTC3787
has adjustable soft-start, a power good
output and maintains ±1% reference voltage accuracy over an operating junction
temperature range of –40°C to 125°C. n
1µF
6V
10µF
25V
4.7µF
6V
PZ1
PZ2
VIN
PGOOD
LTC3588-1
22µH
CAP
SW
VIN2
VOUT
D1
D0
GND
3.3V
1µF
6V
47µF
6V
2.2µF
10V
4.7µF
6V
VIN
PGOOD
CAP LTC3388-3
*
SW
VIN2
VOUT
EN
D1
D0
22µH
GND
STBY
47µF
6V
–3.3V
Figure 3: Piezoelectric energy harvester
with dual ±3.3V outputs
* EXPOSED PAD MUST BE ELECTRICALLY ISOLATED FROM
SYSTEM GROUND AND CONNECTED TO THE –3.3V RAIL.
January 2011 : LT Journal of Analog Innovation | 43
highlights from circuits.linear.com
5V BACKUP SUPPLY FROM SUPERCAP
The LTC3112 is a fixed frequency synchronous buck-boost DC/DC
converter with an extended input and output range. The unique
4-switch, single inductor architecture provides low noise and
seamless operation from input voltages above, below or equal
to the output voltage. With an input range of 2.7V to 15V, the
LTC3112 is well suited to a wide variety of single or multiple cell
battery, backup capacitor or wall adapter source applications.
www.linear.com/3112
VOUT
4.7µH
499k
0.1µF
VIN
15V TO 2V
+
SW1
SW2
BST1
VIN
BST2
VOUT
LTC3112
VCC
1M
0.1µF
680pF
COMP
VOUT
5V/250mA
33k
845k
47µF
22pF
220µF
TANT
FB
PWM/SYNC
22mF
SUPERCAP
499k
47pF
RUN
IOUT
GND
OVP
TO ADC
1V PER AMP
1µF
100pF
10k
42.2k
158k
L1*
10µH
SINGLE CELL POWERED REMOTE WIRELESS SENSOR
The LTC3105 is a high efficiency step-up DC/DC converter that can operate from input voltages as low as 200mV. A 250mV start-up capability and
integrated maximum power point controller (MPPC) enable operation
directly from low voltage, high impedance alternative power sources
such as photovoltaic cells, TEGs (thermoelectric generators) and fuel
cells. A user programmable MPPC set point maximizes the energy
that can be extracted from any power source. Burst Mode operation,
with a proprietary self-adjusting peak current, optimizes converter
efficiency and output voltage ripple over all operating conditions.
www.linear.com/3105
VIN
+
SW
CIN
10µF
–
VOUT
3.3V
VOUT
R1
2.32M
LTC3105
FB
MPPC
COUT
100µF
R2
1.02M
RMPPC
40.2k
XMTR
I/O
OFF ON
PGOOD
CAUX
1µF
2.2V
LDO
SHDN
AUX
2N7000
EN
FBLDO
µC
RPG
499k
VDD
GND
A/D
SENSOR
GPIO
GND
CLDO
4.7µF
* COILCRAFT MSS5131-103MX
5V
FULL BRIDGE ARMATURE CURRENT MONITOR
The LT1999 is a high speed precision current sense amplifier,
designed to monitor bidirectional currents over a wide common
mode range. The LT1999 is offered in three gain options: 10V/V,
20V/V, and 50V/V. With a 2MHz bandwidth and a common mode
input range of –5V to 80V, the LT1999 is suitable for monitoring currents in H-Bridge motor controls, switching power supplies, solenoid
currents, and battery charge currents from full charge to depletion.
www.linear.com/1999
V+
LT1999
VS
V+
2µA
SHDN
4k
V+IN
RS
+
–
V+
0.8k
V–IN
–
+
VOUT
V+
4k
0.8k
V+
5V
SHDN
RG
160k
160k
VREF
0.1µF
0.1µF
L, LT, LTC, LTM, Linear Technology, the Linear logo, Burst Mode, LTSpice, TimerBlox and µModule are registered trademarks, and Hot Swap, LTpowerCAD, PowerPath and are trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
© 2011 Linear Technology Corporation/Printed in U.S.A./49K
Linear Technology Corporation
1630 McCarthy Boulevard, Milpitas, CA 95035
(408) 432-1900
www.linear.com
Cert no. SW-COC-001530