LINEAR TECHNOLOGY MARCH 2009 IN THIS ISSUE… COVER ARTICLE Battery Stack Monitor Extends Life of Li-Ion Batteries in Hybrid Electric Vehicles ..................1 Michael Kultgen and Jon Munson Linear in the News… ...........................2 DESIGN FEATURES DC/DC Converter, Capacitor Charger Takes Inputs from 4.75V to 400V ........9 VOLUME XIX NUMBER 1 Battery Stack Monitor Extends Life of Li-Ion Batteries in Hybrid Electric Vehicles by Michael Kultgen and Jon Munson Robert Milliken and Peter Liu How to Choose a Voltage Reference ...14 Brendan Whelan Introduction 1.2A Monolithic Buck Regulator Shrinks Supply Size and Cost with Programmable Output Current Limit .........................................................20 The cost of running a car on electricity is equivalent to paying $0.75/gallon for gasoline, and if that electricity comes from carbon neutral sources, car owners are saving both money and the environment (gasoline combustion produces 9kg of CO2 per US gallon). Advancements in battery technology (see sidebar), especially with Lithium-based chemistries, hold the greatest promise for converting the worldwide leet of cars to hybrid or fully electric. Tom Sheehan Boost Converters for Keep-Alive Circuits Draw Only 8.5µA of Quiescent Current .........................................................22 Xiaohua Su Industrial/Automotive Step-Down Regulator Accepts 3.6V to 36V and Includes Power-On Reset and Watchdog Timer in 3mm × 3mm QFN ................24 Ramanjot Singh Complete APD Bias Solution in 60mm2 with On-the-Fly Adjustable Current Limit and Adjustable VAPD ...................27 Xin (Shin) Qi DESIGN IDEAS Don’t Want to Hear It? Avoid the Audio Band with PWM LED Dimming at Frequencies Above 20kHz ..................30 + Lithium battery packs offer the highest energy density of any current battery technology, but high performance is not guaranteed simply by design. In real world use, a battery management system (BMS) makes a signiicant difference in the performance and lifetime of Li-Ion batteries—arguably more so than the design of the battery itself. The LTC6802 multicell battery stack monitor is central to any BMS for the continued on page 12-CELL BATTERY MODULE 12-CELL BATTERY MODULE 12-CELL BATTERY MODULE 12-CELL BATTERY MODULE + + + + CURRENT SENSOR Eric Young – – – BATTERY MONITORING & BALANCING BATTERY MONITORING & BALANCING BATTERY MONITORING & BALANCING BATTERY MONITORING & BALANCING DATA BUS DATA BUS DATA BUS DATA BUS DATA BUS DATA BUS DATA BUS DATA BUS BATTERY MONITORING & BALANCING BATTERY MONITORING & BALANCING BATTERY MONITORING & BALANCING BATTERY MONITORING & BALANCING SERVICE SWITCH Eliminate EMI Worries with 2A, 15mm × 9mm × 2.82mm µModule™ Step-Down Regulator ........................33 – David Ng Diode Turn-On Time Induced Failures in Switching Regulators ....................34 CAN Jim Williams and David Beebe µModule Regulator Fits a (Nearly) Complete Buck-Boost Solution in 15mm × 15mm × 2.8mm for 4.5V–36V VIN to 0.8V–34V VOUT ..........39 Judy Sun, Sam Young and Henry Zhang New Device Cameos ...........................41 – HOST CONTROLLER SPI – + 12-CELL BATTERY MODULE – + 12-CELL BATTERY MODULE – + 12-CELL BATTERY MODULE – + 12-CELL BATTERY MODULE Design Tools ......................................43 Sales Offices .....................................44 Figure 1. 96-cell battery pack L, LT, LTC, LTM, Burst Mode, OPTI-LOOP, Over-The-Top and PolyPhase are registered trademarks of Linear Technology Corporation. Adaptive Power, Bat-Track, BodeCAD, C-Load, DirectSense, Easy Drive, FilterCAD, Hot Swap, LinearView, µModule, Micropower SwitcherCAD, Multimode Dimming, No Latency ΔΣ, No Latency Delta-Sigma, No RSENSE, Operational Filter, PanelProtect, PowerPath, PowerSOT, SmartStart, SoftSpan, Stage Shedding, SwitcherCAD, ThinSOT, TimerBlox, True Color PWM, UltraFast and VLDO are trademarks of Linear Technology Corporation. Other product names may be trademarks of the companies that manufacture the products. L LINEAR IN THE NEWS Linear in the News… EDN Highlights Linear for Innovation Awards Linear CEO Comments on Growth Markets EDN magazine in February chose several Linear Technology products as inalists for their annual Innovation Awards, to be announced later this month. And the nominees are: Last month in EE Times, Linear Technology CEO Lothar Maier discussed the challenging market conditions and the bright spots on the horizon: “In these times our customers will continue to invest in new products and new product development. Innovation will return growth to the semiconductor market—speciically to analog. Now is the time to get new products out, to be irst to market and to have products that target emerging growth markets.” He discussed several key markets: q Automotive. “Automotive manufacturers are forecasting automotive electronic content to grow 2–3 times over the next few years, so we will continue to provide new products to the automotive area. In addition, every major automotive manufacturer in the world is now working on hybrid vehicles, which will add even more electronic content in cars. We have just introduced an innovative device, the LTC6802, a highly integrated battery stack monitor that signiicantly eases the design of battery monitoring systems for hybrid/electric vehicles.” q Green Growth Markets. “Products targeted toward energy conservation or energy harvesting will see growth opportunities and are insulated from the current market conditions. Energy costs and environmental concerns, as well as the need to extend battery life for mobile devices, have led to a focus on power optimization. Our energy-eficient products enable customers to convert power more eficiently, consume less power and extend battery life. Our LED drivers enable a new generation of low power lighting for a range of applications, from cars and medical instruments to laptops and ofice lighting. Our eficient analog solutions will help drive innovative cleantech markets such as solar and wind power systems.” q Communications Infrastructure. “Wireless systems continue to produce signiicant market opportunities for products in wireless and network infrastructure. Our high speed data converters and high frequency products are designed into the next generation of cellular basestations. And our Hot Swap™ and Power over Ethernet products are proliferating in networks.” q Industrial. “The broad industrial market continues to provide a solid core of business and is somewhat more insulated from market swings. Linear’s analog products are used in a broad range of industrial systems, including factory automation, industrial process control, medical, instrumentation and security.” Lothar Maier concluded, “Finally, I believe that Linear’s strategy of customer, market and geographic diversity will be a hedge against the current market conditions and will provide the conduit to future growth.” L Best Contributed Article—“High Voltage, Low-Noise DC/DC Converters” by Jim Williams You can ind the article in its entirety on the EDN website at www.edn.com/jimwilliams. Battery ICs Category—LTC6802 Battery Stack Monitor The LTC6802 is a highly integrated multicell battery monitoring IC capable of precisely measuring the voltages of up to 12 series-connected battery cells. Using a novel stacking technique, multiple LTC6802s can be placed in series without optocouplers or isolators. See the cover article of this issue for an overview of this part. Power ICs Category—LTC3642 50mA Synchronous Step-Down Converter The LTC3642 uses a unique high voltage synchronous rectiication design, capable of continuous input voltages of 45V and offers transient protection up to 60V. Its internal synchronous rectiication and its programmable peak current mode control feature enable it to deliver up to 93% eficiency, maximizing battery run time. Power ICs: Modules—LTM4606 Ultralow EMI, 6A DC/DC µModule Regulator The LTM4606 DC/DC µModule™ regulator signiicantly reduces switching regulator noise by attenuating conducted and radiated energy at the source. The µModule device is a complete DC/DC system-in-a-package, including the inductor, controller IC, MOSFETs, input and output capacitors and the compensation circuitry, housed in an enclosed surface-mount plastic package resembling an IC. The LTM4606 reduces switching regulator noise at the source. 2 Linear Technology Magazine • March 2009 DESIGN FEATURES L LTC6802, continued from page large battery stacks common in electric vehicles (EVs) and hybrid electric vehicles (HEVs). Its robust design and high accuracy helps guarantee the performance and lifetime of expensive battery packs. For instance, to meet a 15-year, 5000 charge cycle goal, only a portion (say 40%) of the battery pack’s cellcapacity can be used. Of course, using only 40% of the capacity essentially lowers the energy density of the pack. This is the problem: increasing battery lifetime is traded against the need to use as few kg of batteries as possible—the most expensive component in any EV. Only a well-designed BMS can maximize battery performance and lifetime in the face 200A peak charge and discharge currents. Battery Management System Optimizes Li-Ion Run Time and Lifetime In any battery stack, the more accurately you know state of charge (SOC) of each cell, the more cell capacity you can use while still maximizing cell life. In a laptop computer, gas gauging comes from monitoring cell voltage and counting coulombs in and out of the stack of four to eight cells. Voltage, current, time and temperature are combined in a robust algorithm to give an indication of the SOC. Unfortunately, it’s nearly impossible to count coulombs in a car. The battery drives an electric motor, not a motherboard, so it must handle current spikes of 200A, followed by low level idling. Furthermore, you have from 96 Li-ion Batteries in Electric Vehicles and Hybrids So why aren’t all cars electric? One regenerative braking means the gas Table 1. Energy density comparison reason is energy density. Gasoline engine runs less often and runs at a holds 80 times the energy per kg as higher eficiency, effectively doubling Medium Wh/kg Li-ion batteries (Table 1) and refuels the mpg. Diesel Fuel 12,700 in three minutes, essentially allowing In the 1970s the only available high Gasoline 12,200 indeinite driving. Even a big lithium power battery chemistry was lead pack only gives a passenger car acid, too heavy to reasonably power Li-Ion Battery 150 about a 100-miles after an 8-hour anything larger than a golf cart. Then NiMh Battery 100 charging cycle. To drive a passenger came NiMh batteries, which improved car further than 100 miles you still energy density enough to enable the Lead Acid Battery 25 need a gasoline engine, but even irst commercially successful HEVs, so, batteries improve gas mileage in like the Toyota Prius and Ford Escape. hybrid electric vehicles (HEVs). The peak eficiency of Li-ion batteries take energy density another step forward, the Otto cycle engine is only 30% at high RPMs and the by offering another 50% improvement. The safety of Liaverage eficiency is about 12%. Using batteries to sup- ion was a concern, but new battery technologies like the ply torque during acceleration and recover joules during A123 nanophosphate cell, the EnerDel Spinel-Titanate chemistry, the GS Yuasa EH6 design and others are as safe as NiMh, offer extremely high power (200A peak disMG1 INVERTER BATTERY MG2 INVERTER charge rates), and last 10 to 15 years with proper charge management. By model year 2012, the majority of hybrid cars and trucks will use lithium battery technology. Figure 1 shows a shows a block diagram of the batGASOLINE SILENT ENGINE CHAIN POWER SPLIT tery pack with a BMS, and Figure 2 shows a typical HEV DEVICE power train. The battery pack building block is a 2.5V to 3.9V, 4Ahr to 40Ahr Li-ion cell. 100 to 200 cells are connected in series to bring the battery pack voltage into the hundreds of volts. This DC power source drives a 30kW to 70kW electric motor. The pack voltage is high ELECTRIC MOTOR/ ELECTRIC MOTOR/ so that the average current is low for a given power level. GENERATOR 1 (MG1) GENERATOR 2 (MG2) Lower current reduces I2R power losses, so cables can be smaller, thus reducing weight and cost. The pack REDUCTION should be able to deliver 200A under peak conditions GEARS and be quickly rechargeable. In other words, the battery needs to offer high energy density and high power denAXLES sity, speciications that can be met by Li-ion batteries. FRONT WHEELS Systems for busses and tractor-trailers use up to four parallel packs of 640V each. L DIFFERENTIAL Figure 2. Toyota Prius “split power” hybrid drive train Linear Technology Magazine • March 2009 3 4.5 4.5 4.0 4.0 CELL VOLTAGE (V) to 200 cells in series, in groups of 10 or 12. The cells age at different rates, were manufactured from multiple lots, and vary in temperature. Their capacities diverge constantly. Different cells with the same coulomb count can have wildly different charge levels. That’s why the BMS focuses on cell voltage. If you can accurately measure the voltage of every cell, you can know the cell’s SOC with reasonable accuracy (Figure 3). The trick is to improve the accuracy of the voltage measurement by taking into account temperature effects on battery ESR and capacity. By constantly measuring each cell’s voltage, you keep a running estimation of each cell’s charge level. If some cells are overcharged and some under, they can be balanced by bleeding off charge (passive balancing) or redistributing charge (active balancing). CELL VOLTAGE (V) L DESIGN FEATURES 3.5 3.0 1C 2C 5C 10C 20C 50C 2.5 2.0 1.5 0 3.5 3.0 2.5 –20°C 0°C 30°C 60°C 2.0 1.5 10 20 30 40 50 60 70 80 90 100 DISCHARGE (%) 0 Figure 3. State of charge vs current and temperature for a typical Li-ion cell NEXT 12-CELL PACK ABOVE LTC6802-1 SERIAL DATA TO LTC6802-1 ABOVE DIE TEMP V+ REGISTERS AND CONTROL 12-CELL BATTERY STRING MUX 12-BIT ∆∑ ADC Accurate Monitoring is Key to Raising Battery Performance while Lowering Costs 4 V– EXTERNAL TEMP NEXT 12-CELL PACK BELOW VOLTAGE REFERENCE SERIAL DATA TO LTC6802-1 BELOW 100k 100k NTC Figure 4. Simplified block diagram of the LTC6802 estimation of SOC is accurate to 3%. The BMS must charge cells to no more than 37% (40% – 3%) of their capacity to guarantee the 15-year lifetime. Now consider a monitor IC with 10mV error over similar conditions. In this case, the BMS can only use 32% (40% – 10mV • 1%/1.25mV) of the cells’ capacity and still guarantee a 15-year life. This seemingly negligible increase in measurement error results in a signiicant 14% reduction in the usable capacity. That is, a vehicle requires least 14% more batteries, or 9k 0.30 COST OF TYPICAL BATTERY PACK ($) 0.25 MEASUREMENT ERROR (%) The LTC6802 (Figure 4) is a precision data acquisition IC optimized for measuring the voltage of every cell in a large string series-connected batteries. In the BMS, the LTC6802 does the heavy lifting analog function, passing digital voltage and temperature measurements to the host processor for SOC computation. The LTC6802’s high accuracy, excellent noise rejection, high voltage tolerance, and extensive self-diagnostics make it robust and easy-to-use. The high level of integration means a substantial cost savings for customers when compared to discrete component data acquisition designs. Increasing measurement accuracy reduces battery cost, as illustrated by the following example. Figure 5 shows the typical performance of the LTC6802, where 0.1% total error from –20°C to 60°C translates to 4mV precision for a 3.7V cell. Suppose that to achieve a 15-year battery lifetime, you are limited to 40% of a cell’s capacity per charge cycle, and assume the cell voltage vs charge level of the battery is very lat, e.g., 1.25mV/%SOC. A measurement error of 4mV means the 10 20 30 40 50 60 70 80 90 100 DISCHARGE (%) 7 REPRESENTATIVE UNITS 0.20 0.15 0.10 0.05 0 –0.05 –0.10 –0.15 –0.20 –0.25 –0.30 –50 8k 7k 6k 5k 4k 3k –25 0 25 50 75 TEMPERATURE (°C) 100 125 Figure 5. Typical measurement accuracy vs temperature of seven samples 0 5 20 25 10 15 MEASUREMENT ERROR (mV) 30 Figure 6. High BMS accuracy is important to keeping battery costs in check, as shown in this cost vs measurement error model. Linear Technology Magazine • March 2009 DESIGN FEATURES L TP0610K CELL12 6µs 370V 1M 2.2M LTC6802-1 270V 0 = REF_EN GPIO2 10kHz VSTACK12 0 = CELL1 GPIO1 Figure 8. Inverter noise example WDTB 1M 1M LT1461A-4 10M 1M DNC DNC DNC VIN SD VOUT GND DNC VREG 2N7002 1µF 90.9k 2N7002 V− 4.096V 2.2µF C1 150Ω TP0610K + TP0610K TP0610K 100Ω VDD CH0 CH1 SEL SD LT1636 100nF TC4W53FU – CELL1 COM INH VEE VSS 1M Figure 7. Improving accuracy with calibration Linear Technology Magazine • March 2009 of 3.7V, which we need to measure to 4mV. Breaking the battery stack into 12-cell modules further reduces The LTC6802’s 0.1% total measurement error from –20°C to 60°C translates to 4mV precision for a 3.7V cell. Batteries are expensive. It takes about $4000 worth of batteries to drive 50 miles, so just increasing measurement error to 10mV means $560 in additional cells. This is why BMS designers scrutinize every 0.01% of measurement error. Diagnostic Features of the LTC6802 Improve Robustness Automotive systems require that “no bad cell reading be misinterpreted as a good cell reading.” Two of the more common faults that can cause false readings are open circuits and IC failures. If there is an open circuit in the wiring harness and if there is a ilter capacitor on the ADC input (Figure 11), the capacitor will tend to hold the input voltage at a point midway between the adjacent cells. Some type of open wire detection or cell resistance measuring function is necessary. The LTC6802 includes 100µA current sources to load the cell inputs. The current source will cause large changes in cell readings if there is an open circuit in the harness. 0 0 VCM(IN) = 5VP-P 72dB REJECTION –10 CORRESPONDS TO LESS THAN 1 BIT –20 AT ADC OUTPUT –10 –20 REJECTION (dB) REJECTION (db) at least 14% more weight, cost and electronics to travel an equivalent distance as a vehicle with the more accurate BMS. Batteries are expensive. It takes about $4000 worth of batteries to drive 50 miles, so the increased measurement error means $560 in additional cells. This is why BMS designers scrutinize every 0.01% of measurement error. Figure 6 shows a simple battery cost model as a function of BMS accuracy. Adding a low drift reference, an initial factory calibration, and a periodic self-calibration routine can improve the measurement accuracy of the LTC6802 to 0.03%. For example, in Figure 7 the LT1461A-4 is periodically applied to channel C1. The temperature stable LT1461 measurement is used to correct temperature drift in the LTC6802. The initial error of the LTC6802 and LT1461A is corrected by measuring and storing a calibration reference after board assembly. Inverter noise can seriously interfere with cell voltage measurements. When a 100-cell stack is loaded by an electric motor it can have a 370V open circuit voltage and up to 100V switching transients (Figure 8). Spreading the transient equally over the 100 cells means the top cell has 370V of common mode voltage, 100V of common mode transients, 1V of differential transients and an average DC value the common mode voltage. In a pack like Figure 2, each LTC6802 (one per module) sees up to 12V common mode transients and 1V differential transients per cell. The transients are at the PWM frequency of 10kHz to 20kHz. The LTC6802 has excellent common mode rejection (Figure 9) to eliminate this error term. The SINC2 ilter inherent in the delta-sigma ADC attenuates the differential noise by 40dB (Figure 10). External iltering or measurement averaging can be used to further reduce the differential noise. –30 –40 –30 –40 –50 –50 –60 –60 –70 10 100 1k 10k 100k FREQUENCY (Hz) 1M Figure 9. Cell measurement common mode rejection 10M –70 10 100 1k 10k FREQUENCY (Hz) 100k Figure 10. Cell measurement filtering 5 L DESIGN FEATURES The host controller must be able to run diagnostics on all the modules during normal operation to detect IC failures. If these periodic self-tests fail, then the control algorithm is suspect and the battery pack must be taken off line. The LTC6802 includes a built-in self-test in combination with external support circuits to allow the BMS to completely verify the data acquisition system. See the LTC6802 data sheets for more details. LTC6802-1 C4 B4 CF4 B3 CF3 C2 MUX C1 V– The LTC6802 Isolates Communications from Swings in Ground Potential Breaking a ~100 cell pack into modules makes it easier to integrate the analog circuits. Unfortunately, we are left with the task of getting the data from measurement IC to the host controller when the difference in ground potential exceeds 300V. The LTC6802 can solve this problem in a number of ways, depending on the speciic needs of the application. The LTC6802 comes in two lavors, depending on the desired data communication scheme. The LTC6802-1 offers a built-in stackable serial peripheral interface (SPI) solution designed for easy daisy chaining of the interface. The addressable LTC6802-2 is designed for bus-oriented (parallel) SPI communication, but it can also be used in a parallel-addressable, daisy chained interface for a robust and rela- C3 100µA Figure 11. Current sources help detect open circuits. tively inexpensive solution. All three schemes are described below. SPI Bus Communication with the Addressable LTC6802-2 and Digital Isolators The most straightforward approach is to use a bus communications scheme, with a digital isolator between each module and the host controller. Figure 12 shows a 96-cell pack using eight multicell modules monitored by the LTC6802. The physical layer is a 4-wire SPI bus. An addressing scheme allows the control module to talk to the battery modules separately or in unison. The data buses on the modules are isolated from one another. This is a robust scheme, but it has one major drawback: digital isolators are expensive and require an isolated power supply so that the battery cells don’t have to provide the power to the cell side of the isolator. Daisy Chaining the SPI Interface with the LTC6802-1 The LTC6802-1 provides ixed 1mA signaling between stacked devices to enable easy implementation a daisy chained SPI interface with inexpensive support circuitry. The digital isolators are eliminated as shown in Figure 13. The interface exploits the fact that the positive supply of module “N” is the same voltage as the ground of module “N+1.” A 1mA current is used to transmit data between adjacent modules. Like the analog circuits, the modular approach means the data bus has to deal with a fraction of the total pack voltage. BATTERY MODULE 8 LTC6802 12 Li-Ion SERIES BATTERIES BATTERY MONITOR DIGITAL ISOLATOR BATTERY MODULE 1 CONTROL MODULE GALVANIC ISOLATOR LTC6802 µCONTROLLER 12 Li-Ion SERIES BATTERIES BATTERY MONITOR DIGITAL ISOLATOR SPI CAN CAN TRANSCEIVER TO VEHICLE CAN BUS Figure 12. Using digital isolators to communicate to the LTC6802 6 Linear Technology Magazine • March 2009 DESIGN FEATURES L BATTERY MODULE 8 LTC6802 12 Li-Ion SERIES BATTERIES BATTERY MONITOR CONTROL MODULE BATTERY MODULE 1 GALVANIC ISOLATOR LTC6802 µCONTROLLER 12 Li-Ion SERIES BATTERIES BATTERY MONITOR SPI CAN CAN TRANSCEIVER TO VEHICLE CAN BUS Figure 13. Using the daisy chained SPI to eliminate digital isolators The disadvantage of any pure daisy chain is that a fault in one module results in a loss of communications with all the modules above it in the stack. Also, since there is no galvanic isolation between modules, the interface needs to handle large voltages that occur during fault conditions. For example if the “service switch” in Figure 1 is open and there is a load on the pack then the data bus connection between modules 4 and 5 will see a reverse voltage equal to the total pack voltage (–300V to –400V). The LTC6802 interface relies on external discrete diodes to block the reverse voltage during fault conditions. The Best of Both Worlds: Daisy Chained, Addressable Interface with the LTC6802-2 With inexpensive external circuitry, the LTC6802-2 can also be used in a stacked SPI coniguration like the LTC6802-1, but with more lexibility in the operating parameters. The SPI port of the LTC6802-2 is a 4-wire connection: chip select in (CSBI), clock in (SCKI), data in (SDI), and data out (SDO). The inputs are conventional CMOS levels and the output is an open-drain NMOS. The SDO pin must have an external pull-up current or added resistance suitable for the intended data rate. The IC also provides a versatile always-on 5V output (VREG), which can produce up to Linear Technology Magazine • March 2009 4mA to energize low power auxiliary circuitry. Figure 14 shows a complete stacked LTC6802-2 SPI interface for a 36cell application. The stack can be increased in size by replicating the circuit of the middle IC. In Figure 14, the VREG and V– pins of each stacked IC are used to bias common-base connected transistors to form a signal translation current for each SPI data line. Each LTC6802 can monitor up VBATT LTC6802-2 IC #3 VREG 1M 1.8k WDT 2.2k 2.2k 2.2k NDC7002N ALL NPN: CMPT8099 ALL PNP: CMPT8599 ALL PN: RS07J ALL SCHOTTKY: CMD5H2-3 SDI SCKI CSBI SDO V− LTC6802-2 IC #2 VREG 100Ω 2.2k 2.2k 2.2k 100Ω 2.2k 2.2k 2.2k SDI SCKI CSBI SDO V− LTC6802-2 IC #1 VREG SDI SCKI CSBI SDO V− R12 2.2k CS CK DI DO HOST µP 500kbps MAX DATA RATE Figure 14. Inexpensive SPI daisy chain for parallel-addressed LTC6802-2 7 L DESIGN FEATURES to 12 cell-potentials, which could sum to 60V in certain instances, so the transistors selected for the SPI translation need to have a VCBO over 60V, but they should be the highest available fT to prevent undue slowing of the logic signals. A suitable NPN candidate is the CMPT8099, while the CMPT8599 is its PNP complement, both from Central Semiconductor. These are fast 80V devices (fT > 150MHz). Sending Signals Upwards At the bottom-of-stack IC, the logic signal is furnished by the host connection, be it a microprocessor or an SPI isolation device. By simply pulling down the emitter leg of an NPN having a VREG base potential through a known resistance, a speciic current is formed for a logic low input signal. In the case of the component values shown, the current is about 2mA for a logic low, and conversely, the transistor is essentially turned off with a logic high (~0mA for 5V logic). Since the collector current is nearly identical to the emitter current, the same current pulls on the next higher cascode circuit. Since that next circuit is the same as the irst, the voltage on the upper emitter resistor reproduces that of the bottom circuit logic level for the upper IC. This continues up the daisy chain, eventually terminating at the top potential of the battery stack. Since each IC is provided the same input waveforms, this structure forms a parallel bus from a logical perspective, even though each IC is operating at a different potential in the stack. The NPN transistors at the top IC source the logic current directly from the battery stack. Only small base currents low from any VREG output. The 600V collector diodes provide reverse-voltage protection in the event a battery group interconnection is lost, perhaps during service (these are not required for functionality and could be omitted in some situations). Bringing Data Down the Stack The SDO cascode chain is similar in concept, except the current starts at the top of the stack and lows downward. At the top IC, a PNP transistor 8 with its base connected to the local V– pin has current injected into its emitter by a pullup resistor. Here again, the collector current is essentially identical to the emitter current, and so current flows downward through each successive PNP and terminates into a resistor at the bottom of stack. In this case, the presence of the current in the termination resistor, about 2mA for the component values shown, forms a logic high potential for the host interface. A Schottky diode is connected from each SDO pin to the emitter of a local PNP thereby allowing any LTC6802 on the stack to divert the pullup current to the local V– when outputting a logic low. This effectively turns off the emitter current to the local PNP transistor and all points lower in the stack, so the voltage on the bottom termination resistor then drops to a logic low level. Since each SDO pin can force a low level, this forms a wire-OR function that is equivalent to paralleled connections as far as the host interface is concerned. Note the bottom of stack SDO diode is connected slightly differently; it forms a direct wire-OR at the host interface. Since the LTC6802-2 is designed to use addressed readback commands, this line is properly multiplexed and no inter-IC contention occurs. To eliminate the pull-up current during standby, a general purpose N-channel MOSFET is used to interrupt the top PNP emitter current when the watchdog timer bit goes low. The watchdog timeout will release when clock activity is present, so the SDO line will reactivate as needed. Here again, an NPN is used at the top of stack to ensure the pull-up current comes directly from the battery, rather than loading VREG. Collector diodes are added here as well to provide a high reverse voltage protection capability, plus some added series resistance is included to protect the lower transistor emitters from transient energy (once again, these protection parts don’t add any other functionality to the data transmission and could be omitted in some circumstances). External SPI Advantages Since the LTC6802-2 uses a parallel addressable SPI protocol, the conventional method of connecting multiple devices in a stack is to provide isolation for each SPI connection, then parallel the signals on the host side. Isolators are relatively expensive and often need extra power circuitry, thus adding signiicantly to the total solution cost. The transistor circuitry shown here is quite inexpensive and offers the option to make certain design tradeoffs as well. With the propagation delays involved and desire to keep power fairly low, this circuit as shown still communicates at over 500kbps. Lower SPI currents could be chosen in applications that don’t demand the high data rate by simply raising the resistance values accordingly. The main feature of the transistorized SPI bus is the wide compliance range that is afforded by the unconstrained collector-base operating range of the transistors. In normal operation the VCB ranges from just less than the cells connected to the LTC6802, to some ive volts below that, depending on the logic level transmitted. This becomes important since voltage luctuations on the battery, due to load dynamics or switching transients, affect the VCB of the transistors even though the V+ and ADC cell inputs may be iltered. Some vehicle manufacturers are requiring that a BMS tolerate 1V steps with 200ns rise/fall time per cell in the stack, so this is a 12V waveform edge as seen by the transistors in a typical application. With the low collector capacitance and 2mA logic level of the transistor chain, SPI transmissions remain error free with even this high level of noise. Conclusion EVs and HEVs are here to stay. Inherently safe lithium batteries, which combine energy density, power density, and cycle life, will continue to evolve to improve the performance of these vehicles. Battery management systems using the LTC6802 extract the most driving distance and lifetime from the battery pack while lowering system cost. L Linear Technology Magazine • March 2009 DESIGN FEATURES L DC/DC Converter, Capacitor Charger Takes Inputs from 4.75V to 400V by Robert Milliken and Peter Liu Introduction High voltage power supplies and capacitor chargers are readily found in a number of applications, including professional photolashes, security control systems, pulsed radar systems, satellite communication systems, and explosive detonators. The LT3751 makes it possible for a designer to meet the demanding requirements of these applications, including high reliability, relatively low cost, safe operation, minimal board space and high performance. The LT3751 is a general purpose lyback controller that can be used as either a voltage regulator or as a capacitor charger. The LT3751 operates in boundary-mode, between continuous conduction mode and discontinuous conduction mode. Boundary-mode operation allows for a relatively small transformer and an overall reduced PCB footprint. Boundary-mode also reduces large signal stability issues that could arise from using voltagemode or PWM techniques. Regulation is achieved with a new dual, overlapping modulation technique using both damage. When used as a regulator, the LT3751’s feedback loop is internally compensated to ensure stability. The LT3751 is available in two packages, either a 20-pin exposed pad QFN or a 20-lead exposed pad TSSOP. 2V/DIV GND 250ns/DIV Figure 1. Gate driver waveform in a typical application peak primary current modulation and duty-cycle modulation, drastically reducing audible transformer noise. The LT3751 features many safety and reliability functions, including two sets of undervoltage lockouts (UVLO), two sets of overvoltage lockouts (OVLO), no-load operation, over-temperature lockout (OTLO), internal Zener clamps on all high voltage pins, and a selectable 5.6V or 10.5V internal gate driver voltage clamp (no external components needed). The LT3751 also adds a start-up/shortcircuit protection circuit to protect against transformer or external FET New Gate Driver with Internal Clamp Requires No External Components There are four main concerns when using a gate driver: output current drive capability, peak output voltage, power consumption and propagation delay. The LT3751 is equipped with a 1.5A push-pull main driver, enough to drive +80nC gates. An auxiliary 0.5A PMOS pull-up only driver is also integrated into the LT3751 and is used in parallel with the main driver for VCC voltages of 8V and below. This PMOS driver allows for rail-to-rail operation. Above 8V, the PMOS driver must be deactivated by tying its drain to VCC. Most discrete FETs have a VGS limit of 20V. Driving the FET higher than 20V could cause a short in the internal gate oxide, causing permanent DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY VTRANS 10V TO 24V T1* 1:10 + OFF ON C1 10µF TO µP RVTRANS CHARGE CLAMP RDCM R7 18.2k VCC R8 40.2k LT3751 RVOUT • C4 100µF • VOUT 50V TO 450V + C5 0.47µF VOUT 100V/DIV GND HVGATE LVGATE CSP FAULT M1 VCC R5 6mΩ 1W UVLO1 R2, 475k OVLO1 VCC D2 DONE R1, 154k VTRANS C2 2.2µF s5 R6 40.2k C3 680µF D1 CSN UVLO2 ALL RESISTORS ARE 0805, 1% RESISTORS UNLESS OTHERWISE NOTED D1,D2: VISHAY MURS260 M1: IRF3710Z T1: WURTH 750310349 FB OVLO2 GND RBG R9 4.7nF Y RATED * LIMIT OUTPUT POWER TO 40W FOR 65°C T1 MAX AMBIENT OPERATION IIN(AVG) 2A/DIV 0 VIN = 24V COUT = 100µF 20ms/DIV Figure 3. Isolated high voltage capacitor charger charging waveform Figure 2. Isolated high voltage capacitor charger from 10V to 24V input Linear Technology Magazine • March 2009 9 L DESIGN FEATURES DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY VTRANS 10V TO 24V T1** 1:10 + R6 40.2k C3 680µF OFF ON C1 10µF TO µP RVTRANS CHARGE CLAMP RDCM R7 18.2k VCC R8 40.2k LT3751 RVOUT HVGATE LVGATE CSP R1, 154k UVLO1 R2, 475k OVLO1 VCC D2 • C4 100µF • VOUT 400V ALL RESISTORS ARE 0805, 1% RESISTORS UNLESS OTHERWISE NOTED + C5 0.47µF C4: CDE 380LX101M500J042 C5: TDK CKG57NX7R2J474M D1,D2: VISHAY MURS260 M1: IRF3710Z T1: WURTH 750310349 DONE FAULT VTRANS C2 2.2µF s5 D1 CSN M1 * USE TWO SERIES 1206, 1% RESISTORS FOR R10 R10: 249k s2 VCC R5 6mΩ 1W ** LIMIT OUTPUT POWER TO 40W FOR 65°C T1 MAX R10* AMBIENT OPERATION 499k UVLO2 OVLO2 FB C6 10nF GND RBG R11 1.54k R9 787Ω Figure 4. A 10V to 24V input, 400V regulated power supply damage. To alleviate this issue, the LT3751 has an internal, selectable 5.6V or 10.5V gate driver clamp. No external components are needed, not even a capacitor. Simply tie the CLAMP pin to ground for 10.5V operation or tie to VCC for 5.6V operation. Figure 1 shows the gate driver clamping at 10.5V with a VCC voltage of 24V. Not only does the internal clamp protect the FET from damage, it also reduces the amount of energy injected into the gate. This increases overall eficiency and reduces power consumption in the gate driver circuit. The gate driver overshoot is very minimal, as seen in Figure 1. Placing the external FET closer to the LT3751 HVGATE pin reduces overshoot. High Voltage, Isolated Capacitor Charger from 10V to 24V Input The LT3751 can be conigured as a fully isolated stand-alone capacitor charger using a new differential discontinuous-conduction-mode (DCM) comparator—used to sense the boundary-mode condition—and a new differential output voltage (VOUT) comparator. The differential operation of the DCM comparator and VOUT comparator allow the LT3751 to accurately operate from high voltage input supplies of greater than 400V. Likewise, the LT3751’s DCM comparator and VOUT comparator can work with input supplies down to 4.75V. This accommodates an unmatched range of power sources. Figure 2 shows a high voltage capacitor charger driven from an input supply ranging from 10V to 24V. Only ive resistors are needed to operate the LT3751 as a capacitor charger. The output voltage trip point can be continuously adjusted from 50V to 450V by adjusting R9 given by: 0.98 • N R9 = • R8 VOUT(TRIP) + VDIODE The LT3751 stops charging the output capacitor once the programmed output voltage trip point (VOUT(TRIP)) is reached. The charge cycle is repeated by toggling the CHARGE pin. The maximum charge/discharge rate in 402 90 EFFICIENCY VDRAIN 20V/DIV GND GND IPRIMARY 5A/DIV 0 IPRIMARY 5A/DIV 0 10µs/DIV a. Switching waveform for IOUT = 100mA 401 80 75 VOUT (V) VDRAIN 20V/DIV EFFICIENCY (%) 85 LOAD REGULATION 400 70 65 10µs/DIV b. Switching waveform for IOUT = 10mA 60 0 20 40 60 80 LOAD CURRENT (mA) 399 100 c. Efficiency and load regulation Figure 5. High voltage regulator performance 10 Linear Technology Magazine • March 2009 DESIGN FEATURES L the output capacitor is limited by the temperature rise in the transformer. Limiting the transformer surface temperature in Figure 2 to 65°C with no air low requires the average output power to be ≤40W given by: VOUT AC RIPPLE 10V/DIV IIN(AVG) 20mA/DIV 0 PAVG = 1 C • FREQUENCY • 2 OUT 2VOUT(TRIP) • VRIPPLE – VR2IPPLE ( 2s/DIV ) Figure 6. The LT3751 protecting the output during a no-load condition ≤ 40 W where VOUT(TRIP) is the output trip voltage, VRIPPLE is the ripple voltage on the output node, and frequency is the charge/discharge frequency. Two techniques are used to increase the available output power: increase the airlow across the transformer, or increase the size of the transformer itself. Figure 3 shows the charging waveform and average input current for a 100µF output capacitor charged to 400V in less than 100ms (R9 = 976Ω). For output voltages higher than 450V, the transformer in Figure 2 must be replaced with one having higher primary inductance and a higher turns ratio. Consult the LT3751 data sheet for proper transformer design procedures. High Voltage Regulated Power Supply from 10V to 24V Input The LT3751 can also be used to convert a low voltage supply to a much higher voltage. Placing a resistor divider from the output node to the FB pin and ground causes the LT3751 to operate as a voltage regulator. Figure 4 shows a 400V regulated power supply operating from an input supply range of 10V to 24V. The LT3751 uses a regulation control scheme that drastically reduces audible noise in the transformer and the input and output ceramic bulk capacitors. This is achieved by using an internal 26kHz clock to synchronize the primary winding switch cycles. Within the clock period, the LT3751 modulates both the peak primary current and the number of switching cycles. Figures 5a and 5b show heavy-load and light-load waveforms, respectively, while Figure 5c shows eficiency over most of the operating range for the application in Figure 4. The clock forces at least one switch cycle every period which would overcharge the output capacitor during a no-load condition. The LT3751 handles no-load conditions and protects against over-charging the output node. Figure 6 shows the LT3751 protecting during a no-load condition. Resistors can be added to RVOUT and RBG to add a second layer of protection, or they can be omitted to reduce component count by tying RVOUT and RBG to ground. The trip level for the VOUT comparator is typically set 20% higher than the nominal regulation voltage. If the resistor divider were to fail, the VOUT comparator would disable switching when the output climbed to 20% above nominal. DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY T1*** D1 1:3 F1, 1A C3 100µF 450V + R7 88.7k + 7.5k OFF ON RVTRANS CHARGE RDCM CLAMP VCC C1 10µF LT3751 RVOUT DONE TO µP FAULT R1** 1.5M C4 220µF 550V • R8 137k ×3 R11 14.7k + 17.4k VCC M1 FQB4N80 R2**, 9M OVLO1 CSP R3, 154k VCC UVLO2 R4, 475k CSN OVLO2 4.7nF Y RATED GND RBG R5 1.11k FB C5 0.47µF 630V ALL RESISTORS ARE 0805, 1% RESISTORS UNLESS OTHERWISE NOTED UVLO1 VTRANS + R9 66.5k R10* 208k R13,20Ω HVGATE LVGATE • VOUT 500V R12 68mΩ 1/4W C4: HITACHI PS22L221MSBPF C5: TDK CKG57NX7R2J474M T1: COILCRAFT HA4060-AL D1,D2: VISHAY US1M F1: BUSSMANN PCB-1-R * USE THREE SERIES 1206, 0.1% RESISTORS FOR R6 & R10 R6: 249k ×2 + 127k R10: 66.5k ×2 + 75k ** USE TWO SERIES 1206, 1% RESISTORS FOR R1 & R2 R1: 750k ×2 R2: 4.53M ×2 530 1000 520 850 VOUT,TRIP 700 510 CHARGE TIME 550 500 490 100 CHARGE TIME (ms) VCC 10V TO 24V C2 2.2µF 630V s5 R6* 625k D2 VOUT,TRIP (V) VTRANS 100V TO 400V DC 200 300 INPUT VOLTAGE (V) 400 400 Figure 8. Isolated capacitor charger VOUT(TRIP) and charge time with respect to input voltage *** OUTPUT POWER LIMITED TO 20W FOR 65°C T1 AMBIENT OPERATION Figure 7. A 100V to 400V input, 500V output, isolated capacitor charger Linear Technology Magazine • March 2009 11 L DESIGN FEATURES DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY VTRANS 100V TO 400V DC T1*** D1 1:3 F1, 1A + R7 95.3k C3 100µF OFF ON VCC 10V TO 24V C1 10µF R6* 615k C2 2.2µF s5 RVTRANS CHARGE RDCM CLAMP VCC LT3751 D2 • C4 100µF • VOUT 400V ALL RESISTORS ARE 0805, 1% RESISTORS UNLESS OTHERWISE NOTED + R8* 411k C5 0.47µF C4: CDE 380LX101M500J042 C5: TDK CKG57NX7R2J474M T1: COILCRAFT HA4060-AL D1,D2: VISHAY US1M F1: BUSSMANN PCB-1-R R9 66.5k RVOUT * USE THREE SERIES 1206, 1% RESISTORS FOR R6 & R8 R6: 205k ×3 R8: 137k ×3 DONE TO µP FAULT R13,20Ω HVGATE LVGATE VCC ** USE TWO SERIES 1206, 1% RESISTORS FOR R1, R2 & R11 R1: 750k ×2 R2: 4.53M ×2 R11: 249k ×2 M1 FQB4N80 R1**, 1.5M UVLO1 VTRANS R2**, 9M OVLO1 R3, 154k CSP UVLO2 VCC R4, 475k OVLO2 GND RBG *** OUTPUT POWER LIMITED TO 20W FOR 65°C T1 AMBIENT OPERATION R10 68mΩ ¼W CSN R11** 499k FB R12 1.54k C6 10nF Figure 9. A 100V to 400V input, 400V output, capacitor charger and voltage regulator Note that the FB pin of the LT3751 can also be used for a capacitor charger. The LT3751 operates as a capacitor charger until the FB pin reaches 1.225V, after which the LT3751 operates as a voltage regulator. This keeps the capacitor topped-off until the application needs to use its energy. The output resistor divider forms a leakage path from the output capacitor to ground. When the output voltage droops, the LT3751 feedback circuit will keep the capacitor topped- off with small, low current bursts of charge as shown in Figure 6. High Input Supply Voltage, Isolated Capacitor Charger As mentioned above, the LT3751 differential DCM and VOUT comparators allow the part to accurately work from very high input supply voltages. An ofline capacitor charger, shown in Figure 7, can operate with DC input voltages from 100V to 400V. The transformer provides galvanic isolation from 90 398 OUTPUT VOLTAGE (V) EFFICIENCY (%) 80 70 60 VIN = 100V VIN = 250V VIN = 400V 50 40 0 25 50 397 396 IOUT = 10mA IOUT = 25mA IOUT = 50mA 75 395 100 OUTPUT CURRENT (mA) 200 300 INPUT VOLTAGE (V) a. Overall efficiency b. Line regulation 400 the input supply to output node—no additional magnetics required. Input voltages greater than 80V require the use of resistor dividers on the DCM and VOUT comparators (charger mode only). The accuracy of the VOUT trip threshold is heightened by increasing current IQ through R10 and R11; however, the ratio of R6/R7 should closely match R10/R11 with tolerances approaching 0.1%. A trick is to use resistor arrays to yield the desired ratio. Achieving 0.1% ratio accuracy is not dificult and can reduce the overall cost compared to using individual 0.1% surface mount resistors. Note that the absolute value of the individual resistors is not critical, only the ratio of R6/R7 and R10/R11. The DCM comparator is less critical and can tolerate resistance variations greater than 1%. The 100V to 400VDC input capacitor charger has an overall VOUT(TRIP) accuracy of better than 6% over the entire operating range using 0.1% resistor dividers. Figure 8 shows a typical performance for VOUT(TRIP) and charge time for the circuit in Figure 7. Figure 10. High voltage input and output regulator performance 12 Linear Technology Magazine • March 2009 DESIGN FEATURES L ISOLATION BOUNDARY DANGER HIGH VOLTAGE! OPERATION BY HIGH VOLTAGE TRAINED PERSONNEL ONLY D2 R2, 10Ω T1 • Npb VTRANS 100V TO 200V DC F1, 2A + R1 49.9k 1/2W M1 C2 1µF D1 1 VCC Ns Np • R10, 4.3M 2 RVOUT FAULT HVGATE LVGATE CSP R15 221k R16, 1k OVLO1 Nsb • M2 C5 0.01µF 630V CSN C9 3.3µF R13 5.11Ω 1 R12, 442k GND RBG R8 2.49k 1 2 1 R7 475Ω C8 22nF LT4430 R17 3.16k OC FB D7 U1 FB OVLO2 VIN COMP C10 0.47µF 2 UVLO2 2 U2 D6 VCC R6 40mΩ 1/4W R14 249k VOUT 282V 225mA 2 2 D4 R11, 84.5k 4.7nF Y RATED C7 400µF 330V RDCM DONE UVLO1 VCC + C6 0.1µF 630V R4, 105k LT3751 R9, 2.7M VTRANS D5 • D3 R5, 210k ALL RESISTORS ARE 0805,1% RESISTORS UNLESS OTHERWISE NOTED C7: 330FK400M22X38 D1: 12V ZENER D2: MURS140 D3: P6kE200A D4, D5: STTH112A D6: BAT54 D7: BAS516 M1: IRF830 M2: STB11NM60FD T1: TDK SRW24LQ-UxxH015 (Np:Ns:Npb:Nsb=1:2:0.08:0.08) U1: PS2801-1 U2: LT4430 RVTRANS TO µP 1 C4 1µF 250V 1 s2 R3 210k CHARGE CLAMP OFF ON 1 C1 100pF 1 C3 22µF 350V 1 s2 OPTO GND VCC R18 274Ω 2 1 2 1 Figure 11. Fully isolated, high output voltage regulator High Input Supply Voltage, Non-Isolated Capacitor Charger/Regulator The FB pin of the LT3751 can also be conigured for charging a capacitor from a high input supply voltage. Simply tie a resistor divider from the output node to the FB pin. The resistor dividers on the RVTRANS and RDCM pins can tolerate 5% resistors, and all the RV(OUT) and RBG pin resistors are removed. This lowers the number and the tolerance of required components, reducing board real estate and overall design costs. With the output voltage resistor divider, the circuit in Figure 9 is also a fully functional, high-eficiency voltage regulator with load and line regulation better than 1%. Eficiency and line regulation for the circuit in Figure 9 are shown in Figure 10a and Figure 10b, respectively. Alternatively, a resistor can be tied from VOUT to the OVLO1 pin or OVLO2 pin. This mimics the VOUT comparator, stopping charging once the target voltage is reached. The FB pin is tied to ground. The CHARGE pin must be toggled to initiate another charge sequence, thus the LT3751 operates as a capacitor charger only. Resistor R12 is omitted from Figure 9 and resistor R11 is tied from VOUT directly to OVLO1 or OVLO2. R11 is calculated using the following equation: VDRAIN 100V/DIV VDRAIN 100V/DIV GND IPRIMARY 2A/DIV 0 GND IPRIMARY 2A/DIV 0 R11 = VOUT(TRIP) − 1.225 50µA Note that OVLO1 or OVLO2 will cause the FAULT pin to indicate a fault when the target outpaut voltage, VOUT(TRIP) , is reached. High Voltage Input/Output Regulator with Isolation Using a resistor divider from the output node to the FB pin allows regulation but does not provide galvanic isolation. Two auxiliary windings are added to the transformer in circuit shown in Figure 11 to drive the FB pin, the continued on page 42 20µs/DIV 20µs/DIV a. IOUT = 225mA b. IOUT = 7.1mA Figure 12. Switching waveforms Linear Technology Magazine • March 2009 13 L DESIGN FEATURES How to Choose a Voltage Reference by Brendan Whelan Why Voltage References? 5V It is an analog world. All electronic devices must in some way interact with the “real” world, whether they are in an automobile, microwave oven or cell phone. To do that, electronics must be able to map real world measurements (speed, pressure, length, temperature) to a measurable quantity in the electronics world (voltage). Of course, to measure voltage, you need a standard to measure against. That standard is a voltage reference. The question for any system designer is not whether he needs a voltage reference, but rather, which one? A voltage reference is simply that—a circuit or circuit element that provides a known potential for as long as the circuit requires it. This may be minutes, hours or years. If a product requires information about the world, such VIN 18k LTC1286 1 2 LT1634-4.096 0.1µF 3 4 VREF VCC +IN CLK –IN DOUT GND CS/SHDN 8 5V 7 6 5 0.1µF µC/µP SERIAL INTERFACE Figure 1. Typical use of a voltage reference for an ADC Reference Specifications as battery voltage or current, power consumption, signal size or characteristics, or fault identiication, then the signal in question must be compared to a standard. Each comparator, ADC, DAC, or detection circuit must have a voltage reference in order to do its job (Figure 1). By comparing the signal of interest to a known value, any signal may be quantiied accurately. Voltage references come in many forms and offer different features, but in the end, accuracy and stability are a voltage reference’s most important features, as the main purpose of the reference is to provide a known output voltage. Variation from this known value is an error. Voltage reference specifications usually predict the uncertainty of the reference under Table 1. Specifications for high performance voltage references Temperature Initial Coefficient Accuracy LT1031 5ppm/°C 0.05% VOUT Voltage Noise* Long-Term Drift Package Buried Zener 10V 0.6ppm 15ppm/kHr H 2.5ppm IS Architecture 1.2mA LT1019 5ppm/°C 0.05% 650µA Bandgap 2.5V, 4.5V, 5V, 10V LT1027 5ppm/°C 0.05% 2.2mA Buried Zener 5V 0.6ppm LT1021 5ppm/°C 0.05% 800µA Buried Zener 5V, 7V, 10V 0.6ppm 15ppm/kHr SO-8, PDIP, H LTC6652 5ppm/°C 0.05% 350µA Bandgap 1.25V, 2.048V, 2.5V, 3V, 3.3V, 4.096V, 5V 2.1ppm 60ppm/√kHr MSOP LT1236 5ppm/°C 0.05% 800µA Buried Zener 5V, 10V 0.6ppm 20ppm/kHr SO-8, PDIP LT1461 3ppm/°C 0.04% 35µA Bandgap 2.5V, 3V, 3.3V, 4.096V, 5V 8ppm 60ppm/√kHr SO-8 LT1009 15ppm/°C 0.2% 1.2mA Bandgap 2.5V 20ppm/kHr MSOP-8, SO-8, Z LT1389 20ppm/°C 0.05% 700nA Bandgap 1.25V, 2.5V, 4.096V, 5V 20ppm SO-8 LT1634 10ppm/°C 0.05% 7µA Bandgap 1.25V, 2.5V, 4.096V, 5V 6ppm SO-8, MSOP-8, Z LT1029 20ppm/°C 0.20% 700µA Bandgap 5V LM399 1ppm/°C 2% 15mA Buried Zener 7V LTZ1000 0.05ppm/°C 4% Buried Zener 7.2V SO-8, PDIP 20ppm/ month SO-8, PDIP 20ppm/kHr Z 1ppm 8ppm/√kHr H 0.17ppm 2µV/√kHr H *0.1Hz–10Hz, Peak-to-Peak 14 Linear Technology Magazine • March 2009 DESIGN FEATURES L Initial Accuracy The variance of output voltage as measured at a given temperature, usually 25°C. While the initial output voltage may vary from unit to unit, if it is constant for a given unit, then it can be easily calibrated. Temperature Drift This speciication is the most widely used to evaluate voltage reference performance, as it shows the change in output voltage over temperature. Temperature drift is caused by imperfections and nonlinearities in the circuit elements, and is often nonlinear as a result. For many parts, the temperature drift, TC, speciied in ppm/°C, is the dominant error source. For parts with consistent drift, calibration is possible. A common misconception regarding temperature drift is that it is linear. This leads to assumptions such as “the part will drift a lesser amount over a smaller temperature range.” Often the opposite is true. TC is generally speciied with a “box method” in order to give an understanding of the likely error over the entire operating temperature range. It is a calculated value based only on minimum and maximum values of voltage, and does not take into account the temperatures at which these extrema occur. For voltage references that are very linear over the speciied temperature range, or for those that are not carefully tuned, the worst-case error can be assumed to be proportional to the temperature range. This is because the maximum and minimum output voltages are very likely to be found at the maximum and minimum operating temperatures. However, for very carefully tuned references, often identiied by their very low temperature drift, the nonlinear nature of the reference may dominate. For example, a reference speciied as 100ppm/°C tends to appear quite linear over any temperature range, as the drift due to component mismatches completely obscures the Linear Technology Magazine • March 2009 The best use of the temperature drift speciication is to calculate maximum total error over the speciied temperature range. It is generally inadvisable to calculate errors over unspeciied temperature ranges unless the temperature drift characteristics are well understood. 1.003 OUTPUT VOLTAGE (NORMALIZED) (V) certain conditions using the following deinitions. 10ppm/oC FULL TEMP RANGE “BOX” 1.002 1.001 LT1019 CURVE 1.000 0.999 5ppm/oC 0oC TO 70oC “BOX” 0.998 UNCOMPENSATED “STANDARD” BANDGAP DRIFT CURVE 0.997 –50 –25 50 25 75 0 TEMPERATURE (˚C) 100 125 Figure 2. Voltage reference temperature characteristics inherent nonlinearity. In contrast, the temperature drift of a reference speciied as 5ppm/°C will be dominated by the nonlinearities. Voltage references come in many forms and offer different features, but in the end, accuracy and stability are a voltage reference’s most important features, as the main purpose of the reference is to provide a known output voltage. Variation from this known value is an error. Voltage reference specifications usually predict the uncertainty of the reference under certain conditions. This can be easily seen in the output voltage vs temperature characteristic of Figure 2. Note that there are two possible temperature characteristics represented. An uncompensated bandgap appears as a parabola, with minima at the temperature extrema and maximum in the middle. A temperature compensated bandgap, such as the LT1019, shown here, appears as an “S” shaped curve, with greatest slope near the center of the temperature range. In the latter case, nonlinearity is exacerbated so that the aggregate uncertainty over temperature is reduced. Long Term Stability This is a measure of the tendency of a reference voltage to change over time, independent of other variables. Initial shifts are largely caused by changes in mechanical stress, usually from the difference in expansion rates of the lead frame, die and mold compound. This stress effect tends to have a large initial shift that reduces quickly with time. Initial drift also includes changes in electrical characteristics of the circuit elements, including settling of device characteristics at the atomic level. Longer-term shifts are caused by electrical changes in the circuit elements, often referred to as “aging.” This drift tends to occur at a reduced rate as compared to initial drift, and to further reduce over time. It is therefore often speciied as drift/√khr. Voltage references tend to age more quickly at higher temperatures. Thermal Hysteresis This often-overlooked speciication can also be a dominant source of error. It is mechanical in nature, and is the result of changing die stress due to thermal cycling. Hysteresis can be observed as a change in output voltage at a given temperature after a large temperature cycle. It is independent of temperature coeficient and time drift, and reduces the effectiveness of initial voltage calibration. Most references tend to vary around a nominal output voltage during subsequent temperature cycles, so thermal hysteresis is usually limited to a predictable maximum value. Each manufacturer has their own method for specifying this parameter, so typical values can be misleading. Distribution data, as provided in data sheets such as the LT1790 and LTC6652, is far more useful when estimating output voltage error. 15 L DESIGN FEATURES 5V 4.7M VOUT 1.25V 2.6V b VIN b 18V LT1389-1.25 Figure 3. Shunt voltage reference Other Specifications Additional speciications that may be important, depending on application requirements include: q Voltage Noise q Line Regulation/PSRR q Load Regulation q Dropout Voltage q Supply Range q Supply Current Reference Types The two main types of voltage reference are shunt and series. See Table 2 for a list of Linear Technology series and shunt voltage references. Shunt References The shunt reference is a 2-terminal type, usually designed to work over a speciied range of currents. Though most shunts are of the bandgap type and come in a variety of voltages, they can be thought of and are as simple to use as a Zener diode. The most common circuit ties one terminal of the reference to ground and the other terminal to a resistor. The remaining terminal of the resistor is VOUT = 2.5V 1MF Figure 4. Series voltage reference then tied to a supply. This becomes, in essence, a three terminal circuit. The shared reference/resistor terminal is the output. The resistor must be chosen such that the minimum and maximum currents through the reference are within the speciied range over the entire supply range and load current range. These references are quite easy to design with, provided the supply voltage and load current do not vary much. If either, or both, may change substantially, then the resistor must be chosen to accommodate this variance, often forcing the circuit to dissipate signiicantly more power than required for the nominal case. It can be considered to function like a class A ampliier, in that sense. Advantages of shunt references include simple design, small packages and good stability over wide current and load conditions. In addition, they are easily designed as negative voltage references and can be used with very high supply voltages, as the external resistor holds off most of the potential, or very low supplies, as the output can be as little as a few millivolts below + 7.5k LT1790-2.5 0.1MF the supply. Linear Technology offers shunt products including the LT1004, LT1009, LT1389, LT1634, LM399 and LTZ1000. A typical shunt circuit can be seen in Figure 3. Series References Series references are three (or more) terminal devices. They are more like low dropout (LDO) regulators, so they have many of the same advantages. Most notably, they consume a relatively ixed amount of supply current over a wide range of supply voltages, and they only conduct load current when the load demands it. This makes them ideal for circuits with large changes in supply voltage or load current. They are especially useful in circuits with very large load currents as there is no series resistor between the reference and supply. Series products available from Linear Technology include the LT1460, LT1790, LT1461, LT1021, LT1236, LT1027, LTC6652, LT6660, and many others. Products such as the LT1021 and LT1019 may be operated either as a shunt or a series voltage reference. A series reference circuit is illustrated in Figure 4. Q13 200k Q12 Q4 240mV, +0.8mV/°C Q3 20pF 20pF – + Q11 Q1 50k 600k – + Q10 300k Q5 2.8k 60mV, +0.2mV/°C Q9 Q14 – + VREF 1.235V, 0mV/°C 14.9k Q1 360mV, +1.2mV/°C Q2 135k 82.4k Q8 Q6 575mV, –2.2mV/°C 500k 5007 2.5k 60k – Figure 5. A bandgap circuit is designed for a theoretically zero temperature coefficient. 16 2k Figure 6. A 200mV reference circuit Linear Technology Magazine • March 2009 DESIGN FEATURES L Reference Circuits Linear Technology Magazine • March 2009 Type Series Zener-Based References The buried Zener type reference is a relatively simple design. A Zener (or avalanche) diode has a predictable reverse voltage that is fairly constant over temperature and very constant over time. These diodes are often very low noise and very stable over time if held within a small temperature range, making them useful in applications where changes in the reference voltage must be as small as possible. This stability can be attributed to the relatively small number of components and die area as compared to other types of reference circuits, as well as the careful construction of the Zener element. However, relatively high variances in initial voltage and temperature drift are common. Additional circuitry may be added to compensate these imperfections, or to provide a range of output voltages. Both shunt and series references use Zener diodes. Devices like the LT1021, LT1236 and LT1027 use internal current sources and ampliiers to regulate the Zener voltage and current to increase stability, as well as to provide various output voltages such as 5V, 7V and 10V. This additional circuitry makes the Zener diode more compatible with a wide variety of application circuits, but requires some additional supply headroom and may cause additional error. Alternatively, the LM399 and LTZ1000 use internal heating elements and additional transistors to stabilize the temperature drift of the Zener diode, giving the best combination of temperature and time stability. In addition, these Zener-based products have extraordinarily low noise, giving the best possible performance. The LTZ1000 exhibits 0.05ppm/°C temperature drift, 2µV/√kHr long term stability and 1.2µVP-P noise. To give some perspective, in a laboratory instrument, the total uncertainty in the LTZ1000’s reference voltage due Table 2. Voltage references available from Linear Technology Shunt There are many ways to design a voltage reference IC. Each has speciic advantages and disadvantages. Part Description LT1019 Precision Bandgap LT1021 Precision Low Noise Buried Zener LT1027 Precision 5V Buried Zener LT1031 Precision Low Noise/Low Drift 10V Zener LT1236 Precision Low Noise Buried Zener LT1258 Micropower LDO Bandgap LT1460 Micropower Precision Bandgap LT1461 Micropower Ultra-Precision Bandgap LT1790 Micropower Low Dropout Bandgap LT1798 Micropower LDO Bandgap LT6650 Micropower 400mV/Adjustable Bandgap LTC6652 Precision Low Noise LDO Bandgap LM129 Precision 6.9V Buried Zener LM185 Micropower 1.2V/2.5V Zener LM399 Precision 7V Heated Zener LT1004 Micropower 1.2V/2.5V Bandgap LT1009 Precision 2.5V Bandgap LT1029 5V Bandgap LT1034 Micropower Dual (1.2V Bandgap/7V Zener) LT1389 Nanopower Precision Bandgap LT1634 Micropower Precision Bandgap LTZ1000 Ultra-Precision Heated Zener to noise and temperature would be only about 1.7ppm plus a fraction of 1ppm per month due to aging. Bandgap References While Zener diodes can be used to make very high performance references, they lack lexibility. Speciically, they require supply voltages above 7V and they offer relatively few output voltages. In contrast, bandgap references can produce a wide variety of output voltages with little supply headroom—often less than 100mV. Bandgap references can be designed to provide very precise initial output voltages and low temperature drift, eliminating the need for time-consuming in-application calibration. Bandgap operation is based on a basic characteristic of bipolar junction transistors. Figure 5 shows a simpliied version of the LT1004 circuit, a basic bandgap. It can be shown that a mismatched pair of bipolar junction transistors has a difference in VBE that is proportional to temperature. This difference can be used to create a current that rises linearly with temperature. When this current is driven through a resistor and a transistor, the change over temperature of the baseemitter voltage of the transistor cancels the change in the voltage across the resistor if it is sized properly. While this cancellation is not completely linear, it can be compensated with additional circuitry to yield very low temperature drift. The math behind the basic bandgap voltage reference is interesting in that it combines known temperature coef17 L DESIGN FEATURES icients with unique resistor ratios to produce a voltage reference with theoretically zero temperature drift. Figure 5 shows two transistors scaled so that the emitter area of Q10 is 10-times that of Q11, while Q12 and Q13 hold their collector currents equal. This creates a known voltage between the bases of the two transistors of: ΔVBE = VS LT6700-1 LT6700HV-1 – –INB OUTB + – COMP A OUTA + +INA 18 VS 400mV REFERENCE AREA Q10 kT • ln q AREA Q11 where k is the Boltzmann constant in J/kelvin (1.38 × 10-23), T is temperature in kelvin (273 + T(°C)) and q is the charge of an electron in coulombs (1.6x10-19). At 25°C, kT/q has a value of 25.7mV with a positive temperature coeficient of 86µV/°C. ΔVBE is this voltage times ln(10), or 2.3, for a 25°C voltage of approximately 60mV with a tempco of 0.2mV/°C. Applying this voltage to the 50k resistor tied between the bases creates a current that is proportional to temperature. This current biases a diode, Q14 with a 25°C voltage of 575mV with a –2.2mV/°C temperature coeficient. Resistors are used to create voltage drops with positive tempcos, which are added to the Q14 diode voltage, thus producing a reference voltage potential of approximately 1.235V with theoretically 0mV/°C temperature coeficient. These voltage drops are shown in Figure 5. The balance of the circuit provides bias currents and output drive. Linear Technology produces a wide variety of bandgap references, including the LT1460, a small and inexpensive precision series reference, the LT1389, an ultralow power shunt reference, and the LT1461 and LTC6652, which are very high precision, low drift references. Available output voltages include 1.2V, 1.25V, 2.048V, 2.5V, 3.0V, 3.3V, 4.096V, 4.5V, 5V and 10V. These reference voltages can be provided over a wide range of supplies and load conditions with minimal voltage and current overhead. Products may be very precise, as with the LT1461, LT1019, LTC6652 and LT1790; very small, as with the LT1790 and LT1460 (SOT23), or COMP B GND Figure 7. The LT6700 allows comparisons with thresholds as low as 400mV. LT6660 in a 2mm × 2mm DFN package; or very low power, such as the LT1389, which requires only 800nA. While Zener references often have better performance in terms of noise and long term stability, new bandgap references such as the LTC6652, with 2ppm peak-to-peak noise (0.1Hz to 10Hz) are narrowing the gap. Fractional Bandgap References These are references based on the temperature characteristics of bipolar transistors, but with output voltages that may be as low as a few millivolts. They are useful for very low voltage circuits, especially in comparator applications where the threshold must 1M be less than a conventional bandgap voltage (approximately 1.2V). Figure 6 shows the core circuit from the LM10, which combines elements that are proportional and inversely proportional to temperature in a similar fashion to the normal bandgap reference to obtain a constant 200mV reference. A fractional bandgap usually uses a ΔVBE to generate a current that is proportional to temperature, and a VBE to generate a current that is inversely proportional. These are combined in the proper ratio in a resistor element to generate a temperature-invariant voltage. The size of the resistor may be varied to alter the reference voltage without affecting the temperature characteristic. This differs from a traditional bandgap circuit in that the fractional bandgap circuit combines currents, while the traditional circuits tend to combine voltages, usually a base-emitter voltage and an I•R with opposite TC. Fractional bandgaps like the LM10 circuit are based in part on a subtraction as well. The LT6650 has a 400mV reference of this type, combined with an ampliier. This allows the reference voltage to be altered by changing the gain of the ampliier, and gives a buffered output. Any output voltage from 0.4V to a few millivolts below the supply voltage can be generated with this simple circuit. In a more integrated solution, the LT6700 (Figure 7) and LT6703 combine a VBATT 1.4V (MIN) 3V (NOM) LT6700-3 + 1M 0.1µF ALKALINE AA CELLS + + 1M COMP B VBATT > 1.6V 63.4k – VR = 400mV REFERENCE – VS COMP A VBATT > 2V + 261k MONITOR CONSUMES ~10µA HYSTERESIS IS APPROXIMATELY 2% OF TRIP VOLTAGE Figure 8. Higher thresholds are set by dividing the input voltage. Linear Technology Magazine • March 2009 DESIGN FEATURES L 400mV reference with comparators, and can be used as voltage monitors or window comparators. The 400mV reference allows monitoring of small input signals, which decreases the complexity of monitor circuits and enables monitoring of circuit elements working on very low supplies as well. For larger thresholds, a simple resistor divider may be added (Figure 8). Each of these products is available in a small footprint package (SOT23), consumes low power (less than 10µA) and works on a wide supply range (1.4V to 18V). In addition, the LT6700 is available in a 2mm × 3mm DFN package and the LT6703 is available in a 2mm × 2mm DFN package. Choosing a Reference So, now, with all those options, how do you choose the right reference for your application? Here are a few hints that can narrow the range of options: q Is the supply voltage very high? Choose a shunt. q Does the supply voltage or load current vary widely? Choose a series. q Require high power efficiency? Choose a series. q Figure your real-world temperature range. Linear Technology provides guaranteed speciications and operation over various temperature ranges including 0°C to 70°C, –40°C to 85°C and –40°C to 125°C. q Be realistic about required accuracy. It is important to understand the precision required by the application. This will help identify critical speciications. With the requirement in mind, multiply temperature drift by the speciied temperature range. Add initial accuracy error, thermal hysteresis, and long term drift over the intended product life. Remove any terms that will be factory calibrated or periodically recalibrated. This gives an idea of total accuracy. For the most demanding applications, noise, line regulation and load regulation errors may also be added. As an example, a Linear Technology Magazine • March 2009 reference with 0.1% (1000ppm) initial accuracy error, 25ppm/°C temperature drift over –40°C to 85°C, 200ppm thermal hysteresis, 2ppm peak-to-peak noise and 50ppm/√kHr time drift would have a total uncertainty of over 4300ppm at the time the circuit is built. This uncertainty increases by 50ppm in the irst 1000 hours the circuit is powered. The initial accuracy may be calibrated, reducing the error to 3300ppm + 50ppm • √(t/1000hours). Linear Technology offers a wide variety of voltage reference products. These include both series and shunt references—using Zeners, bandgaps and other schemes. References are available in multiple performance and temperature grades, as well as in nearly every conceivable package type. q What is the real supply range? What is the maximum expected supply voltage? Will there be fault conditions such as battery load dump or hot-swap inductive supply spikes that the reference IC must withstand? This may signiicantly reduce the number of viable choices. q How much power can the reference consume? References tend to fall into a few categories: more than 1mA, ~500µA, <300µA, <50µA, <10µA, <1µA. q How much load current? Will the load draw substantial current or produce current that the reference must sink? Many references can provide only small currents to the load and few can absorb substantial current. The load regulation speciication is a good guide. q How much room do you have? References come in a wide variety of packages, including metal cans, plastic packages (DIP, SOIC, SOT) and very small packages, including the LT6660 in a 2mm × 2mm DFN. There is a widely held view that references in larger package sizes have less error due to mechanical stress than smaller packages. While it is true that some references may give better performance in larger packages, there is evidence that suggests performance difference has little to do directly with the package size. It is more likely that because smaller dice are used for products that are offered in smaller packages, some performance tradeoffs must be made to it the circuit on the die. Usually, the package’s mounting method makes a more signiicant performance difference than the actual package—careful attention to mounting methods and locations can maximize performance. Also, devices with smaller footprints can show reduced stress when a PCB bends compared to devices with larger footprints. This is discussed in detail in application note AN82, “Understanding and Applying Voltage References,” available from Linear Technology. Conclusion Linear Technology offers a wide variety of voltage reference products. These include both series and shunt references designed with Zeners, bandgaps and other types. References are available in multiple performance and temperature grades and nearly every conceivable package type. Products range from the highest precision available to small and inexpensive alternatives. With a vast arsenal of voltage reference products, Linear Technology’s voltage references meet the needs of almost any application. See also Linear Technology’s application note AN82 “Understanding and Applying Voltage References,” available at www.linear.com. L 19 L DESIGN FEATURES 1.2A Monolithic Buck Regulator Shrinks Supply Size and Cost with Programmable Output Current Limit by Tom Sheehan Introduction Power supplies are often overqualiied for their job. This is because power ICs often specify a current limit that is more than twice the rated output current of the device. The power supply components are sized to handle the maximum current that the IC can deliver, even if loads are unlikely to draw that current during normal operation. The components are bigger and more expensive than they need to be. There is, however, an alternative: set an accurate maximum output current on the supply once the real world load is known. Accurately setting the maximum output current reduces the required current rating of the regulator’s power path components, thus replacing big, expensive components with smaller, less expensive ones. A limit on the regulator’s maximum output limits the maximum power dissipation of both the supply and the load, thus reducing the potential for localized heating. Monitoring and controlling the output current also makes for a robust solution, which is able to withstand harsh overload and short circuit conditions. The LT3653 and LT3663 are monolithic step-down switching regulators that have an accurate output current limit programmable from 400mA to 1.2A. The LT3663 is a general purpose high voltage step-down regulator while the LT3653 is designed for use with Linear Technology Bat-Track™ enabled battery chargers and power management ICs (PMICs). The maximum input voltages of 30V (LT3653) or 36V (LT3663) with 60V transient ride through capability are well suited C3 0.1µF 10V BOOST HIGH VOLTAGE INPUT 7.5V TO 30V TRANSIENT TO 60V VIN C1 4.7µF 50V R1 27.4k USB WALL ADAPTER C4 10µF 6.3V R4 6.04k TO µC TO µC C5 0.1µF ILIM VC VBUS SW LT3653 GND VC OVGATE D1 ISENSE HVOK VOUT WALL LTC4098 R2 3.01k SYSTEM LOAD C2 22µF 6.3V L2 3.3µH SW VOUT IDGATE OVSENS D0–D2 CHRG NTC CLPROG L1 4.7µH M1 (OPTIONAL) BAT PROG BATSENS GND SINGLE-CELL Li-Ion R3 1k L1 = TDK, VLCF5020T-4R7NIR7-1 L2 = COILCRAFT, LPS4018-332MLC M1 = VISHAY, Si 2333DS D1 = DIODES INC., DFLS240 SEE THE LTC4098 DATASHEET FOR MORE INFORMATION ON CONFIGURING THE NTC BATTERY TEMPERATURE QUALIFICATION OR REDUCED IDEAL DIODE IMPEDANCE. Figure 1. Charging a single cell Li-ion battery from either a USB input or high voltage input. This solution offers a seamless, highly efficient, low part count approach to dual input charging and PowerPath™ control of a Li-ion battery-powered application. If additional integration is required for more system supplies, the LT3653 can be used in a similar fashion with the LTC3576 PMIC. 20 to automotive, industrial, distributed supply, and wall transformer applications. Programmable Output Current Limit Monolithic switching regulators typically limit the peak switch current to protect the internal switch from being damaged during an overload or short circuit event. The maximum switch peak current limit is typically more than two times the maximum output current rating of the part. While the peak switch current limit prevents overstressing the IC, it does not keep the entire application from overheating during an overload condition. For example, a regulator with an output current rating of 1A is typically capable of providing over 2A at the output. During an output overload condition, the power dissipation of the regulator could more than double, making thermal management more dificult. The LT3653 and LT3663 reduce localized hot spots by controlling the total power dissipation of the application with a programmable, accurate current limit. Conservative design principles call for power path components that are rated for worst-case currents. In the above example, where a 1A part is capable of delivering 2A, the power path components must be sized for greater than 2A, because during an output short circuit or overload the inductor and diode can conduct up to 2A. In contrast, the PowerPath components in LT3653 and LT3663 applications are sized based on the programmed maximum output current limit. Therefore, an application with a 750mA output current limit requires only 750mA rated components. This allows for smaller, lower Linear Technology Magazine • March 2009 DESIGN FEATURES L The LT3653 Plays Well with Bat-Track Battery Chargers The LT3653 is a 1.5MHz constant frequency, current mode control, step-down regulator designed for use with Linear Technology’s Bat-Trackenabled battery charger PowerPath power managers. The LT3653 steps down a high voltage input to power the system load and charge a single-cell Li-ion battery charger. Minimizing the voltage across a linear battery charger increases eficiency. To accomplish this, a BatTrack battery charger controls the LT3653’s VC Pin, overriding the error ampliier. In this way, the output voltage of the LT3653 is regulated by the battery charger to a potential slightly above the battery voltage, typically 300mV. VIN VIN 2.2µF BOOST 0.1µF LT3663 SW ON OFF DIODES, INC. DFLS240 RUN 6.8µH ISENSE ILIM VOUT 28.7k VOUT 59k 22µF FB GND 11k Figure 3. A LT3663 application producing 5V at 1.2A from an input of 7.5V to 36V. The input is capable of handling 60V transients. Linear Technology Magazine • March 2009 6 RILIM = 28.7k 5 OUTPUT VOLTAGE (V) cost devices and a smaller overall application footprint. In early product development, system designers usually don’t know how much current their load will draw. Once they choose a power supply, they are committed. However, with the programmable current limit of the LT3653 and LT3663, once the load has been fully characterized, they can change the output current limit by changing an inexpensive 1% resistor. The output current limit is implemented by monitoring and controlling the average inductor current. When an overcurrent event occurs, the regulator disables the power switch. This robust solution withstands short circuit and overload conditions throughout the entire input voltage range. 4 3 2 1 0 0 0.2 0.4 0.6 0.8 1 OUTPUT CURRENT (A) 1.2 1.4 Figure 2. The LT3663 output current limit at 1.2A Input overvoltage protection allows the LT3653 to handle 60V input transients. The HVOK pin indicates that the internal bias supplies are present and no faults have occurred (i.e., overtemperature and input overvoltage and undervoltage). The LT3653 includes internal compensation, and an internal boost diode to minimize the number of external components. The LT3653 is available in an 8-lead 2mm × 3mm DFN package with an exposed pad. Charging a Single Cell Li-Ion Battery from Either a USB or High Voltage Input Figure 1 shows a LT3653 and LTC4098 application charging a single cell Liion battery from either a USB input or high voltage input. This solution offers a seamless, highly eficient, low part count approach to dual input charging and power path control of a Li-ion battery-powered application. If additional integration is required for more system supplies, the LT3653 can be used in a similar fashion with the LTC3576 PMIC. When a high voltage input is applied, the LT3653 HVOK pin signals the LTC4098 that it is capable of delivering power. The LTC4098 takes control of the LT3653’s VC pin and regulates the output voltage to just above the battery voltage. This BatTrack function optimizes the battery charger eficiency. When present, the high voltage input supplies the battery charge current and the system load current. If the total current increases beyond the LT3653 programmed current limit, the regulator’s output voltage decreases to reduce charge current as the battery charger enters dropout. If the system load continues to increase, the battery charge current irst decreases to zero and then reverses direction to deliver power to the system load, supplementing the LT3653. The transitions between these modes of operation are seamless to the system load. The output current from the LT3653 regulator never exceeds the programmed output current limit. The LT3663 Directly Accepts 36V Inputs The LT3663 is a 1.5MHz constant frequency, current mode control, general purpose, monolithic switching regulator suited for automotive batteries, industrial power supplies, distributed supplies, and wall transformers. The LT3663 includes a low current shutdown mode, input overvoltage and undervoltage lockout, and thermal shutdown. The LT3663 is available in 8-lead (2mm × 3mm) DFN package with exposed pad. An 8-lead MSOP package with exposed pad will be available soon. The LT3663 can also function as a constant current, constant voltage (CC/CV) source to charge a supercapacitor or other energy storage device. The IC operates in constant current mode at the programmed current limit until the capacitor reaches the programmed output voltage. It then operates in a constant voltage mode to maintain that voltage. Figure 2 shows the LT3663 output current limit at 1.2A. For output currents below 1.2A the regulator is in constant voltage mode. When the output current is increased to 1.2A it goes into constant current mode. The output current is maintained at 1.2A from VOUT nominal down to 0V. 7.5V–36V to 5V Buck Regulator with 1.2A Output Current Limit Figure 3 shows a LT3663 application producing 5V at 1.2A from an input of 7.5V to 36V. The input is capable continued on page 29 21 L DESIGN FEATURES Boost Converters for Keep-Alive Circuits Draw Only 8.5µA of by Xiaohua Su Quiescent Current Introduction Industrial remote monitoring systems and keep-alive circuits spend most of their time idle. Many of these systems use batteries, so to maximize run time power losses,even during low power idle modes, must be minimized. Even at no load, power supplies draw some current to produce a regulated voltage for keep-alive circuits. The LT8410/-1 DC/DC boost converter features ultralow quiescent current and integrated high value feedback resistors to minimize the draw on the battery when electronics are idle. An entire boost converter takes very little space, as shown in Figure 1. Ultralow Quiescent Current Low Noise Boost Converter with Output Disconnect When a micropower boost converter is in regulation with no load, the input current depends mainly on two things—the quiescent current (required to keep regulation) and the output feedback resistor value. When the output voltage is high, the output feedback resistor can easily dissipate more power than the quiescent current of the IC. The quiescent current of the LT8410/-1 is a low 8.5µA, while the integrated output feedback resistors have very high values (12.4M/0.4M). This enables the LT8410/-1 to dissipate very little power in regulation at no load. In fact, the LT8410/-1 can regulate a 16V output at no load from 3.6V input with about 30µA of average input current. Figures 2, 3 and 4 show the typical quiescent and input current in regulation with no load. The LT8410/-1 controls power delivery by varying both the peak inductor current and switch off time. This control scheme results in low output voltage ripple as well as high eficiency over a wide load range. As shown in Figure 5, even with a small 0.1µF output capacitor, the output ripple is typically less than 10mV. The part also features output disconnect, which disconnects the output voltage from the input during shutdown. This output disconnect circuit also sets a maximum output current limit, allowing the chip survive output shorts. An Excellent Choice for High Impedance Batteries A power source with high internal impedance, such as a coin cell battery, may show normal output voltage on a voltmeter, but its voltage can collapse under heavy current demands. This makes it incompatible with high Figure 1. The LT8410/-1 is designed to facilitate compact board layout. switch-current DC/DC converters. The LT8410/-1 has an integrated power switch and Schottky diode, and the switch current limits are very low (25mA for the LT8410 and 8mA for the LT8410-1). This low switch current limit enables the LT8410/-1 to operate very eficiently from high impedance sources, such as coin cell batteries, without causing inrush current problems. Figure 6 shows the LT8410-1 charging an electrolytic capacitor. Without any additional external circuitry, the input current for 12 10 1000 6 4 2 AVERAGE INPUT CURRENT (µA) 8 QUIESCENT CURRENT (µA) QUIESCENT CURRENT (µA) VCC = 3.6V 10 8 6 4 2 100 VCC = 3.6V 0 – 40 0 40 80 TEMPERATURE (°C) Figure 2. Quiescent current vs temperature—not switching 22 120 0 0 4 8 12 VCC VOLTAGE (V) Figure 3. Quiescent current vs VCC voltage—not switching 16 10 0 10 20 30 OUTPUT VOLTAGE (V) 40 Figure 4. Average input current in regulation with no load Linear Technology Magazine • March 2009 DESIGN FEATURES L 100µH VIN 2.5V to 16V 2.2µF SW CAP VCC VOUT VOUT = 16V 0.1µF* LT8410 VREF SHDN CHIP ENABLE 604K GND 0.1µF FBP 412K *HIGHER VALUE CAPACITOR IS REQUIRED WHEN THE VIN IS HIGHER THAN 5V 10 100 VIN = 3.6V R1 1.30 • 1 + R2 VIN = 12V 90 8 EFFICIENCY (%) VOUT PEAK-TO-PEAK RIPPLE (mV) the SHDN pin below 0.3V shuts down the part and reduces input current to less than 1µA. When the part is on, and the SHDN pin voltage is close to 1.3V, 0.1µA current lows out of the SHDN pin. A programmable enable voltage can be set up by connecting external resistors as shown in Figure 7. The turn-on voltage for the coniguration is: 0.1µF 6 4 2 VIN = 5V 80 and the turn-off voltage is: VIN = 3.6V 70 R1 (1.24 − R3 • 10 −7 ) • 1 + − (R1• 10 −7 ) R2 60 50 0 0.01 0.1 1 LOAD CURRENT (mA) 40 0.01 10 0.1 1 10 LOAD CURRENT (mA) 100 Figure 5. General purpose bias with wide input voltage and low output voltage ripple the entire charging cycle is less than 8mA. Tiny Footprint with Small Ceramic Capacitors Available in a tiny 8-pin 2mm × 2mm DFN package, the LT8410/-1 is internally compensated and stable for a wide range of output capacitors. For most applications, using 0.1µF output capacitor and 1µF input capacitor is suficient. An optional 0.1µF capacitor at the VREF pin implements a soft-start feature. The combination of small package size and the ability to use small ceramic capacitors enable the VIN 2.5V to 16V LT8410/-1 to it almost anywhere. Figure 1 shows the size of a circuit similar to that shown in Figure 4, illustrating how little board space is required to build a full featured LT8410/-1 application. where R1, R2 and R3 are resistance in Ω. Programming the turn-on/turnoff voltage is particularly useful for applications where high source impedance power sources are used, such as energy harvesting applications. By connecting an external capacitor (typically 47nF to 220nF) to the VREF pin, a soft-start feature can be implemented. When the part is brought continued on page 29 ENABLE VOLTAGE SHDN Pin Comparator and Soft-Start Reset Feature R1 An internal comparator compares the SHDN pin voltage to an internal voltage reference of 1.3V, giving the part a precise turn-on voltage level. The SHDN pin has built-in programmable hysteresis to reject noise and tolerate slowly varying input voltages. Driving R3 CONNECT TO SHDN PIN R2 Figure 7. Programming the enable voltage by using external resistors L1 220µH C1 2.2µF TURN ON/OFF SW CAP VCC VOUT LT8410-1 VREF SHDN GND FBP C2 1.0µF C3 10000µF R1 604k R2 412k C1: 2.2μF, 16V, X5R, 0603 C2: 1.0μF, 25V, X5R, 0603* C3: 10000μF, Electrolytic Capacitor C4: 0.1μF, 16V, X7R, 0402 L1: COILCRAFT LPS3008-224ML * HIGHER CAPACITANCE VALUE IS REQUIRED FOR C2 WHEN THE VIN IS HIGHER THAN 12V SHDN VOLTAGE 2V/DIV C4 0.1µF VOUT = 16V VOUT VOLTAGE 10V/DIV INPUT CURRENT 5mA/DIV INDUCTOR CURRENT 10mA/DIV VIN = 3.6V 20s/DIV Figure 6. Capacitor charger with the LT8410-1 and charging waveforms Linear Technology Magazine • March 2009 23 L DESIGN FEATURES Industrial/Automotive Step-Down Regulator Accepts 3.6V to 36V and Includes Power-On Reset and Watchdog Timer in 3mm × 3mm QFN by Ramanjot Singh Introduction As the number of microprocessors in automotive and industrial applications continues to expand, so does the need for rugged step-down regulators that can operate over a wide input voltage range and withstand high voltage transients and output shorts. Microprocessor-based applications also require supervisory functions, such as power-on reset (POR) and watchdog timing, to ensure high system reliability. The regulator must have high eficiency at light loads to increase battery life. The LT3689 delivers all of these features in tiny 16-pin 3mm × 3mm QFN and 16-pin MSOP packages. Features of the LT3689 Step-Down Regulator The LT3689 employs a constant frequency, current mode architecture to provide 800mA of continuous output current. The part operates from a wide 3.6V to 36V input range and can protect itself from input transients up to 60V. It is internally compensated, which helps to lower the external component count. The switching frequency can be set anywhere between 350kHz and 2.2MHz by tying a resistor from VIN 4.5V TO 36V TRANSIENT TO 60V µP 2.2µF I/O Soft-Start and Output Short Circuit Protection The LT3689 includes a soft-start feature that limits the maximum inrush current during start-up and recovery from fault conditions. The soft-start circuit ramps up the peak switch current limit in approximately 150µs, reducing the peak input current. The DA pin is used to monitor the current in the catch diode. If the catch diode current at the end of switch cycle is higher than the DA current limit then the part delays the switch turn-on until the catch diode current drops below the DA current limit. This protects the LT3689 in the face of inductor current runaway situations, EN/UVLO I/O WDO LT3689 RST CWDT CPOR 68nF tRST = 157ms CPOR OUT BST SW WDI RESET CWDT 10nF tWDU = 182ms tWDL = 5.9ms VIN the R T pin to ground, allowing the designer to optimize component size and eficiency. The switching frequency can also be synchronized to an external clock for noise sensitive applications. An external resistor divider programs the output voltage to any value above the part’s 0.8V reference. Also, the boost diode is integrated into the IC to minimize solution size and cost. Figure 1 shows a typical application of LT3689. GND DA FB RT SYNC 0.1µF 12µH MBRM140 10pF 3.3V 800mA 316k 21k 100k fSW = 700kHz 22µF especially during output overload or short at high switching frequencies with high input voltages and small inductor values. Other protection features such as frequency foldback, cycle-by-cycle current limit, and thermal shutdown together ensure that the part is not damaged by excessive switch currents during startup, overload or short circuit. Pin Selectable Modes of Operation: Low Ripple Burst Mode Operation and Pulse-Skipping Mode Two modes of operation can be selected through the SYNC pin. Applying a logic low to the SYNC pin enables the low ripple Burst Mode® operation, which maintains high eficiency at light loads while keeping output ripple low. In Burst Mode operation, the LT3689 delivers single cycle bursts of current to the output capacitor followed by sleep periods. Between bursts, all circuitry associated with controlling the output switch is shutdown, reducing the VIN pin and OUT pin currents in a typical application to a mere 50µA and 75µA, respectively. As the load current decreases to a no load condition, the percentage of sleep time increases, thus decreasing average input current. A logic high on SYNC disables Burst Mode operation, allowing the part to skip pulses at light loads. The advantage of this pulse-skipping mode over Burst Mode operation is that the part continues to switch at the programmed frequency (set by R T) down to very low load currents, above 15mA at 12VIN in a typical application. Figure 1. LT3689 typical application circuit with reset time set to 157ms and watchdog timeout period set to 182ms 24 Linear Technology Magazine • March 2009 DESIGN FEATURES L The LT3689 can be shutdown by pulling the EN/UVLO pin below 0.3V. In shutdown, quiescent current is less than 0.5µA. The EN/UVLO pin can also be used to perform an accurate undervoltage lockout (UVLO) function. A resistor divider from VIN pin can be used to program the UVLO threshold of the circuit using the 1.26V accurate threshold of the EN/UVLO pin. A 4µA current hysteresis on this pin is also provided to allow the user to program desired voltage hysteresis. The LT3689 also has an internal UVLO that prevents the part from switching if VIN pin ever goes below 3.3V (typical). The part only starts switching when VIN is higher than 3.4V and EN/UVLO pin is above the 1.26V threshold. Low Dropout The LT3689 features low dropout for output voltages above 3V. The minimum operating voltage of the device is determined either by the LT3689’s internal undervoltage lockout or by its maximum duty cycle. Unlike many buck regulators, the LT3689 can extend its duty cycle by staying on for multiple cycles, provided that the boost capacitor is charged above the minimum voltage of 2.5V. Eventually, after several switching cycles, the boost capacitor discharges. Internal circuitry detects this condition and charges the boost capacitor only when needed. Also, a bigger boost capacitor allows even higher duty cycle, allowing extremely low dropout operation. The dropout voltage for a 5V typical application is about 400mV at 200mA load and 900mV at 800mA load. VOUT 2V/DIV RST 2V/DIV tRST = 165ms CPOR = 71.3nF CPOR 1V/DIV 50ms/DIV Figure 2. Power-on reset feature of LT3689 Linear Technology Magazine • March 2009 Power-On Reset (POR) Many microprocessor -based applications powered by the output of a switching regulator must know when the regulator output is ready and stable before the microprocessor starts operating. Likewise, once running, the electronic system must be warned when the regulator output has dropped below a minimum tolerable threshold, such as during overload or shutdown conditions. This is required to prevent unreliable operation and to allow the microprocessor to perform housekeeping operations before power is completely lost. The LT3689’s accurate internal voltage reference and glitch immune precision POR comparator and timer circuit feed these speciic needs of microprocessor-based applications. The switcher’s output voltage must be above 90% of programmed value for its RST pin to remain high (refer to Figure 2). The LT3689 asserts RST during power-up, power-down and brown-out conditions. Once the output voltage rises above the RST threshold, the adjustable reset timer is started and RST is released after the reset timeout period. On power-down, once output voltage drops below RST threshold, RST is held at a logic low. The reset timer is adjustable using an external capacitor. The RST pin has a weak pull up to the OUT pin. The POR comparator is designed to avoid false triggering. High frequency noise on the FB pin can falsely trip RST, particularly when the monitored output is already near the reset threshold. This can cause oscillatory behavior at the RST pin. The traditional way of tackling this problem is to add some DC hysteresis in the comparator input, which changes the threshold point once the output lips. The problem is that the addition of DC hysteresis makes the trip voltage less accurate, since the trip point changes once the output changes. The LT3689 does not use hysteresis. Instead, it performs an integration-like function on transient events at the comparator. In this way the magnitude and duration of the event are both important to the comparator threshold. Figure 3 illustrates the typical transient duration versus comparator overdrive (as a percentage of trip threshold) required to trip the comparator. Selecting the Reset Timing Capacitor The reset timeout period is adjustable in order to accommodate a variety of microprocessor applications. The reset timeout period, tRST, is adjusted by connecting a capacitor between the CPOR pin and ground. The value of this capacitor is determined by: CPOR = tRST • 432 • 10–9 with CPOR in Farads and tRST in seconds. The CPOR value per millisecond of delay can also be expressed as CPOR/ms = 432 (pF/ms). Leaving the CPOR pin unconnected generates a minimum reset timeout of approximately 25µs with 10kΩ pull-up to 5V on RST pin. Maximum reset timeout is limited by the largest available low leakage capacitor. The accuracy of the timeout period will be affected by capacitor leakage (the nominal charging current is 2µA) and capacitor tolerance. A low leakage ceramic capacitor is recommended. Watchdog Modes: Timeout or Window The LT3689 also includes an adjustable watchdog timer that monitors a microprocessor’s activity. If a code execution error occurs in a µP, the watchdog detects the error and sets the WDO pin low. This signal can be used to interrupt a routine or to reset a µP. 800 700 TRANSIENT DURATION (µs) Programmable Undervoltage Lockout 600 500 400 RESET OCCURS ABOVE THE CURVE 300 200 100 0 0.10 1.00 10.00 100.00 POR COMPARATOR OVERDRIVE VOLTAGE AS PERCENTAGE OF RESET THRESHOLD, VRST (%) Figure 3. Typical transient duration vs POR comparator overdrive 25 L DESIGN FEATURES The watchdog is operated either in timeout or window mode (refer to Figure 4). In timeout mode, the microprocessor needs to toggle the WDI pin before the watchdog timer expires to keep the WDO pin high. If the voltage on the WDI pin does not transition during the programmed timeout period then the circuitry pulls WDO low. In window mode, the WDI pin’s negative-going pulses must appear inside a programmed time window to prevent WDO from going low. If more than two falling pulses are registered in the lower boundary period (tWDL), the WDO pin is forced low. The WDO pin also goes low if no negative edge is supplied to the WDI pin within the upper boundary period (tWDU). During a code execution error, the microprocessor outputs WDI pulses that are either too fast or too slow. This condition asserts WDO low and forces the microprocessor to reset the program. In window mode, the WDI signal is bounded by an upper and lower boundary periods for normal operation. The period of the WDI input signal should be longer than the window mode’s lower boundary period and shorter than the upper boundary period to keep WDO high under normal conditions. The window mode’s lower and upper boundary periods have a ixed ratio of 31. These times can be increased or decreased by adjusting an external capacitor on the CWDT pin. In both watchdog modes, when WDO is asserted, the reset timer is enabled. Any WDI pulses that appear while the reset timer is running are ignored. When the reset timer expires, the WDO is allowed to go back high again. Therefore, if no input is applied to the WDI pin then the watchdog circuitry produces a train of pulses on the WDO pin. The high time of this pulse train is equal to the upper boundary period and low time is equal to the reset period. Also, WDO and RST cannot be logic low simultaneously. If WDO is low and there is an undervoltage lockout fault, RST goes low and WDO will go high. The WDE pin allows the user to turn on or off the watchdog function. This 26 WDI WDO tWDU tRST WATCHDOG TIMING (W/T = HIGH), TIMEOUT MODE t < t WDL tRST WDI WDO tRST tWDU WATCHDOG TIMING (W/ T = LOW), WINDOW MODE tRST = PROGRAMMED RESET PERIOD tWDU = WATCHDOG UPPER BOUNDARY PERIOD tWDL = WATCHDOG WINDOW MODE LOWER BOUNDARY PERIOD VUV = OUTPUT VOLTAGE RESET THRESHOLD Figure 4. Watchdog timing diagram feature can be used to reliably program the connected microprocessor in the factory. During factory programming of the microprocessor, WDE pin can be kept high to prevent WDO from toggling and thus prevents WDO from interfering with the microprocessor’s programming procedure. Tying the WDO and RST pins together will generate a reset signal when either the output voltage falls 10% below the regulation value or if there is a watchdog error. Leaving the CWDT pin unconnected generates a minimum watchdog upper boundary period of approximately 200µs with 10kΩ pull-up to 5V on WDO pin. Maximum timeout is limited by the largest available low leakage capacitor. The accuracy of the upper and lower boundary periods is affected by capacitor leakage (the nominal charging current is 2µA) and capacitor tolerance. A low leakage ceramic capacitor is recommended. Selecting the Watchdog Timing Capacitor The wide input range, low quiescent current, supervisory features, robustness and small size of the LT3689 makes it an ideal candidate to power automotive and industrial applications. The part withstands 60V VIN transients and normal operation is guaranteed for max VIN of 36V, and the part is robust against inrush and short circuit conditions. The Burst Mode circuitry provides high eficiencies at light loads. Programmable switching frequency allows the designer to trade off between component size and eficiency. The accurate POR and Watchdog circuitry of LT3689 allows complete supervisory control of a microprocessor connected to the output of the LT3689 switching regulator. L The watchdog upper boundary period is adjustable and can be optimized for software execution. The watchdog upper boundary period is adjusted by connecting a capacitor between the CWDT pin and ground. Given a speciied watchdog upper boundary period, the capacitor is determined by: CWDT = tWDU • 55 (pF/ms) The window mode lower boundary period has a ixed relationship to upper boundary period for a given capacitor. The lower boundary period is related to the upper boundary period by the following: tWDL = 1/31 • tWDU Conclusion Linear Technology Magazine • March 2009 DESIGN FEATURES L Complete APD Bias Solution in 60mm2 with On-the-Fly Adjustable Current By Xin (Shin) Qi Limit and Adjustable VAPD Introduction The overriding factor limiting functionality in iber-optic communication systems is available space. A compact APD (avalanche photo diode) bias solution with a high degree of feature integration is the key to breaking new ground in system size and performance. The LT3571 offers such a solution in a tiny 3mm × 3mm QFN package. The LT3571 combines a current mode step-up DC/DC converter and a high side ixed voltage drop APD current monitor with an integrated 75V power switch and Schottky diode. The combination of a traditional voltage loop and a unique current loop allows customers to set an accurate APD current limit at any given bias voltage. The integrated high side current monitor provides an 8% accurate current that is proportional to the load current, making it possible to adjust the APD bias voltage via the CTRL pin. This feature-rich device makes it possible to produce a single stage boost converter to bias high voltage APDs in only 60mm2. Low Noise APD Bias Supply The gain of the APD is dependent on the bias voltage, so the bias supply must minimize the noise contamination from switching regulators and other sources. Figure 1 shows the LT3571 conigured to produce an ultralow noise power supply for a 45V APD with 2.5mA of load current capability. The MONIN voltage is regulated by the internal voltage reference and the resistor divider made up of R1 and R2. Resistor RSENSE is selected to set the APD current limit at 200mV/1.2RSENSE – 0.2mA. The CTRL pin can override the internal reference, making it possible to optimize the APD bias on the ly to maximize receiver performance. Linear Technology Magazine • March 2009 L1 10µH VIN 5V OFF ON VIN SHDN SW VOUT RSENSE 20Ω VREF C1 1µF CTRL LT3571 MONIN 50V C3 10nF RT SYNC R1 1M FB GND MON R2 20.5k APD RT 12.1k 1MHz R4 49.9Ω C5 10nF R3 10k C2 0.1µF 45V C4 0.1µF L: TDK VLF3010AT – 100MR49 C1: TDK X7R C1608X7R1C105KT C2, C4: MURATA X7R GRM188R72A104KA35 C3: AVX X7R 06031C103K C5: MURATA X7R GRM155R71H103K Figure 1. Low noise APD bias supply When the CTRL pin is connected to a supply above 1V, the output voltage is regulated with feedback at 1V. When driven below 1V, the feedback and the output voltage follow accordingly. The APD pin, the output of the current monitor, provides a voltage to the APD load that is ixed 5V below the MONIN pin. The LT3571 includes a precise current mirror with a factorof-ive attenuation. The proportional current output signal at the MON pin can be used to accurately indicate the APD signal strength. The voltage variance of APD pin voltage is only ±200mV over the entire input current range and the whole temperature range. Figure 2 shows the evaluation board for this topology. The topology uses several ilter capacitors to achieve ultralow noise performance. The capacitor at VOUT pin and the 0.1µF capacitor at the APD pin suppress switching noise. The 10nF feedforward capacitor across the MONIN and FB pins ilters out high frequency internal reference and error ampliier noise. Figure 3 shows the measured switching noise is less than 500µVP–P at 1mA load current. This exceptionally low noise bias voltage 500µV/DIV IAPD = 1mA Figure 2. The LT3571 evaluation board 500ns/DIV Figure 3. AC-coupled noise ripple at APD pin 27 L DESIGN FEATURES gives the APD greater sensitivity and dynamic range. L1 10µH VIN 3.3V Fast APD Current Monitor Transient Response Design efforts in modern communications systems increasingly focus on 10Gbits/s GPON systems, which demand that the transient response of the APD current monitor is less than 100ns for a two-decades-of-magnitude input current step. To meet this challenging requirement, many designers rely on a simple discrete current mirror topology to reduce parasitic capacitance on the signal path, sacriicing monitor accuracy and board space. In contrast, the LT3571’s APD current monitor is carefully designed to provide not only a ixed voltage drop and high accuracy, but also the required fast transient response. Figure 4 shows a compact circuit that responds quickly to current transients. Unlike the ultralow noise topology shown in Figure 1, the ilter capacitor at the APD pin is moved to the MONIN pin. C2, C3 and RSENSE form a π ilter to isolate the APD current monitor from high frequency switching noise. The capacitor at the MON pin is also removed to reduce the transient delay on the measurement path. The transient speed is measured using the same technique described in the Linear Technology Design Note 447 “A Complete Compact APD Bias Solution for a 10GBit/s GPON System.” Figures 5 and 6 show the measured input signal falling transient response and input signal rising transient response, respectively, where the input current levels are 10µA and 1mA. Note that there is an inversion and DC offset present in the measurement. The measurements show a transient response time of less than 100ns, well within the stringent speed demands of the 10Gbits/s GPON system. APD Bias Voltage Temperature Compensation Typically, the APD reverse bias voltage is designed with a compensatory positive temperature coeficient. This can be easily implemented via the CTRL pin of the LT3571—a less complex 28 VIN SHDN OFF ON SW VOUT RSENSE 20Ω VREF C1 1µF CTRL LT3571 55V MONIN R1 1M RT SYNC GND C2 0.1µF FB MON R2 18.2k APD RT 26.1k 500kHz C3 0.1µF 50V 0.5pF FOR TEST PURPOSES, REPLACE APD WITH THIS SIMPLE TEST SETUP APD PIN 4.99k PMBT3904 – 2.5V 4.99k LT1815 1k + MEASURE HERE L1: TDK VLF3010AT-100MR49 0.1µF C1: MURATA X7R GRM21BR71C105KA01B C2, C3: MURATA X7R GRM188R72A104KA35 PWM –VLO –VHI Figure 4. APD bias supply with ultrafast current monitor transient speed PWM GND PWM GND PWM 1V/DIV IAPD = 10µA IAPD = 10µA PWM 1V/DIV IAPD = 1mA IAPD = 1mA OUT 500mV/DIV OUT 500mV/DIV TFD < 100ns TRD < 100ns OUT GND OUT GND 50ns/DIV 50ns/DIV Figure 5. Transient response on input signal falling edge (1mA to 10µA) Figure 6. Transient response on input signal rising edge (10µA to 1mA) L1 15µH VIN 5V R5 30.1k Q2 R7 49.9k R8 36.5k R9 20k OFF ON R6 100k VIN SHDN VREF SW VOUT RSENSE 49.9Ω LT3571 MONIN CTRL RT SYNC C6 0.1µF TEMPERATURE COMPENSATION BLOCK C1 1µF Q1 C3 10nF R1 1M FB GND MON R2 15k APD RT 33.2k 400kHz R4 49.9Ω C5 10nF L1: TDK VLF4012AT – 150MR63 C1: TDK X7R C1608X7R1C105KT C2: MURATA X7R GRM21AR72A224KAC5L C3: AVX X7R 06031C103K C4: MURATA X7R GRM188R72A104KA35 C5: MURATA X7R GRM155R71H103K C6: MURATA X7R GRM155R71A104KA01D 55V R3 10k C2 0.22µF 50V C4 0.1µF Q1, Q2 = PHILIPS PEMT1 Figure 7. Temperature-compensated APD power supply Linear Technology Magazine • March 2009 DESIGN FEATURES L and expensive solution than typical microprocessor-controlled methods. The simplest scheme uses a resistor divider from the VREF pin to the CTRL pin, where the top resistor in the divider is an NTC (negative temperature coeficient) resistor. While simple, this method suffers from nonlinear temperature coeficient of the NTC resistor. A more precise method uses a transistor network as shown in Figure 7. The PTC (Positive Temperature Coeficient) of the CTRL pin voltage is realized by an emitter follower of Q1 and a VBE multiplier of Q2. Assuming: VBE(Q1) = VBE(Q2) = VBE dT = dVBE(Q2) dT = 2mV °C R8 V R7 BE with 60 PTC = 58 56 VAPD (V) dVCTRL R8 2mV = • dT R7 °C 52 50 48 R8 = R7 46 44 42 40 –50 –25 0 50 25 75 TEMPERATURE (°C) 100 125 Figure 8. Temperature response of the circuit shown in Figure 7 R1 = R2 VREF VOUT 2mV VBE + • °C dVOUT dT VBE • dVBE(Q2) dT = 2mV °C Conclusion Given VOUT at room and dVOUT/DT, the R1/R2 and R8/R7 can be calculated as follows 54 = Simulation using LTspice always gives a good starting point. The circuit shown in Figure 7 is designed to have VAPD = 50V (VOUT = 55V) at room and dVAPD/dT = 100mV/°C (dVOUT/dT = 100mV/°C). The measured temperature response is shown in Figure 8, which is very close to the design target. then the CTRL pin voltage is VCTRL = VREF − dVBE(Q1) dT and dVBE(Q1) Resistors R5–R9 are selected to make I(Q1) = I(Q2) ≈ 10µA, and dVOUT 2mV + • VOUT dT °C −1 2mV • VREF °C The LT3571 is a highly integrated, compact solution to APD bias supply design. It provides a useful feature set and the lexibility to meet a variety of challenging requirements, such as low noise, fast transient response speed, and temperature compensation. With a high level of integration and superior performance, the LT3571 is the natural choice for APD bias supply design. L LT840, continued from page 2 out of shutdown, the VREF pin is irst discharged for 70µs with a strong pull down current, and then charged with 10µA to 1.235V. This achieves soft start since the output is proportional to VREF. Full soft-start cycles occur even with short SHDN low pulses since VREF is discharged when the part is enabled. In addition, the LT8410/-1 features a 2.5V to 16V input voltage range, up to 40V output voltage and overvoltage protection for CAP and VOUT. Conclusion The LT8410/-1 is a smart choice for applications which require low quiescent current and low input current. The ultralow quiescent current, combined with high value integrated feedback resistors, keeps the average input current very low, signiicantly 100 LT65/6, continued from page 2 Conclusion 90 EFFICIENCY (%) of handling 60V transients. Figure 4 shows the circuit eficiency at multiple input voltages. The current limit of the application is set to 1.2A, therefore, the power path components are sized to handle 1.2A maximum. To reduce the application footprint, the LT3663 includes internal compensation and a boost diode. The RUN pin, when low, puts the LT3663 into a low current shutdown mode. extending battery operating time. Low current limit internal switches (8mA for the LT8410-1, 25mA for the LT8410) make the part ideal for high impedance sources such as coin cell batteries. The LT8410/-1 is packed with features without compromising performance or ease of use and is available in a tiny 8-pin 2mm × 2mm package. L The accurate programmable output current limit of the LT3653 and LT3663 eliminates localized heating from an output overload, reduces the maximum current requirements on the power components, and makes for a robust power supply solutions. L VIN = 8V VIN = 15V 80 VIN = 30V 70 60 50 40 0.1 0.3 0.5 0.7 0.9 OUTPUT CURRENT (A) 1.1 1.3 Authors can be contacted at (408) 432-1900 Figure 4. Efficiency of the circuit in Figure 3 Linear Technology Magazine • March 2009 29 L DESIGN IDEAS Don’t Want to Hear It? Avoid the Audio Band with PWM LED Dimming at Frequencies Above 20kHz by Eric Young Introduction The requirements of LED drivers become more demanding as application designers exploit the unique characteristics of LEDs. Linear Technology offers a complete portfolio of LED drivers with the performance levels required to meet even the most challenging design requirements. One area where these LED drivers especially excel is in the performance and lexibility of their PWM dimming capabilities. LEDs can be turned on and off rapidly—it takes only nanoseconds to illuminate or extinguish the source. PWM dimming exploits this characteristic to achieve orders of magnitude dimming, even while maintaining a constant output spectrum over the entire dynamic light intensity range. The broad ield of available LED drivers narrows quite a bit when one considers PWM dimming at frequencies above 20kHz. Why 20kHz? Although most LED light designers worry about perceptible licker at PWM frequencies below about 100Hz, in some applications the human eye is not the limiting factor; it is the human ear. The human ear perceives vibrations up to about 20kHz, which in some applications can become the important factor in determining PWM frequency. The versatile LT3755 and LT3756 are members of an elite group VPWM VGATE IL1 5A/DIV ILED 0.5A/DIV 5µs/DIV Figure 2. DCM operation of the boost LED driver in Figure 1 30 VIN 8V TO 18V L1 1.5µH C1 2.2µF x2 25V 499k D1 10µF x2 35V VIN SHDN/UVLO GATE VREF 100k M1 SENSE 255k LT3755 0.01Ω CTRL 499k 75k FB 17.8k PWM ISP INTVCC 100k 0.1Ω OPENLED ISN D2 VC 0.1µF 10k UP TO 8 LEDS 26V PWMOUT SS C2 4.7µF 350mA 22k 470pF GND M2 RT 13k 800kHz L1: COILTRONICS DR125-1R5 D1: ON SEMI MBRS360 M1: VISHAY SILICONIX Si7850DP M2: VISHAY SILICONIX Si2306DS D2: IN4448HWT Figure 1. This 10W boost LED driver stays out of the audio band by achieving 50:1 PWM dimming at 20kHz. Lower PWM frequencies can result in an audible hum as ceramic capacitors vibrate. of LED controllers that can support very high PWM dimming ratios, as much as 50:1, at 20kHz. These controllers support a variety of topologies, including buck mode, boost and buckboost at various power levels. High Performance PWM Dimming The PWM dimming method is straightforward; the LED is driven by a tightly regulated current for a ixed interval in every PWM period. During the off-phase, the current in the LED is zero. During the on-phase, the current is carefully regulated. It is important that the “on” current is consistent, since an LED’s output spectrum is a function of forward current. The duty cycle of the PWM signal corresponds to the dimming value. Although the concept is simple, designing a controller that can achieve this at a high PWM frequency is any- thing but simple. The rise and fall times of the pulsed current should be fast, less than 100ns. Generating a suitable PWM current pulse from an arbitrary input voltage can prove a challenge. This usually requires a high bandwidth DC/DC converter to regulate the current, a storage/ilter capacitor across the LED to provide current during PWM on/off transitions, and a disconnect switch to ensure that the current waveform has sharp turn-on and off edges. Hysteretic converters, while simple to use from the standpoint of closed loop stability, have problems. The slow LED current rise and fall times are one consequence of using a large value inductor to smooth the current through the LED because there is no output capacitor. And since the average current in the LED is related to the ripple current in the inductor, which is in turn sensitive to input voltLinear Technology Magazine • March 2009 DESIGN IDEAS L 1M trollers can be conigured into several different converter circuits to provide a high bandwidth, well regulated output current that can be pulsed at intervals as short as 1µs. 96 92 Discontinuous Conduction Mode Is the Secret to Maximizing PWM Performance 88 84 80 0.0 0.2 1.0 an annoying buzz or hum next to a handheld device containing one of these circuits, then you have observed this effect. The use of a disconnect switch in series with the LED greatly reduces the voltage transient and therefore the hum from the output capacitor. While good design techniques can greatly minimize audible noise for lower PWM frequencies, the elimination of audible emission is not assured so long as PWM frequency is below 20kHz. Many application designers don’t want to tinker with acoustics, preferring instead quiet running circuits that do a reasonable job of PWM dimming. The LT3755 and LT3756 current-mode switching con- The key to short on/off times is for the switching regulator to operate in discontinuous conduction mode (DCM). In this mode, the inductor current always starts from zero at the beginning of each switching period and the peak inductor current is determined by the load and adjusted through the switch duty cycle. In contrast, continuous conduction mode (CCM) maintains a relatively constant switch duty cycle and adjusts the average inductor current to meet the demands of the load. DCM is superior for high performance PWM dimming because it delivers the required energy to the output in a single switching period. This allows the controller to bypass the typical minimum PWM period of 3-4 switching cycles to reach steady state, a familiar requirement of CCM. Operation in DCM places greater demands ISP VIN 0.2Ω VREF 500mA ISN CTRL UP TO 5 LEDS 16V 0.22µF D2 6.2V INTVCC 2200pF OPENLED L1 3.3µH D1 M1 GATE VLED = 16V ILED = 0.5A 96 4.7µF 2x 25V 1M LT3755 4.7µF 100 M2 PWMOUT PWM 0.8 Figure 3. The efficiency of the boost LED driver in Figure 1 is greater than 90%. SHDN/UVLO 68.1k 0.4 0.6 LED CURRENT(A) EFFICIENCY (%) VIN 22V TO 36V 100 EFFICIENCY (%) age transients, the LED light output changes with input supply. In most cases, this method cannot provide acceptable PWM performance. What determines PWM performance? The PWM interval or frequency is determined by the application, and there are several considerations to bear in mind. First, the human eye generally does not perceive licker if the PWM frequency is greater than 120Hz, thus a lower bound on the interval is typically taken to be 8ms. The achievable dimming ratio is a function of the minimum on- and off-times of the current pulse provided by the driver circuit. So an 8µs minimum pulse yields a 1000:1 dimming capability at 120Hz. The 20kHz audible requirement comes about because audible physical vibrations can be introduced to the PC board by the ceramic capacitors, and these caps are ubiquitous in high bandwidth converter circuits because of their low ESR, ruggedness, and long-term reliability. Ceramic capacitors physically change dimension (as well as value) with a change in applied voltage, and rapid voltage transients during the PWM transients cause rapid changes in dimensions that couple vibrations into the boards. If you ever noticed 92 88 84 SENSE SS VC GND FB 22k 0.1µF 470pF RT 13k 800kHz 0.033Ω 2.2µF 2x 50V 80 15 20 25 30 VIN (V) 35 40 L1: TOKO 962BS_3R3M M1: VISHAY SILICONIX Si7850DP M2: VISHAY SILICONIX Si2306DS D1: DIODES, INC SBM540 Figure 4. An 8W buck-mode LED driver with 50:1 PWM dimming at 20kHz and 90% efficiency Linear Technology Magazine • March 2009 31 L DESIGN IDEAS on switching components because the switching components see higher peak currents for a given load. Because of this, a controller is easier to use than a monolithic converter because its maximum switching current can be programmed to the needs of the application, without having to change the application’s features. Operating in DCM does come at a price when compared to CCM: eficiency, input supply range and analog dimming range all suffer some reduction. The ratio of maximumto-minimum input supply range is slightly less than the ratio of the minimum PWM pulse width to the minimum switch on-time. Likewise, provided the input supply is ixed, the maximum analog dimming ratio is the same ratio of minimum PWM pulse to minimum switch on-time. Nevertheless, the beneit of this technique is that minimum PWM period is four to ive times shorter compared with continuous conduction mode. If the application calls for high PWM dimming ratio, DCM mode provides a sure path to achieve that objective. Three application circuits built with LT3755 and shown here demonstrate this technique. ILED 500mA/DIV 3% PWM DUTY 50% PWM DUTY 97% PWM DUTY 5µs/DIV Figure 5. Three PWM dimming settings for the buck mode driver in Figure 4. Even at 33kHz there is no perceptible change in the LED current from minimum to maximum duty cycle. Figure 1 shows a 9W boost converter that regulates 26V of LEDs at a steady 350mA from a supply ranging between 8V and 18V. If the supply is ixed at 12V, the regulator operates at constant switching frequency for LED currents programmed by the CTRL pin between 125mA and 1A (2.4W to 27W). The minimum on-time is 1µs, as is the minimum off-time. The switching waveforms in Figure 2 show the operation at 50% duty cycle, 27V/1A load and 12V supply. Notice the fast rise and fall times of the LED current signal, even at 1A. At maximum load, the GATE pin is 7V for almost 1µs (same as the minimum pulse width) and the inductor current reaches zero before the start of the each GATE pulse, a characteristic of DCM operation. Figure 3 shows the eficiency versus LED current at 12V input, which peaks at just over 90%. Figure 4 shows a buck-mode converter that regulates a 16V LED string at 500mA from a 22V to 36V supply. This circuit has an external chargepump and level shift to drive the gate of an LED disconnect NMOS. This level shift provides much faster rise and fall times than the familiar resistor level shift driving a PMOS, and uses much less current. The scope trace in Figure 5 shows PWM dimming at several duty cycles—it is clear that the output LED current has no perceptible variation as pulse width is smoothly adjusted between the minimum on-time and the minimum off-time. The eficiency of this 8W circuit exceeds 90%. Figure 6 shows a SEPIC converter driving a 1A, 20V LED string from a 12V-to-36V supply. In addition to providing step-up and step-down capability, this circuit is handy because it provides input-output isolation and built in protection from a short to GND on the output. The eficiency of this circuit exceeds 87%. The minimum continued on page 40 L1 1.5µH VIN 10V TO 36V D1 1:1 1M C1 4.7µF 50V 1µF x 2 100V SHDN/UVLO 200k 10µF x2 25V VIN GATE VREF M1 L1B SENSE 0.01Ω GATE CTRL 169k PWM FB LT3755 ISP 10k 0.1Ω INTVCC 100k 1A ISN OPENLED 20W LED STRING D2 PWMOUT SS VC 0.1µF 4.7µF 10k 22k 470pF SWITCH CURRENT (2A/DIV) PEAKS AT CURRENT LIMIT OF 10A GND OUTPUT SHORT CIRCUIT CURRENT (2A/DIV) 5µs/DIV RT 13k 800kHz Figure 7. The SEPIC converter in Figure 6 maintains control during an output fault to GND. M2 L1: COILTRONICS DRQ125-1R5 COUPLED INDUCTOR D1: ON SEMI MBRS360 M1: VISHAY SILICONIX Si7850DP M2: VISHAY SILICONIX Si2306DS Figure 6. A 20W SEPIC LED Driver with 50:1 PWM dimming at 20kHz and output fault protection 32 Linear Technology Magazine • March 2009 DESIGN IDEAS L Eliminate EMI Worries with 2A, 15mm × 9mm × 2.82mm µModule Step-Down Regulator by David Ng Introduction “We failed EMI.” Those three dreaded words strike fear into the hearts and minds of electronics design engineers. There are four words that are even worse: “We failed EMI again.” The psyche of many a seasoned engineer is scarred with dark memories of long days and nights in an EMI lab, struggling with aluminum foil, copper tape, clamp-on ilter beads and inger cuts to ix a design that just won’t keep quiet. A big part of the problem is the necessary profusion of switching power supplies, which contribute signiicantly to the radiated system EMI. The LTM8032 is a DC/DC switching step-down µModule regulator built speciically for low EMI. It is rated for up to 36VIN, 10VOUT at 2A, and features adjustable frequency, synchronization, a power good status lag and soft-start. It is small, measuring only 15mm × 9mm × 2.82mm, integrating the inductor, power stage and controller in a ROHS e3-compliant molded LGA package. 10V/2A Supply Is EN55022 and CSIPR 22 Class B Compliant Like most other µModule regulators, the LTM8032 is easy to use. As shown in Figure 1, all that is needed for a complete power design are the resistors to set the output voltage and operating frequency, and the input and output caps. The LTM8032 is test-proven EN55022 and CSIPR 22 class B complaint, tested in an NRTL 5-meter chamber, set up as shown in the photo given in Figure 2. The LTM8032 is mounted on a circuit board with no bulk capacitance installed. The input and output capacitance are the minimum ceramic values speciied in the data sheet for proper operation. Linear Technology Magazine • March 2009 VIN 5.5VDC TO 36VDC OUT VIN 22µF FIN RUN/SS 2.2µF VOUT 3.3V 2A AUX LTM8032 BIAS SHARE PGOOD RT SYNC GND ADJ 54.9k 78.7k Figure 1. Just two resistors, input and output caps are needed to complete a power supply design with the LTM8032. The LTM8032 is a DC/DC switching step-down µModule regulator built for low EMI. It is rated for up to 36VIN, 10VOUT at 2A, integrating the inductor, power stage and controller in a ROHS e3-compliant molded LGA package. The assembled unit is placed atop an all-wood table. The all-wood construction ensures that the test setup does not shield or shadow noise emanating from the device under test (DUT). The power source, a linear lab grade power supply, is on the loor. The load for the LTM8032, with its heat sink, is also on the table top. Before measuring the emissions from the LTM8032, a baseline measurement is taken to establish the continued on page 8 Figure 2. For EMI testing, the DUT is mounted on a circuit board and placed on a wooden table. The power source is on the floor. 33 L DESIGN IDEAS Diode Turn-On Time Induced Failures in Switching Regulators Never Has So Much Trouble Been Had by So Many with So Few Terminals by Jim Williams and David Beebe +V This article is excerpted from the Linear Technology Application Note AN22 with the same title. +V VIN IC REGULATOR VIN Introduction Most circuit designers are familiar with diode dynamic characteristics such as charge storage, voltage dependent capacitance and reverse recovery time. Less commonly acknowledged and manufacturer speciied is diode forward turn-on time. This parameter describes the time required for a diode to turn on and clamp at its forward voltage drop. Historically, this extremely short time, units of nanoseconds, has been so small that user and vendor alike have essentially ignored it. It is rarely discussed and almost never speciied. Recently, switching regulator clock rate and transition time have become faster, making diode turn-on time a critical issue. Increased clock rates are mandated to achieve smaller magnetics size; decreased transition times somewhat aid overall eficiency but are principally needed to minimize IC heat rise. At clock speeds beyond about 1MHz, transition time losses are the primary source of die heating. A potential dificulty due to diode turn-on time is that the resultant OUTPUT VREG SWITCH PIN IC REGULATOR CONTROL SWITCH SWITCH PIN OUTPUT VREG CONTROL SWITCH REF REF GND GND FEEDBACK NODE FEEDBACK NODE STEP-UP STEP-DOWN Figure 1. Typical voltage step-up/step-down converters. Assumption is diode clamps switch pin voltage excursion to safe limits. transitory “overshoot” voltage across the diode, even when restricted to nanoseconds, can induce overvoltage stress, causing switching regulator IC failure. As such, careful testing is required to qualify a given diode for a particular application to insure reliability. This testing, which assumes low loss surrounding components and layout in the inal application, measures turn-on overshoot voltage due to diode parasitics only. Improper IC BREAKDOWN LIMIT associated component selection and layout will contribute additional overstress terms. Diode Turn-On Time Perspectives Figure 1 shows typical step-up and step-down voltage converters. In both cases, the assumption is that the diode clamps switch pin voltage excursions to safe limits. In the step-up case, this limit is deined by the switch pins PULSE IN tRISE ≤ 2ns AMPLITUDE = 5V + VFWD DIODE ON VOLTAGE 5Ω MEASUREMENT POINT DIODE UNDER TEST DIODE TURN-ON TIME AN122 F02 AN122 F03 Figure 2. Diode forward turn-on time permits transient excursion above nominal diode clamp voltage, potentially exceeding IC breakdown limit. 34 Figure 3. Conceptual method tests diode turn-on time at 1A. Input step must have exceptionally fast, high fidelity transition. Linear Technology Magazine • March 2009 DESIGN IDEAS L PULSE CURRENT AMPLIFIER tRISE = 2ns PULSE GENERATOR tRISE < 1ns OSCILLOSCOPE 1GHz BANDWIDTH tRISE = 350ps TYPICALLY 5V TO 6V, 30ns WIDE 5Ω Z0 PROBE ≈1A DIODE UNDER TEST AN122 F04 Figure 4. Detailed measurement scheme indicates necessary performance parameters for various elements. Subnanosecond rise time pulse generator, 1A, 2ns rise time amplifier and 1GHz oscilloscope are required. VIN = 20V +V + LT1086 22µF + 120Ω +V TYPICAL 17V 22µF * 1k Q1 Q4 1Ω Q5 1Ω +V ADJUST (RISE TIME TRIM) 1k +V PULSE INPUT * EDGE PURITY 100Ω MINIMIZE INDUCTANCE IN ALL PATHS Q2 50Ω 62Ω 2pF TO 12pF EDGE PURITY = 2N3866 OUTPUT +V 5Ω** = 2N3375 * ** = TEN PARALLELED 50Ω RESISTORS * = BYPASS EVERY TRANSISTOR WITH 22µF SANYO OSCON PARALLELED WITH 2.2µF MYLAR Q3 Q6 1Ω AN122 F05 Figure 5. Pulse amplifier includes paralleled, darlington driven RF transistor output stage. Collector voltage adjustment (“rise time trim”) peaks Q4 to Q6 FT, input RC network optimizes output pulse purity. Low inductance layout is mandatory. maximum allowable forward voltage. The step-down case limit is set by the switch pins maximum allowable reverse voltage. Figure 2 indicates the diode requires a inite length of time to clamp at its forward voltage. This forward turnon time permits transient excursions above the nominal diode clamp voltage, potentially exceeding the IC’s breakdown limit. The turn-on time is typically measured in nanoseconds, making observation dificult. A further complication is that the turn-on overshoot occurs at the amplitude extreme of a pulse waveform, precluding high resolution amplitude measurement. These factors must be considered when designing a diode turn-on test method. Linear Technology Magazine • March 2009 Figure 3 shows a conceptual method for testing diode turn-on time. Here, the test is performed at 1A although other currents could be used. A pulse steps 1A into the diode under test via the 5Ω resistor. Turn-on time voltage excursion is measured directly at the diode under test. The igure 1V/DIV 2ns/DIV AN122 F06 Figure 6. Pulse amplifier output into 5Ω. Rise time is 2ns with minimal pulse-top aberrations. 35 36 50Ω 2pF TO 12pF EDGE PURITY 62Ω + 22µF Q3 +V Q2 +V Q1 +V LT1086 * * * Q6 Q5 Q4 1k 1k 22µF +V, TYPICAL 17V 1Ω 1Ω 1Ω DIODE UNDER TEST Z0 PROBE = TEKTRONIX P-6056, 500Ω 5Ω** ≈ 5.5V +V ADJUST (RISETIME TRIM) 120Ω + Figure 7. Complete diode forward turn-on time measurement arrangement includes subnanosecond rise time pulse generator, pulse amplifier, Z0 probe and 1GHz oscilloscope. ADJUST PULSE GENERATOR AMPLITUDE FOR 5.5V AMPLITUDE AT 5Ω RESISTOR ** = TEN PARALLELED 50Ω RESISTORS * = BYPASS EVERY TRANSISTOR WITH 22µF SANYO OSCON PARALLELED WITH 2.2µF MYLAR = 2N3375 = 2N3866 MINIMIZE INDUCTANCE IN ALL PATHS HP-215A PULSE GENERATOR tRISE = 800ps PWIDTH = 30ns 215A EDGE PURITY 100Ω ≈ 6.7V VIN = 20V 7A29 7B15 TEKTRONIX 7104/7A29/7B10/7B15 1GHz (tRISE = 350ps) OSCILLOSCOPE 7A29 7B10 7104 AN122 F05 L DESIGN IDEAS Linear Technology Magazine • March 2009 DESIGN IDEAS L 200mV/DIV 200mV/DIV AN122 F08 2ns/DIV 2ns/DIV Figure 9. “Diode Number 2” peaks ≈750mV before settling in 6ns... > 2x steady state forward voltage. Figure 8. “Diode Number 1” overshoots steady state forward voltage for ≈3.6ns, peaking 200mV. 200mV/DIV AN122 F09 200mV/DIV AN122 F10 2ns/DIV Figure 10. “Diode Number 3” peaks 1V above nominal 400mV VFWD, a 2.5x error. 5ns/DIV AN122 F11 Figure 11. “Diode Number 4” peaks ≈750mV with lengthy (note horizontal 2.5x scale change) tailing towards VFWD value. 200mV/DIV 5ns/DIV AN122 F12 Figure 12. “Diode Number 5” peaks offscale with extended tailing (note horizontal slower scale compared to Figures 8 thru 10). Linear Technology Magazine • March 2009 37 L DESIGN IDEAS is deceptively simple in appearance. In particular, the current step must have an exceptionally fast, high-idelity transition and faithful turn-on time determination requires substantial measurement bandwidth. Detailed Measurement Scheme A more detailed measurement scheme appears in Figure 4. Necessary performance parameters for various elements are called out. A subnanosecond rise time pulse generator, 1A, 2ns rise time ampliier and a 1GHz oscilloscope are required. These speciications represent realistic operating conditions; other currents and rise times can be selected by altering appropriate parameters. The pulse ampliier necessitates careful attention to circuit coniguration and layout. Figure 5 shows the ampliier includes a paralleled, Darlington driven RF transistor output stage. The collector voltage adjustment (“rise time trim”) peaks Q4 to Q6 FT; an input RC network optimizes output pulse purity by slightly retarding input pulse rise time to within ampliier passband. Paralleling allows Q4 to Q6 LTM802, continued from page Diode Testing and Interpreting Results The measurement test ixture, properly equipped and constructed, permits diode turn-on time testing with excellent time and amplitude resolution.5 Figures 8 through 12 show results for ive different diodes from various manufacturers. Figure 8 (Diode Number 1) overshoots steady state forward voltage for 3.6ns, peaking 200mV. This is the best performance of the ive. Figures 9 through 12 show 2A, from the maximum input voltage, 36V. There are two traces in the plot, one for the vertical and horizontal orientations of the test lab’s receiver antenna. As shown in the igure, the LTM8032 easily meets the CISPR 22 class B limits, with 20db of margin for most of the frequency spectrum, with either antenna orientation. 90 90 80 80 70 70 60 50 EN55022 CLASS B LIMIT 40 30 20 10 Conclusion The LTM8032 switching step-down regulator is both easy to use and quiet, meeting the radiated emissions requirements of CISPR22 and EN55022 class B by a wide margin. L Authors can be contacted at (408) 432-1900 EN55022 CLASS B LIMIT 40 30 20 10 0 –10 100 200 300 400 500 600 700 800 900 1000 FREQUENCY (MHz) Notes 1 An alternate pulse generation approach appears in Linear Technology Application Note 22, Appendix F, “Another Way to Do It.” 2 Z0 probes are described in Linear Technology Application Note 22 Appendix C, “About Z0 Probes.” See also References 27 thru 34. 3 The subnanosecond pulse generator requirement is not trivial. See Linear Technology Application Note 22 Appendix B, “Subnanosecond Rise Time Pulse Generators For The Rich and Poor.” 4 See Linear Linear Technology Application Note 22 Appendix E, “Connections, Cables, Adapters, Attenuators, Probes and Picoseconds” for relevant commentary. 5 See Linear Technology Application Note 22 Appendix A, “How Much Bandwidth is Enough?” for discussion on determining necessary measurement bandwidth. 50 –10 0 increasing turn-on amplitude and time which are detailed in the igure captions. In the worst cases, turn-on amplitudes exceed nominal clamp voltage by more than 1V while turn-on times extend for tens of nanoseconds. Figure 12 culminates this unfortunate parade with huge time and amplitude errors. Such errant excursions can and will cause IC regulator breakdown and failure. The lesson here is clear. Diode turn-on time must be characterized and measured in any given application to insure reliability. L 60 0 Figure 3. The baseline measurement of ambient noise in the 5-meter chamber (no devices operating) 38 EMISSIONS LEVEL (dBµV/m) EMISSIONS LEVEL (dBµV/m) amount of ambient noise in the room. Figure 3 shows the noise spectrum in the chamber without any devices running. This can be used to determine the actual noise produced by the DUT. Figure 4 shows the worst case LTM8032 emissions plot, which occurs at maximum power out, 10V at to operate at favorable individual currents, maintaining bandwidth. When the (mildly interactive) edge purity and rise time trims are optimized, Figure 6 indicates the ampliier produces a transcendently clean 2ns rise time output pulse devoid of ringing, alien components or post-transition excursions. Such performance makes diode turn-on time testing practical.1 Figure 7 depicts the complete diode forward turn-on time measurement arrangement. The pulse ampliier, driven by a sub-nanosecond pulse generator, drives the diode under test. A Z0 probe monitors the measurement point and feeds a 1GHz oscilloscope.2, 3, 4 0 100 200 300 400 500 600 700 800 900 1000 FREQUENCY (MHz) Figure 4. The LTM8032 emissions for 20W out, 36VIN Linear Technology Magazine • March 2009 DESIGN IDEAS L µModule Regulator Fits a (Nearly) Complete Buck-Boost Solution in 15mm × 15mm × 2.8mm for 4.5V–36V VIN to 0.8V–34V VOUT Linear Technology offers a number of high eficiency synchronous 4-switch buck-boost DC/DC converter solutions for applications where VOUT falls within the range of VIN. The LTM4605, LTM4607 and LTM4609 µModule regulators are nearly self-contained buck-boost solutions that share pincompatible 15mm × 15mm × 2.8mm packages. The package includes the controller, four power FETs and a number of other discrete components. Only an external inductor, a sensing resistor, a voltage setting resistor and a few input and output capacitors are needed to complete a high eficiency buck-boost converter. Table 1 shows the input voltage, output voltage and current speciications of these three buck-boost µModule regulators. The LTM4609 is the latest addition to this family. It satisies the needs of high output voltage applications with an output range of 0.8V–34V. by Judy Sun, Sam Young and Henry Zhang VIN 10V TO 36V OPTIONAL CLOCK SYNC 4.7µF ×2 50V ON/OFF VIN PLLIN V OUT FCB RUN LTM4609 + 150µF ×2 35V VOUT 30V 3A SW1 SW2 RSENSE L1: SUMIDA CDEP147 SENSE+ 10nF 7mΩ SS SGND SENSE– PGND VFB 2.74k Figure 1. Just a few components form a complete 10V to 36V input, 30V/3A output converter using the LTM4609. As with all Linear Technology µModule regulators, the LTM4609 requires only a few external components to complete a wide input range buck-boost converter. Figure 1 shows a 10V to 36V input, 30V output design. The output current capability is 3A at 10V VIN, and 8A with 36V input. Figure 2 shows the eficiency of this converter, up to 98% in buck mode and 95% in boost mode. The low proile LGA package features low thermal resistance from junction to pin, thus maintaining an acceptable junction temperature even at high output power. The LTM4609’s high VIN = 36V, VOUT = 30V, IOUT = 8A VIN = 24V, VOUT = 30V, IOUT = 6A High Performance with Minimum Component Count 10µF ×2 35V L1 3.4µH 100 95 EFFICIENCY (%) Introduction 36VIN 24VIN 10VIN 90 85 CONTINUOUS CURRENT MODE VOUT = 30V fSW = 275kHz 80 0 2 4 6 LOAD CURRENT (A) 8 10 Figure 2. Efficiency of the 30V buck-boost converter VIN = 12V, VOUT = 30V, IOUT = 3A Figure 3. Thermal-graph taken with the LTM4609 running at different input voltages. The LTM4609 is on the left, the inductor (Sumida CDEP147) is on the right. No heat sink or forced air flow. Ambient temperature = 25°C. Linear Technology Magazine • March 2009 39 L DESIGN IDEAS eficiency combined with its excellent thermal management capability enables it to deliver up to 240W output power without a heat sink or forced airlow. Figure 3 shows the thermalgraphs taken with three different input voltages and loads at 25°C ambient temperature. With 240W output and 36V input, the maximum temperature rise of the LTM4609 is only 52.8°C. L1 L1,L2: FAIR-RITE 2518065007Y6 L2 VIN CBULK 100µF VIN + CIN1 10µF CIN2 10µF LTM4609 GND Figure 4. The LTM4609 µModule regulator with an input π filter. Input Ripple Reduction One way to improve eficiency in a switching DC/DC converter is to minimize the turn-on and turn-off times of the MOSFET—shorter transitions correspond to lower switch losses. However, fast transitions also lead to high frequency switching noise, which can pollute the input power source. For the applications where the input voltage ripple must be limited, a simple LC π ilter can be inserted at the input side to attenuate the high frequency input voltage noise. Figure 4 shows the LTM4609 with an input π ilter. The ilter includes two 10µF low ESR ceramic capacitors and two very small magnetic beads. For lower output power applications, only one magnetic bead is necessary. Figure 5 shows the input ripple reduction with the π ilter. Figure 5a shows the input ripple with 100µF aluminum electrolytic plus 2 × 4.7µF VIN 200mV/DIV VIN 200mV/DIV VIN = 10V VOUT = 30V IOUT = 3A 10µs/DIV CBULK = 100µF CIN1, CIN2 = 4.7µF 5a. Input voltage waveform without the input π filter shown in Figure 4 5b. Input voltage waveform with input π filter as shown in Figure 4 Figure 5. The input π filter shown in Figure 4 effectively reduces the input voltage spike caused by switching action of the MOSFETs. ceramic input capacitors. Figure 5b shows the input ripple with the ilter shown in Figure 4. Both waveforms are measured across the 100µF aluminum capacitor. A 67% reduction in input ripple is obtained with the input π ilter, which requires only two small additional magnetic beads. Table 1. Specification comparison of the LTM4605, LTM4607 and LTM4609 LTM4605 LTM4607 LTM4609 VIN 4.5V ~ 20V 4.5V ~ 36V 4.5V ~ 36V VOUT 0.8V ~ 16V 0.8V ~ 24V 0.8V ~ 34V IOUT 5A (12A in buck mode) 5A (10A in buck mode) 4A (10A in buck mode) Package 10µs/DIV VIN = 10V CBULK = 100µF VOUT = 30V CIN1, CIN2 = 10µF IOUT = 3A L1, L2: FAIR-RITE 2518065007Y6 15mm × 15mm × 2.8mm LGA Conclusion Buck-boost µModule regulators are easy-to-use, high performance solutions for applications where a regulated output voltage sits within the range of the input voltage. The 15mm × 15mm × 2.8mm LTM4609 widens the input/output voltage range of the pin compatible LTM4605 and LTM4607. The advanced package technology, as well as the high eficiency design of the LTM4609, allows it to deliver up to 240W of output power without heat sinks or forced airlow. For applications that require low input voltage ripple, a simple π ilter can be added by inserting one or two small magnetic beads to signiicantly reduce the high frequency input noise. L LT755/56, continued from page 2 PWM on- and off-times are 1µs as with the other circuits. Figure 7 shows the waveforms during a short circuit fault on the output. The input current remains in control as the switch current ramps up to the set limit of 10A, then skips the next few cycles while the current sensed by the LED 40 resistor ramps down to 1.5A. This faulted mode of circuit operation can continue indeinitely without damage to the components. Conclusion The LT3755 and LT3756 offer unparalleled performance for an LED controller generating PWM pulse widths as narrow as 1µs, which enables 50:1 PWM dimming at frequencies above the audible range. Other features include open LED protection, an open LED status indicator, and programmability of the LED current via an analog input. L Linear Technology Magazine • March 2009 NEW DEVICE CAMEOS L New Device Cameos Micropower Low Noise Boost Converter with Output Disconnect Dual 550mA, 1MHz Synchronous Boost Regulator with Output Disconnect in a The LT3495/LT3495B/LT3495-1/ 3mm × 3mm DFN LT3495B-1 is a low noise boost converter with integrated power switch, feedback resistor and output disconnect circuitry. The part controls power delivery by varying both the peak inductor current and switch off-time. This new control scheme results in low output voltage ripple as well as high eficiency over a wide load range. For the LT3495/LT3495-1, the off-time of the switch is not allowed to exceed a ixed level, guaranteeing the switching frequency stays above the audio band for the entire load range. This feature is disabled for the LT3495B/LT3495B-1, which leads to higher eficiency at light load. The difference between the LT3495/LT3495B and LT3495-1/LT3495B-1 is the level of the switch current limit. The LT3495/LT3495B has a typical peak current limit of 650mA while the LT3495-1/LT3495B-1 has a typical peak current limit of 350mA. The LT3495 series has an output disconnect PMOS that blocks the load from the input during shutdown. During normal operation, the maximum current through this PMOS is limited by circuitry inside the chip, which helps the chip survive output shorts. The input voltage range of the LT3495 series is a wide 2.5V to 16V and the output voltage can be up to 40V. In addition, the part is well compensated internally, and can be stable with very small ceramic output capacitors. Other features include low quiescent current (60µA in active mode and 0.1µA in shutdown mode), integrated output dimming, maximum switching on-time and undervoltage lockout. Combining the small ceramic capacitors and space saving 10-pin 3mm × 2mm DFN packages, the LT3495 enables compact solutions for many applications. Linear Technology Magazine • March 2009 The LTC3535 is a dual-channel 1MHz, current mode synchronous boost DC/DC converter with integrated output disconnect and soft-start. The LTC3535’s internal 550mA switches deliver output voltages as high as 5.25V from an input voltage range of 0.7V at start-up/0.5V when running to 5V, making it ideal for single- or multicell alkaline/NiMH as well as Liion/polymer applications. Each of the LTC3535’s channels has it own power input and is completely independent, offering maximum design lexibility. For example, one channel can deliver up to 50mA of continuous output current at 3.3V while the other channel delivers up to 100mA at 1.8V to power a microcontroller from a single alkaline cell. The 1MHz switching frequency minimizes external component sizes while providing up to 94% eficiency. Combined with a compact 3mm × 3mm DFN-12 package, the LTC3535 dual channel boost provides the tiny and eficient solution footprint required in handheld applications. Burst Mode ® operation lowers quiescent current to only 18µA (both channels), providing extended battery run time in handheld applications. The LTC3535 is an ideal part for handheld dual boost applications where small solution size and maximum battery run time are deining factors. Ideal Diode Equipped High Power Battery Charger Handles All Chemistries The LTC4012, LTC4012-1, LTC4012-2 and LTC4012-3 are a family of high power buck battery chargers all in a 20-lead 4mm × 4mm QFN package. Compared to the LTC4009 family of chargers, the 4012 family adds Ideal Diode™ input reverse current input protection and extends the high eficiency to higher current levels. Combined with just a few external components and external termination control, the LTC4012 family facilitates construction of chargers capable of delivering up to 4A to batteries with output power levels approaching 66W in a very small footprint. The LTC4012 family builds upon the proven quasi-constant frequency, constant off-time PWM buck control architecture as found in Linear Technology’s LTC4008. This unique buck topology provides continuous switching with synchronous rectiication even with no load current, critical to preventing audible noise in constant voltage charge termination applications. However, LTC4012 family uses switching NFETs along with an adaptive gate drive to avoid overlap conduction losses. The higher 550kHz switching frequency reduces both the inductor size and output capacitance requirements while offering eficiencies up to 95% or more. If the duty cycle goes below 20% or above 80%, the LTC4012 lowers the switching frequency to avoid pulse skipping that might otherwise begin to occur at 550kHz. Under low dropout conditions requiring high duty cycle operation, the internal watchdog timer prevents the LTC4012 from switching below 25kHz, achieving a maximum duty cycle of 98% without producing audible noise. There is also an input current monitor function that prevents input power overload when the input power is shared with a load. There are four versions of the LTC4012. The LTC4012 and LTC40123 offer a user programmable voltage set point using an external resistor divider allowing for multi-chemistry support. The Li-ion optimized LTC4012-1 and LTC4012-2 support one to four series cells via pin selection. The LTC40121 provides 4.1V/cell charging while the LTC4012-2 produces 4.2V/cell. Output voltage accuracy is typically ±0.5% and a maximum of ±0.8% over temperature. These ICs contain a switch that in shutdown removes voltage divider current drained from the 41 L NEW DEVICE CAMEOS battery whether external or internal. Programming the charge current only requires a single external resistor. The fault management system of the LTC4012 family suspends charging immediately for various conditions. First is battery overvoltage protection, which can occur with the sudden loss of battery load during bulk charge. Second, each IC features internal over-temperature protection to prevent silicon damage during elevated thermal operation. The LTC4012 family has a logic-level shutdown control input and three open-drain status outputs. First is an input current limit (ICL) status lag to tell the system when VIN is running at over 95% of its current capacity. The input current limit accuracy is typically ±3% and a maximum of ±4% over the full operating temperature range. Next is the AC present status, which indicates when VIN is within a valid range for charging under all modes of operation. The last is a charge status output can indicate bulk or C/10 charge states. The control input and status outputs of the LTC4012, along with the analog current monitor output, can be used by the host system to perform necessary preconditioning, charge termination and safety timing functions. 4MHz Synchronous StepDown DC/DC Converter Delivers up to 1.25A from a 3mm × 3mm DFN The LTC3565 is a high eficiency synchronous step-down regulator that can deliver up to 1.25A of continuous output current from a 3mm × 3mm DFN (or MSOP-10E) package. Using a constant frequency of (up to 4MHz) and current mode architecture, the LTC3565 operates from an input voltage range of 2.5V to 5.5V making it ideal for single cell Li-Ion, or multicell Alkaline/NiCad/NiMH applications. It can generate output voltages as low as 0.6V, enabling it to power the latest generation of low voltage DSPs and microcontrollers. An independent RUN pin enables simple turn-on and shutdown. Its switching frequency is user programmable from 400kHz to 4MHz, enabling the designer to optimize eficiency while avoiding critical noise-sensitive frequency bands. The combination of its 3mm × 3mm DFN-10 (or MSOP-10) package and high switching frequency keeps external inductors and capacitors small, providing a very compact, thermally eficient footprint. The LTC3565 uses internal switches with an RDS(ON) of only 0.13Ω (N-Channel lower FET) and 0.15Ω (P-Channel upper FET) to deliver eficiencies as high as 95%. It also utilizes low dropout 100% duty cycle operation to allow output voltages equal to VIN, further extending battery run time. The LTC3565 utilizes Automatic Low Ripple ( < 25mVP–P) Burst Mode® operation to offer only 40µA no load quiescent current. If the application is noise sensitive, Burst Mode operation can be disabled using a lower noise pulse-skipping mode, which still offers only 330µA of quiescent current. The LTC3565 can be synchronized to an external clock throughout its entire frequency range. Other features include ±2% output voltage accuracy and over-temperature protection. L LT75, continued from page Conclusion The ability to run from any input supply voltage ranging from 4.75V to greater than 400V and the abundance of safety features make the LT3751 an excellent choice for high voltage capacitor chargers or high voltage regulated power supplies. In fact, the LT3751 is, for now, the only 42 100 0.5 OUTPUT VOLTAGE ERROR (V) 95 EFFICIENCY (%) LT3751 controller, and the optocoupler on the feedback resistor divider. The auxiliary windings provide the desired galvanic isolation boundary while maintaining an isolated feedback path from the output node to the LT3751 FB pin. Figures 12 and 13 show the regulator’s performance. The fully isolated, high voltage input/output regulator yields over 90% eficiency. Load regulation is excellent as shown in Figure 13b, due mainly to the added gain of the optocoupler circuit. 90 85 80 POUT = 63W POUT = 48W POUT = 25W 75 70 100 120 140 160 180 200 INPUT DC VOLTAGE (V) a. Efficiency 0.25 0 –0.25 –0.5 0 50 100 150 200 250 IOUT (mA) b. Load regulation Figure 13. Fully isolated, high voltage regulator performance boundary-mode capacitor charger controller that can accurately operate from extremely high input voltages. The LT3751 simpliies design by integrating many functions that—due to cost and board real-estate—would otherwise not be realizable. Although several designs are shown here, the LT3751 includes many more features than we can show in one article. We recommended consulting the data sheet or calling the Linear Technology applications engineering department for more in-depth coverage of all available features. L Linear Technology Magazine • March 2009 DESIGN TOOLS L www.linear.com MyLinear (www.linear.com/mylinear) MyLinear is a customizable home page to store your favorite LTC products, categories, product tables, contact information, preferences and more. Creating a MyLinear account allows you to… • Store and update your contact information. No more reentering your address every time you request a sample! • Edit your subscriptions to Linear Insider email newsletter and Linear Technology magazine. • Store your favorite products and categories for future reference. • Store your favorite parametric table. Customize a table by editing columns, ilters and sort criteria and store your settings for future use. • View your sample history and delivery status. Using your MyLinear account is easy. Just visit www.linear.com/mylinear to create your account. Purchase Products Product and Applications Information At www.linear.com you will ind our complete collection of product and applications information available for download. Resources include: Data Sheets — Complete product speciications, applications information and design tips Application Notes — In depth collection of solutions, theory and design tips for a general application area Design Notes — Solution-speciic design ideas and circuit tips LT Chronicle — A monthly look at LTC products for speciic end-markets Product Press Releases — New products are announced constantly Solutions Brochures — Complete solutions for automotive electronics, high speed ADCs, LED drivers, wireless infrastructure, industrial signal chain, handheld, battery charging, and communications and industrial DC/DC conversion applications. Product Selection Purchase products directly from Linear Technology either through the methods below or contact your local Linear sales representative or licensed distributor. The focus of Linear Technology’s website is simple—to get you the information you need quickly and easily. With that goal in mind, we offer several methods of inding the product and applications information you need. Linear Express — Purchase online with credit terms. Linear Express is your new choice for purchasing any quantity of Linear Technology parts. Credit terms are available for qualifying accounts. Minimum order is only $250.00. Call 1-866-546-3271 or email us at [email protected]. Subscribe! Subscribe to Linear Technology magazine at www.linear.com/mylinear Linear Technology Magazine • March 2009 Packaging (www.linear.com/packaging) — Visit our packaging page to view complete information for all of Linear Technology’s package types. Resources include package dimensions and footprints, package cross reference, top markings, material declarations, assembly procedures and more. Quality and Reliability (www.linear.com/quality) — The cornerstone of Linear Technology’s Quality, Reliability & Service (QRS) Program is to achieve 100% customer satisfaction by producing the most technically advanced product with the best quality, on-time delivery and service. Visit our quality and reliability page to view complete reliability data for all of LTC’s products and processes. Also available is complete documentation on assembly and manufacturing lows, quality and environmental certiications, test standards and documentation and failure analysis policies and procedures. Lead Free (www.linear.com/leadfree) — A complete resource for Linear Technology’s Lead (Pb) Free Program and RoHS compliance information. Simulation & Software (www.linear.com/purchase) Credit Card Purchase — Your Linear Technology parts can be shipped almost anywhere in the world with your credit card purchase. Orders up to 500 pieces per item are accepted. You can call (408) 433-5723 or email [email protected] with questions regarding your order. Design Support Part Number and Keyword Search — Search Linear Technology’s entire library of data sheets, Application Notes and Design Notes for a speciic part number or keyword. Sortable Parametric Tables — Any of Linear Technology’s product families can be viewed in table form, allowing the parts to be sorted and iltered by one or many functional parameters. Applications Solutions — View block diagrams for a wide variety of automotive, communcations, industrial and military applications. Click on a functional block to generate a complete list of Linear Technology’s product offerings for that function. Linear Technology offers several powerful simulation tools to aid engineers in designing, testing and troubleshooting their high performance analog designs. LTspice/SwitcherCAD™ III (www.linear.com/swcad) — LTspice / SwitcherCAD III is a powerful SPICE simulator and schematic capture tool speciically designed to speed up and simplify the simulation of switching regulators. LTspice / SwitcherCAD III includes: • Powerful general purpose Spice simulator with schematic capture, waveform viewing, and speed enhancements for switching regulators. • Complete and easy to use schematic capture and waveform viewer. • Macromodels for most of Linear Technology’s switching regulators as well as models for many of Liinear’s high performance linear regulators, op amps, comparators, ilters and more. • Ready to use demonstration circuits for over one hundred of Linear Technology’s most popular products. FilterCAD — FilterCAD 3.0 is a computer-aided design program for creating ilters with Linear Technology’s ilter ICs. Noise Program — This program allows the user to calculate circuit noise using Linear Technology op amps to determine the best op amp for low noise applications. SPICE Macromodel Library — The Library includes Linear op amp SPICE macromodels for use with any SPICE simulation package. 43 SALES OFFICES NORTH AMERICA NORTHERN CALIFORNIA / NEVADA Bay Area 720 Sycamore Dr. Milpitas, CA 95035 Tel: (408) 428-2050 Fax: (408) 432-6331 Sacramento / Nevada 2260 Douglas Blvd., Ste. 280 Roseville, CA 95661 Tel: (916) 787-5210 Fax: (916) 787-0110 ASIA Iowa Tel: (319) 393-5763 Kansas Tel: (913) 829-8844 Minneapolis 7805 Telegraph Rd., Ste. 225 Bloomington, MN 55438 Tel: (952) 903-0605 Fax: (952) 903-0640 Wisconsin Tel: (262) 859-1900 PACIFIC NORTHWEST Denver 7007 Winchester Cir., Ste. 130 Boulder, CO 80301 Tel: (303) 926-0002 Fax: (303) 530-1477 NORTHEAST Boston 15 Research Place North Chelmsford, MA 01863 Tel: (978) 656-4750 Fax: (978) 656-4760 Portland 5005 SW Meadows Rd., Ste. 410 Lake Oswego, OR 97035 Tel: (503) 520-9930 Fax: (503) 520-9929 Connecticut Tel: (860) 228-4104 Salt Lake City Tel: (801) 731-8008 Seattle 2018 156th Ave. NE, Ste. 100 Bellevue, WA 98007 Tel: (425) 748-5010 Fax: (425) 748-5009 SOUTHWEST Los Angeles 21243 Ventura Blvd., Ste. 238 Woodland Hills, CA 91364 Tel: (818) 703-0835 Fax: (818) 703-0517 Orange County 15375 Barranca Pkwy., Ste. A-213 Irvine, CA 92618 Tel: (949) 453-4650 Fax: (949) 453-4765 Phoenix 2085 E. Technology Cir., Ste. 101 Tempe, AZ 85284 Tel: (480) 777-1600 Fax: (480) 838-1104 San Diego 5090 Shoreham Place, Ste. 110 San Diego, CA 92122 Tel: (858) 638-7131 Fax: (858) 638-7231 CENTRAL Chicago 2040 E. Algonquin Rd., Ste. 512 Schaumburg, IL 60173 Tel: (847) 925-0860 Fax: (847) 925-0878 Cleveland 7550 Lucerne Dr., Ste. 106 Middleburg Heights, OH 44130 Tel: (440) 239-0817 Fax: (440) 239-1466 Columbus Tel: (614) 488-4466 Detroit 39111 West Six Mile Road Livonia, MI 48152 Tel: (734) 779-1657 Fax: (734) 779-1658 Indiana Tel: (317) 581-9055 Philadelphia 3220 Tillman Dr., Ste. 120 Bensalem, PA 19020 Tel: (215) 638-9667 Fax: (215) 638-9764 SOUTHEAST Atlanta Tel: (770) 888-8137 Austin 8500 N. Mopac, Ste. 603 Austin, TX 78759 Tel: (512) 795-8000 Fax: (512) 795-0491 Dallas 17000 Dallas Pkwy., Ste. 200 Dallas, TX 75248 Tel: (972) 733-3071 Fax: (972) 380-5138 Fort Lauderdale Tel: (954) 473-1212 Houston 1080 W. Sam Houston Pkwy., Ste. 225 Houston, TX 77043 Tel: (713) 463-5001 Fax: (713) 463-5009 Huntsville Tel: (256) 881-9850 Orlando Tel: (407) 688-7616 Raleigh 15100 Weston Pkwy., Ste. 202 Cary, NC 27513 Tel: (919) 677-0066 Fax: (919) 678-0041 Tampa Tel: (813) 634-9434 EUROPE AUSTRALIA / NEW ZEALAND Linear Technology Corporation 133 Alexander Street Crows Nest NSW 2065 Australia Tel: +61 (0)2 9432 7803 Fax: +61 (0)2 9439 2738 JAPAN Linear Technology KK 8F Shuwa Kioicho Park Bldg. 3-6 Kioicho Chiyoda-ku Tokyo, 102-0094, Japan Tel: +81 (3) 5226-7291 Fax: +81 (3) 5226-0268 CHINA Linear Technology Corp. Ltd. Units 1503-04, Metroplaza Tower 2 223 Hing Fong Road Kwai Fong, N.T., Hong Kong Tel: +852 2428-0303 Fax: +852 2348-0885 Linear Technology KK 6F Kearny Place Honmachi Bldg. 1-6-13 Awaza, Nishi-ku Osaka-shi, 550-0011, Japan Tel: +81 (6) 6533-5880 Fax: +81 (6) 6543-2588 Linear Technology Corp. Ltd. Room 2701, City Gateway No. 398 Cao Xi North Road Shanghai, 200030, PRC Tel: +86 (21) 6375-9478 Fax: +86 (21) 5465-5918 Linear Technology Corp. Ltd. Room 816, 8/F China Electronics Plaza B No. 3 Dan Ling Rd., Hai Dian District Beijing, 100080, PRC Tel: +86 (10) 6801-1080 Fax: +86 (10) 6805-4030 Linear Technology Corp. Ltd. Room 2604, 26/F Excellence Times Square Building 4068 YiTian Road, Futian District Shenzhen, 518048, PRC Tel: +86 755-8236-6088 Fax: +86 755-8236-6008 Linear Technology KK 7F, Sakuradori Ohtsu KT Bldg. 3-20-22 Marunouchi, Naka-ku Nagoya-shi, 460-0002, Japan Tel: +81 (52) 955-0056 Fax: +81 (52) 955-0058 KOREA Linear Technology Korea Co., Ltd. Yundang Building, #1002 Samsung-Dong 144-23 Kangnam-Ku, Seoul 135-090 Korea Tel: +82 (2) 792-1617 Fax: +82 (2) 792-1619 SINGAPORE Linear Technology Pte. Ltd. 507 Yishun Industrial Park A Singapore 768734 Tel: +65 6753-2692 Fax: +65 6752-0108 TAIWAN Linear Technology Corporation 8F-1, 77, Nanking E. Rd., Sec. 3 Taipei, Taiwan Tel: +886 (2) 2505-2622 Fax: +886 (2) 2516-0702 FINLAND Linear Technology AB Teknobulevardi 3-5 P.O. Box 35 FIN-01531 Vantaa Finland Tel: +358 (0)46 712 2171 Fax: +358 (0)46 712 2175 FRANCE Linear Technology S.A.R.L. Parc Tertiaire Silic 2 Rue de la Couture, BP10217 94518 Rungis Cedex France Tel: +33 (1) 56 70 19 90 Fax: +33 (1) 56 70 19 94 GERMANY Linear Technology GmbH Osterfeldstrasse 84, Haus C D-85737 Ismaning Germany Tel: +49 (89) 962455-0 Fax: +49 (89) 963147 Linear Technology GmbH Haselburger Damm 4 D-59387 Ascheberg Germany Tel: +49 (2593) 9516-0 Fax: +49 (2593) 951679 Linear Technology GmbH Jesinger Strasse 65 D-73230 Kirchheim/Teck Germany Tel: +49 (0)7021 80770 Fax: +49 (0)7021 807720 ITALY Linear Technology Italy Srl Orione 3, C.D. Colleoni Via Colleoni, 17 I-20041 Agrate Brianza (MI) Italy Tel: +39 039 596 5080 Fax: +39 039 596 5090 SWEDEN Linear Technology AB Electrum 204 Isafjordsgatan 22 SE-164 40 Kista Sweden Tel: +46 (8) 623 16 00 Fax: +46 (8) 623 16 50 UNITED KINGDOM Linear Technology (UK) Ltd. 3 The Listons, Liston Road Marlow, Buckinghamshire SL7 1FD United Kingdom Tel: +44 (1628) 477066 Fax: +44 (1628) 478153 CANADA Calgary, AB Tel: (403) 455-3577 Montreal, QC Tel: (450) 689-2660 Ottawa, ON Tel: (613) 421-3090 Toronto, ON Tel: (440) 239-0817 Vancouver, BC Tel: (604) 729-1204 Linear Technology Corporation 1630 McCarthy Blvd. Milpitas, CA 95035-7417 1-800-4-LINEAR • 408-432-1900 • 408-434-0507 (fax) www.linear.com To subscribe to Linear Technology magazine, visit www.linear.com/mylinear © 2009 Linear Technology Corporation/Printed in U.S.A./42.6K Linear Technology Magazine • March 2009