CA3160, CA3160A Data Sheet September 1998 4MHz, BiMOS Operational Amplifier with MOSFET Input/CMOS Output The CA3160A and CA3160 are integrated circuit operational amplifiers that combine the advantage of both CMOS and bipolar transistors on a monolithic chip. The CA3160 series are frequency compensated versions of the popular CA3130 series. Gate protected P-Channel MOSFET (PMOS) transistors are used in the input circuit to provide very high input impedance, very low input current, and exceptional speed performance. The use of PMOS field effect transistors in the input stage results in common-mode input voltage capability down to 0.5V below the negative supply terminal, an important attribute in single supply applications. File Number Features • MOSFET Input Stage Provides: - Very High ZI = 1.5TΩ (1.5 x 1012Ω) (Typ) - Very Low II . . . . . . . . . . . . . 5pA (Typ) at 15V Operation . . . . . . . . . . . . . . . . . . . . . . . 2pA (Typ) at 5V Operation • Common-Mode Input Voltage Range Includes Negative Supply Rail; Input Terminals Can Be Swung 0.5V Below Negative Supply Rail • CMOS Output Stage Permits Signal Swing to Either (or Both) Supply Rails Applications • Ground Referenced Single Supply Amplifiers • Fast Sample Hold Amplifiers A complementary symmetry MOS (CMOS) transistor-pair, capable of swinging the output voltage to within 10mV of either supply voltage terminal (at very high values of load impedance), is employed as the output circuit. • Long Duration Timers/Monostables The CA3160 Series circuits operate at supply voltages ranging from 5V to 16V, or ±2.5V to ±8V when using split supplies, and have terminals for adjustment of offset voltage for applications requiring offset null capability. Terminal provisions are also made to permit strobing of the output stage. • Wien-Bridge Oscillators • High Input Impedance Wideband Amplifiers • Voltage Followers (e.g., Follower for Single Supply D/A Converter) • Voltage Controlled Oscillators • Photo Diode Sensor Amplifiers Pinouts CA3160 (METAL CAN) TOP VIEW The CA3160A offers superior input characteristics over those of the CA3160. PART NUMBER TEMP. RANGE (oC) PACKAGE PKG. NO. CA3160AE -55 to 125 8 Ld PDIP E8.3 CA3160E -55 to 125 8 Ld PDIP E8.3 CA3160T -55 to 125 TAB SUPPLEMENTARY COMPENSATION Ordering Information 8 Pin Metal Can 976.3 OFFSET NULL INV. INPUT 1 7 - 2 NON-INV. INPUT STROBE 8 6 + 5 3 4 T8.C V+ OUTPUT OFFSET NULL V- AND CASE CA3160 (PDIP) TOP VIEW OFFSET NULL INV. INPUT NON-INV. INPUT V- 1 2 3 4 + 8 STROBE 7 V+ 6 OUTPUT 5 OFFSET NULL NOTE: CA3160 Series devices have an on-chip frequency compensation network. Supplementary phase compensation or frequency roll-off (if desired) can be connected externally between Terminals 1 and 8. 1 CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures. 1-888-INTERSIL or 321-724-7143 | Copyright © Intersil Corporation 1999 CA3160, CA3160A Absolute Maximum Ratings Thermal Information Supply Voltage (Between V+ and V- Terminals) . . . . . . . . . . . +16V Differential Mode Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . .8V Input Voltage . . . . . . . . . . . . . . . . . . . . . . . . . (V+ +8V) to (V- -0.5V) Input Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1mA Output Short Circuit Duration (Note 2). . . . . . . . . . . . . . . . Indefinite Thermal Resistance (Typical, Note 1) θJA (oC/W) θJC (oC/W) PDIP Package . . . . . . . . . . . . . . . . . . . 110 N/A Metal Can Package . . . . . . . . . . . . . . . 170 85 Maximum Junction Temperature (Metal Can). . . . . . . . . . . . . . .175oC Maximum Junction Temperature (Plastic Package) . . . . . . . .150oC Maximum Storage Temperature Range . . . . . . . . . . -65oC to 150oC Maximum Lead Temperature (Soldering 10s) . . . . . . . . . . . . 300oC Operating Conditions Temperature Range . . . . . . . . . . . . . . . . . . . . . . . . . -55oC to 125oC CAUTION: Stresses above those listed in “Absolute Maximum Ratings” may cause permanent damage to the device. This is a stress only rating and operation of the device at these or any other conditions above those indicated in the operational sections of this specification is not implied. NOTES: 1. θJA is measured with the component mounted on an evaluation PC board in free air. 2. Short Circuit may be applied to ground or to either supply. TA = 25oC, V+ = 15V, V- = 0V, Unless Otherwise Specified Electrical Specifications CA3160 PARAMETER SYMBOL TEST CONDITIONS CA3160A MIN TYP MAX MIN TYP MAX UNITS Input Offset Voltage |VIO| VS = ±7.5V - 6 15 - 2 5 mV Input Offset Current |IIO| VS = ±7.5V - 0.5 30 - 0.5 20 pA II VS = ±7.5V - 5 50 - 5 30 pA 50 320 - 50 320 - kV/V 94 110 - 94 110 - dB CMRR 70 90 - 80 95 - dB VlCR 0 -0.5 to 12 10 0 -0.5 to 12 10 V - 32 320 - 32 150 µV/V 12 13.3 - 12 13.3 - V - 0.002 0.01 - 0.002 0.01 V 14.99 15 - 14.99 15 - V - 0 0.01 - 0 0.01 V Input Current Large-Signal Voltage Gain AOL Common-Mode Rejection Ratio Common-Mode Input-Voltage Range VO = 10VP-P, RL = 2kΩ Power-Supply Rejection Ratio PSRR ∆VIO/∆VS, VS = ±7.5V Maximum Output Voltage VOM+ RL = 2kΩ VOMVOM+ RL = ∞ VOMMaximum Output Current Supply Current (Note 3) IOM+ VO = 0V (Source) 12 22 45 12 22 45 mA IOM- VO = 15V (Sink) 12 20 45 12 20 45 mA VO = 7.5V, R L = ∞ - 10 15 - 10 15 mA VO = 0V, R L = ∞ - 2 3 - 2 3 mA ∆VIO/∆T - 8 - - 6 - µV/oC I+ Input Offset Voltage Temperature Drift For Design Guidance, VSUPPLY = ±7.5V, TA = 25oC, Unless Otherwise Specified Electrical Specifications PARAMETER SYMBOL Input Offset Voltage Adjustment Range TYP TYP UNITS ±22 ±22 mV 1.5 1.5 TΩ 4.3 4.3 pF RS = 1MΩ 40 40 µV RS = 10MΩ 50 50 µV 10kΩ Across Terminals 4 and 5 or Terminals 4 and 1 RI Input Capacitance CI f = 1MHz Equivalent Input Noise Voltage eN BW = 0.2MHz eN 2 CA3160A TEST CONDITIONS Input Resistance Equivalent Input Noise Voltage CA3160 RS = 100Ω 1kHz 72 72 nV/√Hz 10kHz 30 30 nV/√Hz CA3160, CA3160A For Design Guidance, VSUPPLY = ±7.5V, TA = 25oC, Unless Otherwise Specified (Continued) Electrical Specifications PARAMETER SYMBOL TEST CONDITIONS CA3160 CA3160A TYP TYP UNITS Unity Gain Crossover Frequency fT 4 4 MHz Slew Rate SR 10 10 V/µs CL = 25pF, RL = 2kΩ, (Voltage Follower) 0.09 0.09 µs 10 10 % CL = 25pF, RL = 2kΩ, (Voltage Follower) To <0.1%, VIN = 4VP-P 1.8 1.8 µs Transient Response Rise and Fall Time tr Overshoot OS Settling Time tS For Design Guidance, V+ = +5V, V- = 0V, TA = 25oC, Unless Otherwise Specified Electrical Specifications PARAMETER SYMBOL TEST CONDITIONS CA3160 CA3160A TYP TYP UNITS Input Offset Voltage VIO 6 2 mV Input Offset Current IIO 0.1 0.1 pA Il 2 2 pA CMRR 80 90 dB 100 100 kV/V 100 100 dB 0 to 2.8 0 to 2.8 V VO = 5V, RL = ∞ 300 300 µA VO = 2.5V, RL = ∞ 500 500 µA ∆VIO/∆V+ 200 200 µV/V Input Current Common-Mode Rejection Ratio Large Signal Voltage Gain AOL Common-Mode Input Voltage Range VlCR Supply Current I+ Power Supply Rejection Ratio VO = 4VP-P, RL = 5kΩ PSRR NOTE: 3. ICC typically increases by 1.5mA/MHz during operation. Block Diagram 7 200µA 1.35mA 8mA (NOTE 4) 200µA NOTES: 0mA (NOTE 5) BIAS CKT. V+ 4. Total supply voltage (for indicated voltage gains) = 15V with input terminals biased so that Terminal 6 potential is +7.5V above Terminal 4. 5. Total supply voltage (for indicated voltage gains) = 15V with output terminal driven to either supply rail. + 3 OUTPUT AV ≈ 6000X AV ≈ 5X INPUT AV ≈ 30X 6 2 - 4 CC 5 1 8 COMPENSATION (WHEN DESIRED) OFFSET NULL 3 STROBE V- CA3160, CA3160A Schematic Diagram BIAS CURRENT Q1 D1 Z1 8.3V D2 7 “CURRENT SOURCE LOAD” FOR Q11 CURRENT SOURCE FOR Q6 AND Q7 Q2 Q3 Q4 Q5 V+ D3 D4 R1 40kΩ R2 5kΩ INPUT STAGE D5 NON-INV. INPUT 3 2 D7 D6 SECOND STAGE OUTPUT STAGE + Q6 Q7 R3 1kΩ R5 1kΩ 6 30 pF R4 1kΩ Q9 OUTPUT 2kΩ - INV. INPUT Q8 Q12 Q11 Q10 R6 1kΩ 5 1 OFFSET NULL SUPPLEMENTARY COMP IF DESIRED 8 4 STROBING NOTE: Diodes D5 Through D7 Provide Gate Oxide Protection For MOSFET Input Stage. Application Information Circuit Description Refer to the Block Diagram of the CA3160 series CMOS Operational Amplifiers. The input terminals may be operated down to 0.5V below the negative supply rail, and the output can be swung very close to either supply rail in many applications. Consequently, the CA3160 series circuits are ideal for single supply operation. Three class A amplifier stages, having the individual gain capability and current consumption shown in the Block Diagram provide the total gain of the CA3160. A biasing circuit provides two potentials for common use in the first and second stages. Terminals 8 and 1 can be used to supplement the internal phase compensation network if additional phase compensation or frequency roll-off is desired. Terminals 8 and 4 can also be used to strobe the output stage into a low quiescent current state. When Terminal 8 is tied to the negative supply rail (Terminal 4) by mechanical or electrical means, the output potential at Terminal 6 essentially rises to the positive supplyrail potential at Terminal 7. This condition of essentially zero current drain in the output stage under the strobed “OFF” condition can only be achieved when the ohmic load 4 resistance presented to the amplifier is very high (e.g., when the amplifier output is used to drive MOS digital circuits in comparator applications). Input Stage - The circuit of the CA3160 is shown in the Schematic Diagram. It consists of a differential-input stage using PMOS field-effect transistors (Q6, Q7) working into a mirror-pair of bipolar transistors (Q9, Q10) functioning as load resistors together with resistors R3 through R6. The mirrorpair transistors also function as a differential-to-single-ended converter to provide base drive to the second-stage bipolar transistor (Q11). Offset nulling, when desired, can be effected by connecting a 100,000Ω potentiometer across Terminals 1 and 5 and the potentiometer slider arm to Terminal 4. Cascode-connected PMOS transistors Q2, Q4, are the constant-current source for the input stage. The biasing circuit for the constant-current source is subsequently described. The small diodes D5 through D7 provide gate-oxide protection against high-voltage transients, including static electricity during handling for Q6 and Q7. Second-Stage - Most of the voltage gain in the CA3160 is provided by the second amplifier stage, consisting of bipolar CA3160, CA3160A transistor Q11 and its cascode-connected load resistance provided by PMOS transistors Q3 and Q5. The source of bias potentials for these PMOS transistors is described later. Miller Effect compensation (roll off) is accomplished by means of the 30pF capacitor and 2kΩ resistor connected between the base and collector of transistor Q11. These internal components provide sufficient compensation for unity gain operation in most applications. However, additional compensation, if desired, may be used between Terminals 1 and 8. Bias-Source Circuit - At total supply voltages, somewhat above 8.3V, resistor R2 and zener diode Z1 serve to establish a voltage of 8.3V across the series-connected circuit, consisting of resistor R1, diodes D1 through D4, and PMOS transistor Q1. A tap at the junction of resistor R1 and diode D4 provides a gate-bias potential of about 4.5V for PMOS transistors Q4 and Q5 with respect to Terminal 7. A potential of about 2.2V is developed across diode-connected PMOS transistor Q1 with respect to Terminal 7 to provide gate bias for PMOS transistors Q2 and Q3. It should be noted that Q1 is “mirror-connected” to both Q2 and Q3. Since transistors Q1, Q2, Q3 are designed to be identical, the approximately 200µA current in Q1 establishes a similar current in Q2 and Q3 as constant-current sources for both the first and second amplifier stages, respectively. At total supply voltages somewhat less than 8.3V, zener diode Z1 becomes nonconductive and the potential, developed across series-connected R1, D1 - D4, and Q1, varies directly with variations in supply voltage. Consequently, the gate bias for Q4, Q5 and Q2, Q3 varies in accordance with supplyvoltage variations. This variation results in deterioration of the power-supply-rejection ratio (PSRR) at total supply voltages below 8.3V. Operation at total supply voltages below about 4.5V results in seriously degraded performance. Output Stage - The output stage consists of a drain-loaded inverting amplifier using CMOS transistors operating in the Class A mode. When operating into very high resistance loads, the output can be swung within millivolts of either supply rail. Because the output stage is a drain-loaded amplifier, its gain is dependent upon the load impedance. The transfer characteristics of the output stage for a load returned to the negative supply rail are shown in Figure 17. Typical op amp loads are readily driven by the output stage. Because largesignal excursions are non-linear, requiring feedback for good waveform reproduction, transient delays may be encountered. As a voltage follower, the amplifier can achieve 0.01% accuracy levels, including the negative supply rail. Offset Nulling Offset-voltage nulling is usually accomplished with a 100,000Ω potentiometer connected across Terminals 1 and 5 and with the potentiometer slider arm connected to Terminal 4. A fine offset-null adjustment usually can be effected with the slider arm positioned in the mid-point of the potentiometer's total range. 5 Input Current Variation with Common Mode Input Voltage As shown in the Electrical Specifications, the input current for the CA3160 Series Op Amps is typically 5pA at TA = 25oC when Terminals 2 and 3 are at a common-mode potential of +7.5V with respect to negative supply Terminal 4. Figure 23 contains data showing the variation of input current as a function of common-mode input voltage at TA = 25oC. These data show that circuit designers can advantageously exploit these characteristics to design circuits which typically require an input current of less than 1pA, provided the common-mode input voltage does not exceed 2V. As previously noted, the input current is essentially the result of the leakage current through the gate-protection diodes in the input circuit and, therefore, a function of the applied voltage. Although the finite resistance of the glass terminal-to-case insulator of the metal can package also contributes an increment of leakage current, there are useful compensating factors. Because the gateprotection network functions as if it is connected to Terminal 4 potential, and the metal can case of the CA3160 is also internally tied to Terminal 4, input Terminal 3 is essentially “guarded” from spurious leakage currents. Input-Current Variation with Temperature The input current of the CA3160 Series circuits is typically 5pA at 25oC. The major portion of this input current is due to leakage current through the gate-protective diodes in the input circuit. As with any semiconductor junction device, including op amps with a junction-FET input stage, the leakage current approximately doubles for every 10oC increase in temperature. Figure 24 provides data on the typical variation of input bias current as a function of temperature in the CA3160. In applications requiring the lowest practical input current and incremental increases in current because of “warm-up” effects, it is suggested that an appropriate heat sink be used with the CA3160. In addition, when “sinking” or “sourcing” significant output current the chip temperature increases, causing an increase in the input current. In such cases, heat-sinking can also very markedly reduce and stabilize input current variations. Input Offset Voltage (VIO) Variation with DC Bias vs Device Operating Life It is well known that the characteristics of a MOSFET device can change slightly when a DC gate-source bias potential is applied to the device for extended time periods. The magnitude of the change is increased at high temperatures. Users of the CA3160 should be alert to the possible impacts of this effect if the application of the device involves extended operation at high temperatures with a significant differential DC bias voltage applied across Terminals 2 and 3. Figure 25 shows typical data pertinent to shifts in offset voltage encountered with CA3160 devices in metal can packages during life testing. At lower temperatures (metal can and plastic) for example at 85oC, this change in voltage is considerably less. In typical linear applications where the differential voltage is small and symmetrical, these incremental changes are of about the same CA3160, CA3160A magnitude as those encountered in an operational amplifier employing a bipolar transistor input stage. The 2V differential voltage example represents conditions when the amplifier output state is “toggled”, e.g., as in comparator applications. Power Supply Considerations Because the CA3160 is very useful in single supply applications, it is pertinent to review some considerations relating to power supply current consumption under both single and dual supply service. Figures 1A and 1B show the CA3160 connected for both dual and single supply operation. Dual-supply operation: When the output voltage at Terminal 6 is 0V, the currents supplied by the two power supplies are equal. When the gate terminals of Q8 and Q12 are driven increasingly positive with respect to ground, current flow through Q12 (from the negative supply) to the load is increased and current flow through Q8 (from the positive supply) decreases correspondingly. When the gate terminals of Q8 and Q12 are driven increasingly negative with respect to ground, current flow through Q8 is increased and current flow through Q12 is decreased accordingly. Single supply operation: Initially, let it be assumed that the value of RL is very high (or disconnected), and that the inputterminal bias (Terminals 2 and 3) is such that the output terminal (No. 6) voltage is at V+/2, i.e., the voltage-drops across Q8 and Q12 are of equal magnitude. Figure 18 shows typical quiescent supply-current vs supply voltage for the CA3160 operated under these conditions. Since the output stage is operating as a Class A amplifier, the supply current will remain constant under dynamic operating conditions as long as the transistors are operated in the linear portion of their voltage-transfer characteristics (see Figure 17). If either Q8 or Q12 are swung out of their linear regions toward cutoff (a non-linear region), there will be a corresponding reduction in supply-current. In the extreme case, e.g., with Terminal 8 swung down to ground potential (or tied to ground), NMOS transistor Q12 is completely cut off and the supply current to series connected transistors Q8, Q12 goes essentially to zero. The two preceding stages in the CA3160, however, continue to draw modest supply-current (see the lower curve in Figure 18) even though the output stage is strobed off. Figure 1A shows a dual-supply arrangement for the output stage that can also be strobed off, assuming RL = ∞, by pulling the potential of Terminal 8 down to that of Terminal 4. Let it now-be assumed that a load resistance of nominal value (e.g., 2kΩ) is connected between Terminal 6 and ground in the circuit of Figure 1B. Let it further be assumed again that the input-terminal bias (Terminals 2 and 3) is such that the output terminal (No. 6) voltage is at V+/2. Since PMOS transistor Q8 must now supply quiescent current to both RL and transistor Q12, it should be apparent that under these conditions the supply current must increase as an inverse function of the RL magnitude. Figure 20 shows the voltage-drop across PMOS transistor Q8 as a function of load current at several supply 6 voltages. Figure 17 shows the voltage transfer characteristics of the output stage for several values of load resistance. Wideband Noise From the standpoint of low-noise performance considerations, the use of the CA3160 is most advantageous in applications where in the source resistance of the input signal is on the order of 1MΩ or more. In this case, the total input-referred noise voltage is typically only 40µV when the test circuit amplifier of Figure 2 is operated at a total supply voltage of 15V. This value of total input-referred noise remains essentially constant, even though the value of source resistance is raised by an order of magnitude. This characteristic is due to the fact that reactance of the input capacitance becomes a significant factor in shunting the source resistance. It should be noted, however, that for values of source resistance very much greater than 1MΩ, the total noise voltage generated can be dominated by the thermal noise contributions of both the feedback and source resistors. 7 3 V+ + Q8 CA3160 OUTPUT Q12 STAGE 6 RL 2 - 4 V- 8 NEGATIVE SUPPLY FIGURE 1A. DUAL POWER SUPPLY OPERATION V+ 7 3 + Q8 CA3160 OUTPUT Q12 STAGE 6 RL 2 - 4 8 FIGURE 1B. SINGLE POWER SUPPLY OPERATION FIGURE 1. CA3160 OUTPUT STAGE IN DUAL AND SINGLE POWER SUPPLY OPERATION CA3160, CA3160A +7.5V 0.01µF RS 3 1MΩ 7 + 2 - NOISE VOLTAGE OUTPUT 6 CA3160 4 30.1kΩ 0.01 µF -7.5V BW (3dB) = 200kHz TOTAL NOISE VOLTAGE (INPUT REFERRED = 40µV (TYP) 1kΩ FIGURE 2. TEST CIRCUIT AMPLIFIER (30dB GAIN) USED FOR WIDEBAND NOISE MEASUREMENTS Typical Performance Curves +7.5V 0.01µF 3 10kΩ 7 + 6 CA3160 2 - 4 2kΩ 0.01 µF -7.5V 2kΩ BW (-3dB) = 4MHz SR = 10V/µs 25pF SIMULATED LOAD CAPACITANCE 0.1µF FIGURE 3A. Top Trace: Output Bottom Trace: Input FIGURE 3B. SMALL SIGNAL RESPONSE Top Trace: Output Signal Center Trace: Difference Signal 5mV/Div. Bottom Trace: Input Signal FIGURE 3C. INPUT-OUTPUT DIFFERENCE SIGNAL SHOWING SETTLING TIME FIGURE 3. DUAL SUPPLY VOLTAGE FOLLOWER WITH ASSOCIATED WAVEFORMS 7 CA3160, CA3160A Typical Applications +15V Voltage Followers Operational amplifiers with very high input resistances, like the CA3160, are particularly suited to service as voltage followers. Figure 3 shows the circuit of a classical voltage follower, together with pertinent waveforms using the CA3160 in a split-supply configuration. A voltage follower, operated from a single supply, is shown in Figure 4 together with related waveforms. This follower circuit is linear over a wide dynamic range, as illustrated by the reproduction of the output waveform in Figure 4B with inputsignal ramping. The waveforms in Figure 4C show that the follower does not lose its input-to-output phase-sense, even though the input is being swung 7.5V below ground potential. This unique characteristic is an important attribute in both operational amplifier and comparator applications. Figure 4C also shows the manner in which the COS/MOS output stage permits the output signal to swing down to the negative supply-rail potential (i.e., ground in the case shown). The digital-to-analog converter (DAC) circuit, described in the following section, illustrates the practical use of the CA3160 in a single supply voltage follower application. 0.01µF 3 10kΩ 7 + 6 CA3160 2 - 4 5 1 OFFSET ADJUST 2kΩ BW (-3dB) = 4MHz SR = 10V/µs 0.1µF FIGURE 4A. 9-Bit CMOS DAC A typical circuit of a 9-bit Digital-to-Analog Converter (DAC) (see Note 6) is shown in Figure 5. This system combines the concepts of multiple-switch CMOS lCs, a low-cost ladder network of discrete metal-oxide-film resistors, a CA3160 op amp connected as a follower, and an inexpensive monolithic regulator in a simple single power-supply arrangement. An additional feature of the DAC is that it is readily interfaced with CMOS input logic, e.g., 10V logic levels are used in the circuit of Figure 5. The circuit uses an R/2R voltage-ladder network, with the outputpotential obtained directly by terminating the ladder arms at either the positive or the negative power supply terminal. Each CD4007A contains three inverters, each inverter functioning as a single-pole double-throw switch to terminate an arm of the R/2R network at either the positive or negative power-supply terminal. The resistor ladder is an assembly of 1% tolerance metal-oxide film resistors. The five arms requiring the highest accuracy are assembled with series and parallel combinations of 806,000Ω resistors from the same manufacturing lot. A single 15V supply provides a positive bus for the CA3160 follower amplifier and feeds the CA3085 voltage regulator. A “scale-adjust” function is provided by the regulator output control, set to a nominal 10V level in this system. The line-voltage regulation (approximately 0.2%) permits a 9-bit accuracy to be maintained with variations of several volts in the supply. The flexibility afforded by the CMOS building blocks simplifies the design of DAC systems tailored to particular needs. NOTE: 6. “Digital-to-Analog Conversion Using the Intersil CD4007A COS/MOS lC”, Application Note AN6080. 8 Top Trace: Output Bottom Trace: Input FIGURE 4B. OUTPUT WAVEFORM WITH GROUND REFERENCE SINE WAVE INPUT FIGURE 4C. OUTPUT SIGNAL WITH INPUT SIGNAL RAMPING FIGURE 4. SINGLE SUPPLY VOLTAGE FOLLOWER WITH ASSOCIATED WAVEFORMS. (e.g., FOR USE IN SINGLE SUPPLY D/A CONVERTER; SEE FIGURE 9 IN AN6080) CA3160, CA3160A Error-Amplifier in Regulated Power Supplies amplitude (V) and width (T2). Since the output (Terminal 6) of A1 (a CA3130) can swing within about 10mV of either supplyrail, the output pulse amplitude (V) is essentially equal to V+. The average output voltage (EAVG = V T2/T1) is applied to the non-inverting Input terminal of comparator A2 via an integrating network R3, C2. Comparator A2 operates to establish circuit conditions such that EAVG = V1. This circuit condition is accomplished by feeding an output signal from Terminal 6 of A2 through R4, D4 to the inverting terminal (Terminal 2) of A1, thereby adjusting the multivibrator interval, T3. The CA3160 is an ideal choice for error-amplifier service in regulated power supplies since it can function as an erroramplifier when the regulated output voltage is required to approach zero. The circuit shown in Figure 6 uses a CA3160 as an error amplifier in a continuously adjustable 1A power supply. One of the key features of this circuit is its ability to regulate down to the vicinity of 0V with only one DC power supply input. An RC network, connected between the base of the output drive transistor and the input voltage, prevents “turn-on overshoot”, a condition typical of many operational amplifier regulator circuits. As the amplifier becomes operational, this RC network ceases to have any influence on the regulator performance. Voltmeter With High Input Resistance The voltmeter circuit shown in Figure 8 illustrates an application in which a number of the CA3160 characteristics are exploited. Range-switch SW1 is ganged between input and output circuitry to permit selection of the proper output voltage for feedback to Terminal 2 via 10kΩ current-limiting resistor. The circuit is powered by a single 8.4V mercury battery. With zero input signal, the circuit consumes somewhat less than 500µA plus the meter current required to indicate a given voltage. Thus, at full scale input, the total supply current rises to slightly more than 1500µA. Precision Voltage-Controlled Oscillator The circuit diagram of a precision voltage-controlled oscillator is shown in Figure 7. The oscillator operates with a tracking error in the order of 0.02% and a temperature coefficient of 0.01%/oC. A multivibrator (A1) generates pulses of constant 10V LOGIC INPUTS +10.010V 14 LSB 9 8 7 6 3 10 11 6 5 4 3 2 MSB 1 6 3 10 6 3 10 2 CD4007A “SWITCHES” 9 13 1 7 8 5 4 806K 1% CD4007A “SWITCHES” 12 402K 1% 13 1 8 5 200K 1% VOLTAGE REGULATOR +15V 2 62 1 +10.010V CA3085 8 6 3 22.1K 1% 7 + - 4 2µF 25V 1K REGULATED VOLTAGE ADJUST 3.83K 1% 0.001µF 13 12 806K 1% 806K 1% 100K 1% 806K 1% 806K 1% CD4007A “SWITCHES” 1 8 806K 1% 12 5 (2) 806K 1% (4) 806K 1% (8) 806K 1% +15V 750K 1% BIT REQUIRED RATIO-MATCH 1 Standard 2 ±0.1% 3 ±0.2% 4 ±0.4% 5 ±0.8% 6-9 ±1% ABS. PARALLELED RESISTORS OUTPUT + 3 VOLTAGE FOLLOWER CA3160 6 - 4 5 LOAD FIGURE 5. 9-BIT DAC USING CMOS DIGITAL SWITCHES AND CA3160 9 10K 7 1 100K OFFSET NULL 2K 0.1µF 2 CA3160, CA3160A 2N6385 POWER DARLINGTON 3 SHORT-CIRCUIT CURRENT LIMIT ADJUSTMENT 40V INPUT + 1Ω 2 TURN ON DELAY 2.4kΩ 1W 1kΩ 1.5kΩ 1W 100kΩ 1 2N2102 1kΩ 56pF 1N914 7 + + 5µF 10 9 7 3 5 6 4 - 3 CA3160 6 1 2N2102 8 + 100µF 10kΩ + 2kΩ - CA3086 11 2 43kΩ 8 2.2kΩ 100µF 25V OUTPUT 0V TO 35V AT 1A 10kΩ 0.2µF - 5 2 1 12 10kΩ 14 13 8.2kΩ 4 4.7kΩ 1kΩ 50kΩ 100kΩ 0.01µF 62kΩ - - Hum and Noise Output <250µVRMS; Regulation (No Load to Full Load) <0.005%; Input Regulation <0.01%/V FIGURE 6. VOLTAGE REGULATOR CIRCUIT (0.1V TO 35V AT 1A) T2 T3 V VCO CONTROL VOLTAGE (VI) (0V - 10V) (SENSITIVITY = 1kHz/V) fO +15V T1 D1 10K 1M 0.01µF +15V R5 100K 100K 3 R6 100K +15V D2 0.1 µF 7 + MULTIVIBRATOR CA3130 C1 500pF 2 2 EAVG = V T2/T1 6 R3 1M - 3 4 D4 R1 182K R2 10K D1 - D5 = 1N914 D5 COMPARATOR CA3160 + 4 6 5 C2 0.01µF D3 7 - 0.01µF 1 R7 100K R4 3K FIGURE 7. VOLTAGE CONTROLLED OSCILLATOR Function Generator A function generator having a wide tuning range is shown in Figure 9. The adjustment range, in excess of 1,000,000/1, is accomplished by a single potentiometer. Three operational amplifiers are utilized: a CA3160 as a voltage follower, a CA3080 as a high speed comparator, and a second 10 CA3080A as a programmable current source. Three variable capacitors C1, C2, and C3 shape the triangular signal between 500kHz and 1MHz. Capacitors C4, C5, and the trimmer potentiometer in series with C5 maintain essentially constant (+10%) amplitude up to 1MHz. CA3160, CA3160A 300V 300V 100MΩ 100V 100V 30V 30V 1.02 MΩ 10V BATTERY TEST OFF ON 3 POSITION SLIDE SWITCH 9.9kΩ 10V + 500 µF SW1A 3V INPUT SW1B 3V 1V 1V 300V 300V 100V 100V 30V 30V 10V 10V 0-1mA 7 3 + 22MΩ 2.7kΩ - 2 300V 100V 300V 4 820Ω 200Ω 30V 100V 5 1 30V 100kΩ ZERO ADJUST 3V CAL. 500Ω 6 CA3160 0.001µF M BATTERY +9V BATTERY 10V 1V CAL. 10V 3V SW1C 3V 1V 300mV 1V 9.1kΩ 9kΩ 100mV 300mV 100mV 10kΩ SW1D 30mV 900Ω 10mV 30mV 10mV 100Ω FIGURE 8. HIGH INPUT RESISTANCE DC VOLTMETER 20pF 8.2kΩ +7.5V 0.9 - 7pF C1 VOLTAGE-CONTROLLED CURRENT SOURCE 7 3 + 6.2kΩ CA3080A 1kΩ 2 - 1kΩ 6 10-80pF 4 5 C2 2MΩ 3 0.1µF SYMMETRY -7.5V 100kΩ +7.5V 5 + 6 3 4 MIN FREQ.SET -7.5V +7.5V 10kΩ 6.2kΩ 500Ω FREQ ADJUST 500Ω 2 CA3080 -7.5V 0.1µF C4 4 - 60pF 2kΩ HIGH FREQ LEVEL ADJUST FIGURE 9A. 1,000,000/1 SINGLE CONTROL FUNCTION GENERATOR: 1Hz to 1MHz 11 6 + 10kΩ -7.5V MAX FREQ SET 7 10kΩ 4 - 60pF CA3160 C3 2 EXTERNAL SWEEPING INPUT 30kΩ 6.8MΩ 7 -7.5V 4.7kΩ -7.5V 430pF +7.5V HIGH FREQ. SHAPE THRESHOLD DETECTOR +7.5V +7.5V CENTERING 100kΩ BUFFER VOLTAGE FOLLOWER 50kΩ C5 15 - 115pF 2-1N914 CA3160, CA3160A NOTE: A square wave signal modulates the external sweeping input to produce 1Hz and 1MHz, showing the 1,000,000/1 frequency range of the Function Generator. NOTE: The bottom trace is the sweeping signal and the top trace is the actual generator output. The center trace displays the 1MHz signal via delayed oscilloscope triggering of the upper swept output signal. FIGURE 9B. TWO-TONE OUTPUT SIGNAL FROM THE FUNCTION GENERATOR FIGURE 9C. TRIPLE-TRACE OF THE FUNCTION GENERATOR SWEEPING TO 1MHz FIGURE 9. 1,000,000/1 SINGLE CONTROL FUNCTION GENERATOR: 1Hz to 1MHz 5.1kΩ +15V 1N914 470pF STAIRCASE OUTPUT +15V 100 kΩ 100 kΩ 1MΩ 3 100 kΩ + 2 15 - 115pF FREQ ADJUST - 7 7 6 CA3130 2 3 4 - 10kΩ CA3160 1N914 8 +15V +15V STEP HEIGHT ADJUST 4 - 60pF 8.2kΩ 7 6 + CHARGE COMMUTATING NETWORK 1.5 MΩ + 6 CA3130 2kΩ 4 MULTIVIBRATOR 3 +15V 2 - 8 4 INTEGRATOR HYSTERESIS SWITCH MULTIVIBRATOR RETRACE INHIBIT +15mV TO +10V 51kΩ 100kΩ FIGURE 10A. STAIRCASE GENERATOR CIRCUIT Staircase Generator Figure 10 shows a staircase generator circuit utilizing three CMOS operational amplifiers. Two CA3130s are used; one as a multivibrator, the other as a hysteresis switch. The third amplifier, a CA3160, is used as a linear staircase generator. Picoammeter Circuit Figure 11 is a current-to-voltage converter configuration utilizing a CA3160 and CA3140 to provide a picoampere 12 meter for 13pA full scale meter deflection. By placing Terminals 2 and 4 of the CA3160 at ground potential, the CA3160 input is operated in the “guarded mode”. Under this operating condition, even slight leakage resistance present between Terminals 3 and 2 or between Terminals 3 and 4 would result in zero voltage across this leakage resistance, thus substantially reducing the leakage current. If the CA3160 is operated with the same voltage on input Terminals 3 and 2 as on Terminal 4, a further reduction in the CA3160, CA3160A input current to the less than one picoampere level can be achieved as shown in Figure 23. To further enhance the stability of this circuit, the CA3160 can be operated with its output (Terminal 6) near ground, thus markedly reducing the dissipation by reducing the supply current to the device. STAIRCASE OUTPUT 2V STEPS The CA3140 stage serves as a X100 gain stage to provide the required plus and minus output swing for the meter and feedback network. A 100-to-1 voltage divider network consisting of a 9.9kΩ resistor in series with a 100Ω resistor sets the voltage at the 10GΩ resistor (in series with Terminal 3) to ±30mV full-scale deflection. This 30mV signal results from ±3V appearing at the top of the voltage divider network which also drives the meter circuitry. COMPARATOR OSCILLATOR Top Trace: Staircase Output 2V Steps Center Trace: Comparator Bottom Trace: Oscillator By utilizing a switching technique in the meter circuit and in the 9.9kΩ and 100Ω network similar to that used in voltmeter circuit shown in Figure 8, a current range of 3pA to 1nA full scale can be handled with the single 10GΩ resistor. FIGURE 10B. STAIRCASE GENERATOR WAVEFORM FIGURE 10. STAIRCASE GENERATOR CIRCUIT 10GΩ +15V 1MΩ 0.1µF 10pF +15V 7 10MΩ 3 + 7 CA3160 2 6 - 2 10kΩ - CA3140 4 5 6 + 3 5.6kΩ 9.9kΩ 1 4 560kΩ 100kΩ 0.1µF 500Ω 9.1kΩ 100Ω -15V M 500-0-500µA -15V FIGURE 11. CURRENT-TO-VOLTAGE CONVERTER TO PROVIDE A PICOAMMETER WITH ±3pA FULL SCALE DEFLECTION 100kΩ +15V +15V 2200pF 30pF +15V 0.1µF 0.1µF 7 1MΩ 39kΩ 3 + 0.1µF CA3160 - 2 6 2 1N914 8 4 5 7 3 + 0.1µF 2 6 100kΩ 4 100kΩ CA3140 1MΩ 3 27kΩ 4 DROOP ZERO ADJUST 0.1µF 39kΩ 500µA STROBE INPUT SAMPLE = 15V HOLD = 0V 2kΩ FIGURE 12A. SINGLE SUPPLY SAMPLE AND HOLD SYSTEM, INPUT 0V TO 10V 13 6 + 5 8.2kΩ 9.1kΩ 7 CA3080A 1 OFFSET VOLTAGE ADJUST 8.2Ω CA3160, CA3160A SAMPLED OUTPUT SAMPLED OUTPUT 0V- INPUT SIGNAL INPUT 0V- SAMPLING PULSES SAMPLING PULSES Top Trace: Sampled Output Center Trace: Input Signal Bottom Trace: Sampling Pulses Top Trace: Sampled Output Center Trace: Input Signal Bottom Trace: Sampling Pulses FIGURE 12B. SAMPLE AND HOLD WAVEFORM FIGURE 12C. SAMPLE AND HOLD WAVEFORM FIGURE 12. SINGLE SUPPLY SAMPLE AND HOLD SYSTEM, INPUT 0V TO 10V Single Supply Sample-and-Hold System Figure 12 shows a single supply sample-and-hold system using a CA3160 to provide a high input impedance and an input voltage range of 0V to 10V. The output from the input buffer integrator network is coupled to a CA3080A. The CA3080A functions as a strobeable current source for the CA3140 output integrator and storage capacitor. The CA3140 was chosen because of its low output impedance and constant gain-bandwidth product. Pulse “droop” during the hold interval can be reduced to zero by adjusting the 100kΩ bias-voltage potentiometer on the positive input of the CA3140. This zero adjustment sets the CA3080A output voltage at its zero current position. In this sample-and-hold circuit it is essential that the amplifier bias current be reduced to zero to minimize output signal current during the hold mode. Even with 320mV at the amplifier bias circuit terminal (5) at least 1100pA of output current will be available. Wien Bridge Oscillator A simple, single supply Wien Bridge oscillator using a CA3160 is shown in Figure 13. A pair of parallel-connected 1N914 diodes comprise the gain-setting network which standardizes the output voltage at approximately 1.1V. The 500Ω potentiometer is adjusted so that the oscillator will always start and the oscillation will be maintained. Increasing the amplitude of the voltage may lower the threshold level for starting and for sustaining the oscillation, but will introduce more distortion. 14 R1 100kΩ +15V R3 51kΩ C2 51pF 7 3 R2 100kΩ OUTPUT f = 100kHz 2% THD AT 1.1VP-P + CA3160 2 0.1 µF 6 4 2kΩ C1 10-80 pF 2-1N914 0.01µF 680Ω f= 1 2 π √(R1 || R2) C1 R3 C2 500Ω FIGURE 13. SINGLE SUPPLY WEIN BRIDGE OSCILLATOR Operation with Output Stage Power Booster The current sourcing and sinking capability of the CA3160 output stage is easily supplemented to provide power-boost capability. In the circuit of Figure 14, three CMOS transistorpairs in a single CA3600 lC array are shown parallel-connected with the output stage in the CA3160. In the Class A mode of CA3600E shown, a typical device consumes 20mA of supply current at 15V operation. This arrangement boosts the currenthandling capability of the CA3160 output stage by about 2.5X. The amplifier circuit in Figure 14 employs feedback to establish a closed-loop gain of 20dB. The typical largesignal-bandwidth (-3dB) is 190kHz. CA3160, CA3160A +15V 2 14 0.01µF 11 1MΩ - 1µF CA3600 + QP1 QP2 QP3 7 680kΩ 3 + CA3160 INPUT 2 1µF 6 13 1 3 10 - 2kΩ 500µF 8 6 12 50Ω 100mW AT 10% THD 4 8 A = 20dB LARGE SIGNAL BW (-3dB) = 190kHz 5 QN1 QN2 7 QN3 4 9 20kΩ FIGURE 14. CMOS TRANSISTOR ARRAY (CA3600E) CONNECTED AS POWER BOOSTER IN THE OUTPUT STAGE OF THE CA3160 VS = ±7.5V TA = 25oC 100 0 50 100 φ OL 80 150 200 60 40 CL = 30pF RL = 2kΩ 20 0 101 102 103 104 105 106 FREQUENCY (Hz) 107 108 FIGURE 15. OPEN LOOP VOLTAGE GAIN AND PHASE SHIFT vs FREQUENCY 15 150 RL = 2kΩ OPEN LOOP VOLTAGE GAIN (dB) OPEN LOOP VOLTAGE GAIN (dB) 120 OPEN LOOP PHASE (DEGREES) Typical Performance Curves 140 130 120 110 100 90 80 -100 -50 0 50 100 TEMPERATURE (oC) FIGURE 16. OPEN LOOP GAIN vs TEMPERATURE CA3160, CA3160A (Continued) 15.0 17.5 15 QUIESCENT SUPPLY CURRENT (mA) V+ = 15V, V- = 0V TA = 25oC RL = 5kΩ 2kΩ 1kΩ 500Ω 12.5 10 7.5 5 2.5 10.0 5.0 2.5 5 7.5 10 12.5 15 17.5 20 GATE VOLTAGE [TERMINALS 4 AND 8] (V) 2.5 14 TA = -55oC 25oC 125oC 6 4 2 0 0 2 4 6 8 10 12 14 POSITIVE SUPPLY VOLTAGE (V) 16 10 12 14 18 FIGURE 19. QUIESCENT SUPPLY CURRENT vs SUPPLY VOLTAGE 50 10 V- = 0V TA = 25oC 18 V+ = 15V 10V 5V 1 0.1 0.01 0.001 0.001 0.01 0.1 1 10 MAGNITUDE OF LOAD CURRENT (mA) 100 FIGURE 20. VOLTAGE ACROSS PMOS OUTPUT TRANSISTOR (Q8) vs LOAD CURRENT 1000 50 V- = 0V TA = 25oC 16 FIGURE 18. QUIESCENT SUPPLY CURRENT vs SUPPLY VOLTAGE 10 8 8 POSITIVE SUPPLY VOLTAGE (V) VO = V+ / 2 V- = 0 12 6 22.5 FIGURE 17. VOLTAGE TRANSFER CHARACTERISTICS OF CMOS OUTPUT STAGE QUIESCENT SUPPLY CURRENT (mA) HIGH VO = V+ OR LOW VO = V- 0 0 TA = 25oC VS = ±7.5V V+ = 15V 10V 5V 100 1 EN (nV/√Hz) VOLTAGE DROP ACROSS NMOS OUTPUT STAGE TRANSISTOR (Q12) (V) BALANCED VO = V+/2 7.5 0 10 TA = 25oC RL = ∞ V- = 0 12.5 VOLTAGE DROP ACROSS PMOS OUTPUT STAGE TRANSISTOR (Q8) (V) OUTPUT VOLTAGE [TERMS. 4 AND 6] (V) Typical Performance Curves 0.1 10 0.01 0.001 0.001 1 0.01 0.1 1 10 100 MAGNITUDE OF LOAD CURRENT (mA) FIGURE 21. VOLTAGE ACROSS NMOS OUTPUT TRANSISTOR (Q12) vs LOAD CURRENT 16 1 101 102 103 FREQUENCY (Hz) 104 105 FIGURE 22. EQUIVALENT NOISE VOLTAGE vs FREQUENCY CA3160, CA3160A Typical Performance Curves 10 (Continued) 4000 TA = 25oC VS = ±7.5V 1000 5 INPUT CURRENT (pA) INPUT VOLTAGE (V) 7.5 V+ 15V TO 5V 7 2 CA3160 PA 6 3 2.5 0V TO V- -10V 0 -1 0 1 2 3 4 5 6 INPUT CURRENT (pA) 10 8 4 VIN 100 1 -80 -60 -40 -20 7 FIGURE 23. INPUT CURRENT vs COMMON MODE VOLTAGE 0 20 40 60 80 TEMPERATURE (oC) 100 120 140 FIGURE 24. INPUT CURRENT vs TEMPERATURE OFFSET VOLTAGE SHIFT (mV) 7 DIFFERENTIAL DC VOLTAGE (ACROSS TERMINALS 2 AND 3) = 2V OUTPUT STAGE TOGGLED 6 5 TA = 125oC FOR METAL CAN PACKAGES 4 3 2 DIFFERENTIAL DC VOLTAGE (ACROSS TERMINALS 2 AND 3) = 0V OUTPUT VOLTAGE = V+ / 2 1 0 0 500 1000 1500 2000 2500 3000 3500 4000 TIME (HOURS) FIGURE 25. TYPICAL INCREMENTAL OFFSET VOLTAGE SHIFT vs OPERATING LIFE All Intersil semiconductor products are manufactured, assembled and tested under ISO9000 quality systems certification. Intersil semiconductor products are sold by description only. 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