V23N1 - APRIL

April 2013
I N
T H I S
I S S U E
high power controller
drives high power LEDs,
regulates solar cells, and
charges batteries 8
40µA IQ controller operates
from 3.5V–60V VIN,
maintains high efficiency at
light loads 12
quad step-down regulator
with 100% duty cycle
operation withstands 180V
surges 26
Volume 23 Number 1
Active Cell Balancer Extends
Run Time and Lifetime of Large
Series-Connected Battery Stacks
Jim Drew
Large stacks of series-connected battery cells are increasingly
used to power electric vehicles or store energy in wind and solar
power systems. It is not uncommon to have 100 cells connected in
series in an electric vehicle, and even more in energy storage units
for alternative energy systems. Typically, the stack is treated by the
charge-discharge system as a single battery—cells are charged
and discharged as a series stack and the state of charge (SoC) of
each cell depends on its ability to store
and maintain charge. Treating the cell
stack as a single battery composed
of capacity-matched cells can work
well in the short term, but becomes
increasingly inefficient in the long run.
When a battery stack is first constructed, the capacities of its component cells can be well matched, but over
time, individual cells lose capacity at different rates due
to temperature variations and other factors. In a straightforward stack charge-discharge implementation, the cell
with the least capacity—the weakest cell—effectively
limits the run time of the stack. When the stack is charged,
the weakest cell reaches its full charge voltage before
stronger cells, so stronger cells are not charged to capacity. Likewise, when the stack is discharged, the weakest
cell reaches its cutoff voltage sooner, limiting run time.
The LTC®3300-1 balances the states of charge of individual cells in large battery stacks,
increasing
Caption capacity, extending run time and prolonging lifetime of the stack
w w w. li n ea r.com
(continued on page 4)
Linear in the News
In this issue...
COVER STORY
Active Cell Balancer Extends
Run Time and Lifetime of Large
Series-Connected Battery Stacks
Jim Drew
BREAKING GROUND WITH NEW VIDEOS
1
DESIGN FEATURES
High Power Controller Drives High Power LEDs,
Regulates Solar Cells, and Charges Batteries,
Steps Down 60V Inputs
Luke Milner
8
40µA IQ, P-Channel Step-Down Controller
Operates from 60V to 3.5V VIN and Maintains
High Efficiency at Light Loads
Terry Groom
Linear videos fall into three basic categories:
12
15
20
DESIGN IDEAS
What’s New with LTspice IV?
Gabino Alonso
24
Compact Quad Step-Down Regulator with 100%
Duty Cycle Operation Withstands 180V Surges
Jonathan Paolucci
High Efficiency Bidirectional Cell Balancer Maximizes Capacity and Lifetime of Series-Connected
Battery Stacks—Large, series-connected strings of batteries are commonly used
4A Li-Ion Battery Charger Accepts Inputs to 32V
Rick Brewster
Video Design Ideas
These five to eight minute videos cover significant design challenges and solutions.
Following is an example of a recent video topic, one of over 65 produced to date.
Dual, Fast, Step-Down Controller’s External Reference
Input Enables Dynamic Voltage Scaling from 0.4V to
5.5V and 0.3% Total Combined Regulation Accuracy
Shuo Chen and Terry Groom
Linear’s library of video design ideas at www.linear.com has grown to more
than 120 presentations by experienced designers and applications engineers, discussing challenging analog design problems and their solutions. The
videos cover a broad range of topics—from new packaging that improves
the stability of voltage references to energy harvesting applications. The
videos provide focused solutions for power management, data conversion,
signal conditioning, µModule® integration, and wireless sensor networks.
26
µModule Regulator Charges Supercapacitor
Backup Supply, Supporting LDO Outputs
When the Input Supply Fails
Andy Radosevich
28
new product briefs
30
back page circuits
32
in electric vehicles, backup power systems and a wide variety of energy storage
applications. Maximizing the lifetime and ensuring safe usage of such battery stacks requires accurate measurement and balancing of each cells’ state
of charge (SoC). Cell aging occurs in all cells and at different rates due to the
same factors that cause SoC mismatch. Without capacity compensation, the
run time of the battery is limited by the lowest-capacity cell in the stack.
Active balancers such as the LTC3300 have the ability to correct for SoC imbalance
and to compensate for cell-to-cell capacity differences. By efficiently redistributing
charge from mismatched cells, the LTC3300 maximizes the usable capacity of the
battery stack. The LTC3300 provides high current, high efficiency bidirectional cell
balancing for series-connected batteries. Each IC can simultaneously charge or discharge up to six series cells. There is no limit to the height of the stack. Balancerto-balancer communication is achieved through a high noise margin SPI bus, and
Mark Vitunic discusses how to
improve battery stack run time and
lifetime with active cell balancing
2 | April 2013 : LT Journal of Analog Innovation
Linear in the news
numerous safety features ensure reliable,
efficient, high current active balancing.
View at video.linear.com/p4691-146.
Video TechClips
These are brief videos of one to two
minutes, showing a specific aspect of
a design solution. For example, video
TechClips may demonstrate the thermal
properties of a package, the efficiency
of a design solution, or the short circuit protection built into a part. One
example of a video TechClip:
Demonstration of the LTM®4641 µModule regulator’s overcurrent protection capability—The
LTM4641 includes output overcurrent
protection and thermal shutdown.
Once the fault condition has cleared,
the LTM4641 automatically resumes
operation. The output voltage stays
well controlled in all circumstances.
View at video.linear.com/p4656-144.
Video TechClip
demonstration of the
LTM4641 µModule
regulator’s overcurrent
protection capability
Dobkin and focused interviews on such
topics as wireless sensor networks.
In conjunction with publication of the
latest Analog Circuit Design book,
Volume 2, An Immersion in the Black
Art of Analog Design, Bob Dobkin has
completed two video interviews—one
focused on the challenge of analog design
and the other a reminiscence on the late
staff scientist and writer Jim Williams,
who co-edited the book series with Bob
Dobkin. You can view his discussion of
the challenge of analog design at www.
linear.com/designtools/acd_book.php.
LINEAR PRODUCT AWARDS
Several Linear products have recently
received significant industry awards:
Video Interviews
Best of Electronic Design
Interviews cover a broad range of
topics, from interviews with Linear
Executive Chairman Bob Swanson and
CEO Lothar Maier regarding company
direction, to analog technology discussions with Chief Technical Officer Bob
The LTC5800 SmartMesh® system-onchip wireless sensor network solution
received Electronic Design’s 2012 Best
of Wireless Design Award. In their
article on the product award, the editor
stated that the SmartMesh LTC5800 and
LTP5800 product families from Linear
Visit www.linear.com to see
interviews with Bob Dobkin
and other Linear luminaries
Technology’s Dust Networks® product
group “let you build complete wireless
mesh networks for industrial applications. They’re available to implement
WirelessHART and IPv6 networks.”
Electronic Products Product of the Year
In its annual product of the year awards,
Electronic Products magazine selected
Linear’s LT4275 LTPoE++™ powered device
controller as a Product of the Year. In their
award article in the January 2013 issue,
the magazine’s editors stated, “The LT4275
powered-device (PD) interface controllers
are LTPoE++, PoE+, and PoE-compliant and
target applications requiring up to 90W.
The existing PoE+ standard limits the maximum PD power delivery to 25.5W, which
is insufficient for picocells, base stations,
signage, and heated outdoor cameras. The
device expands the power budget to 38.7,
52.7, 70, and 90W power levels to accommodate these applications. They deliver
power to PD loads using just one IC.”
CONFERENCES & EVENTS
Sensors Expo & Conference, Donald E. Stephens
Convention Center, Rosemont, Illinois, June 4–6,
2013, Booth 1020—Linear will showcase
its line of energy harvesting products,
as well as its Dust Networks’ wireless sensor network products. Sam
Nork will make a presentation: “Use
Energy Harvesting to Extend Battery
Life in Wireless Sensor Applications”
at 11:15 am on June 4. More info at
www.sensorsmag.com/sensors-expo. n
April 2013 : LT Journal of Analog Innovation | 3
The LTC3300-1 is a fault-protected controller IC for transformer-based bidirectional
active balancing of multicell battery stacks. Active bidirectional balancing can
transfer charge from the stack to low SoC cells, or transfer charge from high SoC
cells to the stack. In this way, the overall capacity of the stack is improved.
(LTC3300-1 continued from page 1)
The capacity of the stack and its run
time can be improved by balancing the
state of charge between cells within the
stack. Figure 1 shows a simplified schematic of a 12-cell balancer using two
LTC3300-1 cell balancing controllers.
LTC3300-1 IMPROVES BATTERY
STACK RUN TIMES AND LIFETIMES
The LTC3300-1 is a fault-protected controller IC for transformer-based bidirectional
active balancing of multicell battery
stacks. Active bidirectional balancing can
transfer charge from the stack to low
SoC cells, or transfer charge from high
SoC cells to the stack. In this way, the
overall capacity of the stack is improved.
A single LTC3300-1 can balance up to
six series connected cells with a common mode voltage range of up to 36V.
Multiple LTC3300-1 devices can be connected in series, allowing balancing of
long strings of series connected cells.
A unique level shifting SPI-compatible
serial interface allows multiple LTC3300-1
devices to be connected in series without opto-couplers or isolators.
As the stack is charged, weaker cells operate in discharge mode and stronger cells
operate in the charge mode until all cells
reach their full SOC. Likewise, during discharge, weaker cells are operated in charge
mode while stronger cells are operated in
discharge mode until all cells reach their
cutoff voltage. This extends the run time
of the stack, which reduces the number of
charge/discharge cycles and thus extends
the life of the batteries within the stack.
4 | April 2013 : LT Journal of Analog Innovation
With the LTC3300-1, all individual cell balancers can operate simultaneously in any
combination of discharge or charge modes,
even when multiple LTC3300-1 devices are
used. For instance, for a stack of 12 cells,
with two LTC3300-1 devices connected in
series, charge can be transferred from cell
12 to cell 1 in a single time step by discharging cell 12 and charging cell 1. When
compared to other methods of transferring
charge between cells, this single time step
method is the fastest and most efficient.
A single time step can include multiple
balancers in discharge or charge modes
resulting in optimum balance time.
The LTC3300-1 is available in a 48-lead
7mm × 7mm QFN or LQFP package.
HOW TO APPLY THE LTC3300-1
The cell balancer incorporates a boundary
mode synchronous flyback transformer
power stage that is controlled by the
LTC3300-1. There are six sets of control
signals within the LTC3300-1 that control
the gates of the primary side and secondary side NMOS switches and current sense
inputs for each pair of NMOS switches.
The naming convention used for the
LTC3300-1 is that the transformer primary is connected across the battery cell
and the secondary of the transformer is
across the ground reference of the IC to a
point six or more cells up the stack. The
primary side gate signals are referenced
to the next lower cell while the secondary
side gates are referenced to the ground
reference of the IC, the V– exposed pad.
The LTC3300-1 includes fault protection,
including read-back capability, cyclic
redundancy check (CRC) communication error detection, maximum on-time
volt-second clamps and cell or transformer secondary overvoltage shutdown.
CHARGE
SUPPLY
(ICHARGE 1-6)
+
CHARGE
RETURN
(IDISCHARGE 1-6)
LTC3300-1
3
+
•
CHARGE
RETURN
CELL 12
IDISCHARGE
+
CELL 7
CELL 6
•
3
LTC3300-1
•
CHARGE
SUPPLY
Figure 1. Simplified
schematic of how the
LTC3300-1 actively
balances individual cells
in a 12-cell battery stack
ICHARGE
•
+
CELL 1
3
SERIAL
DATA IN
FROM
SYSTEM
CONTROLLER
design features
With the LTC3300-1, all individual cell balancers can operate simultaneously, in
any combination of discharge or charge modes, even when multiple LTC3300‑1
devices are used. For instance for a stack of 12 cells, with two LTC33001 devices connected in series, charge can be transferred from cell 12 to
cell 1 in a single time step by discharging cell 12 and charging cell 1.
During discharge mode (Figure 2) the
primary side NMOS is turned on first and
remains on until the current signal ramps
up to 50mV or the primary max on-time
setting is reached. The flux built up in the
primary side of the flyback transformer
is then transferred to the secondary. The
secondary gate signal turns on the secondary side NMOS, and it remains on until
the secondary current sense signal ramps
down to 0mV or the secondary side max
on-time is reached. The cycle repeats until
the LTC3300-1 is given a command to
stop the discharge mode or encounters a
fault such as a watchdog timer timeout,
a cell undervoltage (2.0V), a cell overvoltage (5.0V) or a transformer secondary
overvoltage caused by a lost connection.
During charge mode (Figure 3) the
secondary is turned on first and remains
of the flyback transformer, the number of
cells within the secondary stack (S) and the
transfer efficiency (η) of the power stage.
on until the secondary current signal
ramps up to 50mV or the secondary max
on-time setting is reached. The flux built
up in the secondary side of the flyback
transformer is then transferred to the
primary. The primary gate signal turns
on the primary side NMOS and it remains
on until the current sense signal ramps
down to 0mV or the primary side max
on-time is reached. The cycle repeats
until the LTC3300-1 is given a command
to stop the charge mode or encounters a
fault such as a watchdog timer timeout,
a cell undervoltage (2.0V), a cell overvoltage (5.0V), or a transformer secondary
overvoltage caused by a lost connection.
RSENSE(PRI) =
50mV
S
•
2 • IDISCHARGE S + T
RSENSE(PRI) =
50mV
S•T
•
ηCHARGE
2 • ICHARGE S + T
The turns ratio of the flyback transformer is selected based on the number
of cells across the secondary winding
and the maximum reflected voltage on
the primary side and secondary side
NMOS switches. For a 12-cell secondary, a
1:2 turns ratio from primary to secondary
provides a good balance between transfer
efficiency and voltage stress on the two
NMOS switches. For a larger number of
cells across the secondary, a higher turns
ratio can be selected and still provide
The average balancing currents are determined by the value of the current sensing
resistors (RSENSE(PRI) and RSENSE(SEC)), the
turns ratio (1:T) from primary to secondary
ICELL
ICELL
ICELL
ICELL
•
•
•
t
956ns
G1S
G1P
ISTACK
LPRI
VSECONDARY
•
VPRIMARY
t
VSECONDARY
956ns
G1S
G1P
t
5.7µs
ISTACK
ISTACK
LPRI
VPRIMARY
t
5.7µs
ISTACK
25.2V
25.2V
I1P
RSENSE(PRI)
25mΩ
I1S
RSENS(SEC)
25mΩ
I1S
I1P
VQ1A(DS)
0V
t
RSENSE(PRI)
25mΩ
VQ1A(DS)
RSENS(SEC)
25mΩ
0V
t
50.4V
50.4V
VQ1B(DS)
VQ1B(DS)
0V
0V
t
t
Figure 2. Discharge mode of a single cell in the stack
Figure 3. Charge mode of a single cell in the stack
April 2013 : LT Journal of Analog Innovation | 5
100
I1S
50mV/DIV
I1P
50mV/DIV
CHARGE TRANSFER EFFICIENCY (%)
I1P
50mV/DIV
I1S
50mV/DIV
PRIMARY
DRAIN
50V/DIV
SECONDARY
DRAIN
50V/DIV
SECONDARY
DRAIN
50V/DIV
PRIMARY
DRAIN
50V/DIV
2µs/DIV
DC2064A DEMO BOARD
ICHARGE = 2.5A
T=2
S = 12
2µs/DIV
DC2064A DEMO BOARD
IDISCHARGE = 2.5A
T=2
S = 12
DC2064A DEMO BOARD
ICHARGE = IDISCHARGE = 2.5A
VCELL = 3.6V
95
CHARGE
DISCHARGE
90
85
80
6
8
10
12
NUMBER OF CELLS (SECONDARY SIDE)
Figure 4. Demonstration circuit DC2064A typical
charge mode waveforms for a 2.5A balance current
Figure 5. Demonstration circuit DC2064A typical
discharge mode waveforms for a 2.5A balance current
Figure 6. Cell balancer efficiency verses the number
of cells across the transformer secondary winding
high transfer efficiency and manageable
voltage stress on the NMOS switches.
shift in cell voltage results in a 10%
shift in the operating frequency.
Once the current sensing resistors and
transformer turns ratio are defined, the
primary inductance of the flyback transformer is determined. To do so, the operation frequency needs to be defined. The
operating frequency is a function of the
cell voltage, the current sensing resistor,
the inductance of the primary, the number
of cells within the stack, and the turns
ratio of the transformer. The operating
frequency is generally set to approximately
150k Hz to reduce interference with other
circuitry that may be in the system and to
yield reasonable circuit component sizes
with high transfer efficiency. The nominal
cell voltage is used in this calculation.
Selection of the NMOS switches is determined by the peak balancing current and
the drain-to-source off-state voltage. The
drain-to-source off-state voltage can be
estimated using the following expressions:
sourced from the boost circuitry, which
gets its energy from C6. All six secondary gate drivers are sourced from the
VREG circuitry. When all six balancers
are operating, the secondary gate drivers present a load current on VREG of:
L PRI =
VCELL • RSENSE(PRI)
S
•
S + T fDISCHARGE • 50mV
L PRI =
VCELL • RSENSE(SEC)
S
•
S + T fCHARGE • 50mV • T
In most designs the average charge
and discharge currents are set
to be equal, which necessitates
RSENSE(SEC) = RSENSE(PRI) • T
As a result, the charge and discharge
frequencies are equal. Note that
the frequency of operation is a linear function of the cell voltage: 10%
6 | April 2013 : LT Journal of Analog Innovation
 S V
VDS(PRI)MIN > VCELL • 1+  + DIODE
T
 T
VDS(SEC)MIN > VCELL • (S + T ) + T • VDIODE
Good design practice recommends that the
MOSFET breakdown rating be 20% higher
than this minimum calculated value to
account for voltage spikes due to leakage
inductance ringing. Some applications may
require a series resistor capacitor snubber
in parallel with the drain and source of the
NMOS switch to reduce the ringing. These
snubber circuits may lower the transfer efficiency but keep the NMOS devices
within their safe operating region.
Additional NMOS parameters that need to
be considered are the total gate charge
(QG) and RDS(ON). The product of total
gate charge and the operating frequency
determines the gate current requirements for the primary and secondary gate
drivers. The primary gate drive for cells
1–5 is sourced from the cell above the
selected cell. Cell 6 primary gate drive is
IV(REG) = 6 • Q G • f
resulting in a power dissipation of:
PV(REG) = ( VC6 – VREG ) • IV(REG)
The primary gate drivers generate
power dissipation in the LTC3300-1 of
PPRI(DRIVE) = 2 • VCELL • 6 • Q G • f
The individual primary and secondary gate drive currents should
be limited to less than 4m A.
Figure 4 shows typical charge mode
waveforms for a 2.5A cell balancer with
a secondary of 12 cells and a transformer
turns ratio of 1:2. The primary inductance is 3µ H, RSENSE(PRI) is 8mΩ, RSENSE(SEC)
is 16mΩ and the cell voltage is 3.6V.
Figure 5 shows the same cell balancer in
discharge mode. Figure 6 shows the cell
balancer efficiency for various numbers
of cells connected to the secondary.
design features
INTERLEAVING SECONDARIES IN AN
18-CELL CONFIGURATION
Large strings of cells can be accomodated
by the LTC3300-1 by interleaving their
secondary windings. Figure 7 shows an
18-cell stack with three LTC3300-1 ICs
connected in series via the SPI-compatible
serial interface. The transformer secondaries of the bottom LTC3300-1 are
connected across (cell 1)– and (cell
12)+ while secondaries of the middle
LTC3300-1 are connected across (cell
6)+ and (cell 18)+. The secondaries of
the top LTC3300-1 are connected across
six cells, (cell 12)+ and (cell 18)+.
The lower two devices have their
BOOST and TOS pins tied to their respective V– pin and BOOST+ pins connected to
the cell above the cell connected to their
respective C6 pins. The top LTC3300-1 has
its BOOST and TOS pins tied to the VREG pin.
A flying capacitor is connected between
the BOOST– and BOOST+ pins along
with a series 6.8Ω resistor and diode
connected from the BOOST+ pin to
cell 6. The VMODE pin of the bottom
LTC3300-1 is tied to its VREG pin while
all other devices have their VMODE pins
tied to their respective V– pins.
0.1µF
6.8Ω
BOOST– BOOST+ C6
C1
CELL 18
•1:1
10µF
10µH
10µH
•
LTC3300-1
G1P
+
I1P
CELL 13
25mΩ
G1S
I1S
25mΩ
VREG
BOOST
V–
BOOST+
C6
TO TRANSFORMER
SECONDARIES OF
BALANCERS 8 TO 12
C1
+
CELL 12
•1:1
10µF
10µH
10µH
•
LTC3300-1
G1P
+
I1P
CELL 7
25mΩ
G1S
I1S
25mΩ
BOOST
V–
BOOST+
C6
TO TRANSFORMER
SECONDARIES OF
BALANCERS 2 TO 6
C1
CONCLUSION
The LTC3300-1 actively balances the state
of charge of individual cells in multicell,
series-connected battery stacks using a
transformer-based bidirectional scheme.
Active balancing extends the run time of
battery stacks, which in turn extends their
lifetimes. The LTC3300-1 integrates gate
drive circuitry and a robust serial interface
with built in watchdog timer, undervoltage
and overvoltage protection in a 48-lead
QFN or LQFP package. Each LTC3300-1
controls up to six cell balancers while
larger stacks can be accommodated with
multiple LTC3300-1 ICs connected in series
using an SPI-compatible serial interface.
+
TO TRANSFORMER
SECONDARIES OF
BALANCERS 14 TO 18
+
CELL 6
•1:1
10µF
10µH
10µH
•
LTC3300-1
G1P
+
I1P
CELL 1
25mΩ
G1S
I1S
25mΩ
BOOST
V–
33001 F05
Figure 7. 18-cell active balancer
Visit www.linear.com/LTC3300-1
for data sheets, demo boards and
other applications information. n
April 2013 : LT Journal of Analog Innovation | 7
High Power Controller Drives High Power LEDs,
Regulates Solar Cells, and Charges Batteries,
Steps Down 60V Inputs
Luke Milner
The best LED drivers accurately regulate LED current for consistent color reproduction
and modulate it rapidly for high contrast dimming. They also recognize and
survive short and open circuits, monitor and report current levels, guard against
overheating, and protect weak power supplies from excessive load currents. A
standard switching converter would require a number of additional expensive
amplifiers, references and passive components to fulfill these responsibilities.
In contrast, the LT®3763 LED driver-controller has these functions built in—reducing BOM costs, saving board space and
improving reliability. The LT3763 is more
than just a high performance LED driver.
Its rich feature set simplifies the design of
other demanding applications, such as safe
charging of a sealed lead-acid batteries, or
maximum power point regulation for a
solar panel, or a combination of both. The
LT3763 performs these tasks with high efficiency, even at input voltages reaching 60V.
prevent overshoot and pulls down the
FAULT pin to mark the occasion.
The LT3763 is designed to provide flickerfree LED dimming as shown in Figure 2.
This is achieved by pulling PWMOUT low
whenever PWM is low and thereby disconnecting the LED, by similarly disconnecting the compensation network at VC , and
resynchronizing internal switching clocks
to the PWM pulse. These maneuvers ensure
that subsequent pulses are identical, that
DRIVING LEDs
Figure 1 shows the LT3763 configured as a
high power LED driver. A potentiometer at
the CTRL1 pin permits manual adjustment
of the regulated LED current from 0 to 20A.
For thermal regulation of the LED current,
a resistor with a negative temperature
coefficient is mounted near the LED and
connected from the CTRL2 pin to GND.
The resistor network at the EN/UVLO pin
programs the LT3763 to shut down if
the input voltage falls to less than 10V.
The resistor network at the FB pin
defines an open-circuit condition as
when the output reaches 6V, and should
that ever happen, the LT3763 automatically reduces the inductor current to
8 | April 2013 : LT Journal of Analog Innovation
the inductor current rises as fast as possible to satisfy the programmed LED current
level, and that the LED light never flickers.
The LT3763 can be configured as in
Figure 3 to deliver 350W with 98% efficiency from a 48V input. An internal
regulator supplies the drivers of the TG and
BG pins with enough power for each to
drive two of the external NMOS power
switches. Higher power applications can
be built by connecting LT3763s in parallel,
Figure 1. A single high power
LED (20A) driver with analog
and PWM dimming
RSENSE_IN
2.5mΩ
VIN
10V TO 30V
REN1
84.5k
RFILTA
1k
REN2
15.4k
CFILT
1µF
RFILTB
1k
RHOT
45.3k
IVINN
VIN
TG
CBOOST
220nF
VREF
LT3763
CTRL2
RNTC
470k
50k
LTspice IV
circuits.linear.com/620
IVINP
EN/UVLO
CREF
2.2µF
CIN2
100µF
CIN1
4.7µF
BOOST
M1
L1
1.5µH
SW
INTVCC
BG
CTRL1
RS
2.5mΩ
VOUT
6V, 20A MAXIMUM
COUT
220µF
×2
RFAULT
47.5kΩ
CVCC
22µF
D1
M2
RSA
10Ω
RSB
10Ω
GND
50Ω
1nF
50Ω
1nF
FBIN
SENSE+
IVINMON
SENSE–
PWMOUT
ISMON
M3
FAULT
PWM
SYNC
RT
RT
82.5k
CS
33nF
FB
SS
CSS
10nF
VC
RC
59k
CC
4.7nF
L1: COILCRAFT XAL1010-152
M1: RENESAS RJK0365
M2: RENESAS RJK0453
M3: IR IRFH6200
RS: VISHAY WSL25122L500FEA
RFB1
47.5k
RFB2
12.1k
design features
The output voltage can be as high as 1.5V less than input voltage, making
it possible to charge three sealed lead-acid batteries in series (up to 45V)
from a 48V supply with the simplicity of a standard buck converter.
small. Once the trickle charge phase is
complete, the charger should allow the
batteries’ voltages to decay to a relaxed
level before finally settling at and holding that final voltage indefinitely.
PWM
10V/DIV
VSW
50V/DIV
IL
5A/DIV
5µs/DIV
Figure 2. PWM dimming performance of the circuit
in Figure 1
so that current is shared equally between
the two controllers. This configuration
also illustrates how the SYNC pin can be
used to synchronize the parallel connected LT3763s to an external clock.
The high output voltage rating of
the LT3763 enables 35V at the output
with the simplicity of a standard buck
converter. The output voltage can be
as high as 1.5V less than input voltage, and the configuration in Figure 4
makes use of this feature to charge three
sealed lead-acid batteries in series
(up to 45V) from a 48V supply.
The combined current and voltage regulation loops on the LT3763, and its LED fault
handling circuitry, nearly make it a
complete battery charger. Only a single
additional transistor is required to form
a complete battery charging system.
The resistor divider at the FB pin has been
designed to program the charging voltage
to 45V. As in the case of an open-circuit,
when the voltage reaches 45V, the LT3763
automatically reduces the current to
prevent overshoot as shown in Figure 5.
When their combined voltage decays to
the newly programmed value, the LT3763
begins switching again and provides a sustaining current necessary to maintain the
output voltage indefinitely. As an added
Figure 3. 350W white LED driver
VIN
48V
REN1
374k
REN2
124k
IVINP
IVINN
VIN
EN/UVLO
TG
VREF
CREF
2.2µF
LT3763
CTRL2
SENSE+
SENSE–
IVINMON
3V
0V
400kHz
RT
200k
RFAULT
100k
CVCC
22µF
M2
×2
FB
CSS
10nF
VC
RC
5k
CC
5nF
VOUT
37V, 10A MAXIMUM
LED1
RFB1
931k
FAULT
SS
RS
5mΩ
CS
1nF
PWMOUT
ISMON
PWM
SYNC
RT
L1
6µH
COUT
10µF
×6
GND
FBIN
M1
×2
SW
BG
CTRL1
CBOOST
220nF
BOOST
INTVCC
INTVCC
CIN2
100µF
CIN1
4.7µF
CHARGING BATTERIES
The battery charger shown in Figure 4,
like all chargers, must be able to precisely
regulate the batteries’ rated charging
current (constant current mode) until the
battery voltages reach the limit set by their
chemistry. The charger must maintain that
voltage (constant voltage mode) without
overshoot until the current drawn by the
trickle-charging batteries becomes very
Subsequently, during trickle charging,
the battery draws less current over time.
When the charging current reduces to
ten percent of the regulated current (C/10
battery specification), the LT3763’s opencircuit fault condition is triggered. The
resulting high-to-low transition at the
FAULT pin is used to turn off the gate of
the added transistor M3 and remove the
resistor RFB3 from the feedback network.
The programmed output voltage is thereby
lowered, and the LT3763 stops switching to
allow the batteries to relax on their own.
LED1: LUMINUSPT-121
L1: COILTRONICS HC2-6R0
M1, M2: RENESAS RJK0851
RS: VISHAY WSL25125L000
RFB2
30.9k
April 2013 : LT Journal of Analog Innovation | 9
The LT3763 is a versatile step-down buck converter that integrates many
complex features essential for LED drivers, solar harvesters and battery
chargers. A PWM driver and current monitors are included with fault
detection, current limiting, input and output voltage regulation.
RSENSE_IN
15mΩ
VIN
48V
RFILTA
1k
CFILT
1µF
IVINP
ENABLE
CREF
2.2µF
RFILTB
1k
IVINN
CIN1
1µF
VIN
EN/UVLO
TG
VREF
LT3763
CTRL2
CBOOST
220nF
BOOST
BG
RFAULT
47.5kΩ
CVCC
22µF
M2
GND
FBIN
SYNC
RT
FB
SS
VC
RC
8.06k
CC
4.7nF
RSB
10Ω
12V
+
12V
+
RFB1
402k
FAULT
CSS
10nF
RSA
10Ω
+
CS
33nF
PWMOUT
ISMON
PWM
VOUT
45V, 3.3A MAXIMUM
RS
15mΩ
12V
SENSE–
IVINMON
RT
82.5k
L1
12µH
COUT
20µF
SENSE+
INTVCC
M1
SW
INTVCC
CTRL1
CIN2
47µF
L1: WÜRTH 74471112
M1, M2: INFINEON BSC100N06LS3
M3: VISHAY VN2222LL
RS: VISHAY WSL2512R0150
RFB2
12.1k
RFB3
178k
M3
Figure 4. 3.3A, six-cell (36V) SLA battery charger
benefit, the FAULT pin transition serves as a
signal that the trickle charging has begun.
A well-designed solar panel power supply requires an intelligent combination
of current and voltage regulation. In an
optimum design, a converter must sense
the voltage on the panel and adjust the
current it draws to maintain the input
voltage at the panel’s maximum power
point. If it draws too much current, the
voltage of the high impedance panel will
collapse. If it draws too little current,
available light energy is essentially wasted.
to sense the input voltage and adjust
the voltage on the current control pin.
The LT3763 includes this function at the
FBIN pin. Simply tie CTRL1 high, to the
2V reference available at VREF, and add a
voltage divider from VIN to FBIN. When the
voltage at FBIN falls to nearly 1.205V, the
internal amplifier automatically overrides
the CTRL1 voltage and reduces the load
current. This regulates the input voltage (the voltage of the solar panel) at the
maximum power point for the panel. The
resistor divider on the FBIN pin is shown
in Figure 6 and can be customized to fit
the requirements of any solar panel.
In many common solutions, a solar panel
controller designer would use an amplifier
In the configuration shown in Figure 6, the
converter can generate whatever inductor
REGULATING SOLAR PANELS
10 | April 2013 : LT Journal of Analog Innovation
current, up to 5A, is required to hold
the panel voltage at 37V. Input voltage feedback is via the voltage divider
at the FBIN pin, which in turn regulates
the inductor current to what is actually necessary to hold the panel at peak
power in any given light condition.
As shown in Figure 7, the process of charging a battery with a solar panel looks very
similar to charging with a low impedance
supply as before. The difference is that
the regulated inductor current (charge
current) is not preset by the designer,
but is instead adjusted on the fly via the
feedback loop regulating input voltage.
This effectively minimizes charge time,
since input power is maximized at all
times, regardless of panel illumination.
Since the LT3763 has the capability of
regulating input voltage and current, as
well as output voltage and current, and
provides a fault flag with C/10, it can easily
be used with a wide variety of solar panels
to charge many different types of batteries.
Figure 5. 36V SLA battery charging cycle
FAULT
10V/DIV
IL
2A/DIV
VOUT
50mV/DIV
AC-COUPLED
50s/DIV
design features
Figure 6. 70W solar energy
harvester with maximum power
point regulation
PANEL VOLTAGE
UP TO 60V
37V VIN REG POINT
RSENSE_IN
10mΩ
D1
RFILTA
1k
D2
CFILT
1µF
IVINP
ENABLE
Dn
CREF
2.2µF
RFILTB
1k
IVINN
TG
VREF
LT3763
CTRL2
In each of the applications presented
here, the LT3763 provides an additional
service by monitoring the input and
output current levels. Voltages across
the IVINP and IVINN pins ranging from
0 to 50mV are amplified with a gain of
20, and the resulting voltage appears
at the IVINMON pin. The voltage at the
ISMON pin is an identical amplification of the voltage across the SENSE+
and SENSE– pins, as shown in Figure 8.
These signals are helpful in systems that
must verify the current provided to LEDs
or measure the efficiency of voltage conversion. They can also help to estimate
the power provided by a solar panel or
to monitor the current trickling into a
charging battery as it decays to zero.
Due to the discontinuous input current
of a step-down buck converter, a lowpass filter is typically necessary at the
IVINP and IVINN pins as shown in Figure 1
and Figure 4. A much smaller filter at the
SENSE+ and SENSE– pins may also be useful
Figure 7. Solar powered SLA battery charging
VREF
CTRL1
FBIN
SENSE–
IVINMON
INTVCC
+
RFAULT
47.5kΩ
CVCC
22µF
M2
RSA
10Ω
12V
RSB
10Ω
SYNC
RT
CS
33nF
PWMOUT
ISMON
PWM
RT
82.5k
VOUT
RS
10mΩ 14V MAXIMUM
GND
SENSE+
RFBIN2
12.1k
L1
12µH
SW
BG
RFBIN1
348k
circuits.linear.com/621
M1
CBOOST
100nF
BOOST
INTVCC
RNTC
470k
LTspice IV
CIN2
100µF
VIN
EN/UVLO
RHOT
45.3k
MONITORING CURRENT LEVELS
CIN1
4.7µF
RFB1
121k
FAULT
FB
SS
CSS
10nF
VC
RC
26.1k
CC
4.7nF
in filtering high frequency noise, but it
is not necessary. Even with these filters,
the monitors are fast enough to track
reasonably short PWM pulses as shown
in Figure 8. Nevertheless, if a designer
is more concerned with average current
levels than instantaneous current levels,
then additional lowpass filters can be easily added to the ISMON and IVINMON pins.
SUMMARY
The LT3763 is a versatile step-down buck
converter that integrates many complex
features essential for not only LED drivers,
L1: COILCRAFT MSS1278-123
M1, M2: INFINEON BSC100N06LS3
M3: VISHAY VN2222LL
RS: VISHAY WSL2512R0100FEA
RFB2
12.1k
RFB3
182k
M3
but solar harvesters and battery chargers as well. A PWM driver and current
monitors are included with fault detection,
current limiting, input and output voltage
regulation. Due to its high voltage rating, all of these features can be utilized to
illuminate long strings of LEDs or charge
stacks of batteries. Available in a 28-lead
TSSOP package, the LT3763 is a compact,
complete, and efficient power system.
Visit www.linear.com/LT3763 for
data sheets, demo boards and other
applications information. n
Figure 8. Current monitor outputs in an LED driver
application with PWM dimming
FAULT
10V/DIV
VIN
50mV/DIV
AC-COUPLED
IL
5A/DIV
IL
2A/DIV
ISMON
500mV/DIV
VOUT
50mV/DIV
IVINMON
200mV/DIV
50s/DIV
100µs/DIV
April 2013 : LT Journal of Analog Innovation | 11
40µA IQ, P-Channel Step-Down Controller Operates from
60V to 3.5V VIN and Maintains High Efficiency at Light Loads
Terry Groom
The LTC3864 is low IQ step-down DC/DC controller. It controls an external
P-channel MOSFET to provide excellent light load efficiency, wide input voltage
range (3.5V–60V) including low dropout operation, reliability and functional simplicity
in an easy-to-use 12-pin package. The LTC3864 is capable of 100% duty factor
operation, allowing continued operation with input supply voltage droop.
HIGH EFFICIENCY
PMOS CONTROLLER
The LTC3864 offers high efficiency at
full and light load by virtue of a strong
0.9Ω turn-on and 2Ω turn-off gate driver
and 40µ A low IQ Burst Mode® operation. Modern automotive always-on
applications often require less than
70µ A total supply current to prevent
battery drain. Burst Mode switching
and a low IQ of only 40µ A allows high
efficiency at these very low currents.
Figure 1 shows a typical efficiency graph,
showing very little decline as the load
current is reduced. Light load efficiency
is achieved in two ways: first by low
frequency Burst Mode switching, and
second by low VIN IQ. In light load Burst
Mode operation, the load current is
supported by multiple switching pulses
generated in a “burst” of activity, with
periods of no switching in between bursts.
12 | April 2013 : LT Journal of Analog Innovation
lockout condition is actually set by the
differential voltage from the VIN pin to
the CAP pin of 3.5V. This voltage is used
to drive the gate of the power FET.
100
90
EFFICIENCY (%)
This combination of features makes it
ideal for automotive applications such
as always-on power in an electronic
control unit (ECU). Low dropout performance is guaranteed down to 3.5V over
the full operating temperature range.
The LTC3864 is offered in automotive
temperature and reliability grade and
has been verified to strict failure mode
and effects analysis, or (FMEA) criteria.
Burst Mode
OPERATION
80
PULSE-SKIPPING
70
60
50
0.01
VIN = 12V
VOUT = 5V
0.1
LOAD CURRENT (A)
1
Figure 1. LTC3864 pulse-skip and Burst Mode
efficiency
This effectively lowers the switching
frequency. Power FET switching losses are
a significant loss component when loads
are light. Reducing the effective operating frequency reduces switching losses
and increases efficiency. The efficiency’s
lower limit is ultimately determined by
the VIN quiescent current, or IQ, of 40µ A,
which enables efficient standby operation in always-on power applications.
WIDE V IN OPERATING RANGE
The LTC3864 has a high voltage PMOS gate
driver capable of operating continuously
up to 60V and down to 3.5V. This input
voltage operating range is guaranteed over
the full temperature range up to a military
grade from –55ºC to 150ºC. The minimum
input voltage operation or undervoltage
The LTC3864’s internal linear regulator
maintains 8V between VIN and CAP. When
VIN is less than 8V, the VCAP regulator
is in dropout and the CAP pin is held at
ground. In this condition, the VIN undervoltage is set by the VIN -to-CAP undervoltage. The LTC3864 guarantees a
3.5V minimum from VIN to CAP to assure
adequate PMOS switch gate voltage. For
low VIN operating conditions, we recommend choosing an external P-channel
MOSFET that has a threshold voltage of
less than 2V to assure adequate overdrive when approaching minimum VIN .
100% DUTY CYCLE OPERATION
The LTC3864 naturally and easily handles
100% duty factor operation with an
external P-channel simply by forcing the
gate on. No boost drive or additional
circuitry is needed. While there is efficiency loss by using a P-channel at high
current as opposed to an N-channel,
the simplicity of the solution makes the
LTC3864 ideally suited for many low and
medium current level applications.
One important function in automotive applications is output voltage
design features
One important function in automotive applications is output
voltage dropout during a cold crank condition. With the
LTC3864, the output simply tracks the input voltage when it
is below the regulation output. The output quickly recovers
to the regulation once the cold crank condition is over.
VBATTERY
12V
VOLTAGE
Figure 2. Typical automotive cold
crank from 12V to below 5V
VOUT
5V
LTC3864’s 100% DUTY CYCLE CAPABILITY ALLOWS
VOUT TO RIDE VIN WITHOUT SIGNIFICANT DROPOUT
TIME
dropout during a cold crank condition. Figure 2 shows how the regulated
5V output drops out and recovers during a cold crank condition. The output
simply tracks the input voltage under
the output regulation voltage. The output quickly recovers to the regulated
5V once the cold crank condition is over.
SOFT-START, FAULT PROTECTION
AND RECOVERY
The LTC3864 includes soft-start, tracking,
fault protection and recovery features
to assure robust operation even under
extreme conditions. The SS pin provides
both soft-start and tracking features.
To set the soft-start ramp-up time,
simply tie a capacitor from the SS pin to
ground and the internal 10µ A charging
current sets the SS voltage ramp from 0
to 0.8V. At 0.8V on the SS pin the output is at the full regulation voltage.
The LTC3864 can track another input
source or supply by overdriving the
10µ A current and forcing the SS pin
input voltage. The output tracks the
SS pin until the signal exceeds 0.8V.
Fault protection features include power
good, undervoltage lockout, short-circuit
recovery and frequency foldback during
start-up and short-circuit conditions.
The LTC3864 includes an internal soft-start
ramping feature, which sets the maximum
output ramping rate under all operational
conditions including short-circuit recovery.
The internal soft-start ramp sets the minimum output voltage ramp time to approximately 650µs. An external capacitor to
the SS pin determines the SS ramp once the
internal minimum of 650µs is exceeded.
The internal soft-start ramp also determines the maximum output voltage
ramp from a short-circuit recovery.
Without this feature, the output recovery would be limited only by current
SHORTCIRCUIT
TRIGGER
SHORT-CIRCUIT REGION
VOUT
5V/DIV
SOFT RECOVERY
FROM SHORT
IL
2A/DIV
500µs/DIV
VIN = 12V, VOUT = 5V
limit. An output recovery rise without
soft-start leads to high transient current
and possible output voltage overshoot.
Figure 3 shows a short-circuit event
including recovery. When the output is
shorted, the output drops near zero and
the current is regulated to the programmed
short-circuit value. The first VOUT rise in
recovery is a result of the energy in the
inductor being transferred to the output once the short is removed. Next, the
internal regulation ramp prevents switching until the ramp exceeds the regulation
point, and then ramps monotonically
once switching begins. Figure 3 shows
a smooth output recovery from a shortcircuit without exceeding current limit
and without output voltage overshoot.
VERIFIED FAILURE MODE AND
EFFECTS ANALYSIS (FMEA)
The LTC3864 is designed to meet the
most stringent automotive requirements and to satisfactorily survive an
FMEA to adjacent-pin short and pin open
operations in a typical configuration.
The purpose of this test is to emulate the
effects of typical PCB defects and determine if they are destructive. In the test,
the LTC3864 was configured for a VIN of
12V and VOUT of 5V with an output load
of 1A. Each pin was then systematically
opened and adjacent pins shorted and
the results measured. In all instances, the
LTC3864 recovered when the FMEA conditions were removed. The results can
be found in the LCT3864 data sheet.
Figure 3. Short-circuit operation including soft
recovery from short
April 2013 : LT Journal of Analog Innovation | 13
SIMPLE AND EASY TO USE
The LTC3864 is a nonsynchronous
PMOS DC/DC controller and can be used in
a variety of low to medium current level
applications. Figure 4 shows a typical
5V output automotive application. This
is a minimum component count solution. Include input and output capacitors,
PMOS switch, nonsynchronous diode,
sense resistor, bias caps and compensation and the application is complete.
CCAP
0.1µF
RRUN
100k
VIN
RITH 9.53k
The LTC3864 fits a wide variety of applications where size and light load efficiency are paramount. The output can be
programmed from 0.8V up to a maximum
of 60V. Output currents typically range
up to 5A depending on the application.
Figure 5 shows 24V output voltage, 750kHz
application with 92% peak efficiency at
1A and greater than 72% at low current
efficiency in Burst Mode operation.
LTC3864
SW
RPGD
100k
FREQ
PGOOD
SGND
CIN1: NICHICON UPJ1J120MDD
D1: DIODES INC SBR3U100LP
L1: TOKO 1217AS-H-100M
MP: FAIRCHILD FDMC5614P
RFB1
80.6k
LTspice IV
circuits.linear.com/622
SUMMARY
The LTC3864 is a versatile, easy-to-use high
voltage PMOS controller with excellent
light load efficiency. Its 40µ A low IQ Burst
Mode operation is suited to applications
where standby light load efficiency is
important such as in always-on power
systems. The 100% duty cycle capability allows the output voltage to ride
through severe input voltage droop such
as in a cold crank condition. The LTC3864
is designed to operate in low VIN droop
conditions where minimum VIN is 3.5V over
the full temperature range. The LTC3864
provides high input voltage capability
CIN2
2.2µF
CAP
MODE/PLLN
CVIN
0.1µF
+
CIN1
33µF
63V
and excellent light load efficiency in a
simple and easy-to-use 12-pin package.
The LTC3864E and LTC3864I versions
operate from –40°C to 125°C junction
temperature. The LTC3864H is guaranteed to operate from a –40°C to 150°C
operating junction temperature. The
LTC3864MP is 100% tri-temperature tested
and guaranteed to operate from –55°C to
150°C operating junction temperature.
Visit www.linear.com/LTC3864 for
data sheets, demo boards and other
applications information. n
VIN
24V TO 60V
RSENSE
50mΩ
SENSE
CITH1
6.8nF
RITH 30.1k
SS
ITH
LTC3864
GATE
CITH2 100pF
RFREQ 97.6k
FREQ
CIN1: NICHICON UPJ1J100MPD
D1: DIODES INC SBR3U100LP
L1: TOKO 1217AS-H-470M
MP: VISHAY/SILICONIX SI7113DN
14 | April 2013 : LT Journal of Analog Innovation
MP
L1
47µH
D1
SGND
RPGD2
768k
VFB
*VOUT FOLLOWS VIN WHEN 3.5V ≤ VIN ≤ 24V
10µF
RFB2
887k
PGOOD
PGND
RFB2
422k
VFB
Figure 4. Typical 5V output automotive application
VIN
CFF
47pF
VOUT*
5V
47µF 2A
×2
*VOUT FOLLOWS VIN WHEN 3.5V ≤ VIN ≤ 5.2V
SEE DROPOUT BEHAVIOR IN TYPICAL PERFORMANCE CHARACTERISTICS
CCAP
0.1µF
circuits.linear.com/623
L1
10µH
D1
RUN
VIN*
5.2V TO 55V
RSENSE
25mΩ
MP
GATE
ITH
CITH2 100pF
Figure 5. 24V to 60V input, 24V/1A
output at 750kHz
LTspice IV
CIN1
12µF
63V
SENSE
PGND
This 5V, 2A output solution achieves an
efficiency of around 90% near maximum
load and maintains this high efficiency
all the way down through Burst Mode
operation at light loads, as shown in
Figure 1. The output voltage is programmed using feedback resistors RFB2
and RFB1 with an optional CFF available to
speed up transient response, if desired.
CVIN
0.1µF
MODE/PLLN
SS
CITH1
3.3nF
+
CAP
RUN
CSS
0.1µF
CIN2
4.7µF
RPGD1
200k
RFB1
30.1k
VOUT*
24V
1A
design features
Dual, Fast, Step-Down Controller’s External Reference Input
Enables Dynamic Voltage Scaling from 0.4V to 5.5V and
0.3% Total Combined Regulation Accuracy
Shuo Chen and Terry Groom
Low voltage, high current
systems require accurate
differential regulation. It is
not uncommon for power
supply rails at or below
0.9V to demand 25A or
more, with fast transients
that look like intermittent
electrical shorts to the
power supply rail. Such
systems typically require
power supply regulation
accuracy of less than 1%
regulated DC and 3% in the
face of input transients.
Increasingly, core processors and other
large scale digital ICs (such as ASICs and
FPGAs) require dynamic voltage scaling—either multiple fixed levels, or a
continuously adjusted reference voltage
using a servo loop—to deliver power
based on the processor demand. The
goal is that the system can keep the
applied power supply at the minimum
voltage necessary for proper operation
based on processing demand to conserve
Figure 1. Channel 2 of the LTC3838-2
regulates to an external reference;
channel 1 to an internal reference.
VOUT1 and VOUT2 allow remote grounds
up to ±500mV and ±200mV, respectively.
VREF2 is also differentially sensed, but
no separate pin for the remote ground
of external reference is required.
PART NUMBER
LTC3838-1/-2 Ch.1
and LTC3838-1 Ch.2
LTC3838-2 Ch.2
(e.g., with ±0.1%
Linear Technology
Voltage References)
REFERENCE
VOLTAGE
OUTPUT
VOLTAGE
TOTAL COMBINED ACCURACY
(GROUND, LINE, LOAD & TEMP)*
0.6V Internal
0.6V to 5.5V
0.6V External
0.6V to 5.5V
< ±0.67% (-40ºC ≤ T A ≤125ºC)
1.5V External
1.5V to 5.5V
< ±0.4% (-40ºC ≤ T A ≤125ºC)
2.5V External
2.5V to 5.5V
< ±0.3%
< ±0.75% (0ºC ≤ T A ≤85ºC)
< ±1% (-40ºC ≤ TA ≤125ºC)
*external resistor divider error not included
Table 1. Output voltage regulation accuracy over remote power ground deviation (up to ±200mV), input voltage
(4.5V to 38V), output current, and temperature
energy. One example is LSI’s adaptive
voltage scaling & optimization (AVSO).
stage. The problem is that soft-start and
many common fault control features such
a overvoltage protection might be sacrificed, depending on the technique used.
The LTC3838-2 is designed to meet
the extreme accuracy requirements
through precision differential output
sensing, and offer dynamic output
voltage scaling using the differential
external reference voltage input.
DUAL DIFFERENTIAL V OUT
ACCURACY THAT MATTERS
To obtain superior regulation accuracy,
power supply designers sometimes bypass
a controller’s internal error amplifier and
instead use a discrete precision reference
and external op amps to control the power
The LTC3838-2 avoids this trade-off by
allowing the use of an external reference
for accuracy while preserving valuable
fault and protection features. With a precision voltage reference (such as LTC6652)
from Linear Technology, or a DAC for programmability, the channel 2 output of the
LTC3838-2 can be tightly regulated from
0.4V to 5.5V in applications with currents
up to 25A per channel. At a very low 0.6V,
the LTC3838-2 is able to achieve a total
VOUT2+
VOUT1+
RFB2
COUT1
LTC3838-2
RDFB2
VOUTSENSE1+ VDFB2+
RDFB1
RFB1
VOUT1–
REMOTELY-SENSED
POWER GROUND 1,
±500mV MAX vs SGND
VOUTSENSE1– VDFB2–
EXTVREF2
VREF2+
+
–
VREF2–
RDFB3 =
RDFB1//RDFB2
TO PROGRAM VOUT2 = VREF2,
COUT2 REMOVE RDFB1AND USE
RDFB3 = RDFB2
VOUT2–
REMOTELY SENSED
POWER GROUND 2,
±200mV MAX vs SGND
REMOTELY SENSED
EXTERNAL REFERENCE GROUND
April 2013 : LT Journal of Analog Innovation | 15
For differential external reference sensing, the LTC3838-2 has only one pin for
external reference input. Channel 2 features a unique feedback amplifier configuration,
which eliminates the need for a separate pin to sense the external reference’s
remote ground. Instead, one additional resistor, equal to the parallel of the two
feedback resistors, is used to connect to the remote ground externally.
combined accuracy of ±4mV, or ±0.67%,
over all operating conditions including line, load, extreme temperature and
remote ground deviation up to ±200mV.
TRACKING DYNAMIC DIFFERENTIAL
EXTERNAL REFERENCE
by scaling feedback with respect to a
fixed lower reference voltage, where
the percentage error does not change.
For example, with an external reference of 2.5V, the total relative tolerance
is less than ±0.3%. The LTC3838-2’s dual
channels can be configured to singleoutput applications using channel 2’s
external reference at such accuracy.
Relative accuracy improves as the reference increases because the absolute
error is a smaller percentage out of a
larger reference voltage. This contrasts
with programming the output voltage
For differential external reference sensing, the LTC3838-2 has only one pin for
external reference input. Channel 2
features a unique feedback amplifier
configuration, which eliminates the need
for a separate pin to sense the external
reference’s remote ground. Instead, one
Figure 2. A LTC3838-2, 300kHz, 2-phase single-output step-down converter with inductor-DCR sense. This application converts a 4.5V to 14V input to a dynamic 0.4V to
2.5V output at 50A.
VIN
4.5V TO 14V
+
CIN2
22µF
×4
CIN1
180µF
2.2Ω
1µF
LTspice IV
0.1µF
4.02k
L1
0.4µH
MT1
+
COUT2
330µF
×2
SENSE1–
SENSE2–
SENSE1+
SENSE2+
BOOST1
BOOST2
0.1µF
TG1
DB2
4.7µF
MB1
DRVCC2
EXTVCC
DRVCC1
INTVCC
BG1
BG2
EXTVREF2
VOUTSENSE1+
VDFB2+
VREF2+
0.4V TO 2.5V 10k
10k
VREF2–
VOUTSENSE1–
PGOOD1
VDFB2–
PGOOD2
TRACK/SS1 TRACK/SS2
ITH1
16 | April 2013 : LT Journal of Analog Innovation
137k
DTR1
RT
SGND
RUN1
L2
0.4µH
VOUT
0.4V TO 2.5V
50A
SW2
PGND
CIN1: SANYO 16SVP180MX
CIN2: MURATA GRM32ER61C226KE20L
COUT1, COUT4: MURATA GRM31CR60J107ME39L
COUT2, COUT3: SANYO 2R5TPE330M9
DB1, DB2: CENTRAL SEMI CMDSH-4ETR
L1, L2: VISHAY IHLP5050FDERR40M01
MT1, MT2: INFINEON BSC050NE2LS
MB1, MB2: INFINEON BSC010NE2LS
4.02k
MT2
TG2
SW1
1µF
16.2k
0.1µF
DB1
2.2Ω
COUT1
100µF
×2
LTC3838-2
0.1µF
16.2k
circuits.linear.com/624
VIN
ITH2
DTR2
PHASMD
MODE/PLLIN
CLKOUT
RUN2
100k
PGOOD
0.01µF
100pF
1000pF
7.5k
MB2
COUT3 +
330µF
×2
COUT4
100µF
×2
design features
In addition to regulation accuracy, the LTC3838-2 offers widebandwidth tracking to a dynamic external reference. Tracking
bandwidth is important in applications like dynamic voltage
scaling because bandwidth determines how quickly the supply
can respond to changes to the programmed external reference.
additional resistor equal to the parallel of
the two feedback resistors is used to connect to the remote ground externally. See
the LTC3838-2 data sheet for an explanation of how this configuration works.
LT3838-1 CONTROLLER: INTERNAL
REFERENCE ON BOTH CHANNELS
quickly the supply can respond to changes
to the programmed external reference.
The LTC3838-1 shares the same functions
as LTC3838-2, except channel 2 of the
LTC3838-1 uses a 0.6V internal reference.
Like its predecessors, the LTC3838 and
LTC3839, both the LTC3838-1 and -2 use the
controlled on-time, valley current mode
architecture, which offers superior regulation during fast load transients without
the typical switching period response delay
of fixed frequency controllers, while still
capable of constant frequency switching
locked to an external 200kHz to 2MHz
clock. They retain all features of the
LTC3838, including the proprietary detect
transient release (DTR), which improves
the transient performance in low output
voltage applications. Like the LTC3838,
both LTC3838-1 and -2 include a full set
of popular features, such as an external
VCC power pin, RSENSE or inductor-DCR current sensing, selectable light load operating
modes, overvoltage protection and current
Figure 3 shows Bode plots from a 350kHz
LTC3838-2 step-down converter compensated to an aggressive bandwidth close
to 1/3 of the switching frequency without sacrificing stability. This allows the
LTC3838-2 to track an external sine wave
of 3.5kHz or 1/100 switching frequency
at full power, without any noticeable
distortion even at the sine wave’s very
high bandwidth start and stop instants
(Figure 4). Careful attention should be
paid to the bandwidth requirements for
any dynamic system. The wide-bandwidth
external-reference-tacking capability, in
addition to high speed transient performance, makes the LTC3838-2 ideally suited
for the most dynamic supply applications.
Figure 2 shows a typical LTC3838-2
application with external reference
input. This 2-phase converter is capable
of producing 50A over a wide ranging
output from 0.4V to 2.5V. For example,
at 1.5V this application can achieve 0.4%
total combined accuracy for all operating conditions. The high accuracy and
superior transient performance make
the LTC3838-2 well suited for the most
demanding processor applications.
In addition to regulation accuracy, the
LTC3838-2 offers wide-bandwidth tracking to a dynamic external reference.
Tracking bandwidth is important in
applications like dynamic voltage scaling because bandwidth determines how
Figure 3. Loop gain and closed-loop Bode plots taken with an OMICRON Lab network analyzer
on VOUT2 of a 350kHz LTC3838-2 step-down converter with external reference (EXTVREF2).
60
90
50
75
PHASE
45
30
10
15
0
–15
–20
–30
10
PHASE
GAIN
VOUT2
1V/DIV
–5
–60
–10
–120
0
–10
–30
0
0
PHASE (deg)
GAIN (dB)
60
20
GAIN
EXTVREF2
1V/DIV
PHASE (deg)
30
60
5
GAIN (dB)
40
Figure 4. The LTC3838-2 tracks a 1V peak-to-peak,
3.5kHz sine wave external reference.
100
FREQUENCY (kHz)
–45
1000
–15
1
10
100
FREQUENCY (kHz)
–180
1000
SW2
10V/DIV
PGOOD2
5V/DIV
100µs/DIV
April 2013 : LT Journal of Analog Innovation | 17
The LTC3838-1/-2 is the ideal choice for power in
applications requiring fast transient performance,
dual accurate differential output regulation, and
external references for increased VOUT accuracy
and programmability down to 0.4V.
limit foldback, soft-start/rail tracking, and
PGOOD and RUN pins for each channel.
In addition to the differential remote output sensing on both channels, a significant
improvement of the LTC3838-1/-2 over the
original LTC3838 is the maximum current
sense threshold voltage (i.e., current limit)
accuracy. Unlike the LTC3838, which has
a continuously variable and two fixed
current limit ranges (VRNG), the LTC3838-2
has a fixed VRNG of 30mV (typical) and
its tolerance over temperature is ±20%,
POWER VIN
3.3V TO 14V
which is much improved. The LTC3838-1
has the same 30mV and an additional
60mV (typical) VRNG setting whose tolerance is also significantly tighter. Refer to
Table 2 for the comparison on the current
limit tolerances and VRNG controls of the
LTC3838-series 2-channel controllers.
VBIAS SUPPLY
5V TO 5.5V
+
CIN1
180µF
LTspice IV
circuits.linear.com/625
CIN2
22µF
×4
2.2Ω
1µF
VIN
L1
0.47µH
M1
SENSE2–
SENSE1+
SENSE2+
BOOST1
BOOST2
TG1
COUT2
330µF
×2
4.7µF
DRVCC1
INTVCC
DRVCC2
EXTVCC
0.01µF
VOUTSENSE1–
VDFB2–
PGOOD2
470pF
137k
ITH1
DTR1
VRNG
RT
SGND
RUN1
ITH2
DTR2
PHASMD
MODE/PLLIN
CLKOUT
RUN2
+
COUT4
100µF
×2
20k //10k
SGND
PGOOD2
100k
0.01µF
TRACK/SS1 TRACK/SS2
47pF
23.2k
20k
PGOOD1
VOUT2
0.9V
20A
10k
VDFB2+
10k
PGOOD1
COUT3
330µF
×2
BG2
VOUTSENSE1+
100k
RS2
L2
0.47µH 0.0015Ω
SW2
BG1
10k
18 | April 2013 : LT Journal of Analog Innovation
M2
DB2
SW1
1µF
100Ω
0.1µF
TG2
PGND
Figure 5. When an external 5V rail is commonly available
to bias up the controller, this LTC3838-1 application
converts a dynamic 3.3V to 14V power input to 20A dual
outputs of 1.2V and 0.9V.
100Ω
DB1
2.2Ω
+
SENSE1–
1nF
0.1µF
RS1
0.0015Ω
LTC3838-1
1nF
100Ω
COUT1
100µF
×2
Figure 5 shows the VIN pin connected
via diode-OR to the VBIAS 5V rail and to
power VIN, 3.3V–14V, rail. This allows
the power VIN rail to dynamically switch
between a higher voltage and a minimum of 3.3V. When operating with the
power VIN supply below 5.5V, this application requires the VBIAS supply to be
present at EXTVCC in order to maintain
The LTC3838-1/-2 controllers require a
minimum VIN pin voltage of 4.5V, but
this does not limit the power input
to 4.5V. For example, many digital
systems have an available regulated
100Ω
VOUT1
1.2V
20A
5V rail, which can be used to bias the
VIN pin and gate drivers, and to efficiently step down inputs less than 4.5V.
47pF
470pF
17.4k
CIN1: SANYO 16SVP180MX
CIN2: MURATA GRM32ER61C226KE20L
COUT1, COUT4: MURATA GRM31CR60J107ME39L
COUT2, COUT3: SANYO 2R5TPE330M9
DB1, DB2: CENTRAL SEMI CMDSH-4ETR
L1, L2: WÜRTH 7443330047
M1, M2: INFINEON BSC0911ND
design features
Using an external reference, the LTC3838-2 can achieve
total accuracy levels as low as 0.3% under all operating
conditions. The external reference feature is designed to
accommodate dynamic voltage scaling and track fast
external reference inputs with minimum distortion.
Table 2. Maximum Current Sense Threshold Voltage Specifications and Range Controls
PART
V RNG = SGND
V RNG = INTV CC
V RNG CONTROL
V RNG PIN(s)
LTC3838 and LTC3839
21mV to 40mV
39mV to 61mV
30mV–200mV continuous & 30mV/50mV fixed
each per channel
LTC3838-1
24mV to 36mV
54mV to 69mV
30mV/60mV fixed
single
LTC3838-2
24mV to 36mV
30mV fixed only
no
DRVCC , INTVCC and VIN pin voltages
needed for the IC to function properly.
The EXTVCC supply is optional when the
power VIN supply is at or above 5.5V.
Note that the power input voltage range
of this application cannot be generalized
for other frequencies and output voltages, and each application that needs a
power input voltage different from the
VIN pin voltage should be tested individually for margin of range in which
the switching nodes (SW1, SW2) phaselock to the clock output (CLKOUT).
SUMMARY
The LTC3838-1/-2 is the ideal choice for
power in applications requiring fast
transient performance, dual accurate differential output regulation, and external
references for increased VOUT accuracy and
programmability down to 0.4V. Compared
to the original LTC3838, the LTC3838-1/
LTC3838-2 offers differential output sensing on both channels, improved current
limit accuracy, and the choice of internal/
external reference. Using an external reference, the LTC3838-2 can achieve accuracy
levels as low as 0.3% under all operating
conditions. The external reference feature is designed to accommodate dynamic voltage scaling
and track fast external reference
inputs with minimum distortion.
The LTC3838-1 and -2 are offered
in 38-pin QFN (5mm × 7mm)
packages with exposed pads for
enhanced thermal performance.
Visit www.linear.com/LTC3838-1
and /LTC3838-2 for data sheets,
demo boards, a variety of applications designs, and for more
information about how:
•a 30ns minimum on-time enables
high step-down ratios, e.g.,
from 38V to 0.8V at 350kHz
•2MHz switching frequency
enables applications with
tiny power components
For More Information…
THE LTC3838/LTC3839, PREDECESSOR TO THE
LTC3838-1/-2:
See the article:
• 2MHz Dual DC/DC Controller Halves Settling
Time of Load Release Transients, Features
0.67% Differential VOUT Accuracy and is
Primed for High Step-Down Ratios in the
LT Journal of Analog Innovation, April 2012
(Volume 22, Number 1).
THE LTC3833 SINGLE-CHANNEL CONTROLLER
The LTC3838 series of dual controllers are based
on and have all features of the single-channel
controller LTC3833. For a full discussion of the
features shared with LTC3833, refer to the cover
article:
• Fast, Accurate Step-Down DC/DC Controller
Converts 24V Directly to 1.8V at 2MHz in the
LT Journal of Analog Innovation, October 2011
(Volume 21, Number 3).
•25A output becomes practical at 2MHz, with 95% peak
efficiency (2V-5V VOUT). n
April 2013 : LT Journal of Analog Innovation | 19
4A Li-Ion Battery Charger Accepts Inputs to 32V
Rick Brewster
Advances in Li-ion battery technologies continue to produce batteries with increased
capacity and energy density. Charge/discharge rate capabilities are also rising,
sometimes to multiple C rates (C is the standard designator for battery capacity
stated in amp-hrs). These technologies are making their way into consumer,
automotive, medical and industrial markets. In most cases, the charger must
be able to recharge multiple sources over a wide range of input voltages.
High capacity/current batteries require
chargers that handle the high currents
safely, efficiently and cost effectively.
Until now, building a safe high current battery charger required the use
of multiple ICs and a host of external
components resulting in expensive and
bulky solutions. The LT3651 integrated
battery charger solves this problem
by supporting charge currents up to
4A and accepting input voltages to 32V.
BATTERY CHARGER FEATURES
Charger safety is a significant concern
as batteries increase in capacity. The
LT3651 includes all of the necessary charge
termination and protection features.
Charge termination methods include
C/10 termination or safety timer termination. Additional protection features
include battery temperature monitoring,
disabling charging of a battery that is
too hot or cold, battery preconditioning
for deeply discharged batteries and bad
battery detection when in timer mode.
The LT3651 provides an additional
PowerPath™ feature that regulates battery charge current in response to total
input supply current. With this feature,
the battery charger current is reduced if
other loads on the input supply increase
their current such that the total input
20 | April 2013 : LT Journal of Analog Innovation
THE CHARGE CYCLE
supply load exceeds a programmed limit.
This allows designers to reduce the input
supply requirements to more efficiently
manage power. This feature can also
be used to enforce a thermal budget by
limiting a set maximum input power.
Li-ion battery charging typically uses a
constant-current/constant-voltage (CC/CV)
charging algorithm. A Li-ion battery is
initially charged with constant current,
generally between 0.5C and 1C, though
newer batteries can use higher rates. As the
battery voltage approaches the full-charge
float voltage, the charger reduces charge
current and transitions into constant
voltage operation. The LT3651 prevents
overcharging of the battery, protecting
the battery against damage. There are
four variants of the LT3651 supporting
4.1V, 4.2V, 8.2V and 8.4V float voltages.
The LT3651 can be programmed via
an external resistor for switching
frequency, average battery charger
current and input current limit (reducing battery charge current to try and
maintain constant input current). An
external capacitor sets timeout period
for timer controlled termination.
The LT3651 operates at high frequency,
reducing inductor and filter component
size. The frequency is user adjustable,
offering the advantage of reduction of
power dissipation at higher voltages
and control of spectral harmonics.
Figure 1. Basic single
cell 4A charger
DCIN
6.5V TO
32V
Si7611DN
100k
The LT3651 combines a synchronous
buck switcher with a battery charger
to efficiently produce high charge current. It provides a CC/CV charging characteristic and adjusts charge current
10V
100k
LTspice IV
circuits.linear.com/626
CLP
VIN
CLN
SHDN
SW
ACPR
FAULT
BOOST
CHRG
LT3651-4.2
SENSE
RT
301k
TIMER
22µF
TO
SYSTEM
LOAD
1µF
1N5819
6.5µH
WÜRTH 744314650
24mΩ
BAT
NTC
ILIM RNG/SS GND
100µF
+
BATTERY
365142 TA01a
design features
Charger safety is a significant concern as batteries increase in
capacity and usage. The LT3651 includes all of the necessary
charge termination and protection features. Charge termination
methods include C/10 termination or safety timer termination.
RIL
16mΩ
SBM540
DCIN
50k
50k
50k
CIN
22µF
1µF
VLOGIC
50k
CLP
SHDN
VIN
SW
CLN
LT3651-8.4
ACPR
TO
CONTROLLER
FAULT
CHRG
Figure 2. 2-cell Li-ion 9V to 32V charger with input
current limit and 3-hour charge timeout
LTspice IV
circuits.linear.com/630
based on battery voltage. During constantcurrent operation, the maximum charge
current provided to the battery is programmable via a sense resistor, up to a
maximum of 4A and adjustable using the
RNG/SS pin. Charge current is internally
reduced as the battery approaches the fullcharge float voltage and the charger transitions to constant voltage charging mode.
A charge cycle terminates by either
charge current level or time. Once terminated, the charger is in a low power
state, which draws about 85µ A from the
input supply and less than 1µ A from
the battery. With both termination
modes, charging is restarted when the
battery voltage drops to 97.5% of the
float voltage (the recharge voltage).
Two pins indicate the charging state.
While charging the CHRG pin actively
sinks current so an LED from a supply
to this pin provides visual indication of
charging. The pin transitions to a high
1µF
CTIMER
0.68µF
1N5819
SENSE
RSENSE
24mΩ
BAT
NTC
ILIM RNG/SS GND
TIMER
0.47µF
impedance upon completion of a charge
cycle. A FAULT pin provides additional
information about charging disruptions
such as a battery out of temperature
range fault or a bad battery fault.
A 4A CHARGER WITH INPUT SHORT
PROTECTION
Figure 1 shows a basic 4A single-cell Li-ion
battery charger that operates from a
6.5V to 32V input. Charging is suspended if
the input supply voltage exceeds 32V, but
the IC can withstand input voltages as high
as 40V without damage. So this application
can be used for charging from different inputs inside the 6.5V to 32V range.
The 4A maximum charge current corresponds to 95mV across the 24mΩ external
sense resistor. This basic design does not
take advantage of the status pins, battery temperature monitoring or safety
timer features. The battery charging cycle
terminates when the battery voltage
approaches 4.2V and the charge current
3.3µH
BOOST
RT
RT
54.9k
TO
SYSTEM
LOAD
CBAT
100µF
NTC B
10k
+
BATTERY
falls to approximately 400m A. A new
charge cycle is automatically initiated
when the battery voltage falls to 4.1V.
A MOSFET is used as a low loss diode to
provide reverse blocking in the event
of an input short. This prevents battery discharge through the charger.
WIDE INPUT RANGE, 2-CELL
CHARGER
Figure 2 shows a 2-cell 9V to 32V charging application. This could be used in an
automotive application where the input
needs to tolerate a wide input voltage.
This application uses the -8.2 or -8.4
option for charging two Li-ion cells at 4A.
This application also uses the input current regulation feature. RIL monitors the
current drawn from the supply that supplies both the charge current and system
load. It is set such that if the combined
input current exceeds 6.3A, charge current is reduced to keep input current from
increasing. Often input supply voltages
April 2013 : LT Journal of Analog Innovation | 21
The LT3651 operates at high frequency, reducing inductor
and filter component size. The frequency is user adjustable
offering the advantage of reduction of power dissipation
at higher voltages and control of spectral harmonics.
In this application, the safety timer is used
for termination, the timer is paused for
the duration of a temperature fault, so a
battery receives a full-duration charging
cycle, even if that cycle is interrupted if
the battery is out of the allowed temperature range. The capacitor on the timer
pin sets the charge time, in this case it is
three hours, so charging continues past
the C/10 charge point. At timeout the
part goes into standby and reduces battery discharge current to less than 1µ A.
The timer also provides for determination of a bad battery. The LT3651 has an
automatic precondition mode, which
gracefully initiates a charging cycle for
deeply discharged batteries. If the battery
voltage is below the precondition threshold of 70% of the float voltage (5.8V for
the -8.4), the maximum charge current
is reduced to 15% of the programmed
maximum (0.15C) until the battery voltage rises past the precondition threshold.
This current is sufficient to activate any
safety circuitry in a battery pack and also
provides a small charge current. If the
battery does not respond to the precondition current and the battery voltage does
not rise past the precondition threshold
after 1/8 of the charge cycle (22.5min in
this application), full-current charge is
not initiated and a battery fault is issued.
22 | April 2013 : LT Journal of Analog Innovation
RIL
100k
180k
SMAZ18
18V
RT
54.9k
CLP
VIN
CLN
SHDN
SW
ACPR
FAULT
BOOST
CHRG
LT3651-4.2
SENSE
RT
SBM540
CIN
22µF
DCIN
5
MAXIMUM CHARGE CURRENT (A)
are relatively constant. For applications where this is true, then the setup in
Figure 2 also limits total input power. For
example, with a 12V input supply total
input power will be limited to about 75W.
1µF
3.3µH
1N5819
RSENSE
24mΩ
BAT
NTC
ILIM RNG/SS GND
TIMER
CBAT
100µF
+
4
3
2
1
BATTERY
0
3k
5
10
15
20
VIN (V)
25
30
35
Figure 3. 4A single cell charger with high voltage current foldback
This application also makes use of an
external NTC resistor in the battery
pack to monitor battery temperature.
Under- and overtemperature protection is
enabled by connecting a 10k NTC thermistor from the part’s NTC pin to ground.
This function suspends a charging cycle
if the temperature of the thermistor is
greater than 40°C or less than 0°C.
back to a controller. While the LT3651
does not need a controller to operate, one
could be used for additional functionality.
The status pins indicate: standby/shutdown; CC/CV charging (>C/10); bad battery
detection and temperature fault. Of course
in other applications an LED could be
placed on these pins for visual indication.
An additional feature of the LT3651 is
the ability to withstand input voltages to
40V, which helps in automotive designs.
The two status pins CHRG and FAULT are
used to communicate charger status
SBM540
TO
SYSTEM
CIN LOAD
22µF
DCIN
SMAZ9V1
9.1V
Figure 4. 4A 2-cell charger with
low voltage current foldback
RT
54.9k
VIN
CLP
CLN
SHDN
SW
ACPR
FAULT
BOOST
CHRG
LT3651-8.4
SENSE
RT
68k
5.1k
BAT
NTC
ILIM RNG/SS GND
TIMER
1µF
1µF
3.3µH
1N5819
RSENSE
24mΩ
CBAT
100µF
+
BATTERY
design features
The LT3651 is a versatile, compact and easy-to-use solution for charging
Li-ion batteries with up to 4A in current and from input supplies up
to 32V (40V ride through). High efficiency, built in safety controls and
compact size make it an easy fit in a wide variety of applications.
When the input voltage exceeds 32V the
output switches are disabled but can
ride out the overvoltage condition.
An input diode is used to protect from
discharging the Li-ion batteries in the
event of an input short. This could be
replaced with a MOSFET as in the previous example to improve efficiency.
MORE OPTIONS
The charge current and input current limit
control pins can also be used to provide
other functionality to a charger application. Figure 3 shows an application where
the charge current is diminished with
increasing DCIN, a useful feature to control
power dissipation of the input source.
SUMMARY
current and can be changed dynamically
to produce additional functionality.
The LT3651 is a versatile, compact and
easy-to-use solution for charging Li-ion
batteries with up to 4A in current and
from input supplies up to 32V (40V ride
through). High efficiency, built-in safety
controls and compact size make it ideal
for a wide variety of applications.
Figure 5 shows an application that offers a
maximum power point control (MPPC) feature that regulates input voltage at a constant voltage. This is useful for solar panel
applications. It makes use of the input current limit amplifier and reconfigures it for
input voltage regulation. The differential
CLP-CLN voltage is used to regulate output
current. The reference is set with a Zener
diode but could be done many ways. The
NPNs are used to buffer the CLN input bias
current. ILIM is shorted to remove the
built-in offset between CLP and CLN. In this
case the input regulation is set for 17V, but
is adjustable with the 100k/61.9k divider.
Visit www.linear.com/LT3651 for
data sheets, demo boards and other
applications information. n
Figure 4 shows an application with
the inverse feature, where charge
current is reduced at lower input
voltage, so in the event a supply voltage drops, less load is drawn.
50k
6.2V
Note in general both the ILIM pin and the
RNG/SS pin provide control over charge
SOLAR
PANEL
INPUT
61.9k
Si7611DN
VIN
100k
10V
10k
90.9k
Figure 5. 4A 2-cell charger with
maximum power point control
12.1k
CLP
VIN
CLN
SHDN
SW
ACPR
FAULT
BOOST
CHRG
LT3651-8.4
SENSE
RT
54.9k
TIMER
LTspice IV
circuits.linear.com/627
100pF
100k
22µF
1µF
3.3µH
1N5819
24mΩ
BAT
NTC
ILIM RNG/SS GND
22µF
+
BATTERY
April 2013 : LT Journal of Analog Innovation | 23
What’s New with LTspice IV?
Gabino Alonso
Follow @LTspice on Twitter for
up-to-date information on models, demo circuits,
events and user tips: www.twitter.com/LTspice
SELECTED DEMO CIRCUITS
LED Drivers
Operational Amplifiers
• LTC3783: Single inductor buckboost LED driver with analog and
PWM dimming (9V–20V to 4× WLEDs at
350m A) www.linear.com/LTC3783
• LT6016: Precision high voltage
high side load current monitor
www.linear.com/LT6016
• LTC6406: Differential amplifier with
impedance matching and level shifting
www.linear.com/LTC6406
High Side Switches
Step-Down Regulators
SELECTED DEVICE MODELS
• LT3763: 70W, solar powered SLA battery
charger with maximum power point
regulation (37V–60V to 14V at 5A)
www.linear.com/LT3763
• LTC3646: High efficiency low quiescent
current step-down converter (7V–40V to
5V at 1A) www.linear.com/LTC3646
• LTM4620A Demo Circuit: High
efficiency single 26A step-down
regulator (4.5V–16V to 1V at 26A)
www.linear.com/LTM4620A
Linear Regulators (LDO)
• LT1185: Negative regulator with
3.5A current limit (6V–16V to –5V at 3A)
www.linear.com/LT1185
• LT1910: Fault protected high side switch
(8V–48V supply) www.linear.com/LT1910
• LTC3864: 60V Low IQ step-down
DC/DC controller with 100% duty cycle
capability www.linear.com/LTC3864
• LTC3890-2: 60V Low IQ, dual, 2-phase
synchronous step-down DC/DC controller
www.linear.com/LTC3890-2
• LTM4620A: Dual 13A or single
26A DC/DC µModule regulator
www.linear.com/LTM4620A
Operational Amplifiers and ADC Driver
Buck-Boost Regulators
• LT6016: Dual/quad 3.2MHz, 0.8V/µs low
power, Over-The-Top® precision op amp
www.linear.com/LT6016
• LT8705: 80V VIN and VOUT synchronous
4-switch buck-boost DC/DC controller
www.linear.com/LT8705
• LT6236: Rail-to-rail output 215MHz,
1.1nV/√Hz op amp/SAR ADC driver
www.linear.com/LT6236
LED Driver
• LTC6090: 140V CMOS rail-to-rail
output, pA input current op amp
www.linear.com/LTC6090
Step-Down Switching Regulators
• LT3504: Quad 40V/1A step-down switching
regulator with 100% duty cycle
operation www.linear.com/LT3504
• LTC3646: 40V, 1A synchronous step-down
converter www.linear.com/LTC3646
• LT3763: 60V high current stepdown LED driver controller
www.linear.com/LT3763
Ideal Diodes and Current Balancing
Controllers
• LTC4353: Dual low voltage ideal diode
controller www.linear.com/LTC4353
• LTC4370: Two-supply diode-OR
current balancing controller
www.linear.com/LTC4370
Battery Chargers
• LTC4009: High efficiency, multi-chemistry
battery charger www.linear.com/4009
What is LTspice IV?
LTspice® IV is a high performance SPICE simulator, schematic capture and waveform viewer designed to speed
the process of power supply design. LTspice IV adds enhancements and models to SPICE, significantly reducing
simulation time compared to typical SPICE simulators, allowing one to view waveforms for most switching
regulators in minutes compared to hours for other SPICE simulators.
LTspice IV is available free from Linear Technology at www.linear.com/LTspice. Included in the download is a
complete working version of LTspice IV, macro models for Linear Technology’s power products, over 200 op amp
models, as well as models for resistors, transistors and MOSFETs.
24 | April 2013 : LT Journal of Analog Innovation
• LTM8061: 32V, 2A µModule
Li-ion/polymer battery charger
www.linear.com/LTM8061 n
design ideas
LTspice HotKeys
Schematic
Symbol
Netlist
Waveform
ESC - Exit Mode
ESC - Exit Mode
M odes
Perform a Small Signal AC Analysis
.DC
F5 – Delete
F5 – Delete
F5 – Delete
F6 – Duplicate
F6 – Duplicate
.ENDS
F7 – Move
F7 – Move
.FOUR
Compute a Fourier Component
F8 – Drag
F8 – Drag
.FUNC
User Defined Functions
F9 – Undo
F9 – Undo
F9 – Undo
F9 – Undo
.FERRET
Download a File Given the URL
Shift+F9 – Redo
Shift+F9 – Redo
Shift+F9 – Redo
Shift+F9 – Redo
.GLOBAL
End of Netlist
.END
End of Subcircuit Definition
Declare Global Nodes
Ctrl+Z – Zoom Area
Ctrl+Z – Zoom Area
Ctrl+Z – Zoom Area
.IC
Set Initial Conditions
Ctrl+B – Zoom Back
Ctrl+B – Zoom Back
Ctrl+B – Zoom Back
.INCLUDE
Include another File
Ctrl+E – Zoom Extents
.LIB
Include a Library
.LOADBIAS
Load a Previously Solved DC Solution
.MEASURE
Evaluate User-Defined Electrical Quantities
Space – Zoom Fit
Vie w
reference table of Hot Keys,
DOT
commands
and more you can download
Annotate the Subcircuit Pin Names on Port currents
Perform a DC
Source
Sweep Analysisflyer at www.linear.com/LTspice.
the
LTspice
.AC
.BACKANNO
F3 – Draw Wire
Pla ce
Simulator Directives - Dot Commands
For a complete
Short Description
Command
Ctrl+G – Toggle Grid
Ctrl+G – Toggle Grid
U – Mark Unncon. Pins
Ctrl+W – Attribute Window
A – Mark Text Anchors
Ctrl+A – Attribute Editor
Ctrl+A – Add Trace
Atl+Click - Power
Ctrl+Y – Vertical Autorange
Ctrl+Click - Attr. Edit
Ctrl+Click - Average
Ctrl+H – Halt Simulation
R – Resistor
R – Rectangle
C – Capacitor
C – Circle
Ctrl+H – Halt Simulation
L – Inductor
L – Line
D – Diode
A – Arc
G – GND
Ctrl+G – Goto Line #
‘0’ - Clear
Ctrl+R – Run Simulation
Ctrl+H – Halt Simulation
Command Line Switches
Flag
Short Description
-ascii
Use ASCII .raw files. (Degrades performance!)
-b
Run in batch mode.
-big or -max Start as a maximized window.
S – Spice Directive
T – Text
T – Text
F2 – Component
F4 – Label Net
Ctrl+E – Mirror
Ctrl+E – Mirror
Ctrl+R – Rotate
Ctrl+R – Rotate
LTspice IV
Keyboard shortcuts are an alternate way to invoke
KEYBOARD SHORTCUTS
one or more commands in LTspice that would
See Demo
otherwise only be accessible by clicking through
www.linear.com/LTspice 1-800-4-LINEAR
the menu or toolbar. You can view these
shortcuts
for the schematic editor by choosing Tools >
Control Panel > Drafting Options and clicking Hot
Keys. Additional Hot Keys are also available for the
Waveform Viewer, Symbol Editor and Netlist Editor.
n
-encrypt
Encrypt a model library.
-FastAccess
Convert a binary .raw file to Fast Access Format.
-netlist
Convert a schematic to a netlist.
-nowine
Prevent use of WINE(Linux) workarounds.
-PCBnetlist
Convert a schematic to a PCB netlist.
-registry
Store user preferences in the registry.
-Run
Start simulating the schematic on open.
Hot
can
be to have up to 7 nodes in subcircuit.
MOSFET’s
-SOIKeysAllow
-uninstall
Executes one step of the uninstallation process.
reprogrammed
by
-wine
Force use of WINE(Linux)
workarounds.
selecting a command
and then pressing the
key or key combination
for the command. For
example, you may want
to reprogram the Undo,
Redo and Duplicate
(Copy) commands to
a more traditional key
combination. To remove
a shortcut, select the
command and press
Delete.
.MODEL
Define a SPICE Model
.NET
Compute Network Parameters in a .AC Analysis
.NODESET
Supply Hints for Initial DC Solution
.NOISE
Perform a Noise Analysis
.OP
Find the DC Operating Point
.OPTIONS
Set Simulator Options
.PARAM
User-Defined Parameters
.SAVE
Limit the Quantity of Saved Data
.SAVEBIAS
Save Operating Point to Disk
.STEP
Parameter Sweeps
.SUBCKT
Define a Subcircuit
.TEMP
Temperature Sweeps
Power User Tip
Find the DC Small-Signal Transfer Function
.TF
.TRAN
Do a Nonlinear Transient Analysis
.WAVE
Write Selected Nodes to a .WAV file
Suffix
Suffix
Constants
f
1e-15
E
2.7182818284590452354
T
1e12
p
1e-12
Pi
3.14159265358979323846
G
1e9
n
1e-9
K
1.3806503e-23
1.602176462e-19
Meg
1e6
u
1e-6
Q
K
1e3
M
1e-3
TRUE 1
Mil
25.4e-6
FALSE 0
0213
UNDOCUMENTED SHORTCUTS
There are also several undocumented
shortcuts in LTspice that may be useful:
Alt + left-click on a label, V(n008), in
the waveform viewer to highlight that
particular net in the schematic editor.
To route wires at an
angle, hold down Ctrl key
as you draw them.
Text with a preceding underscore character,
e.g., “_FAULT” is displayed as an overbar,
active low, digital signal.
Ctrl + Alt + Shift + H temporarily
highlights all hidden text within
the schematic. In this example,
a series resistor and parallel
capacitor are encapsulated and
hidden within C5 to simplify
layout.
Happy simulations!
April 2013 : LT Journal of Analog Innovation | 25
Compact Quad Step-Down Regulator with 100% Duty Cycle
Operation Withstands 180V Surges
Jonathan Paolucci
Automotive, industrial and distributed applications routinely subject step-down
DC/DC converters to a vast assortment of supply voltage transients. High voltage
power spikes and input voltage dips can destroy sensitive circuits and jeopardize
system reliability. To avoid damage, most applications rely on Tranzorbs or protection
circuits that use MOSFETs as pass elements to suppress input voltage transients.
If an N-channel MOSFET is used for this purpose, some means of providing gate
drive above the input rail is necessary to bias the MOSFET on. Generating this bias
is an undesirable complication that most engineers would prefer to avoid.
The LT3504 is a 4-channel monolithic
step-down regulator designed for 100%
duty cycle operation. Its unique architecture makes available a bias voltage,
which is easily adapted to an N-channel
protection scheme, allowing the LT3504
to operate continuously through overvoltage transients and dropouts down
to 3.2V. Among its many features, the
LT3504 includes output voltage tracking
and sequencing, programmable frequency, programmable undervoltage
lockout, and a power good pin to indicate when all outputs are in regulation.
QUAD 1A STEP-DOWN REGULATOR
Figure 1 shows the complete application
circuit for a 4-output, 1A step-down regulator operating over a 3.2V to 30V range.
Q1 provides surge protection to 180V.
An on-chip boost regulator generates
VSUPPLY
3.2V to 30V
SURGE PROTECTION TO 180V
10Ω
Q1
R2
100k
R3
1k
D2
6.8V
SKY
C1
0.1µF
VIN
circuits.linear.com/628
C2
22µF
DA4
FB4
LT3504
L3
8.2µH
DA3
FB3
DA2
FB2
SW1
fSW = 1MHz
GND
DA1
FB1
10µF
10.2k
31.6k
10µF
10.2k
SW2
RT/SYNC
18.2k
53.6k
3.3V/1A
D5 43pF
L2
4.2µH
0.1µF
+
5V/1A
D4 22pF
SW3
EN/UVLO
VIN
VIN
VIN
VIN
RUN/SS1
RUN/SS2
RUN/SS3
RUN/SS4
1µF
×4
Figure 1. Complete quad buck regulator with 180V
surge protection
26 | April 2013 : LT Journal of Analog Innovation
SW4
SW5
L5
10µH
C1: Sanyo 50CE22BS
D1: BZT52C36-7-F
D2: BZT52C6V8-7-F
D3: BAT54-7-F
D4–D7: ON SEMI MBRM140T3
L3, L4: SUMIDA CDRH5D28-8R2 (8.2µH)
L1, L2: CDRH5D28-4R2 (4.2µH)
L5: TAIYO YUDEN CBC2016T100M (10µH)
Q1: FQB34N20L
L4
8.2µH
D3
2.2µF
D1
36V
LTspice IV
a voltage rail (VSKY) that is 5V greater
than the input voltage VIN . Under normal operating conditions (VIN < 33V), the
VSKY rail supplies gate drive to MOSFET Q1,
providing the LT3504 with a low resistance path to VSUPPLY. Additionally,
the VSKY pin supplies base drive for the
switches in each buck converter channel,
which allows for 100% duty cycle and
2.5V/1A
D6 82pF
L1
4.2µH
21.5k
22µF
10.2k
1.8V/1A
D7 100pF
12.7k
10.2k
22µF
design ideas
VIN
50V/DIV
SKY
2V/DIV
VSUPPLY
50V/DIV
VOUT1
1V/DIV
VIN
50V/DIV
VOUT2
1V/DIV
VOUT3
1V/DIV
VOUT4
1V/DIV
VSUPPLY
2V/DIV
VIN
2V/DIV
VOUT1,2,3,4
2V/DIV
20ms/DIV
100ms/DIV
100ms/DIV
Figure 2. Figure 1’s start-up behavior
Figure 3. Figure 1’s dropout performance
Figure 4. Overvoltage protection withstands 180V
surge
eliminates the need for the boost capacitor typically found in buck converters.
cycle-by-cycle peak current limiting, as
well as catch diode current limit sensing, to protect the part and the external
pass device from carrying excessive
current during overload conditions.
during the transient event is approximately half the peak power. As such,
the average power is given by:
Bear in mind that significant power dissipation occurs in Q1 during an overvoltage
event. The MOSFET junction temperature
must be kept below its absolute maximum rating. For the overvoltage transient
shown in Figure 4, MOSFET Q1 conducts
0.5A (1A load on all buck channels) while
withstanding the voltage difference
between VSUPPLY (180V) and VIN (33V). This
results in a peak power of 74W. Since the
overvoltage pulse in Figure 4 is roughly
triangular, average power dissipation
In order to approximate the MOSFET junction temperature rise from an overvoltage transient, one must determine the
MOSFET transient thermal response as
well as the MOSFET power dissipation.
Fortunately, most MOSFET transient
thermal response curves are provided
by the manufacturer (as shown in
Figure 5). For a 400ms pulse duration,
the FQB34N20L MOSFET thermal response
ZθJC (t) is 0.65°C/W. The MOSFET junction temperature rise is given by:
OVERVOLTAGE INPUT TRANSIENT
PROTECTION FOR MULTIPLE
OUTPUTS
Figure 4 shows the LT3504 regulating
all four channels at 1A load through a
180V surge event without interruption.
As the supply voltage rises, Zener diode
D1 clamps Q1’s gate voltage to 36V. The
source-follower configuration prevents
VIN from rising further than about 33V,
well below the LT3504’s 40V maximum
input voltage rating. The LT3504 uses
Figure 5. FQB34N20L MOSFET transient thermal
response
1
ZθJC(t), THERMAL RESPONSE (°C/W)
Start-up behavior is shown in Figure 2.
Resistor R2 pulls up on the gate of Q1,
forcing source-connected VIN to follow
approximately 3V below VSUPPLY. Once
VIN reaches the LT3504’s 3.2V minimum
start-up voltage, the on-chip boost converter immediately regulates the VSKY rail
5V above VIN . Diode D3 and resistor R3
bootstrap Q1’s gate to the VSKY, fully
enhancing Q1. This connects VIN directly
to VSUPPLY through Q1’s low resistance
drain-source path. It should be noted that,
prior to the presence of VSKY, the minimum
input voltage is about 6.2V. However, with
VSKY in regulation and Q1 enhanced, the
minimum run voltage drops to 3.2V, permitting the LT3504 to maintain regulation
through deep input voltage dips. Figure 3
shows all channels operating down to the
LT3504’s 3.2V minimum input voltage.
0.1
0.01
10–3
10–5
SINGLE PULSE
D = 0.5
D = 0.2
D = 0.1
D = 0.05
D = 0.02
D = 0.01
0.1
1
10–4 10–3 0.01
10
t1, SQUARE WAVE PULSE DURATION (s)
PDM
t1
t2
ZθJC(t) = 0.7°C/W MAX
DUTY FACTOR = D = t1/t2
TJM – TC = PDM • ZθJC(t)
PAVG( W) =
1
• PPEAK ( W) = 37 W
2
TRISE (°C) = Z θJC ( t) • PAVG( W) = 24°C
Note that, by properly selecting
MOSFET Q1, it is possible to withstand
even higher input voltage surges.
Consult manufacturer data sheets to
ensure that the MOSFET operates within
its Maximum Safe Operating Area.
INDUCTIVE SPIKE PROTECTION
Input voltage transients, coupled with low
ESR input capacitors, can produce large
inductive spikes, which may damage buck
converters. These high dV/dt events cause
large inrush currents to flow in power
connections and filter capacitors, particularly if parasitic inductance and resistance
(continued on page 29)
April 2013 : LT Journal of Analog Innovation | 27
µModule Regulator Charges Supercapacitor Backup Supply,
Supporting LDO Outputs When the Input Supply Fails
Andy Radosevich
The LTM8001 is a µModule regulator that combines a 5A switching regulator with an
array of five 1.1A low noise LDOs. The switching regulator can be set for constant
current, suitable to charge supercapacitors for power backup. The LTM8001
operates from 6V to 36V inputs. The switching regulator is capable of constant
output voltage or constant output current regulation at switching frequencies from
200kHz to 1MHz. The output of the switching regulator can be adjusted from
1.2V to 24V and the outputs of the LDOs are adjustable from 0V to 24V.
The switching regulator is set to regulate
output current at 5.6A (typical) to provide
a current limit that is above the maximum
output current of 5A. The regulated current level can be easily lowered. The inputs
for three of the LDOs are hardwired to the
output of the switching regulator, but the
input to the remaining bank of two LDOs
is undedicated, so it can be connected to
the switching regulator or elsewhere. The
bias inputs to the LDOs are undedicated
but are separated into two inputs: one
for the bank of three connected to the
switching regulator and the other for the
remaining bank of two LDOs. The outputs
of the LDOs can be operated separately
or paralleled for higher output currents.
Figure 1. The LTM8001 producing
3.3V at 1A and 2.5V, 0.5A
regulated outputs while charging
VIN
a supercapacitor for backup
9V TO 15V
power.
2-OUTPUT REGULATOR WITH
SUPPLY RIDE-THROUGH
SUPERCAPACITOR
Figure 1 shows the LTM8001 in a dual
output application: 3.3V at 1A and
2.5V at 0.5A. This setup also charges
a supercapacitor and draws on the
supercap to support the outputs in
the face of input supply failures.
The switching frequency is 600kHz and
the output voltage of the switching
regulator is 5V when the supercapacitor is fully charged. The input voltage is
from 9V to 15V and the LTM8001 charges
the supercapacitor at 5.6A, typical. The
resistor divider on the RUN pin programs
the circuit to turn on for a 9V or higher
input, but also ensures that the switching
VIN45
10µF
200k
circuits.linear.com/629
RUN
BIAS123
BIAS45 LTM8001
COMP
SS
VREF
ILIM
SYNC
GND
3.3V
1A
VOUT1
LDO 1
SET1
VOUT2
STEP-DOWN LDO 2 SET2
SWITCHING
V
REGULATOR LDO 3 OUT3
SET3
LDO 4
600kHz
3.09k
2.5V
0.5A
VOUT4
SET4
VOUT5
FBO LDO 5 SET5
RT
68.1k
28 | April 2013 : LT Journal of Analog Innovation
Figure 2 shows LDO VBIAS -to-output
dropout voltage vs output current.
According to Figure 2, the bias of the
higher voltage, 3.3V/1A LDO output must
be 1.5V higher than 3.3V, or 4.8V for
proper regulation. This means that the
LDO outputs remain in regulation during the time the supercapacitor voltage
decays 100mV from 4.9V to 4.8V. The
0.07Ω ESR of the PM-5R0V155-R supercapacitor reduces the available voltage from the
supercapacitor from 5V to 4.9V while the
supercapacitor provides 1.5A to the LDOs.
If the supercapacitor is 1.5F and the total
VOUT0
VIN0
48.7k
LTspice IV
regulator remains off when back-fed
by the supercapacitor when there is
an interruption to the input power.
4.7µF
10µF
124k
110k
47µF
5V
1.5F
5V SUPERCAP
PM-5ROV155-R
design ideas
The inputs for three of the LDOs are hardwired to the output of the switching
regulator, but the input to the remaining bank of two LDOs is undedicated, so
it can be connected to the switching regulator or elsewhere. The outputs of the
LDOs can be operated separately or paralleled for higher output currents.
BIAS-TO-OUTPUT DROPOUT VOLTAGE (V)
1.52
the LTM8001 parallels LDOs to distribute
heat and lower operating temperatures.
VIN
10V/DIV
1.50
1.48
VOUT0(SUPERCAP)
2V/DIV
1.46
1.44
1.42
VOUT1,2,3(3.3V)
2V/DIV
1.40
1.38
VOUT4,5(2.5V)
2V/DIV
1.36
1.34
0
200
400
800
600
OUTPUT CURRENT (mA)
1000
500ms/DIV
Figure 2. LDO VBIAS -to-output dropout voltage vs
output current
Figure 3. Supercapacitor power backup system
holds up the 3.3V output for well over 100ms
output current of the LDOs is 1.5A, the
holdup time for the 3.3V LDO output is:
regard to power dissipation, it maximizes
holdup time if the input supply fails.
Power loss is minimized by operating
the LDO with inputs that just meet, and
do not exceed, the bias dropout requirements of the 3.3V LDO. But the supercapacitor voltage must exceed the input
power dropout requirement to meet bias
dropout and holdup requirements. To
mitigate this increased power dissipation,
C
∆V
I
1.5
0.1
=
1.5
= 100ms
3.3V HOLDUP TIME =
Both the LDO bias and LDO input power
are connected to 5V from the supercapacitor. Although 5V is non-optimal with
Holdup time is longer when the supercapacitor provides bias to the LDOs compared to using a conventional capacitor
for that purpose. This avoids detrimental effects of charging a large capacitor
directly with the input voltage. Figure 3
shows that the 3.3V output holdup time
exceeds 100ms when the supercapacitor is charged to 5V and the LDO outputs are 3.3V at 1A and 2.5V at 0.5A.
CONCLUSION
The LTM8001 makes it easy to design a
multiple output voltage regulator circuit
featuring supercapacitor backup power.
It is possible to achieve significant holdup
time without adding large and undesirable capacitance directly to input power.
Visit www.linear.com/LTM8001 for
data sheets, demo boards and other
applications information. n
(LT3504 continued from page 27)
is low. External gate network C1 and D2
limits these inrush currents by controlling
Q1’s gate voltage slew rate. Since VIN follows Q1’s gate voltage, the external gate
network forces VIN to ramp modestly compared to the abrupt input voltage transient
present on VSUPPLY, as shown in Figure 6.
LT3504. During normal operation, the
LT3504’s built-in boost regulator permits
100% switch duty cycle operation and
serves as an excellent MOSFET gate driver.
The LT3504, along with a MOSFET and
gate clamp, provides a transient-robust,
compact multioutput solution.
CONCLUSION
Visit www.linear.com/LT3504 for
data sheets, demo boards and other
applications information. n
The high voltage standoff capability
of the series connected MOSFET blocks
dangerous spikes from reaching the
VSUPPLY
10V/DIV
VIN
10V/DIV
40µs/DIV
Figure 6. Fast VSUPPLY dV/dt is blocked from VIN by
series MOSFET and gate network
April 2013 : LT Journal of Analog Innovation | 29
New Product Briefs
CONTROLLER REPLACES TWO
POWER DIODES WITH MOSFETs TO
CONSERVE POWER AND PCB AREA
The LTC4353 is a dual ideal diode controller that replaces power diodes with
N-channel MOSFETs to save power, voltage drop, and circuit board space. In
high availability redundant supplies and
supply holdup circuits for brownout or
power-down, high current diode-ORs
formed using MOSFETs acting as ideal
diodes are more viable and efficient than
those using Schottky diodes. The LTC4353
joins the LTC4352, an ideal diode controller for a single low voltage supply. With
their unique rapid turn-on feature, both
controllers are well suited for low voltage
applications where limiting the voltage
droop during supply switchover is critical.
The LTC4353 can diode-OR supplies from
0V to 18V. By servoing a 25mV forward
drop across the MOSFET, it provides
a smooth oscillation-free switchover
between supplies. Reverse current through
the MOSFET activates a fast turn-off,
minimizing shoot-through and fault
currents. The LTC4353 can turn on the
external MOSFET within a microsecond,
faster than most ideal diode controllers. It employs a proprietary technique
using an external reservoir capacitor
for the integrated charge pump to provide 1.4A of gate pull-up current. A fast
turn-on curtails the downward excursion of the ORed voltage, averting nuisance resets in low voltage systems.
Enable pins can hold the MOSFET channel off—turning both off reduces device
current consumption. Status outputs
30 | April 2013 : LT Journal of Analog Innovation
indicate when the respective MOSFETs are
on. The LTC4353 is available in a compact
4mm × 3mm, 16-pin leadless DFN and a
16-pin MSOP package and operates over
a –40°C to 85°C temperature range.
PRECISION 50µV OFFSET OP AMP
OPERATES WITH 76V INPUT RANGE
The LT6016 and LT6017 are dual and quad
wide input range operational amplifiers.
These amplifiers combine high precision
with the ruggedness and versatility of
Linear Technology’s unique Over-TheTop® architecture. Input offset voltage
is 50µV max, input bias current is 5n A,
and low frequency noise is 0.5µVP–P,
making these devices suitable for a wide
range of precision industrial, automotive and instrumentation applications.
Over-The-Top inputs provide true
operation well beyond the V+ rail. The
LT6016/LT6017 function normally with
inputs up to 76V above V–, independent of
whether V+ is 3V or 50V. Additional faulttolerant features protect the op amps from
reverse supply conditions (up to –50V at
V+), negative transients (up to –25V at
VIN), and forced output voltage with no
power supplied (up to 50V at VOUT). This
robust architecture is especially useful for
applications where the amplifier is at the
analog interface to another board, and for
high side and low side current sensing.
The LT6016 and LT6017 are fully specified over –40°C to 85°C, –40°C to 125°C,
and –55°C to 150°C temperature ranges.
The dual LT6016 is available in an
8-lead MSOP package; the quad LT6017
in a 6mm × 3mm DFN package.
DUAL OUTPUT SINE WAVE TO LOGIC
CONVERTER UTILIZES SELECTABLE
INPUT FILTERING FOR LOWEST
ADDITIVE JITTER
The LTC6957 is a DC to 300MHz dual output buffer/driver/logic translator, ideal for
converting low frequency sine waves into
low phase noise logic level signals. Prior
solutions were unable to perform this conversion without introducing a significant
amount of jitter. The LTC6957 converts
any DC to 300MHz reference frequency
into dual LVPECL, LVDS or CMOS outputs
with exceptionally low additive jitter
of 45fSRMS (LTC6957-1) over 12kHz to
20MHz integration bandwidth and less
than 150fSRMS total jitter. The device also
features a proprietary, selectable, input
stage bandwidth-limiting feature, which
substantially improves the phase noise for
slow slewing signals by up to 3dB–4dB.
While the LTC6957 can be used to convert
any signal type to a logic level signal,
it particularly excels with sine waves.
The selectable, band-limited input stage
enables optimal conversion of sine waves
with the lowest additive jitter. The device
is ideal for systems that distribute system clock references for board level
synchronization. It can also be used as a
clock driver for analog-to-digital converters (ADCs), digital-to-analog converters
(DACs) or DDS (direct digital synthesis)
ICs with clock rates up to 300MHz.
The LTC6957 is offered in four output logic
signal types: the LTC6957-1 provides two
LVPECL outputs, the LTC6957-2 provides
two LVDS logic outputs, and the LTC6957-3
and LTC6957-4 offer two CMOS or
new product briefs
The LTC6957 converts any DC to 300MHz reference frequency into dual
LVPECL, LVDS or CMOS outputs with exceptionally low additive jitter of
45fSRMS (LTC6957-1) over 12kHz to 20MHz integration bandwidth and less than
150fSRMS total jitter.
complementary CMOS outputs, respectively,
with output skew as low as 2ps (typ). Each
device is available in small RoHS-compliant
12-pin MSOP or 3mm × 3mm DFN packages and can be ordered in industrial and
automotive grades, supporting operating temperature ranges from –40°C to
85°C and –40°C to 125°C, respectively.
WIDE V IN RANGE, LOW NOISE, 250mA
BUCK-BOOST CHARGE PUMP
The LTC3245 is a switched capacitor
buck-boost DC/DC converter that produces a regulated output (3.3V, 5V or
adjustable) from a 2.7V to 38V input.
The device uses switched capacitor
fractional conversion to maintain regulation over a wide range of input voltage.
Internal circuitry automatically selects
the conversion ratio to optimize efficiency as input voltage and load conditions vary. No inductors are required.
The unique constant frequency architecture provides a lower noise output than
conventional charge pump regulators.
To optimize efficiency at the expense of
slightly higher output ripple, the device
has pin-selectable Burst Mode operation. Low operating current (20µ A with
no load, 4µ A in shutdown) and low
external parts count (three small ceramic
capacitors) make the LTC3245 ideal for
low power, space constrained automotive
and industrial applications. The device
is short-circuit and overtemperature
protected, and is available in thermally
enhanced 12-pin MSOP and low profile
3mm × 4mm 12-pin DFN packages.
SPI/DIGITAL OR I 2C µMODULE
ISOLATOR PROVIDES THREE
ISOLATED POWER RAILS
The LTM2883 is a 6-channel SPI/Digital
or I2C digital µModule® isolator with
triple rail regulated power for 3.3V and
5V systems. In industrial systems applications, ground potentials can vary widely,
often exceeding the tolerable range, which
can interrupt communications or even
destroy components. The LTM2883 breaks
ground loops by electrically separating
communication signals, isolating the logic
level interface on each side of an internal
inductive isolation barrier that withstands
a very large common-mode voltage range
up to 2,500VRMS . The LTM2883’s low
EMI isolated DC/DC converter powers the
communications interface and provides
adjustable 5V, +12.5V, and –12.5V supply
outputs, ideal for powering data converters in data acquisition systems. With
2,500VRMS of galvanic isolation, onboard
secondary power and a communications
interface operating at up to 20Mbps, the
LTM2883 requires no external components
and provides a simple µModule solution for isolated data communications.
The LTM2883 is available in two communications interface versions. The LTM2883-I
is I2C compliant at up to 400kHz with bidirectional serial data (SDA) plus clock (SCL)
and three additional isolated CMOS logic
signals that operate at up to 20Mbps. The
LTM2883-S is SPI compliant and offers a
total of six CMOS digital isolator communication channels. All channels operate at
up to 20Mbps and include three forward
direction signals (CS, SCK and SDI) and three
reverse direction signals (SDO, DO1 and
DO2). When configured for SPI, SPI/Digital
or I2C µModule Isolator communications, the maximum clock rate is 8MHz for
unidirectional communication or 4MHz
for round-trip bidirectional operation.
An onboard 2MHz DC/DC converter powers the LTM2883 and allows each of the
three isolated power supply outputs to
source up to 20m A over the full operating temperature range. A logic supply
pin provides direct interfacing with low
voltage microcontrollers down to 1.62V,
and an ON pin enables the LTM2883
to be shut down using less than 10µ A.
Additional features include uninterrupted
communications for common mode
transients greater than 30kV/µs and rugged
±10kV ESD HBM across the isolation barrier.
The LTM2883 is available in 3.3V or
5V supply voltage versions. The LTM2883
is offered in a 15mm × 11.25mm surface
mount BGA package; all integrated circuits
and passive components are housed in
this RoHS-compliant µModule package. n
April 2013 : LT Journal of Analog Innovation | 31
highlights from circuits.linear.com
22nF
SOLAR POWERED CONVERTER WITH MPPC
CHARGES STORAGE CAPACITOR
The LTC3129 is a high efficiency, 200mA buck-boost DC/DC
converter with a wide VIN and VOUT range. It includes an
accurate RUN pin threshold to allow predictable regulator
turn-on and a maximum power point control (MPPC)
capability that ensures maximum power extraction from
nonideal power sources such as photovoltaic panels.
circuits.linear.com/612
BST1 SW1
UVLO = 4.3V
VIN
SW2 BST2
VOUT
VIN
1M
47µF
CERAMIC
VCC
V2: 14.8V Li-Ion
MAIN/SWAPPABLE
COOPER BUSSMANN
PB-5R0V105-R
PGOOD
PGOOD
VCC
NC
NC
GND
IRF7324
1M
FB
PWM
392k
1F
3.09M
RUN
MPPC
PowerFilm
SP4.2-37
SOLAR
MODULE
VOUT
4.8V
+
4.7µF
LTC3129
8.4cm × 3.7cm
V1: 12V
WALL ADAPTER
22nF
4.7µH
2.2µF
PGND
2A
OUTPUT
IRF7324
V3: 12V SLA
BACKUP
IRF7324
V1
VS1 G1
VS2 G2
806k
VS3 G3
VOUT
UV1
1M
39.2k
1M
1M
VALID1
OV1
VALID2
60.4k
VALID3
V2
1.05M
UV2
LTC4417
PRIORITY SWITCHING FROM 12V MAIN SUPPLY
TO 14.8V BATTERY BACKUP
The LTC4417 connects one of three valid power supplies to a common
output based on priority. Priority is defined by pin assignment, with V1
assigned the highest priority and V3 the lowest priority. A power supply
is defined as valid when its voltage has been within its overvoltage (OV)
and undervoltage (UV) window continuously for at least 256ms. If the
highest priority valid input falls out of the OV/UV window, the channel
is immediately disconnected and the next highest priority valid input
is connected to the common output. Two or more LTC4417s can be
cascaded to provide switchover between more than three inputs.
circuits.linear.com/617
LTspice IV
31.6k
circuits.linear.com/617
OV2
68.1k
V3
EN
SHDN
HYS
CAS
698k
UV3
16.9k
OV3
GND
49.9k
12V
CHARGE PUMP 12V TO ±5V SUPPLY
The LTC3260 can supply up to 100mA from the inverted input voltage at its charge
pump output, VOUT. VOUT also serves as the input supply to a negative LDO regulator,
LDO–. The charge pump frequency can be adjusted between 50kHz and 500kHz by a
single external resistor. The MODE pin is used to select between high efficiency Burst
Mode operation or constant frequency mode to satisfy low noise requirements.
circuits.linear.com/611
4.7µF
1µF
LTC3260
EN+
ADJ+
EN–
BYP+
MODE
GND
C+
BYP–
C–
–
ADJ
VOUT
LDO–
4.7µF
10nF
10nF
5V
I + ≤ 50mA
316k LDO
100k
100k
316k
–12V
10µF
LTspice IV
circuits.linear.com/611
LDO+
VIN
|IVOUT| ≤ 100mA – |ILDO–|
RT
4.7µF
–5V
|ILDO–| ≤ 50mA
200k
L, LT, LTC, LTM, Linear Technology, the Linear logo, Burst Mode, Dust Networks, LTspice, Over-The-Top, SmartMesh and µModule are registered trademarks, and PowerPath and LTPoE++ are trademarks of Linear Technology Corporation.
All other trademarks are the property of their respective owners.
© 2013 Linear Technology Corporation/Printed in U.S.A./60.3K
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