TPS65161 www.ti.com SLVS617 – APRIL 2006 BIAS POWER SUPPLY FOR TV AND MONITOR TFT LCD PANELS • • • • • • FEATURES • • • • • • • • 8-V to 14-V Input Voltage Range VS Output Voltage Range up to 19 V 1% Accurate Boost Converter With 2.8-A Switch Current 1.5% accurate 2.3-A Step-Down Converter 500-kHz/750-kHz Fixed Switching Frequency Negative Charge Pump Driver for VGL Positive Charge Pump Driver for VGH Adjustable Sequencing for VGL, VGH Gate Drive Signal to Drive External MOSFET Internal and Adjustable Soft Start Short-Circuit Protection Overvoltage Protection Thermal Shutdown Available in TSSOP-28 Package APPLICATIONS • TFT LCD Displays for Monitor and LCD TV DESCRIPTION The TPS65161 offers a compact power supply solution to provide all four voltages required by thin-film transistor (TFT) LCD panel. With its high current capabilities, the device is ideal for large screen monitor panels and LCD TV applications. TYPICAL APPLICATION C1 2 * 22 μF VGL −5 V/50 mA C16 1 μF 8 12 20 21 22 C6 0.47 μF D2 AVIN EN1 EN2 DRN FBN REF PGND PGND 28 SS 25 DLY1 R3 620 kΩ R4 150 kΩ C8 220 nF 16 9 11 13 24 6 7 SUP FREQ VINB VINB C9 22 nF C10 10 nF SW SW FB OS GND GD DRP FBP Boot SWB NC FBB COMP DLY2 C11 10 nF 4 5 1 3 23 27 Vs 18 V/1.3 A C4 22 pF TPS65161 C3 1 μF D3 C7 0.47 μF D1 SL22 L1 10 μH Vin 12 V R1 825 kΩ C15 470 nF C2 3 * 22 μF R2 56 kΩ D4 GD 10 14 17 18 19 15 C5 D5 0.47 μF VGH 32 V/50 mA R5 560 kΩ 2 26 Cb 100 nF C17 22 nF C13 0.47 μF R6 22 kΩ Vlogic 3.3 V/2.3 A L2 15 μH D6 SL22 R7 2 kΩ C14 10 nF C12 2 * 22 μF R8 1.2 kΩ Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2006, Texas Instruments Incorporated TPS65161 www.ti.com SLVS617 – APRIL 2006 DESCRIPTION (CONTINUED) Compared to the TPS65160 and TPS65160A the device offers a higher output current for the step down converter and allows to connect the positive charge pump supply (SUP) always to the output. Therefore in most applications a simple charge pump doubler can generates VGH reducing external component count. The device can be powered directly from a 12-V input voltage generating the bias voltages VGH and VGL, as well as the source voltage VS and logic voltage for the LCD panels. The device consists of a boost converter to provide the source voltage VS and a step-down converter to provide the logic voltage for the system. A positive and a negative charge-pump driver provide adjustable regulated output voltages VGL and VGH to bias the TFT. Both boost and step-down converters, as well as the charge-pump driver, operate with a fixed switching frequency of 500 kHz or 750 kHz, selectable by the FREQ pin. The TPS65161 includes adjustable power-on sequencing. The device includes safety features like overvoltage protection of the boost converter and short-circuit protection of the buck converter, as well as thermal shutdown. Additionally, the device incorporates a gate drive signal to control an isolation MOSFET switch in series with VS or VGH. See the application circuits at the end of this data sheet. This integrated circuit can be damaged by ESD. Texas Instruments recommends that all integrated circuits be handled with appropriate precautions. Failure to observe proper handling and installation procedures can cause damage. ESD damage can range from subtle performance degradation to complete device failure. Precision integrated circuits may be more susceptible to damage because very small parametric changes could cause the device not to meet its published specifications. ORDERING INFORMATION (1) TA UVLO (typ) OVERVOLTAGE PROTECTION Vs (typ) ORDERING PACKAGE (2) PACKAGE MARKING –40°C to 85°C 6V 20 V TPS65161PWP TSSOP28 (PWP) TPS65161 (1)For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI Web site at www.ti.com. (2) The PWP package is available taped and reeled. Add R-suffix to the device type (TPS65161PWPR) to order the device taped and reeled. The TPS65161PWPR package has quantities of 2000 devices per reel. Without suffix, the TPS65161PWP is shipped in tubes with 50 devices per tube. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (3) UNIT Voltages on pin VIN (4) –0.3 V to 16.5 V Voltages on pin EN1, EN2, FREQ (2) Voltage on pin SW –0.3 V to 15 V (2) 25 V Voltage on pin SWB (2) 20 V Voltages on pin OS, SUP, GD (2) 25 V Continuous power dissipation See Dissipation Rating Table TA Operating junction temperature –40°C to 150°C Tstg Storage temperature range –65°C to 150°C Temperature (soldering, 10 s) (3) (4) 260°C Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. All voltage values are with respect to network ground terminal. DISSIPATION RATINGS (5) 2 PACKAGE RTHJA TA ≤ 25°C POWER RATING TA = 70°C POWER RATING TA = 85°C POWER RATING 28-Pin TSSOP 28°C/W (PowerPAD (5) soldered) 3.57 W 1.96 W 1.42 W See Texas Instruments application report SLMA002 regarding thermal characteristics of the PowerPAD package. Submit Documentation Feedback TPS65161 www.ti.com SLVS617 – APRIL 2006 RECOMMENDED OPERATING CONDITIONS over operating free-air temperature range (unless otherwise noted) MIN NOM MAX Output voltage range of the main boost converter (6) VS Input capacitor at VINB CIN Input capacitor AVIN L VLOGIC CO UNIT 19 µF 1 µF Inductor boost converter (7) 10 Inductor buck converter (2) 15 Output voltage range of the step-down converter VLOGIC V 2x22 µH 1.8 5.0 Output capacitor boost converter 3x22 Output capacitor buck converter 2x22 V µF TA Operating ambient temperature –40 85 °C TJ Operating junction temperature –40 125 °C (6) (7) The maximum output voltage is limited by the overvoltage protection threshold and not be the maximum switch voltage rating. See application section for further information. ELECTRICAL CHARACTERISTICS VIN = 12 V, SUP = VIN, EN1 = EN2 = VIN, VS = 15 V, VLOGIC = 3.3 V, TA = –40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT SUPPLY CURRENT VIN Input voltage range 8 14 V 0.2 2 Quiescent current into AVIN VGH = 2 x VS, Boost converter not switching 0.2 0.5 Quiescent current into VINB VGH = 2 x VS, Buck converter not switching Shutdown current into AVIN EN1 = EN2 = GND 0.1 2 Shutdown current into VINB EN1 = EN2 = GND 0.1 2 Shutdown current into SUP EN1 = EN2 = GND 0.1 4 µA Quiescent current into SUP VGH = 2 x VS 0.2 2 mA VUVLO Undervoltage lockout threshold VIN falling VREF Reference voltage IQIN ISD ISUP Thermal shutdown 1.203 Temperature rising Thermal shutdown hysteresis 6 6.4 1.213 1.223 mA µA V V 155 °C 5 °C LOGIC SIGNALS EN1, EN2, FREQ VIH High-level input voltage EN1, EN2 VIL Low-level input voltage EN1, EN2 VIH High-level input voltage FREQ VIL Low-level input voltage FREQ II Input leakage current 2.0 V 0.8 1.7 V V 0.4 V 0.01 0.1 µA 3.3 4.8 6.2 µA 3.3 4.8 6.2 µA 6 9 12 µA FREQ = high 600 750 900 FREQ = low 400 500 600 EN1 = EN2 = FREQ = GND or VIN CONTROL AND SOFT START DLY1, DLY2, SS IDLY1 Delay1 charge current IDLY2 Delay2 charge current ISS SS charge current VTHRESHOLD = 1.213 V INTERNAL OSCILLATOR fOSC Oscillator frequency Submit Documentation Feedback kHz 3 TPS65161 www.ti.com SLVS617 – APRIL 2006 ELECTRICAL CHARACTERISTICS (continued) VIN = 12 V, SUP = VIN, EN1 = EN2 = VIN, VS = 15 V, VLOGIC = 3.3 V, TA = –40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT 1.136 1.146 1.156 V 10 100 nA BOOST CONVERTER (VS) VS Output voltage range (8) VFB Feedback regulation voltage IFB Feedback input bias current rDS(ON) 19 V N-MOSFET on-resistance (Q1) ISW = 500 mA 100 185 mΩ P-MOSFET on-resistance (Q2) ISW = 200 mA 10 16 Ω 1 A IMAX Maximum P-MOSFET peak switch current ILIM N-MOSFET switch current limit (Q1) Ileak Switch leakage current VSW = 15 V Vovp Overvoltage protection VOUT rising Line regulation 10.6 V ≤ Vin ≤ 11.6 V at 1 mA 2.8 3.5 4.2 A 1 10 µA 19.5 20 21 Load regulation V 0.0008 %/V 0.03 %/A GATE DRIVE (GD) VGD Gate drive threshold (9) VFB rising VOL GD output low voltage I(sink) = 500 µA GD output leakage current VGD = 20 V Vs-12% Vs-8% Vs-4% V 0.3 V 0.05 1 µA 5 V 1.213 1.231 V 10 100 nA 175 300 mΩ 3.2 3.9 A 1 10 µA STEP-DOWN CONVERTER (VLOGIC) VLOGIC Output voltage range VFBB Feedback regulation voltage 1.8 IFBB Feedback input bias current rDS(ON) N-MOSFET on-resistance (Q1) ILIM N-MOSFET switch current limit (Q1) Ileak Switch leakage current VSW = 0 V Line regulation 10.6 V ≤ VIN≤ 11.6 V at 1 mA 1.195 ISW = 500 mA 2.5 Load regulation (8) (9) 4 0.0018 %/V 0.037 %/A The maximum output voltage is limited by the overvoltage protection threshold and not be the maximum switch voltage rating. The GD signal is latched low when the main boost converter output VS is within regulation. The GD signal is reset when the input voltage or enable of the boost converter is cycled low. Submit Documentation Feedback TPS65161 www.ti.com SLVS617 – APRIL 2006 ELECTRICAL CHARACTERISTICS (continued) VIN = 12 V, SUP = VIN, EN1 = EN2 = VIN, VS = 15 V, VLOGIC = 3.3 V, TA = –40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT –2 V –36 0 36 mV 10 100 nA NEGATIVE CHARGE-PUMP VGL VGL Output voltage range VFBN Feedback regulation voltage IFBN Feedback input bias current rDS(ON) Q4 P-Channel switch rDS(ON) VDropN Current sink voltage drop (10) Ω IOUT = 20 mA 4.4 IDRN = 50 mA, VFBN = VFBNnominal –5% 130 190 IDRN = 100 mA, VFBN = VFBNnominal –5% 270 420 1.213 1.238 V 10 100 nA mV POSITIVE CHARGE-PUMP OUTPUT VGH VFBP Feedback regulation voltage IFBP Feedback input bias current rDS(ON) Q3 N-Channel switch rDS(ON) VDropP 1.187 Current source voltage drop (Vsup – VDRP) (3) Ω IOUT = 20 mA 1.1 IDRP = 50 mA, VFBP = VFBPnominal –5% 400 680 IDRP = 100 mA, VFBP = VFBPnominal –5% 850 1600 mV (10) The maximum charge-pump output current is typically half the drive current of the internal current source or current sink. 1 28 SS COMP 2 27 GD OS 3 26 DLY2 SW 4 25 DLY1 SW 5 24 REF PGND 6 23 GND PGND 7 22 AVIN SUP 8 21 VINB EN2 9 20 VINB DRP 10 19 NC DRN 11 18 SWB FREQ 12 17 BOOT FBN 13 16 EN1 FBP 14 15 FBB Thermal PAD (see Note) FB NOTE: The thermally enhanced PowerPAD™ is connected to PGND. Submit Documentation Feedback 5 TPS65161 www.ti.com SLVS617 – APRIL 2006 TERMINAL FUNCTIONS TERMINAL NAME NO. DESCRIPTION SUP 8 I This is the supply pin of the positive charge pump driver and can be connected to the input supply Vin or the output of the main boost converter Vs. This depends mainly on the desired output voltage VGH and numbers of charge pump stages. FREQ 12 I Frequency adjust pin. This pin allows setting the switching frequency with a logic level to 500 kHz = low and 750 kHz = high. AVIN 22 I Analog input voltage of the device. This is the input for the analog circuits of the device and should be bypassed with a 1-µF ceramic capacitor for good filtering. VINB 20, 21 I Power input voltage pin for the buck converter. EN1 16 I This is the enable pin of the buck converter and negative charge pump. When this pin is pulled high, the buck converter starts up, and after a delay time set by DLY1, the negative charge pump comes up. This pin must be terminated and not be left floating. A logic high enables the device and a logic low shuts down the device. EN2 9 I The boost converter starts only with EN1 = high, after the step-down converter is enabled. EN2 is the enable pin of the boost converter and positive charge pump. When this pin is pulled high, the boost converter and positive charge pump starts up after the buck converter is within regulation and a delay time set by DLY2 has passed by. This pin must be terminated and not be left floating. A logic high enables the device and a logic low shuts down the device. DRN 11 O Drive pin of the negative charge pump. FBN 13 I Feedback pin of negative charge pump. REF 24 O Internal reference output typically 1.213 V PGND 6, 7 SS 28 O This pin allows setting the soft-start time for the main boost converter VS. Typically a 22-nF capacitor needs to be connected to this pin to set the soft-start time. DLY1 25 O Connecting a capacitor from this pin to GND allows the setting of the delay time between VLOGIC (step-down converter output high) to VGL during start-up. DLY2 26 O Connecting a capacitor from this pin to GND allows the setting of the delay time between VLOGIC (step-down converter output high) to VS Boost converter and positive charge-pump VGH during start-up. COMP 2 FBB 15 I Feedback pin of the buck converter SWB 18 O Switch pin of the buck converter NC 19 BOOT 17 FBP DRP GD 27 This is the gate drive pin which can be used to control an external MOSFET switch to provide input to output isolation of VS or VGH. See the circuit diagrams at the end of this data sheet. GD is an open-drain output and is latched low as soon as the boost converter is within 8% of its nominal regulated output voltage. GD goes high impedance when the EN2 input voltage is cycled low. GND 23 Analog ground OS 3 I Output sense pin. The OS pin is connected to the internal rectifier switch and overvoltage protection comparator. This pin needs to be connected to the output of the boost converter and cannot be connected to any other voltage rail. Connect a 470-nF capacitor from OS pin to GND to avoid noise coupling into this pin. The PCB trace of the OS pin needs to be wide because it conducts high current. FB 1 I Feedback of the main boost converter generating Vsource (VS). SW 4, 5 I Switch pin of the boost converter generating Vsource (VS). PowerPAD 6 I/O Power ground This is the compensation pin for the main boost converter. A small capacitor and, if required, a resistor is connected to this pin. Not connected I N-channel MOSFET gate drive voltage for the buck converter. Connect a capacitor from the switch node SWB to this pin. 14 I Feedback pin of positive charge pump. 10 O Drive pin of the positive charge pump. The PowerPAD needs to be connected and soldered to power ground (PGND). Submit Documentation Feedback TPS65161 www.ti.com SLVS617 – APRIL 2006 TYPICAL CHARACTERISTICS TABLE OF GRAPHS FIGURE MAIN BOOST CONVERTER (Vs) η Efficiency main boost converter Vs vs Load current VS =15 V,VIN = 12 V 1 rDS(ON) N-channel main switch Q1 vs Input voltage and temperature 2 Soft-start boost converter CSS = 22 nF 3 PWM operation at full-load current 4 PWM operation at light-load current 5 Load transient response 6 STEP-DOWN CONVERTER (Vlogic) η Efficiency main boost converter VS rDS(ON) N-channel main switch Q1 8 PWM operation - continuous mode 9 PWM operation - discontinuous mode 10 Soft start 11 Load transient response 12 vs Load current VLOGIC = 3.3 V,VIN = 12 V 7 SYSTEM PERFORMANCE fosc Oscillation frequency vs Input voltage and temperature 13 Power-up sequencing EN2 connected to VIN 14 Power-up sequencing EN2 enabled separately 15 BOOST CONVERTER EFFICIENCY vs OUTPUT CURRENT BOOST CONVERTER rDS(ON) - N-CHANNEL SWITCH vs TEMPERATURE 100 0.16 90 0.14 rDS(on) − N-Channel Switch − Ω 80 Efficiency − % 70 60 50 40 30 VI = 12 V, VO = 15 V, L = 10 H 20 0.5 1 1.5 IO − Output Current − A 0.12 0.1 0.08 0.06 0.04 0.02 10 0 0 VI = 8 V, VI = 12 V, VI = 14 V 2 0 −40 −20 Figure 1. 0 20 40 60 80 100 120 140 TA − Temperature − C Figure 2. Submit Documentation Feedback 7 TPS65161 www.ti.com SLVS617 – APRIL 2006 SOFT-START BOOST CONVERTER PWM OPERATION BOOST CONVERTER CONTINUOUS MODE VI = 12 V, VO = 15 V/1.5 A VI = 12 V, VO = 15 V/ 1.2 A, C(SS) = 22 nF VSW 10 V/div VO 50 mV/div VS 5 V/div I(Inductor) 1 A/div II 1 A/div 1 s/div 2 ms/div Figure 3. Figure 4. PWM OPERATION BOOST CONVERTER CONTINUOUS MODE: LIGHT LOAD LOAD TRANSIENT RESPONSE BOOST CONVERTER VI = 12 V, VS = 15 V, CO = 3*22 F, C(comp) = 22 nF, L = 6.8 H, FREQ= High VI = 12 V , VO = 10 V/10 mA VSW 10 V/div VS 200 mV/div VO 50 mV/div I(Inductor) 1 A/div IL 500 mA/div 100 s/div 1 µs/div Figure 5. 8 Figure 6. Submit Documentation Feedback TPS65161 www.ti.com SLVS617 – APRIL 2006 EFFICIENCY STEP-DOWN CONVERTER vs LOAD CURRENT STEP-DOWN CONVERTER rDS(ON) - N-CHANNEL SWITCH vs TEMPERATURE 90 0.25 VI = 8 V, VI = 12 V, VI = 14 V r DS(on) − N-Channel Switch − Ω 80 70 Efficiency − % 60 50 40 30 20 VI = 12 V, VO = 3.3 V, L = 15 H 10 0 0 0.5 1 1.5 IO − Output Current − A 2 0.2 0.15 0.1 0.05 0 −40 −20 0 20 40 60 80 100 120 140 TA − Temperature − C Figure 7. Figure 8. STEP-DOWN CONVERTER PWM OPERATION CONTINUOUS MODE STEP-DOWN CONVERTER PWM OPERATION DISCONTINUOUS MODE VI = 12 V, VO = 3.3 V/45 mA VSW 5 V/div VSW 5 V/div VO 20 mV/div VO 20 mV/div VI = 12 V, VO = 3.3 V/1.5 A I(Inductor) 1 A/div I(Inductor) 100 mA/div 500 ns/div 500 ns/div Figure 9. Figure 10. Submit Documentation Feedback 9 TPS65161 www.ti.com SLVS617 – APRIL 2006 SOFT-START STEP-DOWN CONVERTER LOAD TRANSIENT RESPONSE STEP-DOWN CONVERTER VI = 12 V, VO = 3.3 V/1.2 A VO1 100 mV/div VI = 12 V, V(logic) = 3.3 V, CO = 2*22 F, FREQ = High VO 1 V/div IO 270 mA to 1.3 A I(Inductor) 1 A/div 50 s/div 200 s/div Figure 11. Figure 12. SWITCHING FREQUENCY vs TEMPERATURE POWER-UP SEQUENCING EN2 CONNECTED TO VIN 740 Switching Frequency − kHz 735 VI = 8 V, VI = 12 V, VI = 14 V 730 V Logic 2 V/div 725 VGL 5 V/div 720 715 710 VS 5 V/div 705 VGH 10 V/div 700 695 −50 0 50 100 TA − Temperature − C 150 Figure 13. 10 2 ms/div Figure 14. Submit Documentation Feedback TPS65161 www.ti.com SLVS617 – APRIL 2006 POWER-UP SEQUENCING EN2 ENABLED SEPARATELY V Logic 2 V/div VS 5 V/div VGH 5 V/div EN2 2 V/div 1 ms/div Figure 15. Submit Documentation Feedback 11 TPS65161 www.ti.com SLVS617 – APRIL 2006 BLOCK DIAGRAM AVIN SW SW Q2 D SS Bias Vref=1.213 V Thermal Shutdown Sequencing GND FREQ 500 kHz/ 750 kHz Oscillator Clock DLY1 Current Limit and Soft Start DLY2 S AVIN OS OS IDLY Overvoltage Comparator SS Vref Control Logic SS AVIN Vref D S COMP PGND Q1 PGND GM Amplifier Comparator Sawtooth Generator FB VFB 1.154 V Positive Charge Pump OS SUP SUP I DRVP GM Amplifier Low Gain VFB 1.154 Current Control Soft Start Vref 1.2 13 V DRVP Q3 VINB VINB Negative Charge Pump FBP Step-Down Converter Current Control Soft Start DRVN N Regulator 8V BOOT I DRVN D Q3 S AVIN SWB Control Logic FBN NC Current Limit Ref IDLY DLY1 DLY1 Vref Compensation and Soft Start Clock/2 IDLY GD 0.9 V Vref Logic Clock/4 DLY2 DLY2 FBB Error Amplifier Vref Clock 0.6 V Reference Output Clock D Vref 1.2 13 V Clock Select During Short Circuit and Soft Start S EN2 EN1 12 Sawtooth Generator Submit Documentation Feedback VREF TPS65161 www.ti.com SLVS617 – APRIL 2006 DETAILED DESCRIPTION Boost Converter The main boost converter operates in pulse-width modulation (PWM) and at a fixed switching frequency of 500 kHz or 750 kHz set by the FREQ pin. The converter uses an unique fast response, voltage-mode controller scheme with input voltage feedforward. This achieves excellent line and load regulation (0.03%-A load regulation typical) and allows the use of small external components. To add higher flexibility to the selection of external component values, the device uses external loop compensation. Although the boost converter looks like a nonsynchronous boost converter topology operating in discontinuous conduction mode at light load, the TPS65161 maintains continuous conduction even at light-load currents. This is achieved with a novel architecture using an external Schottky diode with an integrated MOSFET in parallel connected between SW and OS. See the Functional Block Diagram. The intention of this MOSFET is to allow the current to go negative that occurs at light-load conditions. For this purpose, a small integrated P-Channel MOSFET with typically 10-Ω rds(on) is sufficient. When the inductor current is positive, the external Schottky diode with the lower forward voltage conducts the current. This causes the converter to operate with a fixed frequency in continuous conduction mode over the entire load current range. This avoids the ringing on the switch pin as seen with standard nonsynchronous boost converter and allows a simpler compensation for the boost converter. Soft Start (Boost Converter) The main boost converter has an adjustable soft start to prevent high inrush current during start-up. The soft-start time is set by the external capacitor connected to the SS pin. The capacitor connected to the SS pin is charged with a constant current that increases the voltage on the SS pin. The internal current limit is proportional to the voltage on the soft-start pin. When the threshold voltage of the internal soft-start comparator is reached, the full current limit is released. The larger the soft-start capacitor value, the longer the soft-start time. Overvoltage Protection of the Boost Converter The main boost converter has an overvoltage protection to protect the main switch Q2 at pin (SW) in case the feedback (FB) pin is floating or shorted to GND. In such an event, the output voltage rises and is monitored with the overvoltage protection comparator over the OS pin. See the functional block diagram. As soon as the comparator trips at typically 23 V, TPS65161, the boost converter turns the N-Channel MOSFET switch off. The output voltage falls below the overvoltage threshold and the converter continues to operate. Frequency Select Pin (FREQ) The frequency select pin (FREQ) allows setting the switching frequency of the entire device to 500 kHz (FREQ = low) or 750 kHz (FREQ = high). A lower switching frequency gives a higher efficiency with a slightly reduced load transient regulation. Thermal Shutdown A thermal shutdown is implemented to prevent damage caused by excessive heat and power dissipation. Typically, the thermal shutdown threshold is 155°C. Step-Down Converter The nonsynchronous step-down converter operates at a fixed switching frequency using a fast response voltage mode topology with input voltage feedforward. This topology allows simple internal compensation, and it is designed to operate with ceramic output capacitors. The converter drives an internal 3.2-A N-channel MOSFET switch. The MOSFET driver is referenced to the switch pin SWB. The N-channel MOSFET requires a gate drive voltage higher than the switch pin to turn the N-Channel MOSFET on. This is accomplished by a bootstrap gate drive circuit running of the step-down converter switch pin. When the switch pin SWB is at ground, the bootstrap capacitor is charged to 8 V. This way, the N-channel gate drive voltage is typically around 8 V. Submit Documentation Feedback 13 TPS65161 www.ti.com SLVS617 – APRIL 2006 DETAILED DESCRIPTION (continued) Soft Start (Step-Down Converter) To avoid high inrush current during start-up, an internal soft start is implemented in the TPS65161. When the step-down converter is enabled over EN1, its reference voltage slowly rises from zero to its power-good threshold of typically 90% of Vref. When the reference voltage reaches this power-good threshold, the error amplifier is released to its normal operation at its normal duty cycle. To further limit the inrush current during soft start, the converter frequency is set to 1/4th of the switching frequency fs and then ½ of fs determined by the comparator that monitors the feedback voltage. See the internal block diagram. Soft start is typically completed within 1 ms. Short-Circuit Protection (Step-Down Converter) To limit the short-circuit current, the device has a cycle-by-cycle current limit. To avoid the short-circuit current rising above the internal current limit when the output is shorted to GND, the switching frequency is reduced as well. This is implemented by two comparators monitoring the feedback voltage. The step-down converter switching frequency is reduced to ½ of fs when the feedback is below 0.9 V and to 1/4th of the switching frequency when the feedback voltage is below 0.6 V. Positive Charge Pump The positive charge pump provides a regulated output voltage set by the external resistor divider. Figure 16 shows an extract of the positive charge-pump driver circuit. The operation of the charge-pump driver can be understood best with Figure 16. During the first cycle, Q3 is turned on and the flying capacitor Cfly charges to the source voltage, Vs. During the next clock cycle, Q3 is turned off and the current source charges the drive pin, DRP, up to the supply voltage, VSUP. Because the flying capacitor voltage sits on top of the drive pin voltage, the maximum output voltage is Vsup+Vs. The SUP pin can be connected either to the input voltage Vin of the TPS65161 or the output voltage of the main boost converter Vs. SUP = Vin or Vs Vs I DRVP Current DRP Cfly VG VGH 23 V/50 mA Control Soft Start Q3 R5 FBP R6 Figure 16. Extract of the Positive Charge-Pump Driver If higher output voltages are required, another charge-pump stage can be added to the output. Setting the output voltage: 14 Submit Documentation Feedback C13 0.47 µF TPS65161 www.ti.com SLVS617 – APRIL 2006 DETAILED DESCRIPTION (continued) V out 1.213 1 R5 R6 R5 R6 Vout V FB 1 R6 V out 1 1.213 Negative Charge Pump The negative charge pump provides a regulated output voltage set by the external resistor divider. The negative charge pump operates similar to the positive charge pump with the difference that it runs from the input voltage VIN. The negative charge pump driver inverts the input voltage. The maximum negative output voltage is VGL = (–VIN)+Vdrop. Vdrop is the voltage drop across the external diodes and internal charge-pump MOSFETs. In case VGL needs to be lower than –VIN, an additional charge-pump stage needs to be added. Setting the output voltage: V out V R3 1.213 V R3 REF R4 R4 |Vout| |Vout| R3 R4 R4 1.213 V REF The lower feedback resistor value, R4, should be in a range between 40 kΩ to 120 kΩ or the overall feedback resistance should be within 500 kΩ to 1 MΩ. Smaller values load the reference too heavily, and larger values may cause stability problems. The negative charge pump requires two external Schottky diodes. The peak current rating of the Schottky diode has to be twice the load current of the output. For a 20-mA output current, the dual-Schottky diode BAV99 is a good choice. Power-On Sequencing (EN1, EN2, DLY1, DLY2) The TPS65161 has an adjustable power-on sequencing set by the capacitors connected to DLY1 and DLY2 and controlled by EN1 and EN2. Pulling EN1 high enables the step-down converter and then the negative charge-pump driver. DLY1 sets the delay time between the step-down converter and negative charge-pump driver. EN2 enables the boost converter and positive charge-pump driver at the same time. DLY2 sets the delay time between the step-down converter Vlogic and the boost converter Vs. This is especially useful to adjust the delay when EN2 is always connected to Vin. If EN2 goes high after the step-down converter is already enabled, then the delay DLY2 starts when EN2 goes high. See Figure 17 and Figure 18. EN2 EN1 DLY2 VGH V s, VGH Vs Vin Vin VLogic Fall Time Depends on Load Current and Feedback Resistor VGL DLY1 GD Figure 17. Power-On Sequencing With EN2 Always High (EN2=Vin) Submit Documentation Feedback 15 TPS65161 www.ti.com SLVS617 – APRIL 2006 DETAILED DESCRIPTION (continued) EN2 EN1 DLY2 VGH Vs VGH ,Vs Vin Vin Fall Time Depends on Load Current and Feedback Resistor VLogic DLY1 VGL GD Figure 18. Power-On Sequencing Using EN1 and EN2 Setting the Delay Times DLY1, DLY2 Connecting an external capacitor to the DLY1 and DLY2 pins sets the delay time. If no delay time is required, these pins can be left open. To set the delay time, the external capacitor connected to DLY1 and DLY2 is charged with a constant current source of typically 4.8 µA. The delay time is terminated when the capacitor voltage has reached the internal reference voltage of Vref = 1.213 V. The external delay capacitor is calculated: 4.8 A td 4.8 A td C with td Desired delay time dly Vref 1.213 V Example for setting a delay time of 2.3 ms: 4.8 A 2.3 ms C 9.4 nF Cdly 10 nF dly 1.213 V Gate Drive Pin (GD) This is an open-drain output that goes low when the boost converter, Vs, is within regulation. The gate drive pin GD remains low until the input voltage or enable EN2 is cycled to ground. Undervoltage Lockout To avoid misoperation of the device at low input voltages, an undervoltage lockout is included which shuts down the device at voltages lower than 6 V. Input Capacitor Selection For good input voltage filtering, low ESR ceramic capacitors are recommended. The TPS65161 has an analog input, AVIN, and two input pins for the buck converter VINB. A 1-µF input capacitor should be connected directly from the AVIN to GND. Two 22-µF ceramic capacitors are connected in parallel from the buck converter input VINB to GND. For better input voltage filtering, the input capacitor values can be increased. See Table 1 and the Application Information section for input capacitor recommendations. 16 Submit Documentation Feedback TPS65161 www.ti.com SLVS617 – APRIL 2006 DETAILED DESCRIPTION (continued) Table 1. Input Capacitor Selection CAPACITOR VOLTAGE RATING COMPONENT SUPPLIER COMMENTS 22 µF/1210 16 V Taiyo Yuden EMK325BY226MM CIN (VINB) 1 µF/1206 16 V Taiyo Yuden EMK316BJ106KL CIN (AVIN) Boost Converter Design Procedure The first step in the design procedure is to verify whether the maximum possible output current of the boost converter supports the specific application requirements. A simple approach is to use the converter efficiency, by taking the efficiency numbers from the provided efficiency curves or to use a worst-case assumption for the expected efficiency, e.g., 80%. 1. Duty Cycle: D 1 Vin Vout Vin 2. Maximum output current: I avg (1 D) lsw Vout 2.8 A with lsw minimum switch current of the TPS65161 (2.8 A). 3. Peak switch current: I swpeak I Vin D out 2 ƒs L 1 D With Isw = converter switch current (minimum switch current limit = 2.8 A) fs = converter switching frequency (typical 500 kHz/750 kHz) L = Selected inductor value η = Estimated converter efficiency (use the number from the efficiency curves or 0.8 as an estimation) The peak switch current is the steady-state peak switch current that the integrated switch, inductor, and external Schottky diode must be able to handle. The calculation must be done for the minimum input voltage where the peak switch current is highest. Inductor Selection (Boost Converter) The TPS65161 operates typically with a 10-µH inductor. Other possible inductor values are 6.8-µH or 22-µH. The main parameter for the inductor selection is the saturation current of the inductor, which should be higher than the peak switch current as previously calculated, with additional margin to cover for heavy load transients. The alternative, more conservative approach, is to choose the inductor with saturation current at least as high as the typical switch current limit of 3.5 A. The second important parameter is the inductor DC resistance. Usually, the lower the DC resistance the higher the efficiency. The efficiency difference between different inductors can vary between 2% to 10%. Possible inductors are shown in Table 2. Table 2. Inductor Selection (Boost Converter) INDUCTOR VALUE DIMENSIONS in mm Isat/DCR 22 µH Coilcraft MSS1038-103NX COMPONENT SUPPLIER 10,2 x 10,2 x 3,6 2.9 A/73 mΩ 22 µH Coilcraft DO3316-103 12,85 x 9,4 x 5,21 3.8 A/38 mΩ 10 µH Sumida CDRH8D43-100 8,3 x 8,3 x 4,5 4.0 A/29 mΩ 10 µH Sumida CDH74-100 7,3 x 8,0 x 5,2 2.75 A/43 mΩ 10 µH Coilcraft MSS1038-103NX 10,2 x 10,2 x 3,6 4.4 A/35 mΩ 6.8 µH Wuerth Elektronik 7447789006 7,3 x 7,3 x 3,2 2.5 A/44 mΩ Submit Documentation Feedback 17 TPS65161 www.ti.com SLVS617 – APRIL 2006 Output Capacitor Selection (Boost Converter) For best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low ESR value and work best with the TPS65161. Usually, three 22-µF ceramic output capacitors in parallel are sufficient for most applications. If a lower voltage drop during load transients is required, more output capacitance can be added. See Table 3 for the selection of the output capacitor. Table 3. Output Capacitor Selection (Boost Converter) CAPACITOR VOLTAGE RATING 22 µF/1812 16 V COMPONENT SUPPLIER Taiyo Yuden EMK432BJ226MM Rectifier Diode Selection (Boost Converter) To achieve high efficiency, a Schottky diode should be used. The reverse voltage rating should be higher than the maximum output voltage of the converter. The average rectified forward-current rating needed for the Schottky diode is calculated as the off-time of the converter times the maximum switch current of the TPS65161: D 1 Vout Vin I avg (1 D) lsw Vin 2.8 A with lsw minimum switch current of the TPS65161 (2.8 A). Vout Usually, a Schottky diode with 2-A maximum average rectified forward-current rating is sufficient for most applications. Secondly, the Schottky rectifier has to be able to dissipate the power. The dissipated power is the average rectified forward current times the diode forward voltage. PD = Iavg x VF = Isw x (1 x D) x VF (with Isw = minimum switch current of the TPS65161 (2.6 A) Table 4. Rectifier Diode Selection (Boost Converter) CURRENT RATING Iavg Vr 3A 20 V 2A 20 V 2A 20 V Vforward RθJA SIZE COMPONENT SUPPLIER 0.36 at 3 A 46°C/W SMC MBRS320, International Rectifier 0.44 V at 3 A 75°C/W SMB SL22, Vishay Semiconductor 0.5 at 2 A 75°C/W SMB SS22, Fairchild Semiconductor Setting the Output Voltage and Selecting the Feedforward Capacitor (Boost Converter) The output voltage is set by the external resistor divider and is calculated as: V out 1.146 V 1 R1 R2 Across the upper resistor, a bypass capacitor is required to achieve a good load transients response and to have a stable converter loop. Together with R1, the bypass capacitor Cff sets a zero in the control loop. Depending on the inductor value, the zero frequency needs to be set. For a 6.8-µH or 10-µH inductor, fz = 10 kHz and for a 22-µH inductor, fz = 7 kHz. 1 1 Cƒƒ 2 ƒ z R1 2 10 kHz R1 A value coming closest to the calculated value should be used. Compensation (COMP) (Boost Converter) The regulator loop can be compensated by adjusting the external components connected to the COMP pin. The COMP pin is the output of the internal transconductance error amplifier. A single capacitor connected to this pin sets the low-frequency gain. Usually, a 22-nF capacitor is sufficient for most of the applications. Adding a series resistor sets an additional zero and increases the high-frequency gain. The following formula calculates at what frequency the resistor increases the high-frequency gain. 18 Submit Documentation Feedback TPS65161 www.ti.com SLVS617 – APRIL 2006 ƒz 1 2 Cc Rc Lower input voltages require a higher gain and therefore a lower compensation capacitor value. Step-Down Converter Design Procedure Setting the Output Voltage The step-down converter uses an external voltage divider to set the output voltage. The output voltage is calculated as: V out 1.213 V 1 R1 R2 with R1 as 1.2 kΩ, and internal reference voltage V(ref)typ = 1.213 V At load current <1 mA, the device operates in discontinuous conduction mode. When the load current is reduced to zero, the output voltage rises slightly above the nominal output voltage. At zero load current, the device skips clock cycles but does not completely stop switching; thus, the output voltage sits slightly higher than the nominal output voltage. Therefore, the lower feedback resistor is selected to be around 1.2 kΩ to always have around 1-mA minimum load current. Selecting the Feedforward Capacitor The feedforward capacitor across the upper feedback resistor divider sets a zero in the converter loop transfer function. For a 15-µH inductor, fz = 8 kHz and when a 22-µH inductor is used, fz = 17 kHz. (Example for the 3.3-V output) Cz 1 1 9.9 nF 10 nF 2 8 kHz R1 2 8 kHz 2k Usually a capacitor value closest to the calculated value is selected. Inductor Selection (Step-Down Converter) The TPS65161 operates typically with a 15-µH inductor value. For high efficiencies the inductor should have a low DC resistance to minimize conduction losses. This needs to be considered when selecting the appropriate inductor. In order to avoid saturation of the inductor, the inductor should be rated at least for the maximum output current of the converter, plus the inductor ripple current that is calculated as: 1 Vout I Vin I I outmax L I Vout Lmax L 2 Lƒ With: f = Switching frequency (750 kHz, 500 kHz minimal) L = Inductor value (typically 15 µH) ∆IL= Peak-to-peak inductor ripple current ILmax = Maximum inductor current The highest inductor current occurs at maximum Vin. A more conservative approach is to select the inductor current rating just for the typical switch current of 3.2 A. Table 5. Inductor Selection (Step-Down Converter) INDUCTOR VALUE COMPONENT SUPPLIER DIMENSIONS in mm Isat/DCR 15 µH Wuerth 7447789115 7,3 x 7,3 x 3,2 1.75 A/100 mΩ 15 µH Sumida CDRH8D28-150 8,3 x 8,3 x 3,0 1.9 A/53 mΩ 15 µH Sumida CDRH8D38-150 8,3 x 8,3 x 4,0 2.3 A/53 mΩ 15 µH Coilcraft MSS1038-153NX 10,2 x 10,2 x 3,6 2.7 A/50 mΩ Submit Documentation Feedback 19 TPS65161 www.ti.com SLVS617 – APRIL 2006 Rectifier Diode Selection (Step-Down Converter) To achieve high efficiency, a Schottky diode should be used. The reverse voltage rating should be higher than the maximum output voltage of the step-down converter. The averaged rectified forward current at which the Schottky diode needs to be rated is calculated as the off-time of the step-down converter times the maximum switch current of the TPS65161: D 1 Vout Vin I avg (1 D) Isw 1 Vout 2.5 A with Isw minimum switch current of the TPS65161 (2.5 A) Vin Usually, a Schottky diode with 1.5-A or 2-A maximum average rectified forward current rating is sufficient for most applications. Secondly, the Schottky rectifier has to be able to dissipate the power. The dissipated power is the average rectified forward current times the diode forward voltage. PD = Iavg x VF = Isw x (1 – D) x VF with Isw = minimum switch current of the TPS65161 (2 A). Table 6. Rectifier Diode Selection (Step-Down Converter) CURRENT RATING Iavg Vr 3A 20 V 2A 20 V 2A 1.5 A Vforward RθJA SIZE COMPONENT SUPPLIER 0.36 V at 3 A 46°C/W SMC MBRS320, International Rectifier 0.44 V at 2 A 75°C/W SMB SL22, Vishay Semiconductor 20 V 0.5 V at 2 A 75°C/W SMB SS22, Fairchild Semiconductor 20 V 0.445 V at 1.0 A 88°C/W SMA SL12, Vishay Semiconductor Output Capacitor Selection (Step-Down Converter) The device is designed to work with ceramic output capacitors. When using a 15-µH inductor, two 22-µF ceramic output capacitors are recommended. More capacitance can be added to improve the load transient response. Table 7. Output Selection (Boost Converter) CAPACITOR VOLTAGE RATING 22 µF/0805 6.3 V COMPONENT SUPPLIER Taiyo Yuden JMK212BJ226MG Layout Consideration The PCB layout is an important step in the power supply design. An incorrect layout could cause converter instability, load regulation problems, noise, and EMI issues. Especially with a switching dc-dc converter at high load currents, too-thin PCB traces can cause significant voltage spikes. Good grounding becomes important as well. If possible, a common ground plane to minimize ground shifts between analog (GND) and power ground (PGND) is recommended. Additionally, the following PCB design layout guidelines are recommended for the TPS65161: 1. Separate the power supply traces for AVIN and VINB, and use separate bypass capacitors. 2. Use a short and wide trace to connect the OS pin to the output of the boost converter. 3. To minimize noise coupling into the OS pin, use a 470-pF bypass capacitor to GND. 4. Place the rectifier diode of the step down converter as close as possible to the SWB pin. 5. Use short traces for the charge-pump drive pins (DRN, DRP) of VGH and VGL because these traces carry switching waveforms. 6. Place a 1-µF bypass capacitor from the SUP pin to GND. 7. Place the flying capacitors as close as possible to the DRP and DRN pin, avoiding a high voltage spike at these pins. 8. Place the Schottky diodes as close as possible to the IC, respective to the flying capacitors connected to the DRP and DRN. 9. Route the feedback network of the negative charge pump away from the drive pin traces (DRN) of the negative charge pump. This avoids parasitic coupling into the feedback network of the negative charge 20 Submit Documentation Feedback TPS65161 www.ti.com SLVS617 – APRIL 2006 pump giving good output voltage accuracy and load regulation. To do this, use the FREQ pin and trace to isolate DRN from FBN. Submit Documentation Feedback 21 TPS65161 www.ti.com SLVS617 – APRIL 2006 APPLICATION INFORMATION C1 2 * 22 μF 8 12 20 21 22 C6 0.47 μF D2 D3 16 9 11 13 24 6 7 SUP FREQ VINB VINB AVIN EN1 EN2 DRN FBN REF PGND PGND 28 SS 25 DLY1 R3 620 kΩ R4 150 kΩ C9 22 nF C8 220 nF SW SW FB OS GND GD DRP FBP Boot SWB NC FBB COMP DLY2 4 5 1 3 23 27 R1 680 kΩ C15 470 nF C2 3 * 22 μF R2 56 kΩ D4 GD C5 10 14 17 18 19 15 D5 0.47 μF VGH 26 V/50 mA R5 909 kΩ 2 26 C13 0.47 μF R6 44.2 kΩ Cb 100 nF Vlogic 3.3 V/2.3 A L2 15 μH C17 22 nF C11 10 nF C10 10 nF Vs 15 V/1.5 A C4 22 pF TPS65161 C3 1 μF C16 1 μF VGL −5 V/50 mA C7 0.47 μF D1 SL22 L1 10 μH Vin 12 V D6 SL22 R7 2 kΩ C14 10 nF C12 2 * 22 μF R8 1.2 kΩ Figure 19. Standard 12-V to 15-V Conversion C1 2 * 22 μF VGL −5 V/50 mA C16 1 μF 8 12 20 21 22 C6 0.47 μF D2 AVIN EN1 EN2 DRN FBN REF PGND PGND 28 SS 25 DLY1 R3 620 kΩ R4 150 kΩ C8 220 nF 16 9 11 13 24 6 7 SUP FREQ VINB VINB C9 22 nF C10 10 nF SW SW FB OS GND GD DRP FBP Boot SWB NC FBB COMP DLY2 C11 10 nF 4 5 1 3 23 27 Vs 18 V/1.3 A C4 22 pF TPS65161 C3 1 μF D3 C7 0.47 μF D1 SL22 L1 10 μH Vin 12 V R1 825 kΩ C15 470 nF C2 3 * 22 μF R2 56 kΩ D4 GD 10 14 17 18 19 15 C5 D5 0.47 μF VGH 32 V/50 mA R5 560 kΩ 2 26 Cb 100 nF C17 22 nF C13 0.47 μF R6 22 kΩ Vlogic 3.3 V/2.3 A L2 15 μH D6 SL22 R7 2 kΩ R8 1.2 kΩ Figure 20. Standard 12-V to 18-V Conversion 22 Submit Documentation Feedback C14 10 nF C12 2 * 22 μF TPS65161 www.ti.com SLVS617 – APRIL 2006 APPLICATION INFORMATION (continued) L1 10 µH Vin 12 V C1 2*22 µF C3 VGL –5 V/50 mA D2 C6 0.47µF D3 C7 0.47 µF REF PGND PGND SS DLY1 R3 620 kΩ R4 150 kΩ C8 220 nF SUP FREQ VINB VINB AVIN EN1 EN2 DRN FBN C9 22 nF C10 10 nF SW SW FB OS GND GD DRP FBP Boot SWB NC FBB COMP DLY2 C11 10 nF Vs 18 V/1.3 A SI2343 R1 C4 22 pF TPS65161 1 µF C16 1 µF D1 SL22 825 kΩ C15 470 nF D4 C19 1 µF GD D5 VGH 32 V/50 mA R5 560 kΩ C17 22nF R9 510 kΩ R10 100 kΩ R2 56 kΩ C5 0.47 µF GD C18 220 nF C2 3* 22 µF C13 0.47 µF R6 22 kΩ Cb 100 nF L2 15 µH D6 SL22 Vlogic 3.3 V/2.3 A C14 R7 2 kΩ 10 nF C12 2*22 µF R8 1.2 kΩ Figure 21. Standard 12-V to 18-V Conversion Using an External Isolation MOSFET to Isolate Vs as well as VGH Submit Documentation Feedback 23 TPS65161 www.ti.com SLVS617 – APRIL 2006 APPLICATION INFORMATION (continued) C1 2 * 22 μF C6 0.47 μF D2 D3 C7 0.47 μF R3 620 kΩ R4 150 kΩ C8 220 nF C4 22 pF TPS65161 C3 1 μF C16 1 μF VGL −5 V/50 mA D1 SL22 L1 6.9 μH Vin 12 V 10% C9 22 nF 8 SUP 12 FREQ 20 VINB 21 VINB 22 AVIN 16 EN1 9 EN2 11 DRN 13 FBN 24 6 REF PGND 7 PGND 28 25 SS DLY1 C10 10 nF SW SW FB OS GND GD DRP FBP Boot SWB NC FBB COMP DLY2 C11 10 nF 4 5 1 3 23 27 10 14 17 18 C15 470 nF Vs 13.5 V/2 A R1 820 kΩ C2 3 * 22 μF R2 75 kΩ D4 GD C5 VGH 23 V/50 mA D5 0.47 μF R5 1 MΩ 19 15 2 26 R6 76 kΩ Cb 100 nF C17 22 nF Vlogic 3.3 V/2.3 A L2 15 μH D6 SL22 R7 2 kΩ R8 1.2 kΩ Figure 22. Standard 12-V to 13.5-V Conversion 24 C13 0.47 μF Submit Documentation Feedback C14 10 nF C12 2 * 22 μF IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. Customers should obtain the latest relevant information before placing orders and should verify that such information is current and complete. All products are sold subject to TI’s terms and conditions of sale supplied at the time of order acknowledgment. TI warrants performance of its hardware products to the specifications applicable at the time of sale in accordance with TI’s standard warranty. Testing and other quality control techniques are used to the extent TI deems necessary to support this warranty. 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