TPS65140-Q1 TPS65145-Q1 www.ti.com ................................................................................................................................................. SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 TRIPLE OUTPUT LCD SUPPLY WITH LINEAR REGULATOR AND POWER GOOD FEATURES 1 • • • • • 2 • • • • • • • • • Qualified For Automotive Applications 2.7-V to 5.8-V Input Voltage Range 1.6-MHz Fixed Switching Frequency Three Independent Adjustable Outputs Main Output up to 15 V With <1% Typical Output Voltage Accuracy Negative Output Voltage Down to -12 V/20 mA Positive Output Voltage up to 30 V/20 mA Auxiliary 3.3-V Linear Regulator Controller Internal Soft Start Internal Power-On Sequencing Fault Detection of all Outputs Thermal Shutdown System Power Good Available in a TSSOP-24 PowerPAD™ Package APPLICATIONS • • • • • TFT LCD Displays for Automobiles – Cluster Panels – Navigation Displays – Passenger Entertainment Modules TFT LCD Displays for Notebooks TFT LCD Displays for Monitors Portable DVD Players Industrial Displays DESCRIPTION The TPS65140/145 offers a compact and small power supply solution to provide all three voltages required by thin film transistor (TFT) LCD displays. The auxiliary linear regulator controller can be used to generate a 3.3-V logic power rail for systems powered by a 5-V supply rail only. The main output Vo1 is a 1.6-MHz fixed frequency PWM boost converter providing the source drive voltage for the LCD display. The device is available in two versions with different internal switch current limits to allow the use of a smaller external inductor when lower output power is required. The TPS65140 has a typical switch current limit of 2.3 A and the TPS65145 has a typical switch current limit of 1.37 A. A fully integrated adjustable charge pump doubler/tripler provides the positive LCD gate drive voltage. An externally adjustable negative charge pump provides the negative gate drive voltage. Due to the high 1.6-MHz switching frequency of the charge pumps, inexpensive and small 220-nF capacitors can be used. Additionally, the TPS65140/145 has a system power good output to indicate when all supply rails are acceptable. For LCD panels powered by 5 V, only the TPS65140/145 has a linear regulator controller using an external transistor to provide a regulated 3.3-V output for the digital circuits. For maximum safety, the entire device goes into shutdown as soon as one of the outputs is out of regulation. The device can be enabled again by toggling the input or the enable (EN) pin to GND. 1 2 Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PowerPAD is a trademark of Texas Instruments. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2004–2008, Texas Instruments Incorporated TPS65140-Q1 TPS65145-Q1 SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 ................................................................................................................................................. www.ti.com TPS65140/45 Vin 2.7 V to 5.8 V Vo1 Up to 15 V / 400 mA Boost Converter Vo3 Up to 30 V / 20 mA Positive Charge Pump Negative Charge Pump Vo2 Up to −12 V / 20 mA Power Good Power Good Vo4 3.3 V Linear Regulator Controller TYPICAL APPLICATION CIRCUIT VI 2.7 V to 5.8 V L1 4.2 µH C3 22 µF TPS65140 C5 VIN C13 10 nF C1 VO2 Up to 12 V/20 mA C6 0.22 µF D2 C12 D3 R3 0.22 µF 0.22 µF VO1 Up to 15 V/350 mA D1 C4 22 µF R1 SW COMP SW GND FB1 EN ENR SUP C1+ C1− C2−/MODE DRV FB3 C2+ C1 0.22 µF R2 VO3 Up to 30 V/20 mA OUT3 FB2 PG REF FB4 PGND BASE C7 0.22 µF R5 PGND GND R4 R6 C11 100 nF Q1 BCP68 VI VO4 3.3 V VI R7 33 kΩ C9 4.7 µF C9 1 µF System Power Good ORDERING INFORMATION (1) TA -40°C to 85°C (1) (2) (3) 2 MINIMUM SWITCH CURRENT LIMIT PACKAGE (2) (3) 3.3 V 1.6 A TPS65140IPWPRQ1 65140IQ1 3.3 V 0.96 A TPS65145IPWPRQ1 65145IQ1 LINEAR REGULATOR OUTPUT VOLTAGE TSSOP PACKAGE MARKING For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI web site at www.ti.com. Package drawings, thermal data, and symbolization are available at www.ti.com/packaging. The PWP and RGE packages are available taped and reeled. Add an R suffix to the device type (TPS65100PWPR) to order the device taped and reeled. The PWPR package has quantities of 2000 devices per reel and the the RGER package has 3000 devices per reel. Without the suffix, the PWP package only, is shipped in tubes with 60 devices per tube. Submit Documentation Feedback Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 TPS65140-Q1 TPS65145-Q1 www.ti.com ................................................................................................................................................. SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) UNIT Voltages on pin VIN (2) Voltages on pin Vo1, SUP, PG -0.3 V to 6 V (2) Voltages on pin EN, MODE, ENR -0.3 V to 15.5 V (2) -0.3 V to VI + 0.3 V Voltage on pin SW (2) 20 V Power good maximum sink current (PG) 1 mA Continuous power dissipation See Dissipation Rating Table Operating junction temperature range -40C to 150C Storage temperature range -65C to 150C Lead temperature (soldering, 10 sec) (1) (2) 260C Stresses beyond those listed under “absolute maximum ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under “recommended operating conditions” is not implied. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability. All voltage values are with respect to network ground terminal. DISSIPATION RATINGS PACKAGE RθJA TA ≤ 25C POWER RATING TA = 70°C POWER RATING TA = 85°C POWER RATING 24-pin TSSOP 30.13°C/W (PWP soldered) 3.3 W 1.83 W 1.32 W RECOMMENDED OPERATING CONDITIONS MIN VIN Input voltage range TYP MAX 2.7 (1) 5.8 UNIT V µH L Inductor TA Operating ambient temperature -40 85 °C TJ Operating junction temperature -40 125 °C (1) 4.7 See the Application Information Section for further information. ELECTRICAL CHARACTERISTICS VIN = 3.3 V, EN = VIN, Vo1 = 10 V, TA= -40°C to 85°C, typical values are at TA = 25C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT SUPPLY CURRENT VIN Input voltage range 5.5 V ENR = GND, Vo3 = 2 × Vo1, Boost converter not switching 2.7 0.7 0.9 mA Vo1 = SUP = 10 V, Vo3 = 2 × Vo1 1.7 2.7 Vo1 = SUP = 10 V, Vo3 = 3 × Vo1 3.9 6 300 800 µA IQIN Quiescent current into VIN IQCharge Charge pump quiescent current into SUP IQEN LDO controller quiescent current into VIN ENR = VIN, EN = GND ISD Shutdown current into VIN EN = ENR = GND 1 10 µA VUVLO Undervoltage lockout threshold VI falling 2.2 2.4 V Thermal shutdown Temperature rising 160 mA °C LOGIC SIGNALS EN, ENR VIH High level input voltage VIL Low level input voltage II Input leakage current 1.5 EN = GND or VIN V 0.01 0.4 V 0.1 µA 15 V MAIN BOOST CONVERTER Vo1 Output voltage range 5 Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 Submit Documentation Feedback 3 TPS65140-Q1 TPS65145-Q1 SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 ................................................................................................................................................. www.ti.com ELECTRICAL CHARACTERISTICS (continued) VIN = 3.3 V, EN = VIN, Vo1 = 10 V, TA= -40°C to 85°C, typical values are at TA = 25C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX 1 UNIT VO1-VIN Minimum input to output voltage difference V VREF Reference voltage 1.205 1.213 1.219 VFB Feedback regulation voltage 1.136 1.146 1.154 IFB Feedback input bias current 10 100 rDS(on) N-MOSFET on-resistance (Q1) Vo1 = 10 V, Isw = 500 mA 195 290 Vo1 = 5 V, Isw = 500 mA 285 420 ILIM N-MOSFET switch current limit (Q1) TPS65140 1.6 2.3 2.8 A TPS65145 0.96 1.37 1.7 A rDS(on) P-MOSFET on-resistance (Q2) Vo1 = 10 V, Isw = 100 mA 9 15 Vo1 = 5 V, Isw = 100 mA 14 22 IMAX Maximum P-MOSFET peak switch current Ileak Switch leakage current fSW Oscillator frequency 1 Vsw = 15 V 1 10 Vsw = 0 V 1 10 0°C ≤ TA ≤ 85°C 1.295 1.6 2.1 -40°C ≤ TA ≤ 85°C 1.191 1.6 2.1 Line regulation 2.7 V ≤ VI ≤ 5.7 V; Iload = 100 mA Load regulation 0 mA ≤ IO ≤ 300 mA V V nA mΩ Ω A µA MHz 0.012 %/V 0.2 %/A NEGATIVE CHARGE PUMP Vo2 Vo2 Output voltage range Vref Reference voltage VFB Feedback regulation voltage IFB Feedback input bias current rDS(on) IO Q8 P-Channel switch rDS(on) Q9 N-Channel switch rDS(on) -2 V 1.205 1.213 1.219 -36 0 36 mV 10 100 nA 4.3 8 2.9 4.4 Ω IO = 20 mA Minimum output current V 20 mA Line regulation 7 V ≤ Vo1 ≤ 15 V, Iload = 10 mA, Vo2 = -5 V 0.09 %/V Load regulation 1 mA ≤ IO ≤ 20 mA, Vo2 = -5 V 0.126 %/mA POSITIVE CHARGE PUMP Vo3 Vo3 Output voltage range 30 V Vref Reference voltage 1.205 1.213 1.219 V VFB Feedback regulation voltage 1.187 1.214 1.238 V IFB Feedback input bias current 10 100 nA Q3 P-Channel switch rDS(on) 9.9 15.5 1.1 1.8 4.6 8.5 1.2 2.2 rDS(on) Q4 N-Channel switch rDS(on) Q5 P-Channel switch rDS(on) Ω IO = 20 mA Q6 N-Channel switch rDS(on) 4 Submit Documentation Feedback Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 TPS65140-Q1 TPS65145-Q1 www.ti.com ................................................................................................................................................. SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 ELECTRICAL CHARACTERISTICS (continued) VIN = 3.3 V, EN = VIN, Vo1 = 10 V, TA= -40°C to 85°C, typical values are at TA = 25C (unless otherwise noted) PARAMETER Vd D1 – D4 Shottky diode forward voltage IO Minimum output current TEST CONDITIONS MIN ID1-D4 = 40 mA TYP MAX UNIT 610 720 mV 20 mA Line regulation 10 V ≤ Vo1 ≤ 15 V, Iload = 10 mA, Vo3 = 27 V 0.56 %/V Load regulation 1 mA ≤ IO ≤ 20 mA, Vo3 = 27 V 0.05 %/mA LINEAR REGULATOR CONTROLLER Vo4 Vo4 IBASE Output voltage 4.5 V ≤ VI ≤ 5.5 V; 10 mA ≤ IO ≤ 500 mA Maximum base drive current VIN-Vo4-VBE≥ 0.5 V (1) Line regulation 4.75 V ≤ VI ≤ 5.5 V, Iload = 500 mA 0.186 Load regulation 1 mA ≤ IO ≤ 500 mA, VI = 5 V 0.064 Start up current Vo4 ≤ 0.8 V VIN-Vo4-VBE≥ 0.75 V (1) 3.2 3.3 13.5 19 20 27 3.4 V mA %/V %/A 11 20 25 mA -12 -8.75% Vo1 -6 V -13 -9.5% Vo2 -5 V -11 -8% Vo3 -5 V SYSTEM POWER GOOD (PG) V(PG, VO1) V(PG, VO2) Power good threshold (2) V(PG, VO3) VOL PG output low voltage I(sink) = 500 A IL PG output leakage current VPG = 5 V (1) (2) 0.001 0.3 V 1 µA With VIN = supply voltage of the TPS65140, Vo4 = output voltage of the regulator, VBE = basis emitter voltage of external transistor. The power good goes high when all three outputs (Vo1, Vo2, Vo3) are above their threshold. The power good goes low as soon as one of the outputs is below their threshold. DEVICE INFORMATION PWP PACKAGE TOP VIEW 1 24 2 23 3 22 4 21 5 6 20 19 7 8 Thermal PAD* FB1 FB4 BASE VIN SW SW PGND PGND SUP PG GND FB3 18 17 9 16 10 15 11 14 12 13 EN ENR COMP FB2 REF GND DRV C1− C1+ C2−/MODE C2+ OUT3 Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 Submit Documentation Feedback 5 TPS65140-Q1 TPS65145-Q1 SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 ................................................................................................................................................. www.ti.com Terminal Functions TERMINAL NAME NO. (PWP) I/O DESCRIPTION VIN 4 I Input voltage pin of the device. EN 24 I Enable pin of the device. This pin should be terminated and not be left floating. A logic high enables the device and a logic low shuts down the device. COMP 22 PG 10 O Open drain output indicating when all outputs Vo1, Vo2, Vo3 are within 10% of their nominal output voltage. The output goes low when one of the outputs falls below 10% of their nominal output voltage. ENR 23 I Enable pin of the linear regulator controller. This pin should be terminated and not be left floating. Logic high enables the regulator and a logic low puts the regulator in shutdown. C1+ 16 C1- 17 DRV 18 O External charge pump driver FB2 21 I Feedback pin of negative charge pump REF 20 O Internal reference output typically 1.23 V FB4 2 I Feedback pin of the linear regulator controller. The linear regulator controller is set to a fixed output voltage of 3.3 V or 3 V depending on the version. BASE 3 O Base drive output for the external transistor GND 11, 19 Compensation pin for the main boost converter. A small capacitor is connected to this pin. Positive terminal of the charge pump flying capacitor Negative terminal of the charge pump flying capacitor Ground PGND 7, 8 FB3 12 I Feedback pin of positive charge pump OUT3 13 O Positive charge pump output C2-/MODE 15 Negative terminal of the charge pump flying capacitor and charge pump MODE pin. If the flying capacitor is connected to this pin, the converter operates in a voltage tripler mode. If the charge pump needs to operate in a voltage doubler mode, the flying capacitor is removed and the C2-/MODE pin needs to be connected to GND. C2+ 14 Positive terminal for the charge pump flying capacitor. If the device runs in voltage doubler mode, this pin needs to be left open. 9 I Supply pin of the positive, negative charge pump, boost converter, and gate drive circuit. This pin needs to be connected to the output of the main boost converter and cannot be connected to any other voltage source. For performance reasons, it is not recommended for a bypass capacitor to be connected directly to this pin. FB1 1 I Feedback pin of the boost converter SW 5, 6 I Switch pin of the boost converter SUP Power ground PowerPAD™/ Thermal Die 6 Submit Documentation Feedback The PowerPAD or exposed thermal die needs to be connected to the power ground pins (PGND). Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 TPS65140-Q1 TPS65145-Q1 www.ti.com ................................................................................................................................................. SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 FUNCTIONAL BLOCK DIAGRAM VIN SW SW Q2 Main boost converter EN FB1 FB2 FB3 Bias Vref = 1.213 V Thermal Shutdown Start−Up Sequencing Undervoltage Detection Overvoltage Detection Short Circuit Protection D S Current Limit and Soft Start 1.6-MHz Oscillator SUP Control Logic Gate Drive Circuit D Q1 COMP S Comparator Sawtooth Generator FB1 VFB 1.146 V SUP FB3 SUP (VO) Positive Charge Pump GM Amplifier Low Gain SUP D VFB 1.146 V Current Control Vref 1.214 V S Q3 C1− Gain Select (Doubler or Tripler Mode) D S Q4 SUP Negative Charge Pump SUP Soft Start D Q8 S DRV D Q7 S D Q9 C1+ Current Control Soft Start SUP Vo3 S D2 D S D1 D4 Q5 C2+ FB2 D Vref 0V D3 Q6 S C2− Reference Output Vref REF 1.213 V Soft Start Iref = 20 mA Short Circuit Detect Vref 1.213 V Vin System Power Good PG FB1 ~1 V D FB2 FB4 D S Linear Regulator Controller Vref 1.213 V Q10 Logic and 1-µs Glitch Filter S FB3 ENR BASE GND Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 GND PGND PGND Submit Documentation Feedback 7 TPS65140-Q1 TPS65145-Q1 SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 ................................................................................................................................................. www.ti.com TYPICAL CHARACTERISTICS Table of Graphs FIGURE Main Boost Converter Efficiency, main boost converter Vo1 vs Load current 1 Efficiency, main boost converter Vo1 vs Load current 2 Efficiency vs Input voltage 3 fsw Switching frequency vs Free-air temperature 4 rDS(on) rDS(on) N-Channel main switch Q1 vs Free-air temperature 5 η PWM operation continuous mode 6 PWM operation, discontinuous (light load) 7 Load transient response, CO = 22 F 8 Load transient response, CO = 2 x 22 F 9 Power-up sequencing 10 Soft start Vo1 11 Negative Charge Pump Imax Vo2 maximum load current vs Output voltage Vo1 12 Positive Charge Pump Imax Vo3 maximum load current vs Output voltage Vo1 (doubler mode) 13 Imax Vo3 Maximum load current vs Output voltage Vo1 (tripler mode) 14 EFFICIENCY vs LOAD CURRENT 100 100 90 90 Vo1 = 15 V 40 30 70 Vo1 = 10 V 60 50 Vo1 = 15 V 40 Vo1 = 6 V 90 Vo2 = 10 V 85 Vo3 = 15 V 80 30 VI = 3.3 V Vo2, Vo3 = No Load, Switching 20 10 10 100 1k IL − Load Current − mA Figure 1. Submit Documentation Feedback 75 VI = 5 V Vo2, Vo3 = No Load, Switching 20 10 1 8 ILoad at Vo1 = 100 mA Vo2, Vo3 = No Load, Switching Efficiency − % Efficiency − % 60 50 100 80 Vo1 = 10 V 70 EFFICIENCY vs INPUT VOLTAGE 95 Vo1 = 6 V 80 Efficiency − % EFFICIENCY vs LOAD CURRENT 1 10 100 1k IL − Load Current − mA 70 2.5 3 3.5 4 4.5 5 5.5 6 VI − Input Voltage − V Figure 2. Figure 3. Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 TPS65140-Q1 TPS65145-Q1 www.ti.com ................................................................................................................................................. SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 SWITCHING FREQUENCY vs FREE-AIR TEMPERATURE rDS(on) N-CHANNEL MAIN SWITCH vs FREE-AIR TEMPERATURE 350 r − N−Channel Main Switch − mΩ DS(on) Switching Frequency − MHz 1.9 1.8 VI = 2.7 V 1.7 VI = 3.3 V 1.6 VI = 5.8 V 1.5 1.4 1.3 −40 PWM OPERATION CONTINUOUS MODE −20 0 20 40 60 80 100 VSW 10 V/div 300 Vo1 = 5 V 250 VO 50 mV/div 200 Vo1 = 10 V 150 IL 1 A/div Vo1 = 15 V 100 −40 −20 0 20 40 60 80 VI = 3.3 V VO = 10 V/300 mA 100 250 ns/div TA − Free-Air Temperature − °C TA − Free-Air Temperature − °C Figure 4. Figure 5. Figure 6. PWM OPERATION AT LIGHT LOAD LOAD TRANSIENT RESPONSE LOAD TRANSIENT RESPONSE VI = 3.3 V Vo1 = 10 V, CO= 2*22 µF Vo1 200 mV/div VSW 10 V/div Vo1 100 mV/div VO 50 mV/div VI = 3.3 V VO = 10 V/10 mA IO 50 mA to 250 mA IL 500 mA/div VI = 3.3 V Vo1 = 10 V, CO= 22 µF IO 50 mA to 250 mA 100 µs/div 100 µs/div 250 ns/div Figure 7. Figure 8. POWER-UP SEQUENCING SOFT START Vo1 Figure 9. Vo2 MAXIMUM LOAD CURRENT 0.20 VI = 3.3 V VO = 10 V, IO = 300 mA Vo2 = −8 V 0.18 I O − Output Current − A Vo1 5 V/div Vo1 5 V/div Vo2 5 V/div Vo3 10 V/div VI = 3.3 V VO = 10 V, II 500 mA/div 0.14 TA = 85°C 0.12 0.10 TA = 25°C 0.08 0.06 0.04 0.02 500 µs/div 500 µs/div TA = −40°C 0.16 0 8.8 9.8 10.8 11.8 12.8 13.8 14.8 Vo1 − Output Voltage − V Figure 10. Figure 11. Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 Figure 12. Submit Documentation Feedback 9 TPS65140-Q1 TPS65145-Q1 SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 ................................................................................................................................................. www.ti.com Vo3 MAXIMUM LOAD CURRENT Vo3 MAXIMUM LOAD CURRENT 0.14 0.12 TA = −40°C Vo3 = 18 V (Doubler Mode) 0.10 TA = −40°C I O − Output Current − A I O − Output Current − A 0.12 TA = 85°C 0.10 TA = 25°C 0.08 0.06 0.04 TA = 25°C 0.08 TA = 85°C 0.06 0.04 0.02 0.02 Vo3 = 28 V (Tripler Mode) 0 9 10 11 12 13 Vo1 − Output Voltage − V Figure 13. 14 15 0 9 10 11 12 13 14 15 Vo1 − Output Voltage − V Figure 14. DETAILED DESCRIPTION The TPS65140/45 consists of a main boost converter operating with a fixed switching frequency of 1.6 MHz to allow for small external components. The boost converter output voltage Vo1 is also the input voltage, connected via the pin SUP, for the positive and negative charge pump. The linear regulator controller is independent from this system with its own enable pin. This allows the linear regulator controller to continue to operate while the other supply rails are disabled or in shutdown due to a fault condition on one of their outputs. Refer to the functional block diagram for more information. Main Boost Converter The main boost converter operates with PWM and a fixed switching frequency of 1.6 MHz. The converter uses a unique fast response, voltage mode controller scheme with input voltage feedforward. This achieves excellent line and load regulation (0.2% A load regulation typical) and allows the use of small external components. To add higher flexibility to the selection of external component values, the device uses external loop compensation. Although the boost converter looks like a nonsynchronous boost converter topology operating in discontinuous mode at light load, the TPS65140/45 maintains continuous conduction even at light load currents. This is achieved with a novel architecture using an external Schottky diode and an integrated MOSFET in parallel connected between SW and SUP (see the functional block diagram). The integrated MOSFET Q2 allows the inductor current to become negative at light load conditions. For this purpose, a small integrated P-channel MOSFET with typically 10 Ω rDS(on) is sufficient. When the inductor current is positive, the external Schottky diode with the lower forward voltage conducts the current. This causes the converter to operate with a fixed frequency in continuous conduction mode over the entire load current range. This avoids the ringing on the switch pin as seen with a standard nonsynchronous boost converter and allows a simpler compensation for the boost converter. Power-Good Output The TPS65140/45 has an open-drain power-good output with a maximum sink capability of 1 mA. The power-good output goes high as soon as the main boost converter Vo1 and the negative and the positive charge pumps are within regulation. The power-good output goes low as soon as one of the outputs is out of regulation. In this case, the device goes into shutdown at the same time. See the electrical characteristics table for the power-good thresholds. Enable and Power-On Sequencing (EN, ENR) The device has two enable pins. These pins should be terminated and not left floating to prevent faulty operation. Pulling the enable pin (EN) high enables the device and starts the power-on sequencing with the main boost converter Vo1 coming up first, then the negative and positive charge pumps. The linear regulator has an independent enable pin (ENR). Pulling this pin low disables the regulator, and pulling this pin high enables this regulator. 10 Submit Documentation Feedback Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 TPS65140-Q1 TPS65145-Q1 www.ti.com ................................................................................................................................................. SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 If the enable pin (EN) is pulled high, the device starts its power-on sequencing. The main boost converter starts up first with its soft start. If the output voltage has reached 91.25% of its output voltage, the negative charge pump comes up next. The negative charge pump starts with a soft start and when the output voltage has reached 91% of the nominal value, the positive charge pump comes up with the soft start. Pulling the enable pin low shuts down the device. Dependent on load current and output capacitance, each of the outputs comes down. Positive Charge Pump The TPS65140/45 has a fully regulated integrated positive charge pump generating Vo3. The input voltage for the charge pump is applied to the SUP pin that is equal to the output of the main boost converter Vo1. The charge pump is capable of supplying a minimum load current of 20 mA. Higher load currents are possible depending on the voltage difference between Vo1 and Vo3. See Figure 13 and Figure 14. Negative Charge Pump The TPS65140/45 has a regulated negative charge pump using two external Schottky diodes. The input voltage for the charge pump is applied to the SUP pin that is connected to the output of the main boost converter Vo1. The charge pump inverts the main boost converter output voltage and is capable of supplying a minimum load current of 20 mA. Higher load currents are possible depending on the voltage difference between Vo1 and Vo2. See Figure 12. Linear Regulator Controller The TPS65140/45 includes a linear regulator controller to generate a 3.3-V rail which is useful when the system is powered from a 5-V supply. The regulator is independent from the other voltage rails of the device and has its own enable (ENR). Soft Start The main boost converter as well as the charge pumps and linear regulator have an internal soft start. This avoids heavy voltage drops at the input voltage rail or at the output of the main boost converter Vo1 during start-up caused by high inrush currents. See Figure 10 and Figure 11. Fault Protection All of the outputs of the TPS65140/45 have short-circuit detection and cause the device to go into shutdown. The main boost converter has overvoltage and undervoltage protection. If the output voltage Vo1 rises above the overvoltage protection threshold of typically 5% of Vo1, then the device stops switching, but remains operational. When the output voltage falls below this threshold, the converter continues operation. When the output voltage falls below the undervoltage protection threshold of typically 8.75% of Vo1, because of a short-circuit condition, the TPS65140/45 goes into shutdown. Because there is a direct pass from the input to the output through the diode, the short-circuit condition remains. If this condition needs to be avoided, a fuse at the input or an output disconnect using a single transistor and resistor is required. The negative and positive charge pumps have an undervoltage lockout (UVLO) to protect the LCD panel of possible latch-up conditions due to a short-circuit condition or faulty operation. When the negative output voltage is typically above 9.5% of its output voltage (closer to ground), then the device enters shutdown. When the positive charge pump output voltage, Vo3, is below 8% typical of its output voltage, the device goes into shutdown. See the fault protection thresholds in the electrical characteristics table. The device is enabled by toggling the enable pin (EN) below 0.4 V or by cycling the input voltage below the UVLO of 1.7 V. The linear regulator reduces the output current to 20 mA typical under a short-circuit condition when the output voltage is typically < 1 V. See the Functional Block Diagram. The linear regulator does not go into shutdown under a short-circuit condition. Thermal Shutdown A thermal shutdown is implemented to prevent damage due to excessive heat and power dissipation. Typically, the thermal shutdown threshold is 160°C. If this temperature is reached, the device goes into shutdown. The device can be enabled by toggling the enable pin to low and back to high or by cycling the input voltage to GND and back to VI again. Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 Submit Documentation Feedback 11 TPS65140-Q1 TPS65145-Q1 SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 ................................................................................................................................................. www.ti.com APPLICATION INFORMATION BOOST CONVERTER DESIGN PROCEDURE The first step in the design procedure is to calculate the maximum possible output current of the main boost converter under certain input and output voltage conditions. Below is an example for a 3.3-V to 10-V conversion: Vin = 3.3 V, Vout = 10 V, Switch voltage drop Vsw = 0.5 V, Schottky diode forward voltage VD = 0.8 V 1. Duty cycle: Vout ) V * V D in + 10 V ) 0.8 V * 3.3 V + 0.73 D+ V out ) V * V sw 10 V ) 0.8 V * 0.5 V D 2. Average inductor current: I I + out + 300 mA + 1.11 A L 1 * 0.73 1*D 3. Inductor peak-to-peak ripple current: Di + L ƪVin * Vswƫ fs L D + (3.3 V * 0.5 V) 0.73 + 304 mA 1.6 MHz 4.2 mH 4. Peak switch current: Di I + I ) L + 1.11 A ) 304 mA + 1.26 A swpeak L 2 2 The integrated switch, the inductor, and the external Schottky diode must be able to handle the peak switch current. The calculated peak switch current has to be equal or lower to the minimum N-MOSFET switch current limit as specified in the electrical characteristics table (1.6 A for the TPS65140 and 0.96 A for the TPS65145). If the peak switch current is higher, then the converter cannot support the required load current. This calculation must be done for the minimum input voltage where the peak switch current is highest. The calculation includes conduction losses like switch rDS(on) (0.5 V) and diode forward drop voltage losses (0.8 V). Additional switching losses, inductor core and winding losses, etc., require a slightly higher peak switch current in the actual application. The above calculation still allows for a good design and component selection. Inductor Selection Several inductors work with the TPS65140. Especially with the external compensation, the performance can be adjusted to the specific application requirements. The main parameter for the inductor selection is the saturation current of the inductor, which should be higher than the peak switch current as calculated above with additional margin to cover for heavy load transients and extreme start-up conditions. Another method is to choose the inductor with a saturation current at least as high as the minimum switch current limit of 1.6 A for the TPS65140 and 0.96 A for the TPS65145. The different switch current limits allow selection of a physically smaller inductor when less output current is required. The second important parameter is the inductor dc resistance. Usually, the lower the dc resistance, the higher the efficiency. However, the inductor dc resistance is not the only parameter determining the efficiency. Especially for a boost converter where the inductor is the energy storage element, the type and material of the inductor influences the efficiency as well. Especially at high switching frequencies of 1.6 MHz, inductor core losses, proximity effects, and skin effects become more important. Usually, an inductor with a larger form factor yields higher efficiency. The efficiency difference between different inductors can vary between 2% to 10%. For the TPS65140, inductor values between 3.3 H and 6.8 H are a good choice but other values can be used as well. Possible inductors are shown in Table 1. 12 Submit Documentation Feedback Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 TPS65140-Q1 TPS65145-Q1 www.ti.com ................................................................................................................................................. SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 Table 1. Inductor Selection DEVICE TPS65140 TPS65145 INDUCTOR VALUE COMPONENT SUPPLIER DIMENSIONS ISAT/DCR 4.7 H Coilcraft DO1813P-472HC 8.89*6.1*5.0 2.6 A/54 mΩ 4.2 H Sumida CDRH5D28 4R2 5.7*5.7*3 2.2 A/23 mΩ 4.7 H Sumida CDC5D23 4R7 6*6*2.5 1.6 A/48 mΩ 3.3 H Wuerth Elektronik 744042003 4.8*4.8*2.0 1.8 A/65 mΩ 4.2 H Sumida CDRH6D12 4R2 6.5*6.5*1.5 1.8 A/60 mΩ 3.3 H Sumida CDRH6D12 3R3 6.5*6.5*1.5 1.9 A/50 mΩ 3.3 H Sumida CDPH4D19 3R3 5.1*5.1*2.0 1.5 A/26 mΩ 3.3 H Coilcraft DO1606T-332 6.5*5.2*2.0 1.4 A/120 mΩ 3.3 H Sumida CDRH2D18/HP 3R3 3.2*3.2*2.0 1.45 A/69 mΩ 4.7 H Wuerth Elektronik 744010004 5.5*3.5*1.0 1.0 A/260 mΩ 3.3 H Coilcraft LPO6610-332M 6.6*5.5*1.0 1.3 A/160 mΩ Output Capacitor Selection For best output voltage filtering, a low ESR output capacitor is recommended. Ceramic capacitors have a low ESR value but depending on the application, tantalum capacitors can be used as well. A 22-F ceramic output capacitor works for most of the applications. Higher capacitor values can be used to improve load transient regulation. See Table 2 for the selection of the output capacitor. The output voltage ripple can be calculated as: I DV out + out Cout ƪ ƫ Ip L 1* ) Ip f s Vout ) V * V d in ESR with: Ip = Peak current as described in the previous section peak current control L = Selected inductor value Iout = Nominal load current fs = Switching frequency Vd = Rectifier diode forward voltage (typically 0.3 V) Cout = Selected output capacitor ESR = Output capacitor ESR value Input Capacitor Selection For good input voltage filtering, low ESR ceramic capacitors are recommended. A 22-F ceramic input capacitor is sufficient for most of applications. For better input voltage filtering, this value can be increased. See Table 2 and the Typical Applications section for input capacitor recommendations. Table 2. Input and Output Capacitors Selection CAPACITOR VOLTAGE RATING COMPONENT SUPPLIER COMMENTS 22 F/1210 16 V Taiyo Yuden EMK325BY226MM CO 22 F/1206 6.3 V Taiyo Yuden JMK316BJ226 CI Rectifier Diode Selection To achieve high efficiency, a Schottky diode should be used. The voltage rating should be higher than the maximum output voltage of the converter. The average forward current should be equal to the average inductor current of the converter. The main parameter influencing the efficiency of the converter is the forward voltage and the reverse leakage current of the diode; both should be as low as possible. Possible diodes are: On Semiconductor MBRM120L, Microsemi UPS120E, and Fairchild Semiconductor MBRS130L. Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 Submit Documentation Feedback 13 TPS65140-Q1 TPS65145-Q1 SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 ................................................................................................................................................. www.ti.com Converter Loop Design and Stability The TPS65140/45 converter loop can be externally compensated and allows access to the internal transconductance error amplifier output at the COMP pin. A small feedforward capacitor across the upper feedback resistor divider speeds up the circuit as well. To test the converter stability and load transient performance of the converter, a load step from 50 mA to 250 mA is applied and the output voltage of the converter is monitored. Applying load steps to the converter output is a good tool to judge the stability of such a boost converter. Design Procedure Quick Steps 1. Select the feedback resistor divider to set the output voltage. 2. Select the feedforward capacitor to place a zero at 50 kHz. 3. Select the compensation capacitor on pin COMP. The smaller the value, the higher the low frequency gain. 4. Use a 50-kΩ potentiometer in series to Cc and monitor Vout during load transients. Fine tune the load transient by adjusting the potentiometer. Select a resistor value that comes closest to the potentiometer resistor value. This needs to be done at the highest ViN and highest load current because stability is most critical at these conditions. Setting the Output Voltage and Selecting the Feedforward Capacitor The output voltage is set by the external resistor divider and is calculated as: V out + 1.146 V ƪ1 ) R1 ƫ R2 Across the upper resistor, a bypass capacitor is required to speed up the circuit during load transients as shown in Figure 15. VO1 Up to 10 V/150 mA D1 C8 6.8 pF SW SW FB1 SUP C2+ C2−/MODE C2 0.22 µF C4 22 µF R1 430 kΩ R2 56 kΩ Figure 15. Feedforward Capacitor Together with R1 the bypass capacitor C8 sets a zero in the control loop at approximately 50 kHz: 1 1 C8 + + 2 p f z R1 2 p 50 kHz R1 A value closest to the calculated value should be used. Larger feedforward capacitor values reduce the load regulation of the converter and cause load steps as shown in Figure 16. 14 Submit Documentation Feedback Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 TPS65140-Q1 TPS65145-Q1 www.ti.com ................................................................................................................................................. SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 Load Step Figure 16. Load Step Caused By A Too Large Feedforward Capacitor Value Compensation The regulator loop can be compensated by adjusting the external components connected to the COMP pin. The COMP pin is connected to the output of the internal transconductance error amplifier. A typical compensation scheme is shown in Figure 17. CC RC 15 kΩ VIN COMP 1 nF Figure 17. Compensation Network The compensation capacitor Cc adjusts the low frequency gain, and the resistor value adjusts the high frequency gain. The following formula calculates at what frequency the resistor increases the high frequency gain. 1 fz + 2 p Cc Rc Lower input voltages require a higher gain and a lower compensation capacitor value. A good start is Cc = 1 nF for a 3.3-V input and Cc = 2.2 nF for a 5-V input. If the device operates over the entire input voltage range from 2.7 V to 5.8 V, a larger compensation capacitor up to 10 nF is recommended. Figure 18 shows the load transient with a larger compensation capacitor, and Figure 19 shows a smaller compensation capacitor. CC = 4.7 nF Figure 18. CC = 4. 7 nF Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 Submit Documentation Feedback 15 TPS65140-Q1 TPS65145-Q1 SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 ................................................................................................................................................. www.ti.com CC = 1 nF Figure 19. CC = 1 nF Lastly, Rc needs to be selected. A good practice is to use a 50-kΩ potentiometer and adjust the potentiometer for the best load transient where no oscillations should occur. These tests have to be done at the highest VIN and highest load current because the converter stability is most critical under these conditions. Figure 20, Figure 21, and Figure 22 show the fine tuning of the loop with Rc. Figure 20. Overcompensated (Damped Oscillation), RC Is Too Large Figure 21. Undercompensated (Loop Is Too Slow), RC Is Too Small 16 Submit Documentation Feedback Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 TPS65140-Q1 TPS65145-Q1 www.ti.com ................................................................................................................................................. SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 Figure 22. Optimum, RC Is Ideal Negative Charge Pump The negative charge pump provides a regulated output voltage by inverting the main output voltage, Vo1. The negative charge pump output voltage is set with external feedback resistors. The maximum load current of the negative charge pump depends on the voltage drop across the external Schottky diodes, the internal on resistance of the charge pump MOSFETS Q8 and Q9, and the impedance of the flying capacitor, C12. When the voltage drop across these components is larger than the voltage difference from Vo1 to Vo2, the charge pump is in drop out, providing the maximum possible output current. Therefore, the higher the voltage difference between Vo1 and Vo2, the higher the possible load current. See Figure 12 for the possible output current versus boost converter voltage Vo1 and the calculations below. Voutmin = -(Vo1 - 2 VD - Io (2 × rDS(on)Q8 + 2 נrDS(on)Q9 + Xcfly)) Setting the output voltage: V out + * V R3 + R4 REF ƪ1 ) R3 ƫ ) VREF + * 1.213 V ƪ1 ) R3 ƫ ) 1.213 V R4 R4 ȱŤVoutŤ ) VREF * 1ȳ+ R4 ȧ ȧ Ȳ VREF ȴ ƪŤ V outŤ ) 1.213 1.213 ƫ *1 The lower feedback resistor value, R4, should be in a range between 40 kΩ to 120 kΩ or the overall feedback resistance should be within 500 kΩ to 1 MΩ. Smaller values load the reference too heavy and larger values may cause stability problems. The negative charge pump requires two external Schottky diodes. The peak current rating of the Schottky diode has to be twice the load current of the output. For a 20-mA output current, the dual Schottky diode BAT54 or similar is a good choice. Positive Charge Pump The positive charge pump can be operated in a voltage doubler mode or a voltage tripler mode depending on the configuration of the C2+ and C2-/MODE pins. Leaving the C2+ pin open and connecting C2-/MODE to GND forces the positive charge pump to operate in a voltage doubler mode. If higher output voltages are required the positive charge pump can be operated as a voltage tripler. To operate the charge pump in the voltage tripler mode, a flying capacitor needs to be connected to C2+ and C2-/MODE. The maximum load current of the positive charge pump depends on the voltage drop across the internal Schottky diodes, the internal on-resistance of the charge pump MOSFETS, and the impedance of the flying capacitor. When the voltage drop across these components is larger than the voltage difference Vo1 נ2 to Vo3 (doubler mode) or Vo1 נ3 to Vo3 (tripler mode), then the charge pump is in dropout, providing the maximum possible output current. Therefore, the higher the voltage difference between Vo1 נ2 (doubler) or Vo1 נ3 (tripler) to Vo3, the higher the possible load current. See Figure 13 and Figure 14 for output current versus boost converter voltage, Vo1, and the following calculations. Voltage doubler: Vo3max = 2 × Vo1 - (2 VD + 2 × Io × (2 × rDS(on)Q5 + rDS(on)Q3 + rDS(on)Q4 + XC1)) Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 Submit Documentation Feedback 17 TPS65140-Q1 TPS65145-Q1 SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 ................................................................................................................................................. www.ti.com Voltage tripler: Vo3max = 3 × Vo1 - (3 × VD + 2 × Io × (3 × rDS(on)Q5 + rDS(on)Q3 + rDS(on)Q4 + XC1 + XC2)) The output voltage is set by the external resistor divider and is calculated as: ƪ1 ) R5 ƫ R6 V out + 1.214 R5 + R6 ƪ Vout V FB ƫ *1 + R6 ƪ V out 1.214 ƫ *1 Linear Regulator Controller The TPS65100/05 includes a linear regulator controller to generate a 3.3-V rail when the system is powered from a 5-V supply. Because an external npn transistor is required, the input voltage of the TPS65140/45 applied to VIN needs to be higher than the output voltage of the regulator. To provide a minimum base drive current of 13.5 mA, a minimum internal voltage drop of 500 mV from VIN to Vbase is required. This can be translated into a minimum input voltage on VIN for a certain output voltage as the following calculation shows: Vinmin = Vo4 + VBE + 0.5 V The base drive current together with the hFE of the external transistor determines the possible output current. Using a standard npn transistor like the BCP68 allows an output current of 1 A and using the BCP54 allows a load current of 337 mA for an input voltage of 5 V. Other transistors can be used as well, depending on the required output current, power dissipation, and PCB space. The device is stable with a 4.7-F ceramic output capacitor. Larger output capacitor values can be used to improve the load transient response when higher load currents are required. Thermal Information An influential component of the thermal performance of a package is board design. To take full advantage of the heat dissipation abilities of the PowerPAD or QFN package with exposed thermal die, a board that acts similar to a heatsink and allows for the use of an exposed (and solderable) deep downset pad should be used. For further information see Texas Instrumens application notes (SLMA002) PowerPAD Thermally Enhanced Package and (SLMA004) Power Pad Made Easy. For the QFN package, see the application report (SLUA271) QFN/SON PCB Attachement. Layout Considerations For all switching power supplies, the layout is an important step in the design, especially at high-peak currents and switching frequencies. If the layout is not carefully designed, the regulator might show stability and EMI problems. Therefore, the traces carrying high-switching currents should be routed first using wide and short traces. The input filter capacitor should be placed as close as possible to the input pin VIN of the IC. See the evaluation module (EVM) for a layout example. 18 Submit Documentation Feedback Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 TPS65140-Q1 TPS65145-Q1 www.ti.com ................................................................................................................................................. SGLS277A – NOVEMBER 2004 – REVISED APRIL 2008 L1 3.3uH Vin 3.3V C3 22uF C13 1n TPS65140 R7 15k VIN C1 0.22u D2 C12 0.22u Vo2 −5 V / 20 mA C6 0.22u Vo1 10V / 150 mA D1 COMP GND EN ENR C1+ C1− DRV R3 620k C2 0.22u C2+ C2−/MODE OUT3 C4 22uF R2 56k Vo3 up to 23V/20mA FB3 PG PGND PGND GND FB2 REF FB4 BASE D3 R1 430 C5 6.8pF SW SW FB1 SUP R5 1M R4 150k C7 0.22u R6 56k C11 220nF Vin R7 33k System Power Good Figure 23. Typical Application, Notebook Supply L1 4.7uH Vin 5.0 V C3 22uF C13 2.2n TPS65140 R7 4.3k VIN C1 0.22u D2 C12 0.22u Vo2 −7 V / 20 mA C6 0.22u Vo1 13.5V / 400 mA D1 D3 R3 750k COMP GND EN ENR C1+ C1− DRV R1 820 C5 3.3pF SW SW FB1 SUP R2 75k C2+ C2−/MODE OUT3 FB2 REF FB4 BASE Vo3 up to 23V/20mA FB3 PG PGND PGND GND R5 1M R4 130k C11 220nF C4 22uF C7 0.22u R6 56k Q1 BCP68 Vin C9 1uF Vo4 3.3V/500mA Vin R7 33k C10 4.7uF System Power Good Figure 24. Typical Application, Monitor Supply Copyright © 2004–2008, Texas Instruments Incorporated Product Folder Link(s): TPS65140-Q1 TPS65145-Q1 Submit Documentation Feedback 19 PACKAGE OPTION ADDENDUM www.ti.com 26-Mar-2012 PACKAGING INFORMATION Orderable Device Status (1) Package Type Package Drawing Pins Package Qty Eco Plan (2) Lead/ Ball Finish MSL Peak Temp TPS65140IPWPRQ1 ACTIVE HTSSOP PWP 24 2000 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM TPS65145IPWPRQ1 ACTIVE HTSSOP PWP 24 2000 Green (RoHS & no Sb/Br) CU NIPDAU Level-1-260C-UNLIM (3) Samples (Requires Login) (1) The marketing status values are defined as follows: ACTIVE: Product device recommended for new designs. LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect. NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design. PREVIEW: Device has been announced but is not in production. Samples may or may not be available. OBSOLETE: TI has discontinued the production of the device. (2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability information and additional product content details. TBD: The Pb-Free/Green conversion plan has not been defined. Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes. Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above. Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight in homogeneous material) (3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature. Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the accuracy of such information. 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OTHER QUALIFIED VERSIONS OF TPS65140-Q1, TPS65145-Q1 : • Catalog: TPS65140, TPS65145 NOTE: Qualified Version Definitions: Addendum-Page 1 PACKAGE OPTION ADDENDUM www.ti.com 26-Mar-2012 • Catalog - TI's standard catalog product Addendum-Page 2 IMPORTANT NOTICE Texas Instruments Incorporated and its subsidiaries (TI) reserve the right to make corrections, modifications, enhancements, improvements, and other changes to its products and services at any time and to discontinue any product or service without notice. Customers should obtain the latest relevant information before placing orders and should verify that such information is current and complete. All products are sold subject to TI’s terms and conditions of sale supplied at the time of order acknowledgment. TI warrants performance of its hardware products to the specifications applicable at the time of sale in accordance with TI’s standard warranty. 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