TPS65165 www.ti.com SLVS723 – DECEMBER 2006 Compact LCD Bias IC With High Speed Amplifiers for TV and Monitor TFT-LCD Panels FEATURES • • • • • • • 2.5 V to 6.0 V Input Voltage Range Vs Output Voltage up to 18 V – 1%-Accurate Boost Converter With 4.5 A Switch Current – 600 kHz Fixed Frequency PWM Operation – Overvoltage Protection – Adjustable Softstart Regulated Positive Charge Pump Converter VGH Integrated Gate Voltage Shaping of VGH Regulated Negative Charge-Pump Driver VGL Adjustable Sequencing for Vs and VGH 3 Integrated High-Speed Operational • • • Amplifiers – 50 MHz 3 db Bandwidth – Slew Rate 45 V/µs – 215 mA Short Circuit Current High Voltage Test Mode (HVS) Thermal Shutdown 40-Pin 5×5-mm QFN Package APPLICATIONS • • LCD Monitor LCD TV Panel DESCRIPTION The TPS65165 is a Compact LCD Bias IC with 3 high-speed operational amplifiers for gamma correction and/or VCOM supply. The device generates all 3 voltage rails for TFT-LCD displays (Vs, VGL and VGH). The device incorporates a high-voltage switch that can be controlled by a logic signal from the timing controller (TCON) to provide the gate-voltage modulation for VGH. If this function is not required, the CTRL pin can be tied high. L1 10uH Vin 2.5V to 6.0V Boost Converter C2P Positive Charge Pump x2 or x3 D C2N SUP SUP SW COMP C11 330 nF FB R4 220 kW RHVS FBP S POUT HVS High Voltage Switch Control CTRL C1P DRN C12 330nF C13 1 mF C14 100 pF C1N POS1 R10 1kW Vs 15V/1.0A C9 100 pF R1 470 kW R2 39 kW R3 10 kW C10 2.2 nF C8 1 mF C7 C5 C6 22 mF 22 mF 22 mF C4 1 mF SW C3 22 mF EN C2 22 mF VIN C1 22 mF D1 R6 16 kW PGND 1 NEG1 PGND OUT1 DRVN 3 2 NEG2 C17 C18 C19 22 nF 22 nF 22 nF AGND BGND OUT3 POS3 SS GDLY ADLY C15 330 nF D2 VGL -5V/50mA D3 FBN OUT2 VGH 27.5V/50mA VGH POS2 R5 300 kW R7 160 kW REF C16 330 nF R8 39 kW C20 220 nF Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. PRODUCTION DATA information is current as of publication date. Products conform to specifications per the terms of the Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. Copyright © 2006, Texas Instruments Incorporated TPS65165 www.ti.com SLVS723 – DECEMBER 2006 The device also features a high-voltage stress test, where the output voltage of VGH is typically set to 30 V, and the output voltage of Vs is programmable to any higher voltage. The high-voltage stress test is enabled by pulling the HVS pin high. Adjustable sequencing is implemented, and can be programmed by selecting the capacitor values connected to ADLY and GDLY. The device consists of a boost converter to provide the source voltage Vs operating at a fixed switching frequency of 600 kHz. A fully integrated positive charge pump, switching automatically between doubler and tripler mode provides an adjustable regulated TFT gate on voltage VGH. A negative charge pump driver provides adjustable regulated output voltage VGL. To minimize external components the charge pumps for VGH and VGL operate at a fixed switching frequency of 1.2 MHz. The device includes safety features like overvoltage protection of the boost converter, short circuit protection of VGH and VGL as well as thermal shutdown. ORDERING INFORMATION (1) (1) TA ORDERING QFN PACKAGE PACKAGE MARKING –40°C to 85°C TPS65165RSBT RSB TPS65165 The RSB package is available taped and reeled and shipped in quantities of 3000 devices per reel. ABSOLUTE MAXIMUM RATINGS over operating free-air temperature range (unless otherwise noted) (1) VALUE UNIT Input voltage range VIN (2) –0.3 V to 7.0 V Voltage range at EN, CTRL, HVS -0.3 to 7.0 V –0.3 V to 7.0 V Voltage on SUP 22 SW 25 POUT, VGH, DRN 32 Peak switch current ESD rating Internally limited V HBM 2 kV MM 200 V CDM 500 V Continuous total power dissipation °C Operating ambient temperature range –40 to 85 °C Storage temperature range –65 to 150 °C Operating virtual junction temperature range TA Tstg (2) See Dissipation Rating Table –40 to 150 TJ (1) V Stresses beyond those listed under absolute maximum ratings may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated under recommended operating conditions is not implied. Exposure to absolute–maximum–rated conditions for extended periods may affect device reliability. All voltage values are with respect to network ground terminal. DISSIPATION RATINGS (1) (1) PACKAGE RθJA TA < 25°C POWER RATING TA = 70°C POWER RATING TA = 85°C POWER RATING 40-pin QFN 30 C/W 3.3 W 1.8 W 1.3 W Exposed thermal die is soldered to the PCB using thermal vias. Refer to Texas Instruments Application report (SLUA271) QFN/SON PCB Attachment. RECOMMENDED OPERATING CONDITIONS MIN 2 NOM MAX UNIT VIN Input voltage range 2.5 6.0 V TA Operation ambient temperature –40 85 °C TJ Operating junction temperature –40 125 °C Submit Documentation Feedback TPS65165 www.ti.com SLVS723 – DECEMBER 2006 ELECTRICAL CHARACTERISTICS VIN=5.0V, Vs=15V, HVS=low, EN=CTRL=high, TA = –40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted) PARAMETER TEST CONDITIONS MIN TYP MAX UNIT SUPPLY CURRENT VIN Input voltage range IQVIN No load quiescent current into VIN Device not switching IQSUP No load quiescent current into SUP Device not switching Shutdown current into VIN Vin=6V 1 Shutdown current into SUP Vin=6V, SUP = Vin-0.5V 8 ISD VUVLO 2.5 Under-voltage lockout threshold Thermal shutdown 6.0 V 1 1.5 mA 20 25 mA µA Vin rising 2.3 2.5 V Vin falling 2.2 2.3 V Temperature rising 155 °C 5 °C Thermal shutdown hysteresis LOGIC SIGNALS EN, CTRL, HVS Vth Threshold voltage II Input leakage current Vin = 2.5 V to 6.0 V 0.4 1.4 V ±0.01 ±0.1 µA 28.5 30 31.5 V 0.5 1 1.5 kΩ 100 nA 18 V HIGH VOLTAGE STRESS TEST (HVS) VPOUT Positive charge pump output voltage HVS = high RHVS RHVS pull down resistance HVS = high, Vin = 2.5 V to 6.0 V, IHVS = 100 µA IRHVS RHVS leackage current HVS = low, VRHVS = 1.5 V MAIN BOOST CONVERTER Vs Vs Output voltage range VFB Feedback regulation voltage 7 IFB Feedback input bias current VFB = 1.146 N-MOSFET on-resistance (Q1) Vs = 15 V, ISW = 500 mA, Vs = 7 V, ISW = 500 mA 75 140 P-MOSFET on-resistance (Q2) Vs = 15 V; ISW = 100 mA, Vs = 7 V; ISW = 100 mA 10 16 Ω 1 A 6.6 A 10 µA 1.136 RDS(ON) IMAX Maximum P-MOSFET peak switch current (Q2) ILIM N-MOSFET switch current limit (Q1) Ileak Switch leakage current VSW = 15 V Vovp Output overvoltage protection FB = GND, Vout rising fOSC Oscilator frequency Line regulation Vin=3.0V to 6.0V, Iout=100mA Load regulation Iout=100mA to 700mA, Vin=5.0V 1.146 1.154 V 100 nA mΩ 4.4 5.5 19.5 20 21 V 480 600 720 kHz 0.045 %/V 0.23 %/A NEGATIVE CHARGE PUMP VGL VGL Output voltage range VFB Feedback regulation voltage IFB Feedback input bias current VFB = 0 V Vref Reference voltage VIN = 2.5 V to 6 V, IREF = 10µ A RDSon Q7 P-Channel switch RDSon IDRVN = 40 mA 4.4 IDRN = 40 mA, VFBN = VFBNnominal– 5% 130 300 mV IDRN = 100 mA, VFBN = VFBNnominal– 5% 280 450 mV 30 V 1.214 1.238 V 100 nA VDropN Current sink voltage drop (1) –48 1.205 0 1.213 –2 V 48 mV 100 nA 1.219 V Ω POSITIVE CHARGE PUMP (POUT) VPOUT Output voltage range VFB Feedback regulation voltage CTRL = GND, VGH = open IFB Feedback input bias current FBO = 1.214 V (1) 1.187 The maximum charge pump output current is half the drive current IDRNof the internal current source or sink Submit Documentation Feedback 3 TPS65165 www.ti.com SLVS723 – DECEMBER 2006 ELECTRICAL CHARACTERISTICS (continued) VIN=5.0V, Vs=15V, HVS=low, EN=CTRL=high, TA = –40°C to 85°C, typical values are at TA = 25°C (unless otherwise noted) PARAMETER Vd D1–D4 Schottky diode forward voltage TEST CONDITIONS MIN ID1-D4 = 40 mA TYP MAX UNIT 610 800 mV Doubler Mode (x2); IPOUT = 20 mA 94 Ω Doubler Mode (x2); IPOUT = 50 mA 63 Ω Tripler Mode (x3); IPOUT = 20 mA 141 Ω Tripler Mode (x3); IPOUT = 50 mA 94 Ω POUT to VGH RDSon CTRL = high, POUT = 27 V, I = 20 mA 8.5 16 DRN to VGH RDSon CTRL = low, VDRN = 5 V, I = 20 mA 38 62 IDRN DRN input current CTRL = low, VVGH = VDRN 10 µA tdly CTRL to VGH propagation delay CTRL = high to low, POUT = 27 V, VDRN = GND 125 ns RVGH VGH pull down resistance EN = low, I = 20 mA 1 kΩ Reff Effective output resistance HIGH VOLTAGE SWITCH VGH RDSon Ω Ω CONTROL AND SOFTSTART ADLY, GDLY, SS IADLY Drive current into delay capacitor ADLY VADLY = 1.213 V 3.5 4.8 6.2 µA IGDLY Drive current into delay capacitor GDLY VADLY = 1.213 V 3.5 4.8 6.2 µA ISS SS charge current VSS = 0 V 2.8 4.5 6.2 µA –15 3 18 mV 0 3 µA Vs V OPERATIONAL AMPLIFIERS 1, 2, 3 4 Vos Input offset voltage VCM = Vs/2 IB Input bias current VCM = Vs/2 VCM Common mode input voltage range CMRR Common mode rejection ratio VCM = 7.5 V 55 AVOL Open loop gain 0.5 V ≤ Vout ≤ 14.5 V, No load 50 VOL Output voltage swing low IOUT = 10 mA VOH Output voltage swing high IOUT = 10 mA Isc Short circuit current IO Output current PSRR Power supply rejection ratio 80 dB SR Slew rate AV = 1, VIN = 2 Vpp 45 V/µs BW – 3dB Bandwidth AV = 1, VOUT = 50 mVpp, Output High Impedance 50 MHz GBWP Gain bandwidth product 26 MHz Roff Pull down resistor 10 kΩ 0 VOUT = 7.5V , Input offset voltage 10 mV Submit Documentation Feedback 75 dB dB 100 200 mV 100 Vs–200 mV 120 215 mA 90 170 mA TPS65165 www.ti.com SLVS723 – DECEMBER 2006 C1N C1P POUT FBP DRN VGH FB COMP RHVS NC PIN ASSIGNMENT 40 39 38 37 36 35 34 33 32 31 C2P 2 29 SW BGND 3 28 PGND SUP 4 27 PGND POS1 5 26 NC NEG1 6 25 EN OUT1 7 24 VIN OUT2 8 23 AGND NEG2 9 22 REF POS2 10 11 21 FBN 12 13 14 15 ADLY GDLY SW NC 30 OUT3 1 POS3 C2N Exposed Thermal Die* 17 18 19 20 HVS DRVN NC SS CTRL 16 NOTE: The exposed thermal die is connected to AGNG. NC pin is internally not connected. TERMINAL FUNCTIONS TERMINAL NAME NO. I/O DESCRIPTION C2N 1 I/O Negative terminal of the flying capacitor for the positive charge pump. Typically a 330nF flying capacitor is required. C2P 2 I/O Positive terminal of the flying capacitor for the positive charge pump. Typically a 330nF flying capacitor is required. BGND 3 SUP 4 I Supply input for the operational amplifier and charge pump stages. Connect to the main output Vs, with a 1-µF bypass capacitor. POS1 5 I Non-inverting input of Operational Amplifier 1. NEG1 6 I Inverting input of the Operational Amplifier 1. OUT1 7 O Output of Operational Amplifier 1. When the device is disabled, the output is pulled to GND via a 1-kΩ resistor. OUT2 8 O Output of Operational Amplifier 2. When the device is disabled, the output is pulled to GND via a 1-kΩ resistor. NEG2 9 I Inverting input of Operational Amplifier 2. POS2 10 I Non-inverting input of Operational Amplifier 2. POS3 11 I Non-inverting input of Operational Amplifier 3. OUT3 12 O Output of Operational Amplifier 3. When the device is disabled, the output is pulled to GND via a 1-kΩ resistor. NC Low noise ground for the operational amplifier. 13, 19, 26, 31 Not connected. These pin can be connected to GND to improve the thermal resistance of the package. ADLY 14 O Adjustable EN high-to-start-up delay of the main boost converter, negative and positive charge pump. Connect a capacitor from this pin to GND to set the desired delay time. (See SETTING THE DELAY TIMES ADLY, GDLY) GDLY 15 O Adjustable EN high-to-enable delay of the high-voltage switch Q8 (gate voltage shaping). Connect a capacitor from this pin to GND to set the desired delay time. (See SETTING THE DELAY TIMES ADLY, GDLY) CTRL 16 I Logic control input for the internal high voltage switch (gate voltage shaping). Submit Documentation Feedback 5 TPS65165 www.ti.com SLVS723 – DECEMBER 2006 PIN ASSIGNMENT (continued) TERMINAL FUNCTIONS (continued) TERMINAL NAME 6 NO. I/O DESCRIPTION HVS 17 I Logic control input to enable High Voltage Stress Test. With HVS=low the high voltage stress test is disabled. With HVS=high the high voltage stress test is enabled. DRVN 18 I/O Drive pin for the negative charge pump converter generating VGL. Using a single stage charge pump inverts the voltage present at the main boost converter Vs and regulates it down to the desired voltage programmed by the feedback divider. SS 20 O Softstart for the main boost converter generating Vs. Connect a capacitor to this pin to set the softstart time. FBN 21 I Feedback of the negative charge pump converter. REF 22 O Reference output. Connect a 220-nF capacitor directly from REF pin to AGND to minimize possible noise coupling into the reference of the IC. AGND 23 I Analog Ground , positive and negative charge pump ground. VIN 24 I Supply pin for the IC. Bypass this pin with a 1-µF capacitor directly to GND. EN 25 I Enable pin of the IC. EN=high enables the IC. EN= low disables the IC. This pin must be terminated. PGND 27, 28 SW 29, 30 I/O Power Ground for the boost converter. RHVS 32 I This resistor sets the voltage of the boost converter Vs when the High Voltage Stress test is enabled. (HVS=high). With HVS=high the RHVS pin is pulled to GND which sets the output voltage for the boost converter. When HVS is disabled (HVS=low) the RHVS pin is high impedance. COMP 33 O Compensation for the regulation loop of the boost converter generating Vs. FB 34 I Feedback of the boost converter generating Vs. VGH 35 O This is the output voltage of the internal high voltage switch, controlled by the CTRL signal. DRN 36 I/O Connect the discharge resistor for the Gate voltage shaping to this pin. FBP 37 I This is the feedback for the positive charge pump converter generating VGH POUT 38 O Output of the positive charge pump which is internally connected to the high voltage switch Q2. Connect a 1-µF output capacitor to this pin as well as the feedback divider to set the output voltage for the positive charge pump respectively for VGH. C1P 39 I/O Positive terminal of the flying capacitor for the positive charge pump. Typically a 330-nF flying capacitor is required. C1N 40 I/O Negative terminal of the flying capacitor for the positive charge pump. Typically a 330-nF flying capacitor is required. Switch pin of the boost regulator generating Vs Submit Documentation Feedback TPS65165 www.ti.com SLVS723 – DECEMBER 2006 BLOCK DIAGRAM L1 10uH Vin 2.5V to 6.0V Boost Converter C2P Positive Charge Pump x2 or x3 D C2N SUP SUP SW COMP C11 330 nF FB R4 220 kW RHVS FBP S POUT HVS High Voltage Switch Control CTRL C1P DRN C12 330nF C13 1 mF C14 100 pF C1N POS1 PGND OUT1 DRVN 3 2 NEG2 C15 330 nF D2 VGL -5V/50mA D3 C17 C18 C19 22 nF 22 nF 22 nF AGND BGND OUT3 POS3 SS GDLY FBN OUT2 VGH 27.5V/50mA VGH POS2 R5 300 kW R6 16 kW PGND 1 NEG1 ADLY R10 1kW Vs 15V/1.0A C9 100 pF R1 470 kW R2 39 kW R3 10 kW C10 2.2 nF C8 1 mF C7 C5 C6 22 mF 22 mF 22 mF C4 1 mF SW C3 22 mF EN C2 22 mF VIN C1 22 mF D1 R7 160 kW REF C16 330 nF R8 39 kW C20 220 nF Submit Documentation Feedback 7 TPS65165 www.ti.com SLVS723 – DECEMBER 2006 TYPICAL CHARACTERISTICS TABLE OF GRAPHS FIGURE MAIN BOOST CONVERTER (Vs) η Efficiency, main boost converter Vs vs Load current Figure 1 PWM operation at nominal load current Figure 2 PWM operation at light load current Figure 3 Load transient response Figure 4 Softstart boost converter VS Figure 5 Overvoltage protection Figure 6 SYSTEM FUCTIONALITY Power-on sequencing Figure 7 Gate voltage shaping VGH Figure 8 NEGATIVE CHARGE PUMP DRIVER VGL vs load current Figure 9 vs load current (doubler mode) Figure 10 Input to output offset voltage vs Opamp 1 load current Figure 11 Input to output offset voltage vs Opamp 2 load current Figure 12 Input to output offset voltage vs Opamp 3 load current Figure 13 POSITIVE CHARGE PUMP VGH VCOM BUFFERS EFFICIENCY, MAIN BOOST CONVERTER vs LOAD CURRENT 100 EFFICIENCY – % 90 VS=15 VGH=VGL=no load, switching OpAmp operating, no load PWM OPERATION NOMINAL LOAD CURRENT VSW 10 V / div VIN = 5.0 V VO 50 mV / div 80 VIN = 3.3 V 70 IL 1A / div VIN = 2.7 V 60 50 VIN = 5.0 V VO = 15 V / 700 mA 40 0 200 400 IOUT 600 – mA 800 1000 Figure 1. 8 400 ns / div Figure 2. Submit Documentation Feedback TPS65165 www.ti.com SLVS723 – DECEMBER 2006 PWM OPERATION AT LIGHT LOAD CURRENT VSW 10 V / div LOAD TRANSIENT RESPONSE VO1 200 mV / div VO 50 mV / div VIN = 5.0 V VS = 15 V VIN = 5.0 V VO = 15 V / 50 mA IL 100 mA to 600 mA IL 1A / div 100 ms / div 400 ns / div VS 10 V / div Figure 3. Figure 4. SOFTSTART BOOST CONVERTER - VS OVERVOLTAGE PROTECTION VIN = 5.0 V VO = 15 V IOUT = 100 mA CSS = 100 nF VIN = 5.0 V VO = 15 V / 100 mA VSW 10 V / div VO 2 V / div 16 V offset VIN 5V / div IL 500 mA / div IL 2A / div 2.0 ms / div 40 ms / div Figure 5. Figure 6. POWER-ON SEQUENCING GATE VOLTAGE SHAPING - VGH VIN 5 V / div CTRL 5V / div VS 10 V / div VGH 10 V / div VGH 20 V / div VGL 5 V / div VIN = 5.0 V VO = 15 V ADLY = 22 nF GDLY = 47nF DRN = 1.5 kW to VS VGH = 1 nF capacitive load to represent panel 4.0 ms / div 2.0 ms / div Figure 7. Figure 8. Submit Documentation Feedback 9 TPS65165 www.ti.com SLVS723 – DECEMBER 2006 VGH vs LOAD CURRENT - DOUBLER STAGE VGL vs LOAD CURRENT 24.6 –4.80 VS = 15 V VGL = 5 V VS = 15 V VGH = 24 V COUT = 2 ´ 1 mF / 35 V 24.4 –4.85 24.2 24.0 –4.90 TA = –40°C TA = 25°C TA = 85°C VGH – V VGL – V 23.8 TA = –40°C TA = 25°C TA = 85°C –4.95 –5.00 23.6 23.4 23.2 23.0 –5.05 22.8 22.6 –5.10 10 20 30 40 50 60 70 80 90 0 100 50 60 70 80 90 INPUT TO OUTPUT OFFSET VOLTAGE vs OPAMP 1 LOAD CURRENT INPUT TO OUTPUT OFFSET VOLTAGE vs OPAMP 2 LOAD CURRENT 15 10 10 5 5 0 IOUT = –80 IOUT = –50 IOUT = –20 IOUT = –10 IOUT = no load IOUT = 10 IOUT = 20 IOUT = 50 IOUT = 80 –5 –10 Unity Gain Buffer VS = 15 V –20 2 3 4 5 6 7 8 9 –15 Unity Gain Buffer VS = 15 V –20 0 1 2 3 4 5 6 7 8 9 VIN – V Figure 11. Figure 12. INPUT TO OUTPUT OFFSET VOLTAGE vs OPAMP 3 LOAD CURRENT 20 15 10 5 VOFFSET – V IOUT = –80 IOUT = –50 IOUT = –20 IOUT = –10 IOUT = no load IOUT = 10 IOUT = 20 IOUT = 50 IOUT = 80 –5 –10 10 11 12 13 14 15 0 IOUT = –80 IOUT = –50 IOUT = –20 IOUT = –10 IOUT = no load IOUT = 10 IOUT = 20 IOUT = 50 IOUT = 80 –5 –10 –15 Unity Gain Buffer VS = 15 V –20 0 1 2 3 4 5 6 7 8 9 100 0 VIN – V 10 11 12 13 14 15 VIN – V Figure 13. 10 40 Figure 10. 15 1 30 Figure 9. 20 0 20 IOUT – mA 20 –15 10 IOUT – mA VOFFSET – V VOFFSET – V 0 Submit Documentation Feedback 10 11 12 13 14 15 TPS65165 www.ti.com SLVS723 – DECEMBER 2006 APPLICATION INFORMATION SUP UVLO Undervoltage lockout 2.3V typ Vref 1.213V VIN AGND Regulator 1 Clamps at 13.6V Thermal Shutdown latch 155deg°C typ Regulator 2 4.8V typ Driver supply Driver Logic supply 30uA REF Vin Voltage clamp 5.8V max EN Ichg Vref ADLY Start Boost converter, negative and positive charge pump 3.5k Vin Voltage clamp 5.8V max Ichg GDLY Vref Enable Gate voltage shaping block 3.5k Figure 14. Control Block TPS65165 THERMAL SHUTDOWN The thermal-shutdown feature prevents damage from excessive heat and power dissipation. Typically the thermal-shutdown threshold is 155°C. When the device enters thermal-shutdown then the device does not restart automatically. The device can only be restarted by cycling the input voltage below its undervoltage-lockout threshold or by cycling the enable EN to ground. Submit Documentation Feedback 11 TPS65165 www.ti.com SLVS723 – DECEMBER 2006 APPLICATION INFORMATION (continued) UNDERVOLTAGE LOCKOUT To avoid device malfunction at low input voltages, an undervoltage lockout is included which enables the device only when the input voltage exceeds 2.3 V. REFERENCE OUTPUT, REF The device provides a reference output that is used to regulate the negative charge pump. In order to have a stable reference voltage, a 220-nF bypass capacitor is required, connected directly from REF to AGND. The reference output has a current capability of 30 µA which should not be exceeded. Because of this, the feedback resistor value from FBN to REF should not be smaller than 40 kΩ. START-UP SEQUENCING Start-up sequencing can be controlled by adjusting the delay times ADLY and GDLY. After the delay time set by ADLY passed by, the boost converter, negative and positive charge pumps start at the same time. VGH will only go high once the delay time, set by GDLY passed by and the signal applied to CTRL is high. EN VGH With CTRL = High GDLY ADLY Fall time depends output capacitor value and load current Vs VGL Figure 15. Power-On Sequencing SETTING THE DELAY TIMES ADLY, GDLY Connecting an external capacitor to the ADLY and GDLY pin sets the delay time. If no delay time is required these pins can be left open. To set the delay time, the external capacitor connected to ADLY and GDLY is charged with a constant current source (typically 5 µA). The delay time is terminated when the capacitor voltage has reached the internal reference voltage of Vref = 1.213 V. The external delay capacitor is calculated by: 5 mA td 5 mA td C dly + + Vref 1.213V (1) with td = Desired delay time 12 Submit Documentation Feedback TPS65165 www.ti.com SLVS723 – DECEMBER 2006 APPLICATION INFORMATION (continued) BOOST CONVERTER The TPS65165 boost converter block is shown in Figure 16. The boost converter operates with PWM (Pulse Width Modulation) and a fixed switching frequency of 600 kHz. The converter uses a unique fast-response, voltage-mode controller scheme with input voltage feedforward. This achieves excellent line and load regulation (0.2%/A load regulation typical) and allows the use of small external components. To increase the flexibility in the selection of external component values, the device uses external loop compensation. Although the boost converter looks like a non-synchronous boost converter topology operating in discontinuous conduction mode, at light loads, the TPS65165 maintains continuous conduction even at minimal load currents. This is achieved with a novel architecture using an external Schottky diode with an integrated MOSFET in parallel connected between SW pin and the SUP pin. The purpose of this MOSFET is to allow the current to go below ground, which is the case at light load conditions. For this purpose, a small integrated P-Channel MOSFET (Q2) with a typical RDSON of 10 Ω is sufficient. When the inductor current is positive, the external Schottky diode with the lower forward voltage carries the current. This causes the converter to operate with a fixed frequency in continuous-conduction mode over the entire load-current range. This avoids ringing on the switch pin as seen with a typical non-synchronous boost converter, and allows a simpler compensation network. SW VIN SW Softstart Vref Q2 SS 600 kHz Oszillator 70Ohm SUP Current limit and Soft Start EN Comparator Control Logic Q1 COMP GM Amplifier PGND Sawtooth Generator FB VFB 1.154V Overvoltage Comparator OVP GM Amplifier Low Gain SUP PGND Vref VFB 1.154 RHVS HVS Figure 16. Boost Converter Block TPS65165 SOFTSTART BOOST CONVERTER To minimize inrush current during start-up, an external capacitor connected to the softstart pin (SS) is used to slowly ramp up the internal current limit of the boost converter. The larger the capacitor, the slower the ramp-up of the current limit, and the longer the softstart time. A 22-nF capacitor is usually sufficient for typical applications. Submit Documentation Feedback 13 TPS65165 www.ti.com SLVS723 – DECEMBER 2006 APPLICATION INFORMATION (continued) HIGH VOLTAGE STRESS TEST BOOST CONVERTER AND POSITIVE CHARGE PUMP The TPS65165 incorporates a high voltage stress test where the output voltage of the boost converter Vs and the positive charge pump POUT are set to a higher output voltage compared to the nominal programmed output voltage. The High Voltage Stress test is enabled by pulling the HVS pin to high. With HVS=high the voltage on POUT, respectively VGH is regulated to a fixed output voltage of 30 V. The boost converter Vs is programmed to a higher voltage determined by the resistor connected to RHVS. With HVS=high the RHVS pin is pulled to GND which sets the voltage for the boost converter during the High Voltage Stress Test. The output voltage for the boost converter during high voltage stress test is calculated as: R1 ) R2 ø R4 R1 ) R2 ø R4 Vs HVS + VFB + 1.146V R2 ø R4 R2 ø R4 R4 + ǒ R1 VsHVS VFB Ǔ *1 R2 R2 * R1 (2) With: VsHVS = Boost converter output voltage with HVS=high VFB = 1.146V R4 = Resistor connected to pin RHVS OVERVOLTAGE PROTECTION BOOST CONVERTER The boost converter has an integrated overvoltage-protection circuit to prevent the switch voltage from exceeding the absolute maximum switch voltage rating in the event of a system fault. The device protects itself if the feedback pin is shorted to ground by clamping the boost-converter output voltage to 20 V. To implement the overvoltage protection, the overvoltage comparator shown in Figure 16 monitors the output voltage via the SUP pin. When the output voltage exceeds the overvoltage threshold of typically 20 V, the device stops switching until the output voltage drops below the comparator threshold again. The typical waveform is shown in Figure 6. INPUT CAPACITOR SELECTION VIN, SUP Low-ESR ceramic capacitors are recommended for good input-voltage filtering. The TPS65165 has an analog input (VIN) and a power supply input (SUP) powering all the internal rails, including the operational amplifiers. 1-µF bypass capacitors are required as close as possible from VIN to GND, and from SUP to GND. Depending on the overall load current, two or three 22-µF input capacitors are required. For better input-voltage filtering, the input capacitor values can be increased. Refer to Table 1 and typical applications for input capacitor recommendations. Table 1. Input Capacitor Selection 14 CAPACITOR VOLTAGE RATING COMPONENT SUPPLIER 22µF/1210 6.3V Taiyo Yuden Cin 1µF/1206 6.3V Taiyo Yuden Bypass AVIN, SUP Submit Documentation Feedback COMMENTS TPS65165 www.ti.com SLVS723 – DECEMBER 2006 BOOST CONVERTER DESIGN PROCEDURE The first step in the design procedure is to verify whether the maximum possible output current of the boost converter supports the specific application requirements. To simplify the calculation the fastest approach is to estimate the converter efficiency, by taking the efficiency numbers from the provided efficiency curves or to use a worst case assumption for the expected efficiency, e.g., 80%. With the efficiency number it is possible to calculate the steady state values of the application. 1. Converter Duty Cycle: Vin h D +1* Vout (3) 2. Maximum output current: ǒ Ǔ Iout + Isw * Vin D 2 ƒs L (1 * D) (4) 3. Peak switch current: I I swpeak + Vin D ) out 2 ƒs L I * D (5) With: Isw = converter switch current (minimum switch current limit=4.4 A) fs = converter switching frequency (typical 600kHz) L = Selected inductor value η = Estimated converter efficiency (use the number from the efficiency curves or 0.8 as an estimation) The peak switch current is the steady state peak switch current the integrated switch, inductor and external Schottky diode has to be rated for. The calculation must be done for the minimum input voltage where the peak switch current is highest. INDUCTOR SELECTION The TPS65165 typically operates with a 10-µH inductor. The main parameter for inductor selection is the inductor saturation current. This should be higher than the peak switch current as calculated in Equation 5, with additional margin for heavy load transients. An alternative, more conservative approach is to choose the inductor with saturation current at least as high as the typical switch current limit of 5.5 A. The second important parameter is the inductor DC resistance. Usually the lower the DC resistance the higher the efficiency of the converter. The choice of an inductor can affect converter efficiency by as much as 10%. Possible inductors are shown in Table 2. Table 2. Inductor Selection Boost Converter INDUCTOR VALUE COMPONENT SUPPLIER DIMENSIONS IN mm Isat/DCR 10 µH Sumida CDRH8D38-100 8.3×8.3×4.0 3.0 A / 38 mΩ 10 µH Wuerth 744066100 10×10×3.8 4.0 A / 25 mΩ 10 µH Coilcraft DO3316P-103 12.95×9.4×5.51 3.8 A / 3838 mΩ Submit Documentation Feedback 15 TPS65165 www.ti.com SLVS723 – DECEMBER 2006 OUTPUT CAPACITOR SELECTION For best output voltage filtering, a low-ESR output capacitor is recommended. Ceramic capacitors have a low ESR value and work best with the TPS65165. Three 22µF or six 10uF ceramic output capacitors in parallel are sufficient for most applications. More capacitors can be added to improve the load transient regulation. Refer to Table 3 for details on selecting output capacitors. Table 3. Output Capacitor Selection CAPACITOR VOLTAGE RATING COMPONENT SUPPLIER 22 µF / 1812 16 V Taiyo Yuden EMK432BJ226MM Rectifier diode selection To achieve high efficiency, a Schottky diode should be used. The reverse voltage rating should be higher than the maximum output voltage of the converter. The current rating for the Schottky diode is calculated as the off time of the converter times the typical switch current of the TPS65165: I avg + (1 * D) Isw + Vin 5.5A Vout (6) where Isw = the typical switch current of the TPS65165 (5.5 A) A Schottky diode with 2-A maximum average rectified forward current rating is sufficient for most applications. The Schottky rectifier must have adequate power dissipation. The dissipated power is the average rectified forward current times the diode forward voltage. Vin P D + I avg VF + Isw (1 * D) Vsw VF Vout (7) where Isw = typical switch current of the TPS65165 (5.5 A) Table 4. Rectifier Diode Selection (Boost Converter) CURRENT RATING Iavg Vr Vforward RθJA SIZE COMPONENT SUPPLIER 3A 20 V 0.36 V at 3 A 46°C/W SMC MBRS320, International Rectifier 2A 20 V 0.44 V at 2 A 75°C/W SMB SL22, Vishay Semiconductor 2A 20 V 0.5 V at 2 A 75°C/W SMB SS22, Fairchild Semiconductor SETTING THE OUTPUT VOLTAGE AND SELECTING THE FEEDFORWARD CAPACITOR The output voltage is set by the external resistor-divider value, and is calculated as: V out + 1.146V ǒ1 ) R1 Ǔ R2 (8) Across the upper resistor, a bypass capacitor is required to speed up the circuit during load transients. The capacitor value is caluculated as: 1 1 Cƒƒ + + 2 p ƒ z R1 2 p 4000 R1 (9) A standard value nearest to the calculated value should be used. COMPENSATION (COMP) The regulator loop can be compensated by adjusting the external components connected to the COMP pin. The COMP pin is the output of the internal transconductance error amplifier. A single capacitor connected to this pin sets the low-frequency gain. A 2.2-nF capacitor is sufficient for most of the applications. Adding a series resistor sets an additional zero and increases the high-frequency gain. Equation 10 calculates the frequency where the resistor increases the high frequency gain. 16 Submit Documentation Feedback TPS65165 www.ti.com SLVS723 – DECEMBER 2006 ƒz + 2 p 1 Cc Rc (10) Lower input voltages require a higher gain, and therefore a lower compensation-capacitor value. Refer to the typical applications for the appropriate component selection. POSITIVE CHARGE PUMP The fully-integrated positive charge pump automatically switches its gain between doubler and tripler mode. As shown in Figure 17, the input voltage of the positive charge pump is the SUP pin, that is connected to the output of the main boost converter (Vs). The charge pump requires two 330-nF flying capacitors and a 1-µF output capacitance for stable operation. The positive charge pump also supports the high-voltage stress test by pulling the HVS pin high. This programs the output voltage to a fixed output voltage of 30 V by using the internal voltage divider as shown in Figure 17. During normal operation the HVS pin is pulled low, and the output voltage is programmed with the external voltage divider. ǒ1 ) R5 Ǔ R6 V out + 1.213V R5 + R6 ǒ Ǔ Vout * 1 + R6 V FB ǒ Ǔ Vout *1 1.213 (11) To minimize noise and leakage-current sensitivity, we recommend a value of approximately 20 kΩ for the lower feedback divider resistor R6. A 100-pF feedforward capacitor across the upper feedback resistor R5 is typically required. FBP 600 kHz POUT f clockx 2 1.2MHz SU P HVS select Q3 Vref 1 .213 V HVS Control Logic Automatic Gain select (doubler or tripler mode ) C1 N Q4 Softstart C 1P Q6 SUP =Vs POUT D3 D0 I DRVP D1 C 2P D2 Q5 C2 N AGND Figure 17. Positive Charge Pump Block TPS65165 Submit Documentation Feedback 17 TPS65165 www.ti.com SLVS723 – DECEMBER 2006 NEGATIVE CHARGE PUMP DRIVER The negative charge pump provides a regulated output voltage set by the external resistor divider. It inverts the voltage applied to the SUP pin (the boost-converter output voltage), and regulates it to the programmed voltage. SUP 600 kHz Q7 DRVN f clock x2 1 .2MHz Control Logic Softstart I DRVN AGND FBN Vref 0V Figure 18. Negative Charge Pump Block TPS65165 The output voltage is VGL = (–Vs) + VDROP. VDROP is the voltage drop across the external diodes and internal charge pump MOSFETs. Setting the output voltage: V out + *VREF R7 + *1.213V R8 R7 + R8 ŤV outŤ VREF + R8 R7 R8 (12) ŤV outŤ 1.213 (13) Since the reference-output driver current should typically not exceed 30 µA, the lower feedback-resistor value R8 should be in a range of 40 kΩ to 120 kΩ. The negative charge pump requires two external Schottky diodes. The peak current rating of the Schottky diode must be twice the load current of the output. For a 20-mA output current, the dual Schottky diode BAT54 is a good choice. HIGH VOLTAGE SWITCH CONTROL (Gate Voltage Shaping) For correct operation of this block it is not recommended to connect an output capacitor to VGH. If the output shows higher output ripple voltage than expected then the output capacitor value on POUT needs to be increased instead. The device has an integrated high-voltage switch to provide gate-voltage modulation of VGH. If this feature is not required, then CTRL pin must be pulled high or connected to VIN. When the device is disabled or the input voltage is below the undervoltage lockout (UVLO), both switches (Q4 and Q5) are off, and VGH is discharge by a 1-kΩ resistor over Q8, as shown in Figure 19. 18 Submit Documentation Feedback TPS65165 www.ti.com SLVS723 – DECEMBER 2006 Power Good FB Power Good FBP Power Good FBN UVLO EN POUT CTRL Vref GDLY Q4 3.5 k I DLY EN VGH Control Voltage clamp 5 . 8 V max CTRL=high Q4 on, Q5 off CTRL=low Q4 off, Q5 on EN=low Q4 and Q5 off, Q8 on 1 kW Q5 Vin Q8 DRN Vs R10 10k R10 1k R10 10k Option 1 Option 2 Figure 19. High Voltage Switch (Gate Voltage Shaping) Block TPS65165 To implement gate-voltage shaping, the control signal from the LCD timing controller (TCON) is connected to the CTRL pin. CTRL is activated when the device is enabled, the input voltage is above the undervoltage lockout, all the output voltages (Vs, VGL, VGH) are in regulation, and the delay time is set by the GDLY pin passed by. As soon as one of the outputs is pulled below its Power Good level, Q4 and Q5 are turned off and VGH is discharged via a 1-kΩ resistor over Q8. Submit Documentation Feedback 19 TPS65165 www.ti.com SLVS723 – DECEMBER 2006 With CTRL = high, Q4 is turned on and the charge pump output voltage is present at VGH. When the CTRL pin is pulled low, Q4 is turned off and Q5 is turned on, discharging VGH. The slope and time for discharging VGH is determined by the LCD capacitance and the termination on DRN. An additional output capacitor is not recommended on VGH. There are basically two options available to terminate the DRN pin, depending on the LCD capacitance and required overall converter efficiency. VH VGH VL CTRL toff T Figure 20. High Voltage Switch (Gate Voltage Shaping) Timing Diagram Option 1 in Figure 19 draws no current from Vs, and is therefore better in terms of converter efficiency. The voltage level VL (the discharge level of VGH) is determined by the LCD capacitance, the resistor connected to DRN and the off time, toff. The lower the resistor value connected to DRN, the lower the discharge voltage level VL. Option 2 in Figure 19 constantly draws current from Vs due to the voltage divider connected to Vs. The advantage of this solution is that the low-level voltage VL is given by the voltage divider, assuming the feedback resistor values are small, allowing the LCD capacitance to discharge during toff. This solution is not recommended for very large display panels because the feedback divider resistor values must be too low, drawing too much current from Vs. 20 Submit Documentation Feedback TPS65165 www.ti.com SLVS723 – DECEMBER 2006 Operational Amplifier 1, 2 and 3 The TPS65165 has three integrated operational amplifiers. Opamp 3 is already configured as a standard buffer as shown in Figure 21. The operational amplifiers can be used as a gamma correction buffer or as a VCOM buffer. SUP Series Resistor for Driving POS1 Capacitive Loads 3.3R EN 1 10k NEG1 OUT1 10k 10nF POS2 2 NEG2 10k 3 OUT3 BGND POS3 OUT2 Figure 21. Operational Amplifier Block The opamp power supply pin is the SUP pin connected to the boost converter Vs. To achieve good performance and minimize the output noise, a 1-µF bypass capacitor is required directly from the SUP pin to ground. When the device is disabled, the opamp outputs are pulled low via a 10kΩ resistor. The opamps are not designed to drive capacitive loads; therefore it is not recommended to connect a capacitor directly to the opamp outputs. If capacitive loads are driven we recommend using a series resistor at the output to provide stable operation. With a 3.3-Ω series resistor, a capacitive load of 10 nF can be driven, which is usually sufficient for typical LCD applications. Operational Amplifier Termination The TPS65165 has three integrated operational amplifiers. For some applications, not all of the amplifiers are used. To minimize device quiescent current the terminals need to be terminated. For the unity gain buffer, OpAmp 3 the positive terminal is connected to GND and the output is left open. For OpAmp 2 and 3 the negative terminal is connected to the OpAmp output and the positive terminal is connected to GND. Using such a termination minimizes device quiescent current and correct functionality of the device. Submit Documentation Feedback 21 TPS65165 www.ti.com SLVS723 – DECEMBER 2006 PCB Layout Design Guidelines: 1. Place the power components outlined in bold first on the PCB. 2. Rout the traces outlined in bold with wide PCB traces 3. Place a 1-µF bypass capacitor directly from the Vin pin to GND since this is the supply pin for internal circuits. 4. Place a 1-µF bypass capacitor directly from the SUP pin to GND since this is the supply pin for internal circuits. 5. Use a short and wide trace to connect the SUP pin to the output of the boost converter Vs. 6. Place the 220-nF reference capacitor directly from REF to AGND close to the IC pins. 7. The feedback resistor for the negative charge pump between FBN and REF needs to be >40kΩ. 8. Use short traces for the charge pump drive pin (DRVN) of VGL because the traces carry switching waveforms. 9. Place the flying capacitors as close as possible to the C1P, C1N and C2P, C2N pin. 10. Solder the Power Pad of the QFN package to GND and use thermal vias to lower the thermal resistance L1 10 mH Vin 5.0 V C11 C10 2.2 nF 330nF 29 30 33 2 COMP FB C2P RHVS C2N FBP HVS POUT 1 17 16 36 5 R10 1kW R2 39 kW 4 SUP 24 25 SW R3 10 kW 6 7 CTRL C1P DRN C1N TPS65165 POS1 PGND NEG1 PGND OUT1 VGH 10 POS2 9 DRVN NEG2 14 15 20 C17 C18 C19 22 nF 22 nF 22 nF 11 3 12 AGND OUT3 BGND POS3 SS GDLY OUT2 ADLY FBN 8 Vs 15V/1.0A C9 100 pF R1 470 kW C8 1 mF C7 C5 C6 22 mF 22 mF 22 mF C4 1 mF SW C3 22 mF EN C2 22 mF VIN C1 22 mF D1 23 REF 34 R4 220 kW 32 37 38 39 C13 1 mF C12 40 330nF 27 VGH 27.5V/50mA 35 18 21 C15 330 nF D2 VGL -5V/50mA D3 22 R7 160 kW R8 39 kW C20 220 nF Submit Documentation Feedback R5 300 kW R6 16 kW 28 Figure 22. Layout Recommendation 22 C14 100 pF C16 330 nF TPS65165 www.ti.com SLVS723 – DECEMBER 2006 TYPICAL APPLICATION L1 10 mH Vin 5.0 V C11 C10 2.2 nF 330nF 29 30 33 2 COMP FB C2P RHVS C2N FBP HVS POUT 1 17 16 36 5 6 7 CTRL C1P DRN C1N TPS65165 POS1 PGND NEG1 PGND OUT1 VGH 10 POS2 9 DRVN NEG2 14 15 20 11 3 12 C17 C18 C19 22 nF 22 nF 22 nF AGND OUT3 BGND POS3 OUT2 SS 8 GDLY FBN ADLY R10 1kW R2 39 kW 4 SUP 24 25 SW R3 10 kW Vs 15V/1.0 A C9 100 pF R1 470 kW C8 1 mF C7 C5 C6 22 mF 22 mF 22 mF C4 1 mF SW C3 22 mF EN C2 22 mF VIN C1 22 mF D1 23 REF 34 R4 220 kW 32 37 38 39 C13 1 mF C12 40 330 nF C14 100 pF 27 R6 16 kW 28 VGH 27.5 V/50 mA 35 18 21 22 R5 300 kW C15 330 nF D2 VGL -5 V/50 mA D3 R7 160 kW C16 330 nF R8 39 kW C20 220 nF Figure 23. 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