INTERSIL ISL6236A

ISL6236A
®
Data Sheet
March 18, 2008
High-Efficiency, Quad-Output, Main Power
Supply Controllers for Notebook
Computers
The ISL6236A dual step-down, switch-mode power-supply
(SMPS) controller generates logic-supply voltages in
battery-powered systems. The ISL6236A includes two
pulse-width modulation (PWM) controllers, 5V/3.3V and
1.5V/1.05V. The output of SMPS1 can also be adjusted from
0.7V to 5.5V. The SMPS2 output can be adjusted from 0V to
2.5V by setting REFIN2 voltage. An optional external charge
pump can be monitored through SECFB. This device features
a linear regulator providing 3.3V/5V, or adjustable from 0.7V to
4.5V output via LDOREFIN. The linear regulator provides up
to 100mA output current with automatic linear-regulator
bootstrapping to the BYP input. When in switchover, the LDO
output can source up to 200mA. The ISL6236A includes
on-board power-up sequencing, the power-good (POK)
outputs, digital soft-start, and internal soft-stop output
discharge that prevents negative voltages on shutdown.
A constant on-time PWM control scheme operates without
sense resistors and provides 100ns response to load
transients while maintaining a relatively constant switching
frequency. The unique ultrasonic pulse-skipping mode
maintains the switching frequency above 25kHz, which
eliminates noise in audio applications. Other features include
pulse skipping, which maximizes efficiency in light-load
applications, and fixed-frequency PWM mode, which reduces
RF interference in sensitive applications.
Ordering Information
PART
NUMBER
(Note)
ISL6236AIRZ
Features
• Wide Input Voltage Range 4.5V to 25V
• Dual Fixed 1.05V/3.3V and 1.5V/5.0V Outputs or
Adjustable 0.7V to 5.5V (SMPS1) and 0V to 2.5V
(SMPS2), ±1.5% Accuracy
• Secondary Feedback Input (Maintains Charge Pump
Voltage)
• 1.7ms Digital Soft-Start and Independent Shutdown
• Fixed 3.3V/5.0V, or Adjustable Output 0.7V to 4.5V,
±1.5% (LDO): 200mA
• 3.3V Reference Voltage ±2.0%: 5mA
• 2.0V Reference Voltage ±1.0%: 50µA
• Constant ON-TIME Control with 100ns Load-Step
Response
• Frequency Selectable
• rDS(ON) Current Sensing
• Programmable Current Limit with Foldback Capability
• Selectable PWM, Skip or Ultrasonic Mode
• BOOT Voltage Monitor with Automatic Refresh
• Independent POK1 and POK2 Comparators
• Soft-Start with Pre-Biased Output and Soft-Stop
• Independent ENABLE
• High Efficiency - up to 97%
• Very High Light Load Efficiency (Skip Mode)
TEMP
RANGE
(°C)
PART
MARKING
FN6453.3
PACKAGE
(Pb-free)
PKG.
DWG. #
ISL6236 AIRZ -40 to +100 32 Ld 5x5 QFN L32.5x5B
ISL6236AIRZ-T* ISL6236 AIRZ -40 to +100 32 Ld 5x5 QFN L32.5x5B
(Tape and Reel)
*Please refer to TB347 for details on reel specifications.
NOTE: These Intersil Pb-free plastic packaged products employ special
Pb-free material sets; molding compounds/die attach materials and 100%
matte tin plate PLUS ANNEAL - e3 termination finish, which is RoHS
compliant and compatible with both SnPb and Pb-free soldering operations.
Intersil Pb-free products are MSL classified at Pb-free peak reflow
temperatures that meet or exceed the Pb-free requirements of IPC/JEDEC
J STD-020.
• 5mW Quiescent Power Dissipation
• Thermal Shutdown
• Extremely Low Component Count
• Pb-Free (RoHS Compliant)
Applications
• Notebook and Sub-Notebook Computers
• PDAs and Mobile Communication Devices
• 3-Cell and 4-Cell Li+ Battery-Powered Devices
• DDR1, DDR2, and DDR3 Power Supplies
• Graphic Cards
• Game Consoles
• Telecommunications
1
CAUTION: These devices are sensitive to electrostatic discharge; follow proper IC Handling Procedures.
1-888-INTERSIL or 1-888-468-3774 | Intersil (and design) is a registered trademark of Intersil Americas Inc.
Copyright Intersil Americas Inc. 2007, 2008. All Rights Reserved
All other trademarks mentioned are the property of their respective owners.
ISL6236A
Pinout
2
REFIN2
ILIM2
OUT2
SKIP
POK2
EN2
UGATE2
PHASE2
ISL6236A
(32 LD 5x5 QFN)
TOP VIEW
32
31
30
29
28
27
26
25
22 PGND
EN LDO
4
21 GND
VREF3
5
20 SECFB
VIN
6
19 PVCC
LDO
7
18 LGATE1
LDOREFIN
8
17 BOOT1
9
10
11
12
13
14
15
16
PHASE1
3
UGATE1
VCC
EN1
23 LGATE2
POK1
2
ILIM1
TON
FB1
24 BOOT2
OUT1
1
BYP
REF
FN6453.3
March 18, 2008
ISL6236A
Absolute Voltage Ratings
Thermal Information
VIN, EN LDO to GND. . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +27V
BOOT to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +33V
BOOT to PHASE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +6V
VCC, EN, SKIP, TON, PVCC, POK to GND . . . . . . . . . -0.3V to +6V
LDO, FB1, REFIN2, LDOREFIN to GND . . . -0.3V to (VCC + 0.3V)
OUT, SECFB, VREF3, REF to GND . . . . . . . . -0.3V to (VCC + 0.3V
UGATE to PHASE . . . . . . . . . . . . . . . . . . . . -0.3V to (PVCC + 0.3V)
ILIM to GND . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to (VCC + 0.3V)
LGATE, BYP to GND . . . . . . . . . . . . . . . . . . -0.3V to (PVCC + 0.3V)
PGND to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . -0.3V to +0.3V
LDO, REF, VREF3 Short Circuit to GND . . . . . . . . . . . . Continuous
VCC Short Circuit to GND . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1s
LDO Current (Internal Regulator) Continuous . . . . . . . . . . . . 100mA
LDO Current (Switched Over to OUT1) Continuous . . . . . . +200mA
Thermal Resistance (Typical)
θJA (°C/W)
θJC (°CW)
32 Ld QFN (Notes 1, 2) . . . . . . . . . . . . 32
3.0
Operating Temperature Range . . . . . . . . . . . . . . . .-40°C to +100°C
Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +150°C
Storage Temperature Range . . . . . . . . . . . . . . . . . .-65°C to +150°C
Pb-free reflow profile . . . . . . . . . . . . . . . . . . . . . . . . . .see link below
http://www.intersil.com/pbfree/Pb-FreeReflow.asp
CAUTION: Do not operate at or near the maximum ratings listed for extended periods of time. Exposure to such conditions may adversely impact product reliability and
result in failures not covered by warranty.
NOTES:
1. θJA is measured in free air with the component mounted on a high effective thermal conductivity test board with “direct attach” features. See
Tech Brief TB379.
2. For θJC, the “case temp” location is the center of the exposed metal pad on the package underside.
3. Limits established by characterization and are not production tested.
4. Parts are 100% tested at +25°C. Temperature limits established by characterization and are not production tested.
Electrical Specifications
No load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V, EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V,
VEN_LDO = 5V, TA = -40°C to +100°C, unless otherwise noted. Typical values are at TA = +25°C.
PARAMETER
CONDITIONS
MIN
(Note 4)
TYP
MAX
(Note 4)
UNITS
MAIN SMPS CONTROLLERS
VIN Input Voltage Range
LDO in regulation
5.5
25
V
VIN = LDO, VOUT1 <4.43V
4.5
5.5
V
3.3V Output Voltage in Fixed Mode
VIN = 4.5V to 25V, REFIN2 > (VCC - 1V), SKIP = 5V
3.285
3.330
3.375
V
1.05V Output Voltage in Fixed Mode
VIN = 4.5V to 25V, 3.0 < REFIN2 < (VCC - 1.1V),
SKIP = 5V
1.038
1.05
1.062
V
1.5V Output Voltage in Fixed Mode
VIN = 4.5V to 25V, FB1 = VCC, SKIP = 5V
1.482
1.500
1.518
V
5V Output Voltage in Fixed Mode
VIN = 5.5V to 25V, FB1 = GND, SKIP = 5V
4.975
5.050
5.125
V
FB1 in Output Adjustable Mode
VIN = 4.5V to 25V
0.693
0.700
0.707
V
REFIN2 in Output Adjustable Mode
VIN = 4.5V to 25V
0.7
2.50
V
SECFB Voltage
VIN = 4.5V to 25V
1.920
2.080
V
SMPS1 Output Voltage Adjust Range
SMPS1
0.70
5.50
V
SMPS2 Output Voltage Adjust Range
SMPS2
0.50
2.50
V
SMPS2 Output Voltage Accuracy
(Referred for REFIN2)
REFIN2 = 0.7V to 2.5V, SKIP = VCC
-1.0
1.0
%
DC Load Regulation
Either SMPS, SKIP = VCC, 0A to 5A
-0.1
%
Either SMPS, SKIP = REF, 0A to 5A
-1.7
%
Either SMPS, SKIP = GND, 0A to 5A
-1.5
%
0.005
%/V
Line Regulation
Either SMPS, 6V < VIN < 24V
Current-Limit Current Source
Temperature = +25°C
ILIM Adjustment Range
4.75
0.2
3
2.00
5
5.25
µA
2
V
FN6453.3
March 18, 2008
ISL6236A
Electrical Specifications
No load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V, EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V,
VEN_LDO = 5V, TA = -40°C to +100°C, unless otherwise noted. Typical values are at TA = +25°C. (Continued)
PARAMETER
MIN
(Note 4)
TYP
MAX
(Note 4)
UNITS
93
100
107
mV
VILIM = 0.5V
40
50
60
mV
VILIM = 1V
93
100
107
mV
VILIM = 2V
185
200
215
mV
CONDITIONS
Current-Limit Threshold (Positive, Default)
ILIM = VCC, GND - PHASE
(No temperature compensation)
Current-Limit Threshold
(Positive, Adjustable)
GND - PHASE
Zero-Current Threshold
SKIP = GND, REF, or OPEN, GND - PHASE
Current-Limit Threshold (Negative, Default)
SKIP = VCC, GND - PHASE
Soft-Start Ramp Time
Zero to full limit
Operating Frequency
(VtON = GND), SKIP = VCC
On-Time Pulse Width
3
mV
-120
mV
1.7
ms
SMPS 1
400
kHz
SMPS 2
500
kHz
(VtON = REF or OPEN),
SKIP = VCC
SMPS 1
400
kHz
SMPS 2
300
kHz
(VtON = VCC), SKIP = VCC
SMPS 1
200
kHz
SMPS 2
300
kHz
VtON = GND (400kHz/500kHz)
Minimum Off-Time
Maximum Duty Cycle
VOUT1 = 5.00V
0.895
1.052
1.209
µs
VOUT2 = 3.33V
0.475
0.555
0.635
µs
VtON = REF or OPEN
(400kHz/300kHz)
VOUT1 = 5.05V
0.895
1.052
1.209
µs
VOUT2 = 3.33V
0.833
0.925
1.017
µs
VtON = VCC (200kHz/300kHz)
VOUT1 = 5.05V
1.895
2.105
2.315
µs
VOUT2 = 3.33V
0.833
0.925
1.017
µs
TA = -40°C to +100°C
200
300
425
ns
TA = -40°C to +85°C
200
300
410
ns
VtON = GND
VOUT1 = 5.05V
88
%
VOUT2 = 3.33V
85
%
VOUT1 = 5.05V
88
%
VOUT2 = 3.33V
91
%
VOUT1 = 5.05V
94
%
VOUT2 = 3.33V
91
%
25
37
kHz
BYP = GND, 5.5V < VIN < 25V, LDOREFIN < 0.3V,
0 < ILDO < 100mA
4.925
5.000
5.075
V
BYP = GND, 4.5V < VIN < 25V, LDOREFIN > (VCC - 1V),
0 < ILDO < 100mA
3.250
3.300
3.350
V
BYP = GND, 4.5V < VIN < 25V, LDOREFIN = 2V,
0 < ILDO < 100mA
3.94
4.00
4.06
V
VIN = 4.5V to 25V, VLDOREFIN = 0.35V to 0.5V
±2.5
%
VIN = 4.5V to 25V, VLDOREFIN = 0.5V to 2V
±1.5
%
VIN = 5.5V to 25V, VLDOREFIN = 2V to 2.25V
±1.5
%
2.25
V
VtON = REF or OPEN
VtON = VCC
Ultrasonic SKIP Operating Frequency
SKIP = REF or OPEN
INTERNAL REGULATOR AND REFERENCE
LDO Output Voltage
LDO Output Accuracy in Adjustable Mode
VLDO = 2 x VLDOREFIN
LDOREFIN Input Range
4
0.35
FN6453.3
March 18, 2008
ISL6236A
Electrical Specifications
No load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V, EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V,
VEN_LDO = 5V, TA = -40°C to +100°C, unless otherwise noted. Typical values are at TA = +25°C. (Continued)
MAX
(Note 4)
UNITS
100
mA
LDO Output Current During Switchover to 5V BYP = 5V, VIN = 5.5V to 25V, LDOREFIN < 0.3V
200
mA
LDO Output Current During Switchover
to 3.3V
BYP = 3.3V, VIN = 4.5V to 25V, LDOREFIN > (VCC - 1V)
100
mA
LDO Short-Circuit Current
LDO = GND, BYP = GND
200
400
mA
Undervoltage-Lockout Fault Threshold
Rising edge of PVCC
4.35
4.5
V
PARAMETER
CONDITIONS
LDO Output Current
MIN
(Note 4)
TYP
BYP = GND, VIN = 4.5V to 25V (Note 3)
Falling edge of PVCC
3.9
4.05
V
LDO 5V Bootstrap Switch Threshold to BYP Rising edge at BYP regulation point
LDOREFIN = GND
4.53
4.68
4.83
V
LDO 3.3V Bootstrap Switch Threshold to
BYP
Rising edge at BYP regulation point
LDOREFIN = VCC
3.0
3.1
3.2
V
LDO 5V Bootstrap Switch Equivalent
Resistance
LDO to BYP, BYP = 5V, LDOREFIN > (VCC - 1V)
(Note 3)
0.7
1.5
Ω
LDO 3.3V Bootstrap Switch Equivalent
Resistance
LDO to BYP, BYP = 3.3V, LDOREFIN < 0.3V (Note 3)
1.5
3.0
Ω
VREF3 Output Voltage
No external load, VCC > 4.5V
3.235
3.300
3.365
V
No external load, VCC < 4.0V
3.220
3.300
3.380
V
VREF3 Load Regulation
0 < ILOAD < 5mA
10
VREF3 Current Limit
VREF3 = GND
10
17
mA
REF Output Voltage
No external load
2.000
2.020
V
REF Load Regulation
0 < ILOAD < 50µA
REF Sink Current
REF in regulation
VIN Operating Supply Current
Both SMPSs on, FB1 = SKIP = GND, REFIN2 = VCC
VOUT1 = BYP = 5.3V, VOUT2 = 3.5V
25
50
µA
VIN Standby Supply Current
VIN = 5.5V to 25V, both SMPSs off, EN LDO = VCC
180
250
µA
VIN Shutdown Supply Current
VIN = 4.5V to 25V, EN1 = EN2 = EN LDO = 0V
20
30
µA
Quiescent Power Consumption
Both SMPSs on, FB1 = SKIP = GND, REFIN2 = VCC,
VOUT1 = BYP = 5.3V, VOUT2 = 3.5V
5
7
mW
1.980
mV
10
mV
10
µA
FAULT DETECTION
Overvoltage Trip Threshold
FB1 with respect to nominal regulation point
+8
+11
+14
%
REFIN2 with respect to nominal regulation point
+12
+16
+20
%
Overvoltage Fault Propagation Delay
FB1 or REFIN2 delay with 50mV overdrive
POK_ Threshold
FB1 or REFIN2 with respect to nominal output, falling
edge, typical hysteresis = 1%
POK_ Propagation Delay
Falling edge, 50mV overdrive
POK_ Output Low Voltage
ISINK = 4mA
POK_ Leakage Current
High state, forced to 5.5V
10
-12
-9
µs
-6
10
Thermal-Shutdown Threshold
%
µs
0.2
V
1
µA
+150
°C
5
%
Out-Of-Bound Threshold
FB1 or REFIN2 with respect to nominal output voltage
Output Undervoltage Shutdown Threshold
FB1 or REFIN2 with respect to nominal output voltage
65
70
75
%
Output Undervoltage Shutdown Blanking
Time
From EN signal
10
20
30
ms
5
FN6453.3
March 18, 2008
ISL6236A
Electrical Specifications
No load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V, EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V,
VEN_LDO = 5V, TA = -40°C to +100°C, unless otherwise noted. Typical values are at TA = +25°C. (Continued)
PARAMETER
CONDITIONS
MIN
(Note 4)
TYP
MAX
(Note 4)
UNITS
0.3
V
INPUTS AND OUTPUTS
FB1 Input Voltage
Low level
High level
REFIN2 Input Voltage
LDOREFIN Input Voltage
VCC - 1.0
V
OUT2 Dynamic Range, VOUT2 = VREFIN2
0.5
2.50
V
Fixed OUT2 = 1.05V
3.0
VCC - 1.1
V
Fixed OUT2 = 3.3V
VCC - 1.0
V
Fixed LDO = 5V
LDO Dynamic Range, VLDO = 2 x VLDOREFIN
Fixed LDO = 3.3V
SKIP Input Voltage
0.35
Float level (ULTRASONIC SKIP)
1.7
High level (PWM)
2.4
Input Leakage Current
V
Float level
1.7
High level
2.4
0.8
V
2.3
V
V
0.8
V
2.3
V
V
Clear fault level/SMPS off level
EN LDO Input Voltage
2.25
V
Low level
EN1, EN2 Input Voltage
V
VCC - 1.0
Low level (SKIP)
TON Input Voltage
0.30
0.8
V
2.3
V
Delay start level
1.7
SMPS on level
2.4
Rising edge
1.2
1.6
2.0
V
Falling edge
0.94
1.00
1.06
V
-1
+1
µA
-0.1
+0.1
µA
-1
+1
µA
VFB1 = VSECFB = 0V or 5V
-0.2
+0.2
µA
VREFIN = 0V or 2.5V
-0.2
+0.2
µA
VLDOREFIN = 0V or 2.75V
-0.2
+0.2
µA
0.8
V
500
nA
VtON = 0V or 5V
VEN = VEN LDO = 0V or 5V
VSKIP = 0V or 5V
V
INTERNAL BOOT DIODE
VD Forward Voltage
PVCC - VBOOT, IF = 10mA
IBOOT LEAKAGE Leakage Current
VBOOT = 30V, PHASE = 25V, PVCC = 5V
0.65
MOSFET DRIVERS
UGATE Gate-Driver Sink/Source Current
UGATE1, UGATE2 forced to 2V
2
A
LGATE Gate-Driver Source Current
LGATE1 (source), LGATE2 (source), forced to 2V
1.7
A
LGATE Gate-Driver Sink Current
LGATE1 (sink), LGATE2 (sink), forced to 2V
3.3
A
UGATE Gate-Driver ON-resistance
BOOT_ - PHASE_ forced to 5V (Note 3)
1.5
4.0
Ω
LGATE Gate-Driver ON-resistance
LGATE, high state (pull-up) (Note 3)
2.2
5.0
Ω
LGATE, low state (pull-down) (Note 3)
0.6
1.5
Ω
Dead Time
OUT1, OUT2 Discharge ON-resistance
6
LGATE Rising
15
20
35
ns
UGATE Rising
20
30
50
ns
25
40
Ω
FN6453.3
March 18, 2008
ISL6236A
Pin Descriptions
PIN
NAME
1
REF
2V Reference Output. Bypass to GND with a 0.1µF (min) capacitor. REF can source up to 50µA for external loads.
Loading REF degrades FB and output accuracy according to the REF load-regulation error.
2
TON
Frequency Select Input. Connect to GND for 400kHz/500kHz operation. Connect to REF (or leave OPEN) for
400kHz/300kHz operation. Connect to VCC for 200kHz/300kHz operation (5V/3.3V SMPS switching frequencies,
respectively).
3
VCC
Analog Supply Voltage Input for PWM Core. Bypass to GND with a 1µF ceramic capacitor.
4
EN LDO
LDO Enable Input. The LDO is enabled if EN LDO is within logic high level and disabled if EN LDO is less than the logic
low level.
5
VREF3
3.3V Reference Output. VREF3 can source up to 5mA for external loads. Bypass to GND with a 0.01µF capacitor if
loaded. Leave open if there is no load.
6
VIN
Power-Supply Input. VIN is used for the constant-on-time PWM on-time one-shot circuits. VIN is also used to power the
linear regulators. The linear regulators are powered by SMPS1 if OUT1 is set greater than 4.78V and BYP is tied to
OUT1. Connect VIN to the battery input and bypass with a 1µF capacitor.
7
LDO
Linear-Regulator Output. LDO can provide a total of 100mA external loads. The LDO regulates at 5V If LDOREFIN is
connected to GND. When the LDO is set at 5V and BYP is within 5V switchover threshold, the internal regulator shuts
down and the LDO output pin connects to BYP through a 0.7Ω switch. The LDO regulates at 3.3V if LDOREFIN is
connected to VCC. When the LDO is set at 3.3V and BYP is within the 3.3V switchover threshold, the internal regulator
shuts down and the LDO output pin connects to BYP through a 1.5Ω switch. Bypass LDO output with a minimum of
4.7µF ceramic.
8
FUNCTION
LDOREFIN LDO Reference Input. Connect LDOREFIN to GND for fixed 5V operation. Connect LDOREFIN to VCC for fixed 3.3V
operation. LDOREFIN can be used to program LDO output voltage from 0.7V to 4.5V. LDO output is two times the
voltage of LDOREFIN. There is no switchover in adjustable mode.
9
BYP
BYP is the switchover source voltage for the LDO when LDOREFIN is connected to GND or VCC. Connect BYP to 5V
if LDOREFIN is tied to GND. Connect BYP to 3.3V if LDOREFIN is tied to VCC.
10
OUT1
SMPS1 Output Voltage-Sense Input. Connect to the SMPS1 output. OUT1 is an input to the Constant on-time-PWM
on-time one-shot circuit. It also serves as the SMPS1 feedback input in fixed-voltage mode.
11
FB1
SMPS1 Feedback Input. Connect FB1 to GND for fixed 5V operation. Connect FB1 to VCC for fixed 1.5V operation.
Connect FB1 to a resistive voltage-divider from OUT1 to GND to adjust the output from 0.7V to 5.5V.
12
ILIM1
SMPS1 Current-Limit Adjustment. The GND-PHASE1 current-limit threshold is 1/10th the voltage seen at ILIM1 over a
0.2V to 2V range. There is an internal 5µA current source from VCC to ILIM1. Connect ILIM1 to REF for a fixed 200mV
threshold. The logic current limit threshold is default to 100mV value if ILIM1 is higher than VCC - 1V.
13
POK1
SMPS1 Power-Good Open-Drain Output. POK1 is low when the SMPS1 output voltage is more than 10% below the
normal regulation point or during soft-start. POK1 is high impedance when the output is in regulation and the soft-start
circuit has terminated. POK1 is low in shutdown.
14
EN1
SMPS1 Enable Input. The SMPS1 is enabled if EN1 is greater than the logic high level and disabled if EN1 is less than
the logic low level. If EN1 is connected to REF, the SMPS1 starts after the SMPS2 reaches regulation (delay start). Drive
EN1 below 0.8V to clear fault level and reset the fault latches.
15
UGATE1
High-Side MOSFET Floating Gate-Driver Output for SMPS1. UGATE1 swings between PHASE1 and BOOT1.
16
PHASE1
Inductor Connection for SMPS1. PHASE1 is the internal lower supply rail for the UGATE1 high-side gate driver.
PHASE1 is the current-sense input for the SMPS1.
17
BOOT1
Boost Flying Capacitor Connection for SMPS1. Connect to an external capacitor according to the “Typical Application
Circuits” on page 19 (Figures 66, 67 and 68). See “MOSFET Gate Drivers (UGATE, LGATE)” on page 29.
18
LGATE1
SMPS1 Synchronous-Rectifier Gate-Drive Output. LGATE1 swings between GND and PVCC.
19
PVCC
PVCC is the supply voltage for the low-side MOSFET driver LGATE. Connect a 5V power source to the PVCC pin and
bypass to PGND with a 1µF MLCC ceramic capacitor. Refer to Figure 70 - A switch connects PVCC to VCC with 10Ω
when in normal operation and is disconnected when in shutdown mode. An external 10Ω resistor from PVCC to VCC
is prohibited as it will create a leakage path from VIN to GND in shutdown mode.
20
SECFB
The SECFB is used to monitor the optional external 14V charge pump. Connect a resistive voltage-divider from 14V
charge pump output to GND to detect the output. If SECFB drops below the threshold voltage, LGATE1 turns on for
300ns. This will refresh the external charge pump driven by LGATE1 without over-discharging the output voltage.
7
FN6453.3
March 18, 2008
ISL6236A
Pin Descriptions (Continued)
PIN
NAME
FUNCTION
21
GND
22
PGND
23
LGATE2
SMPS2 Synchronous-Rectifier Gate-Drive Output. LGATE2 swings between GND and PVCC.
24
BOOT2
Boost Flying Capacitor Connection for SMPS2. Connect to an external capacitor according to the “Typical Application
Circuits” on page 19 (Figures 66, 67 and 68). See “MOSFET Gate Drivers (UGATE, LGATE)” on page 29.
25
PHASE2
Inductor Connection for SMPS2. PHASE2 is the internal lower supply rail for the UGATE2 high-side gate driver.
PHASE2 is the current-sense input for the SMPS2.
26
UGATE2
High-Side MOSFET Floating Gate-Driver Output for SMPS2. UGATE1 swings between PHASE2 and BOOT2.
27
EN2
SMPS2 Enable Input. The SMPS2 is enabled if EN2 is greater than the logic high level and disabled if EN2 is less than
the logic low level. If EN2 is connected to REF, the SMPS2 starts after the SMPS1 reaches regulation (delay start). Drive
EN2 below 0.8V to clear fault level and reset the fault latches.
28
POK2
SMP2 Power-Good Open-Drain Output. POK2 is low when the SMPS2 output voltage is more than 10% below the
normal regulation point or during soft-start. POK2 is high impedance when the output is in regulation and the soft-start
circuit has terminated. POK2 is low in shutdown.
29
SKIP
Low-Noise Mode Control. Connect SKIP to GND for normal Idle-Mode (pulse-skipping) operation or to VCC for PWM
mode (fixed frequency). Connect to REF or leave floating for ultrasonic skip mode operation.
30
OUT2
SMPS2 Output Voltage-Sense Input. Connect to the SMPS2 output. OUT2 is an input to the Constant on-time-PWM
on-time one-shot circuit. It also serves as the SMPS2 feedback input in fixed-voltage mode.
31
ILIM2
SMPS2 Current-Limit Adjustment. The GND-PHASE1 current-limit threshold is 1/10th the voltage seen at ILIM2 over a
0.2V to 2V range. There is an internal 5µA current source from VCC to ILIM2. Connect ILIM2 to REF for a fixed 200mV.
The logic current limit threshold is default to 100mV value if ILIM2 is higher than VCC - 1V.
32
REFIN2
Output voltage control for SMPS2. Connect REFIN2 to VCC for fixed 3.3V. Connect REFIN2 to VREF3 for fixed 1.05V.
REFIN2 can be used to program SMPS2 output voltage from 0.5V to 2.50V. SMPS2 output voltage is 0V if
REFIN2 < 0.5V.
Analog Ground for both SMPS and LDO. Connect externally to the underside of the exposed pad.
Power Ground for SMPS controller. Connect PGND externally to the underside of the exposed pad.
Typical Performance Curves
12VIN ULTRA SKIP MODE
25VIN SKIP MODE
25VIN PWM MODE
25VIN ULTRA SKIP MODE
100
100
90
90
80
80
70
70
EFFICIENCY (%)
EFFICIENCY (%)
7VIN SKIP MODE
7VIN PWM MODE
7VIN ULTRA SKIP MODE
12VIN SKIP MODE
12VIN PWM MODE
Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C.
60
50
40
30
60
50
40
30
20
20
10
10
0
0.001
0.010
0.100
OUTPUT LOAD (A)
1.000
10.000
FIGURE 1. VOUT2 = 1.05V EFFICIENCY vs LOAD (300kHz)
8
12VIN ULTRA SKIP MODE
25VIN SKIP MODE
25VIN PWM MODE
25VIN ULTRA SKIP MODE
7VIN SKIP MODE
7VIN PWM MODE
7VIN ULTRA SKIP MODE
12VIN SKIP MODE
12VIN PWM MODE
0
0.001
0.010
0.100
1.000
10.000
OUTPUT LOAD (A)
FIGURE 2. VOUT1 = 1.5V EFFICIENCY vs LOAD (200kHz)
FN6453.3
March 18, 2008
ISL6236A
Typical Performance Curves
7VIN SKIP MODE
7VIN PWM MODE
7VIN ULTRA SKIP MODE
12VIN SKIP MODE
12VIN PWM MODE
12VIN ULTRA SKIP MODE
25VIN SKIP MODE
25VIN PWM MODE
25VIN ULTRA SKIP MODE
90
80
80
70
60
50
40
30
50
40
30
20
10
0.100
OUTPUT LOAD (A)
1.000
0
0.001
10.000
1.070
7VIN SKIP MODE
7VIN PWM MODE
7VIN ULTRA SKIP MODE
12VIN SKIP MODE
12VIN PWM MODE
0.010
12VIN ULTRA SKIP MODE
25VIN SKIP MODE
25VIN PWM MODE
25VIN ULTRA SKIP MODE
1.540
10.000
7VIN SKIP MODE
7VIN PWM MODE
7VIN ULTRA SKIP MODE
12VIN SKIP MODE
12VIN PWM MODE
12VIN ULTRA SKIP MODE
25VIN SKIP MODE
25VIN PWM MODE
25VIN ULTRA SKIP MODE
1.535
OUTPUT VOLTAGE (V)
1.066
1.064
1.062
1.060
1.058
1.056
1.054
1.530
1.525
1.520
1.515
1.510
1.505
1.052
1.050
0.001
0.010
0.100
1.000
1.500
0.001
10.000
0.010
FIGURE 5. VOUT2 = 1.05V REGULATION vs LOAD (300kHz)
7VIN SKIP MODE
7VIN PWM MODE
7VIN ULTRA SKIP MODE
12VIN SKIP MODE
12VIN PWM MODE
1.000
10.000
FIGURE 6. VOUT1 = 1.5V REGULATION vs LOAD (200kHz)
12VIN ULTRA SKIP MODE
25VIN SKIP MODE
25VIN PWM MODE
25VIN ULTRA SKIP MODE
5.16
7VIN SKIP MODE
7VIN PWM MODE
7VIN ULTRA SKIP MODE
12VIN SKIP MODE
12VIN PWM MODE
12VIN ULTRA SKIP MODE
25VIN SKIP MODE
25VIN PWM MODE
25VIN ULTRA SKIP MODE
5.14
OUTPUT VOLTAGE (V)
3.37
3.36
3.35
3.34
3.33
3.32
3.31
0.001
0.100
OUTPUT LOAD (A)
OUTPUT LOAD (A)
OUTPUT VOLTAGE (V)
1.000
FIGURE 4. VOUT1 = 5V EFFICIENCY vs LOAD (400kHz)
1.068
3.38
0.100
OUTPUT LOAD (A)
FIGURE 3. VOUT2 = 3.3V EFFICIENCY vs LOAD (500kHz)
OUTPUT VOLTAGE (V)
60
10
0.010
12VIN ULTRA SKIP MODE
25VIN SKIP MODE
25VIN PWM MODE
25VIN ULTRA SKIP MODE
70
20
0
0.001
7VIN SKIP MODE
7VIN PWM MODE
7VIN ULTRA SKIP MODE
12VIN SKIP MODE
12VIN PWM MODE
100
90
EFFICIENCY (%)
EFFICIENCY (%)
100
Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C. (Continued)
5.12
5.10
5.08
5.06
5.04
5.02
0.010
0.100
OUTPUT LOAD (A)
1.000
10.000
FIGURE 7. VOUT2 = 3.3V REGULATION vs LOAD (500kHz)
9
5.00
0.001
0.010
0.100
OUTPUT LOAD (A)
1.000
10.000
FIGURE 8. VOUT1 = 5V REGULATION vs LOAD (400kHz)
FN6453.3
March 18, 2008
ISL6236A
Typical Performance Curves
POWER DISSIPATION (W)
POWER DISSIPATION (W)
2.5
2.0
1.5
1.0
0.5
0.0
0.001
0.010
0.100
1.000
2.0
1.5
1.0
0.5
0.0
0.001
10.000
0.010
OUTPUT LOAD (A)
FIGURE 9. VOUT2 = 1.05V POWER DISSIPATION vs LOAD
(300kHz)
3.0
2.5
2.0
1.5
1.0
0.5
0.0
0.001
0.010
0.100
OUTPUT LOAD (A)
1.000
10.000
3.0
2.5
2.0
1.5
1.0
0.5
0.0
0.001
10.000
FIGURE 11. VOUT2 = 3.3V POWER DISSIPATION vs LOAD
(500kHz)
0.010
0.100
OUTPUT LOAD (A)
1.000
10.000
FIGURE 12. VOUT1 = 5V POWER DISSIPATION vs LOAD
(400kHz)
1.064
1.068
NO LOAD PWM
1.066
OUTPUT VOLTAGE (V)
1.062
OUTPUT VOLTAGE (V)
1.000
12VIN ULTRA SKIP MODE
25VIN SKIP MODE
25VIN PWM MODE
25VIN ULTRA SKIP MODE
7VIN SKIP MODE
7VIN PWM MODE
7VIN ULTRA SKIP MODE
12VIN SKIP MODE
12VIN PWM MODE
3.5
POWER DISSIPATION (W)
POWER DISSIPATION (W)
3.5
0.100
OUTPUT LOAD (A)
FIGURE 10. VOUT1 = 1.5V POWER DISSIPATION vs LOAD
(200kHz)
12VIN ULTRA SKIP MODE
25VIN SKIP MODE
25VIN PWM MODE
25VIN ULTRA SKIP MODE
7VIN SKIP MODE
7VIN PWM MODE
7VIN ULTRA SKIP MODE
12VIN SKIP MODE
12VIN PWM MODE
12VIN ULTRA SKIP MODE
25VIN SKIP MODE
25VIN PWM MODE
25VIN ULTRA SKIP MODE
7VIN SKIP MODE
7VIN PWM MODE
7VIN ULTRA SKIP MODE
12VIN SKIP MODE
12VIN PWM MODE
12VIN ULTRA SKIP MODE
25VIN SKIP MODE
25VIN PWM MODE
25VIN ULTRA SKIP MODE
7VIN SKIP MODE
7VIN PWM MODE
7VIN ULTRA SKIP MODE
12VIN SKIP MODE
12VIN PWM MODE
2.5
Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C. (Continued)
1.060
1.058
1.056
MID LOAD PWM
1.054
1.052
MAX LOAD PWM
1.050
1.064
1.062
NO LOAD PWM
1.060
1.058
1.056
1.054
MID LOAD PWM
1.052
1.050
1.048
5
7
9
11
13
15
17
19
21
23
25
INPUT VOLTAGE (V)
FIGURE 13. VOUT2 = 1.05V OUTPUT VOLTAGE REGULATION
vs VIN (PWM MODE)
10
1.048
MAX
MAX LOAD
LOAD PWM
PWM
5
7
9
11
13
15
17
19
21
23
25
INPUT VOLTAGE (V)
FIGURE 14. VOUT2 = 1.05V OUTPUT VOLTAGE REGULATION
vs VIN (SKIP MODE)
FN6453.3
March 18, 2008
ISL6236A
Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C. (Continued)
1.518
1.530
1.516
1.525
NO LOAD PWM
1.514
1.512
MID LOAD PWM
1.510
MAX LOAD PWM
1.508
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
Typical Performance Curves
1.506
1.504
5
7
9
11
13
15
17
19
INPUT VOLTAGE (V)
21
23
1.510
MAX LOAD PWM
1.505
5
7
9
11 13
15
17
19
INPUT VOLTAGE (V)
21
23
25
FIGURE 16. VOUT1 = 1.5V OUTPUT VOLTAGE REGULATION
vs VIN (SKIP MODE)
3.340
3.380
3.370
3.335
NO LOAD PWM
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
MID LOAD PWM
1.515
1.500
25
FIGURE 15. VOUT1 = 1.5V OUTPUT VOLTAGE REGULATION
vs VIN (PWM MODE)
NO LOAD PWM
1.520
3.330
3.325
3.320
MID LOAD PWM
3.315
3.360
NO LOAD PWM
3.350
MAX LOAD PWM
3.340
3.330
3.320
MID LOAD PWM
3.310
MAX LOAD PWM
3.310
7
9
11
13
15
17
19
21
23
3.300
25
7
9
11
INPUT VOLTAGE (V)
13
15
17
19
21
23
25
INPUT VOLTAGE (V)
FIGURE 17. VOUT2 = 3.3V OUTPUT VOLTAGE REGULATION
vs VIN (PWM MODE)
FIGURE 18. VOUT2 = 3.3V OUTPUT VOLTAGE REGULATION
vs VIN (SKIP MODE)
5.065
5.14
5.060
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
NO LOAD PWM
MAX LOAD PWM
5.055
MID LOAD PWM
5.050
5.045
5.12
5.10
NO LOAD PWM
5.08
MID LOAD PWM
5.06
5.04
5.040
7
9
11
13
15
17
19
INPUT VOLTAGE (V)
21
23
25
FIGURE 19. VOUT1 = 5V OUTPUT VOLTAGE REGULATION vs
VIN (PWM MODE)
11
MAX LOAD PWM
5.02
7
9
11
13
15
17
19
INPUT VOLTAGE (V)
21
23
25
FIGURE 20. VOUT1 = 5V OUTPUT VOLTAGE REGULATION vs
VIN (SKIP MODE)
FN6453.3
March 18, 2008
ISL6236A
Typical Performance Curves
Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C. (Continued)
50
300
45
40
200
RIPPLE (mV)
FREQUENCY (kHz)
250
PWM
150
100
35
30
10
0.010
0.100
OUTPUT LOAD (A)
SKIP
5
SKIP
0
0.001
1.000
0
0.001
10.000
FIGURE 21. VOUT2 = 1.05V FREQUENCY vs LOAD
0.010
0.100
OUTPUT LOAD (A)
10.000
50
45
PWM
200
40
PWM
RIPPLE (mV)
35
150
100
ULTRA-SKIP
50
30
25
SKIP
20 ULTRA-SKIP
15
10
SKIP
0
0.001
5
0.010
0.100
OUTPUT LOAD (A)
1.000
0
0.001
10.000
FIGURE 23. VOUT1 = 1.5V FREQUENCY vs LOAD
0.010
0.100
OUTPUT LOAD (A)
1.000
10.000
FIGURE 24. VOUT1 = 1.5V RIPPLE vs LOAD
14
600
PWM
PWM
12
500
10
RIPPLE (mV)
400
300
200
100
1.000
FIGURE 22. VOUT2 = 1.05V RIPPLE vs LOAD
250
FREQUENCY (kHz)
ULTRA-SKIP
20
15
ULTRA-SKIP
50
FREQUENCY (kHz)
PWM
25
ULTRA-SKIP
SKIP
6
4
ULTRA-SKIP
2
SKIP
0
0.001
8
0.010
0.100
OUTPUT LOAD (A)
1.000
FIGURE 25. VOUT2 = 3.3V FREQUENCY vs LOAD
12
10.000
0
0.001
0.010
0.100
OUTPUT LOAD (A)
1.000
10.000
FIGURE 26. VOUT2 = 3.3V RIPPLE vs LOAD
FN6453.3
March 18, 2008
ISL6236A
Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C. (Continued)
450
40
400
35
PWM
350
30
PWM
300
RIPPLE (mV)
FREQUENCY (kHz)
Typical Performance Curves
250
200
150
20
SKIP
15
10
ULTRA-SKIP
100
ULTRA-SKIP
25
5
50
SKIP
0
0.001
0.010
0.100
OUTPUT LOAD (A)
1.000
0
0.001
10.000
FIGURE 27. VOUT1 = 5V FREQUENCY vs LOAD
10.000
3.30
OUTPUT VOLTAGE (V)
BYP = 0V
5.00
4.98
4.96
4.94
4.92
4.90
BYP = 5V
4.88
3.25
BYP = 0V
3.20
3.15
BYP = 3.3V
3.10
3.05
4.86
4.84
0
50
100
OUTPUT LOAD (mA)
150
3.00
200
0
FIGURE 29. LDO OUTPUT 5V vs LOAD
50
100
OUTPUT LOAD (mA)
150
200
FIGURE 30. LDO OUTPUT 3.3V vs LOAD
3.5
15.5
3.0
15.0
OUTPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
1.000
3.35
5.02
OUTPUT VOLTAGE (V)
0.100
OUTPUT LOAD (A)
FIGURE 28. VOUT1 = 5V RIPPLE vs LOAD
5.04
2.5
2.0
1.5
1.0
0.5
0
0.010
0
2
6
4
OUTPUT LOAD (mA)
FIGURE 31. VREF3 vs LOAD
13
8
10
14.5
14.0
13.5
13.0
12.5
0
2
4
5
10
8
OUTPUT LOAD (mA)
FIGURE 32. CHARGE PUMP vs LOAD (PWM)
FN6453.3
March 18, 2008
ISL6236A
Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C. (Continued)
50
1400
45
1200
INPUT CURRENT (µA)
INPUT CURRENT (mA)
Typical Performance Curves
40
35
30
25
20
800
600
400
200
7
9
11
13
15
17
19
INPUT VOLTAGE (V)
21
23
0.0
25
FIGURE 33. PWM NO LOAD INPUT CURRENT vs VIN
(EN = EN2 = EN LDO = VCC)
7
9
11
13
15
17
19
INPUT VOLTAGE (V)
21
23
25
FIGURE 34. SKIP NO LOAD INPUT CURRENT vs VIN
(EN1 = EN2 = EN LDO = VCC)
177.5
26.5
177.0
26.0
176.5
25.5
INPUT CURRENT (µA)
INPUT CURRENT (µA)
1000
176.0
175.5
175.0
174.5
174.0
173.5
25.0
24.5
24.0
23.5
23.0
22.5
173.0
7
9
11
13
15
17
19
INPUT VOLTAGE (V)
21
23
25
FIGURE 35. STANDBY INPUT CURRENT vs VIN
(EN = EN2 = 0, EN LDO = VCC)
22.0
7
9
11
13
15
17
19
INPUT VOLTAGE (V)
21
23
25
FIGURE 36. SHUTDOWN INPUT CURRENT vs VIN
(EN = EN2 = EN LDO = 0)
VREF3 500mV/DIV
EN1 5V/DIV
VOUT1 2V/DIV
LDO 1V/DIV
CP 5V/DIV
REF 1V/DIV
IL1 2A/DIV
POK1 2V/DIV
FIGURE 37. REF, VREF3, LDO = 5V, CP, NO LOAD
14
FIGURE 38. START-UP VOUT1 = 5V (NO LOAD, SKIP MODE)
FN6453.3
March 18, 2008
ISL6236A
Typical Performance Curves
Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C. (Continued)
EN1 5V/DIV
EN1 5V/DIV
VOUT1 2V/DIV
VOUT1 2V/DIV
IL1 5A/DIV
IL1 2A/DIV
POK1 2V/DIV
POK1 2V/DIV
FIGURE 39. START-UP VOUT1 = 5V (NO LOAD, PWM MODE)
FIGURE 40. START-UP VOUT1 = 5V (FULL LOAD, PWM MODE)
EN2 5V/DIV
EN2 5V/DIV
VOUT2 2V/DIV
VOUT2 2V/DIV
IL2 2A/DIV
IL2 2A/DIV
POK2 2V/DIV
FIGURE 41. START-UP VOUT2 = 3.3V (NO LOAD, SKIP MODE)
POK2 2V/DIV
FIGURE 42. START-UP VOUT1 = 3.3V (NO LOAD, PWM MODE)
EN2 5V/DIV
EN2 5V/DIV
VOUT2 2V/DIV
VOUT2 2V/DIV
VOUT1 2V/DIV
IL2 5A/DIV
POK2 5V/DIV
POK2 2V/DIV
POK1 5V/DIV
FIGURE 43. START-UP VOUT1 = 3.3V (FULL LOAD,
PWM MODE)
15
FIGURE 44. DELAYED START-UP (VOUT1 = 5V, VOUT2 = 3.3V,
EN1 = REF)
FN6453.3
March 18, 2008
ISL6236A
Typical Performance Curves
Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C. (Continued)
EN1 5V/DIV
EN1 5V/DIV
VOUT2 2V/DIV
VOUT1 2V/DIV
VOUT2 2V/DIV
VOUT1 2V/DIV
POK1 5V/DIV
POK1 OR POK2 5V/DIV
POK2 5V/DIV
FIGURE 45. DELAYED START-UP (VOUT1 = 5V, VOUT2 = 3.3V,
EN2 = REF)
LGATE1 5V/DIV
FIGURE 46. SHUTDOWN (VOUT1 = 5V, VOUT2 = 3.3V,
EN2 = REF)
LGATE1 5V/DIV
VOUT1 RIPPLE 50mV/DIV
VOUT1 RIPPLE 100mV/DIV
IL1 5A/DIV
IL1 5A/DIV
VOUT2 RIPPLE 50mV/DIV
VOUT2 RIPPLE 50mV/DIV
FIGURE 47. LOAD TRANSIENT VOUT1 = 5V
LGATE1 5V/DIV
FIGURE 48. LOAD TRANSIENT VOUT1 = 5V (SKIP)
LGATE2 5V/DIV
VOUT1 RIPPLE 20mV/DIV
IL2 5A/DIV
VOUT1 RIPPLE 20mV/DIV
VOUT2 RIPPLE 50mV/DIV
IL2 5A/DIV
VOUT2 RIPPLE 50mV/DIV
FIGURE 49. LOAD TRANSIENT VOUT1 = 3.3V (PWM)
16
FIGURE 50. LOAD TRANSIENT VOUT1 = 3.3V (SKIP)
FN6453.3
March 18, 2008
ISL6236A
Typical Performance Curves
Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C. (Continued)
VOUT RIPPLE 20mV/DIV
VOUT RIPPLE 20mV/DIV
LDO 1V/DIV
VOUT2 0.5V/DIV
REFIN2 0.5V/DIV
LDOREFIN 0.5V/DIV
LDO RIPPLE 50mV/DIV
VOUT2 RIPPLE 50mV/DIV
FIGURE 51. VOUT2 TRACKING TO REFIN2
FIGURE 52. LDO TRACKING TO LDOREFIN
EN1 5V/DIV
EN1 5V/DIV
VOUT1 0.5V/DIV
VOUT1 0.5V/DIV
IL1 2A/DIV
IL1 2A/DIV
POK1 2V/DIV
POK1 2V/DIV
FIGURE 53. START-UP VOUT1 = 1.5V (NO LOAD, SKIP MODE)
EN1 5V/DIV
FIGURE 54. START-UP VOUT1 = 1.5V (NO LOAD, PWM MODE)
EN2 5V/DIV
VOUT1 0.5V/DIV
VOUT2 0.5V/DIV
IL1 5A/DIV
IL2 2A/DIV
POK2 2V/DIV
POK1 2V/DIV
FIGURE 55. START-UP VOUT1 = 1.5V (FULL LOAD,
PWM MODE)
17
FIGURE 56. START-UP VOUT2 = 1.05V (NO LOAD,
SKIP MODE)
FN6453.3
March 18, 2008
ISL6236A
Typical Performance Curves
Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C. (Continued)
EN2 5V/DIV
EN2 5V/DIV
VOUT2 0.5V/DIV
VOUT2 0.5V/DIV
IL2 2A/DIV
IL2 2A/DIV
POK2 2V/DIV
POK2 2V/DIV
FIGURE 57. START-UP VOUT1 = 1.05V (NO LOAD,
PWM MODE)
FIGURE 58. START-UP VOUT1 = 1.05V (FULL LOAD,
PWM MODE)
VOUT1 2V/DIV
EN2 5V/DIV
EN1 500mV/DIV
VOUT2 0.5V/DIV
VOUT2 500mV/DIV
VOUT1 2V/DIV
POK2 5V/DIV
POK1 5V/DIV
POK1 5V/DIV
POK2 5V/DIV
FIGURE 59. DELAYED START-UP (VOUT1 = 1.5V,
VOUT2 = 1.05V, EN1 = REF)
FIGURE 60. DELAYED START-UP (VOUT1 = 1.5V,
VOUT2 = 1.05V, EN2 = REF)
LGATE1 5V/DIV
EN1 5V/DIV
VOUT1 RIPPLE 50mV/DIV
VOUT2 2V/DIV
VOUT1 2V/DIV
IL1 5A/DIV
POK1 OR POK2 5V/DIV
FIGURE 61. SHUTDOWN (VOUT1 = 1.5V, VOUT2 = 1.05V,
EN2 = REF)
18
VOUT2 RIPPLE 20mV/DIV
FIGURE 62. LOAD TRANSIENT VOUT1 = 1.5V (PWM)
FN6453.3
March 18, 2008
ISL6236A
Typical Performance Curves
Circuit of Figures 66, 67 and 68, no load on LDO, OUT1, OUT2, VREF3, and REF, VIN = 12V,
EN2 = EN1 = VCC, VBYP = 5V, PVCC = 5V, VEN_LDO = 5V, TA = -40°C to +100°C, unless
otherwise noted. Typical values are at TA = +25°C. (Continued)
LGATE1 5V/DIV
LGATE2 5V/DIV
VOUT1 RIPPLE 20mV/DIV
VOUT1 RIPPLE 50mV/DIV
IL1 5A/DIV
IL1 5A/DIV
VOUT2 RIPPLE 20mV/DIV
VOUT2 RIPPLE 20mV/DIV
FIGURE 63. LOAD TRANSIENT VOUT1 = 1.5V (SKIP)
FIGURE 64. LOAD TRANSIENT VOUT1 = 1.05V (PWM)
LGATE2 5V/DIV
VOUT1 RIPPLE 20mV/DIV
IL2 5A/DIV
VOUT2 RIPPLE 20mV/DIV
FIGURE 65. LOAD TRANSIENT VOUT1 = 1.05V (SKIP)
Typical Application Circuits
The typical application circuits (Figures 66, 67 and 68) generate
the 5V/7A, 3.3V/11A, 1.25V/5A, dynamic voltage/10A, 1.5V/5A,
1.05V/5A and external 14V charge pump main supplies in a
notebook computer. The ISL6236A is also equipped with a
secondary feedback, SECFB, used to monitor the output of the
14V charge pump. In an event when the 14V drops below its
threshold voltage, SECFB comparator will turn on LGATE1 for
300ns. This will refresh an external 14V charge pump without
overcharging the output voltage. The input supply range is
5.5V to 25V.
Detailed Description
The ISL6236A dual-buck, BiCMOS, switch-mode power
supply controller generates logic supply voltages for
notebook computers. The ISL6236A is designed primarily for
battery-powered applications where high efficiency and low
quiescent supply current are critical. The ISL6236A provides
19
a pin-selectable switching frequency, allowing operation for
200kHz/300kHz, 400kHz/300kHz, or 400kHz/500kHz on the
SMPSs.
Light-load efficiency is enhanced by automatic Idle-Mode
operation, a variable-frequency pulse-skipping mode that
reduces transition and gate-charge losses. Each step-down,
power-switching circuit consists of 2 N-Channel MOSFETs, a
rectifier, and an LC output filter. The output voltage is the
average AC voltage at the switching node, which is
regulated by changing the duty cycle of the MOSFET
switches. The gate-drive signal to the N-Channel high-side
MOSFET must exceed the battery voltage, and is provided
by a flying-capacitor boost circuit that uses a 100nF
capacitor connected to BOOT.
Both SMPS1 and SMPS2 PWM controllers consist of a triple
mode feedback network and multiplexer, a multi-input PWM
comparator, high-side and low-side gate drivers and logic. In
FN6453.3
March 18, 2008
ISL6236A
addition, SMPS2 can also use REFIN2 to track its output from
0.5V to 2.50V. The ISL6236A contains fault-protection circuits
that monitor the main PWM outputs for undervoltage and
overvoltage conditions. A power-on sequence block controls
the power-up timing of the main PWMs and monitors the
outputs for undervoltage faults. The ISL6236A includes an
adjustable low drop-out linear regulator. The bias generator
blocks include the linear regulator, 3.3V precision reference, 2V
precision reference and automatic bootstrap switchover circuit.
The synchronous-switch gate drivers are directly powered
from PVCC, while the high-side switch gate drivers are
indirectly powered from PVCC through an external capacitor
and an internal Schottky diode boost circuit.
An automatic bootstrap circuit turns off the LDO linear
regulator and powers the device from BYP if LDOREFIN is
set to GND or VCC. See Table 1.
TABLE 1. LDO OUTPUT VOLTAGE TABLE
LDO VOLTAGE
CONDITIONS
COMMENT
as measured by the VIN input and proportional to the output
voltage. This algorithm results in a nearly constant switching
frequency despite the lack of a fixed-frequency clock
generator. The benefit of a constant switching frequency is
that the frequency can be selected to avoid noise-sensitive
frequency regions, as shown in Equation 1:
K ( V OUT + I LOAD ⋅ r DS ( ON ) ( LOWERQ ) )
t ON = ---------------------------------------------------------------------------------------------------------V IN
(EQ. 1)
See Table 2 for approximate K- factors. Switching frequency
increases as a function of load current due to the increasing
drop across the synchronous rectifier, which causes a faster
inductor-current discharge ramp. On-times translate only
roughly to switching frequencies. The on-times established
in the “Electrical Specifications” table on page 4 are
influenced by switching delays in the external high-side
power MOSFET. Also, the dead-time effect increases the
effective on-time, reducing the switching frequency. It occurs
only in PWM mode (SKIP = VCC) and during dynamic output
voltage transitions when the inductor current reverses at
light or negative load currents. With reversed inductor
current, the inductor's EMF causes PHASE to go high earlier
than normal, extending the on-time by a period equal to the
UGATE-rising dead time.
VOLTAGE at BYP
LDOREFIN < 0.3V,
BYP > 4.63V
Internal LDO is
disabled.
VOLTAGE at BYP
LDOREFIN > VCC - 1V,
BYP > 3V
Internal LDO is
disabled.
5V
LDOREFIN < 0.3V,
BYP < 4.63V
Internal LDO is
active.
3.3V
LDOREFIN > VCC - 1V,
BYP < 3V
Internal LDO is
active.
SMPS
2 x LDOREFIN
0.35V < LDOREFIN < 2.25V
Internal LDO is
active.
(tON = GND, REF,
or OPEN), VOUT1
400
2.5
±10
(tON = GND),
VOUT2
500
2.0
±10
(tON = VCC),
VOUT1
200
5.0
±10
(tON = VCC, REF,
or OPEN), VOUT2
300
3.3
±10
FREE-RUNNING, CONSTANT ON-TIME PWM
CONTROLLER WITH INPUT FEED-FORWARD
The constant on-time PWM control architecture is a
pseudo-fixed-frequency, constant on-time, current-mode
type with voltage feed forward. The constant on-time PWM
control architecture relies on the output ripple voltage to
provide the PWM ramp signal; thus the output filter
capacitor's ESR acts as a current-feedback resistor. The
high-side switch on-time is determined by a one-shot whose
period is inversely proportional to input voltage and directly
proportional to output voltage. Another one-shot sets a
minimum off-time (300ns typ). The on-time one-shot triggers
when the following conditions are met: the error comparator's
output is high, the synchronous rectifier current is below the
current-limit threshold, and the minimum off time one-shot
has timed out. The controller utilizes the valley point of the
output ripple to regulate and determine the off-time.
ON-TIME ONE-SHOT (tON)
Each PWM core includes a one-shot that sets the high-side
switch on-time for each controller. Each fast, low-jitter,
adjustable one-shot includes circuitry that varies the on-time
in response to battery and output voltage. The high-side
switch on-time is inversely proportional to the battery voltage
20
TABLE 2. APPROXIMATE K-FACTOR ERRORS
APPROXIMATE
SWITCHING
K-FACTOR
FREQUENCY K-FACTOR
ERROR (%)
(kHz)
(µs)
For loads above the critical conduction point, the actual
switching frequency is:
V OUT + V DROP1
f = ------------------------------------------------------t ON ( V IN + V DROP2 )
(EQ. 2)
where:
• VDROP1 is the sum of the parasitic voltage drops in the
inductor discharge path, including synchronous rectifier,
inductor and PC board resistances
• VDROP2 is the sum of the parasitic voltage drops in the
charging path, including high-side switch, inductor and PC
board resistances
• tON is the on-time calculated by the ISL6236A
FN6453.3
March 18, 2008
ISL6236A
VIN: 5.5V TO 25V
5V
C5
1µF
C8
1µF
PVCC
VCC
VIN
C10
10µF
NC
LDO
GND
LDOREFIN
BOOT1
BOOT2
UGATE1
UGATE2
PHASE1
PHASE2
LGATE1
LGATE2
C1
10
10µF
Q3a
SI4816BDY
C9
0.1µF
OUT1 – PCI-e
L1: 3.3µH
1.25V/5A
C4
0.22µF
Q3b
C11
330µF
9mΩ
6.3V
PGND
EN1
OUT2
OUT2-GFX
L2: 2.2µH TRACK REFIN2/10A
Q2
IRF7832
C2
2 x 330µF
4mΩ
6.3V
VCC
R1
7.87kΩ
5V
R2
10kΩ
OUT1
Q1
IRF7821
BYP
FB1 TIED TO GND = 5V
FB1 TIED TO VCC = 1.5V
R3
200kΩ
ISL6236A
EN2
FB1
AGND
REFIN2
ILIM1
ILIM2
SKIP
VREF3
GND
EN LDO
VCC
SECFB
VCC
TON
VCC
2 BITS
DAC
REFIN2: DYNAMIC 0V TO 2.5V
REFIN2 TIED TO VREF3 = 1.05V
REFIN2 TIED TO VCC = 3.3V
+
+
R5
200kΩ
C3
OPEN
C7
0.1µF
REF
VCC
DROOP
+
VCC
R4
200kΩ
R6
200kΩ
POK1
POK2
PAD
FREQUENCY-DEPENDENT COMPONENTS
tON = VCC
1.25V/1.05V SMPS
SWITCHING
FREQUENCY
200kHz/300kHz
L1
3.3µH
L2
2.7µH
C2
2 x 330µF
C11
330µF
FIGURE 66. ISL6236A TYPICAL DYNAMIC GFX APPLICATION CIRCUIT
21
FN6453.3
March 18, 2008
ISL6236A
VIN: 5.5V TO 25V
5V
C5
1µF
LDOREFIN TIED TO GND = 5V
LDOREFIN TIED TO VCC = 3.3V
LDO
C8
1µF
PVCC
VCC
VIN
C10
10µF
SI4816BDY
LDO
VCC
LDOREFIN
BOOT1
BOOT2
UGATE1
UGATE2
C1
10
10µF
C6
F
4.7µF
Q3a
OUT1
1.5V/5A
C9
0.1µF
L1: 3.3µH
PHASE1
PHASE2
LGATE1
LGATE2
Q1a
C4
0.22µF
Q3b
C11
330µF
9mΩ
6.3V
OUT1
PGND
EN1
OUT2
OUT2
L2: 2.2µF 1.05V/5A
Q1b SI4816BDY
C2
330µF
4mΩ
6.3V
VCC
3.3V
VCC
FB1 TIED TO GND = 5V
FB1 TIED TO VCC = 1.5V
BYP
ISL6236A
VCC
EN2
REFIN2: DYNAMIC 0V TO 2.5V
REFIN2 TIED TO VREF3 = 1.05V
VREF3 REFIN2 TIED TO VCC = 3.3V
FB1
R3
200kΩ
REFIN2
AGND
ILIM1
ILIM2
SKIP
VREF3
R5
200kΩ
ON
EN LDO
REF
OFF
VCC
SECFB
VCC
TON
C3
0.01µF
C7
0.1µF
VCC
R4
200kΩ
VCC
R6
200kΩ
POK1
POK2
PAD
FREQUENCY-DEPENDENT COMPONENTS
tON = VCC
1.5V/1.05V SMPS
SWITCHING
FREQUENCY
200kHz/300kHz
L1
3.3µH
L2
2.7µH
C2
330µF
C11
330µF
FIGURE 67. ISL6236A TYPICAL SYSTEM REGULATOR APPLICATION CIRCUIT WITHOUT CHARGE PUMP
22
FN6453.3
March 18, 2008
ISL6236A
VIN: 4.5V TO 5.5V*
C5
1µF
R7
10Ω
LDOREFIN TIED TO GND = 5V
LDOREFIN TIED TO VCC = 3.3V
LDO
C8
1µF
PVCC
VCC
VIN
C10
10µF
SI4816BDY
LDO
LDOREFIN
BOOT1
BOOT2
UGATE1
UGATE2
VCC
C1
10
10µF
C6
F
4.7µF
Q3a
OUT1
1.5V/5A
C11
330µF
9mΩ
6.3V
C9
0.1µF
L1: 3.3µH
PHASE1
PHASE2
LGATE1
LGATE2
Q1a
C4
0.22µF
Q3b
OUT1
PGND
EN1
OUT2
OUT2
L2: 2.2µF 1.05V/5A
Q1b SI4816BDY
C2
330µF
4mΩ
6.3V
VCC
3.3V
VCC
FB1 TIED TO GND = 5V
FB1 TIED TO VCC = 1.5V
R3
200kΩ
ISL6236A
BYP
VCC
EN2
FB1
REFIN2
AGND
ILIM1
ILIM2
SKIP
VREF3
R5
200kΩ
C3
0.01µF
C7
0.1µF
ON
EN LDO
REF
OFF
VCC
SECFB
VCC
TON
REFIN2: DYNAMIC 0 TO 2.5V
REFIN2 TIED TO VREF3 = 1.05V
VREF3 REFIN2 TIED TO VCC = 3.3V
VCC
R4
200kΩ
VCC
R6
200kΩ
POK1
POK2
PAD
*NOTE: To prevent noise in high loading conditions, place a 10Ω resistor between VIN and PVCC. Additional
electrolytic bulk decoupling can also be used for this purpose
FREQUENCY-DEPENDENT COMPONENTS
tON = VCC
1.5V/1.05V SMPS
SWITCHING
FREQUENCY
200kHz/300kHz
L1
3.3µH
L2
2.7µH
C2
330µF
C11
330µF
FIGURE 68. ISL6236A TYPICAL LOW INPUT VOLTAGE SYSTEM REGULATOR APPLICATION CIRCUIT WITHOUT CHARGE PUMP
23
FN6453.3
March 18, 2008
ISL6236A
VIN: 5.5V TO 25V
C5
1µF
PVCC
LDOREFIN TIED TO GND = 5V
LDOREFIN TIED TO VCC = 3.3V
LDO
VCC
LDO
VIN
C10
10µF
Q3
IRF7807V
OUT1
5V/7A
C9
0.1µF
L1: 4.7µH
Q4
IRF7811AV
C11
330µF
9mΩ
6.3V
BOOT1
BOOT2
UGATE1
UGATE2
PHASE1
PHASE2
LGATE1
LGATE2
D3
EN1
C8
0.1µF
C12
D1a
0.1µF
D1b
CP
14V/10mA
ISL6236A
ILIM1
ILIM2
SKIP
VREF3
REF
OFF
D2
C15
0.1µF
OUT2
3.3V/11A
C2
330µF
9mΩ
4V
VCC
REFIN2: DYNAMIC 0V TO 2V
REFIN2 TIED TO VREF3 = 1.05V
REFIN2 TIED TO VCC = 3.3V
R5
150kΩ
EN LDO
SECFB
R1
200kΩ
VCC
REFIN2
AGND
ON
D2b
L2: 4.7µH
Q2
IRF7832
EN2
FB1 TIED TO GND = 5V
FB1 TIED TO VCC = 1.5V
C14
0.1µF
D2a
C4
0.1µF
FB1
R3
200kΩ
Q1
IRF7821
OUT2
BYP
D1
C1
10µF
10
PGND
OUT1
VCC
C6
4.7µF
LDOREFIN
C3
OPEN
C7
0.1µF
VCC
VCC
R4
200kΩ
R6
200kΩ
POK1
GND
R2
39.2kΩ
TON
POK2
PAD
FREQUENCY-DEPENDENT COMPONENTS
5V/3.3V SMPS
SWITCHING
FREQUENCY
tON = VCC
tON = REF
(OR OPEN)
tON = GND
200kHz/300kHz
400kHz/300kHz
400kHz/500kHz
L1
6.8µH
6.8µH
4.7µH
L2
7.6µH
4.7µH
4.7µH
C2
2x470µF
2x330µF
2x330µF
C11
330µF
330µF
330µF
FIGURE 69. ISL6236A TYPICAL SYSTEM REGULATOR APPLICATION CIRCUIT WITH 14V CHARGE PUMP
24
FN6453.3
March 18, 2008
ISL6236A
TON
SKIP
BOOT1
BOOT2
UGATE1
UGATE2
PHASE2
PHASE1
PVCC
PVCC
SMPS1
SYNCHRONOUS
PWM BUCK
CONTROLLER
LGATE1
GND
ILIM1
EN1
SMPS2
SYNCHRONOUS
PWM BUCK
CONTROLLER
LGATE2
PGND
ILIM2
EN2
SECFB
FB1
POK1
OUT1
REFIN2
POK2
OUT1
BYP
OUT2
OUT2
POK2
+
-
SW THRESHOLD
POK1
LDO
LDO
VCC
INTERNAL
LOGIC
LDOREFIN
10Ω
VIN
PVCC
EN LDO
EN1
EN2
POWER-ON
SEQUENCE
SQUENCE
CLEAR FAULT
CLEAR
FAULT
LATCH
LATCH
VREF3
VREF3
THERMAL
THERMAL
SHUTDOWN
SHUTDOWN
REF
REF
FIGURE 70. DETAILED FUNCTIONAL DIAGRAM ISL6236A
25
FN6453.3
March 18, 2008
ISL6236A
tON
MIN. tOFF
Q
TRIG
ONE SHOT
VIN
+
Q
R Q
OUT
TO UGATE DRIVER
S Q
Q
+
+
+
+
+
ILIM
REFIN2 (SMPS2)
VREF
COMP
SLOPE COMP
BOOT
UV
DETECT
+
5µA
VCC
BOOT
+
TO LGATE DRIVER
Â
Σ
+
PHASE
Q
S Q
+
OUT
R Q
Q
ONE-SHOT
SKIP
SECFB
+
+
PGOOD
+
FB
DECODER
OV LATCH
2V
SMSP1 ONLY
0.9VREF
FB
1.1VREF
UV LATCH
+
0.7VREF
FAULT
FAULT
LATCH
LATCH
LOGIC
20ms
BLANKING
FIGURE 71. PWM CONTROLLER (ONE SIDE ONLY)
26
FN6453.3
March 18, 2008
ISL6236A
Automatic Pulse-Skipping Switchover
(Idle Mode)
In Idle Mode (SKIP = GND), an inherent automatic
switchover to PFM takes place at light loads. This switchover
is affected by a comparator that truncates the low-side
switch on-time at the inductor current's zero crossing. This
mechanism causes the threshold between pulse-skipping
PFM and nonskipping PWM operation to coincide with the
boundary between continuous and discontinuous
inductor-current operation (also known as the critical
conduction point):
K ⋅ V OUT V IN – V OUT
I LOAD ( SKIP ) = ------------------------ -------------------------------2⋅L
V IN
(EQ. 3)
where K is the on-time scale factor (see “ON-TIME ONESHOT (tON)” on page 20). The load-current level at which
PFM/PWM crossover occurs, ILOAD(SKIP), is equal to half
the peak-to-peak ripple current, which is a function of the
inductor value (Figure 72). For example, in the ISL6236A
typical application circuit with VOUT1 = 5V, VIN = 12V,
L = 7.6µH, and K = 5µs, switchover to pulse-skipping
operation occurs at ILOAD = 0.96A or about on-fifth full load.
The crossover point occurs at an even lower value if a
swinging (soft-saturation) inductor is used.
ΔI
=
V IN -V OUT
t
L
INDUCTOR CURRENT
I PEAK
ILOAD = IPEAK/2
0
ON-TIME
TIME
FIGURE 72. ULTRASONIC CURRENT WAVEFORMS
The switching waveforms may appear noisy and
asynchronous when light loading causes pulse-skipping
operation, but this is a normal operating condition that
results in high light-load efficiency. Trade-offs in PFM noise
vs light-load efficiency are made by varying the inductor
value. Generally, low inductor values produce a broader
efficiency vs load curve, while higher values result in higher
full-load efficiency (assuming that the coil resistance remains
fixed) and less output voltage ripple. Penalties for using
higher inductor values include larger physical size and
degraded load-transient response (especially at low
input-voltage levels).
conduction, the output voltage has a DC regulation higher
than the trip level by 50% of the ripple. In discontinuous
conduction (SKIP = GND, light load), the output voltage has
a DC regulation higher than the trip level by approximately
1.0% due to slope compensation.
Forced-PWM Mode
The low-noise, forced-PWM (SKIP = VCC) mode disables
the zero-crossing comparator, which controls the low-side
switch on-time. Disabling the zero-crossing detector causes
the low-side, gate-drive waveform to become the
complement of the high-side, gate-drive waveform. The
inductor current reverses at light loads as the PWM loop
strives to maintain a duty ratio of VOUT/VIN. The benefit of
forced-PWM mode is to keep the switching frequency fairly
constant, but it comes at a cost: the no-load battery current
can be 10mA to 50mA, depending on switching frequency
and the external MOSFETs.
Forced-PWM mode is most useful for reducing
audio-frequency noise, improving load-transient response,
providing sink-current capability for dynamic output voltage
adjustment, and improving the cross-regulation of
multiple-output applications that use a flyback transformer or
coupled inductor.
Enhanced Ultrasonic Mode
(25kHz (min) Pulse Skipping)
Leaving SKIP unconnected or connecting SKIP to REF
activates a unique pulse-skipping mode with a minimum
switching frequency of 25kHz. This ultrasonic pulse-skipping
mode eliminates audio-frequency modulation that would
otherwise be present when a lightly loaded controller
automatically skips pulses. In ultrasonic mode, the controller
automatically transitions to fixed-frequency PWM operation
when the load reaches the same critical conduction point
(ILOAD(SKIP)).
An ultrasonic pulse occurs when the controller detects that
no switching has occurred within the last 20µs. Once
triggered, the ultrasonic controller pulls LGATE high, turning
on the low-side MOSFET to induce a negative inductor
current. After FB drops below the regulation point, the
controller turns off the low-side MOSFET (LGATE pulled low)
and triggers a constant on-time (UGATE driven high). When
the on-time has expired, the controller re-enables the
low-side MOSFET until the controller detects that the
inductor current dropped below the zero-crossing threshold.
Starting with a LGATE pulse greatly reduces the peak output
voltage when compared to starting with a UGATE pulse, as
long as VFB < VREF, LGATE is off and UGATE is on, similar
to pure SKIP mode.
DC output accuracy specifications refer to the trip level of the
error comparator. When the inductor is in continuous
27
FN6453.3
March 18, 2008
ISL6236A
D2a. Finally, C14 charges C15 thru D2b when LGATE1
switched to high. CP output voltage is shown in Equation 4:
40µs (MAX)
INDUCTOR
CURRENT
CP = V OUT1 + 2 ⋅ V LGATE1 – 4 ⋅ V D
(EQ. 4)
where:
• VLGATE1 is the peak voltage of the LGATE1 driver
0A
FB<REG.POINT
FB<Reg.Point
) )
ON-TIME (tON)ON
FIGURE 73. ULTRASONIC CURRENT WAVEFORMS
Reference and Linear Regulators (VREF3,
REF, LDO and 14V Charge Pump)
The 3.3V reference (VREF3) is accurate to ±1.5%
over-temperature, making VREF3 useful as a precision
system reference. VREF3 can supply up to 5mA for external
loads. Bypass VREF3 to GND with a 0.01µF capacitor.
Leave it open if there is no load.
The 2V reference (REF) is accurate to ±1% over- temperature,
also making REF useful as a precision system reference.
Bypass REF to GND with a 0.1µF (min) capacitor. REF can
supply up to 50µA for external loads.
An internal regulator produces a fixed 5V
(LDOREFIN < 0.2V) or 3.3V (LDOREFIN > VCC - 1V). In an
adjustable mode, the LDO output can be set from 0.7V to
4.5V. The LDO output voltage is equal to two times the
LDOREFIN voltage. The LDO regulator can supply up to
100mA for external loads. Bypass LDO with a minimum
4.7µF ceramic capacitor. When the LDOREFIN < 0.2V and
BYP voltage is 5V, the LDO bootstrap-switchover to an
internal 0.7Ω P-Channel MOSFET switch connects BYP to
LDO pin while simultaneously shutting down the internal
linear regulator. These actions bootstrap the device,
powering the loads from the BYP input voltages, rather than
through internal linear regulators from the battery. Similarly,
when the BYP = 3.3V and LDOREFIN = VCC, the LDO
bootstrap-switchover to an internal 1.5Ω P-Channel
MOSFET switch connects BYP to LDO pin while
simultaneously shutting down the internal linear regulator.
No switchover action in adjustable mode.
In Figure 68, the external 14V charge pump is driven by
LGATE1. When LGATE1 is low, D1a charged C8 sourced
from OUT1. C8 voltage is equal to OUT1 minus a diode
drop. When LGATE1 transitions to high, the charges from C8
will transfer to C12 through D1b and charge it to VLGATE1
plus VC8. As LGATE1 transitions low on the next cycle, C12
will charge C14 to its voltage minus a diode drop through
• VD is the forward diode dropped across the Schottkys
SECFB is used to monitor the charge pump through the
resistive divider. In an event when SECFB dropped below
2V, the detection circuit force the highside MOSFET
(SMPS1) off and the lowside MOSFET (SMPS1) on for
300ns to allow CP to recharge and SECFB rise above 2V. In
the event of an overload on CP where SECFB cannot reach
more than 2V, the monitor will be deactivated. Special care
should be taken to ensure enough normal voltage ripple on
each cycle as to prevent CP shut-down. The SECFB pin has
~17mV of hysteresis, so the ripple should be enough to bring
the SECFB voltage above the threshold by ~3x the
hysteresis, or (2V + 3*17mV) = 2.051V. Reducing the CP
decoupling capacitor and placing a small ceramic capacitor
(10pF to 47pF) in parallel with the upper leg of the SECFB
resistor feedback network (R1 of Figure 68), will also
increase the robustness of the charge pump.
Current-Limit Circuit (ILIM ) with rDS(ON)
Temperature Compensation
The current-limit circuit employs a "valley" current-sensing
algorithm. The ISL6236A uses the ON-resistance of the
synchronous rectifier as a current-sensing element. If the
magnitude of the current-sense signal at PHASE is above
the current-limit threshold, the PWM is not allowed to initiate
a new cycle. The actual peak current is greater than the
current-limit threshold by an amount equal to the inductor
ripple current. Therefore, the exact current-limit
characteristic and maximum load capability are a function of
the current-limit threshold, inductor value and input and
output voltage.
I PEAK
INDUCTOR CURRENT
ZERO-CROSSING
Zero-Crossing
DETECTION
Detection
I LOAD
ΔI
I LIMIT
I LOAD(MAX)
ILIM (VAL) = I
LOAD -
ΔI
2
TIME
FIGURE 74. “VALLEY” CURRENT LIMIT THRESHOLD POINT
28
FN6453.3
March 18, 2008
ISL6236A
For lower power dissipation, the ISL6236A uses the
ON-resistance of the synchronous rectifier as the
current-sense element. Use the worst-case maximum value
for rDS(ON) from the MOSFET data sheet. Add some margin
for the rise in rDS(ON) with temperature. A good general rule
is to allow 0.5% additional resistance for each °C of
temperature rise. The ISL6236A controller has a built-in 5µA
current source as shown in Figure 75. Place the hottest
power MOSEFTs as close to the IC as possible for best
thermal coupling. The current limit varies with the
ON-resistance of the synchronous rectifier. When combined
with the undervoltage-protection circuit, this current-limit
method is effective in almost every circumstance.
5V
5V
BOOT
10Ω
10
UGATE
VIN
Q1
C BOOT
OUT
PHASE
ISL88732
ISL88733
ISL6236A
ISL6236A
ISL88734
FIGURE 76. REDUCING THE SWITCHING-NODE RISE TIME
ILIM
+
5µA
+
RILIM
VILIM
9R
VCC
TO CURRENT
LIMIT LOGIC
R
FIGURE 75. CURRENT LIMIT BLOCK DIAGRAM
A negative current limit prevents excessive reverse inductor
currents when VOUT sinks current. The negative
current-limit threshold is set to approximately 120% of the
positive current limit and therefore tracks the positive current
limit when ILIM is adjusted. The current-limit threshold is
adjusted with an external resistor for ISL6236A at ILIM. The
current-limit threshold adjustment range is from 20mV to
200mV. In the adjustable mode, the current-limit threshold
voltage is 1/10th the voltage at ILIM. The voltage at ILIM pin
is the product of 5µA*RILIM. The threshold defaults to
100mV when ILIM is connected to VCC. The logic threshold
for switch-over to the 100mV default value is approximately
VCC -1V.
The PC board layout guidelines should be carefully
observed to ensure that noise and DC errors do not corrupt
the current-sense signals at PHASE.
MOSFET Gate Drivers (UGATE, LGATE)
The UGATE and LGATE gate drivers sink 2.0A and 3.3A
respectively of gate drive, ensuring robust gate drive for
high-current applications. The UGATE floating high-side
MOSFET drivers are powered by diode-capacitor charge
pumps at BOOT. The LGATE synchronous-rectifier drivers
are powered by PVCC.
29
The internal pull-down transistors that drive LGATE low have
a 0.6Ω typical ON-resistance. These low ON-resistance
pull-down transistors prevent LGATE from being pulled up
during the fast rise time of the inductor nodes due to
capacitive coupling from the drain to the gate of the low-side
synchronous-rectifier MOSFETs. However, for high-current
applications, some combinations of high- and low-side
MOSFETs may cause excessive gate-drain coupling, which
leads to poor efficiency and EMI-producing shoot-through
currents. Adding a 1Ω resistor in series with BOOT
increases the turn-on time of the high-side MOSFETs at the
expense of efficiency, without degrading the turn-off time
(see Figure 76).
Adaptive dead-time circuits monitor the LGATE and UGATE
drivers and prevent either FET from turning on until the other
is fully off. This algorithm allows operation without shootthrough with a wide range of MOSFETs, minimizing delays
and maintaining efficiency. There must be low-resistance,
low-inductance paths from the gate drivers to the MOSFET
gates for the adaptive dead-time circuit to work properly.
Otherwise, the sense circuitry interprets the MOSFET gate
as "off" when there is actually charge left on the gate. Use
very short, wide traces measuring 10 to 20 squares (50 mils
to 100 mils wide if the MOSFET is 1” from the device).
Boost-Supply Capacitor Selection (Buck)
The boost capacitor should be 0.1µF to 4.7µF, depending on
the input and output voltages, external components, and PC
board layout. The boost capacitance should be as large as
possible to prevent it from charging to excessive voltage, but
small enough to adequately charge during the minimum
low-side MOSFET conduction time, which happens at
maximum operating duty cycle (this occurs at minimum input
voltage). The minimum gate to source voltage (VGS(MIN)) is
determined by:
C BOOT
V GS ( MIN ) = PVCC ⋅ --------------------------------------C BOOT + C GS
(EQ. 5)
FN6453.3
March 18, 2008
ISL6236A
where:
OVERVOLTAGE PROTECTION
• PVCC is 5V
In pure skip mode, the converter frequency can be very low
with little to no output loading. This produces very long off
times, where leakage can bleed down the BOOT capacitor
voltage. If the voltage falls too low, the converter may not be
able to turn on UGATE when the output voltage falls to the
reference. To prevent this, the ISL6236A monitors the BOOT
capacitor voltage, and if it falls below 3V, it initiates an
LGATE pulse, which will refresh the BOOT voltage.
When the output voltage of VOUT1 is 11% (16% for VOUT2)
above the set voltage, the overvoltage fault protection
activates. This latches on the synchronous rectifier MOSFET
with 100% duty cycle, rapidly discharging the output
capacitor until the negative current limit is achieved. Once
negative current limit is met, UGATE is turned on for a
minimum on-time, followed by another LGATE pulse until
negative current limit. This effectively regulates the
discharge current at the negative current limit in an effort to
prevent excessively large negative currents that cause
potentially damaging negative voltages on the load. Once an
overvoltage fault condition is set, it can only be reset by
toggling SHDN, EN, or cycling VIN (POR).
POR, UVLO and Internal Digital Soft-Start
UNDERVOLTAGE PROTECTION
Power-on reset (POR) occurs when VIN rises above
approximately 3V, resetting the undervoltage, overvoltage,
and thermal-shutdown fault latches. PVCC undervoltage
lockout (UVLO) circuitry inhibits switching when PVCC is
below 4V. LGATE is low during UVLO. The output voltages
begin to ramp up once PVCC exceeds its 4V UVLO and REF
is in regulation. The internal digital soft-start timer begins to
ramp up the maximum-allowed current limit during start-up.
The 1.7ms ramp occurs in five steps. The step size are 20%,
40%, 60%, 80% and 100% of the positive current limit value.
When the output voltage drops below 70% of its regulation
voltage for at least 100µs, the controller sets the fault latch
and begins the discharge mode (see “Shutdown Mode” and
“Discharge Mode” on page 30). UVP is ignored for at least
20ms (typical), after start-up or after a rising edge on EN.
Toggle EN or cycle VIN (POR) to clear the undervoltage fault
latch and restart the controller. UVP only applies to the buck
outputs.
• CGS is the gate capacitance of the high-side MOSFET
Boost-Supply Refresh Monitor
Power-Good Output (POK)
The POK comparator continuously monitors both output
voltages for undervoltage conditions. POK is actively held
low in shutdown, standby, and soft-start. POK1 releases and
digital soft-start terminates when VOUT1 outputs reach the
error-comparator threshold. POK1 goes low if VOUT1 output
turns off or is 10% below its nominal regulation point. POK1
is a true open-drain output. Likewise, POK2 is used to
monitor VOUT2.
Fault Protection
THERMAL PROTECTION
The ISL6236A has thermal shutdown to protect the devices
from overheating. Thermal shutdown occurs when the die
temperature exceeds +150°C. All internal circuitry shuts
down during thermal shutdown. The ISL6236A may trigger
thermal shutdown if LDO is not bootstrapped from OUT
while applying a high input voltage on VIN and drawing the
maximum current (including short circuit) from LDO. Even if
LDO is bootstrapped from OUT, overloading the LDO causes
large power dissipation on the bootstrap switches, which
may result in thermal shutdown. Cycling EN, EN LDO, or
VIN (POR) ends the thermal-shutdown state.
Discharge Mode (Soft-Stop)
The ISL6236A provides overvoltage/undervoltage fault
protection in the buck controllers. Once activated, the
controller continuously monitors the output for undervoltage
and overvoltage fault conditions.
OUT-OF-BOUND CONDITION
When the output voltage is 5% above the set voltage, the
out-of-bound condition activates. LGATE turns on until
output reaches within regulation. Once the output is within
regulation, the controller will operate as normal. It is the "first
line of defense" before OVP. The output voltage ripple must
be sized low enough as to not nuisance trip the OOB
threshold. The equations in “Output Capacitor Selection” on
page 33 should be used to size the output voltage ripple
below 3% of the nominal output voltage set point.
30
When a transition to standby or shutdown mode occurs, or
the output undervoltage fault latch is set, the outputs
discharge to GND through an internal 25Ω switch. The
reference remains active to provide an accurate threshold
and to provide overvoltage protection.
Shutdown Mode
The ISL6236A SMPS1, SMPS2 and LDO have independent
enabling control. Drive EN1, EN2 and EN LDO below the
precise input falling-edge trip level to place the ISL6236A in
its low-power shutdown state. The ISL6236A consumes only
20µA of quiescent current while in shutdown. When
shutdown mode activates, the 3.3V VREF3 remain on. Both
SMPS outputs are discharged to 0V through a 25Ω switch.
FN6453.3
March 18, 2008
ISL6236A
Power-Up Sequencing and On/Off Controls (EN )
second SMPS remains on until the first SMPS turns off, the
device shuts down, a fault occurs or PVCC goes into
undervoltage lockout. Both supplies begin their power-down
sequence immediately when the first supply turns off. Driving
EN below 0.8V clears the overvoltage, undervoltage and
thermal fault latches.
EN1 and EN2 control SMPS power-up sequencing. EN1 or
EN2 rising above 2.4V enables the respective outputs. EN1
or EN2 falling below 1.6V disables the respective outputs.
Connecting EN1 or EN2 to REF will force its outputs off while
the other output is below regulation. The sequenced SMPS
will start once the other SMPS reaches regulation. The
TABLE 3. OPERATING-MODE TRUTH TABLE
MODE
CONDITION
COMMENT
Power-Up
PVCC < UVLO threshold.
Transitions to discharge mode after a VIN POR and after REF becomes valid. LDO,
VREF3 and REF remain active.
Run
EN LDO = high, EN1 or EN2
enabled.
Normal operation
Overvoltage
Protection
Either output > 111% (VOUT1) or
116% (VOUT2) of nominal level.
LGATE is forced high. LDO, VREF3 and REF active. Exited by a VIN POR, or by
toggling EN1 or EN2.
Undervoltage
Protection
Either output < 70% of nominal after
20ms time-out expires and output is
enabled.
The internal 25Ω switch turns on. LDO, VREF3 and REF are active. Exited by a VIN
POR or by toggling EN1 or EN2.
Discharge
Either SMPS output is still high in
either standby mode or shutdown
mode
Discharge switch (25Ω) connects OUT to GND. One output may still run while the
other is in discharge mode. Activates when PVCC is in UVLO, or transition to UVLO,
standby, or shutdown has begun. LDO, VREF3 and REF active.
Standby
EN1, EN2 < start-up threshold, EN
LDO = High
LDO, VREF3 and REF active.
Shutdown
EN1, EN2, EN LDO = low
Discharge switch (25Ω) connects OUT to PGND. All circuitry off except VREF3.
Thermal Shutdown
TJ > +150°C
All circuitry off. Exited by VIN POR or cycling EN. VREF3 remain active.
TABLE 4. SHUTDOWN AND STANDBY CONTROL LOGIS
VEN LDO
VEN1 (V)
VEN2 (V)
LDO
SMPS1
SMPS2
Low
Low
Low
Off
Off
Off
“>2.5” → High
Low
Low
On
Off
Off
“>2.5” → High
High
High
On
On
On
“>2.5” → High
High
Low
On
On
Off
“>2.5” → High
Low
High
On
Off
On
“>2.5” → High
High
REF
On
On
On (after SMPS1 is up)
“>2.5” → High
REF
High
On
On (after SMPS2 is up)
On
31
FN6453.3
March 18, 2008
ISL6236A
Adjustable-Output Feedback (Dual-Mode FB)
Connect FB1 to GND to enable the fixed 5V or tie FB1 to
VCC to set the fixed 1.5V output. Connect a resistive
voltage-divider at FB1 between OUT1 and GND to adjust the
respective output voltage between 0.7V and 5.5V (see
Figure 77). Choose R2 to be approximately 10k and solve for
R1 using Equation 6.
⎛ V OUT1
⎞
R 1 = R 2 ⋅ ⎜ ------------------- – 1⎟
⎝ V FB1
⎠
(EQ. 6)
condition. The minimum practical inductor value is one
that causes the circuit to operate at critical conduction
(where the inductor current just touches zero with every
cycle at maximum load). Inductor values lower than this
grant no further size-reduction benefit.
The ISL6236A pulse-skipping algorithm (SKIP = GND)
initiates skip mode at the critical conduction point, so the
inductor's operating point also determines the load
current at which PWM/PFM switchover occurs. The
optimum LIR point is usually found between 25% and
50% ripple current.
where VFB1 = 0.7V nominal.
Likewise, connect REFIN2 to VCC to enable the fixed 3.3V
or tie REFIN2 to VREF3 to set the fixed 1.05V output. Set
REFIN2 from 0V to 2.50V for SMPS2 tracking mode (see
Figure 78).
VR
R 3 = R 4 ⋅ ⎛ ------------------- – 1⎞
⎝V
⎠
OUT2
(EQ. 7)
VIN
Q3
UGATE1
OUT1
ISL6236A
Q4
LGATE1
where:
• VR = 2V nominal (if tied to REF)
R1
OUT1
or
FB1
• VR = 3.3V nominal (if tied to VREF3)
R2
Design Procedure
Establish the input voltage range and maximum load current
before choosing an inductor and its associated ripple current
ratio (LIR). The following four factors dictate the rest of the
design:
1. Input Voltage Range. The maximum value (VIN(MAX))
must accommodate the maximum AC adapter voltage.
The minimum value (VIN(MIN)) must account for the
lowest input voltage after drops due to connectors, fuses
and battery selector switches. Lower input voltages result
in better efficiency.
2. Maximum Load Current. The peak load current
(ILOAD(MAX)) determines the instantaneous component
stress and filtering requirements and thus drives output
capacitor selection, inductor saturation rating and the
design of the current-limit circuit. The continuous load
current (ILOAD) determines the thermal stress and drives
the selection of input capacitors, MOSFETs and other
critical heat-contributing components.
3. Switching Frequency. This choice determines the basic
trade-off between size and efficiency. The optimal
frequency is largely a function of maximum input voltage
and MOSFET switching losses.
4. Inductor Ripple Current Ratio (LIR). LIR is the ratio of
the peak-peak ripple current to the average inductor
current. Size and efficiency trade-offs must be
considered when setting the inductor ripple current ratio.
Low inductor values cause large ripple currents, resulting
in the smallest size, but poor efficiency and high output
noise. Also, total output ripple above 3.5% of the output
regulation will cause controller to trigger out-of-bound
32
FIGURE 77. SETTING VOUT1 WITH A RESISTOR DIVIDER
VIN
Q1
UGATE2
UGATE2
UGATE
ISL88732
OUT2
ISL6236A
ISL88733
ISL88734
LGATE2
LGATE
LGATE2
Q2
OUT2
VOUT
OUT2
VR
FB
REFIN2
REFIN2
R3
R4
FIGURE 78. SETTING VOUT2 WITH A VOLTAGE DIVIDER FOR
TRACKING
FN6453.3
March 18, 2008
ISL6236A
Inductor Selection
The switching frequency (on-time) and operating point
(% ripple or LIR) determine the inductor value using
Equation 8:
V OUT_ ( V IN + V OUT_ )
L = --------------------------------------------------------------------V IN ⋅ f ⋅ LIR ⋅ I LOAD ( MAX )
(EQ. 8)
Example: ILOAD(MAX) = 5A, VIN = 12V, VOUT2 = 5V,
f = 200kHz, 35% ripple current or LIR = 0.35:
5V ( 12V – 5V )
L = ----------------------------------------------------------------- = 8.3μH
12V ⋅ 200kHz ⋅ 0.35 ⋅ 5A
(EQ. 9)
(EQ. 10)
The inductor ripple current also impacts transient response
performance, especially at low VIN - VOUT differences. Low
inductor values allow the inductor current to slew faster,
replenishing charge removed from the output filter capacitors
by a sudden load step. The peak amplitude of the output
transient (VSAG) is also a function of the maximum duty
factor, which can be calculated from the on-time and
minimum off-time:
⎞⎞
⎛ ⎛ V OUT_
2
( ΔI LOAD ( MAX ) ) ⋅ L ⎜ K ⎜ ------------------- + t OFF ( MIN )⎟ ⎟
V
⎠⎠
⎝ ⎝
IN
VSAG = ---------------------------------------------------------------------------------------------------------------------------⎛ V IN – V OUT⎞
2 ⋅ C OUT ⋅ V OUT K ⎜ --------------------------------⎟ - t
V IN
⎝
⎠ OFF ( MIN )
(EQ. 11)
where minimum off-time = 0.35µs (max) and K is from
Table 2.
Determining the Current Limit
The minimum current-limit threshold must be great enough
to support the maximum load current when the current limit
is at the minimum tolerance value. The valley of the inductor
current occurs at ILOAD(MAX) minus half of the ripple
current; therefore:
I LIMIT ( LOW ) > I LOAD ( MAX ) – [ ( LIR ⁄ 2 ) ⋅ I LOAD ( MAX ) ]
I LIMIT ( LOW ) = ( 25mV ) ⁄ ( ( 5mΩ × 1.2 ) > 5A – ( 0.35 ⁄ 2 )5A )
(EQ. 13)
4.17A > 4.12A
(EQ. 14)
4.17A is greater than the valley current of 4.12A, so the
circuit can easily deliver the full-rated 5A using the 30mV
nominal current-limit threshold voltage.
Output Capacitor Selection
Find a low-loss inductor having the lowest possible DC
resistance that fits in the allotted dimensions. Ferrite cores
are often the best choice. The core must be large enough
not to saturate at the peak inductor current (IPEAK) shown in
Equation 10:
IPEAK = I LOAD ( MAX ) + [ ( LIR ⁄ 2 ) ⋅ I LOAD ( MAX ) ]
Examining the 5A circuit example with a maximum
rDS(ON) = 5mΩ at room temperature. At +125°C reveals the
following:
(EQ. 12)
where: ILIMIT(LOW) = minimum current-limit threshold
voltage divided by the rDS(ON) of Q2/Q4.
Use the worst-case maximum value for rDS(ON) from the
MOSFET Q2/Q4 data sheet and add some margin for the
rise in rDS(ON) with temperature. A good general rule is to
allow 0.2% additional resistance for each °C of temperature
rise.
The output filter capacitor must have low enough equivalent
series resistance (ESR) to meet output ripple and
load-transient requirements, yet have high enough ESR to
satisfy stability requirements. The output capacitance must
also be high enough to absorb the inductor energy while
transitioning from full-load to no-load conditions without
tripping the overvoltage fault latch. In applications where the
output is subject to large load transients, the output
capacitor's size depends on how much ESR is needed to
prevent the output from dipping too low under a load
transient. Ignoring the sag due to finite capacitance:
V DIP
R SER ≤ ---------------------------------I LOAD ( MAX )
(EQ. 15)
where VDIP is the maximum-tolerable transient voltage drop.
In non-CPU applications, the output capacitor's size
depends on how much ESR is needed to maintain an
acceptable level of output voltage ripple:
VP – P
R ESR ≤ ----------------------------------------------L IR ⋅ I LOAD ( MAX )
(EQ. 16)
where VP-P is the peak-to-peak output voltage ripple. The
actual capacitance value required relates to the physical size
needed to achieve low ESR, as well as to the chemistry of
the capacitor technology. Thus, the capacitor is usually
selected by ESR and voltage rating rather than by
capacitance value (this is true of tantalum, OS-CON, and
other electrolytic-type capacitors).
When using low-capacity filter capacitors such as polymer
types, capacitor size is usually determined by the capacity
required to prevent VSAG and VSOAR from tripping the
undervoltage and overvoltage fault latches during load
transients in ultrasonic mode.
For low input-to-output voltage differentials (VIN/ VOUT < 2),
additional output capacitance is required to maintain stability
and good efficiency in ultrasonic mode. The amount of
overshoot due to stored inductor energy can be calculated
as shown in Equation 17:
2
I PEAK ⋅ L
V SOAR = -----------------------------------------------2 ⋅ C OUT ⋅ V OUT_
(EQ. 17)
where IPEAK is the peak inductor current.
33
FN6453.3
March 18, 2008
ISL6236A
Input Capacitor Selection
The input capacitors must meet the input-ripple-current
(IRMS) requirement imposed by the switching current. The
ISL6236A dual switching regulator operates at different
frequencies. This interleaves the current pulses drawn by
the two switches and reduces the overlap time where they
add together. The input RMS current is much smaller in
comparison than with both SMPSs operating in phase. The
input RMS current varies with load and the input voltage.
The maximum input capacitor RMS current for a single
SMPS is given by Equation 18:
⎛ V OUT ( V IN – V OUT_ )⎞
I RMS ≈ I LOAD ⎜ ------------------------------------------------------------⎟
V IN
⎝
⎠
(EQ. 18)
When V IN = 2 ⋅ V OUT_ ( D = 50% ) , IRMS has maximum
current of I LOAD ⁄ 2 .
The ESR of the input-capacitor is important for determining
capacitor power dissipation. All the power (IRMS2 x ESR)
heats up the capacitor and reduces efficiency. Nontantalum
chemistries (ceramic or OS-CON) are preferred due to their
low ESR and resilience to power-up surge currents. Choose
input capacitors that exhibit less than +10°C temperature
rise at the RMS input current for optimal circuit longevity.
Place the drains of the high-side switches close to each
other to share common input bypass capacitors.
Power MOSFET Selection
Most of the following MOSFET guidelines focus on the
challenge of obtaining high load-current capability (>5A)
when using high-voltage (>20V) AC adapters. Low-current
applications usually require less attention.
Choose a high-side MOSFET (Q1/Q3) that has conduction
losses equal to the switching losses at the typical battery
voltage for maximum efficiency. Ensure that the conduction
losses at the minimum input voltage do not exceed the
package thermal limits or violate the overall thermal budget.
Ensure that conduction losses plus switching losses at the
maximum input voltage do not exceed the package ratings
or violate the overall thermal budget.
Choose a synchronous rectifier (Q2/Q4) with the lowest
possible rDS(ON). Ensure the gate is not pulled up by the
high-side switch turning on due to parasitic drain-to-gate
capacitance, causing cross-conduction problems. Switching
losses are not an issue for the synchronous rectifier in the
buck topology since it is a zero-voltage switched device
when using the buck topology.
MOSFET Power Dissipation
Worst-case conduction losses occur at the duty-factor
extremes. For the high-side MOSFET, the worst-case power
dissipation (PD) due to the MOSFET's rDS(ON) occurs at the
minimum battery voltage, as shown in Equation 19:
34
⎛ V OUT_ ⎞
2
PD ( Q H Resistance ) = ⎜ ------------------------⎟ ( I LOAD ) ⋅ r DS ( ON )
⎝ V IN ( MIN )⎠
(EQ. 19)
Generally, a small high-side MOSFET reduces switching
losses at high input voltage. However, the rDS(ON) required
to stay within package power-dissipation limits often limits
how small the MOSFET can be. The optimum situation
occurs when the switching (AC) losses equal the conduction
(rDS(ON)) losses.
Switching losses in the high-side MOSFET can become an
insidious heat problem when maximum battery voltage is
applied, due to the squared term in the CV2f switching-loss
equation. Reconsider the high-side MOSFET chosen for
adequate rDS(ON) at low battery voltages if it becomes
extraordinarily hot when subjected to VIN(MAX).
Calculating the power dissipation in NH (Q1/Q3) due to
switching losses is difficult since it must allow for quantifying
factors that influence the turn-on and turn-off times. These
factors include the internal gate resistance, gate charge,
threshold voltage, source inductance, and PC board layout
characteristics. The following switching-loss calculation
provides only a very rough estimate and is no substitute for
bench evaluation, preferably including verification using a
thermocouple mounted on NH (Q1/Q3):
2 ⎛ C RSS ⋅ f SW ⋅ I LOAD⎞
PD ( Q H Switching ) = ( V IN ( MAX ) ) ⎜ -----------------------------------------------------⎟
I GATE
⎝
⎠
(EQ. 20)
where CRSS is the reverse transfer capacitance of QH
(Q1/Q3) and IGATE is the peak gate-drive source/sink
current.
For the synchronous rectifier, the worst-case power
dissipation always occurs at maximum battery voltage:
V OUT ⎞
⎛
2
PD ( Q L ) = ⎜ 1 – --------------------------⎟ I LOAD ⋅ r DS ( ON )
V
⎝
IN ( MAX )⎠
(EQ. 21)
The absolute worst case for MOSFET power dissipation
occurs under heavy overloads that are greater than
ILOAD(MAX) but are not quite high enough to exceed the
current limit and cause the fault latch to trip. To protect
against this possibility, "overdesign" the circuit to tolerate:
I LOAD = I LIMIT ( HIGH ) + ( ( LIR ) ⁄ 2 ) ⋅ I LOAD ( MAX )
(EQ. 22)
where ILIMIT(HIGH) is the maximum valley current allowed
by the current-limit circuit, including threshold tolerance and
resistance variation.
Rectifier Selection
Current circulates from ground to the junction of both
MOSFETs and the inductor when the high-side switch is off.
As a consequence, the polarity of the switching node is
negative with respect to ground. This voltage is
approximately -0.7V (a diode drop) at both transition edges
FN6453.3
March 18, 2008
ISL6236A
while both switches are off (dead time). The drop is
I L ⋅ r DS ( ON ) when the low-side switch conducts.
The rectifier is a clamp across the synchronous rectifier that
catches the negative inductor swing during the dead time
between turning the high-side MOSFET off and the
synchronous rectifier on. The MOSFETs incorporate a
high-speed silicon body diode as an adequate clamp diode if
efficiency is not of primary importance. Place a Schottky
diode in parallel with the body diode to reduce the forward
voltage drop and prevent the Q2/Q4 MOSFET body diodes
from turning on during the dead time. Typically, the external
diode improves the efficiency by 1% to 2%. Use a Schottky
diode with a DC current rating equal to one-third of the load
current. For example, use an MBR0530 (500mA-rated) type
for loads up to 1.5A, a 1N5817 type for loads up to 3A, or a
1N5821 type for loads up to 10A. The rectifier's rated
reverse breakdown voltage must be at least equal to the
maximum input voltage, preferably with a 20% derating
factor.
Applications Information
The output voltage-adjust range for continuous-conduction
operation is restricted by the nonadjustable 350ns (max)
minimum off-time one-shot. Use the slower 5V SMPS for the
higher of the two output voltages for best dropout
performance in adjustable feedback mode. The duty-factor
limit must be calculated using worst-case values for on- and
off-times, when working with low input voltages.
Manufacturing tolerances and internal propagation delays
introduce an error to the tON K-factor. Also, keep in mind that
transient-response performance of buck regulators operated
close to dropout is poor, and bulk output capacitance must
often be added (see Equation 11 on page 33).
The absolute point of dropout occurs when the inductor
current ramps down during the minimum off-time (ΔIDOWN)
as much as it ramps up during the on-time ( ΔIUP). The ratio
h = ΔIUP/ΔIDOWN indicates the ability to slew the inductor
current higher in response to increased load, and must
always be greater than 1. As h approaches 1, the absolute
minimum dropout point, the inductor current is less able to
increase during each switching cycle and VSAG greatly
increases unless additional output capacitance is used.
A reasonable minimum value for h is 1.5, but this can be
adjusted up or down to allow trade-offs between VSAG,
output capacitance and minimum operating voltage. For a
given value of h, the minimum operating voltage can be
calculated in Equation 23:
35
Operating frequency must be reduced or h must be
increased and output capacitance added to obtain an
acceptable VSAG if calculated VIN(MIN) is greater than the
required minimum input voltage. Calculate VSAG to be sure
of adequate transient response if operation near dropout is
anticipated.
Dropout Design Example:
ISL6236A: With VOUT2 = 5V, fsw = 400kHz, K = 2.25µs,
tOFF(MIN) = 350ns, VDROP1 = VDROP2 = 100mV, and
h = 1.5, the minimum VIN is:
( 5V + 0.1V )
V IN ( MIN ) = ---------------------------------------------- + 0.1V – 0.1V = 6.65V
0.35μs ⋅ 1.5
1 – ⎛ -------------------------------⎞
⎝ 2.25μs ⎠
(EQ. 24)
Calculating with h = 1 yields:
Dropout Performance
( V OUT_ + V DROP )
V IN ( MIN ) = --------------------------------------------------- + V DROP2 – V DROP1
t OFF ( MIN ) ⋅ h
1 – ⎛ ------------------------------------⎞
⎝
⎠
K
where VDROP1 and VDROP2 are the parasitic voltage drops
in the discharge and charge paths (see “ON-TIME ONESHOT (tON)” on page 20), tOFF(MIN) is from “Electrical
Specifications” table on page 6 and K is taken from Table 2.
The absolute minimum input voltage is calculated with h = 1.
(EQ. 23)
( 5V + 0.1V )
V IN ( MIN ) = ----------------------------------------- + 0.1V – 0.1V = 6.04V
0.35μs ⋅ 1
1 – ⎛ --------------------------⎞
⎝ 2.25μs ⎠
(EQ. 25)
Therefore, VIN must be greater than 6.65V. A practical input
voltage with reasonable output capacitance would be 7.5V.
PC Board Layout Guidelines
Careful PC board layout is critical to achieve minimal switching
losses and clean, stable operation. This is especially true when
multiple converters are on the same PC board where one
circuit can affect the other. Refer to the ISL6236 Evaluation Kit
Application Notes (AN1271 and AN1272) for a specific layout
example.
Mount all of the power components on the top side of the board
with their ground terminals flush against one another, if
possible. Follow these guidelines for good PC board layout:
• Isolate the power components on the top side from the
sensitive analog components on the bottom side with a
ground shield. Use a separate PGND plane under the
OUT1 and OUT2 sides (called PGND1 and PGND2). Avoid
the introduction of AC currents into the PGND1 and PGND2
ground planes. Run the power plane ground currents on the
top side only, if possible.
• Use a star ground connection on the power plane to
minimize the crosstalk between OUT1 and OUT2.
• Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation.
• Keep the power traces and load connections short. This
practice is essential for high efficiency. Using thick copper
PC boards (2oz vs 1oz) can enhance full-load efficiency
by 1% or more. Correctly routing PC board traces must be
FN6453.3
March 18, 2008
ISL6236A
approached in terms of fractions of centimeters, where a
single mΩ of excess trace resistance causes a
measurable efficiency penalty.
• PHASE (ISL6236A) and GND connections to the
synchronous rectifiers for current limiting must be made
using Kelvin-sense connections to guarantee the
current-limit accuracy with 8 Ld SO MOSFETs. This is best
done by routing power to the MOSFETs from outside using
the top copper layer, while connecting PHASE traces
inside (underneath) the MOSFETs.
• When trade-offs in trace lengths must be made, it is
preferable to allow the inductor charging path to be made
longer than the discharge path. For example, it is better to
allow some extra distance between the input capacitors
and the high-side MOSFET than to allow distance
between the inductor and the synchronous rectifier or
between the inductor and the output filter capacitor.
• Ensure that the OUT connection to COUT is short and
direct. However, in some cases it may be desirable to
deliberately introduce some trace length between the OUT
connector node and the output filter capacitor.
• Route high-speed switching nodes (BOOT, UGATE,
PHASE, and LGATE ) away from sensitive analog areas
(REF, ILIM, and FB ). Use PGND1 and PGND2 as an EMI
shield to keep radiated switching noise away from the IC's
feedback divider and analog bypass capacitors.
• Make all pin-strap control input connections (SKIP, ILIM,
etc.) to GND or VCC of the device.
Layout Procedure
Place the power components first with ground terminals
adjacent (Q2/Q4 source, CIN, COUT ). If possible, make all
these connections on the top layer with wide, copper-filled
areas.
Group the gate-drive components (BOOT capacitor, VIN
bypass capacitor) together near the controller device.
Make the DC/DC controller ground connections as follows:
1. Near the device, create a small analog ground plane.
2. Connect the small analog ground plane to GND and use
the plane for the ground connection for the REF and VCC
bypass capacitors, FB dividers and ILIM resistors (if any).
3. Create another small ground island for PGND and use
the plane for the VIN bypass capacitor, placed very close
to the device.
4. Connect the GND and PGND planes together at the
metal tab under device.
On the board's top side (power planes), make a star ground
to minimize crosstalk between the two sides. The top-side
star ground is a star connection of the input capacitors and
synchronous rectifiers. Keep the resistance low between the
star ground and the source of the synchronous rectifiers for
accurate current limit. Connect the top-side star ground
(used for MOSFET, input, and output capacitors) to the small
island with a single short, wide connection (preferably just a
via). Create PGND islands on the layer just below the
top-side layer (refer to ISL6236 Evaluation Kit Application
Notes, AN1271 and AN1272 for an example) to act as an EMI
shield if multiple layers are available (highly recommended).
Connect each of these individually to the star ground via,
which connects the top side to the PGND plane. Add one
more solid ground plane under the device to act as an
additional shield, and also connect the solid ground plane to
the star ground via.
Connect the output power planes (VCORE and system
ground planes) directly to the output filter capacitor positive
and negative terminals with multiple vias.
Mount the controller IC adjacent to the synchronous rectifier
MOSFETs close to the hottest spot, preferably on the back
side in order to keep UGATE, GND, and the LGATE gate
drive lines short and wide. The LGATE gate trace must be
short and wide, measuring 50 mils to 100 mils wide if the
MOSFET is 1” from the controller device.
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Intersil products are sold by description only. Intersil Corporation reserves the right to make changes in circuit design, software and/or specifications at any time without
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36
FN6453.3
March 18, 2008
ISL6236A
Package Outline Drawing
L32.5x5B
32 LEAD QUAD FLAT NO-LEAD PLASTIC PACKAGE
Rev 2, 11/07
4X 3.5
5.00
28X 0.50
A
B
6
PIN 1
INDEX AREA
6
PIN #1 INDEX AREA
32
25
1
5.00
24
3 .30 ± 0 . 15
17
(4X)
8
0.15
9
16
TOP VIEW
0.10 M C A B
+ 0.07
32X 0.40 ± 0.10
4 32X 0.23 - 0.05
BOTTOM VIEW
SEE DETAIL "X"
0.10 C
0 . 90 ± 0.1
C
BASE PLANE
SEATING PLANE
0.08 C
( 4. 80 TYP )
(
( 28X 0 . 5 )
SIDE VIEW
3. 30 )
(32X 0 . 23 )
C
0 . 2 REF
5
( 32X 0 . 60)
0 . 00 MIN.
0 . 05 MAX.
DETAIL "X"
TYPICAL RECOMMENDED LAND PATTERN
NOTES:
1. Dimensions are in millimeters.
Dimensions in ( ) for Reference Only.
2. Dimensioning and tolerancing conform to AMSE Y14.5m-1994.
3. Unless otherwise specified, tolerance : Decimal ± 0.05
4. Dimension b applies to the metallized terminal and is measured
between 0.15mm and 0.30mm from the terminal tip.
5. Tiebar shown (if present) is a non-functional feature.
6. The configuration of the pin #1 identifier is optional, but must be
located within the zone indicated. The pin #1 identifier may be
either a mold or mark feature.
37
FN6453.3
March 18, 2008