MAXIM MAX1999EEI

19-2187; Rev 1; 9/04
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
The MAX1777/MAX1977/MAX1999 dual step-down,
switch-mode power-supply (SMPS) controllers generate
logic-supply voltages in battery-powered systems. The
MAX1777/MAX1977/MAX1999 include two pulse-width
modulation (PWM) controllers, adjustable from 2V to 5.5V
or fixed at 5V and 3.3V. These devices feature two linear
regulators providing 5V and 3.3V always-on outputs.
Each linear regulator provides up to 100mA output current with automatic linear regulator bootstrapping to the
main SMPS outputs. The MAX1777/MAX1977/MAX1999
include on-board power-up sequencing, a power-good
(PGOOD) output, digital soft-start, and internal soft-stop
output discharge that prevents negative voltages on shutdown.
Maxim’s proprietary Quick-PWM™ quick-response, constant on-time PWM control scheme operates without
sense resistors and provides 100ns response to load
transients while maintaining a relatively constant switching frequency. The unique ultrasonic pulse-skipping
mode maintains the switching frequency above 25kHz,
which eliminates noise in audio applications. Other features include pulse skipping, which maximizes efficiency
in light-load applications, and fixed-frequency PWM
mode, which reduces RF interference in sensitive applications.
The MAX1777 features a 200kHz/5V and 300kHz/3.3V
SMPS for highest efficiency, while the MAX1977 features
a 400kHz/5V and 500kHz/3.3V SMPS for “thin and light”
applications. The MAX1999 provides a pin-selectable
switching frequency, allowing either 200kHz/300kHz or
400kHz/500kHz operation of the 5V/3.3V SMPSs, respectively. The MAX1777/MAX1977/MAX1999 are available in
28-pin QSOP packages and operate over the extended
temperature range (-40°C to +85°C). The MAX1777/
MAX1977/MAX1999 are available in lead-free packages.
Features
♦ No Current-Sense Resistor Needed (MAX1999)
♦ Accurate Current Sense with Current-Sense
Resistor (MAX1777/MAX1977)
♦ 1.5% Output Voltage Accuracy
♦ 3.3V and 5V 100mA Bootstrapped Linear
Regulators
♦ Internal Soft-Start and Soft-Stop Output
Discharge
♦ Quick-PWM with 100ns Load Step Response
♦ 3.3V and 5V Fixed or Adjustable Outputs
(Dual Mode™)
♦ 4.5V to 24V Input Voltage Range
♦ Ultrasonic Pulse-Skipping Mode (25kHz min)
♦ Power-Good (PGOOD) Signal
♦ Overvoltage Protection Enable/Disable
Ordering Information
PART
TEMP RANGE
MAX1777EEI
PINPACKAGE
-40°C to +85°C 28 QSOP
MAX1777EEI+ -40°C to +85°C 28 QSOP
200kHz/300kHz
-40°C to +85°C 28 QSOP
400kHz/500kHz
MAX1977EEI+ -40°C to +85°C 28 QSOP
400kHz/500kHz
MAX1977EEI
-40°C to +85°C 28 QSOP
200kHz/300kHz or
400kHz/500kHz
MAX1999EEI+ -40°C to +85°C 28 QSOP
200kHz/300kHz or
400kHz/500kHz
MAX1999EEI
+Denotes lead-free package.
Pin Configurations
TOP VIEW
N.C. 1
Applications
Notebook and Subnotebook Computers
PDAs and Mobile Communication Devices
3- and 4-Cell Li+ Battery-Powered Devices
Quick-PWM and Dual Mode are trademarks of Maxim
Integrated Products, Inc.
Pin Configurations continued at end of data sheet.
5V/3V
SWITCHING
FREQUENCY
200kHz/300kHz
28 BST3
PGOOD 2
27 LX3
ON3 3
26 DH3
ON5 4
25 LDO3
ILIM3 5
SHDN 6
24 DL3
MAX1999
23 GND
FB3 7
22 OUT3
REF 8
21 OUT5
FB5 9
20 V+
PRO 10
19 DL5
ILIM5 11
18 LDO5
SKIP 12
17 VCC
TON 13
16 DH5
BST5 14
15 LX5
QSOP
________________________________________________________________ Maxim Integrated Products
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
1
MAX1777/MAX1977/MAX1999
General Description
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
ABSOLUTE MAXIMUM RATINGS
V+, SHDN to GND ..................................................-0.3V to +25V
BST_ to GND ..........................................................-0.3V to +30V
LX_ to BST_ ..............................................................-6V to +0.3V
CS_ to GND (MAX1777/MAX1977 only)......................-2V to +6V
VCC, LDO5, LDO3, OUT3, OUT5, ON3, ON5, REF,
FB3, FB5, SKIP, PRO, PGOOD to GND ...............-0.3V to +6V
DH3 to LX3 ..............................................-0.3V to (VBST3 + 0.3V)
DH5 to LX5 ..............................................-0.3V to (VBST5 + 0.3V)
ILIM3, ILIM5 to GND...................................-0.3V to (VCC + 0.3V)
DL3, DL5 to GND....................................-0.3V to (VLDO5 + 0.3V)
TON to GND (MAX1999 only)...................................-0.3V to +6V
LDO3, LDO5, REF Short Circuit to GND ....................Momentary
LDO3 Current (Internal Regulator) Continuous ..............+100mA
LDO3 Current (Switched Over to OUT3) Continuous.....+200mA
LDO5 Current (Internal Regulator) Continuous ..............+100mA
LDO5 Current (Switched Over to OUT5) Continuous.....+200mA
Continuous Power Dissipation
28-Pin QSOP (derate 10.8mW/°C above +70°C).........860mW
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature ......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, V SHDN = 5V,
TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
MAIN SMPS CONTROLLERS
V+ Input Voltage Range
3.3V Output Voltage in
Fixed Mode
5V Output Voltage in Fixed Mode
LDO5 in regulation
V+ = LDO5, VOUT5 < 4.43V
V+ = 6V to 24V, FB3 = GND, V SKIP = 5V
V+ = 6V to 24V, FB5 = GND, V SKIP = 5V,
MAX1777/MAX1999 (TON = VCC)
6
24
4.5
5.5
V
3.285
3.330
3.375
V
4.975
5.050
5.125
V
1.975
2.00
2.025
V
V+ = 7V to 24V, FB5 = GND, V SKIP = 5V,
MAX1977/MAX1999 (TON = GND)
Output Voltage in
Adjustable Mode
V+ = 6V to 24V, either SMPS
Output Voltage Adjust Range
Either SMPS
2.0
5.5
V
FB3, FB5 Adjustable-Mode
Threshold Voltage
Dual-mode comparator
0.1
0.2
V
DC Load Regulation
Either SMPS, V SKIP = 5V, 0 to 5A
-0.1
Either SMPS, SKIP = GND, 0 to 5A
-1.5
Either SMPS, V SKIP = 2V, 0 to 5A
-1.7
Line Regulation
Either SMPS, 6V < V+ < 24V
Current-Limit Threshold
(Positive, Default)
ILIM_ = VCC, GND - CS_ (MAX1777/MAX1977),
GND - LX_ (MAX1999)
Current-Limit Threshold
(Positive, Adjustable)
GND - CS_
(MAX1777/MAX1977),
GND - LX_ (MAX1999)
0.005
%/V
93
100
107
40
50
60
VILIM_ = 1V
93
100
107
VILIM_ = 2V
185
200
215
VILIM_ = 0.5V
Zero-Current Threshold
SKIP = GND, ILIM_ = VCC, GND - CS_
(MAX1777/MAX1977), GND - LX_ (MAX1999)
Current-Limit Threshold
(Negative, Default)
SKIP = ILIM_ = VCC, GND - CS_ (MAX1777/MAX1977),
GND - LX_ (MAX1999)
Soft-Start Ramp Time
Zero to full limit
2
%
mV
mV
3
mV
-120
mV
1.7
ms
_______________________________________________________________________________________
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, V SHDN = 5V,
TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER
Operating Frequency
CONDITIONS
5V SMPS
200
3.3V SMPS
300
MAX1977 or MAX1999
(VTON = 0), SKIP = VCC
5V SMPS
400
3.3V SMPS
500
MAX
25
36
VOUT5 = 5.05V
1.895
2.105
2.315
VOUT3 = 3.33V
0.833
0.925
1.017
MAX1977 or MAX1999
(VTON = 0)
VOUT5 = 5.05V
0.895
1.052
1.209
VOUT3 = 3.33V
0.475
0.555
0.635
250
300
350
MAX1777 or MAX1999
(VTON = 5V)
VOUT5 = 5.05V
94
VOUT3 = 3.33V
91
MAX1977 or MAX1999
(VTON = 0)
VOUT5 = 5.05V
88
VOUT3 = 3.33V
85
INTERNAL REGULATOR AND REFERENCE
LDO5 Output Voltage
ON3 = ON5 = GND, 6V < V+ < 24V, 0 < ILDO5 < 100mA
4.90
5.00
UNITS
kHz
MAX1777 or MAX1999
(VTON = 5V)
Minimum Off-Time
Maximum Duty Cycle
TYP
MAX1777 or MAX1999
(VTON = 5V), SKIP = VCC
SKIP = REF
On-Time Pulse Width
MIN
µs
ns
%
5.10
LDO5 = GND
LDO5 Undervoltage Lockout
Fault Threshold
Falling edge of LDO5, hysteresis = 1%
3.7
4.0
4.3
V
LDO5 Bootstrap Switch Threshold
Falling edge of OUT5, rising edge at OUT5 regulation
point
4.43
4.56
4.69
V
LDO5 Bootstrap
Switch Resistance
LDO5 to OUT5, VOUT5 = 5V
1.4
3.2
Ω
3.35
3.42
LDO3 Output Voltage
ON3 = ON5 = GND, 6V < V+ < 24V, 0 < ILDO3 < 100mA
LDO3 Short-Circuit Current
LDO3 = GND
LDO3 Bootstrap Switch Threshold
Falling edge of OUT3, rising edge at OUT3 regulation
point
LDO3 Bootstrap Switch
Resistance
LDO3 to OUT3, VOUT3 = 3.2V
REF Output Voltage
No external load
REF Load Regulation
0 < ILOAD < 50µA
REF Sink Current
REF in regulation
190
V
LDO5 Short-Circuit Current
3.28
mA
180
2.80
1.980
V
mA
2.91
3.02
V
1.5
3.5
Ω
2.000
2.020
V
10
mV
10
µA
V+ Operating Supply Current
LDO5 switched over to OUT5, 5V SMPS
25
50
µA
V+ Standby Supply Current
V+ = 6V to 24V, both SMPSs off, includes ISHDN
150
250
µA
V+ Shutdown Supply Current
V+ = 4.5V to 24V
6
15
µA
Quiescent Power Consumption
Both SMPSs on, FB3 = FB5 = SKIP = GND, VOUT3 =
3.5V, VOUT5 = 5.3V
3
4.5
mW
+11
+14
%
FAULT DETECTION
Overvoltage Trip Threshold
FB3 or FB5 with respect to nominal regulation point
+8
_______________________________________________________________________________________
3
MAX1777/MAX1977/MAX1999
ELECTRICAL CHARACTERISTICS (continued)
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, V SHDN = 5V,
TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER
CONDITIONS
Overvoltage Fault
Propagation Delay
FB3 or FB5 delay with 50mV overdrive
PGOOD Threshold
FB3 or FB5 with respect to nominal output, falling edge,
typical hysteresis = 1%
PGOOD Propagation Delay
Falling edge, 50mV overdrive
PGOOD Output Low Voltage
ISINK = 4mA
PGOOD Leakage Current
High state, forced to 5.5V
MIN
TYP
MAX
10
-12
-9.5
µs
-7
10
Thermal Shutdown Threshold
UNITS
%
µs
0.3
V
1
µA
o
160
C
Output Undervoltage
Shutdown Threshold
FB3 or FB5 with respect to nominal output voltage
65
70
75
%
Output Undervoltage
Shutdown Blanking Time
From ON_ signal
10
22
35
ms
-200
+40
+200
nA
INPUTS AND OUTPUTS
Feedback Input Leakage Current
PRO Input Voltage
VFB3 = VFB5 = 2.2V
Low level
High level
0.6
1.5
Low level
SKIP Input Voltage
TON Input Voltage
0.8
Float level
1.7
High level
2.4
2.3
Low level
High level
0.8
2.4
Clear fault level/SMPS off level
ON3, ON5 Input Voltage
Input Leakage Current
SHDN Input Trip Level
DH_ Gate-Driver
Sink/Source Current
V
V
0.8
Delay start level
1.7
2.3
SMPS on level
2.4
V PRO or VTON = 0 or 5V
-1
+1
VON_ = 0 or 5V
-2
+2
V SKIP = 0 or 5V
-1
+1
V SHDN = 0 or 24V
-1
+1
VCS_ = 0 or 5V
-2
+2
VILIM3, VILIM5 = 0 or 2V
-0.2
Rising edge
1.2
1.6
2.0
Falling edge
0.96
1.00
1.04
DH3, DH5 forced to 2V
V
V
µA
+0.2
2
V
A
DL_ Gate-Driver Source Current
DL3 (source), DL5 (source), forced to 2V
1.7
A
DL_ Gate-Driver Sink Current
DL3 (sink), DL5 (sink), forced to 2V
3.3
A
DH_ Gate-Driver On-Resistance
DL_ Gate-Driver On-Resistance
4
BST - LX_ forced to 5V
1.5
4.0
DL_, high state (pullup)
2.2
5.0
DL_, low state (pulldown)
0.6
1.5
_______________________________________________________________________________________
Ω
Ω
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, V SHDN = 5V,
TA = 0°C to +85°C, unless otherwise noted. Typical values are at TA = +25°C.)
PARAMETER
CONDITIONS
MIN
OUT3, OUT5 Discharge-Mode
On-Resistance
OUT3, OUT5 Discharge-Mode
Synchronous Rectifier
Turn-On Level
0.2
TYP
MAX
UNITS
12
40
Ω
0.3
0.4
V
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = V CC, V SHDN = 5V,
TA = -40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
MAIN SMPS CONTROLLERS
V+ Input Voltage Range
3.3V Output Voltage in
Fixed Mode
5V Output Voltage in Fixed Mode
LDO5 in regulation
6
24
V+ = LDO5, VOUT5 < 4.41V
4.5
5.5
V+ = 6V to 24V, FB3 = GND, V SKIP = 5V
3.27
3.39
V
4.95
5.15
V
V+ = 6V to 24V, FB5 = GND, V SKIP = 5V,
MAX1777/MAX1999 (TON = VCC)
V
V+ = 7V to 24V, FB5 = GND, V SKIP = 5V,
MAX1977/MAX1999 (TON = GND)
Output Voltage in
Adjustable Mode
V+ = 6V to 24V, either SMPS
1.97
2.03
V
Output Voltage Adjust Range
Either SMPS
2.0
5.5
V
FB3, FB5 Adjustable-Mode
Threshold Voltage
Dual-mode comparator
0.1
0.2
V
Current-Limit Threshold
(Positive, Default)
ILIM_ = VCC, GND - CS_ (MAX1777/MAX1977),
GND - LX_ (MAX1999)
90
110
mV
GND - CS_
(MAX1777/MAX1977),
GND - LX_ (MAX1999)
40
60
Current-Limit Threshold
(Positive, Adjustable)
VILIM_ = 1V
90
110
VILIM_ = 2V
180
220
On-Time Pulse Width
VILIM_ = 0.5V
mV
MAX1777 or MAX1999
(VTON = 5V)
VOUT5 = 5.05V
1.895
2.315
VOUT3 = 3.33V
0.833
1.017
MAX1977 or MAX1999
(VTON = 0)
VOUT5 = 5.05V
0.895
1.209
VOUT3 = 3.33V
0.475
0.635
200
400
ns
Minimum Off-Time
µs
INTERNAL REGULATOR AND REFERENCE
LDO5 Output Voltage
ON3 = ON5 = GND, 6V < V+ < 24V, 0 < ILDO5 < 100mA
4.90
5.10
V
LDO5 Undervoltage
Lockout Fault Threshold
Falling edge of LDO5, hysteresis = 1%
3.7
4.3
V
_______________________________________________________________________________________
5
MAX1777/MAX1977/MAX1999
ELECTRICAL CHARACTERISTICS (continued)
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12.0.V, ON3 = ON5 = VCC, V SHDN = 5V,
TA = -40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER
CONDITIONS
MIN
TYP
MAX
UNITS
4.69
V
3.2
Ω
LDO5 Bootstrap Switch Threshold
Falling edge of OUT5, rising edge at OUT5 regulation
point
LDO5 Bootstrap
Switch Resistance
LDO5 to OUT5, VOUT5 = 5V
LDO3 Output Voltage
ON3 = ON5 = GND, 6V < V+ < 24V, 0 < ILDO3 < 100mA
3.27
3.43
V
LDO3 Bootstrap Switch Threshold
Falling edge of OUT3, rising edge at OUT3 regulation
point
2.80
3.02
V
LDO3 Bootstrap
Switch Resistance
LDO3 to OUT3, VOUT3 = 3.2V
3.5
Ω
REF Output Voltage
No external load
REF Load Regulation
0 < ILOAD < 50µA
4.43
1.975
2.025
V
10
mV
REF Sink Current
REF in regulation
V+ Operating Supply Current
LDO5 switched over to OUT5, 5V SMPS
10
50
µA
µA
V+ Standby Supply Current
V+ = 6V to 24V, both SMPSs off, includes ISHDN
300
µA
V+ Shutdown Supply Current
V+ = 4.5V to 24V
15
µA
Quiescent Power Consumption
Both SMPSs on, FB3 = FB5 = SKIP = GND, VOUT3 =
3.5V, VOUT5 = 5.3V
4.5
mW
FAULT DETECTION
Overvoltage Trip Threshold
FB3 or FB5 with respect to nominal regulation point
+8
+14
%
PGOOD Threshold
FB3 or FB5 with respect to nominal output, falling edge,
typical hysteresis = 1%
-12
-7
%
PGOOD Output Low Voltage
ISINK = 4mA
PGOOD Leakage Current
High state, forced to 5.5V
Output Undervoltage
Shutdown Threshold
FB3 or FB5 with respect to nominal output voltage
Output Undervoltage
Shutdown Blanking Time
From ON_ signal
0.3
V
1
µA
65
75
%
10
40
ms
-200
+200
nA
INPUTS AND OUTPUTS
Feedback Input Leakage Current
PRO Input Voltage
VFB3 = VFB5 = 2.2V
Low level
High level
0.6
1.5
Low level
SKIP Input Voltage
TON Input Voltage
6
0.8
Float level
1.7
High level
2.4
Low level
High level
V
2.3
0.8
2.4
_______________________________________________________________________________________
V
V
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12.0.V, ON3 = ON5 = VCC, V SHDN = 5V,
TA = -40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER
CONDITIONS
MIN
Clear fault level/SMPS off level
ON3, ON5 Input Voltage
Input Leakage Current
DH_ Gate-Driver On-Resistance
DL_ Gate-Driver On-Resistance
MAX
1.7
SMPS on level
2.4
V PRO or VTON = 0 or 5V
-1
+1
VON_ = 0 or 5V
-1
+1
V SKIP = 0 or 5V
-2
+2
V SHDN = 0 or 24V
-1
+1
2.3
-2
+2
VILIM3, VILIM5 = 0 or 2V
-0.2
+0.2
Rising edge
1.2
2.0
Falling edge
0.96
1.04
BST - LX_ forced to 5V
4.0
DL_, high state (pullup)
5.0
DL_, low state (pulldown)
1.5
OUT3, OUT5 Discharge-Mode
On-Resistance
OUT3, OUT5 Discharge-Mode
Synchronous Rectifier
Turn-On Level
UNITS
0.8
Delay start level
VCS_ = 0 or 5V
SHDN Input Trip Level
TYP
0.2
V
µA
V
Ω
Ω
40
Ω
0.4
V
Note 1: Specifications to -40°C are guaranteed by design, not production tested.
_______________________________________________________________________________________
7
MAX1777/MAX1977/MAX1999
ELECTRICAL CHARACTERISTICS (continued)
Typical Operating Characteristics
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, SHDN = V+,
RCS = 7mΩ, VILIM_ = 0.5V, TA = +25°C, unless otherwise noted.)
5V OUTPUT EFFICIENCY
vs. LOAD CURRENT (MAX1977)
E
B
F
D
50
40
A: IDLE MODE, VIN = 7V
B: IDLE MODE, VIN = 12V
C: IDLE MODE, VIN = 24V
D: PWM MODE, VIN = 7V
E: PWM MODE, VIN = 12V
F: PWM MODE, VIN = 24V
30
20
10
E
B
60
F
D
50
40
A: IDLE MODE, VIN = 7V
B: IDLE MODE, VIN = 12V
C: IDLE MODE, VIN = 24V
D: PWM MODE, VIN = 7V
E: PWM MODE, VIN = 12V
F: PWM MODE, VIN = 24V
20
10
0.01
0.1
1
20
0.01
0.1
1
10
0.01
0.1
1
10
20
10
F
0
0.001
0.01
1
34
MAX1977
32
30
28
MAX1777
26
0.37
0.34
0.31
0.28
0.22
MAX1777
0.19
0.16
22
0.13
20
MAX1977
0.25
24
0.10
7
10
MAX1777 toc06
36
BATTERY CURRENT (mA)
A: IDLE MODE, VIN = 7V
B: IDLE MODE, VIN = 12V
C: IDLE MODE, VIN = 24V
D: PWM MODE, VIN = 7V
E: PWM MODE, VIN = 12V
F: PWM MODE, VIN = 24V
0.1
38
BATTERY CURRENT (mA)
D
B
0.40
MAX1777 toc05
40
MAX1777 toc04
E
10
13
16
19
22
25
7
10
13
16
19
22
25
LOAD CURRENT (A)
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
STANDBY INPUT CURRENT
vs. INPUT VOLTAGE
SHUTDOWN INPUT CURRENT
vs. INPUT VOLTAGE
5V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT (MAX1777)
186
184
MAX1777
180
178
MAX1977
174
10.0
9.5
9.0
8.5
MAX1777
8.0
MAX1977
7.5
7.0
6.5
6.0
10
13
16
19
INPUT VOLTAGE (V)
22
25
200
PWM MODE
175
150
125
100
75
50
IDLE MODE
0
5.0
170
VIN = 7V
225
25
5.5
172
250
SWITCHING FREQUENCY (kHz)
188
SHUTDOWN INPUT CURRENT (µA)
MAX1777 toc07
190
7
0.001
IDLE-MODE NO-LOAD BATTERY CURRENT
vs. INPUT VOLTAGE
50
176
F
10
A: IDLE MODE, VIN = 7V
B: IDLE MODE, VIN = 12V
C: IDLE MODE, VIN = 24V
D: PWM MODE, VIN = 7V
E: PWM MODE, VIN = 12V
F: PWM MODE, VIN = 24V
PWM NO-LOAD BATTERY CURRENT
vs. INPUT VOLTAGE
C
182
B
3.3V OUTPUT EFFICIENCY
vs. LOAD CURRENT (MAX1977)
60
30
40
LOAD CURRENT (A)
A
40
D
LOAD CURRENT (A)
90
70
E
50
LOAD CURRENT (A)
100
80
C
60
0
0.001
10
A
70
30
0
0.001
EFFICIENCY (%)
80
C
70
30
0
8
90
MAX1777 toc09
60
A
80
MAX1777 toc08
EFFICIENCY (%)
70
90
EFFICIENCY (%)
C
100
MAX1777 toc02
A
80
EFFICIENCY (%)
90
100
MAX1777 toc01
100
3.3V OUTPUT EFFICIENCY
vs. LOAD CURRENT (MAX1777)
MAX1777 toc03
5V OUTPUT EFFICIENCY
vs. LOAD CURRENT (MAX1777)
STANDBY INPUT CURRENT (µA)
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
7
10
13
16
19
INPUT VOLTAGE (V)
22
25
0.001
0.01
0.1
LOAD CURRENT (A)
_______________________________________________________________________________________
1
10
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
240
200
160
ULTRASONIC MODE
120
80
175
150
125
100
75
50
0
0.01
0.1
1
10
PWM MODE
280
240
200
160
120
80
ULTRASONIC MODE
PFM MODE
PFM MODE
0
0.001
0.01
0.1
1
10
0.001
0.01
0.1
1
10
LOAD CURRENT (A)
LOAD CURRENT (A)
5V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT (MAX1977)
3.3V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT (MAX1977)
5V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT (MAX1977)
250
200
150
100
500
450
400
350
300
ULTRASONIC MODE
250
200
150
500
100
0.001
0.01
0.1
1
0
0.001
10
PFM MODE
0.01
LOAD CURRENT (A)
VIN = 24V
550
PWM MODE
350
300
250
200
ULTRASONIC MODE
0
0.001
5.14
5.13
1
10
IDLE MODE
0.1
LOAD CURRENT (A)
200
ULTRASONIC MODE
150
100
PFM MODE
0.001
0.01
1
1
10
OUT3 VOLTAGE REGULATION
vs. LOAD CURRENT
IDLE MODE
ULTRASONIC MODE
5.11
5.10
5.09
5.08
3.390
3.385
ULTRASONIC
3.380
3.375
IDLE MODE
3.370
3.365
3.360
3.355
3.350
FORCED PWM
3.345
3.340
FORCED PWM
3.335
5.05
10
0.1
LOAD CURRENT (A)
5.12
5.06
0.01
0.1
5.07
100
50
250
0
5.15
OUTPUT VOLTAGE (V)
500
150
300
OUT5 VOLTAGE REGULATION
vs. LOAD CURRENT
MAX1777 toc16
600
400
PWM MODE
350
LOAD CURRENT (A)
3.3V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT (MAX1977)
450
400
50
50
IDLE MODE
0
OUTPUT VOLTAGE (V)
50
VIN = 24V
450
MAX1777 toc18
300
VIN = 7V
PWM MODE
SWITCHING FREQUENCY (kHz)
350
MAX1777 toc14
550
SWITCHING FREQUENCY (kHz)
PWM MODE
400
600
MAX1777 toc13
VIN = 7V
450
MAX1777 toc15
LOAD CURRENT (A)
500
SWITCHING FREQUENCY (kHz)
320
40
0
0.001
SWITCHING FREQUENCY (kHz)
ULTRASONIC MODE
25
VIN = 24V
360
MAX1777 toc12
200
PFM MODE
40
PWM MODE
400
SWITCHING FREQUENCY (kHz)
PWM MODE
280
VIN = 24V
225
3.3V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT (MAX1777)
MAX1777 toc17
SWITCHING FREQUENCY (kHz)
320
250
MAX1777 toc11
VIN = 7V
360
5V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT (MAX1777)
SWITCHING FREQUENCY (kHz)
400
MAX1777 toc10
3.3V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT (MAX1777)
0.001
0.01
0.1
LOAD CURRENT (A)
1
10
3.330
0.001
0.01
0.1
1
10
LOAD CURRENT (A)
_______________________________________________________________________________________
9
MAX1777/MAX1977/MAX1999
Typical Operating Characteristics (continued)
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, SHDN = V+,
RCS = 7mΩ, VILIM_ = 0.5V, TA = +25°C, unless otherwise noted.)
Typical Operating Characteristics (continued)
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, SHDN = V+,
RCS = 7mΩ, VILIM_ = 0.5V, TA = +25°C, unless otherwise noted.)
4.97
4.96
4.95
3.348
3.346
2.003
3.344
2.002
3.342
2.001
3.340
2.000
3.338
1.999
3.336
1.998
3.334
1.997
3.332
1.996
1.995
3.330
0
10 20 30 40 50 60 70 80 90 100
0
-10 0 10 20 30 40 50 60 70 80 90 100
10 20 30 40 50 60 70 80 90 100
LDO5 OUTPUT CURRENT (mA)
LDO3 OUTPUT CURRENT (mA)
IREF (µA)
REF, LDO3, AND LDO5 POWER-UP
DELAYED START WAVEFORMS
(ON3 = REF)
DELAYED START WAVEFORMS
(ON5 = REF)
MAX1777 toc23
MAX1777 toc22
MAX1777 toc24
5V
12V
5V
ON5
2V/div
V+
5V/div
0
3.3V
0
3.3V
LDO3
2V/div
0
5V
LDO5
5V/div
REF
2V/div
0
2V
0
OUT3
2V/div
0
5V
0
5V
0
0
200µs/div
SHUTDOWN WAVEFORMS
MAX1777/MAX1999 (TON = VCC)
5V PWM-MODE
LOAD TRANSIENT RESPONSE
MAX1777 toc26
IL5
5A/div
IL3
5A/div
5A
0
3.3V
OUT3
5V/div
0
5V
OUT5
5V/div
0
200µs/div
OUT5
2V/div
200µs/div
MAX1777 toc25
0
OUT3
2V/div
OUT5
2V/div
SOFT-START WAVEFORMS
5A
ON3
2V/div
0
3.3V
4ms/div
10
2.004
VREF (V)
4.98
2.005
MAX1777 toc20
4.99
3.350
LDO3 OUTPUT VOLTAGE (V)
MAX1777 toc19
5.00
REFERENCE VOLTAGE
vs. OUTPUT CURRENT
LDO3 REGULATOR OUTPUT VOLTAGE
vs. OUTPUT CURRENT
MAX1777 toc21
LDO5 REGULATOR OUTPUT VOLTAGE
vs. OUTPUT CURRENT
LDO5 OUTPUT VOLTAGE (V)
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
5V
MAX1777 toc27
ON3
5V/div
0
3.3V
0
OUT3
5V/div
5V
OUT5
5V/div
0
5V
SWITCHING
VOUT,
ACCOUPLED
100mV/div
5V
4A
INDUCTOR
CURRENT
2A/div
1A
5V
DL3
5V/div
0
10ms/div
DL5
5V/div
0
20µs/div
______________________________________________________________________________________
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, SHDN = V+,
RCS = 7mΩ, VILIM_ = 0.5V, TA = +25°C, unless otherwise noted.)
MAX1977/MAX1999 (TON = GND)
3.3V PWM-MODE
LOAD TRANSIENT RESPONSE
MAX1777/MAX1999 (TON = VCC)
3.3V PWM-MODE
LOAD TRANSIENT RESPONSE
MAX1977/MAX1999 (TON = GND)
5V PWM-MODE
LOAD TRANSIENT RESPONSE
MAX1777 toc28
MAX1777 toc29
VOUT,
ACCOUPLED
100mV/div
5V
4A
1A
VOUT,
ACCOUPLED
100mV/div
3.3V
INDUCTOR
CURRENT
2A/div
5V
MAX1777 toc30
4A
INDUCTOR
CURRENT
2A/div
1A
5V
DL5
5V/div
0
0
5V ULTRASONIC EFFICIENCY
vs. LOAD CURRENT (TON = GND)
VIN = 24V
50
40
80
VIN = 24V
70
60
50
40
30
30
20
20
10
10
0.01
0.1
1
0
0.001
10
0.01
0.1
1
LOAD CURRENT (A)
LOAD CURRENT (A)
3.3V ULTRASONIC EFFICIENCY
vs. LOAD CURRENT (TON = VCC)
3.3V ULTRASONIC EFFICIENCY
vs. LOAD CURRENT (TON = GND)
90
80
80
EFFICIENCY (%)
VIN = 7V
60
VIN = 12V
50
40
VIN = 24V
30
70
50
30
20
10
LOAD CURRENT (A)
VIN = 24V
40
10
0.1
VIN = 12V
60
20
0.01
VIN = 7V
90
10
MAX1777 toc34
100
MAX1777 toc33
100
EFFICIENCY (%)
VIN = 12V
90
EFFICIENCY (%)
EFFICIENCY (%)
100
VIN = 12V
60
0
0.001
DL3
5V/div
0
MAX1777 toc32
90
70
1A
10µs/div
MAX1777 toc31
100
0
0.001
INDUCTOR
CURRENT
2A/div
20µs/div
5V ULTRASONIC EFFICIENCY
vs. LOAD CURRENT (TON = VCC)
70
4A
5V
DL3
5V/div
10µs/div
80
VOUT,
ACCOUPLED
100mV/div
3.3V
1
10
0
0.001
0.01
0.1
1
10
LOAD CURRENT (A)
______________________________________________________________________________________
11
MAX1777/MAX1977/MAX1999
Typical Operating Characteristics (continued)
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
Pin Description
PIN
MAX1777
MAX1999
MAX1977
12
NAME
1
—
CS3
—
1
N.C.
2
2
PGOOD
3
3
ON3
4
4
ON5
5
5
ILIM3
6
6
SHDN
7
7
FB3
8
8
REF
9
9
FB5
10
10
PRO
11
11
ILIM5
12
12
SKIP
FUNCTION
3.3V SMPS Current-Sense Input. Connect CS3 to a current-sensing resistor from the source of
the synchronous rectifier to GND. The voltage at ILIM3 determines the current-limit threshold
(see the Current-Limit (ILIM) Circuit section).
No Connection. Not internally connected.
Power-Good Output. PGOOD is an open-drain output that is pulled low if either output is
disabled or is more than 10% below its nominal value.
3.3V SMPS Enable Input. The 3.3V SMPS is enabled if ON3 is greater than the SMPS on level
and disabled if ON3 is less than the SMPS off level. If ON3 is connected to REF, the 3.3V
SMPS starts after the 5V SMPS reaches regulation (delay start). Drive ON3 below the clear
fault level to reset the fault latches.
5V SMPS Enable Input. The 5V SMPS is enabled if ON5 is greater than the SMPS on level and
disabled if ON5 is less than the SMPS off level. If ON5 is connected to REF, the 5V SMPS
starts after the 3.3V SMPS reaches regulation (delay start). Drive ON5 below the clear fault
level to reset the fault latches.
3.3V SMPS Current-Limit Adjustment. The GND-LX current-limit threshold defaults to 100mV if
ILIM3 is tied to VCC. In adjustable mode, the current-limit threshold is 1/10th the voltage seen
at ILIM3 over a 0.5V to 3V range. The logic threshold for switchover to the 100mV default value
is approximately VCC - 1V. Connect ILIM3 to REF for a fixed 200mV threshold.
Shutdown Control Input. The device enters its 6µA supply current shutdown mode if
V SHDN is less than the SHDN input falling edge trip level and does not restart until V SHDN is
greater than the SHDN input rising edge trip level. Connect SHDN to V+ for automatic startup.
SHDN can be connected to V+ through a resistive voltage-divider to implement a
programmable undervoltage lockout.
3.3V SMPS Feedback Input. Connect FB3 to GND for fixed 3.3V operation. Connect FB3 to a
resistive voltage-divider from OUT3 to GND to adjust the output from 2V to 5.5V.
2V Reference Output. Bypass to GND with a 0.22µF (min) capacitor. REF can source up to
100µA for external loads. Loading REF degrades FB_ and output accuracy according to the
REF load-regulation error.
5V SMPS Feedback Input. Connect FB5 to GND for fixed 5V operation. Connect FB5 to a
resistive voltage-divider from OUT5 to GND to adjust the output from 2V to 5.5V.
Overvoltage and Undervoltage Fault Protection Enable/Disable. Connect PRO to VCC to
disable undervoltage and overvoltage protection. Connect PRO to GND to enable
undervoltage and overvoltage protection (see the Fault Protection section).
5V SMPS Current-Limit Adjustment. The GND-LX current-limit threshold defaults to 100mV if
ILIM5 is tied to VCC. In adjustable mode, the current-limit threshold is 1/10th the voltage seen
at ILIM5 over a 0.5V to 3V range. The logic threshold for switchover to the 100mV default value
is approximately VCC - 1V. Connect ILIM5 to REF for a fixed 200mV threshold.
Low-Noise Mode Control. Connect SKIP to GND for normal idle-mode (pulse-skipping)
operation or to VCC for PWM mode (fixed frequency). Connect to REF or leave floating for
ultrasonic mode (pulse skipping, 25kHz minimum).
______________________________________________________________________________________
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
PIN
MAX1777
MAX1999
MAX1977
NAME
13
—
CS5
—
13
TON
14
14
BST5
15
15
LX5
16
16
DH5
17
17
VCC
18
18
LDO5
19
19
DL5
20
20
V+
21
21
OUT5
22
22
OUT3
23
24
23
24
GND
DL3
25
25
LDO3
26
26
DH3
27
27
LX3
28
28
BST3
FUNCTION
5V SMPS Current-Sense Input. Connect CS5 to a current-sensing resistor from the source of
the synchronous rectifier to GND. The voltage at ILIM5 determines the current-limit threshold
(see the Current-Limit Circuit section).
Frequency Select Input. Connect to VCC for 200kHz/300kHz operation and to GND for
400kHz/500kHz operation (5V/3.3V SMPS switching frequencies, respectively).
Boost Flying Capacitor Connection for 5V SMPS. Connect to an external capacitor and diode
according to the Typical Application Circuits (Figure 1 and Figure 2). See the MOSFET Gate
Drivers (DH_, DL_) section.
Inductor Connection for 5V SMPS. LX5 is the internal lower supply rail for the DH5 high-side
gate driver. LX5 is the current-sense input for the 5V SMPS (MAX1999 only).
High-Side MOSFET Floating Gate-Driver Output for 5V SMPS. DH5 swings from LX5 to BST5.
Analog Supply Voltage Input for PWM Core. Connect VCC to the system supply voltage with a
series 50Ω resistor. Bypass to GND with a 1µF ceramic capacitor.
5V Linear-Regulator Output. LDO5 is the gate-driver supply for the external MOSFETs. LDO5
can provide a total of 100mA, including MOSFET gate-drive requirements and external loads.
The internal load depends on the choice of MOSFET and switching frequency (see the
Reference and Linear Regulators (REF, LDO5, and LDO3) section). If OUT5 is greater than the
LDO5 bootstrap switch threshold, the LDO5 regulator shuts down and the LDO5 pin connects
to OUT5 through a 1.4Ω switch. Bypass LDO5 with a minimum of 4.7µF. Use an additional 1µF
per 5mA of load.
5V SMPS Synchronous Rectifier Gate-Drive Output. DL5 swings between GND and LDO5.
Power-Supply Input. V+ powers the LDO5/LDO3 linear regulators and is also used for the
Quick-PWM on-time one-shot circuits. Connect V+ to the battery input through a 4Ω resistor
and bypass with a 4.7µF capacitor.
5V SMPS Output Voltage-Sense Input. Connect to the 5V SMPS output. OUT5 is an input to the
Quick-PWM on-time one-shot circuit. It also serves as the 5V feedback input in fixed-voltage
mode. If OUT5 is greater than the LDO5 bootstrap-switch threshold, the LDO5 linear regulator
shuts down and LDO5 connects to OUT5 through a 1.4Ω switch.
3.3V SMPS Output Voltage-Sense Input. Connect to the 3.3V SMPS output. OUT3 is an input to
the Quick-PWM on-time one-shot circuit. It also serves as the 3V feedback input in fixedvoltage mode. If OUT3 is greater than the LDO3 bootstrap-switch threshold, the LDO3 linear
regulator shuts down and LDO3 connects to OUT3 through a 1.5Ω switch.
Analog and Power Ground
3.3V SMPS Synchronous-Rectifier Gate-Drive Output. DL3 swings between GND and LDO5.
3.3V Linear-Regulator Output. LDO3 can provide a total of 100mA to external loads. If OUT3 is
greater than the LDO3 bootstrap-switch threshold, the LDO3 regulator shuts down and the
LDO3 pin connects to OUT3 through a 1.5Ω switch. Bypass LDO3 with a minimum of 4.7µF.
Use an additional 1µF per 5mA of load.
High-Side MOSFET Floating Gate-Driver Output for 3.3V SMPS. DH3 swings from LX3 to BST3.
Inductor Connection for 3.3V SMPS. LX3 is the current-sense input for the 3.3V SMPS
(MAX1999 only).
Boost Flying Capacitor Connection for 3.3V SMPS. Connect to an external capacitor and diode
according to the Typical Application Circuits (Figure 1 and Figure 2). See the MOSFET Gate
Drivers (DH_, DL_) section.
______________________________________________________________________________________
13
MAX1777/MAX1977/MAX1999
Pin Description (continued)
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
Typical Application Circuit
Table 1. Component Suppliers
The typical application circuits (Figures 1 and 2) generate the 5V/5A and 3.3V/5A main supplies in a notebook
computer. The input supply range is 7V to 24V. Table 1
lists component suppliers.
Detailed Description
The MAX1777/MAX1977/MAX1999 dual-buck, BiCMOS,
switch-mode power-supply controllers generate logic
supply voltages for notebook computers. The
MAX1777/MAX1977/MAX1999 are designed primarily
for battery-powered applications where high-efficiency
and low-quiescent supply current are critical.
The MAX1777 is optimized for highest efficiency with a
5V/200kHz SMPS and a 3.3V/300kHz SMPS, while the
MANUFACTURER
PHONE
FACTORY FAX
Central Semiconductor
516-435-1110
516-435-1824
Dale-Vishay
402-564-3131
402-563-6418
Fairchild
408-721-2181
408-721-1635
International Rectifier
310-322-3331
310-322-3332
NIEC (Nihon)
805-843-7500
847-843-2798
Sanyo
619-661-6835
619-661-1055
Sprague
603-224-1961
603-224-1430
Sumida
847-956-0666
847-956-0702
Taiyo Yuden
408-573-4150
408-573-4159
TDK
847-390-4461
847-390-4405
VIN 7V TO 24V
5V ALWAYS ON
1µF
4.7µF
50Ω
4Ω
LDO5
1µF
VCC
0.1µF
BST5
10Ω
DH5
0.1µF
L5
N3
FDS6612A
DH3
MAX1777/
MAX1977
0.1µF
LX5
L3
LX3
C5
3.3V
C3
D3
EP10QY03
N2
IRF7811AV
DL5
DL3
CS5
CS3
OUT3
OUT5
RCS5
20mΩ
FB5
N4
IRF7811AV
VCC
FB3
100kΩ
ON
OFF
SHDN
VCC
ON5
REF
ON3
D2
EP10QY03
RCS3
20mΩ
PGOOD
SKIP
FREQUENCY-DEPENDENT COMPONENTS
MAX1777
MAX1977
5V/3.3V SMPS
SWITCHING FREQUENCY
200kHz/300kHz
400kHz/500kHz
L3
4.7µH
3.0µH
GND
REF
PRO
LDO3
1MΩ
3.3V ALWAYS ON
L5
7.6µH
5.6µH
C3
470µF
220µF
C5
330µF
150µF
0.22µF
4.7µF
Figure 1. MAX1777/MAX1977 Typical Application Circuit
14
10µF
BST3
10Ω
5V
10µF
CMPSH-3A
1/2
D1
N1
FDS6612A
1/2
D1
ILIM5
V+
4.7µF
10µF
ILIM3
______________________________________________________________________________________
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
MAX1777/MAX1977/MAX1999
VIN 7V TO 24V
5V ALWAYS ON
1µF
50Ω
4Ω
LDO5
1µF
VCC
1/2
D1
ILIM5
V+
4.7µF
10µF
ILIM3
0.1µF
BST5
BST3
DH5
DH3
10Ω
10µF
10Ω
N1
FDS6612A
MAX1999
N3
FDS6612A
0.1µF
0.1µF
L5
LX5
L3
LX3
470pF*
C5
10µF
CMPSH-3A
1/2
D1
5V
4.7µF
SEE
TABLE
TON
D3
EP10QY03
N2
IRF7811AV
DL5
DL3
OUT5
OUT3
FB5
470pF*
N4
IRF7811AV
3.3V
C3
D2
EP10QY03
VCC
FB3
100kΩ
ON
SHDN
OFF
VCC
ON5
REF
ON3
PGOOD
SKIP
GND
REF
PRO
LDO3
1MΩ
3.3V ALWAYS ON
0.22µF
*OPTIONAL CAPACITANCE BETWEEN
LX AND PGND (CLOSE TO THE IC) ONLY
REQUIRED FOR ULTRASONIC MODE
4.7µF
FREQUENCY-DEPENDENT COMPONENTS
TON = VCC
TON = GND
5V/3.3V SMPS
SWITCHING FREQUENCY
200kHz/300kHz
400kHz/500kHz
L3
4.7µH
3.0µH
L5
7.6µH
5.6µH
C3
470µF
220µF
C5
330µF
150µF
Figure 2. MAX1999 Typical Application Circuit
______________________________________________________________________________________
15
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
MAX1977 is optimized for “thin and light” applications with a
5V/400kHz SMPS and a 3.3V/500kHz SMPS. The MAX1999
provides a pin-selectable switching frequency, allowing
either 200kHz/300kHz or 400kHz/500kHz operation of the
5V/3.3V SMPSs, respectively.
Light-load efficiency is enhanced by automatic Idle
Mode™ operation, a variable-frequency pulse-skipping
mode that reduces transition and gate-charge losses.
Each step-down, power-switching circuit consists of two
N-channel MOSFETs, a rectifier, and an LC output filter.
The output voltage is the average AC voltage at the
switching node, which is regulated by changing the duty
cycle of the MOSFET switches. The gate-drive signal to
the N-channel high-side MOSFET must exceed the
battery voltage, and is provided by a flying-capacitor
boost circuit that uses a 100nF capacitor connected to
BST_.
V+
PGOOD
MAX1777/
MAX1977/
MAX1999
PGOOD3
PGOOD5
TON
(MAX1999 ONLY)
BST3
BST5
DH3
DH5
LX3
3.3V
SMPS PWM
CONTROLLER
LDO5
LX5
5V
SMPS PWM
CONTROLLER
LDO5
DL5
DL3
CS3
(MAX1777/
MAX1977)
CS5
(MAX1777/
MAX1977)
ILIM5
ILIM3
FB3
FB5
OUT3
OUT5
EN3
LDO3
4.56V
2.91V
EN5
5V
LINEAR
REG
3V
LINEAR
REG
LDO5
VCC
ON3
ON5
POWER-ON SEQUENCE/
CLEAR FAULT LATCH
THERMAL
SHUTDOWN
SHDN
2V
REFERENCE
PRO
REF
GND
Figure 3. Detailed Functional Diagram
Idle Mode is a trademark of Maxim Integrated Products, Inc.
16
______________________________________________________________________________________
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
controls the power-up timing of the main PWMs and monitors the outputs for undervoltage faults. The
MAX1777/MAX1977/MAX1999 include 5V and 3.3V linear
regulators. Bias generator blocks include the 5V (LDO5)
linear regulator, 2V precision reference, and automatic
bootstrap switchover circuit.
V+
OUT
TON (MAX1999)
ON-TIME
COMPUTE
tOFF
TRIG
Q
ONE SHOT
tON
TRIG
Q
R
TO DH_ DRIVER
ONE SHOT
Q
REF
S
ERROR
AMPLIFIER
ILIM_
CURRENT
LIMIT
Σ
TO DL_ DRIVER
CS_ (MAX1777/1977)
LX_ (MAX1999)
ZERO
CROSSING
S
Q
R
SKIP
OUT_
PGOOD
0.9 ✕ VREF
FB_
OV_FAULT
UV_FAULT
FAULT
LATCH
1.1 ✕ VREF
0.15V
PRO
20ms
BLANKING
0.7✕VREF
Figure 4. PWM Controller (One Side Only)
______________________________________________________________________________________
17
MAX1777/MAX1977/MAX1999
Each PWM controller consists of a Dual Mode feedback
network and multiplexer, a multi-input PWM comparator,
high-side and low-side gate drivers, and logic. The
MAX1777/MAX1977/MAX1999 contain fault-protection circuits that monitor the main PWM outputs for undervoltage
and overvoltage conditions. A power-on sequence block
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
These internal blocks are not powered directly from the
battery. Instead, the 5V (LDO5) linear regulator steps
down the battery voltage to supply both internal circuitry and the gate drivers. The synchronous-switch gate
drivers are directly powered from LDO5, while the highside switch gate drivers are indirectly powered from
LDO5 through an external diode-capacitor boost circuit. An automatic bootstrap circuit turns off the 5V linear regulator and powers the device from OUT5 when
OUT5 is above 4.56V.
Free-Running, Constant On-Time PWM
Controller with Input Feed Forward
The Quick-PWM control architecture is a pseudo-fixedfrequency, constant on-time, current-mode type with
voltage feedforward. The Quick-PWM control architecture relies on the output ripple voltage to provide the
PWM ramp signal, thus the output filter capacitor’s ESR
acts as a current-feedback resistor. The high-side
switch on-time is determined by a one-shot whose period is inversely proportional to input voltage and directly
proportional to output voltage. Another one-shot sets a
minimum off-time (300ns typ). The on-time one-shot
triggers when the following conditions are met: the error
comparator is low, the synchronous rectifier current is
below the current-limit threshold, and the minimum offtime one-shot has timed out.
On-Time One-Shot (tON)
Each PWM core includes a one-shot that sets the highside switch on-time for each controller. Each fast, lowjitter, adjustable one-shot includes circuitry that varies
the on-time in response to battery and output voltage.
The high-side switch on-time is inversely proportional to
the battery voltage as measured by the V+ input, and
proportional to the output voltage. This algorithm results
in a nearly constant switching frequency despite the
lack of a fixed-frequency clock generator. The benefit
of a constant switching frequency is the frequency can
be selected to avoid noise-sensitive frequency regions:
tON = K (VOUT + 0.075V) / V+
See Table 2 for approximate K-factors. The constant
0.075V is an approximation to account for the expected
drop across the synchronous-rectifier switch. Switching
frequency increases as a function of load current due
to the increasing drop across the synchronous rectifier,
which causes a faster inductor-current discharge ramp.
On-times translate only roughly to switching frequencies. The on-times guaranteed in the Electrical
Characteristics are influenced by switching delays in
the external high-side power MOSFET. Also, the deadtime effect increases the effective on-time, reducing the
switching frequency. It occurs only in PWM mode (SKIP
= VCC) and during dynamic output voltage transitions
when the inductor current reverses at light or negative
load currents. With reversed inductor current, the
inductor’s EMF causes LX to go high earlier than normal, extending the on-time by a period equal to the DHrising dead time.
For loads above the critical conduction point, the actual
switching frequency is:
f=
VOUT + VDROP1
t ON ( V + + VDROP2 )
where VDROP1 is the sum of the parasitic voltage drops
in the inductor discharge path, including synchronous
rectifier, inductor, and PC board resistances; VDROP2 is
the sum of the parasitic voltage drops in the charging
path, including high-side switch, inductor, and PC
board resistances, and tON is the on-time calculated by
the MAX1777/MAX1977/MAX1999.
Automatic Pulse-Skipping Switchover
(Idle Mode)
In Idle Mode (SKIP = GND), an inherent automatic
switchover to PFM takes place at light loads. This
switchover is affected by a comparator that truncates
the low-side switch on-time at the inductor current’s
zero crossing. This mechanism causes the threshold
between pulse-skipping PFM and nonskipping PWM
operation to coincide with the boundary between con-
Table 2. Approximate K-Factor Errors
SMPS
MAX1777/MAX1999 (tON = VCC), 5V
MAX1777/MAX1999 (tON = VCC), 3.3V
MAX1977/MAX1999 (tON = GND), 5V
MAX1977/MAX1999 (tON = GND), 3.3V
18
SWITCHING
FREQUENCY (kHz)
200
300
400
500
K-FACTOR (µs)
5.0
3.3
2.5
2.0
APPROXIMATE KFACTOR ERROR (%)
±10
±10
±10
±10
______________________________________________________________________________________
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
ILOAD(SKIP) =
K × VOUT _ ⎛ V + − VOUT _ ⎞
⎜
⎟
2×L
V+
⎝
⎠
where K is the on-time scale factor (see the On-Time
One-Shot (tON) section). The load-current level at which
PFM/PWM crossover occurs, ILOAD(SKIP), is equal to 1/2
the peak-to-peak ripple current, which is a function of the
inductor value (Figure 5). For example, in the MAX1777
typical application circuit with VOUT2 = 5V, V+ = 12V,
L = 7.6µH, and K = 5µs, switchover to pulse-skipping
operation occurs at ILOAD = 0.96A or about 1/5 full load.
The crossover point occurs at an even lower value if a
swinging (soft-saturation) inductor is used.
The switching waveforms may appear noisy and asynchronous when light loading causes pulse-skipping
operation, but this is a normal operating condition that
results in high light-load efficiency. Trade-offs in PFM
noise vs. light-load efficiency are made by varying the
inductor value. Generally, low inductor values produce
a broader efficiency vs. load curve, while higher values
result in higher full-load efficiency (assuming that the
coil resistance remains fixed) and less output voltage
ripple. Penalties for using higher inductor values
include larger physical size and degraded load-transient response (especially at low input voltage levels).
DC output accuracy specifications refer to the trip level of
the error comparator. When the inductor is in continuous
conduction, the output voltage has a DC regulation higher than the trip level by 50% of the ripple. In discontinuous conduction (SKIP = GND, light load), the output
voltage has a DC regulation higher than the trip level by
approximately 1.5% due to slope compensation.
∆i
V+ - VOUT
=
∆t
L
Forced-PWM Mode
The low-noise, forced-PWM (SKIP = VCC) mode disables the zero-crossing comparator, which controls the
low-side switch on-time. Disabling the zero-crossing
detector causes the low-side, gate-drive waveform to
become the complement of the high-side, gate-drive
waveform. The inductor current reverses at light loads
as the PWM loop strives to maintain a duty ratio of
VOUT/V+. The benefit of forced-PWM mode is to keep
the switching frequency fairly constant, but it comes at
a cost: the no-load battery current can be 10mA to
50mA, depending on switching frequency and the
external MOSFETs.
Forced-PWM mode is most useful for reducing audiofrequency noise, improving load-transient response,
providing sink-current capability for dynamic output
voltage adjustment, and improving the cross-regulation
of multiple-output applications that use a flyback transformer or coupled inductor.
Minimum 25kHz Pulse-Skipping Mode
(Ultrasonic Mode)
Leaving SKIP unconnected or connecting SKIP to REF
activates a pulse-skipping mode with a minimum
switching frequency of 25kHz. This ultrasonic pulseskipping mode reduces audio-frequency modulation of
the power supply that may occur in Idle Mode at very
light loads. The transition to fixed-frequency PWM operation is automatic and occurs at the same point as in
Idle Mode. Ultrasonic pulse skipping occurs if no
switching has taken place within the last 28µs. DL_
turns on to induce a regulated negative current in the
inductor. DH_ turns on when the inductor current
reaches the regulated negative current limit. Starting
with a DL_ pulse greatly reduces the ripple current
when compared to starting with a DH_ pulse (Idle
Mode). The output voltage level determines the negative current limit.
Calculate the negative ultrasonic current-limit threshold
with the following equation:
-IPEAK
INDUCTOR CURRENT
VNEGUS = ILX × RON =
ILOAD = IPEAK/2
0 ON-TIME
(VREF − VFB )
VILIM _
× 0.467V
where VFB > VREF, and RON is the on-resistance of the
synchronous rectifier (MAX1999) or the current-sense
resistor value (MAX1777/MAX1977).
TIME
Figure 5. Pulse- Skipping/Discontinuous Crossover Point
______________________________________________________________________________________
19
MAX1777/MAX1977/MAX1999
tinuous and discontinuous inductor-current operation
(also known as the critical conduction point):
Reference and Linear Regulators
(REF, LDO5, and LDO3)
The 2V reference (REF) is accurate to ±1% over temperature, making REF useful as a precision system reference. Bypass REF to GND with a 0.22µF minimum
capacitor. REF can supply up to 100µA for external
loads. However, if extremely accurate specifications for
both the main output voltages and REF are essential,
avoid loading REF. Loading REF reduces the LDO5,
LDO3, OUT5, and OUT3 output voltages slightly,
because of the reference load-regulation error.
Two internal regulators produce 5V (LDO5) and
3.3V(LDO3). LDO5 provides gate drive for the external
MOSFETs and powers the PWM controller, logic, reference, and other blocks within the device. The LDO5
regulator supplies a total of 100mA for internal and
external loads, including MOSFET gate drive, which
typically varies from 10mA to 50mA, depending on
switching frequency and the external MOSFETs. LDO3
supplies up to 100mA for external loads. Bypass LDO5
and LDO3 with a minimum of 4.7µF load, use an additional 1µF per 5mA of internal and external load.
When the 5V main output voltage is above the LDO5
bootstrap-switchover threshold, an internal 1.4Ω P-channel MOSFET switch connects OUT5 to LDO5, while simultaneously shutting down the LDO5 linear regulator.
Similarly, when the 3.3V main output voltage is above the
LDO3 bootstrap-switchover threshold, an internal 1.5Ω
P-channel MOSFET switch connects OUT3 to LDO3,
while simultaneously shutting down the LDO3 linear regulator. These actions bootstrap the device, powering the
internal circuitry and external loads from the output SMPS
voltages, rather than through linear regulators from the
battery. Bootstrapping reduces power dissipation due to
gate charge and quiescent losses by providing power
from a 90%-efficient switch-mode source, rather than
from a much-less-efficient linear regulator.
Current Limit Circuit (ILIM_)
The current-limit circuit employs a “valley” current-sensing algorithm. The MAX1999 uses the on-resistance of
the synchronous rectifier, while the MAX1777/MAX19777
uses a discrete resistor in series with the source of the
synchronous rectifier as a current-sensing element. If the
magnitude of the current-sense signal at CS_
(MAX1777/MAX1977) / LX_ (MAX1999) is above the current-limit threshold, the PWM is not allowed to initiate a
new cycle (Figure 6). The actual peak current is greater
than the current-limit threshold by an amount equal to the
inductor ripple current. Therefore, the exact current-limit
characteristic and maximum load capability are a function of the current-limit threshold, inductor value, and
input and output voltage.
For the MAX1777/MAX1977, connect CS_ to the junction
of the synchronous rectifier source and a current-sense
resistor to GND. With a current-limit threshold of 100mV,
the accuracy is approximately ±7%. Using a lower current-sense threshold results in less accuracy. The current-sense resistor only dissipates power when the
synchronous rectifier is on.
For lower power dissipation, the MAX1999 uses the onresistance of the synchronous rectifier as the currentsense element (Figure 7). Use the worst-case maximum
value for RDS(ON) from the MOSFET data sheet, and
add some margin for the rise in RDS(ON) with temperature. A good general rule is to allow 0.5% additional
resistance for each °C of temperature rise. The current
limit varies with the on-resistance of the synchronous
rectifier. The reward for this uncertainty is robust, lossless overcurrent sensing. When combined with the
undervoltage protection circuit, this current-limit
method is effective in almost every circumstance.
V+
MAX1999
-IPEAK
ILOAD
INDUCTOR CURRENT
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
ILIMIT
0
TIME
Figure 6. “Valley” Current-Limit Threshold Point
20
DH_
LX_
OUT_
DL_
Figure 7. Current Sensing Using RDS(ON) of Synchronous
Rectifier
______________________________________________________________________________________
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
MOSFET Gate Drivers (DH_, DL_)
V+
MAX1777
MAX1977
DH_
LX_
OUT_
DL_
CS_
Figure 8. Current Sensing Using Sense Resistor
(MAX1777/MAX1977)
V+
MAX1777
MAX1977
Carefully observe the PC board layout guidelines to
ensure that noise and DC errors do not corrupt the current-sense signals at CS_. Mount or place the device
close to the synchronous rectifier or sense resistor
(whichever is used) with short, direct traces, making a
Kelvin sense connection to the sense resistor. The current-sense accuracy of Figure 8 is degraded if the
Schottky diode conducts during the synchronous rectifier on-time. To ensure that all current passes through
the sense resistor, connect the Schottky diode in parallel with only the synchronous recifier (Figure 9) if the
voltage drop across the synchronous rectifier and
sense resistor exceeds the Schottky diode’s forward
voltage. Note that at high temperatures, the on-resistance of the synchronous rectifier increases, and the
forward voltage of the Schottky diode decreases.
The DH_ and DL_ gate drivers sink 2.0A and 3.3A
respectively of gate drive, ensuring robust gate drive for
high-current applications. The DH_ floating high-side
MOSFET drivers are powered by diode-capacitor charge
pumps at BST_. The DL_ synchronous-rectifier drivers are
powered by LDO5.
The internal pulldown transistors that drive DL_ low
have a 0.6Ω typical on-resistance. These low on-resistance pulldown transistors prevent DL_ from being
pulled up during the fast rise time of the inductor nodes
due to capacitive coupling from the drain to the gate of
the low-side synchronous-rectifier MOSFETs. However,
for high-current applications, some combinations of
high- and low-side MOSFETS may cause excessive
gate-drain coupling, which leads to poor efficiency and
EMI-producing shoot-through currents. Adding a resistor in series with BST_ increases the turn-on time of the
high-side MOSFETs at the expense of efficiency, without
degrading the turn-off time (Figure 10).
DH_
5V
LX_
VIN
OUT_
BST
10Ω
DL_
DH
CS_
LX
MAX1777
MAX1977
MAX1999
Figure 9. More Accurate Current Sensing with Adjusted
Schottky Connection
Figure 10. Reducing the Switching-Node Rise Time
______________________________________________________________________________________
21
MAX1777/MAX1977/MAX1999
A negative current limit prevents excessive reverse
inductor currents when VOUT sinks current. The negative current-limit threshold is set to approximately 120%
of the positive current limit and therefore tracks the
positive current limit when ILIM_ is adjusted.
The current-limit threshold is adjusted with an external
voltage-divider at ILIM_. The current-limit threshold
adjustment range is from 50mV to 300mV. In the
adjustable mode, the current-limit threshold voltage is
precisely 1/10th the voltage at ILIM_. The threshold
defaults to 100mV when ILIM_ is connected to VCC.
The logic threshold for switchover to the 100mV default
value is approximately VCC - 1V.
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
Adaptive dead-time circuits monitor the DL_ and DH_
drivers and prevent either FET from turning on until the
other is fully off. This algorithm allows operation without
shoot-through with a wide range of MOSFETs, minimizing delays and maintaining efficiency. There must be
low-resistance, low-inductance paths from the gate drivers to the MOSFET gates for the adaptive dead-time circuit to work properly. Otherwise, the sense circuitry
interprets the MOSFET gate as “off” when there is actually charge left on the gate. Use very short, wide traces
measuring 10 to 20 squares (50mils to 100mils wide if
the MOSFET is 1in from the device).
POR, UVLO, and Internal Digital
Soft-Start
Power-on reset (POR) occurs when V+ rises above
approximately 1V, resetting the undervoltage, overvoltage, and thermal-shutdown fault latches. LDO5
undervoltage lockout (UVLO) circuitry inhibits switching
when LDO5 is below 4V. DL_ is low if PRO is disabled;
DL_ is high if PRO is enabled. The output voltages
begin to ramp up as LDO5 rises above 4V. The internal
digital soft-start timer begins to ramp up the maximum
allowed current limit during startup. The 1.7ms ramp
occurs in five steps: 20%, 40%, 60%, 80%, and 100%.
Power-Good Output (PGOOD)
The PGOOD comparator continuously monitors both output voltages for undervoltage conditions. PGOOD is
actively held low in shutdown, standby, and soft-start.
PGOOD releases and digital soft-start terminates when
both outputs reach the error-comparator threshold.
PGOOD goes low if either output turns off or is 10%
below its nominal regulation point. PGOOD is a true
open-drain output. Note that PGOOD is independent of
the state of PRO.
Fault Protection
The MAX1777/MAX1977/MAX1999 provide over/undervoltage fault protection. Drive PRO low to activate fault
protection. Drive PRO high to disable fault protection.
Once activated, the devices continuously monitor for
both undervoltage and overvoltage conditions.
Overvoltage Protection
When the output voltage is 11% above the set voltage,
the overvoltage fault protection activates. The synchronous rectifier turns on 100% and the high-side MOSFET
turns off. This rapidly discharges the output capacitors,
decreasing the output voltage. The output voltage may
dip below ground. For loads that cannot tolerate a negative voltage, place a power Schottky diode across the
output to act as a reverse-polarity clamp. In practical
applications, there is a fuse between the power source
22
(battery) and the external high-side switches. If the
overvoltage condition is caused by a short in the highside switch, turning the synchronous rectifier on 100%
creates an electrical short between the battery and
GND, blowing the fuse and disconnecting the battery
from the output. Once an overvoltage fault condition is
set, it can only be reset by toggling SHDN, ON_, or
cycling V+ (POR).
Undervoltage Protection
When the output voltage is 30% below the set voltage
for over 22ms (undervoltage shutdown blanking time),
the undervoltage fault protection activates. Both SMPSs
stop switching. The two outputs start to discharge (see
the Discharge Mode (Soft-Stop) section). When the output voltage drops to 0.3V, the synchronous rectifiers
turn on, clamping the outputs to GND. Toggle SHDN,
ON_, or cycle V+ (POR) to clear the undervoltage fault
latch.
Thermal Protection
The MAX1777/MAX1977/MAX1999 have thermal shutdown to protect the devices from overheating. Thermal
shutdown occurs when the die temperature exceeds
+160°C. All internal circuitry shuts down during thermal
shutdown. The MAX1777/MAX1977/MAX1999 may trigger thermal shutdown if LDO_ is not bootstrapped from
OUT_ while applying a high input voltage on V+ and
drawing the maximum current (including short circuit)
from LDO_. Even if LDO_ is bootstrapped from OUT_,
overloading the LDO_ causes large power dissipation
on the bootstrap switches, which may result in thermal
shutdown. Cycling SHDN, ON3, or ON5, or a V+ (POR)
ends the thermal shutdown state.
Discharge Mode (Soft-Stop)
When PRO is low, and a transition to standby or shutdown mode occurs, or the output undervoltage fault
latch is set, the outputs discharge to GND through an
internal 12Ω switch, until the output voltages decrease
to 0.3V. The reference remains active to provide an
accurate threshold and to provide overvoltage protection. When both SMPS outputs discharge to 0.3V, the
DL_ synchronous rectifier drivers are forced high. The
synchronous rectifier drivers clamp the SMPS outputs
to GND. When PRO is high, the SMPS outputs do not
discharge, and the DL_ synchronous rectifier drivers
remain low.
Shutdown Mode
Drive SHDN below the precise SHDN input falling-edge
trip level to place the MAX1777/MAX1977/MAX1999 in
its low-power shutdown state. The MAX1777/MAX1977/
MAX1999 consume only 6µA of quiescent current while
______________________________________________________________________________________
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
MODE
CONDITION
COMMENT
Power-Up
LDO5 < UVLO threshold
Transitions to discharge mode after a V+ POR and after REF becomes valid.
LDO5, LDO3, REF remain active. DL_ is active if PRO is low.
Run
SHDN = high, ON3 or ON5
enabled
Normal operation
Overvoltage
Protection
Either output > 111% of
nominal level, PRO = low
DL_ is forced high. LDO3, LDO5 active. Exited by a V+ POR or by toggling
SHDN, ON3, or ON5.
Undervoltage
Protection
Either output < 70% of
nominal after 22ms timeout expires and output is
enabled, PRO = low
If PRO is low, DL_ is forced high after discharge mode terminates. LDO3,
LDO5 active. Exited by a V+ POR or by toggling SHDN, ON3, or ON5.
Discharge
PRO is low and either
SMPS output is still high in
either standby mode or
shutdown mode
Discharge switch (12Ω) connects OUT_ to PGND. One output may still run
while the other is in discharge mode. Activates when LDO_ is in UVLO, or
transition to UVLO, standby, or shutdown has begun. LDO3, LDO5 active.
Standby
ON5, ON3 < startup
threshold, SHDN = high
DL_ stays high if PRO is low. LDO3, LDO5 active.
Shutdown
SHDN = low
All circuitry off
Thermal Shutdown
TJ > +160°C
All circuitry off. Exited by V+ POR or cycling SHDN, ON3, or ON5.
Power-Up Sequencing and
On/Off Controls (ON3, ON5)
in shutdown mode. When shutdown mode activates, the
reference turns off, making the threshold to exit shutdown inaccurate. To guarantee startup, drive SHDN
above 2V (SHDN input rising-edge trip level). For automatic shutdown and startup, connect SHDN to V+. If
PRO is low, both SMPS outputs are discharged to 0.3V
through a 12Ω switch before entering true shutdown.
The accurate 1V falling-edge threshold on SHDN can be
used to detect a specific analog voltage level and shut
the device down. Once in shutdown, the 1.6V risingedge threshold activates, providing sufficient hysteresis
for most applications. For additional hysteresis, the
undervoltage threshold can be made dependent on
REF or LDO_, which go to 0V in shutdown.
ON3 and ON5 control SMPS power-up sequencing.
ON3 or ON5 rising above 2.4V enables the respective
outputs. ON3 or ON5 falling below 1.6V disables the
respective outputs.
Connecting ON3 or ON5 to REF forces the respective
outputs off while the other output is below regulation and
starts after that output regulates. The second SMPS
remains on until the first SMPS turns off, the device shuts
down, a fault occurs, or LDO5 goes into undervoltage
lockout. Both supplies begin their power-down sequence
immediately when the first supply turns off. Driving ON_
Table 4. Power-Up Sequencing
SHDN (V)
VON3 (V)
VON5 (V)
LDO5
LDO3
5V SMPS
3V SMPS
< 1.0
X
X
Off
Off
Off
Off
> 2.4
< 1.6
< 1.6
On
On
Off
Off
> 2.4
> 2.4
> 2.4
On
On
On
On
> 2.4
> 2.4
< 1.6
On
On
Off
On
> 2.4
< 1.6
> 2.4
On
On
On
Off
> 2.4
> 2.4
REF
On
On
On (after 3V
SMPS is up)
On
> 2.4
REF
> 2.4
On
On
On
On (after 5V
SMPS is up)
______________________________________________________________________________________
23
MAX1777/MAX1977/MAX1999
Table 3. Operating Mode Truth Table
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
below 0.8V clears the overvoltage, undervoltage, and
thermal fault latches.
Adjustable-Output Feedback
(Dual-Mode FB)
Connect FB_ to GND to enable the fixed, preset SMPS
output voltages (3.3V and 5V). Connect a resistive voltage-divider at FB_ between OUT_ and GND to adjust
the respective output voltage between 2V and 5.5V
(Figure 11). Choose R2 to be about 10kΩ and solve for
R1 using the equation:
⎛V
⎞
OUT _
R1 = R2 × ⎜
− 1⎟
⎜ V
⎟
⎝ FB
⎠
where VFB = 2V nominal.
When using the adjustable output mode, set the 3.3V
SMPS lower than the 5V SMPS. LDO5 connects to OUT5
through an internal switch only when OUT5 is above the
LDO5 bootstrap-switch threshold (4.56V). LDO3 connects to OUT3 through an internal switch only when
OUT3 is above the LDO3 bootstrap switch threshold
(2.91V). Bootstrapping is most effective when the fixed
output voltages are used. Once LDO_ is bootstrapped
from OUT_, the internal linear regulator turns off. This
reduces internal power dissipation and improves efficiency when LDO_ is powered with a high input voltage.
V+
DH_
VOUT_
MAX1777
MAX1977
MAX1999
DL_
GND
R1
OUT_
FB_
R2
Design Procedure
Establish the input voltage range and maximum load
current before choosing an inductor and its associated
ripple-current ratio (LIR). The following four factors dictate the rest of the design:
1) Input Voltage Range. The maximum value (V+(MAX))
must accommodate the maximum AC adapter voltage. The minimum value (V+(MIN)) must account for
the lowest input voltage after drops due to connectors, fuses, and battery selector switches. Lower input
voltages result in better efficiency.
2) Maximum Load Current. The peak load current
(ILOAD(MAX)) determines the instantaneous component stress and filtering requirements, and thus drives output capacitor selection, inductor saturation
rating, and the design of the current-limit circuit.
The continuous load current (ILOAD) determines the
thermal stress and drives the selection of input
capacitors, MOSFETs, and other critical heat-contributing components.
3) Switching Frequency. This choice determines the
basic trade-off between size and efficiency. The
optimal frequency is largely a function of maximum
input voltage and MOSFET switching losses. The
MAX1777 has a nominal switching frequency of
200kHz for the 5V SMPS and 300kHz for the 3.3V
SMPS. The MAX1977 has a nominal switching frequency of 400kHz for the 5V SMPS and 500kHz for
the 3.3V SMPS. The MAX1999 has pin-selectable
switching frequency.
4) Inductor Ripple Current Ratio (LIR). LIR is the
ratio of the peak-peak ripple current to the average
inductor current. Size and efficiency trade-offs must
be considered when setting the inductor ripple current ratio. Low inductor values cause large ripple
currents, resulting in the smallest size, but poor efficiency and high output noise. The minimum practical inductor value is one that causes the circuit to
operate at critical conduction (where the inductor
current just touches zero with every cycle at maximum load). Inductor values lower than this grant no
further size reduction benefit.
The MAX1777/MAX1977/MAX1999s’ pulse-skipping
algorithm (SKIP = GND) initiates skip mode at the
critical conduction point. So the inductor’s operating point also determines the load current at which
PWM/PFM switchover occurs. The optimum point is
usually found between 20% and 50% ripple current.
Figure 11. Setting VOUT_ with a Resistor-Divider
24
______________________________________________________________________________________
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
L =
(
VOUT _ V + − VOUT _
)
V + × f × LIR × ILOAD(MAX)
Example: ILOAD(MAX) = 5A, V+ = 12V, VOUT5 = 5V, f =
200kHz, 35% ripple current or LIR = 0.35:
L=
(
5V 12V − 5V
)
12V × 200kHz × 0.35 × 5A
= 8.3µH
Find a low-loss inductor having the lowest possible DC
resistance that fits in the allotted dimensions. Ferrite cores
are often the best choice. The core must be large enough
not to saturate at the peak inductor current (IPEAK):
IPEAK = ILOAD(MAX) + [(LIR/2) x ILOAD(MAX)]
The inductor ripple current also impacts transientresponse performance, especially at low V+ - VOUT_
difference. Low inductor values allow the inductor current to slew faster, replenishing charge removed from
the output filter capacitors by a sudden load step. The
peak amplitude of the output transient (VSAG) is also a
function of the maximum duty factor, which can be calculated from the on-time and minimum off-time:
VSAG =
(∆ILOAD(MAX) )
2
⎛ V
⎞
OUT _
× L ⎜K
+ t OFF(MIN) ⎟
⎜
⎟
V+
⎝
⎠
⎡ ⎛ V+− V
⎤
⎞
OUT _
− t OFF(MIN)
2 × COUT × VOUT _ ⎢K⎜
⎟
⎟
⎢ ⎜⎝
V+
⎠
⎣
⎦
where minimum off-time = 0.350µs (max) and K is from
Table 2.
Determining the Current Limit
The minimum current-limit threshold must be great
enough to support the maximum load current when the
current limit is at the minimum tolerance value. The valley of the inductor current occurs at ILOAD(MAX) minus
half of the ripple current; therefore,
ILIMIT(LOW) > ILOAD(MAX) - [(LIR / 2) x ILOAD(MAX)]
where ILIMIT(LOW) = minimum current-limit threshold
voltage divided by the RDS(ON) of N2/N4 (MAX1999).
For the MAX1777/MAX1977/MAX1999, the minimum
current-limit threshold voltage is 93mV (ILIM_ = VCC).
Use the worst-case maximum value for RDS(ON) from
the MOSFET N2/N4 data sheet and add some margin
for the rise in RDS(ON) with temperature. A good gener-
al rule is to allow 0.5% additional resistance for each °C
of temperature rise.
Examining the 5A circuit example with a maximum
RDS(ON) = 12mΩ at high temperature reveals the following:
ILIMIT(LOW) = 93mV / 12mΩ > 5A - (0.35 / 2) 5A
7.75A > 4.125A
7.75A is greater than the valley current of 4.125A, so
the circuit can easily deliver the full-rated 5A using the
fixed 100mV nominal current-limit threshold voltage.
Connect the source of the synchronous rectifier to a
current-sense resistor to GND (MAX1777/MAX1977),
and connect CS_ to that junction to set the current limit
for the device. The MAX1777/MAX1977/MAX1999 limit
the current with the sense resistor instead of the
RDS(ON) of N2/N4. The maximum value of the sense
resistor can be calculated with the equation
ILIM_ = 93mV / RSENSE
Output Capacitor Selection
The output filter capacitor must have low enough equivalent series resistance (ESR) to meet output ripple and
load-transient requirements, yet have high enough ESR
to satisfy stability requirements. The output capacitance must also be high enough to absorb the inductor
energy while transitioning from full-load to no-load conditions without tripping the overvoltage fault latch. In
applications where the output is subject to large load
transients, the output capacitor’s size depends on how
much ESR is needed to prevent the output from dipping too low under a load transient. Ignoring the sag
due to finite capacitance:
RESR ≤
VDIP
ILOAD(MAX)
where VDIP is the maximum tolerable transient voltage
drop. In non-CPU applications, the output capacitor’s
size depends on how much ESR is needed to maintain
an acceptable level of output voltage ripple:
RESR ≤
VP−P
LIR × ILOAD(MAX)
where VP-P is the peak-to-peak output voltage ripple.
The actual capacitance value required relates to the
physical size needed to achieve low ESR, as well as to
the chemistry of the capacitor technology. Thus, the
capacitor is usually selected by ESR and voltage rating
rather than by capacitance value (this is true of tantalum, OS-CON, and other electrolytic-type capacitors).
______________________________________________________________________________________
25
MAX1777/MAX1977/MAX1999
Inductor Selection
The switching frequency (on-time) and operating point
(% ripple or LIR) determine the inductor value as follows:
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
When using low-capacity filter capacitors such as
polymer types, capacitor size is usually determined by
the capacity required to prevent VSAG and VSOAR from
tripping the undervoltage and overvoltage fault latches
during load transients in ultrasonic mode .
For low input-to-output voltage differentials (VIN/VOUT<2),
additional output capacitance is required to maintain stability and good efficiency in ultrasonic mode.
The amount of overshoot due to stored inductor energy
can be calculated as:
VSOAR =
L × IPEAK 2
2 × C × VOUT _
where IPEAK is the peak inductor current.
Stability Considerations
Stability is determined by the value of the ESR zero
(fESR) relative to the switching frequency (f). The point
of instability is given by the following equation:
f
fESR ≤
π
where:
fESR =
1
2 × π × RESR × COUT
For a typical 300kHz application, the ESR zero frequency must be well below 95kHz, preferably below 50kHz.
Low-ESR capacitors (especially polymer or tantalum),
in widespread use at the time of publication, typically
have ESR zero frequencies lower than of 30kHz. In the
design example used for inductor selection, the ESR
needed to support a specified ripple voltage is found
by the equation:
ESR =
VRIPPLE(P−P)
LIR × ILOAD
where LIR is the inductor ripple current ratio and ILOAD
is the average DC load. Using a LIR = 0.35 and an
average load current of 5A, the ESR needed to support
50mVP-P ripple is 28mΩ.
Do not place high-value ceramic capacitors directly
across the fast-feedback inputs (OUT_ to GND for internal feedback, FB_ divider point for external feedback)
without taking precautions to ensure stability. Large
ceramic capacitors can have a high-ESR zero frequency
and cause erratic, unstable operation. Adding a discrete
resistor or placing the capacitors a couple of inches
downstream from the junction of the inductor and OUT_
may improve stability.
26
Unstable operation manifests itself in two related but
distinctly different ways: double pulsing and fast-feedback loop instability. Noise on the output or insufficient
ESR may cause double pulsing. Insufficient ESR does
not allow the amplitude of the voltage ramp in the output
signal to be large enough. The error comparator mistakenly triggers a new cycle immediately after the 350ns
minimum off-time period has expired. Double pulsing
results in increased output ripple, and can indicate the
presence of loop instability caused by insufficient ESR.
Loop instability results in oscillations or ringing at the
output after line or load perturbations, causing the output voltage to fall below the tolerance limit.
The easiest method for checking stability is to apply a
very fast zero-to-max load transient (refer to the
MAX1999 EV kit data sheet) and observe the output
voltage-ripple envelope for overshoot and ringing.
Monitoring the inductor current with an AC current
probe may also provide some insight. Do not allow
more than one cycle of ringing of under- or overshoot
after the initial step response.
Input Capacitor Selection
The input capacitors must meet the input ripple current
(IRMS) requirement imposed by the switching current.
The MAX1777/MAX1977/MAX1999 dual switching regulators operate at different frequencies. This interleaves
the current pulses drawn by the two switches and
reduces the overlap time where they add together. The
input RMS current is much smaller in comparison than
with both SMPSs operating in phase. The input RMS current varies with load and the input voltage.
The maximum input capacitor RMS current for a single
SMPS is given by:
(
⎛
⎜ VOUT _ V + − VOUT _
IRMS ≈ ILOAD ⎜
V+
⎜
⎝
) ⎞⎟
⎟
⎟
⎠
when V+ = 2 x VOUT_(D = 50%), IRMS has maximum
current of ILOAD/2.
The ESR of the input capacitor is important for determining capacitor power dissipation. All the power
(I2RMS x ESR) heats up the capacitor and reduces efficiency. Nontantalum chemistries (ceramic or OS-CON)
are preferred due to their low ESR and resilience to
power-up surge currents. Choose input capacitors that
exhibit less than +10°C temperature rise at the RMS
input current for optimal circuit longevity. Place the
drains of the high-side switches close to each other to
share common input bypass capacitors.
______________________________________________________________________________________
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
Choose a synchronous rectifier (N2/N4) with the lowest
possible RDS(ON). Ensure the gate is not pulled up by the
high-side switch turning on due to parasitic drain-to-gate
capacitance, causing cross-conduction problems.
Switching losses are not an issue for the synchronous
rectifier in the buck topology, since it is a zero-voltage
switched device when using the buck topology.
MOSFET Power Dissipation
Worst-case conduction losses occur at the duty factor
extremes. For the high-side MOSFET, the worst-case
power dissipation (PD) due to the MOSFET’s RDS(ON)
occurs at minimum battery voltage:
⎛ VOUT _ ⎞
PD (N1/ N3 resis tan ce) = ⎜
⎟×
⎝ V + (MIN) ⎠
ILOAD2 × RDS(ON)
tance, and PC board layout characteristics. The following switching loss calculation provides only a very
rough estimate and is no substitute for bench evaluation, preferably including verification using a thermocouple mounted on N1/N3:
(
)
PD N1/ N3 switching =
⎞
⎛
2
⎜ CRSS × V + (MAX) × f × ILOAD ⎟
⎟
⎜
IGATE
⎟
⎜
⎠
⎝
where C RSS is the reverse transfer capacitance of
N1/N3 and IGATE is the peak gate-drive source/sink
current.
For the synchronous rectifier, the worst-case power dissipation always occurs at maximum battery voltage:
⎛
VOUT _ ⎞
2
PD (N2 / N4) = ⎜1−
⎟ × ILOAD × RDS
⎝ V + (MAX ) ⎠
The absolute worst case for MOSFET power dissipation
occurs under heavy overloads that are greater than
ILOAD(MAX) but are not quite high enough to exceed
the current limit and cause the fault latch to trip. To
protect against this possibility, “overdesign” the circuit
to tolerate:
ILOAD = ILIMIT(HIGH) + (LIR / 2 ) x ILOAD(MAX)
where I LIMIT(HIGH) is the maximum valley current
allowed by the current-limit circuit, including threshold
tolerance and resistance variation.
Rectifier Selection
Generally, a small high-side MOSFET reduces switching losses at high input voltage. However, the RDS(ON)
required to stay within package power-dissipation limits
often limits how small the MOSFET can be. The optimum situation occurs when the switching (AC) losses
equal the conduction (RDS(ON)) losses.
Switching losses in the high-side MOSFET can become
an insidious heat problem when maximum battery voltage is applied, due to the squared term in the CV2 ✕ f
switching loss equation. Reconsider the high-side
MOSFET chosen for adequate RDS(ON) at low battery
voltages if it becomes extraordinarily hot when subjected to V+(MAX).
Calculating the power dissipation in N1/N3 due to
switching losses is difficult since it must allow for quantifying factors that influence the turn-on and turn-off
times. These factors include the internal gate resistance, gate charge, threshold voltage, source induc-
Current circulates from ground to the junction of both
MOSFETs and the inductor when the high-side switch is
off. As a consequence, the polarity of the switching
node is negative with respect to ground. This voltage is
approximately -0.7V (a diode drop) at both transition
edges while both switches are off (dead time). The drop
is IL x RDS(ON) when the low-side switch conducts.
The rectifier is a clamp across the synchronous rectifier
that catches the negative inductor swing during the dead
time between turning the high-side MOSFET off and the
synchronous rectifier on. The MOSFETs incorporate a
high-speed silicon body diode as an adequate clamp
diode if efficiency is not of primary importance. Place a
Schottky diode in parallel with the body diode to reduce
the forward voltage drop and prevent the N2/N4 MOSFET
body diodes from turning on during the dead time.
Typically, the external diode improves the efficiency by
1% to 2%. Use a Schottky diode with a DC current rating
equal to one-third of the load current. For example, use
______________________________________________________________________________________
27
MAX1777/MAX1977/MAX1999
Power MOSFET Selection
Most of the following MOSFET guidelines focus on the
challenge of obtaining high load-current capability
(>5A) when using high-voltage (>20V) AC adapters.
Low-current applications usually require less attention.
Choose a high-side MOSFET (N1/N3) that has conduction losses equal to the switching losses at the typical
battery voltage for maximum efficiency. Ensure that the
conduction losses at the minimum input voltage do not
exceed the package thermal limits or violate the overall
thermal budget. Ensure that conduction losses plus
switching losses at the maximum input voltage do not
exceed the package ratings or violate the overall thermal budget.
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
an MBR0530 (500mA-rated) type for loads up to 1.5A, a
1N5819 type for loads up to 3A, or a 1N5822 type for
loads up to 10A. The rectifier’s rated reverse breakdown
voltage must be at least equal to the maximum input voltage, preferably with a 20% derating factor.
Boost Supply Diode
A signal diode, such as a 1N4148, works well in most
applications. Use a small (20mA) Schottky diode for
slightly improved efficiency and dropout characteristics, if the input voltage can go below 6V. Do not use
large power diodes, such as 1N5817 or 1N4001, since
high-junction capacitance can force LDO5 to excessive
voltages.
Applications Information
Dropout Performance
The output voltage-adjust range for continuous-conduction operation is restricted by the nonadjustable 350ns
(max) minimum off-time one-shot. Use the slower 5V
SMPS for the higher of the two output voltages for best
dropout performance in adjustable feedback mode. The
duty-factor limit must be calculated using worst-case values for on- and off-times, when working with low input
voltages. Manufacturing tolerances and internal propagation delays introduce an error to the tON K factor. Also,
keep in mind that transient response performance of
buck regulators operated close to dropout is poor, and
bulk output capacitance must often be added (see the
VSAG equation in the Output Capacitor Selection section).
The absolute point of dropout occurs when the inductor
current ramps down during the minimum off-time
(∆IDOWN) as much as it ramps up during the on-time
(∆IUP). The ratio h = ∆IUP/∆IDOWN indicates the ability to
slew the inductor current higher in response to
increased load, and must always be greater than 1. As
h approaches 1, the absolute minimum dropout point,
the inductor current is less able to increase during each
switching cycle and VSAG greatly increases unless
additional output capacitance is used.
A reasonable minimum value for h is 1.5, but this may
be adjusted up or down to allow tradeoffs between
V SAG , output capacitance, and minimum operating
voltage. For a given value of h, the minimum operating
voltage can be calculated as:
V + (MIN) =
28
(VOUT _ + VDROP1)
⎞
⎛t
OFF (MIN) × h
⎟
1− ⎜
⎟
⎜
K
⎠
⎝
where VDROP1 and VDROP2 are the parasitic voltage
drops in the discharge and charge paths (see the OnTime One-Shot section), tOFF(MIN) is from the EC table,
and K is taken from Table 2. The absolute minimum
input voltage is calculated with h = 1.
Operating frequency must be reduced or h must be
increased and output capacitance added to obtain an
acceptable VSAG if calculated V+(MIN) is greater than
the required minimum input voltage. Calculate VSAG to
be sure of adequate transient response if operation
near dropout is anticipated.
Dropout Design Example
MAX1977: With VOUT5 = 5V, fsw = 400kHz, K = 2.25µs,
tOFF(MIN) = 350ns, VDROP1 = VDROP2 = 100mV, and h = 1.5,
the minimum V+ is:
V + (MIN) =
(5V + 0.1V)
⎛ 0.35µs × 1.5 ⎞
1− ⎜
⎟
2.25µs ⎠
⎝
+ 0.1V − 0.1V = 6.65V
Calculating with h = 1 yields:
V + (MIN) =
(5V + 0.1V)
⎛ 0.35µs × 1⎞
1− ⎜
⎟
⎝ 2.25µs ⎠
+ 0.1V − 0.1V = 6.04 V
Therefore, V+ must be greater than 6.65V. A practical
input voltage with reasonable output capacitance
would be 7.5V.
MAX1658/
MAX1659
V+
LDO
DH_
MAX1777
MAX1977
MAX1999
DL_
T1
10µH
1:2.2
+ VDROP2 − VDROP1
T1 = TRANSPOWER TECHNOLOGIES TTI-5870
Figure 12. Transformer-Coupled Secondary Output
______________________________________________________________________________________
12V
POSITIVE
SECONDARY
OUTPUT
5V
MAIN
OUTPUT
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
A coupled inductor or transformer can be substituted for
the inductor in the 5V or 3.3V SMPS to create an auxiliary
output (Figure 12). The MAX1777/MAX1977/MAX1999 are
particularly well suited for such applications because they
can be configured in ultrasonic or forced-PWM mode to
ensure good load regulation when the main supplies are
lightly loaded. An additional postregulation circuit can be
used to improve load regulation and limit output current.
The power requirements of the auxiliary supply must be
considered in the design of the main output. The transformer must be designed to deliver the required current
in both the primary and the secondary outputs with the
proper turns ratio and inductance. The power ratings of
the synchronous rectifier MOSFETs and the current limit
in the MAX1777/MAX1977/MAX1999 must also be
adjusted accordingly. Extremes of low input-output differentials, widely different output loading levels, and high
turns ratios can further complicate the design due to parasitic transformer parameters such as interwinding
capacitance, secondary resistance, and leakage inductance. Power from the main and secondary outputs is
combined to get an equivalent current referred to the
main output. Use this total current to determine the current limit (see the Determining the Current Limit section):
I TOTAL = PTOTAL / VOUT
where ITOTAL is the equivalent output current referred
to the main output, and PTOTAL is the sum of the output
power from both the main output and the secondary
output:
LPRIMARY =
N=
VOUT (VIN(MAX) − VOUT
VIN(MAX) × ƒ × I TOTAL × LIR
VSEC + VFWD
VOUT(MIN) + VRECT
where LPRIMARY is the primary inductance, N is the
transformer turns ratio, VSEC is the minimum required
rectified secondary voltage, VFWD is the forward drop
across the secondary rectifier, VOUT(MIN) is the minimum
value of the main output voltage, and VRECT is the onstate voltage drop across the synchronous rectifier
MOSFET. The transformer secondary return is often connected to the main output voltage instead of ground in
order to reduce the necessary turns ratio. In this case,
subtract V OUT from the secondary voltage (V SEC –
VOUT) in the transformer turns ratio equation above.
The secondary diode in coupled-inductor applications
must withstand flyback voltages greater than 60V, which
usually rules out most Schottky rectifiers. Common silicon rectifiers, such as the 1N4001, are also prohibited
because they are too slow. This often makes fast silicon
rectifiers such as the MURS120 the only choice. The flyback voltage across the rectifier is related to the VIN VOUT difference, according to the transformer turns ratio:
VFLYBACK = VSEC + (VIN - VOUT) ✕ N
where N is the transformer turns ratio (secondary windings/primary windings), VSEC is the maximum secondary
DC output voltage, and VOUT is the primary (main) output voltage. If the secondary winding is returned to VOUT
instead of ground, subtract VOUT from VFLYBACK in the
equation above. The diode’s reverse breakdown voltage
rating must also accommodate any ringing due to leakage inductance. The diode’s current rating should be at
least twice the DC load current on the secondary output.
The optional linear postregulator must be selected to
deliver the required load current from the transformer’s
rectified DC output. The linear regulator should be configured to run close to dropout to minimize power dissipation and should have good output accuracy under
those conditions. Input and output capacitors are chosen to meet line regulation, stability, and transient
requirements. There are a wide variety of linear regulators appropriate for this application; consult the specific
linear-regulator data sheet for details.
Widely different output loads effect load regulation. In
particular, when the secondary output is left unloaded
while the main output is fully loaded, the secondary output capacitor may become overcharged by the leakage
inductance, reaching voltages much higher than intended. In this case, a minimum load or overvoltage protection may be required on the secondary output to protect
any device connected to this output.
PC Board Layout Guidelines
Careful PC board layout is critical to achieve minimal
switching losses and clean, stable operation. This is
especially true when multiple converters are on the
same PC board where one circuit can affect the other.
The switching power stages require particular attention
(Figure 13). Refer to the MAX1999 EV kit data sheet for
a specific layout example.
Mount all of the power components on the top side of
the board with their ground terminals flush against one
another, if possible. Follow these guidelines for good
PC board layout:
• Isolate the power components on the top side from
the sensitive analog components on the bottom side
with a ground shield. Use a separate PGND plane
under the OUT3 and OUT5 sides (called PGND3 and
______________________________________________________________________________________
29
MAX1777/MAX1977/MAX1999
Use of Coupled Inductors to Create
Auxiliary Outputs
USE PGND PLANE TO:
USE AGND PLANE TO:
- BYPASS LDO_
- BYPASS VCC AND REF
- TERMINATE EXTERNAL FB
- CONNECT PGND TO THE TOPSIDE STAR GROUND
DIVIDER (IF USED)
OUT5
VIA BETWEEN POWER
- TERMINATE RILIM
AND ANALOG GROUND
(IF USED)
AGND
VIA TO OUT5
- PIN-STRAP CONTROL
INPUTS
PGND
L1
VIA TO LX5
CONNECT PGND TO AGND
BENEATH THE CONTROLLER AT
ONE POINT ONLY AS SHOWN.
C3
C4
VIA TO OUT3
C1
N4
C2
N3
VIAS TO GROUND
OUT3
N2
D2
ANALOG GROUND
PLANE ON INNER LAYER
VIA TO PGND
GROUND
D1
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
L2
N1
V+
VIA TO LX3
NOTE: EXAMPLE SHOWN IS FOR DUAL N-CHANNEL MOSFET.
Figure 13. PC Board Layout Example
PGND5). Avoid the introduction of AC currents into
the PGND3 and PGND5 ground planes. Run the
power plane ground currents on the top side only, if
possible.
• Use a star ground connection on the power plane to
minimize the crosstalk between OUT3 and OUT5.
• Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable, jitter-free operation.
• Keep the power traces and load connections short.
This practice is essential for high efficiency. Using
thick copper PC boards (2oz vs. 1oz) can enhance
full-load efficiency by 1% or more. Correctly routing
PC board traces must be approached in terms of
fractions of centimeters, where a single milliohm of
excess trace resistance causes a measurable efficiency penalty.
• CS_ (MAX1777/MAX1977) / LX_ (MAX1999) and GND
connections to the synchronous rectifiers for current
limiting must be made using Kelvin sense connections to guarantee the current-limit accuracy. With 8pin SO MOSFETs, this is best done by routing power
to the MOSFETs from outside using the top copper
layer, while connecting CS_/LX_ traces inside (underneath) the MOSFETs.
• When trade-offs in trace lengths must be made, it is
preferable to allow the inductor charging path to be
made longer than the discharge path. For example,
is better to allow some extra distance between the
input capacitors and the high-side MOSFET than to
30
allow distance between the inductor and the synchronous rectifier or between the inductor and the
output filter capacitor.
• Ensure that the OUT_ connection to COUT_ is short and
direct. However, in some cases it may be desirable to
deliberately introduce some trace length between the
OUT_ connector node and the output filter capacitor
(see the Stability Considerations section).
• Route high-speed switching nodes (BST_, DH_, LX_,
and DL_) away from sensitive analog areas (REF,
ILIM_, and FB_). Use PGND3 and PGND5 as an EMI
shield to keep radiated switching noise away from the
IC’s feedback divider and analog bypass capacitors.
• Make all pin-strap control input connections (SKIP,
ILIM_, etc.) to GND or VCC of the device.
Layout Procedure
1) Place the power components first with ground terminals adjacent (N2/N4 source, CIN_, COUT_, D1
anode). If possible, make all these connections on
the top layer with wide, copper-filled areas.
2) Mount the controller IC adjacent to the synchronous
rectifier MOSFETs, preferably on the back side in
order to keep DH_, GND, and the DL_ gate drive
lines short and wide. The DL_ gate trace must be
short and wide measuring 50mils to 100mils wide if
the MOSFET is 1in from the controller device.
3) Group the gate-drive component (BST_ diode and
capacitor, V+ bypass capacitor) together near the
controller device.
______________________________________________________________________________________
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
5) On the board’s top side (power planes), make a
star ground to minimize crosstalk between the two
sides. The top-side star ground is a star connection
of the input capacitors and synchronous rectifiers.
Keep the resistance low between the star ground
and the source of the synchronous rectifiers for
accurate current limit. Connect the top-side star
ground (used for MOSFET, input, and output
capacitors) to the small island with a single short,
wide connection (preferably just a via).
Create PGND islands on the layer just below the
top-side layer (refer to the MAX1777 EV kit for an
example) to act as an EMI shield if multiple layers
are available (highly recommended). Connect each
of these individually to the star ground via, which
connects the top side to the PGND plane. Add one
more solid ground plane under the device to act as
an additional shield, and also connect the solid
ground plane to the star ground via.
Pin Configurations (continued)
TOP VIEW
CS3 1
28 BST3
PGOOD 2
27 LX3
ON3 3
26 DH3
ON5 4
25 LDO3
ILIM3 5
SHDN 6
24 DL3
MAX1777
MAX1977
23 GND
FB3 7
22 OUT3
REF 8
21 OUT5
FB5 9
20 V+
PRO 10
19 DL5
ILIM5 11
18 LDO5
SKIP 12
17 VCC
CS5 13
16 DH5
BST5 14
15 LX5
QSOP
Chip Information
TRANSISTOR COUNT: 8335
PROCESS: BiCMOS
6) Connect the output power planes (VCORE and system
ground planes) directly to the output filter capacitor
positive and negative terminals with multiple vias.
______________________________________________________________________________________
31
MAX1777/MAX1977/MAX1999
4) Make the DC-DC controller ground connections as
follows: near the device, create a small analog
ground plane. Connect the small analog ground
plane to GND (Figure 13) and use the plane for the
ground connection for the REF and VCC bypass
capacitors, FB dividers, and ILIM resistors (if any).
Create another small ground island for PGND, and
use the plane for the V+ bypass capacitor, placed
very close to the device. Connect the AGND and
PGND planes together at the GND pin of the device.
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
QSOP.EPS
MAX1777/MAX1977/MAX1999
High-Efficiency, Quad Output, Main PowerSupply Controllers for Notebook Computers
PACKAGE OUTLINE, QSOP .150", .025" LEAD PITCH
21-0055
E
1
1
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