AN3358 Application note Turbo2 600 V diodes: optimized solutions for PFC and other applications Introduction In a switched mode power supply, there are a great number of electronic functions where 600 V ultrafast diodes are used. Each diode has a specific function. In one application a parameter can be critical but secondary in another. A rectifier manufacturer who wants to propose an optimized solution for each function needs to develop several families with different trade-offs (mainly between the forward voltage VF and reverse recovery charge Qrr). STMicroelectronics’ Turbo2 600 V ultrafast diodes offer three different families in order to offer an optimal solution for each application. After some general information about this new technology, a discussion of the PFC application, working in continuous mode, transition mode and fixed-off-time, is presented. In the case of continuous mode operation, hard switching and soft switching conditions are considered. Some other conventional functions are also touched upon. September 2011 Doc ID 018581 Rev 1 1/18 www.st.com Contents AN3358 Contents 1 2 General information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 1.1 Technology information . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 1.2 VF, Qrr trade-off for the three families . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 3 1.3 Platinum doping and low leakage current . . . . . . . . . . . . . . . . . . . . . . . . . 3 Main applications of 600 V ultrafast diodes . . . . . . . . . . . . . . . . . . . . . . 5 2.1 2.2 Power factor corrector applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 2.1.1 Boost diode in PFC working in continuous mode . . . . . . . . . . . . . . . . . . 5 2.1.2 Boost diode in PFC working in transition mode . . . . . . . . . . . . . . . . . . . 10 2.1.3 Boost diode working in fixed-off-time (FOT) PFC . . . . . . . . . . . . . . . . . 12 Other applications . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 13 3 Conclusion . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 16 4 References . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 5 Revision history . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 17 2/18 Doc ID 018581 Rev 1 AN3358 General information 1 General information 1.1 Technology information The Turbo2 families are manufactured using simple rules to insure high quality and reliability. These diodes are planar structures on epitaxial layers. The wafers are thus subjected to reduced mechanical stress for planar diodes compared to mesa ones. The use of epitaxial layers makes the VF/trr trade-off independent of the wafer thickness, the contrary of homogenous diodes. These properties make the manufacturing of large diameter wafers possible. So the wafers benefit from state-of-the-art technology on recent equipment. Epitaxial diodes, which present good drift area thickness, are particularly suitable for diodes up to 600 V and exhibit a significantly superior VF/trr trade-off. The lifetime control of the carriers for the Turbo2 diodes is obtained through platinum (Pt) doping. Pt doping is required for high junction temperature applications because it results in low reverse current at elevated temperature and, in this way, presents a low thermal runaway risk. 1.2 VF, Qrr trade-off for the three families The three families are: STTHxxR06 (R stands for rapid with low Qrr), STTHxx06 (medium VF and Qrr), and STTHxxL06 (Low forward voltage). Figure 1 shows where a trade-off occurs in three operational areas. A technology using gold doping is also shown. Figure 1. VF - Qrr trade-off for an 8 A diode 1.7 VF (V) Typical values R Family 1.2 IF = 8 A VR = 400 V dIF/dt = 200 A/µs Tj = 125 °C Gold doping Medium Family Platinum doping L Family Qrr (nC) 0.7 0 1.3 200 400 600 800 1000 1200 1400 1600 Platinum doping and low leakage current Figure 2 shows the trade-off between leakage current IR and Qrr in several operational areas. The faster the diode, the higher the IR is. This rule is true for both gold and platinum doping. For the same Qrr, IR is approximately 100 times lower with platinum doping. The corresponding “R” family with gold doping would have a high maximum leakage current (18 mA at 125 °C and 400 V). As shown later in this Application note, with such a leakage current thermal instability can be reached for operating junction temperatures higher than 125 °C in a conventional application. Doc ID 018581 Rev 1 3/18 General information AN3358 It will also be shown that IR is also a critical parameter for diodes in axial and SMD packages. Figure 2. IR - Qrr trade-off in several operational areas for an 8 A diode 100000 IRmax (µA) IF = 8 A VR = 400 V dIF/dt = 200 A/µs Tj = 125 °C 10000 1000 R Family Gold doping Medium Family 100 L Family 10 Platinum doping Qrr typ (nC) 1 0 4/18 500 1000 Doc ID 018581 Rev 1 1500 2000 AN3358 2 Main applications of 600 V ultrafast diodes Main applications of 600 V ultrafast diodes This section discusses the trade-offs in a common application. Boost power factor corrector (PFC) will be widely covered since it is a major application. A typical PFC circuit is shown in Figure 3. 2.1 Power factor corrector applications Figure 3. Boost power factor corrector circuit Vmains L Dboost Vout IRM Vgate 2.1.1 Boost diode in PFC working in continuous mode Hard switching conditions PFC applications are mainly designed in continuous mode when the power is greater than 200 W. In such an application, it is well known that the greatest losses due to the diode are the switching losses in the transistor (Pontr) when it turns on. The reverse recovery current (IRM) of the boost diode flows into the MOSFET (Figure 3). Consequently, the best choice in most cases is the “R” family. Switching losses due to IRM depend mainly on two parameters: the operating junction temperature Tj and the mains voltage Vmains. Figure 4 and Figure 5 show that the switching losses for STTH8R06 quickly increase when Tj increases and when Vmains decreases. These curves are drawn with a software tool realized by these authors. If the PFC only works on 240 V mains, with a low operating junction temperature, switching losses will be less critical and the best trade-off could be the intermediate trade-off: STTHxx06. However, most PFCs are designed to work in a wide mains voltage range (85 V-264 V) with an operating junction temperature (in the worst case) close to 100 °C. The “R” family will be the family usually recommended. Doc ID 018581 Rev 1 5/18 Main applications of 600 V ultrafast diodes Figure 4. AN3358 Switching losses versus Tj at turn off of the diode 14 Poff diode + Pontr due to the diode (W) STTH8R06D 12 10 Vmains = 90 V dI/dt = 400 A/µs L = 0.5 mH Fsw = 100 kHz VOUT = 400 V POUT = 400 W 8 6 4 2 Tj (°C) 0 0 Figure 5. 25 50 75 100 125 150 175 Switching losses versus Vmains at turn off of the diode 12 Poff diode + Pontr due to the diode (W) STTH8R06D 10 8 Tj = 100 °C dI/dt = 400 A/µs L = 0.5 mH Fsw = 100 kHz VOUT = 400 V POUT = 400 W 6 4 2 Vmains (V) 0 0 6/18 50 100 150 Doc ID 018581 Rev 1 200 250 300 AN3358 Main applications of 600 V ultrafast diodes Tjmax before thermal runaway The maximum junction temperature Tjmax before thermal runaway can be calculated using Equation 1, Equation 2 then Equation 3. Equation 1 δ=1- 2 Vmains peak π VOUT Equation 2 IR(VOUT, Tjmax) = 1 VOUT · δ · c Rth(j-a) Equation 3 Tjmax = 125 + 1 c · loge ( IR(VOUT, Tjmax) IRmax(VOUT, 125 °C) ( Where: ● δ is the average duty cycle of the blocking time of the diode given by Equation 3. ● VOUT is the output voltage. ● c is a constant with units of °C-1. Each diode has its own “c” coefficient depending on the technology of the diode and the reverse voltage VR applied. It can be determined from Equation 3 for two values of leakage current corresponding to application reverse voltage Vout, for example: IR(Vout,100 °C) and IR(Vout,125 °C). ● Rth(j-a) is the thermal resistance between junction and ambient (heatsink + diode). With the following conditions: VOUT = 400 V, c ≈ 0.055 °C-1 (for the “R” family) Vmains = 85 V, δ = 0.8, Rth(j-a) = 10 °C/W Figure 2 gives IRmax(400 V, 125 °C) = 215 µA for an 8 A “R” family diode and 17 mA for the equivalent diode in gold doping. Equation 2 and Equation 3 give Tjmax = 184 °C for Turbo2 and 104 °C for the equivalent diode in gold doping. Soft switching condition Designers can use a number of techniques to turn on the MOSFET in soft switching conditions and reduce the switching losses due to IRM. Figure 6 and Figure 7 show two solutions, widely used with the associated waveforms during switching time. In the non-dissipative circuit Figure 6, the smaller transistor T2 turns on before the main one T1. The dl/dt when Dboost turns off is controlled by Lr (dl/dt = Vout/Lr), and T1 turns on at zero current. Consequently, the switching-on losses will be close to zero. With this circuit, the reverse recovery current of the boost diode is less critical. The best choice, following the application conditions (switching frequency, Lr…) will be “the intermediate” or the “L” trade-off. Doc ID 018581 Rev 1 7/18 Main applications of 600 V ultrafast diodes Figure 6. AN3358 Non-dissipative soft switching solution Vmains Dboost L Vout Lr Dr Cr T1 T2 T1, T2 T1on I DBoost I0 t T 2on ⎛ VOUT ⎞ ⎜ L r ⎟⎠ ⎝ I Lr t I RM I 0+I RM+I res I 0+IRM I Dr t VC r t t I T1 t I RM+I res 8/18 Doc ID 018581 Rev 1 AN3358 Main applications of 600 V ultrafast diodes The topology shown in Figure 7 is more simple but more dissipative than that in Figure 6. The waveforms in Figure 7 show the MOSFET turning on at zero current, thus reducing the switching losses. When the diode turns off, the Lr inductor is charged with the reverse recovery current of the boost diode. This energy will be dissipated in the resistor. The higher IRM is the higher the losses in the resistor are. In this application IRM is more critical than in the previous one. The best choice for the boost diode trade-off will be “R” or medium family depending on the application conditions. Figure 7. Dissipative soft switching solution Dr Vmains L VRC DBoost VOUT Lr VDS T 450 V 20 A IT 250 V 10 A VDS 0A 0V -250 V 180.0 60 V IDBoost IRM Qrr 180.1 180.2 180.3 180.4 t (µs) 180.5 180.6 180.7 VRC IDr 40 V -10 A 180.8 20 A 10 A IDBoost 20 V 0A 0V 180.0 ILr 180.1 180.2 180.3 t (µs) 180.4 180.5 180.6 180.7 -8 A 180.8 Another very interesting alternative soft switching solution is described in the application note AN3276, “ST solution for efficiency improvement in PFC applications, back current circuit (BC2)”. AN3276 presents a patented soft switching circuit from STMicroelectronics offering performance similar to that of SiC Schottky diodes. Doc ID 018581 Rev 1 9/18 Main applications of 600 V ultrafast diodes 2.1.2 AN3358 Boost diode in PFC working in transition mode The transition mode (TM) is widely used for low power PFC (<200 W). The particularity of this control mode working between continuous and transition mode is a simple control and a few external components. This control mode results in variable frequency operation and a constant on time of the MOSFET. Consequently, the current flowing through the Boost inductor is triangular (Figure 8). It increases through the MOSFET following the slope defined by Vmains/L, and decreases through the diode following a low dl/dt given in Equation 4. Equation 4 dI dt = Vmains - VOUT L In this case dI/dt may have a value up to 0, the necessary condition for the next cycle. Figure 8. Inductor current waveform and MOSFET timing IPK = 2 2x POUT V mains Average input current Inductor current Vmains L Vmains-Vout L T= N 1 = 2 · Fmains FSW On MOSFET Off ton fixed TSW variable The ZCD circuit (zero current detection) turns on the MOSFET at zero current, avoiding high switching losses in the MOSFET due to the recovery charge of the diode. Unlike the continuous mode, the Qrr of the diode it is not the key parameter any more. In the transition mode, the main losses of the diode are due to the forward voltage. It is then possible to optimize the VF parameter to the detriment of Qrr, due to the low dl/dtoff of the diode (<1 A/µs) fixed by the inductor. 10/18 Doc ID 018581 Rev 1 AN3358 Main applications of 600 V ultrafast diodes Nevertheless, an accurate study at switch-off of the diode shows that the Qrr parameter cannot be indefinitely relaxed. Figure 9 highlights this phase when the current of the diode reaches 0, and shows that this time is composed of 3 phases: ● Phase 1 [t0,t1]: The diode is open. There is a resonant circuit between the equivalent capacitance (Cds MOS + Cj diode) and the boost inductance, which has as its initial condition the IRM of the diode. ● Phase 2 [t1,t2]: VDS reaches 0 and the body diode of the MOSFET enters in conduction and the current linearly increases through the VF of the body diode. ● Phase 3 at t2: The ZCD circuit turns the MOSFET on and the current continues to linearly rise through the RDS(on). Figure 9. Switch-off comparison between STTH1L06 and a slower diode t 0 t1 t2 IRM Vds V ds Slower diode Vgrille IDiode IMos STTH1L06 It can be observed that the dead time (t0,t2) increases with the IRM of the diode. This time a negative current flows through the power MOSFET and is the source of additional losses. This duration depends on the slope (versus Vmains, L) and also on the IRM of the diode (the initial condition of phase 1). During this time there is no power transferred to the load. In this way, with a very slow diode, the sum of the losses due to high IRM cannot be negligible compared to these of the conduction losses. Therefore, there is a limit for Qrr. This limit appears for the full range PFC at 110 V. In this condition the current in the power MOSFET takes more time to reach 0 (maximum dead time). The maximum Qrr of the “L” family has been optimized taking these considerations into account. According to the application conditions (Pout, Vmains, dI/dtmax, Fsw, Tj), the medium trade-off could be also considered. The optimum choice between low forward voltage trade-off (STTHxxL06) and the medium trade-off (STTHxx06) could be determined by efficiency measurement. In transition mode a diode with a small current rating is used. It is generally a small package (axial or SMD packages) with high thermal resistances. Consequently, the junction Doc ID 018581 Rev 1 11/18 Main applications of 600 V ultrafast diodes AN3358 temperature of the diode, which is mainly fixed by the conduction losses, can be high. Equation 2 in Section 2.1 shows that the thermal resistance is a critical parameter for the thermal runaway limit. Table 1 compares the thermal runaway limit between Turbo2 and a gold-doped diode working in a transition mode PFC in the following conditions: Rth(j-a) = 75 °C/W, c ≈ 0.072 °C-1, VOUT = 400 V, Vmains = 85 V, δ = 0.808 Table 1. Tjmax comparison between Turbo2 and gold doping diode STTH3L06 Gold Doping IRmax 125 °C, 400 V 15 µA 1.5 mA Tjmax before thermal runaway limit is reached. 176 °C 112 °C This comparison shows that gold-doped diodes are limited in high temperature. There is no thermal runaway risk when Turbo2 uses platinum doping. For all these reasons, in most cases, the “L” family is recommended for the PFC application working in transition mode. 2.1.3 Boost diode working in fixed-off-time (FOT) PFC In this third PFC operating mode, instead of maintaining the on-time fixed, such as TM PFC, the Toff is kept constant and the Ton is free to be changed in order to modulate the power drained from the source according to the load. This modulation method, is described in the Application note AN1792, “Design of fixed-offtime controller PFC pre-regulators with L6562”. As shown in Figure 10 in FOT mode, the PFC works in DCM and CCM modes along the line semi period. Figure 10. Inductor, switch and diode currents in a CCM FOT-controlled PFC stage CCM Inductor current peak envelope ILpk DCM DCM Low frequency inductor current Switch current Diode current ON OFF Switch θt 12/18 OFF OFF Doc ID 018581 Rev 1 π−θt AN3358 Main applications of 600 V ultrafast diodes In this operating mode, according to the application conditions the optimal diode will be the medium trade-off (VF/QRR) or the rapid trade-off (“R” family). The designer should make some measurements of efficiency to confirm the good trade-off diode in its application. 2.2 Other applications There are numerous other electronic functions, where 600 V ultrafast diodes are used. For example, rectification, demagnetization, snubber, bootstrapping, clamping, or East-West correction in a horizontal deflection circuit for TV or monitor (Figure 11). Figure 11. Traditional applications of 600 V ultrafast diodes Clamping diode Demagnetization diode Snubber diode Bootstrap diode Modulator diode in horizontal deflection circuit Doc ID 018581 Rev 1 13/18 Main applications of 600 V ultrafast diodes AN3358 It is not possible in this document to analyze each function in detail. We will focus on the clamping function used in flyback converters. The function of the clamping circuit is to protect the MOSFET against the overvoltage due to the energy in the leakage inductance of the transformer. The associated waveforms are represented in Figure 12. Figure 12. 600 V ultrafast diode waveforms in clamping function Vmains VLf m or DR VC VIN VOUT VDCL DCL VDS Breakdown voltage of the MOS transistor VDS VDS=V IN+VC+V FP VL f V IN+(VOUT/m) VIN VCEsat ID CL IDR VDCL ID CL IDR Qrr V FP V Rmax=V IN+VC+S pike 14/18 Doc ID 018581 Rev 1 AN3358 Main applications of 600 V ultrafast diodes When the MOSFET turns off, the inductive circuit opens and an overvoltage VLf appears in addition to the voltage across the primary inductor VOUT/m. The effect of this overvoltage turns on the clamping diode. Thus, the drain voltage is equal to VDS = VIN + VC +VFP. VFP is the peak forward voltage across the 600 V diode. VC is a DC voltage realized either by an RC circuit in parallel or by a clamping diode such as a Transil™. The first key parameter of the diode is VFP. VDS has to be lower than the breakdown voltage of the MOSFET. If VFP is too high the designer may be obliged to choose a higher voltage MOSFET (for example 800 V instead of 600 V). To avoid thermal runaway problems a low value of leakage current is necessary as the diode is normally a 1 A device in an SMD or axial package. A low IR will also contribute to the reduction of consumption in stand-by mode. The forward voltage is not a critical parameter because the diode conducts about ten nanoseconds every switching period. When the clamping voltage is made with a Transil, it is generally better to use an ultrafast type diode. When an RC solution is used, the capacitance is discharged through the reverse recovery current of the diode, thus reducing the losses in the resistor. The Turbo2 technology, which allows low leakage current and low peak forward voltage, is well suited for this application. The best trade-off with a Transil, will be the “R” or the medium family. With an RC solution the choice will generally be between the “L” and the “medium” families. TM: Transil is a trademark of STMicroelectronics Doc ID 018581 Rev 1 15/18 Conclusion 3 AN3358 Conclusion This Application note presents the main applications of the 600 V ultrafast diodes. These applications are numerous, each requiring a slightly different trade-off among the diode parameters. In order to propose an optimized solution for each one, three trade-offs are proposed by STMicroelectronics. There are some general rules to define the right trade-off. For example, the “R” family for PFC working in continuous mode and hard switching condition and the “L” family for PFC working in transition mode. However, there are also applications for which a deeper study will be necessary. An important benefit of the platinum doping implemented in the Turbo2 technology resides in the use of the diodes at high junction temperature without thermal runaway risk in normal prescribed condition of use (<175 °C). 16/18 Doc ID 018581 Rev 1 AN3358 4 References References [1] ST Application note AN628, “Designing a high power factor switching preregulator with the l4981 continuous mode” [2] PCIM, Nuremburg, 2000 “New solution to optimize diode recovery in PFC boost converter”, B. Rivet. [3] ST Application note AN667, “Designing a high power factor switching preregulator with the l6560 transition mode” [4] ST Application note AN966, “Enhanced transition mode power factor corrector” [5] ST Application note AN1792, “Design of fixed-off-time controller PFC preregulator with the L6562” [6] ST Application note AN3276, “ST solution for efficiency improvement in PFC applications, back current circuit (BC2)” 5 Revision history Table 2. Document revision history Date Revision 14- Sep-2011 1 Changes First issue Doc ID 018581 Rev 1 17/18 AN3358 Please Read Carefully: Information in this document is provided solely in connection with ST products. STMicroelectronics NV and its subsidiaries (“ST”) reserve the right to make changes, corrections, modifications or improvements, to this document, and the products and services described herein at any time, without notice. All ST products are sold pursuant to ST’s terms and conditions of sale. Purchasers are solely responsible for the choice, selection and use of the ST products and services described herein, and ST assumes no liability whatsoever relating to the choice, selection or use of the ST products and services described herein. No license, express or implied, by estoppel or otherwise, to any intellectual property rights is granted under this document. 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