APPLICATION NOTE A 500W HIGH POWER FACTOR WITH THE L4981A CONTINUOUS MODE IC The widespread use of passive AC/DC off-line converters causes low power factor and high line current harmonic distortion. To reduce these phenomena and to comply with relevant regulatory agency requirements , designers are employing active power factor correction in their off-line SMPS applications. This paper describes a practical, low cost and easy to implement 500W power factor corrected application that employs the L4981A Continuous Mode PFC IC. INTRODUCTION Reduction of line current harmonic distortion and improvement of power factor is of great concern to many designers of off-line switched mode power supplies. This concern has been motivated by present and impending regulatory requirements regarding line current harmonics. The reasons for improving power factor and reducing line current harmonic distortion are well known and understood. Active power factor correction using the boost topology and operating in the continuous inductor current control mode is an excellent method to comply with these requirements and is well accepted in the industry. This paper will present a practical power factor corrected design for a 500 Watt output and universal mains input application. The detailed derivations of all power, IC biasing and control component values and types will be shown. The evaluation results from an actual working demoboard will be presented as well as several relevant oscillograms. DESIGN SPECIFICATIONS The design specifications given below are realized by the implementation of a functional demoboard. The design target specifications are as follows: - Universal mains input AC voltage Virms = 88Vac to 264Vac, 60/50Hz - DC regulated output voltage Vout = 400Vdc - Full load output ripple voltage ∆Vripple = ±8V - Rated output power Pout = 500W - Maximum output overvoltage Vomax = 450V - Switching frequency fsw = 80kHz - Maximum inductor current ripple ∆IL = 23% - Input power factor PF > 0.99 - Input line current total harmonic distortion <5% AN827/1297 To meet these specifications, the selection of component values and material types is very important. The next sections will describe the component selection criteria along with some critical derivations. For detailed explanations on the controller operation and pin description, refer to Application Note AN628 Designing A High Power Factor Switching Preregulator With The L4981 Continuous Mode [1] and the corresponding Datasheet L4981A/B Power Factor Corrector [2]. POWER COMPONENTS SELECTION The power component values and types are derived and selected in the next section. Please refer to Figure 2, 500 Watt Demoboard Schematic. Input Diode Bridge The input diode bridge, D1, can be a standard slow-recovery type. The selection criteria include the maximum peak reverse breakdown voltage, maximum forward average current, maximum surge current and thermal considerations. Maximum peak reverse voltage: Vprv = Virmsmax ⋅ √ 2 ⋅ 1.2 (safety margin) = = 264V ⋅ √2 ⋅ 1.2 = 448V Therefore use a 600V rated diode. Maximum forward average current: Irmsmax = Ifave = POUT 500 = = 6.31A Vrms min ⋅ n 88 ⋅ 0.9 Irmsmax ⋅ √ 2 6.31 ⋅ √ 2 = = 2.84A π π The thermal considerations require the Ifave rating to be significantly higher than the value calculated. The part chosen has a Ifave of 25A. Addi1/16 APPLICATION NOTE tionally, a small heatsink is required to keep the case temperature within specification. Maximum surge current: There is a significant inrush current at start-up due to the large value bulk capacitor, C6, at the output. There is minimal impedance from the mains to this capacitor, thus at the peak of the input voltage waveform a large inrush current exists. This inrush current can be significantly reduced by some means of current limiting such as an NTC or triac/resistor combination. The input bridge diode’s maximum surge current rating must not be exceeded. This demoboard has a low cost and simple NTC for current inrush limiting. The efficiency can be improved by using the triac/resistor scheme, however the cost and complexity increases. Input Fuse The input fuse, F1, must open during severe current overloads without tripping during the transient inrush current condition or during normal operation. The fuse must have a current rating above the maximum continuous current (6.3Arms) that occurs at the low line voltage (88V). The fuse chosen for this demoboard has a continuous current rating of 10A/250VAC. Input Filter Capacitor The input filter capacitor, C3, is placed across the diode bridge output. This capacitor must smooth the high frequency ripple and must sustain the maximum instantaneous input voltage. In a typical application an EMI filter will be placed between the mains and the PFC circuit. This demoboard does not have the EMI filter except for this input capacitor. However, the evaluation results listed in Table 1 were made with an EMI filter placed between the mains input and the PFC circuit. The design of the EMI filter is not described here. The value of the input filter capacitor can be calculated as follows: Irms Cin > Kr 2 ⋅ π ⋅ fsw ⋅ r ⋅ Vrms min 6.31 Cin > 0.25 ⋅ = 0.59µF. 2 ⋅ π ⋅ 80k ⋅ 0.06 ⋅ 88 Where: Kr is the current ripple coefficient r = 0.02 to 0.08 The maximum value of this capacitor is limited to avoid line current distortion. The value chosen for this demoboard is 0.68µF. Output Bulk Capacitor The choice of the output bulk capacitor, C6, de2/16 pends on the electrical parameters that affect the filter performance and also on the subsequent application. Capacitance Value: The value shall be chosen to limit the output voltage ripple according to the following formula: Assume low ESR and ∆Vripple= ±8V Cout = Pout Pout = = 207µF 2π ⋅ 2f ⋅ ∆VO ⋅ VO 2π ⋅ 120 ⋅8 ⋅ 400 The value chosen is 330uf to ensure that the maximum specified voltage ripple is not exceeded. Although the ESR does not normally affect the voltage ripple, it has to be considered for the power losses due to the line and switching frequency ripple currents. It is important to verify that the low and high frequency ripple currents do not exceed the manufacturer’s specified ratings at the operating case temperature. Capacitors may be connected in parallel to decrease the equivalent ESR and to increase the ripple current handling capability. If a specific hold-up time is required, that is the capacitor has to deliver the supply voltage for a specified time and for a specified dropout voltage, then the capacitor value will be determined by the following equation: Cout = 2 ⋅ Pout ⋅thold Vo min 2 − Vop min 2 Where: Pout is the maximum output power Vomin is the minimum output voltage at max. load Vopmin is the minimum operating voltage before "power fail" detection thold is the required hold-up time Voltage Rating: The capacitor output voltage rating should not be exceeded under worst case conditions. The minimum voltage rating is calculated as follows: Vcap > Vout + ∆Vripple + Vmargin = 400 + 8 + 40 = 448V Where: Vout is the nominal regulated DC output voltage ∆Vripple is the ac voltage superimposed on the regulated DC output voltage ∆Vmargin is the allowance for tolerances in Vout and additional margin before OVP intervention The capacitor chosen has a voltage rating of 450VDC. The overvoltage trip level of Pin 3 (OVP) must be set below 450VDC. APPLICATION NOTE Power Mosfet The power mosfet, Q1, is used as the active switch due to its high frequency capability, ability to be driven directly from the controller and availability. The main criteria for its selection include the drain to source breakdown voltage (BVdss), delivered power and temperature considerations. Voltage Rating: The power mosfet has to sustain the maximum boosted output dc voltage according to the following equation: BVdss > Vout + ∆Vripple + Vmargin = 400 + 8 + 40 = 448V The power mosfet chosen has a BVdss of 500V. Power Rating: The main parameters to consider are Rdson and the thermal characteristics of the package and heatsink. The main losses in the power mosfet are the conduction and switching losses. The switching losses can be separated into two quantities, capacitive and crossover losses. The switching losses are dependent on the mosfet current di/dt. The maximum conduction (on-state) power losses can be calculated according to the following equations: IQrmsmax = = P out η⋅√ 2 V irms 500 ⁄ 0.9 ⋅ 2 ⋅ 88 √ ⋅ min √ min 16 ⋅ √ 2 ⋅ V irms 2– 3π ⋅ V out = 16 ⋅ √ 2 ⋅ 88 √ 2– 3 π ⋅ 400 IQrmsmax = 5.42A Ponmax = IQrms2max ⋅ R(DS)on max = 5.422 ⋅ 0.54 = 15.86W Where: IQrmsmax is the max. power mosfet rms current Virmsmin is the min. specified rms input voltage R(DS)on typ. = 0.27Ω at 25°C at 10A, VGS = 10V R(DS)on max = 0.54Ω at 100°C The capacitive switching losses at turn-on are calculated as follows: 1 Pcapacitive = (3.3 ⋅ Coss ⋅ Vout1.5 + Cext ⋅ Vout2) ⋅ 2 ⋅ fsw = 2W Where: Coss = 650pF is the mosfet drain capacitance at 25V Cext = 100pF is the equivalent stray capacitance of the layout and external parts The estimated crossover switching losses (turnon and turn-off) are calculated as follows: Pcrossover = Vout ⋅ IQrms ⋅ fsw ⋅ tcr + Prec = = 400 ⋅ 5.42 ⋅ 80k ⋅ 40ns + 1.5 = 8.43W Where: tcr is the crossover time Prec is the boost diode recovery power loss contribution To reduce the turn-off losses in the mosfet, an RCD turn-off snubber has been employed. The capacitor value is calculated as follows: C11 = IQ1pk ⋅ trise 8.92 ⋅ 40ns = = 892pF ∆Vout 400 Therefore, use C11 = 820pF, 1000VDC rating The resistors, R23-24, must dissipate the energy stored in the snubber capacitor upon turn-on of the power mosfet. The capacitor must fully discharge during the switching cycle. The time constant of the RC combination is determined as follows: R ≤ 1 1 ⋅ = 1524 10 fSW ⋅ C11 The power dissipated in the resistors, R23-24, is calculated as follows: 1 1 C11 ⋅ Vout2 ⋅ fsw = ⋅ 820pF ⋅ 4002 ⋅ 2 2 ⋅ 80k = 5.25W Pdiss = Therefore, use R23 = R24 = 510Ω, 3W rating. The power mosfet chosen is the SGS-THOMSON Part Number STW20NA50. This part has a BVdss = 500V, RDSon = 0.27Ω, and is in a TO-247 package. In order to keep the junction temperature at a safe level, the mosfet is attached to an AAVID Heatsink Part Number 61085 with a thermal resistance of 3.0°C/W. This will keep the mosfet junction temperature at a safe level at worst case conditions, low-line input voltage (88V) and full load (500W). The thermal resistance of the heatsink may need to decrease depending upon the ambient temperature, type of enclosure (vented or non-vented) and the method of cooling (natural or forced convection). Boost Diode The main criteria for the selection of the boost diode, D2, include the repetitive peak reverse breakdown voltage (Vrrm), average forward current (Ifave), reverse recovery time (trr) and thermal considerations. 3/16 APPLICATION NOTE Voltage Rating: The voltage rating of the boost diode is determined by the same equation as for the power mosfet. The value chosen is Vrrm = 600V. Current Rating: The power losses in the boost diode consist of the conduction and switching losses. The switching losses are a function of the reverse recovery time (trr) and output voltage (Vout) . The switching losses are negligible compared to the conduction losses if a suitable ultra fast recovery diode is chosen. The conduction power losses can be calculated as follows: Iout = Pout 500 = = 1.25A Vout 400 IDrms = Pin 2 V in rms min √ 2 ⋅ Vin rms min 16 ⋅ √ √ = 3.24A 3 ⋅π⋅V out ⋅ Rd = 1.15 ⋅ 1.25 + Pcond = Vto ⋅ Iout + 2 +3.24 ⋅ 0.043 = 1.89W IDrms2 Where: Vto = 1.15V is the threshold voltage of the diode Rd = 0.043Ω is the diode differential resistance The diode must sustain the average output current and also keep the power losses to a minimum in order to keep the diode junction temperature within acceptable limits. The switching losses can be significantly reduced if an ultra-fast diode is employed. Since this circuit operates in the continuous current mode, the mosfet has to recover the boost diode minority carrier charge at turn-on. Thus, a diode with a small reverse recover time, trr, must be used. This circuit employs the SGSTHOMSON Turboswitch Diode Part Number STTA806D. This part offers the best solution for the continuous current mode operation due to its very fast reverse recovery time, 25ns typical. This part has a breakdown voltage rating (Vrrm) of Vin (rms) 4/16 Vin(peak) IL(rms) Iin (rms) 600V, average forward current rating (Ifave) of 8A and reverse recovery time (trr) of 25ns. The diode is attached to the same heatsink as the power mosfet, Q1. The STTA806D is non-isolated thus requiring a thermal insulator with good heat transfer characteristics. The STTA806DI is an isolated package and can be attached directly to the heatsink. Silicone thermal grease may be applied to improve the thermal contact between the diode and heatsink. Boost Inductor The boost inductor, T1, design starts with defining the minimum inductance value, L, to limit the high frequency current ripple, ∆IL. The next step is to define the number of turns, air gap length, ferrite core geometry, size and type for the specified power level. Finally, the wire size and type are determined. In the continuous mode approach, the acceptable current ripple factor, Kr, can be considered between 10% to 35%. For this design, the maximum specified current ripple factor is 23%. The maximum current ripple occurs when the peak of the input voltage is equal to Vout/2. ∆ILmax Vout 400 = = 2.50A 4 ⋅ fSW ⋅ L 4 ⋅ 80k ⋅ 0.5mH Occurs at Vinpk = Vout/2 = 200V; Vinrms = 141V Vinpk (Vout − Vinpk ) For all other input voltages Vout ⋅ fsw ⋅ L ∆IL 2 ⋅ Pin √ 2 ⋅ ILrms = Kr = ; ILpk = √ 2 ⋅ ILpk Vinrms ∆IL = The minimum boost inductor value can be calculated as follows: Lmin = Vout 400 = = 0.5mH 4 ⋅ fsw ⋅ ∆ILmax 4 ⋅ 80kHz ⋅ 2.50 The Table shown below relates the current ripple to the input voltage. IL(peak) Current Ripple Kr 88 124 6.31 8.92 2.13 0.119 120 170 4.63 6.55 2.44 0.186 141 199 3.94 5.57 2.50 0.224 180 255 3.09 4.37 2.31 0.264 200 283 2.78 3.93 2.07 0.263 220 311 2.53 3.58 1.73 0.242 240 339 2.31 3.27 1.29 0.197 264 373 2.10 2.97 0.63 0.106 APPLICATION NOTE The number of turns, N, can be calculated according to the following formula: N= L ⋅ ILpk 0.5mH ⋅ 8.92mA = = 59 Turns Aeff ⋅ B max 211 ⋅ 10−6 m2 ⋅ 0.36T Where: L is the calculated inductance value to limit the ripple current, ∆IL. ILpk is the worst case inductor current occurring at low-line input voltage (88V) Aeff is the effective cross-sectional area of the core Bmax is the maximum allowable flux density of the core The air gap is determined by referring to the magnetic core manufacturer’s AL vs. air gap curves. The air gap needed for the specified inductance, turns and core type is found to be 2.8mm in the center post. To approximate the minimum core size needed for the conversion, the following equation may be used: Volume ≥ K ⋅ L [ILpk ⋅ (ILpk + ∆IL)] Where K is the specific energy constant that depends on the ratio of the gap length (lgap) and the effective length (leff) of the core set and the maximum ∆B swing. Practically, K can be estimated as follows: K = 11.5 leff 114 = 11.5 ⋅ = 468 lgap 2.8 Thus, we have the following calculation for the minimum core set volume in cm3: Volume ≥ 468 ⋅ 0.5 ⋅ 10-3 [8.92 ⋅ (8.92 + 2.5)] = 23.8 cm3. The core chosen for this design is an ETD geometry ferrite core set with the following characteristics: Core type ETD4916A Effective core volume = 24.0 cm3. Effective magnetic path length = 114 mm Effective core area = 211 mm2 Ferrite material is 3C85 or equivalent Np = 59T Ns = 5T The ETD geometry has the following advantages: 1. Round center post for ease of winding 2. Commercially available from Philips, Siemens, Thomson, Magnetics, etc.. 3. Increased winding area 4. The center leg area is equal to the sum of the areas of the two external legs. The legs are working with the same flux density The wire size is determined by the maximum copper losses allowed and available winding area. For this design the wire size selected was 30AWG, 30 strand Litz. An auxiliary winding is used to supply power to the controller. The number of turns was determined experimentally to be 5. The worst case conditions for the auxiliary winding power supply voltage are at low-line input voltage (88V) and full load (500Watts) and at high-line input voltage (264V) and light-load. The auxiliary winding must supply sufficient voltage to prevent turn-off (UVLO) during normal operation and also must not supply excessive voltage causing burn-out of the controller. CoilCraft Part Number R4849-A meets the above specifications and is available. IC BIASING AND CONTROL COMPONENTS SELECTION The IC biasing and control component values are derived and selected in the next section. Please refer to Figure 2, 500 Watt Demoboard Schematic. Pin 1 P-GND (Power stage ground) This pin should be connected to the source of the power mosfet, Q1, with a short length and wide copper trace on the printed circuit board to minimize the copper trace resistance and inductance. Refer to Figure 3, 500 Watt Demoboard printed circuit board layout. Pin 2 IPK (Overcurrent protection input) In order to obtain a very precise overcurrent protection trip level, R12 and R13 are calculated as follows: Iaux = R12 = Vref 5.1 = = 1mA R13 5.1k Rsense ⋅ Ipeak 0.033 ⋅ 17 = = 561Ω Iaux 0.001 Use R12 = 562 ohms, R13 = 5.1k The peak current threshold is set at 17A and Rsense is chosen as 0.033 ohms. Pin 3 OVP (Overvoltage protection input) The overvoltage protection trip level is determined by the voltage divider across the output bulk capacitor, C6. The resistor values R11, R21 and R22 are calculated as follows: R21 + R22 Vout + ∆Vout 400 + 47 909k + 909k = −1 = −1 = Vref R11 5.1 21k 5/16 APPLICATION NOTE Where ∆Vout = 47V is the maximum overvoltage limit. The overvoltage limit selection is dependent upon the voltage rating of the output bulk capacitor (450VDC) and the power mosfet (500BVdss). Care must be taken that the level is not set too low, thus causing false tripping of the OVP. Pin 4 IAC (AC current input) This pin must be connected through resistors R1 and R2 to the rectified line to drive the multiplier with a current IIAC proportional to the instantaneous line voltage as shown below: IIAC (88V) = Vinpk 2 ⋅ 88 √ = = 77µA R1 + R2 806k + 806k IIAC(264V) = 2 ⋅ 264 √ = 231µA 806k + 806k Thus IIAC ranges from 77µA to 231µA. The relationship between IIAC and multiplier output current, Imult, is described in section Pin 8 (MULTOUT). Pin 5 CA-OUT (Current amplifier output) The current amplifier output delivers its signal to the PWM comparator. An external network defines the suitable loop gain to process the multiplier output and the inductor current signals. To avoid oscillation problems, the maximum inductor downslope (Vout/L) must be lower than the oscillator ramp-slope (Vsrp*fsw). The current amplifier high frequency gain can be described as follows: Gca = Vsrp ⋅ fsw ⋅ L 5.0 ⋅ 80k ⋅ 0.5m R15 +1 ≤ = R14 400 ⋅ 0.033 Vout ⋅ Rsense Where: Vsrp = 5.0V is the oscillator ramp peak-peak voltage Gca is the current amplifier gain fsw = 80kHz is the switching frequency Rsense = 0.033Ω is the parallel combination of R30-32 Thus, use R14=R16=2.7k, and R15=36K. To define the value of the compensation capacitor, C9, it is useful to consider the open loop current gain, defined by the ratio of the voltage across the sense resistor and the current amplifier output voltage. The crossover frequency is given by the following equation: fc = fsw 80k = = 12.7kHz 2⋅π 2⋅π To ensure a good phase margin, the zero fre6/16 quency, fz, should equal approximately fc/2. fsw 1 = therefore, 4π 2 ⋅ π ⋅ C9 ⋅ R15 2 = 692pF C9 = R15 ⋅ fsw fz = Use C9 = 680pF Pin 6 LFF (Load feed-forward input) This pin allows the modification of the multiplier output current proportionally to the load in order to improve the load transient response time. This function is not used in this circuit and the pin is connected to VREF. Pin 7 VRMS (Voltage input) This function is very useful for universal input mains applications to compensate the gain variation related to the input voltage change. This pin is connected through an external network to the rectified line input. The best control is achieved when the VRMS voltage level is in the range of 1.5 to 5.5V. To avoid the rectified mains line ripple (2f), a two pole low-pass filter is realized with R3-R6 and C12. The lowest pole is set near 3Hz and the highest pole near 13 Hz to reduce the gain to -80dB at 100 Hz. R3 Vrmspin7 = Vrmsline + + +R6 R3 R4 R5 fpole1 = 1 = 3.66Hz (R5 + R6) ⋅ C2 fpole2 = 1 = 12.6Hz R4 ⋅ C1 Where: R3 = 33kΩ, R4 = 360kΩ, R5 = R6 = 620kΩ, C1 = C2 = 220nF At 88 Vrms, Vpin7 = 1.78 Vrms At 264 Vrms, Vpin7 = 5.33 Vrms Gain at 2f (100Hz) = -80dB For single mains operation, this pin can be connected directly to Vref (pin 11) or to ground and the RC network can be removed. If connected to ground, the Vrms multiplier input is clamped at 1.5V. APPLICATION NOTE Pin 8 MULT-OUT (Output of the Multiplier) This pin delivers the current Imult that is used to fix the reference voltage for the current amplifier. Pin 8 is connected through R14 to the negative side of the sense resistor, R30-32, to sum the (IL ⋅ Rs) and the (Imult ⋅ R14) signals, where IL is the inductor current. The sum is the error voltage signal at the current amplifier non-inverting input. The multiplier output current is determined by the equation given below: Imult = 0.37 ⋅ IAC ⋅ (Vva−out – 1.28V) ⋅ (0.8 ⋅ Vlff –1.28V) = IIAC ⋅ Vrms2 = (vva−out – 1.28V) Vrms2 Where: Vva-out = Error amplifier output voltage range Vlff = Vref = 5.1V if not used for load feed-forward Vrms = Voltage at pin 7 IIAC = Input current at pin 4 To optimize the multiplier biasing for each application, the relationships between Imult and other input signals are reported in the Designing A High Power Factor Switching Preregulator With The L4981 Continuous Mode Application Note [1], Figures 13a-13h. Pin 9 ISENSE (Current amplifier inverting input) This pin is the current amplifier inverting input. It is externally connected to the network described at CA-OUT (pin 5). Note that R14=R16=2.7k have the same value because of the high impedance feedback network. The sense resistors, R30-R32, have a combined resistance of 0.033 ohms. The low value is chosen to minimize the power losses since the total inductor current flows through this resistor. The value must be large enough to provide a good signal to noise ratio signal to the current amplifier. can deliver up to 10mA for external circuit needs such as the fast start-up power supply circuit as described in Pin 19. Pin 12 SS (Soft start) This feature avoids current overload through the power mosfet during the ramp-up of the output boosted voltage. An internal switch discharges the capacitor if an output overvoltage (OVP) or a VCC undervoltage (UVLO) is detected. The voltage at the soft-start pin acts on the output of the error amplifier and the soft start time is calculated as follows: tss = Css Vva−out 5.1V = 1µF = 51ms Iss 100µA Where: Css = C8 = 1µF Vva-out = 5.1V is the typical error amplifier voltage swing Iss is the internal soft start current generator Pin 13 Vva-out (Error amplifier output) To ensure system stability, the compensation network must be designed with sufficient phase margin. Additionally, the system must not regulate the twice mains frequency output ripple voltage in order to avoid line current distortion. The compensation capacitor, C10, can be calculated as follows: C10 > ∆Vout 1 = Ka 4 ⋅ π ⋅ fmains ⋅ (R9 +R10) ⋅ Gea (R9 + R10) Where: R9 + R10 are the resistors from the output voltage feedback resistor divider Gea is the small signal gain of the error amplifier ∆Vout is the maximum output voltage ripple 1 1 Ka = for 50 Hz and for 60 Hz mains fre60 72 quency 1 8 ⋅ = 162nF, therefore use standard 60 824k value 220nF C10 > Pin 10 SGND (Signal ground) This pin should be connected close to the reference voltage filter capacitor (C7). Refer to Figure 3, 500 Watt Demoboard printed circuit board layout. Pin 11 VREF (Voltage reference) An external capacitor filter of 1uF, C7, should be connected from pin 11 (Vref) to ground. This reference voltage of 5.1V is externally available and The voltage open loop gain contains two poles at the origin, causing stability problems. This can be avoided by shifting the error amplifier pole from the origin to near the crossover frequency. This can be accomplished by placing a resistor, R19, in parallel with the compensation capacitor, C10. The crossover frequency is calculated as follows: 7/16 APPLICATION NOTE Pout 1 = fc = √ V ⋅ ∆V ⋅ 2π ⋅ C 2π ⋅ (R9 + R10) ⋅ C10 out ea out = 500 1 √ = 11.77Hz 400 ⋅ 3.82 ⋅ 2π ⋅ 330 µF 2π ⋅ 824k ⋅ 220nF R19 ≥ 1 = 83.4k 2π ⋅ fc ⋅C10 Use R19 = 120k to increase error amplifier dc gain. Pin 14 VFEED (Error amplifier input) This pin is the error amplifier inverting input. This pin is connected to the resistor divider connected across the boosted output voltage to provide regulation. The boosted output voltage is specified at 400VDC. The resistor divider network is calculated as follows: 400 R9 + R10 824k Vout = = −1 = −1 10.6k Vref 5.1 R20 Use R9 = R10 = 412k Pin 15 P-UVLO (Programmable supply undervoltage threshold) This pin may be used to modify the turn-on and turn-off power supply thresholds. This circuit does not employ this feature and the pin is left floating. The typical turn-on threshold is 15.5V and the turn-off threshold is 10V. Pin 16 SYNC (In/Out synchronization) This function allows for synchronization in master or slave mode with other circuits in the system. This demoboard does not use this function and the pin is left floating. Pin 17 ROSC (Oscillator resistor) Pin 18 COSC (Oscillator capacitor) These pins determine the oscillator frequency of the circuit. A resistor, R17, is connected from pin 17 to ground. A capacitor, C4, is connected from pin 18 to ground. The operating frequency is calculated as follows: fSW = 2.44 2.44 = = 80kHz approx. Rosc ⋅ Cosc 30.1k ⋅ 1n Pin 19 VCC (Supply voltage input) The IC must be supplied with a very low current, 0.3mA typical, during start-up. The turn-on threshold is 15.5V typical with 5.5 Volts typical of hysteresis. The start-up current is provided by 8/16 the resistor/capacitor network driven off the rectified line voltage. A fast start-up circuit is employed to quickly turn on the IC and reduce power consumption in the start-up resistor, R28. The capacitor, C12, has a value of 220uF to ensure sufficient hold-up time to allow the auxiliary winding to provide voltage after initial start-up. The fast start-up is realized with Q2, Q3, R25, R26, R27, R28, D5 and C12. The fast start-up circuit is turned-off when the controller turn-on threshold is reached and Vref forward biases Q2, pulling the gate of Q3 to ground. The auxiliary winding on the main boost inductor provides the normal operating voltage for the controller. The voltage induced on this winding is rectified by diodes D7-D10. Resistor R29 provides current limiting and zener D6 regulates the supply voltage to 18 Volts. Pin 20 GDRV (Gate driver output) The output of this pin is internally clamped at 15V to prevent breakdown of the power mosfet gate oxide. A resistor, R18, of 15Ω is placed in series with the gate of the power mosfet to avoid overshoot and limit the di/dt of the switch. A 1N4148 diode, D3, is connected to the gate to provide fast turn-off of the power mosfet. EVALUATION RESULTS The 500W demoboard has been evaluated for the following parameters: PF (power factor), % THD (percent total harmonic distortion), H3..H7 (percentage of current’s nth harmonic amplitude), Vout (output voltage) and efficiency (n). The test configuration and test results are shown below: Test Set-Up and Equipment AC POWER SOURCE LARCET 3KW PM1200 AC POWER ANALYSER EMI FILTER PFC L4981 DEMO LOAD Table 1: 500W Demoboard Evaluation Results Vin f Pi (Vrms) (Hz) (W) 88 PF THD H3 H5 H7 Vout Po η (%) (%) (%) (%) (V) (W) (%) 60 560 99.9 2.9 1.3 1.7 1.2 402 490 87.5 110 60 543 99.9 2.8 1.4 1.8 1.3 403 492 90.6 220 50 525 99.8 3.3 1 2.4 1.1 406 499 95.1 270 50 523 99.8 3.4 1 2.6 1.1 408 504 96.3 APPLICATION NOTE source and the demoboard under test, while the efficiency has been calculated without the filter contribution. EMI/RFI FILTER The harmonic content measurement was made with the EMI/RFI filter interposed between the AC Figure 1: EMI/RFI Test Filter T1 T2 LINE C1 PFC C EARTH D94IN052 Part List of the Figure 2. Part Des. Description Vendor’s Part # Fuse F1 Fuse, 3AG Fast Acting 10A, 250VAC Digi-Key #F127-ND Fuse Clip C1 3AG Fuse Clips Met. Poly. Film Cap., 0.22µF, 100V Panasonic ECQ-E1224KF Digi-Key #F048-ND Digi-Key #EF1224 C2 C3 Met. Poly. Film Cap., 0.22µF, 100V Panasonic ECQ-E1224KF Met. Poly. Film, .68uF, 250VAC, Panasonic ECQU2A684MV Digi-Key #EF1224 Digi-Key #P4615-ND C4 Polyester Cap., .001µF, 50V, Panasonic ECQ-B1H102JF Digi-Key #P4551-ND C5 Polyester Cap., .012µF, 50V Panasonic ECQ-B1H123JF Digi-Key #P4583-ND C6 C7 Alum. Electrolytic Cap., 330µF, 450VDC, 85 Deg. C Electrolytic Cap., 1.0µF, 63V, Panasonic ECE-A1JU010,85Deg C Digi-Key#P6443-ND Digi-Key #P6275-ND C8 C9 Electrolytic Cap., 1.0µF, 63V, Panasonic ECE-A1JU010,85Deg C Polyester Cap. 680pfd., 50V, Panasonic ECQ-B1H681JF Digi-Key #P6275-ND Digi-Key # P4580-ND C10 C11 C12 Met. Poly. Film Cap., 0.22µF, 100V Panasonic ECQ-E1224KF Ceramic Capacitor, 820pfd.,1000VDC Electrolytic Cap., 220µF, 25V,Panasonic ECE-A1EU101,85Deg C Digi-Key #EF1224 Digi-Key #P4127-ND Digi-Key #P6240-ND D1 D2 Diode Bridge, 600V, 25A STTA806D/DI,, 600V, 8A, Isolated TO220AC Package Digi-Key #MB256-ND SGS-THOMSON STTA806D/DI D3 D4 Switching Diode, 1N4148, 100V Fast Recovery Diode, STTB406, 600V, 4A Digi-Key #1N4148CT-ND SGS-THOMSON STTB406 D5 D6 Zener Diode, 22V, 1/2W, DO-35 Package Zener Diode, 18V, 1/2W, DO-35Package Digi-Key #1N5251BCT-ND Digi-Key #1N5248BCT-ND D7 D8 Fast Recovery Rectifier Diode, 100V, 1.5A Fast Recovery Rectifier Diode, 100V, 1.5A SGS-THOMSON BYW-100-100 SGS-THOMSON BYW-100-100 D9 D10 Fast Recovery Rectifier Diode, 100V, 1.5A Fast Recovery Rectifier Diode, 100V, 1.5A SGS-THOMSON BYW-100-100 SGS-THOMSON BYW-100-100 R1 R2 Metal Film Res., 806K, 1/4W, 1% Metal Film Res., 806K, 1/4W, 1% Digi-Key #806KXBK-ND Digi-Key #806KXBK-ND R3 Carbon Film Res., 33K, 1/4W, 5% Digi-Key #33KQBK-ND 9/16 APPLICATION NOTE Part List of the Figure 2 (continued) Part Des. Description Vendor’s Part # R4 R5 Carbon Film Res., 360k, 1/4W, 5% Carbon Film Res., 620k, 1/4W, 5% Digi-Key #360KQBK-ND Digi-Key #620KQBK-ND R6 R9 Carbon Film Res., 620k, 1/4W, 5% Metal Film Res., 412k, 1/4W, 1% Digi-Key #620KQBK-ND Digi-Key #412KXBK-ND R10 Metal Film Res., 412k, 1/4W, 1% Digi-Key #412KXBK-ND R11 Metal Film Res., 21k, 1/4W, 1% Digi-Key #21.0KXBK-ND R12 Metal film Res., 562, 1/4W, 1% Digi-Key #562XBK-ND R13 Metal Film Res., 5.11k, 1/4W, 1% Digi-Key #5.11KXBK-ND R14 Carbon Film Res., 2.7k, 1/4W, 5% Digi-Key #2.7KQBK-ND R15 Carbon Film Res., 36k, 1/4W, 5% Digi-Key #36KQBK-ND R16 Carbon Film Res., 2.7k, 1/4W, 5% Digi-Key #2.7KQBK-ND R17 R18 Metal Film Res., 30.1k, 1/4W, 1% Carbon Film Res., 15 ohms, 1/4W, 5% Digi-Key #30.1KXBK-ND Digi-Key #15QBK-ND R19 Carbon Film Res., 120k, 1/4W, 5% Digi-Key #120KQBK-ND R20 Metal Film Res., 10.7k, 1/4W, 1% Digi-Key # 10.7KXBK-ND R21 R22 Metal Film Res., 909k, 1/4W, 1% Metal Film Res., 909k, 1/4W, 1% Digi-Key #909KXBK-ND Digi-Key #909KXBK-ND R23 Metal Oxide Resistor, 510 ohms, 3 Watts, 5% Digi-Key#P510W-3BK-ND R24 Metal Oxide Resistor, 510 ohms, 3 Watts, 5% Digi-Key#P510W-3BK-ND R25 R26 Carbon Film Resistor, 10k, 1/4W, 5% Carbon Film Resistor, 1.1M, 1/4W, 5% Digi-Key #10KQBK-ND Digi-Key #1.1MQBK-ND R27 R28 Carbon Film Resistor, 1.1M, 1/4W, 5% Carbon Film Res., 10k, 1/2W, 5% Digi-Key #1.1MQBK-ND Digi-Key #10KH-ND R29 R30 Carbon Film Resistor, 33 ohms, 1/2W, 5% 3 Watt, non-inductive 0.1 ohms, Type LO-3-.010 Digi-Key #33H-ND Newark #96F3616 R31 R32 3 Watt, non-inductive 0.1 ohms, Type LO-3-.010 3 Watt, non-inductive 0.1 ohms, Type LO-3-.010 Newark #96F3616 Newark #96F3616 NTC 1 20 Ga (0.8mm) Jumper Wire 22 Ga Jumper NTC 2 20 Ga. (0.8mm) Jumper Wire 22 Ga Jumper Heatsink 1 Heatsink 2 PCB 1 AAVID type 61085, 1.5Deg C/W/3in., 1.5" length Bridge Diode attachable heatsink FR-4 Material AAVID #61085 datogliere CALS 95 001_A T1 Standoffs Coilcraft Part# R4849-A 0.5mH Aluminum Hex Standoff 0.375", 4-40 Thread CoilCraft ’Part # R4849-A Newark#89F1949 Q1 Q2 STW20NA50, 500V, 20A, 2.7 ohms TO-247 NPN transistor high speed, 30V, .8A, TO-18 Package SGS-THOMSON STW20NA50 SGS-THOMSON 2N2222 Q3 J1 N-Channel Mosfet, STK2N50, 500V, 2A, SOT-82 3 Pole, 15A, Terminal Block SGS-THOMSON STK2N50 Newark #93F7182 J2 U1 3 Pole, 15A, Terminal Block L4981A, PFC IC Newark #93F7182 SGS-THOMSON L4981A IC Socket Misc. 20 Pin DIP Socket, Gold Pin and Clip Mounting screws, nuts, insulators Digi-Key #ED56203-ND 10/16 - D1 BRIDGE - + NTC1 C3 0.68µF 250VAC + NTC2 + C1 220nF 100V R12 562 1% R2 806K 1% C2 220nF 100V R14 2.7K 5% R13 5.11K 1% 15 9 R8 D5 22V 1/4W R32 0.1Ω 3W R31 0.1Ω 3W R30 0.1Ω 3W R16 2.7K 5% R15 36K C9 5% 0.68nF 50V PUVLO R7 Q2 2N2222 8 5 2 4 7 16 CAOUT MOUT IPK IAC VRMS S/FM R25 10K 5% R3 33K 5% R26 1.1M 5% 18 17 C4 1nF 50V 12 C8 1µF 63V D7 20 3 13 C7 1µF 63V T1 C10 220nF 100V D10 D8 GDRV R18 15Ω 5% D3 1N4148 OVP VAOUT R19 120K 5% 11 6 10 1 14 D6 18V 1/4W BYW100-100 D9 VFEED R29 33 1/4W SSC R17 30.1K 1% U1 L4981A 19 VCC C12 220µF 25V Q3 STK2N50 COSC R1 806K 1% ISENSE R28 10K 1/4W ROSC R27 1.1M 5% LFF R6 620K 5% VREF R5 620K 5% SGND R4 360K 5% PGND F1 10A Q1 STW20NA50 R24 510Ω 3W R23 510Ω 3W C11 820pF 1000V D2 STTA806DI D4 STTB-406 + - Vi (88 to 264V) C5 12nF 50V D95IN258B R11 21K 1% R22 909K 1% R10 412K 1% R20 10.7K 1% R21 909K 1% R9 412K 1% C6 330µF 450V Vout APPLICATION NOTE Figure 2: 500W Demoboard Schematic 11/16 APPLICATION NOTE Figure 3: 500W Demoboard Printed Circuit Board Layout 12/16 - D1 BRIDGE - + NTC1 R12 360 1% R2 200K 1% R14 6.2K 5% R13 5.11K 1% 15 9 R8 D5 22V 1/4W R32 120mΩ 2W R31 120mΩ 2W R30 120mΩ 2W R16 6.2K 5% R15 47K C9 5% 0.47nF 50V PUVLO R7 Q2 2N2222 8 5 2 4 7 16 CAOUT MOUT IPK IAC VRMS S/FM R25 10K 5% Trasformer T1: core: ETD 39 x 20 x 13 / gap ≈1.8mm. Primary Inductance =0.25mH 44 Turns 15 x AWG29. Secondary = 5 Turns C3 470nF 250VAC + NTC2 + 18 17 C4 1.2nF 60V R17 24K 1% SSC R29 33 1/4W U1 L4981A 19 VCC C12 220µF 25V COSC R26 1.1M 5% ISENSE Q3 STK2N50 ROSC R1 200K 1% C8 1µF 63V 14 20 3 13 C7 1µF 63V T1 C10 120nF 100V D10 D8 GDRV R18 15Ω 5% D3 1N4148 OVP VAOUT R19 220K 5% 11 6 10 1 VFEED D6 18V 1/4W 12 D7 BYW100-100 D9 LFF R28 10K 2W VREF R27 1.1M 5% SGND F1 10A PGND R24 680Ω 2W R23 680Ω 2W C11 330pF 250V D2 BYT08P-400 C5 12nF 50V D95IN284C R11 15K 1% R22 360K 1% R10 330K 1% R20 15K 1% R21 360K 1% R9 330K 1% C6 680µF 315V Vout=230V An Application Program named Designing PFC [3] is available for the designer. This program allows the designer to make changes to the input/output design specifications and calculates and selects the component values and types. For example, this program can easily convert this de- Q1 STP9N30 D4 STTB-406 + - Vi (88 to 132V) APPLICATION NOTE sign to single mains operation (120 or 240 Volts). The results are presented in two screens, the schematic and parts list, and may be sent to a printer for a hardcopy for future reference. Two solutions at 110Vac (fig. 4) and 220Vac (fig. 5) are shown below. Figure 4: 400W/230V; Vin =110V ±20%. 13/16 - D1 BRIDGE - + R12 390 1% R2 510K 1% R14 7.5K 5% R13 5.11K 1% 15 9 R8 D5 22V 1/4W R32 120mΩ 2W R31 120mΩ 2W R30 120mΩ 2W R16 7.5K 5% R15 36K C9 5% 0.47nF 50V PUVLO R7 Q2 2N2222 8 5 2 4 7 16 CAOUT MOUT IPK IAC VRMS S/FM R25 10K 5% Trasformer T1: core: ETD 44 x 22 x 14 / gap =2mm. Primary Inductance =0.43mH 53 Turns 10 x 0.36mm Secondary = 5 Turns NTC1 C3 220nF 250VAC + NTC2 18 17 C4 1.2nF 60V R17 24K 1% SSC R29 33 1/4W U1 L4981A 19 VCC C12 220µF 25V COSC + ISENSE Q3 STK2N50 ROSC R26 1.1M 5% C8 1µF 63V 14 20 3 13 C7 1µF 63V T1 C10 220nF 100V D10 D8 GDRV R18 15Ω 5% D3 1N4148 OVP VAOUT R19 130K 5% 11 6 10 1 VFEED D6 22V 1/4W 12 D7 BYW100-100 D9 LFF R28 10K 4W VREF R1 510K 1% SGND F1 10A PGND R27 1.1M 5% Q1 STH14N50/FI R24 510Ω 3W R23 510Ω 3W C11 820pF 1000V D2 STTA806DI D4 STTB-406 + 14/16 - Vi (176 to 264V) C5 12nF 50V D95IN283C R11 21K 1% R22 909K 1% R10 412K 1% R20 10.7K 1% R21 909K 1% R9 412K 1% C6 470µF 450V Vout=400V APPLICATION NOTE Figure 5: 800W/400V; Vin = 220V ±20%. APPLICATION NOTE REFERENCES [1] G. Comandatore and U. Moriconi, Application Note 628 Designing A High Power Factor Switching Preregulator With The L4981 Continuous Mode, SGS-THOMSON Microelectronics, Inc., May, 1994. [2] Datasheet Power Factor Corrector, SGSTHOMSON Microelectronics, Inc., May, 1994. [3] Designing PFC Application Program, SGSTHOMSON Microelectronics, Inc., April, 1995. 15/16 APPLICATION NOTE Information furnished is believed to be accurate and reliable. However, SGS-THOMSON Microelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No license is granted by implication or otherwise under any patent or patent rights of SGS-THOMSON Microelectronics. Specification mentioned in this publication are subject to change without notice. This publication supersedes and replaces all information previously supplied. SGSTHOMSON Microelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of SGS-THOMSON Microelectronics. © 1997 SGS-THOMSON Microelectronics – Printed in Italy – All Rights Reserved SGS-THOMSON Microelectronics GROUP OF COMPANIES Australia - Brazil - Canada - China - France - Germany - Italy - Japan - Korea - Malaysia - Malta - Morocco - The Netherlands Singapore - Spain - Sweden - Switzerland - Taiwan - Thailand - United Kingdom - U.S.A. 16/16