MIC285 513 4 45V 4A Syn nchronous Buck Reg gulator Gen neral Desc cription Featu ures The MIC28513 is i a synchro onous step-d down switching regullator with inte ernal power sw witches capable of providing up to o 4A output current from a wide inpu ut supply range from 4.6V to 45V. The output voltage v is adju ustable down to 0.8V with a gua aranteed accuracy of ±1% %. A consta ant switcching frequen ncy can be prrogrammed from f 200kHz to 680kkHz. The MIC28513’s Hy yper Speed Control™ and Hype erLight Load® architecture es allow for high VIN (lo ow VOUT) operation and ultra-fas st transient response r wh hile reduccing the requ uired output capacitance. In addition to fore-mentioned attributes, a the e MIC28513--1’s HyperLig ght Load d architecture provides very y good light lo oad efficiency y. The MIC28513 offers o a full suite s of featu ures to ensu ure prote ection unde er fault co onditions. These T include unde ervoltage lock kout to ensu ure proper operation und der powe er sag condittions, internal soft-start to o reduce inrush curre ent, foldback current limit, “hiccup” mo ode short-circ cuit prote ection, and the ermal shutdow wn. Datasheets and support s docu umentation arre available on Micre el’s web site at: a www.micre el.com. 4.6V V to 45V operrating input vo oltage supply Up tto 4A output ccurrent Integ grated high-sside and low-sside N-channe el MOSFETs Hyp perLight Load (MIC28513-1) and H Hyper Speed d Con ntrol (MIC28513-2) architeccture Ena able input, pow wer good (PG GOOD) outputt grammable current limit an nd foldback ““hiccup” mode e Prog shorrt circuit prote ection Builtt-in 5V regula ator for single supply opera ation Adju ustable 200kH Hz to 680kHz switching fre equency Fixe ed 5ms soft-sttart Interrnal compenssation and the ermal shutdow wn. The rmally enhanced 24-pin 3m mm × 4mm FCQFN ature range off –40°C to +125°C Juncction tempera Appllications Indu ustrial power ssupplies Disttributed supply regulation wer supplies Bas e station pow Wal l transformer regulation h-voltage sing gle board systems High Typ pical Application Efficiency (V VIN =12V) vs. Output Curre ent MIC28513-1 100.0 5.0V 3.3V EFFICIENCY (%) 90.0 80.0 70.0 FSW = 30 00kHz 60.0 50.0 0.01 1 0.1 1 10 0 OUTPUT CURRENT (A) Hype er Speed Control is a trademark of o Micrel, Inc. Hype erLight Load is a registered trade emark of Micrel, Inc. I Micrel Inc. • 2180 Fortune Driv ve • San Jose, CA C 95131 • USA • tel +1 (408) 94 44-0800 • fax + 1 (408) 474-1000 0 • http://www.m micrel.com Marcch 25, 2015 Revision 1.2 Micre el, Inc. MIC28513 3 Ord dering Info ormation Archite ecture Packa age(1) Junctio on Temperaturre Range Lead Finish MIC C28513-1YFL HyperLight Load 24-Pin 2 3mm × 4mm FCQFN – –40°C to +125°°C Pb-Free MIC C28513-2YFL Hyper Spee ed Control 24-Pin 2 3mm × 4mm FCQFN – –40°C to +125°°C Pb-Free Partt Number Note: 1. FC CQFN is a lead-ffree package. Pb b-free lead finish is Matte Tin. Pin Configuration 24-Pin 3mm m × 4mm FCQF FN (FL) (Top View) Marcch 25, 2015 2 Revision 1.2 2 Micrel, Inc. MIC28513 Pin Description Pin Number Pin Name 1 DL 2 PGND 3 DH Pin Function Low-side gate drive. Internal low-side power MOSFET gate connection. This pin must be left unconnected or floating. PGND is the return path for the low-side driver circuit. Connect to the source of low-side MOSFET (PGND, Pin 10, 11, 22, 23, and 26) through a low impedance path. High-side gate drive. Internal high-side power MOSFET gate connection. This pin must be left unconnected or floating. PVIN Power input voltage. The PVIN pins supply power to the internal power switch. Connect all PVIN pins together and locally bypass with ceramic capacitors. The positive terminal of the input capacitor should be placed as close as possible to the PVIN pins; the negative terminal of the input capacitor should be placed as close as possible to the PGND pins 10,11, 22, 23, and 26. 5 LX The LX pin is the return path for the high-side driver circuit. Connect the negative terminal of the bootstrap capacitor directly to this pin. Also connect this pin to the SW pins 12, 21, and 27, with a low impedance path. The controller monitors voltages on this pin and the PGND for zero current detection. 6 BST Bootstrap pin. This pin provides bootstrap supply for the high-side gate driver circuit. Connect a 0.1µF capacitor and an optional resistor in series from the LX (Pin 5) to the BST pin. 10, 11, 22, 23, 26 (26 is ePad) PGND Power ground. These pins are connected to the source of the low-side MOSFET. They are the return path for the step-down regulator power stage and should be tied together. The negative terminal of the input decoupling capacitor should be placed as close as possible to these pins. 12, 21, 27 (27 is ePad) SW 13 AGND 14 FB 15 PGOOD 16 EN Enable input. The EN pin enables the regulator. When the pin is pulled below the threshold, the regulator will shut-down to an ultra-low current state. A precise threshold voltage allows the pin to operate as an accurate UVLO. Do not tie EN to VDD. 17 VIN Supply voltage for the internal LDO. The VIN operating voltage range is from 4.6V to 45V. A ceramic capacitor from VIN to AGND is required for decoupling. The decoupling capacitor should be placed as close as possible to the supply pin. 18 ILIM Currrent limit setting. Connect a resistor from this pin to the SW pin node to allow for accurate current-limit-sense programming of the internal low-side power MOSFET. 19 VDD Internal +5V linear regulator: VDD is the internal supply bus for the IC. Connect to an external 1µF bypass capacitor. When VIN is less than 5.5V, this regulator operates in drop-out mode. Connect VDD to VIN. 20 PVDD A 5V supply input for the low-side N-channel MOSFET driver circuit that can be tied to VDD externally. A 1μF ceramic capacitor from PVDD to PGND is recommended for decoupling. 24 FREQ Switching frequency adjust pin. Connect this pin to VIN to operate at 600kHz. Place a resistor divider network from VIN to the FREQ pin to program the switching frequency. 4, 7, 8, 9, 25 (25 is ePad) March 25, 2015 Switch node. The SW pins are the internal power switch outputs. These pins should be tied together and connected to the output inductor. Analog ground. The analog ground for VDD and the control circuitry. The analog ground return path should be separate from the power ground (PGND) return path. Feedback input. The FB pin sets the regulated output voltage relative to the internal reference. This pin is connected to a resistor divider from the regulated output such that the FB pin is at 0.8V when the output is at the desired voltage. The power good output is an open drain output requiring an external pull-up resistor to external bias. This pin is a high impedance open circuit when the voltage at FB pin is higher than 90% of the feedback reference voltage (typically 0.8V). 3 Revision 1.2 Micrel, Inc. MIC28513 Absolute Maximum Ratings(2) Operating Ratings(3) PVIN, VIN to PGND ........................................ 0.3V to 50V VDD, PVDD to PGND ................................ ……0.3V to 6V VBST to VSW, VLX ........ …………………..…………0.3V to 6V VBST to PGND .......................................... 0.3V to (VIN + 6V) VSW, VLX to PGND ............................... 0.3V to (VIN + 0.3V) VFREQ, VILIM, VEN to AGND .................... 0.3V to (VIN + 0.3V) VLX, VFB, VPG, VFREQ, VILIM, VEN to AGND ................................ 0.3V to (VDD + 0.3V) PGND to AGND ………………......................0.3V to +0.3V Junction Temperature (TJ) ....................................... +150C Storage Temperature (TS) ......................... 65C to +150C Lead Temperature (soldering, 10s) ............................ 300C ESD HBM Rating(4)...................................................... 1.5kV ESD MM Rating(4) ......................................................... 150V Supply Voltage (PVIN, VIN)............................. +4.6V to +45V Enable Input (VEN) .................................................. 0V to VIN VSW, VFEQ, VILIM, VEN ............................................... 0V to VIN Junction Temperature (TJ) ........................ –40°C to +125°C Junction Thermal Resistance FQFN (JA) ......................................................... 30°C/W Electrical Characteristics(5) VIN = 12V; TA = 25°C, bold values indicate –40°C≤ TA ≤ +125°C, unless noted. Parameter Condition Min. Typ. Max. Units 45 V Power Supply Input 4.6 Input Voltage Range (PVIN, VIN) Quiescent Supply Current Shutdown Supply Current VFB = 1.5V (MIC28513-1) 0.4 0.75 VFB = 1.5V (MIC28513-2) 0.7 1.5 SW = unconnected, VEN = 0V 0.1 10 µA mA VDD Supply VDD Output Voltage VIN =7V to 45V, IVDD = 10mA 4.8 5.2 5.4 V VDD UVLO Threshold VVDD rising 3.8 4.2 4.6 V VDD UVLO Hysteresis 400 Load Regulation @ 40mA mV 0.6 2 4 % 25°C (±1%) 0.792 0.8 0.808 -40°C ≤ TJ ≤ 125°C (±2%) 0.784 0.8 0.816 5 500 Reference Feedback Reference Voltage FB Bias Current VFB = 0.8V V nA Enable Control 1.8 EN Logic Level High V 0.6 EN Logic Level Low EN Hysteresis EN Bias Current 200 VEN = 12V 5 V mV 40 µA Notes: 2. Exceeding the absolute maximum ratings may damage the device. 3. The device is not guaranteed to function outside its operating ratings. 4. Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5kΩ in series with 100pF. 5. Specification for packaged product only. March 25, 2015 4 Revision 1.2 Micrel, Inc. MIC28513 Electrical Characteristics(5) (Continued) VIN = 12V; TA = 25°C, bold values indicate –40°C≤ TA ≤ +125°C, unless noted. Parameter Condition Min. Typ. Max. VFREQ = VIN 450 680 800 Units Oscillator Switching Frequency VFREQ = 50% VIN 340 Maximum Duty Cycle Minimum Duty Cycle VFB > 0.8V Minimum Off-Time 110 kHz 85 % 0 % 200 270 ns Internal MOSFETs High-Side NMOS On-Resistance 37 m Low-Side NMOS On-Resistance 20 m Short Circuit Protection Current-Limit Threshold VFB = 0.79V –30 –14 0 mV Short-Circuit Threshold VFB = 0V –24 –7 8 mV Current-Limit Source Current VFB = 0.79V 50 70 90 µA Short-Circuit Source Current VFB = 0V 25 36 43 µA 50 µA 95 %VOUT Leakage SW, BST Leakage Current Power Good 85 Power Good Threshold Voltage Sweep VFB from low to high 90 Power Good Hysteresis Sweep VFB from high to low 6 %VOUT Power Good Delay Time Sweep VFB from low to high 100 µs Power Good Low Voltage VFB < 90% x VNOM, IPGOOD = 1mA 70 TJ rising 160 °C 15 °C 5 ms 200 mV Thermal Protection Overtemperature Shutdown Overtemperature Shutdown Hysteresis Soft Start Soft Start Time March 25, 2015 5 Revision 1.2 Micrel, Inc. MIC28513 Typical Characteristics Switching Frequency vs. Output Current MIC28513-1 Feedback Voltage vs. Temperature MIC28513-1 0.812 350 300 250 200 150 100 0.808 FEEBACK VOLTAGE (V) VIN = 12V VOUT = 5V FEEBACK VOLTAGE (V) SWITCHING FREQUENCY (kHz) 400 Feedback Voltage vs. Temperature MIC28513-2 0.808 0.804 0.800 VIN = 12V VOUT = 5.0V IOUT = 0A 0.796 50 0.0 0.1 1.0 -25 0.776 0 25 50 75 100 -50 125 -25 TEMPERATURE (°C) OUTPUT CURRENT (A) VDD Voltage vs. Input Voltage OUTPUT VOLTAGE ERROR (%) VDD THRESHOLD (V) VOUT = 5.0V 5.0 IDD = 40mA 4.5 50 75 100 125 1.0 Rising IDD = 10mA 25 Line Regulation Error (VOUT vs. VIN) 5.0 5.5 0 TEMPERATURE (°C) VDD UVLO Threshold vs. Temperature MIC28513-1 6.0 VDD VOLTAGE (V) 0.784 0.760 -50 10.0 0.792 0.768 0.792 0 VIN = 12V VOUT = 5.0V IOUT = 0A 0.800 4.8 Falling 4.6 4.4 VIN =12V IOUT = 0A 4.2 0.8 VOUT = 5.0V IOUT = 2A FSW = 300kHz 0.6 0.4 0.2 0.0 -0.2 -0.4 -0.6 -0.8 -1.0 5 4.0 4.0 5 10 15 20 25 30 35 40 -50 45 -25 25 50 75 100 1.4 FALLING 0.8 0.6 0.4 HYSTERESIS 25 1.6 1.2 0.8 0.4 10 15 20 25 30 INPUT VOLTAGE (V) March 25, 2015 35 40 45 40 45 16 14 12 10 8 6 VOUT = 5V IOUT = 0A FSW = 300kHz 4 2 0.0 5 35 18 0.2 0.0 30 20 VOUT = 5V IOUT = 0A FSW = 300kHz SUPPLY CURRENT (mA) SUPPLY CURRENT (mA) RISING 20 VIN Operating Supply Current vs. Input Voltage MIC28513-2 2.0 1.0 15 INPUT VOLTAGE (V) VIN Operating Supply Current vs. Input Voltage MIC28513-1 Enable Threshold vs. Input Voltage 1.2 10 125 TEMPERATURE (°C) INPUT VOLTAGE (V) ENABLE THRESHOLD (V) 0 0 5 10 15 20 25 30 INPUT VOLTAGE (V) 6 35 40 45 5 10 15 20 25 30 35 40 45 INPUT VOLTAGE (V) Revision 1.2 Micrel, Inc. MIC28513 Typical Characteristics (Continued) Output Voltage vs. Output Current MIC28513-2 4.95 4.90 VIN = 12V VOUT = 5.0V FSW = 300kHz 4.85 4.80 450 10 400 350 300 250 200 VIN = 12V VOUT = 5V 150 0.5 1.0 1.5 2.0 2.5 3.0 3.5 0.5 OUTPUT CURRENT (A) 1.0 1.5 2.0 2.5 3.0 3.5 -50 4.0 0 80.0 70.0 FSW = 300kHz 100 50.0 0.01 125 0.1 TEMPERATURE (°C) FSW = 300kHz 1 FSW = 300kHz 50 0.01 10 0.1 10 1 10 OUTPUT CURRENT (A) IC Power Dissipation vs. Output Current (VIN = 24V) 2.5 VIN =12V fSW = 300kHz 1.2 1.0 0.8 5.0V 3.3V 0.6 0.4 0.2 0.0 0.1 OUTPUT CURRENT (A) March 25, 2015 1 IC POWER DISSIPATION (W) 70 IC POWER DISSIPATION (W) 80 50 0.01 70 60 1.4 5.0V 3.3V 125 80 IC Power Dissipation vs. Output Current (VIN = 12V) 100 60 100 5.0V 3.3V OUTPUT CURRENT (A) Efficiency (VIN = 36V) vs. Output Current MIC28513-1 90 75 90 60.0 2 75 50 100 EFFICIENCY (%) EFFICIENCY (%) 4 50 25 Efficiency (VIN = 24V) vs. Output Current MIC28513-1 90.0 6 25 0 5.0V 3.3V 8 0 -25 TEMPERATURE (°C) 100.0 VIN =12V VOUT = 5.0V FSW = 300kHz -25 VIN =12V VOUT = 5.0V FSW = 300kHz Efficiency (VIN = 12V) vs. Output Current MIC28513-1 12 -50 4 OUTPUT CURRENT (A) Output Peak Current Limit vs. Temperature MIC28513-2 10 6 0 0.0 4.0 8 2 100 0.0 CURRENT LIMIT (A) CURRENT LIMIT (A) SWITCHING FREQUENCY (kHz) OUTPUT VOLTAGE (V) 12 500 5.00 EFFICIENCY (%) Output Peak Current Limit vs. Temperature MIC28513-1 Switching Frequency vs. Output Current MIC28513-2 2.0 VIN =24V fSW = 300kHz 1.5 1.0 5.0V 3.3V 0.5 0.0 0.0 0.5 1.0 1.5 2.0 2.5 3.0 OUTPUT CURRENT (A) 7 3.5 4.0 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 OUTPUT CURRENT (A) Revision 1.2 4.0 Micrel, Inc. MIC28513 Typical Characteristics (Continued) 2.5 VIN = 36V fSW = 300kHz 2.0 1.5 5.0V 3.3V 1.0 0.5 4.5 4.5 4.0 4.0 3.5 5.0V 3.3V 3.0 VIN = 12V fSW = 300kHz TJMAX = 125°C ΘJA = 30°C/W 2.5 2.0 1.5 1.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 3.5 3.0 VIN = 24V fSW = 300kHz TJMAX = 125°C ΘJA = 30°C/W 2.5 2.0 1.5 1.0 0.0 0.0 0.0 5.0V 3.3V 0.5 0.5 0.0 25 40 OUTPUT CURRENT (A) 55 70 85 100 25 Efficiency (VIN = 12V) vs. Output Current MIC28513-2 100 100 4.0 90 90 3.5 80 2.5 2.0 VIN = 36V fSW = 300kHz TJMAX = 125°C ΘJA = 30°C/W 1.5 1.0 60 50 40 FSW = 300kHz 20 0.0 25 40 55 70 85 100 AMBIENT TEMPERATURE (°C) 10 0.01 85 100 70 5.0V 3.3V 60 50 40 30 30 0.5 70 80 5.0V 3.3V 70 EFFICIENCY (%) EFFICIENCY (%) 5.0V 3.3V 55 Efficiency (VIN = 24V) vs. Output Current MIC28513-2 4.5 3.0 40 AMBIENT TEMPERATURE (°C) AMBIENT TEMPERATURE (°C) 36V Input Thermal Derating OUTPUT CURRENT (A) OUTPUT CURRENT (A) OUTPUT CURRENT (A) IC POWER DISSIPATION (W) 3.0 24V Input Thermal Derating 12V Input Thermal Derating IC Power Dissipation vs. Output Current (VIN = 36V) 0.1 1 OUTPUT CURRENT (A) FSW = 300kHz 20 10 10 0.01 0.1 1 OUTPUT CURRENT (A) Efficiency (VIN = 36V) vs. Output Current MIC28513-2 100 90 EFFICIENCY (%) 80 70 5.0V 3.3V 60 50 40 30 FSW = 300kHz 20 10 0.01 0.1 1 10 OUTPUT CURRENT (A) March 25, 2015 8 Revision 1.2 10 Micre el, Inc. MIC28513 3 Fun nctional Characteris stics Marcch 25, 2015 9 Revision 1.2 2 Micre el, Inc. MIC28513 3 Fun nctional Characteris stics (Con ntinued) Marcch 25, 2015 10 Revision 1.2 2 Micre el, Inc. MIC28513 3 Fun nctional Characteris stics (Con ntinued) Marcch 25, 2015 11 Revision 1.2 2 Micre el, Inc. MIC28513 3 Fun nctional Characteris stics (Con ntinued) Marcch 25, 2015 12 Revision 1.2 2 Micre el, Inc. MIC28513 3 Fun nctional Characteris stics (Con ntinued) Marcch 25, 2015 13 Revision 1.2 2 Micre el, Inc. MIC28513 3 Fun nctional Diagram Marcch 25, 2015 14 Revision 1.2 2 Micre el, Inc. MIC28513 3 not recomme ended to use e MIC28513 with an OFF FIt is n time cclose to tOFF(m during stea ady-state ope eration. min) Fun nctional Description n The MIC28513 is an adaptive on-time sync chronous buc ck regullator with integrated high-side and a low-side MOS SFETs suitable for high-in nput voltage to low-outpu ut voltage conversion applications. It is design ned to operate over a wide input voltage range, from 4.6V to 45V, which is suitable for auto omotive and industrial app plications. The outpu ut is adjustab ble with an ex xternal resistive divider. An adap ptive on-time control schem me is employed to obtain a constant switchin ng frequency in continuou us conduction mode e and reduce ed switching frequency f in discontinuous opera ation mode to improv ve light loa ad efficiency y. Overrcurrent prote ection is imp plemented by y sensing the low-sside MOSFET T’s RDS(ON). The device fea atures interna al softstart, enable, UVLO, U and th hermal shutdo own. The adaptive ON N-time contrrol scheme results in a consttant switchin ng frequencyy in the MIC C28513. The e actua al ON-time a and resulting g switching ffrequency will vary with the diffferent rising and falling times of the e extern nal MOSFET Ts. Also, the minimum tON results in a lowerr switching fre equency in hig gh VIN to VOUTT applications s. Durin ng load tran nsients, the switching frequency is chang ged due to th e varying OF FF-time. Figurre 1 shows th he allowable rrange of the o output voltage e versu us the input vvoltage. The minimum outtput voltage is 0.8V which is lim mited by the e reference voltage. The e maxim mum output voltage is 24 4V which is llimited by the e intern nal circuitry. ory of Operattion Theo As illustrated in the Functio onal Diagram m, the outpu ut voltage is sense ed by the fe eedback (FB) pin via the voltage divider R1 R and R2, and compare ed to a 0.8V V refere ence voltage VREF at the e error compa arator through a lo ow-gain transconductance e (gm) amp plifier. If the feedb back voltage decreases and a the amplifier output is below w 0.8V, then n the error comparator c will w trigger the contrrol logic and generate an ON-time perriod. The ON Ntime period lengtth is predete ermined by the “Fixed tON O Estim mator” circuitry y. t ON(ESTIMATED D) VOUT VIN fSW W Eq. 1 Figurre 1. Allowable e Output Volta age Range vs.. Input Voltage e wherre VOUT is the e output volta age, VIN is the e power stage inputt voltage, and d fSW is the switching s freq quency. At the end of the ON-tim me period, th he internal high-side drive er turnss off the high-side MOSFE ET and the lo ow-side drive er turnss on the low w-side MOSFET. The OF FF-time period lengtth depends up pon the feedb back voltage in i most cases s. When the feedback voltage de ecreases and d the output of o 8V, the ON-ttime period is the gm amplifier is below 0.8 trigge ered and the OFF-time pe eriod ends. If the OFF-time perio od determined d by the feed dback voltage e is less than the m minimum OF FF-time tOFF(m min), which is about 200ns (typ.)), the MIC28513 control logic l will app ply the tOFF(min) instead. tOFF(min) is s required to maintain eno ough energy in the boost capa acitor (CBST) to drive the t high-side MOS SFET. To ill ustrate the ccontrol loop o operation, botth the steady ystate and load tran nsient scenarrios will be analyzed. Figurre 2 shows th he MIC28513 3 control loop timing during g stead dy-state ope eration. Durin ng steady-sttate, the gm ampliifier senses the feedbackk voltage rip pple, which is propo ortional to the e output volta age ripple and d the inducto or curre nt ripple, to trrigger the ON N-time period.. The ON-time e is pre edetermined b by the tON estimator. The termination of o the O OFF-time is co ontrolled by th he feedback vvoltage. At the e valleyy of the feedb back voltage ripple, which h occurs when n VFB fa alls below VRREF, the OFF period ends and the nex xt ON-tiime period iis triggered through the control logic circuiitry. The m maximum dutty cycle is obttained from: D MAX 1 t OFF O (MIN) fSW Marcch 25, 2015 Eq. 2 15 Revision 1.2 2 Micre el, Inc. 3 MIC28513 e true current-mode contrrol, the MIC28 8513 uses the e Unlike outpu ut voltage rip pple to trigger an ON-time e period. The e outpu ut voltage riipple is proportional to the inducto or curre nt ripple if th he ESR of the e output capacitor is large e enoug gh. The MIC C28513 contro ol loop has the advantage e of elim minating the n need for slope e compensation. In orrder to meet stability req quirements, th he MIC28513 3 feedb back voltage ripple shou uld be in ph hase with the e inducctor current rip pple and larg ge enough to be sensed by y the gm amplifier and the error comparator. The e recom mmended fee edback voltage ripple is 20mV~100mV. If a low-ESR ou utput capacittor is selectted, then the e feedb back voltage rripple may be e too small to be sensed by y the g m amplifier a and the erro or comparatorr. Also, if the e ESR of the outp put capacitorr is very low w, the outpu ut voltag ge ripple and d the feedba ack voltage rripple are no ot necesssarily in pha ase with the inductor currrent ripple. In n these e cases, ripple injection is required to e ensure prope er opera ation. Please refer to the Ripple Injecttion section in n Appliccation Inform mation for mo ore details ab bout the ripple e injecttion technique e. Figure 2.. MIC28513 Co ontrol Loop Timing Figurre 3 shows th he operation of the MIC28 8513 during a load transient. The T output voltage v drops s due to the sudden load incre ease, which causes the VFB to be les ss than VREF. This will w cause the error comparrator to trigge er an O ON-time perio od. At the end d of the ON-ttime period, a minim mum OFF-tim me tOFF(min) is generated to o charge CBSST since e the feedbac ck voltage is still below VREF. Then, the next ON-time periiod is triggere ed due to the low feedbac ck voltage. Thereforre, the switc ching freque ency change es durin ng the load tra ansient, but returns r to the nominal fixed frequ uency once th he output has s stabilized att the new load curre ent level. With the varying g duty cycle and switching frequ uency, the outtput recovery y time is fast and a the outpu ut voltage deviation is small in MIC28513 conv verter. Disco ontinuous M Mode (MIC285 513-1 Only) In co ontinuous mode, the ind ductor curre ent is always greatter than zero; however, at light loads th he MIC28513 31 is able to forcce the inducctor current tto operate in n disco ontinuous mo ode. Discontin nuous mode is where the e inducctor current fa falls to zero, as indicated d by trace (IL) show wn in Figure 4. During thiis period, the e efficiency is s optim mized by shuttting down all the non-esssential circuits and m minimizing the e supply currrent. The MIC C28513 wakes up a and turns on n the high-sside MOSFE ET when the e feedb back voltage VFB drops bellow 0.8V. The MIC28513-1 has a zero crossing comparator tha at monittors the inducctor current b by sensing the e voltage drop p acrosss the low-sid de MOSFET during its O ON-time. If the e VFB > 0.8V and d the inducttor current goes slightly y negattive, then th he MIC28513 3-1 automattically powers down n most of the IC circuitry a and goes into o a low-powe er mode e. Once e the MIC285 513-1 goes into discontiinuous mode e, both DH and DL are low, which turns off the high-side e and l ow-side MOS SFETs. The lload current iis supplied by y the o output capacittors and VOUTT drops. If the e drop of VOUT cause es VFB to go o below VREFF, then all th he circuits will wake e up into norrmal continuo ous mode. F First, the bias curre nts of most circuitss reduced during the e disco ontinuous mo ode are resto ored, then a tON pulse is trigge ered before tthe drivers arre turned on to avoid any y possiible glitches. Finally, the high-side drriver is turned d on. Figure 4 sshows the control loo op timing in n disco ontinuous mod de. Figure 3. MIC28513 Load Transient Re esponse Marcch 25, 2015 16 Revision 1.2 2 Micre el, Inc. 3 MIC28513 Curre ent Limit The M MIC28513 usses the RDS(OON) of the inte ernal low-side e powe er MOSFET to o sense overrcurrent condiitions. In each h switch hing cycle, the inducto or current iss sensed by y monittoring the low w-side MOSF FET during itts ON period d. The ssensed voltag ge V(ILIM) is compared w with the powe er groun nd (PGND) after a blanking time off 150ns. The e voltag ge drop of th he resistor RILIM is compared with the e low-sside MOSFET T voltage dro op to set the e over-curren nt trip le evel. The sma all capacitor cconnected fro om ILIM pin to o PGND D can be add ded to filter tthe switching node ringing g, allow wing a betterr short limit measureme ent. The time e consttant created by RILIM and the filter cap pacitor should d uch less than be mu n the minimum m off time. The over currentt limit can b be programm med by using g Equa ation 3. RILIM M ICLIM 0.5 IL(PP) R DS(ON) VCL Eq. 3 ICL Wherre ICLIM = desired current limit Figure 4. MIC28513-1 Control C Loop Mode M (Discontinuous Mode) RDS(OON) = on-resisttance of the lo ow-side MOS SFET VCL = current-limit threshold (typical abso olute value is 14mV V) Durin ng discontinu uous mode, the bias current of mos st circuits are reduc ced. As a res sult, the total power supply curre ent during dis scontinuous mode m is only about 450μA A, allow wing the MIC C28513-1 to achieve high h efficiency in light load applicatiions. ICL = ccurrent-limit ssource curren nt (typical valu ue is 80µA) ∆IL(PPP) = inductor ccurrent peak-tto-peak. Use Equation 4 to o calcu ulate the inducctor ripple currrent. VDD Regulator The MIC28513 provides p a 5V regulated VDD to bia as intern nal circuitry fo or VIN ranging from 5.5V to 45V. When VIN iss less than 5.5V, VDD sho ould be tied to o the VIN pins to byypass the internal linear reg gulator. IL(PP) Soft--Start Soft-start reduces s the powerr supply inrush current at a startu up by controlling the outp put voltage riise time while the o output capacittor charges. Eq. 4 The MOSFET RDS(ON) varies 30% to o 40% with h tempe erature; therrefore, it is rrecommended to use the e RDS(OON) at max jun nction temperature with 20% margin to o calcu ulate RILIM in E Equation 3. The M MIC28513 im mplements an internal digita al soft-start by ramp ping up the 0.8V 0 referenc ce voltage (VREF) from 0 to 100% % in about 5m ms with 9.7m mV steps. This controls the outpu ut voltage ratte of rise at turn on, minimizing inrush curre ent and eliminating outputt voltage ove ershoot. Once the ssoft-start cycle ends, the related r circuittry is disabled to red duce current consumption. Marcch 25, 2015 VOU UT VIN(MAX ) VOUT VIN(MAX ) fSW W L In ca ase of a harrd short, the current limitt threshold is s folded d down to a allow an indefinite hard short on the e outpu ut without any destructive e effect. It is mandatory to o make e sure that th he inductor ccurrent used to charge the e outpu ut capacitor during soft-sta art is under th he folded shorrt limit; otherwise the supply will go into hicccup mode and d may n not finish the soft-start succcessfully. 17 Revision 1.2 2 Micrel, Inc. MIC28513 Power Good (PG) The Power Good (PG) pin is an open drain output which indicates logic high when the output is nominally 90% of its steady state voltage. MOSFET Gate Drive The Functional Diagram shows a bootstrap circuit, consisting of DBST, CBST and RBST. This circuit supplies energy to the high-side drive circuit. Capacitor CBST is charged, while the low-side MOSFET is on, and the voltage on the SW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the high-side MOSFET turns on, the voltage on the SW pin increases to approximately VIN. Diode DBST is reverse-biased and CBST floats high while continuing to bias the high-side gate driver. The bias current of the high-side driver is less than 10mA so a 0.1μF to 1μF is sufficient to hold the gate voltage with minimal droop for the power stroke (highside switching) cycle, i.e. ∆BST = 10mA × 1.25μs/0.1μF = 125mV. When the low-side MOSFET is turned back on, CBST is then recharged through the boost diode. A 30Ω resistor RBST, which is in series with the BST pin, is required to slow down the turn-on time of the high-side Nchannel MOSFET. March 25, 2015 18 Revision 1.2 Micre el, Inc. MIC28513 3 App plication Informatio on Outp put Voltage Setting S Comp ponents The MIC28513 re equires two resistors r to set s the outpu ut voltage as shown in Figure 5. VIN N MIC28513 R19 9 FR REQ R17 7 GND Figure 6. S Switching Freq quency Adjus stment Figure 5. 5 Voltage Divider Configura ation ation 7 gives the estimated switching fre equency. Equa The o output voltage e is determine ed by Equatio on 5. R1 VOUT VFB 1 R2 7 R17 fSW f0 R17 R19 Eq. 7 Eq. 5 Wherre f0 = switching fre equency whe en R17 is o open, 600kHz z typica ally. Where: VFB = 0.8V. Figurre 7 shows th he switch freq quency versu us the resisto or R17 w when R19 = 1 100kΩ. A typ pical value of o R1 used on o the standa ard evaluation board d is 10kΩ. If R1 is too larg ge, it may allo ow noise to be introd duced into th he voltage fe eedback loop p. If R1 is too small, it will decre ease the effic ciency of the power supply y, espe ecially at light loads. Once R1 is selecte ed, R2 can be calcu ulated using Equation E 6. R2 VFB R1 VOUT VFB Eq. 6 Setting the Switc ching Freque ency The MIC28513 sw witching frequency can be adjusted by b changing the resis stor divider ne etwork from VIN. V Figure 7 7. Switching Frrequency vs. R17 Marcch 25, 2015 19 Revision 1.2 2 Micrel, Inc. MIC28513 The winding resistance must be minimized, although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by using Equation 11. Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by: L VOUT VIN(MAX ) VOUT VIN(MAX ) IL(PP) fSW PL(Cu) = IL(RMS)2 × DCR The resistance of the copper wire, DCR, increases with the temperature. The value of the winding resistance used should be at the operating temperature. Eq. 8 DCR(HT) = DCR20C × (1 + 0.0042 × (TH T20C)) Eq. 12 Where: Where: TH = temperature of the wire under full load fSW = switching frequency T20C = ambient temperature L(PP) = The peak-to-peak inductor current ripple, Typically 20% of the maximum output current. DCR(20C) = room temperature winding resistance (usually specified by the manufacturer). In the continuous conduction mode, the peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. IL(PK ) IOUT 0.5 IL(PP) Output Capacitor Selection The type of the output capacitor is usually determined by its equivalent series resistance (ESR). Voltage and RMS current capability are also important factors in selecting an output capacitor. Recommended capacitor types are ceramic, tantalum, low-ESR aluminum electrolytic, OSCON and POSCAP. For high ESR electrolytic capacitors, ESR is the main cause of the output ripple. The output capacitor ESR also affects the control loop from a stability point of view. For a low ESR ceramic output capacitor, ripple is dominated by the reactive impedance. Eq. 9 The RMS inductor current is used to calculate the I2R losses in the inductor. 2 IL(RMS) I OUT(MAX ) I2L(PP) I2 The maximum value of ESR is calculated with Equation 13. Eq. 10 ESR COUT Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC28513 requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used, but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. March 25, 2015 Eq. 11 VOUT(PP) IL(PP) Eq. 13 Where: ΔVOUT(pp) = peak-to-peak output voltage ripple ∆IL(PP) = peak-to-peak inductor current ripple 20 Revision 1.2 Micrel, Inc. MIC28513 The total output ripple is a combination of the ESR and output capacitance. The total ripple is calculated by Equation 14. 2 IL(PP) IL(PP) ESRCOUT VOUT(PP) C OUT fSW 8 Input Capacitor Selection The input capacitor for the power stage input VIN should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. A tantalum input capacitor’s voltage rating should be at least two times the maximum input voltage to maximize reliability. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de-rating. The input voltage ripple will primarily depend on the input capacitor’s ESR. The peak input current is equal to the peak inductor current, so: 2 Eq. 14 Where D = duty cycle COUT = output capacitance value fSW = switching frequency. VIN IL(PK ) ESR CIN As described in the “Theory of Operation” subsection in the Functional Description section, the MIC28513 requires at least 20mV peak-to-peak ripple at the FB pin for the gm amplifier and the error comparator to operate properly. Also, the ripple on FB pin should be in phase with the inductor current. Therefore, the output voltage ripple caused by the output capacitors value should be much smaller than the ripple caused by the output capacitor ESR. If low-ESR capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provide the enough feedback voltage ripple. Please refer to the “Ripple Injection” sub-section for more details. The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor current ripple is low: ICIN(RMS) IOUT(MAX ) D 1 D IL(PP) 12 Eq. 18 The power dissipated in the input capacitor is: The voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or OS-CON. The output capacitor RMS current is calculated by Equation 15. ICOUT(RMS) Eq. 17 PDISS(CIN) I2 CIN(RMS) ESR CIN Eq. 19 Eq. 15 The power dissipated in the output capacitor is calculated using Equation 16. PDISS(COUT ) I2 COUT (RMS) ESR COUT March 25, 2015 Eq. 16 21 Revision 1.2 Micre el, Inc. MIC28513 3 Ripp ple Injection The VFB ripple required for proper ope eration of the MIC2 28513’s gm am mplifier and error e compara ator is 20mV to 100m mV. Howeverr, the output voltage ripple is generally desig gned as 1% to 2% of the t output voltage. v If the feedb back voltage ripple is so small that the gm amplifie er and error comparrator can’t se ense it, then the t MIC28513 will lo ose control an nd the output voltage is no ot regulated. In orderr to have some s amoun nt of VFB rip pple, a ripple injecttion method is applied for low output voltage ripple applications. Figu ure 8. Enough Ripple at FB The applications s are divide ed into three situations according to the amount a of the feedback volltage ripple: 1. E Enough ripple e at the feedback voltage due d to the la arge ESR of the t output capacitors. A As shown in Figure F 8, the converter c is stable without a any ripple inje ection. The fee edback voltag ge ripple is: 2. R2 ESR COUT IL(PP C P) R1 R 2 VFB(PP) Eq. 20 Figure e 9. Inadequatte Ripple at FB B W Where: ∆ ∆IL(pp) is the peak-to-pea ak value of the inducto or ccurrent ripple. 3. Inadequate rip pple at the fe eedback volta age due to the ssmall ESR of the output ca apacitors. T The output voltage v ripple e is fed into o the FB pin through a feed-forward cap pacitor, Cff in this situation n, a as shown in n Figure 9. The typical Cff value is d determined by y the following g equation. R1 CFF 0 10 fSW W Eq. 21 Figurre 10. Invisible e Ripple at FB B W With the feed--forward capa acitor, the fee edback voltage rripple is very close c to the output o voltage e ripple. VFB(PP) ES SR COUT IL(PP) Eq. 22 4. V Virtually no riipple at the FB pin voltag ge due to the vvery low ESR of the outputt capacitors. Marcch 25, 2015 22 Revision 1.2 2 Micrel, Inc. MIC28513 In this situation, the output voltage ripple is less than 20mV. Therefore, additional ripple is injected into the FB pin from the switching node SW via a resistor Rinj and a capacitor Cinj, as shown in Figure 10. The injected ripple is: ∆VFB(pp) = VIN × K div × D × (1 - D) × K div R1//R2 Rinj R1//R2 1 fSW × τ The process of sizing the ripple injection resistor and capacitors is as follows. 1. Select Cff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. Typical choice of Cff is 1nF to 100nF if R1 and R2 are in kΩ range. 2. Select Rinj according to the expected feedback voltage ripple using Equation 26. Eq. 23 K div Eq. 24 ∆VFB(pp) VIN fSW τ D (1- D) Eq. 26 Then the value of Rinj is obtained using: Where: VIN = power stage input voltage R inj (R1//R2) ( D = duty cycle fSW = switching frequency τ = (R1//R2//RINJ) × CFF K div 1) Eq. 27 3. Select Cinj as 100nF, which can be considered a short for a wide range of frequencies. It is assumed in Equation 23 and Equation 24 that the time constant associated with Cff must be much greater than the switching period. 1 T 1 fSW 1 Eq. 25 If the voltage divider resistors R1 and R2 are in the kΩ range, a Cff of 1nF to 100nF can easily satisfy the large time constant requirements. Also, a 100nF injection capacitor Cinj is used in order to be considered as short for a wide range of the frequencies. March 25, 2015 23 Revision 1.2 Micrel, Inc. MIC28513 PCB Layout Guidelines PCB Layout is critical to achieve reliable, stable and efficient performance. A ground plane is required to control EMI and minimize the inductance in power, signal and return paths. If a Tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. The following figures optimized from small form factor point of view shows top and bottom layer of a four layer PCB. It is recommended to use mid layer 1 as a continuous ground plane. In “Hot-Plug” applications, a Tantalum or Electrolytic bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is suddenly applied. SW Node Do not route any digital lines underneath or close to the SW node. Keep the switch node (SW) away from the feedback (FB) pin. Output Capacitor Use a copper island to connect the output capacitor ground terminal to the input capacitor ground terminal. Phase margin will change as the output capacitor value and ESR changes. Contact the factory if the output capacitor is different from what is shown in the BOM. The feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. Sensing a long high-current load trace can degrade the DC load regulation. Figure 11. Top and Bottom Layer of a Four Layer Board Thermal Measurements Measuring the IC’s case temperature is recommended to insure it is within its operating limits. Although this might seem like a very elementary task, it is easy to get erroneous results. The most common mistake is to use the standard thermal couple that comes with a thermal meter. This thermal couple wire gauge is large, typically 22 gauge, and behaves like a heatsink, resulting in a lower case measurement. The following guidelines should be followed to ensure the proper operation of the MIC28513 converter. IC The analog ground pin (AGND) must be connected directly to the ground planes. Do not route the AGND pin to the PGND pin on the top layer. Place the IC close to the point of load (POL). Use copper planes to route the input and output power lines. Analog and power grounds should be kept separate and connected at only one location. Two methods of temperature measurement are using a smaller thermal couple wire or an infrared thermometer. If a thermal couple wire is used, it must be constructed of 36 gauge wire or higher than (smaller wire size) to minimize the wire heat-sinking effect. In addition, the thermal couple tip must be covered in either thermal grease or thermal glue to make sure that the thermal couple junction is making good contact with the case of the IC. Omega brand thermal couple (5SC-TT-K-36-36) is adequate for most applications. Input Capacitor Place the input capacitors on the same side of the board and as close to the PVIN and PGND pins as possible. Place several vias to the ground plane close to the input capacitor ground terminal. Use either X7R or X5R dielectric input capacitors. Do not use Y5V or Z5U type capacitors. Do not replace the ceramic input capacitor with any other type of capacitor. Any type of capacitor can be placed in parallel with the input capacitor. March 25, 2015 Wherever possible, an infrared thermometer is recommended. The measurement spot size of most infrared thermometers is too large for an accurate reading on a small form factor ICs. However, a IR thermometer from Optris has a 1mm spot size, which makes it a good choice for measuring the hottest point on the case. An optional stand makes it easy to hold the beam on the IC for long periods of time. 24 Revision 1.2 Micre el, Inc. MIC28513 3 MIC C28513 Ev valuation Board B Sch hematic Bill of Materials Item m Part Number Man nufacturer Niichicon (6) Qty. D Description C1 UVZ2A3 330MPD 3 33µF/100V 20% % Radial Aluminum Capacito or 1 C2, C3 12061Z4 475KAT2A AVX(7) 4 4.7µF/100V, X7 7S, Size 1206 Ceramic Capa acitor 2 C4, C7 C1608X X7R1A225K080 0AC TDK(8) 2 2.2µF/10V, X7R R, Size 0603 C Ceramic Capaccitor 2 O Open NA 0 0.1µF/10V, X7R R, Size 0603 C Ceramic Capaccitor 2 0 0.47µF/100V, X X7R, Size 0805 5 Ceramic Cap pacitor 1 0 0.1µF/100V, X7 7R, Size 0603 Ceramic Capa acitor 2 C5, C13 C6, C16 C9 C10 0, C17 C0603C C104K8RACTU U GRM21B BR72A474KA7 73 08051C4 474KAT2A GRM188 8R72A104KA3 35D Kemet K (9) (10) Murata M AVX Murata C11 C12 2 CGA3E2 2X7R1H471K C14 4, C15 GRM32E ER71A476KE1 15L O Open NA TDK 4 470pF/50V, X7 7R, Size 0603 C Ceramic Capaccitor 1 Murata 4 47µF/10V, X7R R, Size 1210 Ceramic Capacitor 2 C18 8 O Open NA C19 9 O Open NA C20 0 O Open NA C21 O Open 1 Notes s: 6. Niichicon: www.nic chicon.co.jp/engliish. 7. AV VX: www.avx.com m. 8. TD DK: www.tdk.com m. 9. Ke emet: www.keme et.com. 10. Murata: www.mura ata.com. Marcch 25, 2015 25 Revision 1.2 2 Micrel, Inc. MIC28513 Bill of Materials (Continued) Item D1 Part Number BAT46W-TP D3 MMSZ5231B-7-F J1, J7, J8, J10, J11, J12, J16, J17, J18 77311-118-02LF L1 XAL7030-682MED R1 CRCW060310K0FKEA Manufacturer Description Qty. (11) 100V Small Signal Schottky Diode, SOD123 (12) 5.1V/500MW SOD123 Zener Diode NA CONN HEADER 2POS VERT T/H 9 6.8µH, 10.7A sat current 1 10.0kΩ, Size 0603, 1% Resistor 1 MCC Diode FCI(13) Coilcraft(14) Vishay Dale (15) 1 R2 Open NA R9 Open NA R10 CRCW06033K24FKEA Vishay Dale 3.24kΩ, Size 0603, 1% Resistor 1 R11 CRCW06031K91FKEA Vishay Dale 1.91kΩ, Size 0603, 1% Resistor 1 R14, R15 CRCW06030000FKEA Vishay Dale 0.0 Ω, Size 0603, Resistor Jumper 2 R26 CRCW06030000FKEA Vishay Dale 0Ω, Size 0603, Resistor Jumper NA R16, R19, R17,R3 CRCW0603100K0FKEA Vishay Dale 100kΩ, Size 0603, 1% Resistor 4 R25 CRCW0603100K0FKEA Vishay Dale 100kΩ, Size 0603, 1% Resistor NA R18 CRCW06031K00JNEA Vishay Dale 1.0kΩ, Size 0603, 5% Resistor 1 R20, R21 CRCW060349R9FKEA Vishay Dale 49.9Ω, Size 0603, 1% Resistor 2 R22 CRCW06031K82FKEA Vishay Dale 1.82kΩ, Size 0603, 1% Resistor 1 R23 CRCW08051R21FKEA Vishay Dale 1.21Ω, Size 0805, 1% Resistor 1 R24 CRCW060310R0FKEA Vishay Dale 10.0Ω, Size 0603, 1% Resistor 1 TP1 TP2 Open TP7 TP14 77311-118-02LF FCI CONN HEADER 2POS VERT T/H 1 TP8 TP13 77311-118-02LF FCI CONN HEADER 2POS VERT T/H 1 TP17 TP18 77311-118-02LF FCI CONN HEADER 2POS VERT T/H 1 TP9, TP10, TP11, TP12 1502 Keystone Electronics(16) Testpoint Turret, .090 4 U1 MIC28513-1YFL Micrel Inc.(17) 45V 4A Synchronous Buck Regulator 1 Notes: 11. MCC: www.mccsemi.com. 12. Diode: www.diodes.com. 13. FCI: www.fciconnect.com. 14. Coilcraft: www.coilcraft.com. 15. Vishay Dale: www.vishay.com. 16. Keystone Electronics: www.keyelco.com. 17. Micrel Inc.: www.micrel.com. March 25, 2015 26 Revision 1.2 Micrel, Inc. MIC28513 MIC28513 Evaluation Board Layout Top Layer Mid Layer 1 March 25, 2015 27 Revision 1.2 Micrel, Inc. MIC28513 MIC28513 Evaluation Board Layout (Continued) Mid Layer 2 Bottom Layer March 25, 2015 28 Revision 1.2 Micre el, Inc. MIC28513 3 Pac ckage Info ormation and a Recom mmended d Land Patttern(18) 24-Pin 3mm m × 4mm FCQF FN (FL) Note: 18. Pa ackage information is correct as of o the publication n date. For updattes and most currrent information, go to www.micrel.com. Marcch 25, 2015 29 Revision 1.2 2 Micrel, Inc. MIC28513 MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com Micrel, Inc. is a leading global manufacturer of IC solutions for the worldwide high-performance linear and power, LAN, and timing & communications markets. The Company’s products include advanced mixed-signal, analog & power semiconductors; high-performance communication, clock management, MEMs-based clock oscillators & crystal-less clock generators, Ethernet switches, and physical layer transceiver ICs. Company customers include leading manufacturers of enterprise, consumer, industrial, mobile, telecommunications, automotive, and computer products. Corporation headquarters and state-of-the-art wafer fabrication facilities are located in San Jose, CA, with regional sales and support offices and advanced technology design centers situated throughout the Americas, Europe, and Asia. Additionally, the Company maintains an extensive network of distributors and reps worldwide. Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this datasheet. This information is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual property rights is granted by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability whatsoever, and Micrel disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties relating to fitness for a particular purpose, merchantability, or infringement of any patent, copyright, or other intellectual property right. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. © 2014 Micrel, Incorporated. March 25, 2015 30 Revision 1.2