70V 2A Synchronous Buck Regulator

MIC285
512
7 0V 2A Syn
nchronous Buck Reg
gulator
Gen
neral Desc
cription
Featu
ures
The MIC28512 is
i a synchro
onous step-d
down switching
regullator with inte
ernal power sw
witches capable of providing
up to
o 2A output current from a wide inpu
ut supply range
from 4.6V to 70V. The output voltage
v
is adju
ustable down to
0.8V with a gua
aranteed accuracy of ±1%
%. A consta
ant
switcching frequen
ncy can be prrogrammed from
f
200kHz to
680kkHz. The Hy
yper Speed Control™ and
a
HyperLig
ght
Load
d® architecturres of the MIC28512 allo
ow for high VIN
(low VOUT) opera
ation and ultra-fast trans
sient response
while
e reducing the required
d output capacitance and
providing very goo
od light load efficiency.
e




The MIC28512 offfers a full su
uite of protecttion features to
ensure protection
ns under fault conditions. These include
unde
er-voltage loc
ckout to ensu
ure proper operation
o
und
der
powe
er sag condittions, internal soft-start to
o reduce inrush
curre
ent, fold-back current limit,, “hiccup” mo
ode short-circ
cuit
prote
ection, and the
ermal shutdow
wn.
Datasheets and support
s
docu
umentation arre available on
Micre
el’s web site at:
a www.micre
el.com.








4.6V
V to 70V operrating input vo
oltage supply
Up tto 2A output ccurrent
Integ
grated high-sside and low-sside N-channe
el MOSFETs
Hyp
perLight Load (MIC28512-1) and H
Hyper Speed
d
Con
ntrol (MIC28512-2) architeccture
Ena
able input and
d power good (PGOOD) ou
utput
grammable current limit an
nd foldback ““hiccup” mode
e
Prog
shorrt-circuit prote
ection
Builtt-in 5V regula
ator for single-supply opera
ation
Adju
ustable 200kH
Hz to 680kHz switching fre
equency
Fixe
ed 5ms soft-sttart
Interrnal compenssation and the
ermal shutdow
wn
The rmally enhanced 24-pin
n 3mm × 4
4mm FCQFN
N
packkage
Juncction tempera
ature range off –40°C to +125°C
Appllications





Indu
ustrial power ssupplies
Disttributed supply regulation
wer supplies
Bas e station pow
Wal l transformer regulation
h-voltage sing
gle board systems
High
Typ
pical Application
Efficiency
y (VIN =12V)
vs. Output Currrent MIC28512-1
100
0
5.0V
V
3.3V
V
90
0
EFFICIENCY (%)
80
0
70
0
60
0
50
0
40
0
30
0
FSW = 300kHz
20
0
10
0
0.01
0.1
1
10
OUTPUT CURRENT (A)
Hype
er Speed Control is a trademark of
o Micrel, Inc.
Hype
erLight Load is a registered trade
emark of Micrel, Inc.
I
Micrel Inc. • 2180 Fortune Driv
ve • San Jose, CA
C 95131 • USA • tel +1 (408) 94
44-0800 • fax + 1 (408) 474-1000
0 • http://www.m
micrel.com
Marcch 25, 2015
Revision 1.2
Micre
el, Inc.
MIC28512
2
Ord
dering Info
ormation
Arch
hitecture
Pa
ackage(1)
Junctiion Temperatu
ure Range
Lead Finish
MIC
C28512-1YFL
HyperL
Light Load
24-Pin 3mm
m × 4mm FCQ FN
–40°C to +125
5°C
Pb-Free
MIC
C28512-2YFL
Hyper Sp
peed Control
24-Pin 3mm
m × 4mm FCQ FN
–40°C to +125
5°C
Pb-Free
Partt Number
Note:
1. FC
CQFN is a lead-ffree package. Pb
b-free lead finish is Matte Tin.
Pin Configuration
24-Pin 3mm
m × 4mm FCQF
FN (FL)
(Top View)
Marcch 25, 2015
2
Revision 1.2
2
Micrel, Inc.
MIC28512
Pin Description
Pin Number
Pin Name
1
DL
2
PGND
3
DH
4, 7, 8, 9, 25
(25 is ePad)
PVIN
Power Input Voltage. The PVIN pins supply power to the internal power switch. Connect all PVIN
pins together and locally bypass with ceramic capacitors. The positive terminal of the input
capacitor should be placed as close as possible to the PVIN pins; the negative terminal of the
input capacitor should be placed as close as possible to the PGND pins 10,11, 22, 23, and 26.
5
LX
The LX pin is the return path for the high-side driver circuit. Connect the negative terminal of the
bootstrap capacitor directly to this pin. Also connect this pin to the SW pins 12, 21, and 27, with a
low impedance path. The controller monitors voltages on this pin and the PGND for zero current
detection.
6
BST
Bootstrap Pin. This pin provides bootstrap supply for the high-side gate driver circuit. Connect a
0.1µF capacitor and an optional resistor in series from the LX (Pin 5) to the BST.
10, 11, 22,
23, 26
(26 is ePad)
PGND
Power Ground. These pins are connected to the source of the low-side MOSFET. They are the
return path for the step-down regulator power stage and should be tied together. The negative
terminal of the input decoupling capacitor should be placed as close as possible to these pins.
12, 21, 27
(27 is ePad)
SW
13
AGND
14
FB
Feedback Input. The FB pin sets the regulated output voltage relative to the internal reference.
This pin is connected to a resistor divider from the regulated output such that the FB pin is at 0.8V
when the output is at the desired voltage.
15
PGOOD
Power Good. The power good output is an open drain output requiring an external pull-up resistor
to external bias. This pin is a high impedance open circuit when the voltage at FB pin is higher
than 90% of the feedback reference voltage (typically 0.8V).
16
EN
Enable Input. The EN pin enables the regulator. When the pin is pulled below the threshold, the
regulator will shut down to an ultra-low current state. A precise threshold voltage allows the pin to
operate as an accurate UVLO. Do not tie EN to VDD.
17
VIN
Supply voltage for the internal LDO. The VIN operating voltage range is from 4.6V to 70V. A
ceramic capacitor from VIN to AGND is required for decoupling. The decoupling capacitor should
be placed as close as possible to the supply pin.
18
ILIM
Current Limit Setting. Connect a resistor from this pin to the SW pin node to allow for accurate
current limit sensing programming of the internal low-side power MOSFET.
19
VDD
Internal +5V Linear Regulator. VDD is the internal supply bus for the IC. Connect to an external
1µF bypass capacitor. When VIN is less than 5.5V, this regulator operates in drop-out mode.
Connect VDD to VIN.
20
PVDD
A 5V supply input for the low-side N-channel MOSFET driver circuit that can be tied to VDD
externally. A 1μF ceramic capacitor from PVDD to PGND is recommended for decoupling.
24
FREQ
Switching Frequency Adjust pin. Connect this pin to VIN to operate at 680kHz. Place a resistor
divider network from VIN to the FREQ pin to program the switching frequency.
March 25, 2015
Pin Function
Low-side gate drive: Internal low-side power MOSFET gate connection. This pin must be left
unconnected or floating.
PGND is the return path for the low-side driver circuit. Connect to the source of low-side
MOSFET’s (PGND, Pin 10, 11, 22, 23, and 27) through a low impedance path.
High-side gate drive: Internal high-side power MOSFET gate connection. This pin must be left
unconnected, or floating.
Switch Node. The SW pins are the internal power switch outputs. These pins should be tied
together and connected to the output inductor.
Analog Ground. Signal ground for VDD and the control circuitry. The signal ground return path
should be separate from the power ground (PGND) return path.
3
Revision 1.2
Micrel, Inc.
MIC28512
Absolute Maximum Ratings(2)
Operating Ratings(3)
PVIN, VIN to PGND ........................................ 0.3V to 76V
VDD, PVDD to PGND ....................................... 0.3V to 6V
VBST to VSW, VLX ................................................. 0.3V to 6V
VBST to PGND …………………..…………0.3V to (VIN +6V)
VSW,VLX to PGND ............................... 0.3V to (VIN +0.3V)
VFREQ, VILIM, VEN to AGND .................. 0.3V to (VIN +0.3V)
VFB, VPG, to AGND ............................. 0.3V to (VDD+ +0.3V)
PGND to AGND ............................................ 0.3V to +0.3V
Junction Temperature (TJ) ....................................... +150C
Storage Temperature (TS) ......................... 65C to 150C
Lead Temperature (soldering, 10s) ............................ 300C
ESD HBM Rating(4)...................................................... 1.5kV
ESD MM Rating(4) ......................................................... 150V
Supply Voltage (PVIN, VIN)............................. +4.6V to +70V
Enable Input (VEN) .................................................. 0V to VIN
VSW, VFEQ, VILIM, VEN ............................................... 0V to VIN
Junction Temperature (TJ) ........................ –40°C to +125°C
Junction Thermal Resistance
FQFN (JA) ......................................................... 30°C/W
Electrical Characteristics(5)
VIN = 12V; TA = 25°C, bold values indicate –40°C≤ TA ≤ +125°C, unless noted.
Parameter
Condition
Min.
Typ.
Max.
Units
70
V
Power Supply Input
Input Voltage Range (PVIN, VIN)
Quiescent Supply Current
Shutdown Supply Current
4.6
VFB = 1.5V (MIC28512-1)
0.4
0.75
VFB = 1.5V (MIC28512-2)
0.7
1.5
SW = unconnected, VEN = 0V
0.1
10
µA
mA
VDD Supply
VDD Output Voltage
VIN =7V to 70V, IVDD = 10mA
4.8
5.2
5.4
V
VDD UVLO Threshold
VVDD rising
3.8
4.2
4.6
V
VDD UVLO Hysteresis
400
Load Regulation @40mA
mV
0.6
2
4
%
25ºC (±1.0%)
0.792
0.8
0.808
-40°C ≤ TJ ≤ 125°C (±2%)
0.784
0.8
0.816
5
500
Reference
Feedback Reference Voltage
FB Bias Current
VFB = 0.8V
V
nA
Enable Control
EN Logic Level High
V
1.8
EN Logic Level Low
0.6
EN Hysteresis
EN Bias Current
200
VEN = 12V
23
V
mV
40
µA
Notes:
2. Exceeding the absolute maximum ratings may damage the device.
3. The device is not guaranteed to function outside its operating ratings.
4. Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5kΩ in series with 100pF.
5. Specification for packaged product only.
March 25, 2015
4
Revision 1.2
Micrel, Inc.
MIC28512
Electrical Characteristics(5) (Continued)
VIN = 12V; TA = 25°C, bold values indicate –40°C≤ TA ≤ +125°C, unless noted.
Parameter
Condition
Min.
Typ.
Max.
VFREQ = VIN
450
680
800
Units
Oscillator
Switching Frequency
VFREQ = 50%VIN
340
Maximum Duty Cycle
Minimum Duty Cycle
VFB>0.8V
Minimum Off-time
110
kHz
85
%
0
%
200
270
ns
Internal MOSFETs
High-Side NMOS On-Resistance
77
m
Low-Side NMOS On-Resistance
43
m
Short Circuit Protection
Current-Limit Threshold
VFB = 0.79V
-30
-14
0
mV
Short-Circuit Threshold
VFB = 0V
-24
-7
8
mV
Current-Limit Source Current
VFB = 0.79V
50
70
90
µA
Short-Circuit Source Current
VFB = 0V
25
36
43
µA
50
µA
95
%VOUT
Leakage
SW, BST Leakage Current
Power Good
Power Good Threshold Voltage
Sweep VFB from Low to High
Power Good Hysteresis
Sweep VFB from High to Low
6
%VOUT
Power Good Delay Time
Sweep VFB from Low to High
100
µs
Power Good Low Voltage
VFB < 90% x VNOM, IPGOOD = 1mA
70
TJ Rising
160
°C
15
°C
5
ms
85
90
200
mV
Thermal Protection
Over-Temperature Shutdown
Over-Temperature Shutdown
Hysteresis
Soft Start
Soft Start Time
March 25, 2015
5
Revision 1.2
Micrel, Inc.
MIC28512
Typical Characteristics
VIN Shutdown Current
vs. Input Voltage
VIN Operating Supply Current
vs. Input Voltage MIC28512-1
4.0
3.0
2.0
1.0
ENABLE THRESHOLD (V)
Rising
VOUT = 5V
IOUT = 0A
FSW = 300kHz
SHUTDOWN CURRENT (mA)
30.0
25.0
20.0
15.0
10.0
5.0
5 10 15 20 25 30 35 40 45 50 55 60 65 70
18.0
VIN UVLO Threshold
vs. Temperature
6.0
44.0
57.0
CURRENT LIMIT (A)
5.0
4.5
4.3
Falling
3.9
3.7
INPUT VOLTAGE (V)
Feedback Voltage
vs. Temperature
0.812
4.0
3.0
2.0
0.0
3.3
0
25
50
75
100
-50
125
-25
TEMPERATURE (°C)
Output Voltage
vs. Input Voltage
5.0
4.9
VOUT = 5V
IOUT = 1A
FSW = 300kHz
4.8
ENABLE THRESHOLD (V)
5.1
0
25
50
75
100
5 10 15 20 25 30 35 40 45 50 55 60 65 70
INPUT VOLTAGE (V)
March 25, 2015
0.800
-50
1.3
1.2
Falling
1.0
0.9
0.8
0.7
VIN = 12V
VDD = 5V
0.6
-50
-25
0
25
50
75
TEMPERATURE (°C)
6
25
50
75
100
125
600
Rising
1.1
0
Switching Frequency
vs. Output Current
1.5
1.4
-25
TEMPERATURE (°C)
0.5
4.7
0.804
0.792
125
Enable Threshold
vs. Temperature
1.6
5.2
0.808
TEMPERATURE (°C)
5.3
VIN = 12V
VOUT = 5.0V
IOUT = 0A
0.796
1.0
VIN =12V
IOUT = 0A
-25
Hyst
5 10 15 20 25 30 35 40 45 50 55 60 65 70
70.0
FEEDBACK VOLTAGE (V)
4.7
VIN THRESHOLD (V)
31.0
VIN = 12V
VOUT = 5.0V
FSW = 300kHz
Rising
-50
0.30
Output Peak Current Limit
vs. Temperature
4.9
3.5
0.60
INPUT VOLTAGE (V)
INPUT VOLTAGE (V)
4.1
Falling
0.90
0.00
0.0
0.0
1.20
5.0
SWITCHING FREQUENCY (kHz)
SUPPLY CURRENT (mA)
1.50
35.0
5.0
OUTPUT VOLTAGE (V)
Enable Threshold vs.
Input Voltage
100
125
550
500
450
400
350
300
250
200
VIN = 12V
VOUT = 5V
150
100
0.0
0.5
1.0
1.5
2.0
OUTPUT CURRENT (A)
Revision 1.2
Micrel, Inc.
MIC28512
Typical Characteristics (Continued)
Output Voltage
vs. Output Current
Efficiency (VIN =12V)
vs. Output Current MIC28512-1
5.2
100
100
5.0V
3.3V
4.9
70
60
50
40
0.0
0.5
1.0
1.5
10
0.01
OUTPUT CURRENT (A)
0.1
1
20.0
SUPPLY CURRENT (mA)
5.0V
3.3V
70
60
50
40
30
FSW = 300kHz
16.0
14.0
12.0
10.0
8.0
6.0
4.0
VOUT = 5V
IOUT = 0A
FSW = 300kHz
2.0
1
10
0.804
0.800
0.796
0.792
-50
100
5.2
100
5.0V
3.3V
5.0
70
60
50
40
30
4.9
4.8
March 25, 2015
75
100
125
2.0
10
0.01
Efficiency (VIN =24V)
vs. Output Current MIC28512-2
5.0V
3.3V
70
60
50
40
30
FSW = 300kHz
20
VIN = 12V
VOUT = 5V
OUTPUT CURRENT (A)
50
80
EFFICIENCY (%)
EFFICIENCY (%)
5.1
1.5
25
90
80
1.0
0
TEMPERATURE (°C)
Efficiency (VIN =12V)
vs. Output Current MIC28512-2
90
0.5
-25
INPUT VOLTAGE (V)
Output Voltage
vs. Output Current MIC28512-2
0.0
10
VIN = 12V
VOUT = 5.0V
IOUT = 0A
5 10 15 20 25 30 35 40 45 50 55 60 65 70
OUTPUT CURRENT (A)
1
0.808
0.0
0.1
0.1
Feedback Voltage
vs. Temperature MIC28512-2
0.812
18.0
10
0.01
FSW = 300kHz
OUTPUT CURRENT (A)
VIN Operating Supply Current
vs. Input Voltage MIC28512-2
90
20
40
OUTPUT CURRENT (A)
Efficiency (VIN =48V)
vs. Output Current MIC28512-1
80
50
10
0.01
10
FEEDBACK VOLTAGE (V)
100
60
20
FSW = 300kHz
20
2.0
70
30
30
VIN = 12V
VOUT = 5V
5.0V
3.3V
80
EFFICIENCY (%)
5.0
4.8
EFFICIENCY (%)
90
80
5.1
EFFICIENCY (%)
OUTPUT VOLTAGE (V)
90
OUTPUT VOLTAGE (V)
Efficiency (VIN =24V)
vs. Output Current MIC28512-1
0.1
1
OUTPUT CURRENT (A)
7
FSW = 300kHz
20
10
10
0.01
0.1
1
10
OUTPUT CURRENT (A)
Revision 1.2
Micrel, Inc.
MIC28512
Typical Characteristics (Continued)
1.0
90
5.0V
3.3V
70
60
50
40
30
FSW = 300kHz
20
10
0.01
VIN =12V
fSW = 300kHz
0.8
0.6
0.4
5.0V
3.3V
0.2
0.0
0.1
1
0.5
1
1.5
5.0V
3.3V
0.6
0.4
0.2
2
0
0.5
OUTPUT CURRENT (A)
Vin =48V
=24V
fSW = 300kHz
5.0V
3.3V
1.2
0.8
1.5
2
24V Input Thermal Derating
MIC28512-1
2.5
2.0
5.0V
3.3V
1.5
Vin =24V
=12V
fSW = 300kHz
Tjmax =125°C
Θja = 30°C/W
40°C/W
1.0
1
OUTPUT CURRENT (A)
12V Input Thermal Derating
2.5
2.4
1.6
0.8
OUTPUT CURRENT (A)
IC Power Dissipation
vs. Output Current
2.0
VIN =24V
Vin
=24V
fSW = 300kHz
1.0
0.0
0
10
OUTPUT CURRENT (A)
OUTPUT CURRENT (A)
EFFICIENCY (%)
80
1.2
IC POWER DISSIPATION (W)
IC POWER DISSIPATION (W)
100
IC POWER DISSIPATION (W)
IC Power Dissipation
vs. Output Current
IC Power Dissipation
vs. Output Current
Efficiency (VIN =48V)
vs. Output Current MIC28512-2
0.5
2.0
5.0V
3.3V
1.5
Vin =24V
fSW = 300kHz
Tjmax =125°C
Θja = 30°C/W
1.0
0.5
0.4
0.0
0.0
0.0
0
0.5
1
1.5
25
2
40
55
70
85
100
AMBIENT TEMPERATURE (°C)
25
40
55
70
85
100
AMBIENT TEMPERATURE (°C)
OUTPUT CURRENT (A)
48V Input Thermal Derating
MIC28512-1
Switching Frequency
vs. Input Voltage (MIC2812-1)
600
SWITCHING FREQUENCY (KHz)
OUTPUT CURRENT (A)
2.5
2.0
5.0V
3.3V
1.5
Vin =48V
fSW = 300kHz
Tjmax =125°C
Θja = 30°C/W
1.0
0.5
0.0
25
40
55
70
85
AMBIENT TEMPERATURE (°C)
100
590
580
570
560
550
540
530
520
510
500
7.0
16.0
25.0
34.0
43.0
52.0
61.0
70.0
INPUT VOLTAGE (V)
March 25, 2015
8
Revision 1.2
Micre
el, Inc.
MIC28512
2
Fun
nctional Characteris
stics
Marcch 25, 2015
9
Revision 1.2
2
Micre
el, Inc.
MIC28512
2
Fun
nctional Characteris
stics (Con
ntinued)
Marcch 25, 2015
10
Revision 1.2
2
Micre
el, Inc.
MIC28512
2
Fun
nctional Characteris
stics (Con
ntinued)
Marcch 25, 2015
11
Revision 1.2
2
Micre
el, Inc.
MIC28512
2
Fun
nctional Characteris
stics (Con
ntinued)
Marcch 25, 2015
12
Revision 1.2
2
Micre
el, Inc.
MIC28512
2
Fun
nctional Characteris
stics (Con
ntinued)
Marcch 25, 2015
13
Revision 1.2
2
Micre
el, Inc.
MIC28512
2
Fun
nctional Characteris
stics (Con
ntinued)
Marcch 25, 2015
14
Revision 1.2
2
Micre
el, Inc.
MIC28512
2
Fun
nctional Diagram
Marcch 25, 2015
15
Revision 1.2
2
Micrel, Inc.
MIC28512
It is not recommended to use MIC28512 with an OFFtime close to tOFF(min) during steady-state operation.
Functional Description
The MIC28512 is an adaptive on-time synchronous buck
regulator with integrated high-side and low-side
MOSFETs suitable for high-input voltage to low-output
voltage conversion applications. It is designed to operate
over a wide input voltage range (4.6V to 70V) which is
suitable for automotive and industrial applications. The
output is adjustable with an external resistive divider. An
adaptive on-time control scheme is employed to produce
a constant switching frequency in continuous-conduction
mode and reduced switching frequency in discontinuousoperation mode, improving light load efficiency.
Overcurrent protection is implemented by sensing lowside MOSFET’s RDS(ON). The device features internal softstart, enable, UVLO, and thermal shutdown.
The adaptive ON-time control scheme results in a
constant switching frequency in the MIC28512. The
actual ON-time and resulting switching frequency will
vary with the different rising and falling times of the
external MOSFETs. Also, the minimum tON results in a
lower switching frequency in high VIN to VOUT applications.
During load transients, the switching frequency is
changed due to the varying OFF-time.
Figure 1 shows the allowable range of the output voltage
versus the input voltage. The minimum output voltage is
0.8V which is limited by the reference voltage. The
maximum output voltage is 24V which is limited by the
internal circuitry.
Theory of Operation
As illustrated in the Functional Diagram, the output
voltage of the MIC28512 is sensed by the feedback (FB)
pin via the voltage dividers R1 and R2, and compared to
a 0.8V reference voltage VREF at the error comparator
through a low-gain transconductance (gM) amplifier. If the
feedback voltage decreases and the amplifier output is
below 0.8V, the error comparator will trigger the control
logic and generate an ON-time period. The ON-time
period length is predetermined by the “Fixed tON
Estimator” circuitry:
t ON(ESTIMATED ) 
VOUT
VIN  fSW
Output Voltage Rrange
vs. Input Voltage
OUTPUIT VOLTAGE (V)
30
Eq. 1
25
Fsw = 600kHz
20
Fsw = 400kHz
Fsw = 200kHz
15
ALLOWABLE RANGE
10
0.8V (MINIMUM)
5
0
5
18
31
44
57
70
INPUT VOLTAGE (V)
where VOUT is the output voltage, VIN is the power stage
input voltage, and fSW is the switching frequency.
Figure 1. Allowable Output Voltage Range vs. Input Voltage
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in most
cases. When the feedback voltage decreases and the
output of the gM amplifier is below 0.8V, the ON-time
period is triggered and the OFF-time period ends. If the
OFF-time period determined by the feedback voltage is
less than the minimum OFF-time tOFF(min), which is about
200ns (typical), the MIC28512 control logic will apply the
tOFF(min) instead. The tOFF(min) is required to maintain
enough energy in the boost capacitor (CBST) to drive the
high-side MOSFET.
To illustrate the control loop operation, both the steadystate and load transient scenarios will be analyzed.
Figure 2 shows the MIC28512 control loop timing during
steady-state operation. During steady-state, the gM
amplifier senses the feedback voltage ripple, which is
proportional to the output voltage ripple and the inductor
current ripple, to trigger the ON-time period. The ON-time
is predetermined by the tON estimator. The termination of
the OFF-time is controlled by the feedback voltage. At the
valley of the feedback voltage ripple, which occurs when
VFB falls below VREF, the OFF period ends and the next
ON-time period is triggered through the control logic
circuitry.
The maximum duty cycle is obtained from:
DMAX  1  t OFF(MIN)  fSW
March 25, 2015
Eq. 2
16
Revision 1.2
MIC28512
2
Micre
el, Inc.
e true current-mode contrrol, the MIC28
8512 uses the
e
Unlike
outpu
ut voltage rip
pple to trigger an ON-time
e period. The
e
outpu
ut voltage riipple is proportional to the inducto
or
curre nt ripple if th
he ESR of the
e output capacitor is large
e
enoug
gh. The MIC
C28512 contro
ol loop has the advantage
e
of elim
minating the n
need for slope
e compensation.
In orrder to meet stability req
quirements, th
he MIC28512
2
feedb
back voltage ripple shou
uld be in ph
hase with the
e
inducctor current rip
pple and larg
ge enough to be sensed by
y
the gm amplifier and the error comparator. The
e
recom
mmended fee
edback voltage ripple is 20mV~100mV.
If a low-ESR ou
utput capacittor is selectted, then the
e
feedb
back voltage rripple may be
e too small to be sensed by
y
the g m amplifier a
and the erro
or comparatorr. Also, if the
e
ESR of the outp
put capacitorr is very low
w, the outpu
ut
voltag
ge ripple and
d the feedba
ack voltage rripple are no
ot
necesssarily in pha
ase with the inductor currrent ripple. In
n
these
e cases, ripple injection is required to e
ensure prope
er
opera
ation. Please refer to the ““Ripple Injectiion” section in
n
Appliccation Inform
mation for mo
ore details ab
bout the ripple
e
injecttion technique
e.
Figure 2.. MIC28512 Co
ontrol Loop Timing
Figurre 3 shows th
he operation of the MIC28
8512 during a
load transient. The
T
output voltage
v
drops
s due to the
ease, which causes the VFB to be les
ss
sudden load incre
than VREF. This will
w cause the error comparrator to trigge
er
an O
ON-time perio
od. At the end
d of the ON-ttime period, a
minim
mum OFF-tim
me tOFF(min) is generated to
o charge CBSST
since
e the feedbac
ck voltage is still below VREF. Then, the
next ON-time periiod is triggere
ed due to the low feedbac
ck
voltage. Thereforre, the switc
ching freque
ency change
es
durin
ng the load tra
ansient, but returns
r
to the nominal fixed
frequ
uency once th
he output has
s stabilized att the new load
curre
ent level. With the varying
g duty cycle and switching
frequ
uency, the outtput recovery
y time is fast and
a the outpu
ut
voltage deviation is small in MIC28512 conv
verter.
Disco
ontinuous M
Mode (MIC285
512-1 Only)
ductor curre
In co
ontinuous mode, the ind
ent is always
greatter than zero; however, at light loads th
he MIC28512
21 is able to forcce the inducctor current tto operate in
n
disco
ontinuous mo
ode. Discontin
nuous mode is where the
e
inducctor current fa
alls to zero, a
as indicated b
by trace (IL), is
show
wn in Figure 4. During thiis period, the
e efficiency is
s
optim
mized by shuttting down all the non-esssential circuits
and minimizing tthe supply ccurrent. The MIC28512-1
wake
es up and turn
ns on the hig
gh-side MOSF
FET when the
e
feedb
back voltage VFB drops bellow 0.8V.
The MIC28512-1 has a zero crossing comparator tha
at
monittors the inducctor current b
by sensing the
e voltage drop
p
acrosss the low-sid
de MOSFET during its O
ON-time. If the
e
VFB > 0.8V and the
e inductor currrent goes slig
ghtly negative
e,
then tthe MIC28512-1 automatically powers down most of
o
the IC
C circuitry and
d goes into a low-power m
mode.
512-1 goes into discontiinuous mode
Once
e the MIC285
e,
both DH and DL are low, which turns off the high-side
e
and l ow-side MOS
SFETs. The lload current iis supplied by
y
the o
output capacittors and VOUTT drops. If the
e drop of VOUT
cause
es VFB to go
o below VREFF, then all th
he circuits will
wake
e up into norrmal continuo
ous mode. F
First, the bias
curre nts of most circuitss reduced during the
e
disco
ontinuous mo
ode are resto
ored, then a tON pulse is
trigge
ered before tthe drivers arre turned on to avoid any
y
possiible glitches. Finally, the high-side drriver is turned
d
on. Figure 4 sshows the control loo
op timing in
n
disco
ontinuous mod
de.
Figure 3. MIC28512 Load Transient Re
esponse
Marcch 25, 2015
17
Revision 1.2
2
Micre
el, Inc.
2
MIC28512
Curre
ent Limit
The M
MIC28512 usses the RDS(OON) of the inte
ernal low-side
e
powe
er MOSFET to
o sense overrcurrent condiitions. In each
h
switch
hing cycle, the inducto
or current iss sensed by
y
monittoring the low
w-side MOSF
FET during itts ON period
d.
The ssensed voltag
ge V(ILIM) is compared w
with the powe
er
groun
nd (PGND) after a blanking time off 150ns. The
e
voltag
ge drop of th
he resistor RILIM is compared with the
e
low-sside MOSFET
T voltage dro
op to set the
e over-curren
nt
trip le
evel. The sma
all capacitor cconnected fro
om ILIM pin to
o
PGND
D can be add
ded to filter tthe switching node ringing
g,
allow
wing a betterr short limit measureme
ent. The time
e
consttant created by RILIM and the filter cap
pacitor should
d
uch less than
be mu
n the minimum
m off time.
The over currentt limit can b
be programm
med by using
g
Equa
ation 3.
R ILIM
M 
ICLIM  0.5  IL(PP)  R DS(ON)  VCL
ICL
Figure 4. MIC28512-1 Control
C
Loop Mode
M
(Discontinuous Mode)
Eq. 3
Wherre:
Durin
ng discontinu
uous mode, the bias current of mos
st
circuits are reduc
ced. As a res
sult, the total power supply
curre
ent during dis
scontinuous mode
m
is only about 450μA
A,
allow
wing the MIC
C28512-1 to achieve high
h efficiency in
light load applicatiions.
ICLIM = Desired currrent limit.
RDS(OON) = On-ressistance of llow-side pow
wer MOSFET
T
40mΩ
Ω (typical).
VCL = Current-lim
mit threshold
d 14mV (typ
pical absolute
e
value
e). See the Ele
ectrical Chara
acteristics(5) ta
able.
VDD Regulator
The MIC28512 provides
p
a 5V regulated VDD to bia
as
intern
nal circuitry VIN ranging fro
om 5.5V to 70V.
7
When VIN
I
is lesss than 5.5V, VDD should
d be tied to th
he VIN pins to
bypa
ass the interna
al linear regulator.
ICL = Current-limit source curre
ent 70µA (typ
pical). See the
e
Electr
trical Characte
eristics(5) table
e.
∆IL(PPP) = Inductor ccurrent peak-to-peak (use Equation 4 to
o
calcu
ulate the inducctor ripple currrent).
Soft--Start
Soft-start reduces
s the powerr supply inrush current at
a
up by controlling the outp
put voltage riise time while
startu
the o
output capacittor charges.
The p
peak-to-peak inductor currrent ripple is:
 IL(PP) 
The M
MIC28512 im
mplements an internal digita
al soft-start by
ramp
ping up the 0.8V
0
referenc
ce voltage (VREF) from 0 to
100%
% in about 5m
ms with 9.7m
mV steps. This controls the
outpu
ut voltage ratte of rise at turn on, minimizing inrush
curre
ent and eliminating outputt voltage ove
ershoot. Once
the ssoft-start cycle ends, the related
r
circuittry is disabled
to red
duce current consumption.

VOU
UT  VIN(MAX )  VOUT
VIN(MAX )  fSW
W L

Eq. 4
The MOSFET R
RDS(ON) va
aries 30% tto 40% with
h
tempe
erature; therrefore, it is rrecommended to use the
e
RDS(OON) at max jun
nction temperature with 20% margin to
o
calcu
ulate RILIM in:
In ca
ase of a harrd short, the current limitt threshold is
s
folded
d down to a
allow an indefinite hard short on the
e
outpu
ut without any destructive
e effect. It is mandatory to
o
make
e sure that th
he inductor ccurrent used to charge the
e
outpu
ut capacitor during soft-sta
art is under th
he folded shorrt
limit; otherwise, th
he supply will go into hicccup mode and
d
may n
not finish the soft-start succcessfully.
Marcch 25, 2015
18
Revision 1.2
2
Micrel, Inc.
MIC28512
Power Good (PG)
The power good (PG) pin is an open drain output which
indicates logic high when the output is nominally 90% of
its steady state voltage.
MOSFET Gate Drive
The Functional Diagram shows a bootstrap circuit,
consisting of DBST, CBST and RBST. This circuit supplies
energy to the high-side drive circuit. Capacitor CBST is
charged, while the low-side MOSFET is on, and the
voltage on the SW pin is approximately 0V. When the
high-side MOSFET driver is turned on, energy from CBST
is used to turn the MOSFET on. As the high-side
MOSFET turns on, the voltage on the SW pin increases
to approximately VIN. Diode DBST is reverse-biased and
CBST floats high while continuing to bias the high-side
gate driver. The bias current of the high-side driver is less
than 10mA so a 0.1μF to 1μF is sufficient to hold the gate
voltage with minimal droop for the power stroke (highside switching) cycle, i.e. ∆BST = 10mA × 1.25μs/0.1μF =
125mV. When the low-side MOSFET is turned back on,
CBST is then recharged through the boost diode. A 30Ω
resistor RBST, which is in series with the BST pin, is
required to slow down the turn-on time of the high-side Nchannel MOSFET.
March 25, 2015
19
Revision 1.2
Micre
el, Inc.
MIC28512
2
App
plication Informatio
on
VIN
N
Outp
put Voltage Setting
S
Comp
ponents
The MIC28512 re
equires two resistors
r
to set
s the outpu
ut
voltage as shown in Figure 5.
MIC28512
R3
FR EQ
R4
GND
Figure 6. S
Switching Freq
quency Adjus
stment
g
The following formula gives the estimatted switching
frequ ency.
Figure 5.
5 Voltage Divider Configura
ation
The o
output voltage
e is determine
ed by Equatio
on 5.
R1 

VOUT  VFB   1 

R
2

 R4 
fSW  f0  

 R3  R 4 
Eq. 5 Eq. 7 Wherre:
f0 = switching frrequency wh
hen R4 is o
open, 680kHz
z
typica
ally.
Where:
VFB = 0.8V.
Figurre 7 shows th
he switch freq
quency versu
us the resisto
or
R4 w
when R3 = 100
0kΩA.
A typ
pical value of
o R1 used on
o the standa
ard evaluation
board
d is 10kΩ. If R1 is too larg
ge, it may allo
ow noise to be
introd
duced into th
he voltage fe
eedback loop
p. If R1 is too
small, it will decre
ease the effic
ciency of the power supply
y,
espe
ecially at light loads. Once R1 is selecte
ed, R2 can be
calcu
ulated using Equation
E
6.
R2 
VFB
F  R1
VOU
UT  VFB
Eq. 6
Setting the Switc
ching Freque
ency
The MIC28512 switching
s
fre
equency can be adjusted
between 200kHz and 680kHz
z by changing the resisto
or
divide
er network fro
om VIN.
.
Figure 7
7. Switching F
Frequency vs. R4
Marcch 25, 2015
20
Revision 1.2
2
Micrel, Inc.
MIC28512
The winding resistance must be minimized, although this
usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of the
core and copper losses. At higher output loads, the core
losses are usually insignificant and can be ignored. At
lower output currents, the core losses can be a significant
contributor. Core loss information is usually available
from the magnetics vendor. Copper loss in the inductor is
calculated by using Equation 11:
Inductor Selection
Values for inductance, peak, and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine the
peak-to-peak inductor ripple current. Generally, higher
inductance values are used with higher input voltages.
Larger peak-to-peak ripple currents will increase the
power dissipation in the inductor and MOSFETs. Larger
output ripple currents will also require more output
capacitance to smooth out the larger ripple current.
Smaller peak-to-peak ripple currents require a larger
inductance value and therefore a larger and more
expensive inductor. A good compromise between size,
loss and cost is to set the inductor ripple current to be
equal to 20% of the maximum output current. The
inductance value is calculated by:
L

VOUT  VIN(MAX )  VOUT
VIN(MAX )  IL(PP)  fSW

PL(Cu) = IL(RMS)2 × DCR
Eq. 11
The resistance of the copper wire, DCR, increases with
the temperature. The value of the winding resistance
used should be at the operating temperature.
DCR(HT) = DCR20C × (1 + 0.0042 × (TH  T20C))
Eq. 8
Eq. 12
Where:
Where:
TH = temperature of the wire under full load.
fSW = switching frequency.
T20C = ambient temperature.
L(PP) = The peak-to-peak inductor current ripple,
Typically 20% of the maximum output current.
DCR(20C) = room temperature winding resistance (usually
specified by the manufacturer).
In the continuous conduction mode, the peak inductor
current is equal to the average output current plus one
half of the peak-to-peak inductor current ripple.
IL(PK )  IOUT  0.5  IL(PP)
Output Capacitor Selection
The type of the output capacitor is usually determined by
its equivalent series resistance (ESR). Voltage and RMS
current capability are also important factors in selecting
an output capacitor. Recommended capacitor types are
ceramic, tantalum, low-ESR aluminum electrolytic, OSCON and POSCAP. For high ESR electrolytic capacitors,
ESR is the main cause of the output ripple. The output
capacitor ESR also affects the control loop from a
stability point of view. For a low ESR ceramic output
capacitor, ripple is dominated by the reactive impedance.
Eq. 9
The RMS inductor current is used to calculate the I2R
losses in the inductor.
IL(RMS)  I2 OUT(MAX ) 
I2L(PP)
I
2
The maximum value of ESR is calculated with Equation
13.
Eq. 10
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance. The
high frequency operation of the MIC28512 requires the
use of ferrite materials for all but the most cost sensitive
applications. Lower cost iron powder cores may be used,
but the increase in core loss will reduce the efficiency of
the power supply. This is especially noticeable at low
output power. The winding resistance decreases
efficiency at the higher output current levels.
March 25, 2015
ESR COUT 
VOUT(PP)
IL(PP)
Eq. 13
Where:
ΔVOUT(pp) = peak-to-peak output voltage ripple.
∆IL(PP) = peak-to-peak inductor current ripple.
21
Revision 1.2
Micrel, Inc.
MIC28512
The total output ripple is a combination of the ESR and
output capacitance. The total ripple is calculated by
Equation 14.
 2  IL(PP)

  IL(PP)  ESR COUT
VOUT (PP)  
C

 OUT  f SW  8 

2
Input Capacitor Selection
The input capacitor for the power stage input VIN should
be selected for ripple current rating and voltage rating.
Tantalum input capacitors may fail when subjected to
high inrush currents, caused by turning the input supply
on. A tantalum input capacitor’s voltage rating should be
at least two times the maximum input voltage to
maximize reliability. Aluminum electrolytic, OS-CON, and
multilayer polymer film capacitors can handle the higher
inrush currents without voltage de-rating. The input
voltage ripple will primarily depend on the input
capacitor’s ESR. The peak input current is equal to the
peak inductor current, so:
Eq. 14
Where:
D = duty cycle
COUT = output capacitance value
fSW = switching frequency.
VIN  IL(PK )  ESR CIN
As described in the “Theory of Operation” section in the
Functional Description section, the MIC28512 requires at
least 20mV peak-to-peak ripple at the FB pin for the gm
amplifier and the error comparator to operate properly.
Also, the ripple on FB pin should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic capacitors,
are selected as the output capacitors, a ripple injection
method should be applied to provide the enough
feedback voltage ripple. Please refer to the “Ripple
Injection” section for more details.
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming the
peak-to-peak inductor current ripple is low
ICIN(RMS)  IOUT(MAX )  D  1  D 
IL(PP )
12
PDISS(CIN)  I2 CIN(RMS)  ESR CIN
March 25, 2015
Eq. 19
Ripple Injection
The VFB ripple required for proper operation of the
MIC28512’s gm amplifier and error comparator is 20mV to
100mV. However, the output voltage ripple is generally
designed as 1% to 2% of the output voltage. If the
feedback voltage ripple is so small that the gm amplifier
and error comparator can’t sense it, then the MIC28512
will lose control and the output voltage is not regulated. In
order to have some amount of VFB ripple, a ripple
injection method is applied for low output voltage ripple
applications.
Eq. 15
The power dissipated in the output capacitor is calculated
using Equation 16.
PDISS(COUT )  I2 COUT (RMS)  ESR COUT
Eq. 18
The power dissipated in the input capacitor is:
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated by Equation 15.
ICOUT(RMS ) 
Eq. 17
Eq. 16
22
Revision 1.2
Micre
el, Inc.
MIC28512
2
The applications
s are divide
ed into three situations
according to the amount
a
of the feedback volltage ripple:
1. E
Enough ripple
e at the feedback voltage due
d to the
la
arge ESR of the
t output capacitors.
A
As shown in Figure
F
8, the converter is stable withou
ut
a
any ripple inje
ection. The fee
edback voltag
ge ripple is:
R2
 ESR COUT
 IL(PP
C
P)
R1  R2
VFB(PP) 
Eq. 20
2
Figu
ure 8. Enough Ripple at FB
W
Where:
∆
∆IL(pp) is the peak-to-pea
ak value of the inducto
or
ccurrent ripple.
2. Inadequate rip
pple at the fe
eedback volta
age due to the
ssmall ESR of the output ca
apacitors.
T
The output voltage
v
ripple
e is fed into
o the FB pin
through a feed-forward cap
pacitor, Cff in this situation
n,
a
as shown in
n Figure 9. The typical Cff value is
d
determined by
y:
Figure
e 9. Inadequatte Ripple at FB
B
R1  CFF 
0
10
fSW
W
Eq. 21
2
W
With the feed--forward capa
acitor, the fee
edback voltage
rripple is very close
c
to the output
o
voltage
e ripple.
VFB(PP)  ES
SR COUT  IL(PP)
2
Eq. 22
3. V
Virtually no riipple at the FB pin voltag
ge due to the
vvery low ESR of the outputt capacitors.
Marcch 25, 2015
Figurre 10. Invisible
e Ripple at FB
B
23
Revision 1.2
2
Micrel, Inc.
MIC28512
In this situation, the output voltage ripple is less than
20mV. Therefore, additional ripple is injected into the FB
pin from the switching node SW via a resistor Rinj and a
capacitor Cinj, as shown in Figure 10. The injected ripple
is:
∆VFB(pp)  VIN  K div  D  (1- D) 
K div 
1
fSW  
R1//R2
R inj  R1//R2
The process of sizing the ripple injection resistor and
capacitors is:
Step 1. Select Cff to feed all output ripples into the
feedback pin and make sure the large time constant
assumption is satisfied. Typical choice of Cff is 1nF to
100nF if R1 and R2 are in kΩ range.
Step 2. Select Rinj according to the expected feedback
voltage ripple using Equation 26.
Eq. 23
K div 
Eq. 24
∆VFB(pp)
VIN

fSW  
D  (1- D)
Eq. 26
Then the value of Rinj is obtained using:
Where:
VIN = power stage input voltage
Rinj  (R1//R2)  (
D = duty cycle
fSW = switching frequency
τ = (R1//R2//RINJ) × CFF
In
∆VFB(pp)  VIN  K div  D  (1- D) 
K div
 1)
Eq. 27
Step 3. Select CINJ as 100nF, which can be considered a
short for a wide range of frequencies.
1
fSW  
1
and
R1//R2
K div 
R inj  R1//R2
, it is assumed that the time constant
associated with CFF must be much greater than the
switching period:
1
fSW  

T
 1

Eq. 25
If the voltage divider resistors R1 and R2 are in the kΩ
range, a Cff of 1nF to 100nF can easily satisfy the large
time constant requirements. Also, a 100nF injection
capacitor Cinj is used in order to be considered as short
for a wide range of the frequencies.
March 25, 2015
24
Revision 1.2
Micrel, Inc.
MIC28512
Input Capacitor
PCB Layout Guidelines
Warning: To minimize EMI and output noise, follow
these layout recommendations.
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power, signal
and return paths.
Figure 11 is optimized from small form factor point of
view shows top and bottom layer of a four layer PCB. It is
recommended to use mid layer 1 as a continuous ground
plane.

Place the input capacitors on the same side of the
board and as close to the PVIN and PGND pins as
possible.

Place several vias to the ground plane close to the
input capacitor ground terminal.

Use either X7R or X5R dielectric input capacitors. Do
not use Y5V or Z5U type capacitors.

Do not replace the ceramic input capacitor with any
other type of capacitor. Any type of capacitor can be
placed in parallel with the input capacitor.

If a Tantalum input capacitor is placed in parallel with
the input capacitor, it must be recommended for
switching regulator applications and the operating
voltage must be derated by 50%.

In “Hot-Plug” applications, a Tantalum or Electrolytic
bypass capacitor must be used to limit the overvoltage spike seen on the input supply with power is
suddenly applied.
SW Node

Do not route any digital lines underneath or close to
the SW node.

Keep the switch node (SW) away from the feedback
(FB) pin.
Output Capacitor

Use a copper island to connect the output capacitor
ground terminal to the input capacitor ground
terminal.

Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in the
BOM.

The feedback trace should be separate from the
power trace and connected as close as possible to
the output capacitor. Sensing a long high-current load
trace can degrade the DC load regulation.
Figure 11. Top and Bottom Layer of a Four Layer Board
The following guidelines should be followed to ensure the
proper operation of the MIC28512 converter:
IC

The analog ground pin (AGND) must be connected
directly to the ground planes. Do not route the AGND
pin to the PGND pin on the top layer.

Place the IC close to the point of load (POL).

Use copper planes to route the input and output
power lines.

Analog and power grounds should be kept separate
and connected at only one location.
March 25, 2015
25
Revision 1.2
Micrel, Inc.
MIC28512
Thermal Measurements
Measuring the IC’s case temperature is recommended to
insure it is within its operating limits. Although this might
seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heatsink, resulting in a
lower case measurement.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared thermometer. If
a thermal couple wire is used, it must be constructed of
36 gauge wire or higher than (smaller wire size) to
minimize the wire heat-sinking effect. In addition, the
thermal couple tip must be covered in either thermal
grease or thermal glue to make sure that the thermal
couple junction is making good contact with the case of
the IC. Omega brand thermal couple (5SC-TT-K-36-36) is
adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, a IR
thermometer from Optris has a 1mm spot size, which
makes it a good choice for measuring the hottest point on
the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
March 25, 2015
26
Revision 1.2
Micre
el, Inc.
MIC28512
2
MIC
C28512 Ev
valuation Board
B
Sch
hematic
Bill of Materials
Item
m
C1
C2, C3
C4, C7
Part Number
UVZ2A3
330MPD
12061Z4
475KAT2A
C1608X
X7R1A225K080
0AC
Man
nufacturer
Niichicon
(6)
C9
C10
0, C17
C0603C
C104K8RACTU
U
GRM21B
BR72A474KA7
73
08051C4
474KAT2A
GRM188
8R72A104KA3
35D
3
33µF/100V 20%
% Radial Aluminum Capacito
or
1
4
4.7µF/100V, X7
7S, Size 1206 Ceramic Capa
acitor
2
(8)
2
2.2µF/10V, X7R
R, Size 0603 C
Ceramic Capaccitor
2
O
OPEN
NA
0
0.1µF/10V, X7R
R, Size 0603 C
Ceramic Capaccitor
2
0
0.47µF/100V, X
X7R, Size 0805
5 Ceramic Cap
pacitor
1
0
0.1µF/100V, X7
7R, Size 0603 Ceramic Capa
acitor
2
AVX
TDK
(9)
Kemet
K
(10)
Murata
M
AVX
Murata
C11
C12
2
CGA3E2
2X7R1H102K
C14
4, C15
GRM32E
ER71A476KE1
15L
Qty.
(7)
C5, C13
C6, C16
D
Description
O
OPEN
NA
TDK
1
1nF/50V, X7R, Size 0603 Ceramic Capacito
or
1
Murata
4
47µF/10V, X7R
R, Size 1210 Ceramic Capacitor
2
Notes
s:
6. Niichicon: www.nic
chicon.co.jp/engliish.
7. AV
VX: www.avx.com
m.
8. TD
DK: www.tdk.com
m.
9. Ke
emet: www.keme
et.com.
10. Murata: www.mura
ata.com.
Marcch 25, 2015
27
Revision 1.2
2
Micrel, Inc.
MIC28512
Bill of Materials (Continued)
Item
Part Number
Manufacturer
Description
Qty.
C18
Open
NA
C19
Open
NA
C20
Open
NA
Open
NA
C21
D1
BAT46W-TP
(11)
MCC
100V Small Signal Schottky Diode, SOD123
D3
Open
J1, J7, J8,
J10, J11, J12,
J16, J17, J18
77311-118-02LF
L1
XAL7030-682MED
R1
CRCW060310K0FKEA
FCI
(13)
Coilcraft(14)
Vishay Dale
(15)
1
NA
CONN HEADER 2POS VERT T/H
9
8.2µH, 10.2A sat current
1
10.0kΩ, Size 0603, 1% Resistor
1
R2
OPEN
NA
R9
OPEN
NA
R10
CRCW06033K24FKEA
Vishay Dale
3.24kΩ, Size 0603, 1% Resistor
1
R11
CRCW06031K91FKEA
Vishay Dale
1.91kΩ, Size 0603, 1% Resistor
1
R14, R15
CRCW06030000FKEA
Vishay Dale
0.0 Ω, Size 0603, Resistor Jumper
2
CRCW0603100K0FKEA
Vishay Dale
100kΩ, Size 0603, 1% Resistor
R26
R16, R19, R17,R3
Open
R25
Open
NA
4
NA
R18
CRCW06031K00JNEA
Vishay Dale
1.0kΩ, Size 0603, 5% Resistor
1
R20, R21
CRCW060349R9FKEA
Vishay Dale
49.9Ω, Size 0603, 1% Resistor
2
R22
CRCW06031K74FKEA
Vishay Dale
2.21kΩ, Size 0603, 1% Resistor
1
R23
CRCW08051R21FKEA
Vishay Dale
1.21Ω, Size 0805, 1% Resistor
1
R24
CRCW060340R0FKEA
Vishay Dale
40.0Ω, Size 0603, 1% Resistor
1
TP1  TP2
OPEN
TP7  TP14
77311-118-02LF
FCI
CONN HEADER 2POS VERT T/H
1
TP8  TP13
77311-118-02LF
FCI
CONN HEADER 2POS VERT T/H
1
TP17  TP18
77311-118-02LF
FCI
CONN HEADER 2POS VERT T/H
1
TP9, TP10,
TP11, TP12
1502
Keystone
(16)
Electronics
Testpoint Turret, .090
4
U1
MIC28512-1YFL
Micrel. Inc.(17)
70VIN, 2A Synchronous Buck Regulator
1
Notes:
11. MCC: www.mccsemi.com.
12. Diode: www.diodes.com.
13. FCI: www.fciconnect.com.
14. Coilcraft: www.coilcraft.com.
15. Vishay Dale: www.vishay.com.
16. Keystone Electronics: www.keystone.com.
17. Micrel, Inc.: www.micrel.com.
March 25, 2015
28
Revision 1.2
Micrel, Inc.
MIC28512
MIC28512 Evaluation Board
Top Layer
Mid Layer 1
March 25, 2015
29
Revision 1.2
Micrel, Inc.
MIC28512
MIC28512 Evaluation Board (Continued)
Mid Layer 2
Bottom Layer
March 25, 2015
30
Revision 1.2
Micre
el, Inc.
MIC28512
2
Pac
ckage Info
ormation and
a Recom
mmended
d Land Patttern (18)
24-Pin 3mm
m × 4mm FCQF
FN (FL)
Note:
18. Pa
ackage information is correct as of
o the publication
n date. For updattes and most currrent information, go to www.micrel.com.
Marcch 25, 2015
31
Revision 1.2
2
Micrel, Inc.
MIC28512
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
Micrel, Inc. is a leading global manufacturer of IC solutions for the worldwide high-performance linear and power, LAN, and timing & communications
markets. The Company’s products include advanced mixed-signal, analog & power semiconductors; high-performance communication, clock
management, MEMs-based clock oscillators & crystal-less clock generators, Ethernet switches, and physical layer transceiver ICs. Company customers
include leading manufacturers of enterprise, consumer, industrial, mobile, telecommunications, automotive, and computer products.
Corporation headquarters and state-of-the-art wafer fabrication facilities are located in San Jose, CA, with regional sales and support offices and advanced
technology design centers situated throughout the Americas, Europe, and Asia. Additionally, the Company maintains an extensive network of distributors
and reps worldwide.
Micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this datasheet. This information
is not intended as a warranty and Micrel does not assume responsibility for its use. Micrel reserves the right to change circuitry, specifications and
descriptions at any time without notice. No license, whether express, implied, arising by estoppel or otherwise, to any intellectual property rights is granted
by this document. Except as provided in Micrel’s terms and conditions of sale for such products, Micrel assumes no liability whatsoever, and Micrel
disclaims any express or implied warranty relating to the sale and/or use of Micrel products including liability or warranties relating to fitness for a particular
purpose, merchantability, or infringement of any patent, copyright, or other intellectual property right.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can
reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into
the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser’s
use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully indemnify
Micrel for any damages resulting from such use or sale.
© 2014 Micrel, Incorporated.
March 25, 2015
32
Revision 1.2